U.S. patent number RE30,586 [Application Number 06/008,775] was granted by the patent office on 1981-04-21 for solid-state regulated voltage supply.
This patent grant is currently assigned to Analog Devices, Incorporated. Invention is credited to Adrian P. Brokaw.
United States Patent |
RE30,586 |
Brokaw |
April 21, 1981 |
Solid-state regulated voltage supply
Abstract
A solid-state (IC) regulated voltage supply compensated for
effects of changes in temperature comprising first and second
transistors operated at different current densities. Associated
circuitry develops a voltage proportional to the .DELTA.V.sub.BE of
the two transistors and having a positive temperature coefficient.
This voltage is connected in series with the V.sub.BE voltage of
one of the two transistors, having a negative temperature
coefficient, to produce a resultant voltage with nearly zero
temperature coefficient. A feedback circuit responsive to current
flow through the two transistors automatically adjusts the base
voltages to maintain a predetermined ratio of current density for
the two transistors. Other embodiments provide higher-level DC
outputs and compensation for base current flow.
Inventors: |
Brokaw; Adrian P. (Burlington,
MA) |
Assignee: |
Analog Devices, Incorporated
(Norwood, MA)
|
Family
ID: |
21733596 |
Appl.
No.: |
06/008,775 |
Filed: |
February 2, 1979 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
Issue Date |
|
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799760 |
May 23, 1977 |
3887863 |
Jun 3, 1975 |
|
Reissue of: |
419616 |
Nov 28, 1973 |
03887863 |
Jun 3, 1975 |
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Current U.S.
Class: |
323/314 |
Current CPC
Class: |
G05F
3/30 (20130101) |
Current International
Class: |
G05F
3/08 (20060101); G05F 3/30 (20060101); G05F
001/48 () |
Field of
Search: |
;307/296R,297
;323/1,4,16,19,22T,23,25 |
References Cited
[Referenced By]
U.S. Patent Documents
Other References
Kuijk, "A Precision Reference Voltage Source", IEEE Journal of
Solid State Circuits, vol. SC-8, No. 3, pp. 222-226, Jun. 1973.
.
Brokaw, "A Simple Three-Terminal IC Band Gap Reference", IEEE
Journal of Solid State Circuits, vol. SC-9, No. 6, pp. 388-393,
Dec. 1974..
|
Primary Examiner: Pellinen; A. D.
Attorney, Agent or Firm: Parmelee, Johnson, Bollinger &
Bramblett
Parent Case Text
.Iadd.This is a continuation of U.S. Pat. application Ser. No.
799,760 filed May 23, 1977, which is a Reissue application of U.S.
Pat. No. 3,887,638 dated June 3, 1975. .Iaddend.
Claims
I claim:
1. A solid-state temperature-compensated voltage supply
comprising:
first and second transistors;
a resistor connected between the emitter of said first transistor
and the emitter of said second transistor,
circuit means for furnishing supply voltage to said two transistors
to develop current flow therethrough with the current through said
first transistor also flowing through said resistor;
means for sensing the magnitudes of the respective currents flowing
through said two transistors;
voltage-control means responsive to the currents sensed by said
sensing means and operable to adjust the base potentials of said
transistors to maintain the magnitudes of said transistor currents
at levels which provide a predetermined non-unity ratio of current
densities within the two transistors and thereby cause the current
through said resistor to vary positively with respect to
temperature of said two transistors;
means for developing a first voltage proportional to said resistor
current and for combining said first voltage with a second voltage
which varies negatively with respect to temperature to produce a
combined voltage having minimal overall variation with respect to
temperature; and
output means coupled to said last-named means and including an
output terminal providing an output voltage proportional to said
combined voltage.
2. Apparatus as in claim 1, wherein said voltage-control means
comprises a high-gain amplifier serving as a comparator responsive
to signals proportional to said transistor currents to produce an
output signal corresponding to the difference between said signals
proportional to said currents; and
means coupling a voltage proportional to said output signal to the
bases of said transistors to drive the base potentials to values
providing the desired ratio of current densities in said
transistors.
3. Apparatus as in claim 2, including a voltage-dividing network
coupled in the output of said amplifier and having a network
terminal providing a voltage which is a predetermined fraction of
the amplifier output; and
means coupling said network terminal to the bases of said
transistors.
4. Apparatus as in claim 2, wherein said sensing means comprises
first and second load resistors connected in the collector circuits
of said transistors, respectively.
5. Apparatus as in claim 1, wherein said transistor bases are
connected together to provide equal base potentials.
6. Apparatus as in claim 1, wherein said transistor bases are
coupled together by resistor means to compensate for the effects of
change in base current.
7. A solid-state temperature-compensated voltage supply
comprising:
first and second transistors;
positive and negative voltage lines;
means coupling one of said voltage lines to the transistor
collectors;
a resistor connected in a circuit between the emitter of said first
transistor and the other of said voltage lines, to carry the
current flowing through said first transistor;
means connecting the emitter of said second transistor to the end
of said resistor which is remote from said first transistor
emitter;
means for sensing the magnitudes of the respective currents flowing
through said first and second transistors;
voltage-control means responsive to said transistor currents and
operable to adjust the base potentials of said transistors to
maintain the magnitudes of said transistor currents at levels which
provide a predetermined non-unity ratio of current densities within
the two transistors and thereby cause the current through said
resistor to vary positively with respect to temperature of said two
transistors;
means for developing a first voltage proportional to said resistor
current and for combining said first voltage with a second voltage
which varies negatively with respect to temperature to produce a
combined voltage having minimal variation with respect to
temperture; and
output means coupled to said last-named means and including an
output terminal providing an output voltage proportional to said
combined voltage.
8. Apparatus as in claim 7, including first and second load
resistors in the collector circuits of said transistors,
respectively, to develop voltage drops proportional to the
transistor currents flowing therethrough;
an amplifier having its input terminals connected to said collector
circuits respectively to receive therefrom voltages proportional to
the corresponding collector currents; and
means connecting the output of said amplifier to the bases of said
transistors to apply thereto a voltage proportional to the
amplifier output to drive the base potentials to a value providing
a null voltage at the input of said amplifier.
9. Apparatus as in claim 8, including a voltage-divider network
connected to the output of said amplifier; and
means coupling an intermediate point of said network to said
transistor bases to provide thereto a control voltage which is a
predetermined fraction of the amplifier output.
10. A solid-state regulated-voltage supply comprising:
first and second transistors;
positive and negative supply voltage lines;
means coupling one of said supply voltage lines to the collectors
of said two transistors;
first and second resistors connected in series between the emitter
of said first transistor and the other of said supply voltage lines
to carry the current flowing through said first transistor;
means connecting the emitter of said second transistor to the
junction between said first and second resistors, whereby said
second resistor also carries the current flowing through said
second transistor;
means establishing a predetermined relationship between the base
potentials of said two transistors;
circuit means for establishing different current densities in said
two transistors with the ratio of current densities being set at a
predetermined value to cause the currents through said resistors to
vary positively with respect to temperature; and
output circuit means connected to the base of said second
transistor for developing at an output terminal an output voltage
proportional to the voltage across said second resistor combined
serially with the V.sub.BE voltage of said second transistor.
11. In a voltage supply of the type comprising means to produce a
first voltage having a positive temperature coefficient for
combination with a second voltage having a negative temperature
coefficient so as to develop a combined voltage having a
substantially reduced overall temperature coefficient;
the improvement in said means for producing said first voltage
having a positive temperature coefficient which comprises;
first and second transistors arranged to conduct respective
currents therethrough;
means connecting the bases of said two transistors together to
provide for tracking of the base potentials;
sensing means coupled to both of said transistors and responsive to
said currents passing therethrough;
voltage-control means coupled to said sensing means and having an
output circuit for producing a control voltage responsive to the
change in the relative levels of said transistor currents;
means connecting said output circuit to the base of a least one of
said transistors for automatically adjusting the base voltage
thereof responsive to said control voltage so as to maintain the
ratio of said transistor currents at a value which provides a
non-unity ratio of current densities within said transistors;
and
means connected to the emitters of both of said transistors to
produce a voltage proportional to the difference in base-to-emitter
voltage of said two transistors to serve as said first voltage
having a positive temperature coefficient.
12. A voltage supply as claimed in claim 11, wherein said
voltage-control means comprises a high-gain amplifier producing
said control voltage at its output.
13. A voltage supply as claimed in claim 12, including means
coupling to said one transistor base a voltage proportional to the
output voltage of said amplifier.
14. A voltage supply as claimed in claim 13, wherein said coupling
means comprises voltage-dividing means to couple to said transistor
base a voltage which is a pre-set fraction of the amplifier output
voltage.
15. A voltage supply as claimed in claim 11, wherein the emitters
of said two transistors have substantially different areas.
16. A voltage supply as claimed in claim 15, wherein said
transistor currents are maintained equal.
17. A solid-state temperature-compensated voltage supply
comprising:
first and second transistors arranged to conduct respective
currents;
voltage means to provide base voltage to said transistor bases to
produce current densities therein having a non-unity ratio;
circuit means including resistance means connected to the emitters
of said two transistors to develop a first voltage proportional to
the difference in base-to-emitter voltages of said transistors and
to apply said first voltage to the emitter of said second
transistor;
means coupling the bases of said two transistors together to
provide for tracking of the base potentials;
an output terminal; and
means coupling said output terminal to the base of said second
transistor to provide at said output terminal an output voltage
proportional to said first voltage combined with the
base-to-emitter voltage of said second transistor.
18. A voltage supply as claimed in claim 17, wherein said voltage
means comprises an amplifier the input of which is coupled to said
two transistors to receive signals therefrom corresponding to the
transistor currents; and
means coupling the output of said amplifier to the bases of said
two transistors to automatically maintain the base potentials at
the value which produces the required transistor currents to
maintain the transistor current densities at the desired non-unity
ratio.
19. A voltage supply as claimed in claim 18, wherein said coupling
means comprises a voltage dividing network arranged to apply to the
base of said second transistor a voltage which is a predetermined
fraction of the amplifier output voltage, whereby said amplifier
output serves as said output terminal developing an output voltage
which is greater than the base voltage at said second transistor.
.Iadd. 20. A solid-state band-gap reference circuit comprising:
first and second transistors each having a base, emitter and
collector;
positive and negative supply leads;
first and second sensing resistor means connected between said
positive supply lead and said collectors respectively;
third and fourth series resistor means connected between the
emitter of said first transistor and said negative supply lead;
means connecting the emitter of said second transistor to a
junction between said third and fourth resistor means;
means coupling said transistor bases together to provide for
tracking of the base voltages thereof;
means to apply a control voltage to said bases;
said resistor means being arranged to set the currents through said
transistors at levels providing a non-unity ratio of current
densities therein; and
amplifier means having a pair of input terminals coupled
respectively to said first and second resistor means to develop an
amplifier input voltage responsive to variations in the relative
magnitude of the currents flowing through said two transistors
respectively, whereby the magnitude of the output of said amplifier
reflects a comparison between the base voltage of said transistors
and a predetermined band-gap reference voltage developed as a
composite voltage consisting of a first voltage proportional to the
difference in base-to-emitter voltages of said two transistors, and
a second voltage corresponding to the base-to-emitter voltage of
said second transistor. .Iaddend. .Iadd. 21. A solid-state band-gap
reference device comprising:
first and second transistors having their bases coupled
together;
means for supplying commonly controllable and tracking base
voltages to said two transistors;
means coupled to said transistors for developing therethrough
currents of magnitude providing a non-unity ratio of current
densities in said transistors;
means for producing a first voltage responsive to the difference
between the base-to-emitter voltages of said two transistors;
means to connect said first voltage to the emitter of one of said
transistors, said first voltage and the base-to-emitter voltage of
said one transistor defining a composite voltage compensated for
temperature;
means to sense the changes in relative magnitudes of current
through said two transistors responsive to changes in said commonly
controllable and tracking base voltages of said two transistors;
and
an amplifier coupled to said sensing means to produce an output the
magnitude of which reflects a comparison of said changes in current
through said two transistors produced by changes in said commonly
controllable base voltages. .Iaddend.
Description
BACKGROUND OF THE INVENTION
1. Field of the Invention
This invention relates to regulated DC voltage supplies. More
particularly, this invention relates to solid-state (IC) regulators
capable of maintaining a substantially constant DC output voltage
in the face of temperature variations.
2. Description of the Prior Art
Conventional prior-art regulated voltage supplies commonly have
included an internal reference source and an error amplifier
arranged to compare the reference voltage with a pre-set fraction
of the regulated DC output voltage. The output of the error
amplifier is directed to a control element, such as a controllable
impedance or the like, arranged to adjust the output DC voltage so
as to maintain the two compared voltages equal. Fluctuations in the
DC output voltage are thereby reduced.
In transistorized voltage-regulator circuits, the reference source
typically has been a Zener diode. However, as is known in the art,
Zener diodes have certain inherent characteristics which
undesirably restrict the capability of a voltage regulator. An
alternative type of solid-state regulator has been developed which
does not use a Zener diode reference, relying instead on certain
temperature-dependent characteristics of the base-to-emitter
voltage (V.sub.BE) of a transistor.
U.S. Pat. No. 3,617,859 discloses a circuit of the latter type
which includes a diode-connected transistor operated at one current
density, and a second transistor operated at a different current
density. These two transistors are interconnected with associated
circuitry so as to develop a voltage proportional to the difference
in the respective base-to-emitter voltages (.DELTA.V.sub.BE). This
difference voltage has a positive temperature-coefficient (TC), and
is connected in series with the V.sub.BE voltage of a third
transistor, having a negative TC, to produce a composite resultant
voltage which serves as the output of the regulator. Since the
temperature coefficients of the two individual voltages are of
opposite sign, the output voltage can be made relatively
insensitive to temperature variations by proper choice of certain
parameters.
Although such regulators based on the V.sub.BE characteristic of
transistors have significant advantages, the circuit arrangements
proposed and used heretofore suffer from serious limitations. It is
a principal object of the present invention to provide a
solid-state voltage regulator which avoids or significantly
minimizes such limitations of prior art regulators.
SUMMARY OF THE INVENTION
In an exemplary embodiment of the present invention, to be
described in detail hereinbelow, there is provided a two-transistor
voltage-regulator circuit wherein the ratio of current densities of
the two transistors is automatically controlled to a predetermined
value (different from unity) by a negative feedback arrangement. A
voltage corresponding to the .DELTA.V.sub.BE of the two transistors
is developed, having a positive TC, and this voltage is connected
in series with the V.sub.BE voltage of one of the two transistors,
having a negative TC. The circuit parameters are selected so that
the resultant combined voltage has a very low temperature
coefficient. The regulator of this invention provides important
advantages over previous regulators, as will be outlined
hereinbelow in describing specific embodiments of the
invention.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a circuit diagram of one embodiment of the present
invention;
FIG. 2 is a circuit diagram of a modified arrangement to provide
higher regulated output voltages;
FIG. 3 shows a further circuit arrangement modified to reduce base
current effects; and
FIG. 4 is a circuit diagram of a voltage reference source including
means to establish bias levels and to provide current limiting.
DESCRIPTION OF PREFERRED EMBODIMENTS
Referring now to FIG. 1, there is shown a circuit diagram
representing basic components and interconnections of a regulated
voltage source in accordance with the present invention. The
circuit includes a pair of transistors Q.sub.1 and Q.sub.2 which
are supplied with operating voltages by positive and negative
voltage lines 10 and 12. The emitter of the left-hand transistor
Q.sub.2 is coupled through two series-connected resistors R.sub.2
and R.sub.1 to the negative voltage line 12, and the emitter of the
other transistor Q.sub.1 is connected to the common junction 18
between the two resistors.
The invention proceeds on the concept of (1) developing a first
voltage, having a positive temperature coefficient (TC), (2)
combining that voltage serially with a second voltage having a
negative TC, and (3) relating the two temperature coefficients in a
complementary sense such that the resultant composite voltage has a
very low TC, approximately zero. To develop the positive TC
voltage, the two transistors Q.sub.1 and Q.sub.2 are controllably
operated at markedly different current densities (i.e., referring
to density of current flowing through the emitters), and a voltage
is produced proportional to the difference in the two transistor
base-to-emitter voltages, referred to as .DELTA.V.sub.BE.
In the specific embodiment disclosed herein, transistor Q.sub.2 is
operated at a smaller current density than the other transistor
Q.sub.1. Such difference in current densities can be produced (1)
by using identical transistors operating at unequal currents, (2)
by using transistors having unequal emitter areas operating at
equal currents, or (3) by some combination of the latter two
arrangements. Simply by way of example, in the described embodiment
the emitter areas of the transistors Q.sub.1 and Q.sub.2 are
specified as A and nA respectively, with n being greater than one,
and the currents through the two transistors are equal.
The transistor currents are forced to be equal by a negative
feedback arrangement comprising current-sensing means in the form
of equal-resistance load resistors R.sub.L1 and R.sub.L2 in the
collector circuits of transistors Q.sub.1 and Q.sub.2. These load
resistors develop voltages which are proportional to the respective
collector currents, and which are directed by leads 24, 26 to the
input terminals of a high-gain operational amplifier 28. The output
of this amplifier is connected to a common base line of the two
transistors Q.sub.1 and Q.sub.2, and also to an output terminal 30
presenting the DC output voltage of the regulator. The amplifier 28
drives the common base line until the voltage drops across the load
resistors R.sub.L1 and R.sub.L2 are equal, thereby forcing the
transistor collector currents to be equal. By using well matched
transistors Q.sub.1 and Q.sub.2, the emitter currents also will be
equal.
Since the bases of transistors Q.sub.1 and Q.sub.2 are connected
together, the difference voltage .DELTA..sub.VBE will appear across
emitter resistor R.sub.2, and the current through that resistor
thus will be directly proportional to .DELTA.V.sub.BE. The series
connected resistor R.sub.1 also carries this emitter current, and
additionally carries the emitter current of the second transistor
Q.sub.1. Since the latter emitter current is directly proportional
to the first emitter current (actually equal, in this particular
example), it will be evident that the total current through
resistor R.sub.1, and hence the voltage across that resistor, also
will be directly proportional to .DELTA.V.sub.BE.
It has previously been established that, for two transistors
operating at different current densities, the difference in
base-to-emitter voltage is given by:
where T is absolute temperature, k is Boltzman's constant, q is the
charge of an electron, and J.sub.1 /J.sub.2 is the ratio of the
transistor current densities. Accordingly, the voltage developed
across resistor R.sub.1 is independent of absolute emitter current,
and is a linear function of absolute temperature with a positive
temperature coefficient.
As is evident from the circuit lay-out, the voltage across resistor
R.sub.1 is in series with the V.sub.BE voltage of transistor
Q.sub.1 and the resultant composite voltage constitutes the DC
output voltage on terminal 30. Since V.sub.BE has a negative
temperature coefficient, changes in that voltage with temperature
tend to complement the positive TC changes in the voltage across
resistor R.sub.1.
To approach zero TC, the output voltage at the regulator terminal
30, with respect to the negative voltage line 12, should be set
approximately to the value of the energy band-gap voltage
(V.sub.GO), extrapolated to 0.degree.. For silicon, this
extrapolated voltage is 1.205V. A slightly higher voltage produces
superior results. It can be shown mathematically, based on certain
reasonable assumptions, that for zero TC the output voltage should
be set at:
where m.congruent.1.5 and To is the nominal operating
temperature.
This voltage V.sub.OUT can be adjusted to the desired value by
proper selection of resistor R.sub.1, such that the resistive
voltage drop complements the V.sub.BE of Q.sub.1 to optimize the
total (sum) voltage for zero TC.
When the DC output voltage (V.sub.OUT) at terminal 30 drops below
the pre-established optimal level, the ratio of collector currents
I.sub.2 /I.sub.1 is larger than the ratio of load resistors
R.sub.L1 /R.sub.L2 (i.e., larger than one), so that the input to
amplifier 28 is positive. This causes the amplifier output to
increase, so as to return the voltage V.sub.OUT back up to the
optimal level. If the DC output rises above optimal, the feedback
action of amplifier 28 will have the opposite effect. Thus the
voltage-control circuit continuously holds the DC output voltage at
the proper level to provide a very low overall temperature
coefficient, close to zero.
In some applications, DC output voltages higher than the energy
band-gap voltage may be required. FIG. 2 shows an arrangement for
that purpose. The basic operation of this circuit is similar to
that of FIG. 1, and like reference numerals are used throughout for
corresponding elements. However, FIG. 2 differs in that the output
of amplifier 28 is connected to a voltage-dividing network
comprising two series-connected resistors hR.sub.3 and R.sub.3. The
common junction terminal 32 of these resistors provides a voltage
which is a predetermined fraction of V.sub.OUT, and this voltage is
directed to the commonly connected bases of transistor Q.sub.1 and
Q.sub.2.
As before, the amplifier 28 drives the transistor bases until their
collector currents are equal. By proper selection of circuit
parameters, the reference voltage (V.sub.REF) at this stable point
can be set to be optimum for achieving zero TC. The output voltage
(V.sub.OUT) then will be some predetermined multiple of V.sub.REF,
specifically (h+1).multidot.V.sub.REF.
This arrangement of FIG. 2 provides a quite accurate result. It is
degraded only a small amount due to the base current of the
transistors. This base current is relatively low, and in any event
the positive TC of the transistor beta tends to act with the
positive TC of the emitter current to stabilize the base current
and reduce any drift.
Where further reduction of such small drift effects may be
desirable, a controlled beta PNP may be used to reflect the base
current from a pair matched to Q.sub.1 and Q.sub.2 and connected in
cascode with them into the base of Q.sub.1 and Q.sub.2.
Alternatively, the base of transistors Q.sub.1 and Q.sub.2 can be
connected together through a resistor R.sub.4, as illustrated in
FIG. 3. Here, the voltage across resistor R.sub.2 is no longer
.DELTA.V.sub.BE, since the bases are no longer at the same voltage.
However, it can be shown that this arrangement may, with matched
betas, produce the basic regulation of the FIG. 2 embodiment, but
with reduced drift due to base current, providing R.sub.4 is
selected such that:
To take into account the possible effects of base spreading
resistance of the two transistors, further analysis indicates that
the base-connecting resistor R.sub.4 should be selected such
that:
where R.sub.b1 and R.sub.b2 are the base spreading resistances of
transistors Q.sub.1 and Q.sub.2 ; C.sub.1 =I.sub.e1 /I.sub.e2
(emitter currents of Q.sub.1 and Q.sub.2); and C.sub.2 =R.sub.1
/R.sub.2
The above-derived expression for R.sub.4 also indicates that the
use of a base-connecting resistor may be helpful in the basic
circuit configuration of FIG. 1. For R.sub.4 to be zero, R.sub.b2
must be larger than R.sub.b1 ; typically, however, Q.sub.2 is the
larger transistor with a very low base resistance, and design
considerations thus suggest that the base resistance of Q.sub.1
should be minimized. It may be possible to correct for the effect
with a pinched base resistor in series with the large
transistor.
Voltage-regulated supplies in accordance with the present invention
have a number of important and beneficial features. Foremost, such
voltage supplies provide a highly stable output voltage in the face
of changing ambient temperature. Only two matched active elements
are required, rather than three as in the above-identified U.S.
Pat. No. 3,617,859. Advantageously, the reference voltage in the
disclosed circuits appears in the control loop at a point with a
high impedance, so that it can readily be driven. Moreover, the
reference voltage may be multiplied as desired to produce output
voltages higher than the band-gap voltage, by means of a single
control loop, and without stacking junctions. In the FIG. 2
configuration, the reference voltage can first be adjusted to
minimize temperature coefficient, and then the output voltage can
separately be adjusted to a predetermined voltage without affecting
the temperature coefficient. The basic circuit is convenient to
trim by adjusting a single resistor (R.sub.1). Finite beta and beta
drift does not result in uncorrectable errors; only beta matching
is required.
Referring now to FIG. 4, there is shown a voltage reference source
including transistors Q.sub.1 and Q.sub.2 used to establish the
reference voltage in the manner generally as described hereinabove.
In this circuit, these transistors are driven so that they will
operate at equal collector currents. Neglecting R23, for the
moment, the bases of these transistors are driven from the circuit
output by the voltage divider consisting of R31 and R24. The output
current is provided by Darlington-connected transistors Q.sub.4 and
Q.sub.7, which draw operating current from the input voltage
terminal. The base of Q.sub.4 is driven by a bias current from
Q.sub.18.
The circuit output voltage is controlled by adjusting the base
voltage of Q.sub.4, so that Q.sub.4 and Q.sub.7 form a voltage
follower. A voltage drop provided by Q.sub.3 approximately matches
the V.sub.BE of Q.sub.4, with R27 and Q.sub.15 providing a voltage
drop matching other circuit voltages. The base voltage of Q.sub.4
is controlled by the emitter follower Q.sub.12 which is driven by
Q.sub.1 and Q.sub.14.
In operation the collector current of Q.sub.2 drives the base of
Q.sub.11 negative. Acting as an emitter follower, Q.sub.11 turns on
Q.sub.13 and drives it until its collector current approximately
equals the collector current of Q.sub.2. The base of Q.sub.13
connects to Q.sub.14, a matching transistor. Since R25 and R26 are
also matched, the collector current of Q.sub.14 will approximately
equal that of Q.sub.13 and hence of Q.sub.2. If the collector
current of Q.sub.2 exceeds the collector current of Q.sub.1,
Q.sub.14 will drive the base of Q.sub.12 positive. Alternatively,
if the collector current of Q.sub.1 exceeds the collector current
of Q.sub.2, it will also exceed that of Q.sub.14 and will,
therefore, drive the base of Q.sub.12 negative. The circuit output
voltage will follow the base voltage at Q.sub.12 as previously
explained.
The emitter area of Q.sub.2 is eight times larger than that of
Q.sub.1. When the voltage at the base of Q.sub.1 and Q.sub.2 is
low, the current through R21 and R22 is low. The resulting voltage
drop across R22 will be low, and the base-emitter voltages of
Q.sub.1 and Q.sub.2 will be nearly equal. As a result of the area
ratio mismatch the emitter current in Q.sub.2 will be nearly eight
times the current in Q.sub.1. This current mismatch will cause
Q.sub.14 to drive the base of Q.sub.12, and, thereby, the output -
positive.
If the base voltage applied to Q.sub.1 and Q.sub.2 is made larger,
the current through R21 and R22 will also be larger. At a
sufficiently high base voltage the voltage drop across R22 will
limit the current in Q.sub.2, and it will drop below the current in
Q.sub.1. The excess collector current in Q.sub.1 will drive the
base of Q.sub.12 negative, and with it the circuit output.
Between these two extremes of base voltage there will be a voltage
at which the collector currents of Q.sub.1 and Q.sub.2 are equal.
At this voltage the current in Q.sub.14 will balance the current in
Q.sub.1 and the base of Q.sub.12 will be held at a voltage which
maintains the circuit output and the Q.sub.1 - Q.sub.2 base voltage
constant. Changes in output loading or other disturbances which
tend to change the output voltage will change the voltage on the
bases of Q.sub.1 and Q.sub.2. This will disturb their collector
current balance so as to drive Q.sub.12 to restore the output
voltage. This control loop forcing the collector currents of
Q.sub.1 and Q.sub.2 to be equal satisfies the condition, previously
described, to hold constant C.sub.1 =1.
With the collector currents of Q.sub.1 and Q.sub.2 forced to be
equal, the voltage drop across R22 will be (kT/q) ln J.sub.1
/J.sub.2 =(kT/Q) ln 8. The current in R21 will be just twice that
in R22 so that the voltage across R21 will be proportional to the
drop across R22. Therefore, the voltage at the base of Q.sub.1
which results in the balance condition is the sum of the V.sub.BE
of Q.sub.1 and the temperature-dependent voltage on R21. This
voltage is set (by selecting the ratio of R21 and R22) so that this
voltage is just above the bandgap voltage and satisfies the
conditions previously outlined for zero temperature
coefficient.
The stabilized base voltage of Q.sub.1 is a fraction of the circuit
output voltage determined by R31 and R24. The output voltage is,
therefore, a temperature stable multiple of the bandgap voltage
determined by the resistor ratio. The interbase resistor R23
corrects for the offset and drift due to base current flow in R31.
It also corrects for the base spreading resistance of Q.sub.1, as
previously noted.
The voltage divider R28 and R29 is connected across the circuit
output voltage. It is selected to have a Thevenin equivalent output
voltage which differs from the circuit output voltage by the
bandgap voltage. The equivalent resistance at the divider output is
set at twice the resistance of R21. Transistor Q.sub.5 is designed
to match Q.sub.1. As a result of the equivalent voltage and
resistance applied across its base and emitter, its emitter and
collector currents will be approximately equal to those of Q.sub.1.
This current drives the common base of Q.sub.16 and Q.sub.17, a
matched transistor pair. The matched emitter resistors, R32 and
R33, force the emitter currents of Q.sub.16 and Q.sub.17 to be
equal and raise the output impedance of Q.sub.16. This current
mirror "reflects" the collector current of Q.sub.5 down through
Q.sub.3, R27 and Q.sub.15. A small fraction of this current drives
the base of Q.sub.4 which in turn drives Q.sub.7 and also supplies
the current for Q.sub.13 and Q.sub.14. Since the current in the
Q.sub.5, Q.sub.17, Q.sub.16 and Q.sub.3 path approximates the
current in Q.sub.2, it is approximately half the current in
Q.sub.13 and Q.sub.14 combined. This combined current is the
majority of the emitter current in Q.sub.4. By making the emitter
area of Q.sub.4 twice that of Q.sub.3, the current densities and
hence the base-emitter voltages of Q.sub.3 and Q.sub.4 are made
nearly equal. Therefore, the voltage at the top of R27
approximately equals the voltage applied to R25 and R26. The
currents in R25, R26 and R27 are approximately equal so that the
voltage drops across them are approximately equal. Similarly,
Q.sub.15 is sized so that its emitter current density approximates
that of Q.sub.13 and Q.sub.14. In this way the base voltage of
Q.sub.15 is made nearly equal to the base voltage of Q.sub.13 and
Q.sub.14. This equality is translated through the base-emitter
voltage of the matched transistors Q.sub.11 and Q.sub.12 to the
collectors of Q.sub.2 and Q.sub.1 . This keeps the collector
voltages of these transistors approximately equal at all
temperatures and bias conditions. This minimizes problems resulting
from different base width modulation in Q.sub.1 and Q.sub.2 which
might result from unbalanced collector voltage.
The bias voltage stabilization also keeps the free collector
voltage of Q.sub.15 nearly equal to the base voltage. This helps to
insure an equal split of the current in the forced beta transistor
Q.sub.15 (beta=1). This current split ensures equal emitter
currents in Q.sub.11 and Q.sub.12, thereby minimizing errors due to
differences between their base currents.
The circuit as described so far would have a stable "off" state.
The epitaxial layer FET portion of Q.sub.5 eliminates this
possibility. The FET provides a small starting current that turns
on the circuit when voltage is applied. Although it diverts some of
the current from R28, it has only a small effect on the current
delivered to Q.sub.17. This total current is determined largely by
the voltage drop across the equivalent R28, R29 resistance. The
slight change in Q.sub.5 V.sub.BE which results from the diverted
current is a small fraction of the total voltage applied to R28 and
R29.
The frequency stability of the output control loop is established
by C36. This capacitance rolls off the open-loop gain to unity
below the frequency at which excess phase shift in the PNP's might
cause instability.
Output overload protection is provided by Q.sub.6 and R30. The
output current flows through R30 and produces a small voltage drop
across it. In the event of overload, this voltage will rise and
drive Q.sub.6 on. As Q.sub.6 comes on it will divert the drive
current from the base of Q.sub.4 into the load. As a result, the
output current is limited to that necessary to drive Q.sub.6 on by
way of R30.
The overall circuit consists of a current input amplifier which
operated the control loop stabilizing the reference voltage. The
amplifier input circuit, Q.sub.13 and Q.sub.14, is bootstrapped to
the regulated output. This bootstrap connection minimizes the
effects of power supply voltage variation on the amplifier which
improves the overall supply voltage rejection of the circuit.
Although several preferred embodiments of the invention have been
described hereinabove in detail, it is desired to emphasize that
such details have been disclosed for the purpose of illustrating
the nature of the invention, and should not be considered as
necessarily limiting of the invention which can be expressed in
many modified forms to meet particular requirements.
* * * * *