U.S. patent number RE39,076 [Application Number 08/588,301] was granted by the patent office on 2006-04-25 for apparatus and method for controlling brushless electric motors and position encoders and indicating position thereof.
This patent grant is currently assigned to Papst Licensing GmbH & Co. KG. Invention is credited to Micheal Hermann, Uwe Muller, Johann von der Heide.
United States Patent |
RE39,076 |
von der Heide , et
al. |
April 25, 2006 |
Apparatus and method for controlling brushless electric motors and
position encoders and indicating position thereof
Abstract
A brushless direct current motor is energized by a power stage.
Switches of the power stage are selectively switched by an
electronic control device. In order to determine the correct
commutation pattern, without using rotor position indicators such
as Hall effect devices, the control device is arranged to sample
the power fed to the motor. The samples, each responsive to a test
signal or normal energization pulses received by the motor, in a
ordered sequence define an analogue function whose parameters can
be determined by mathematical analysis. The analysis applied to the
samples derives the necessary commutation information and the
energization signals in a rapid manner by employing a vector or
complex number addition based on functions dependent on variable
inductance in the operation of the motor.
Inventors: |
von der Heide; Johann
(Schramberg, DE), Muller; Uwe (Eisenbach,
DE), Hermann; Micheal (Villingen-Schwenningen,
DE) |
Assignee: |
Papst Licensing GmbH & Co.
KG (Spaichingen, DE)
|
Family
ID: |
25881494 |
Appl.
No.: |
08/588,301 |
Filed: |
May 31, 1990 |
PCT
Filed: |
May 31, 1990 |
PCT No.: |
PCT/DE90/00411 |
371(c)(1),(2),(4) Date: |
November 29, 1991 |
PCT
Pub. No.: |
WO90/15473 |
PCT
Pub. Date: |
December 13, 1990 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
Issue Date |
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Reissue of: |
07777283 |
Nov 29, 1991 |
05280222 |
Jan 18, 1994 |
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Foreign Application Priority Data
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Jun 1, 1989 [DE] |
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3917924 |
Dec 23, 1989 [DE] |
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3942755 |
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Current U.S.
Class: |
318/400.4 |
Current CPC
Class: |
H02P
6/006 (20130101); H02P 6/18 (20130101); H02P
6/185 (20130101); H02P 6/22 (20130101); H02P
25/089 (20160201) |
Current International
Class: |
H02P
7/00 (20060101) |
Field of
Search: |
;318/254,138,434,701,799-812,815,511 ;382/806,815,807,909 |
References Cited
[Referenced By]
U.S. Patent Documents
Foreign Patent Documents
Other References
Transactions on Industry Applications, vol. 1A-21, No. 3, Article:
Microcomputer Control for Sensorless Brushless Motor, Published
May/Jun., 1985. cited by other .
Schroedl, M. and Stefan, T., Algorithm for Mathematical Detection .
. . Antriebssysteme fuer die Geraete- und Kraftfahrzeugtechnik, Bad
Nauheim, Germany, VDE-Verlag Berlin-Offfenbach, pp. 48-54 (1988).
cited by other .
Schroedl, M., "Detection of the Rotor Position of a Permanent
Magnet Synchronous Machine at Standstill", proc. ICEM, Pisu
(Italy), pp. 195-197 (1988). cited by other .
Schroedl, M., "Operation of the Permanent Magnet Synchronous
Machine Without a Mechanical Sensor", Proc. IEE International
Conference on Power Electronics and Variable Speed Devices, London,
pp. 51-56 (1990). cited by other .
P.P. Acarnley et al., "Detection of Rotor Position in Stepping and
Switched Motors by Monitoring of Current Wave forms," IEEE
Transactions on Industrial Electronics, vol. IE-32, No. 3, Aug.,
1985, pp. 215-222. cited by other.
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Primary Examiner: Salata; Johatham
Attorney, Agent or Firm: Welsh & Katz, Ltd.
Claims
We claim:
1. Apparatus for controlling a brushless electric motor having
motor coils, the apparatus comprising: a power stage having a
plurality of switching means, and output terminals for supplying
power to the motor coils of a motor to be controlled; and control
means coupled to the power stage and arranged to apply commutation
signals to the power stage to selectively switch said switching
means whereby power is selectively applied to said motor by way of
said switching means, and wherein the control means is arranged to
receive sequence of samples of a component of the power fed to the
motor, the control means further comprising means for determining
parameters of a function depending on variable inductance in the
motor and defined by the sequence of samples, including means for
performing a vector addition or a complex number addition for each
set of a plurality of the samples, wherein the parameter
determining means operates repeatedly for multiple successive
applications of power to the motor and from the parameters
determines position-dependent and/or inductance-dependent
electrical motor values.
2. The apparatus according to claim 1, wherein the parameter
determining means includes means for determining parameters of
inductance of at least one motor winding as position dependent
electrical motor values.
3. The apparatus according to claim 2, further including a rotor
and a magnetic circuit in the motor affected by and affecting the
at least one motor winding, the magnetic circuit being of a type to
approach magnetic saturation during a rotor movement of 360.degree.
el.
4. The apparatus according to claim 2 or claim 3, wherein the motor
comprises a plurality of permanent magnets having pole faces that
have magnetic fluxes that vary sufficiently across the pole faces
to provide rotor-position-dependent variations of inductance of the
at least one motor winding.
5. The apparatus according to claim 2 or claim 3, wherein a
preliminary current is supplied to the one winding to produce a
pre-magnetization state.
6. The apparatus according to claim 2 or claim 3, wherein the
current path to the one winding is arranged to facilitate the
determination of the inductance of the one winding.
7. The apparatus according to claim 2 or claim 3, wherein the
control means including means for determining the parameters of
inductances of at least two windings as functions of rotor
position, determining a combined function of the inductance-rotor
position functions, and electronically storing the combined
function as a table of values.
8. The apparatus according to claim 7, including means for
interpolating between values in the table of values to provide
further values for storage in the table of values.
9. The apparatus according to claim 8, wherein the means for
interpolating provides a spline interpolation.
10. The apparatus according to claim 8, wherein the means for
interpolating provides an interpolation based on a sin-cos series
expansion.
11. The apparatus according to claim 8, wherein the control means
adjusts the application of commutation signals in response to a
determination of rotor position for which the decision criterion is
the similarity of functions, which functions are based on at least
one of measured energization current pulses to the winding and on
the function defined by the electronically stored table of
values.
12. The apparatus according to claim 11, wherein the similarity of
functions is based upon the method known as sum of least squares or
the method known as sum of smallest absolute amounts.
13. The apparatus according to claim 2 or claim 3, wherein the
position dependent electrical motor values are applied to determine
rotor position, the control means including means for multiplying
the individual measured current values by associated directional
components thereof and adding the products on a directional basis
to assist in the determination of position.
14. The apparatus according to claim 13, wherein the directional
components have directions separated by so-called integral angles,
which are respective ones of the angles 0=360.degree./n, wherein
n=1, 2, 3 etc.
15. The apparatus according to claim 14, wherein the addition of
products on a directional basis is a vector addition.
16. The apparatus according to claim 13, wherein the control means
establishes a test mode including planned, incremental movements of
the rotor, and the determining and electronically storing means
determines and electronically stores the associated directional
components of the currents.
17. The apparatus according to claim 12, wherein the control means
sets a desired rotor position, compares it with the determined
rotor position to provide an error signal, and applies the error
signal via proportional-integral-derivative negative feedback to
reduce the error signal.
18. The apparatus according to claim 11, wherein the control means
schedules a start-up mode for the motor according to a schedule in
which the period duration of the applied current pulses is
shortened progressively to approach a normal operating mode.
19. The apparatus according to claim 18, wherein the control means
provides a linear shortening of the period duration of the applied
current pulses.
20. The apparatus according to claim 13, wherein the control means,
in determining rotor position, determines a phase of a fundamental
oscillation of at least some of the functions, which are repeatedly
determined.
21. The apparatus according to claim 20, wherein the control means
determines the amplitude and phase of the fundamental oscillation
by a centroid analysis.
22. The apparatus according to claim 11, wherein the control means
further adjusts the application of commutation signals and the
determination of rotor position based up an electronically stored
table of correction values that are specific to the brushless
electric motor.
23. The apparatus according to claim 11, wherein, for the
application of commutation signals, the control means includes, in
series toward the motor, a commutation logic unit, a commutation
control unit, a current detection unit, a counter, an attenuator, a
voltage comparison unit, and a signal delay unit, the current
detection unit including current sensors and a sample and hold
device, the voltage comparison unit including inputs for voltages
from the coils, the control means further including an input to
which an interrupt request may be applied.
24. The apparatus according to claim 23, wherein the motor includes
at least three coils and the control means provides at least three
commutation signals.
25. The apparatus according to claim 11, wherein the control means,
in determining rotor position, determines a phase of a fundamental
oscillation of at least some of the functions, which are repeatedly
determined.
26. The apparatus according to claim 7, wherein the brushless
electric motor is operated at no load.
27. The apparatus according to claim 13, wherein the brushless
electric motor is operated at no load.
28. The apparatus according to claim 20, wherein the control means
determines phases and amplitudes of oscillatory functions of the at
least some functions by Fourier transform analysis.
29. The apparatus according to claim 17, comprising additional
control means that provides a desired rotor position, compares it
with the determined rotor position to provide an error signal, and
applies the error signal via proportional-integral-derivative
negative feedback to reduce the error signal.
.Iadd.30. A brushless direct current motor, comprising: a stator
including at least three stator coils; a rotor which is mounted for
rotation about an axis with respect to said stator and which
includes a permanent magnetic member, wherein an air gap is defined
between adjacent surface of said permanent magnetic member and said
at least three stator coils; a plurality of switches operatively
electrically connected to said stator coils, said switches allowing
power to be supplied to said stator coils during operation of said
motor; and a switch controller electrically connected to said
plurality of switches, said switch controller applying commutation
signals to said plurality of switches to selectively switch said
switches thereby selectively applying power to said stator coils
during operation of said motor, said switch controller including, a
first circuit mounted stationary with respect to said stator and
operatively connected to said stator coils, said first circuit
generating a sequence of samples of a component of power fed to
said stator coils which are generally representative of the
rotational position of said rotor with respect to said stator, and
a second circuit electrically connected to said first circuit,
wherein said second circuit determines parameters of a function
based upon a variable inductance in the motor and said sequence of
samples, and wherein said second circuit generates said commutation
signals based upon said parameters of a function..Iaddend.
.Iadd.31. The brushless direct current motor of claim 30, wherein
said second circuit further comprises a memory electrically
connected to said first and second circuits, said memory storing
said sequence of samples to allow an adjustable offset in time to
be applied to said commutation signals..Iaddend.
.Iadd.32. A brushless direct current motor, comprising: a stator
including at least three stator coils; a rotor which is mounted for
rotation about an axis with respect to said stator and which
includes a permanent magnetic member, wherein an air gap is defined
between adjacent surface of said permanent magnetic member and said
at least three stator coils; a plurality of switches operatively
electrically connected to said stator coils, said switches allowing
power to be supplied to said stator coils during operation of said
motor; and a switch controller electrically connected to said
plurality of switches, said switch controller applying commutation
signals to said plurality of switches to selectively switch said
switches thereby selectively applying power to said stator coils
during operation of said motor, said switch controller including, a
first circuit mounted stationary with respect to said stator and
operatively connected to said stator coils, said first circuit
generating a sequence of samples of a component of power fed to
said stator coils which are generally representative of the
rotational position of said rotor with respect to said stator, and
a second circuit electrically connected to said first circuit,
wherein said second circuit includes a memory electrically
connected to said first circuit, said memory storing said sequence
of samples to allow an adjustable offset in time to be applied to
said commutation signals..Iaddend.
.Iadd.33. The brushless direct current motor of 32 wherein said
second circuit determines parameters of a function based upon a
variable inductance in the motor and said sequence of samples, and
wherein said second circuit generates said commutation signals
based upon said parameters of a function..Iaddend.
Description
CROSS-REFERENCE TO RELATED APPLICATION
The invention of this application is related to that of our
copending patent application Ser. No. 07/805,154, filed Dec. 11,
1991.
BACKGROUND OF THE INVENTION
1. Field of the Invention
The invention relates to a motor or a position indicator (position
sensor) comprising an apparatus or method for controlling a
brushless electric motor. The term motor is here understood to mean
also e.g. a linear motor, whose position is to be detected. Both a
rotary motor and a linear motor has as its movable part a
rotor.
2. Description of the Related Art
Different types of position sensors are known and are e.g. used for
determining the position of motor elements. In most cases there is
an associated linear acting principle for a rotary acting
principle. The most important classes of such position sensors or
indicators are based on the following:
Position sensors with direct contact with the movable element:
a) Mechanical contacts.
Position sensors with contactless operating means: a) optoelectric
position sensors: light barriers, differential photosensors,
optical interference sensors b) magnetoelectronic (galvanomagnetic)
position sensors: Hall effect sensors (based on Hall generators),
Wiegand effect sensors, magnetostrictive sensors, magnetoresistive
sensors c) electromagnetic position sensors: resolvers, eddy
current sensors, differential transformers d) electrical position
sensors: capacitively acting position sensors, microwave distance
measuring equipment.
This list only represents the most important classes and only
constitutes a small selection of the sensors used in practice for
rotary or linear measuring functions. Hereinafter the position
determination or detection processes are described to the extent
that they are used both in linear or rotary acting electric
motors.
Reference is subsequently made to the electrical angular position.
The latter is measured in electrical degrees, e.g. 20.degree. el.
In the case of a bipolar motor with a number of poles p=1, the
electrical degrees of the rotor position correspond to the
mechanical degrees. In the case of a four pole motor (p=2), on
rotating by 360.degree. mech, the rotor passes through in all
p.times.360.degree. el, i.e. 720.degree. el. etc.
The described measuring methods agree within a range of 360.degree.
el, i.e. they determine the electrical angular position of the
rotor. For example, with a four-pole motor (p=2), the measured
electrical angular position 90.degree. el represents a rotor
position of 90.degree. el or (90+360).degree. el, which is
unimportant for the commutation control.
Besides methods, which carry out a position determination in motors
through external sensors (e.g. using Hall effect sensors, light
barriers and magnetoresistive sensors), there are methods whose
function is to determine the position of a motor by components
within the motor. These methods can essentially be subdivided into
two classes.
In a first class use is made of the generator characteristics of
motor coils and there is a dynamic position determination. As a
function of the level or integral of the generated voltage over
time, conclusions can be drawn regarding the rotor position.
This class is characterized by different use fields and widely
varying embodiments. As an example of the relevant prior art
reference is made to the following patent literature: U.S. Pat. No.
4,495,450 EP-B 0171635 DE-OS 27 52 880 DE-PS 30 13 550 EP-A
0316077
It is unimportant whether the magnetic field of the rotor is
generated by permanent magnets or by electric magnets. To the
extent that reference is made here and hereinafter to permanent
magnets, it always includes electric magnets, unless indicated
otherwise.
In the same way it is possible to functionally interchange rotating
and stationary parts of the electric motor, i.e. the invention is
suitable for internal rotor-type motors, external rotor-type
motors, motors with a planar air gap, motors in which the permanent
magnetic rotor rotates and the motor winding is stationary or
conversely motors in which the motor winding rotates and is e.g.
supplied across slip rings and the permanent magnetic part is
stationary and other motor construction types, in the manner
illustrated by examples hereinafter.
In a second class the position determination methods are based on
the fact that the motors in question are not ideal, i.e. despite
constant, distinct electrical quantities, there are normally
variable positional and angular dependencies of the mechanical
quantities.
Thus, in this case interest is attached to those physical effects,
which are based on the position-dependent change of electrical or
electromagnetic parameters of the motor and which can be
established by checking with electrical means only. Such a check or
measurement of interesting parameters can e.g. be performed in that
one or more suitable additional measuring coils are fitted at one
or more points within the motor. As two-terminal networks, the
coils have complex resistors, which are characterized by ohmic
resistances and inductance values and said complex resistors are
normally also frequency-dependent.
When choosing a suitable fitting point within the motor one or more
of the indicated quantities are dependent on the position taken up
by the rotor in the case of linear and rotary motors.
Thus, e.g. by the very precise determination of the inductance of
the measuring coils, conclusions can be drawn concerning the rotor
position, at least with respect to the angular position, as
explained hereinbefore. It is disadvantageous that in place of an
externally fitted sensor, there must be a sensor in the form of at
least one additional measuring coil within the motor.
If there are several such measuring coils, then apart from the
complex input resistance and complex output resistance,
additionally the coupling ratio of two coils can be used as a
measured quantity. There are then several possibilities for
measuring the position-dependent quantities in combination. It is
also possible to use different frequencies or frequency
mixtures.
A method of this type is e.g. described in German patent 26 16 552.
It is advantageous that the inductances of the coils can be such
that without difficulty higher frequency test signals can be
used.
In a technically improved procedure no additional fitting of
measurement coils takes place within the motor. Instead of this
concomitant use is made of the inductance values of the inductances
of the motor, i.e. the strands (phases) or coils of the motor for
the measurement process to be performed.
As no additional constructional expenditure apart from the
necessary sensor electronics is required in the last-mentioned
sensor class, all the members of this class have the advantage of
potentially high cost advantages, whilst simultaneously improving
the reliability of the motor.
Examples of such arrangements are disclosed by EP-A-251785 and
EP-A-171635. In the arrangement according to EP-A-251785
essentially only the additional power stages are required for
controlling the motor phases, as well as special components as
electronics for controlling the motor. However, the circuit suffers
from the disadvantage that a range of 360.degree. el to be measured
is split up into two undistinguishable ranges of in each case
180.degree. el. This is due to the fact that symmetrically to the
angle of 0.degree. el or 180.degree. el, in the following range of
180.degree. to 360.degree. el there are equally large function
values, i.e. it is impossible to know whether a basic value of
0.degree. or 180.degree. el must be added to the calculated rotor
position (FIG. 22A). Moreover the precision of the position
determination is limited to approximately .+-.60.degree. el.
An arrangement and a method for obviating this problem are
described in a work by Watanabe, Isii and Fugi (IECON 87, pp
228-234), whose solution is based on the fact that depending on the
pole position of the permanent magnets of the rotor an individual
stator coil through the permanent magnetic premagnetization has a
current passed through it either more rapidly, or more slowly in
the case of a reverse pole position or current application
direction.
This is based on the non-linear magnetization characteristic of the
participating soft magnetic coil cause or the magnetic circuit and
the resulting inductance change of the coil through which current
flows. This circuit gives a rotor position determination within a
range of 360.degree. el without ambiguity.
However, here again the position determination precision is not
adequate and the typically attainable precision is limited to
.+-.15.degree. el. However, this precision is sufficient in order
to be able to rotate an appropriate motor in planned manner. Above
a certain angular speed the voltages induced in the coils are so
high, that they can be used for the "dynamic" determination of the
commutation times and an acceleration to higher angular speeds is
possible. This speed-controlled interchange between dynamic and
static position detection is also described in EP-A-251785 and the
aforementioned article by Watanabe, Isii and Fuji.
Under suitable conditions, through position-dependent variation of
the coil inductances, it is also possible to use the transformer
coupling ratio between two coils for determining the position of
the rotor. Apart from the amplitude behaviour analysis, a time
behaviour analysis can be carried out. The latter method is
described in Japanese patent 60-207489 and by Ishii and Watanabe,
Mem. Fac. Eng., Osaka City University, vol. 26, pp 57-65, 1985.
For their performance, all the aforementioned methods normally
require the use of comprehensive electronic aids.
Methods of this type are in particular used in commutatorless
direct current electric motors. As a result of their simple
construction and the resulting advantages concerning reliability
and efficiency, they are subject to wide-spread use. Essential
components of such commutatorless motors are magnets, coils (with
or without an iron core), position sensors and control and power
electronics. For controlling such motors, use is normally made of
galvano-magnetically acting position sensors, e.g. Hall generators
and the working magnets of the motor, i.e. its rotor magnets are
used as the controlling magnets for said position sensors. However,
in such cases the reliability is limited by that of the sensor
components, their feedline and by the operating temperature of the
sensors and the control magnets.
It is therefore advantageous for improving the reliability of such
devices to make no use of external position sensors. In order to
carry out a position determination with the components of the
motor, it is consequently necessary to concomitantly use as sensor
elements belonging to the system, the motor coils--in parallel to
their function of driving the motor. A simultaneously obtained
advantage is that through the omission of external sensor elements,
e.g. the Hall generators, it is possible to save space and
consequently smaller structures are obtained. For the operation of
the motor without external position sensors, the motor is
appropriately alternately brought into the state of a sensor, the
state of a motor (actuator), the state of a sensor, etc.
However, it is relatively difficult to detect the position when the
motor is stopped. The difficulties more particularly consist of
generating sufficiently precise position information within the
shortest measurement time and signal processing time from a minimum
of measurement information. In a simple case it is sufficient to
fix with a reduced precision only the commutation times necessary
for changing the current application conditions for the motor. As a
function of the motor operation requirements, it may be sufficient
to determine the rotor position with an accuracy of approximately
.+-.10.degree.. However, in other cases it is e.g. necessary for
motor regulating or control purposes to obtain extremely accurate
measured values in a very short measurement time, so that a desired
speed correction or position check is obtained. This sets increased
demands on the power of the associated electronics and software
used. Another difficulty in the known methods is to exclude
type-typical phase errors of .+-.180.degree. el in the case of a
rotor position determination. Such errors mean that if the rotor is
further rotated by 180.degree., 360.degree. el, etc., the typical
position information remains largely unchanged compared with the
original position.
SUMMARY OF THE INVENTION
Therefore a problem of the invention is to provide accurate,
inexpensive and insensitive sensors for rotary and linear
movements. Another problem of the invention is to ensure reliable
motor operation through reliable commutation processes. Another
problem of the invention is to bring about a certain starting
direction of such motors.
These problems are solved according to the invention by the
measures given in claim 1 and the following claims. The invention
makes it possible to increase the usable speed range compared with
the prior art. According to the invention using microelectronics
means and the relevant software, it is possible to obtain an
improved accuracy and reduce the relative errors during position
determination.
BRIEF DESCRIPTION OF THE DRAWINGS
The invention is described in greater detail hereinafter relative
to non-limitative embodiments and the attached drawings, wherein
show:
FIG. 1 The basic structure of the control electronics for a
motor.
FIG. 2 The basic structure of a commutating processor.
FIG. 3 The non-linear behaviour for ferromagnetic core materials
for coils and relative permeabilities/inductances.
FIG. 4 The field pattern for a d.c. motor in no-load operation.
FIG. 5 The field pattern of a d.c. motor when current is
applied.
FIG. 6 The diagrammatic construction of a reluctance motor with six
coils and a rotor with quad symmetry.
FIG. 7 A 3-T-armature with three coils and two magnetic poles.
FIG. 8 A stator of an external rotor-type motor with stator plates
and stator phases.
FIG. 9 The field pattern of an internal rotor-type motor with four
poles and six stator phases.
FIG. 10 The motor according to FIG. 9 for illustrating the magnetic
flux in different stator elements.
FIG. 11A A further flux pattern for a non-symmetrical rotor
position of the motor of FIG. 9.
FIG. 11B The construction of the rotor for a flat motor with a
different construction of the individual rotor poles.
FIG. 11C The construction of a rotor as in FIGS. 8 to 11A with
different magnetization of individual rotor poles.
FIG. 11D A motor construction with a different construction of the
air gap under individual stator poles.
FIG. 11E Diagrammatically variants for producing saturation in the
stator iron.
FIG. 12 The current application pattern for measured value
obtention.
FIG. 13 The derivation of a mean value from several
measurements.
FIG. 14 The linear arrangement of several position-dependent mean
values.
FIG. 15 The position comparison between two phase-displaced
curves.
FIG. 16a Different methods for carrying out the position comparison
of the 16C two phase-displaced curves.
FIGS 17A The spatial representation of a measurement result
comprising and 17B three individual measurements and its
evaluation.
FIG. 18 The bidimensionally oriented representation of a
measurement result with three individual measurements and the
evaluation thereof.
FIG. 19A A diagram for the current application to three coils by
means of six switches.
FIG. 19B The resulting current flow in the coils of FIG. 19A for
two closed switches.
FIG. 19C A simple current application sequence and six associated
current paths.
FIG. 19D A diagram for determining the fundamental wave of a
function and in fact graphically with six values m1 to m6
(corresponding to a motor with three or six phases L1 to L6, as
shown e.g. in FIG. 9).
FIG. 20 The representation of a vector sum, calculated according to
the above calculating specification of FIG. 19D.
FIG. 21 A flow diagram for phase and amplitude determination of the
first harmonic of a function determined by six interpolation
nodes.
FIG. 21A The approximate and periodically continued curve pattern
by six interpolation nodes m1 and m6 and the phase position of the
associated first harmonic.
FIG. 22A The representation for inductance-dependent values
relative 22B & 22C to the electrical angle.
FIG. 23 The path of a correction function for the association of an
actual value with a characteristic value of the rotor position.
FIG. 24 A flow diagram for the current application to a motor or
position sensor.
FIG. 25 A diagram of the different possibilities for controlling
the start.
FIG. 26 A flow diagram showing the sequence of the measurement of
the rotor position alternating with the current application for a
motor.
DESCRIPTION OF THE PREFERRED EMBODIMENTS
The inventive idea of solving the set problem comprises the
following parts. a) A microprocessor is used. The novel current
application necessary for measurement purposes is controlled by the
microprocessor. b) The measurement and novel evaluation of the
measured values is also controlled by the microprocessor with
associated interface circuits.
It must be borne in mind that cost-favourable microprocessors only
have a low calculating speed and cannot normally match the problem
to be solved. According to the invention for solving the specific
problem the control and measurement processes to be performed can
at least proportionally be processed by a special additional
processor. The basic structure of such an inventive apparatus is
shown in FIG. 1. Apart from the actual motor 10, it comprises an
associated current application and output stage unit 26, a
commutating processor 100 and the controlling or regulating
microprocessor 23. Hereinafter the term processor is used instead
of microprocessor. As shown in FIG. 1, the motor 10 can e.g. be a
three-phase, commutatorless, d.c. motor. However, the invention is
not restricted to such motors and, as shown in subsequent drawings,
can also be advantageously used with other motor constructions. If
necessary, the invention also covers completely renouncing motor
operation and the use of the motor exclusively in sensor operation
for determining positions. That is, the motor is operated at
essentially no load as a position indicator.
The output stage unit 26 essentially comprises a switching unit for
supplying current to the individual motor phases.
It is approximately constructed in conventional technology as an
arrangement of e.g. three transistor half-bridges. Such circuits
are e.g. known from DE-OS 31 06 856. Other embodiments, e.g. with
thyristor elements or SCR elements are also conceivable. The output
stage unit 26 has several control inputs (261 to 266) for operating
the not shown, electronic switches contained therein. These
switches supply the motor with current 10 via the individual phases
in accordance with the signal combination of the switch inputs 261
to 266.
The resulting total motor current is measured by means of a random
current sensor, e.g. the position resistor 1011 in FIG. 2 and via
the current sensor output 267 is supplied as a measurement signal
to the input 101 of the commutating processor 100.
Other elements, which are not shown in the drawings, can e.g.
comprise a regulating or control unit, which ensures that a
predetermined current intensity of a predetermined level is applied
to the individual phases of the motor 10, e.g. by using
conventional, linearly acting controller stages. However, it is
also possible to use conventional switching controllers for this
purpose.
The function of the commutating processor 100 is to accept current
sensor signals 267 and voltage signals 111, 112 and 113 of the
three motor phases L1, L2, L3 (FIG. 2) and to calculate commutating
signals therefrom. In addition, the commutating processor 100 has
analog and digital control and communication connections 102 to 110
to the controlling processor 23. The commutating processor 100 can
alone ensure a correct operation of the motor 10, i.e. without the
microprocessor 23, but in interplay with the output stage unit
26.
It is also possible in a processor-controlled operation to carry
out planned movement sequences of the motor or, whilst renouncing
motor operation, to only carry out a position detection. For this
purpose, the interplay of processor 23, commutating processor 100
and output stage unit 26 is necessary. By means of the sample and
hold signal line 105, one function of the processor 23 is to
establish times at which current sensor signals 267 are to be
determined, said signals being supplied to the processor 23 across
the current line 102. If necessary, the commutating processor 100
can also require the microprocessor 23 across a line 104 to accept
a measured current value or perform other processor activities. The
essential function of the processor 23 is to calculate the correct
position of the rotor of the motor 10 with the aid of current
sensor data obtained in this way. The thus calculated data can be
displayed e.g. on a display, but can also be supplied for further
processing to a controlling computer system. This positional
information can be supplied across the position lines 108 to 110 to
the commutating processor 100, which in turn can carry out a
correct commutation of the motor 10 on the basis of this
information.
As can be gathered from FIG. 2, the commutating processor 100
essentially comprises the following unit. The central component of
the commutating processor 100 is the commutating logic 1002, which
calculates from external, predetermined position information fed in
via the line 108-110, or from coil voltages, which are to be
measured via lines 111 to 113, firstly the commutating time, i.e.
the first time at which a commutation of the motor currents is to
take place. For this purpose a commutating switching signal or
control signal is generated, which is supplied via the signal line
261 to 266 to the output stage unit 26. In place of the direct
transmission of switching signals, such signals can naturally also
be transmitted in coded form.
The commutating logic 1002 is also influenced by a commutating
control unit 1005, which can react to different external signals.
Such signals can e.g. be the start mode signal 107 or the
microprocessor presence signal 106. Exception signals C1 to C4 can
also influence the control unit 1005, which is also influenced by
signal voltages, which can be derived from the coil voltages of the
motor 10 applied to inputs 111 to 113.
For evaluating such coil voltages, it is either possible to use
coil voltages from on-load operation or coil voltages from the
current-applied or energized operation can be investigated. For
this purpose an attenuator 1008 is provided, together with a
voltage comparison unit 1007 and a signal delay unit 1006.
Thus, the signals from phases L1 to L3 firstly reach the attenuator
108, pass from there to the voltage comparison unit 1007 and
finally from there to the signal delay unit 1006, where they are
passed on to the commutating logic 1002 and the commutating control
unit 1005. Thus, on reaching a sufficiently high induced voltage
(emf) in the phases L1 to L3, switching takes place to commutation
by the induced voltage.
Another device of the commutating processor 100 is the current
detection unit 1009, whose function is to detect, amplify and
prepare the current sensor signals 267. This takes place in that at
given times the current sensor signals 267 are amplified and
stored. In this way rapidly varying current sequences can,
so-to-speak, be frozen and supplied to the microprocessor 23 across
the current output and the current signal line 102 for further
processing. The processor 23 then has sufficient time to evaluate
these current signals, because the stored signals do not initially
change further. In addition to the internal switching means, the
commutating processor 100 normally has external switching means,
such as resistors and capacitors, which are not shown in FIGS. 1
and 2.
Further details of the commutating processor 100 are a starting
oscillator 1004 and a counter 1003 for the random production of
preset times or preset commutating patterns. The actual function of
the commutating processor 100, without the associated
microprocessor 23, is, as stated, the evaluation of the induced
voltages (emf) at the motor phases L1 to L3, such as are generated
e.g. under no-load conditions.
However, this operation is only practicable, if the motor rotor is
already moving. Even a position measurement process using these
signals requires a moving rotor for correct operation using
them.
In the case of a motor stoppage, either an unidirectional starting
movement is to be initiated, or by means of a special current
application to the motor phases L1 to L3 and measurement of the
individual phase currents, the necessary measurement information
must be obtained. The investigation of the path of the phase
currents can vary widely and the simplest possibility consists of
evaluating the time pattern of the motor sum current.
It can also be advantageous according to the invention to carry out
a current application for several stator phases for measurement
purposes and to record the current intensities at different times.
As a function of the intensity of the processors used for this
purpose, i.e. the microprocessor 23 and/or the commutating
processor 100, either only one or more current values at specific
times and in particular current maxima are recorded, or times
associated with clearly defined current thresholds are recorded, or
one or more current paths are recorded on a time and intensity
basis by means of several measurement points in each case. The thus
recorded values constitute implicit numerical values for the sought
and rotor position-dependent inductances or mutually inductances of
the motor phases. Under appropriate conditions these can be numeral
values for transformer coupling ratios.
As shown in FIG. 3, a position information obtention by appropriate
current application to stator phases is brought about in that
through a current application of differing levels the iron
components of a magnetic circuit belonging to the coils of this
phase are brought close to or at saturation. Thus, there is a
change to the overall inductance of a participating coil, because
according to the coil formula L=n.sup.2.mu..mu..sub.0F/d under
otherwise constant conditions generally only the .mu. changes and
consequently any change to the relative permeability leads to a
proportional change to the inductance L. As is the case with iron,
the non-linear connection between the field strength H and magnetic
flux density (induction) B can be used for establishing
current-dependent, effective inductances of the participating
stator coils.
However, the inductance of these stator coils is not only
influenced by the coil current level, but also by the polarity of
the permanent magnet of a rotor, said permanent magnets facing the
stator coils. This is due to the fact that through the permanent
magnets, there is a premagnetization of the coil cores, so that as
a function of the polarity of the coil current there is a
saturation of the coil core either faster or slower than in the
neutral case, i.e. without permanent magnetic premagnetization. In
the case of a conventional design of magnetic circuits of motors of
a general type an attempt is generally made to keep the magnetic
flux density in the iron parts well below saturation, so as to keep
the stray flux components low. As the change to the stator coil
inductances caused by nominal current application and moving the
rotor is normally only a few percent, it is important for
troublefree measurement data determination for position detection
purposes to use additional measures to bring about a maximum change
to the inductance over a motor movement range of 360.degree.
el.
For this purpose it is e.g. possible to modify the motor magnetic
circuit. An optimization with a view to a large variation of the
coil inductances as a function of the rotor position can possibly
lead to a reduction of the maximum motor efficiency.
Thus, according to the invention it is advantageous to apply
current to several and not a single coil. It is also advantageous
to apply a relatively high current compared with normal motor
energization and for a short time, i.e. a few dozen microseconds
(e.g. 10 to 2000 microseconds).
Correspondingly it is advantageous according to the invention to
apply current to a combination of coils in a way not normally used
for drive purposes. If such a current application is time-limited,
then there is no need to fear significant motor efficiency losses
in motor operation.
For example, in an embodiment of the invention it is possible to
proceed in such a way that two stator coils of a three-phase motor
are briefly supplied with a very high current and intermediately
the third stator coil is supplied with a high, short current pulse.
The current-time information obtained in this way normally yields
more than comparable information obtained with conventional means
and prior art methods.
In order to obtain the indicated differences in the inductance, as
stated hereinbefore, it is useful to supply current to some of the
coils present in such a way that their ferromagnetic cores are
close to or at saturation. An additional premagnetization by
permanent magnets within the magnetic circuit or circuits can, as
stated, by helpful, but is not a prerequisite.
In the case of a motor, FIG. 22A shows the path of a coil
inductance over the rotor rotation angle. As a function of the
motor construction, there are variations to the inductance of a few
to several percent. As can be gathered from FIG. 22A, in such a
motor type this function is periodic with a period of 180.degree.
el.
In the presence of permanent magnetic premagnetization said curve
becomes asymmetrical (FIG. 22B) and then has a period of
360.degree. el, so that clear measurements are possible in this
angular range (360.degree. el). The path of the curve in FIG. 22B
is dependent on the strength or intensity of the premagnetization
(through a permanent magnet) and can be further amplified by
additional current application to further coils. The coils to which
the current is applied in this way are partly saturated.
FIG. 22C shows the same conditions as in FIG. 22B, i.e. the
inductance plotted in polar coordinates and a function of the motor
rotation angle.
This path is similar for each of the motor coils. However, compared
with an adjacent phase, there is a corresponding phase displacement
of 360.degree. elm, in which n is the number of motor phases. This
means the following connection found according to the invention. If
a motor e.g. has six phases and all of these successively carry out
almost simultaneously inductance measurements, the measured values
obtained also define a curve, which can also be plotted in polar
coordinates (FIG. 22C). Independently of the rotor position, all
curves determined in this way have the same basic shape. However,
they are rotated about a phase angle compared with a standardized
representation, i.e. the curve associated with a basic position.
Simultaneously this phase angle is surprisingly also a measure of
the rotor position. It is clear that the dependence of the
inductance of a phase on the rotor position should be as great as
possible, so as to be able to perform a clear and unambiguous
measurement.
An additional inventive possibility for bringing about this
dependence comprises designing rotor permanent magnets in such a
way that there is a non-uniformly high effective field strength of
these permanent magnets. This can e.g. take place in that on
magnetizing the rotor a non-uniformly high magnetization is carried
out or, after magnetization, there is a planned demagnetization of
individual magnetic poles.
Another possibility consists of following one combination of
equally strong north and south pole rotor magnetic poles by another
identical combination, but whose effective field strength is
reduced by a constant amount.
These connections and associations are further explained relative
to FIGS. 11B and 11C. In FIG. 11C e.g. the rotor magnetic poles
501, 502 are more strongly magnetized than the rotor magnetic poles
503 and 504.
In place of differently strongly magnetized permanent magnets of
the rotor, alternatively or additionally it is possible to use an
arrangement of varyingly large air gaps 1121, 1122, cf. FIG.
11D.
The same effect is obtained when using an arrangement of magnetic
partial resistors, which have a different size. This can e.g. be
set up in that the ferromagnetic coil cores are given a different
size or in that yoke elements of the magnetic circuits are given a
different magnetic resistance, either by varying the length or
cross-section, or by the insertion of air gaps. The embodiment of
FIG. 11E represents a linear motor, which has as the rotor element
a permanent magnet with north poles 1101 and south poles 1102.
Opposite the magnetic poles are provided the stator unit 1103 with
stator pole shoes 1104. By means of specially shaped stator webs in
elongated form (1105) or short form (1106), the stator pole shoes
are connected to the return plates 1108.
The stator webs are wound with corresponding coils 1107. Compared
with the normal cross-section, the return plates 1103 can have a
reduced cross-section 1109 or an increased cross-section 1110. It
is also appropriate to insert an air gap 1111. The advantage of
these arrangements is that through the varyingly high magnetization
within the magnetic circuit, an additional, position-dependent
information (encoding) is introduced into the motor or position
sensor. This permits a more precise and geometrically more
extensive position determination, which allows an absolute position
determination within the full rotation angle of 360.degree. el. The
inventive procedure, according to which by high and pulse-like
current application certain ferromagnetic components of the
magnetic circuit are brought close to or into magnetic saturation,
naturally presupposes that the permanent magnets are provided with
a sufficiently high coercive field strength, because there must be
no demagnetization of the magnets as a result of the current
application for position measurement purposes. Thus, when using
alnico magnets, particular attention must be paid to this point.
When using modern magnetic materials, such as e.g. rare
earth-cobalt materials or neodymium-iron-boron materials no
problems occur in this connection.
Before calculating an actual position of the rotor, it is firstly
necessary to obtain a sufficiently large number of measured values.
It is appropriate to separate the determination of the measured
values and the associated evaluation, so that measured value
determination takes place as quickly as possible. The faster the
motor rotor moves, the greater the attention which must be paid to
this requirement. A mixing of measured values from different
positions must also be expected, so that a clear reconstruction can
only be obtained with difficulty or maybe impossible. In the
evaluation phase time restrictions of this type are not as
critical. However, it is favourable to limit the evaluation phase
to a time minimum, so as to obtain maximum comprehensive
measurement data through a maximum number of measured values per
time unit.
Although other procedures are possible, the following is the most
appropriate for determining the measured values. Current is applied
to a predetermined stator winding with the aim of obtaining high
magnetization of its ferromagnetic core. The current in this stator
winding is recorded at several times. This current flow is e.g.
designated 11 in the pulse sequence 1 in FIG. 12. For example,
besides continuous current recording, only a single measurement
point M1 may be observed. For this purpose the associated current
value is determined, e.g. following a predetermined time difference
t1-t0. It is alternatively possible at a predetermined current
value, e.g. half the maximum current value, to determine the time
requirement until the current has risen from 0 to this value. (The
current I1 is connected in at t o and measured at t 1). In parallel
or with a slight time displacement, i.e. in the range of
approximately 10 to 1000 microseconds, this process is repeated in
a further stator coil, without the first stator coil being
disconnected.
FIG. 12 shows this time connection for the current application to a
selected number of three coils, whose current is designated I1, I2
and I3. If there are further coils, it is possible to use for a
more extensive information obtention, one or more additional
processes of the same type on a fourth or further stator coils.
After current has been supplied to all the stator coils in
question, disconnection either takes place completely and
simultaneously or, as shown in FIG. 12 for the pulse sequences 1,
2, and 3, there is a clearly defined disconnection procedure. Also
in these two cases it is possible to perform time and/or amplitude
measurements, e.g. the amplitudes M4, M5 and M6 at the times t4, t5
and t6.
Thus, it is e.g. possible to switch the coils in currentless manner
in the reverse order compared with the connection frequency. In
this way in a first pulse sequence 1 is obtained a first set of
measured values, which in the simplest case consists of a single
value. If two measured values are obtained per coil, as shown in
FIG. 12, then for a motor with three coils there are six measured
values M1 to M6. It is also naturally possible to obtain additional
measured points from rising and falling slopes and to allow them to
influence the further calculation. Normally attempts are made to
select a few representative measured values from the plurality
thereof.
With an individual, first pulse sequence it is not generally
possible to carry out a position determination and further measured
value sets must be obtained for this purpose. The associated
procedure is illustrated in FIG. 2 with the current application
curves of pulse sequences 2 and 3. This constitutes a similar
process to that for pulse sequence 1, but the current application
commences with a stator coil close to the coil 1.
For a motor with three coils, current application can e.g. commence
with coil 2 and shortly thereafter current application to coil 3
and then coil 1 takes place. Correspondingly to the current
application process during the pulse train 1, also during the pulse
train 2 the current application to the individual coils is broken
off at given times. In a pulse train 3 the procedure is repeated
again, but beginning with coil 3, followed by coil 1 and then coil
2. If there are more than three coils, then more complicated
application sequences can be performed. Therefore the specification
for the quasi-simultaneous current application to further coils
will correspondingly relate to coils having comparable spacings
with respect to the next coil to which current is to be applied.
Thus, after the start of current application to one coil, current
can be applied to a directly adjacent coil.
The second current application process can also relate to a coil
which is at some distance from the coil to which current was first
applied. A fixing of the most useful coil current application
sequence must take place individually as a function of the motor
type. In order to obtain complete data the process is repeated
several times. Current application begins in each case with a
different stator coil. Current obtention is complete if there is a
cyclically produced measured value set on the basis of the same
diagram for all the motor stator coils.
However, it is also possible to carry out a simpler position
determination with reduced precision. For this purpose it is
sufficient to have a smaller data quantity, which only comprises
measured value sets emanating from fewer stator coils than the
number of the latter actually present in the motor or position
sensor. For three-phase motors with e.g. six stator coils, it is
e.g. merely sufficient to analyze three measured value sets, it
being advantageous if said sets are obtained for every other
coil.
After a predetermined number of measured value sets has been
obtained, evaluation is carried out (as a function of the
efficiency of the evaluation unit evaluation can naturally also
take place during the obtention of the measured value sets). As a
function of the necessary speed with which the simply structured
data sets are to be evaluated, for the measured value analysis it
is preferable to use a special microprocessor or controller 23,
whose program is tailor-made to the position detection problem to
be solved, but which can also be used for regular motor operation.
For this purpose it is e.g. possible to use a standard processor 23
with an associated, externally positioned expander circuit or
commutating processor 100.
Another possibility consists of combining units 23 and 100, so that
a standard processor unit is present on the same silicon surface,
which is supplemented by customer-specific additional functional
units.
For mean value formation, the measured values determined in a pulse
train, e.g. 1 in FIG. 12 are appropriately combined to form a
single, representative measured value. As the measurement accuracy
of the individual measured values M1 to M6 can be of varying
levels, it is appropriate to carry out a weighting in the case of
the necessary mean value formation. This can e.g. consist of
calculating a mean value m1 from the measured values M1 to M6
according to the following or a similar calculating rule (FIG. 13),
the weighting factors having to be individually determined, e.g.:
m1=(0.8*M1+0.3*M2+0.1*M3+0.7*M4+0.3*M5-+0.1*M6)/6
A similar mean value formation with the same coefficients can be
used for the pulse train 2 on the measured values M21 to M26, which
gives a mean value m2. The same applies with respect to the pulse
train 3, where the measured values M31 to M36 must be multiplied
with the same coefficients, which gives a mean value m3. The mean
values m1 to m3 implicitly contain the sought position information,
which must still be calculated therefrom in explicit form. FIGS. 14
and 15 (lower half) show in linear from the amplitudes of the
calculated mean values m1-m3. In the specific case it can also be:
m1=M1, m2=M21, m3=M31, etc.
If there are further pulse trains, e.g. not shown pulse trains 4,
5, 6, etc., then the associated mean values m4, m5, m6, etc. are to
be determined by comparable procedures.
As the evaluation of values determined in this way has surprisingly
shown, through each measured value multiplet obtained in this way
it is possible to place a substantially identical periodic curve K,
as shown e.g. in FIG. 14 or by reference 161 or 163 in FIG. 15, but
whose phase position must be varied until the measured values, here
e.g. m1 to m3, are located on the curve (FIG. 15 and FIG. 16A). The
measure of the periodicity for this curve is designated 144 in FIG.
14. The measure of said phase displacement constitutes a
characteristic value for the actual motor rotor position.
One form of determining six values m1 to m6 is shown in FIGS. 19A
and 19B. Three motor coils L1, L2, L3 with terminals R, S and T are
supplied with current via a switch set SW1 to SW6 (reference
numerals 178 to 183). In each case one of the switches SW1 to SW6
and one of the switches SW4 to SW6 is simultaneously closed. Thus,
there is a current flow from the supply line 175 across the coils
and the current sensor 177 to the ground potential line 176, the
magnitude of the current being determined via the sensor line 184.
The resulting current flows are defined in FIG. 19B. In the
crossing points of rows and columns, the current flow is given for
in each case two switches and in this way six switching
possibilities are provided. A current application for determining
the rotor position then takes place e.g. in accordance with FIG.
19C.
Firstly SW1 and SW5 are used for producing the current flow i1
which, rises in accordance with a rising slope 190, whose intensity
is measured after the time interval 193 and gives a value m1. On
reaching a predetermined current application time the current is
interrupted by opening the switches, which leads to a disconnection
slope 191. This procedure with relatively fast connection and
disconnection times is repeated according to the other switch
combinations shown in FIG. 19B for five further current application
i2 to i6 and in each case following the time interval 193 the
associated value m2, m3, . . . , m6 is determined and buffer stored
for further evaluation. Although this process is somewhat more
complicated than the obtaining of only three values m1, m2 and m3,
the rotor position determination precision is increased. As stated,
the aim is to place a known curve through the measured values
obtained in this way and the phase is to be displaced in such a way
that the measured values are located on or as close as possible to
the curve.
This phase displacement can be mathematically determined in several
different ways. One possibility is e.g. to initially determine a
curve K, which interconnects the measured values in an optimum
manner. This procedure has the advantage that on comparing a curve
found with a reference curve not only a few, but a random number of
points can be compared with one another (FIG. 14). In this way the
individual points of a curve found can be displaced in fine steps
until both curves to be compared have an almost congruent
position.
In order to generate a very smooth curve from a measured value set
with e.g. three or six or further mean value points, different
known calculating rules can be used. One of these consists of using
mean values m1-m n obtained as input variables for a Fourier
analysis. These mean values are multiplied with suitable, complex
coefficients, e.g. the nth unit cubes and are subject to complex
summation. The process is performed individually in per se known
manner for all the spectral components. These spectral components
are sine or cosine terms, by means of which it is possible to
construct a relatively smooth connecting function through the
measurement points. The precise functional representation of the
connecting curve is, in known manner, obtained by a so-called
inverse transformation.
However, according to the invention it is advantageous in
accordance with relevant known calculating instructions, to
represent a so-called spline function, which represents an
optimized continuous connection between all the points of the
measured values set. The generation of such functions differs
compared with the first-mentioned process in that the spectrum
obtained is modified in a planned manner, or the functions are
approximated e.g. by third degree polynominals. In this way a
limited number of points n1 . . . n m of a mean value set is
transferred into a random number of points of matching spline
functions.
For carrying out the position-characteristic determination
comparison is necessary between the reference curve 161 and the
actual curve 163 defined by the actual measured values (FIG. 15).
This is performed in the following way. Both curves, referred to
hereinafter as measurement function and reference function, are
periodically supplemented outside the periodicity range 144 by
others having the length of said range 144. The two curves are
checked for correlation, i.e. coincidence or similarity, with the
aid of a conventional calculating process.
According to the invention this most appropriately takes place by
the use of the method of least squares (FIG. 16B), where the
measurement curve is 163, the reference curve 161 and their maxima
165 or 166 have the spacing d, 167. For an agreed number of
measurement and reference function values, the difference between
in each case two associated function values is determined and
squared (curve 164). The result is added to a sum total, i.e. the
curve 164 is integrated between 0 and .times.1. After this
calculation has been performed for each function value pair, the
similarity is detected by means of the sum total. If the sum found,
i.e. the integral of the curve 164, is zero, then identical
functions are present and the phase difference is zero. Therefore
small sum values represent a large similarity of the functions,
whereas large sum values represent small similarities of the
functions.
It is obvious that in the case of the aforementioned summation,
summation or integrated summation must take place over a standard
interval 0 to .times.1, which is at least as long as that defined
e.g. by the reference function.
The position-characteristic determination is now carried out in
such a way that the first summation process is followed by others.
However, one of the two functions, including their supplements or
additions on both sides, are successively further displaced by a
phase angle xi 114 (FIG. 16C). For each phase displacement having a
minimum summation result, the correlation is at a maximum, so that
a numerical value for the phase displacement found in this way can
be used as the rotor position characteristic.
For example, in FIG. 16C with the phase difference xi 4, an
approximately complete coincidence of the curves 161 and 163
exists.
Another process consists of reciprocally displacing the reference
and comparison functions until the sum of the absolute quantities
of the differences of a number of function values of the reference
and comparison function is as small as possible, this process being
illustrated in FIG. 16A. The two functions to be compared have in
each case a maximum 165 or 166 and the phase displacement is
designated d. Through the function differences surfaces are
generated and the resulting overall surface also constitutes a
measure for the congruence of both functions.
As in the previous case, a minimum comparison amount, namely the
integral of the function 168 between 0.degree. and 360.degree. el.
amounts to a good coincidence of the curves. With a complete
coincidence the difference surface amount is 0. The numerical
values of the rotor position obtained in this way are clearly more
accurate than the determination of a maximum value of different
interpolation nodes of the function, as described in EU-A-251785.
However, the calculating expenditure and effort for obtaining this
improved precision is greater, because firstly it is necessary to
determine the spline function values for interpolation nodes and
then the optimum curve position must be determined by iterative
curve comparison operations.
Both processes can be combined to solve a pattern detection
problem. However, according to the invention there are further
solutions for this problem.
Another such process is based on the centroid analysis method. For
this purpose the values m1, m2, etc. are represented in graphic
form, so that a surface is produced in the form of an irregular or
regular polygon. In a particularly simple case a position
characteristic is to be calculated from three values only, i.e. the
mean values m1, m2 and m3.
The associated inventive rules for calculating a position
characteristic in this case comprise the three measured values m1,
m2, m3 being plotted as axis portions in a three-dimensional
cartesian coordinate system. Such a coordinate system, which is
spanned by the three unit vectors i (801), j (802) and k (803), is
shown in FIG. 17A. The latter also shows the space diagonal 804
passing through the origin and the point with the coordinates i, j,
k, i.e. the space diagonal is defined by the origin of the
coordinate system with the associated unit vectors i, j and k (801,
802, 803) and the vector sum i+j+k.
A first, second and third measured value m1, m2, m3 of a trio of
values is in each case plotted in the direction i, j or k. In this
way a triangle 81 is defined, whose (surface) centroid S, 82, as a
function of the trio of measured values to be used as a basis, is
of varying form, because in the case of a different position of the
rotor position to be determined different trios of measured values
have to be evaluated. This surface centroid S can be determined by
drawing or also by calculation/mathematically and is located on a
curve 810 (FIG. 17B), which essentially passes in cycloid manner
around the space diagonal 804 and which is shown on a larger scale
in FIG. 17B. The space diagonal 804 passes through the triangle 81
in a centre Z, reference 83.
Together with the centroid 82, which varies as a function of the
measured values, the centre 83 defines a straight line 804 located
on the triangle 801, based on a straight line 806 between the
center 83 and an angle point or corner, e.g. m1 of the triangle 81,
a clearly defined angle 807 in the plane of the triangle. This
angle 807 is used as the position characteristic for the rotor
position and further processed.
In place of the three-dimensional representation form, said three
measured values can also be evaluated in a bidimensionally oriented
representation (FIG. 18). For this purpose the three axes of space
are projected onto three lines, which emanate from the origin of an
x-y coordinate system and in each case have an angle difference of
120.degree.. Starting from the origin, distances are plotted in the
direction of these axes and are proportional to the mean values m1,
m2, m3. In this way a triangle is spanned, whose centroid S does
not necessarily coincide with the origin 1802 of the three lines.
In known manner, this centroid can be constructed by drawing or
mathematically. As a function of the shape and position of the
generated triangle, a line between the centroid S and the origin of
the three axes will assume a different angle 1801 with respect to
one of these three axes.
For determining the position of the rotor of an electric motor use
is directly made of this angle. The distance between the centroid
and the origin only plays a minor part and can optionally be used
for correcting the rotor position determination.
The position of the centroid can be determined in known manner by
construction of the median or alternatively using the centroid
formula, which must here be applied e.g. to three equal-weight
elements m1, m2 and m3. According to this calculating rule the
portions in the X and Y direction (FIG. 18) can be separately
calculated.
A closer consideration of this process reveals the identity of the
calculating form with a further process.
As in FIG. 18 only the angular position 1801 of the centroid S with
respect to a line 1803 emanating from the origin 1802 is sought, in
the case of the mathematical determination of the centroid there is
no need for the otherwise necessary division, in this case by
three, so that for determining a X component of the centroid
direction it is merely sufficient to add the X values of the angle
points of this triangle, which also applies with respect to the Y
values thereof. Put differently, this means that the vectors m1, m2
and m3, passing from the center 1802 to the angles of the triangle,
must be added up and that the resulting vector of the vector sum
indicator the direction which substantially coincides with the
sought electrical rotor position.
In addition, the aforementioned centroid analysis process can be
looked upon as a special case for all processes based on Fourier
analyses and transformations. Also in the case of such analyses and
transformations a corresponding vector summation is carried out for
the values for the amplitude and phase of the first harmonic of a
function defined by several interpolation nodes.
Together with FIG. 20, FIG. 19D shows a corresponding example for
the evaluation of e.g. three or six stator coils, for which the six
values m1 to m6 are to be determined in accordance with the
procedures represented in FIGS. 12 and 13. As explained, m1=M1,
m2=M21, m3=M31, etc. FIG. 19D shows the "complex plane", in which
the x axis corresponds to the real part and the y axis to the
imaginary part. FIG. 20 can also be looked upon as a complex
plane.
Apart from the coordinates of the complex plane, FIG. 19D shows all
six vectors v1 to v6 of the sixth unit cubes, the latter being
understood to mean a solution x, which fulfils the set problem x
high n=1.
The first harmonic, i.e. the sinusoidal fundamental wave, which is
associated with a predetermined number of measured values of a
periodic function, is calculated in amplitude and phase by the
addition of associated complex numbers. This addition is carried
out in much the same way as the addition of vectors and is e.g.
represented in FIG. 20 for a so-called 6-point DFT (discreet
Fourier transformation). If there are fewer or more measured values
(measurement points), correspondingly modified procedures apply, as
are known from the theory of Fourier transformations.
In a first calculating step of said procedure firstly the measured
values, which are generally present as real quantities, e.g. as
measured current values, are to be converted into the correct
complex numbers, as is illustrated in FIG. 19D.
If there are e.g. six values m1 to m6, then on the unit circle in
the complex plane firstly the six so-called sixth unit cubes v1,
v2, v3, v4, v5, v6 are plotted, for which the equation (v.sub.k)
6=1 is fulfilled, in which k=1, 2, 3, 4, 5, 6. The angle phi o
shown in FIG. 19D is 60.degree., so that said unit cubes can also
be represented according to the formula exp(k*i*phi o), in which k
has a value of 1 to 6. In a further step for the six values m1 to
m6 a linear stretching of the unit cubes takes place in a
successive order and in each case by the particular factor m1 to
m6, so that the points m1 to m6 of FIG. 19D are obtained in the
complex plane. The vectors emanating from the coordinate origin up
to the said points m1 to m6 are shown in FIG. 20, but in a
different form.
The expression "stretching the unit cubes" is understood to mean
that they are multiplied by the value m1 to m6. For example, in
FIG. 19D the value m1 is greater than 1 and therefore the
particular vector m1 is greater than 1 (from the amount
standpoint). Therefore m3 is smaller than 1 and the particular
vector m3 becomes smaller than 1. The vectors m1, m2, m3, m4, m5,
m6 are given the reference numerals 201 to 206 in FIG. 20. To
illustrate their vector sum they are also strung together in
chain-like manner.
In order to illustrate the associated calculating rule for the
arithmetic summation of these vectors their x and y parts are shown
and designated x1, x2, y2, x3, y3 etc. The x component of the
resulting vector is designated x7 and is formed from the x parts
shown, whilst the y component is formed from the y parts and is
designated y7. The resulting vector has the end point Z 0, which
can once again be looked upon as a complex number with the real
part x7 and the imaginary part y7. The amount R and the phase phi
of said complex number can be determined according to known
calculation rules from the real part and the imaginary part, e.g.
the amount of R is equal to the square root of the sum of the
squares of the real part and the imaginary part, i.e.
##EQU00001##
An associated calculating rule for the processing of the values m1
to m6 by a digital computer, e.g. a microprocessor, is given as a
flow diagram in FIG. 21. As is conventional with such processes,
numerous other procedures are possible.
In a first step S40 are known values m1 to m6 determined by another
program part are filed in a memory, e.g. a stack memory. In the
following steps S41 and S42 different quantities are given initial
values and in particular the storage locations with the allocation
addresses SUM_R and SUM_I are set to zero and form the memories for
the real and imaginary parts. In the following steps S43 to S58
values are added or subtracted in these memories, being derived
from the values m1 to m6. Thus, in step S43 and S44 initially the
value m1 is retrieved from the stack memory and then added to the
content of the memory SUM_R, which then assumes the value m1.
In step S45 the value m2 is obtained from the stack memory and is
divided by two. This result is added to the content of the memory
SUM_R and multiplied by the factor 3 and then added to the content
of the memory SUM_I. This process is carried out in similar form in
the following steps and in steps S51 and S52 the division by the
factor two is omitted and the procedure is much as in steps S43 and
S44, but m4 must be subtracted from the content of the memory
SUM_R. In the other cases it is firstly necessary to carry out a
division by two in order to obtain subtitles, which are
subtractively or additively added to the memory SUM_R, or to the
memory SUM_I following multiplication by the factor 3. This
totalling up or down of the memories SUM_R, SUM_I are then
concluded with the step S58.
The values in memories SUM_R and SUM_I consequently represent the x
component x7 of the vector R in FIG. 20 and also the y component y7
of the same vector.
The further calculating steps of FIG. 21 are intended to determine
the angle of this vector relative to the x axis and also its
amount. According to steps S59 to S64 of FIG. 21 for determining
the amount initially the contents of the memories SUM_R and SUM_I
are individually squared and then from the squares is formed the
sum. From this sum is drawn the square root and this value
represents the sought amount of the vector and is therefore stored
in the storage location designated BETRAG or amount. For
calculating the phase angle phi relative to the x axis, in steps
S65 to S68 firstly the contents of the memories SUM_R and SUM_I are
obtained and the sign of these values is noted. In accordance with
known trigonometric procedures the sought main angle value phi is
calculated from the quotient of SUM_I/SUM_R and the arc tan
function used thereon. The final angular value is then determined
in accordance with known transformations on the basis of the stored
sign values within a range 0.degree. to 360.degree..
The above facts can also be described in shortened form by the
summation formula Z.sub.0=.epsilon.m.sub.kexp(i*k*phi.sub.o) and
the sum Z.sub.0 consequently gives through its real and imaginary
parts the amount and phase of the first harmonic or fundamental
wave, i.e. the fundamental wave of a periodic function defined by
the measured values m1 to m6.
The last-described process can be compared with calculation
procedures and processes of discreet Fourier transformation
(DFT).
This is illustrated by FIG. 21A, where at A) are shown the values
m1 to m6 in the form of 6 interpolation nodes of a periodic
function 230, which are subsequently repeated. Comparison shows
that the last-described process can be looked upon as DFT for the
determination of the amount R and phase phi o of the first harmonic
of this function 230, which is also defined by equidistant
interpolation nodes and which are repeated outside the
interpolation nodes in periodic manner, i.e. continued in the same
form.
FIG. 21A shows a B) the determined first harmonic with its
amplitude R and its phase angle phi o, which essentially gives the
rotor position and which is e.g. calculated according to the flow
diagram of FIG. 21.
As has already been stated, the angle or phase phi of this first
harmonic graphically represented in FIGS. 20 and 21A and
calculatable according to FIG. 21 is of interest and represents the
sought position characteristic for the motor rotor position.
It is obvious that in place of the indicated example similar
mathematical procedures can be used, as are known from the theory
of Fourier transformations and in general integral transformations.
In particular in the case of a larger number of interpolation
nodes, it is possible to use a related algorithm, which is known as
fast Fourier transformation (FFT) and is suitably for saving
mathematical effort. As a function of the process, it may be
necessary to respect restricting conditions, which are well known
to the Expert in this field.
On comparing the described process for determining the sought phase
position of a function, it becomes clear that the processes based
on integral transformations or on centroid analyses are
characterized by a significant mathematical or calculating time
saving. For a simple determination of the position of a rotor, the
amount of R is not required (FIG. 20 or S64, FIG. 21).
A check on this value regarding the exceeding of previously
determined upper and lower limits, can be used to establish whether
there is an overall plausible calculating or mathematical result
for the determination of the angle phi or whether it is necessary
to assume any type of error influence. If the latter need not be
feared, the amount can also be used for correction during the
determination of the rotor position, as explained hereinafter.
For the precise determination of the position of the rotor with
relative errors in the percentage range a substantially linear
association or connection is necessary between the actual angle or
position of the rotor and the calculated angle or position value.
As the measurement is based from the standpoint of its principle on
non-linear physical effects, the initially attainable results are
usually not linear and can therefore only be referred to as
position characteristics. According to the inventive idea it is
advantageous by means of a table previously determined in an
experiment to utilize correction values, which can be also plotted
in a diagram or graph, as shown in FIG. 23.
The latter shows the cartesian coordinates "position actual value"
and "position characteristic value". The functional association of
the actual and characteristic values is given by the assignment
curve 72, as shown by the example of the abscissa and ordinate
values of the function value 71. The assignment curve 72 generally
does not have a linear path and consequently differs from a
straight line 73 with optimum matching to the function values. The
deviations of individual interpolation nodes of the function 72,
represented on the example of the paths 71 to 76, can be kept in a
correction table. According to a known mathematical process, one or
more matching lines can be implied through a number of
interpolation nodes.
For defining a matching line 73, which need not necessarily be
characterized by a best matching to the interpolation nodes, as is
known it is sufficient to have the coordinate intersections 74 and
75.
For each position value determined according to one of the already
indicated processes, it is possible to calculate an actual position
value using such a correction table and conventional interpolation
processes. In place of a correction table, it is possible to give
for different motor types a correction formula, which is defined by
one or more sine terms and optionally associated phase terms.
In practice, maximum correction values of a few to a few dozen
degrees el can occur. A typical value represents a sinusoidally
varying deviation, which assumes a maximum of approximately
20.degree. el and which can be reduced approximately by a factor of
10 by the indicated correction calculation.
In a further refinement of the correction measures it is naturally
also possible to introduce further parameters into the
calculations, such as e.g. temperature values of the motor coils or
the magnet.
Additional correction information can be determined from the
previously not considered higher harmonics of the fundamental
measured value sequences, provided that correspondingly large
numbers of measurement points are available.
In addition, a further correction function of this type can be
derived from the amplitude of the first harmonic. An additional
correction function or table is required for this procedure and the
sought rotor position is then calculated from the angle phi (FIG.
20) and an additive or multiplicative correction value, which is
directly dependent on the amount (amplitude) of the first harmonic.
Correction functions or tables are to be fixed by an experiment
carried out beforehand.
With respect to the commutation strategy for sensor-free motors,
the decision sequence of how the motor commutation is to be carried
out, is illustrated by the flow diagram of FIG. 24.
It is firstly necessary to establish whether or not the stepping
motor can be started, because the entry into a basic position can
take place by an inadmissible rotation direction (FIG. 24, S1). If
stepping motor operation is permitted, then e.g. a stepping pulse
is supplied (FIG. 24, S2), which increments the counter 1003 (FIG.
2). In step S3 the commutation logic 1002 produces a new
commutation pattern, which is supplied to the output stage unit 26,
which in a step S4 ensures the energizing of the motor. In the
following step S5, it is established by emf observation whether the
rotation direction is correct. If this is the case, in step S6 the
rotation direction can be maintained, otherwise commutation
sequences must be ensured which bring about a rotation direction
change. This sequence is initially repeated until a first speed
limit is reached. If alternatively the motor start in the correct
rotation direction is immediately necessary, steps S7 to S9 are
performed. Thus, by the rotor position determined in step S7, as is
also the case in the regular operation of a commutatorless d.c.
motor, it is possible in one step S8 to generate and emit the
necessary correction pattern, which brings the motor into the
correct rotation during the step S9. A decision is made in the step
S10 as to whether this process must be repeated or whether a speed
limit has been reached.
On reaching this first speed limit, through steps S11 to S14 a
further motor acceleration is ensured. This is achieved in that
after detecting the zero passages of the motor counter-emf, at an
appropriate time and in accordance with a new-calculated
commutation pattern, power is supplied to the motor. In this way
the motor is run to the maximum attainable speed.
The motor speed or position is regulated or controlled according to
the inventive process in the following way. The precise
determination of the position of a rotor using the inventive
process can be performed with a sufficiently high repetition
frequency, e.g. 100 times per second, in order to record deviations
from a desired motor rotor position and to keep same as small as
possible by an active counteraction (control principle).
As the position values can only be determined at discreet times,
said control may only be a scanning control. Numerous publications
are known in this field.
For scanning controls the attainable control quantity is influenced
by the scanning time and also the different system time constants,
as well as by the control algorithm used.
It is therefore a further object of the invention to use the
position determination of the rotor according to one of the
indicated processes in order to either carry out a speed or a
position control of the rotor. Preferably use is made of a
controller structure with PID algorithm or also a state controller
algorithm.
For control purposes it is also possible to previously provide a
program or externally preset the course of the speed or sequence of
positions to be assumed by the motor rotor as a function of
time.
Another inventive process for controlling the speed comprises only
carrying out one rotor position measurement and then carrying out
the current application steps. The duration of the latter will
generally depend on when the next motor commutation is to be
carried out and will constantly decrease on accelerating the
motor.
According to the inventive idea an advantageous process when
calculating the necessary pulse shortening is to take account of
the number of already performed current application cycles.
Generally, with increasing rotor speed, the pulses are
shortened.
In place of the dependence of the pulse shortening on a number of
current application pulses as an implicit measure of the time, such
a dependence can also be linear and explicitly dependent on
time.
According to the invention, according to FIG. 25 another
possibility consists of making the current application period
duration linearly dependent on the obtained overall angle or
position, e.g. (starting with a relatively large value for the
current application time), in accordance with the already attained
position, the current application time is continuously and linearly
(131) reduced, up to e.g. a minimum value, below which no further
drop must take place.
It is also possible to perform these procedures in non-linear form.
This can in particular be brought about in the form of a
logarithmic or a hyperbolic dependence (132). If necessary, more
complicated functional dependencies can be used.
As a function of the motor type and the coupled load, particularly
with a resiliently coupled load, it can be advantageous to reduce
the current application (tau) according to a tabulated dependence
(FIG. 25, 133), i.e. the necessary current application times are no
longer calculated on the basis of a mathematical expression, but
are instead filed in table form, e.g. in a memory component. With
increasing rotor angle or time, in each case new current
application time information is obtained from the memory component.
This process is performed until a minimum current application time
is reached, which is kept constant, so that no further accesses to
the memory component are needed (FIG. 25, value 134). The
above-described process is shown in the form of a structogram in
FIG. 26.
Beginning with a starting step S20, in which there is in particular
an initial allocation of initial values to variable memory values,
in step S21 a time counter is set to zero and then periodically
increments its count automatically and in quasi-continuous manner,
i.e. with a predetermined short cycle time.
With the aid of the previously described processes, in the
following step S22 and without the aid of external sensors, at
least the electrical position is determined. The following steps
relate to the running-up phase, in which the kinetic energy of the
motor constantly increases. As a function of the already determined
position value, in step S23 a suitable commutation signal is
generated and is used in step S24 for motor current application, in
such a way that the rotor acceleration takes place in the desired
direction. However, the current is initially not interrupted.
Immediately thereafter in step S25 the actual time t is determined,
by means of which in the next step S26 the provided current
application time tau is determined according to a predetermined
calculation rule or according to a predetermined table stored in
the processor. Waiting takes place in the next step S27 until the
current application time determined in step S26 has elapsed, so
that in step S28 the motor current can initially be switched
off.
As the acceleration phase only lasts for a certain time, a check is
carried out in step S29 to establish whether the provided
acceleration time t LIM has been exceeded or not. If this is not
the case, a continuation occurs with step S23 and so on. However,
if the time t LIM has been exceeded, then the program is branched
into the constant speed phase and a further commutation signal is
supplied in step S30. This step is followed by the step S31 which,
as a result of the generated commutation signal, ensures the
associated correct current application to the motor.
In the following step S32, waiting takes place in the same way as
in step S27 to establish if the motor current is to be temporarily
switched out again, which is possible after a time corresponding to
tau o time units, the constant quantity tau o being known to the
processor.
The temporary disconnection then takes place in step S33. A check
is made in the following step S34 to establish whether any
condition has led to the generation of a stop signal. If this is
not the case, there is again branching to step S30, in which a new
commutation signal is supplied, etc. If this is not the case, i.e.
when a stop signal is present, branching takes place to the end of
the program, i.e. step S35. No further motor current is supplied
and the motor stops.
In the case of a speed control it is also possible in forced manner
to carry out a change of the position determination mode from
static, i.e. at low speeds and per determination of the phase
position of a fundamental wave, to dynamic procedure by means of
counter-emf or vice versa. According to the invention, this change
can take place directly after a predetermined number of current
application cycles.
The prerequisite for the last-mentioned operating procedures is
naturally that there are adequate known and reproducible friction
ratios and moments of inertia of the motor and the equipment driven
by the latter, so that there can be no premature change and the
dynamic position determination mode can function in error-free
manner.
The use of the invention is not restricted to standard
commutatorless d.c. motors, such as are preferably used with 4 or 8
magnetic poles and e.g. 6 or 12 coils (FIGS. 8, 9 and 10). It is
also possible to extend the embodiment to all motors, whose
inductance varies in position-dependent manner, e.g. simple motors
with so-called 3-T-armatures (FIGS. 4, 5 and 7) and in particular
all types of reluctance motors with or without permanent magnets
(FIG. 6).
The magnetic conditions are more particularly shown in FIGS. 4 and
5. FIG. 4 firstly shows a conventional 3-T-armature 44, which is
located in the magnetic field of a permanent magnetic pole pair
with north pole 40 and south pole 41 and which is rotatable about
the centre 46. In the currentless case (FIG. 4) the leg, which is
nearer to the north pole 40 than the legs 43 and 44 to the south
pole 41, is exposed to a higher magnetic flux density (10 flux
lines) than the two other legs (approximately 5 flux lines).
It is clear that a coil 47, as shown in FIG. 5, increases the
magnetization of leg 42 when current flows through and an iron part
of said leg can be saturated with less current than if no
premagnetization existed or only existed to a slight extent through
the permanent magnets 40, 41, as is the case with leg 43 in FIG. 5.
However, correspondingly a higher current application to such a
coil 47 is required, in order to achieve a magnetic saturation of
the leg 42 in the opposite direction. As current application
processes of this type are subject to a first time constant due to
the initial inductance present, the time required for magnetization
up to saturation with a desired polarity will differ.
Similar considerations apply in the case of current application in
a series connection of coils 47 and 48. As a function of the
current polarity, in particular leg 42 will be saturated more
slowly or rapidly, which also leads to a change to the effective
inductance of this coil. It is clear that these effects can have a
different action as a function of the angular position of the
armature 44, but according to the rotational symmetry of the
armature, following a rotation of 120.degree. mech, will be
repeated for the next coil.
FIG. 7 shows a simple spatial arrangement of such a 3-T-armature
motor. The contacting of the coils 47 and 48 can take place
directly with the armature 44 stationary and the magnets 40, 41
rotating. In the case of fixed magnets and rotating armature, the
coils are appropriately supplied with current via slip rings (not
shown in FIG. 7).
A similar action can be obtained with the arrangement according to
FIG. 6 (reluctance motor). A dynamo iron stator 60 with six
distinct stator poles 62 acts on a movable rotor 61, which can also
be made entirely from iron.
On the stator pole 62 there are coils L1 to L6, which can be
supplied with d.c. voltage of random polarity. As the rotor 61 is
firstly non-magnetic, comparable conditions to those of FIG. 4 can
be produced, in that the coils L1 and L4 are connected in such a
way that the indicated south poles of the rotor 61 are obtained.
The simultaneously obtained north poles of the rotor with the south
poles produced by the same magnetization process then face those
stator poles on which the coils L2 and L3 or L5 and L6 are
located.
The iron cores of these coils are consequently also premagnetized
and consequently influence the current path through these coils, if
they are supplied with short test pulses for position measurement
purposes. Such test pulses can naturally also be of different
electrical polarity and it is also possible to use coil
combinations as premagnetization devices, particularly in series
and parallel connection.
A similar configuration to FIG. 7 is shown in FIG. 8, but in which
a fixed armature 44' has twice as many stator poles, which are
provided with associated coils L1 to L6. The number of permanent
magnetic poles of the rotor 55 is also doubled in FIG. 8 and the
said poles of the magnet 50 are interconnected in interruption-free
manner. This magnet is held by an iron yoke 51, which
simultaneously leads to an improvement of the magnetic circuit.
A comparable arrangement with rotating, but internal magnets
(internal rotors) is shown in FIGS. 9 and 10, which additionally
illustrate the magnetic conditions under no-load conditions of the
motor. The stator 915 also has six distinct poles with in each case
associated coils L1 to L6. The magnets 912 of the rotor are located
on an iron yoke 913 and can rotate about the centre 914.
The stator poles with the coils L1, L2, L4 and L5 detect the full
magnetic flux of the permanent magnets 912. They can consequently
be easily saturated by additional current application to the said
coils than is possible with current application to coil L3 or L6 in
the represented rotor position. This is illustrated by the section
lines 910 and 911 in FIG. 10. As can be seen the magnetic flux Phi
1 is represented by a larger number of magnetic flux lines, which
cross the section line 910, whereas there is no significant
magnetic flux (Phi 2) at the section line 911.
Naturally, these conditions are also dependent on the angular
position of the rotor, as shown in FIG. 11A, where the rotor is
further rotated clockwise by a few degrees compared with FIG.
10.
Whereas the flux through the section line 910 has not significantly
changed, the flux through the section line 909 has been
significantly reduced. Therefore the associated coil L2 (FIG. 9)
must be supplied with more current to bring about a magnetic
saturation in this stator pole.
A corresponding embodiment of a rotor with four magnetic poles 501
to 504 is shown in FIG. 11C. Poles 501 and 502 are more strongly
magnetized than poles 503 and 504. Those stator poles which face
the magnetic poles 501 and 502 consequently receive a higher
pre-magnetization and can consequently be brought into magnetic
saturation in an easier manner, i.e. with less current.
However, if said premagnetization has to be overcome by oppositely
acting current application, naturally higher currents must be used
than with those stator coils which face the permanent magnetic
poles 503 and 504.
As a result of this asymmetry in the magnetization having a
periodicity of 360.degree., clear conclusions can be drawn
regarding the rotor position over the entire mechanical full angle
of 360.degree..
A surface structure of the rotating permanent magnet, as shown in
FIG. 11B, has a set back surface 116 on one or more poles, e.g. the
north pole 117.
As a result of this geometrical asymmetry, an asymmetry is also
produced in the magnetic flux, so that as a result of this
asymmetry it is possible to clearly determine the rotor position
over the entire mechanical angle.
Similar facts are illustrated by FIG. 11D. The permanent magnet
with the north pole 1123 and south pole 1124 is rotatable about the
centre 1127 and faces the stator poles 1126 of the stator 1125. The
air gap 1122 is larger than the air gap 1121 and other, not shown
air gaps, as in the aforementioned case, in this way a
mechanical/magnetic asymmetry is produced, which allows a clear
determination of the rotor position over the full mechanical
angle.
* * * * *