U.S. patent number RE35,535 [Application Number 08/435,856] was granted by the patent office on 1997-06-17 for broadband acoustic doppler current profiler.
This patent grant is currently assigned to Rowe, Deines Instruments Incorporated. Invention is credited to Blair H. Brumley, Ramon G. Cabrera, Kent L. Deines, Eugene A. Terray.
United States Patent |
RE35,535 |
Brumley , et al. |
June 17, 1997 |
**Please see images for:
( Certificate of Correction ) ** |
Broadband acoustic doppler current profiler
Abstract
A system and method for measuring current velocities using
coded-pulse broadband acoustic signals. Autocorrelation of two
phase coded pulses which are in the water during a single
transmission cycle is used to calculate a Doppler frequency. The
effective result is current profilers having improved profiling
range and spatio-temporal resolution.
Inventors: |
Brumley; Blair H. (San Diego,
CA), Deines; Kent L. (Poway, CA), Cabrera; Ramon G.
(San Diego, CA), Terray; Eugene A. (Woods Hole, MA) |
Assignee: |
Rowe, Deines Instruments
Incorporated (San Diego, CA)
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Family
ID: |
27080293 |
Appl.
No.: |
08/435,856 |
Filed: |
May 4, 1995 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
Issue Date |
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588469 |
Sep 26, 1990 |
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Reissue of: |
851269 |
Mar 13, 1992 |
05208785 |
May 4, 1993 |
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Current U.S.
Class: |
367/90;
73/170.13 |
Current CPC
Class: |
G01S
15/582 (20130101); G01S 15/8959 (20130101) |
Current International
Class: |
G01S
15/89 (20060101); G01S 15/58 (20060101); G01S
15/00 (20060101); G01S 015/58 () |
Field of
Search: |
;367/89,90,91
;342/104,117 ;364/565 ;128/661.07 ;73/170.13,861.18 |
References Cited
[Referenced By]
U.S. Patent Documents
Foreign Patent Documents
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A2170807 |
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Sep 1973 |
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FR |
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54-131971 |
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Oct 1979 |
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JP |
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Other References
Doviak, R.J., et al. (1976) Resolution of pulse-doppler radar range
and velocity ambiguities in severe storms pp. 15-22. .
"Electronic Communication Series, Radar Technology (Part II),"
edited by Electronic Communication Society, first edition, issued
on Apr. 30, 1968, pp. 119 to 124. .
E-mail from Yasuo Muramatsu dated Jun. 21, 1996 regarding cited
Japanese reference "Radar Technology," (see above reference B).
.
* English translation of Claim 1 of Japanese Patent Publication No.
54-131971. .
Miller, et al., "A Covariance Approach to Spectral Moment
Estimation", IEEE Transactions on Information Theory, Sep. 1972,
pp. 588-596. .
Cabrera, et al., "Development of a Practical Coherent Acoustic
Doppler Current Profiler", pp. 93-97, Proceedings of Oceans, 1987.
.
Brumley, et al., "Performance of a Broadband Acoustic Doppler
Current Profiler". .
Foster, et al., "Flow Velocity Profile via Time-Domain Correlation:
Error Analysis and Computer Simulation", IEEE Transactions on
Ultrasonics, vol. 37, No. 2, May 1990, pp. 164-175. .
Embree, et al., "Volumetric Blood Flow via Time-Domain Correlation:
Experimental Verification", IEEE Transactions on Ultrasonics, vol.
37, No. 2, May 1990, pp. 176-189. .
RD Instruments, "Pulse-to-Pulse Coherent Doppler Sonar Development,
Phase I Report", Apr. 25, 1986. .
Rowe, Francis D., "Pulse-to-Pulse Coherent Doppler Sonar
Development (Broadband ADCP), Phase II Final Report", Jan. 16,
1990. .
RD Instruments, "Non-Disclosure Agreement", Apr. 28, 1987. .
Brumley, et al., "Doppler Profiler Observations at Stellwagen
Bank", Aug. 2, 1989. .
Rowe, Francis D., "Acoustic Doppler Velocity Profiler,
Characterization Parameters", May 15, 1989. .
Rowe, Francis D., "Letters", Oct. 5, 1989. .
Lhermitte, Roger, "Doppler Sonar Observation of Tidal Flow". .
Lhermitte, Roger, "Water Velocity and Turbulence Measurements by
Pulse Coherent Doppler Sonar". .
Lhermitte, Roger, "Observations of Water Flow with High Resolution
Doppler Sonar". .
Dickey, Jr., Frank, and Edward, John, "Velocity Measurement Using
Correlation Sonar". .
Preliminary Specification, Signetics NE/SA604A High-Performance
Low-Power FM IF System, Dec. 1988..
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Primary Examiner: Pihulic; Daniel T.
Attorney, Agent or Firm: Knobbe, Martens, Olson &
Bear
Parent Case Text
This application is a continuation of application Ser. No.
07/588,469, filed Sep. 26, 1990 now abandoned.
Claims
What is claimed is:
1. A velocity measuring system, comprising:
a transmitting transducer for producing a beam pointing in a
direction along which a phase change is measured;
pulse transmission means for providing a pulse train to the
transmitting transducer, the pulse train comprising at least first
and second emitted pulses having a predetermined pulse
separation;
mean for complex sampling in echo return of the emitted pulses
received by a receiving transducer so as to provide a first set of
complex samples;
means for displaying the first set of complex samples by a selected
time lag thereby producing a second set of complex samples;
means for computing a measured value of complex correlation using
at least a portion of the first set of complex samples and at least
a portion of the second set of samples; and
means for obtaining a velocity measurement based on a Doppler
frequency calculated from the complex correlation.
2. The system defined in claim 1, wherein each pulse is coded.
3. The system defined in claim 2, wherein the code comprises phase
coding.
4. The system deemed in claim 3, wherein the phase coding includes
0.degree.and 180.degree.phase codes.
5. The system defined in claim 1, wherein the transmitting
transducer and the receiving transducer are together a unitary
transducer.
6. The system defined in claim 1, wherein the transmitting
transducer and the receiving transducer are separate
structures.
7. The system defined in claim 1, wherein the beam emits over a
water column so that a plurality of velocity measurements are made
at varying depths.
8. The system defined in claim 1, wherein the beam is an acoustic
signal.
9. The system defined in claim 1, wherein the lag time is equal to
the length of a selected pulse.
10. The system defined in claim 1, wherein the pulses are the same
length.
11. The system defined in claim 1, wherein the lag time includes a
time interval when the transducer is not transmitting. .[.12. A
method of deriving a measure of the relative velocity of a signal
source-sensor combination and a field of scatterers separated
therefrom by a medium through which the signal is propagated,
wherein the source-sensor combination includes a transducer, the
method comprising the steps of:
energizing the transducer to emit into the medium towards the field
of scatterers a signal comprising a plurality of pulses having a
predetermined separation, wherein the pulses include at least a
first and second pulse;
sensing a signal comprising a set of echo returns of the pulses
reflected from the field of scatterers;
complex sampling the echo return signal at preselected sampling
intervals so as to provide a first set of complex samples;
delaying the first set of complex samples by a predetermined lag
time so as to provide a second set of complex samples;
presenting to a complex correlator the first and second sets of
samples wherein the fast set consists primarily of echoes from the
first pulse and the second set consisting primarily of echoes from
the second pulse, and forming as outputs the complex products of
members of the first set with complex conjugates of members of the
second set, to produce a complex correlation value; and
obtaining a Doppler frequency based on the phase change calculated
from the complex correlation value..]..[.13. The method of claim
12, additionally comprising the step of calculating a velocity
component of the scatterers using the Doppler frequency..]..[.14.
The method of claim 12, wherein the pulses are coded..]..[.15. The
method of claim 12, wherein the step of delayed samples comprises
storing the first set of samples in a memory..]..[.16. A current
profiler, comprising:
a transducer for transmitting and a transducer for receiving an
acoustic signal having fast and second coded pulses separated by a
preselected lag time wherein the coded pulses are in water together
for at least a portion of a selected time interval;
a sampler for sampling quadrature components of a received signal
over a time interval which depends on the selected range cell,
wherein the received signal comprises echoes of the coded
pulses;
means for calculating the autocorrelation of the sampled quadrature
components;
means for determining the Doppler frequency of the received signal
comprising means for calculating the phase change of the acoustic
signal from the autocorrelation result; and
means for calculating a velocity using the Doppler frequency..].17.
.[.The.]. .Iadd.A .Iaddend.current profiler.Iadd.,
.Iaddend..[.defined in claim 16.]. .Iadd.comprising:
a transducer for transmitting and a transducer for receiving an
acoustic signal having first and second coded pulses separated by a
preselected lag time wherein the coded pulses are in water together
for at least a portion of a selected time interval;
a sampler for sampling quadrature components of a received signal
over a time interval which depends on the selected range cell,
wherein the received signal comprises echoes of the coded
pulses;
means for calculating the autocorrelation of the sampled quadrature
components;
means for determining the Doppler frequency of the received signal
comprising means for calculating the phase change of the acoustic
signal from the autocorrelation result; and
means for calculating a velocity using the Doppler
frequency.Iaddend., wherein a plurality of transducers are
configured such that a plurality of
orthogonal velocity components can be obtained. 18. The current
profiler defined in claim 17, wherein the transducers are arranged
in a Janus
configuration. 19. The current profiler defined in claim .[.16.].
.Iadd.17.Iaddend., wherein the sampler comprises control means
for
limiting voltage offsets. 20. The current profiler defined in claim
.[.16.]. .Iadd.17.Iaddend., wherein the velocity calculating means
comprises means for normalizing the velocity to fixed earth
reference
coordinates. 21. The current profiler defined in claim .[.16.].
.Iadd.17.Iaddend., wherein the selected time interval depends on a
round-trip time between scatterers at a selected range and the
current
profiler. 22. The current profiler defined in claim .[.16.].
.Iadd.17.Iaddend., wherein the transmitting and receiving
transducers are
together a unitary transducer. 23. The current profiler defined in
claim .[.16.]. .Iadd.17.Iaddend., wherein the transmitting and
receiving
transducers are separate structures. .[.24. A Doppler sonar system
for providing velocity measurements, comprising:
a transducer;
a pulse generator communicating to the transducer two or more coded
pulses of preselected length separated by a preselected lag
time;
a complex sampling circuit connected to the transducer so as to
provide quadrature components of a received echo signal; and
a processor including autocorrelation means for generating the
autocorrelation between a first set of quadrature samples and a
second set of quadrature samples delayed by the lag time, the
processor also including means for obtaining a velocity measurement
based on a Doppler
frequency calculated from the autocorrelation..]..[.25. The system
defined in claim 24, wherein the autocorrelation means comprises a
digital
signal processor..].26. .[.The.]. .Iadd.A Doppler sonar
.Iaddend.system .[.defined in claim 24.]. .Iadd.for providing
velocity measurements, comprising:
a transducer;
a pulse generator communicating to the transducer two or more coded
pulses of preselected length separated by a preselected lag
time;
a complex sampling circuit connected to the transducer so as to
provide quadrature components of a received echo signal; and
a processor including autocorrelation means for generating the
autocorrelation between a first set of quadrature samples and a
second set of quadrature samples delayed by the lag time, the
processor also including means for obtaining a velocity measurement
based on a Doppler frequency calculated from the
autocorrelation.Iaddend., wherein a plurality of transducers
receive independently transmitted signals so as
to measure a plurality of orthogonal velocity components. 27. The
system defined in claim 26, wherein the transducers are configured
to generate
non-interfering signals. 28. The system defined in claim .[.24.].
.Iadd.26.Iaddend., wherein the coded pulses include one or more
code elements, each code element defined by a predetermined portion
of a carrier signal which is phase modulated according to one of a
preselected
set of phase codings. 29. The system defined in claim .[.24.].
.Iadd.26.Iaddend., wherein the lag time is selected according to
a
predetermined accuracy of range-velocity resolution. 30. The system
defined in claim .[.24.]. .Iadd.26.Iaddend., wherein the coded
pulse length is selected according to a predetermined accuracy of
range
resolution. 31. The system defined in claim .[.24.].
.Iadd.26.Iaddend.,
additionally comprising means for measuring amplitude. 32. The
system defined in claim 31, wherein the amplitude measuring means
comprises means
for determining backscatter strength. 33. The system defined in
claim 31, wherein the amplitude measuring means comprises means for
determining
particle concentration. 34. The system defined in claim 31, wherein
the
amplitude measuring means comprise means for measuring particle
flux. 35. The system defined in claim 28, wherein the code elements
of a second
coded pulse are inverted with respect to a first coded pulse. 36.
The system defined in claim 35, wherein the code elements are
selected so that:
(1) the code has zero autocorrelation at one lag time to each side
of the sidepeak;
(2) the code has less than a preselected number of sidelobes near
the sidepeak; and
(3) pairs of Golay codes are used in successive coded pulses.
.Iadd.37. The system defined in claim 26, wherein the
autocorrelation means comprises a digital signal processor.
.Iaddend.
Description
BACKGROUND OF THE INVENTION
1. Field of the Invention
The present invention relates to velocity measurement system and,
more particularly, to acoustic Doppler current profilers.
2. Description of the Prior Art
A current profiler is a type of sonar system that is used to
remotely measure water velocity over varying ranges. Current
profilers are used in freshwater environments such as rivers, lakes
and estuaries, as well as in saltwater environments such as the
ocean, for studying the effects of current velocities. The
measurement of accurate current velocities is important in such
diverse fields as weather prediction, biological studies of
nutrients, environmental studies of sewage dispersion, and
commercial exploration for natural resources, including oil.
Typically, current profilers are used to measure current velocities
in a vertical column of water for each depth "cell" of water up to
a maximum range, thus producing a "profile" of water velocities.
The general profiler system includes a transducer to generate
pulses of sound (which when downconverted to human hearing
frequencies sound like "pings") that backscatter as echoes from
plankton, small particles, and small-scale inhomogeneities in the
water. The received sound has a Doppler frequency shift
proportionate to the relative velocity between the scatters and the
transducer.
The physics for determining a single velocity vector component
(V.sub.x) from such a Doppler frequency shift may be concisely
stated by the following equation: ##EQU1## In equation (1), c is
the velocity of sound in water, about 1500 meters/second. Thus, by
knowing the transmitted sound frequency, f.sub.T, and declination
angle of the transmitter transducer, .theta., and measuring the
received frequency from a single, narrowband pulse, the Doppler
frequency shift, f.sub.D, determines one velocity vector component.
Relative velocity of the measured horizontal "slice", or depth
cell, is determined by subtracting out a measurement of vessel
earth reference velocity, V.sub.e. Earth reference velocity can be
measured by pinging the ocean bottom whenever it comes within sonar
range or by a navigation system such as LORAN or GPS. FIGS. 1a and
1b show example current profiles where North and East current
velocities (V.sub.x, V.sub.y) are shown as a function of depth
cells.
Commercial current profilers are typically configured as an
assembly of four diverging transducers, spaced at 90.degree.
azimuth intervals from one another around the electronics housing.
This transducer arrangement is known in the technology as the Janus
configuration. A three beam system permits measurements of three
velocity components, V.sub.x, V.sub.y and V.sub.z (identified
respectively as u, v, w in oceanographic literature) under the
assumption that currents are uniform in the plane perpendicular to
the transducers mutual axis. However, four beams are often used for
redundancy and reliability. The current profiler system may be
attached to the hull of a vessel, remain on stationary buoys, or be
moored to the ocean floor as is a current profiler 100 shown in
FIG. 2.
One class of current profilers now in use, so-called
"pulse-incoherent" systems, measure mean current profiles over
ranges of hundreds of meters. These pulsed sonars use a
pulse-to-pulse incoherent method to derive current velocity.
Profilers characterized by pulse-incoherent processing use the
echoes from each pulse independently, measuring phase changes over
a fraction of the pulse duration to determine the Doppler frequency
shift, i.e., f.sub.D =.theta./T, where .theta. is a phase change
calculated from performing an autocorrelation on a received
waveform at an autocorrelation lag T period. To avoid confusion it
can be stated that the received signal is coherent during the short
lag time over which phase change is detected; the term "incoherent"
refers only to the fact that coherence need not be maintained
between pulses.
Current profilers are subject to trade-offs among a variety of
factors, including maximum profiling range and temporal, spacial
(the size of the depth cell), and velocity resolution. Temporal
resolution refers to the time required to achieve a velocity
estimate with the required degree of accuracy. In typical
applications, a current profiler will make a series of measurements
which are then averaged together to produce a single velocity
estimate with an acceptable level of velocity variance, or squared
error.
For many applications, the resulting combination of profiling range
and resolution is satisfactory to produce useful results. Often
bias is more of a concern than the variance in observations. Bias
is the difference between measured velocity and actual velocity. It
is caused, for example, by asymmetries in band limiting system
components. Measurement bias remains even after long-term averaging
has reduced variance to a predetermined acceptable limit. For
instance, bias dominance is typically found in measuring
large-scale features such as those found at temperature and
salinity interfaces.
For other applications, though, the range and resolution of
pulse-incoherent systems is inadequate. These applications require
the study of oceanic dynamics such as internal waves, small scale
turbulence, sharp scale frontal regions delineating jets, meanders,
and eddies. Using a visual analogy, the pictures produced of such
structures by a pulse-incoherent system are two blurred to be of
any use.
The primary limitations of existing pulse-incoherent systems are
threefold. First, many seconds or minutes of averaging are required
to produce acceptable statistical errors in mean velocity
measurement. Second, for traditional applications, depth cell
resolution is limited to one meter or greater. Third, small scale
turbulence measurement is not possible due to fundamental
limitations of incoherent echo processing, namely, because the
turbulence produced velocities that change too quickly for the
possible combinations of velocity variance and time to average
measurements.
Conventional pulse-incoherent systems estimate the Doppler shift
from either the pulse change per unit time or the shaft in spectral
peak of a single pulse echo. The transmitted waveform is typically
a periodic pulse train characterized by a pulse repetition interval
(PRI). Thus, to provide for a round-trip visit (including echo
time) to the particles, or scatterers, in a given depth cell, the
maximum profiling range or depth is one-half the PRI times c. The
received echoes are placed in memory bins defined by "time-gating"
the receiving signals, i.e., echoes received at time t.sub.n come
from scatterers located at a distance 1/2ct.sub.n. The width of the
gate is usually matched to the pulse length, T, giving a range
resolution of 1/2cT. The velocity (v) of the scatterers in a
particular cell is related to the Doppler shift f.sub.D by the
following equation:
where .pi. is the acoustic wavelength (for example, .lambda.=0.5 cm
at 300 kHz).
Pulse-incoherent systems are significantly affected by noise. A
theoretical lower bound on the variance of the Doppler frequency
estimate from a single pulse is given by the Cramer-Rao bound,
which for an unbiased estimator is approximated by the following
equation for the standard deviation (.sigma..sub.D) of the Doppler
frequency:
where SNR is the signal-to-noise ratio of the Doppler shifted echo
pulse. Applying equations (2) and (3), the corresponding error
(.sigma..sub.r) in the radial velocity (along the beam) estimate is
given by the following equation:
Therefore, for a given carrier frequency, which depends on the
transducer, the minimum velocity error per ping achievable is
inversely proportional to the length of the transmitted pulse. It
can be shown that the variance, or squared error, grows
quadratically toward smaller SNR, and tends to a constant in the
limit of zero noise (a large SNR). Thus, conventional
pulse-incoherent Doppler systems perform well above an SNR of
roughly 10 db where the variance is relatively constant.
Neglecting noise, it is evident that the product of range
resolution, 1/2cT, and velocity error per ping, .sigma..sub.r from
equation (4), is proportional to the acoustic wave-length,
.lambda., and is independent of the pulse length. This range
resolution-velocity error trade-off is the most serious limitation
of pulse-incoherent system, and is directly responsible for the
widely recognized long averaging times required to control the
absolute velocity error.
As an example of averaging time with a pulse-incoherent current
profiler consider a 300 kHz carrier frequency profiling over a
water column of 300 meters which is measured at depth cells of 1
meter, and pinging twice a second. Further assume a monostatic
system wherein the transmitter and receiver circuits share the same
transducer. The range resolution of 1 m means that the pulse length
T is 1.33 ms. The velocity error per ping can be found from
equations (1) and (2) to be about 30 cm/s. To reduce the standard
deviation in the estimate of radial velocity to 1 cm/s, requires
about 30.sup.2 or 900 pings, which at two pings per second requires
that velocity estimates be averaged over about 71/2 minutes.
Pulse-coherent Doppler current profilers have been developed which
improve the velocity measurement accuracy over pulse-incoherent
current profilers by a factor on the order of 100. These sonar
systems profile current velocities over ranges of several meters,
but they are seriously limited in application by small velocity
dynamic ranges which are ultimately caused by velocity ambiguity
effects inherent to pulse-coherent techniques.
For general transmit waveforms, the range-velocity uncertainty
(defined by rearranging equation (4) such that the left-hand side
of the equation is the product of .sigma..sub.r T) is inversely
proportional to the time-bandwidth product of the signal,
determined by the signal decorrelation time (e.g., the time that
the echo is in the water causing the echo to lose enough energy so
that it can not be correlated with itself) and pulse bandwidth.
Signal decorrelation time is related to equations (7-9) below as
well as to a drop in the SNR due to noise. The basic premise behind
the pulse-coherent approach is to increase this product by
transmitting a series of short pulses, in which phase coherence is
maintained over the transmitted sequence. The time between pulses
is adjusted to minimize ping-to-ping interference. A given range
cell is ensonified by successive pulses, so that after
demodulation, the received signal (sampled by time-gating) is a
discrete representation of the Doppler return from that particular
range. The Doppler frequency of this signal can then be estimated
by a variety of techniques, including spectral analysis, or the
"pulse-pair" algorithm (see, e.g., "A Covariance Approach to
Spectral Moment Estimation", Kenneth S. Miller and Marvin M.
Rochwarger, IEEE Trans. Info. Theory, Sep., 1972).
Velocity error for independent pulse pairs has been analyzed. It
can be shown that the pulse pair estimator is a maximum likelihood
estimator (i.e., the estimator having the highest probability of
being correct), and in the limit of large SNR, the Doppler velocity
error per pair is given by the following:
where B is the Doppler bandwidth in Hz. (2.pi.B).sup.-1 is the
decorrelation time, assuming a Gaussian correlation function
exp(-1/2(2.pi..tau.B).sup.2) where is time lag. Typical values of B
imply an error per root ping (the square root of the variance per
number of pings which are included in the average) between 0.1 and
2.5 cm/s, depending on conditions. In the more general case where
successive pairs are correlated, the velocity error is a
complicated function of pulse spacing, Doppler bandwidth, and the
signal to noise ratio.
Since a transmit pulse need only contain a few cycles of the
carrier, range resolutions on the order of 5-10 cm are easily
attainable (for example, 10 cycles at 300 kHz corresponds to a 2.5
cm pulse length, where the velocity is calculated as c/2 to account
for round-trip time). However, despite their outstanding range
resolution capabilities, because pulse-coherent systems are
sampled, velocities are aliased about the Nyquist frequency of the
sampling. This means that samples 2.pi. radians apart in phase are
indistinguishable, which leads to the well-known "range-velocity"
ambiguity presented in the equation below:
where R.sub.max is the maximum profiling range of the system and
V.sub.max is the maximum velocity resolution. Thus, for a given
transmission frequency and desired velocity resolution, a
pulse-coherent system is limited in profiling range. Although the
ambiguity can be improved by using a non-periodic pulse tram,
experience has shown that a factor of order five improvement is a
practical limit. As a consequence, conventional pulse-coherent
systems have been limited to relatively short ranges, of order tens
of meters.
As is well-known in the technology, the autocorrelation function is
used to measure the dependence of a received waveform at time t
with the received wave-form delayed by a lag time, and the result
is used in calculating the Doppler frequency. In pulse-incoherent
Doppler, the correlation time of the signal is primarily determined
by the pulse width. Pulse-coherent systems, besides being dependent
on pulse width, are also sensitive to various changes in scatterer
movement. These phenomenon cause a narrowing of the autocorrelation
function, or equivalently, a broadening of the Doppler spectral
peak. There are three principal sources of spectral broadening:
finite residence time, turbulence within the sample volume, and
beam divergence.
With respect to residence time between successive pulses some
particles will have moved out of the sample volume while new
particles will have been introduced. Since the new particles enter
with random phases, the signal will completely decorrelate over a
"residence time" of order d/U, where d is a measure of the size of
the range cell, and U is the relative velocity between the beam and
the scatterers.
Another source of spectral broadening is sample volume turbulence.
Turbulent eddies with spatial scales on the order of the sample
volume or smaller cause the scatterers to have a distribution of
velocities.
Finally, beam divergence contributes to spectral broadening. This
effect is analogous to the turbulence broadening except that the
diversity in scatterer velocity within the sample volume is caused
by the small variation across the beam of the angle between the
velocity vector and the normal to the transducer.
The contributions of these three effects to the Doppler spectral
broadening can be estimated as follows:
where d is the half-power scattering volume width,
.vertline.u.vertline. is the magnitude of the relative velocity
between the beam and the scatters, .epsilon. is the turbulent
energy dissipation rate, .DELTA..theta. is the two-way, half-power
beamwidth, and U.sub.c is the cross-beam velocity component. The
total Doppler bandwidth (B) is the root-mean-square (RMS) of the
individual contributions: B=(B.sub.r.sup.2 +B.sub.r.sup.2
+B.sub.d.sup.2).sup.1/2.
In summary, pulse-coherent systems are hampered by a limited
profiling range, often just tens of meters. Further, interference
between the transmit pulses creates instability: instability to the
point where the system produces either very, good or very bad
velocity measurements with no in-between.
Accordingly, an acoustic Doppler current profiler overcoming
limitations such as those described above would readily find
application over the entire range of shipboard, fixed-mounted, and
moored deployments. Among the possible applications is that of
weather prediction wherein the dynamics of cold and warm water
mixing remains a difficult and important problem requiring greater
spatio-temporal resolutions for large profiling ranges.
In addition, an entirely new set of short spacial and temporal
current measurement scales would be made accessible to remote
sensing instruments. These measurement include internal waves,
small scale turbulence, sharp scale frontal regions, delineating
jets, meanders, eddies, and other large scale structures in the
ocean. An improved current profiler would achieve a current
profiling range comparable to that of existing incoherent acoustic
Doppler profilers, but realize a factor of about 100 improvement in
the variance of single-pulse velocity estimates.
Lastly, it would be desirable to provide a current profiler with a
fast velocity response, i.e., a decrease in averaging time. Such a
fast response will improve horizontal spacial resolution if the
current profiler is mounted on a moving ship. For example, a
current profiler which could average velocity measurements over
one-tenth of a mile in the time now required to average over five
miles would be a valuable improvement over present technology.
SUMMARY OF THE INVENTION
The above-mentioned needs are satisfied by the present invention
which includes a system and method for measuring velocities using
coded-pulse broadband acoustic signals. The present invention
allows greater range-velocity resolutions over greater profiling
ranges with less velocity averaging time than has heretofore been
achieved. The present invention samples quadrature components of a
received signal which are used to calculate phase change over a
time interval as a Doppler frequency. One or more relative velocity
components can be transformed into absolute velocity components
using fixed earth reference coordinates.
The present invention includes a velocity measuring system
comprising a transmitting transducer for producing a beam pointing
in a direction along which a phase change is measured. The velocity
measuring system further pulse transmission means for providing a
pulse tram to the transmitting transducer, the pulse train
comprising at least first and second emitted pulses having a
predetermined pulse separation. The velocity measuring system also
includes a means for complex sampling an echo return of the emitted
pulses received by a receiving transducer so as to provide a first
set of complex samples, In addition, the velocity measuring system
includes a means for delaying the first set of complex samples by a
selected time lag thereby producing a second set of complex
samples. The velocity measuring system also includes a means for
computing a measured value of complex correlation using at least a
portion of the first set of complex samples and at least a portion
of the second set of complex samples. Lastly, the velocity
measuring system includes a means for deriving a velocity component
from the complex correlation.
Another aspects of the present invention is coding each pulse using
a technique such as phase coding. One preferred embodiment uses
0.degree. and 180.degree. phase codes. The pulses may have a lag
time that is equal to the pulse length, Also, the lag time may
include a time interval when the transducer is not
transmitting.
To measure multiple orthogonal velocity components a number of
transducers may be configured in either monostatic or bistatic
configurations. In a current profiler embodiment, the beam produced
by each transducer is an acoustic, signal and a plurality of
velocity measurements are made over a plurality of depth cells so
as to form a current profile of a water column. Amplitude
measurements may also be made so as to determine backscatter
strength, particle concentration and particle flux.
These and other objects and features of the present invention will
become more fully apparent from the following description and
appended claims taken in conjunction with the accompanying
drawings.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1a is a scatter diagram of an exemplary current profile
showing the East velocity vector plotted as a function of
depth;
FIG. 1b is a scatter diagram of an exemplary current profile
showing the North velocity vector plotted as a function of
depth;
FIG. 2 is a perspective view of a current profiler, having a Janus
configuration of transducers, moored to the ocean floor;
FIG. 3 is a pulse diagram providing a comparison between the pulses
transmitted by various current profilers including a
pulse-incoherent Doppler system, a pulse-coherent Doppler system, a
broadband Doppler system and a coded-pulse Doppler system, the
latter two belonging to the present invention;
FIGS. 4a,b,c are sets of coded-pulse diagrams illustrating
exemplary transmission codes of the present invention;
FIG. 5 is a side elevational view of one preferred mechanical
assembly for a current profiler of the present invention;
FIG. 6 is a top plan view of the current profiler shown in FIG.
5;
FIG. 7 is a block diagram of one preferred embodiment of the
electronics for a current profiler of the present invention;
and
FIG. 8 is a block diagram of one preferred embodiment of the
sampling module shown in FIG. 7.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
Reference is now made to the drawings wherein like numerals refer
to like parts throughout.
FIGS. 1a and 1b were introduced in the "Backgound of the Invention"
section above. The exemplary current velocity profile depicted in
the scatter diagrams of FIGS. 1a and 1b is the type of information
that is also the objective of the current profiler of the present
invention. However, the present invention provides greater accuracy
in current velocity measurements, at greater ranges, than has
heretofore been possible.
FIG. 2 illustrates a current profiler 100 which is semipermanently
moored to the ocean floor 102, In this type of profiler deployment,
a record of current profiles is typically stored in a non-volatile
memory (not shown) located inside the current profiler 100.
The current profiler 100, as shown in FIG. 2, generates a set of
acoustic beams 104a,b,c,d which emanate from transducers. The
current profiler 100 is upward looking, that is, the acoustic beams
104 are directed vertically towards the ocean surface. Each beam
104 "illuminates" a water column which can be decomposed into
horizontal dices known as range, or depth, cells such as the cell
indicated at 106. By appropriate transmission and reception of
sound pulses, the phase shift between pulse echoes is calculated.
The phase shift is then step-by-step transformed into a Doppler
frequency, a velocity along the beam 104, and then one or more
orthogonal current velocity components such as those indicated at
108a,b. The current, profiler 100 may be deployed in other ways
than that shown in FIG. 2 including, for example, various
combinations of downward, upward or other angled looking, on fixed
or moving platforms, or on surface, bottom, or mid-depth
moorings.
FIG. 3 presents in schematic form a number of different Doppler
measurement techniques used in acoustic Doppler current profilers
(ADCPs), including the broadband Doppler and coded-pulse broadband
Doppler methods of the present invention. In the first technique, a
pulse-incoherent ADCP transducer 120 is shown generating a pulse
122 at a time t. The single transmitted pulse 122 is sized to match
the associated depth cell. After passing through a depth cell, the
pulse 122 is shown at a time t plus a time equal to the length of
the pulse (Lpulse), having moved to a new location as indicated at
124.
The pulse 122 may generate an echo (not shown) at each depth cell
depending upon the density of scatterers at each depth. Measurement
of current velocity at the desired depth cell is based upon a
predetermined lag time between transmission of the pulse and
reception of the desired echo. A pulse-incoherent ADCP measures
current velocity by measuring the Doppler shift in the frequency of
the returning echo. The Doppler frequency is indirectly calculated
from the difference in phase between two different samples of the
received signal.
In FIG. 3, a pulse-coherent ADCP Transducer 126 is shown emitting a
pulse 128. The pulse 128 is a shorter duration (greater depth
resolution) than the pulse 122 of the pulse-incoherent system. Like
the pulse-incoherent Doppler system, the echo from each single
pulse is allowed to return to the transducer 126 before the next
pulse 130 is transmitted. However, unlike a pulse-incoherent
system, the fundamental measurement of a pulse-incoherent system is
the phase change between two successive echoes at the same
depth.
FIG. 3 also illustrates pulses that are generated by a broadband
ADCP transducer 132 of the present invention. The broadband method
differs from either the pulse-incoherent or pulse-coherent methods
in that the broadband method utilizes two (or more) pulses in the
beam (or the equivalent thereof) at the same time such as the
pulses indicated at 134a and 134b. In FIG. 3, the pulses are
separated by a lag time, L1, equal to the pulse separation. After
traveling some distance and echoing back to the transducer 132, the
phase change between the pulse echoes at the same range is measured
using an autocorrelation function.
Unlike the pulse-coherent method, the maximum profiling range of
the broadband current profiler is not limited to the pulse
repetition interval. The pulse length, or width, is typically much
shorter than the depth cell size which results in a large
time-bandwidth product (hence the term "broadband").
The present invention also includes a coded-pulse broadband ADCP
which is characterized by pulses shown in FIG. 3. A transducer 138
generals a pulse pair 140a,b that propagates through the water as
shown, for example, by the later pulses 141a,b. Each pulse 140
includes four equal-sized code elements 142a,b,c,d that each
comprise one or more cycles (or portions thereof) of the
transmitted acoustic waveform. The code elements 142 represent
phase codings such that each element is either at 0 or 180 degrees
of phase. While only two coded-pulses are shown in FIG. 3, the
method can be generalized to include more than two pulses.
For a coded-pulse ADCP, measurement of phase change is identical to
that of the broadband method previously discussed. In addition,
however, the pseudo-random phase coding is applied to the pulses
allowing longer pulses to be used without decreasing the
band-width. Longer pulses increase the echo power thus delaying the
signal decorrelation to greater ranges and extending the useful
profiling range of the system. The coded pulses may be as large as
the size of the depth cell. If the pulse separation or lag time L1
is equal to the pulse length, the pulses are combined into a
single, continuous-coded transmission.
FIG. 4 shows three examples of "ideal" coded pulses having
different length that may be generated by the coded-pulse broadband
system of the present invention. Each diagram (FIGS. 4a,b,c)
corresponds to one pulse, or ping. The actual waveforms that are
injected in the water are somewhat different than those portrayed
in FIG. 4 due to the finite bandwidth of the transducers and the
power amplifier. Therefore, in the corresponding actual waveforms
there is a short recovery time after phase reversals.
FIG. 4a includes three different representations of a sequence of
code elements generally indicated at 144a-j. The first code
representation is a transmit waveform generally indicated at 146.
Each code element 144 is a collection of four cycles of the carrier
signal. Phase shifts of 180 degrees my occur between code elements
144 as, for example, shown by the transition between the code
elements 144a and 144b. The exemplary pulse of FIG. 4a has M=10
code elements 144 wherein the first five code elements 144a-e are
inverted and repeated by the last five code elements 144f-j so as
to essentially combine two pulses in the continuous waveform 146.
Inverting a second pulse, such as code elements 144f-j, may be
useful in reducing noise bias.
Thus, for the waveform 146, an autocorrelation function (as is
further discussed below) is performed on the first five elements
144a-e and the last five elements 144f-j after inversion using a
lag time equal to the time to transmit five code elements. In the
typical case, the number of code elements for a particular
application will be matched to the size of the depth cell.
The pulse coding can also be represented in binary form as shown by
a code sequence generally indicated at 147 in FIG. 4a. The code
sequence 147 is based on each code element 144 being deemed by two
bits. The two bit code is shown in Table 1 below.
TABLE 1 ______________________________________ B.sub.1 B.sub.0
Phase ______________________________________ 0 x off 1 0 0 degrees
1 1 180 degrees ______________________________________
In Table 1, the most significant bit (B.sub.1) indicates whether
the transmitter is on (1) or off (0) for the duration of the code
element 144. The least significant bit (B.sub.o) indicates the
phase, 0.degree.(0) or 180.degree.(1), of the code element 144. The
character "x", in Table 1, is a "don't care" state.
The code sequence 147 shows the decimal equivalent of the binary
code. The code element 144a, for example, is defined in the code
sequence 147 as "2" meaning that the transmitter is on and the code
element 144a is 0 degrees phase. A phase waveform 148 presents the
same fundamental information as the transmit waveform 146 and code
sequence 147 but it is expressed in the form of a square-wave.
FIG. 4b shows a coded-pulse that differs from that of FIG. 4a in
that the pulse is twice as long (M=20). The first ten code elements
144 of the pulse in FIG. 4b are the same as the code elements 144
of FIG. 4a. The last ten code elements 144' are simply a repetition
of the first ten. Thus, the two pulses 144, 144' are combined in a
single transmit waveform having a lag time equal to the time to
transmit ten code elements.
FIG. 4c shows a coded-pulse that differs from that of FIG. 4b in
that the pulse is longer (M=30) due to a ten code element dead-time
placed between the two sets of ten transmitted code elements 144,
144'. Thus, the lag time is equal to twenty code elements. The
error in the Doppler frequency is inversely proportional to the
pulse separation. The range resolution is determined by the length
of the coded pulse.
In a presently preferred embodiment of the velocity measurement
system, the code is carefully chosen so as to eliminate a from
central peak and sidelobe noise in the autocorrelation function.
Central peak noise is effectively eliminated by inverting the
second pulse, e.g., as shown in FIG. 4a in half of the transmitted
pings. The following steps are taken to eliminate sidelobe noise:
(1) a code is used that has zero autocorrelation at one lag time to
each side of the sidepeak (where phase measurements are made), (2)
a code is used that has minimal sidelobes near the sidepeak, which
are arranged symmetrically around the sidepeak, and (3) pairs of
complementary, or Golay, codes are used on successive pings so that
biases will cancel with averaging.
The pulse separation, or lag time L1, determines accuracy of
range-velocity resolution with shorter lag time meaning grater
resolution. It is even possible to make the lag time less than the
length of a single coded pulse by transmitting pulses that overlap
in one or more code elements. For example, using letters of the
alphabet to represent code elements, the sequence "ABABA" would
allow two pulses "ABA" having a length of three code elements to be
transmitted with a lag time equal to the time to transmit two code
elements.
A skilled technologist will thus understand and appreciate that
there are trade-offs in choosing the proper code, code length and
pulse separation of a multi-pulse waveform that will depend on the
particular application of the present invention.
Hereinafter, both the broadband ADCP and coded-pulse broadband ADCP
systems and methods will generally be referred to as the broadband
ADCP unless otherwise indicated.
FIG. 5 shows a mechanical assembly, generally indicated at 150,
that houses and protects the electronics (FIG. 7) necessary to
implement the broadband ADCP of the present invention. The
mechanical assembly 150 includes a set of four transducers
152a,b,c,d arranged in a Janus configuration. The mechanical
assembly 150 may, of course, host other numbers and configurations
of transducers than the four transducers shown in FIG. 5. The
transducers 152 include piezoelectric ceramic plates that are
encapsulated in a protective covering of various materials.
The transducers 152 are typically manufactured so that each
operates at a particular frequency chosen from a suitable range of
frequencies such as, for example, 75, 150, 300, 600 and 1200
kilohertz. Low-frequency transducers are commonly used in open
ocean applications where a long profiling range is desirable.
High-frequency transducers, on the other hand, are used in shallow
water applications where depth resolution, as characterized by the
size of a depth cell, and finer spatial and temporal scales are
important. The transducers 152 are manufactured to be easily
substitutable on the current profiler assembly 150 so that the
proper acoustic frequency can be used to achieve the desired
combination of profiling range and velocity resolution, which may
vary from one velocity profiling experiment to another. A top plan
view of the transducers 152 is illustrated in FIG. 6.
In FIG. 5, the transducers 152 are connected to one end of a
cylindrical pressure vessel 154 whereto acoustic transmitting,
receiving and processing electronics are contained. The transducers
152 are positioned at 90.degree.intervals of azimuth around the
periphery of the pressure vessel 154 in a Janus configuration. To
achieve multiple degrees of freedom in calculating orthogonal
components of velocity, the transducers 152 are canted outward from
the longitudinal axis or the pressure vessel 154. The mechanical
assembly 150 is conveniently positioned in the water by connecting
one or more cables and/or buoys to a pair of mounting lugs 156a,b
located on the side of the pressure vessel 154.
An I/O connector 158 is located at the other end of the pressure
vessel 154. The I/O connector 158 is connected to a transmission
cable (not shown) for measurements wherein post-processing of
current profiles in real-time is desired. Otherwise, the current
velocities may be stored on a recording media (not shown) such as,
for example, magnetic tape or electrically erasable programmable
read-only memory (EEPROM), optionally configured in the electronics
of the pressure vessel 154.
With reference now to FIG. 7, a block diagram shows a presently
preferred embodiment of the electronics in the coded-pulse
broadband ADCP. The electronics can be functionally partitioned
into a front-end transducer assembly 160 that receives acoustic
signals, and an electronics assembly 162 that coordinates
transmitting and receiving, and performs signal processing. Because
the transducer assembly 160 is specifically matched to the
transducers 152, whenever the transducers 152 are changed, the
entire transducer assembly 160 is replaced.
Referring First to the transducer assembly 160 shown in FIG. 7, the
transducers 152 are each electrically connected to one of a set of
tuning and transmit-receive (T/R) switch circuits 164a,b,c,d. In
the presently preferred embodiment of the tuning section of the
circuit 164, the primary to a transformer (not shown) is connected
to the two leads of the transducer 152 to resonate out the
capacitance of the transducer. One side of the transformer
secondary is connected to a series LC circuit (not shown) that is
tuned to the frequency of the transducer 152. This forms the tuning
section of the circuit 164.
A circuit comprising a diode network and transmitter transformer
(not shown) is used to cancel the effect of the capacitor in the LC
circuit allowing the transmission signal to be fed to the
transducer 152. The transmit/receive section of the circuit 164
thus allows the transducers 152 to either receive or transmit
pulses at any given time. It will be recognized by one skilled in
the relevant technology that although the embodiment of the
broadband ADCP shown and described herein is a monostatic system,
requiring time-sharing of a single set of transducers, a bistatic
implementation is also possible that would not require a
transmit-receive section in the circuit 164.
A coded-pulse transmission is initiated by a microcomputer 166. In
one presently preferred embodiment, the microcomputer 166 includes
a CMOS 68000 microprocessor available from a number of vendors
including Motorola. A user specifiable set of parameters, including
the number of cycles per code element and the code length, is
stored in a ROM in the microcomputer 166. The microcomputer 166
transfers the waveform specific parameters across a digital bus 168
to a timing generator 170. Under the control of the microcomputer
166, the timing generator 170 controls a coder transmitter 172 to
generate the appropriate pair of coded-pulses, including dead-time.
The coded-pulses are amplified by a power amplifier 174 and are
eventually transmitted into the water by the transducers 152 as a
coded acoustic waveform.
During some user specified blanking interval, when no pulses are
transmitted, echo pulse received from the transducers 152 are fed
from the tuning and T/R switch circuits 164 to a set of
preamplifiers 178a,b,c,d. In a preferred embodiment, the
preamplifers 178 are differential amplifiers having one side of the
amplifier 178 tied to ground. The received signal, which is the sum
of the transmitted signal and noise, is amplified by the
differential amplifier. The amplified signals are fed from the
preamplifers 178 to a set of receiver amplifiers 180a,b,c,d. The
preamplifers 178 allow the gain of the combined amplifier set 178
and 180 to be accurately controlled although another embodiment
could combine the two amplifiers 178, 180.
In one preferred embodiment, the receiver amplifiers 180 each
include a Signetics SA604A semiconductor chip. Although designed
for intermediate frequency conversion applications, the two
amplifiers (not shown) of the SA604A chip happen to operate over
the anticipated frequency range of the current profiler. The
amplifiers are connected in series to the output of each
preamplifier 178. The signal strength of the echo is also made
available to the system by the receiver amplifiers 180, for
example, from the pin 5, RSSI output of the SA604A chip. In one
preferred embodiment, the signal strength is digitized and recorded
for later processing.
The signal strength signal can be calibrated for use in measuring
backscatter strength, particle concentration and particle flux. For
example, one application of this type of measurement is in dredging
operations where signal strength is used in determining sediment
concentration and vertical flux in plumes created by dumping
spoil.
The output signals of the receiver amplifiers 180 are fed to a set
of in-phase mixers 182a,b,c,d and a set of quadrature mixers
183a,b,c,d. The mixers 182, 183 form the product of the received
signal and the output of the quadrature generator 184. More
specifically, the mixers 182, 183 are used to heterodyne the
received signal so as to translate the carrier signal into a DC
signal (where the carrier signal includes an in-phase [cosine] and
quadrature [sine] signal, collectively called quadrature signals).
In the present embodiment, the mixers 182, 183 are implemented as
two 74HC4053 triple two-channel analog multiplexer/demultiplexer
chips such as those supplied by Signetics. The quadrature signals
are received by the mixers 182, 183 from a quadrature generator
184.
The quadrature generator 184, of the preferred embodiment,
comprises a pair of D flip-flops (not shown) that are connected in
series. The inverted output Q' of second flip-flop is fed back into
the input D of the first flip-flop. In operation, the quadrature
generator 184 receives an oscillator signal from the timing
generator 170. The oscillator signal is fed into the clock input of
two D flip-flops. The in-phase signal is thus sampled from the
inverted output Q' of the second flip-flop and the quadrature
signal is sampled from the noninverted output Q of the first
flip-flop. The quadrature signals are then fed from the quadrature
generator 184 to the mixers 182, 183.
The mixers 182,183 feed their respective amplified quadrature
signals to a set of programmable low-pass filters 188a,b,c,d and
189a,b,c,d. The low-pass filters 188 are programmed by a controller
192 to pass the sideband frequencies, e.g., up to 20% of the easier
frequency, corresponding to the phase modulation of the coded
pulse. The filtered quadrature signals output from the low-pass
filters 188, 189 (labeled as cosine and sine channels) are fed into
a sampling module 194 which is discussed in more detail below with
reference to FIG. 8.
The function of the sampling module 194, in FIG. 7, is controlled
by the controller 192 and the timing generator 170. A receive cycle
is initiated by the timing generator 170 at a time after the last
element of a code sequence, e.g., the code elements 144 (FIG. 4),
has been transmitted. After a user programmable delay, to permit
the recovery of the receiver electronics in the transducer assembly
160, the timing generator 170 produces a train of sampling strobes
that trigger analog-to-digital converters in the sampling module
194. In the preferred embodiment, the sampling module 194 outputs
four samples of four bits of digit data per word (16 bits)
transferred across the digital bus 168. This is so since the
sampling module 194 is allocated on two separately addressable
boards, each board servicing two of the transducers 152. Thus, each
sample bit corresponds to one sample of one quadrature component of
the waveform received by one of the four transducers 152. The
digital data is transferred to a digital signal processor (DSP) 196
across the digital bus 168. In the preferred embodiment, the
digital bus 168 is a custom, asynchronous bus having sixteen data
lines (BD0-BD15) and twelve address lines (BA1-BA12). The preferred
digital bus 168 can transfer data at speeds up to 400 ns per word
which is primarily limited to the transfer rates of the DSP 196 and
microcomputer 166.
The DSP 196 calculates the autocorrelation function (R(h)) of the
received signal at a predetermined lag corresponding to the number
of code elements in the first pulse. To calculate this function the
DSP 196 applies the following equation, independently, for each of
the four cosine-sine pairs output by the sampling module 194:
##EQU2## where
h is a predetermined lag represented by an integer sample
number;
j is integer sample numbers within a depth cell of interest;
cosine and sine is data sampled from cosine and sine channels (such
as from the low-pass filters 188, 189 in FIG. 7)
i=(-1).sup.1/2 ;
S.sub.j =cos.sub.j +sin.sub.j i; and
S* denotes the complex conjugate of S.
As an example of this calculation, consider Table 2, below, having
the set of cosine-sine samples numbered 123 to 139 for a given
depth cell and lag h=3. (In this example it is assumed, only for
the purpose of simplifying calculations, that the sample number, j,
and the angle, in radians, of the cosine-sine sample at time j, are
one and the same.) The "cosine" and "sine" columns contain data
representing analog values that could be output by the sampling
module 194. The "products" columns contain the products as defined
in the summation of equation (10).
TABLE 2 ______________________________________ Sample Products
Number Cosine Sine Real Imaginary
______________________________________ 123 -0.88796 -0.45990
-0.98999 0.141120 124 -0.09277 -0.99568 -0.98999 0.141120 125
0.787714 -0.61604 -0.98999 0.141120 126 0.943984 0.329990 -0.98999
0.141120 127 0.232359 0.972630 -0.98999 0.141120 128 -0.69289
0.721037 -0.98999 0.141120 129 -0.98110 -0.19347 -0.98999 0.141120
130 -0.36729 -0.93010 -0.98999 0.141120 131 0.584208 -0.81160
-0.98999 0.141120 132 0.998590 0.053083 -0.98999 0.141120 133
0.494872 0.868965 -0.98999 0.141120 134 -0.46382 0.885924 -0.98999
0.141120 135 -0.99608 0.088368 -0.98999 0.141120 136 -0.61254
-0.79043 -0.98999 0.141120 137 0.334165 -0.94251 138 0.973648
-0.22805 139 0.717964 0.696080
______________________________________
For example, the real product shown opposite the sample number
j=123 is obtained as follows: ##EQU3## Note that for a first
product j+h=126. There is no samples beyond j=139. Therefore,
products beyond j=136 cannot be calculated since, at those samples,
j+h is greater than 139.
After the products are calculated they are summed and output by the
DSP 196. For example, using the products in Table 2, the output
value is -13.8598+1.975680i. In the presently preferred embodiment,
resolution has been sacrificed for speed and each sample value is
represented by one bit. However, it can be shown that only half the
information available in the cosine-sine information is lost by
using this method.
In this way, the DSP 196 can perform a fast multiply by
exclusive-oring two 16-bit data words received from the cosine-sine
channel via the sampling module 194. The digital representation of
(0,1) is interpreted by the DSP 196 as (-1,+1). Once the multiplies
are performed, the summation of products is accomplished using a
look-up table stored in EPROM. The presently preferred
configuration of the DSP 196 makes use of Texas Instruments
TMS320E15 16-bit, digital signal processor chip.
The complex number representation of each autocorrelation result is
transferred from the DSP 196 across the digital bus 168 to the
microcomputer 166. For linear systems, the Doppler frequency
f.sub.D is calculated as follows: ##EQU4## where
f.sub.D is the Doppler frequency of the echo;
I is the imaginary part of the complex number;
R is the real part of the complex number;
h is the lag used to calculate the autocorrelation; and
T is the time between samples.
For a hardlimiting system, such as the one shown and described
herein, the microcomputer 166 uses the following Doppler frequency
equation: ##EQU5## In addition, the microcomputer 166 uses
normalized values of I and R in equation (11b) by dividing each by
the autocorrelation at zero lag, i.e., the normalized
autocorrelation function must be used. Note that for linear systems
the normalization step cancels in the division I/R and therefore is
unnecessary. In one alternative embodiment, the microcomputer 166
calculates orthogonal velocity components based on equation (11)
and then translate these velocities to earth reference values,
e.g., subtracting out the components of velocity generated by the
ship. In another embodiment, the Doppler frequency and/or other
intermediate calculations can be forwarded to a conveying vessel
via the I/O port 158 (FIG. 5). In yet another embodiment of the
current profiler electronics, the Doppler frequency results can be
stored in a recording media such as EEPROM that would be added on
to the digital bus 168.
It will be appreciated by one skilled in the relevant technology
that the DSP 196 is an optional element of the electronics assembly
162, and that the operations there specified may be carried out in
the microcomputer 166.
FIG. 8 shows a block diagram of a portion of the sampling module
194. The portion shown corresponds to the circuitry necessary to
sample processed signals received from two of the four transducers
152. A digital-to-analog (D/A) converter 210, preferably a PM7226
chip, receives a threshold control word from either the DSP 196 or
microcomputer 166.
The resulting analog signals are fed to a set of comparators
212a,b,c,d. The quadrature signals from the low-pass filters 188a,b
and 189a,b are also fed to the comparators 212, which compare the
threshold signals to the quadrature signals. The comparators 212
are preferably implemented with high-speed CMOS circuits such as,
for example, the TLC374 comparator chips distributed by Texas
Instruments. The shift register 214, preferably formed from four
74HC4094 8 bit shift register chips, takes the output of the
comparators 212 at a time specified by the timing generator 170
(FIG. 7) and converts the samples to a 16-bit parallel word for
storage in the FIFO 216.
Once the shift register 214 is full, its output is strobed into a
first-in-first-out (FIFO) buffer 216, preferably including four 4
bit.times.16 word 74HC40105 chips. The stored samples, up to 128,
are then accessed by the DSP 196 via the digital bus 168. Thus, the
DSP 196 does not have to continually read the shift register 214
since samples are buffered in the FIFO 216.
The basic idea behind the s ample module is that random scatterers
cause the echo signal to generate as many highs as lows which
should produce a zero mean signal. The DSP 196 or microcomputer 166
performs a statistical calculation to determine if the samples ar
zero mean and, if not, then a new threshold control word is written
to the D/A converter 210. This threshold manipulation eliminates
voltage offsets likely caused by the circuitry, including such
components as the low-pass filter 188, 189 (FIG. 7), comparators
212 and shift registers 214.
It will now be appreciated that the present at invention provides a
means for measuring current velocity that offers improvements in
the combination of profiling range and spatio-temporal resolution.
The improvements stem from the use of a phase coded acoustic signal
and autocorrelation processing to measure the Doppler shift between
two pulses that are generated in a single transmission cycle.
Although the application of the present invention for measuring
velocity disclosed herein relates to current profiling, it shall be
understood that other velocity measurement applications would
likely benefit from the present invention including the following:
bottom tracking to determine vessel velocity, airborne object
velocity measurements using radar, blood flow measurement and
sewage and water velocities inside of pipes.
While the above detailed description has shown, described and
pointed out the fundamental novel features of the invention as
applied to various embodiments, it will be understood that various
omissions and substitutions and changes in the form and details of
the device illustrated may be made by those skilled in the art,
without departing from the spirit of the invention.
* * * * *