U.S. patent number RE34,810 [Application Number 06/552,632] was granted by the patent office on 1994-12-27 for fast acting phase shifting apparatus for use in digital sampling systems.
This patent grant is currently assigned to Ampex Corporation. Invention is credited to Maurice G. Lemaine, Leonard A. Pasdera.
United States Patent |
RE34,810 |
Lemaine , et al. |
December 27, 1994 |
Fast acting phase shifting apparatus for use in digital sampling
systems
Abstract
A method and apparatus is disclosed for controlling the sampling
of a composite analog color video signal so that the analog signal
is sampled at precise locations relative to the phase of the color
subcarrier of the color video signal in the absence of modulation.
The invention thereby enables a digitized signal having a constant
phase relationship relative to the unmodulated subcarrier phase to
be provided which can be advantageously used in magnetic recording
and reproducing apparatus. Since the constancy of the location of
the samples relative to the phase of the color burst is maintained
prior to recording, there is no phase error that would pose a
problem during reproducing of the signal. The apparatus stores a
set of reference samples of a selected number of cycles of the
color burst interval of the color video signal and generates an
output clock signal that is phase synchronized with the reference
samples, subsequently stores sets of comparison samples of color
burst cycles from subsequent horizontal video lines using the
generated clock signal and detects any error in the phase location
thereof. The apparatus adjusts the phase of the generated output
clock signal in response to any detected error signal and stores a
new set of reference samples in the event that the error detected
is beyond predetermined limits.
Inventors: |
Lemaine; Maurice G. (Redwood
City, CA), Pasdera; Leonard A. (San Carlos, CA) |
Assignee: |
Ampex Corporation (Redwood
City, CA)
|
Family
ID: |
22374633 |
Appl.
No.: |
06/552,632 |
Filed: |
November 16, 1983 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
Issue Date |
|
Reissue of: |
117752 |
Feb 1, 1980 |
04301466 |
Nov 17, 1981 |
|
|
Current U.S.
Class: |
348/472;
348/505 |
Current CPC
Class: |
H04N
9/64 (20130101); H04N 9/808 (20130101); H04N
9/888 (20130101) |
Current International
Class: |
H04N
9/808 (20060101); H04N 9/64 (20060101); H04N
011/06 () |
Field of
Search: |
;358/13,19,311,314,320,326,472,505,311,314,320,326 ;328/139 |
References Cited
[Referenced By]
U.S. Patent Documents
Primary Examiner: Coles, Sr.; Edward L.
Assistant Examiner: Stoll; Thomas L.
Attorney, Agent or Firm: Schumacher; Robert K. Samples;
Kenneth H. Mossino; Ralph L.
Claims
What is claimed is:
1. A method of sampling a composite analog color television signal
whereby the samples are taken at precise locations in the
television signal relative to the phase of the color burst thereof,
comprising the steps of:
storing reference samples obtained from color burst cycles of a
horizontal television line at times determined by a phase stable
signal and generating an output clock signal that is phase
synchronized with said reference samples;
sampling the color burst of subsequent horizontal television lines
at times determined by said output clock signal and storing a set
of comparison samples of color burst cycles;
examining each set of stored comparison samples of color burst
cycles and detecting a change in the sample phase location thereof
and providing an error signal indicative of the magnitude of the
change;
adjusting the phase of the generated output clock signal in
response to the error signal when the magnitude of the phase change
is within a predetermined limit; and,
storing new reference samples of color burst cycles when said
magnitude of phase change exceeds said predetermined limit and
generating a new output clock signal that is phase synchronized
with said new reference samples.
2. A method as defined in claim 1 wherein the television signal is
sampled at a rate that is three times the frequency of the color
subcarrier of said analog color television signal.
3. A method as defined in claim 2 wherein the television signal is
sampled at phase locations for each cycle of the subcarrier,
relative to the 0.degree., 120.degree. and 240.degree. phase
locations of the color burst.
4. A method as defined in claim 1 wherein said comparison samples
that are stored and examined are taken from color burst cycles
occurring during the next succeeding horizontal television line
following the line from which said reference samples are taken and
thereafter during alternate succeeding horizontal television
lines.
5. A method of sampling a composite analog color television signal
whereby the samples are taken in precise locations in the
television signal relative to the phase of the color burst thereof,
comprising the steps of:
storing reference samples obtained from color burst cycles of a
horizontal television line at times determined by a phase stable
signal and generating an output clock signal that is phase
synchronized with said reference samples;
sampling the color burst of subsequent horizontal lines at times
determined by said output clock signal and storing a set of
comparison samples of color burst cycles;
examining each set of stored comparison samples of color burst
cycles and detecting a change in the sample phase location thereof
and providing an error signal indicative of the magnitude of the
phase change; and
adjusting the phase of the generated output clock signal in
response to the error signal.
6. A method of sampling a composite analog color television signal
whereby the samples are taken in precise locations in the
television signal relative to the phase of the color burst thereof,
comprising the steps of:
storing reference samples obtained from color burst cycles of a
horizontal television line at times determined by a phase stable
signal and generating an output clock signal that is phase
synchronized with said reference samples;
sampling the color burst of subsequent horizontal television lines
at times determined by said output clock signal and storing a set
of comparison samples of color burst cycles;
examining each set of stored comparison samples of color burst
cycles and detecting a change in the sample phase location thereof;
and,
storing new reference samples of color burst cycles when a phase
change is detected and generating a new output clock signal that is
phase synchronized with said new reference samples.
7. A method of sampling a composite analog color television signal
whereby the samples are taken in precise locations in the
television signal relative to the phase of the color burst thereof,
comprising the steps of:
storing reference samples obtained from color burst cycles of a
horizontal television line at times determined by a phase stable
signal and generating an output clock signal that is phase
synchronized with said reference samples;
sampling the color burst of subsequent horizontal television lines
at times determined by said output clock signal and storing a set
of comparison samples of color burst cycles;
examining each set of stored comparison samples of color burst
cycles and detecting a change in the sample phase location thereof;
and,
storing new reference samples of color burst cycles when a phase
change that exceeds a predetermined magnitude is detected and
generating a new output clock signal that is phase synchronized
with said new reference samples.
8. Apparatus for generating an output clock signal that is phase
synchronized to the color burst of a composite analog color
television signal having a color subcarrier component for use in
sampling the analog color television signal, comprising:
means for sampling said analog color television signal at times
determined by a clock signal having a sampling rate that is a
multiple of the frequency of said color subcarrier component and
generating samples of said analog color television signal;
means for storing reference samples of color burst cycles of a
horizontal line of the color television signal generated by said
sampling means clocked by a phase stable clock signal and
generating said output clock signal that is phase synchronized with
said reference samples;
means for storing sets of comparison samples of color burst cycles
from subsequent horizontal lines of said color television signal
generated by said sampling means clocked by said output clock
signal;
means for detecting a change between the phase of each set of
stored comparison samples and the phase of the reference samples
and generating an error signal indicative of the magnitude of any
phase change;
means for adjusting the phase of the generated output clock signal
in response to the error signal generated when the magnitude of the
phase change is within predetermined limits; and,
means for activating said reference sample storing means to store
new reference samples of burst cycles in response to the magnitude
of said error signal being outside of said predetermined limits and
thereby generate a new output clock signal that is phase
synchronized with said new reference samples.
9. Apparatus as defined in claim 8 wherein said means for storing
said sets of comparison samples is controlled to store said samples
from color burst cycles that occur during subsequent and
alternately succeeding horizontal lines.
10. Apparatus as defined in claim 8 wherein said sampling rate is
three times the frequency of the color subcarrier component of said
analog color television signal.
11. Apparatus as defined in claim 10 wherein said sampling means
samples said analog color television signal at phase locations
corresponding to the 0.degree., 120.degree. and 240.degree. phase
locations of said color burst.
12. Apparatus as defined in claim 11 wherein said phase change
detecting means comprises:
means for arithmatically adding the values of said 120.degree. and
240.degree. sample locations of said color burst to provide a
resultant value, and
means for comparing said resultant value with the value of said
0.degree. sample location of said color burst and providing an
error signal having a magnitude that is proportional to the
difference therebetween.
13. Apparatus for controlling the phase locations of the sampling
of a composite analog color television signal which is sampled rate
that is a multiple of the frequency of the color subcarrier
component of said analog color television signal, comprising:
means for sampling said analog color television signal at said
sampling rate and at specific locations determined by a clock
signal applied thereto and generating samples of said analog color
television signal;
first means for storing a set of reference samples of color burst
cycles of a horizontal line generated by said sampling means when
the clock signal applied thereto is phase stable and generating an
output clock signal therefrom that is phase synchronized with said
reference samples;
means for adjusting the phase of said generated output clock signal
in response to phase error signals being applied thereto;
second means for storing a set of comparison samples of said color
burst cycles from the color burst of subsequent horizontal lines
generated by said sampling means when the clock signal applied
thereto is the phase adjusted output clock signal from said phase
adjusting means;
means coupled to said second sample storing means for examining the
comparison samples stored therein and for generating phase error
signals in response to the detection of any phase differences
between the phase of said reference samples and the phase of said
comparison samples and applying said phase error signals to said
phase adjusting means;
means for monitoring said error signals and for activating said
first storing means to store a new set of reference samples of
color burst cycles in the event the error signal exceeds a
predetermined limit.
14. Apparatus as defined in claim 13 wherein said sampling means
samples said analog color television signal at the phase locations
corresponding to the 0.degree., 120.degree. and 240.degree. phase
locations of said color burst.
15. Apparatus as defined in claim 14 wherein said examining and
error signal generating means comprises:
means for arithmatically adding the values of said comparison
samples taken at 120.degree. and 240.degree. sample locations of
said color burst to obtain a first resultant value, and
means for comparing said first resultant value with the value of
said comparison sample taken at 0.degree. sample location of said
color burst and providing an error signal when said first resultant
value and said sample value of said 0.degree. sample location are
unequal.
16. Apparatus as defined in claim 15 wherein said error signal
monitoring means comprises level detecting means adapted to provide
an activating signal to said first storing means when the magnitude
of said error signal exceeds said predetermined limit.
17. Apparatus for generating an output clock signal that is phase
synchronized to the color burst of a composite analog color
television signal having a color subcarrier component for use in
sampling the analog color television signal, comprising:
means for sampling said analog color television signal at times
determined by a clock signal having a sampling rate that is a
multiple of the frequency of the color subcarrier component and
generating samples of said analog color television signal;
.[.means for storing reference samples of color burst cycles of a
horizontal line of the television signal generated by said sampling
means clocked by a phase stable clock signal and generating said
output clock signal that is phase synchronized with said reference
samples;.].
means for .[.storing sets of.]. .Iadd.generating
.Iaddend.comparison samples of color burst cycles from
.[.subsequent.]. horizontal lines of the television signal
generated by said sampling means clocked by said output clock
signal;
means for detecting a change between the phase of .[.each set of
stored.]. .Iadd.the .Iaddend.comparison samples and the phase of
the .[.reference samples.]. .Iadd.output clock signal .Iaddend.and
generating an error signal indicative of the magnitude of any phase
change; and
means for adjusting the phase of the generated output clock signal
in response to the error signal.
18. Apparatus for generating an output clock signal that is phase
synchronized to the color burst of a composite analog color
television signal having a color subcarrier component for use in
sampling the analog color television signal, comprising:
means for sampling said analog color television signal at times
determined by a clock signal having a sampling rate that is a
multiple of the frequency of the color subcarrier component and
generating samples of said analog color television signal;
means for storing reference samples of color burst cycles of a
horizontal line of the television signal generated by said sampling
means clocked by a phase stable clock signal and generating said
output clock signal that is phase synchronized with said reference
samples;
means for storing sets of comparison samples of color burst cycles
from subsequent horizontal lines of the television signal generated
by said sampling means clocked by said output clock signal;
means for detecting a change between the phase of each set of
stored comparison samples and the phase of the reference samples;
and
means for activating said reference sample storing means to store
new reference samples of burst cycles in response to the detection
of a phase change and thereby generate a new output clock signal
that is phase synchronized with said new reference samples.
19. Apparatus for generating an output clock signal that is phase
synchronized to the color burst of a composite analog color
television signal having a color subcarrier component for use in
sampling the analog color television signal, comprising:
means for sampling said analog color television signal at times
determined by a clock signal having a sampling rate that is a
multiple of the frequency of the color subcarrier component and
generating samples of said analog color television signal;
means for storing reference samples of color burst cycles of a
horizontal line of the television signal generated by said sampling
means clocked by a phase stable clock signal and generating said
output clock signal that is phase synchronized with said reference
samples;
means for storing sets of comparison samples of color burst cycles
from subsequent horizontal lines of the television signal generated
by said sampling means clocked by said output clock signal;
means for detecting a change between the phase of each set of
stored comparison samples and the phase of the reference samples;
and
means for activating said reference sample storing means to store
new reference samples of burst cycles in response to the detection
of a phase change that exceeds a predetermined magnitude and
thereby generate a new output clock signal that is phase
synchronized with said new reference samples.
20. A method of sampling an information signal having a time-base
synchronizing component of a known frequency whereby the samples
are taken in precise locations relative to the synchronizing
component thereof, comprising the steps of:
storing reference samples obtained from said synchronizing
component at times determined by a phase stable clock signal and
generating an output clock signal that is phase synchronized with
said reference samples; periodically sampling the synchronizing
component at times determined by said output clock signal to obtain
a set of comparison samples thereof;
examining each set of comparison samples and detecting a change in
the sample phase location thereof and providing an error signal
indicative of the magnitude of the change;
adjusting the phase of the generated output clock signal in
response to the error signal when the magnitude of the phase change
is within a predetermined limit; and
storing new reference samples of said synchronizing component when
said magnitude of phase change exceeds said predetermined limit and
generating a new output clock signal that is phase synchronized
with said new reference samples.
21. A method of sampling an information signal having a time-base
synchronizing component of a known frequency whereby the samples
are taken in precise locations relative to the synchronizing
component thereof, comprising the steps of:
storing reference samples obtained from said synchronizing
component at times determined by a phase stable clock signal and
generating an output clock signal that is phase synchronized with
said reference samples;
periodically sampling the synchronizing component at times
determined by said output clock signal to obtain a set of
comparison samples thereof;
examining each set of comparison samples and detecting a change in
the sample phase location thereof and providing an error signal
indicative of the magnitude of the change; and
adjusting the phase of the generated output clock signal in
response to the error signal.
22. A method of sampling an information signal having a time-base
synchronizing component of a known frequency whereby the samples
are taken in precise locations relative to the synchronizing
component thereof, comprising the steps of:
storing reference samples obtained from said synchronizing
component at times detemined by a phase stable clock signal and
generating an output clock signal that is phase synchronized with
said reference samples;
periodically sampling the synchronizing component at times
determined by said output clock signal to obtain a set of
comparison samples thereof;
examining each set of comparison samples and detecting a change in
the sample phase location thereof; and
storing new reference samples of said synchronizing component when
a change is detected and generating a new output clock signal that
is phase synchronized with said new reference samples.
23. A method of sampling an information signal having a time-base
synchronizing component of a known frequency whereby the samples
are taken in precise locations relative to the synchronizing
component thereof, comprising the steps of:
storing reference samples obtained from said synchronizing
component at times determined by a phase stable clock signal and
generating an output clock signal that is phase synchronized with
said reference samples;
periodically sampling the synchronizing component at times
determined by said output clock signal to obtain a set of
comparison samples thereof;
examining each set of comparison samples and detecting a change in
the sample phase location thereof; and
storing new reference samples of said synchronizing component when
a phase change that exceeds a predetermined magnitude is detected
and generating a new output clock signal that is phase synchronized
with said new reference samples.
24. Apparatus for generating an output clock signal that is phase
synchronized to a time-base synchronizing component of a known
frequency associated with an information signal for use in sampling
the information signal, comprising:
means for sampling said synchronizing component at times determined
by a clock signal having a sampling rate that is a multiple of the
frequency of said synchronizing component and generating samples
thereof;
means for storing reference samples of said synchronizing component
generated by said sampling means clocked by a phase stable clock
signal and generating said output clock signal that is phase
synchronized with said reference samples;
means for storing sets of comparison samples of said synchronizing
components generated by said sampling means clocked by said output
clock signal;
means for detecting a change between the phase of the comparison
samples and the phase of the reference samples and generating any
error signal indicative of the magnitude of an phase change;
means for adjusting the phase of the generated output clock signal
in response to the error signal generated when the magnitude of the
phase change is within predetermined limits; and
means for activating said reference sample storing means to store
new reference samples of said synchronizing component in response
to the magnitude of said error signal being outside of said
predetermined limits and thereby generate a new output clock signal
that is phase synchronized with said new reference samples.
25. Apparatus for generating an output clock signal that is phase
synchronized to a time-base synchronizing component of a known
frequency associated with an information signal for use in sampling
the information signal, comprising:
means for sampling said synchronizing component at times determined
by .[.a.]. .Iadd.said output .Iaddend.clock signal having a
sampling rate that is a multiple of the frequency of said
synchronizing component and generating samples thereof;
.[.means for storing reference samples of said synchronizing
component generated by said sampling means clocked by a phase
stable clock signal and generating said output clock signal that is
phase synchronized with said reference samples.].
means for .[.storing sets of.]. .Iadd.generating
.Iaddend.comparison samples of said synchronizing component
generated by said sampling means clocked by said output clock
signal;
means for detecting a change between the phase of the comparison
samples and the phase of the said .[.reference samples.].
.Iadd.output clock signal .Iaddend.and generating an error signal
indicative of the magnitude of any phase change; and
means for adjusting the phase of the generated output clock signal
in response to the error signal.
26. Apparatus for generating an output clock signal that is phase
synchronized to a time-base synchronizing component of a known
frequency associated with an information signal for use in sampling
the information signal, comprising:
means for sampling said synchronizing component at times determined
by a clock signal having a sampling rate that is a multiple of the
frequency of said synchronizing component and generating samples
thereof;
means for storing reference samples of said synchronizing component
generated by said sampling means clocked by a phase stable clock
signal and generating said output clock signal that is phase
synchronized with said reference samples;
means for storing sets of comparison samples of said synchronizing
component generated by said sampling means clocked by said output
clock signal;
means for detecting a change between the phase of the comparison
samples and the phase of the reference samples; and
means for activating said reference samples storing means to store
new reference samples of said synchronizing component when a phase
change is detected and thereby generate a new output clock signal
that is phase synchronized with said new reference samples.
27. Apparatus for generating an output clock signal that is phase
synchronized to a time-base synchronizing component of a known
frequency associated with an information signal for use in sampling
the information signal, comprising:
means for sampling said synchronizing component at times determined
by a clock signal having a sampling rate that is a multiple of the
frequency of said synchronizing component and generating samples
thereof;
means for storing reference samples of said synchronizing component
generated by said sampling means clocked by a phase stable clock
signal and generating said output clock signal that is phase
synchronized with said reference samples;
means for storing sets of comparison samples of said synchronizing
component generated by said sampling means clocked by said output
clock signal;
means for detecting a change between the phase of the comparison
samples and the phase of the reference samples; and
means for activating said reference sample storing means to store
new reference samples of said synchronizing component in response
to the detection of a phase change that exceeds a predetermined
magnitude and thereby generate a new output clock signal that is
phase synchronized with said new reference samples. .Iadd.
28. In a system for digitizing a color television signal wherein
the color television signal is sampled with a sampling signal whose
rate is a multiple of the burst signal frequency and each sample is
converted to a corresponding digital signal, a method for
controlling the phase of the sampling signal comprising the steps
of
obtaining at least three samples of the burst signal including in
said color television signal;
determining a reference sampling phase as a function of at least
two of said samples by deriving a calculated sample level from said
at least two samples;
sensing a phase error between the actual sampling phase of said
sampling signal and said reference sampling phase by detecting the
difference between said calculated sample level and the level of a
predetermined one of said at least three samples; and
adjusting the phase of said sampling signal in accordance with said
detected difference in a direction to reduce said detected
difference to a null value. .Iaddend. .Iadd.29. The method of claim
28 wherein said predetermined sample differs from both of said at
least two samples. .Iaddend. .Iadd.30. The method of claim 29
wherein the rate of said sampling signal is an even multiple of
twice the burst signal frequency.
.Iaddend. .Iadd.31. The method of claim 28 wherein said step of
adjusting the phase of said sampling signal comprises
generating a control voltage, changing the magnitude of said
control voltage by an incremental amount during each period of said
burst signal, the change being a function of whether the level of
said predetermined one sample exceeds or is less than said
calculated sample level;
and adjusting the phase of said sampling signal in accordance with
said control voltage. .Iaddend. .Iadd.32. A method of determining
whether the burst signal of a color television signal is sampled in
a predetermined phase relationship, comprising the steps of
sampling said burst signal at a rate which is a multiple of the
burst signal frequency;
determining a theoretical sample level of said burst signal as a
function of a plurality of the actual sample levels thereof, said
theoretical sample level representing a sample level that would be
obtained if said burst signal is sampled in said predetermined
phase relationship; and
detecting the difference between said theoretical sample level and
a predetermined one of said actual sample levels. .Iaddend.
.Iadd.33. The method of claim 32 wherein said step of determining a
theoretical sample level of said burst signal comprises
obtaining the difference between two actual sample levels; and
multiplying said difference by a predetermined constant. .Iaddend.
.Iadd.34. Apparatus for use in a digital encoder for a color
television signal, comprising
sampling pulse generating means for generating sampling pulses with
a rate that is a multiple of the burst signal frequency of said
color television signal;
sampling means responsive to said sampling pulses for sampling said
color television signal, including said burst signal, and for
digitizing each sample;
theoretical sample level producing means for producing a
theoretical sample level of said burst signal as a function of at
least two actual samples thereof, said theoretical sample level
representing a sample that would be obtained if said burst signal
is sampled by sampling pulses which exhibit a predetermined phase
with respect thereof;
phase difference determining means for determining the difference
between the actual phase of said sampling pulses and said
predetermined phase in accordance with the difference between said
theoretical sample level and an actual sample; and
adjusting means responsive to said determined phase difference to
adjust the phase of said sampling pulses so as to reduce said
determined phase
difference to a null value. .Iaddend. .Iadd.35. The apparatus of
claim 34 wherein
the rate of said sampling pulses is an even multiple of twice the
burst signal frequency. .Iaddend. .Iadd.36. The apparatus of claim
35 wherein said adjusting means comprises
control voltage generating means for generating a control voltage
to control the phase of said sampling pulses; and
means for changing said control voltage in accordance with said
difference between said theoretical sample level and an actual
sample. .Iaddend. .Iadd.37. The apparatus of claim 36 wherein said
means for changing said control voltage comprises
means for increasing said control voltage by an incremental amount
during a period of said burst signal when said theoretical sample
level exceeds said actual sample; and
means for decreasing said control voltage by said incremental
amount during a period of said burst signal when said theoretical
sample level is less
than said actual sample. .Iaddend. .Iadd.38. The apparatus of claim
37 further comprising capacitance means; and wherein
said means for increasing said control voltage comprises charging
means for charging said capacitance means for a predetermined
duration, and
said means for decreasing said control voltage comprises discharge
means for discharging said capacitance means for said predetermined
duration. .Iaddend. .Iadd.39. Apparatus for determining if the
burst signal of a color television signal is sampled at a
predetermined phase angle thereof, comprising sampling means for
sampling said burst signal at a rate that is a multiple of the
burst signal frequency; theoretical sample level producing means
for producing a theoretical sample level of said burst signal as a
function of a plurality of the actual sample levels thereof, said
theoretical sample level representing a sample that would be
obtained if said burst signal is sampled at said predetermined
phase angle; and determining means responsive to said theoretical
sample level and a predetermined one of said actual sample levels
for determining whether an actual sample of said burst signal is
produced at said predetermined phase
angle. .Iaddend. .Iadd.40. In a system for digitizing a color
television signal wherein the color television signal is sampled
with a sampling signal whose rate is a multiple of the burst signal
frequency and each sample is converted to a corresponding digital
signal, a method for controlling the phase of the sampling signal
comprising the steps of obtaining at least three samples of the
burst signal included in said color television signal; determining
a sampling phase as a function of at least two of said samples by
deriving a calculated sample level from said at least two samples;
sensing a phase error between the actual sampling phase of said
sampling signal and said sampling phase by detecting the difference
between said calculated sample level and the level of a
predetermined one of said at least three samples; and adjusting the
phase of said sampling signal in accordance with said detected
difference in a direction to reduce said detected difference to a
null value. .Iaddend. .Iadd.41. The method of claim 40 wherein said
predetermined sample differs
from both of said at least two samples. .Iaddend. .Iadd.42. The
method of claim 41 wherein the rate of said sampling signal is an
even multiple of twice the burst signal frequency. .Iaddend.
.Iadd.43. The method of claim 1 wherein said step of adjusting the
phase of said sampling signal comprises generating a control
voltage, changing the magnitude of said control voltage by an
incremental amount during each period of said burst signal, the
change being a function of whether the level of said predetermined
one sample exceeds or is less than said calculated sample level;
and adjusting the phase of said sampling signal in accordance with
said control voltage. .Iaddend. .Iadd.44. A method of determining
whether the burst signal of a color television signal is sampled in
a predetermined phase relationship, comprising the steps of
sampling said burst signal at a rate which is a multiple of the
burst signal frequency; determining a sample level of said burst
signal as a function of a plurality of the actual sample levels
thereof, said sample level representing a sample level that would
be obtained if said burst signal is sampled in said predetermined
phase relationship; and detecting the difference between said
sample level and a predetermined one of said actual sample levels.
.Iaddend. .Iadd.45. The method of claim 44 wherein said step of
determining a sample level of said burst signal comprises obtaining
the difference between two actual sample levels; and multiplying
said difference by a predetermined constant. .Iaddend. .Iadd.46.
Apparatus for use in a digital encoder for a color television
signal, comprising sampling pulse generating means for generating
sampling pulses with a rate that is a multiple of the burst signal
frequency of said color television signal; sampling means
responsive to said sampling pulses for sampling said color
television signal, including said burst signal, and for digitizing
each sample; sample level producing means for producing a sample
level of said burst signal as a function of at least two actual
samples thereof, said sample level representing a sample that would
be obtained if said burst signal is sampled by sampling pulses
which exhibit a predetermined phase with respect thereof; phase
difference determining means for determining the difference between
the actual phase of said sampling pulses and said predetermined
phase in accordance with the difference between said sample level
and an actual sample; and adjusting means responsive to said
determined phase difference to adjust the phase of said sampling
pulses so as to reduce said determined phase difference to a null
value. .Iaddend. .Iadd.47. The apparatus of claim 46 wherein the
rate of said sampling pulses is an even multiple of twice the burst
signal frequency. .Iaddend. .Iadd.48. The apparatus of claim 46
wherein said adjusting means comprises control voltage generating
means for generating a control voltage to control the phase of said
sampling pulses; and means for changing said control voltage in
accordance with said difference between said sample level and an
actual sample. .Iaddend. .Iadd.49. The apparatus of claim 48
wherein said means for changing said control voltage comprises
means for increasing said control voltage by an incremental amount
during a period of said burst signal when said sample level exceeds
said actual sample; and means for decreasing said control voltage
by said incremental amount during a period of said burst signal
when said sample level is less than said actual sample. .Iaddend.
.Iadd.50. The apparatus of claim 49 further comprising capacitance
means; and wherein said means for increasing said control voltage
comprises charging means for charging said capacitance means for a
predetermined duration, and said means for decreasing said control
voltage comprises discharge means for discharging said capacitance
means for said predetermined duration. .Iaddend. .Iadd.51.
Apparatus for determining if the burst signal of a color television
signal is sampled at a predetermined phase angle thereof,
comprising sampling means for sampling said burst signal at a rate
that is a multiple of the burst signal frequency; sample level
producing means for producing a sample level of said burst signal
as a function of a plurality of the actual sample levels thereof,
said sample level representing a sample that would be obtained if
said burst signal is sampled at said predetermined phase angle; and
determining means responsive to said sample level and a
predetermined one of said actual sample levels for determining
whether an actual sample of said burst signal is produced at
said
predetermined phase angle. .Iaddend. .Iadd.52. Apparatus as set
forth in claim 17 which further includes:
means for storing reference samples of color burst cycles of a
horizontal line of the television signal generated by said sampling
means and clocked by a phase stable clock signal and for generating
said output clock signal phase synchronized with said reference
samples. .Iaddend. .Iadd.53. Apparatus as set forth in claim 52
wherein:
said means for generating comparison samples stores sets of
comparison samples of color burst cycles from subsequent horizontal
lines; and
said detecting means detects a change between the phase of each set
of stored comparison samples and the phase of the reference
samples. .Iaddend. .Iadd.54. An apparatus as set forth in claim 25
which further includes:
means for storing reference samples of said synchronizing component
generated by said sampling means clocked by a phase stable clock
signal and for generating said output clock signal phase
synchronized with said reference samples. .Iaddend. .Iadd.55. An
apparatus as set forth in claim 54 wherein:
said means for generating comparison samples stores sets of
comparison samples of said synchronizing component; and
said detecting means detects a change between the phase of the
comparison samples and the phase of the reference samples.
.Iaddend. .Iadd.56. A method of controlling the phase of a clock
signal which effects sampling of a color television signal having a
color burst signal, the clock signal having a desired sampling
phase relative to the color burst signal, said method comprising
the steps of:
sampling a plurality of values of the color burst signal at the
actual phase of the sampling clock relative to the color burst
signal;
generating an error signal as a function of a plurality of said
sampled values which is indicative of a difference between the
actual sampling phase and the desired sampling phase; and
adjusting the phase of the clock signal in response to said error
signal to
eliminate the phase difference. .Iaddend. .Iadd.57. A method as set
forth in claim 56 wherein the step of adjusting the phase of the
clock signal includes the step of:
adjusting the phase of the clock signal in response to said error
signal to eliminate the phase difference when the error signal is
within a selected range; and
rephasing the clock signal such that it is phase synchronized with
a reference when the error signal is outside of said selected
range. .Iaddend. .Iadd.58. A method of sampling a color television
signal having a color burst signal which obtains sample values of
the color television signal relative to a selected phase of the
color burst signal, the sampling of the color television signal
producing sample values which are a function of the actual sampling
phase of a sampling clock signal relative to the color burst
signal, the sampling clock signal having a desired sampling phase
producing the sample values of the color television signal at the
selected phase, said method comprising the steps of:
generating the sampling clock signal with a controllable phase;
separating samples of the color burst signal from the samples of
the color television signal to obtain sample values thereof which
are determined by the actual sampling phase of the sampling
clock;
determining, from a plurality of sample values of the color burst
signal, the actual sampling phase of the sampling clock signal;
and
controlling the phase of the sampling clock signal in accordance
with the determined sampling phase to adjust the sampling phase to
the desired
sampling phase. .Iaddend. .Iadd.59. The method as set forth in
claim 58 further comprising the steps of:
converting each obtained sample value to a digital representation
prior to the step of determining the sampling phase. .Iaddend.
.Iadd.60. The method of claim 58 wherein the step of controlling
the phase of the sampling clock signal includes the steps of:
controlling the phase of the sampling clock signal in accordance
with the determined sampling phase to adjust the sampling phase to
the desired sampling phase when the difference between the
determined sampling phase and the desired sampling phase is within
a selected range; and
rephasing the sampling clock signal such that it is phase
synchronized to a reference when the difference between the
determined sampling phase and the desired sampling phase is outside
of said selected range. .Iaddend. .Iadd.61. A method for sampling
an analog signal having a reference signal component by controlling
the phase of a sampling clock signal to a desired sampling phase
relative to the reference signal component, said method
comprising:
sampling the reference signal with the sampling clock to produce
sample values thereof;
determining from the sample values of said reference signal the
actual phase relation between said sampling clock signal and the
reference signal component;
determining from the sample values of said reference signal the
desired phase relation between said sampling clock signal and the
reference signal component;
generating an error signal indicative of a difference between said
actual sampling phase and said desired sampling phase; and
adjusting the phase of said sampling clock signal in accordance
with the error signal to eliminate the difference between the
actual phase and the
desired phase. .Iaddend. .Iadd.62. A method as set forth in claim
61 wherein the sampled signal is a composite television signal and
the reference signal component is the color burst synchronizing
signal of the composite signal. .Iaddend. .Iadd.63. A method as set
forth in claim 61 which further includes the step of:
converting the sampled analog signal into digital values at times
based upon the sampling clock. .Iaddend. .Iadd.64. A method as set
forth in claim 63 wherein said step of determining the actual phase
includes the step of:
utilizing one of said sampled reference signal values as an
indication of the actual phase relation between said sampling clock
signal and said reference signal. .Iaddend. .Iadd.65. A method as
set forth in claim 64 wherein said step of determining the desired
phase includes the step of:
utilizing at least one other of said sampled reference signal
values to calculate a value for said one reference sample value
which is an indication of the desired phase relation between said
sampling clock signal and said reference signal. .Iaddend.
.Iadd.66. A method as set forth in claim 65 wherein said step of
generating said error signal includes the step of:
differentially combining said actual phase indication value and
said desired phase indication value. .Iaddend.
Description
CROSS REFERENCE TO RELATED APPLICATIONS
Digital Recording and Reproducing Apparatus, Ser. No. 117,428,
filed Feb. 1, 1980, by Maurice G. Lemoine and Leonard A.
Pasdera.
High Frequency Digital PCM Decoding Apparatus, Ser. No. 117,881,
filed Feb. 1, 1980, by Maurice G. Lemoine and Leonard A.
Pasdera.
Parity Checking Circuitry for Use in Multi-Bit Cell PCM Recording
and Reproducing Apparatus, Ser. No. 117,745, filed Feb. 1, 1980, by
Maurice G. Lemoine.
Apparatus for Providing Drop-Out Compensation in Recording and
Reproducing Systems, Ser. No. 117,422, filed Feb. 1, 1980, by
Maurice G. Lemoine and Leonard A. Pasdera.
BACKGROUND AND FIELD OF THE INVENTION
The present invention generally relates to a method and apparatus
for taking samples of an analog television signal that is
particularly suited for use in a magnetic recording and reproducing
apparatus.
As is comprehensively set forth in our copending patent application
Ser. No. 117,428, a digital recording and reproducing apparatus is
disclosed which has many significant advantages compared to most
commercial quality television broadcasting recording and
reproducing apparatus which record and reproduce FM signals. In our
aforementioned application, pluse code modulated digital video
signals are recorded and reproduced. The digital recording and
reproducing apparatus has, when compared to FM recording and
reproducing, the desirable attributes of virtually eliminating
banding and Moire of any nature from any cause, reducing chroma and
luminance noise to a value better than -54 db, relaxing mechanical
tolerances for quadrature adjustments in a quadruplex format
recording apparatus by a factor of about 100 and providing an
inherent time base error that is no greater than about 1/2
nanosecond.
The present invention also enables the aforementioned recording and
reproducing apparatus to eliminate the problem of cycle hops which
is present in conventional video tape recorders, for the reason
that the decision for making the zero crossing selection for
sampling the television signal is precisely determined with respect
to the location of a horizontal synchronization pulse which is
generated and which is precisely defined with respect to color
burst phase and therefore the phase of the unmodulated color
subcarrier and this is done before recording so that consistent
reliable reproducing can be achieved.
The present invention is also broadly applicable to information
signals other than a composite color television signal provided
that the information signal is of the type which has a time-base
synchronizing component. Such signals are used in wideband
telemetry systems wherein the wideband telemetry signal has a pilot
signal that represents the time-base synchronizing component.
OBJECTS OF THE INVENTION
Accordingly, it is an object of the present invention to provide an
improved method and apparatus for sampling an information signal
having a time-base synchronizing component.
It is also an object of the present invention to provide an
improved method and apparatus for sampling the composite analog
color television signal.
Another object of the present invention is to provide an improved
method and apparatus for sampling and digitizing the composite
analog color television signal in a manner whereby the samples are
taken at precise locations relative to the phase of the unmodulated
color subcarrier.
Yet another important object is to provide a digital sampling
method and apparatus of the foregoing type which is extremely fast
acting in that it can rapidly adjust to take samples in the precise
locations relative to the phase of the unmodulated color
subcarrier, even when a new input television signal is provided
having a radically different phase relationship from a prior input
signal.
Still another object of the present invention is to provide a
digital sampling method and apparatus of the foregoing type which
can make adjustments in the sampling locations to compensate for
small phase error that occur during operation, in addition to being
capable of rapidly adjusting the sampling when an extremely large
error is experienced, such as would occur during a "wild switch" or
the like.
Other objects and advantages will become apparent upon reading the
following detailed description, while referring to the attached
drawings.
DESCRIPTION OF THE DRAWINGS
FIG. 1 is a system block diagram for the digital recording and
reproducing apparatus that is described herein.
FIG. 2 is a simplified end view of a rotatable head wheel carrying
a plurality of transducing heads that can be incorporated in the
apparatus described herein.
FIG. 3 is a simplified plan view of a segment of a magnetic tape,
generally illustrating the quadruples recording format including
the transversely recorded television signal data tracks and the
longitudinally recorded cue, control and audio tracks.
FIGS. 4a and 4b are timing diagrams which illustrate the
relationship to the timing sequences that occur during operation of
portions of the apparatus described herein during a recording
operation.
FIGS. 5a and 5b are timing diagrams which illustrate the
relationship of the timing sequences that occur during operation of
portions of the apparatus described herein during a reproducing
operation.
FIG. 6 illustrates the relation of a single line of a color
television signal with the horizontal sync pulse and the color
burst interval contained in the horizontal blanking interval,
together with the relative timing of digital synchronizing
information that is inserted in a portion of the horizontal
blanking interval for each line.
FIG. 7 is an electrical schematic diagram of circuitry for
controlling the random access memories during a recording
operation.
FIG. 8 is an electrical schematic diagram of circuitry for
controlling the random access memories during a reproducing
operation.
FIG. 9 is an electrical schematic diagram of logic and clock
generating circuitry that is used for controlling the operation of
the random access memories during either a recording or reproducing
operation.
FIG. 10 is an electrical schematic diagram of additional circuitry
that is used for controlling the random access memories during a
reproducing operation.
FIG. 11 is a functional block diagram of circuitry that is used to
adjust the phase relation of the sampling of the analog color
television signal, so that the samples are taken at proper
locations with respect to the phase of the color subcarrier of the
composite color television signal.
FIG. 12 is a functional block diagram of circuitry for inserting
the digital synchronizing sequence that is added in during the
horizontal blanking interval as shown in FIG. 6, lines (2), (3) and
(4).
FIG. 13 is an electrical schematic diagram of one of the random
access memories, with portions removed for purpose of drawing
economy.
FIG. 14a and 14b together comprises an electrical schematic diagram
of the 24-to-8 bit converter, parallel-to-serial converter and
encoder.
FIG. 15 is an electrical schematic diagram of one of the
preamplifying circuits of the apparatus described herein.
FIGS. 16a and 16b together comprise an electrical schematic diagram
of one of the equalizer circuits that is used to correct for
inter-symbol interference of the off-tape reproduce signal.
FIG. 16c(1) and (2) are graphs illustrating the waveform and depth
of recording respectively, of a portion of a data stream.
FIGS. 17a and 17b together comprise an electrical schematic diagram
of decoder circuitry, drop-out processing circuitry, off-tape clock
acquisition circuitry and serial-to-parallel converting
circuitry.
FIGS. 18a and 18b together comprise an electrical schematic diagram
of an 8-to-24 bit converter, a 2-to-1 switch, identification number
decoders, drop-out processing circuitry and parity checking
circuitry.
FIGS. 19a and 19b together comprise an electrical schematic diagram
of one form of specific circuitry that can be used to carry out the
operation of the block diagram of FIG. 11.
FIGS. 20a, 20b, 20c, 20d, 20e, 20f and 20g comprise the electrical
schematic diagrams of circuitry that can be used to carry out the
operation of the block diagram of FIG. 12.
FIG. 21 is an electrical schematic diagram of a 2-to-1 switch.
FIG. 22 is a diagram illustrating a single cycle of color
subcarrier and the proper phase relation when sampling is correctly
performed, together with a single subcarrier cycle shown in phantom
with the sampling being performed at incorrect phase locations.
FIG. 23 is a block diagram of a portion of the dropout compensator
that can be used in the present apparatus.
FIG. 24 is a group of diagrams illustrating the relationship of the
timing sequences that occur during operation of one embodiment of
the drop-out compensator.
FIGS. 25a and 25b together comprise an electrical schematic diagram
of a portion of the memory control circuitry for the memory of the
drop-out compensator shown in FIGS. 26A and 26B.
FIGS. 26a and 26b together comprise an electrical schematic diagram
of the memory of the drop-out compensator.
FIGS. 27a and 27b together comprise the switching circuitry that
can be used to carry out the operation of the drop-out compensator
shown in FIG. 23.
FIG. 28 is a block diagram illustrating the servo control system of
the recording and reproducing apparatus described herein.
DETAILED DESCRIPTION OF THE APPARATUS
Turning now to the drawings, and particularly the functional block
diagram of FIG. 1 which broadly illustrates the recording and
reproducing apparatus described herein, it is shown to have a
number of blocks that are interconnected with broad solid lines
that are intended to illustrate the signal flow path during a
recording operation, together with broad cross hatched lines which
illustrate the signal path during a reproducing operation. The
relatively thin lines reflect control signals, clock signals and
other signals which do not specifically define the signal flow path
of the video signals. It should be understood that the width of the
lines are not intended to reflect the number of separate parallel
conductors or lines that exist and, as will be fully explained
herein, the signal path may be comprised of a single line serial
data, or eight bits of parallel data or 24 bits of parallel data.
The apparatus will be broadly described in conjunction with the
block diagram of FIG. 1, first during a recording process and
subsequently for a reproducing process, although certain blocks are
utilized during both processes and may be described with respect to
both processes when they are initially introduced.
The composite analog color television input signal is applied via
line 30 to an input processing circuit 32 which performs various
functions with respect to the signal, such as DC clamping,
filtering, stripping the horizontal sync signals from the composite
signal and the like, and the processed signal is then applied via
line 34 to an analog-to-digital converter 36. The input processing
circuitry 32 will not be described in detail inasmuch as it is
disclosed in the Digital Time Base Corrector, Model No. TBC-800,
manufactured by Ampex Corporation of Redwood City, California, the
assignee of the present invention. The specific electrical
schematic diagrams of the input processing circuitry are shown in
Schematic Nos. 1374104 and 1374156 which respectively appear on
pages 3-5/6 and 3-21/22 of the TBC-800 Digital Time Base Corrector
Catalog No. 7896382-02 issued Oct., 1975. The schematic circuit
diagrams are incorporated by reference herein.
The clamped and horizontal sync stripped analog color television
signal from the input processing circuitry 32 is applied via line
34 to the analog-to-digital converter 36 which is operable to
convert the signal to an eight bit binary coded signal format that
is applied via 8 parallel lines 38 to a digital synchronization
sequence adder 40. The analog-to-digital converter samples the
analog color television signal at a rate that is preferably three
times the frequency of the subcarrier component of the composite
color television signal, although it could sample the signal at a
higher rate of four times subcarrier. With respect to NTSC
television signal format systems, the frequency of the subcarrier
is approximately 3.58 MHz and for PAL and SECAM color television
signal format systems, the subcarrier frequency is approximately
4.45 MHz. Thus, the sampling rate for NTSC systems is preferably
three times the 3.58 MHz subcarrier frequency or approximately 10.7
MHz, while PAL and SECAM systems would utilize a sampling rate of
about 13.3 MHz.
The clock used to control the sampling that is performed by the
analog-to-digital converter 36 is generated by clock generator and
burst store circuitry 42 that is capable of performing phase
shifting of the sampling clock so that the samples are always taken
of the analog color television signal at precise positions relative
to the phase of the color burst component and, more specifically,
on the positive going zero crossing or 0.degree. phase position
with respect to the blanking level, the 120.degree. and 240.degree.
phase positions. In this regard, it should be understood that the
0.degree., 120.degree. and 240.degree. phase positions refer to the
burst of subcarrier cycles occurring during the horizontal blanking
interval and that although the sampling oviously continues during
the video information interval of the color television signal; the
reference to the 0.degree., 120.degree. and 240.degree. positions
is only relevant during the presence of burst. By precisely
controlling the sampling so that they coincide with these phase
positions, several advantages result during subsequent operations
of the apparatus, including the significant advantage of making the
apparatus during reproducing not required to measure subcarrier
phase changes as in FM recording apparatus time base correctors. A
stable reference subcarrier signal (from the broadcasting station
reference, for example) is applied to the clock generator via line
44 and the clock generator and burst store circuitry 42 is
interconnected to the A/D converter 36 via lines 46. As will be
described in detail, the burst store portion of the circuitry 42
interacts with a burst store associated with the A/D converter 36
to phase shift the clock signal as required so that analog color
television signal is always sampled at the same phase positions.
This is accomplished by examining the samples of the burst obtained
from the input video signal every other horizontal line as a result
of sampling by the clock signal derived from previously stored
burst samples until it is determined that the phase of sampling the
incoming burst has changed, whereupon the burst store from which
the sampling clock signal is derived is updated or refreshed to
provide a new "standard" for generating the sampling clock signal.
After a phase adjustment has been performed, the burst store of the
A/D converter 36 is not refreshed until the circuitry 42 detects
that the phase relation of the incoming analog color television
signal has changed sufficiently to require new burst information to
be stored in the burst store of the A/D converter 36 for the
purpose of rephasing the sampling. As will be comprehensively
described herein, the clock generator and A/D converter 36 burst
store is extremely fast acting and can completely rephase the
sampling in less than the time of a single television line, after
the refresh decision has been made. If a "wild switch" occurs in
the input signal whereby it has a radically different phase
relation relative to the signal that was present before the switch,
the decision to rephase the sampling will be made within a few
lines and the A/D converter 36 burst store will be rephased within
the next television line.
The digital samples that are taken by the A/C converter 36 are
applied in the form of an eight bit parallel digital word on eight
lines to the digital synchronization sequence adder 40 which
inserts digital synchronizing and other information in a portion of
the horizontal blanking interval for the purpose of providing the
necessary synchronization information that is used during the
recording and reproducing operations. Although the digital words
are supplied via lines 38 to the sequence adder 40, they may also
be provided on lines 39 which can be supplied by another apparatus
such as would be used in the editing process, for example. It
should be appreciated that there is no precise phase relationship
between the horizontal synchronization pulse and the phase of the
subcarrier of the composite analog color television signal in
commonly used television signal systems. It is for this reason that
the horizontal synchronization pulse has been stripped and will be
subsequently reconstructed at the output. However, when the
horizontal synchronization pulses are removed, there must be some
means of determining the active video information on a line-by-line
basis and the digital synchronization sequence adder circuitry 40
performs this operation by inserting information into the data
stream. With the digital synchronization information added to the
digital samples of the video data interval of the television
signal, it forms a processed color television signal, which is
applied via lines 48 to circuits 50 and 52, each of which has an
8-to-24 bit converter as well as a 2-to-1 switch for applying
either of two inputs to the output thereof. During recording, the
signals on line 48 are applied to the output and during
reproducing, the signals appearing on the reproduce signal paths
146 or 148 are applied to the output. The 8-to-24 bit converter
merely converts three successive eight bit words into one 24 bit
parallel word for processing through random access memories and may
be uncessary if the particular memories used in the apparatus are
sufficiently fast to process information at the eight bit rate. In
this regard, it should be appreciated that converting three 8 bit
words to one 24 bit word enables the data to be clocked at one
third of the clock rate of the 8 bit data. The data from the
circuits 50 and 52 are respectively applied via lines 54 and 56 to
a group of random access memories as shown. The block diagram is
also shown to have the signal flow path from the switches 50 and 52
during reproducing extending to the memories; it should be
appreciated that only one group of lines are used for this
interconnection, i.e., the signal path during recording uses the
same conductors as the signal path during reproducing.
The lines 54 from the circuit 50 extend to random access memories
60 and 62 which are identified as RAM 1 and RAM 3, respectively,
and the lines 56 extend to memories 64 and 66 which are identified
as RAM 2 and RAM 4, respectively. Since the operation of the
memories 60-66 will be described in detail with respect to the
timing diagrams shown in FIGS. 4a, 4b, 5a and 5b in terms of the
writing and reading of data with respect thereto, the use of the
identification "RAM 1" or "RAM 4" will be predominantly used in the
interest of clarity when the timing diagrams are discussed. The
output of memories 60 and 62 are applied via lines 70 to a 24-to-8
bit converter 72, and in a similar manner, the outputs of the
memories 64 and 66 are applied via lines 74 to a 24-to-8 bit
converter 76. It should be appreciated that if the memories are
capable of handling data at the 8 bit word rate, then the 24-to-8
bit converter would obviously be unnecessary. The outputs of the
converters 72 and 76 are applied via respective lines 78 and 80 to
circuits 82 and 84 which add a parity information bit, convert the
parallel 8 bit information to serial data and encode the same using
a pulse code modulation scheme that encodes the data in an
advantageous coded format that can be characterized as a DC free,
self-clocking nonreturn to zero format. The encoded data from
circuit 82 is applied via line 86 to amplifiers 88 and 90 which
have their output lines 92 and 94, respectively, extending to
transducing heads 96 which are designated 1, 3, 5 and 7 for reasons
that will be hereinafter explained. The parity and encoding
circuitry 84 has its output on line 96 similarly extending to
amplifiers 98 and 100, which respectively have output lines 102 and
104 which extend to transducing heads 106 which are designated 2,
4, 6 and 8. As is evident from the drawing, the transducing heads
96 record the encoded data from one signal channel while the
transducing heads 106 record the encoded data from the second
channel.
In this regard, reference is made to FIG. 2 which shows transducing
heads designated 1 through 8 being mounted on a head wheel 108 in a
manner whereby the heads are equally spaced around the
circumference thereof in a common axial plane. The signals that are
applied to the transducing heads are recorded on the magnetic tape
when recording current is applied to them and they are in contact
with the tape. By utilizing 8 heads rather than the customary four
heads for conventional quadruplex recorders, two heads can be
simultaneously recording on two separate tracks. Thus, one set of
four heads will record data from one channel while the other set
records data from the second. Such an arrangement is described in
U.S. Pat. No. 3,497,634 by Damron et al. entitled Wide Band
Instrumentation Rotary Head System Using Redundant Recording and
Reproducing. As the title implies, the 8 heads in the Damron et al.
patent are used for the purpose of redundant recording as opposed
to that which is disclosed herein, namely, recording two channels
of separate information at the same time.
Referring again to FIG. 1, the operation of the block diagram
during reproducing will now be described with respect to the block
diagram, it being understood that the reproducing signal flow path
is shown by the wider cross-hatched lines. The transducing heads 96
and 106 apply signals to preamplifiers 109 which amplify the
recovered signal and forward it to two 2-to-1 switches 110 and 112
which select the appropriate signals from the preamplifiers and
apply them to the respective output lines 114 and 116 which extend
to respective equalizers and drop-out processing circuits 118 and
120. The outputs 124 and 126 of the equalizers extend through
switches 128 and 130 which are adapted to switch the output of
either equalizer 118 or 120 to the input lines 132 and 134 that
extend to decoder, drop-out processing, clock acquisition and
deserializing circuits 138 and 140. Since two channels of
information are being reproduced, and as will be hereinafter
described, each channel simultaneously processes successive lines
of processed television signal information, the reversing of the
two channels of information during playback would have the effect
of reversing the vertical location of adjacent pairs of horizontal
lines and would therefore produce a somewhere garbled video
picture. For this reason, the switches 128 and 130 can apply the
output of either equalizer 118 or 120 to either decoder circuit 138
or 140. The position of switches 128 and 130 is controlled by a
control signal that extends from the reproduce memory control logic
circuitry (FIG. 10) via line 142, which signal is determined by the
line identification signal detected by the decoding circuitry
contained in the circuit 52.
After the respective circuits 138 and 140 have decoded the data,
performed parity checking to determine if errors are present in the
data, acquired clocks from the data itself for use during
reproducing and have converted the serial data to parallel data,
i.e., converted the serial data back to 8 bit parallel digital
words, the data is applied on lines 146 and 148 to the circuits 50
and 52, respectively, for application to the memories 60-66 as
shown. The data is then read out of the memories 60 and 62 on line
150 that extends to a 2-to-1 switch 152 and the data from memories
64 and 66 is also applied to the switch 152 via line 154. The
switch 152 selects the data from either of the lines 150 and 154
and applies it on line 156 to a drop-out compensator 160 which is
adapted to insert information in the data stream to compensate for
missing, errors or other defects that have been detected in the
data during reproducing. In the event the drop-out compensator 160
comprises a two line delay, it inserts a data word that occurred at
the same relative location along the horizontal video line, but
which occurred two lines earlier and therefore four horizontal line
positions earlier in the video raster, which is relatively
representative of the information that has been lost in the data
stream. In this regard, the NTSC 525 line television picture has
approximately 570 eight bit samples in the video data portion of
each line and, since the second previous line has information that
is of the same subcarrier phase and in most instances is relatively
close in content to the actual video information in the line being
replace, insertion of that digital word in the data stream for the
defective information does not introduce noticeable disturbances in
the video information in most instances. However, for more accurate
compensation, the drop-out compensator 160 is constructed to
comprise a 262 line delay (for a NTSC system apparatus) and insert
the data word that occurred in the previous field. This results in
a more accurate compensation for defective data, since the inserted
data is one line position away in the 525 line television raster
from the defective data and, while the inserted information
occurred 1/60 of a second prior to the defective information, it
appears to viewer upon display to be nearly identical.
During operation of the apparatus, if the data from the 2-to-1
switch 152 has not been detected as being either lost, erroneous or
otherwise defective, it is passed via line 156 to a switch 162 that
has a movable contact 164 placed in the lower position 2 and the
data passes through the switch 162 to the digital-to-analog
converter 170 via line 166. In the event the data is determined to
be defective, the switch is controlled to have the movable contact
in position 1 where it receives data from the drop-out compensator
160 via line 168. By switching between positions 1 and 2, either
current data or replacement data from the drop-out compensator 160
is passed to the D/A converter 170.
To control the operation of the switch as well as the drop-out
compensator 160, a control line 174 is provided. The signals on
line 174 effectively place switch 162 in position 2 when the data
has been determined to be lost or in error through the detection of
an RF dropout or a parity error, respectively, as will be described
in detail herein. Line 174 also extends to the drop-out compensator
160 for controlling certain aspects of its operation, particularly
the storing or writing of data therein. Since it is desired to only
substitute reasonably good data from the drop-out compensator, it
should be appreciated that the storing of bad data into the
compensator 160 could result in bad data being applied by the
switch 162 at a later time. For this reason, the signals on line
174 which operate the switch 162 also inhibit the writing of lost
erroneous data in the compensator 160.
The two line delay drop-out compensators will not be shown or
described in detail herein, since it can comprise the two line
delay circuitry that is contained in the Ampex Corporation Digital
Time Base Corrector No. TBC-800, the schematic diagram of which is
shown on Schematic No. 1374060 on page 3-91/92 of the Catalog No.
7896382-02 issued October, 1975. This schematic is for an NTSC
system and a companion schematic for a PAL-SECAM circuit is shown
in the same catalog, Schematic No. 1374064 located on page 3-97/98.
It should be appreciated that the 262 line delay drop-out
compensator 160 is for an NTSC system and that a drop-out
compensator that essentially stores a full field of information for
a PAL or SECAM system would require a 312 line delay and
180.degree. chroma phase inverter.
After the data stream has undergone drop-out compensation, it is
applied via the switch 162 and line 166 to the digital-to-analog
converter 170 which converts the 8 bit digital words to an analog
signal using conventional circuitry such as disclosed in the Ampex
Corporation Digital Time Base Corrector Model No. TBC-800. The
digital data on line 166 can also be applied to a separate 24-to-8
bit converter 173 to provide an 8 bit word on line 175 that can be
interfaced to another apparatus for editing purposes. The schematic
diagram for the digital-to-analog converter is shown in Schematic
No. 1374068 located in page 3-105/106 of the Catalog No. 7896382-02
issued October, 1975 and such schematic is incorporated by
reference herein.
After the data has been converted to an analog signal, it is
applied via line 184 to output processing circuitry 186 which
provides the proper DC level to the analog signal, filters it,
equalizes the amplitude, provides black clipping and inserts
horizontal sync, subcarrier color burst, vertical sync and
equalizer pulses to the signal so that a complete composite analog
color television signal is present at the output on line 188 as is
desired. The specific schematics shown in the output processing
circuitry 186 are not shown herein and may be supplied by
conventional circuitry shown in the video output circuitry for the
Ampex Corporation Digital Time Base Corrector Model No. TBC-800.
The schematics for this circuitry are shown on page 3-115/116,
Schematic No. 1374224 of Ampex Catalog No. 7896382-02 issued
October, 1975. Use of the specifically identified schematic
diagrams referred to herein are incorporated by reference
herein.
The reference video from the station is also applied via line 190
to a sync generator 192 that provides a reference clock signal via
line 194 to a clock generator and switching circuitry 196 that is
used to supply various clocks on lines indicated generally at 198
for use by the circuits throughout the block diagram of FIG. 1.
Also, logic and servo feedback circuitry 20 is operatively
connected to the servo control circuits for driving the tape and
head wheel and receives tape transport servo signals from, for
example, the tachometers operatively associated with the tape drive
capstan and rotating head wheel as will be described further
hereinbelow. Moreover, editor and master record and playback mode
control signals are applied to the circuitry 200 which provides
control signals to the clock generator and switcher 196 for
controlling the operation of the recording and reproducing
apparatus disclosed herein.
While the foregoing description of FIG. 1 has provided a general
description of the operation of the apparatus in terms of the
signal paths during recording and reproducing and of the general
operations that are carried out by the circuitry shown herein, what
has not been described is the relative timing of the reproduce and
recording operations, other than in a very general way in that the
composite color television signal applied at the input 30 during
recording operations and the color television signal provided at
the output on line 188 during reproducing operations are intended
to be real time data, i.e., the signal is continuous and
synchronous with the station reference and has the basic timing in
terms of horizontal and vertical synchronization pulses, subcarrier
frequency and the like. However, the processing of the digital
signal that is recorded on the magnetic tap is done so as to time
expand the data to minimize the effect of tape imperfections on the
recorded signal. Stated in other words, the signal is recorded on
tape at a slower clock rate than the real time clock rate, but is
recorded on two channels rather than a single channel so that no
information is lost.
Referring again to FIG. 1 and viewing the entire apparatus from an
overview perspective, the recording and reproducing can be broadly
described as occurring in four separate steps, i.e., the processed
digital color television signal is (1) written into the memories
RAM 1 through RAM 4 at a real time clock rate, (2) read-out of the
memories at a slower rate but on two separate channels and
recorded, (3) reproduced from the tape on the two channels and
written into the memories at the slower rate, and (4) read-out of
the memories at the faster real time rate and combined into a
single channel so as to reproduce the color television signal at
the real time rate. From the foregoing, it should be appreciated
that the random access memories or any other memory device which
can be written into and read from are used during both the record
and reproducing operations and have data written into them at a
fast rate which is readout at a slower rate during recording and
have the data written into them at the slower rate and read-out at
a faster rate during reproducing.
With respect to the record operation, and referring to FIG. 4a in
conjunction with FIG. 1, keeping in mind that the input data on
line 48 is applied via circuits 50 and 52 to each of the four
memories RAM 1 through RAM 4, the data is selectively written into
the memories and read therefrom on a television line-by-line basis,
with each memory being capable of storing the data for a processed
television line. Thus, the television signal on line 48 can be
considered to be comprised of successive groups of four lines of
data which are selectively written into the memories on a
line-by-line basis. With respect to the order of the writing of the
lines of data, and referring to FIG. 4a, the first line is written
into RAM 1 followed by writing line 2 data into RAM 2, line 3 data
in RAM 3 and finally line 4 data in RAM 4. It should be apparent
that RAMs 1 and 3 are operatively connected together as are RAMs 2
and 4 and that the data is written into the RAMs at a real time
rate. As is also shown in FIG. 4a, the line 1 and line 2 data is
simultaneously read from RAMs 1 and 2 at a slower or time expanded
rate as depicted by the longer lines in the timing diagram of FIG.
4a, with the reading of the information from RAMs 1 and 2 occurring
during the writing of lines 3 and 4 into RAMs 3 and 4. Similarly,
the reading of the line 3 and line 4 data from RAM 3 and RAM 4
occurs while subsequently occurring line 1 and line 2 data is being
written into RAM 1 and RAM 2. Thus, it should be realized that
writing into the memories during the recording operation occurs at
a real time rate and reading of the data out of the memory occurs
at the slower, time expanded rate and that none of the RAMs can
have a reading and writing operation occurring simultaneously.
Moreover, line 1 and line 2 data are applied to the separate
channels and the simultaneous reading on line 3 and line 4 data
from RAM 3 and RAM 4 occurs on the separate channels as well. The
writing of data into the memories is done at a clock rate that is
derived from the video signal itself and the clock that is used to
read the data from the memories at the slower rate is the timing
signal used by the system following the memories to control the
signal processing operations and is generated by circuitry in the
encoder 82.
During reproducing, the relative timing of the read and write
operations of the memories can be easily understood by referring to
FIG. 5a in conjunction with the block diagram of FIG. 1 wherein the
line 1 and line 2 data is simultaneously written into RAM 1 and RAM
2 at the time expanded, slower rate followed by simultaneously
writing line 3 and line 4 data into RAM 3 and RAM 4 at the same
lower rate. While writing is occurring in RAM 3 and RAM 4, the line
1 and line 2 data is sequentially read at the faster real time rate
from respective RAM 1 and RAM 2 and reading of the line 3 and line
4 data occurs sequentially from RAM 3 and RAM 4 at the faster real
time rate during the simultaneous writing of line 1 and line 2 data
into the RAM 1 and RAM 2. Thus, the output of the RAMs provide the
correct sequence of lines of data at the faster real time rate even
though the data is written into the memories at the time expanded,
slower rate and none of the memories simultaneously read or write.
The clock that controls the writing of the data into the memories
is generated by the decoder circuitry and is acquired from the data
itself. The clock for reading the data from the memories is
synchronized to the station reference and is labeled the reference
clock which is, of course, at real time.
With the general understanding of the timing for the writing and
reading operations of the random access memories during recording
and reproducing as has been described, the actual data that is
recorded on and reproduced from the magnetic tape will be described
before the detailed timing diagrams of FIGS. 4b and 5b will be
described. In this regard, reference is made to FIG. 6 which
illustrates the processed television signal data that will be
recorded for each horizontal line of the television picture and is
shown for an NTSC system as opposed to a PAL or SECAM system. Thus,
referring to FIG. 6 (1), there is shown a complete horizontal line
which has 227.5 cycles of color subcarrier (SC), with the first
portion shown to the left comprising the horizontal blanking
interval, followed by the active video portion which has about 190
cycles of subcarrier occurring during this time. As is well known,
the composite analog color television signal has the horizontal
sync pulse at the beginning of each television line followed by a
burst of about eight or eleven cycles of the subcarrier frequency
signal before the active video information occurs. In FIG. 6(1),
the horizontal sync and burst cycles are shown in the dotted
representation in the horizontal blanking interval which is shown
to have a duration equal to 37 cycles of subcarrier.
As previously mentioned, the horizontal sync signal and the burst
of subcarrier are removed from the composite color television
signal by the digital synchronizing adder circuitry 40, and the
apparatus described herein is adapted to insert the digital
synchronizing information within this time period. The requisite
information is written within the horizontal blanking interval in a
time that is significantly less than the duration of the complete
horizontal blanking interval, with the writing of the data delayed
at the beginning of each horizontal line interval for a period
equal to about 25 cycles of subcarrier to be placed in the last 12
cycles subcarrier interval of the horizontal blanking interval. It
should be appreciated that the delay is shown in the drawing to be
equal to 25 cycles of the color subcarrier. However, the signal
that controls the writing of the data into memory is actually
delayed 25.5 cycles and the write signal is synchronized to write
12 cycles of synchronizing sequence followed by 190 cycles of
active video information for each line and this total of 202 cycles
forms the processed television signal line interval that is always
written into memory. The remaining 25.8 cycles are disregarded. It
should be appreciated that the digital synchronization sequence may
be determined to be somewhat greater or smaller than 12 cycles of
subcarrier and also that the number of subcarrier cycles of the
active video interval of each television line may be somewhat
greater than 190. However, the total of the active video interval,
synchronization sequence and the delay must equal 227.5 for each
horizontal television line. The synchronizing information inserted
into the television line provides significantly more information
than was provided by the horizontal sync and color burst, as will
become apparent. Thus, as is shown in FIG. 6(1), writing data into
the random access memories is delayed for a period during the
beginning of each horizontal line corresponding to approximately 25
cycles of subcarrier, and during the final 12 cycles subcarrier
period of the horizontal blanking interval, the digital
synchronizing sequence is added to the data stream, this being
accomplished by the digital synchronization sequence adder
circuitry 40. The digital synchronization sequence together with
the video information interval of the television line is then
written into memory as processed television line information, with
the video information interval extending for a time period equal to
190 cycles of subcarrier.
Since the input analog color television signal was preferably
sampled at a rate of three times the subcarrier frequency, 570
eight bit digital samples are present for the video interval
portion of each television line. This data, in addition to the
added synchronization data sequence, appears on line 48 for writing
into one of the memories RAM 1 through RAM 4.
It should also be appreciated that the 25 cycle subcarrier delay in
writing the processed television signal information into memory
provides a time interval during every line interval where data is
not written in memory, which means that this time interval can be
subsequently used to perform head switching and time base
correction. Stated in other words, since the delay occurs before
writing of the information is begun during recording, and also
during reproducing when the processed television signal data is
again written into the memories, there will necessarily be a
commensurate delay that can be used to advantage before reading the
date from memories to reconstruct the line-by-line sequence of the
television signal.
The digital synchronizing information that is inserted within the
latter portion of the horizontal blanking interval contains
clocking information, frame and field identification information as
well as information that identifies whether the line is an odd or
even line.
The servo systems which control the rotation of the head wheel 108
carrying the transducing heads and the transport of the magnetic
tape are generally conventional and are described herein below with
respect to the block diagram of FIG. 28. During recording, the head
wheel and transport servo systems use a horizontal line interval
related signal, which in the apparatus described herein is a H/64
signal derived from the input television signal by the input
processor 32, and this signal is used to control the rotation of
the head wheel 108 whereby the head wheel rotation and capstan or
tape transport are locked together. During reproducing, the
identification signal is used to provide horizontal line
synchronizing information and a vertical synchronizing related
signal is used to provide information of deriving a vertical
synchronizing signal and for color framing. In apparatus designed
for the NTSC color television format, the information added by the
sequence adder circuitry 40 contains the actual line interval
number for each line interval in the four field sequence, i.e., the
line intervals are numbered from 1 to 1050.
During the vertical interval following each fourth field of the
four field sequency of an NTSC color television signal, the
circuitry 40 inserts a series of unique digital words into the
active video interval of line interval 1050. It is this series of
words that are used by the servo systems to derive vertical sync to
perform proper color framing.
Referring to FIG. 6(2), which is an expanded representation of the
horizontal blanking interval, the write delay of 25 cycles of
subcarrier is shown at the left, followed by an interval of 12
cycles of subcarrier during which the digital synchronization
sequence is added. There are nine cycles of clock sequency
preceding an identification No. 1 or "ID 1" clock cycle, which is
followed by a framing information cycle and subsequently by an
identification No. 2 or "ID 2" cycle. The ID 1 and ID 2 information
results in several advantages during subsequent operations of the
apparatus, including the significant advantage of making the
apparatus greatly immune to cycle hops that are prevalent in FM
recording apparatus. This advantage is due to the synchronization
of the horizontal line to subcarrier phase being determined prior
to recording, which is contained in the nine cycle clock sequences
and ID 1 and ID 2 information. Each of the nine cycles of clock
sequence comprise what is shown in the left portion of the expanded
line FIG. 6(3) and specifically comprises the binary coded numbers
0, 0 and 5. The binary representation of a clock sequence cycle is
also shown in the left portion of FIG. 6(4) and comprises two
series of eight bits of low level for the zeros with the binary
coded, number 5 having the 2.sup.0 bit and 2.sup.2 bit high and the
2' bit low, which is the binary number for the decimal number 5. As
will be shown herein, a parity bit is also added to the data,
which, when the sequence is serialized, causes the sequence to
appear as 24 successive zeros, followed by the sequence "101". This
is used in decoding upon reproduction to identify the word sync as
will be described herein. The cycle that is marked ID 1 includes
three samples of a particular number, such as the digital
representation for two in the event that the video line is an odd
numbered line and the digital representation for twenty in the
event that it is an even line. Similarly, the cycle marked ID 2 may
contain the digital representation for ten, for example, for an odd
line and the digital representation for forty for an even line.
Thus, four separate numbers are provided in the ID 1 and ID 2
cycles with the numbers effectively identifying whether a line is
even or odd.
In the eleventh cycle located between the ID 1 and ID 2, framing
information can be provided so that the apparatus can instantly
have the information that will indicate the field and frame in
which the line is located. In this regard, the NTSC system contains
a four field sequence and the information contained in the framing
cell can identify whether it is the first or second field of either
the first or second frame of the full four field sequence.
Moreover, since a four field sequence would necessarily include
1,050 television lines of information, the particular line of the
four fields of lines may be provided, i.e., the number 256 may be
provided which would indicate that the first line of the first
field of the second frame is identified. The line number as well as
other information is added as shown in the right portion of FIG.
6(3) and comprises three words labeled A, B and C. The number 1050
requires 11binary bits and, for a PAL system having a total of 2500
lines in a color frame sequence, a total of 12 bits is required.
These bits are separated so that the first 6 most significant bits
are contained in word A, followed by the 6 least significant bits
in word B. Word C can contain 3 bits of data which identifies such
information as a NTSC, PAL, SECAM system, whether it is color or a
monochrome system, for example. Three other bits can be used to
identify the field number in the full sequence. While the exact
line number would also provide the field number, a less
sophisticated apparatus or a portable apparatus may utilize only
the field number rather than the actual line number. The last bit
in each of words A, B and C are high, so that a consecutive zero
counter will not be able to detect an incorrect word
synchronization as will be described in detail herein. By providing
this information, the exact color framing and line identification
is available on a line-by-line basis, which information can be
advantageously used in an editing operation. Thus, in the time
period of 12 cycles of color subcarrier, considerably more
information is provided in the recorded television signal than is
present in the entire horizontal interval of the analog color
television signal.
As has been previously mentioned, the data in the memories is
read-out for recording on the two channels comprised of lines 70
and 74 at a slower rate than the rate in which the data is written
into the memories. Since the sampline rate of the A/D converter 36
is a multiple of the subcarrier frequency, preferably 3 SC
(approximately 10.7 MHz), the data on lines 48 is at a 10.7 MHz
rate. However, by virtue of being converted from 8 bits of parallel
data to 24 bits of parallel data, the effective rate in which the
data is written into the memory during recording is at the
subcarrier frequency of approximately 3.58 MHz. The slower rate in
which the data is read from the memories onto lines 70 and 74 is
approximately 1.6 MHz. However, the precise frequency in which this
is done will now be discussed in conjunction with FIG. 6(1), which
shows that the active video interval of the horizontal line
together with the 12 subcarrier cycles of digital synchronizing
sequence synchronizing sequence information. The data associated
with each SC cycle of the 12 subcarrier cycles of the digital
synchronizing sequence and the following video data interval are
read from the memories as 24 bits of parallel data using 202 cycles
of the 1.6 MHz clock, whereby the single line of processed
television information is read from the memories and recorded over
a time corresponding to two horizontal line intervals. With this
frequency being chosen, the frequency at which data in each channel
must be recorded is as follows:
The 9 bits per sample reflect the addition of a parity bit to the 8
bit data word. Since the 9 bit data word, before being serialized
by the serializing and encoding circuitry 82 and 84 is in parallel,
the frequency of the data will be the about 42.90733711 MHz divided
by 9 or 4.767481901 MHz. However, the recorded data read from the
memories during reproducing is at a rate corresponding to 27 bits
of parallel data (providing for the addition of 3 parity bits to
the 24 bit word read from the memories) rather than 9 bits and,
accordingly, the frequency in which the data is read from memories
will be 4.767481901 MHz divided by 3 or 1.589160634 MHz which will
hereinafter be referred to as 1.6 MHz. The foregoing calculations
of the frequencies are for a NTSC system rather than a PAL or SECAM
system which would necessarily involve different frequencies which
can be similarly calculated but which will not be presented herein.
It should also be apparent that if the data is read from the
memories for recording using the 1.6 MHz clock, the same clock
frequency will be used during reproducing to write the data into
the memories and the subcarrier frequency of 3.58 MHz will
similarly be used to read the data therefrom for application to the
switch 152.
With the above description of the clock frequencies that are used
during writing and reading from the memory, together with the
operational sequence of writing and reading data into and out of
the memories during the record and reproducing operations for the
apparatus described herein with respect to FIGS. 4a and 5a and the
digital information and the timing relation of the digital
information relative to the processed television signal described
with respect to FIG. 6 in mind, the specific operation of the
random access memories will now be described in more detail in
conjunction with FIGS. 4b and 5b.
Turning initially to the record process and referring to FIG.
4b(3), there is illustrated a series of four consecutive television
lines with the horizontal blanking interval being shown as a low
level and the active video information interval being shown as a
high level. Lines 4b(1) and 4b(2) respectively illustrate the
horizontal sync rate divided by 4 and by 2, (H/4 and H/2). As
previously mentioned with respect to the description of FIG. 6, the
initial portion of the horizontal blanking interval is effectively
discarded by delaying the writing of the digital information into
the memories, with the delay being equal to approximately 25 cycles
of subcarrier. FIG. 4b(4) illustrates the reset pulses that occur
for the purpose of causing a counter to be reset which controls the
writing of data into the memories. FIGS. 4b(5), 4b(10), 4b(7) and
4b(12) respectively show the timing for writing data into RAMs 1-4
in the time sequence that has been described with respect to FIG.
4a. Thus, the write enable control signals to the respective
memories enable writing to occur when they are low and reading to
occur when they are high. Similarly, the memory select lines
control whether the outputs of the four memories RAM 1-RAM 4 can be
applied to the output lines, it being realized that the memories
are connected in pairs. Effectively, the data from a memory is
gated to the output line when its corresponding memory select line
is high. FIGS. 4b(6), 4b(11), 4b(8) and 4b(113) respectively
illustrate the timing for the memory select lines for memories RAM
1-RAM 4.
For reading the data from the memories, FIG. 4b(9) shows reset
pulses occurring for each two lines with the left reset pulse
resetting RAM 3 and RAM 4 and the subsequently occurring reset
pulse resetting RAM 1 and RAM 2 so that the data for each line can
be read-out at the 1.6 MHz clock rate. In this regard, it should be
recalled that RAM 1 and RAM 2 are simultaneously read onto two
separate channels as are RAM 3 and RAM 4. The reset pulses for
reading the memories is delayed to occur during the discarded
horizontal blanking interval for the purpose of insuring that all
data is written into the respective memories during the write
operation. The dotted lines shown in FIGS. 4b(6), 4b(8), 4b(11) and
4b(13) are intended to illustrate the timing sequence during
operation of the apparatus in the EE mode which is a test mode
where the data is processed through the memories from the input 30
to the output 188 without recording or reproducing the data. The
input television signal is processed through memory directly to the
output using a real time 3.58 MHz clock and the time required to
read the data from memory corresponds to the time required to write
the data therein.
With respect to the operation of the random access memories RAM 1
through RAM 4 during reproducing operations, as had been broadly
described with respect to FIG. 5a, the more detailed operation is
shown in the timing diagrams of FIG. 5b, which includes the
equivalent of four successive video lines in FIG. 5b(3), a H/4
signal on line 5b(1), as well as a tachometer reset pulse on line
5b(2), which occurs for each revolution of the head wheel 108
carrying the eight heads. Since each transducing head writes a
total of eight lines of processed television signal information per
pass on the video tape, and there are eight heads on the head wheel
as shown in FIG. 2, the tachometer reset pulse will occur every 64
lines. A read reset pulse occurs in the latter part of the
horizontal interval, as shown by comparing line FIG. 5b(4) with
FIG. 5b(3), with the read reset pulse being timed to correspond
with the delay that occurs in writing the information from the
memories during recording operations, the reset pulse appearing so
as to read only the ID 1, ID 2, and framing information that is
present in the digital synchronizing sequence that was added during
the latter portion of the horizontal interval and the following
video data interval. As has been described with respect to the
block diagram of FIG. 1, the output from RAM 1 and RAM 3 appears on
line 150 while the output of RAM 2 and RAM 4 appears on line 154,
with both lines being connected to the 2-to-1 switch 152 which
alternately switches the data from the two lines onto line 156
which is connected to the drop-out compensator 160 or 162,
depending upon the one in use. The signal for switching the 2-to-1
switch 152 comes from the clock generator and switcher circuitry
196 and the timing diagram for the control to the 2-to-1 switch
appears on FIG. 5b(5), which switches at the beginning of the read
reset pulse so as to receive a full line of processed television
signal data from either line 150 or 154 and alternates between the
two. FIGS. 5b(8), 5b(9), 5b(14) and 5b(15) illustrate pulses which
are used by the memory control logic 200 to reset the memories for
writing the data into them. As is shown in the middle portion of
FIGS. 5b(14) and 5b(15), the first reset occurs after nine cycles
of the 1.6 MHz clock and the second pulse appears after 11 cycles
of the clock. These pulses are used by the reproduce memory control
logic and timing circuitry contained in the logic and servo
feedback circuit 200 and the clock generator and switcher circuit
196 to inhibit the memories from writing the nine cycles of clock
sequence includes in digital synchronizing information that is
inserted in the processed television signal during the record
operation, as discussed with respect to FIG. 6(2). The nine cycles
of clock sequence are added to the digital synchronization sequence
to enable detection of the "101" word sync and recovery of the
correctly phased clock from the data during reproducing operations,
which occurs in the decoder circuitry 138 and 140 located before
the inputs of the memories 60-66. Since that occurs before the
memories, it is unnecessary to write the clock sequence into the
memory during reproducing operations and it is therefore not done.
However, the timing of the memory control write pulses effectively
write the ID 1, framing information and ID 2 data into memory at
predetermined memory address locations. Then, using read reset
pulses that are timed relative to a station reference, the memories
are read from predetermined address locations and the recovered
data is correctly timed.
FIGS. 5b(6), 5b(12), 5b(10) and 5b(16) are the timing diagrams for
selecting RAMs 1 through 4, respectively, while FIGS. 5b(7),
5b(13), 5b(11) and 5b(17) illustrate the write enable signals which
permit reading and writing operations to be performed with respect
to the memories RAMs 1 through 4, respectively. The duration of the
read and write operations shown in 5b are similar to, but are time
reversed relative to the corresponding diagrams previously
described with respect to FIG. 4b, it being understood that during
reproducing, writing of the data occurs at the slower 1.6 MHz rate
while reading thereof is at the faster 3.58 MHz rate in contrast to
the writing at 3.58 MHz and reading at the 1.6 MHz rate during
recording.
In accordance with an important aspect of the apparatus described
herein, and referring briefly to the block diagram of FIG. 1, the
sampling of the analog color television signal by the A/D converter
36 is done at a rate of three samples per subcarrier cycle, which
for the NTSC system is at a rate of about 10.7 MHz and is
controlled by a clock signal received over line 46. Referring to
FIG. 22, which illustrates a single cycle of subcarrier, the
television signal is sampled at phase locations relative to the
zero phase crossing point, the 120.degree. phase point and the
240.degree. phase point of the color burst time and the timing of
the sampling is controlled so as to obtain samples throughout the
television signal from locations that are precisely defined
relative to the phase of the color burst contained in the signal
that is to be recorded. By so doing, the subsequent recording and
reporducing can be performed in a manner whereby phase shifting of
the subcarrier will not complicate the operation of the apparatus
for reliable recovery of the color television signal information.
In this regard, and as previously mentioned, the phase of the color
subcarrier is not synchronized with respect to horizontal sync
pulse in a NTSC composite video signal. The clock generator and
burst store circuitry 42 interacts with the analog-to-digital
converter 36 so as to provide accurate sampling that is synchronous
with respect to subcarrier in the manner whereby samples are taken
precisely at the zero phase crossing point, the 120.degree. phase
and 240.degree. phase points relative to the color burst. The clock
signal that controls the time of sampling of the analog color
television signal is phase adjusted so that the sampling always
occurs at the aforesaid phase points. As will be described herein,
in the event that a "wild switch" occurs wherein the input line 30
is switched from one source of color television signals to another
unsynchronized source which provides a signal with a radically
different subcarrier phase, the circuitry 42 can very rapidly
rephase the sampling so that samples are accurately taken at the 0,
120.degree. and 240.degree. phase points as is desired.
To provide the phase adjustment of the sampling clock so as to
maintain the desired timing of the sampling relative to the color
burst, reference is made to the block diagram shown in FIG. 11
which broadly illustrates the operation of the clock generator and
burst store circuitry 42 in conjunction with the analog-to-digital
converter 36. After the A/D converter 36 has sampled the television
signal information and the obtained samples encoded into 8 bit
digital words, the digital samples are applied to line 220 which is
applied to a burst data gate 222 that is controlled by a gate
control line 224 so that the samples of the color burst cycles are
gated through to line 226 for application to either a first burst
store 228 or a second burst store 230. The first burst store 228 is
adapted to receive and store the samples representative of five
cycles of burst and utilizes this data for generating a 3.58 MHz
clock that is phase synchronized to color burst, hence, also phased
for the input signal to be processed for recording. The burst data
is clocked into the first burst store 228 using a reference clock
signal applied over line 44 from station reference or the like, the
only requirements for this clock being that it be a phase stable
clock signal and essentially frequency stable relative to the color
subcarrier of the input television signal. The output of burst
store 228 appears on line 234 which is applied to a phase shifter
236 that controls the phase shifting of the generated clock
signals, which for the apparatus described herein are at a rate of
3.58 MHz and 10.7 MHz. These clock signals appear on lines 238 and
239, respectively, and are used to control the sampling of the
input signal and clocking of the resulting data into the random
access memories RAM 1 through RAM 4 during the record process.
The second burst store 230 is also adapted to receive and store the
samples representative of a few cycles of the burst of the signal
using the derived clock on line 238 to effect the generation and
storage of the burst samples. The signal from the second burst
store 230 is applied via line 240 to a zero crossing detector and
error corrector 242, which examines the samples of the burst and
determines whether the zero phase samples is actually occurring on
the zero crossing point of the color burst and whether the other
samples taken during the color burst cycle are similarly correctly
taken. If there is an error in the location of the sampling points,
it appears as a signal on line 244 which is applied to the phase
shifter 236 as well as to a limit detector 246. The limit detector
246 determines the amount of error that is present in the actual
sampling points compared to the desired sampling points and, if the
error is outside of a predetermined limit, issues a command on line
248 to cause the first burst store 228 to refresh itself, i.e., to
store a new set of samples from the incoming burst on line 226. The
new set of burst cycle samples are obtained from the A/D converter
36 by sampling the incoming color burst at times determined by the
reference clock. At times other than the refreshing of the first
burst store 238, the A/D converter 36 is clocked by the 10.7 MHz
derived clock signal on line 239. The output of the error corrector
242 also provides a signal to the phase shifter 236 for rephasing
the clock signals on line 234 so that the derived record clock
signals on lines 238 and 239 are correctly phased and thereby
corrects for slow or minor drifts of the sampling phase points that
can occur.
It should be appreciated that the circuitry shown in the block
diagram of FIG. 11 is particularly adapted for use with a color
television information signal having color burst cycles which
function as a time-base synchronizing component of the information
signal. However, the circuitry of FIG. 11 can be used to provide a
phase adjustable clock signal for sampling other types of
information signals, provided they have periodically occurring
intervals of a time-base synchronizing component. It should also be
appreciated that if the phase adjusting circuitry was used in
apparatus where the slow or minor drifts in phase were not
particularly critical, the aspect of its operation where the
shifting is performed by the phase shifter 236 may not be required
and in such event, only a refreshing of the first burst store need
be done when the phase error exceeds a predetermined limit. On the
other hand, if the phase adjusting circuitry is used in apparatus
that seldom experiences fast or large phase changes, the phase
shifter 236 may desirably be employed to make the corrections of
the slow or minor drifts, and the circuitry would not include the
limit detector 246 to refresh the burst store 228.
The error correcting signals on line 244 are intended and are
coupled to control the phase shifter 236 to correct for slow
moderate errors in the sampling of the signal relative to the
precise desired sampling points and the phase shifter 236 is not
operable to make corrections for large fast errors that are outside
of the predetermined limit that is detected by the limit detector
246. Large changes in the phase of the color burst, for example, as
a consequence of a wild switch, are corrected by the operation of
the limit detector 246, which issues a command on line 248 for
causing the first burst store 228 to receive a new series of
reference samples for generating the record clock signals that
appear on lines 234 and 239.
An important aspect of the phase shifting circuitry shown in FIG.
11 is the interaction of the two burst stores 228 and 230 and the
ability of the circuitry to rapidly correct for errors that may be
present. In this regard, the operation of the first burst store 228
is such that it receives five cycles of burst and stores this
information, indefinitely, using the stable reference clock on line
44 to write the burst samples into the memory of the burst store.
The 3.58 MHz clock signal that is generated from the burst samples
stored in burst store 228 is employed by the A/D converter 36 to
sample the input television signal and the first burst store 228 is
not refreshed every line or even every second line, but is kept
indefinitely until the phase of the burst on line 226 is determined
to be outside of the predetermined limits. The operation of the
circuitry is such that the burst cycles will not be simultaneously
written into both burst stores 228 and 230. If the first burst
store 228 is given a command to store the samples of the burst,
burst store 230 will be inhibited from storing the samples unitl
the next successive horizontal line of burst occurs. The reference
clock is used to sample the burst in the A/D converter 36 and store
the burst samples in the first burst store 228 and the derived 10.7
MHz output clock on line 239 is used to sample the burst in the A/D
converter 36 and store the burst samples in the second burst store
230. If the phase of the incoming burst changes from line-to-line
by an amount that is outside of the predetermined limits, the
sequence would be to sample the burst of a television line and
refresh the first burst store 228, using the reference 10.7 MHz
clock, use the derived 10.7 MHz clock on line 239 to sample the
burst of the next or second television line and store the burst
samples in the second burst store 230. If the phase of the burst on
the second line was outside of the predetermined error limit from
the burst of the first line, a new command would cause the first
burst store 228 to refresh itself again on the third television
line, creating a different phase clock on line 239, which is used
to sample the burst of the fourth television line and store the
samples in the second burst store 230. Once the phase of the
incoming burst on line 226 settles down and is relatively constant,
so as to not be outside of the predetermined phase error limits,
the first burst store 228 would not be refreshed and minor phase
corrections would be accomplished by the error corrector circuitry
242 applying error correcting signals over line 244 to the phase
shifter 236.
The detailed circuitry that can be used to carry out the operation
of the block diagram shown in FIG. 11 is illustrated in FIGS. 19a
and 19b which together comprise the schematic electrical diagrams
for this circuitry. However, it should be appreciated that the
burst data gate as well as the clock generator of the first burst
store 228 shown in FIG. 10 is not shown in detail herein, inasmuch
as it is identical to circuitry shown in electrical schematics for
the TBC-800 Digital Time Base Corrector of Ampex Corporation. The
clock generator is shown on Schematic No. 1374028 sheets 1 and 2
contained in Catalog No. 7896382-02 issued October, 1975 for the
TBC-800 and is incorporated by reference herein. The phase shifter
236 is merely added after the 3.58 MHz filter and before the tape
3.58 limiter shown on sheet 2 of Schematic No. 1374028 and the
horizontal line between the inductor L30 and the resistor R101.
Since the remainder of the circuitry of that schematic produces
3.58 and 10.7 MHz square waves, the phase shifting that is
performed by the phase shifter 236 simultaneously adjusts the phase
of both of these signals which are used for clocking the A/D
converter 36 and for the record clocks elsewhere in the circuitry.
Moreover, the first burst store 228 is not incorporated herein as
much as it is essentially identical to the burst store of the
TBC-800 by Ampex Corporation and is shown on Schematic No. 1374044
sheets 1 and 2 of the Catalog No. 7896382-02 issued October, 1975
wherein sheet 2 of the schematic shows the 8 bit word input being
applied to random access memories A36 and A37 which are adapted to
store 15 samples comprising five cycles of burst which are used by
its clock generator to generate a 3.58 MHz clock that is
synchronous with the samples stored therein. A burst store control
signal is applied on input terminals 81 and 82 that pass through a
resistor and inverter with the output of the inverter A41 pin 12
supplying a burst store command at an H/2 rate, hence, for every
second burst, which is applied to the input line 254 shown in FIG.
19a. This burst store command is derived from that used in the
first burst store 228 by dividing such command used by the first
burst store by two. The burst store command causes the second burst
store 230 to load samples of burst using the derived 10.7 MHz
record clock received over line 239 from the first burst store 228,
as will be described in detail hereinafter. As has been described
with respect to the block diagram of FIG. 11, in the event that the
first burst store 228 is to be refreshed, then a resemble inhibit
control signal on line 248 is removed to allow the burst store 228
to receive a write enable signal and, thereby, be loaded. This
inhibit control signal is applied to the clear input of a flip-flop
labeled A45 on the lower portion of sheet 1 of Schematic No.
1374044 to permit the burst store comprised of the random access
memories A36 and A37 to load 15 new samples comprising five cycles
of the burst.
Returning to FIG. 19a, the derived record 3.58 and 10.7 MHz clocks
received from the phase shifting circuitry via lines 238 and 239,
respectively, whereby three samples of a single cycle of burst from
the A/D converter 36, in the form of eight bits of data appearing
on lines 226, are stored in random access memories 230 forming the
second burst store. The flip-flops indicated generally at 256
reclock the burst store command signal on line 254 with the derived
record 3.58 MHz clock signal to identify the zero crossing sample
and provide delays so that the three samples of burst cycle that
are written into memory are taken from the center of the burst
sample interval rather than the start or end of it. During the
writing of the three burst samples into the memories 230, the
address generator controller 258 is clocked by the retimed MHz
clock received over line line 239 to issue write address signals
over output lines 260, which are connected to the address line
inputs of the memories 230. In addition, the flip-flops 256 apply a
gating signal to the NAND gate 237 lasting for an interval of three
10 MHz clock cycles to cause it to issue a write enabling command
of comparable interval to the memories 230. The memories 230 are
responsive to these signals to store three successive burst samples
at the 10.7 MHz rate. After the three samples of the single burst
cycle have been written into the memories, the address generator
controller 258 disables the NAND gate 237 after the last of the
three write addresses have been provided, thereby, preventing the
further storing of samples present on lines 226.
The stored samples are then read from the memory at a substantially
slower rate via output lines 264 into a digital-to-analog converter
266. The converter responsively provides an analog value on line
268 that is applied to a multiplexing switch 270 (FIG. 19b), which
applies the three successively occurring analog values from line
268 successively on to lines 272, 274 and 276 according to the
address signals placed on address lines 278 by a memory read
address generator 20 (FIG. 19a). The memory read address generator
280 together with a number of monostable multivibrators or
one-shots, forming a gated clock signal generator indicated at 282,
provide timing and read address signals so that each of the three
successive stored samples are read from the memories 230 onto lines
264 and the resulting analog values provided by the converter 266
are applied successively to the respective output lines 272, 274
and 276 (FIG. 19b) of the multiplexing switch 270. The application
of the analog values on line 268 occurs for a time equal to about 2
microseconds with the successive analog voltage values represented
by the three successive samples charging respective capacitors 284,
286 and 288, which define sample and hold circuits for the analog
values of the three samples. The reading of the stored three
samples of the single color burst cycle is initiated by the gate
signal provided by the flip-flops 256. The gate signal activates a
one-shot 241 to cause the shift register forming the address
generator 280 to activate the lines 278 and 279 to apply read
address signals to the memories 230 and the multiplexing switch
270, respectively. The address generator 280 is cleared in response
to the gate signal to remove the inhibit applied to the line 285
that extends to the string of one shots indicated generally at 282
and, thereby, enable the one-shots to generate clock signals that
are applied to the clock input, C1, of the address generator 280.
The address generator 280 activates the lines 278 and 279 by
shifting a high logic state signal (resulting from its being
cleared) successively onto its outputs QA-QD in response to the
clock signals provided by the string of one shots. The generator
280 cooperates with the time delay circuit indicated generally at
281 and the address generator 258 to provide the proper sequence of
read address signals to the memories 230. The gate signal provided
by the one-shot 256 is also coupled to the load input of the
address generator 258 and places the generator in a condition
whereby it is unresponsive to the 10.7 MHz clock signal and any
signals on its inputs A-C are coupled directly to its outputs that
are connected to the address lines 260. The address lines 278
extending to the multiplexing switch 270 are activated by the
address generator for directing the successively received analog
values of the samples to the proper output line 272-276. The
multiplexing switch 270 is enabled to transfer the analog values by
the coupling of a sampling control signal via line 283 to the
inhibit input of the switch 270. The sampling signal is generated
by the one-shots 282 to occur a selected interval after each
activation of one of the outputs QA-QD of the shift register 280 so
that the A/D converter 266 has adequate time to convert each
digital sample to an analog value for application to the
multiplexing switch 270 before the switch is addressed. The clock
generator and burst store circuitry 42 has one horizontal line
interval to detect and correct any changes that may occur in the
locations of the sampling points of the burst. Therefore, the one
shots 282 is arranged to provide the clock pulses to the address
generator 280 and the sampling control signal to the multiplexing
switch 270 during such one television line interval so that the
rephasing of the clock signals employed to effect the sampling of
the following television line interval is accomplished before its
arrival at the input of the A/D converter 36. Termination of the
reading of the samples from the memories 230 is accomplished by
deactivating the one-shot clock generator 282 by activating the QD
output of the shift register generator 280 after the sequence of
read address have been provided.
The value of the most positive sample appears at output line 290 of
operational amplifier 292, the value of the most negative sample
appears on output line 294 of operational amplifier 296 and the
analog value of the zero crossing sample appears on line 298 which
is the output of operational amplifier 300. The most positive and
most negative values on lines 290 and 294 are arithmatically
subtracted with one another by being connected together through
resistors 302 and 304 with the difference appearing on line 306
that provides one input to a comparator 308, the other input of
which is supplied by line 298.
The manner in which the zero crossing detector 242 determines
whether samples are being taken at the precise zero phase crossing
point, the 120.degree. and 240.degree. phase points can be easily
understood by referring to FIG. 22 which shows sampling points at
the 0, 120.degree. and 240.degree. phase points with respect to the
single cycle of color burst depicted by the solid line. By applying
the analog value of the three samples to the operational amplifiers
292, 296 and 300, the value of the most positive sample, i.e., the
120.degree. phase sample will appear on line 290 and the negative
sample will appear on line 294 which, when they are arithmatically
subtracted from one another, will equal zero since the magnitude L1
will equal the magnitude L2. Thus, the value on line 306 will be
zero when the samples are taken at the precise 120.degree. and
240.degree. phase locations. Similarly, the zero crossing value
will appear on line 294 and the comparator 308 will compare zero
with zero and produce no DC error correcting voltage.
However, in the event the sampling is not being performed on the
precise desired locations as depicted, for example, by the dotted
representation of a cycle of color burst in FIG. 22, then the
difference between L3 and L4 will result in a voltage on line 306
applied to the comparator 308 and the zero crossing sample will
also have a value that is negative as opposed to zero, which will
be applied to the other input of comparator 308 and a resulting DC
error correcting voltage will be produced on line 310. Thus, by
using one or more combinations of three successive samples, an
error correcting voltage can be generated that will be used to
rephase the 3.58 MHz clock that is used for performing the actual
sampling by the A/D converter 36 and to control other circuit
components during the recording process. The error voltage produced
by the comparator 308 on output line 310 is then applied to a
buffer operational amplifier 312 and provides an error correcting
signal on line 244 which is connected to a monostable multivibrator
or one-shot 316.
As shown in FIG. 19b, the line 234 originates in the clock
generator portion of the Time Base Corrector Model No. TBC-800 as
previously mentioned and the signal on line 234 is an analog
voltage at a frequency of 3.58 MHz. It is applied to a comparator
318 which produces a square wave that is applied to a one-shot 320
that positions the square wave signal and applies it to the
one-shot 316. The error voltage on line 244 modulates the length of
the output of the multivibrator 316 on line 324 and thereby phase
adjusts the 3.58 MHz signal. This phase adjusted 3.58 MHz signal is
applied to another monostable multivibrator 326 which produces a
square wave. Subsequent circuit components indicated generally at
327 effectively convert the square wave to a sine wave on line 328
which is again converted to a square wave by other circuitry in the
clock generator of the TBC-800 and which appears on line 238. It
should be appreciated that conversion from a square wave to sine
wave and the converse is easily accomplished and the reason that
the output signal from the multivibrator 326 is converted to a sine
wave is that the clock generator uses the sine wave to produce a
synchronized 10.7 MHz signal in the reference clock generator of
the TBC-800 and the phase shifting that is performed by the
circuitry 236 will therefore simultaneously phase shift the 3.58 as
well as the 10.7 MHz signals.
The error voltage from the amplifier 308 appearing on line 310 is
also extended downwardly to the limit detector 246 which monitors
the voltage levels and provides a signal on line 330 that is
applied to a flip-flop 332 having an output line 248 which extends
to the circuitry of the TBC-800 which controls the operation of the
first burst store 228. When the line 248 is low, it inhibits the
application of the write enable signal to the memory of the burst
store, thereby inhibiting the refreshing of the first burst store
228. This occurs when the voltage on line 310 is within a
predetermined limit. A new series of samples are loaded into the
burst store 228 when line 248 is high as a result of the voltage on
line 310 being outside the predetermined limit.
As described hereinabove, the second burst store 230 is controlled
to receive samples of the color burst associated with every second
horizontal line interval of the input color television signal. This
simplifies the circuitry required to construct the second burst
store. However, the second burst store 230 could be arranged to
receive and process the samples of color burst associated with each
horizontal line interval of the color television signal for the
purpose of correcting the phase of the clock signals provided on
lines 238 and 239 for effecting the sampling of the color
television signal.
With respect to the digital synchronization sequence that is
combined with the video data interval by the adder circuitry 40 to
form the processed television signal, as has been broadly described
in conjunction with the block diagram of FIG. 1, and referring to
the timing diagrams of FIG. 6, the circuitry that inserts the
digital synchronization sequence will now be described in
conjunction with a block diagram shown in FIG. 12.
The video digital data from the A/D converter 36 appears in the
form of eight lines of parallel digital information on lines 38
which are applied to one set of inputs of a 2-to-1 switch 340,
which has another set of inputs 342 upon which the digital
synchronization sequence is applied. The switch 340 selects either
the set of input lines 38 or 342 and passes the data from one set
or the other to lines 48 which extend to the circuits 50 and 52.
The switch 340 is controlled by a signal on line 344 which is
controlled by a clock sequence generator 346. The digital
synchronization sequence adder circuitry 40 has a composite sync
signal applied on line 348 which originates at the input processing
circuitry 32 and the composite sync is separated by a sync
separator circuit 350, which provides the vertical sync signal on
output line 352 and horizontal synchronization signals on line 354.
Both of these separated signals are applied to a field decode and
logic circuit 356 and the H horizontal synchronizing signals also
are applied to a 1050 counter and logic circuit 358 as well as to a
subcarrier phase to horizontal sync synchronization circuit
360.
Since the NTSC four field sequence contains a total of 1,050
horizontal lines, the H sync being applied to the 1050 counter
logic enables it to provide unique output signals on lines 364,
366, 368 and 370, which correspond to the first line of each field
and which are applied to the field decode and logic circuitry 356
to enable it to provide signals on a frame identification output
line 372 as well as on a field identification output line 374.
These lines extend to a programmable read only memory (PROM) and
signal generator 376 as well as back to the 1050 counter and logic
circuitry 358. Line 370 from the 1050 counter and logic 358 is also
applied to the PROM and signal generator 376 so as to identify the
start of each four field NTSC sequence. A signal on line 375 is
also applied to the AND 345 and is effective to provide a control
signal thereto that is delayed for the horizontal line interval and
is active for the duration of the active video interval which
results in the application of a unique digital word being
successively asserted on the data stream each 1050th line, i.e.,
every fourth field, for use by the serve related circuitry 200.
Also, eleven lines 377 and 379, which provide the actual horizontal
video line number of the 1050 counter 358, extend to the PROM and
signal generator 376 for insertion into the synchronization
sequence. The synchronization circuitry 360 is effective to
synchronize the subcarrier phase to horizontal sync and provides a
reset pulse on line 378 that resets a 455 counter and programmable
read only memory (PROM) 380, the counter of which has a terminal
count equal to the number of subcarrier cycles in two video lines,
it being understood that there are 227.5 cycles of 3.58 subcarrier
in each video line for an NTSC system.
The counter and PROM 380 are operable to generate basic timing
signals for controlling an address counter 382 as well as the clock
sequence generator 346 for inserting the digital synchronization
sequence into the digital color television signal during the
appropriate part of the horizontal interval and, thereby, form the
processed color television signal. The PROM circuitry and 455
counter 380 also provide signals on line 384 which specify whether
a line is an even or an odd television line and line 384 is
connected to the field decode and logic circuitry 356, the PROM and
signal generator 376 and to the synchronization circuitry 360. The
455 counter and PROM circuitry 380 also provide clock sequence
signals on line 385, sync word control signals on line 386 and a
sequence end signal on line 387, all of which are applied to
control the operation of the clock sequence generator 346.
Additionally, the 455 counter and PROM circuitry 380 provides a
window of one subcarrier cycle on line 388 which is applied to the
synchronization circuitry 360 for use in synchronizing the
subcarrier phase to the horizontal sync signal. The 455 counter and
PROM circuitry 380 also provide various 3.58 MHz related control
signals that are applied to switching circuitry 196 for supplying
he record 3.58 clock to the memory RAM 1 through RAM 4 using the
record 3.58 MHz signal that is derived from the phase shift clock
generator and burst store circuitry 42 that has been described with
respect to the block diagram of FIG. 11. The 455 counter and PROM
380 control the address generator 382 which addresses, via lines
390, the PROM signal generator 376 that generates the ID 1 and ID 2
sequences in the tenth and twelveth cycles (labeled Nos. 9 and 11
and in the specific circuitry herein) of the digital synchronizing
sequence, as well as the framing information contained in the
eleventh cycle thereof. Moreover, it generates the binary coded
number 5 which is used in the "005" clock sequence contained in the
first nine cycles of the synchronization sequence, all of which
have been described herein with respect to FIG. 6. The actual
generation of the 005 sequence is accomplished by the PROM and
signal generator 376 together with the clock sequence generator
346, with the latter generating zeros at the appropriate times and
the PROM signal generator 376 generating the number 5 where it is
to be inserted. As will be appreciated from the ensuing description
thereof, the PROM and signal generator 376 could be used to
generate the entire "005" sequence if desired.
The specific circuitry that can be used to carry out the operation
of the block diagram shown in FIG. 12 is illustrated in FIGS. 20a,
20b, 20c, 20d, 20e, 20f and 20g, each of which contains circuitry
that comprises one or more of the blocks of FIG. 12 and which are
interconnected with the illustrated lines between the blocks.
Moreover, the schematic circuits specifically illustrated in the
particular FIG. 20 drawing are identified adjacent the
corresponding block thereof in FIG. 12. The operation of the
circuitry will now be broadly described in conjunction with the
specific schematic diagrams.
Turning initially to FIG. 20a, the composite sync signal is applied
at input line 348 and is used to trigger a monostable multivibrator
400 which has complementary outputs on lines 354 which provide the
horizontal rate and horizontal sync signals. The composite sync
signal is also applied to vertical sync integrator circuits
indicated generally at 402 which is connected to a vertical
synchronization counter 404 that has an output line 352 which
generates a vertical sync signal at the fourth broad pulse of the
vertical sync signal.
Turning to FIG. 20b, the vertical sync and horizontal rate signals
are applied via lines 352 and 354, together with the even or odd
line information on line 384 to a video field decoder 408 which
includes a pair of flip-flops 410 that have output lines that are
connected to logic gates, indicated generally at 412, which provide
steering information identifying the four fields of an NTSC
sequence, with the outputs of these gates being true for a short 2
microscope pulse during preselected lines of each of the fields.
Thus, the outputs of the logic gates 412 are applied to another set
of NAND gates 414 which, together with lines 364, 366, 368 and 370
from the 1050 counter and logic circuitry 358 provide steering and
thereby insures that the information is synchronized. The logic
gates 414 selectively either clear or preset flip-flops 416 and 418
which have respective output lines 372 and 374 which provide the
frame and field identification information for the PROM and signal
generator 376. The circuitry of FIG. 20b also provides bit loading
numbers as well as a video load signal on lines 375 that are
applied to the 1050 counter and logic circuitry 358.
With respect to the 1050 counter and logic circuitry shown in FIG.
20c, the frame and field information lines 372 and 374, and the
horizontal sync clock line 354 are connected, together with the
video load and but load lines 375 to a 1050 counter 422 which has
selected output lines 424 that extend to logic circuitry indicated
generally at 426. Also, the entire 12 lines of the counter,
comprising the 6 most significant bit lines 377 and the 6 least
significant bit lines 379 are connected to 4-to-1 switches
associated with the circuitry shown in FIG. 20f as will be
described herein. The logic circuitry 426 has four lines 427 that
are connected to flip-flops integrated circuit and the signals
applied via lines 427 are clocked through the flip-flops 428 and
provide the signals on lines 364, 366, 368 and 370, which identify
the horizontal lines 788, 263, 526 and 1051, respectively, which
are the first lines of each field in a four field NTSC sequence.
The flip-flops 428 merely reclock the signals from the logic 426 in
accordance with the horizontal rate being applied on line 430 from
a monostable multivibrator 432 that is triggered by the H rate
signal on line 354. The outputs on lines 364, 366, 368 and 370 are
maintained true only for the duration of the corresponding line
occurrence. Line 370 is also connected to a monostable
multivibrator 436 which has an output line 438 to a NAND gate 440
which is enabled by the video load line 375 which causes the
counter to be reset or reloaded when it has reached the terminal
count of 1050.
With respect to the 455 counter and PROM circuitry 380 shown in
FIG. 20d, a reset pulse on line 378 is applied to a counter 450
which has a terminal count of 455 and which is reset by the reset
pulse which is synchronized on the proper odd line as determined by
the synchronization circuitry 360. The counter 450 is clocked by a
record 3.58 MHz clock on line 238 and has output lines 452 which
control a programmable read only memory (PROM) 454 having output
lines 456, 458, 460 and 462 on which true signals are asserted at
the proper addresses in accordance with the program in the memory
at the addresses determined by the signals from the counter on
lines 452. The output lines of the PROM 454 are clocked through the
flip-flops 464 and provide signals on output lines 466, 468, 386,
472, 385 and 388, which extend to various locations of the
circuitry including the clock sequence generator 346 as well as the
PROM and signal generator 376, address generator 382 and the
synchronization circuitry 360. More specifically, line 456 from the
PROM 454 provides a load pulse which is clocked through the
flip-flops 464 with the Q output line 466 providing a load control
to the counter 450, while the Q output 468 clocks a second D
flip-flop 476 which provides the even or odd identification
information for a particular television line on output lines 384
and 478. Line 478 is also extended back to an address input of the
455 counter 450 and indexes the counter to alternately load the
number 246 and 247 on successive television lines so that at the
end of two lines, 455 counts will be produced which correspond to
the total number of whole subcarrier cycles that occur in two
television lines. Line 458 from the PROM 454 is clocked through the
D flip-flop 464 and provides a clock sequence signal on line 385.
The Q output line 472 is connected to a monostable multivibrator
480 and D flip-flop 482 and provides a sequence end signal on line
387 that is supplied to the clock sequence generator 346. Line 460
from the PROM 454 is clocked through the flip-flop 464 and provides
a sync word control signal on line 386 that is applied to the clock
sequence generator 346 as well as the address generator 382 that
controls the PROM signal generator 376. The output line 462 from
the PROM 454 is clocked through a flip-flop 464 and provides a
window of one subcarrier cycle on line 388 which is applied to the
synchronization circuit 360.
With respect to the PROM signal generator 376, and referring to
FIG. 20f, the frame and field information on lines 372 and 374,
respectively, are applied to the progrmmable read only memories
PROM 376 together with the line 384 that identifies whether a
television line is an even or an odd numbered line and this
information is applied to three addresses of the PROM 376. Other
address information is generated by a sequence address generator
480 which is clocked by the 3.58 MHz clock on line 238 and is
cleared by the sync word control signal on line 386. The address
counter 480 has output lines 482 that extend to four address inputs
of the PROM 376 and together with a signal generated by line number
1050, being applied to line 370 and sequenced through two
monostable multivibrators 483 and 484, is asserted on line 486 that
is also applied to one of the address lines of the PROM 376. The
first multivibrator 483 delays the triggering of the second
multivibrator 484 until the horizontal blanking interval has ended
and then the multivibrator asserts an active signal on line 486 for
a period corresponding to the video interval. This results in the
unique word from the circuit 376 to be inserted into the data
stream during the active video for one line of each four fields for
use by the servo to obtain vertical synchronizing information. The
output information from the PROM 376 appears on lines 488 which are
clocked through D flip-flops 490 and provide eight bits of
information on lines 341 that are connected to the 4-to-1 switch
491.
The information that is supplied by the PROM and signal generator
376 contains the ID 1 and ID 2 information in the tenth and twelfth
cycle locations of the twelve cycle sequence, as well as the frame
and field information in the eleventh cycle. In tis regard, on odd
television lines, the ID 1 is the binary coded decimal number 2 and
the ID 2 is the binary coded decimal number 10. Similarly, for even
numbered telvision lines, the ID 1 is the binary coded decimal
number 20 and ID 2 is the binary coded decimal number 40. The
framing information identifies which frame, whether it is the first
or second frame of the NTSC sequence as well as the first or second
field thereof. By utilizing both the frame and field information,
the specific field of the four field sequence can be determined on
a line-by-line basis. As previously mentioned, the horizontal line
number of the lines for a full four field sequence (or a full 8
field sequence for the PAL or SECAM system) is preferably inserted
in the eleventh cycle of the digital synchronization sequence and
is done by selective operation of the 4-to-1 switches 491. In this
regard, lines 341 supply the data from the PROM 376 and is passed
through the switches 491 except during the eleventh cycle when the
framing information is asserted. This is accomplished by
selectively controlling the switches 491 to sequentially pass the
data for word A from lines 377, the data for word B from lines 379
and the data for word C from the remaining input lines 381
generated by circuitry that is not shown.
To control the switching of the switches 491, the clock sequence
signal on line 385 is used to trigger a monostable multivibrators
493 at the end of the clocking sequence, i.e., at the end of the
first 9 cycles of the synchronization sequence shown in FIG. 6(2).
The monostable multivibrator 493 provides a delay equal to one
cycle of the sequence, specifically the cycle containing ID 1 and
then triggers a second monostable multivibrator 497 which provides
a one cycle duration pulse on lines 499 that steers flip-flops 501
and 503 to synchronize the address control signals on lines 505 and
507 extending to the address data selectors 491 with the input
data. The flip-flops 501 and 503 have output lines 505 and 507
extending to the 4-to-1 switches 491 and generate the addresses for
sequentially selecting lines 377, 379 and 381 during the eleventh
cycle and then selects lines 341 for the twelfth cell containing ID
2 and maintains this address until the end of the next clock
sequence occurring at the next horizontal line. The flip-flops are
clocked by the record 10.7 MHz clock on line 239 so that the three
words A, B and C can be inserted in the single cycle of the
sequence that occurs at the rate of 3.58 MHz.
The PROM 376 also generates the binary coded number 5 that is used
in the nine cycles of clock sequence previously described with
respect to FIG. 6. After the data has been clocked through the
flip-flops 490 using the 3.58 MHz clock applied via line 238, the
data on lines 342 is applied to 2-to-1 switches 340 which are shown
in FIG. 20g.
As shown therein, the switches either select lines 342 or lines 348
and presents the data from the selected lines on output lines 492
and the data is reclocked by D flip-flops 495 and appears on lines
48 that extend to the switches 50 and 52 shown in FIG. 1. It should
be appreciated that the flip-flops 495 are clocked using the record
10.7 MHz clock signal that is applied on line 239 that extends to
the clock input of the flip-flop 495, whereas the data from the
PROMs 376 is presented using a clock rate of 3.58 MHz. Thus, if the
data presented by the PROM has a duration of one cycle of the 3.58
MHz clock, then it will be clocked onto the lines 48 three times
using the 10.7 MHz clock. Thus, the ID 1 and ID 2 information is
repeated three times in the data stream on line 48. However, with
respect to the "005" clock sequence described with respect to FIG.
6, the number 5 is only asserted on lines 492 by the switch 340
during the final cycle of 10.7 or, stated in other words, during
the last 1/3 cycle of the 3.58 clock interval. This is accomplished
by using line 496 to enable only the number 5 to be asserted on
lines 492 during this desired time period. When line 496 is at a
high level, then the switch 340 provides zeros at all output lines
492 and the D flip-flop 494, which is controlled by clock sequence
generator 346, is caused to provide this level during the first 2/3
of each cycle of subcarrier during the nine cycles where the "005"
clock sequence is to be generated. The sequence end signal on line
387 disables the flip-flop 494 at the end of the nine cycles of the
clock sequence. The 2-to-1 switch 340 otherwise selects between the
lines 342 and lines 348 by the control of select line 498 which,
when low, selects lines 348 and when high, selects line 342. The
line 498 is controlled by a flip-flop 500 and is preset by the
clock sequence signal on line 385 and is clocked by line 502 that
is connected to a monostable multivibrator 504 that is triggered by
a sync word control signal on line 386.
The circuitry of FIG. 20g also performs another function that
effectively protects the word synchronization detection circuitry
in the decoders 138 and 140. In this regard, the word
synchronization is detected by detecting the "005" sequence, which
comprises 24 consecutive 0's followed by the logical states 101.
Because this "005" sequence is provided during the synchronization
sequence, it should only be detected during this time and the
circuitry of FIG. 20g prevents this sequence from occurring at any
time other than during the synchronization sequence. This is
accomplished by forcing the least significant bit of the 8 bit
digital words to a logical 1 state any time the words contain all
logical 0's during the active video portion of the data stream,
i.e., at any time other than during the synchronization sequence.
This is accomplished by a NAND gate 508 having the data lines 38
applied to the inputs and providing an output signal that is
applied to the D input of a flip-flop 509 when all 0's are present
on the lines 38. A line 511 from the flip-flop 500 effectively
disables the flip-flop 509 during the synchronization sequence so
that a logical 1 will not be asserted during the time when the
consecutive 0's are to be present. However, during the time when
the active video is occurring, whenever all logical 0's are present
on the video lines 38, the flip-flop 509 will provide an output
signal on line 515 which presets a flip-flop 517 and forces it to a
logical 1 as is desired.
The remaining portion of the block diagram shown in FIG. 12 for
which specific circuitry has not been described concerns the
synchronization circuitry 360 shown in FIG. 20e which provides the
reset signal to the 455 counter and PROM 380 at the proper time by
insuring that the subcarrier phase is synchronized to horizontal
sync. Stated in other words, the circuitry shown in FIG. 20e
determines that the phase of the subcarrier is synchronized with
respect to horizontal sync by insuring that the H sync is phased to
occur in the middle of a subcarrier cycle. The circuitry
essentially establishes the even or odd relation of the lines by
making a decision with respect to the location of the horizontal
sync relative to subcarrier and thereafter maintaining the
relationship so that the odd designated lines are always odd and
even lines are always even. The circuitry thereby defines whether a
line is even or odd and maintains that relationship throughout the
recording of the data so that no problems with respect to this
relationship will exist during subsequent reproducing.
To accomplish this decision making and referring to FIG. 20e, the
horizontal sync signal from the sync separator 350 is applied via
line 354 to a centering monostable multivibrator 510 which is
capable of moving the phase of the horizontal sync forward or
backward as a result of controlling the conduction of a transistor
512 which can vary the pulse width of the output of the one-shot
510. The output of the monostable multivibrator 510 appears on line
513 which is applied to another monostable multivibrator 514 that
asserts a relatively narrow pulse on line 516 which is directly
connected to a NAND gate 518 and also via line 519 and a number of
components 520 while generate a propagation delay. When the signal
designating a line as being even or odd appearing on line 384 is
also applied to the NAND gate 518, the gate 518 asserts an
extremely narrow pulse of 20-30 nanoseconds on line 522 which
clocks a flip-flop 524 to which the D input is supplied by the one
cycle of subcarrier via line 388. The even or odd defining signal
on line 384 is synchronized to the subcarrier and is also applied
via inverter 526 to one input of a NAND gate 527 which has other
inputs supplied by the line 516 and line 519 from the propagation
delay 520 so that the NAND gate 527 also produces a narrow 20-30
nanosecond pulse on line 528 which is inverted by inverter 530 and
is applied via line 532 to a clock input of a second flip-flop 534,
the D input of which is also supplied by the line 388. Thus, the
flip-flops 524 and 534 are clocked by signals that are synchronized
to H rate which provide timing signals on lines 536 and 538 which
are clocked into D flip-flop 540 and 542 using the subcarrier
synchronized signal on line 384 and provide four possible
conditions at the outputs of the flip-flops 540 and 542, i.e., one
or both of the clocks applied via lines 532 and 522 may be inside
or outside of the window. The logic and other circuitry indicated
generally at 544 examine these possible conditions and provide a
signal on line 546 which effectively controls conduction of the
transistor 512 to advance or retard the H sync position to clearly
select one cycle of subcarrier in the middle of which the
horizontal sync is to be located. The 3.58 clock signal on line 238
clocks a flip-flop 550 which has the D input supplied via line 552
from the monostable multivibrator 514. The output 558 of the
flip-flop 550 is coupled through a series of components 554, which
provide a propagation delay, to one input of a NAND gate 556, which
has a second input which is directly supplied by line 558. The NAND
gate 556 generates a narrow pulse on line 560 from the signal
provided by flip-flop 550, which enables NAND gate 562 to generate
the reset pulse that is placed on line 378 when the signal on line
564 is activated by the circuitry 544. Thus, the reset pulse occurs
at a time that is precisely in the middle of a subcarrier cycle and
thereby always resets the 455 counter at the proper time on an odd
line.
The processed television signal, containing the digital
synchronization sequence, is applied on the eight lines 48 that
extend to the switches 50 and 52, one of which is shown in detail
in FIGS. 18a and 18b which together comprise an electrical
schematic circuit diagram of the switch 52 and the line
identification decode circuitry that is used to control the
switches 128 and 130 via line 142, from logic circuitry 200.
Turning initially to FIG. 18a, the eight lines 48 containing the
data to be recorded is applied to one set of inputs of a 2-to-one
switch 580, which selects between lines 48 or the sets of lines 148
carrying the reproduced data from the decoder, dropout processing,
clock acquisition and deserializing circuitry 140. The lines 148
have MECL level signals which are converted to TTL levels by
circuits indicated generally at 582 and all of the inputs except
for the parity bit are applied to the alternate terminals of the
2-to-1 switches 580. During recording, the lines 48 are selected
and during reproducing the lines 148 are selected. In this regard,
it should be appreciated that the entire circuitry shown in FIGS.
18a and 18b is duplicated and that one set of lines from the
decoder circuits in one of the channels consist of lines 146 while
the lines from the decoder circuit of the other channel consists of
lines 148. The selection of either set of input lines to the 2-to-1
switch 580 is controlled by a line 586 which is controlled by logic
in response to the selection of either a recording or reproducing
operation. When the level on line 586 is low, the lines 48 carrying
the processed television signal to be recorded are selected and the
signal is passed through the switch 580 for eventual application to
the memories RAM 2 and RAM 4. When the level is high, the
reproduced processed television signal received from the decoder
and passed through the switch 580 for eventual application to the
memories.
The data lines 148 also include a parity bit line, but it is not
applied to the 2-to-1 switch but is rather connected directly to an
input of a shift register 584. The 2-to-1 switch 580 also has clock
inputs which include 1.6 MHz and 4.8 MHz reproduce clocks received
from the decoder via lines 590 and 1328 and lines 1332 and 594,
respectively and 3.58 MHz and 10.7 MHz record clocks received from
input clock generator circuit (FIG. 11) via lines 238 and 592 and
lines 239 and 596, respectively. As previously described with
respect to the block diagram in FIG. 1, the clock rate of the 8-bit
parallel data that is received on lines 48 by the 2-to-1 switch 580
for writing into the random access memories 60-66 during the record
operation is essentially at the sampling rate of 10.7 MHz while the
9-bit parallel data that is received from the decoders on lines 146
or 148 during the reproduce operation is at the rate of 4.8 MHz.
The received data is transmitted to the memories 60-66 as 24-bit
parallel data at a 3.58 MHz rate during record operations and at a
1.6 MHz rate during reproduce operations. The four clocks are
applied to the 2-to-1 switch 580 which selects between the 3.58 MHz
and 10.7 MHz record clocks or the 1.6 MHz and 4.8 MHz reproduce
clocks. Thus, one of these sets of clocks, i.e., record or
reproduce clocks, appears on line 598 and 600 and are used to
control the timing of the components of the circuitry shown in
FIGS. 18a and 18b. More specifically, the clock on line 600
controls the shift register 548 and a series of shift registers 602
which have input lines 604 comprising the data from the 2-to-1
switch 580. Each of the shift registers 602 and 584 receives three
consecutive bits of data and transfers them to output lines 606
which comprise 24 bits of data. Three output lines 608 from a
parity check circuit are also added to the 24 bits of information
and the lines 606 and 608 are applied to a series of D flip-flops
610 which reclock the data using the record 3.58 MHz signal on line
612 that is connected to line 598 via a pulse shaping monostable
multivibrator 614. The outputs of the flip-flops 610 are lines 56
which are the input lines to the memories RAM 2 and RAM 4. It
should be understood as previously mentioned, that while the block
diagram in FIG. 1 illustrates the record and reproduce paths as
separate paths, the actual conductors are the same, by virtue of
the 2-to-1 switch 50. The two paths shown in the block diagram were
illustrated in that manner for the sake of clearly identifying the
data flow during both operations.
The foregoing description of FIGS. 18a and 18b complete the circuit
operation that occurs during a recording operation, but as is
evident from the drawing, other circuitry is included therein which
comes into operation during reproducing and which will now be
described. With the input lines 148 being converted to TTL levels,
these lines are applied through jumpers 615 to the 2-to-1 switches
and also extend downwardly and to the right to FIG. 18b where they
are connected to a series of switches 614, 616, 618 and 620 which
are set to decode the appropriate identification number so as to
satisfy NAND gates 622, 624, 626 and 628 which respectively provide
a true output when the respective ID numbers 2, 20, 10 and 40 are
present in the reproduced data at the input line 148. The outputs
of the NAND gates pass through switches 630 and 632 and present
respective signals on lines 634 and 636 when the ID 1 and ID 2
numbers have been decoded. The signals on lines 634 and 636 are
applied to the logic circuitry 200 which will be hereinafter
described. Since the circuitry of FIGS. 18a and 18b will be
duplicated, the switches 630 and 632 will be set in one position
for one of the circuits and in the other for the duplicate
circuitry. Since each of the signal channels contains either only
even video lines and the other contains only odd lines, the
switches 630 and 632 can be appropriately set to decode the numbers
2 and 10 or 20 and 40.
With respect to the use of parity in the apparatus to provide an
indication whether the data has been accurately recorded and
reproduced, the circuitry shown in FIGS. 18a and 18b performs
parity checking and provides an error signal that commands the
drop-out compensator to insert data at the location in the data
stream where the data is indicated to be missing or incorrect. It
should be recalled that the parity bit is added in the data stream
by the encoder circuitry 82 before the data is recorded. During
reproducing, the signal from the decoder and other circuitry 140
includes a parity bit data which is applied to the shift register
584 and for three successive 8 bit words, provides the most
significant bit parity bit on line 640, the second most significant
bit parity bit on line 642 and the third and fourth most
significant bit parity bit on line 646, which are respectively
connected to parity checkers 648, 640 and 652. The output lines 606
from the shift registers 602, as previously mentioned, contain the
bit data for three successive samples and the most significant bit
data from three successive samples of the data stream is applied to
the parity checker 648. Similarly, the data of three successive
samples of the second most significant bit are applied to the
parity checker 650 and the data of three successive samples of both
the third and fourth most significant bits are applied to the
parity checker 652.
The logical state of parity bit is selectively added as either a
logical 1 or logical 0 so that for three successive samples,
including the parity bit, an even number of logical ones (no ones
is considered even) obtains, and the parity checkers 648, 650 and
652 merely process the data applied thereto and provide a true
signal on outputs 654, 656 and 658 is an even number of ones is
received. The signals are respectively applied to AND gates 660,
662 and 664. Also, all three of the output lines are applied to
another AND gate 666. If all outputs are true, AND gate 666
provides a high true output on line 668 which enables the other AND
gates 660, 662 and 664 in addition to providing a true signal that
is clocked through the flip-flops 610 to provide a signal on line
670 that extends to logic circuitry indicated generally at 672, the
operation of which will be described hereinafter. If even one of
the parity checkers detects a parity error, then all parity
channels are forced to provide the same indication, by virtue of
line 668 disabling the AND gates 660, 662 and 664. The outputs of
AND gates 660, 662 and 664 comprise the lines 608 which are clocked
through the flip-flop 610 and provide signals for use by the
drop-out compensator to specify that one or more of the first four
most significant bits of three successive samples contains a parity
error or that a RF drop-out has occurred and that other data should
be inserted therefor.
The parity error signal on line 670 is applied to circuit 672 which
effectively integrates the error signal by determining if it
exceeds about four closely located groups of three samples. If so,
it triggers a monostable multivibrator 673 having an output line
674 which is applied to OR gate 675, the output of which is applied
via line 676 to the AND gates 660, 662 and 664 and disables them
for a longer time than is actually indicated by the parity checker
outputs, i.e., for another 3 to 6 samples. This is to safeguard
against the possibility that random noise could generate a true
parity check in a series of bad cycles of data and thereby extends
the duration of the parity error signals on lines 608. If random
noise which generated a true parity output would be allowed to pass
onto lines 608, the bad video data which parity falsely indicated
as being good would cause either a flash or a black hole in the
displayed video image. While random noise would not generate a
significant number of true parity indications, the circuitry 672
disables such occurrence during the presence of a series of
detected parity errors.
In accordance with another aspect of the circuitry shown in FIGS.
18a and 18b, in the event that the decoder circuitry 138 or 140
detects an RF drop-out, for example, when information is not
reproduced due to an imperfection in the tape or the like, a
drop-out indicative signal is generated and applied to line 677,
which is converted to TTL levels and then applied to the circuitry
672 shown in FIG. 18b. The signal on line 677 is applied to gate
678 and its output is applied via line 679 to the gate 675 which
forces a parity error signal on to line 676. The signal on line 677
also triggers a monostable multivibrator 681, which has output line
680 that is also applied to the OR gate 675. The output provided by
the multivibrator 681 extends the duration of the drop-out and the
forced parity error signal beyond its actual length, i.e., another
six or nine samples for example, to permit internal clocks and the
like to resettle after the drop-out has terminated. The signal on
line 677 also provides a composite drop-out output signal on line
682 which is extended to logic circuitry 200 and essentially
precludes that circuitry from processing the ID 1 and ID 2 signals
for acquiring word sync. The H/8 signal applied to line 686 extends
to circuitry shown generally at 688 which provides an error rate of
the number of parity and drop-out induced errors that are
occurring. Since the H/8 signal is the rate at which head switching
occurs, and during this time period the errors should not be
counted since they are not a true indication of the error rate
occurring in the active video signal.
The generation of the drop-out signal provided on line 682 is
inhibited during the synchronizing sequence interval by the
sequence window signal provided on line 1270 (FIG. 18a) by the
circuitry of FIG. 10. The sequence window signal triggers a
one-shot 601 to set the following D latch 603 to place on its
output lines 605 and 607 inhibit signals that are coupled to the
circuitry to inhibit the generation of the drop-out signal. The
inhibit condition remains on lines 605 and 607 until the composite
ID signal is provided on line 1726 by the circuitry of FIG. 10. The
composite ID signal is delayed by delay means so that the inhibit
condition is removed from the lines 605 and 607 by resetting the D
latch 603 just before the beginning of the video interval portion
of the processed television line.
The 27 bits of data on a parallel lines 56 are applied to the
respective memories RAM 2 and RAM 4 for writing the data therein.
Each of the random access memories RAM 1 through RAM 4 comprises
specific circuitry, portions of which are shown in detail in FIG.
13. Those portions not shown in FIG. 13 are merely redundant of the
general design of the circuitry. The input lines 54 or 56 are
separated into three groups of nine lines, each group of which
extends to a 256 bit random access memory integrated circuit 800 of
which only 6 of the total of 27 are shown. Each set of the lines 54
or 56 is connected to the input terminal of the memory circuitry
800 as shown. Similarly, each of the memory circuits 800 has an
output line 802 that extends to a tri-state gate 804 having an
output line that is either line 70, 75, 150 or 154 depending upon
which RAM is identified. However, the single output lines from each
of the memory circuits 800 extends to the 2-to-1 switch 152, as
well as to the 24-to-8 bit converters 72. Since the memories are
connected to operate in pairs, i.e., memories RAM 1 and RAM 3 have
their inputs and outputs interconnected as do memories RAM 2 and
RAM 4, the tri-state NAND gates 804 effectively isolate the
individual memory circuits 800 from output lines when they are not
enabled so that only the outputs from individual memory circuits
800 for one of the random access memories, such as RAM 1 or RAM 3,
for example, will be asserted onto the output lines 70 or 74.
Control lines 806, which have inverters therein as shown, enable
and disable the tri-state NAND gates 804 at the appropriate times
as shown and described with respect to the timing diagrams of FIGS.
4b and 5b. A write enable signal on line 808 is applied to a
monostable multivibrator 810 which can be adjusted to position the
write pulse with respect to the data and output line 812 is
connected to the write enable input of each of the memory
integrated circuits 800. The level of output line 812 controls
whether write or read operation can occur with respect to the
memory. In the absence of a high write pulse on line 812, the
memories are in a condition to read data from storage. When the
write pulse is placed on the 812, the memories are conditioned to
write data into storage for the duration of the write pulse. The
timing for the write enable signal is shown for each of the random
access memories RAM 1 through RAM 4 in FIGS. 4b and 5b.
Each of the memory circuits 800 is addressed via eight address
lines 814 which are controlled by an address generator 816 so that,
for any address generated by the address generator 816, all of the
individual random access memory integrated circuits 800 will have
the identical address being accessed. Thus, for the 27 bits of data
that is input, one bit will be appropriately written into or read
out of one of the memory circuits 800 for each address that is
generated by the address generator 816. While only two of the
address lines from the address generator 816 are shown to be
actually connected in the drawing, it should be understood that the
other six lines are similarly connected to the remaining address
lines that are shown adjacent the memory circuits 800. The address
generator 816 is clocked by clock line 818 from a monostable
multivibrator 820 that is used to properly time the clocking with
respect to the data on the input lines 54 and 56.
A clock signal applied on line 822 is used to trigger the
monostable multivibrator 820 with a clock that is determined by the
mode of operation, i.e., whether it is writing or reading during a
recording operation or writing or reading during a reproducing
operation. The clock is either a 3.58 MHz or 1.6 MHz clock and both
of these frequency clocks originate from one of two sources. During
a record operation, the data is written into the memories 800 at a
rate of 3.58 MHz under the control of the record clocks provided by
the clock generator circuitry 42. The data to be recorded is read
from the memories at a rate of 1.6 MHz determined by a clock signal
provided by the encoder circuitry 82. During a reproduce operation,
the data is written into memory at the lower 1.6 MHz rate
determined by a clock signal that originates from the decoder
circuitry 138 or 140. The reproduced data is read from the memories
at the rate of 3.58 MHz determined by a clock signal obtained from
and synchronized to station reference. The clock on line 822 also
is applied to trigger a monostable multivibrator 824 to properly
time the write pulses with respect to the data that is present on
the input line 54 or 56.
The address generator 816 is controlled during record and reproduce
operations by reset signals placed on a line 830. The reset signals
reset the counter 816 to zero and thereby insure that the data is
written at address zero at the beginning of the digital
synchronization sequence. The reset signal on line 830 originates
at the logic circuitry 200. During reproducing or playback, the ID
1 and ID 2 control signals appear on lines 832 and 834,
respectively, which are inverted and applied to a NAND gate 836
with line 834 being inverted again and applied to one address input
of the address generator 816 so as to load it with the proper load
number for writing data into the memories. A read reset signal on
line 838 from the control logic 200, generates a load signal for
loading the address generator 816 to begin reading the data from
the memory at the proper time.
During record operations, the data read from the random access
memories RAM 1 through RAM 4 is asserted on lines 70 and 74 that
extend to respective 24-to-8 bit converters 72 and 76, one of which
is shown in the electrical schematic diagram of FIG. 14a, the
24-to-8 bit converter being the circuitry shown to the left of the
generally vertical dotted line. The data on lines 70 or 74 is
applied to a series of D flip-flops, indicated generally at 850,
which reclocks the data using a 1.6 MHz clock signal on line 852
that is generated by the encoder circuitry shown generally at 900
in FIGS. 14a and 14b. The data that is clocked through the
flip-flops 850 appears on lines 854, which extend to a number of
parallel-to-serial shift registers 856 which are loaded by a 1.6
MHz clock signal on line 858. The data from the input lines 854 is
sequentially clocked out on lines 860 at a three times faster rate
determined by a 4.8 MHz clock signal generated by the encoder
circuitry 900 and placed on line 862 coupled to the output clock
terminal of each of the shift registers 856. Thus, the 24 bits of
data being asserted on the input lines 854 is converted to 8 bits
of data that is transferred at a rate that is three times faster.
The data on lines 860 is passed through jumpers 861 and then
through gates 863 and is applied to another parallel-to-serial
shift register 864 which has an output line 868 that contains the
serialized NRZ data on the input lines 866. The jumpers can be used
to change the order of the data bits so that the three most
significant bits are not adjacent one another and would therefore
not be adjacent one another in the serial data after being
converted to serial data. This would decrease the vulnerability to
losing all of the most significant bits due to a drop-out having a
duration of 2-to-4 bits. If the order of the data is changed, it
must be similarly changed back to its proper order during
reproducing through the use of the jumpers 615 in the circuitry 50
and 52 (FIG. 18a) as should be understood. The clock rate of the
data on the input lines 866 is 4.8 MHz as previously mentioned and
comprises 8 bits of data at this rate. To provide a serial output,
the data is clocked onto line 868 using a clock signal that is nine
times faster than the 4.8 MHz clock signal, i.e., approximately 43
MHz. The clock rate is nine rather than eight times faster because
of the addition of a parity bit to each 8 bit word being asserted
on input line 870, which originates from parity generating
circuitry that will now be described.
The most significant bit, the second, third and fourth most
significant bits for three consecutive data words are applied to
parity generating circuits 872, 874 and 876, in addition to being
applied to the shift registers 856. Thus, the three of the lines
854 which are applied to the parity generator 872 comprise the most
significant bits of three successive samples. Similarly, the three
lines that are input to the parity generator 874 comprise the
second most significant bits for three successive samples and the
six lines that are applied to the parity generator 876 comprise the
third and fourth most significant bits for three successive
samples. The parity generators examine the data on the inputs and
assert a low level on each of the output lines 878 in the event
that an even number of logical ones occurs in data that is applied
to the corresponding parity generator. The three lines 876 are
reclocked by the 1.6 MHz clock on line 880 so as to provide the
data on lines 882 that are connected to a parallel-to-serial shift
register 884. The shift register 884 is clocked by the 4.8 MHz
clock on line 886 so that the parity bit from each of the lines 882
is serially asserted on output line 870 that extends to the
parallel-to-serial shift register 864. The parity generating
circuitry that is shown and described in detail herein is one type
of parity that conveniently can be employed in the apparatus.
However, it should be understood that the particular significant
bits that are examined need not be from the three successive
samples but may be from three individual samples that are not
successive. However, three successive samples are most convenient
because they are simultaneously present in the parallel presence of
three successive eight bit data words.
The frequencies used by the circuitry, i.e., the 43 MHz clock, the
4.8 MHz clock and the 1.6 MHz clock are produced by an 86 MHz
oscillator, indicated generally at 890, that provides the basic
timing reference for the operation of the encoder 900. The
oscillator 890 provides an output signal on line 892 which is
applied to level and shaping circuitry 894 to generate the 85 MHz
signal on line 896, as well as line 898, with the 86 MHz clock
signal line 896 being used to reclock the serialized data after it
has been encoded by encoder 900 in a format that will be
hereinafter discussed. The 86 MHz signal on line 898 is applied to
a pair of divide-by-2 dividers 902 and 904, the latter of which
produces an approximately 43 MHz signal having complementary phases
on lines 906 and 908. The complementary phases 43 MHz signals are
applied through pulse narrowing logic circuitry 909 and 910 to
provide very narrow pulses of opposite phase at the 43 MHz clock
rate on lines 911 and 912 which are used by the encoder 900. The
divide-by-2 divider 902 has its output connected to the first of
three successive divide-by-3 dividers 914 which are used to
generate a 1.6 MHz clock on line 916, a TTL level 1.6 MHz clock on
line 852 and a 4.8 MHz clock on line 862.
The serialized nonreturn-to-zero (NRZ) data being clocked at a rate
of 43 MHz on line 868 is applied to the encoder 900 which encodes
the data into a Miller "squared" channel code, which is a
self-clocking, DC-free, type of code. The DC-free code avoids the
introduction of any possible DC component into the encoded data as
a result of a preponderance of one logical state over a period of
time. Because the record and reproduce apparatus does not transmit
at DC, the presence of a DC component in the encoded data to be
recorded can introduce errors in the recovery of the data during
the reproducing. In this regard, reference is made to U.S. Pat. No.
4,027,335 by Jerry W. Miller issued May 31, 1977 and entitled
"DC-Free Encoding for Data Transmission System", assigned to the
same assignee as the present invention. As is comprehensively
described therein, the coded format can be characterized as a
self-clocking format, which provides for transmitting binary data
over an information channel of limited bandwidth and
signal-to-noise ratio where the data is transmitted in a
self-clocking format that is DC-free.
In limited bandwidth information channels which do not transmit at
DC, binary waveforms suffer distortions of zero-crossing location
which cannot be totally removed by means of linear response
compensation networks, particularly, at the high data rates
characteristic of this apparatus. These distortions are commonly
referred to as base-line wander and act to reduce the effective
signal-to-noise ratio and modify the zero-crossings of the signals
and thus degrade bit recovery reliability of the decoder. A common
transmission format or channel data code that is utilized in
recording and reproducing systems is disclosed in Miller U.S. Pat.
No. 3,108,261 issued Oct. 22, 1963. In the Miller code, logical 1's
are represented by signal transitions at a particular location,
i.e., preferably at mid-cell, and logical 0's are presented by
signal transitions at a particular earlier location, i.e., near the
leading edge of the bit cell. The Miller format suppresses any
transition occurring at the beginning of a one bit interval
following an interval having a transition at its center. Asymmetry
of the waveform generated by these rules can introduce DC into the
encoded signal. The so-called Miller "squared" code used in the
present apparatus effectively eliminates the DC content of the
original Miller format and does so without requiring either large
memory or the necessity of a clock rate change in the encoding and
decoding operations. As is described in the aforementioned Miller
U.S. Pat. No. 4,027,335 directed to the Miller "squared" format,
the data stream can be viewed as a concatenation of variable length
sequences of three types: (a) sequences of the form 1111--111
having any number of logical 1's but no logical 03 s: (b) sequences
of the form 0111--1110 having any odd number of consecutive 1's or
no 1's, with 0's occurring in the first and last positions; and (c)
sequences of the form 0111--111 having any even number of
consecutive 13 s preceded by a 0. The sequences of the type (c)
occur only if the first bit of the next following sequence is a 0.
Sequences of type (a) and (b) are encoded according to the code
rules described in the U.S. Pat. No. 3,108,261. The sequence of
type (c) is encoded according to the code rules that are described
in the U.S. Pat. No. 3,108,261 for all bits except that last
logical 1, and for this 1, the transition is simply suppressed. By
this suppression, the type (c) sequence viewed in isolation is made
to appear the same as the type (b) sequence, i.e., the final
logical 1 looks like a logical 0.
By definition, the type (c) sequence is followed immediately by a
logical 0 at the beginning of the next sequence. No transition is
allowed to separate the type (c) sequence from the following 0.
Therefore, the spatial coding is distinctive for decoding purposes
and the decoder must merely recognize that when a normally encoded
logical 1 is followed by 2 bit intervals with no transitions, then
a logical 1 and logical 0 should be provided successively during
these intervals. All other transition sequences are decoded as
according to the Miller code disclosed in the U.S. Pat. No.
3,108,261. Thus, the output on line 86 from the encoder 900
provides the serialized encoded data in the Miller "squared" format
that is applied to the amplifiers 88 and 90, for example, and the
amplified signal is then forwarded to the transducing heads for
recording on the magnetic tape.
During reproducing, the transducing head 96 carried by the head
wheel 108 reproduce the signals recorded on the tracks and apply
them to preamplifiers 109, one of which is shown in detail in FIG.
15. The input lines 950 are connected to rotary transformers of
conventional design and the derived signal is amplified and appears
on output lines 111 that are connected to the 2-to-1 switches 110
and 112, which selectively connect one of the lines 109 to output
114 or 116 extending to the equalizers 118 or 120.
With respect to the specific circuitry that can be used to perform
this switching and equalization, reference is made to FIGS. 16a and
16b, which together comprise the electrical schematic circuitry
that can be used to carry out these circuit operations. Referring
to FIG. 16a, the output of the preamplifiers 109 appears on lines
111, which are shown to extend to diode switches 970 and 972 that
are respectively controlled by head switching signals applied to
lines 974 and 976. The signals from one of the preamplifiers is
passed through the associated switch at the proper time and appears
at the line 114 that represents the input to the equalizer, which
is shown on the remainder of the drawings of FIGS. 16a and 16b.
Line 114 is applied to an amplifier indicated generally at 978
which is connected to a 6 db per octave increasing response
controller 980 that includes a low frequency compensator 982 and a
high frequency compensator 984, both of which compensate for the
nonconstant amplitude-frequency response of the reproducing heads.
As is well known in the art, the output voltage of a reproduce head
and preamplifier combination rises at low frequencies at a rate of
6 db per octave, levels off at mid-band frequencies and falls at
high frequencies. Consequently, if an overall flat amplitude
response of the play-back signal is to be obtained, it is necessary
for the equalizer to boost the amplitude at both the low and high
frequencies. To effect the boost, the circuitry 980 is applied to
an amplifier and line driver 990 which in turn is connected to a
low pass filter 992 having a cut off frequency slightly above the
half data rate. i.e.,. 21.5 MHz in the present apparatus. The
amplifier and line driver 990 and filter 992 are designed to
minimize the effect of any high frequency noise present on the
off-tape signal. The low pass filter 992 is connected to a phase
equalizer 994 which drives a second line driver 996 (FIG. 16b). The
line driver 996 has an output line 998 that is connected to a
balanced modulator circuit, indicated generally at 1000, as well as
to a delay line 1002 that is connected to another balanced
modulator circuit 1004 (FIG. 16b) as well as to a second delay line
1006 that extends to a third balanced modulator 1008. The outputs
of the balanced modulators 1000, 1004 and 1008 appear on respective
lines 1010, 1012 and 1014 (FIG. 16b) which are connected at a
common summing point 1016. The summing point 1016 represents the
input of an amplifier 1018 that is connected through a transformer
1020 to a limiter 1022 which provides the equalized output on line
1024. A circuit indicated generally at 1026 detects the presence of
an RF drop-out in the recovered signal and provides a drop-out line
1028.
The circuitry between the output of the line driver 996 and the
output of 1024 of the equalizer compensates for inter-symbol
interference of the Miller "squared" data stream that occurs during
reproducing. Inter-symbol interference can broadly be described as
a distortion of the location of the zero-crossings in the signal
which occur in the data stream, and which are distorted due to the
effect of prior and subsequently occurring signal transitions. In
other words, the zero-crossing point for a subject transition may
be phase advanced or retarded to differing degrees depending upon
what occurred immediately prior to or after the zero-crossing point
of interest. While it is, at first impression, some-what unusual to
suggest that a future transition can affect a present transition,
it must be realized that transitions are the result of the
transducing head recording and reproducing signals on and from the
magnetic tape or other medium and that three successive transitions
are in a sense the past, present and future transitions and that
magnetic influence can occur from either adjacent transition while
the transducing heads are operating. Referring to FIG. 16c(1), a
relatively long wavelength 1030 having three data cells between
transitions is shown which is followed by two successive shorter
wavelengths 1032 and 1034 which have only one data cell between
transitions. As shown in FIG. 16c(2), it is well known that the
depth of recording for the signals shown in FIG. 16c(1) are greater
for longer wavelengths, i.e., low frequency, than for short
wavelengths. Thus, the amplitude is greater for the portion 1036
associated with the longer wavelength 1030 than for either of the
portions 1038 and 1040 associated with the shorter wavelength. This
depth of recording will therefore distort the location of the
zero-crossing point from the transition of the long wavelength to
the short wavelength, i.e., the zero-crossing point 1042 shown in
FIG. 16c(1) and the distortion will affect the amplitude response
as well as the phase response, although the phase response will be
more significantly affected. The long wavelength transition may be
phase retarded as shown by the dotted line and have a zero-crossing
point at location 1044 or phase advanced as shown by the dotted
line and have a zero-crossing point at location 1046.
The circuitry located between the output line 998 of the line
driver 996 and the summing point 1016 corrects for distortion by
algebraically adding correcting signals that are proportional in
amplitude and phase displaced relative to the signal that occurred
prior in time as well as a signal that occurs later in time. This
is accomplished by (a) applying the signal on line 998 through the
first delay line 1002 to balanced modulator 1004 which provides an
output signal that is delayed from arriving at the summing point
1016 by a first predetermined time corresponding to a nominal value
of 11/2 data cells; (b) applying the signal through the first delay
line 1002 and also through a second delay line 1006 to a balanced
modulator 1008 which provides an output signal on line 1014 to the
summing point 1016 which is delayed by a greater amount which
nominally is about 3 data cells; and (c) applying the signal
directly to the balanced modulator 1000 which provides an output
signal on line 1010 that is applied to the summing junction 1016
prior to either of the outputs on lines 1012 and 1014. For a given
sample in the signal that is present on line 998 at a given time,
it will be processed through the balanced modulators and delay
lines and will reach the summing point 1016 at three successive
points in time as would samples that occurred immediately before
and after the subject sample. Thus, by forwarding the signals
through the delay lines and balanced modulators, the effect is to
phase modify the instant sample with the immediately preceding and
succeeding samples. The predominate signal in terms of amplitude is
the signal from the balanced modulator 1004 and the outputs from
the other balanced modulators 1000 and 1008 are proportionally
smaller in amplitude and are algebraically added to the predominate
signal to correct for errors in the zero-crossing portion of the
predominate signal. Referring again to FIG. 16c(1), by adding a
component signal that is phase advanced as shown at point 1046,
compensation for the phase retardation of the zero-crossing point
shown at point 1044 can be made so that the resulting zero-crossing
point is correctly shifted to the location identified as point
1040.
With respect to the operation of the balanced modulators, and
referring specifically to the balanced modulator 1004 shown in FIG.
16a, there is a constant current source represented by the
transistor 1050 which provides a current on line 1052 that extends
to the emitters of transistors 1054 and 1056. The total current is
divided and flows through the two paths and the current that flows
to the transistor 1056 is equal to the total current less the
current that is flowing to the transistor 1054. The base of
transistor 1054 is connected to a variable resistor 1058 that can
be adjusted to control the output of the balanced modulator circuit
1004. The current flowing through each of the transistors 1054 and
1056 effectively control the gain of the transistors 1060a, 1060b,
1062a and 1062b. Since the collectors of transistors 1060a and
1062b are connected together and are oppositely phased, if the
current flowing through transistors 1054 and 1056 are equal, then
the gain for transistors 1060a and 1062b will be equal and the
current on line 1064 will be zero which will cause transistor 1066
to be nonconducting and provide a zero output on line 1012.
However, if they are unequal, there will be a current that varies
in phase depending upon which transistor 1062a or 1062b is
conducting. The input signal from the delay line 1002 is applied to
the base of transistors 1060a and 1062a which will be reflected at
the output on line 1012 that will be some proportion of the
amplitude of the input signal and also phase shifted in accordance
with the preset adjustment of the variable resistor 1058.
Since the other balanced modulators operate substantially
similarly, it can be seen that the outputs therefrom can be
amplitude adjusted and some proportion of the amplitude of the
input signal can be added to compensate for the inter-symbol
interference that is present on the data. The amplitude of the
added signal generally varies between about 10 to about 15%, but
may approach about 30%. In any event, the amplitude should be that
which is necessary to adequately perform the compensation. In this
regard, the balanced modulator 1000 has a transistor corresponding
to transistor 1054 in the balanced modulator 1004 controlled by
line 1070 and a similar transistor of the balanced modulator 1008
is controlled by a line 1072, both of which are extended to
variable current sources that can be controlled by an operator who
can adjust the balanced modulator to vary the phase and amplitude
compensation in a manner whereby the inter-symbol interference is
minimized.
The equalized data, still encoded in the Miller squared code, is
applied on lines 124 and 126 that are connected to two switches 128
and 130, which are adapted to select the outputs of either
equalizer and apply the same to one of the decoder, drop-out
processing, clock acquisition and deserializing circuits 138 or 140
via lines 132 and 134. The switches 128 and 130 are adapted to
reverse the equalizer outputs if such is necessary in the event
that the consecutive lines that are being decoded are reversed
relative to recording which would ultimately produce a garbled
display of the video image, as previously mentioned. The switches
128 and 130 are controlled by a signal on line 142 that is
generated by the logic circuitry 200.
Specific circuitry that can be used to decode the Miller squared
encoded data, recover the clocks from the self-clocking data,
provide a drop-out processing as well as deserialize the data and
convert it back to a 9 bit parallel data is shown in FIGS. 17a and
17b, which together comprise an electrical schematic diagram of
this circuitry. Thus, the Miller squared data is input on lines 132
(in MECL form) which occurs essentially at a 43 Mbit rate, since
transitions can occur at both the beginning and the middle of the
bit cells with the bit cells being at a 43 Mbit rate. While the
data is in MECL form at the input, it should be appreciated that
the circuitry could be modified to accept the Miller squared data
in a form whereby the logic signal transitions are pulses which
occur at the beginning or middle of bit cells. Thus, one of the
complementary outputs of the last stage of a three stage limiter
1100 is applied to a series of three exclusive-OR (EXCL-OR) gates
1102, which generate a pulse on output line 1104 at each
zero-crossing point. The generated pulses are applied to a narrow
band pass filter 1106 and subsequently input to a limiter 1108,
which generates a square wave. The output of the limiter appears on
line 1110 as well as on a line 1112, with line 1112 extending to
another filter 1114 which is also a narrow band pass filter and the
output of the filter 1114 is applied to another limiter 1116
followed by another narrow band pass filter 1118 and yet another
limiter 1120 so as to produce the 86 MHz square wave on line 1122
that is connected to a buffer 1124 having complementary outputs.
One of the complementary outputs is applied to buffer 1126 which
provides the 86 MHz clock on line 139 that can be used by the
companion decoded as previously discussed with respect to the block
diagram of FIG. 1. The narrow band pass filters of the clock
extracting circuitry have a band pass of approximately 2 MHz.
In the event a RF drop-out occurred on one of the channels, then
the 86 MHz clock from the other decoder is used to clock the
circuitry so as to be able to retain the proper data word
synchronization and thereby be able to immediately recover the data
when the drop-out terminates. Since it is extremely unlikely that
drop-outs would simultaneously occur in both channels, the
probability is high that the 86 MHz clock can be recovered by one
or the other of the decoders for use in clocking the circuitry.
The series of limiters and narrow band pass filters successively
provide a more accurate 86 MHz clock that is used to clock the data
that is being received on the lines 132. The complementary output
of the first limiter stage 1100 contains the coded data and is
applied via line 1128 to a delay means, indicated generally at
1130, which is tapped by line 1132 and applied to the D input of a
flip-flop 1134 that is clocked by line 1110. Thus, the encoded data
output by the flip-flop 1134 on line 1136 is reclocked by a
recovered clock from the data itself and thereby removes some
errors that may be present due to propagation and timing delays
that are present in the extremely high rate 86 Mbit data. The line
1136 containing the reclocked data is also applied to a D flip-flop
1138 which is clocked by the refined 86 MHz clock signal on line
1140 that is output by a buffer 1142 which has one input supplied
by the buffer 1124. The flip-flop 1138 reclocks the data a second
time and thereby removes virtually all errors that would be present
due to propagation and other timing delays. The reclocked data
appears on line 1144 and is applied to three EXCL-OR gates 1146,
1148 and 1150, two of which provide a narrow pulse on respective
output lines 1152 and 1154 for each transition that occurs in the
data itself.
The other output of buffer 1142 is applied to a buffer 1160, which
has one output clocking a divide-by-2 flip-flop 1162 while the
other output line 1164 is applied to a buffer 1166. The output of
the divide-by-2 flip-flop 1162 is a 43 MHz signal on line 1170
which is passed through buffer 1172, is thereafter filtered by a
filter 1174. The filter 1174 forms part of a flywheel circuit which
is operable to maintain the clock at the same phase by resisting
any instantaneous variation or change of phase of the signal, due
to the delay characteristics of the filtering. The phase of the 43
MHz clock would not change until several cycles of a different
phased signal occurs. The output of the filter circuit 1174 appears
on line 1178 which is passed through a buffer 1180 to another
buffer 1182 having output line 1184 containing the 43 MHz clock
which is used to clock a shift register comprising D flip-flops
1186, 1188, 1190, 1192 and 1194. The complementary output of the
buffer 1182 is applied to OR gate 1196 which has output line 1198
that is used to clock a divide-by-9 divider indicated generally at
1200. The divide-by-9 divider 1200 is formed by 4 flip-flops wired
to issue an output on line 1316 for every nine 43 MHz clock signals
received over line 1184. The above description generally comprises
the extent of the clock generation that is used to decode the
Miller "squared" coded data and these clocks are used to clock the
circuitry shown in the drawing, thereby utilizing clocks that are
derived from the data stream itself.
To decode the Miller "squared" coded data, and referring to FIG.
17a, it is recalled that the EXCL-OR gate 1146 produces a pulse for
every data transition, whether it occurs in the middle of a bit
cell or at the beginning thereof. The pulses are applied via line
1152 to gate 1204 which has another input line 1206 supplied by
gate 1208 that is clocked by line 1184. The gate 1204 essentially
functions as a logical 1 detector and provides a true high output
pulse on line 1210 whenever a logical 1 is detected and the line
1210 effectively sets the flip-flop 1186 in the first stage of the
shift register with a logical "1". The successive flip-flops
comprising the shift register are clocked by the 43 MHz clock
signal to propagate the logical "1" state therethrough. In
accordance with the Miller "squared" coded rules used to decode the
encoded data, certain logical "1's" are suppressed in the data
stream so as to remove the DC component therefrom. To detect the
presence of such a suppressed logical "1", the output line 1154
from the EXCL-OR gate 1150 produces a short pulse at each
transition, which is passed through the buffer 1214 and provides a
reset pulse on line 1216 whenever a transition occurs. An 8 bit
counter comprised of three flip-flops 1218, 1220 and 1222 are
adapted to provide an output signal on line 1224 when they reach a
count of five or more, it being appreciated that the 8 bit counter
is clocked by an 86 MHz clock via line 1164, buffer 1166 and line
1226. The count of five intervals of the 86 MHz clock corresponds
to 21/2 cells of the 43 Mbit signal which is detected and indicates
that a logical 1 had been suppressed during the encoding process.
If a transition occurs before five counts of the 86 MHz clock, then
the counter will be reset whenever the transition has occurred.
When the counter provides an output signal on line 1224, it is
applied through the gate circuitry 1228 to generate a narrow pulse
on output line 1230, which is applied to the set input of the
flip-flop 1190 of the shift register and thereby inserts a logical
1 at the proper time where it had been suppressed during the
encoding process. The output of the final flip-flop 1194 in the
shift register appears on line 1232, which carries the decoded
nonreturn-to-zero data that is applied to a serial-to-parallel
shift register 1234. This shift register generates the 8 parallel
bits of data on lines 1236 that are applied to respective
flip-flops 1238 having output lines 146 or 148 that extend to the
circuits 50 and 52. The data on line 1232 is also applied to a D
flip-flop 1240 which is clocked by line 1242 which is at the sync
word rate and is timed so as to obtain the parity bit which is
placed on output line 1244. The sync word rate related signal on
line 1242 occurs at the rate of 4.8 MHz and is also used to clock
the flip-flop 1238 containing the bits of the parallel data.
In addition to acquiring the clock signals from the encoded data,
decoding the Miller "squared" encoded data into NRZ data, the
circuitry of FIGS. 17a, and 17b also operates to acquire the word
synchronization, i.e., identify the proper 9 bits of serialized
data that include the 8 bits of a single sample, together with the
appropriate parity bit, and the word sync detection is accomplished
by detecting the digital synchronization sequence that was added by
the sequence adder 40 during the recording process. More
specifically, the "005" sequence, when serialized and after parity
has been added, will appear as 24 consecutive zeros followed by the
sequence "101". Referring again to the EXCL-OR gate 1150 shown in
FIG. 17a, its output line 1154 is also applied to a buffer 1250,
which has an output line 1252 upoh which a pulse appears during
each transition of the data stream. The signals on line 1252
effectively reset a pair of flip-flops 1254 and 1256 which,
together with four successive gates and buffers, 1258, 1260, 1262
and 1264, detect the occurrence of the digital sequence " 101".
However, the "101" sequence could easily occur at various locations
in the active video data interval of the processed television
signal and for this reason, an input line 1270 has a sequence
window signal that is only true during the time in which the "005"
sequence is occurring, i.e., for a period of about 4 to 5
microseconds during each horizontal line, and this signal on line
1270 is applied to gate 1272 having an output line 1274 that is
connected to OR gate 1276 which in turn is connected to OR gate
1278 via line 1280. The sequence window signal is generated by the
circuitry of FIG. 10. An output line 1279 enables the gate 1264
only during the sequence window so that the true signal on output
lines 1286 and 1288 from the gate 1264 can only occur for a "101"
sequence detection during the presence of the sequence window. This
line 1286 is used to steer the divide-by-2 divider 1162 (FIG. 17b)
so that it is reset at the proper time to maintain 43 MHz clock
phase correct and to acquire bit synchronization. The other output
of the NAND gate 1264, i.e., line 1288, is applied to NAND gate
1290 which provides a signal on output line 1292 provided the other
input line 1294 has been enabled. Since the "101" sequence detector
is driven by a clock signal on line 1126 (via buffer 1166 and line
1164) which is obtained from the data stream itself, it is always
correctly phased with respect to the data stream. The detector will
always detect a "101" sequence if it is present provided it is
enabled and this occurs during the sequence window. The gate 1290
is enabled only when the occurrence of 20 successive zeros in the
bit stream is detected which legitimately occurs during the digital
synchronization "005" sequence and this occurs prior to the "101"
detection as would be expected.
To detect the occurrence of 20 successive zeros and referring to
FIG. 17b, a counter, indicated generally at 1296, examines the data
being shifted through the shift register, particularly, the data
appearing on the output of the flip-flop 1192 which operates to
reset the counter in the event that a logical 1 appears. The
counter 1296 is clocked by the 43 MHz clock on line 1298
originating from a buffer 1300. The counter provides an output
signal on line 1302 when 20 consecutive zeros have occurred and
this signal triggers a monostable multivibrator 1304 (FIG. 17a)
which provides a signal on line 1306 that is transmitted through
NAND gate 1308 in the event that the gate has been enabled by a
true signal on line 1310, which occurs during the occurrence of the
sequence window. If the NAND gate 1308 is enabled, then the
enabling signal is provided on line 1294 for enabling the gate
1290. The true signal on line 1292 therefore occurs in response to
the detection of the "101" sequence during the sequence window
which occurs during the horizontal blanking interval of every
processed television line and provides the word synchronization
signal on line 1292 that is applied to OR gate 1314 (FIG. 17b),
which has output line 1316 connected to the reset of the
divide-by-9 divider 1200. The output of the divider 1200 appears on
line 1318 which is connected to OR gate 1320 which has the effect
of resetting itself every 9 counts of the clock as well and,
thereby, adapt the four flip-flops forming the counter 1200 to a
divide-by-9 counter. The output line 1316 of the gate 1314 also
extends to the clock input of a monostable multivibrator 1322 which
has an output 1324 that clocks a divide-by-3 divider indicated
generally at 1326 which produces an output of 1.6 MHz decoder clock
on line 1328. Line 1324 carries a signal that is a 43 MHz clock
divided by 9, or 4.8 MHz, which extends through buffer 1330 and
produces a 4.8 MHz decoder clock signal on line 1332. The line 1324
is also coupled by the buffer 1334 having output line 1242 which
carries the 4.8 MHz clock which clocks the flip-flop 1238. The
lines 1328 and 1332 comprise the decoder clocks that are used to
clock the random access memories RAM 1 through RAM 4, as well as
the circuits 50 and 52 during the reproducing operation as
previously described.
The output of the divide-by-9 counter is also applied via line 1338
to a flywheel circuit, indicated generally at 1340, which is
operable to prevent any sudden step in the word synchronization and
is adapted to provide a recurring 4.8 MHz at its output on line
1342 for 30 to 40 cycles of word sync. The signal on line 1342 is
applied to a flip-flop 1344 that triggers a monostable
multivibrator 1346 via line 1348. The monostable multivibrator 1346
merely properly times the signal and has an output on line 1350
which is coupled to a differentiating circuit comprised of delays
1352 and 1354 and gate 1356 which produces a very narrow pulse on
line 1358. The pulse activates the gate 1360 during the sequence
window when line 1364 is active, which provides an output on line
1362 that will activate the OR gate 1314 for resetting the
divide-by-9 counter in the event the "101" sequence detector output
on line 1292 is not present for some reason, such as a drop-out or
the like. Thus, the divide-by-9 counter will be properly reset by
either the "101" sequence detector, or the flywheel reset circuitry
just described even if a clock pulse on line 1198 is temporarily
lost. An important effect of the circuit operation is to maintain
the sync word at a relatively constant rate over several tens of
cycles and not change it due to a loss of a clock count or for the
loss of a few occurrences of the "101" detection and the like.
In accordance with another aspect of the operation of the circuitry
shown in FIGS. 17a and 17b, each of the decoders is adapted to
provide the 86 MHz clock to the other, with the one shown in FIG.
17b providing the 86 MHz clock on line 139 and the present
illustrated decoder similarly receiving the 86 MHz clock from the
other decoder on line 141 shown at the lower left of FIG. 17a. This
is to compensate for a drop-out that may occur in the RF channel to
one of the decoders and, if such occurs, the clock from the other
channel can be used to maintain clocking of the circuitry so as to
retain the sync word timing. This allows a clock signal to be
maintained so that the clock from the subject channel can be
reacquired easily upon the reoccurrence of the signal after the
drop-out has ended. It should be appreciated that while the
detection of the occurrence of an RF drop-out provides an
indication of the absence of the clock signal, indications other
than the detection of the loss of the RF signal may be conveniently
used to cause the clock signal from the other channel to be
used.
The detected RF drop-out from the equalizer 118 is applied on line
1028 to a buffer 1370, the output of which is applied to a first
integrator stage, indicted generally at 1372, which is reclocked by
flip-flop 1374 that is clocked by line 1376 from the buffer 1172
providing the 86 MHz clock. The output of the flip-flop 1374
appears on line 1378 extending to one input of a gate 1390, which
has the other input supplied by line 1380 that originates from an
OR gate 1382. The input to the gate 1382 is supplied via buffer
1384 and a monostable multivibrator 1386 that is triggered by line
1388 which has an H/8 signal, i.e., the head switching signal, so
that a drop-out indication will not be generated during this time.
This signal prevents switching to the other channel clock during
the head switch caused drop-out. Either of the input lines 1378 and
1380 enables the OR gate 1390 and provides a signal on output line
1392 which extends to the output flip-flop 1238 to reset the same,
and thereby provide a drop-out indication on output line 146 and
for 148 which are used by the circuitry 52 and eventually the
drop-out compensator 160. The other output of the NAND gate 1390 is
applied via line 1394 to a second integrator indicated generally at
1396 which integrates the drop-out signals and thereby effectively
confirms the presence of an actual drop-out. The integrated signal
is in turn connected to a flip-flop 1398 that is connected to a
stretching circuit 1400. The stretching circuit 1400 has output
line 1402 that is connected to the reset terminals of a flip-flop
1414 which has output line 1416 that enables gate 1418 to pass the
86 Mhz signal from the other decoder for use in clocking the
present decoder circuitry. The stretching circuitry is effective to
hold the drop-out indication for a predetermined time beyond the
duration of the actual dropout so as to be sure that the RF signal
has fully returned and the 86 MHz clock from the present decoder
has been acquired before it is again used.
Thus, when the drop-out signal occurs, a delayed pulse appears on
line 1402 which resets the flip-flop 1414 and after the drop-out
terminates, a pulse appears on line 1404, although the latter is
extended by the stretching circuit 1400, and is applied to gate
1406 which provides an output signal on line 1408 that provides one
input to gate 1410, the other of which is supplied by line 1412.
The output line 1412 of the gate 1410 sets the flip-flop 1414 and
its output line 1416 then disables NAND gate 1418 so that the 86
MHz clock on the other input line 1420 can no longer be clocked
therethrough. However, before returning the operation of the
present decoder to the clock derived by the present decoder from
the data stream it receives, it is necessary to confirm that it is
bit synchronized, i.e., that the 43 MHz clock used to clock the
circuitry is properly synchronized to decode the logical ones in
the middle of a data cell. Since the 43 MHz clock is derived by
dividing the 86 MHz clock by two, the divider 1162 that performs
the division is reset at the proper time. This is accomplished by a
gate 1419 having input lines 1402 and 1416 being enabled for a time
period of about 6 to 12 words occurring between the time of the
actual termination of the RF drop-out and the termination of the
stretched drop-out and the gate provides a signal on line 1421
which is applied to gate 1278 producing a signal on line 1279 which
enables the "101" detector. When this is done, the occurrence of
any "101" sequence in active video or in the synchronizing sequence
will provide a reset pulse on line 1286 that resets the flip-flop
1162 and properly synchronizes the 43 MHz clock. The 43 MHz clock
on line 1420 originates from a divide-by-2 divider 1422 that is
clocked the 86 MHz clock on line 1424 from a buffer 1426 that has
its input supplied by line 149 carrying the 86 MHz clock from the
other decoder. When the line 1416 enables the gate 1418, the 43 MHz
clock appears on an output line 1430 which extends to the clock
input of the divide-by-9 divider 1200 and therefore supplies the
clock in place of that which had been supplied on line 1198 but
which is not present due to the drop-out on the channel having the
data on line 132. The divide-by-2 divider 1422 is essentially reset
by line 1432 that is clocked by the divide-by9 divider 1200 which
effectively switches the clock from the other decoder into the
subject decoder at the proper time with respect to the operation of
the divider. Thus, through the above described operation, each
decoder effectively acquires the clock frequency from the Miller
squared encoded data during normal operation and also receives and
uses the acquired clock from the other decoder in the event of a
drop-out occurring in the subject channel, thereby insuring that
the basic word synchronization is maintained during drop-out.
The control of the operation of the random access memories RAM 1
through RAM 4 shown in the block diagram of FIG. 1 is accomplished
by the clock generator and switcher circuitry 196 and logic
circuitry 200, the detailed circuitry of which is shown in FIGS. 7,
8, 9 and 10.
Turning initially to the logic and clock circuitry of the memory
control circuitry as shown in FIG. 9, this portion of the circuitry
is adapted to apply the appropriate clocks to the memories RAM 1
through RAM 4 depending upon whether a recording or reproducing
operation is occurring. Thus, from external switches controlled by
an operator, four input lines 1450, 1452, 1454 and 1456 are
operable to place the apparatus in one of four modes, i.e., the
play mode, the record mode, an EE mode and a test mode. During the
EE operation, the data is merely written into the memories the
record mode, an EE mode and a test mode. During the EE operation,
the data is merely written into the memories and thereafter read
out of them using the same clock, bypassing the actual recording
and reproducing operations, which essentially provides a test of
this part of the circuitry. These four lines together with a test
select line 1458, which selects either one pair of interconnected
random access memories, i.e., RAM 1 and RAM 3 or the other set,
i.e., RAM 2 and RAM 4, together with an even or odd level from a
programmable read only memory 1600 (FIG. 7) on line 1460, which is
used during testing modes, is applied through various logic
circuitry to provide the appropriate signals and clocks for use in
controlling the memories. The level of the signal provided on line
1460 during normal record and reproduce operating modes is selected
to enable the memory control circuitry to function as described
herein to provide the needed memory control signals.
The 1.6 MHz clock from the decoder 138 or 140 is applied to the
circuitry on line 1328 and this clock is used to write the data
into the memory during a reproducing process. The clock on line
1328 is changed from a MECL level to TTL level by the converter
1462 and is applied to successive monostable multivibrators 1464
and 1466 which adjust the phase of the clock. The monostable
multivibrator 1464 has an output line 1468 labeled IDENT clock
which extends to the identification processing circuitry of the
memory control circuitry shown in FIG. 10. The output of the
monostable multivibrator 1466 is applied through line 1470 to AND
gate 1472 that is enabled by line 1474 which is high during a
reproducing or play operation. Line 1474 also enables gate 1476
which has the reference 3.58 MHz clock on the other input thereof
for use in reading the data from memory during reproducing.
Similarly, AND gate 1478 is enabled during a recording process via
line 1480 and the record 3.58 MHz clock signal will be gated
through the gate 1478 for use in writing data into the memory
during a record process.
The 1.6 MHz clock from the encoder 82 appears on line 916 which is
similarly converted from MECL level to TTL level by a converter
1482 is retimed by two monostable multivibrators 1484 and provides
the properly phased 1.6 MHz clock on line 1486 that is used to read
data from memory during a recording operation, except in an EE mode
where a 3.58 MHz clock on line 1488 is utilized. Gates 1490, 1492
and 1494 effectively gate either of these clock frequencies onto
line 1496 that is applied to gate 1498 that is enabled during a
recording operation. Thus, the AND gates 1472 and 1498 effectively
select either a 1.6 MHz clock from the two sources and utilizes the
decoder 1.6 MHz clock for writing the off tape data into the
memories during reproducing or the encoder 1.6 MHz clock for
reading the data from memories during a recording operation. One of
these clocks is applied on line 1500 which is steered through
logic, indicated generally at 1502, and supplies the clocks on
lines 822 to the memories. It should be appreciated that the
circuitry shown in FIG. 9 is duplicated and for one of the circuits
the line 822 would supply the clock for memory RAM 1 and the
duplicate thereof would supply the clock for the memory RAM 2.
Similarly, the other line 822 for one of the circuits would supply
the clock for memory RAM 3 while the duplicate thereof would supply
the clock for memory RAM 4. Other similar designations 1/2 and 3/4
in other drawings reflect similar usage. The gates 1476 and 1478
effectively select either the record or the reference 3.58 MHz
clock for application to line 1508 and is gated through the
steering logic 1502 to supply these frequency clocks on lines 822
when they are required. In this regard, the reference 3.58 MHz
clock would be used to read the data from the memories in a
reproducing process and the record 3.58 MHz clock would be used to
write the data into the memories during a recording operation. The
steering logic 1502 is also controlled by additional steering
logic, indicated generally at 1510, together with inverters 1512.
The inputs to the logic 1510 are supplied by the lines 1474 and
1480, which reflect whether the apparatus is in a record or
reproduce mode, together with write enable signals on lines 1514,
1516, 1518 and 1520. The write enable signals on lines 1514 and
1518 are supplied by a read only memory 1600 (FIG. 7) that is
programmed to supply the appropriate write enable signals during a
recording operation and the signals on lines 1516 and 1520 are
provided by another read only memory 1816 (FIG. 8) that is
programmed to supply the write enable signals during a reproducing
operation. Thus, the steering logic 1510 and 1502 together with the
inverters 1512 select the proper clocks at the proper time for
carrying out the writing and reading of the random access memories
RAM 1 through RAM 4 during the recording and reproducing operations
in the manner that has been described with respect to the timing
diagrams shown in FIGS. 4b and 5b. The write enable lines 1514
through 1520 are also applied to a 2-to-1 switch 1522 which has
memory select inputs on lines 1524, 1528 and 1530 that are supplied
by the same read only memories (1600 and 1816) that supply the
write enable signals. Lines 1524 and 1528 are used to supply the
memory select signals during a recording operation, while lines
1526 and 1530 supply the memory select signals during a reproducing
operation. A signal on line 1474 controls the switch 1522 and
effectively selects the appropriate write enable and memory select
lines during a recording and reproducing operation and provides the
signals on output lines 806 and 808 that are connected to the
memory circuitry show in FIG. 13. It should be appreciated that
only one memory has been shown in FIG. 13 and that one of the RAM
select lines 806 as well as one of the write enable lines 808 shown
in FIG. 13 would be connected to either the line 806a or b, of the
circuitry shown in FIG. 9, depending upon the identity of the
representative memory shown in FIG. 13 as should be readily
understood.
Other signals that are produced by the circuitry shown in FIG. 9
are provided on lines 1534, 1536, 1538 and 1540 which indicate that
the EE, test, play and record modes are in process and these
signals are applied to other of the memory control circuit for
control thereof as will be hereinafter described. Similarly, a head
switch control signal is provided on line 1542 that is high during
a reproducing operation and in a similar manner, a record current
signal on line 1544 is used by other of the memory control circuit
and it is high during a recording operation. The line 586 is used
to control the 8-to-24 bit converters 50 and 52 and is high during
a reproducing operation and controls the selection of either the
1.6 MHz or 3.58 MHz clock for clocking the data through the
converter. Similarly, a control line 1546 is used to switch the
encoder on or off by controlling a relay that turns on the 86 MHz
oscillator portion of the encoder during a recording operation and
disables it during reproducing. The circuitry also provides a
signal on line 1550 for controlling the operation of the 2-to-1
switch 152 to select the output of the proper pair of the random
access memories during reproducing as well as during the EE mode.
Since the switching of the 2-to-1 switch occurs at a line-by-line
rate, a H/2 signal that is synchronized to the record clocks is
applied on line 1552 to a D flip-flop 1554 that is clocked by a H
rate clock on line 1556 that is synchronized to the record clocks
and is phase coherent with the 3.58 record clock. The H/2 rate
signal on line 1550 for controlling the 2-to-1 switch is used
during reproducing and is supplied by a D flip-flop 1558 that has a
H/2 signal applied on line 1560 by the address generator 1882 (FIG.
8) and is clocked by line 1562 from the monostable multivibrator
1780 (FIG. 8).
To control the memories during the recording operation, circuitry
shown in FIG. 7 provides the proper write enable and memory select
signals for controlling the memories in accordance with the timing
diagram shown in FIG. 4b and also provides signals for controlling
the record current for the transducing heads for recording the
signals on tape. In contrast to head switch that is done in the
reproducing process, record current is applied to the transducing
heads and effectively enables them to record the data on tape. As
previously mentioned, the record current is sequentially applied to
the eight heads in numerical order as they are identified in FIG.
2. Each head records eight video lines per pass across the tape and
two heads are always simultaneously recording. Since the heads are
equally spaced around the circumference of the head wheel, when
head number 1 is half way across the tape, record current will be
applied to head number 2. As the head wheel continues to rotate,
record current will be applied to head current 3 at the time the
record current is removed from head 1.
With respect to the circuitry shown in FIG. 7, the record clock
frequency of 3.58 MHz is applied on input line 238 which is used to
clock a counter 1570 that operates with selecting circuits 1572 and
1574 to provide a load signal on line 1576 which loads a
preselected number so that the counter operates as a 25 cycle
counter which corresponds to the amount of delay that is desired in
the horizontal blanking interval before writing of the digital
synchronization sequence begins. A horizontal sync signal on line
385 from the 455 counter and PROM 380 (FIG. 12) is applied to a
monostable multivibrator 1578 which properly times the H sync
signal so as to provide an output on line 1580 which clears the
counter at the proper time, i.e., at the start of the blanking
interval. The selector 1574 has output line 1582 which is applied
to a flip-flop 1584 at the terminal count of 25 and provides a
pulse on output line 1586 that is properly positioned by monostable
multivibrators 1588 and 1590, the latter of which has output line
1592 that is sent through steering logic indicated generally at
1594 that provides the write reset pulses on lines 830 for
resetting the appropriate one of the memories RAM 1 through RAM 4.
Read reset pulses are also generated by the steering logic 1594.
The 455 counter and PROM 380 (FIG. 12) supplies a 7.5 KHz odd/even
line identifying signal on line 384 that is inverted and provided
to one input of a NAND gate 1571. The second input of the NAND 1571
receives an enabling signal from the D flip-flop 1608 in response
to the output from the selector 1574 on line 1610 at the end of the
aforementioned 25 SC cycle interval. The NAND gate 1571 provides a
pulse on its output 1573, which is coupled by a series of
monostable multivibrators 1575, indicated generally at 1575, to one
input of each of the NAND gates 1577 and 1579. The other inputs of
the NAND gates are supplied by the address line 1581 from the
address counter 1636. This address line is high when memories RAM 1
and RAM 2 are selected for reading and is low when memories RAM 3
and RAM 4 are selected for reading. Thus, the NAND gates 1577 and
1579 are selectively gated by the memory select signal on line 1581
to pass the H/2 rate pulses received from the NAND gate 1571 to the
steering logic 1594, which responsively provides the read reset
pulses to the memory selected for reading. In this regard, the
entire circuitry shown in FIG. 7 is duplicated and the output
designations 1/2 correspond to the same uses as was described with
respect to the circuitry shown in FIG. 9.
To provide the write enable and memory select signals, a
programmable read only memory 1600 is provided and it has four
output lines 1602, each of which is applied to a D flip-flop 1604
which is clocked by a line 1606 having a horizontal rate clock
thereon and the outputs of the D flip-flops 1604 provide the write
enable and memory select signals as shown. The clock line 1606
originates from the flip-flop 1608 that is clocked by the 3.58 MHz
clock but which has its D input supplied by line 1610 that occurs
at a horizontal rate. The signals for providing the record current
are also generated by a programmable read only memory 1612 which
has output lines 1614 that are clocked by flip-flops 1616 and
provide signals on lines 1618 that are gated through gates 1620
onto lines 1622 which are connected to one input of NAND gates 1624
which are enabled by line 1544 when a recording operation is in
progress. Thus, the outputs of these gates appear on lines 1626
which extend to the various record current sources associated with
the appropriate transducing head.
The read only memories 1600 and 1612 are addressed by address lines
1630, line 1552, the EE mode control line 1534, and line 1632 which
is alternately low and high for odd and even numbered video lines.
The line 1632 is low for one of the duplicate sets of circuitry
shown in FIG. 7, i.e., the circuit that controls memories RAM 1 and
RAM 3 and the duplicate circuitry has this line high since it
controls the memories RAM 2 and RAM 4. The other addresses are
controlled by the operation of an address counter 1636 which
generates signals on output lines 1630 for accessing the proper
information for generating the appropriate memory select, write
enable and record current control signals in accordance with the
timing diagram shown in FIG. 4b. The address controller 1636 is
essentially a 5 bit or 32 cycle counter which is cleared by a
signal placed on line 1638 by the output of a monostable
multivibrator 1640. The monostable multivibrator 1640 is triggered
by a signal on line 1643 that is connected to a servo control
circuit (FIG. 28), which provides a processed H/64 tach reset pulse
for every rotation of the head wheel, it being realized that for
each rotation of the head wheel there will be 64 lines of data
recorded on tape. By synchronizing the counter 1636 with this head
wheel, the proper head will have record current applied at the
proper time.
To control the operation of the random access memory RAM 1 through
RAM 4 during reproducing, in addition to the circuitry described in
FIGS. 7 and 9, circuitry particularly adapted for use in
controlling this aspect of the operation of the memories is shown
in FIGS. 8 and 10. As has been previously mentioned, the digital
synchronization sequence that is added prior to every video line
includes the ID 1 and ID 2 numbers which are used during the
reproducing process to properly time the operation of the memory
with respect of the data that is to be written in the memories. As
recalled from the discussion of the synchronization sequence adder,
each of the numbers ID 1 and ID 2 is written three times in
succession within each cycle of subcrrier and the circuitry of FIG.
10 is adapted to process the ID 1 and ID 2 numbers that are decoded
by the identification number decoders contained within the 8-to-24
bit converter circuits 50 and 52, to insure that they are valid.
Since the identification numbers effectively determine the
horizontal sync position during reproducing, it is important that
they be reliable or the resulting picture would be horizontally
displaced for those lines in which the identification information
may be bad. In this regard, the ID 1 and ID 2 signals are applied
via lines 634 and 636, respectively, together with a composite
drop-out signal on line 682 which will enable NAND gates 1640 and
1642, if there has not been a detected composite drop-out, so that
the three successive ID 1 and ID 2 pulses will be gated through the
respective gates onto lines 1644 and 1646, respectively. Each of
the lines 1644 and 1646 is applied to integrators, indicated
generally at 1648 and 1650, which are operable to integrate the
pulses and provide an output on lines 1652 and 1654, respectively,
if two out of three of the three successive identification pulses
occur. The lines 1652 and 1654 are applied to flip-flops 1656 and
1658 which are clocked by clock line 1660, which is obtained from a
1.6 MHz clock received on line 1468 that is derived from the
reproduced data by the decoder and retimed by the memory control
logic and clock circuitry illustrated in FIG. 9. The 1.6 MHz clock
is derived from the reproduced data to be coherent with the data.
The identification pulses are therefore reclocked by this clock
signal and appear on lines 1662 and 1664. The 1.6 MHz clock on line
1468 is applied to two monostable multivibrators 1668 and 1670 for
timing the clock signal and the output of the monostable
multivibrator 1668 is applied to a second retiming monostable
multivibrator 1672 that supplies a 1.6 MHz clock on line 1674 and
is used to clock a 202 count counter. The output of the monostable
multivibrator 1670 on line 1660 is also applied to a divide-by-2
divider 1676 having output lines 1680 and 1678 which, respectively,
extend through inverters to the integrators 1648 and 1650. More
specifically, line 1680 is applied through an inverter 1682 to line
1684 and is also applied to an inverter 1686 which is connected to
line 1688 that is connected to the intergrator 1650. Similarly,
line 1678 is applied to line 1690 via inverter 1692, as well as to
line 1694 via inverter 1696.
With respect to the operation of the integrator 1648, which is
substantially identical to the operation of the integrator 1650,
the ID 1 pulses on line 1644 are applied through inverters 1700 and
1702 which provide separate parallel paths on lines 1704 and 1706
which are respectively connected to capacitors 1708 and 1710. As
previously mentioned, the presence of any two of three successive
pulses will provide an output from one of two voltage comparators
1712 and 1714 is such occurs. The divide-by-2 divider 1676
alternately changes the level on lines 1690 and 1684 to alternately
discharge the capacitors 1708 and 1710, thereby permitting one of
the capacitors to be charging during the presence of the set of
three ID pulses while the other is discharging. During the presence
of the next sets of ID 1 pulses, the other capacitor is charged
while the first is being discharged. If any two of the three
successive ID 1 pulses are present, then the appropriate one of the
voltage comparators 1712 and 1714 will provide an output level on
line 1652 that confirms the presence of the identification ID 1
pulses. The integrator 1650 operates in the same manner with
respect to detecting the ID 2 pulses.
The reclocking flip-flops 1656 and 1658, which reclock the detected
ID 1 and ID 2 pulses, also have output lines 1720 and 1722, both of
which are applied to and NAND gate 1724 which provides a signal on
line 1726 that indicates the presence of detected ID 1 and ID 2
pulses. This signal is sent to the 8-to-24 bit converter and 2-to-1
switch circuitry 50 to 52 which, when not present, has the effect
of commanding that circuit to provide a signal on the parity
channels which will cause the drop-out compensators to insert a
whole line of information rather than using the data stream data,
for the reason that the absence of the detection of the
identification pulses indicates that the horizontal timing may be
incorrect and the entire line may be horizontally displaced which
would disrupt the video image.
Lines 1720 and 1722 also extend to an integrator, indicated
generally at 1732, which detects whether the signals from each of
the channels is inverted or not and provides a signal on line 142
which is low when they are correct. This line effectively controls
the operation of the switches 128 and 130 shown in the block
diagram of FIG. 1. A H/2 play signal is applied on line 1560 by the
address generator 1882 (FIG. 8) which triggers a monostable
multivibrator 1740 which has an output line 1742 that is connected
to a flip-flop 1744 having an output connected to a positioning
monostable multivibrator 1746 that is triggered by an H rate pulse
provided on line 1750 by the monostable multivibrator 1776 (FIG.
8). The output of the monostable multivibrator 1746 is applied to
another monostable multivibrator 1752 which provides an output of
proper duration on line 1754 which is gated through gate 1756 to
provide a flywheel window signal on line 1758 that is used by the
reproducing memory control circuitry shown in FIG. 8.
The operation of the monostable multivibrator 1740 also clocks a
flip-flop 1760 which triggers a monostable multivibrator 1762 and
provides the sequence window signal on line 1270 that is applied to
the decoders 138 and 140 which are used to decode the occurrence of
the synchronization sequence during reproducing as has been
described.
Turning now to the circuitry shown in FIG. 8, it generates the
memory select and write enable signals for operating the random
access memories RAM 1 through RAM 4 during the reproducing
operation, as well as supplies the ID 1 and ID 2 pulses to the
memories. It also generates head switching signals for switching
between the outputs of the preamplifiers so as to apply the proper
output to the equalizers. A reference 3.58 MHz clock signal is
applied on input line 190 which is used to clock a counter 1772
that is loaded by a signal on line 1750 that originates from a
monostable multivibrator 1776 that is triggered by a station
reference H rate signal on line 1777. The output of the counter
appears on line 1778 that is supplied to a monostable multivibrator
1780 which has an output line 1782 that is gated through NAND gates
1784 and 1786 to provide the read address signal for the RAM
address circuitry on lines 838. The NAND gates 1784 are enabled via
line 1538 when the apparatus is in the play or reproducing
operational mode and the signal is alternately gated through gates
1784 and 1786 by lines 1526 and 1530 so as to apply the read pulse
to either memory RAM 1 or RAM 3. In this regard, the circuitry
shown in FIG. 8 is also duplicated and the duplicate circuitry
would control memories RAM 2 and RAM 4. The counter 1772 merely
delays the occurrence of the H rate pulse on line 1778 for the
proper time to have the memory set in its proper position to read
the data therefrom in accordance with the timing diagram shown in
FIG. 5b. The ID 1 and ID 2 pulses are applied to the NAND gates
1790 and 1792 via lines 1664 and 1662, respectively, and the gates
are enabled during reproducing by a signal on line 1538. The
circuitry, indicated generally at 1794, merely provides much
narrower ID pulses than were present on input lines 1664 and 1662
and these pulses are applied via line 1796 and 1798 to steering
logic comprising gates 1800, 1802, 1804, 1806 and 1808, together
with inverter 1810. The outputs of the gates 1802 through 1808
provide the identification pulses on output pulses 832 and 834 as
shown. The NAND gates 1802 through 1808 are enabled by signals on
lines 1812 and 1814 which are two of the outputs from a read only
memory 1816 which controls the steering of the identification
pulses to the proper memory RAM 1 or RAM 3 or, in the case of the
duplicate circuitry, the memory RAM 2 or RAM 4.
As previously explained with respect to the timing diagram of FIG.
6, there are 202 twenty-four bit and twenty-seven bit words that
are written in and read from memory during recording and
reproducing, respectively, and the 202 cycles represent 190 cycles
of active video information together with 12 cycles containing the
digital synchronization sequence. When the data is to be written
into memory during reproducing, it is written using a 1.6 MHz
clock. The 1.6 MHz clock is applied to the circuitry of FIG. 8 via
line 1674 extending from the memory control circuitry illustrated
in FIG. 10 and is used to clock a divide-by-202 divider 1820 which
functions as a 202 cycle counter. At the terminal count of 201 (0
through 201 equals 202 cycles), the four output lines 1822 from the
divider are applied to a number 201 decoder 1824, which provides a
signal on line 1826 to a flip-flop 1828 that is clocked using the
1.6 MHz clock. The output of the flip-flop 1828 is applied to
another flip-flop 1830 via line 1832 and has its Q output on line
1834 connected to a NAND gate 1836, the other input of which is
applied by line 1838 from the flip-flop 1828. The gate 1836
produces a clear pulse on line 1840 that clears the counter 1820.
The presence of the ID 1 signal is effective to load the counter
1820 via line 1842 with the number 9 and the presence of ID 2 has
the effect of loading the counter with the number 11 via line 1844.
This has the effect of disregarding writing the digital
synchronization sequence into memory during reproducing since it is
no longer needed for any further processing and the ID pulses
effectively synchronize the 202 counter to the data that is
present. However, in the event that the ID pulses are missing, then
the 202 counter will continue to run through its 202 cycles and two
of the output lines of the counter are applied to the monostable
multivibrators 1846 and 1848, which have their outputs connected to
a NAND gate 1850. The NAND gate 1850 decodes the number 8 and
provides a signal on line 1852 which is gated through a flip-flop
1854 if the flywheel window signal on line 1758 is present at that
time. If it is, then a signal is provided on line 1856, which is
clocked through another flip-flop 1858, provided line 1538 is high,
which occurs when the apparatus is in the reproduce operation. The
signal at the output of the flip-flop 1858 is passed through gate
1860 so as to provide a "flywheel" ID 1 signal on line 1862 which
extends to the NAND gate 1800 and will provide the ID 1 signal to
the memory. This has the effect of filling in the ID 1 when it is
not present from the off-tape information.
The read only memory 1816 has output lines 1864 and 1866 in
addition to output lines 1812 and 1814 and these four output lines
are clocked through D flip-flops 1868 at the H rate to provide the
memory select and write enable signals on lines 1526, 1516, 1530
and 1520 that are used to control the memories during a reproducing
operation. In addition to the programmable read only memory 1816,
another read only memory 1870 is provided and has output lines 1872
which are clocked through D flip-flops 1874 and onto line 1876
which extend to one input of NAND gates 1878 that are enabled by
line 1542 when the apparatus is in the reproducing operational
mode. The signals are then gated onto the output lines 974 and 976
for switching the outputs of the preamplifier into the appropriate
equalizer. The addressing of the read only memories 1816 and 1870
are provided by address lines 1880 which, together with line 1460
access the information of these read only memories. The address
signals on lines 1880 are provided by an address generator 1882
that is essentially a 64 cycle counter that is clocked at H rate by
line 1886 from the counter 1772 and is cleared by a signal on line
1888 that is the output of a NAND gate 1890. The record/reproducing
steering signal on line 1642 originates from the servo control
board and occurs as a single pulse for each rotation of the head
wheel or at a 64 line rate. The signal on line 1643 provided by the
servo control circuit (FIG. 28) is applied to a gate 1892 which is
enabled during a reproducing operation and applies the signal on
line 1894 which extends to the gate 1890 and has the effect of
synchronizing the address counter to the rotation of the head wheel
so that the proper head switching occurs during operation. One of
the address lines 1880 provides the H/2 play signal and is
specifically identified as line 1560.
During reproducing, the data that is read from the memories is
applied to the 2-to-1 switch 152, a portion of which is shown in
detail in FIG. 21. The lines 150 and 154 are applied to the 2-to-1
switch 152 and if the even lines are to be applied to the output
lines 156, the control line 1550 (from FIG. 9) is high which
selects the signals from lines 154 and when the signal on line 1550
is low, the switch selects the signals from line 150. As is evident
from the drawing, only eight of the total 27 lines have been
specifically illustrated.
Turning now to one specific embodiment of the dropout compensator
160 that has been described with respect to the block diagram of
FIG. 1 for the entire system, reference is made to FIG. 23 which
illustrates a block diagram of the drop-out compensator 160
together with a downstream 2-to-1 data selector switch 162. As
shown in FIG. 23, there are 24 bits of parallel data on lines 156
which are applied to a memory 1900 as well as to a 21/2 cycle (of
the 3.58 MHz clock) delay circuit 1902 that effectively delays the
application of the data to the 2-to-1 switch 162 via lines 1904 for
the purpose of compensating for internal delays that are inherent
in the operation of the memory 1900. The information indicating the
existence of a drop-out is also applied via the three parallel
lines 156 to a similar 21/2 cycle delay circuit 1096 and to a
select control circuit 1908 that is operable to select either the
video data interval received over line 1904 or the output of the
memory 1900 appearing on lines 1910. The select control circuit
1908 controls the 2-to-1 switch 162 via line 1909, passes the data
from the memory 1900 whenever a drop-out or parity error occurs and
provides the data that occurred 262 lines or a multiple thereof
prior to the data in which the drop-out is indicated, so that
erroneous active video data will not be passed through the 2-to-1
switch 162 onto the output lines 1911. The output lines 1911 are
applied to a latch 1912 that is clocked by a 3.58 MHz clock signal
on line 1914 that is provided by an output monostable multivibrator
1916 that properly positions the output data. This clock signal is
obtained from line 1918 which is provided by a monostable
multivibrator 1920 that properly positions a 3.58 MHz clock signal
on line 1922, which is synchronized with the subcarrier and
provided by the clock generator circuitry 196. The output of the
delay circuit 1906 is provided on line 1924 that extends to the
select control circuitry 1908 for the purpose of providing the
appropriate command to the 2-to-1 switch and the select control
circuitry 1908 has an output line 1926 that extends to the memory
1900 and precludes it from writing bad data therein whenever a
drop-out or parity error is present. The lines 1924 are also
applied to a latch 1928 that is clocked by the 3.58 MHz clock
signal on line 1914 and provides an output on line 1930 that may be
used for other circuitry not shown.
The drop-out compensator shown and described herein has the
advantage of a recirculating compensator in the sense that the data
that is stored in the memory 1900 represents only nondefective data
and therefore only nondefective data is available to be read and
applied to the output lines 166. During operation, if a drop-out or
parity error is detected, the memory is inhibited from writing the
defective data at that time. If another drop-out or parity error
occurs 262 lines later, the memory will again be inhibited from
writing and will read the data that occurred and was written 524
lines previously, i.e., a multiple of 262 lines previously. As soon
as nondefective data is present for the memory address locations
corresponding to where writing had been inhibited it will, of
course, be written into the memory 1900.
The 21/2 cycle delay circuits 1902 and 1906 compensate for the
inherent delay of 21/2 cycles that is provided by the particular
memory circuitry 1900, which effectively reads the video data and
then immediately writes data. During operation of the memory,
reading continually occurs even though a drop-out is present which
would preclude writing data therein during operation. Even though
writing is inhibited during the presence of a drop-out, the memory
1900 is operated in a manner whereby reading occurs after the
inhibited write cycle. Reading from the memory 1900 occurs 21/2
cycles subsequently of any writing operation and it is for this
reason that the 21/2 cycle delay is interposed in the data lines
156 containing the video data. The select control circuitry 1908 is
also adapted to inhibit writing of the memory when an operator
controlled field bypass line 1932 from the drop-out compensator
memory control is active, as well as when a switch inhibit line
1934, also from the drop-out compensator memory control, is active.
The switch inhibit line inhibits writing into the drop-out
compensator memory during the vertical blanking interval as well as
during the horizontal blanking interval because there is no active
video information during these times, and the capacity of the
memory can be accordingly decreased. It should be appreciated that
the drop-out compensator is intended to insert data from the
previous field in the event that the active video data is either
missing or is incorrect and the purpose of the compensator is to
correct the video image and has no purpose with respect to the
horizontal and vertical synchronization signals. Accordingly, the
switch inhibit line 1934 effectively disables writing into the
memory 1900 during the horizontal and vertical intervals.
One embodiment of specific circuitry that can be used to carry out
the operation of the block diagram of FIG. 23 is shown in FIGS.
26a, 26b, 27a and 27b in conjunction with timing diagrams shown in
FIG. 24. The circuits illustrated in those figures received various
control signal inputs from the drop-out compensator memory control
illustrated in FIGS. 25a and 25b, which will be described in detail
hereinbelow. Turning initially to the data switching portion of the
circuitry shown in FIGS. 27a and 27b, the 24 video data lines 156
are applied to the 21/2 cycle delay circuits 1902 which comprise
four flip-flops for each line that is in a single package and
function as a shift register with the output of each shift register
being applied to the 2-to-1 switch 162 via lines 1904. Similarly,
the 24 data lines 1910 from the memory are applied directly to the
2-to-1 switch 162 as shown. Referring to FIG. 27b, the switch
inhibit line 1934 is applied to an AND gate 1940 which has output
line 1909 for controlling the operation of the 2-to-1 switch 162.
Similarly, the operator controlled field bypass line 1932 is
applied to an AND gate 1942 that has output line 1944 connected
through an inverter 146 to line 1948 that extends to the AND gate
1940. The framing line 1950 from the servo (FIG. 28) is also
connected to the AND gate 1942 and effectively inhibits the
insertion of data from the drop-out compensator when the servo
system is attempting to properly frame the tape and the magnetic
transducing heads are crossing tracks during the active video. In
this regard, when line 1909 is low, the data from lines 1910 is
selected by the 2-to-1 switch 162 and when it is high, the data
from lines 1904 is selected.
Turning now to the random access memory associated with the
drop-out compensator 160, it should be understood that the
particular embodiment shown in the block diagram of FIG. 23
represents essentially a 262 line delay and one embodiment of a
memory that can be used therein is shown in FIGS. 26a and 26b which
together comprise a single electrical schematic diagram. The
circuitry for operating the memory shown in FIGS. 26a and 26b is
shown in FIGS. 25a and 25b and will be subsequently described. It
should be understood that the particular memory shown in the
drawings is representative and can be replaced by other memory
devices that could operate faster and have greater capacity so as
to reduce or eliminate much of the complexity and timing
requirements that are present and which will be described. In the
memory illustrated in FIGS. 26a and 26b, there are 72 separate
integrated circuits, each having a capacity of 4,096 bits and, as
is well known, integrated circuits have significantly greater
capacity are now available, which would simplify many of the
switching and control circuitry that will be described herein. In
this regard, the memory 1900 has a total capacity of about 295,000
bits and the detailed circuitry shown in FIGS. 26a and 26b
represents only 1/4 of the total. As previously mentioned, there
are 24 data lines and the circuitry shown in FIGS. 26a and 26b is
operable to provide memory for data for 6 of the 24 lines. The
operating speed of the memory is less than the 3.58 MHz data rate,
which requires grouping of the data into data words that are
operated upon by the slower memory integrated circuits. The data
words are sequentially applied to latches and then presented to the
memories in groups of four words so that the memories operate on
the data at about 1/4 the data rate of 3.58 MHz, which is
compatible, with their speed capability.
More specifically, with respect to the circuitry shown in FIGS. 26a
and 26b, six of the 24 data lines 156 are applied to four
integrated circuit latches 1956, which are flip-flops that operate
to latch the data therein for subsequent processing by the memory
1900. A data selector 1958 is adapted to control the latching of
the data into the appropriate one of the latches 1956 at the
appropriate time which is controlled by two byte select lines 1960
together with a data input strobe signal on line 1962. The two byte
select lines 1960 control the selector 1958 so that it selectively
activates one of the four output lines 1964 to strobe the data into
one of the latches 1956. During operation, the data on lines 156
occurs at the 3.58 MHz data rate and the byte select control lines
1960 are activated at the 3.58 MHz rate so as to sequentially latch
the six bits of data for four consecutive words into the four
latches 1956 so that after four cycles of the 3.58 MHz clock, 24
bits are loaded into the latches 1956 for subsequent writing into
the memory 1900. As is shown in the drawings, the memory 1900
comprises 72 individual integrated circuits 1966, each of which
provides 4,096 bits of randon addressable memory with the 72
integrated circuits being grouped into three groups of 24
integrated circuits arranged in vertical rows as shown. Each of the
output lines, such as the line 1968 from each of the latches 1956
extends to three of the memories 1966 so that depending on which
group is activated, the data on the line 1968 can be selectively
written into any one of the memories 1966 of the three respective
groups. Similarly, output lines 1970 from the individual memories
are interconnected and extend to respective output latches 1972
shown in FIG. 26b. Thus, depending upon which group of memories
1966 are read, the read data appears on line 1970 which is latched
into the latches 1972 when the signal on a data output strobe line
1974 is true. The outputs of the latches 1972 appear on lines 1976
which extend to 4-to-1 data selector switches 1978 that are
controlled by output byte select lines 1980 to assert the data from
one of four possible lines 1976 to the corresponding output line
1910. The output byte select lines 1980 are switched at the 3.58
MHz rate so that the six output lines 1910 are provided with data
at the same rate as the data is applied at the input on lines 156,
even though the actual processing of the data through the memory
occurs at a rate that is 1/4 the input and output data rate.
Each of the individual random access memories 1966 has six address
lines 1986 as well as a write enable line 1988, a group select line
1990, a row address strobe line 1992 and the column address strobe
line 1994. The addresses are applied to the address lines 1986 in
two steps, i.e., row address signals are applied to the six address
lines, followed by column address signals on the same lines. The
row is addressed when the row address strobe 1992 is applied and
the column is addressed when the column address strobe signal is
applied to line 1994. Thus, the group 1, group 2 or group 3
memories 1966 are written into or read from when the group select
lines 1990 for the appropriate group are true. Circuitry is also
shown in the lower portion in FIGS. 26a and 26b for controlling the
memories 1966. Group select lines 1996 are applied to a selector
circuit 1998 having three output lines 2000, any one of which is
active at one time for selecting one of the groups of the memories
1966. The lines 2000 also supply one input of NAND gates 2002, the
other input of which is provided by lines 2004 which respectively
control the refreshing of the respective groups of memories with
the output of the gates 2002 being applied to NAND gates 2006, the
other input of which is supplied by line 2008 which has the read
address strobe signal. The output of the gates 2006 provides the
row address strobe signal on line 2010 that occurs for only one
group at any one time. A column address strobe signal on line 2012
provides column address strobe signals on lines 2014 that occur
simultaneously for each group. Similarly, a write enable command on
line 2016 provides write enable commands on lines 2018 that are
applied to each group of memories. Because of the manner in which
the internal circuitry of the memories operate, only the row
address strobe signal needs to be selectively applied so that only
one group of memories is selected. After one group has received the
row address strobe signal, column address strobe and write enable
commands for the nonselected groups are ineffective to cause them
to operate. Turning to FIG. 26b, address lines 2020 are connected
to simultaneously provide address signals on line 2022, 2024 and
2026 which extend to the three groups of memories 1966.
Prior to describing the circuitry of FIGS. 25a and 25b which
generate the input signals to the memory circuitry shown in FIGS.
26a and 26b, reference is made to the timing diagrams of FIG. 24
which illustrate the timing sequences for writing and reading data
into and out of the memories.
It should be understood that the entire data from each video field
is not written into the memory because of several reasons, one of
which is that writing all of the information necessarily includes
data that is not useful in correcting the active video and thereby
represents a waste of memory capacity. Moreover, it is not desired
to compensate for drop-outs of data that are used by the servo
mechanisms, since this can easily cause more problems than are
solved and fly wheel circuits and the like which have been
previously described are adequate to control the servo operation.
Accordingly, it is only desired to write data for the active video
information and consequently, the data that occurs during the
vertical interval of about 20 lines is not written into memory, nor
is any data written into memory during substantial portions of the
horizontal interval. Thus, data for 196 cycles of subcarrier for
each active video line comprises the totality of the data that is
written into memory, this amount essentially providing the active
video information of 190 cycles, plus three cycles at each end of
the line, which provides some allowance that insures that all of
the active video information is written into memory. Thus, in terms
of 24 parallel lines of data occurring at a 3.58 MHz rate, wherein
the 24 bits comprise three samples per subcarrier cycle, there will
be 194 24 bit words per processed television line that are written
into memory. Referring to FIG. 24(2), words 1 through 4 are
specifically illustrated and it is understood that 196 24 bit words
will be present in each line. As previously described with respect
to the memory shown in FIG. 26a, the words are multiplexed for
operation by the memory 1900 so that the 196 words per line are
written into memory using 49 memory cycles, i.e., the data is
written into memory and read from memory using 96 bit words at 1/4
the 3.58 MHz rate and the timing diagrams shown in FIG. 24
illustrate the manner in which groups of four words are processed
by memory. The input byte select signals are shown in FIGS. 24(3)
and 24(4) which together generate the two bit binary code for
multiplexing the words into the appropriate latches 1956 (FIG.
26a), and FIGS. 24(13) and 24(14) illustrate the output byte select
signals for reading the information from the 4-to-1 switches 1978
(FIG. 26b). The address for the memory integrated circuits 1966 is
selected by addressing the rows using a six bit address word on the
address lines followed by a column address on the same address
lines and FIG. 24(7) illustrates the row address strobe followed by
the column address strobe in FIG. 24(8). The timing shown in FIG.
24(7) through FIG. 24(11) are in nanoseconds and represent basic
tolerances which permit the memory to operate within its timing
capability to produce valid information. The end of the CAS pulse
effectively initiates the read cycle, the data being valid within
165 nanoseconds of the end of the CAS pulse, as shown in FIG.
24(11). The occurrence of the next output data strobe (FIG. 24(12))
then latches the data from memory, and as shown, the time period
from the beginning of word 1 being written into memory and the
first opportunity it can be read from memory represents a 21/2
cycle delay as shown at the bottom of the drawing. As is evident
from FIGS. 24(7) and 24(8), the addresses are maintained for a
period of four words and after reading occurs, writing is performed
as shown by the occurrence of the write enable pulse shown in FIGS.
24(10) which occurs after the fourth word has been latched into the
latches 1956. If a drop-out occurs during the presence of any one
of the four words, then writing is inhibited and the data in the
memory is not updated.
As previously mentioned, the data during the 20 lines of vertical
interval is not written into the memory so that only 242 lines
forming the video data interval are written into memory rather than
the total of 262.5 lines forming a television field. By providing
four lines on each end of the vertical interval to provide a
centering tolerance, a capacity of 250 lines need only be required
to provide an actual effective delay of 262 lines. Accordingly,
when writing is to be performed, the memory is inhibited until line
17 of the field, at which time the memory is activated and 250
lines are then written into the memory before it has been inhibited
for 13 additional lines whereupon the second field of a frame will
be written into memory beginning with line 279. It is important
that the memory begin on an odd line for the subsequent field if it
had begun on an odd line in the initial field. Thus, as has been
described, when line 17 of the first field is the first line to be
written, writing line 279 of the second field conforms to this
requirement which is necessary in order for the proper phase of the
subcarrier to be maintained.
To carry out the operation of the memory in accordance with the
timing requirements that have been described, the circuitry shown
in FIGS. 25a and 25b operates to provide the necessary signals that
are used by the circuitry shown in FIGS. 26a and 26b to operate the
input latches, memory circuits, output latches and other circuit
components. Turning initially to FIG. 25a, a station reference
vertical signal is applied on line 2030 that is connected to the
input of a positioning monostable multivibrator 2032, the output of
which is connected to the input of another monostable multivibrator
2034 that is connected via line 2036 to the input of a third
monostable multivibrator 2038 and a NAND gate 2040. The other input
of the NAND gate 2040 is supplied by a frame signal on line 372
from the digital synchronization sequence adder circuitry 40. Line
372 is also connected to a NAND gate 2044 that has as its other
input line 2046 that is supplied by the monostable multivibrator
2038. The outputs of the gates 2040 and 2044 are connected,
respectively, to the two inputs of a gate 2046 that produces a
single pulse on line 2048 that occurs at the first line of each
field, and this pulse is used to begin the field start sequencing
that is used by other circuitry as will be described.
A reference horizontal sync pulse that is synchronized with
subcarrier is applied on line 2050 which is properly positioned by
the cascaded monostable multivibrators 2052 and 2054, with the
output line 2056 of the latter extending to counters 2058 which
operate to provide a predetermined delay that is approximately four
or five cycles of subcarrier. The delayed pulse appears on line
2060 and is also applied to cascaded monostable multivibrators 2062
and 2064, the former of which properly positions the delayed pulse
while the latter provides a pulse having a 140 nanosecond pulse
width. The output line 2068 of the monostable multivibrator 2064 is
connected to a gate 2066 so that the single pulse that is produced
on line 2048 is passed at the proper time relative to horizontal
sync and produces a field start signal on line 2070 as well as a
start signal line on line 2072.
The field start signal on line 2070 effectively clears the address
counter that addresses the memory circuits 1900. A reference 3.58
MHz clock on line 2073 is gated through gates indicated generally
at 2074 and provides a clock signal on line 2076 for use by the
counters 2058 and also as the input to a monostable multivibrator
2078 that positions the phase of the clock and provides a rephased
3.58 MHz clock signal on lines 2080 and 2082 that control the
remainder of the circuitry shown in FIGS. 25a and 25b. More
specifically, the line 2080 is applied to a pair of flip-flops 2084
which are connected to function as a divide-by-4 counter and
generate the input byte select signals on lines 1960. The
flip-flops 2084 are reset by line 2072 so as to synchronize the
word counter every 49 counts, i.e., at the beginning of the video
interval portion of every line. The output lines of the flip-flops
2084 are also decoded by NAND gates 2086 and 2088 and produce
signals on lines 2090 and 2092 which comprise the main clocking for
the reading and writing operations that are done by the memory. The
signal on line 2090 comprises a pulse that occurs at the first word
of the four word sequences and the signal on line 2092 comprises
the write clock and occurs at the fourth word of every four word
sequence. The output line 2082 from the monostable multivibrator
2078 is used to trigger a monostable multivibrator 2094 that is
used to properly position the input strobe and output line 2096
triggers a monostable multivibrator 2098 that provides a 60
nanosecond output pulse on line 2100 that extends to a gate 2102
that provides the data input strobe signal on line 1962. Similarly,
the Q output line 2106 of the monostable multivibrator 2094 extends
to a monostable multivibrator 2108 that properly positions the
output strobe signal and output line 2110 triggers a monostable
multivibrator 2112 which provides a 60 nanosecond pulse on line
2114 that clocks flip-flops 2116 and 2118 which have output lines
1980 for generating the output byte select signals. Line 2114 also
extends to a NAND gate 2120 which, together with outputs from the
flip-flops 2084, generate the output strobe signal on line
1974.
The line start signal on line 2072 is also applied to a 49 count
counter 2122 to load the same and the counter 2122 is clocked by
line 2092 which has a pulse every fourth word when the gate 2088 is
enabled. When the 49 count memory counter 2122 reaches the terminal
count, then the signal on line 2124 disables gate 2086 as well as
gate 2088 until the video interval portion of the next television
line is received. The signal on line 2124 also clocks a 250 line
counter 2126 that has output line 2128 extending to a flip-flop
2130. The flip-flop 2130 has output lines 2132 and 2134, the former
of which extends to one input of a gate 2136, the other input of
which is supplied by line 2138 from a flip-flop 2140 that is
clocked by output strobe line 1974. The signal on line 2138
provides line blanking whereas the signal on line 2152 provides
field blanking of either 12 or 13 lines and the output of the gate
2136 is provided on line 2142 that is inverted and provides the
switch inhibit signal on line 1934 (see FIG. 27b).
If a drop-out has been detected and a drop-out command signal has
been generated on line 1926 which extends to a flip-flop 2144, the
3.58 Mhz output strobe signal on line 1974 will clock the drop-out
command signal on line 1926 through the flip-flop 2144 onto line
2146. The passed through drop-out command signal clears a flip-flop
2148 and its output line 2150 will have a signal that is gated
through gate 2152 and gate 2154 to provide a drop-out disable
signal on line 2156 that will disable gate 2158 and preclude the
write enable signal from being asserted on line 2016. Thus, if a
drop-out appears for any one of the four words, the write enable
will not be asserted, which will prohibit bad data from being
written into the memory. The signals on line 2090 which occur every
fourth word also trigger a monostable multivibrator 2160 which
properly positions the signal and its output is connected to
another monostable multivibrator 2162 that provides a 150
nanosecond pulse on line 2164. The Q output line 2166 of the
monostable multivibrator 2162 is applied to clock at flip-flop 2168
as well as to clear input of a flip-flop 2170. If the drop-out
inhibiting signal is not present on line 2156, then the signal on
line 2164 will be gated through gate 2158 and produce the write
enable signal on line 2016 at the proper time after the fourth word
has been written into the input latches 1956. Line 2090 is also
applied to a monostable multivibrator 2174 and triggers the same to
provide the RAS start on line 2176, which clocks a flip-flop 2178
that provides the RAS pulse on line 2008. The output line 2176 also
triggers a monostable multivibrator 2180 which has an output line
2182 that clocks a flip-flop 2184 that produces the CAS pulse on
line 2012. Line 2176 also triggers yet another monostable
multivibrator 2186 that has an output which clocks the flip-flop
2170 to change the address from one set of six inputs to the other
set of six inputs via line 2188 that is the select line for a pair
of 2-to-1 integrated circuits 2190 containing 2-to-1 switches. The
switches have six output lines 2020 that are connected to the
address inputs of the memory chips 1966. The addresses are provided
by address generators 2192 which has 12 output lines 2194 that are
connected to the 2-to-1 switches 2190 and the address generator
2192 is clocked by line 2164 which is incremented every fourth word
in the manner that has been described with respect to FIG. 24. A
line 2196 from the address generator 2192 is applied to the clock
input of a flip-flop 2198 that cooperates with a flip-flop 2200 to
generate the block select signals on line 1996 for selecting the
proper group of rows of memory as previously described. The field
start signal on line 2070 clears the address generator 2192 as well
as the flip-flops 2198 and 2200 at the beginning of every
field.
Referring to FIG. 28, there is exemplified a servo system of
generally conventional capstan and head wheel servo loops 3020 and
3022, respectively, employed to maintain synchronous control of the
tape movement and of the head wheel rotation during the record and
reproduce operations. Typical of servo loops 3020, 3022 are those
described in the Ampex Corporation, AVR-1 Videotape Recorder,
Operation and Maintenance Manual, catalog No. 1809214, issued July,
1976, particularly in pages 6-4 through 6-31 and 6-45 through 6-84,
which description is incorporated by reference herein.
As previously described, the usual off-tape horizontal and vertical
sync information typically used to provide servo control during
reproduce operations are not available. Instead, the servo system
is controlled by employing the horizontal line interval related
signal extracted from the reproduced data, i.e., the unique digital
word series at line interval 1050, that is inserted in the stream
of processed television data during the record operation by the
sequence adder circuitry 40 of previous mention shown in, for
example, FIGS. 1 and 12.
In FIG. 28, the conventional head wheel tach pulses and 246 Hz
(NTSC standard) control track signal off tape, are applied to a
phase comparator 3028 via lines 3024, 3026 respectively. The output
of the latter is applied to a differential amplifier 3030 (which
performs a comparison) via a playback contact of a playback-frame
bias switch 3032. The frame bias contact of the switch 3032 is
coupled to a fixed frame bias source 3034. The second input to the
amplifier 3030 is coupled to a fixed reference voltage 3036. The
switch 3032 is controlled by a signal on a line 1950 from playback
circuitry within the logic and servo feedback circuit 200 of
previous mention. A voltage controlled oscillator 3040 is coupled
to the output of the differential amplifier 3030 and thence to a
playback contact of a switch 3042, whose record contact is coupled
to the H/64 reference signal on a line 3044 derived from a
horizontal (H) reference signal on a line 3066, further discussed
below. Switch 3042 is in turn coupled to the capstan servo loop
3020.
In generally conventional fashion, during the record mode the
capstan and head wheel servos 3020, 3022 are locked together in
response to the H/64 reference signal on line 3044.
In the reproduce mode, the series of unique digital words, which
identify the frames to derive vertical sync, are extracted via, for
example, a vertical pulse decoder 3046 in the converter/switch
circuit 52, which may be similar to the decoding gates 622, 624 of
FIG. 18b. The extracted series of digital words are fed via a line
3048 (corresponding to lines 634, 636 of FIG. 1) to capstan and
head wheel coincidence gates 3050, 3052 respectively. The latter
gates also receive the frame reference (FR. REF.) sync signal from
the sync generator 192 of previous mention via a line 3054. Gate
3050 is coupled to a AND gate 3056 via an AND gate 3058 and an
inverter 3060, wherein AND gate 3056 is also coupled to the head
wheel coincidence gate 3052. AND gate 3058 also is coupled to a
pulse detector circuit 3026 which detects the presence of the
frame-identifying unique digital words on the line 3048.
The AND gate 3056 in turn is coupled to an AND gate 3064 which also
receives the horizontal reference (H-ref) sync signal from the sync
generator 192 via a line 3066. A divide-by-64 (.div.64) divider
3068 is coupled to the AND gate 3064, and provides the H/64 signal
to control the servo loop of the head wheel servo 3022.
During the reproduce process, the series of unique digital words
which identifies line one of the first of the fields of the
four-field sequence in the NTSC format, are compared with the frame
reference signal. When the capstan gate 3050 detects that the tape
is not properly synchronized with the frame reference, AND gate
3058 provides a logic level, on line 1950 which activate the switch
3032 to connect the amplifier 3030 to the fixed frame bias source
3034 which, in turn, runs the capstan off frequency to properly
position the tape with respect to the frame reference. The capstan
coincidence gate 3050 then detects the tape sync condition, the
switch 3032 is returned to the playback position, and the capstan
is locked to the head wheel tach.
If the head wheel coincidence gate 3052 detects that the head wheel
is not properly synchronized with the frame reference signal, it
generates additional pulses which are fed to the .div.64 divider
3068 to drive the head wheel into proper sync via the head wheel
servo loop 3022. When the head wheel achieves sync with the frame
reference signal, the head wheel servo is locked to the horizontal
reference related H/64 signal and the servo system is color framed
to provide synchronous reproduction of the processed television
signal.
The pulse detector 3062 detects the presence of the unique digital
words and prevents eratic operation of the servos in the absence of
the frame-identifying digital words.
From the foregoing description, it is apparent that several read
only memories are included in the apparatus and the programming for
these memories is shown in the following chart. The memories are
all of the type which have four output lines and the output code is
in hexadecimal format which is well known. For each of the read
only memories herein, the addresses are specified together with the
hexadecimal output that is generated at the corresponding
address.
______________________________________ CHART I
______________________________________ Outputs-Addresses for ROM
376 (top) 4-46, 110, 174, 238 8-44, 108, 172, 236 0-0-12, 14,
32-43, 64-76, 78, 96-107, 128-140, 142, 160-171. 192-204, 206,
224-235 Outputs-Addresses for ROM 376 (bottom) 1-46, 110, 174, 238
2-44, 108, 172, 236 4-12, 76, 140, 204 5-14, 78, 142, 206 A-0-11,
32-43, 64-75, 96-107, 128-139, 160-171, 192-203 Outputs-Addresses
for ROM 454 5-255 9-13-24 B-25-27 D-1-12, 248-254 E-217
Output-Addresses for ROM 1600 3-2, 6, 10, 14, 18, 22, 26, 30, 34,
38, 42, 46, 50, 54, 58, 62, 67, 71, 75, 79, 83, 87, 91, 95, 99,
103, 107, 111, 115, 119, 123, 127, 130, 134, 138, 142, 146, 150,
154, 158, 162, 166, 170, 174, 178, 182, 186, 190, 195, 199,203,
207, 211, 215, 219, 223, 227, 231, 235, 239, 243, 247, 251, 255
5-129, 131, 133, 135, 137, 139, 141, 143, 145, 147, 149, 151, 153,
155, 157, 159, 161, 163, 165, 167, 169, 171, 173, 175, 177, 179,
181, 183, 185, 187, 189, 191, 192, 194, 196, 198, 200, 202, 204,
206, 208, 210,212, 214, 216, 218, 220, 222, 224, 226, 228, 230,
232, 234, 236, 238, 240, 242, 244, 246, 248, 250, 252, 254 7-3, 7,
11, 15, 19, 23, 27, 31, 35, 39, 43, 47, 51, 55, 59, 63, 66, 70, 74,
78, 82, 86, 90, 94, 98, 102, 106, 110, 114, 118, 122, 126 C-0, 4,
8, 12, 16, 20, 24, 28, 32, 36, 40, 44, 48, 52, 56, 60, 65, 69, 73,
77, 81, 85, 89, 93, 97, 101, 105, 109, 113, 117, 121, 125, 128,
132, 136, 140, 144, 148, 152, 156, 160, 164, 168, 172, 176, 180,
184, 188, 193, 197, 201, 205, 209, 213, 217, 221, 225, 229, 233,
237, 241, 245, 249, 253 D-1, 5, 9, 13, 17, 21, 25, 29, 33, 37, 41,
45, 49, 53, 57, 61, 64, 68, 72, 76, 80, 84, 88, 92, 96, 100, 104,
108, 112, 116, 120, 124 Outputs-Addresses for ROM 1816 2-1, 5, 9,
13, 17, 21, 25, 29, 33, 37, 41, 45, 49, 53, 57, 61, 64, 68, 72, 76,
80, 84, 88, 92, 96, 100, 104, 108, 112, 116, 120, 124, 128, 132,
136, 140, 144, 148, 152, 156, 160, 164, 168, 172, 176, 180, 184,
188, 193, 197, 201, 205, 209, 213, 217, 221, 225, 229, 233, 237,
241, 245, 249, 253 3-0, 4, 8, 12, 16, 20, 24, 28, 32, 36, 40, 44,
48, 52, 56, 60, 65, 69, 73, 77, 81, 85, 89, 93, 97, 101, 105, 109,
113, 117, 121, 125, 129, 133, 137, 141, 145, 149, 153, 157, 161,
165, 169, 173, 177, 181, 185, 189, 192, 196, 200, 204, 208, 212,
216, 220, 224, 228, 232, 236, 240, 244, 248, 252 8-3, 7, 11, 15,
19, 23, 27, 31, 35, 39, 43, 47, 52, 56, 69, 72, 77, 81, 85, 89, 93,
97, 191, 105, 109, 113, 117, 121, 125, 60, 130, 134, 138, 142, 146,
150, 154, 158, 162, 166, 170, 174, 65, 178, 182, 186, 190, 195,
199, 203, 207, 211, 215, 219, 223, 227, 231, 235, 239, 243, 247,
251, 255 C-2, 6, 10, 14, 18, 22, 26, 30, 34, 38, 42, 46, 50, 54,
58, 62, 67, 71, 75, 79, 83, 87, 91, 95, 99, 103, 107, 111, 115,
119, 123, 127, 131, 135, 139, 143, 147, 151, 155, 159, 163, 167,
171, 175, 179, 183, 187, 191, 194, 198, 202, 206, 210, 214, 218,
222, 226, 230, 234, 238, 242, 246, 250, 254 Outputs-Addresses for
ROM 1612 7-48-71, 120-127 B-32-47, 104-119 D-16-31, 88-103 E-0-15,
72-87 Outputs-Addresses for ROM 1870 7-48-71, 120-127 B-32-47,
104-119 D-16-31, 88-103 E-0-15, 72-87
______________________________________
The specific electrical schematic diagrams also contain a large
number of integrated circuits, and these integrated circuits where
appropriate, include the model number in parentheses, utilizing
model numbers from well known sources of such components. Where
such model numbers are provided, the pin numbers are also shown
adjacent thereto. For typical flip-flop circuits, monostable
multivibrator circuits, AND gates, NAND gates, OR gates, NOR gates,
inverters and the like, such components are well known and for this
reason, neither model numbers nor pin numbers for them have been
provided.
From the foregoing detailed description, it should be understood
that a recording and reproducing apparatus of superior design has
been described and illustrated which offers many significant
advantages over present commercial FM recording and reproducing
systems. The use of digital data throughout the recording and
reproducing processing provide extraordinarily reliable operation
even at the significantly higher frequency at which the information
is being clocked, recorded and reproduced. The system utilizes only
two channels and operates at a clock rate of about 43 Mbits which
is significantly faster than comparable FM recorders and represents
a marked improvement in the state of the art. Furthermore, the
apparatus has been described as arranged to employ quadruplex type
record and reproduce apparatus. It should be appreciated that other
types of record and reproduce apparatus can be employed as well.
The characteristics of other record and reproduce apparatus may
alter the timing and control of the signal processing circuitry
because of the nature of the operation of such apparatus. However,
the nature of and the manner of making such alterations will be
readily apparent to those skilled in the art. Also, the apparatus
has been described as arranged to receive and process analog color
television signals. Should it be desired to employ the apparatus to
process, record and reproduce other signals, such as digital data
signals, component television signals and monochrome television
signals, it would be necessary only to modify the input processing
circuitry 32, the analog-to-digital converter 36 and clock
generator and burst store circuitry 42, as well as the timing and
control of the signal processing circuitry, to adapt the signal
processing circuitry to the characteristics of the signals to be
processed. In addition, those skilled in the art will appreciate
that other forms of digital storage devices, for example, shift
registers, can be utilized to perform the operations of the
memories 60-66. While the apparatus has been described as arranged
to record and reproduce color television signals at a rate less
than real time, if the conservation of magnetic recording media is
not an important consideration, the record and reproduce operations
can be performed at the input data rate. However, by still
discarding a portion of the horizontal blanking interval of each
television line, or other periodic synchronization interval
associated with other data signals, the time base correction
feature is retained, although the apparatus is modified to record
and reproduce at the input data rate.
It is of course understood that although preferred embodiments of
the present invention have been illustrated and described, various
modification, alternatives and equivalents thereof will become
apparent to those skilled in the art and, accordingly, the scope of
the present invention should be defined only by the appended claims
and equivalents thereof.
Various features of the invention are set forth in the following
claims.
* * * * *