U.S. patent number 9,500,727 [Application Number 13/868,014] was granted by the patent office on 2016-11-22 for system and method for control of rf circuits for use with an mri system.
This patent grant is currently assigned to Regents of the University of Minnesota. The grantee listed for this patent is Anand Gopinath, Sung-Min Sohn, John Thomas Vaughan, Jr.. Invention is credited to Anand Gopinath, Sung-Min Sohn, John Thomas Vaughan, Jr..
United States Patent |
9,500,727 |
Sohn , et al. |
November 22, 2016 |
**Please see images for:
( Certificate of Correction ) ** |
System and method for control of RF circuits for use with an MRI
system
Abstract
A system and method for automatically adjusting electrical
performance of a radio frequency (RF) coil assembly of a magnetic
resonance imaging (MRI) system during a medical imaging process of
a subject to control changes in loading conditions of the RF coil
caused by the subject during the medical imaging process.
Inventors: |
Sohn; Sung-Min (St Paul,
MN), Vaughan, Jr.; John Thomas (Stillwater, MN),
Gopinath; Anand (Wayzata, MN) |
Applicant: |
Name |
City |
State |
Country |
Type |
Sohn; Sung-Min
Vaughan, Jr.; John Thomas
Gopinath; Anand |
St Paul
Stillwater
Wayzata |
MN
MN
MN |
US
US
US |
|
|
Assignee: |
Regents of the University of
Minnesota (Minneapolis, MN)
|
Family
ID: |
49476697 |
Appl.
No.: |
13/868,014 |
Filed: |
April 22, 2013 |
Prior Publication Data
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Document
Identifier |
Publication Date |
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US 20130285659 A1 |
Oct 31, 2013 |
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Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
Issue Date |
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61636085 |
Apr 20, 2012 |
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Current U.S.
Class: |
1/1 |
Current CPC
Class: |
G01R
33/54 (20130101); G01R 33/3628 (20130101); G01R
33/3456 (20130101); G01R 33/34092 (20130101) |
Current International
Class: |
G01V
3/00 (20060101); G01R 33/34 (20060101); G01R
33/54 (20060101); G01R 33/36 (20060101); G01R
33/345 (20060101) |
Field of
Search: |
;324/318,322 |
References Cited
[Referenced By]
U.S. Patent Documents
Foreign Patent Documents
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2042145 |
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Jan 1992 |
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CA |
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2005111645 |
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Nov 2005 |
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2006014260 |
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Feb 2006 |
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WO |
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2006121949 |
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Nov 2006 |
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WO |
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2008064365 |
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May 2008 |
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WO |
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2010045457 |
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Apr 2010 |
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WO |
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Other References
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Primary Examiner: Arana; Louis
Attorney, Agent or Firm: Quarles & Brady LLP
Parent Case Text
CROSS REFERENCE TO RELATED APPLICATIONS
This application is based on, claims priority to, and incorporates
herein by reference in its entirety U.S. Provisional Application
Ser. No. 61/636,085, filed Apr. 20, 2012, and entitled "SYSTEM AND
METHOD FOR AUTOMATIC TUNING OF RF COIL CIRCUITS FOR USE WITH AN MRI
SYSTEM."
Claims
The invention claimed is:
1. A radio-frequency (RF) system for use with a magnetic resonance
imaging (MRI) system during a imaging process of a subject, the RF
system comprising: a radio frequency (RF) element configured to at
least one of transmit RF energy to and receive RF energy from the
subject during the medical imaging process; an array of reactive
components coupled to the RF element and configured to adjust at
least one of an impedance transformation and a frequency tuning
associated with the RF element; a mismatch detector circuit
configured to measure a reflected signal from the RF element using
a coupler including a coupling port and an isolated port separated
by a desired phase shift providing isolation therebetween; and a
feedback circuit configured to receive an indication of the
reflected signal from the mismatch detector circuit and
automatically determine at least one of an impedance adjustment and
a frequency tuning to be implemented by adjusting the array of
reactive components based on the reflected signal to effectuate
transmit-receive isolation.
2. The system of claim 1 wherein the mismatch detector includes a
directional coupler configured to sense the reflected signal.
3. The coil assembly of claim 1 further comprising a control system
configured to receive the at least one of the impedance adjustment
and the frequency tuning from the feedback circuit and change a
total reactance of the array of reactive components based
thereon.
4. The system of claim 3 further comprising a diode driver system
configured to be controlled by the control system to change the
total reactance of the array of reactive components.
5. The system of claim 1 wherein the array of reactive components
forms part of a Pi matching circuit.
6. A method for automatically controlling operation of a radio
frequency (RF) element for use with a magnetic resonance imaging
(MRI) system, the method comprising the steps of: (a) determining
at least one of a frequency tuning need and an impedance matching
mismatch; (b) determining a desired performance of the RF element
including a desired isolation between transmit and receive
functionality of the RF element to be achieved by reducing the at
least one of a frequency tuning need and an impedance matching
mismatch; (c) adjusting at least one reactive component coupled to
the RF element to reduce the at least one of frequency tuning need
and impedance matching mismatch based on the desired performance of
the RF coil element and achieve the desired isolation between
transmit and receive functionality of the RF element; and (d)
repeating step (c) during an MRI process using the RF coil element
to control the at least one of frequency tuning need and impedance
matching mismatch and the desired isolation based on the desired
performance of the RF element despite loading condition changes
during the MRI process.
7. The method of claim 6 wherein step (c) includes automatically
changing a resonance frequency of the RF element by adjusting a
reactance of the at least one reactive component and the desired
performance includes matching the resonance frequency of the RF
element to a desired Larmor frequency.
8. The method of claim 6 wherein the RF element includes a
transmission line element and the at least one reactive component
includes at least one capacitor coupled thereto and configured to
adjust at least one of tuned condition and a matched condition of
the RF element.
9. The method of claim 6 wherein step (c) includes automatically
changing an impedance matching condition of the RF element by
adjusting a capacitance of a reactive component and the desired
performance includes matching to a desired loading condition of the
RF element.
10. The method of claim 6 wherein step (b) includes automatically
determining a reduced reflection coefficient at a desired Larmor
frequency to determine the at least one of a frequency tuning need
and an impedance matching mismatch.
11. The method of claim 6 further comprising adjusting at least one
reactive component coupled to the RF coil element based on changes
in operation of the RF coil between a transmit operational mode and
a receive operational mode.
12. A system for automatically adjusting electrical performance of
a radio frequency (RF) system of a magnetic resonance imaging (MRI)
system during a medical imaging process of a subject, the system
comprising: an adjustment circuit coupled between the MRI system
and the RF system; a coupler providing isolation between the MRI
system and the RF system; a frequency detector circuit configured
to measure a resonance frequency from the RF system; a feedback
circuit configured to receive an indication of the resonance
frequency from the frequency detector circuit and determine at
least one of a frequency tuning need and a impedance matching
mismatch using the resonance frequency; and a control circuit
configured to control operation of the adjustment circuit to
implement at least one of frequency tuning adjustments and
impedance matching adjustments to control changes in the resonance
frequency at least caused by changes in loading conditions of the
RF system caused by the subject during the medical imaging
process.
13. The system of claim 12 wherein the feedback circuit includes a
directional coupler configured to sense a reflected signal from the
RF system.
14. The system of claim 12 wherein the adjustment circuit includes
an array of reactive components coupled to the RF system and
wherein the control circuit is configured to change a reactive
parameter of the adjustment circuit to control changes in the
resonance frequency from the RF system at least caused by changes
in loading conditions of the RF system caused by the subject during
the medical imaging process.
15. The system of claim 14 wherein the array of reactive components
forms part of a Pi matching circuit.
16. The system of claim 12 further comprising a diode driver system
configured to be controlled by the control circuit to control
operation of the adjustment circuit.
17. The system of claim 12 wherein the medical imaging process
includes at least one of a continuous mode sweep imaging with
Fourier transformation (cSWIFT) and a continuous steering resonance
over the object (cSTEREO) imaging process.
18. The system of claim 12 wherein the control circuit is
configured to control operation of the adjustment circuit to
automatically control changes induced by changing in loading
conditions of the RF system caused by the subject during the
medical imaging process.
19. The system of claim 12 wherein the RF system includes a
transmission line element.
20. The system of claim 12 wherein the adjustment circuit,
frequency detector circuit, feedback circuit, and control circuit
form a coil element control circuit and the RF system further
comprises a plurality of coil element with associated control
circuits, each control circuit configured to connect to a
respective coil element of the RF system to control changes induced
by changes in loading conditions of each coil element in the RF
system caused by the subject during the medical imaging
process.
21. The system of claim 12 wherein the RF system includes at least
one of an RF coil and an RF antenna.
22. A radio-frequency (RF) system for use with a magnetic resonance
imaging (MRI) system during a imaging process of a subject, the RF
system comprising: a radio frequency (RF) element configured to
transmit RF energy to and receive RF energy from the subject during
the medical imaging process; an array of reactive components
coupled to the RF element to perform a parallel imaging process; a
diode system configured to provide electrical isolation between
components of the RF stem and adjust parameters of the reactive
components to perform at least one of an impedance matching and a
frequency tuning of the RF element; and a control circuit
configured to control the diode system to adjust parameters of the
reactive components.
23. The system of claim 22 further comprising a feedback circuit
including a directional coupler configured to measure a reflected
signal from the RF element and wherein the control circuit is
configured to use the measure of the reflected signal to
automatically control the diode system to adjust parameters of the
reactive components.
24. The system of claim 22 wherein the array of reactive components
forms part of a Pi matching circuit.
Description
STATEMENT REGARDING FEDERALLY FUNDED RESEARCH
N/A.
BACKGROUND OF THE INVENTION
The field of the invention relates to systems and methods for
magnetic resonance imaging ("MRI"). More particularly, the present
invention relates to systems and methods controlling a radio
frequency (RF) circuit for use with an MRI system.
When a substance such as human tissue is subjected to a uniform
magnetic field (polarizing field B.sub.0) applied along, for
example, a z axis of a Cartesian coordinate system, the individual
magnetic moments of the spins in the tissue attempt to align with
this polarizing field, but process about it in random order at
their characteristic Larmor frequency. If the substance, or tissue,
is subjected to a magnetic field (excitation field B.sub.1) that is
in the x-y plane and that is near the Larmor frequency, the net
aligned moment, Mz, may be rotated, or "tipped", into the x-y plane
to produce a net transverse magnetic moment M.sub.t. A NMR signal
is emitted by the excited spins after the excitation signal B.sub.1
is terminated, this signal may be received and processed to form an
image or produce a spectrum.
When utilizing these signals to produce images, magnetic field
gradients (G.sub.x, G.sub.y and G.sub.z) are employed. Typically,
the region to be imaged is scanned by a sequence of measurement
cycles in which these gradients vary according to the particular
localization method being used. The resulting set of received NMR
signals are digitized and processed to reconstruct the image using
one of many well known reconstruction techniques.
Radio frequency antennas, or coils are used to produce the
excitation field B.sub.1 and other RF magnetic fields in the
subject being examined. Such coils are also used to receive the
relatively weak NMR signals that are produced in the subject. Such
coils may be so-called "whole body" coils that are large enough to
produce a magnetic field for a human subject or, they can be much
smaller "local" coils that are designed for specific clinical
applications such as head imaging, knee imaging, wrist imaging, and
the like. Local coils may be either volume coils or surface
coils.
The aforementioned polarizing magnetic field is a common metric
upon which standard systems are differentiated. Standard magnetic
field strengths include 1.5 Tesla (T), 3 T, as well as those of
lesser and greater strength. Increased magnetic field strength
brings better signal-to-noise ratio (SNR), higher resolution, and
improved contrast and, therefore, experimental system use
ultra-high-fields of 7 T, 9.4 T and 11.74 T.
MR Imaging at higher magnetic fields strengths, including the
above-referenced ultra-high-fields, presents certain challenges in
RF coil circuit design. The common RF transverse electromagnetic
(TEM) coil design has widely used microstrip transmission line as
elements that inductively couple to the human anatomy at Larmor
frequencies of up to 500 MHz (11.74 T). As used herein RF coil, RF
antenna, microstrip, and the like all refer generally to electrical
elements or "RF elements" and are used herein. As shown in FIG. 1A,
the general RF structure for an MRI system is illustrated and
includes an RF coil element 10 on a dielectric substrate 12, for
example Teflon, wireless receiver components 14, wireless
transmitter components 16, an RF switch 18, and capacitors 20.
RF coils 10 for use with a microstrip line provide advantages,
including distributed coil circuit, high sensitivity due to high Q,
and relatively-simple structure. However, this high sensitivity
also creates a critical disadvantage in the form of a loading
(body) effect. As illustrated in FIG. 1B, the resonance frequency
and quality factor (Q) are changed from location 22 to location 24
due to impedance mismatch when different human body weight, shape,
and tissue composition are loaded. As illustrated, resonance
frequency shifts down from the Larmor frequency determined by the
strength of magnetic field (B.sub.0) because of the coupling
between RF coil and human anatomy.
The loading effect needs to be taken into consideration by way of a
tuning procedure after the body comes into the MRI scanner and it
is unpredictable. In general, referring again to FIG. 1A, tuning of
the RF coil system entails adjusting the capacitors 20. A first
capacitor is called the matching capacitor (C.sub.m) connected in
series and another capacitor called the tuning capacitor (C.sub.t)
connected in parallel. The matching capacitor C.sub.m matches the
impedance of the RF coil together with the effects of human anatomy
to the source and power amplifiers. The tuning capacitor C.sub.t
holds the resonance frequency (which is the Larmor frequency) of
the RF coil element, which is determined by the magnetic field
strength (B.sub.0) by:
.omega..gamma..times..times..gamma..times..times..times.
##EQU00001##
In current practice essentially all coils operating in transmit,
receive, and transceiver modes in MRI applications operate in fixed
tuned and fixed impedance matched conditions. The isolation between
coil elements, between transmit and receive coils, or modes is also
fixed. Coils are designed, manufactured, and used to be "one size
fits all". However all human body loads to which coils are applied
are not the same size, are not the same shape, are not in the same
position and, therefore, do not present the same electrical load to
the fixed coils. Because these coils with fixed tune, match, and
isolation conditions cannot be adjusted by any existing practical
means, suboptimal coil performance is the consequence. Reflected
power lost to impedance mismatch, attenuated power lost to
off-resonance transmission and reception, field distortions, and
power loss to coil-to-coil and T/R mode coupling (lack of
isolation) renders images with lower signal-to noise ratios, lower
homogeneity, more RF artifacts, and higher specific absorption
rates.
These problems have been tolerated at lower field strengths, such
as 1.5 T and below, because the longer wavelengths for the Lower
Larmor frequencies produce fields with stronger penetration and
higher uniformity, attributes which compensate somewhat for the
problems of ignoring coil tuning, matching and isolation. However
at higher B.sub.0 fields and B.sub.1 frequencies, coil tuning and
matching becomes more critical for the reasons given above. And
while, at lower frequencies, single monolithic resonators can be
used to generate uniform excitation fields, safe and successful
images at higher fields increasingly benefit from multi-channel
transmit, receive, and transceiver coils. Multichannel coils give
the ability to adjust the B.sub.1 field in any or all of the phase,
magnitude, frequency, space, or time domains to facilitate B.sub.1
field optimization over a field of interest. Each channel of a
multi-channel coil must be tuned, matched, and isolated. Also, each
channel should, ideally, be tuned, matched, and isolated per
patient or other load. That is, it would be beneficial if each
channel were tuned, matched, and/or isolated for each patient, or,
even better, be tuned, matched and/or isolated dynamically to track
patients' movements over the imaging process, be it course physical
movements of the body, or be it breathing, heartbeat, or other
physiological motion. Given that receivers of up to 64 channels and
transmitters of up to 16 channels are being delivered with MRI
systems today, manual adjustment of tune, match, and/or isolation
capacitances per channel is impractical for either clinical or
research applications. That is, practically speaking, to make such
adjustments, the operator of the MRI scanner would need to adjust
the capacitances of these capacitors 20 by hand. It is a major
obstacle to the application of these coils to the MRI system.
Therefore, it would be desirable to have a system and method for
providing and operating an RF system within an MR imaging process
that does not require cumbersome tuning, matching, and adjustments
thereto that varying substantially with operational characteristics
of the MR system and the subject being imaged.
SUMMARY OF THE INVENTION
The present invention overcomes the aforementioned drawbacks by
providing a system and method for tuning, matching, and/or
isolating a radio frequency (RF) system for use in an MRI system.
Such control may include electronically adjusting, even without the
need to manually intervene, the resonance frequency, impedance
matching, transmit-receive isolation, and the like of a loaded RF
circuit. In some configurations, the system and method can detect
operational variables, such as changes due to loading conditions,
and automatically tune circuitry associated with the RF coil to
compensate for the detected operational variables.
In accordance with one aspect of the invention, a method for
automatically tuning a radio frequency (RF) element for use with a
magnetic resonance imaging (MRI) system is provided. The method
includes measuring a reflected signal of the RF coil, determining
an adjusted condition, and adjusting at least one reactive
component based on the adjusted condition. Components may refer to
individual circuit elements or circuits themselves.
In accordance with another aspect of the present invention, a
radio-frequency (RF) system is disclosed for use with a magnetic
resonance imaging (MRI) system during a imaging process of a
subject. The RF system includes a radio frequency (RF) element
configured to at least one of transmit RF energy to and receive RF
energy from the subject during the medical imaging process and an
array of reactive components coupled to the RF element and
configured to adjust at least one of an impedance transformation
and a frequency tuning associated with the RF element. The RF
system also includes a mismatch detector circuit configured to
measure a reflected signal from the RF element and a feedback
circuit configured to receive an indication of the reflected signal
from the mismatch detector circuit and automatically determine at
least one of an impedance adjustment and a frequency tuning to be
implemented by adjusting the array of reactive components based on
the reflected signal.
In accordance with yet another aspect of the invention, a method is
disclosed for automatically controlling operation of a radio
frequency (RF) element for use with a magnetic resonance imaging
(MRI) system. The method includes the steps of (a) determining at
least one of a frequency tuning need and an impedance matching
mismatch and (b) determining a desired performance of the RF
element to be achieved by reducing the at least one of a frequency
tuning need and an impedance matching mismatch. The method also
includes (c) adjusting at least one reactive component coupled to
the RF element to reduce the at least one of frequency tuning need
and impedance matching mismatch based on the desired performance of
the RF coil element and (d) repeating step (c) during an MRI
process using the RF coil element to control the at least one of
frequency tuning need and impedance matching mismatch based on the
desired performance of the RF element despite loading condition
changes during the MRI process.
In accordance with still another aspect of the invention, a system
is disclosed for automatically adjusting electrical performance of
a radio frequency (RF) system of a magnetic resonance imaging (MRI)
system during a medical imaging process of a subject. The system
includes an adjustment circuit coupled between the MRI system and
the RF system and a frequency detector circuit configured to
measure a resonance frequency from the RF system. The system also
includes a feedback circuit configured to receive an indication of
the resonance frequency from the frequency detector circuit and
determine at least one of a frequency tuning need and a impedance
matching mismatch using the resonance frequency and a control
circuit configured to control operation of the adjustment circuit
to implement at least one of frequency tuning adjustments and
impedance matching adjustments to control changes in the resonance
frequency at least caused by changes in loading conditions of the
RF system caused by the subject during the medical imaging
process.
In accordance with yet another aspect of the invention, a
radio-frequency (RF) system is disclosed for use with a magnetic
resonance imaging (MRI) system during a imaging process of a
subject. The RF system includes a radio frequency (RF) element
configured to at least one of transmit RF energy to and receive RF
energy from the subject during the medical imaging process, an
array of reactive components coupled to the RF element, and a diode
system configured to adjust parameters of the reactive components
to perform at least one of an impedance matching and a frequency
tuning of the RF element. The system also includes a control
circuit configured to control the diode system to adjust parameters
of the reactive components.
The foregoing and other advantages of the invention will appear
from the following description. In the description, reference is
made to the accompanying drawings which form a part hereof, and in
which there is shown by way of illustration embodiments of the
invention. Such embodiments do not necessarily represent the full
scope of the invention, however, and reference is made therefore to
the claims and herein for interpreting the scope of the
invention.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1A is a schematic representation of a traditional radio
frequency (RF) coil system for use with an magnetic resonance (MR)
system to performing an imaging process.
FIG. 1B is a graph illustrating the loading effect on RF coil
systems such as illustrated in FIG. 1A.
FIG. 2 is a block diagram of an exemplary magnetic resonance
imaging ("MRI") system for use with the present invention.
FIG. 3 is a block diagram of an example of a radio frequency ("RF")
system that may form part of the MRI system of FIG. 2.
FIG. 4 is a partial, perspective view of an RF antenna system in
accordance with the present invention.
FIG. 5 is a simplified model of a subsection of the system of FIG.
4.
FIG. 6 is a schematic illustration of a RF coil system in
accordance with the present invention.
FIGS. 7A and 7B are Smith charts showing an open circuit state
associated with the system of FIG. 6 and a mismatched region from
the different impedance on Smith chart results from various
characteristics of human anatomies and reflection coefficients that
have resonance frequencies, respectively.
FIG. 8 is a schematic diagram of an example of an RF coil and
automatic tuning system in accordance with the present
invention.
FIGS. 9A and 9B are graphs illustrating matching and tuning
processes in accordance with the present invention.
FIG. 10 is a schematic diagram illustrating two options for
matching networks in accordance with the present invention.
FIG. 11A is a schematic diagram of matching and tuning network
applied capacitor array using an L network.
FIG. 11B is a schematic diagram of matching and tuning network
applied capacitor array using a Pi network.
FIG. 12A is a schematic diagram of a Pi network in a matching
network coupled to a RF coil element.
FIG. 12B Smith chart illustrating impedance matching and tuning
steps for use with the system of FIG. 12A.
FIG. 13 is a schematic diagram with the automatic matching and
tuning unit supporting a multi-channel RF coil.
FIG. 14 is a flow chart setting for the steps of an exemplary
process for automatically matching and tuning an RF coil or antenna
in accordance with the present invention.
FIG. 15 is another flow chart setting for the steps of an exemplary
process for automatically matching and tuning an RF coil or antenna
in accordance with the present invention.
FIG. 16 is a flow chart setting forth the steps of a processes that
can be implemented using a system, such as illustrated in FIG. 13
that includes a Pi matching circuit and other components.
DESCRIPTION OF THE INVENTION
Referring particularly now to FIG. 2, an example of a magnetic
resonance imaging ("MRI") system 100 is illustrated. The
workstation 102 includes a processor 108, such as a commercially
available programmable machine running a commercially available
operating system. The workstation 102 provides the operator
interface that enables scan prescriptions to be entered into the
MRI system 100. The workstation 102 is coupled to four servers: a
pulse sequence server 110; a data acquisition server 112; a data
processing server 114; and a data store server 116. The workstation
102 and each server 110, 112, 114, and 116 are connected to
communicate with each other.
The pulse sequence server 110 functions in response to instructions
downloaded from the workstation 102 to operate a gradient system
118 and a radiofrequency ("RF") system 120. Gradient waveforms
necessary to perform the prescribed scan are produced and applied
to the gradient system 118, which excites gradient coils in an
assembly 122 to produce the magnetic field gradients, and used for
position encoding MR signals. The gradient coil assembly 122 forms
part of a magnet assembly 124 that includes a polarizing magnet 126
and a whole-body RF coil 128.
RF waveforms are applied to the RF coil 128, or a separate local
coil (not shown in FIG. 2), by the RF system 120 to perform the
prescribed magnetic resonance pulse sequence. Responsive MR signals
detected by the RF coil 128, or a separate local coil (not shown in
FIG. 2), are received by the RF system 120, amplified, demodulated,
filtered, and digitized under direction of commands produced by the
pulse sequence server 110. The RF system 120 includes an RF
transmitter for producing a wide variety of RF pulses used in MR
pulse sequences. The RF transmitter is responsive to the scan
prescription and direction from the pulse sequence server 110 to
produce RF pulses of the desired frequency, phase, and pulse
amplitude waveform. The generated RF pulses may be applied to the
whole body RF coil 128 or to one or more local coils or coil arrays
(not shown in FIG. 2).
The RF system 120 also includes one or more RF receiver channels.
Each RF receiver channel includes an RF preamplifier that amplifies
the MR signal received by the coil 128 to which it is connected,
and a detector that detects and digitizes the I and Q quadrature
components of the received MR signal. The magnitude of the received
MR signal may thus be determined at any sampled point by the square
root of the sum of the squares of the and components: M= {square
root over (I.sup.2+Q.sup.2)} Eqn. (2);
and the phase of the received MR signal may also be determined:
.phi..function..times. ##EQU00002##
The pulse sequence server 110 also optionally receives patient data
from a physiological acquisition controller 130. The controller 130
receives signals from a number of different sensors connected to
the patient, such as electrocardiograph ("ECG") signals from
electrodes, or respiratory signals from a bellows or other
respiratory monitoring device. Such signals are typically used by
the pulse sequence server 110 to synchronize, or "gate," the
performance of the scan with the subject's heart beat or
respiration.
The pulse sequence server 110 also connects to a scan room
interface circuit 132 that receives signals from various sensors
associated with the condition of the patient and the magnet system.
It is also through the scan room interface circuit 132 that a
patient positioning system 134 receives commands to move the
patient to desired positions during the scan.
The digitized MR signal samples produced by the RF system 120 are
received by the data acquisition server 112. The data acquisition
server 112 operates in response to instructions downloaded from the
workstation 102 to receive the real-time MR data and provide buffer
storage, such that no data is lost by data overrun. In some scans,
the data acquisition server 112 does little more than pass the
acquired MR data to the data processor server 114. However, in
scans that require information derived from acquired MR data to
control the further performance of the scan, the data acquisition
server 112 is programmed to produce such information and convey it
to the pulse sequence server 110. For example, during prescans, MR
data is acquired and used to calibrate the pulse sequence performed
by the pulse sequence server 110. Also, navigator signals may be
acquired during a scan and used to adjust the operating parameters
of the RF system 120 or the gradient system 118, or to control the
view order in which k-space is sampled. By way of example, the data
acquisition server 112 acquires MR data and processes it in
real-time to produce information that may be used to control the
scan.
The data processing server 114 receives MR data from the data
acquisition server 112 and processes it in accordance with
instructions downloaded from the workstation 102. Such processing
may include, for example: Fourier transformation of raw k-space MR
data to produce two or three-dimensional images; the application of
filters to a reconstructed image; the performance of a
backprojection image reconstruction of acquired MR data; the
generation of functional MR images; and the calculation of motion
or flow images.
Images reconstructed by the data processing server 114 are conveyed
back to the workstation 102 where they are stored. Real-time images
are stored in a data base memory cache (not shown in FIG. 2), from
which they may be output to operator display 112 or a display 136
that is located near the magnet assembly 124 for use by attending
physicians. Batch mode images or selected real time images are
stored in a host database on disc storage 138. When such images
have been reconstructed and transferred to storage, the data
processing server 114 notifies the data store server 116 on the
workstation 102. The workstation 102 may be used by an operator to
archive the images, produce films, or send the images via a network
to other facilities.
As shown in FIG. 2, the radiofrequency ("RF") system 120 may be
connected to the whole body RF coil 128, or, as shown in FIG. 3, a
transmission section of the RF system 120 may connect to one or
more transmit channels 202 of an RF coil array 204 and a receiver
section of the RF system 120 may connect to one or more receiver
channels 206 of the RF coil array 204. The transmit channels 202
and the receiver channels 206 are connected to the RF coil array
204 by way of one or more transmit/receive ("T/R") switches 208.
Though illustrated as having multiple transmit channels 202 and
multiple receiver channels 206 connected to multiple
transmit/receive switches 208, the present invention is not limited
to traditional or parallel imaging systems. Also, the receiver
channel 206 may also be an assembly of coils separate from the
transmit coil array. In such a configuration, the T/R switches 208
are not needed. The transmit coil elements are detuned or otherwise
rendered dysfunctional during the receive operation, and the
receiver coil elements are similarly detuned or otherwise rendered
dysfunctional during operation of the transmit coils. Such detuning
may be accomplished with appropriate control logic signals.
Referring particularly to FIG. 3, the RF system 120 includes one or
more transmit channels 202 that produce a prescribed RF
electromagnetic field. The base, or carrier, frequency of this RF
field is produced under control of a frequency synthesizer 210 that
receives a set of digital signals from the pulse sequence server
110. These digital signals indicate the frequency, amplitude, and
phase of the RF carrier signal produced at an output 212. The RF
carrier is applied to a modulator and, if necessary, an up
converter 214 where its amplitude and phase is modulated in
response to a signal, R(t), also received from the pulse sequence
server 110. The signal, R(t), defines the envelope of the RF pulse
to be produced and is produced by sequentially reading out a series
of stored digital values. These stored digital values may be
changed to enable any desired RF pulse envelope to be produced.
The magnitude of the RF pulse produced at output 216 is attenuated
by an attenuator circuit 218 that receives a digital command from
the pulse sequence server 110. The phase of the RF pulse may also
be altered using phase shifters (not shown). The modulated RF
pulses are then applied to a power amplifier 220 that drives one
element of the RF coil array 204, or several such elements that are
electrically coupled. Multiple transmit channels then drive other
elements of the multichannel transmit coil array.
The MR signal produced by the subject is picked up by the RF coil
array 202 and applied to the inputs of the set of receiver channels
206. A preamplifier 222 in each receiver channel 206 amplifies the
signal, which is then attenuated, if necessary, by a receiver
attenuator 224 by an amount determined by a digital attenuation
signal received from the pulse sequence server 110. The received
signal is at or around the Larmor frequency, and this high
frequency signal may be down converted in a two step process by a
down converter 226. In an example of such a process, the down
converter 226 first mixes the MR signal with the carrier signal on
line 212 and then mixes the resulting difference signal with a
reference signal on line 228 that is produced by a reference
frequency generator 230. The MR signal is applied to the input of
an analog-to-digital ("A/D") converter 232 that samples and
digitizes the analog signal. As an alternative to down conversion
of the high frequency signal, the received analog signal can also
be detected directly with an appropriately fast analog-to-digital
("A/D") converter and/or with appropriate undersampling. The
sampled and digitized signal may then be applied to a digital
detector and signal processor 234 that produces in-phase (I) and
quadrature (Q) values corresponding to the received signal. The
resulting stream of digitized I and Q values of the received signal
are output to the data acquisition server 112. In addition to
generating the reference signal on line 228, the reference
frequency generator 230 also generates a sampling signal on line
236 that is applied to the ND converter 232.
The above-described RF coils can be formed using transmission line
elements. One example of such a transmission line element is a
microstrip line and, as will be described, such microstrip lines
can be advantageously utilized with the present invention. However,
other elements may be likewise utilized, such as strip lines,
coaxial cable, and the like.
The fundamental resonance frequency of a single microstrip
resonator can be modeled by:
.times..times..times..times..times..times..times. ##EQU00003##
where c is the speed of light in free space, L is the physical
length of the microstrip element and .di-elect cons..sub.eff is the
effective dielectric constant of the microstrip line. Therefore,
the one important parameter in determining the microstrip resonance
frequency is L, which is inversely proportional of f.
The characteristics of a single microstrip resonator can be
described as follows. Referring to FIG. 4, a microstrip line coil
300 can be formed by a low-loss dielectric substrate (.di-elect
cons..sub.r) 302 between a microstrip line 304 and a ground plane
306, where h is the height between the microstrip line 304 and
ground plane 306 and w is the width of microstrip line 304. A
simplified model of microstrip line 304 depicted in FIG. 4 is
illustrated in FIG. 5, where Z.sub.0 and .beta. are the
characteristic impedance and propagation constant, respectively, of
the microstrip line 304. Due to its specific semi-open transmission
line structure, substantial electromagnetic energy is stored in the
area near the strip conductor line. This results in reduced
radiation losses and preserving current uniformity in circuits
exceeding one tenth the wavelength of its carrier signal at high
fields. A rule of thumb for any resonant circuit in the electronics
industry is: "when circuit length exceeds 0.1.lamda., use a
transmission line."
The characteristic impedance, Z.sub.0, and propagation constant,
.beta., of microstrip line 304 resonator element can be calculated
as follows:
.di-elect
cons..times..times..times..times..times.<.times..times..time-
s..times..times..times..pi..times..times..function..times..times..times..t-
imes..times..mu..times..OMEGA..times. ##EQU00004## in free space.
The system 300 has asymmetric structure between the microstrip line
304 and ground plane 306 as shown in FIG. 4. Therefore, the
effective dielectric constant, .di-elect cons..sub.eff, of the
system 300 instead of the relative dielectric constant, .di-elect
cons..sub.r, of the substrate has to be considered in
characterizing different parameters related to the microstrip
resonance element. The expression of effective dielectric constant
and the characteristic impedance in the unloaded case is given by
Eqns. 5 and 6, above.
If there is a distributed capacitive load on the microstrip line
system for a shorter length of microstrip line, the characteristic
impedance and other parameters should be modified as:
.times..times..times..times..function..times..times..DELTA..times..times.-
.times..DELTA..times..times..times..times..times. ##EQU00005##
That is, CF is a correction factor according to capacitance loaded,
C.sub.f, capacitance of microstrip line per unit length, C.sub.0,
and the effective length, L.sub.eff, considered fringing effect. If
.lamda..sub.0 is the free-space wavelength, the wavelength of the
wave component along the microstrip is given by:
.lamda..lamda..times. ##EQU00006##
In the RF coil design, a TEM resonator is useful and has the best
B1 field distribution if the length of the microstrip line is
closing to the half-wavelength. When the half-wavelength TEM
resonator element has open-circuit terminations, the maximum
voltage occurs at the ends and the maximum current occurs at the
center of the microstrip line. This length, however, is impractical
to build a practical RF coil. Therefore, a capacitive termination
method can be used to reduce the electric length resulting in the
reduction of a practical size of a TEM resonator, and it also
provides more uniform B1 field distribution. The input impedance of
the capacitively terminated microstrip resonator is given by:
.times..function..times..times..times..times..times..times..function..bet-
a..times..times..function..function..times..times..times..times..function.-
.beta..times..times..times. ##EQU00007##
Where .beta.=(2.pi./.lamda.) is the phase constant, l is the length
of the microstrip line, Z.sub.0 is the characteristic impedance of
the microstrip line, Z.sub.C.sub.1 is 1/j.omega.C.sub.1, and
Z.sub.C.sub.2 is 1/j.omega.C.sub.2.
To meet the resonance condition (Z.sub.in.fwdarw..infin.), a
denominator should become an infinite. In case both capacitances
are identical (C=C.sub.1=C.sub.2), the capacitance value can be
derived from equation 13, such that:
.function..beta..times..times..omega..times..times..function..beta..times-
..times..times. ##EQU00008##
From equation (14), a tuning capacitor value can be estimated when
the dimensions of the microstrip line associated with .beta., l,
and Z.sub.0 are first fixed.
The impedance matching and frequency tuning of resonator element of
an RF coil based on microstrip line can be described as follows.
The use of a terms "matching" and "tuning" is often confused, or
the terms are used interchangeably. As follows hereafter,
"matching" refers to impedance matching, and "tuning" refers to
frequency tuning in the RF coil analysis.
The roles of the impedance matching is to deliver the maximum power
from a source (power amplifier) to a load (RF coil) for RF
transmitting, and improve the signal-to-noise (SNR) from a load (RF
coil) to a receiver (Low noise pre-amplifier) during RF receiving.
In addition, the impedance matching can protect RF devices (passive
and active elements, e.g. capacitors or RF T/R switch circuits)
from the reflected high RF power (typical a few Watt range in RF
coils) that builds standing waves containing the phase and
amplitude (i.e. voltage or current). The amplitude of waves can be
either subtracted or added due to the different phases. If the
standing wave with the maximum of amplitudes is positioned and
applied to a certain device, the device may be destroyed.
The frequency tuning is for adjusting the resonance frequency
rather than the impedance matching. The process of this frequency
tuning may affect the impedance matching condition, but the impact
is certainly lower than the one of the impedance matching
process.
Referring to FIG. 6, the microstrip element 304 may be connected to
a total of three capacitors, including a matching capacitor
(C.sub.m), a tuning capacitor (C.sub.t), and a capacitor with fixed
capacitance (C.sub.f) for reducing size of microstrip line 304.
With these capacitors and as will be described in further detail,
impedance matching is achieved as illustrated the Smith chart from
open circuit state, as illustrated in FIG. 7A. Referring to FIGS.
6-7B, the input impedance (Z.sub.in) of RF coil element 300
including the matching capacitor (C.sub.m) and tuning capacitors
(C.sub.t) may preferably be placed on the center of Smith chart for
impedance matching. The trace of impedance rotates about two third
of outer circle from open status with almost lossless property and
a fixed value capacitor, C.sub.f, takes a share of this trace. In
practice, a starting point for impedance matching may be determined
by the results of impedance of the combination of both the fixed
capacitance (C.sub.f) value and the physical length of microstrip
line 304 with the characteristic impedance. The well-chosen value
of tuning capacitor (C.sub.t) moves input impedance to unit circle
along outer circle, R=0, and then impedance travels unity circle by
that of matching capacitor (C.sub.m) toward the center of the Smith
chart.
However, the input impedance (Z.sub.in) of RF coil 300 changes when
human anatomy enters the MRI scanner and a portion of the human
anatomy is arranged proximate to the microstrip line 304.
Specifically, the RF coil 300 inductively or capacitively couples
with the human anatomy, which lowers the resonant frequency of the
RF coil element 300. FIG. 7B illustrates the mismatched region from
the different impedance on Smith chart results from various
characteristics of human anatomies and reflection coefficients that
have resonance frequencies, which corresponds to a drop in Qs.
Experimental results showing matched and mismatched reflection
coefficients illustrate that the body loading effect seriously
reduces the ability of power transfer and causes the RF coil to
fail to excite the human anatomy to the level anticipated by the
clinician using the prescribed pulse sequence. As a result, in
clinical practice, the images are degraded. To compensate, the
operator of the MRI scanner should tune capacitances of matching
and tuning capacitors (C.sub.m and C.sub.t); however, this process
remains a major obstacle to the application of these coils to
varied MRI systems and particularly to ultra-high-field MRI
systems.
Referring to FIG. 8, the present invention provides an automatic
tuning structure. Specifically, FIG. 8 provides a schematic
illustration of an example configuration of an automatic tuning
system 400 in accordance with the present invention that includes a
matching network 401. As illustrated, the above-described RF coil
element 300 may form one part of a multi-channel RF coil 402, such
as a multi-channel head coil. Such multi-channel head coil may
commonly have 8 channels and may have more or less channels. The
following components of the matching network 401 may be formed into
an array of detection/tuning sub-systems, for example, such that
each channel of the coil 402 is automatically tuned.
The system 400 includes an array of reactive components that form a
matching array 403. As illustrated, the reactive components may
include an electrically controlled capacitor array. However, the
reactive components may be inductive components in combination with
our instead of such capacitive components. That is, as one of
ordinary skill in the art will appreciate, reactive components or
elements may be represented by capacitive components or elements,
inductive components or elements, combinations thereof, and/or
combinations of capacitors, inductors, and other components. Thus,
reference to capacitors, likewise, can be construed to represent
reactive components and, thereby, inductive components.
The system 400 also includes a mismatch detection circuit 404 using
an RF directional coupler 406 and an automatic tuning control and
reactive-array switching driver block 408. The directional coupler
406 provides the capability of sensing the reflected signal, such
as reflected power, in RF signal path. The reflected signal coupled
from the main RF signal path is amplified and converted to a
constant voltage level by an envelope detector 410, that includes
PIN diodes 412, a capacitor (or inductor) 414, and a resistor 416.
A reference voltage (Vref) is determined and fixed at an output of
the envelope detector 410 under initial matched condition and the
difference between this reference voltage (Vref) and the loaded
output of the envelope detector 410 is generated in an operational
amplifier 418. It is noted that the envelope detector 410 that is
to measure the reflected signal, such a reflected power, can be
replaced with the general power detector integrated circuits
available in the commercial market. Eventually, the difference
output becomes large as the impedance moves away from the center of
Smith chart. The output of the mismatch detection block 404 may be
an analog signal depending on load (human body) characteristics and
is converted to the digital domain to compare and determine the
optimal capacitances (and/or inductances) of the matching array
block 403 in the automatic tuning control block 408. In the
automatic tuning control block 408, the measured mismatch
information may be processed to compare with current values and
previous values. For example, the automatic tuning control bock 408
may include a reactance-array switching driver to turn on or off in
each branch among the capacitor (and/or inductor) arrays and this
driver may be designed to apply to the MRI system under high RF
power, for example, 1 KW, switching. For example, the driver may be
designed to meet the specification of -30V to turn off one branch
in the capacitor (and/or inductor) array to decrease the
capacitances (or inductances) and the driver generates over 200 mA
to turn on one branch in reactance-array to increase
capacitances.
Within this exemplary architecture or other architectures or
configurations (both digital and analog), the automatic tuning
algorithm of the present invention can operate as will be
described. That is, while an exemplary circuit design and
configuration is described, it is contemplated that the present
tuning algorithm may be performed using various hardware
configurations. However, the following description will be made
with initial reference to the exemplary configuration provided with
respect to FIG. 8.
The operation of impedance tuning using traditional tuning schemes
has two capacitances to tune, C.sub.t and C.sub.m. Within this
structure, from initial unloaded matching conditions, the purpose
of C.sub.t tuning is to match the resonance frequency to Larmor
frequency as automatic tuning control block 408 decreases
capacitance using switch combination capacitors in the matching
array block 4. Referring to FIG. 9A, this matching step alone does
not create the overall matching condition although the resonance
frequency of the RF coil element is at the Larmor frequency. The
C.sub.t value is determined by the automatic tuning control block
408 during C.sub.t the tuning step and C.sub.m is tuned, as shown
in FIG. 9B for fine tuning. Adjusting C.sub.m cannot much change
the resonance frequency, only the amplitude of the reflection
coefficient is sharply changed. This means that C.sub.m tuning is
useful for fine tuning the impedance point on the Smith chart, and
moves it to the center after C.sub.m tuning moves it from away the
center to the nearby center.
FIG. 6 and associated description provides an inductance matching
network circuit with, for example, two capacitors representing the
reactance components. As described with respect to FIGS. 7A and 7B,
the input impedance (Zin) of the RF coil element including matching
(Cm) and tuning (Ct) capacitors should be placed on the center of
Smith chart for the impedance matching. The trace of impedance
rotates about two third of outer circle from the open status with
almost lossless property, and a fixed value capacitor (Cf) takes a
share of this trace. As also described, there are two different
dominant functions (tuning and matching), and two processes are not
independent. However, both effects can be seen as the same in the
reflection coefficient, but a degree of the influence is
significantly different.
As also described above with respect to FIGS. 9A and 9B, it is
clear that the adjustment of the tuning capacitor (Ct) has priority
because the error in the tuning capacitor's manipulation can
detrimentally influence the process. There are various
configurations to build matching network circuits. An L matching
network, in general, is popular in the RF coil design. Two
capacitors are employed. One dominantly plays a role in the
impedance matching process, and another is in charge of the tuning
process. A discrete inductor element is barely used in RF coil
designs due to the loss and radiation issues. This simple L
matching network successfully has accomplished the matching and
tuning in the existing RF coils, but it can be problematic when
additional functions are desired.
As such, a microstrip Pi matching network can be used to reduce the
physical size and increase the performance supporting reactance
arrays of the above-described automatic matching and tuning unit.
Before describing the integration of a Pi matching circuit with the
reactance arrays, the fundamentals of Pi matching circuit without
reactance arrays can be considered.
Referring to FIG. 10, as described above, each element in a
multi-channel coil can be viewed as a half-wavelength transmission
line 300, fore-shortened by adding capacitors shunt in the form of
capacitor C.sub.f and the matching network 401. The matching
network 401 may be created by a variety of specific circuit
configurations. For example, as described above and as illustrated
by a first circuit 430, the RF coil element 300 is terminated to
the two trimmer capacitors, the matching capacitor (C.sub.m) and
tuning capacitor (C.sub.t), at one end and the fixed value
capacitor (C.sub.f) at the other end. This creates a distributed LC
resonant circuit whose natural frequency can be adjusted with the
trimmer capacitors to accommodate changes in the loading of the
microstrip line by the sample, the human body.
However, a second, Pi matching circuit 432 may be used that
includes two capacitors (C.sub.phi1 and C.sub.phi2) connected to
ground in parallel and microstrip line (M-line) between the
capacitors (C.sub.phi1 and C.sub.phi2) to obtain a desired
inductance. As mentioned the Pi matching circuit 432 can be
advantageously applied when additional functions are required
because L matching for adding function blocks like capacitor
(and/or inductor) arrays for the auto-tuning and matching become
bulky and large losses experienced in the main signal path. That
is, as illustrated in FIG. 11A, the signal path from "Input" to
"Output" of the first circuit 430 can become unwieldy as additional
functionality is added to the circuit design. Since the substrate
has relatively high loss tangent (.delta.), in the range of, for
example, .delta.=0.0021, in dielectric material, the microstrip
line is also lossy. However, series connected components (e.g.
capacitors and/or inductors and PIN diodes) on a microstrip line
can be a more significant loss and impedance mismatch term in a RF
resonant circuit. Equivalent series resistance (ESR) of capacitors
and/inductors used in the illustrated design is about 0.3.OMEGA. to
0.5.OMEGA.. Because the quality factor (Q) of capacitors is
smaller, and equivalent series resistance becomes larger as the
operating frequency is higher, the series connected capacitor is
disadvantageous at higher operating frequencies. The size and
physical construction of a capacitor also may disturb the
electromagnetic field distribution on a transmission line as to
cause an impedance bump or impedance mismatch at that point.
This, and the series resistance internal to the capacitor, changes
the impedance of the line at the input to the capacitor. Moreover,
the range of capacitances of a series connection in L-matching is
about 1 pF to 5 pF generally. This is an unfavorably small range of
values to implement the capacitor array due to parasitic
capacitances and inductances.
On the other hand, as illustrated in FIG. 11B, the signal path from
"Input" to "Output" of the Pi matching circuit 432, however,
doesn't have any component on the main signal line, so it can
reduce the physical size from the signal input to output.
Therefore, there is not any impedance bump or connection loss in
the microstrip Pi matching network circuits.
Referring to FIG. 12A, a coil element 300 and matching network 401
is illustrated, where the matching network 401 includes the Pi
matching circuit 432. FIG. 12B shows a Smith chart illustrating the
steps of impedance matching and tuning using the matching network
401 including the Pi matching circuit 432. Capacitor values were
calculated as well as the dimensions of microstrip line that will
serve as an inductor in this Pi matching network 432 are directly
dependent on the choice of the dimensions of the microstrip
resonator element. The Pi matching network can significantly reduce
sheath current compared to L matching network due to the balanced
circuit configuration, thereby, making it easier to adjust matching
and tuning capacitances when a load is changed.
Referring to FIG. 13, the above-described systems can be
incorporated into a fully electric controlled stand-alone system
500 for automatic frequency tuning and impedance matching of the RF
coil. This system 500 includes the RF coupler 406, the Pi matching
circuit 432 with PIN diode/driver 412, an RF switch and power
detector 502, and an FPGA-based control system 504 coupled between
the RF coil 402 and the MRI system 100, such as the console 102 and
RF system 120.
Referring to the PIN diode driver 412, switches that control the
path of the RF high power for MRI applications (64 MHz.about.500
MHz) are the most common application of PIN diodes. PIN diodes are
three layer diodes, formed of a heavily doped P-type layer (anode)
and a heavily doped N-type layer (cathode) separated by a virtually
undoped intrinsic layer. Under forward bias, charge carriers from
the P and the N layers are forced into the intrinsic layer, which
reduces its RF impedance. When a reverse bias, voltage bias is
applied across the PIN diodes, all free charge carriers are removed
from the intrinsic layer, thereby causing its RF impedance to
increase. This variable RF impedance versus DC bias allows the
diode to be used in RF switching circuits, in which the PIN diode
is either heavily forward-biased or reverse-biased. Therefore, the
PIN diode is essentially a variable resistor. A simple PIN diode
switch circuit can provide reasonably low insertion loss (I.sub.L)
depending on the series resistance (R.sub.S) of the PIN diode. This
is one of the issues in the RF coil design with electrically
controlled reactance array because many PIN diodes are used, and
the overall loss must be controlled. Another contribution of PIN
diodes is high isolation (ISO) property in the reverse bias state.
It keeps the constant capacitances and/or inductances according to
the combination of the PIN diode switches to turn on or off
capacitor and/or inductor branches. Overall, the maximum RF power
rate considered as the limiting factor is defined by:
.times..times..function..times..times..times..times..function..times..pi.-
.times..times..times..times..times..times..times..times..times..function..-
times. ##EQU00009##
where Z.sub.0 is the characteristic impedance (typically 50.OMEGA.)
of the input power source, f is the operating frequency, C.sub.t is
the diode total capacitance, and P.sub.av is the maximum available
power, V.sub.g.sup.2/4Z.sub.0 (watt). These equations are under the
matched condition. In addition, a peak RF voltage and current
applied at the PIN diode switch is given by:
.times..function..times..times..times..function..times.
##EQU00010##
PIN diodes, like most diodes, are nonlinear devices in their
response characteristics, and as a result, they produce harmonics
and intermodulation distortion (IMD). Fortunately, these products
are usually at very low levels in a PIN diode switch because the
diodes themselves are either in a saturated, forward-biased
condition or are reversed-biased.
As mentioned above, components to be used in MRI systems are
subjected to severe requirements, such as high power handling
capability (.about.kW range), non-magnetic property. Thus, it is
desirable to select a non-magnetic property, the high power
capability, and the low series loss component. For example, this
diode may be designed for 53 dBm RF continuous wave input power,
150 mA forward DC current driving, -800V reverse DC voltage, 0.7 pF
maximum total capacitance, and 0.8.OMEGA. maximum series
resistance.
Regarding the driver for the PIN diodes, a fundamental property of
PIN diodes is their ability to control large RF signals with much
lower values of DC current and voltage using two states of PIN
diode driving circuits in which either a forward bias current or a
reverse bias voltage. Therefore, a pull up driving circuit for the
positive bias (+V) and a pull down driving circuit for the negative
bias (-V) can be used. Both driving circuits may be toggled between
+V and -V without the overlapped time. In reverse bias condition,
the instantaneous voltage (both RF and DC bias) across the PIN
diode should not exceed its reverse breakdown voltage supplied by
manufacturers. If the RF voltage swing exceeds this voltage, the
driver should have sufficient reverse bias current capability to
achieve the desired switching speed, but it should also provide the
excess reverse current required during the high power RF pulse.
Under this reverse biased leakage condition, the PIN diode may heat
appreciably causing an increase in the leakage current. If the
leakage current is large enough, thermal runaway will cause the PIN
diode to be destroyed.
In many applications, high applied reverse bias voltages are often
problematic part to implement. Fortunately, the practical reverse
bias does not require a full reverse bias (Vbias) condition to keep
the safe switching region. That is, the requirement of the reverse
bias voltage is less than the peak RF voltage, and the relationship
between the reverse bias voltage and the intrinsic layer of PIN
diode can be readily determined.
Turning now to the RF power monitoring system 502, the main control
circuit uses the information made by sampling RF power from the
main RF signal path, and manipulates matching and tuning capacitor
(and/or inductor) arrays based on PIN diode switch circuits to find
the optimal switching combination. In the power measurement, the
first term for the reflected power measurement is a reflection
coefficient, usually denoted by the symbol gamma (.GAMMA.), is
given by:
.GAMMA..times. ##EQU00011##
This represents the ratio of the reflected signal voltage to the
forwarded signal voltage, and also expressed by the impedances
between the load and source. Some RF power is reflected and .GAMMA.
becomes greater than zero if Z.sub.in is not matched the impedance
of the RF source (power amplifier), typically 50.OMEGA..
Two more useful expressions are commonly used to describe the
reflection measurement: VSWR (Voltage Standing Wave Ratio) and
Return Loss (RL):
.GAMMA..GAMMA..times..times..times..times..GAMMA..times..times..times..ti-
mes. ##EQU00012##
The ratio of the maximum to minimum voltage is known as VSWR, and a
measure of how well matched a RF source (power amplifier) is to a
RF coil element with a simple unit. The return loss is the
measurement in dB of the ratio between the forward and reflected
power. For example, a RF coil element with a VSWR of 2:1 would have
a reflection coefficient of 0.33, a return loss of 9.5 dB, and
about 11 percent of power loss. If 1 kW (60 dBm) is applied to this
RF coil element, the return loss would be 9.5 dB. Therefore, 111
watts would be reflected and the rest 889 watts would be
transmitted to generate B1 fields. In this case, the reflected 111
watts should not be ignored, and an impedance matching can be used
to reduce the reflection power. In general, RF coils of MRI system
should have at least VSWR of 1.2 (.apprxeq.-20 dB in the reflection
coefficient) to keep the proper matching condition. The VSWR can be
checked on the Smith chart by S11 measurements in a network
analyzer. The ratio of Vmax to Vmin becomes larger as the
reflection coefficient increases. That is, if the ratio of Vmax to
Vmin is one, then there is no variation in VSWR, and the impedance
of the RF coil is perfectly matched to the RF source. If the ratio
of Vmax to Vmin is greater than unity, then there is a signal
fluctuation that makes the VSWR. In the power measurement circuit
respect, these VSWR signals are used to generate DC output voltages
with a rectifier circuit. In other words, the RF power detection
circuit offers the reference level when VSWR=1, and the DC output
of this circuit is a linear function with the VSWR levels.
For power measurement, RF input signals (i.e. VSWR signals in case
of the power monitoring circuit) enter the power detect circuit
that may include a diode (typically Schottky diode), a reactive
component (capacitor and/or inductor), and a resistor. Schottky
diodes are characterized by fast switching times, low forward
voltage drop, and low junction capacitance. This Schottky diode
detector is a basic simple rectifier circuit which produces an
almost DC output current or voltage that is proportional to the
magnitude of the alternating input signal. In this circuit, the
Schottky diode rectifies the input alternating signal and charges
the output capacitor.
Regarding the design of the coupler 406, it may tap off a sample
signal of the input power without significantly affecting the main
signal path. An example of a coupler is provided in FIG. 14. The
coupler 406 may have, for example, 4 ports, and some parameters are
defined as follows:
.times..times..times..times..times..times..times..times..function..times.-
.times..times..times..times..times..times..times..times..function..times..-
times..times..times..times..times..times..times..function..times.
##EQU00013##
The coupling factor is the ratio of the coupled output power to the
input power. This is a negative number in dB, and the fundamental
specification of couplers. In scattering matrix, this is S13=S31 in
the forward direction and S24=S42 in the reverse direction. The
directivity is the ratio of the power out of the coupling port to
the power out of the isolation port, in dB. This is how effective
the coupler has the independency between the coupled and an
isolated port. Higher directivity is better performance of
couplers. The isolation is the ratio of the input power to the
power out of the isolation port, in dB. It indicates S14=S41 in the
forward direction and S23=S32 in the reverse direction. In
practice, it is not practical to build a perfect coupler, and some
amounts of unintended power exist in the ports.
To support a power monitoring circuit, directivity is useful
parameter. That is, both the coupled port and the isolated port
must be separated with a proper phase shift (e.g. 90 degrees). In
fact, Directivity=Isolation-Coupling factor in dB by equation
(20).about.equation (22), and a finite isolation is the reason for
limited directivity. Power detector circuits measure the reflected
signal indicating the reflected power by measuring the output of a
bi-directional coupler at the coupled port. This output can be
affected by the coupling factor, and the relative amplitude/phase
difference between the reflected signal and signal present due to
directivity make very complicated signals in the bi-directional
coupler. Thus, it is hard to measure the pure power component from
the coupler. The high accuracy of power measurement circuits can be
obtained with a high directivity coupler.
In the coupler design of the automatic matching and tuning unit,
there are some desirable constraints that include a main signal
line carrying high power RF signals, a small size to fit a RF coil
dimension, and a good coupled signal generation at the coupled
port. Since the operating frequency is 300 MHz at 7 T, .lamda./4
wavelength for a coupler design is about 25 cm. To meet these
constraints, the fundamental concepts have been modified and a new
coupler with lumped elements has been designed without affecting a
main signal line but with high directivity as shown in FIG. 15.
This coupler is adapted to monitor the RF power resulting from the
RF coil's impedance mismatch condition. In this example, the
coupled output of this coupler is connected to the input of the
aforementioned power measurement circuit and then the DC voltage
output is generated according to the coupled RF power level
depending on the loading condition. That is, a heavy load condition
makes voltage output higher.
In the overall operation of the system 500, there are two general
steps. The first step is an automatic tuning/match procedure with
the moderate power level, such as less than 20 dBm, and the second
step is the same with a normal MRI scanner operation with the high
power (up to 1 kW) to take MR images. In the first step, the output
of RF power detector represents the reflected power level through
the RF coupler for the power monitoring circuit at the input of the
RF coil. From this information, the main control and decision block
decides the optimum condition that is the impedance is matched to
50.OMEGA., and the frequency is tuned to the Larmor frequency,
regardless of the load (patient) conditions. The reactance-arrays
with PIN diodes are built with the microstrip Pi matching circuit
432, and the control system 504 electrically drives PIN diodes to
turn on or off through PIN diode driver circuits 412 during
searching the optimum impedance matching/tuning condition. Once the
control system 504 keeps the final results, the RF switch turns off
502 all circuits except for the essential part to drive PIN diodes.
The purpose of this step is to protect the circuitry operated with
the low power domain. As a result, this system 500 automatically
works for the impedance matching and tuning and, thus the
time-consuming manual re-tuning/matching is not required.
Referring to FIG. 15, a flowchart is provided setting forth steps
of an example of a method for frequency tuning and impedance
matching in accordance with the present invention, for example,
using a system such as described with respect to FIG. 8. As will be
described, variations on the system such as described with respect
to FIG. 8, for example, such as described with respect to FIG. 13
can utilize similar methods.
Referring to FIG. 15, the process starts under unloaded conditions
at process block 600 and the output of mismatch detection (V.sub.d)
is stored at process block 602 under those initial matched
condition without human anatomy. The output of mismatch detection
(V.sub.d) is compared to a threshold (Vthr) at decision block 604.
If V.sub.d is less than Vthr, the tuning is already matched.
However, if V.sub.d is greater than Vthr, C.sub.t tuning and
C.sub.m tuning are performed. As illustrated, this tuning may be
performed sequentially, beginning by selecting a new C.sub.t value
at process block 606 for the C.sub.t tuning steps; however, other
tuning steps or orders of steps may be used. At decision block 608,
the desired switch combination of C.sub.t array block for new
C.sub.t values are selected until the output of mismatch detection
(V.sub.d) is minimal, whereby a new C.sub.t value is updated at
process block 610. The first in the C.sub.m tuning steps begins by
selecting a new C.sub.m value at process bock 612. Decision block
614 causes the minimum output to be searched like C.sub.t tuning
plus monitoring the output level related to the threshold to
determine a matching condition. If the output of mismatch detection
satisfies the minimum and the threshold condition, the automatic
tuning control block decides that an overall matching condition is
met, stores the C.sub.m value at process block 616 and retains this
desired state of match at process block 618.
Referring to FIG. 16, the above-described processes can likewise be
implemented using a system, such as illustrated in FIG. 13 that
includes a Pi matching circuit and other components. Generally, a
process for frequency tuning and impedance matching begins by
detecting a need or mismatch, such as indicated at process block
700. Upon detecting the need or mismatch, at process block 702,
capacitor C.sub.ph1, such as illustrated in FIG. 13, can be
adjusted and, likewise, at process block 704, capacitor C.sub.ph2
can be adjusted. Notably, the adjustments made with reference to
process block 702 and 704 can be done in series or parallel. With
adjusting completed to provide the desired frequency tuning and
impedance matching, the system is controlled, at process block 706,
to maintain the system performance at the desired frequency tuning
and impedance matching.
It can be difficult to obtain high quality images in
ultra-high-field MRI system unless impedance matching is fulfilled.
The present invention provides a system and method for automatic
tuning of RF coil circuits using a mismatch detector, matching
array block, and control block. It provides fast tuning and
matching procedures, robust performance regardless of anatomy
types, and a structure that can be integrated a small board or
semiconductor chip.
The above-described systems and methods provide automatic and
highly-accurate frequency tuning and impedance matching. However,
the above-described systems and methods can also be used to perform
other tasks. For example, the above-described systems and methods
can be used to isolate transmit and receive operations of an RF
system, for example, when performing imaging techniques, where it
could be advantageous to dynamically adjust the phase angle to
maintain quadrature operation for continuous transmit and receive
techniques, such as a continuous steering resonance over the object
(cSTEREO) or continuous mode sweep imaging with Fourier
transformation (cSWIFT) for imaging and spectroscopy studies, such
as described in Idiyatullin, Djaudat, et al. "Continuous SWIFT."
Journal of Magnetic Resonance (2012), which is incorporated herein
by reference in its entirety. Resolution in the millisecond range
is desirable to compensate for motion, such as respiration and the
present invention is designed to advantageously provide automatic
control capabilities, such as described above.
The present invention has been described in terms of one or more
preferred embodiments, and it should be appreciated that many
equivalents, alternatives, variations, and modifications, aside
from those expressly stated, are possible and within the scope of
the invention. Therefore, the invention should not be limited to a
particular described embodiment.
* * * * *