U.S. patent number 9,479,199 [Application Number 14/930,139] was granted by the patent office on 2016-10-25 for low-cost receiver using integrated inductors.
This patent grant is currently assigned to SILICON LABORATORIES, INC.. The grantee listed for this patent is Silicon Laboratories Inc.. Invention is credited to Mustafa H. Koroglu, Yu Su.
United States Patent |
9,479,199 |
Koroglu , et al. |
October 25, 2016 |
Low-cost receiver using integrated inductors
Abstract
A receiver includes a first amplifier having an input for
receiving a radio frequency (RF) signal, and an output for
providing an amplified RF signal, a switch section for selectively
switching the RF signal onto one of a plurality of nodes, and a
filter section comprising a plurality of filters coupled to
respective ones of the plurality of nodes. A first filter of the
plurality of filters comprises a first variable capacitor coupled
in parallel with an inductance leg between a corresponding one of
the plurality of nodes and a power supply voltage terminal, wherein
the first variable capacitor has a capacitance that varies in
response to a tuning signal, and the inductance leg comprises a
first inductor in series with an effective resistance, wherein the
effective resistance has a value related to an upper cutoff
frequency to be tuned by the first filter.
Inventors: |
Koroglu; Mustafa H. (Austin,
TX), Su; Yu (Austin, TX) |
Applicant: |
Name |
City |
State |
Country |
Type |
Silicon Laboratories Inc. |
Austin |
TX |
US |
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Assignee: |
SILICON LABORATORIES, INC.
(Austin, TX)
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Family
ID: |
48608116 |
Appl.
No.: |
14/930,139 |
Filed: |
November 2, 2015 |
Prior Publication Data
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Document
Identifier |
Publication Date |
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US 20160056845 A1 |
Feb 25, 2016 |
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Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
Issue Date |
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14612232 |
Feb 2, 2015 |
9209838 |
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13468824 |
Mar 17, 2015 |
8983417 |
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13342548 |
May 13, 2014 |
8725103 |
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Current U.S.
Class: |
1/1 |
Current CPC
Class: |
H03H
11/1291 (20130101); H04B 1/18 (20130101); H04B
1/005 (20130101); H03H 7/0169 (20130101); H04B
1/16 (20130101); H03J 3/10 (20130101); H04B
1/1638 (20130101); H03F 3/19 (20130101); H03H
7/46 (20130101); H03F 1/565 (20130101); H03F
3/193 (20130101); H03H 2210/033 (20130101); H03F
2200/294 (20130101); H03H 7/09 (20130101); H03H
2210/025 (20130101); H03F 2200/451 (20130101); H03J
2200/32 (20130101); H03F 2200/171 (20130101) |
Current International
Class: |
H04B
1/18 (20060101); H03J 3/10 (20060101); H04B
1/00 (20060101); H03H 7/01 (20060101); H04B
1/16 (20060101); H03H 7/46 (20060101); H03H
11/12 (20060101); H03F 3/19 (20060101); H03F
1/56 (20060101); H03F 3/193 (20060101); H03H
7/09 (20060101) |
References Cited
[Referenced By]
U.S. Patent Documents
Other References
"Applications of Switched-Capacitor Circuits in Active Filters and
Instrumentation Amplifiers," Dr. William R. Grise, Department of
IET, Morehead State University, Technology Interface, vol. 3 No. 3,
Fall 1999, ISSN# 1523-9926. cited by applicant .
"Novel LC Pseudo Switched Capacitor Filter Suited for Wireless RF
Applications," Ahmed El Oualkadi et al., IEICE Electronics Express,
vol. 2, No. 8, Apr. 2005, pp. 286-291. cited by applicant .
Actions on the Merits for U.S. Appl. No. 14/612,232, filed Feb. 2,
2015 which is the parent application of 1052-3108; Actions on the
Merits for U.S. Appl. No. 13/468,824, filed May 10, 2012; and
Actions on the Merits for U.S. Appl. No. 13/342,548, filed Jan. 3,
2012. cited by applicant.
|
Primary Examiner: Le; Lana N
Attorney, Agent or Firm: Polansky & Associates, P.L.L.C
Polansky; Paul J.
Parent Case Text
This application is a division of prior application Ser. No.
14/612,232, filed Feb. 2, 2015, which is a division of prior
application Ser. No. 13/468,824, filed May 10, 2012, now U.S. Pat.
No. 8,983,417, which is a continuation-in-part of prior application
Ser. No. 13/342,548, filed Jan. 3, 2012, now U.S. Pat. No.
8,725,103, entitled "Receiver Including a Tracking Filter,"
invented by Mustafa H. Koroglu and Yu Su.
Claims
What is claimed is:
1. A receiver comprising: a first amplifier having an input for
receiving a radio frequency (RF) signal, and an output for
providing an amplified RF signal; a switch section for selectively
switching the RF signal onto a selected one of a plurality of
nodes; and a filter section comprising a plurality of filters
coupled to respective ones of the plurality of nodes, wherein a
first filter of the plurality of filters comprises a first variable
capacitor coupled in parallel with an inductance leg between a
corresponding one of the plurality of nodes and a power supply
voltage terminal, wherein the first variable capacitor has a
capacitance that varies in response to a tuning signal, and the
inductance leg comprises a first inductor in series with an
effective resistance, wherein the effective resistance has a value
related to an upper cutoff frequency to be tuned by the first
filter.
2. The receiver of claim 1, further comprising: a resistor having a
first terminal coupled to the output of the first amplifier, and a
second terminal coupled to a first terminal of the inductor,
wherein the effective resistance of the inductance leg is equal to
a resistance of the resistor plus a parasitic resistance of the
inductor.
3. The receiver of claim 1, wherein the first variable capacitor
comprises: a fixed capacitor having a first terminal coupled to the
output of the first amplifier, and a second terminal coupled to the
power supply voltage terminal; a binary capacitor having a first
terminal coupled to the output of the first amplifier, a second
terminal coupled to the power supply voltage terminal, and a
control terminal for receiving a binary coded signal; and a
thermometer capacitor having a first terminal coupled to the output
of the first amplifier, a second terminal coupled to the power
supply voltage terminal, and a control terminal for receiving a
thermometer coded signal.
4. The receiver of claim 1, further comprising: a second amplifier
having an input coupled to the output of the first amplifier, and
first and second outputs forming a differential signal pair; a
second variable capacitor having a first terminal coupled to the
first output of the second amplifier, and a second terminal coupled
to the second output of the second amplifier, and a tuning input
terminal; and a second inductor having a first terminal having a
first terminal coupled to the first output of the second amplifier,
and a second terminal coupled to the second output of the second
amplifier.
5. The receiver of claim 1, wherein the plurality of filters
comprises: a second filter having an input selectively coupled to
the output of the first amplifier through the switch section for
attenuating components of the output of the first amplifier below
the upper cutoff frequency of the first filter and above an upper
cutoff frequency of the second filter; and a third filter having an
input selectively coupled to the output of the first amplifier
through the switch section for attenuating components of the output
of the first amplifier below the upper cutoff frequency of the
second filter.
6. The receiver of claim 5, wherein the upper cutoff frequency of
the first filter is approximately equal to 190 megahertz (MHz).
7. The receiver of claim 6, wherein the third filter comprises: a
third variable capacitor including a first terminal coupled to the
output of the first amplifier, and a second terminal coupled to a
second power supply voltage terminal; and a transformer including a
primary winding having a first terminal coupled to the first
terminal of the third variable capacitor and a second terminal
coupled to a third power supply voltage terminal, and a secondary
winding having a first terminal and a second terminal for providing
a differential filtered radio frequency signal thereto.
8. The receiver of claim 7, wherein the upper cutoff frequency of
the second filter is approximately equal to 470 MHz.
9. The receiver of claim 1, wherein the RF signal comprises a
television signal.
10. The receiver of claim 9, wherein the switch section selectively
switches the RF signal onto a selected one of the plurality of
nodes in response to a desired channel.
11. A method comprising: receiving a radio frequency (RF) input
signal; amplifying the RF input signal to form an amplified RF
input signal; selectively switching the amplified RF input signal
to one of a plurality of nodes; filtering the amplified RF signal
using a selected one of a plurality of filters coupled to
respective ones of the plurality of nodes; and in response to
switching the amplified RF signal onto a first of the plurality of
nodes, filtering the amplified RF input signal using a first filter
of the plurality of filters, wherein the first filter comprises a
first variable capacitor coupled in parallel with an inductance leg
between a corresponding one of the plurality of nodes and a power
supply voltage terminal, wherein the first variable capacitor has a
capacitance that varies in response to a tuning signal, and the
inductance leg comprising a first inductor in series with an
effective resistance, wherein the effective resistance has a value
related to an upper cutoff frequency to be tuned by the first
filter.
12. The method of claim 11, further comprising selectively
switching the amplified RF input signal to one of the plurality of
nodes corresponding to a desired frequency band.
13. The method of claim 12 further comprising: selectively
switching the amplified RF signal onto a second node coupled to a
second filter, wherein the second filter attenuates components of
the amplified RF signal below the upper cutoff frequency of the
first filter and above an upper cutoff frequency of the second
filter; and selectively switching the amplified RF signal onto a
third node coupled to a third filter, wherein the third filter
attenuates components of the amplified RF signal below the upper
cutoff frequency of the second filter.
14. The method of claim 13, wherein receiving the RF signal further
comprises receiving a television signal.
15. The method of claim 14, wherein selectively switching the
amplified RF input signal to one of the plurality of nodes
comprises selectively switching the amplified RF input signal to
one of the plurality of nodes in response to a desired channel.
16. The method of claim 11, further comprising: forming the
effective resistance by coupling a resistor in series with the
first inductor, wherein the effective resistance of the inductance
leg is equal to a resistance of the resistor plus a parasitic
resistance of the inductor.
17. The method of claim 11, further comprising forming the first
variable capacitor with a parallel combination of: a fixed
capacitor; a binary capacitor having a control terminal for
receiving a binary coded signal; and a thermometer capacitor having
a control terminal for receiving a thermometer coded signal.
18. The method of claim 11, wherein filtering the amplified RF
input signal using the first filter of the plurality of filters
further comprises: further amplifying the amplified signal; and
further filtering the further amplified signal using a second
variable capacitor in parallel with a second inductor.
19. The method of claim 18, further comprising forming the second
variable capacitor with a parallel combination of: a fixed
capacitor; a binary capacitor having a control terminal for
receiving a binary coded signal; and a thermometer capacitor having
a control terminal for receiving a thermometer coded signal.
20. The method of claim 18, wherein: the further amplifying the
amplified signal comprises further amplifying the amplified signal
to provide the further amplified signal as a differential signal;
and the further filtering comprises differentially filtering the
differential signal using the second variable capacitor in parallel
with the second inductor.
Description
FIELD OF THE DISCLOSURE
The present disclosure is generally related to receiver circuits,
and more particularly to receiver circuits configurable to receive
ultra high frequency and very high frequency broadcast radio
frequency signals.
BACKGROUND
Receiver circuits are used for both terrestrial and cable reception
within televisions, digital video recorders, video cassette
records, set-top box devices (such as cable and satellite tuners),
frequency modulation (FM) radios, models, and other electronic
devices. Some smart phones also utilize receiver circuits for
television reception. In general, such receiver circuits include a
tuner that selects a narrowband signal from within a wide or
broad-band signal having multiple channels. The tuner includes
bandpass filters, amplifiers, and mixer circuits for selecting a
desired channel and for rejecting unwanted channels, noise and
interference.
Television signals can be broadcast at a variety of different
frequencies, including ultra-high frequency (UHF), very high
frequency (VHF), and high frequency (HF) frequency bands. The
International Telecommunications Union (ITU) defines the UHF
frequency range as encompassing electromagnetic waves between 300
MHz and 3 GHz. VHF occupies frequencies within a range from
approximately 30 MHz to 300 MHz, and HF occupies frequencies within
a range from approximately 3 MHz to 30 MHz.
For a conventional tracking filter, multiple bandpass filters are
included to cover the range of possible frequencies, each tuned to
a limited portion of the overall bandwidth of the receiver. For
each filter, a large tuned impedance is desirable to minimize the
noise figure; however, the tuned impedance is typically a function
of the frequency and the size of the inductor, which size can limit
the frequency range for the particular filter.
Recently developed integrated circuit processes have allowed
inductors to be built with sufficiently large inductance to be
integrated with conventional signal processing circuitry on a
common complementary-metal-oxide-semiconductor (CMOS) chip. However
the inductors are relatively large, and to increase the inductance
it may be necessary to either increase the size or decrease the
quality factor of the inductor. A new receiver that is able to
relax this tradeoff would be desirable.
BRIEF DESCRIPTION OF THE DRAWINGS
The present disclosure may be better understood, and its numerous
features and advantages made apparent to those skilled in the art
by referencing the accompanying drawings, in which:
FIG. 1 is a block diagram of an embodiment of a system including a
receiver circuit having a tracking filter configurable to receive
VHF.sub.LO, VHF.sub.HI, and UHF broadcast signals.
FIG. 2 is a circuit diagram of a representative example of a
receiver including conventional tracking filter having a separate
bandpass filter tuned for each frequency band.
FIG. 3 is a block diagram of an embodiment of a portion of a
receiver including a tracking bandpass filter having a VHF.sub.LO
filter, a VHF.sub.HI filter, and a UHF filter.
FIG. 4 is a diagram of an embodiment of the VHF.sub.LO filter of
FIG. 3.
FIG. 5 is a diagram of an embodiment of the UHF filter of FIG.
3.
FIG. 6 is an embodiment of a small signal model of the UHF Filter
of FIG. 5.
FIG. 7 is a block diagram of a low-cost television receiver
according to an embodiment of the present invention.
FIG. 8 illustrates in schematic form the VHF.sub.LO filter of FIG.
7.
FIG. 9 illustrates a frequency domain graph useful in understanding
the operation of the VHF.sub.LO filter of FIG. 7.
The use of the same reference symbols in different drawings
indicates similar or identical items.
DETAILED DESCRIPTION
Embodiments of a receiver circuit are described below that include
a low noise amplifier for receiving a radio frequency signal and
including an output coupled to an input of a tracking filter
configured to receive very high frequency (VHF.sub.LO and
VHF.sub.HI) and UHF broadcast signals. As used herein, VHF.sub.LO
signals refer to radio frequency signals in a range of
approximately 50 to 190 MHz, VHF.sub.HI signals refer to radio
frequency signals in a range of approximately 190 to 470 MHz, and
UHF signals refer to radio frequency signals in a range of
approximately 470 to 860 MHz. The VHF.sub.LO portion of the
tracking filter includes a low pass filter for receiving VHF.sub.LO
broadcast signals and that utilizes inductive peaking to extend an
upper frequency range (or limit) of the low-pass filter. The
VHF.sub.LO portion of the tracking filter includes a
single-to-differential amplifier having an input coupled to the
output of the low noise amplifier, and having a first output and a
second output. The VHF.sub.LO portion of the tracking filter
further includes a band-pass filter coupled between the first and
second outputs, which band-pass filter can be an
inductive-capacitive filter. The UHF portion of the tracking filter
uses a transformer in place of a single-to-differential amplifier.
The transformer includes a primary winding coupled to the output of
the low noise amplifier and a secondary winding magnetically
coupled to the primary winding. The secondary winding is configured
to include more turns than the primary winding in order to provide
a signal gain between the primary and secondary windings. An
example of one possible embodiment of a receiver circuit having a
filter configured for VHF.sub.LO, VHF.sub.HI, and UHF reception is
described below with respect to FIG. 1.
FIG. 1 is a block diagram of an embodiment of a system 100
including a receiver circuit 102 having a tracking filter 110
configurable to receive VHF.sub.LO, VHF.sub.HI, and UHF broadcast
signals. System 100 includes a signal source 104 coupled to
receiver circuit 102 through an input terminal or pad 106. Signal
source 104 can be an antenna, a coaxial cable, or other signal
source for delivering radio frequency broadcast signals to pad 106.
Receiver circuit 102 includes a front end circuit 123, which
includes a low-noise amplifier (LNA) 108 having an input coupled to
pad 106 and an output coupled to an input of a tracking filter 110.
Tracking filter 110 includes a control input coupled to
microcontroller unit 124 and an output coupled to an input of a
mixer 112.
Mixer 112 includes inputs for receiving an oscillator signal from a
local oscillator 128. Mixer 112 further includes an output for
providing an in-phase signal to a programmable gain amplifier (PGA)
114 and an output for providing a quadrature signal to PGA 116. PGA
114 includes an output coupled to an analog-to-digital converter
(ADC) 118, which has an output coupled to an input of a digital
filter 122. PGA 116 includes an output coupled to ADC 120, which
has an output coupled to an input of digital filter 122.
Front end circuit 123 also includes a radio frequency automatic
gain control (AGC) circuit 126 including an input coupled to an
output of MCU 124 and an output coupled to an input of LNA 108. MCU
124 is also coupled to local oscillator 128 to control the
frequency of the local oscillator signals. Front end circuit 123
further includes an intermediate frequency (IF) AGC circuit 130
including a control input coupled to an output of MCU 124 and
outputs coupled to PGAs 114 and 116. Front end circuit 123 also
includes a low-IF circuit 132 having a control input coupled to an
output of MCU 124 and includes outputs coupled to inputs of digital
filter 122. MCU 124 is also coupled to memory 148, which may store
instructions and/or data that can be used by MCU 124 to control
operation of the front end circuit 123.
Receiver 102 further includes an analog television demodulator 134
having an input coupled to an output of digital filter 122 and an
output coupled to an input of an output interface 136. Receiver 102
also includes a digital video broadcast terrestrial/cable (DVB-T/C)
demodulator 138 having an input coupled to the output of digital
filter 122, an input coupled to an output of digital signal
processor (DSP) 144, and an output coupled to an input of equalizer
140. Equalizer 140 includes an input coupled to an output of DSP
144, and an output coupled to an input of a forward error
correction (FEC) circuit 142, which has a second input coupled to
an output of DSP 144 and an output coupled to an input of output
interface 136. DSP 144 includes a control input coupled to an
output of a control interface 146, which may be coupled to a host
system for receiving control and other signals. Control interface
146 also includes an output coupled to MCU 124.
Digital filter 122 provides the output signals to tuner circuitry,
including analog television demodulator 134 and DVB-T/C demodulator
138, which demodulate the video signal. Analog television
demodulator 134 provides the demodulated output to output interface
136, which may be coupled to an audio/video system on a chip or
other multimedia circuit. DVB-T/C demodulator 138 provides the
output to equalizer 140, which adjusts the relative strength of
selected frequencies within the demodulated output signal and
provides the adjusted signal to FEC 142. FEC 142 uses forward error
correction to correct signal errors and provides the corrected
signal to output interface 136.
In this example, tracking filter 110 includes a low-pass filter
portion for receiving and filtering VHF.sub.LO broadcast signals, a
second filter portion for receiving and filtering VHF.sub.HI
broadcast signals, and a third filter portion for receiving and
filtering UHF broadcast signals. In an example, the low-pass filter
portion includes a single-to-differential amplifier that includes
an inductor/capacitor bandpass filter at the outputs and uses
inductive peaking to extend the bandwidth range. The third portion
of the filter eliminates the single-to-differential amplifier and
replaces it with a transformer having a primary winding with a
relatively high quality factor (Q) and a secondary winding with a
much lower Q. The transformer can provide approximately a 3 decibel
(3 dB) gain. By introducing a gain between the primary and second
windings, any impact of limited linearity of the output of LNA 108
on overall system linearity is also reduced.
By adjusting the VHF.sub.LO and UHF filter portions, the overall
number of inductive-capacitive (LC) filters through the tracking
filter 110 can be reduced. In an example, the frequency range over
which the receiver 102 operates can be divided into an upper range,
a mid-range, and a low range of frequencies, thereby reducing the
overall complexity of the tracking filter.
In conventional tracking filters, the input impedance of each
filter is often designed to be large to minimize the noise figure.
To achieve the high tuned impedance, the size of the inductor is
increased, which limits the maximum tuned frequency for each
filter. Accordingly, a number of filters are included to provide
filtering across the frequency range. A representative example of a
receiver including a conventional tracking filter is described
below with respect to FIG. 2.
FIG. 2 is a circuit diagram of a representative example of a
receiver 200 including a conventional tracking filter 201 having a
separate bandpass filter tuned for each frequency band. Receiver
200 includes LNA 108 having an input for receiving a radio
frequency input signal (RF.sub.IN) and an output coupled to an
input of tracking filter 201.
Tracking filter 201 can include any number (N) of signal paths. In
the illustrated example includes a first signal path including an
inductor 202, a variable capacitor 204, a single-to-differential
amplifier 206, and a resistor 208. Inductor 202 includes a first
terminal coupled to the input of tracking filter 201 and a second
terminal coupled to ground. Capacitor 204 includes a first
electrode coupled to the input of tracking filter 201 and a second
electrode coupled to ground. Single-to-differential amplifier 206
includes an input coupled to the input of tracking filter 201, a
first output coupled to a first terminal of resistor 208 and a
second output coupled to a second terminal of resistor 208.
Tracking filter 201 further includes a second signal path including
an inductor 212, a variable capacitor 214, a single-to-differential
amplifier 216, and a resistor 218. Inductor 212 includes a first
terminal coupled to the input of tracking filter 201 and a second
terminal coupled to ground. Capacitor 214 includes a first
electrode coupled to the input of tracking filter 201 and a second
electrode coupled to ground. Single-to-differential amplifier 216
includes an input coupled to the input of tracking filter 201, a
first output coupled to a first terminal of resistor 218 and a
second output coupled to a second terminal of resistor 218.
Tracking filter 201 also includes an N-th signal path including an
inductor 222, a variable capacitor 224, a single-to-differential
amplifier 226, and a resistor 228. Inductor 222 includes a first
terminal coupled to the input of tracking filter 201 and a second
terminal coupled to ground. Capacitor 224 includes a first
electrode coupled to the input of tracking filter 201 and a second
electrode coupled to ground. Single-to-differential amplifier 226
includes an input coupled to the input of tracking filter 201, a
first output coupled to a first terminal of resistor 228 and a
second output coupled to a second terminal of resistor 228.
In an example, resistors 208, 218, and 228 are approximately ninety
ohm resistors and inductors 202, 212, and 222 have widths and
lengths of approximately 600 .mu.m.times.600 .mu.m. The input
impedances (R.sub.P1, R.sub.P2, . . . , and R.sub.PN) for each of
the signals paths (1 through N) is large enough to achieve a low
noise factor. For some receivers, the input impedances between 400
Ohms and 800 Ohms. For tuned inductive LC filters, the tuned
impedance is given by Rp=w.sub.0*L*Q where w.sub.0 is the resonant
frequency, L is the inductance, Q is the quality factor, and Rp is
the input impedance. For on-chip inductors, the quality factor (Q)
is substantially constant for a given area and Q is proportional to
the square root of the inductor area. To synthesize the desired
impedance, the size of inductors 202, 212, and 222 is increased to
improve Q or the inductance value is increased to avoid increased
inductor area. However, large inductance values limit the maximum
tuned frequency according to the following equation:
.omega..times..times. ##EQU00001## where the variable (C.sub.T)
represents a total parasitic capacitance. Unfortunately, this
limitation increases the number of filters that are included in
tracking filter 201 in order to provide the desired channel
tuning.
To reduce the number of signal paths and thus the number of LC
filters, tracking filter 110 of FIG. 1 extends each transfer
function to cover a wider frequency band, reducing the number of
signal paths and thus reducing the complexity of the circuit. An
example of a tracking filter that provides three transfer functions
is described below with respect to FIG. 3.
FIG. 3 is a block diagram of an embodiment of a portion of a
receiver 300 including a tracking filter 110 having a VHF.sub.LO
filter 316, a VHF.sub.HI filter 318, and a UHF filter 320. Receiver
300 includes LNA 108 having an input for receiving a radio
frequency input signal (RF.sub.IN) and an output coupled to a
source of a transistor 302. Transistor 302 includes a gate coupled
to an output of an MCU 304, and a drain coupled to an input of
tracking filter 110. Tracking filter 110 includes a transistor 306
having a source coupled to the input of tracking filter 110, a gate
coupled to MCU 304, and a drain coupled to an input of VHF.sub.LO
filter 316, which is implemented as a low pass filter. Tracking
filter 110 further includes a transistor 308 having a source
coupled to the input of tracking filter 110, a gate coupled to MCU
304, and a drain coupled to an input of VHF.sub.HI filter 318.
Tracking filter 110 also includes a transistor 310 having a source
coupled to the input of tracking filter 110, a gate coupled to MCU
304, and a drain coupled to an input of UHF filter 320.
During operation, MCU 304 selectively enables the signal path
between LNA 108 and one of the filters 316, 318, and 320. In an
example, MCU 304 activates transistors 302 and 306 to provide the
output of LNA 108 to the input of VHF.sub.LO filter 316. In this
instance, the VHF.sub.LO filter 316 is a low-pass filter that
passes frequencies within the tuning range from approximately 50
MHz to approximately 190 MHz. There are no television channels
below a frequency of 50 MHz, thus VHF.sub.LO filter 316 can be
configured to low pass filter operation. Toward the higher end of
the range (i.e., as the frequency approaches approximately 190 MHz,
the bandwidth of the VHF.sub.LO filter 316 is extended by using
inductive peaking to extend the tuning range. An example of the
VHF.sub.LO filter 316 illustrating one possible circuit
configuration is described below with respect to FIG. 4.
FIG. 4 is a diagram of an embodiment of a receiver 400, such as
receiver 300 in FIG. 3, including the VHF.sub.LO filter 316.
VHF.sub.LO filter 316 includes an inductor 402 and a resistor 403.
Resistor 403 has a first terminal coupled to the source of
transistor 306 and has a second terminal coupled to a first
terminal of inductor 402, which has a second terminal coupled to
ground. In this instance, resistor 403 represents the series
resistance of inductor 402. VHF.sub.LO filter 316 further includes
a variable capacitor 404 having a first electrode coupled to the
drain of transistor 306 and a second electrode coupled to ground.
VHF.sub.LO filter 316 also includes a single-to-differential
amplifier 406 having a first output and a second output. Inductors
408 and 409 are connected in series between the first and second
outputs. Additionally, a variable capacitor 410 includes a first
electrode coupled to the first output and a second electrode
coupled to the second output. The first and second outputs may be
coupled to a load, such as a ninety ohm resistance.
In this example, the input impedance of VHF.sub.LO filter 316 is
mostly provided by the series resistance of the inductor 402 at low
frequencies. At higher frequencies, inductor 402 extends the
bandwidth through inductive peaking. The quality factor
requirements are relaxed substantially making it possible to use a
smaller inductor. However, the smaller inductance and
correspondingly smaller input impedance sacrifices some rejection
at the higher and the lower side of the frequency band. To improve
rejection, VHF.sub.LO filter 316 includes an inductive/capacitive
(LC) bandpass filter at the outputs of the single-to-differential
amplifier 406 using inductors 408 and 409 and variable capacitor
410. Instead of using a single 600 .mu.m.times.600 .mu.m inductor,
VHF.sub.LOfilter 316 includes three 150 .mu.m.times.150 .mu.m
inductors, reducing the overall circuit area while extending the
frequency range of VHF.sub.LO filter 316 relative to filters in
tracking filter 201 in FIG. 2.
In the above example, VHF.sub.LO filter 316 from FIG. 3 uses an LC
filter on the output of the single-to-differential amplifier to
provide additional filtering. As the impedance level at the
single-to-differential amplifier output is much smaller (about 90
Ohms), small-size, low-Q inductors can be used to implement the
bandpass filter at the output of the single-to-differential
amplifier. Thus, a high Q bandpass filter with a very large
inductor is replaced by two lower-Q LC filters using much smaller
inductors resulting in a reduced overall inductor area. The UHF
filter 320 of FIG. 3 eliminates the active single-to-differential
amplifier, reducing power consumption and improving the noise
figure and linearity of the overall system. In place of the
single-to-differential amplifier, UHF filter 320 introduces a
transformer with a gain and with different Q factors between the
primary and secondary inductive windings. By introducing a gain
between the primary and secondary, the impact of the limited
linearity of LNA 108 on overall system linearity is reduced. An
example of a UHF filter 320 that uses a transformer instead of a
single-to-differential amplifier is described below with respect to
FIG. 5.
FIG. 5 is a diagram of an embodiment of a receiver 500 including
the UHF filter 320 of FIG. 3. In this example, UHF filter 320
includes a variable capacitor 502 and an transformer 504. Variable
capacitor 502 includes a first electrode coupled to the drain of
transistor 310 and a second electrode coupled to ground.
Transformer 504 includes a primary inductive winding 506 having a
first terminal coupled to the drain of transistor 310 and a second
terminal coupled to a supply voltage (such as a voltage
approximately midway between the supply voltage rails, labeled
"V.sub.MID"). Transformer 504 further includes a secondary
inductive winding 508 having a first terminal and a second
terminal, which can be coupled to mixer 112, and including a center
tap coupled to ground.
In the illustrated example, the primary inductive winding 506 has a
higher Q than the Q of the secondary inductive winding 508.
Additionally, the transformer 504 introduces a gain from the
primary inductive winding 506 to the secondary inductive winding
508. In one instance, the gain is approximately 3 dB. In an
example, the secondary inductive winding 508 of transformer 504 has
more turns than the primary inductive winding 506, providing the
gain.
In a particular embodiment, the primary inductive winding 506 is
formed from approximately 3.4 .mu.m of high quality copper (Cu),
such as Cu M8, and approximately 2.8 .mu.m of aluminum (Al). In
this embodiment, the secondary inductive winding 508 is formed from
lower quality metal, such as Cu M4 or M5 and each has approximately
0.2 .mu.m of Cu. The primary and secondary inductive windings 506
and 508 can be patterned in redistribution layers of the
semiconductor substrate. The secondary inductive winding 508 is
formed from lower metals to improve the self-resonant frequency
response of transformer 504. Further, the coupling coefficient is
very strong (approximately k=1). A small signal model of an example
of the UHF filter 320 is described below with respect to FIG.
6.
FIG. 6 is an embodiment of a small signal model 600 of the UHF
Filter of FIG. 5. Small signal model 600 includes an input voltage
terminal 602 for receive an input voltage (v.sub.in) and an output
voltage terminal 604 for providing an output voltage (v.sub.out).
Small signal model 600 includes variable capacitor 606 coupled
between input voltage terminal 602 and ground. Variable capacitor
606 may represent variable capacitor 502 in FIG. 5 as well as a
parasitic capacitance of the primary winding 506 of transformer 504
in FIG. 5. Small signal model 600 further includes an inductor 608
and a resistor 610 connected in series between input voltage
terminal 602 and ground. Secondary winding 508 of transformer 504
in FIG. 5 is modeled as an inductor 614 and a resistor 616
connected in series between output voltage terminal 604 and ground.
Additionally, a secondary capacitor 618 is coupled between output
voltage terminal 604 and ground.
Assuming that the primary inductive winding 506 is tuned to a
resonant frequency (.omega..sub.0) while the secondary inductive
winding 508 is not tuned, resonant frequency of the secondary
winding (.omega..sub.s) is much greater than the resonant frequency
(.omega..sub.0) as follows:
.omega..times..times..times. ##EQU00002## where L.sub.614
represents the inductance of inductor 614 and C.sub.618 represents
the capacitance of capacitor 618. The coupling between the primary
and the secondary inductive windings 506 and 508 is very strong (k
is approximately equal to 1). The current into the secondary
winding (i.sub.s) is determined according to the following
equation:
.times..times..omega..times..times..times..times..omega..times..times..ti-
mes..times..omega..times..times..times..times..omega..times..times..times.-
.omega..times..times..times..times..omega..times..times..times..times.
##EQU00003## wherein R.sub.616 represents the resistance of
resistor 616, the frequency (.omega.) represents the input
frequency, and the variable (M) is determined by the following
equation: M=k {square root over (L.sub.608L.sub.614)} (4)
Since k=1, for frequencies around the resonant frequency
(.omega..sub.0), the current into the secondary winding 508 can be
determined according to the following equation:
.times..omega..times..times..times..omega..times..times..times.
##EQU00004## where i.sub.s is the current flowing in the secondary
inductive winding 508 and i.sub.p is the current flowing in the
primary inductive winding 506.
In the illustrated example, the input voltage (v.sub.in) is
determined by the following equation:
.times..times..times..times..times..times..times..omega..times..times..ti-
mes..omega..times..times..times..omega..times..times..times..times.
##EQU00005## and the effective impedance (Z.sub.eff) looking into
the primary winding of the transformer is determined as
follows:
.times..times..omega..times..times..omega..times..times..times..times..om-
ega..times..times..times..times..omega..times..times..times.
##EQU00006##
In the equation 7, the factor j.omega.L.sub.608+R.sub.610 is the
impedance due to the primary inductive winding 506, and the
remainder of the equation is the impedance due to the secondary
inductive winding 508. Assuming that the factor
.omega.R.sub.616C.sub.618 is much smaller than 1, the effective
impedance (Z.sub.eff) simplifies as follows:
Z.sub.eff=j.omega.L.sub.608+R.sub.610+.omega..sup.2M.sup.2j.omega.C.sub.6-
18(1-j.omega.R.sub.616C.sub.618)=j.omega.(L.sub.608+.omega..sup.2M.sup.2C.-
sub.618)+R.sub.610+.omega..sup.4M.sup.2C.sub.618.sup.2R.sub.616.sup.2
(8)
The effective impedance due to the inductance (L.sub.eff) is
determined according to the following equation:
.omega..times..times..function..omega..times..times..function..omega..ome-
ga. ##EQU00007##
The effective impedance due to the resistance (R.sub.eff) is
determined according to the following equation:
.omega..times..times..times..omega..omega..times..times.
##EQU00008##
In the above examples, if the ratio of the resonant frequency
(.omega..sub.0) to the frequency (.omega..sub.s) is kept small, the
secondary inductive winding 508 has limited effect on the tuned
frequency of primary inductive winding 506. Additionally, by
keeping this ratio small, the effect of the secondary inductive
winding on the quality factor (Q) is also kept small. For example,
if the ratio is approximately 1/3.sup.rd, and the resistance
(R.sub.616) is approximately 10 times the resistance of resistor
610, and if the inductances of inductors 608 and 614 are
approximately equal, then the effective quality factor (Q.sub.eff)
can be determined by the following equation:
.apprxeq..times..omega..times..times. ##EQU00009##
Further, on the secondary side, the output voltage (v.sub.out)
relates to the input voltage (v.sub.in) according to the following
equation:
.times..times..times. ##EQU00010## Equation 12 demonstrates a
conversion gain in transformer 504. At higher frequencies, it can
be demonstrated that the secondary inductive winding 508 acts as a
second order inductive/capacitive (LC) low-pass filter on top of
the LC bandpass characteristics of the primary inductive winding
506.
Thus, the single-ended to differential conversion provided by a
single-to-differential amplifier in the prior art can be replaced
by performing the conversion magnetically with a transformer 504
having a secondary inductive winding 508 with a low quality factor.
The effect of the secondary inductive winding 508 on the quality
factor of the primary inductive winding 506 is proportional to the
fourth power of the frequency. Accordingly, the quality factor and
hence the gain of the LNA 108 decreases with increased tuned
frequency, which helps to reduce the gain variation in the transfer
function of the UHF filter 320. Further, as discussed, it is
possible to implement a conversion gain of more than 0 dB by
adjusting a number of turns on the secondary inductive winding 508
to provide the gain. By introducing such a gain, the transformer
504 operates to reduce the signal level at the output of LNA 108,
improving the in-band third-order intercept point (IIP3) of the LNA
108 by providing the gain in the transformer 504.
In conjunction with the circuits described above with respect to
FIGS. 1-6, a receiver is disclosed that includes a tracking filter
having a reduced number of LC filters relative to conventional
tracking filters for television reception. In particular, the
tracking filter divides the frequency range into three bands
(VHF.sub.LO, VHF.sub.HI, and UHF), and provides a low-pass filter
for the VHF.sub.LO frequencies, a bandpass filter for the
VHF.sub.HI frequencies, and a transformer-based bandpass filter for
UHF broadcast signals.
FIG. 7 is a block diagram of a low-cost television receiver 700
according to an embodiment of the present invention. Receiver 700
includes generally a low noise amplifier (LNA) 710, a switch
section 720, a filter section 730, a mixer and combiner 740, an
amplifier section 750, an analog-to-digital converter (ADC) section
760, a digital signal processor and demodulator section 770, and a
microcontroller 780.
LNA 710 has an input for receiving input signal RF.sub.IN, and an
output for providing an amplified RF.sub.IN signal. Switch section
720 includes three N-channel transistors 722, 724, and 726.
Transistor 722 has a first source/drain terminal connected to the
output of LNA 710, a gate, and a second source-drain terminal.
Transistor 724 has a first source/drain terminal connected to the
output of LNA 710, a gate, and a second source-drain terminal.
Transistor 726 has a first source/drain terminal connected to the
output of LNA 710, a gate, and a second source-drain terminal.
Filter section 730 includes a first filter labeled "VHF.sub.LO
FILTER" 732, a second filter labeled "VHF.sub.HI FILTER" 734, and a
third filter labeled "UHF FILTER" 736. Filter 732 has an input
connected to the second terminal of transistor 722, a tuning input,
and an output for providing a filtered differential signal. Filter
734 has an input connected to the second terminal of transistor
724, a tuning input, and an output for providing a filtered
differential signal. Filter 736 has an input connected to the
second terminal of transistor 726, a tuning input, and an output
for providing a filtered differential signal.
Mixer and combiner 740 has inputs connected to respective outputs
of filters 732, 734, and 736, a first output for providing a
differential in-phase signal labeled "I", and a second output for
providing a differential quadrature signal labeled "Q".
Amplifier section 750 includes programmable gain amplifiers (PGAs)
752 and 754. PGA 752 has a differential input coupled to the first
output of mixer and combiner 740, a gain control input, and a
differential output. PGA 754 has a differential input coupled to
the second output of mixer and combiner 740, a gain control input,
and a differential output.
ADC section 760 includes ADCs 762 and 764. ADC 762 has a
differential input connected to the output of PGA 752, and a
multi-bit digital output. ADC 764 has a differential input
connected to the output of PGA 754, and a multi-bit digital
output.
Digital signal processor and demodulator 770 has a first input
connected to the output of ADC 762, a second input connected to the
output of ADC 764, and an output for providing a signal labeled
"TV.sub.OUT".
MCU 780 has a first output connected to the gate of transistor 722,
a second output connected to the gate of transistor 724, a third
output connected to the gate of transistor 726, and a set of
outputs not specifically shown in FIG. 7 connected to various other
blocks as will be described further below.
In operation, receiver 700 is part of a highly integrated low-cost
television receiver. LNA 710 receives the RF.sub.IN signal from an
antenna or other signal source (not shown in FIG. 7), and provides
the amplified RF.sub.IN signal to a selected one of three signal
processing paths depending on the frequency band of the desired
channel. In response to a tuning input (not shown in FIG. 7), MCU
780 under the control of firmware selects the appropriate signal
processing path by activating one of the signals driving the gates
of transistors 722, 724, and 726. In receiver 700, if the desired
channel falls in the 50-190 MHz range, then MCU 780 activates the
signal to the gate of transistor 722 and receiver 700 forms
TV.sub.OUT from the amplified RF.sub.IN signal using VHF.sub.LO
filter 732. If the desired channel falls in the 190-470 MHz range,
then MCU 780 activates the signal to the gate of transistor 724 and
receiver 700 forms TV.sub.OUT from the amplified RF.sub.IN signal
using VHF.sub.HI filter 734. Filter 734 is a parallel resonant LC
filter and includes a single-ended to differential converter. If
the desired channel falls in the 470-860 MHz range, then MCU 780
activates the signal to the gate of transistor 726 and receiver 700
forms TV.sub.OUT from the amplified RF.sub.IN signal using UHF
filter 736. Filter 736 is implemented as described with respect to
FIGS. 5 and 6 above.
Receiver 700 is suitable for very low cost applications by using
several techniques. First, receiver 700 is fully integrated onto a
single integrated circuit chip without expensive external
components by using on-chip inductors now available in
state-of-the-art CMOS processes. Second, filter section 730 uses
only three bands instead of the five bands that receiver 200 of
FIG. 2 uses. Third, UHF filter 736 uses an on-chip transformer in
conjunction with a variable capacitor to accomplish both bandpass
filtering as well as single-ended to differential conversion, thus
saving circuit area required by an additional amplifier as well as
improving the dynamic range of receiver 700.
Fourth, VHF.sub.LO filter 732 uses a low-Q RLC filter and a low-Q
LC bandpass filter that are smaller in area than a conventional
parallel resonant LC filter, while providing most of the benefits
of such a filter. The construction and characteristics of this
filter will now be described.
FIG. 8 illustrates in schematic form VHF.sub.LO filter 732 of FIG.
7. Filter 732 includes generally an RLC lowpass filter 800, an
amplifier 830, and a bandpass filter 840. Filter 800 includes a
variable capacitor 810 and an inductance leg 820. Variable
capacitor 810 includes a fixed capacitor 812, a binary-coded
capacitor 814, and a thermometer-coded capacitor 816. Fixed
capacitor 812 has a first terminal for receiving the amplified
RF.sub.IN signal, and a second terminal connected to ground.
Binary-coded capacitor 814 has a first terminal for receiving the
amplified RF.sub.IN signal, a second terminal connected to ground,
and a tuning input terminal for receiving a binary-coded tuning
signal. Thermometer-coded capacitor 816 has a first terminal for
receiving the amplified RF.sub.IN signal, a second terminal
connected to ground, and a tuning input terminal for receiving a
thermometer-coded tuning signal. Inductance leg 820 includes a
resistor 822 and an inductor 824. Resistor 822 has a first terminal
for receiving the amplified RF.sub.IN signal, and a second
terminal. Inductor 824 has a first terminal connected to the second
terminal of resistor 822, and a second terminal connected to
V.sub.MID.
Amplifier 830 has a first input terminal for receiving the
amplified RF.sub.IN signal, a second input terminal connected to a
voltage labeled "V.sub.AG", and first and second output terminals
for providing a differential output signal pair. V.sub.AG is an
analog ground voltage that is nominally halfway between the
more-positive and ground.
Bandpass filter 840 includes a variable capacitor 850, and a
differential inductor 860. Variable capacitor 850 includes two
fixed capacitors 851 and 852, two binary-coded capacitors 853 and
844, and two thermometer-coded capacitors 855 and 856. Capacitor
851 has a first terminal connected to the first output terminal of
amplifier 850, and a second terminal connected to ground. Capacitor
852 has a first terminal connected to ground, and a second terminal
connected to the second output terminal of amplifier 830. Capacitor
853 has a first terminal connected to the first output terminal of
amplifier 830, a second terminal connected to ground, and a tuning
input terminal for receiving a binary-coded tuning signal.
Capacitor 854 has a first terminal connected to ground, a second
terminal connected to the second output terminal of amplifier 830,
and a tuning input terminal for receiving the binary-coded tuning
signal. Capacitor 855 has a first terminal connected to the first
output terminal of amplifier 830, a second terminal connected to
ground, and a tuning input terminal for receiving a
thermometer-coded tuning signal. Capacitor 856 has a first terminal
connected to ground, a second terminal connected to the second
output terminal of amplifier 830, and a tuning input terminal for
receiving a thermometer-coded tuning signal.
Differential inductor 860 includes inductors 862 and 864. Inductor
862 has a first terminal connected to the first output terminal of
amplifier 830, and a second terminal connected to V.sub.AG.
Inductor 864 has a first terminal connected to V.sub.AG, and a
second terminal connected to the second output terminal of
amplifier 830.
In operation, filter 732 operates as a band filter and single-ended
to differential converter for the amplified RF.sub.IN signal when
the user selects a channel in the VHF.sub.LO band from 50 MHz to
190 MHz. However instead of merely providing a variable cutoff
frequency within this band, lowpass filter 732 provides some
selectivity in a narrower frequency band centered around the
desired channel. Thus lowpass filter 732 provides some of the
benefit of a bandpass filter without requiring high-quality (and
thus large) on-chip inductors. At the same time, filter 732
significantly attenuates higher frequency components which helps
maintain good signal-to-noise ratio that might otherwise cause
parasitic mixing of unwanted energy into the passband.
The resonant frequency of filter 800 ("f.sub.R") is given by:
.times..times..pi..times..times..times..times. ##EQU00011## in
which L represents the inductance of inductor 824, C represents the
capacitance of variable capacitor 810, and R represents the
resistance of resistor 822. The 3 dB bandwidth around f.sub.R
("BW") is given by:
.times..times..times..times..pi. ##EQU00012##
TABLE I below shows the values of the various components used in
RLC filter 800 for the VHF.sub.LO filter band:
TABLE-US-00001 TABLE I Element Value Fixed capacitor 812 1.63 pF
Binary coded capacitor 814 15 .times. 9.4 fF Thermometer coded
capacitor 816 31 .times. 150 fF Resistor 822 396 .OMEGA. Inductor
824 465 nH
in which pF represents picoFarads, fF represents femtoFarads,
.OMEGA. represents ohms, and nH represents nanoHenrys. Note that
the 396.OMEGA. resistance of resistor 822 is formed by 345.OMEGA.
of parasitic resistance of inductor 824 along with a separate
54.OMEGA. resistor.
TABLE II below shows the values of the various components in
bandpass filter 840 used for the VHF.sub.LO filter band:
TABLE-US-00002 TABLE II Element Value Fixed capacitors 851 and 852
10 pF Binary coded capacitors 853 and 854 15 .times. 150 fF
Thermometer coded capacitors 855 and 856 31 .times. 2.4 pF
Parasitic resistance of inductors 862 and 864 28 .OMEGA. Inductors
862 and 864 86 nH
Using a currently available 0.55 micron CMOS manufacturing process
with copper metallization and eight available metal layers, in
VHF.sub.LO filter 732, inductor 824 can be formed in a die surface
area of 150 microns (.mu.m).times.150 .mu.m, and inductor 860 in a
die surface area of 280 .mu.m.times.160 .mu.m. However as noted
above the inductor in the VHF.sub.LO bandpass filter of receiver
200 of FIG. 2 requires a die surface area of approximately 600
.mu.m.times.600 .mu.m. Thus the lower-Q inductors used in
VHF.sub.LO filter 732 require only about 19% of the die surface
area of the higher-Q inductor required for the VHF.sub.LO bandpass
filter of FIG. 2. Filter 732 actually takes advantage of the
parasitic resistance in inductor 824 to reduce the size of the
other components for a given f.sub.R. Thus filter 732 achieves most
of the benefit of a parallel resonant bandpass filter but with
significantly reduced die area and helps receiver 700 achieve low
cost.
FIG. 9 illustrates a frequency domain graph useful in understanding
the operation of the VHF.sub.LO filter of FIG. 7. In FIG. 9, the
vertical axis represents gain in decibels (dB), whereas the
horizontal axis represents frequency on a logarithmic scale in Hz.
A first waveform 910 corresponds to the frequency response of RLC
filter 800 and is relatively flat but with a small dip of about 2-3
dB below f.sub.R, exhibits a relatively constant dropoff with
increasing frequency for frequencies above f.sub.R, and has a peak
at resonant frequency f.sub.R. The peak is small for f.sub.R near
the low end of the VHF.sub.LO band, but increases to about 6 dB for
f.sub.R at the high end of the VHF.sub.LO band (the case
illustrated in FIG. 9).
A second waveform 920 corresponds to the frequency response of
bandpass filter 840, and is substantially flat for lower
frequencies below f.sub.R, exhibits a relatively constant dropoff
with increasing frequency for frequencies above f.sub.R, and has a
peak of about 4-6 dB at resonant frequency f.sub.R. The peak is
around 4 dB for f.sub.R near the low end of the VHF.sub.LO band,
and increases to about 6 dB for f.sub.R at the high end of the
VHF.sub.LO band.
A third waveform 930 corresponds to the overall frequency response
to VHF.sub.LO filter 732, and is relatively flat with a small dip
for frequencies below f.sub.R, exhibits a relatively constant
dropoff with increasing frequency for frequencies above f.sub.R,
and has a peak of 4-10 dB at resonant frequency f.sub.R. The peak
is around 4 dB for f.sub.R near the low end of the VHF.sub.LO band,
and increases to about 10 dB for f.sub.R at the high end of the
VHF.sub.LO band. Thus the addition of bandpass filter 840 ensures
some selectivity for frequencies near f.sub.R for frequencies at
the low end of the VHF.sub.LO band.
Thus receiver 700 is highly integrated and achieves low cost by
using on-chip inductors. The size of the inductor in the VHF.sub.LO
band can be made small by implementing a single-ended filter as an
RLC filter in which the resistance is formed in part by the
parasitic resistance of an on-chip inductor, and in part by a
separate integrated resistor. Receiver 700 adds an additional
parallel resonant LC filter bandpass with a slightly higher quality
inductor to provide additional selectivity in the passband.
The above-disclosed subject matter is to be considered
illustrative, and not restrictive, and the appended claims are
intended to cover all such modifications, enhancements, and other
embodiments that fall within the true scope of the claims. Thus, to
the maximum extent allowed by law, the scope of the present
invention is to be determined by the broadest permissible
interpretation of the following claims and their equivalents, and
shall not be restricted or limited by the foregoing detailed
description.
* * * * *