U.S. patent number 7,859,450 [Application Number 12/200,024] was granted by the patent office on 2010-12-28 for detection and ranging appartus and detection and ranging method.
This patent grant is currently assigned to Fujitsu Limited, Fujitsu Ten Limited. Invention is credited to Yasuyuki Kondo, Katsuyuki Ohguchi, Kazuo Shirakawa.
United States Patent |
7,859,450 |
Shirakawa , et al. |
December 28, 2010 |
Detection and ranging appartus and detection and ranging method
Abstract
In a detection and ranging apparatus that performs
direction-of-arrival estimation using a sensor array and that
enlarges an effective aperture using a plurality of transmitting
sensors, adverse effects associated with time division switching
are eliminated, achieving high-accuracy measurement. A transmitter
wave is spread in modulators by using mutually orthogonal codes,
and the resulting transmitter waves are radiated from two
transmitting sensors. Signals received by receiving sensors are
each split by a splitter into two parts, which are then
respectively despread in a demodulator by using the same codes as
those used in the transmitter.
Inventors: |
Shirakawa; Kazuo (Kawasaki,
JP), Ohguchi; Katsuyuki (Hyogo, JP), Kondo;
Yasuyuki (Hyogo, JP) |
Assignee: |
Fujitsu Limited (Kawasaki,
JP)
Fujitsu Ten Limited (Kobe, JP)
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Family
ID: |
40010661 |
Appl.
No.: |
12/200,024 |
Filed: |
August 28, 2008 |
Prior Publication Data
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Document
Identifier |
Publication Date |
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US 20090079617 A1 |
Mar 26, 2009 |
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Foreign Application Priority Data
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Sep 26, 2007 [JP] |
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2007-249822 |
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Current U.S.
Class: |
342/147; 342/70;
375/130 |
Current CPC
Class: |
G01S
13/325 (20130101); G01S 13/87 (20130101); G01S
13/34 (20130101); G01S 7/032 (20130101); G01S
2013/0254 (20130101); G01S 13/931 (20130101) |
Current International
Class: |
G01S
13/00 (20060101); H04B 1/00 (20060101) |
Field of
Search: |
;342/70-72,118,128-139,147 ;340/901-904,435-437
;375/130,135,138,146,269,279,308 |
References Cited
[Referenced By]
U.S. Patent Documents
Foreign Patent Documents
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20 24 557 |
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Jan 1980 |
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GB |
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2024557 |
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Jan 1980 |
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GB |
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2000155171 |
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Jun 2000 |
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JP |
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2001237755 |
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Aug 2001 |
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JP |
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2004264067 |
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Sep 2004 |
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JP |
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2005257384 |
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Sep 2005 |
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JP |
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200629858 |
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Feb 2006 |
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JP |
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200698181 |
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Apr 2006 |
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JP |
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2007-199085 |
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Aug 2007 |
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JP |
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Other References
European Search Report with Written Opinion and the Annex to the
European Search Report on European Patent Application No.
EP08163216.8-2220/2045612. Dated Mar. 17, 2009. cited by other
.
Replacement European Search Report with written opinion for
corresponding European Patent Application No. 08163216.8 issued by
the European Patent Office on Oct. 28, 2009. cited by other .
Kees N. et al.; "Improvement of Angular Resolution of a
Millimeterwave Imaging System by Transmitter Location
Multiplexing"; IEEE MTT-S International Orlando, FL IEEE, US; pp.
969-972; XP010141425; May 16, 1995; [Ref.: Replacement ERS dated
Oct. 28, 2009]. cited by other.
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Primary Examiner: Tarcza; Thomas H
Assistant Examiner: Bythrow; Peter
Attorney, Agent or Firm: Myers Wolin, LLC
Claims
The invention claimed is:
1. A detection and ranging apparatus comprising: spreaders for
generating spread transmitter waves; transmitting sensor elements
for transmitting the spread transmitter waves; receiving sensor
elements for receiving reflected transmitter waves from targets;
wherein said spreaders are configured as M spreaders for receiving
transmitter waves from an oscillator and generating M spread
transmitter waves by spreading the transmitter waves using mutually
orthogonal M orthogonal codes, where M is an integer not smaller
than 2; said transmitting sensor elements are configured as M
transmitter sensors for transmitting out said M spread transmitter
waves; said receiving sensor elements are configured as N receiving
sensor elements separately from the transmitting sensor elements,
where N is an integer not smaller than 2, and enlarge an effective
aperture thereof by a combination with said transmitting sensor
elements; a receiving control part having units for receiving, from
the N receiving sensor elements, the N signals of the reflected
transmitter waves which have transmitted from the M transmitter
sensor elements at the same time, demodulating by a demodulating
unit the received N signals, wherein said receiving control part is
operative to split by splitting unit each of the N received signals
into M parts and generate M.times.N despread outputs by despreading
at despreading means the M parts using the M orthogonal codes given
from a code generator so that the MxN despread outputs are
processed in parallel; a direction-of-arrival estimating part for
estimating, based on said M.times.N despread outputs, directions of
arrival of the reflected transmitter waves from the targets; and a
processor for applying a control input to the oscillator for
switching the oscillator output to operate the detection and
ranging apparatus as at least one of FM-CW radar and SS radar.
2. The detection and ranging apparatus according to claim 1,
wherein said M spreaders each include a BPSK modulator for
spreading said transmitter wave by binary phase-shift keying (BPSK)
said transmitter wave with a corresponding one of said orthogonal
codes.
3. The detection and ranging apparatus according to claim 1,
further comprising a transmitter wave generator for selectively
generating one of a carrier wave frequency-modulated by a
triangular wave and an unmodulated carrier wave, as said
transmitter wave.
4. The detection and ranging apparatus according to claim 1,
wherein said N receiving sensor elements are arranged at equally
spaced intervals, and said M transmitting sensor elements are two
transmitting sensor elements arranged so as to flank said N
receiving sensor elements.
5. The detection and ranging apparatus according to claim 4,
further comprising a phase shifter for adjusting the phase of a
radio wave to be radiated from at least one of said two
transmitting sensor elements.
6. The detection and ranging apparatus according to claim 1,
wherein said N receiving sensor elements are arranged at equally
spaced intervals, and said M transmitting sensor elements are
arranged one spaced apart from another by a distance that is N
times as great as the distance by which said N receiving sensor
elements are spaced apart.
7. A detection and ranging method of detecting a direction of
arrival of a signal, comprising: generating M spread transmitter
waves by receiving transmitter waves from an oscillator and
spreading the transmitter waves using mutually orthogonal M
orthogonal codes at M spreaders, where M is an integer not smaller
than 2; transmitting out said M spread transmitter waves from M
transmitter sensors; receiving, at N receiving sensor elements, the
N received signals of the reflected transmitter waves which have
transmitted from the M transmitter sensors at the same time, while
enlarging an effective aperture of the receiving antenna by the
combination with the M transmitting antennas; demodulating the N
received signals; splitting each of the N received signals into M
parts; generating M.times.N despread outputs by dispreading the M
parts using said M orthogonal codes; processing the M.times.N
despread outputs in parallel; estimating, based on said MxN
despread outputs, a direction of arrival of the reflected
transmitter waves from targets; and controlling said oscillator to
operate the detection and ranging apparatus as at least one of
FM-CW radar and SS radar.
8. A detection and ranging apparatus comprising: M spreaders for
generating M spread transmitter waves by spreading a transmitter
wave using mutually orthogonal M orthogonal codes, where M is an
integer not smaller than 2; M transmitting sensor elements for
transmitting out said M spread transmitter waves; N receiving
sensor elements, where N is an integer not smaller than 2; a
receiving control part for splitting each of N received signals
obtained at said N receiving sensor elements into M parts, and for
generating M.times.N despread outputs by dispreading the M parts
using said M orthogonal codes; and a direction-of-arrival
estimating part for estimating, based on said M.times.N despread
outputs, directions of arrival of reflected signals arriving from a
plurality of targets, wherein said M orthogonal codes includes a
first orthogonal code and a second orthogonal code having a higher
chip rate than said first orthogonal code, and at least the
transmitter wave spread by said first orthogonal code is a carrier
wave frequency-modulated by a triangular wave, said apparatus
further comprising a first distance calculating part for taking as
an input the despread output produced by despreading with said
first orthogonal code, and for calculating a distance to a target
from the frequency of said despread output in an upsweep section of
said triangular wave and the frequency of said despread output in a
downsweep section of said triangular wave, and a second distance
calculating part for taking as an input the despread output
produced by despreading with said second orthogonal code, and for
calculating the distance to said target from the phase of said
second orthogonal code used for said despreading.
9. The detection and ranging apparatus according to claim 8,
further comprising: a correcting part for correcting the distance
calculated by said first distance calculating part, based on a
difference between the distances calculated by said first and
second distance calculating parts for the same target.
Description
BACKGROUND OF THE INVENTION
1. Field of the Invention
The present invention relates to a detection and ranging apparatus
and detection and ranging method having a function for estimating
the direction of arrival of a signal by using a sensor array and a
direction-of-arrival estimation method.
2. Description of the Related Art
As an example of such a detection and ranging apparatus, consider a
radar apparatus capable of estimating all three quantities, i.e.,
relative distance, relative velocity, and angular direction of a
target. In a radar, a sensor refers to an antenna. FM-CW radar is a
type of radar that projects forward a transmitter wave
frequency-modulated by a triangular wave or the like, and receives
a reflected wave from a target, mixes it in a mixer with a portion
of the transmitted signal to obtain a baseband signal containing
information on the target, and calculates the distance and relative
velocity of the target from the baseband signal, and this type of
radar is commercially implemented as automotive radar, etc.
One possible method for determining the direction in which the
target is located is to use an array antenna having a plurality of
antenna elements to receive the reflected wave from the target, and
to determine the direction of the target by applying a
direction-of-arrival estimation method such as a known beam former
method to the received signal.
In a direction-of-arrival estimation method using an array antenna,
for example, in the beam former method that scans the main lobe of
the array antenna in a given direction and determines the direction
where the output power is the greatest as being the direction of
arrival, the beam width of the main lobe determines the angular
resolution, and therefore, if it is desired to increase the
resolution so that the directions of many targets can be
determined, the aperture length of the array must be increased by
increasing the number of antenna elements. The same can be said of
the minimum norm (Min-Norm) method that determines the direction of
arrival from the eigenvalue and eigenvector of the correlation
matrix of the array's received signal, and its extended algorithms
such as MUSIC (MUltiple SIgnal Classification) and ESPRIT
(Estimation of Signal Parameters via Rotational Invariance
Techniques), i.e. in these methods also, since the dimension of the
correlation matrix, i.e., the number of antenna elements,
determines the number of targets that can be detected, the number
of antenna elements must be increased in order to be able to
determine the directions of many targets.
However, in the case of a radar apparatus such as an automotive
radar where severe constraints are imposed on the mounting
dimensions of the antenna, it has been difficult to increase the
number of antenna elements without compromising reception
power.
JP 2006-98181A and JP 2000-155171A each propose a method that
enlarges the effective aperture by using a plurality of
transmitting antennas.
In this case, for each received reflected wave, the transmitting
antenna that transmitted the original radiowave must be identified.
In the above patent documents, each one of the plurality of
transmitting antennas is selected for use in time division fashion
by using a switch so that the transmitting antenna can be
identified for each received wave. Further, on the receiver side
each one of the plurality of receiving antennas is selected in time
division fashion by using a switch, in order to reduce the cost by
reducing the amount of RF circuitry.
However, switching from one antenna to another using a switch
involves adverse effects such as degradation in signal and
reduction in detection range. Furthermore, since it does not follow
that the waves transmitted out from the plurality of transmitting
antennas and returned by reflection are received simultaneously by
the plurality of receiving antennas, there arises the problem that
a time shift and a phase shift occur. This problem can be
alleviated by increasing the switching speed, but there is
naturally a limit to it.
Further, in JP 2001-237755A, which relates to a different field
than the present invention, i.e., to a so-called phase monopulse
scheme in which radiowaves transmitted from two antennas at a base
station in a mobile communication system are received by one
antenna at a mobile station to determine the direction based on the
difference between the arriving phases, it is described that the
radiowaves are spread using mutually orthogonal spreading codes so
that the transmitting antennas that transmitted the respective
radiowaves can be discriminated at the receiving end.
SUMMARY OF THE INVENTION
Accordingly, it is an object of the present invention to provide a
detection and ranging apparatus that uses a plurality of
transmitting sensors in order to enlarge the effective aperture of
a sensor array having a plurality of sensor elements, wherein
provisions are made to eliminate the adverse effects associated
with the time division switching and thereby achieve a
high-accuracy measurement.
The above object is achieved by a detection and ranging apparatus
comprising: M spreaders for generating M spread transmitter waves
by spreading a transmitter wave using mutually orthogonal M
orthogonal codes, where M is an integer not smaller than 2; M
transmitting sensor elements for transmitting out the M spread
transmitter waves; N receiving sensor elements, where N is an
integer not smaller than 2; receiving control means for splitting
each of N received signals obtained at the N receiving sensor
elements into M parts, and for generating M.times.N despread
outputs by despreading the split received signals using the M
orthogonal codes; and direction-of-arrival estimating means for
estimating, based on the M.times.N despread outputs, directions of
arrival of reflected signals arriving from a plurality of
targets.
The above object is also achieved by a detection and ranging method
for detecting a direction of arrival of a signal, wherein signals
to be input to M transmitting sensor elements are respectively
spread by mutually orthogonal M orthogonal codes and transmitted
out simultaneously from the M transmitting sensor elements, where M
is an integer not smaller than 2, and signals output from N
receiving sensor elements are each split into M parts, which are
then respectively despread using the M orthogonal codes.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a block diagram of a radar apparatus according to one
embodiment of the present invention;
FIG. 2 is a diagram showing one configuration example of a
modulator 16 (16.sub.1, 16.sub.2) in FIG. 1;
FIG. 3 is a diagram showing one configuration example of a
demodulator 26 (26.sub.11 to 26.sub.2N) in FIG. 1;
FIG. 4 is a diagram explaining the operation of the apparatus of
FIG. 1;
FIG. 5 is a schematic diagram showing output signals of respective
ports in FIG. 4;
FIG. 6 is a block diagram of a radar apparatus according to
according to one modified example; and
FIG. 7 is a block diagram showing an alternative example of antenna
configuration.
DESCRIPTION OF THE PREFERRED EMBODIMENTS
FIG. 1 shows the configuration of a radar apparatus according to an
embodiment. The example shown in FIG. 1, similarly to the one shown
in JP 2006-98181A, uses a receiving array antenna comprising N
antenna elements A.sub.R1 to A.sub.RN and a transmitting array
antenna comprising two (M=2) antenna elements A.sub.T1 and A.sub.T2
arranged on both sides of the receiving array antenna.
An oscillator module 10 includes an oscillator 12 for generating a
baseband signal such as a triangular wave and a voltage-controlled
RF (Radio Frequency) oscillator 14 whose frequency is controlled by
the output of the oscillator 12, and generates a transmitter wave
frequency-modulated by the triangular wave. A code generator 20
generates mutually orthogonal two PN codes 1 and 2. The transmitter
wave frequency-modulated by the triangular wave is directly spread
by binary phase-shift keying (BPSK) using the PN codes 1 and 2 in
modulators 16.sub.1 and 16.sub.2, and the resulting transmitter
waves are fed via power amplifiers 18.sub.1 and 18.sub.2 to the
antennas A.sub.T1 and A.sub.T2 for transmission.
N signals received by the antenna elements A.sub.R1 to A.sub.RN are
fed via low-noise amplifiers 20.sub.1 to 20.sub.N to mixers
22.sub.1 to 22.sub.N where each signal is downconverted using the
transmitter wave before spreading that is output from the
oscillator module 10. The outputs of the N mixers 22.sub.1 to
22.sub.N are each split by a corresponding one of splitters
24.sub.1 to 24.sub.N into two parts, and are despread in 2N
demodulators 26.sub.11 to 26.sub.2N by using the same PN codes 1
and 2 as those used at the transmitter end. The 2N despread results
from the demodulators 26.sub.11 to 26.sub.2N are converted by A/D
converters 28.sub.11 to 28.sub.2N into digital signals which are
input to a signal processing unit 30.
As will be described later, reflected waves of the transmitter wave
transmitted from the antenna element A.sub.T1 are received by the
antenna elements A.sub.R1 to A.sub.RN and downconverted by the
mixers 22.sub.1 to 22.sub.N, and the demodulators 26.sub.11 to
26.sub.1N despread them by the code 1; this means that a baseband
signal of a conventional radar having a transmitting antenna at the
position of A.sub.T1 is output from the demodulators 26.sub.11 to
26.sub.1N. Similarly, reflected waves of the transmitter wave
transmitted from the antenna element A.sub.T2 are received by the
antenna elements A.sub.R1 to A.sub.RN and downconverted by the
mixers 22.sub.1 to 22.sub.N, and the demodulators 26.sub.21 to
26.sub.2N despread them by the code 2; this means that a baseband
signal of a conventional radar having a transmitting antenna at the
position of A.sub.T2 is output from the demodulators 26.sub.21 to
26.sub.2N. Accordingly, the result when reflected waves of the
transmitter wave transmitted from the antenna element A.sub.T1 are
received by the antenna elements A.sub.R1 to A.sub.RN and the
result when reflected waves of the transmitter wave transmitted
from the antenna element A.sub.T2 are received by the antenna
elements A.sub.R1 to A.sub.RN are obtained at the same time and
without interruption.
The signal processing unit 30 applies FFT (Fast Fourier
Transformation) to these signals and calculates the distance and
the relative velocity with respect to a target from the peak
frequencies obtained by FFT in the upsweep and downsweep sections
of the triangular wave. Further, the direction of each target is
determined by applying a direction-of-arrival estimation method
such as the beam former method at a target presence indicating
frequency position obtained by FFT. If it is necessary to estimate
the number of targets before the direction estimation, a
number-of-targets estimation method based on such as AIC (Akaike's
Information Criterion) is used as the preprocessing.
FIG. 2 shows one configuration example of the modulator 16.sub.1,
16.sub.2. The high-frequency signal fed via a transmission line 40
is split by a T-shaped transmission line 42 into two parts, and one
is inverted in sign by an inverter 44 and thus shifted in phase by
180.degree. and is thereafter supplied to a transmission line 46,
while the other is supplied directly to a transmission line 56. If
the DC potential of the transmission line 46 is at a high level,
diodes 48, 50, 52, and 54 are forward biased, and the transmission
line 46 is put in a short-circuited condition, thus blocking the
transmission of the high-frequency wave, but if the DC potential is
at a low level, the diodes 48, 50, 52, and 54 are reverse biased,
and the transmission line 46 is put in a through-line condition,
thus allowing the high-frequency wave to transmit therethrough. The
same principle applies to the transmission line 56.
The logic value (for example, 0 or 1) S of the PN code generated by
the code generator is converted to a suitable value as a bias
voltage and applied as the DC potential to the transmission line
46, and the inverted logic value is applied to the transmission
line 56. Accordingly, either the high-frequency signal whose phase
is not shifted or the high-frequency signal whose phase is shifted
by 180.degree. is selected for output according to the value of the
PN code.
FIG. 3 shows one configuration example of the demodulator 26
(26.sub.11 to 26.sub.2N). The received signal downconverted by the
mixer 22 (22.sub.1 to 22.sub.N) using the transmitter wave before
spreading is input to the demodulator 26. In other words, the
so-called beat signal as produced in conventional FM-CW radar and
BPSK-modulated by the PN code is input here. This signal is
multiplied in a multiplier 58 by a PN code as a .+-.1 bipolar
signal, and the product is integrated over a period Tf by an
integrator 60. The integration period Tf is set equal to the chip
duration of the PN code multiplied by the code length, and the chip
duration and the code length are chosen so that the integration
period Tf becomes sufficiently shorter than the period of the beat
signal. It is desirable that, to reduce the influence of the
filtering effect by the code such as described later, the chip rate
(the reciprocal of the chip duration) of the PN code be made
greater than the maximum delay time to be measured, in particular
when the main objective is to enlarge the aperture, and that the
chip rate be set to a value sufficiently smaller than the width of
the triangular wave frequency modulation in order to prevent the
bandwidth of the transmitter wave from being expanded by the
spreading to such an extent that interference is caused to other
apparatus.
In the demodulator, since the integrator 60 is a component part for
computing correlation for each code component contained in the
incoming signal, if the PN code applied to the multiplier 58
matches one of PN codes contained in the incoming signal, and if
the phase of the code also matches between them, the output of the
integrator 60 for this code component takes a maximum value 1 (when
normalized to non-spread signal power), while the output for the
code component, if the phase does not match, is -1/(code length)
(of course, generally 0 in the case of an orthogonal code). In view
of this, the PN code is scanned by a variable delay device 62 by
changing the phase of the PN code at time intervals of one chip
(actually, it is changed at intervals of about 1/3 chip, but to
simplify explanation, it is assumed to be changed at intervals of
one chip), and when a correlation value exceeding a suitably set
threshold value is detected by a decision-making device 64 provided
at the output of the integrator 60, the variable delay device 62
stops the scanning, thus accomplishing synchronization acquisition.
A counter 66 counts the number of pulses from the variable delay
device 62, and outputs a delay index indicating the amount of delay
introduced by the variable delay device 62; as will be described
later, the distance to the target can also be calculated from the
delay index after the synchronization acquisition. Once the
synchronization is established, a switch 65 is turned on by a
control signal from the decision-making device 64, and the beat
signal before being spread by the PN code is output from the
demodulator.
Since the N paralleled demodulators 26 are used for the same code,
the time required to accomplish synchronization acquisition can be
reduced to 1/N if the initial values of the delay amounts are set
to 0, Tf/N, 2Tf/N, . . . , (N-1)Tf/N, respectively, where Tf
represents the scan range (usually, the integration interval).
As an alternative configuration example of the demodulator, a
matched filter, SAW convolver, etc., may be used. Of course, a
series/parallel synchronization circuit, a DLL (Delay Locked Loop),
or the like may be used as the synchronization acquisition
circuit.
By constructing the transmitter IC from a GaAs HEMT, HBT, or like
device that achieves high output power and the receiver IC from a
CMOS or like device that has low output power but is well matched
to the digital signal processing system that follows, the overall
performance of the apparatus can be enhanced.
FIG. 4 is a diagram showing the essential configuration extracted
from FIG. 1. The transmitting and receiving antenna elements are
arranged along the X axis in a rectangular coordinate system, and
the receiving array antenna comprises the N antenna elements
A.sub.R1 to A.sub.RN one spaced apart from another by a distance d
along the X axis in the positive direction thereof starting from
the origin, while the transmitting array antenna comprises the two
(M=2) antenna elements A.sub.T1 and A.sub.T2 located outwardly of
the outermost receiving antenna elements.
Turning back to FIG. 1, the system reference signal generated by
the oscillator module 10 is denoted by V.sub.S(t), the modulating
signal for A.sub.T1, generated by the code generator 20 that
generates the mutually orthogonal codes, is denoted by V.sub.C1(t),
and the modulating signal for A.sub.T2 likewise generated is
denoted by V.sub.C2(t). Further, for convenience, it is assumed
that V.sub.C(t).ident.V.sub.C1(t)+V.sub.C2(t); then, probe signals
from the respective transmitting antenna elements are given as
V.sub.T1(t)=V.sub.S(t)V.sub.C1(t) and
V.sub.T2(t)=V.sub.S(t)V.sub.C2(t), respectively. If PN code
sequences [C.sub.1.sup.1, . . . , C.sub.Q1.sup.1] and
[C.sub.1.sup.2, . . . , C.sub.Q2.sup.2] of code lengths Q1 and Q2,
for example, are assigned as the orthogonal codes to A.sub.T1 and
A.sub.T2, respectively, V.sub.C1(t) and V.sub.C2(t) are
specifically expressed as shown below in relation to the pulse
waveform p(t) that carries the code and the pulse chip durations
T.sub.C1 and T.sub.C2.
.times..times..function..infin..infin..times..function..times..times..tim-
es..function..times..times..function..infin..infin..times..function..times-
..times..times..function..times..times. ##EQU00001##
Referring again to FIG. 4, if there are L independent targets
within the detection range of the apparatus, and if the m-th target
(m=1 to L) is located at a relative line-of-sight distance d.sub.m
and at an angle .theta..sub.m (with the positive part of the y-axis
as the starting point, angles measured in the clockwise direction
are taken as positive), then the RF demodulated echo signal,
X.sub.m(t), is expressed as shown below. The, .tau..sub.m is the
delay time, which is given as .tau..sub.m=2d.sub.m/C.sub.0, where
C.sub.0 is the velocity of light.
x.sub.m(t)=v.sub.S(t)v.sub.S*(t-.tau..sub.m)v.sub.C*(t-.tau..sub.m)
(3)
Since the phase difference of X.sub.m(t) arriving at the k-th
receiving antenna element A.sub.Rk (k=1 to N) relative to
X.sub.m(t) arriving at A.sub.R1 is expressed as
.PHI..times..pi..lamda..times..times..times..times..function..theta.
##EQU00002## the baseband signal, V.sub.1k(t), obtained by
demodulating the output of A.sub.Rk by the PN code signal
V.sub.C1(t) assigned to A.sub.T1, is expressed by the following
equation together with a noise signal n.sub.k(t).
.times..function..times..times..function..times..function..times..functio-
n..PHI..function. ##EQU00003##
For simplicity, noting only the process for demodulating the m-th
signal component in the above equation by the PN code, the
demodulation process is expressed as
.times..times..function..times..function..tau..times..times..times..funct-
ion..times..times..times..function..tau..times..times..function..times..ti-
mes..times..function..tau..times..infin..infin..times..infin..infin..times-
..function..times..times..times..times..times..times..times..times..functi-
on..times..times..times..function..times..times..tau..times..infin..infin.-
.times..infin..infin..times..function..times..times..times..times..times..-
times..times..times..function..times..times..times..function..times..times-
..tau. ##EQU00004## Since the demodulator in FIG. 1 has the
configuration shown, for example, in FIG. 3, and performs the
integration by multiplying the incoming signal with the
demodulating PN code signal while varying the amount of shift,
xT.sub.C1, of the PN code signal, if the transmitted code signal
and the demodulating code signal, including its delayed version,
are in phase, the first term on the right-hand side of the equation
(6) is, for example, 1 for each integration interval (if out of
phase, -1/(code length)). On the other hand, since {C.sup.1} and
{C.sup.2} are orthogonal to each other, the second term on the
right-hand side of the equation (6) is always 0.
Then, setting n.sub.k(t).ident.V.sub.C(t)n.sub.k(t) and
X.sub.m(t).ident.V.sub.S(t)V*.sub.S(t-.tau..sub.m), the baseband
signal associated with A.sub.T1, after demodulation, is given
as
.times..function..times..function..times..function..PHI..function.
##EQU00005## and the spatial phase of vector
V.sub.1(t)=[V.sub.11(t), . . . ,V.sub.1N(t)].sup.T constructed by
arranging these baseband signals forms the equiphase surface 1 in
FIG. 4.
Likewise, the phase difference of X.sub.m(t) arriving at A.sub.Rk
relative to X.sub.m(t) arriving at A.sub.RN, the reference antenna
in this case being spaced (N-1)d away from the reference antenna in
the above case, is expressed as
.PHI..times..pi..lamda..function..times..times..times..function..theta..t-
imes..pi..lamda..times..times..times..times..function..theta.
##EQU00006## Therefore, similarly to the above case, the baseband
signal, V.sub.2k(t), obtained by demodulating the output of
A.sub.Rk by the PN code signal V.sub.C2(t) assigned to A.sub.T2, is
given as
.times..function..times..function..times..function..PHI..function.
##EQU00007## and the spatial phase of V.sub.2(t)=[V.sub.21(t), . .
. , V.sub.2N(t)].sup.T constructed by arranging these baseband
signals forms the equiphase surface 2 in FIG. 4.
Accordingly, if the variation of the target angle that occurs
during the demodulation with the PN code is sufficiently small, and
if the system is stable to the shifting of the phase origin (to
maintain the similarity of electromagnetic coupling, etc., the
reference position of the receiving antenna element with respect to
the transmitting antenna element is rotationally symmetrical), then
by the synthetic aperture using the extended signal vector defined
by the following equation, the present invention can achieve an
effective aperture of 2Nd with the physical aperture of Nd for the
time interval longer than the time required for all the 2N
demodulators 26 to accomplish synchronization acquisition.
v(t)=[v.sub.1(t), v.sub.2(t)].sup.T (10)
FIG. 5 is a timing chart schematically illustrating the output
signal of each port, for example, when the signal from the
oscillator module 10 is a carrier wave frequency-modulated by a
triangular wave; here, it can be seen that an array signal vector
having an aperture of 2Nd is obtained during one period T.sub.FM of
the FM modulation input.
The amount of delay indicated by the delay index output from the
counter 66 in the demodulator 26 shown in FIG. 3, after the
synchronization acquisition, corresponds to the distance to the
target. In other words, when the delay index output from the
counter 66 is denoted by m, the chip duration of the PN code by Tc,
the distance to the target by d, and the velocity of light by c,
since the relation mT.sub.C=2d/c holds, the distance to the target
can also be calculated from d=cmT.sub.C/2 Accordingly, when
employing FM-CW as the basic system, if T.sub.C is made too small,
the determination of the distance by the PN code is done first,
that is, the signal from a target located at a specific distance is
selected and passed to the subsequent processing stage (stated
another way, this is equivalent to filtering the incoming signal
based on the distance). Therefore, if the main objective is to
enlarge the antenna aperture, it is desirable to set T.sub.C not
shorter than the maximum delay time to be measured (which is
determined by the maximum detection range). In other words, while
the main object to be achieved by the present invention is to
enlarge the antenna aperture in real time by multiplexing the
physical aperture in code space, the filtering effect which may be
an attendant effect is also an essential feature of the present
invention.
From another standpoint, if the chip rate of the PN code is
increased, the distance can be measured with good accuracy, but on
the other hand, the bandwidth of the transmitter wave also expands,
which is disadvantageous when it comes to measuring distances at
long range where interference with other apparatus' becomes a
problem. In view of this, preferably, the same apparatus can be
used as SS (Spread Spectrum) radar which measures the distance to
the target by the same method as described above by stopping the
triangular modulation. In this case, relative velocity can also be
measured by determining the Doppler frequency by applying FFT as in
the case of FM-CW radar.
For example, if the control input to the voltage-controlled RF
oscillator 14 in the oscillator module 10 in FIG. 1 is switched
suitably or in time division fashion under instruction from the CPU
31 in FIG. 1, the apparatus can be used not only as FM-CW radar but
also as SS radar. In this case, if the chip rate of the PN code
being generated by the code generator 20 is switched in
synchronized fashion, it becomes possible to use the apparatus as
FM-CW radar to measure targets at long range and as SS radar to
measure targets at short range with high accuracy.
Further, if the oscillator module 10 is made to select the
transmitter wave frequency-modulated by the triangular wave, and
the code generator 20 is made to generate the PN codes for the
respective modulators 16.sub.1 and 16.sub.2 by changing the chip
rate between them (of course, while maintaining orthogonality
between them), the range measuring by FM-CW radar and the range
measuring by SS radar can be performed at the same time, though the
effective aperture remains at Nd.
When switching the operation between FM-CW radar and SS radar in
time division fashion, or when simultaneously operating FM-CW radar
and SS radar, the distances, d.sub.FM and d.sub.SS, measured for
the same target by the respective radars can be used to calculate
the range measuring error .delta.d of the lower accuracy FM-CW
radar, i.e., .delta.d=d.sub.FM-d.sub.SS and d.sub.FM can be
corrected using d.sub.SS.
For example, if the vehicle traveling ahead of the
apparatus-equipped vehicle is radiating backward a radar wave
spread by the same code as the code used in the apparatus-equipped
vehicle, the radar wave will be directly picked up by the receiving
antenna elements A.sub.R1 to A.sub.RN, causing interference and
resulting in an inability to make accurate measurement. If this
happens, the transmitting antenna elements A.sub.T1 and A.sub.T2
are caused to stop transmitting, and the outputs of the
demodulators 26.sub.11 to 26.sub.2N are checked to determine the
code that the vehicle traveling head is using; then, by changing
the code used in the apparatus-equipped vehicle by an instruction
from the CPU 31, the interference can be avoided.
FIG. 6 shows the configuration of a radar apparatus according to a
modified example of the radar apparatus shown in FIG. 1. The same
component elements as those in FIG. 1 are designated by the same
reference numerals, and the description thereof will not be
repeated here.
A phase shifter 70 is provided between the modulator 16.sub.2 and
the power amplifier 18.sub.2 (or between the modulator 16.sub.1 and
the power amplifier 18.sub.1). In a normal mode in which the
direction of the target is determined by using the method described
thus far, the phase shifter 70 is rendered inoperative with its
phase shift amount set to zero, but is made operative in a tracking
mode in which target tracking is performed by directing the
transmit beam to the target after the direction of the target has
been determined in the normal mode. In the tracking mode, the phase
shift amount .phi. necessary to direct the transmit beam in the
direction .theta. of the target to be tracked is determined based
on the target's direction .theta. determined in the normal mode,
and the CPU 31 sets the phase shifter 70 accordingly. With the two
antenna elements outputting transmitter waves shifted in phase
difference by .phi., the transmitter waves are directed in the
specific direction .theta. thus making it easier to track the
target of interest. After the tracking is started, the phase shift
amount .phi. is updated using the target's direction .theta.
estimated from the signals received by the antenna elements
A.sub.R1 to A.sub.RN, and the updated phase shift amount .phi. is
fed back to the phase shifter 70 under the control of the CPU 31.
It is desirable that the switching between the normal mode and the
tracking mode be performed by calculating the degree of danger
according to the velocity and position of the target.
In the array antenna configuration shown in FIGS. 1 and 4 in which
the receiving antenna elements are arranged at equally spaced
intervals with two transmitting antenna elements placed on both
sides thereof, the received data associated with one transmitting
antenna and the received data associated with the other
transmitting antenna are in a rotational invariance relationship
with respect to each other (i.e., they can be regarded as one
linear antenna). Therefore, the apparatus of the present invention
can be used advantageously when super resolution angle measurement
techniques such as the ESPRIT algorithm using the rotational
invariance relationship are applied to the direction-of-arrival
estimation in the signal processing unit 30.
However, the present invention is not limited to the antenna
configuration described above, but can also be applied to an
antenna configuration such as described in JP 2000-155171A in
which, as shown in FIG. 7, a plurality of transmitting antenna
elements A.sub.T1 to A.sub.TM (M=3 in the figure) are placed on one
side of the array of receiving antenna elements A.sub.R1 to
A.sub.RN. In this configuration, by spacing the M transmitting
antenna elements A.sub.T1 to A.sub.TM apart by a distance d.sub.T
that is N times as great as the distance d.sub.R by which the
receiving antenna elements are spaced apart, that is, by setting
d.sub.T=Nd.sub.R, the effective aperture of the receiving antenna
can be increased by M times.
In FIG. 7, the code generator 20 generates mutually orthogonal M PN
codes (M=3 in the figure) and supplies them to M modulators
16.sub.1 to 16.sub.M. The signals received by the receiving antenna
elements A.sub.R1 to A.sub.RN are respectively split by splitters
24.sub.1 to 24.sub.N into M parts, and supplied to demodulators
26.sub.11 to 26.sub.MN (M=3 in the figure).
* * * * *