U.S. patent number 7,711,329 [Application Number 10/792,171] was granted by the patent office on 2010-05-04 for adaptive filter for transmit leakage signal rejection.
This patent grant is currently assigned to QUALCOMM, Incorporated. Invention is credited to Vladimir Aparin, Gary John Ballantyne, Charles J. Persico.
United States Patent |
7,711,329 |
Aparin , et al. |
May 4, 2010 |
Adaptive filter for transmit leakage signal rejection
Abstract
An adaptive filter suitable for fabrication on an RF integrated
circuit and used for transmit (TX) leakage rejection in a wireless
full-duplex communication system is described. The adaptive filter
includes a summer and an adaptive estimator. The summer receives an
input signal having a TX leakage signal and an estimator signal
having an estimate of the TX leakage signal, subtracts the
estimator signal from the input signal, and provides an output
signal having the TX leakage signal attenuated. The adaptive
estimator receives the output signal and a reference signal having
a version of the transmit signal, estimates the TX leakage signal
in the input signal based on the output signal and the reference
signal, and provides the estimator signal. The adaptive estimator
may utilize an LMS algorithm to minimize a mean square error
between the TX leakage signal in the input signal and the TX
leakage signal estimate in the estimator signal.
Inventors: |
Aparin; Vladimir (San Diego,
CA), Ballantyne; Gary John (Christchurch, NZ),
Persico; Charles J. (Rancho Santa Fe, CA) |
Assignee: |
QUALCOMM, Incorporated (San
Diego, CA)
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Family
ID: |
34576850 |
Appl.
No.: |
10/792,171 |
Filed: |
March 2, 2004 |
Prior Publication Data
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Document
Identifier |
Publication Date |
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US 20050107051 A1 |
May 19, 2005 |
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Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
Issue Date |
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60519561 |
Nov 12, 2003 |
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Current U.S.
Class: |
455/83;
455/556.1; 455/24 |
Current CPC
Class: |
H04B
1/525 (20130101) |
Current International
Class: |
H04B
1/44 (20060101) |
Field of
Search: |
;455/83,115.1,295,296,310,311,556.1,26,63.1,278.1,24 |
References Cited
[Referenced By]
U.S. Patent Documents
Foreign Patent Documents
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2001-047138 |
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Jun 2001 |
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KR |
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0154290 |
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Jul 2001 |
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WO |
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Other References
International Search Report--PCT/US04/038343, International Search
Authority European Patent Office, Mar. 4, 2005. cited by other
.
Written Opinion PCT/US2004/038343, International Search Authority
European Patent Office May 15, 2007. cited by other .
International Preliminary Report on
Patentability--PCT/US04/038343--IPEA/US--Feb. 9, 2009. cited by
other.
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Primary Examiner: Maung; Nay A
Assistant Examiner: Nguyen; Tuan H
Attorney, Agent or Firm: Loomis; Timothy F. Mobarhan;
Ramin
Parent Case Text
This application claims the benefit of provisional U.S. Application
Ser. No. 60/519,561, entitled "Adaptive Filtering of TX Leakage in
CDMA Receivers," filed Nov. 12, 2003.
Claims
What is claimed is:
1. An integrated circuit comprising: a summer operative to receive
an input signal having a transmit leakage signal and to receive an
estimator signal having an estimate of the transmit leakage signal,
to subtract the estimator signal from the input signal, and to
provide an output signal having the transmit leakage signal
attenuated, wherein the transmit leakage signal corresponds to a
portion of a modulated signal being transmitted in a wireless
full-duplex communication system; and an estimator operative to
receive the output signal and a reference signal having a version
of the modulated signal, to estimate the transmit leakage signal in
the input signal based on the output signal and the reference
signal, and to provide the estimator signal having the estimate of
the transmit leakage signal.
2. The integrated circuit of claim 1, further comprising: a low
noise amplifier (LNA) operative to amplify a receiver input signal
and provide the input signal.
3. The integrated circuit of claim 1, further comprising: a low
noise amplifier (LNA) operative to amplify the output signal and
provide an amplified signal for frequency downconversion to
baseband.
4. The integrated circuit of claim 1, further comprising: a mixer
operative to frequency downconvert the output signal with a local
oscillator (LO) signal and provide a downconverted signal.
5. The integrated circuit of claim 1, wherein the estimator
utilizes a least mean squared (LMS) algorithm to minimize a mean
square error (MSE) between the transmit leakage signal in the input
signal and the estimate of the transmit leakage signal in the
estimator signal.
6. The integrated circuit of claim 1, wherein the estimator
comprises a first multiplier operative to multiply the output
signal with an in-phase reference signal and to provide a first
in-phase signal, a first integrator operative to integrate the
first in-phase signal and to provide a second in-phase signal, a
second multiplier operative to multiply the second in-phase signal
with the in-phase reference signal or a quadrature reference signal
and to provide a third in-phase signal, wherein the in-phase and
quadrature reference signals are generated from the reference
signal, a third multiplier operative to multiply the output signal
with the quadrature reference signal and to provide a first
quadrature signal, a second integrator operative to integrate the
first quadrature signal and to provide a second quadrature signal,
and a fourth multiplier operative to multiply the second quadrature
signal with the in-phase or quadrature reference signal and to
provide a third quadrature signal, and wherein the estimator signal
is obtained by summing the third in-phase signal and the third
quadrature signal.
7. The integrated circuit of claim 6, further comprising: a
quadrature splitter operative to receive the reference signal and
provide the in-phase reference signal and the quadrature reference
signal.
8. The integrated circuit of claim 6, wherein the second multiplier
is operative to multiply the second in-phase signal with the
quadrature reference signal, and wherein the fourth multiplier is
operative to multiply the second quadrature signal with the
in-phase reference signal.
9. The integrated circuit of claim 6, wherein the second multiplier
is operative to multiply the second in-phase signal with the
inphase reference signal, and wherein the fourth multiplier is
operative to multiply the second quadrature signal with the
quadrature reference signal.
10. The integrated circuit of claim 6, wherein the first, second,
third, and fourth multipliers are implemented with mixers, and
wherein the in-phase and quadrature reference signals are used as
local oscillator (LO) signals for the mixers.
11. The integrated circuit of claim 6, wherein the estimator
further comprises a first lowpass filter coupled between the first
integrator and the second multiplier, and a second lowpass filter
coupled between the second integrator and the fourth
multiplier.
12. The integrated circuit of claim 11, wherein the first and
second lowpass filters are single-pole lowpass filters.
13. The integrated circuit of claim 6, further comprising: switches
operable to reset outputs of the first and second integrators prior
to enabling the estimator.
14. The integrated circuit of claim 6, wherein the first through
fourth multipliers and the first and second integrators are
implemented with a differential circuit design.
15. The integrated circuit of claim 1, wherein the estimator is
operable to derive a set of weight values based on a training
burst, and to use the set of weight values to estimate the transmit
leakage signal in the input signal.
16. The integrated circuit of claim 1, wherein the estimator
provides at least 30 dB of rejection of the transmit leakage
signal.
17. A wireless device in a wireless full-duplex communication
system, comprising: a low noise amplifier (LNA) operative to
amplify a receiver input signal and to provide an input signal
having a transmit leakage signal, wherein the transmit leakage
signal corresponds to a portion of a modulated signal being
transmitted; an adaptive filter operative to receive the input
signal and a reference signal having a version of the modulated
signal, to generate an estimator signal having an estimate of the
transmit leakage signal based on an output signal and the reference
signal, and to subtract the estimator signal from the input signal
to obtain the output signal having the transmit leakage signal
attenuated; and a mixer operative to receive and frequency
downconvert the output signal with a local oscillator (LO) signal
and to provide a downconverted signal.
18. The wireless device of claim 17, wherein the wireless
full-duplex communication system is a Code Division Multiple Access
(CDMA) system.
19. The wireless device of claim 17, wherein the adaptive filter
utilizes a least mean squared (LMS) algorithm to minimize a mean
square error (MSE) between the transmit leakage signal in the input
signal and the estimate of the transmit leakage signal in the
estimator signal.
20. The wireless device of claim 17, wherein the adaptive filter
comprises a first multiplier operative to multiply the output
signal with an in-phase reference signal and to provide a first
in-phase signal, a first integrator operative to integrate the
first in-phase signal and to provide a second in-phase signal, a
second multiplier operative to multiply the second in-phase signal
with the in-phase reference signal or a quadrature reference signal
and to provide a third in-phase signal, wherein the in-phase and
quadrature reference signals are generated from the reference
signal, a third multiplier operative to multiply the output signal
with the quadrature reference signal and to provide a first
quadrature signal, a second integrator operative to integrate the
first quadrature signal and to provide a second quadrature signal,
a fourth multiplier operative to multiply the second quadrature
signal with the in-phase or quadrature reference signal and to
provide a third quadrature signal, and wherein the estimator signal
is obtained by summing the third in-phase signal and the third
quadrature signal, and a summer operative to subtract the estimator
signal from the input signal and to provide the output signal.
21. An apparatus in a wireless full-duplex communication system,
comprising: means for subtracting an estimator signal from an input
signal and providing an output signal, the input signal having a
transmit leakage signal, the estimator signal having an estimate of
the transmit leakage signal, and the output signal having the
transmit leakage signal attenuated, wherein the transmit leakage
signal corresponds to a portion of a modulated signal being
transmitted; and means for estimating the transmit leakage signal
in the input signal based on the output signal and a reference
signal and providing the estimator signal, the reference signal
having a version of the modulated signal.
22. The apparatus of claim 21, wherein transmit leakage signal in
the input signal is estimated based on a least mean squared (LMS)
algorithm to minimize a mean square error (MSE) between the
transmit leakage signal in the input signal and the estimate of the
transmit leakage signal.
23. The apparatus of claim 21, wherein the means for estimating the
transmit leakage signal in the input signal comprises means for
multiplying the output signal with an in-phase reference signal to
obtain a first in-phase signal, means for integrating the first
in-phase signal to obtain a second in-phase signal, means for
multiplying the second in-phase signal with the in-phase reference
signal or a quadrature reference signal to obtain a third in-phase
signal, wherein the in-phase and quadrature reference signals are
generated from the reference signal, means for multiplying the
output signal with the quadrature reference signal to obtain a
first quadrature signal, means for integrating the first quadrature
signal to obtain a second quadrature signal, means for multiplying
the second quadrature signal with the in-phase or quadrature
reference signal to obtain a third quadrature signal, and means for
summing the third in-phase signal and the third quadrature signal
to obtain the estimator signal.
24. The apparatus of claim 23, wherein the means for estimating the
transmit leakage signal in the input signal farther comprises means
for filtering the second in-phase signal to obtain a filtered
second in-phase signal, and wherein the filtered second in-phase
signal is multiplied with the in-phase or quadrature reference
signal to obtain the third in-phase signal, and means for filtering
the second quadrature signal to obtain a filtered second quadrature
signal, and wherein the filtered second quadrature signal is
multiplied with the in-phase or quadrature reference signal to
obtain the third quadrature signal.
25. The apparatus of claim 23, further comprising: means for
resetting the second in-phase signal and the second quadrature
signal to known values.
26. A method of suppressing transmit leakage signal in a wireless
full-duplex communication system, comprising: subtracting an
estimator signal from an input signal to obtain an output signal,
the input signal having a transmit leakage signal, the estimator
signal having an estimate of the transmit leakage signal, and the
output signal having the transmit leakage signal attenuated,
wherein the transmit leakage signal is a portion of a modulated
signal being transmitted; and estimating the transmit leakage
signal in the input signal based on the output signal and a
reference signal having a version of the modulated signal and
providing the estimator signal having the estimate of the transmit
leakage signal.
27. The method of claim 26, wherein transmit leakage signal in the
input signal is estimated based on a least mean squared (LMS)
algorithm to minimize a mean square error (MSE) between the
transmit leakage signal in the input signal and the estimate of the
transmit leakage signal.
28. The method of claim 26, wherein the estimating the transmit
leakage signal comprises multiplying the output signal with an
in-phase reference signal to obtain a first in-phase signal,
integrating the first in-phase signal to obtain a second in-phase
signal, multiplying the second in-phase signal with the in-phase
reference signal or a quadrature reference signal to obtain a third
in-phase signal, wherein the in-phase and quadrature reference
signals are generated from the reference signal, multiplying the
output signal with the quadrature reference signal to obtain a
first quadrature signal, integrating the first quadrature signal to
obtain a second quadrature signal, multiplying the second
quadrature signal with the in-phase or quadrature reference signal
to obtain a third quadrature signal, and summing the third in-phase
signal and the third quadrature signal to obtain the estimator
signal.
Description
BACKGROUND
I. Field
The present invention relates generally to electronics, and more
specifically to techniques for mitigating the deleterious effects
of a transmit (TX) leakage signal in a wireless full-duplex
communication system.
II. Background
A wireless device in a wireless full-duplex communication system
can simultaneously transmit and receive data for two-way
communication. One such full-duplex system is a Code Division
Multiple Access (CDMA) system. On the transmit path, a transmitter
within the wireless device (1) modulates data onto a radio
frequency (RF) carrier signal to generate an RF modulated signal
and (2) amplifies the RF modulated signal to obtain a transmit
signal having the proper signal level. The transmit signal is
routed via a duplexer and transmitted from an antenna to one or
more base stations. On the receive path, a receiver within the
wireless device (1) obtains a received signal via the antenna and
duplexer and (2) amplifies, filters, and frequency downconverts the
received signal to obtain baseband signals, which are further
processed to recover data transmitted by the base station(s).
For a full-duplex wireless device, the RF circuitry in the receiver
is often subjected to interference from the transmitter. For
example, a portion of the transmit signal typically leaks from the
duplexer to the receiver, and the leaked signal (which is commonly
referred to as a "TX leakage" signal or a "TX feed-through" signal)
may cause interference to a desired signal within the received
signal. Since the transmit signal and the desired signal typically
reside in two different frequency bands, the TX leakage signal can
normally be filtered out and does not pose a problem in itself.
However, the TX leakage signal may interact with a "jammer" (which
is a large amplitude undesired signal close in frequency to the
desired signal) to generate "cross modulation" distortion
components on both sides of the jammer, as described below.
Distortion components that fall within the signal band of the
desired signal and which are not filtered out act as additional
noise that may degrade performance.
A surface acoustic wave (SAW) filter is often used to filter out
the TX leakage signal and mitigate its deleterious effects. The use
of a SAW filter for TX leakage rejection is undesirable for several
reasons. First, the SAW filter is normally a discrete component
that is not fabricated on an RF integrated circuit (RFIC) and thus
occupies space on a circuit board. Second, the SAW filter typically
requires other discrete components for input and output impedance
matching. Third, the SAW filter and its impedance matching
circuitry increase the cost of the wireless device.
There is therefore a need in the art for techniques to mitigate the
deleterious effects of a TX leakage signal without using a SAW
filter.
SUMMARY
Adaptive filters that can attenuate a TX leakage signal in wireless
full-duplex communication systems (e.g., a CDMA system) are
described herein. An adaptive filter may be fabricated on an RFIC,
along with other circuit blocks for a receiver such as a low noise
amplifier (LNA) for amplification, a mixer for frequency
downconversion, and so on. The adaptive filters can avoid the
disadvantages described above for SAW filters.
In an embodiment, an adaptive filter suitable for use for TX
leakage rejection includes a summer and an adaptive estimator. The
summer receives an input signal having TX leakage signal and an
estimator signal having an estimate of the TX leakage signal,
subtracts the estimator signal from the input signal, and provides
an output signal having the TX leakage signal attenuated. The
adaptive estimator receives the output signal and a reference
signal having a portion or version of the signal being transmitted,
estimates the TX leakage signal in the input signal based on the
output signal and the reference signal, and provides the estimator
signal having the TX leakage signal estimate.
The adaptive estimator may utilize a least mean squared (LMS)
algorithm to minimize a mean square error (MSE) between the TX
leakage signal in the input signal and the TX leakage signal
estimate in the estimator signal. In this case, the adaptive
estimator may include (1) a first multiplier that multiplies the
output signal with an in-phase reference signal and provides a
first in-phase signal, (2) a first integrator that integrates the
first in-phase signal and provides a second in-phase signal, (3) a
second multiplier that multiplies the second in-phase signal with
either the in-phase reference signal or a quadrature reference
signal and provides a third in-phase signal, (4) a third multiplier
that multiplies the output signal with the quadrature reference
signal and provides a first quadrature signal, (5) a second
integrator that integrates the first quadrature signal and provides
a second quadrature signal, and (6) a fourth multiplier that
multiplies the second quadrature signal with either the in-phase or
quadrature reference signal and provides a third quadrature signal,
and (7) a summer that sums the third in-phase signal and the third
quadrature signal and provides the estimator signal. The
multipliers may be implemented with mixers. The adaptive estimator
may further include other circuit blocks/elements for improved
performance, as described below. A quadrature splitter receives the
reference signal and provides the in-phase and quadrature reference
signals for the adaptive estimator.
Various aspects and embodiments of the invention are described in
further detail below.
BRIEF DESCRIPTION OF THE DRAWINGS
The features and nature of the present invention will become more
apparent from the detailed description set forth below when taken
in conjunction with the drawings in which like reference characters
identify correspondingly throughout and wherein:
FIG. 1 shows an RF portion of a wireless device;
FIGS. 2A through 2C show signals at various points in a receiver
within the wireless device;
FIG. 3 shows an implementation of the receiver with an RF SAW
filter;
FIG. 4 shows the RF portion of a wireless device with an adaptive
filter for TX leakage rejection;
FIGS. 5 and 6 show two embodiments of the adaptive filter;
FIGS. 7 and 8 show two more detailed embodiments of the adaptive
filter;
FIG. 9 shows a simplified model of the adaptive filter for
stability analysis;
FIG. 10 shows the frequency responses of the adaptive filter for
three different damping factors; and
FIGS. 11A through 11D show the generation of cross modulation
distortion due to double mixing within the adaptive filter.
DETAILED DESCRIPTION
The word "exemplary" is used herein to mean "serving as an example,
instance, or illustration." Any embodiment or design described
herein as "exemplary" is not necessarily to be construed as
preferred or advantageous over other embodiments or designs.
The adaptive filters described herein may be used for various
wireless full-duplex communication systems. These adaptive filters
may also be used for various frequency bands such as a cellular
band from 824 to 894 MHz, a Personal Communication System (PCS)
band from 1850 to 1990 MHz, a Digital Cellular System (DCS) band
from 1710 to 1880 MHz, an International Mobile
Telecommunications-2000 (IMT-2000) band from 1920 to 2170 MHz, and
so on. For clarity, the following description is for the cellular
band, which includes (1) an uplink frequency band from 824 to 849
MHz and (2) a downlink frequency band from 869 to 894 MHz. The
uplink and downlink frequency bands are transmit (TX) and receive
(RX) frequency bands, respectively, for a wireless device.
FIG. 1 shows a block diagram of an RF portion of a wireless device
100. On the transmit path, a power amplifier (PA) 112 within a
transmitter 110 receives and amplifies a TX modulated signal and
provides a transmit signal. The transmit signal is routed through a
duplexer 116 and transmitted via an antenna 118 to one or more base
stations. A portion of the transmit signal also couples or leaks
through duplexer 116 to the receive path. The amount of TX leakage
is dependent on the isolation between the transmit and receive
ports of the duplexer, which may be approximately 50 dB for a SAW
duplexer at the cellular band. A lower TX-RX isolation results in
higher level of TX leakage.
On the receive path, a received signal containing a desired signal
and possibly a jammer is received via antenna 118, routed through
duplexer 116, and provided to an LNA 122 within a receiver 120. LNA
122 also receives a TX leakage signal from the transmit path,
amplifies the receiver input signal at its input, and provides an
amplified RF signal, x(t). A filter 130 receives and filters the
amplified RF signal to remove out of band signal components (e.g.,
the TX leakage signal) and provides a filtered RF signal, y(t). A
mixer 132 receives and frequency downconverts the filtered RF
signal with a local oscillator (LO) signal and provides a
downconverted signal.
FIG. 2A shows the received signal, which includes a desired signal
212 and a jammer 214. Jammer 214 is an undesired signal and may
correspond to, for example, a signal transmitted by a nearby base
station in an Advanced Mobile Phone Service (AMPS) system. The
jammer may have an amplitude that is much higher than that of the
desired signal and may be located close in frequency to the desired
signal.
FIG. 2B shows the signal at the input of LNA 122. This signal
contains desired signal 212 and jammer 214 in the received signal
as well as a TX leakage signal 216 from the transmit path. The TX
leakage signal may have a large amplitude relative to the desired
signal because the transmit signal is often much larger in
amplitude than the desired signal.
FIG. 2C shows the signal at the output of mixer 132. Non-linearity
in LNA 122 and mixer 132 can cause the modulation on TX leakage
signal 216 to be transferred to (narrowband) jammer 214, which then
results in a widened spectrum 218 around the jammer. This spectral
widening is referred to as cross modulation and is described in
detail below. As shown in FIG. 2C, a portion 220 of widened
spectrum 218 (which is shown with shading) may fall within the
desired signal band. Portion 220 acts as additional noise that
degrades the performance of the wireless device. This noise further
degrades the receiver sensitivity so that the smallest desired
signal that can be reliably detected by the receiver needs to have
a larger amplitude.
FIG. 3 shows a schematic diagram of a conventional implementation
of receiver 120 with an RF SAW filter 330. SAW filter 330 has
various desirable characteristics such as sharp transition band
edges and large attenuation of out-of-band signal components. SAW
filter 330 is often used to reject the TX leakage signal at the
input of mixer 132, which then reduces the amount of cross
modulation distortion generated by the mixer.
LNA 122 couples to SAW filter 330 via an input impedance matching
network 310 formed by a resistor 312, inductors 314 and 316, and a
capacitor 318. SAW filter 330 couples to mixer 132 via an output
impedance matching network 340 formed by capacitors 342, 344, and
346 and inductors 348 and 350. A capacitor 320 provides filtering
of the power supply, VCC, for LNA 122.
The use of an RF SAW filter for TX leakage signal filtering has
several disadvantages. First, if LNA 122 and mixer 132 are
implemented within a single RFIC for reduced cost and improved
reliability, then SAW filter 330 is implemented off-chip and
requires three IC package pins for interface to the LNA and mixer.
Second, SAW filter 330 and the discrete components for matching
networks 310 and 340 require extra board space and further add cost
to the wireless device. Third, the insertion losses of SAW filter
330 and matching networks 310 and 340 degrade the cascaded gain and
noise figure of the receiver.
An adaptive filter may be used to reject the TX leakage signal and
avoid the disadvantages described above for the SAW filter. The
adaptive filter may be implemented on an RFIC (e.g., the same RFIC
used for the LNA and mixer) so that no additional board space is
needed for external components and cost is reduced. The adaptive
filter may be designed to achieve the desired rejection of the TX
leakage signal and consume low power.
FIG. 4 shows a block diagram of the RF portion of a wireless device
400 with an adaptive filter 430 for TX leakage rejection. On the
transmit path, a TX modulated signal is amplified by a power
amplifier 412 within a transmitter 410, routed through a duplexer
416, and transmitted via an antenna 418 to one or more base
stations. A coupler 414 receives the transmit signal from power
amplifier 412 and provides a portion of this transmit signal as a
reference signal, r(t).
On the receive path, a received signal is received via antenna 418,
routed through duplexer 416, and provided to an LNA 422 within a
receiver 420. LNA 422 also receives the TX leakage signal from the
transmit path, amplifies the signal at its input, and provides an
amplified RF signal, x(t). Adaptive filter 430 receives and filters
the amplified RF signal to attenuate/reject the TX leakage signal
and provides a filtered RF signal, y(t). A mixer 432 frequency
downconverts the filtered RF signal with an LO signal and provides
a downconverted signal.
In general, adaptive filter 430 may be located at any point on the
received path prior to mixer 432. For example, adaptive filter 430
may be placed either before or after LNA 422. Improved noise
performance can typically be achieved with adaptive filter 430
placed after LNA 422.
FIG. 5 shows a block diagram of an adaptive filter 430a, which is
an embodiment of adaptive filter 430 within receiver 420. Adaptive
filter 430a generates an estimate of the TX leakage signal, e(t),
based on the r(t) reference signal and further subtracts the TX
leakage signal estimate from the x(t) signal to obtain the y(t)
signal for mixer 432. The x(t) signal is also referred to as the
filter input signal, and the y(t) signal is also referred to as the
filter output signal.
For the embodiment shown in FIG. 5, adaptive filter 430a utilizes
an LMS algorithm to minimize the mean square error between the TX
leakage signal in the filter input signal and the TX leakage signal
estimate. Adaptive filter 430a includes a quadrature splitter 508,
an LMS adaptive estimator 510a, and a summer 540. Quadrature
splitter 508 receives the reference signal, r(t), and provides an
in-phase reference signal, i(t), and a quadrature reference signal,
q(t). The i(t) and q(t) signals respectively contain the in-phase
and quadrature components of the reference signal, with the i(t)
signal leading the q(t) signal by 90.degree..
LMS estimator 510a includes an in-phase section 520a, a quadrature
section 520b, and a summer 530. Within in-phase section 520a, a
multiplier 522a receives and multiplies the i(t) signal with the
y(t) signal and provides an m.sub.i(t) signal, which is
m.sub.i(t)=y(t)i(t). An integrator 524a receives and integrates the
m.sub.i(t) signal and provides an in-phase integrated signal,
w.sub.i(t). A multiplier 528a receives and multiplies the i(t)
signal with the w.sub.i(t) signal and provides a z.sub.i(t) signal,
which is z.sub.i(t)=w.sub.i(t)i(t). Similarly, within quadrature
section 520b, a multiplier 522b receives and multiplies the q(t)
signal with the y(t) signal and provides an m.sub.q(t) signal,
which is m.sub.q(t)=y(t)q(t). An integrator 524b receives and
integrates the m.sub.q(t) signal and provides a quadrature
integrated signal, w.sub.q(t). A multiplier 528b receives and
multiplies the q(t) signal with the w.sub.q(t) signal and provides
a z.sub.q(t) signal, which is z.sub.q(t)=w.sub.q(t)q(t).
Multipliers 522a, 522b, 528a, and 528b are four-quadrant
multipliers. Summer 530 receives and sums the z.sub.i(t) and
z.sub.q(t) signals and provides an estimator signal, e(t), which
contains the TX leakage signal estimate obtained based on the LMS
algorithm. The w.sub.i(t) and w.sub.q(t) signals are effectively
weights used for estimating the TX leakage signal.
Summer 540 receives the estimator signal, e(t), from LMS estimator
510a and the filter input signal, x(t), which contains the received
signal as well as the TX leakage signal. Summer 540 subtracts the
estimator signal from the filter input signal and provides the
filter output signal, y(t).
For the LMS algorithm, the estimator signal from LMS estimator 510a
may be expressed as:
.function..mu. .function..intg..tau..times..function..tau.
.function..tau..times..times.d.tau..function..intg..tau..times..function.-
.tau. .function..tau..times..times.d.tau..times..times.
##EQU00001## where .mu. is the unity-gain angular frequency of LMS
estimator 510a, which is the angular frequency at which the overall
gain from the output of summer 540 to the inverting input of summer
540 is equal to one. The parameter .mu. includes the gains of all
circuit blocks in the feedback loop from the output to the
inverting input of summer 540 and is given in units of
rad/sec/V.sup.2. Equation (1) assumes that the integrators are
ideal with a single pole at DC.
The filter output signal from adaptive 430a may be expressed
as:
.function..function..function..function..mu..function..intg..tau..times..-
function..tau.
.function..tau..times..times.d.tau..function..intg..tau..times..function.-
.tau. .function..tau..times..times.d.tau..times..times.
##EQU00002## The filter output signal, y(t), is often referred to
as an error signal. For simplicity, the following analysis assumes
that the x(t) signal contains only the TX leakage signal. The TX
leakage signal and the in-phase and quadrature reference signals
may also be assumed to be sinusoids with the following form:
x(t)=Asin(.omega.t+.phi.), i(t)=Bsin(.omega.t), and
q(t)=Bcos(.omega.t), Eq (3) where A is the amplitude of the TX
leakage signal; .phi. is a random angle of the TX leakage signal; B
is the amplitude of the r(t) reference signal; and .omega. is the
angular frequency of the transmit signal and the reference signal.
A frequency f and its angular frequency .omega. are related by a
factor of 2.pi., or .omega.=2.pi.f. Equation (2) may be converted
to a linear second-order ordinary differential equation with the
signals shown in equation (3), as follows:
d.times..function.d.mu..times.d.function.d.omega..function..times..times.
##EQU00003##
Equation (4) may be solved using Laplace transform, as follows:
s.sup.2Y(s)-sy(0)-y'(0)+.mu.B.sup.2[sY(s)-y(0)]+.omega..sup.2Y(s)=0,
Eq (5) where y(0) and y'(0) are initial conditions for y(t) and d
y(t)/dt, respectively. If no reference signal is applied for
t.ltoreq.0 (i.e., i(t)=0 and q(t)=0 for t.ltoreq.0), then y(t)=x(t)
for t.ltoreq.0, and the initial conditions may be expressed as:
y(0)=x(0)=Asin(.phi.), and y'(0)=x'(0)=A.omega.cos(.phi.). Eq (6)
With the initial conditions as shown in equation (6), the Laplace
transform of the adaptive filter output, y(t), may be expressed
as:
.function..function..PHI..omega..function..PHI..times..zeta..omega..funct-
ion..PHI..times..zeta..omega..omega..times..times. ##EQU00004##
where .zeta. is a damping factor, which is
.zeta.=.mu.B.sup.2/(2.omega.). The Laplace transform of the
adaptive filter input, x(t), may be expressed as:
.function..function..PHI..omega..function..PHI..omega..times..times.
##EQU00005##
The transfer function of adaptive 430a may then be expressed
as:
.function..function..function..function..PHI..omega..function..PHI..times-
..zeta..omega..function..PHI..function..PHI..omega..function..PHI..omega..-
times..zeta..omega..omega..times..times..times. ##EQU00006## where
s=j.omega..sub.x and .omega..sub.x is a variable for angular
frequency.
FIG. 10 shows the frequency responses of adaptive 430a for three
different damping factors. The frequency responses are given for a
TX leakage signal composed of a single tone at frequency of
.omega./2.pi.=835 MHz. Plot 1012 in FIG. 10 shows the frequency
response for a damping factor of .zeta.=0.001 (under-damping),
which has the narrowest notch and the least amount of attenuation
in the RX frequency band of 869 to 894 MHz for the cellular band.
Plots 1014 and 1016 in FIG. 10 show the frequency responses for
damping factors of .zeta.=0.01 and .zeta.=0.1, respectively. As the
damping factor increases, the notch widens and the amount of
attenuation in the RX frequency band increases. An ideal adaptive
filter can achieve infinite attenuation of the TX leakage signal.
The amount of TX leakage attenuation achieved by a practical
adaptive filter is dependent on imperfections in the adaptive
filter, as described below.
The inverse Laplace transform of Y(s) in equation (7) may be
expressed as:
.function.e.zeta..omega..times..times..function..PHI..zeta..function..PHI-
..zeta..function..omega..times..times..times..zeta..function..PHI..functio-
n..omega..times..times..times..zeta..times..times. ##EQU00007## The
exponential term e.sup.-.zeta..omega.t controls the settling time
and thus the convergence speed of the LMS algorithm. Since the
damping factor .zeta. needs to be much smaller than one (i.e.,
.zeta.<<1) to reduce filter distortion and attenuation, as
shown in FIG. 10, equation (10) may be simplified as follows:
y(t).apprxeq.x(t)e.sup.-.zeta..omega.t. Eq (11) Equation (11)
indicates that the filter output signal is simply an exponentially
decaying version of the filter input signal. For 30 dBc of TX
leakage rejection, e.sup.-.zeta..omega.t=10.sup.-30/20, and the
settling time may be expressed as:
.times..function. .zeta..omega..times..times. ##EQU00008##
Adaptive 430a generates cross modulation distortion even if all of
the circuit blocks of the adaptive filter are perfectly linear. The
cross modulation distortion is generated by the frequency mixing
function of multipliers 522 and 528, as illustrated in FIGS. 11A
through 11D.
FIG. 11A shows a case in which the filter input signal, x(t),
contains TX leakage signal 1112 centered at frequency f.sub.TX and
a single-tone jammer 1114 located at frequency f.sub.J. For this
example, the jammer frequency is close to the signal band of the
desired signal and f.sub.J-f.sub.TX.apprxeq.45 MHz, which is the
separation between the TX and RX frequency bands for the cellular
band.
FIG. 11B shows the signal components at the output of each of
multipliers 522a and 522b. Signal component 1122 at DC and signal
component 1126 at 2f.sub.TX are generated by the mixing between the
TX leakage signal and the i(t) and q(t) reference signals. Signal
component 1124 at f.sub.J-f.sub.TX and signal component 1128 at
f.sub.J+f.sub.TX are generated by the mixing between the jammer and
the reference signals.
FIG. 11C shows the signal components at the output of each of
integrators 524a and 524b. For this analysis, each integrator 524
has an ideal transfer function, which is a single pole at DC.
Signal component 1124 at f.sub.J-f.sub.TX is attenuated by a
particular amount, and signal components 1126 and 1128 at the
higher frequencies are attenuated by larger amounts and are
negligible. Signal component 1124 represents an undesired component
that contains the convolved spectra of the jammer and the transmit
signal.
FIG. 11D shows the signal components at the output of adaptive
filter 430a. Signal component 1144 centered at f.sub.J is generated
by the mixing of signal component 1124 centered at f.sub.J-f.sub.TX
and the reference signals centered at f.sub.TX. The double mixing
actions of multipliers 522a/522b and multipliers 528a/528b result
in the transmit signal component centered at f.sub.TX being
transferred to the jammer frequency f.sub.J. Signal component 1144
represents the cross modulation distortion that is added to the
filter input signal by summer 540. The filter output signal
contains an attenuated/rejected TX leakage signal 1112, an
unattenuated jammer 1114, and signal component 1144.
The cross modulation distortion generated by adaptive 430a may be
analyzed by a triple beat distortion. For the analysis, the
transmit signal (and thus the reference signal) contains two
closely spaced tones at frequencies of f.sub.TX.+-..DELTA.f/2. The
filter input signal contains (1) the TX leakage signal with the two
transmit tones and (2) an inband single-tone jammer at a frequency
of f.sub.J. If the adaptive filter completely rejects the TX
leakage signal such that the filter output signal, y(t), contains
only the jammer, then its triple beat distortion, d(t), may be
derived as:
.function..mu..omega..omega..function..function..function..function..func-
tion..omega..omega..+-..DELTA..omega..times..times. ##EQU00009##
where
i(t)=B[sin((.omega.-.DELTA..omega./2)t)+sin((.omega.+.DELTA..omega./2)t)]-
;
q(t)=B[cos((.omega.-.DELTA..omega./2)t)+cos((.omega.+.DELTA..omega./2)t)-
]; and y(t)=Ccos(.omega..sub.Jt), where C is the amplitude of the
jammer. Equation (13) indicates that two triple beat distortion
terms are generated at frequencies of f.sub.J.+-..DELTA.f, as
described above for FIGS. 11A through 11D.
A triple beat rejection ratio (TBRR) is defined as the ratio of the
jammer amplitude to the amplitude of the cross modulation
distortion. The TBRR may be obtained by performing simple
trigonometric manipulations on equation (13) and taking the ratio
of the jammer amplitude to the triple beat distortion amplitude.
The TBRR may be expressed as:
.times..function..omega..omega..mu..times..times..function..times..zeta..-
times..omega..omega..times..times. ##EQU00010## where
.zeta.=.mu.B.sup.2/(2.omega..sub.TX) is the damping factor.
Equation (14) indicates that a TBRR of 68 dBc may be obtained with
a damping factor of .zeta..ltoreq.8.1.times.10.sup.-6 for
f.sub.TX=849 MHz and f.sub.J=894 MHz. The settling time is 81
.mu.sec with this damping factor.
FIG. 6 shows a block diagram of an adaptive filter 430b, which is
another embodiment of adaptive filter 430 within receiver 420.
Adaptive filter 430b includes an additional pole used to (1) reduce
the amplitude of the cross modulation distortion generated by the
adaptive filter (2) achieve faster LMS algorithm convergence, and
(3) shorten the settling time.
Adaptive filter 430b includes an LMS adaptive estimator 510b and
summer 540. LMS estimator 510b includes all of the circuit blocks
for LMS estimator 510a in FIG. 5. LMS estimator 510b further
includes (1) a single-pole or first-order lowpass filter (LPF) 526a
placed between the output of integrator 524a and the input of
multiplier 528a and (2) a single-pole lowpass filter 526b placed
between the output of integrator 524b and the input of multiplier
528b. Each lowpass filter 526 may be implemented, for example, with
an RC lowpass network composed of a series resistor and a shunt
capacitor to circuit ground. The frequency of the single pole is
selected such that the adaptive filter is unconditionally stable.
As an example, with the additional pole placed at 318 KHz, the
damping factor may be increased to .zeta.=1.times.10.sup.-4, the
settling time may be reduced to 6 .mu.sec, and the TBRR may be
improved to better than 80 dBc. All of these improvements are
achieved with an unconditionally stable adaptive filter. Lowpass
filters of higher order and/or with poles placed at different
frequencies may also be used for lowpass filters 526a and 526b.
An ideal adaptive filter provides infinite rejection of the TX
leakage signal so that the filter output signal contains no TX
leakage signal. However, various imperfections in a
practical/realizable adaptive filter limit the amount of TX leakage
rejection that may be achieved. Such imperfections may include, for
example, finite gain for the integrators and non-zero DC offsets
for the circuit blocks of the LMS estimator.
A TX leakage rejection ratio (TXRR) is the ratio of the TX leakage
signal power at the adaptive filter output to the TX leakage signal
power at the adaptive filter input. The TXRR requirement for
adaptive filter 430 is dependent on various factors such as, for
example, (1) the maximum TX leakage signal power expected at the
output of LNA 422 and (2) the maximum acceptable TX leakage signal
power at the input of mixer 432. It can be shown that an adaptive
filter with a TXRR of approximately 30 dB can provide performance
comparable to that achieved by a receiver with an RF SAW filter
(e.g., the receiver shown in FIG. 3). In general, the TXRR
requirement for the adaptive filter is dependent on various
factors, such as those noted above and possibly other factors.
Adaptive filters 430a and 430b have similar TXRR performance. The
actual TXRR achieved by adaptive 430a is dependent on various
factors such as, for example, (1) the overall gain of the
integrators and multipliers and (2) the DC offsets of the
multipliers and integrators. An inadequate overall gain limits the
TXRR that can be achieved by the adaptive filter. The overall gain
is thus selected such that the required TXRR can be achieved and is
appropriately distributed among the integrator and multipliers.
DC offsets can also adversely affect the TXRR performance of
adaptive filter 430a. Multipliers 522a, 522b, 528a, and 528b
typically have DC responses due to imbalances on the two inputs.
Integrator 524a and 524b have systematic as well as random input DC
offsets. The DC offsets introduce an error that reduces the amount
of TX leakage rejection by the filter. Also, due to their large DC
gains, the integrators may be saturated initially by the combined
DC offsets. Once saturated, the integrators have very low gains
that result in a long settling time for the adaptive filter. To
prevent saturation due to DC offsets, the output of each integrator
may be reset (e.g., by shorting together the differential output of
each integrator) prior to enabling the adaptive filter and then
released thereafter.
Various techniques may be used to achieve low combined DC offsets
for the in-phase and quadrature paths. The combined DC offsets may
be reduced by: Increasing the gain of multipliers 522a and 522b and
reducing their DC offsets; Increasing the reference signal power
(i.e., increase B); and/or Using dynamic offset cancellation
techniques such as chopper stabilization and/or auto-zeroing
techniques.
The multiplier gain may be increased by converting multipliers 522a
and 522b into mixers and using the in-phase and quadrature
reference signals as strong LO signals. The high gain of the mixers
(e.g., approximately 50 dB for an exemplary mixer design) can
significantly reduce the DC offset contribution of the integrators.
The output DC offset of the mixers is low due to the inherent
chopping action of the mixers.
Chopper stabilization techniques may be able to achieve low input
DC offset voltages (e.g., below 10 .mu.V). Auto-zeroing techniques,
such as correlated double sampling techniques, typically increase
the noise floor, which may then contaminate the RX frequency band.
The auto-zeroing techniques should thus be used with care.
Adaptive filter 430 inherently introduces additional noise that
degrades the noise figure of the receiver. Adaptive filter 430 may
be designed to minimize noise contribution by using various circuit
design techniques known in the art. This way, system requirements
can be met even with the additional noise contribution from
adaptive filter 430.
Adaptive filter 430 is a feedback system and is unstable if the
total phase delay along the feedback loop is 180.degree. and the
loop gain is greater than one. For an ideal adaptive filter, the
only delay along the feedback loop is 90.degree. introduced by the
integrators. For a practical adaptive filter, a delay is introduced
by each circuit block within the adaptive filter.
FIG. 7 shows a block diagram of an adaptive filter 430c, which is a
more detailed embodiment of adaptive filter 430. A quadrature
splitter 708 receives the reference signal, r(t), and provides a
differential in-phase reference signal, i'(t), and a differential
quadrature reference signal, q'(t).
A pre-amplifier 718 receives and amplifies the filter input signal,
x(t), and provides a differential output signal, y'(t), to an
in-phase section 720a and a quadrature section 720b. Within
in-phase section 720a, a multiplier 722a receives and multiplies
the y'(t) signal with the i'(t) signal and provides a differential
m.sub.i'(t) signal. An integrator 724a receives and integrates the
m.sub.i'(t) signal and provides a differential w.sub.i'(t) signal.
Integrator 724a is implemented with an amplifier and two capacitors
coupled between the differential output and the differential input
of the amplifier, as shown in FIG. 7. A multiplier 728a receives
and multiplies the w.sub.i'(t) signal with the i'(t) signal and
provides a z.sub.i'(t) signal. Within quadrature section 720b, a
multiplier 722b, an integrator 724b, and a multiplier 728b
similarly process the y'(t) signal with the q'(t) signal and
provide a z.sub.q'(t) signal. The z.sub.i'(t) and z.sub.q'(t)
signals are current outputs and may be combined by tying these
outputs together to obtain the estimator signal, e(t). The e(t)
signal is subtracted from the x(t) signal by tying these signals
together at the input of pre-amplifier 718. For the circuit
embodiment shown in FIG. 7, the summer is just a node labeled by a
black dot at the input of pre-amplifier 718, and the input of the
adaptive filter is also its output (i.e., y(t)=x(t)).
Pre-amplifier 718 has a delay of .DELTA..phi..sub.1 at the
frequency of the TX leakage signal. Multipliers 722a and 722b each
has a delay of .DELTA..phi..sub.2 due to unequal delays of the RF
and LO inputs. Multipliers 728a and 728b each has a delay of
.DELTA..phi..sub.3 from the reference signal to the multiplier
output at the frequency of the TX leakage signal. The total delay
.DELTA..phi. for adaptive filter 430c may be computed as:
.DELTA..phi.=.DELTA..phi..sub.1+.DELTA..phi..sub.2+.DELTA..phi..sub.3.
FIG. 9 shows a simplified model 900 for the adaptive filter which
is suitable for stability analysis. A summer 912 receives and
subtracts the output of an integrator 916 from the filter input
signal, V.sub.in, and provides the filter output signal, V.sub.out.
A delay element 914 delays the V.sub.out, signal by a delay of
.DELTA..phi.. Integrator 916 integrates the delayed signal with a
transfer function of G.sub.o/(s/p+1). The transfer function between
the filter output signal to the filter input signal may be
expressed as:
e.DELTA..phi..times..times. ##EQU00011## A step response for
equation (15) may be expressed as:
V.sub.out(t).apprxeq.e.sup.-(1+G.sup.o.sup.e.sup.j.DELTA..phi..sup.)pt=e.-
sup.-(1+G.sup.o.sup.cos(.DELTA..phi.))pt. Eq (16) Equation (16)
indicates that the filter output signal is an oscillatory signal
having an exponential decay of e.sup.-pt due to the integrator pole
at p. The presence of the delay .DELTA..phi. introduces
oscillations in the filter output signal. The amplitude of the
oscillations can either decay or grow depending on the delay
.DELTA..phi.. It can be shown that the adaptive filter is (1)
stable if .DELTA..phi. is in the range of -90.degree. to
+90.degree. and (2) unstable if |.DELTA..phi.| exceeds 90.degree..
For example, if .DELTA..phi..sub.1=40.degree.,
.DELTA..phi..sub.2=0.degree., and .DELTA..phi..sub.3=60.degree.,
then .DELTA..phi.=100.degree. and the adaptive filter
oscillates.
FIG. 8 shows a block diagram of an adaptive filter 430d, which is
another more detailed embodiment of adaptive filter 430. Adaptive
filter 430d utilizes an architecture that can compensate for phase
delays (e.g., .DELTA..phi..sub.1 and .DELTA..phi..sub.3) at RF
frequencies. Adaptive filter 430d includes all of the circuit
blocks of adaptive filter 430c in FIG. 7. However, adaptive filter
430d uses different reference signals for the two multipliers in
each of sections 720a and 720b. For in-phase section 720a,
multiplier 722a is driven by the i'(t) signal and multiplier 728a
is driven by the q'(t) signal (instead of the i'(t) signal). For
quadrature section 720b, multiplier 722b is driven by the q'(t)
signal and multiplier 728b is driven by the i'(t) signal (instead
of the q'(t) signal). The LO signals for multipliers 728a and 728b
thus lead the LO signals for multipliers 722a and 722b,
respectively, by 90.degree.. The total delay .DELTA..phi. is also
correspondingly reduced by 90.degree.. For the example described
above, the total delay is .DELTA..phi.=10.degree. instead of
100.degree., and the adaptive filter is stable.
The adaptive filters shown in FIGS. 5 and 6 may be implemented in
various manners. Two exemplary implementations are shown in FIGS. 7
and 8. The circuit blocks for the adaptive filters may also be
implemented in various manners. For example, the multipliers may be
implemented with mixers, summers may be implemented by tying
current outputs together, and so on. The adaptive filters may also
be implemented with differential or single-ended circuit designs.
FIGS. 7 and 8 show exemplary differential designs for adaptive 430a
in FIG. 5. A differential design may provide certain advantages
over a single-ended design such as better immunity to noise.
The adaptive filters described herein utilize an LMS adaptive
estimator to estimate the TX leakage signal. Other types of
estimators may also be used to estimate the TX leakage signal, and
this is within the scope of the invention. For example, the
transmit signal may be stepped across the TX frequency band and
weight values w.sub.i and w.sub.q may be determined for the
in-phase and quadrature sections, respectively, and used in place
of w.sub.i(t) and w.sub.q(.sup.t) in FIG. 5 to estimate the TX
leakage signal.
The adaptive filters may also be trained in various manners. For
example, an adaptive filter may be enabled at the beginning of a
training burst (which contains a known training signal) and weight
values may be derived based on this burst. The weight values may
thereafter be fixed and used for estimating the TX leakage signal
during the signaling interval. The weight values may be updated
whenever the training bursts are available. To speed up
convergence, the conditions for the integrators may be determined
and stored prior to tuning away from an RF channel, and the
integrators may be initialized with the stored conditions the next
time this RF channel is selected.
The adaptive filters described herein may also be used for various
systems and applications. For example, the adaptive filters may be
used in wireless full-duplex communication systems such as cellular
systems, OFDM systems, orthogonal frequency division multiple
access (OFDMA) systems, multiple-input multiple-output (MIMO)
systems, wireless local area networks (LANs), and so on. The
full-duplex cellular systems include CDMA system and some versions
of Global System for Mobile Communications (GSM) systems, and the
CDMA systems include IS-95, IS-2000, IS-856, and Wideband-CDMA
(W-CDMA) systems. The adaptive filters may be used for a wireless
device as well as a base station in a wireless full-duplex
communication system.
The adaptive filters described herein may be implemented within an
integrated circuit (IC), an RF integrated circuit, an application
specific integrated circuit (ASIC), or other electronic units
designed to perform the functions described herein. The adaptive
filters may also be fabricated with various IC process technologies
such as complementary metal oxide semiconductor (CMOS), bipolar
junction transistor (BJT), bipolar-CMOS (BiCMOS), silicon germanium
(SiGe), gallium arsenide (GaAs), and so on.
The previous description of the disclosed embodiments is provided
to enable any person skilled in the art to make or use the present
invention. Various modifications to these embodiments will be
readily apparent to those skilled in the art, and the generic
principles defined herein may be applied to other embodiments
without departing from the spirit or scope of the invention. Thus,
the present invention is not intended to be limited to the
embodiments shown herein but is to be accorded the widest scope
consistent with the principles and novel features disclosed
herein.
* * * * *