U.S. patent number 7,570,497 [Application Number 11/449,486] was granted by the patent office on 2009-08-04 for discontinuous quasi-resonant forward converter.
This patent grant is currently assigned to Cambridge Semiconductor. Invention is credited to David Robert Coulson, Russell Jacques.
United States Patent |
7,570,497 |
Jacques , et al. |
August 4, 2009 |
**Please see images for:
( Certificate of Correction ) ** |
Discontinuous quasi-resonant forward converter
Abstract
A discontinuous resonant forward power converter including a
controller having an output coupled to a controllable switch and
which is configured to control the switch such that a voltage
waveform on a secondary winding of a transformer of the converter
has a first portion during which the switch is on and current flows
into an output node of the converter which is coupled to the output
rectifier and to a smoothing capacitor, and which has a second
substantially resonant portion during which the switch and an
output rectifier are both off. Substantially no current flows into
the output node during the second portion of said voltage
waveform.
Inventors: |
Jacques; Russell (Burton Green,
GB), Coulson; David Robert (Comberton,
GB) |
Assignee: |
Cambridge Semiconductor
(Cambridge, GB)
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Family
ID: |
36687744 |
Appl.
No.: |
11/449,486 |
Filed: |
June 8, 2006 |
Prior Publication Data
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Document
Identifier |
Publication Date |
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US 20070274108 A1 |
Nov 29, 2007 |
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Foreign Application Priority Data
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May 26, 2006 [GB] |
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0610422.8 |
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Current U.S.
Class: |
363/21.03;
363/21.09; 363/21.08 |
Current CPC
Class: |
H02M
3/33507 (20130101); Y02B 70/10 (20130101); Y02B
70/1433 (20130101) |
Current International
Class: |
H02M
3/335 (20060101) |
Field of
Search: |
;363/16,21.03,21.08,21.1,21.04,21.07,21.09 |
References Cited
[Referenced By]
U.S. Patent Documents
Foreign Patent Documents
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0 055 064 |
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Jun 1982 |
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EP |
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0 074 399 |
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Aug 1988 |
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EP |
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0 658 968 |
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Jun 1995 |
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EP |
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1 156 580 |
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Nov 2001 |
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EP |
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1 432 108 |
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Jun 2004 |
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EP |
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1 508 961 |
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Feb 2005 |
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EP |
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2 151 822 |
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Jul 1985 |
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GB |
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05292741 |
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May 1993 |
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JP |
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09182424 |
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Jul 1997 |
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JP |
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2002345236 |
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Nov 2002 |
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JP |
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Other References
Shukla J et al: "A novel active resonant snubber for single switch
unity power factor three-phase diode rectifiers" Power Electronics
Specialists Conference, 2004. PESC 04. 2004 IEEE 35.sup.TH Annual
Aachen, Germany Jun. 20-25, 2004, Piscataway, NJ, USA, IEEE, US,
Jun. 20, 2004, pp. 3818-3823vol. 5, XP010738323 ISBN: 0-7903-8399-0
paragraph [Introduction]; figure 1. cited by other .
International Search Report for corresponding PCT/GB2007/050276.
cited by other .
Search Report Under Section 17 for corresponding GB0610422.8,
completed Mar. 5, 2007. cited by other .
International Search Report for corresponding PCT/GB2007/050276,
completed Sep. 1, 2008 by M. Marannino. cited by other .
UK Search Report for corresponding GB0811374.8 completed Sep. 30,
2008. cited by other .
UK Search Report for corresponding GB0811279.9 completed Oct. 2,
2008. cited by other .
EPODOC/EPO JP529741, Feb. 10, 2008. cited by other .
EPODOC/EPO JP918424, Feb. 10, 2008. cited by other .
EPODOC/EPO JP2002345236, Sep. 29, 2008. cited by other.
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Primary Examiner: Riley; Shawn
Claims
We claim:
1. A discontinuous resonant forward converter for converting an
input dc voltage to an output dc voltage, the converter comprising:
first and second dc inputs; a transformer having primary and
secondary windings with matched polarities; a controllable switch
for switching power from said dc inputs through said primary
winding of said transformer, said controllable switch and said
primary winding of said transformer being coupled in series between
said first and second dc voltage inputs; first and second dc
voltage outputs; a rectifier coupled to said secondary winding of
said transformer, said rectifier and said secondary winding of said
transformer being coupled in series between said first and second
dc voltage outputs; a smoothing capacitor having a first connection
coupled to receive dc power from said rectifier at a first
connection node, said first connection node being coupled to said
first dc voltage output, said smoothing capacitor having a second
connection coupled to said second dc voltage output; a controller
having an output coupled to said controllable switch and being
configured to control said switch such that a voltage waveform on
said secondary winding has a first portion during which said switch
is on and current flows into said first connection node to provide
a secondary loading current, and second portion during which said
switch and said rectifier are both off; wherein substantially no
current flows into said first connection node during said second
portion of said voltage waveform; a resonant circuit to demagnetize
the transformer during the second portion, the resonant circuit
including magnetising inductance (Lmag) and leakage inductance
(Lleak) of said transformer representing a level of energy stored
in said transformer, a resonant characteristic of said resonant
circuit varies in response to changes in energy stored in said
transformer, which in turn is dependent on said secondary loading
current just before turn off of said switch; and wherein said
controller is configured to receive at least one input signal
associated with energy stored in said transformer and control an on
pulse width of said switch dependent on energy stored in said
transformer such that said on pulse width is adjusted to maintain
resonance with varying said secondary loading current.
2. A forward converter as claimed in claim 1 wherein a connection
between said rectifier and said first connection node has an
inductance, and wherein substantially no current flows in said
inductance during said second portion of said waveform.
3. A forward converter as claimed in claim 1 lacking a freewheeling
rectifier coupled to said secondary winding of said
transformer.
4. A forward converter as claimed in claim 1 wherein said
controller is configured to provide a switch drive signal to said
controller output, said switch drive signal having an on period for
controlling said switch to switch power to said primary winding on
and an off period for controlling said switch to switch power to
said primary winding off, and wherein said controller is configured
to use a first control signal to start said drive signal on period
and a second control signal to start said drive signal off
period.
5. A forward converter is claimed in claim 4 wherein one or both of
said first and second control signals is responsive to said level
of stored energy in said transformer due to changes in said
secondary loading current to vary a timing of said start of said
drive signal off period.
6. A forward converter as claimed in claim 4 wherein said first
control signal is responsive to a voltage on said primary or on an
auxiliary winding of said transformer.
7. A forward converter as claimed in claim 6 wherein said first
control signal is responsive to a value of substantially zero volts
on said primary winding node connected to said switch of said
transformer to start said drive signal on period.
8. A forward converter as claimed in claim 6 wherein said first
control signal is responsive to a prediction of a value of
substantially zero volts on said primary winding of said
transformer to start said drive signal on period.
9. A forward converter as claimed in claim 5 wherein said drive
signal off period is sufficiently long for said resonant portion of
said secondary winding voltage waveform to include substantially
half a cycle of substantially sinusoidal resonance, wherein said
first control signal identifies an end of said half cycle, and
wherein said controller is configured to start said drive signal on
period following a delay from said end of said half cycle.
10. A forward converter as claimed in claim 4 wherein said first
control signal is responsive to a voltage on an auxiliary winding
of said transformer.
11. A forward converter as claimed in claim 4 wherein said
controller includes a current limiting system and a system for
controlling said switch drive signal such that, responsive to
implementation of current limiting, a frequency of said switch
drive signal is increased.
12. A forward converter as claimed in claim 1 wherein said
controller includes a system for controlling said output to said
switch such that, during start-up of said converter, a frequency of
a drive signal to said switch is increased.
13. A forward converter as claimed in claim 1 wherein said
controller includes a system for controlling said output to said
switch such that, during start-up of said converter, said converter
operates in a non-resonant mode.
14. A forward converter as claimed in claim 1 wherein said
controller comprises a single integrated circuit.
15. A controller for controlling a discontinuous forward converter
for converting an input dc voltage to an output dc voltage, the
converter comprising: first and second dc inputs; a transformer
having primary and secondary windings with matched polarities; a
controllable switch for switching power from said dc inputs through
said primary winding of said transformer, said controllable switch
and said primary winding of said transformer, being coupled in
series between said first and second dc voltage inputs; first and
second do voltage outputs; a rectifier coupled to said secondary
winding of said transformer, said rectifier and said secondary
winding of said transformer being coupled in series between said
first and second dc voltage outputs; and a smoothing capacitor
having a first connection coupled to receive dc power from said
rectifier at a first connection node, said first connection node
being coupled to said first dc voltage output, said smoothing
capacitor having a second connection coupled to said second dc
voltage output; and a controller having an output coupled to said
controllable switch and is configured to control said switch such
that a voltage waveform on said secondary winding has a first
portion during which said switch is on and current flows into said
first connection node to provide a secondary loading current, and
second portion during which said switch and said rectifier are both
off; and wherein substantially no current flows into said first
connection node during said second portion of said voltage
waveform; a resonant circuit to demagnetize the transformer during
the second portion, the resonant circuit including magnetising
inductance (Lmag) and leakage inductance (Lleak) of said
transformer representing a level of energy stored in said
transformer, a resonant characteristic of said resonant circuit
varies in response to changes in energy stored in said transformer,
which in turn is dependent on said secondary loading current just
before turn off of said switch; and wherein said controller is
configured to receive at least one input signal associated with
energy stored in said transformer and control an on pulse width of
said switch dependent on energy stored in said transformer such
that said on pulse width is adjusted to maintain resonance with
varying said secondary loading current.
16. An integrated circuit including the controller of claim 15.
17. A method of controlling a forward converter, the converter
comprising: first and second dc inputs; a transformer having
primary and secondary windings with matched polarities; a
controllable switch for switching power from said dc inputs through
said primary winding of said transformer, said controllable switch
and said primary winding of said transformer being coupled in
series between said first and second dc voltage inputs; first and
second dc voltage outputs; a rectifier coupled to said secondary
winding of said transformer, said rectifier and said secondary
winding of said transformer being coupled in series between said
first and second dc voltage outputs; and a smoothing capacitor
having a first connection coupled to receive dc power from said
rectifier at a first connection node, said first connection node
being coupled to said first dc voltage output, said smoothing
capacitor having a second connection coupled to said second dc
voltage output; the method comprising controlling said controllable
switch such that a voltage waveform on said secondary winding has a
first portion during which said switch is on and current flows to
provide a secondary loading current into said first connection
node, and second portion during which said switch and said
rectifier are both off; and wherein substantially no current flows
into said first connection node during said second portion of said
voltage waveform; wherein the converter comprises a resonant
circuit to demagnetize the transformer during the second portion,
the resonant circuit including magnetising inductance (Lmag) and
leakage inductance (Lleak) of said transformer representing a level
of energy stored in said transformer, a resonant characteristic of
said resonant circuit varies in response to changes in energy
stored in said transformer, which in turn is dependent on said
secondary loading current just before turn off of said switch; and
wherein the method further comprises: receiving at least one input
signal associated with energy stored in said transformer;
controlling an on pulse width of said switch dependent on energy
stored in said transformer such that said on pulse width is
adjusted to maintain resonance with varying said secondary loading
current.
18. A discontinuous resonant forward converter as claimed in claim
1 wherein said controllable switch is a bipolar transistor minority
carrier device having a collector terminal coupled to said primary
winding of said transformer.
19. A discontinuous resonant forward converter as claimed in claim
1 wherein said at least one input signal is a current sense signal
of a primary current through the transformer wherein said secondary
load current approximately matches the primary current.
Description
FIELD OF THE INVENTION
This invention generally relates to forward power converters, and
more particularly to improved systems and methods for operating
such converters, and to controllers for implementing these systems
and methods.
BACKGROUND TO THE INVENTION
FIG. 1 (which is taken from U.S. Pat. No. 4,688,160) shows an
example of a forward power converter comprising a dc input 101, 102
coupled to the primary winding 109 of a transformer 110. The
primary winding 109 is connected in series with a switching device
105, here a bipolar transistor, which switches on and off, during
an on period building up magnetising flux in the primary winding
109, which drives a current in a secondary winding 111 of the
transformer. Unlike a so-called flyback converter, in a forward
converter the primary and secondary windings have matched
polarities, as indicated by the dots on the windings in FIG. 1. The
output from the transformer 110 is rectified by a rectifier 114 and
smoothed by a smoothing capacitor 119 to provide a dc output 121,
122. When switch 105 is off the core of the transformer is "reset"
allowing the magnetising flux to return to its initial state. In
the example of FIG. 1 (U.S. Pat. No. 4,688,160) this is performed
by resonant action between the magnetising inductance of
transformer 110 and a capacitor 113 shunting diode 114, returning
energy to the input voltage source.
The circuit of FIG. 1 includes a large output choke 117 between
rectifier 114 and smoothing capacitor 119, and a freewheeling or
"flyback" diode 115 across the series combination of choke 117 and
smoothing capacitor 119. This is because when the switch 105 is
turned off, because the primary and secondary windings have the
same sense, rectifier 114 immediately becomes non-conducting. The
function of the freewheeling diode 115 is to allow the choke 117 to
maintain a continuous output current into output node "X" when
switch 105 is off by providing a path for this current.
FIG. 1 shows a conventional, continuous forward converter. There
are many other prior art documents describing such converters,
including, for example, U.S. Pat. Nos. 4,415,959; 6,760,236;
6,304,463; 6,252,781; and the reference design SLUA276 for the
Texas Instruments UCC38C42. In some of these later circuits the
secondary side diodes are replaced by synchronous rectifiers
embodied in MOS transistors. Other background prior art can be
found in U.S. Pat. No. 4,788,634 which describes a resonant forward
converter in which natural self-inductance of the transformer in
parallel with the transformer provides a resonant "ring" so that
the switching circuit can be self-resonant; and U.S. 2005/0270809
(WO 2004/057745) which describes use of an auxiliary transformer in
a current limiting circuit.
The inventors have recognised that improved operation, such as
improved regulation and start-up may be achieved by use of switch
control in a discontinuous current flow mode.
SUMMARY OF THE INVENTION
According to a first aspect of the invention there is therefore
provided a discontinuous resonant forward converter for converting
an input dc voltage to an output dc voltage, the converter
comprising: first and second dc inputs; a transformer having
primary and secondary windings with matched polarities; a
controllable switch for switching power from said dc inputs through
said primary winding of said transformer, said controllable switch
and said primary winding of said transformer being coupled in
series between said first and second dc voltage inputs; first and
second dc voltage outputs; a rectifier coupled to said secondary
winding of said transformer, said rectifier and said secondary
winding of said transformer being coupled in series between said
first and second dc voltage outputs; a smoothing capacitor having a
first connection coupled to receive dc power from said rectifier at
a first connection node, said first connection node being coupled
to said first dc voltage output, said smoothing capacitor having a
second connection coupled to said second dc voltage output; and a
controller having an output coupled to said controllable switch and
being configured to control said switch such that a voltage
waveform on said secondary winding has a first portion during which
said switch is on and current flows into said first connection
node, and second substantially resonant portion during which said
switch and said rectifier are both off; and wherein substantially
no current flows into said first connection node during said second
portion of said voltage waveform.
A connection between the rectifier and the first connection node
may include a small inductor but substantially no current flows in
this inductance during the second, resonant portion of the
waveform. There is, however, no need for a large choke as used in
the continuous forward converters mentioned in the introduction. If
an inductor is present this may have a value of less than 10%, 5%,
2%, 1% or 0.1% of the primary side magnetising inductance of the
transformer. Embodiments of the forward converter lack a
freewheeling rectifier coupled to the secondary winding. Adding a
small amount of output inductance can assist in implementing the
current limit function (as described later), as well as regulation
and start-up providing that the converter remains
discontinuous.
As described above, a forward converter according to an embodiment
of the invention employs controlled switching of the switch rather
than relying on self-oscillation to directly control the power
switch. The inventors have nonetheless recognised that embodiments
of the forward converter may tolerate variations in component
parameters, in particular the inductance of the transformer and the
capacitance of the resonant capacitor, both of which contribute to
variations in the resonant frequency. The inventors have recognised
that in a discontinuous resonant forward converter,
counter-intuitively a substantially fixed frequency oscillator may
be employed and nonetheless achieve robustness with respect to
component value variations, which in turn facilitates the use of
components with larger tolerances and hence reduced cost
implementations.
In particular, broadly speaking in implementations of the forward
converter the voltage waveform on the secondary winding may be
approximately divided into three (in general not equal) periods,
the first portion of the voltage waveform described above occupying
one of these three periods, the other two periods being occupied by
the second, substantially resonant portion of the voltage waveform
and then a period of "dead time" while the switch voltage is close
to zero volts. (The waveform is clamped on the primary side and, in
embodiments with a bipolar transistor switch described later, in
fact is clamped at one diode drop away from zero volts, because
there is an intrinsic diode in the switch). When the secondary side
voltage waveform is at substantially zero volts, this is a good
time at which to turn on the primary side switch again, to reduce
electromagnetic interference (EMI) and to achieve good efficiency.
There is, however, a relatively long period during which this
voltage waveform remains at substantially zero volts, and thus the
frequency of the oscillator may be selected to allow for some
variation in the duration of the half cycle of resonance. If the
transformer voltage is already at substantially zero volts
switching on the switch does not cause any significant voltage
change.
Thus by controlling the drive to the switch so that the off period
is sufficiently long the switch does not switch on again until the
half cycle of sinusoidal resonance is over and the waveform is
approximately at zero volts. Preferably, such a substantially fixed
frequency oscillator has a duty cycle (on period as a percentage of
the total period) of less than 70%. Eventually the secondary
winding voltage waveform begins to sinusoidally ring again and it
is therefore preferable that the off period of the duty cycle is
sufficiently short to avoid this region of the waveform.
Preferably, therefore, the duty cycle of the oscillator is such
that the percentage of the on period of the total period is greater
than 5%, preferably greater than 10%, in embodiments greater than
30%. In general the duty cycle can vary depending on the design
and, in mains powered embodiments, on the mains input voltage. For
example a 110V version may operate with an on duty cycle in the
region of 60% to 70% whilst a 250V version may have an on duty
cycle of 30% to 40%; other designs may have a duty cycle in the
region of 40% to 60%, for example approximately 50%.
Additionally or alternatively one or both of pulse width modulation
(PWM) and pulse frequency modulation (PFM) may be employed whilst
maintaining discontinuous resonance. This facilitates regulation of
the output of the forward converter, in particular under varying
load conditions. However preferably the switch controller is
configured to control the switch by employing a pair of control
signals, a first to turn the switch on, and a second to turn the
switch off. This facilitates implementation of a range of different
control strategies, optionally in the context of PWM and/or PFM. In
particular this facilitates the implementation of zero volts
switching (ZVS), and of over current protection (OCP) in the
context of a discontinuous resonant forward converter.
Thus in some preferred embodiments a first control signal controls
the switch on and a second control signal controls the switch off.
The first control signal may, in embodiments, be responsive to
detection of a substantially zero volt condition on the primary
winding voltage. This may either be used to turn the power switch
on immediately, or after a delay. Alternatively a voltage from an
auxiliary winding of the transformer may be employed. More
generally the sensed voltage may be compared with a reference level
rather than necessarily zero volts. In some preferred embodiments a
non-zero reference point, say 50 volts, on the sensed waveform is
detected and used to predict the time at which the sensed voltage
goes to zero, or close to zero (the time is approximately known
since the reference level is known). This can facilitate
implementation of the controller. In a further alternative the
power switch may be turned on in response to a sensed current
through the primary winding and switch, for example sensed by
measuring a voltage drop across a current sensing resistor. This
may be employed, for example, to delay the turn on of the power
switch.
The second control signal, which turns the power switch off, may
also be responsive to one or more of a number of different
variables. For example in a simple embodiment the second control
signal may control the switch off after a time delay from the first
control signal controlling the power switch on. Optionally this
time delay may be variable, and in this way pulse width modulation
may be implemented. The pulse width may be responsive to, for
example, a voltage on the primary or on an auxiliary winding of the
transformer and/or to a sensed voltage on the secondary side of the
forward converter. It will be recognised that, in general, the
voltage on the primary (or secondary) winding of the transformer
may be sensed either directly or indirectly. In general, the second
control signal may be responsive to any sensed primary or secondary
side voltage or current.
In some particularly preferred embodiments the second control
signal implements an over current protection (OCP) function by
substantially immediately switching the power switch off when an
over current condition is detected, for example via voltage sensed
from a sensor such as a current sensing resistor in series with the
switch. This can be used to implement a cycle-by-cycle OCP and
facilitates a rapid response when a switch current greater than a
threshold level is detected.
In some preferred embodiments the controller implements a current
limiting mode which includes increased frequency operation. Thus in
embodiments the controller increases a frequency of the drive
signal when current limiting. This can help to avoid a runaway
process (as described later) which, for certain types of load can
cause the output voltage to continuously fall. In embodiments a
threshold current for current limiting is adjusted in response to
changing the frequency or pulse width of the drive signal, and in
particular the threshold current may be increased as the drive
pulse width is reduced or the drive frequency is increased. As
mentioned above, the output side of the forward converter may
include a small inductance and still operate in a discontinuous
resonant mode, and the inclusion of such an inductance can help to
limit (regulate) the output current in overload, in particular by
facilitating regulation of the output current as described above.
In embodiments this inductance may be provided by leakage or
parasitic inductance in the circuit, in particular leakage
inductance of the transformer. In embodiments the transformer may
be configured to provide a leakage inductance to contribute to a
desired value of output inductance for the forward converter.
One particular difficulty, often encountered with forward power
converter designs and particularly acute in a converter without
series inductance, is ensuring reliable start-up. This is because
at start-up the forward converter output effectively appears as a
short circuit which can potentially damage the power switch or,
where a current limiting arrangement is in place, which can trigger
the current limiting and hence prevent the output voltage from
reaching its proper value. Embodiments of the forward converter we
describe, which employ an arrangement to turn the power switch on
and off in a controlled manner facilitate management of a start-up
of the forward converter and, in particular, enable a frequency of
a drive signal to the power switch to be increased at start-up.
This takes the converter out of its resonant mode of operation at
start-up and enables more power to be transferred to the output
whilst still protecting the power switch. The start-up condition
may straightforwardly be detected on the primary side of the
forward converter or, indirectly, by making use of a current
sense/limit system in the forward converter.
In some particularly preferred embodiments the above-described
controller is implemented in a single integrated circuit. This IC
may implement one or more of a range of controlled strategies as
described above. In some preferred implementations of the IC,
however, the power switch is left off the chip, for
flexibility.
Thus in another aspect the invention provides a controller for
controlling a forward converter for converting an input dc voltage
to an output dc voltage, the converter comprising: first and second
dc inputs; a transformer having primary and secondary windings with
matched polarities; a controllable switch for switching power from
said dc inputs through said primary winding of said transformer,
said controllable switch and said primary winding of said
transformer, being coupled in series between said first and second
dc voltage inputs; first and second dc voltage outputs; a rectifier
coupled to said secondary winding of said transformer, said
rectifier and said secondary winding of said transformer being
coupled in series between said first and second dc voltage outputs;
and a smoothing capacitor having a first connection coupled to
receive dc power from said rectifier at a first connection node,
said first connection node being coupled to said first dc voltage
output, said smoothing capacitor having a second connection coupled
to said second dc voltage output; and wherein said controller has
an output coupled to said controllable switch and is configured to
control said switch such that a voltage waveform on said secondary
winding has a first portion during which said switch is on and
current flows into said first connection node, and second
substantially resonant portion during which said switch and said
rectifier are both off; and wherein substantially no current flows
into said first connection node during said second portion of said
voltage waveform; whereby said forward converter is controllable by
said controller to operate in a discontinuous forward voltage
conversion mode.
Preferably the controller is implemented on a single integrated
circuit as described above.
In further aspects the invention provides a method of controlling a
forward converter as described above to operate in a discontinuous
resonant mode by controlling the controllable (power) switch such
that a voltage waveform on the secondary winding has a first
portion during which the switch is on and current flows into the
first connection node and the second substantially resonant portion
during which the switch is off and wherein substantially no current
flows into the first connection node during the second portion of
the voltage waveform.
A controller as described above may be implemented in either
analogue or digital circuitry. Thus, where the controller is
implemented mainly or wholly in digital circuitry the invention
further provides a carrier medium carrying processor control code
such as RTL (Register Transfer Level) or SystemC defining hardware
to implement the controller.
The skilled person will understand that a discontinuous resonant
forward converter as described above may be implemented using a
range of circuit topologies including, but not limited to, those
described later. The transformer, for example, may comprise an
auto-transformer.
BRIEF DESCRIPTION OF THE DRAWINGS
These and other aspects of the invention will now be further
described, by way of example only, with reference to the
accompanying figures in which:
FIG. 1 shows an example of a forward converter according to the
prior art.
FIGS. 2a and 2b show, respectively, an embodiment of a
discontinuous resonant forward converter, and an example timing and
control arrangement for the converter of FIG. 2a;
FIGS. 3a and 3b show example waveforms of the forward converter of
FIG. 2a during operation from a 170V dc input supply providing
output currents of, respectively, 1 A and 2 A;
FIG. 4 shows an equivalent circuit model of a forward converter
power supply according to an embodiment of the invention;
FIGS. 5a to 5d show alternative topologies for a forward converter
according to an embodiment of the invention;
FIGS. 6a to 6c show examples of using an auxiliary winding to reset
a transformer of a forward converter according to an embodiment of
the invention.
FIGS. 7a and 7b show waveforms for a forward converter respectively
without and with high frequency control during start-up;
FIGS. 8a to 8c show, respectively, input sensing connections for an
embodiment of a discontinuous resonant forward converter according
to the invention. A forward converter in, respectively, overload
and no load conditions;
FIGS. 9a to 9c show examples of, respectively, late, early and
target timings for waveforms of a forward converter; and
FIGS. 10a and 10b show, respectively regulation of a forward
converter using secondary side feedback, and a multiphase forward
converter circuit.
DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS
Broadly speaking we will describe techniques for implementing a
resonant discontinuous forward converter (RDFC) which employ a
control system to turn a power switch of the RDFC on and off in a
controlled manner. As previously described, the control system may
operate in an uncontrolled, fixed frequency mode or the control
system may sense from one or more inputs and decide when to turn
the power switch on and off responsive to this sensing, for example
to implement pulse width and/or frequency modulation. This
facilitates regulation of the RDFC which, in detail, may be
performed using a range of algorithms. One technique uses the
control system to operate the RDFC to compensate for circuit
variables and to operate in a zero voltage switching (ZVS) mode.
The converter may also control the switching frequency during
start-up and/or current limit in order to protect the power switch
and increase the energy transferred to the load. The control system
is preferably implemented using a control IC (integrated
circuit).
As mentioned above, the RDFC operates without a freewheeling or
flyback diode, and with or without an output inductor. However, if
present the output inductor is sufficiently small to ensure that
the forward converter operates in a discontinuous mode and
substantially resonantly that is at or close to resonance.
Referring now to FIG. 2a, this shows an embodiment of discontinuous
resonant forward converter 200 according to the invention. FIG. 2b
shows an example timing and control system 210 for the forward
converter of FIG. 2a.
Referring to FIG. 2a, this shows a fully resonant discontinuous
mode forward converter 200 with a dc input 202 coupled to the
primary winding 204 of a transformer 206, connected in series with
a power switch 212. A resonant capacitor 214 is connected across
the primary winding of the transformer and the dc input 202 is
provided with a smoothing capacitor 216. On the output side of the
forward converter a secondary winding 208 of the transformer
provides power to a pair of dc output terminals 218 via a rectifier
220. A smoothing capacitor 222 is connected across the dc output
terminals 218 and an output node at the junction of rectifier 220,
smoothing capacitor 222 and a connection to one of the dc output
terminals 218 is denoted "X". The current into node X, which flows
to either or both of the smoothing capacitor 222 and output 218, is
discontinuous, by contrast with the circuit shown in FIG. 1.
The switch 212 may comprise a bipolar or MOS transistor such as a
MOSFET or IGBT, or some other device. The rectifier 220 may be
implemented as a diode or by means of a MOS transistor. The
resonant capacitor 214 may either comprise a discrete component, or
may be entirely provided by parasitic capacitance, or may comprise
a combination of the two.
The switch 212 is controlled by a controller 210 comprising a
timing control module 210a and a switch control module 210b, the
timing control module providing switch on and switch off signals
210c to the switch control module 210b. The timing control module
may have one or more sense inputs, such as a voltage sense input
and a current sense input as illustrated, or such sensing may be
omitted and the timing control module 210a may operate
substantially independently of any sensed condition of the forward
converter circuit.
Where voltage sensing is employed the voltage on the primary
winding of the transformer may be sensed, either directly or
indirectly. For example the voltage may be sensed as shown by means
of a connection to a junction between the primary winding and
switch; alternatively, for example, a sensing voltage may be
derived from an auxiliary winding of the transformer (not shown in
FIG. 2a). Where current sensing is employed this may be
conveniently implemented by sensing the voltage across a current
sense resistor.
In operation the circuit of FIG. 2a converts the input dc voltage,
typically relatively high, to an output dc voltage, typically in a
range suitable for consumer electronic devices, for example between
around 5V and 20V. In some preferred implementations, the dc output
is isolated from the dc input, as shown in FIG. 2a; in other
implementations secondary side feedback may be employed, in which
case an opto-isolator may be included to provide isolation between
the primary and secondary sides of the forward converter.
In general forward converters have a number of advantages including
relatively small size and low cost: However conventionally they
have been difficult to regulate and the components, particularly
the switch, have been prone to failure under some load conditions
and at start-up. Theoretically they have a good efficiency because
they may be operated in resonant mode although the conventional
freewheeling or flyback diode can prevent resonance from being
achieved. Further, conventionally resonance is achieved by careful
choice of component values allowing self-resonance, but this
entails the use of components with a tight tolerance, which is
costly and increases the difficulty of manufacture.
The arrangements we describe employ a controller 210 to control the
timing of the switch 212 on and off, and this allows a variety of
advantageous techniques to be employed. Thus we describe below how
the forward converter of FIG. 2a can be made to operate reliably
over a range of component values, how current limiting and start-up
control can be employed (which both help to achieve reliable
operation and which help to protect switch 212), and how switching
timing can be controlled in a discontinuous resonant mode to
achieve regulation.
FIG. 2b illustrates an example implementation of the controller 210
of FIG. 2a. A comparator 250 compares a sensed voltage with a
reference voltage, for example zero volts, to provide a control
signal 252 to a switch control unit 256 to control switch 212 on.
The output of comparator 250 is also provided to a timer 258 which
begins timing an on pulse width. When the timer times out a signal
is provided on a second control line 254 to switch control unit 256
to control switch 212 off. Switch control unit 256 may comprise,
for example, a set-reset latch together with interface circuitry
for driving the base of a bipolar transistor and/or the gate of an
MOS transistor. Preferably the circuit also includes an OR gate 260
with an input 262 from an over current protection line. This may be
generated by a comparing a current sense input with a reference
level defining a threshold for current limiting. When the over
current protection input 262 becomes active the switch control unit
256 is immediately controlled to switch 212 off, thus implementing
cycle-by-cycle current limit control.
FIGS. 3a and 3b show example waveforms illustrating the operation
of the forward converter of FIG. 2. In these figures (and similar
later figures) waveform 300 indicates the drive voltage on the base
of a bipolar transistor switch, waveform 302 shows a collector
current, which is substantially equal to a current through the
primary winding 204 of transformer 206 of the forward converter
200. The primary side current thus controls the flux in transformer
206 and hence also the secondary side current. Waveform 304 shows a
voltage on the collector terminal of the bipolar transistor switch;
when the switch is open this voltage, which is equal to the voltage
on the primary winding 204 of transformer 206, is reflected on the
secondary winding 208 of the transformer. When the switch is closed
the current in the primary side of the transformer drives a current
in the secondary side, thus charging smoothing capacitor 222 via
rectifier 220; when the switch is open the primary side of the
forward converter ceases to drive the secondary side and power is
supplied to output terminals 218 from smoothing capacitor 222 (and
diode 220 is off). In the waveforms of 3a and 3b the scale for
waveform 300 is 500 mV per division, for waveform 302 is mA per
division and for waveform 304 is 100V per division. In FIG. 3a the
frequency of the drive waveform is approximately 59 KHz; in FIG. 3b
the drive waveform has a frequency of approximately 48.4 KHz; Close
inspection of waveform 300 reveals that the switch-off of the drive
signal is not completely clean, which is due to the characteristics
of the bipolar switch; waveforms 302 and 304 correspond.
FIG. 4 shows an equivalent circuit for the discontinuous resonant
forward converter of FIG. 2a. This shows the parasitic capacitance
(C.sub.p) of the bipolar transistor switch, output rectifier and
transformer, as well as the resonant capacitor (C.sub.r), a
magnetising inductance (Lmag) which represents energy stored in a
transformer and a leakage inductance (Lleak) which represents
leakage inductance between the primary and secondary windings of
the transformer (because some flux lines leak linking the primary
and secondary coils so that they behave similarly to an inductor).
Generally, but not necessarily, C.sub.r is much greater than
C.sub.p. In operation Lmag keeps the primary current flowing into
C.sub.r causing resonance, and the secondary current approximately
matches the primary current. The leakage inductance provides a
degree of current limiting, in particular helping to reduce
overload at start-up when the smoothing capacitor can effectively
appear as a short circuit.
FIGS. 5a to 5d show alternative topological configurations for the
resonant discontinuous forward converter. In FIG. 5a the resonant
capacitor is coupled across the switch (in this example, shown as a
bipolar transistor switch). In FIG. 5b the resonant capacitor is on
the output side of the converter, more particularly, connected
across the secondary winding of the transformer. In FIG. 5c a small
inductor is explicitly included in series with the output
rectifier. FIG. 5d illustrates a configuration of the forward
converter in which an auto-transformer is employed.
In embodiments the transformer is reset by the resonant portion of
the transformer waveform: to demagnetise the transformer the
magnetisation current discharges into the resonant capacitor and
discharges resonantly. Additionally or alternatively the
transformer may be reset by means of an auxiliary winding coupled
in series with a rectifier. FIG. 6a shows an example of such a
reset circuit in which a primary side auxiliary winding of the
transformer has an opposite or inverted polarity compared with the
primary and secondary windings of the transformer. During the off
period of the switch a diode in series with the auxiliary winding
becomes forward biased and conducts power back to the dc input (so
that the technique is non-dissipative). FIGS. 6b and 6c show
alternative configurations in which the auxiliary winding is placed
on the secondary side of the transformer and (again) has an
opposite polarity to the primary and secondary windings (the diodes
on the secondary side are connected to opposite ends of the
windings). In these examples the auxiliary winding is connected in
series with a rectifier and across the secondary winding and
rectifier, and optionally inductor, in the output side of the
forward converter.
Referring again to FIGS. 2 and 3, it can be seen by comparing
waveforms 304 and 300 that there is a short period after waveform
304 goes to substantially zero volts before the drive signal 300 to
the bipolar transistor switch turns the switch on. If there were
sufficient delays further resonance would eventually be seen in
waveform 304 but nonetheless it can be appreciated that there is a
range of periods during which the switch may be once again turned
on and thus the switch may be controlled by detecting a
substantially zero voltage level of waveform 304 after its resonant
half cycle and then waiting for a delay (which may be zero) before
turning the switch on. This tolerance in the operation of the
circuit allows the switching timing (more specifically, the switch
off time) to be sufficiently long to cope with a range of resonant
frequencies, and hence resonant component values.
We next consider start-up of the forward converter. On start-up the
output of the power supply appears as a short circuit. Unlike
continuous forward converters, which employ a flyback diode,
depending upon the load present on the RDFC insufficient energy may
be transferred to the output of the converter to charge the output
capacitor. This is particularly a problem where current limiting is
employed since very high currents can appear on the primary side of
the transformer and the current limiting can activate to switch off
the drive signal which can have the consequence that, with certain
loads, the output capacitor may not be charged.
FIG. 7a illustrates this difficulty showing that, with current
limiting, during start-up the output (voltage) of the power supply
may not rise up to its correct value. Inspection of the collector
voltage waveform also reveals that there is a non-zero component to
this when the switch is off (because the secondary side output is
reflected in reverse) and this non-zero collector voltage may be
sensed in order to identify this start-up condition, as well as
current limit, overload and short circuit if desired.
In preferred embodiments of the discontinuous resonant forward
converter, the forward converter is controlled to operate in an
increased frequency mode at start-up, for example at 5 or 10 times
a normal frequency of operation. This may be implemented by means
of a simple oscillator selected at start-up or the collector
voltage may be sensed and used to control the switch on to invoke a
higher frequency mode of operation. Operating the RDFC at an
increased frequency increases the charge transferred to the output
whilst still protecting the power switch. FIG. 7b illustrates this
increased frequency operation (the time divisions are shorter than
those shown in FIG. 7a) and it can be seen that the output voltage
in this high-frequency start-up mode has an upward trend. Over time
the output voltage increases to a normal operating output voltage
for the forward converter.
We next describe current limiting systems for an embodiment of a
discontinuous resonant forward converter according to the
invention.
Once the RDFC has started up and achieved steady state operation,
it operates in a resonant mode with an output (voltage) that tracks
the input (voltage). However if an overload is applied, in
particular when operating at a fixed frequency, the output current
and hence the switch current will increase significantly and the
circuit may be damaged. It is therefore desirable to sense the
switch current in the RDFC and the controller we describe enables
the drive to be shortened to control the drive current in an
overload condition.
FIG. 8a shows an embodiment of an RDFC which includes a controller
with current sense terminals (Si) as well as collector voltage (Sc)
and dc input voltage (Sdc) sense inputs. FIG. 8b illustrates
waveforms of a forward converter during overload, in which it can
be seen that the collector voltage waveform is no longer properly
resonant (the first portion of the half cycle having been
truncated) because of loading by the output circuit. For
comparison, FIG. 8c shows a forward converter under no-load
conditions.
We have described above how over current protection may be
implemented. However there are situations in which a fixed current
limit converter can reduce the power transferred to the output,
this in turn reducing the output voltage, which increases the
output current, which can result in the converter output voltage
falling significantly, even when the load is removed. In this
situation it is possible that the forward converter may not
recover. To address this one or more of a number of strategies may
be employed. For example an increased frequency re-start may be
employed, effectively as described above, to bring the output
voltage back up to its normal operating level. Additionally or
alternatively an output side inductance may be employed and/or the
leakage inductance of the transformer may be controlled (generally
allowed to increase) in order to provide a current limiting effect.
Also, the current limit may be varied, increasing the current limit
as the pulse width reduces. This latter strategy, in particular, is
described in more detail later.
In more detail, in some applications, such as a constant current
load, the output voltage may enter a state in which it continuously
falls and in which the power supply is not able to deliver full
power. By increasing the frequency in a similar manner to that
described above during start-up the power delivered to the load can
be increased, thus increasing the output voltage. In this way it is
also possible to regulate whilst in current limit at a reduced
output voltage; the leakage and/or a series inductance may also be
employed to drop a part of the output voltage across this
inductance.
When the forward converter is operating in a current limited mode
it is nonetheless possible to regulate the output current by
increasing the allowable switch current as the pulse width is
reduced. This can be achieved safely in an RDFC according to an
embodiment of the invention because the risk of damage to the
converter is reduced with reducing pulse width. Combining this with
the leakage inductance of the transformer and/or a series output
inductance enables the output current to be regulated as the output
voltage falls. Thus, broadly speaking the effect is that an
increasing pulse width results in a reduced current limit.
We now discuss further techniques which can be employed to
compensate for the use of components with relatively wide
tolerances. It is difficult to manufacture a power transformer with
a tight tolerance primary magnetising inductance. One technique is
to clean and glue the cores, but this is expensive. A tight
tolerance resonant capacitor is also expensive. We have previously
described how a fixed frequency oscillator in the controller can be
employed together with a suitable choice of duty cycle to
compensate for increased tolerances in these components. Another
technique comprises compensating for tolerances by controlling the
switch so that it turns on during the zero voltage phase of the
primary (voltage) waveform. As previously described, there is a
dead time while the switch voltage is at approximately zero volts
(in practice the voltage may be slightly below ground potential).
In zero voltage switching (ZFS) embodiments of the controller, the
power switch is turned on during this time interval. FIGS. 9a to 9c
illustrate different example timings for the switch drive with
respect to the collector voltage waveform.
Referring to FIG. 9a, this shows an example in which zero voltage
switching is achieved but in which the switch is turned on later
than ideal. However this is preferable to turning the switch on too
early, as shown in FIG. 9b, which can result in non-zero voltage
switching, which causes switching losses and electromagnetic
interference (EMI). FIG. 9c shows a preferred timing of the switch
drive with respect to the collector voltage waveform.
A preferred timing of FIG. 9c can be achieved by sensing when the
collector voltage is at zero volts and turning the switch on in
response to this, either as soon as the collector voltage has
fallen to zero, or a short time after the voltage has reached zero,
or just as the collector voltage starts to rise again. The timing
of FIG. 9c illustrates that of a "perfect" resonant switch, with
the switch turning on just as the collector voltage reaches
zero.
We next discuss regulation of the output voltage of an RDFC. In
general the regulation can be poor due to relatively high leakage
inductance and component (winding) resistances. The result of this
is that as more load is applied to the converter, the output
voltage falls. Further an RDFC can have difficulty in compensating
for variations in input voltage and, in general, the output voltage
tracks the input voltage. This can be a particular problem in
forward converters run off a grid mains supply because the mains
voltage can often vary significantly. However embodiments of the
controller described above are suitable for implementation of one
or both of pulse width and pulse frequency control in order to
regulate the output voltage of an RDFC. More particularly,
increasing the pulse width and/or increasing the frequency during
either or both of low input and high load conditions can improve
regulation.
FIG. 10a shows another technique which may be employed for output
voltage regulation. In this arrangement an input voltage converter,
either an ac-to-dc or a dc-to-dc converter is used to provide a dc
input power supply to the forward converter, and this is controlled
by feedback from the secondary side of the forward converter. In
order to regulate the output voltage. The input converter may
comprise a boost or buck or PFC (Power Factor Correction)
stage.
FIG. 10b illustrates the use of two power transformers in a
multiphase configuration to improve output regulation. In the
arrangement of FIG. 10b the switches are controlled so that each
switch is only driven on when the other switch is off, creating
complementary, but non-overlapping, drive waveforms. This technique
is useful for small size forward converters according to
embodiments of the invention operating at relatively high
frequencies where regulation may be poor.
Broadly speaking we have described resonant discontinuous forward
converters which employ a controller to analyse one or more inputs
and determine turn-on and turn-off times for a power switch,
providing a drive signal accordingly (although in simple
embodiments a substantially fixed frequency/duty cycle drive may be
employed). In embodiments the pulse width and/or frequency is
adjusted in accordance with the resonance circuit in order to
alleviate tolerance issues in the resonant components, either using
sensing signals input to the controller or by means of a
free-running oscillator. Preferably, to ensure that the maximum
energy is passed through the RDFC without significantly
compromising the resonant behaviour and increasing losses or EMI
the controller is configured to implement zero (switch) voltage
switching. Preferably the controller is configured to terminate an
on-pulse when an over current condition is detected, in order to
protect the circuit (switch) and/or load. Preferably embodiments of
the RDFC employ an increased frequency during start-up and/or
current limit in order to assist the output voltage rise. Either or
both of PWM and PFM techniques may be employed in order to improve
load and line regulation.
No doubt many other effective alternatives will occur to the
skilled person. It will be understood that the invention is not
limited to the described embodiments and encompasses modifications
apparent to those skilled in the art lying within the spirit and
scope of the claims appended hereto.
* * * * *