U.S. patent number 7,492,812 [Application Number 11/209,627] was granted by the patent office on 2009-02-17 for rfid transceiver device.
This patent grant is currently assigned to Fujitsu Limited. Invention is credited to Yusuke Kawasaki, Osamu Kuroda, Teruhisa Ninomiya, Yoshinori Tanaka.
United States Patent |
7,492,812 |
Ninomiya , et al. |
February 17, 2009 |
**Please see images for:
( Certificate of Correction ) ** |
RFID transceiver device
Abstract
An RFID transceiver device is capable of high sensitivity
reception, by the reduction of noise, irrespective of the distance
to the tag. The RFID transceiver device includes a delay circuit
between a local oscillation circuit and a demodulation circuit,
wherein the amount of delay of the delay circuit is set to a
magnitude corresponding to the path difference between the path of
leakage, via a duplexer into the demodulation circuit, of
transmission signal output from the local oscillation circuit for
transmission, and the path of direct input of the local oscillation
signal from the local oscillation circuit to the demodulation
circuit.
Inventors: |
Ninomiya; Teruhisa (Kawasaki,
JP), Kawasaki; Yusuke (Inagi, JP), Kuroda;
Osamu (Kawasaki, JP), Tanaka; Yoshinori
(Kawasaki, JP) |
Assignee: |
Fujitsu Limited (Kawasaki,
JP)
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Family
ID: |
36675932 |
Appl.
No.: |
11/209,627 |
Filed: |
August 24, 2005 |
Prior Publication Data
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Document
Identifier |
Publication Date |
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US 20060229032 A1 |
Oct 12, 2006 |
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Foreign Application Priority Data
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Apr 8, 2005 [JP] |
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2005-111531 |
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Current U.S.
Class: |
375/219 |
Current CPC
Class: |
G06K
7/0008 (20130101); H04B 1/525 (20130101) |
Current International
Class: |
H04L
5/16 (20060101) |
Field of
Search: |
;375/219,222 |
References Cited
[Referenced By]
U.S. Patent Documents
Foreign Patent Documents
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1-1710-727 |
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Jul 2006 |
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EP |
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2-300-318 |
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Oct 1996 |
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GB |
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2003-174388 |
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Jun 2003 |
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JP |
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Primary Examiner: Bocure; Tesfaldet
Attorney, Agent or Firm: Hanify & King, P.C.
Claims
What is claimed is:
1. An RFID transceiver device comprising: a local oscillation
circuit that generates a local oscillation signal; a demodulation
circuit that demodulates a reception signal using the frequency of
the local oscillation signal output from said local oscillation
circuit; a transmission circuit that modulates, amplifies and
transmits a local oscillation signal output from said local
oscillation circuit; a duplexer that supplies to a transceiving
antenna the transmission signal from said transmission circuit and
branches the reception signal received by said transceiving antenna
to said demodulation circuit; and a delay circuit between said
local oscillation circuit and said demodulation circuit, wherein
the amount of delay of said delay circuit is set to a magnitude
corresponding to the path difference between the path of the
leakage, via said duplexer into said demodulation circuit, of the
transmission signal that is output from said local oscillation
circuit for transmission, and the path of direct input of the local
oscillation signal from said local oscillation circuit to said
demodulation circuit.
2. The RFID transceiver device according to claim 1, further
comprising a control and processing circuit that detects noise
level in accordance with the output of said demodulation circuit,
wherein said control and processing circuit controls the delay
amount of said delay circuit on the basis of said detected noise
level.
3. An RFID transceiver device comprising: a local oscillation
circuit that generates a local oscillation signal; a demodulation
circuit that demodulates a reception signal using the frequency of
the local oscillation signal output from said local oscillation
circuit; a modulation circuit that modulates the local oscillation
signal output from said local oscillation circuit; a duplexer that
supplies to a transceiving antenna the transmission signal output
from said modulation circuit and branches the reception signal
received by said transceiving antenna to said demodulation circuit;
and a path for supplying the transmission signal output from said
modulation circuit as the local oscillation signal to said
demodulation circuit, wherein the amount of delay on said path and
the amount of delay on the path of the leakage of the transmitted
carrier signal that is input to said demodulation circuit through
said duplexer are set to the same magnitude.
4. The RFID transceiver device according to any of claim 1 further
comprising: a transceiving antenna; and a delay circuit that
connects said transceiving antenna to said duplexer, wherein the
amount of delay of the delay circuit is set such that the impedance
of said transceiving antenna seen from said duplexer is
substantially equal to the characteristic impedance.
5. The RFID transceiver device according to claim 2, wherein
detection of the noise level in said performed in a condition in
which command transmission from said transmission circuit to the
tag is halted.
Description
CROSS-REFERENCE TO RELATED APPLICATIONS
This application is based upon and claims the benefit of priority
from the prior Japanese Patent Application No. 2005-111531, Filed
on Apr. 8, 2005, the entire contents of which are incorporated
herein by reference.
BACKGROUND OF THE INVENTION
1. Field of the Invention
The present invention relates to an RFID transceiver device in an
RFID system, and more particularly to an RFID transceiver device in
which receiver noise is improved.
2. Description of the Related Art
As shown in FIG. 1, in an RFID system, a carrier signal is
transmitted (P1) from an interrogator constituted by an RFID
transceiver device 1 to a transponder for example an IC tag 2. The
IC tag 2 modulates the received carrier signal with information
data and sends this back to the RFID transceiver device 1 by
reflection (back scattering). The RFID transceiver device 1
acquires information data by demodulating the reflected signal.
FIG. 2 shows an example of the configuration of an RFID transceiver
device. This RFID transceiver device is connected to a data
processing device, not shown, through an external interface I/F. A
control and processing circuit 10 controls a local oscillation
circuit 11 to generate a local oscillation signal corresponding to
each channel.
The local oscillation signal that is generated from the local
oscillation circuit 11 is modulated and amplified in power by a
transmission circuit 12 before being emitted from an antenna 16
through a duplexer 13. The local oscillation signal is additionally
supplied to a demodulation circuit in a reception circuit 14 and
the demodulation circuit outputs information data by demodulating
the reflected signal from the IC tag 2.
It is undesirable from the point of view of cost and size that
separate antennas should be provided respectively for transmission
and reception in FIG. 2, so a transceiving antenna 16 as shown in
FIG. 2 may be employed for the RFID transceiver device 1.
Furthermore, since, in the case where the IC tag 2 is a passive
tag, the operating power (power source energy) is obtained from the
electromagnetic wave transmitted by the RFID transceiver device 1,
the RFID transceiver device 1 needs to have large transmission
power. In contrast, since the response transmission from the IC tag
2 is performed by reflection (back scattering), its power is very
weak in comparison with the power of the electromagnetic wave
transmitted by the RFID transceiver device. Thus, the RFID
transceiver device 1 whose communications partner is a passive IC
tag needs to have high output power in order to supply power source
energy to the IC tag 2 and, at the same time, must be provided with
a high sensitivity reception capability, since the back-scattered
signal from the passive IC tag is very weak.
When a transceiving antenna 16 is employed, in order to isolate the
transmission and reception signal, a duplexer (typically
constituted by a circulator or coupler) 13 is provided; however, as
mentioned above, the energy of the transmission signal is large, so
leakage 15 of the transmission signal is generated, of a level that
depends on the degree of isolation achieved by the duplexer 13.
Also, as shown in FIG. 3, apart from the component 15a that passes
through the duplexer with attenuation, the leakage components of
the transmission signal include a component 15b that is reflected
by the power feed terminal of the antenna 16. Also, as shown in
FIG. 4, if the transmission and reception signal frequencies f1 and
f2 are different, as they are in the case of for example a mobile
telephone terminal, isolation of transmission and reception can be
achieved by providing bandpass filters 12a, 14a as well as the
duplexer 13. However, in the case of an RFID system, as shown in
FIG. 3, the frequency of transmission (carrier signal) and
reception (tag reflection signal) are the same, so isolation using
filters is not possible. For the above reasons, phase noise of the
carrier signal appears to be detected at the output of the
demodulation circuit constituting the reception circuit 14. Also,
due to the problem of saturation caused by the leakage of the
transmission signal, low noise amplification cannot be performed
upstream of the demodulation circuit.
The mechanism by which such phase noise is detected will further be
described with reference to the drawings. FIG. 5 is an example of a
specific configuration of the transmission circuit 12 and reception
circuit 14 of the RFID transceiver device 1 shown in FIG. 2.
FIG. 6 is a view showing the input signal of the demodulation
circuit 14b constituting the reception circuit 14. The input
signals of the demodulation circuit 14b are the local oscillation
signal 17 (FIG. 6A) from the local oscillation circuit 11 and the
leakage component 15 (FIG. 6B) of the transmission signal including
the component 15a that is transmitted, with attenuation, through
the duplexer 13 and the reflected signal 15b from the antenna power
feed terminal.
Consequently, assuming that the operation of the demodulation
circuit 14b is multiplicative, when the higher order component is
discarded, the output of the demodulation circuit 14b may be
expressed by the expression (1).
.function..omega..times..times..function..times..function..omega..functio-
n..tau..function..tau..times..function..omega..times..times..function..fun-
ction..tau. ##EQU00001##
The term that determines the magnitude of the phase noise component
in the output of the demodulation circuit in expression (1) i.e.
P[t]-P[t-.tau.]
is 0 when .tau.=0. In contrast, it increases with increasing .tau.
if the phase noise is time-correlated.
However, in the above expression, cos[.omega.t+P[t]]
is the local oscillation signal (FIG. 6A) from the local
oscillation circuit 11, and cos[.omega.(t-.tau.)+P[t-.tau.]]
is the leakage 15 (FIG. 6B) of the transmission signal.
On the other hand, the phase noise component is expressed by
.function..intg..times..times.d.times..intg..times..function..times..func-
tion..times..times.d ##EQU00002##
where g[t]
is the input noise of the VCO of the local oscillation circuit 11
and h[t]
is the frequency characteristics (loop filter characteristics) of
the VCO input stage. Thus the phase noise component has a
correlation with time.
From the above relationship, it can be seen that, if the paths to
the output of the demodulation circuit 14b are respectively
different for the local oscillation signal from the local
oscillation circuit 11 and for the leakage 15 of the transmission
signal, so that there is a time difference between the paths to the
demodulation circuit as shown in FIG. 6, the correlation of the
leakage 15 (FIG. 6B) of the transmission signal and the local
oscillation signal (FIG. 6A) becomes smaller as the path time
difference .tau. becomes larger: as a result, the noise component
that is output from the demodulation circuit 14b also becomes
larger. FIG. 7 is a graph showing the relationship between the path
time difference and the noise level (relative value). From the
graph of FIG. 7, it can be understood that the detected phase noise
level becomes larger as the path time difference .tau. becomes
larger and if there is no path time difference, the phase noise
component is substantially cancelled out.
Laid-open Japanese Patent Application No. 2003-174388 may be
mentioned as prior art. This Laid-open Japanese Patent Application
No.2003-174388 mentions that the phase noise possessed by the
carrier itself that is transmitted from the interrogator and the
phase noise of the PLL oscillation circuit that is involved in
synchronous detection appear in the demodulation signal and
adversely affect reception sensitivity. The object of the invention
set out in Laid-open Japanese Patent Application No. 2003-174388
referred to above is to prevent lowering of the reception
sensitivity in synchronous detection by the interrogator.
However, the invention set out in Laid-open Japanese Patent
Application No. 2003-174388 referred to above is an arrangement in
which the phase of the local signal LO is corrected using the
response signal from the tag as a reference. Such a configuration
is effective in systems in which there is substantially no change
in the amplitude/phase of the response signal and the leakage of
the transmission signal i.e. systems in which the frequency is low,
at about 13.56 MHz, and in which the distance to the transponder is
small, at about 30 cm.
However, in the case where an RFID transceiver device and IC tag
are employed with a distance of a few m in the UHF band (860 MHz to
960 MHz) or higher frequency bands, phase variations of 10 or more
times 360.degree. are experienced, depending on the distance.
SUMMARY OF THE INVENTION
An object of the present invention is therefore to provide an RFID
transceiver device that makes possible noise reduction even under
conditions in which the invention set out in Laid-open Japanese
Patent Application No. 2003-174388 referred to above cannot be
used, irrespective of the distance to the tag and which thus makes
possible high sensitivity reception.
According to a first aspect, an RFID transceiver device whereby the
above object of the present invention is achieved is characterized
in that it comprises: a local oscillation circuit that generates a
local oscillation signal; a demodulation circuit that demodulates a
reception signal using the frequency of the local oscillation
signal output from the local oscillation circuit; a transmission
circuit that modulates and amplifies and transmits a local
oscillation signal output from the local oscillation circuit; a
duplexer that supplies to a transceiving antenna the transmission
signal from the transmission circuit and that branches the
reception signal received by the transceiving antenna to the
demodulation circuit; and, in addition, a delay circuit between the
local oscillation circuit and the demodulation circuit, the amount
of delay of the delay circuit being set to a magnitude
corresponding to the path difference between the path of the
leakage through the duplexer into the demodulation circuit, of the
transmission signal output from the local oscillation circuit for
transmission and the path of the direct input from the local
oscillation circuit to the demodulation circuit.
According to a second aspect, in the first aspect, an RFID
transceiver device whereby the above object of the present
invention is achieved, is characterized in that it further
comprises a control and processing circuit that detects noise level
in accordance with the output of the demodulation circuit and the
control and processing circuit performs feedback control of the
delay amount of the delay circuit in response to the noise level
that is detected.
According to a third aspect, an RFID transceiver device whereby the
above object of the present invention is achieved is characterized
in that it comprises: a local oscillation circuit that generates a
local oscillation signal; a demodulation circuit that demodulates a
reception signal using the frequency of the local oscillation
signal output from the local oscillation circuit; a modulation
circuit that modulates the local oscillation signal that is output
from the local oscillation circuit; a duplexer that supplies to a
transceiving antenna the transmission signal output from the
modulation circuit and that branches the reception signal received
by the transceiving antenna to the demodulation circuit; and, in
addition, the amount of delay on the path whereby the local
oscillation signal that is output from the modulation circuit is
supplied to the demodulation circuit being set identical to the
amount of delay on the path of the leakage of the transmission
signal that is input to the demodulation circuit through the
duplexer.
According to a fourth aspect, an RFID transceiver device whereby
the above object of the present invention is achieved is
characterized in that, in any of the first to the third aspects, it
further comprises a transceiving antenna and a delay circuit that
connects the transceiving antenna and the duplexer, the amount of
delay of this delay circuit being set such that the impedance of
the transceiving antenna seen from the duplexer is substantially
equal to the characteristic impedance.
According to a fifth aspect, an RFID transceiver device whereby the
above object of the present invention is achieved is characterized
in that, in the second aspect, detection of the noise level in the
control and processing circuit is performed in a condition in which
command transmission from the transmission circuit to the tag is
halted.
According to the present invention, an RFID transceiver device is
obtained wherein lowering of the noise level can be achieved
irrespective of the distance to the tag and consequently high
sensitivity reception can be achieved and wherein system
stabilization can be achieved at a frequency in the UHF band or
above.
Characteristic features of the present invention will become
additionally apparent from the embodiments of the present invention
described below with reference to the drawings.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a view given in explanation of an RFID system;
FIG. 2 is a block diagram of an RFID transceiver device;
FIG. 3 is a view given in explanation of the leakage component of
the transmission signal and reflection at the antenna power feed
terminal;
FIG. 4 is a view given in explanation of the leakage component of
the transmission signal and reflection at the antenna power feed
terminal in the case where the up and down frequencies are
different;
FIG. 5 is a view showing a specific example of the configuration of
a transmission circuit and a reception circuit of the RFID
transceiver device 1 shown in FIG. 2;
FIG. 6 is a view showing the input signal of the demodulation
circuit constituting the reception circuit;
FIG. 7 is a graph showing the relationship between the path time
difference and the noise level (relative value);
FIG. 8 is a block diagram of a first embodiment of an RFID
transceiver device according to the present invention;
FIG. 9 is a block diagram of a second embodiment of an RFID
transceiver device according to the present invention;
FIG. 10 is a block diagram of a third embodiment of an RFID
transceiver device according to the present invention;
FIG. 11 is a schematic diagram of the demodulation circuit 14b in
FIG. 10;
FIG. 12 shows the processing flow in respect of the delay circuit
18 of the control and processing circuit 10 in the case where a
demodulation circuit according to FIG. 11 is employed;
FIGS. 13A and 13B are a view given in explanation of control of the
amount of delay in the delay circuit 18;
FIG. 14 is a view given in further explanation of the path time
difference and the noise level (relative value) illustrated in FIG.
7;
FIG. 15 is a graph found for the path time difference and noise
level (relative value), for the sum (I.sup.2+Q.sup.2) of the noise
of the I and Q channels;
FIG. 16 is a view showing yet a further embodiment of the present
invention;
FIG. 17 is an embodiment of the delay circuit 20;
FIG. 18 is a graph showing the relationship between the antenna
impedance Z and line length l; and
FIG. 19 is an example extending the embodiment of FIG. 16.
DESCRIPTION OF THE PREFERRED EMBODIMENTS
Embodiments of the present invention are described below with
reference to the drawings. It should be noted that the embodiments
are given to assist understanding of the present invention and the
technical scope of the present invention is not restricted
thereto.
FIG. 8 is a block diagram of a first embodiment of an RFID
transceiver device according to the present invention. In
comparison with the prior art configuration shown in FIG. 5, a
characteristic feature of this embodiment is the provision of a
delay circuit 18 between the local oscillation circuit 11 and
demodulation circuit 14b.
In FIG. 8, a phase difference is produced because the path of the
leakage signal 15 whereby the transmission signal that is
transmitted through the modulation circuit 12b and power amplifier
12c from the local oscillation circuit 11 leaks from the duplexer
13 to the demodulation circuit 14b is longer than the path of the
local oscillation signal 17 that is directly supplied to the
demodulation circuit 14b from the local oscillation circuit 11.
Consequently, a characteristic feature in FIG. 8 is that the path
of the leakage signal 15 is equalized by using a delay circuit 18
to provide a delay on the path of the local oscillation signal 17
that is directly supplied to the demodulation circuit 14b from the
local oscillation circuit 11. In this way, as shown in FIG. 7, the
same situation is produced as when the path difference is "0" and
the relative noise level can thus be reduced to a minimum.
FIG. 9 is a block diagram of a second embodiment of an RFID
transceiver device according to the present invention. A
characteristic feature of the second embodiment is that the circuit
configuration is such that the path of the local oscillation signal
17 that is supplied from the local oscillation circuit 11 to the
demodulation circuit 14b is substantially equal to the path of the
leakage signal 15 from the local oscillation circuit 11 via the
duplexer 13.
Specifically, instead of supplying the local oscillation signal
directly from the local oscillation circuit 11 to the demodulation
circuit 14b, a coupling circuit 12d is provided downstream of the
power amplifier 12c so as to reduce the difference with regard to
the path of the local oscillation signal 17 supplied to the
demodulation circuit 14b and the leakage signal 15 via the duplexer
13. Additionally, the amount of delay corresponding to the minute
difference of the existing paths is adjusted by providing a
trimming circuit 18 between the coupling circuit 12d and the
trimming circuit 14b.
Thus, in the embodiment of FIG. 9 also, phase noise can be reduced
since the paths of the local oscillation signal 17 supplied to the
demodulation circuit 14b and of the leakage signal 15 via the
duplexer 13 can be made substantially the same.
FIG. 10 shows a third embodiment. A characteristic feature of this
embodiment is that the amount of delay of the delay circuit 18 in
the first embodiment of FIG. 8 can be suitably controlled in
accordance with the noise level that is received and detected.
Specifically, the output of the demodulation circuit 14b is input
to an A/D converter 14d through an amplifier 14c and low-pass
filter 14 to be converted to a digital signal. The digital output
of the A/D converter 14d is input to the control and processing
circuit 10, which evaluates the noise level.
The control and processing circuit 10 supplies a correction amount
control signal 19 depending on the noise level to the delay circuit
18 so as to provide a corresponding amount of delay. In this way it
is possible to control the delay amount appropriately in accordance
with the noise level. The control and processing circuit 10 may be
configured with a suppliment of a correspondence table between the
correction level and correction amount control signal in order to
perform such control.
FIG. 11 is an example of the configuration of the demodulation
circuit 14b in FIG. 10. FIG. 12 shows the processing flow in
respect of the delay circuit 18 of the control and processing
circuit 10 when the demodulation circuit of FIG. 11 is employed.
Also, FIG. 13 is a view given in explanation of the delay amount
control in the delay circuit 18.
We shall now return to the description of FIG. 11. The demodulation
circuit 14b comprises a quadrature separation circuit 141 that
separates the reception signal RX into a mutually orthogonal I
channel signal and Q channel signal, a multiplier 142 that
multiplies the I channel signal and the local oscillation signal
(LO) 17 that is output from the local oscillation circuit 11, and a
multiplier 143 that multiplies the Q channel signal by the local
oscillation signal that is output from the local oscillation
circuit 11 and the phase of which is shifted by 90.degree. through
the phase shifter 144.
In FIG. 12, when the amount of delay of the delay circuit 18 is set
by performing calibration, this is performed (step S1) in a
condition with transmission of commands from the RFID transmission
device to the IC tag disabled. The control and processing circuit
10 finds (step S2) the initial power P1 (=I.sup.2+Q.sup.2) of the
demodulation circuit output by inputting the I and Q channel
demodulation signals that are output from the demodulation circuit
14b.
When calibration of the delay circuit 18 is performed as described
above, the power (I.sup.2+Q.sup.2) is thus found in a condition in
which transmission of commands from the RFID transmission device to
the IC tag is disabled and therefore corresponds to the noise level
resulting from the path difference.
Next, the delay amount .tau. of the delay circuit 18 is increased
by .DELTA. .tau.1 (step S3). The output power P2 (=I.sup.2+Q.sup.2)
of the demodulation circuit 14b at this point is then found (step
S4). The power P2 when this delay amount .tau. is increased by
.DELTA. .tau.1 and the initial power P1 are then compared (step
S5). In this power comparison, if P2<P1 (Yes in step S5), as the
path of the local oscillation signal (LO) 17 that is directly input
to the demodulation circuit 14b becomes larger, its difference from
the path of the leakage component that arrives via the duplexer 13
becomes smaller, indicating that the noise level becomes
smaller.
FIG. 13 shows the relationship between this path difference and
noise level. In addition, FIG. 13A shows the characteristic whereby
the noise level increases or decreases centered on the point
corresponding to a phase difference .lamda./2 of the path
difference; in this figure, the target value for feedback control
is a minimum noise level.
FIG. 13B represents the path difference of FIG. 13A extended in the
positive and negative directions and shows the direction of control
towards the target value at which the noise level is a minimum,
when the amount of delay of the delay circuit 18 is controlled in a
range in which the path difference is smaller than the path
difference corresponding to a phase difference .lamda./2.
It can be understood that, in the step S5 referred to above,
P2<P1 corresponds for example to the direction I of control in
FIG. 13B.
Furthermore, returning to FIG. 12, if P2<P1 (Yes in step S5),
P1=P2 is set (step S6) and processing returns to step S3, in which
a further delay amount .DELTA. .tau.1 is additionally set; the
processing of step S4 and the subsequent steps is then
continued.
In contrast, if, in step S5, P2>P1 (No in step S5), the delay
amount .tau. is set (step S7) in the direction such as to be
reduced by an amount .DELTA. .tau.2 (<.DELTA. .tau.1). Next, the
power P2 (=I.sup.2+Q.sup.2) of the demodulation circuit output at
this point is found (step S8). The power P2 when this delay amount
.tau. is reduced by .DELTA. .tau.2 and the initial power P1 are
compared (step S9).
In this power comparison, if P2<P1 (Yes in step S9), as the
direct path from the local oscillation circuit 11 to the
demodulation circuit 14b becomes smaller, its difference from the
path of the leakage component 15 through the duplexer 13 becomes
smaller, showing that the noise level decreases. This corresponds
to the direction II of control in FIG. 13B described above.
Consequently, in order to achieve convergence to the target value,
P1 is substituted by P2 (step S10) and, returning to step S7, the
delay amount is further reduced by .DELTA. .tau.2, and the
processing of step S7 and the subsequent steps is continued
with.
The control range (see FIG. 13A) of the above path difference will
now be examined. In the case of quadrature modulation, as shown in
FIG. 11, setting to the optimum value must be performed by the
combination of the I and Q channels. FIG. 14 is a view given in
further explanation of the path time difference and the noise level
(relative value) illustrated in FIG. 7; since the I channel and the
Q channel have a 90.degree. phase difference, if optimization is
effected in respect of one channel, the noise level of the other
channel becomes larger.
For example, in FIG. 14, even if the I channel is set to -85 dB,
since the Q channel has a phase difference of 90.degree. with
respect to the I channel, its noise level becomes larger, at -40 dB
(see FIG. 14, A). Consequently, control is performed such as to
achieve an optimum value in regard to the combination of I and Q
(see FIG. 14, B).
FIG. 15 is a graph in which the path time difference and the noise
level (relative value) are found in respect of the sum
(I.sup.2+Q.sup.2) of the noise of the I and Q channels. In FIG. 15,
when for example the path time difference is 0.5 (path difference
.lamda./2), and an improvement in the noise level of 10 dB can be
achieved by correcting the path time difference to 0.2 (path
difference .lamda./5) by controlling the amount of delay of the
delay circuit 18 (see FIG. 15, II.fwdarw.I).
Next, FIG. 16 is a view showing a further embodiment of the present
invention. A characteristic feature of this embodiment is that a
delay circuit 20 is provided between the duplexer 13 and the power
feed terminal of the antenna 16. In this case, the port allocation
of the duplexer is such that the TX terminal communicates with the
ANT terminal i.e. TX terminal.fwdarw.ANT terminal, but the TX
terminal.fwdarw.RX terminal is blocked, and coupling is effected
from the ANT terminal.fwdarw.RX terminal.
In this embodiment, as shown in FIG. 17, the delay circuit 20
comprises a delay line DL of line length l arranged between the
duplexer 13 and the antenna (load Z.sub.L in the Figure). The line
length l between the antenna 16 and the duplexer 13 can be adjusted
by means of this delay line DL.
The impedance Z of the antenna seen from the duplexer, for a line
length l, antenna load impedance Z.sub.L and line characteristic
impedance Z.sub.0 is as follows.
.times..times..times..times..beta..times..times..times..times..times..bet-
a..times..times..times..times..times..beta..times..times..times..times..ti-
mes..beta..times..times. ##EQU00003##
FIG. 18 is a graph showing the relationship between the impedance Z
of the antenna seen from the duplexer in FIG. 17 and line length l.
The line length is normalized in terms of the wavelength .lamda.
and the impedance Z is calculated assuming Z.sub.0=50 .OMEGA.,
Z.sub.L=47 .OMEGA.. It can be understood from FIG. 18 that the
antenna terminal impedance Z can be varied by varying the line
length l.
On the other hand, the amount of the leakage of the transmission
signal changes depending on the antenna impedance Z. Consequently,
the amount of leakage can be controlled by adjusting the line
length l to the power feed terminal by means for example of a delay
line DL inserted as a delay circuit 20. The degree of coupling of
the duplexer 13 (transmission (TX) terminal.fwdarw.reception (RX)
terminal) therefore theoretically becomes infinitely small (0) if
the antenna terminal impedance coincides with the characteristic
impedance Z.sub.0. However, in an actual circuit, the limit is
about -40 dB.
Thus, in this embodiment of the present invention, the amount of
the leakage of the transmission signal to the demodulation circuit
can be reduced by providing a delay circuit 20 and adjusting the
line length of the delay circuit 20 so as to make the impedance Z
of the antenna seen from the duplexer approach more closely to the
characteristic impedance.
It should be noted that the configuration in which a delay circuit
20 is provided between the transceiving antenna 16 and duplexer 13
in FIG. 16 can also be applied in the embodiments of FIG. 5, FIG. 8
and FIG. 9 described above.
FIG. 19 is an example extending the embodiment of FIG. 16. When a
common RFID transceiver device is employed with a plurality of IC
tags, connection is changed over between a plurality of antennas
16a to 16d using a switch 21.
The principles of the embodiment of FIG. 16 can be applied in this
embodiment also. Specifically, delay circuits 22a to 22d are
inserted between the switch 21 and the antennas 16a to 16d.
The leakage of the transmission signal to the demodulation circuit
14b can be minimized by making the corresponding antenna terminal
impedance approach the characteristic impedance by adjusting the
line lengths in the respective delay circuits 22a to 22d.
As described above with reference to the drawings, according to the
present invention, an RFID transceiver device is provided that is
capable of high sensitivity reception, by the reduction of noise,
irrespective of the distance to the tag. This makes it possible to
configure an RFID system of high reliability.
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