U.S. patent number 7,362,251 [Application Number 11/436,348] was granted by the patent office on 2008-04-22 for method and system for digital to analog conversion for power amplifier driver amplitude modulation.
This patent grant is currently assigned to Broadcom Corporation. Invention is credited to Henrik Jensen, Alireza Zolfaghari.
United States Patent |
7,362,251 |
Jensen , et al. |
April 22, 2008 |
Method and system for digital to analog conversion for power
amplifier driver amplitude modulation
Abstract
Aspects of a method and system for digital to analog conversion
for power amplifier driver amplitude modulation are presented.
Various aspects of the system may include circuitry that enables
oversampling, within a single integrated circuit device, of each of
a plurality of samples in a digital baseband signal. The circuitry
may enable reduction of a number of bits, i.e., coarse
quantization, in each of the oversampled plurality of samples so as
to cause displacement of the quantization noise that occurred as a
result of the coarse quantization. A subsequent signal may be
generated based on the oversampled signal. The circuitry may enable
the subsequent signal to be low-pass filtered utilizing filter
circuitry in the single integrated circuit device, thereby
attenuating the quantization noise displaced into the higher
frequency range of the oversampled signal.
Inventors: |
Jensen; Henrik (Long Beach,
CA), Zolfaghari; Alireza (Irvine, CA) |
Assignee: |
Broadcom Corporation (Irvine,
CA)
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Family
ID: |
38442182 |
Appl.
No.: |
11/436,348 |
Filed: |
May 18, 2006 |
Prior Publication Data
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Document
Identifier |
Publication Date |
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US 20070268168 A1 |
Nov 22, 2007 |
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Current U.S.
Class: |
341/143 |
Current CPC
Class: |
H03M
7/3008 (20130101); H04L 27/361 (20130101); H03M
1/0673 (20130101); H03M 3/502 (20130101); H03M
7/3026 (20130101); H03M 7/3042 (20130101) |
Current International
Class: |
H03M
3/00 (20060101) |
Field of
Search: |
;341/143,144,152,118 |
References Cited
[Referenced By]
U.S. Patent Documents
Foreign Patent Documents
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1235403 |
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Aug 2002 |
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EP |
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249874 |
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Feb 2006 |
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TW |
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Other References
Jau J K et al: Polar Modulation-based RF Power Amplifiers with
Enhanced Envelope Processing Technique; Microwave Conference, 2004;
34th European; Amsterdam, The Netherlands; Oct. 13, 2004,
Piscataway, NJ, IEEE; pp. 1317-1320; XP010788343. cited by other
.
Yves Geerts, Michel Steyaert; Design of Multi-bit Delta-Sigma A/D
Converters; Jan. 1, 2002; Kluwer, Netherlands, XP002449461. cited
by other .
Nosworthy et al; Delta-Sigma Data Converters, Theory, Design and
Simulation, Section 8.3.3; pp. 251-264; 1997; XP002082100. cited by
other .
Mucahit Kozak, Izzet Kale: Oversampled Delta-Sigma Modulators; Jan.
1, 2003; Kluwer, Netherlands; XP002449462. cited by other.
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Primary Examiner: Young; Brian
Attorney, Agent or Firm: McAndrews, Held & Malloy,
Ltd.
Claims
What is claimed is:
1. A method for amplitude modulation in a wireless communication
system, the method comprising: oversampling, within a single
integrated circuit device, each of a plurality of samples in a
baseband signal, wherein each of said plurality of samples in said
baseband signal comprises an M-bit binary data word, that is
converted to at least one of a plurality of K signal levels where K
and M represent numbers, wherein said value of K is less than
2.sup.M; reducing a number of bits in each of said oversampled
plurality of samples so as to cause displacement of quantization
noise that occurred as a result of said oversampling; and low-pass
filtering a subsequent signal comprising said reduced number of
bits using filter circuitry integrated within said single
integrated circuit device.
2. The method according to claim 1, comprising generating said
subsequent signal comprising a current one of said plurality of K
signal levels, followed by a zero signal level, followed by a
subsequent one of said plurality of K signal levels.
3. The method according to claim 2, comprising converting each of
said at least one of said plurality of K signal levels to a
corresponding N-bit binary data word, wherein N represents a
number.
4. The method according to claim 3, wherein one of: a lowest and a
highest, value for said N-bit binary data word corresponds to one
of: a highest and a lowest, value for one of said plurality of K
signal levels.
5. The method according to claim 3, wherein a value for said N-bit
binary data word that is greater than a lowest value and less than
a highest value for said N-bit binary data word corresponds to a
value for one of said plurality of K signal levels that is greater
than a lowest level and less than a highest level for said one of
said plurality of K signal levels.
6. The method according to claim 3, comprising generating switching
control bits based on said N-bit binary data word and bits
generated based on a pseudo random bit sequence.
7. The method according to claim 6, wherein said switching control
bits are differentially encoded.
8. The method according to claim 7, comprising selecting one of
said plurality of K signal levels based on values for positive
polarity bits in said differentially encoded switching control
bits.
9. The method according to claim 7, comprising generating a zero
signal level based on values for negative polarity bits in said
differentially encoded switching control bits.
10. The method according to claim 7, comprising generating said
subsequent signal comprising one of said plurality of K signal
levels based on current values for positive polarity bits in said
differentially encoded switching control bits, followed by a zero
signal level based on values for negative polarity bits in said
differentially encoded switching control bits, followed by a
subsequent one of said plurality of K signal levels based on
subsequent values for said positive polarity bits in said
differentially encoded switching control bits.
11. The method according to claim 10, wherein said current values
for said positive polarity bits are binary complements to
corresponding subsequent values for said positive polarity bits
based on said pseudo random bit sequence.
12. A system for amplitude modulation in a wireless communication
system, the system comprising: circuitry that enables oversampling,
within a single integrated circuit device, of each of a plurality
of samples in a baseband signal, wherein each of said plurality of
samples in said baseband signal comprises an M-bit binary data
word, that is converted to at least one of a plurality of K signal
levels where K and M represent numbers, wherein said value of K is
less than 2.sup.M; said circuitry enables reduction of a number of
bits in each of said oversampled plurality of samples so as to
cause displacement of quantization noise that occurred as a result
of said oversampling; and said circuitry enables low-pass filtering
of a subsequent signal comprising said reduced number of bits using
filter circuitry integrated within said single integrated circuit
device.
13. The system according to claim 12, wherein said circuitry
enables generation of said subsequent signal comprising a current
one of said plurality of K signal levels, followed by a zero signal
level, followed by a subsequent one of said plurality of K signal
levels.
14. The system according to claim 13, wherein said circuitry
enables conversion of each of said at least one of said plurality
of K signal levels to a corresponding N-bit binary data word,
wherein N represents a number.
15. The system according to claim 14, wherein one of: a lowest and
a highest, value for said N-bit binary data word corresponds to one
of: a highest and a lowest, value for one of said plurality of K
signal levels.
16. The system according to claim 14, wherein a value for said
N-bit binary data word that is greater than a lowest value and less
than a highest value for said N-bit binary data word corresponds to
a value for one of said plurality of K signal levels that is
greater than a lowest level and less than a highest level for said
one of said plurality of K signal levels.
17. The system according to claim 14, wherein said circuitry
enables generation of switching control bits based on said N-bit
binary data word and bits generated based on a pseudo random bit
sequence.
18. The system according to claim 17, wherein said switching
control bits are differentially encoded.
19. The system according to claim 18, wherein said circuitry
enables selection of one of said plurality of K signal levels based
on values for positive polarity bits in said differentially encoded
switching control bits.
20. The system according to claim 18, wherein said circuitry
enables generation of a zero signal level based on values for
negative polarity bits in said differentially encoded switching
control bits.
21. The system according to claim 18, wherein said circuitry
enables generation of said subsequent signal comprising one of said
plurality of K signal levels based on current values for positive
polarity bits in said differentially encoded switching control
bits, followed by a zero signal level based on values for negative
polarity bits in said differentially encoded switching control
bits, followed by a subsequent one of said plurality of K signal
levels based on subsequent values for said positive polarity bits
in said differentially encoded switching control bits.
22. The system according to claim 21, wherein said current values
for said positive polarity bits are binary complements to
corresponding subsequent values for said positive polarity bits
based on said pseudo random bit sequence.
Description
CROSS-REFERENCE TO RELATED APPLICATIONS/INCORPORATION BY
REFERENCE
NOT APPLICABLE.
FIELD OF THE INVENTION
Certain embodiments of the invention relate to wireless
communications. More specifically, certain embodiments of the
invention relate to a method and system for digital to analog
conversion for power amplifier driver amplitude modulation.
BACKGROUND OF THE INVENTION
Mobile communications have changed the way people communicate and
mobile phones have been transformed from a luxury item to an
essential part of every day life. The use of mobile phones today is
dictated by social situations, rather than hampered by location or
technology. While voice connections fulfill the basic need to
communicate, and wireless voice and data connections continue to
filter even further into the fabric of every day life, various
integrated mobile multimedia applications, utilizing wireless
and/or wired networks, may be the next step in the mobile
communication revolution.
Third generation (3G) cellular networks offering various high speed
access technologies and mobile telephones that have been
specifically designed to utilize these technologies, fulfill
demands for integrated multimedia applications supporting TV and
audio applications utilizing advanced compression standards,
high-resolution gaming applications, musical interfaces, peripheral
interface support, etc. The processing requirements are being
increased as chip designers take advantage of compression and
higher bandwidths to transmit more information. 3G wireless
applications support bit rates from 384 kilobits (Kbits)/second to
2 megabits (Mbits)/second, allowing chip designers to provide
wireless systems with multimedia capabilities, superior quality,
reduced interference, and a wider coverage area.
As mobile multimedia services grow in popularity and usage, factors
such as power consumption, cost efficient optimization of network
capacity and quality of service (QoS) continue to be even more
essential to cellular operators than it is today. These factors may
be achieved with careful network planning and operation,
improvements in transmission methods, and advances in receiver
techniques and chip integration solutions. To this end, carriers
need technologies that will allow them to increase downlink
throughput for the mobile multimedia applications support and, in
turn, offer advanced QoS capabilities and speeds for consumers of
mobile multimedia application services. Currently, mobile
multimedia processors may not fully utilize system-on-a-chip (SoC)
integration for advanced total system solution for today's mobile
handsets. For example, conventional mobile processors may utilize a
plurality of hardware accelerators to enable a variety of
multimedia applications, which significantly increases power
consumption, implementation complexity, mobile processor real
estate, and ultimately terminal size.
Some mobile communications technologies, for example the global
system for mobile communications (GSM), general packet radio
service (GPRS), and enhanced data rates for GSM evolution (EDGE)
may utilize polar modulation. Polar modulation may comprise
converting a signal from a representation that utilizes in-phase
(I), and quadrature phase (Q) components, to a corresponding
representation that utilizes magnitude (.rho.) and phase (.phi.)
components. Quantization noise may be introduced as a result of the
conversion from the I and Q signal representation to the .rho. and
.phi. signal representation. Consequently, at least a portion of
the components in the .rho. and .phi. signal representation may be
filtered.
While some conventional polar modulation transceiver designs may
comprise circuitry that enables conversion from the I and Q signal
representation to the .rho. and .phi. signal representation to be
performed within a single integrated circuit device, or chip, the
characteristics of the filtering circuitry may result in designs in
which the filtering circuitry being located within a separate chip
or off-chip as a discrete component filter.
Further limitations and disadvantages of conventional and
traditional approaches will become apparent to one of skill in the
art, through comparison of such systems with some aspects of the
present invention as set forth in the remainder of the present
application with reference to the drawings.
BRIEF SUMMARY OF THE INVENTION
A system and/or method is provided for digital to analog conversion
for power amplifier driver amplitude modulation, substantially as
shown in and/or described in connection with at least one of the
figures, as set forth more completely in the claims.
These and other advantages, aspects and novel features of the
present invention, as well as details of an illustrated embodiment
thereof, will be more fully understood from the following
description and drawings.
BRIEF DESCRIPTION OF SEVERAL VIEWS OF THE DRAWINGS
FIG. 1A is a block diagram illustrating an exemplary mobile
terminal, in accordance with an embodiment of the invention.
FIG. 1B is a block diagram illustrating an exemplary polar
modulation transmitter, which may be utilized in connection with an
embodiment of the invention.
FIG. 2 is a block diagram of an exemplary system for digital to
analog conversion for power amplifier driver amplitude modulation,
in accordance with an embodiment of the invention.
FIG. 3 is a block diagram of an exemplary system for delta sigma
modulation, in accordance with an embodiment of the invention.
FIG. 4 is a block diagram of an exemplary system for high pass
shaped dither generation, in accordance with an embodiment of the
invention.
FIG. 5 is a block diagram of an exemplary system for generating
control signals for digital to analog conversion, in accordance
with an embodiment of the invention.
FIG. 6 is a diagram of an exemplary system for dynamic element
matching, in accordance with an embodiment of the invention.
FIG. 7 is a diagram of an exemplary system for pseudo random bit
sequence generation, in accordance with an embodiment of the
invention.
FIG. 8 is a diagram of an exemplary circuit for digital to analog
conversion, in accordance with an embodiment of the invention.
FIG. 9 is a flowchart illustrating exemplary steps for digital to
analog conversion for power amplifier driver amplitude modulation,
in accordance with an embodiment of the invention.
DETAILED DESCRIPTION OF THE INVENTION
Certain embodiments of the invention may be found in a method and
system for digital to analog conversion for power amplifier driver
amplitude modulation. One aspect of the invention comprises
circuitry that enables amplitude control for a signal generated in
a polar amplitude transmitter. The system may comprise circuitry in
a single integrated circuit device, or chip, that enables a
conversion of a baseband digital signal to an oversampled
subsequent signal. The oversampling may enable the subsequent
signal to be filtered by filtering circuitry contained within the
single integrated circuit device. The filtered subsequent signal
may be utilized to control a power amplifier. The power amplifier
may receive a phase modulated signal and generate an amplified
signal. The power amplifier may vary an amplitude of the amplified
signal based on the filtered subsequent signal.
FIG. 1A is a block diagram illustrating an exemplary mobile
terminal, in accordance with an embodiment of the invention.
Referring to FIG. 1A, there is shown a mobile terminal 120 that may
comprise an RF receiver 123a, an RF transmitter 123b, a digital
baseband processor 129, a processor 125, and a memory 127. A
receive antenna 121a may be communicatively coupled to the RF
receiver 123a. A transmit antenna 121b may be communicatively
coupled to the RF transmitter 123b.
The RF receiver 123a may comprise suitable logic, circuitry, and/or
code that may enable processing of received RF signals. The RF
receiver 123a may enable receiving RF signals in a plurality of
frequency bands. The RF receiver 123a may enable receiving signals
in cellular frequency bands, for example, GSM, GPRS, and/or EDGE.
Each frequency band supported by the RF receiver 123a may have a
corresponding front-end circuit for handling low noise
amplification and down conversion operations, for example.
The RF receiver 123a may down convert the received RF signal to a
baseband frequency signal that comprises an in-phase (I) component
and a quadrature (Q) component. In some instances, the RF receiver
123a may enable analog-to-digital conversion of the baseband signal
components before transferring the components to the digital
baseband processor 129.
The digital baseband processor 129 may comprise suitable logic,
circuitry, and/or code that may enable processing and/or handling
of baseband frequency signals. In this regard, the digital baseband
processor 129 may process or handle signals received from the RF
receiver 123a and/or signals to be transferred to the RF
transmitter 123b. The digital baseband processor 129 may also
provide control and/or feedback information to the RF receiver 123a
and to the RF transmitter 123b based on information from the
processed signals. The digital baseband processor 129 may
communicate information and/or data from the processed signals to
the processor 125 and/or to the memory 127. Moreover, the digital
baseband processor 129 may receive information from the processor
125 and/or to the memory 127, which may be processed and
transferred to the RF transmitter 123b.
The RF transmitter 123b may comprise suitable logic, circuitry,
and/or code that may enable processing of RF signals for
transmission via a wireless medium. The RF transmitter 123b may
enable transmission of RF signals in a plurality of frequency
bands. Moreover, the RF transmitter 123b may enable transmission of
signals in cellular frequency bands, for example. Each frequency
band supported by the RF transmitter 123b may have a corresponding
front-end circuit for handling amplification and/or up conversion
operations, for example.
The RF transmitter 123b may up convert the baseband frequency
signal comprising l/Q components to an RF signal. In some
instances, the RF transmitter 123b may enable digital-to-analog
conversion of the baseband signal components received from the
digital baseband processor 129 before up conversion.
The processor 125 may comprise suitable logic, circuitry, and/or
code that may enable control and/or data processing operations for
the mobile terminal 120. The processor 125 may be utilized to
control at least a portion of the RF receiver 123a, the RF
transmitter 123b, the digital baseband processor 129, and/or the
memory 127. In this regard, the processor 125 may generate at least
one signal for controlling operations within the mobile terminal
120. The processor 125 may also enable execution of applications
that may be utilized by the mobile terminal 120.
The memory 127 may comprise suitable logic, circuitry, and/or code
that may enable storage of data and/or other information utilized
by the mobile terminal 120. For example, the memory 127 may be
utilized for storing processed data generated by the digital
baseband processor 129 and/or the processor 125. The memory 127 may
also be utilized to store information, such as configuration
information, that may be utilized to control the operation of at
least one block in the mobile terminal 120.
FIG. 1B is a block diagram illustrating an exemplary polar
modulation transmitter, which may be utilized in connection with an
embodiment of the invention. Referring to FIG. 1B, there is shown a
digital baseband processor 132, a reference frequency block 134, a
power amplifier 152, a bandpass filter 138, a transmit antenna
121b, a phase locked loop (PLL) 156, a preamplifier 144, and a
Cartesian to polar conversion block 158. The polar modulation
transmitter, as illustrated in FIG. 1B, may be part of a mobile
terminal, such as the mobile terminal 120 in FIG. 1A, for
example.
The reference frequency block 134 may comprise suitable logic,
circuitry, and/or code that may enable generation of local
oscillator (LO) and/or carrier frequency signals. The reference
frequency block 134 may comprise a crystal, which may be utilized
for generating the LO signals.
The digital baseband processor 132 may comprise suitable logic,
circuitry, and/or code that may enable processing and/or handling
of baseband frequency signals. The digital baseband processor may
generate a digital baseband signal comprising in-phase (I) and
quadrature phase (Q) components. The digital baseband signal may
comprise a plurality of samples and each sample may comprise a
plurality of bits, for example 12 bits. The samples within the
digital baseband signal may occur at a sampling rate, for example
13 MHz. Each sample in the baseband digital signal, which may
represent a plurality of signal levels, for example 2.sup.12, or
4,096, signal levels.
The bandpass filter 138 may comprise suitable logic, circuitry,
and/or code that may enable generation of an output signal by
processing and input signal to attenuate input signal amplitudes
for a range of electromagnetic frequencies below a low frequency,
f.sub.LOW, and above a high frequency, f.sub.HIGH. The range of
frequencies that are greater than or equal to f.sub.LOW and less
than or equal to f.sub.HIGH may comprise a pass band.
The preamplifier 144 may comprise suitable logic, circuitry, and/or
code that may enable generation of an output signal whose signal
level comprises a fixed or variable attenuation in comparison to a
signal level associated with a corresponding input signal.
The power amplifier 152 may comprise suitable logic, circuitry,
and/or code that may enable generation of an output signal, based
on an input signal, with sufficient electrical power, that
amplitude associated with the output signal may be maintained when
the output signal is applied to an electrical load. The power
amplifier 152 may be characterized by a linear operation when a
change in amplitude for the input signal corresponds to a
proportional change in amplitude for the output signal. The gain
associated with the power amplifier 152 may be variable based on a
received gain control input signal.
The Cartesian to polar conversion block 158 may comprise suitable
logic, circuitry, and/or code that may enable generation of
magnitude and phase components corresponding to a received input
signal. The Cartesian to polar conversion block 158 may receive a
digital baseband signal comprising I and Q components. The
Cartesian to polar conversion block 158 may generate a
representation of the digital baseband signal that comprises a
magnitude (.rho.) component, and a phase (.phi.) component. The
magnitude component may represent amplitude modulated signal
component, and the phase component may represent a phase modulated
signal component.
In various embodiments of the invention, circuitry utilized in
connection with the Cartesian to polar conversion block 158 may
generate a digital subsequent signal that represents an
oversampling of the digital baseband signal, while generating the
.rho. component. The oversampling may result in one sample within
the digital baseband signal corresponding to a plurality of samples
within the digital subsequent signal. The sampling rate for the
digital subsequent signal may be greater than the corresponding
sampling rate for the digital baseband signal. For example, while
the sampling rate for an exemplary digital baseband signal may be
13 MHz, the sampling rate for the digital subsequent signal may be
400 MHz. Each sample in the digital baseband signal, which may
represent a plurality of signal levels, for example 2.sup.12, may
be converted to a sample in a digital subsequent signal that
comprises about 2 bits, which may represent fewer signal levels,
for example 3. The exemplary 3 levels may comprise a level 0, level
1, and a level 2, for example. The signal levels may correspond to
voltage levels in an analog signal. Consequently, each sample in
the digital subsequent signal may be represented by a plurality of
bits that may be fewer in number than the corresponding plurality
of bits contained within a sample in the baseband digital signal.
For example, approximately 2 bits may represent each sample in the
subsequent signal. The reduction in the number of bits per sample
may enable digital to analog conversion within a single integrated
circuit device that processes approximately 2 bits per sample at a
rate of 400 million samples per second, for example.
The conversion from a digital baseband signal, which comprises
12-bit samples, to an analog subsequent signal, which comprises
corresponding values for 1 of 3 signal levels selected based on the
12 bit samples, may introduce quantization noise. Various
embodiments of the invention may comprise circuitry that modulates
the digital subsequent signal in a way that biases the quantization
noise to high frequency components within the analog subsequent
signal. During digital to analog conversion, in which a digital
subsequent signal comprising a plurality of 2 bit digital samples
is converted to an analog signal comprising a corresponding 1 of 3
signal levels for example, the analog signal may be low pass
filtered to attenuate the introduced quantization noise.
The oversampling of the digital baseband signal may enable
implementation of low pass filter circuitry that may be
characterized by a relatively high cutoff frequency, f.sub.HIGH,
for example 1 MHz. The relatively high cutoff frequency may enable
practical integration of the low pass filter circuitry within the
single integrated circuit device. In some conventional power
amplifier driver circuitry, the low pass filter circuitry may
require large capacitive value components resulting in the low pass
filter circuitry being implemented within a separate integrated
circuit device, or in discrete component form outside the
integrated circuit.
The fractional-N synthesizer 156 may comprise suitable logic,
circuitry, and/or code that may enable utilization of the phase
modulated signal component to generate a synthesized RF signal. The
fractional-N synthesizer 156 may generate the synthesized RF signal
based on an input IF signal. The fractional-N synthesizer 156 may
generate a change in signal level and/or frequency for the
synthesized RF signal based on a corresponding change in the input
IF signal. The fractional-N synthesizer 156 may perform calibration
and pre-distortion procedures to equalize the corresponding change
across a range of frequencies, substantially as described for the
fractional-N synthesizer 142.
In operation, the digital baseband processor 132 may provide a
baseband signal comprising I and Q signal components. The I and Q
components may be communicated to the Cartesian to polar conversion
block 158. The Cartesian to polar conversion block 158 may generate
magnitude (.rho.) and phase (.phi.) signal components, which
correspond to the received I and Q, signal components. The phase
signal component may be communicated to the PLL 156. The PLL 156
may utilize the phase signal component, and the LO signal and/or
carrier frequency signal from the reference frequency block 134, to
generate an RF synthesized signal. The frequency associated with
the RF synthesized signal may be based on the carrier frequency
derived from an input signal received from the reference frequency
block 134.
The preamplifier 144 may modify the amplitude associated with the
RF synthesized signal. The amplitude modified RF synthesized signal
may comprise an output RF synthesized signal. The power amplifier
136 may modify the amplitude associated with the output RF
synthesized signal. The power amplifier 136 may modify the
amplitude associated with the output RF synthesized signal based on
the magnitude component signal, received from the Cartesian to
polar conversion block 158. The output RF synthesized signal may
comprise signal components that span a range of frequencies. The
bandpass filter 138 may band limit the amplified output RF
synthesized signal by reducing signal levels for signal components
associated with frequencies that are not within the pass band for
the bandpass filter 138. The transmit antenna 121b may enable the
band limited signal to be transmitted via a wireless medium.
FIG. 2 is a block diagram of an exemplary system for digital to
analog conversion for power amplifier driver amplitude modulation,
in accordance with an embodiment of the invention. Referring to
FIG. 2, there is shown a single integrated circuit device 200. The
single integrated circuit device 200 may comprise a pseudo random
bit sequence generator 202, a delta sigma modulation quantizer 204,
a binary encoder 206, a dynamic element matching block 208, and a
digital to analog converter (DAC) block 210.
The pseudo random bit sequence generator 202 may comprise suitable
logic, circuitry, and/or code that may enable generation of a
plurality of bits. Taken as a group, the binary values for the
plurality of bits may represent a pseudo random sequence. The
pseudo random sequence may be generated based on a seed value.
The delta sigma modulation quantizer 204 may comprise suitable
logic, circuitry, and/or code that may enable generation of a K
level output signal wherein K may represent a number. In an
exemplary embodiment of the invention K=3. The delta sigma
modulation quantizer 204 may generate the K level output signal
based on a digital baseband signal comprising a plurality of
samples, and on a pseudo random bit sequence. Each sample in the
digital baseband signal may comprise a plurality of M bits. In an
exemplary digital baseband signal M=12. The samples within the
digital baseband signal may occur at a sampling rate, for example
13 MHz. Each sample in the K level output signal may comprise a
plurality of levels, for example 3 levels. The exemplary 3 levels
may comprise a level 0, a level 1, and a level 2. The 3 levels may
be represented by a corresponding plurality of bits, for example
1.5 bits, or approximately 2 bits. The samples within the K level
output signal may occur at a sampling rate, for example 400
MHz.
The delta sigma modulation quantizer 204 may compute a difference
value based on an addition value, based on a current sample from
the digital baseband signal, and a value based on a current sample
from the K level output signal. The computed difference value may
be added to a value based on a subsequent sample from the digital
baseband signal, and to a dithering value based on the pseudo
random bit sequence.
The delta sigma modulation quantizer 204 may generate a K level
output signal based on a plurality of computed difference values.
An exemplary K level output signal may comprise a current sample,
whose value may be chosen from one of the K levels based on a
current difference value, followed by a succeeding sample, whose
value is level 0, followed by a succeeding sample, whose value may
be chosen from one of the K levels based on a subsequent difference
value. In various embodiments of the invention, the exemplary K
level output signal may be generated based on return to zero
quantization.
During the conversion from M bit samples in a digital baseband
signal, to corresponding levels in a K level output signal,
quantization noise may be introduced. The delta sigma modulation
quantizer 204 may generate a high pass shaped dithering signal,
comprising a plurality of dithering values, based on the pseudo
random bit sequence. The high pass shaped dither signal, when added
during the addition function, may cause the quantization noise
present in the K level output signal to be highly uncorrelated with
the input signal and thus well modeled as an additive "white" noise
source. The delta sigma modulation quantizer 204 may cause the
quantization noise to occur predominantly at the high end of the
frequency spectrum associated with the output signal. The
displacement of the quantization noise to the high end of the
frequency spectrum may facilitate filtering of the quantization
noise by utilizing a low pass filter. In various embodiments of the
invention, the low pass filter circuitry and the delta sigma
modulation quantizer circuitry 204 may be integrated within a
single integrated circuit (IC) device.
The binary encoder 206 may comprise suitable logic, circuitry,
and/or code that may generate a digital subsequent signal based on
conversion of individual samples from a K level input signal to
corresponding N bit binary samples, wherein N may represent a
number. Each binary sample may be represented as a binary tuple
comprising {bit N, bit N-1, . . . , bit 1}, for example. The value
of N may be such that 2.sup.N.gtoreq.K. In an exemplary embodiment
of the invention M=3, and N=2, wherein the correspondence between
levels and bit values may be as follows: level 0 corresponds to the
bit tuple {0, 0}, level 1 corresponds to the bit tuple {1, 0}, and
level 2 corresponds to the bit tuple {1, 1}.
The dynamic element matching block 208 may comprise suitable logic,
circuitry, and/or code that may enable generation of switching
control bits by conversion of bit values in a received digital
signal, comprising N bit binary samples, based on a pseudo random
bit sequence. The dynamic element matching block 208 may generate L
switching control bits based on each N bit binary sample, for
example. In an exemplary embodiment of the invention, L=2, and N=2.
The dynamic element matching block 208 may receive a sample in a
received digital signal that comprises a bit tuple {1, 0} and
generate switching control bits represented by a bit tuple {1, 0}
based on a current value in the pseudo random bit sequence. The
dynamic element matching block 208 may receive a sample in a
received digital signal that comprises a bit tuple {1, 0} and
generate switching control bits represented by a bit tuple {0, 1}
based on a subsequent value in the pseudo random bit sequence.
The DAC block 210 may comprise suitable logic, circuitry, and/or
code that may enable generation of an analog output signal based on
a received digital signal. The digital signal may comprise a
plurality of N bit binary samples. The DAC block 210 may convert
the binary samples to a corresponding one of K signal levels that
may be utilized to generate an analog output signal. In an
exemplary embodiment of the invention K=3, and N=2, wherein the
correspondence between bit values and levels may be as follows: the
bit tuple {0, 0} corresponds to signal level 0, the bit tuples {0,
1} and {1, 0} correspond to signal level 1, and the bit tuple {1,
1} corresponds to signal level 2.
In various embodiments of the invention, the signal level in the
analog output signal may correspond to a voltage level that may be
generated by activating transistor circuitry. At least a portion of
the transistor circuitry may be activated and/or deactivated based
on individual bit values in a current N bit binary sample. The
activated transistor circuitry may initiate current flow through
output resistor and capacitor circuitry. The signal level in the
analog output signal may be based on a voltage level associated
with the output resistor and capacitor circuitry.
The performance characteristics of a transistor may vary based on
process variations that may occur during integrated circuit
manufacture. This may be referred to as a mismatch. Consequently,
current flow through one transistor in the chip may differ from
current flow through another nominally identical transistor in the
chip, for a given input voltage level applied to each transistor.
As a result, a voltage level corresponding to a signal level in the
analog output signal may vary depending upon which transistors were
activated and/or deactivated within the DAC circuitry. The
variation may be referred to as mismatch noise, which may be
introduced into the analog output signal based on the mismatch of
transistors within the chip.
In various embodiments of the invention, a level of mismatch noise
in the analog output signal may be attenuated based on a method
referred to as dynamic element matching. In dynamic element
matching, a voltage level corresponding to a signal level in the
analog output signal may be generated by alternating which
transistors are activated or deactivated at different time instants
when the signal level is to be generated. In an exemplary
embodiment of the invention, at a given time instant a signal level
1 may be generated based on a received bit tuple {0, 1}. This bit
tuple may result in an activation of a first transistor, and a
deactivation of a second transistor. A corresponding voltage level
may be generated for the output signal corresponding to the signal
level 1. At a subsequent time instant, a signal level 1 may be
generated based on a received bit tuple {1, 0}. This subsequent bit
tuple may result in deactivation of the first transistor, and
activation of the second transistor. A corresponding subsequent
voltage level may be generated for the output signal corresponding
to the signal level 1 at the subsequent time instant. When measured
over a time duration spanning a plurality of binary samples,
dynamic element matching may enable a reduction of in-band mismatch
noise when compared to some conventional circuit designs for
digital to analog conversion.
In operation, the pseudo random bit sequence generator 202 may
generate a pseudo random sequence. The pseudo random sequence may
be utilized by the delta sigma modulation quantizer 204, and the
dynamic element matching block 208. The delta sigma modulation
quantizer 204 may receive a digital baseband signal and generate a
K level output signal. The binary encoder 206 may convert the K
level output signal to a digital subsequent signal comprising N bit
binary samples. An N bit tuple value may correspond to one level
among the K levels in the output signal.
The dynamic element matching block 208 may receive the digital
subsequent signal from the binary encoder 206. The dynamic element
matching block 208 may generate switching control bits by modifying
bits in the digital subsequent signal based on the received pseudo
random sequence. The modified bits may comprise binary complements
of the corresponding bits received in the digital subsequent
signal. The dynamic element matching block 208 may modify bits in
the received digital subsequent signal when the N bit sample
comprises a data word whose value may be greater than the lowest
value and less than the highest value for the N bit data word. For
example, in an exemplary embodiment of the invention wherein N=2, a
lowest value may be equal to 0, corresponding to a bit tuple {0,
0}, and a highest value may be equal to 3, corresponding to a bit
tuple {1, 1}. The dynamic element matching block may modify bits in
a received digital subsequent word when the N bit data word is
equal to 1, corresponding to a bit tuple {0, 1}, or 2,
corresponding to a bit tuple {1, 0}. Based on the pseudo random bit
sequence, the dynamic element matching block 208 may modify a bit
tuple {0, 1} to generate a bit tuple {1, 0}, or vice versa.
The DAC block 210 may receive switching control bits from the
dynamic element matching block 208. The DAC block may utilize the
switching control block to perform digital to analog conversion for
a signal level represented by the switching control bits. The DAC
block 210 may utilize the switching control bits to activate and/or
deactivate transistor circuitry that may result in a corresponding
voltage level. The voltage level may be utilized to generate an
analog output signal. The analog output signal from the DAC block
may be utilized generate a magnitude component as illustrated for
the Cartesian to polar conversion block 158 (FIG. 1B).
FIG. 3 is a block diagram of an exemplary system for delta sigma
modulation, in accordance with an embodiment of the invention. The
function for the system for delta sigma modulation as illustrated
in FIG. 3 may be substantially as described for the delta sigma
modulation quantizer 204 (FIG. 2). Referring to FIG. 3, there is
shown a dithering block 302, a binary generation block 304, a
complementary binary generation block 306, a multiplexer 308, an
adder block 310, a time delay block 312, a difference block 314,
and a return to zero quantizer 316.
The dithering block 302 may comprise suitable logic, circuitry,
and/or code that may enable generation of dithering selection bits
from a received pseudo random bit sequence. The dithering selection
bits may be utilized to enable generation of a high pass shaped
dither signal.
The binary generation block 304 may comprise suitable logic,
circuitry, and/or code that may enable generation of bits based on
a seed number. In an exemplary embodiment of the invention, a seed
number value of 2.sup.-4 may be utilized.
The complementary binary generation block 306 may comprise suitable
logic, circuitry, and/or code that may enable generation of bits
based on a seed number. The bits may comprise a two's complement,
or one's complement representation, for example. In an exemplary
embodiment of the invention, a seed number value of -2.sup.-4 may
be utilized.
The multiplexer 308 may comprise suitable logic, circuitry, and/or
code that may enable generation of at least one output signal based
on a plurality of input signals. Input signals may be selected for
output based on a selector input comprising one or more bits. In an
exemplary embodiment of the invention, the multiplexer 308 may
comprise a 1 bit select input, two input signals, and one output
signal. One of the two input signals may be associated with a
binary value 0, while the other input signal may be associated with
a binary value 1. The input signal associated with the binary value
0 may be selected for output when the select input receives an
input with a binary value 0. The input signal associated with the
binary value 1 may be selected for output when the select input
receives an input with a binary value 1. The output signal from the
multiplexer 308 may comprise a high pass shaped dither signal.
The adder block 310 may comprise suitable logic, circuitry, and/or
code that may enable generation of an addition value. The addition
value may be computed based on a received digital baseband signal,
a high pass shaped dither signal, and a difference signal.
The time delay block 312 may comprise suitable logic, circuitry,
and/or code that may enable generation of an output signal based on
an input signal, wherein a value for the output signal at a current
time instant corresponds to a value for the input signal at a
previous time instant. The difference block 314 may comprise
suitable logic, circuitry, and/or code that may enable generation
of a difference value based on a current addition value and on a
value for a current sample from an output signal. The return to
zero quantizer 316 may comprise suitable logic, circuitry, and/or
code that may enable generation of a K level output signal based on
values of computed addition values.
In operation, the dithering block 302 may receive bits from a
pseudo random bit sequence. The pseudo random bit sequence may be
utilized to generate dithering selection bits. The dithering
selection bits may be output and utilized as input to the
multiplexer 308. Based on a value for a current dithering selection
bit, the multiplexer 308 may select an input from the binary
generation block 304 or the complementary binary generation block
306. Based on the selection, the multiplexer 308 may generate a
high pass shaped dither signal.
The adder block 310 may generate a current addition value based on
a current sample from a digital baseband signal, a current value
from a high pass shaped dither signal, and a time delayed
difference value received from the time delay block 312. The
difference block 314 may compute a current difference value based
on a current addition value and a current sample value from an
output signal generated by the return to zero quantizer 316. The
return to zero quantizer 316 may generate a value for a current
sample in the output signal based on a current addition value.
FIG. 4 is a block diagram of an exemplary system for high pass
shaped dither generation, in accordance with an embodiment of the
invention. The system for high pass shaped dither, as illustrated
in FIG. 4, may enable generation of dithering selection bits, which
may be utilized for generating a high pass shaped dither signal as
described for the delta sigma modulation quantizer 204 (FIG. 2),
and the dithering block 302 (FIG. 3). Referring to FIG. 4, there is
shown a logical negation block 402, a plurality of logical AND
blocks 404 and 406, a plurality of JK flip flop blocks 408 and 414,
and a plurality of logical OR blocks 410 and 412.
The logical negation block 402 may comprise suitable logic,
circuitry, and/or code that may enable generation of an output
signal that comprises binary values that are binary complements of
corresponding binary values in an input signal.
The logical AND block 404 may comprise suitable logic, circuitry,
and/or code that may enable generation of a binary output value
that is based on a logical AND of current binary values for each of
a plurality of input signals. In an exemplary embodiment of the
invention, the logical AND block 404 may comprise 2 inputs and 1
output. The logical AND block 406 may be substantially as described
for the logical AND block 404.
The JK flip-flop block 408 may comprise suitable logic, circuitry,
and/or code that may enable generation of complementary binary
outputs, Q and .about.Q based on a binary J input signal and a
binary K input signal. A current binary value for the J input
signal of logical HIGH may enable the corresponding binary value
for the Q output signal to be set to a binary value of logical
HIGH. The corresponding binary value for the .about.Q output may be
set to a binary value of logical LOW. A current binary value for
the K input signal of logical HIGH may enable the corresponding
binary value for the Q output to be set to a binary value of
logical LOW. The corresponding value for the .about.Q output may be
set to a binary value of logical HIGH. The JK flip-flop 414 may be
substantially as described for the JK flip flop 408.
The logical OR block 410 may comprise suitable logic, circuitry,
and/or code that may enable generation of a binary output value
that is based on a logical OR of current binary values for each of
a plurality of input signals. In an exemplary embodiment of the
invention, the logical OR block 410 may comprise 2 inputs and 1
output. The logical OR block 412 may be substantially as described
for the logical OR block 410.
In operation, the logical negation block 402 may receive bits from
a pseudo random sequence and generate an output sequence that
comprises bits that are binary complements to corresponding bits in
the pseudo random sequence. The logical AND block 404 may receive
bits from the pseudo random sequence and from the Q output from the
JK flip-flop 414. The logical AND block 406 may receive bits from
the logical negation block 402 and from the .about.Q output from
the JK flip-flop 414. The output from the logical AND block 404 may
be coupled to the J input of the JK flip-flop 408. The output from
the logical AND block 406 may be coupled to the K input of the JK
flip-flop 408. The Q output from the JK flip-flop 408 may be
coupled to an input of the logical OR block 412. The .about.Q
output from the JK flip-flop 408 may be coupled to an input of the
logical OR block 410. The logical OR block 410 may also receive
bits from the pseudo random sequence. The logical OR block 412 may
also receive bits from the logical negation block 402. The output
from the logical OR block 410 may be coupled to the J input of the
JK flip-flop 414. The output from the logical OR block 412 may be
coupled to the K input of the JK flip-flop 414. The Q output from
the JK flip-flop 414 may comprise generated dithering selection
bits.
FIG. 5 is a block diagram of an exemplary system for generating
control signals for digital to analog conversion, in accordance
with an embodiment of the invention. The system for generating
control signals for digital to analog conversion, as illustrated in
FIG. 5, may be substantially as described for the dynamic element
matching block 208 (FIG. 2). Referring to FIG. 5, there is shown a
logical exclusive or (XOR) block 502, a dynamic element matching
logic block 504, a plurality of logical negative AND blocks 506,
508, 510, 512, and 514, and a plurality of time delay blocks 516
and 518. The plurality of time delay blocks 516 and 518 may be
substantially as described for the time delay block 312 (FIG.
3).
The dynamic element matching logic block 504 may comprise suitable
logic, circuitry, and/or code that may enable generation of
switching control bits. The switching control bits may comprise
complementary binary outputs O and .about.O from the dynamic
element matching block 504. A current binary value for the .about.O
output may comprise a complementary value in comparison to a
current binary value for the O output. The complementary binary
outputs O and .about.O may be generated based on a corresponding
binary E input and a binary R input.
The logical XOR block 502 may comprise suitable logic, circuitry,
and/or code that may enable generation of a binary output value
based on a logical XOR of current binary values for each of a
plurality of input signals. In an exemplary embodiment of the
invention, the logical XOR block 502 may comprise 2 inputs and 1
output.
The logical NAND block 506 may comprise suitable logic, circuitry,
and/or code that may enable generation of a binary output value
that is based on a logical negated AND, or NAND, of current binary
values for each of a plurality of input signals. In an exemplary
embodiment of the invention, the logical NAND block 506 may
comprise 2 inputs and 1 output. The logical NAND blocks 508, 510,
512, and 514 may be substantially as described for the logical NAND
block 506.
In operation the logical XOR block 502 may receive inputs from
signals Bit2_In and Bit1_In. The input signals may correspond to
samples in a received digital signal from the binary encoder 206
(FIG. 2), wherein each sample may be represented by a bit tuple
{bit 2, bit 1}. The bit value Bit2_In may correspond to bit 2
within the tuple, and the bit value Bit1_In may correspond to the
bit 1 within the tuple. The output from the logical XOR block 502
may be coupled to an input of the dynamic element matching logic
block 504. The dynamic element matching logic block 504 may also
receive an input from the pseudo random bit sequence. The logical
NAND block 506 may receive input from signals Bit2_in and Bit1_In.
The logical NAND block 508 may receive input from the output of the
logical XOR block 502, and from the O output from the dynamic
element matching logic block 504. The logical NAND block 510 may
receive input from the output of the logical XNOR block 502 and
from the .about.O output from the dynamic element matching logic
block 504. The logical NAND block 512 may receive input from the
output of the logical NAND block 506 and from the output of the
logical NAND block 508. The logical NAND block 514 may receive
input from the output of the logical NAND block 506 and from the
output of the logical NAND block 510. The time delay block 516 may
generate bit 2 of a switching control bit tuple based on a time
delayed version of the output from the logical NAND gate 512. The
time delay block 518 may generate bit 1 of a switching control bit
tuple based on a time delayed version of the output from the
logical NAND gate 514.
FIG. 6 is a diagram of an exemplary system for dynamic element
matching, in accordance with an embodiment of the invention. The
system for dynamic element matching, as illustrated in FIG. 6, may
be substantially as described for the dynamic element matching
block 208 (FIG. 2), and presents a detailed view of the circuitry
associated with the dynamic element matching logic block 504 (FIG.
5). Referring to FIG. 6, there is shown a logical negation block
602, a plurality of logical AND blocks 604, 606, 614, and 616, a
plurality of JK flip flop blocks 608 and 618, and a plurality of
logical OR blocks 610 and 612. The logical negation block 602 may
be substantially as described for the logical negation block 402
(FIG. 4). The logical OR blocks 610 and 612 may be substantially as
described for the logical OR block 410. The logical AND blocks 604,
606, 614 and 616 may be substantially as described for the logical
AND block 404. In an exemplary embodiment of the invention, the
logical AND blocks 604 and 606 may each comprise 3 inputs and 1
output. The JK flip-flop blocks 608 and 618 may be substantially as
described for the JK flip flop 408.
In operation, the logical negation block 602 may receive bits from
a pseudo random sequence and generate an output sequence that
comprises bits that are binary complements to corresponding bits in
the pseudo random sequence. The logical AND block 604 may receive
bits from the pseudo random sequence, from the Q output from the JK
flip flop 618, and an input generated based on a logical XOR of
bits in the bit tuple {bit 2, bit 1} from the binary encoder 206
(FIG. 2). The logical AND block 606 may receive bits from the
logical negation block 402, from the .about.Q output from the JK
flip flop 618, and the input generated based on a logical XOR of
bits in the bit tuple {bit 2, bit 1} from the binary encoder
206.
The output from the logical AND block 604 may be coupled to the J
input of the JK flip-flop 608. The output from the logical AND
block 606 may be coupled to the K input of the JK flip-flop 608.
The Q output from the JK flip-flop 608 may be coupled to an input
of the logical OR block 612. The .about.Q output from the JK
flip-flop 608 may be coupled to an input of the logical OR block
610. The logical OR block 610 may also receive bits from the pseudo
random sequence. The logical OR block 612 may also receive bits
from the logical negation block 602. The output from the logical OR
block 610 may be coupled to an input of the logical AND block 614.
The logical AND block 614, and logical AND block 616, may each also
receive the input generated based on a logical XOR of bits in the
bit tuple {bit 2, bit 1} from the binary encoder 206.
The output from the logical AND block 614 may be coupled to the J
input of the JK flip-flop 618. The output from the logical AND
block 616 may be coupled to the K input of the JK flip-flop 618.
The Q output from the JK flip-flop 618 may correspond to the O
output from the dynamic element matching logic block 504. The
.about.Q output from the JK flip-flop 618 may correspond to the
.about.O output from the dynamic element matching logic block
504.
FIG. 7 is a diagram of an exemplary system for pseudo random bit
sequence generation, in accordance with an embodiment of the
invention. The system for pseudo random bit sequence generation, as
illustrated in FIG. 7, may be substantially as described for the
pseudo random bit sequence generator 202 (FIG. 2). Referring to
FIG. 7, there is shown a logical XOR block 702, and a plurality of
time delay blocks 704, 706, 708, 710, 712, 714, 716, 718, 720, 722,
724, 726, 728, 730, 732, 734, 736, 738, 740, 742, 744, 746, 748,
750, 752, 754, 756, 758, 760, 762, and 764. The logical XOR block
702 may be substantially as described for the logical XOR block 502
(FIG. 5). The plurality of timing delay blocks 704, 706, . . . ,
764 may each be substantially as described for the timing delay
block 312 (FIG. 3).
At the start of operation, each of the timing delay blocks 704,
706, . . . , 764 may comprise a corresponding initial bit value.
Collectively among the plurality of timing delay blocks, the
corresponding plurality of initial bit values may represent a seed
number. The logical XOR block 702 may receive an input from the
output of the timing delay block 708, and an input from the output
of the timing delay block 764. The output from the logical XOR
block 702 may be coupled to an input of the timing delay block
704.
The output from the timing delay block 704 may be coupled to an
input of the timing delay block 706. The output from the timing
delay block 706 may be coupled to an input of the timing delay
block 708. The output from the timing delay block 708 may be
coupled to an input of the timing delay block 710. The output from
the timing delay block 710 may be coupled to an input of the timing
delay block 712. The output from the timing delay block 712 may be
coupled to an input of the timing delay block 714. The output from
the timing delay block 714 may be coupled to an input of the timing
delay block 716. The output from the timing delay block 716 may be
coupled to an input of the timing delay block 718. The output from
the timing delay block 718 may be coupled to an input of the timing
delay block 720.
The output from the timing delay block 720 may be coupled to an
input of the timing delay block 722. The output from the timing
delay block 722 may be coupled to an input of the timing delay
block 724. The output from the timing delay block 724 may be
coupled to an input of the timing delay block 726. The output from
the timing delay block 726 may be coupled to an input of the timing
delay block 728. The output from the timing delay block 728 may be
coupled to an input of the timing delay block 730. The output from
the timing delay block 730 may be coupled to an input of the timing
delay block 732. The output from the timing delay block 732 may be
coupled to an input of the timing delay block 734. The output from
the timing delay block 734 may be coupled to an input of the timing
delay block 736. The output from the timing delay block 736 may be
coupled to an input of the timing delay block 738. The output from
the timing delay block 738 may be coupled to an input of the timing
delay block 740.
The output from the timing delay block 740 may be coupled to an
input of the timing delay block 742. The output from the timing
delay block 742 may be coupled to an input of the timing delay
block 744. The output from the timing delay block 744 may be
coupled to an input of the timing delay block 746. The output from
the timing delay block 746 may be coupled to an input of the timing
delay block 748. The output from the timing delay block 748 may be
coupled to an input of the timing delay block 750. The output from
the timing delay block 750 may be coupled to an input of the timing
delay block 752. The output from the timing delay block 752 may be
coupled to an input of the timing delay block 754. The output from
the timing delay block 754 may be coupled to an input of the timing
delay block 756. The output from the timing delay block 756 may be
coupled to an input of the timing delay block 758. The output from
the timing delay block 758 may be coupled to an input of the timing
delay block 760. The output from the timing delay block 760 may be
coupled to an input of the timing delay block 762. The output from
the timing delay block 762 may be coupled to an input of the timing
delay block 764. The output from the timing delay block 764 may
correspond to the output from the pseudo random bit sequence
generator 202.
FIG. 8 is a diagram of an exemplary circuit for digital to analog
conversion, in accordance with an embodiment of the invention. The
system for digital to analog conversion, as illustrated in FIG. 8,
may be substantially as described for the DAC block 210 (FIG. 2).
Referring to FIG. 8, there is shown a plurality of resistors 802,
and 804, a plurality of capacitors 806, and 808, a plurality of
transistors 810, 812, 814, and 816, and a plurality of current
sources 818 and 820. The transistors 810, 812, 814, and 816 may
utilized any among a plurality of technologies, for example various
metal oxide silicon (MOS) technologies such as complementary MOS
(CMOS), n-channel MOS (NMOS), p-channel MOS (PMOS), or junction
field effect transistor (JFET), or bipolar technologies, for
example.
The resistors 802 and 804, and capacitors 806 and 808 may form low
pass filter circuitry, which is integrated within a single
integrated circuit device. The low pass filter circuitry may be
utilized to filter an analog signal generated by the DAC block 210.
The filtered analog signal, as measured at the signal point labeled
Out in FIG. 8, may correspond to an output from the DAC block 210.
The filtered analog signal may comprise a magnitude component
output from the Cartesian to polar conversion block 158. The
magnitude component may be utilized to control a gain of the power
amplifier 152. The cutoff frequency f.sub.HIGH for the low pass
filter may be determined based on the capacitive values for the
capacitors 806 and 808, and on values of the resistors 802 and
804.
The inputs I1p and I1n may correspond to a differentially encoded
signal corresponding to the Bit1_In input to the DAC block 210. The
input I1p may represent a positive polarity bit in the
differentially encoded signal corresponding to the Bit1_In input.
The input I1n may represent a negative polarity bit in the
differentially encoded signal corresponding to the Bit1_In input.
The inputs I2p and I2n may correspond to differentially encoded
values corresponding to the Bit2_In input to the DAC block 210. The
input I2p may represent a positive polarity bit in the
differentially encoded signal corresponding to the Bit2_In input.
The input I2n may represent a negative polarity bit in the
differentially encoded signal corresponding to the Bit2_In input.
The Bit1_In input to the DAC block 210 may correspond to the
Bit1_Out output from the dynamic element matching block 208. The
Bit2_In input to the DAC block 210 may correspond to the Bit2_Out
output from the dynamic element matching block 208.
The Bit2_In input may correspond to bit 2 in a bit tuple {bit 2,
bit 1} generated by the dynamic element matching block 208. The
Bit1_In input may correspond to bit 1 in the bit tuple. The
positive polarity bit I2p may correspond to the bit 2 equal to a
logic HIGH value. The positive polarity bit I2p may equal a HIGH
signal level when bit 2 equals a binary HIGH value, for example.
The negative polarity bit I2n may correspond to the bit 2 equal to
a logic LOW value. The negative polarity bit I2n may equal a HIGH
signal level when bit 2 equals a logic LOW value, for example. The
positive polarity bit I1p may correspond to the bit 1 equal to a
logic HIGH value. The positive polarity bit I1p may equal a HIGH
signal level when bit 1 equals a logic HIGH value, for example. The
negative polarity bit I1n may correspond to the bit 1 equal to a
logic LOW value. The negative polarity bit I1n may equal a HIGH
signal level when bit 1 equals a binary LOW value, for example.
Based on signal levels for the input I1n, the transistor 810 may be
enabled to enter an activated conducting state. In the activated
conducting state, the transistor 810 may enable a current
conduction path between the supply voltage source, labeled Vdd, and
ground, labeled Gnd, via the current source 818. When the input I1n
enables the transistor 810 to enter an active conducting state, the
signal level for the input I1p may enable the transistor 812 to
enter a deactivated nonconducting state. In the deactivated
nonconducting state the transistor 812 may disable a current
conduction path between the supply voltage source and ground, via
the current source 818.
Similarly, based on signal levels for the input I2n, the transistor
816 may be enabled to enter an activated conducting state. In the
activated conducting state, the transistor 816 may enable a current
conduction path between the supply voltage source, and ground, via
the current source 820. When the input I2n enables the transistor
816 to enter an active conducting state, the signal level for the
input I2p may enable the transistor 814 to enter a deactivated
nonconducting state. In the deactivated nonconducting state the
transistor 814 may disable a current conduction path between the
supply voltage source and ground, via the current source 820.
When inputs I1n and I2n enable the corresponding transistors 810
and 816 to enter activated conducting states, a current flow may be
approximately equal to 0 through the resistors 802 and 804, for
example. The corresponding voltage measured at the Out signal label
may correspond to a level 0 signal level, in an exemplary
embodiment of the invention. When inputs I1p and I2p enable the
corresponding transistors 812 and 814 to enter activated conducting
states, an increased level of current flow may be enabled through
the resistors 802 and 804 in comparison to the condition in which
the transistors 812 and 814 are in deactivated nonconducting
states. The corresponding voltage measured at the Out signal label
may correspond to the level 2 signal level, in an exemplary
embodiment of the invention. The filter circuitry may limit a rate
of change in the voltage level as measured at the Out signal label
based on the cutoff frequency f.sub.HIGH.
When input I1p enables the transistor 812 to enter an activated
conducting state, while the input I2p enables the transistor 814 to
enter an deactivated nonconducting state, the corresponding voltage
as measured at the Out signal label may correspond to a level 1
signal level, in an exemplary embodiment of the invention. This
voltage may be referred to as a {0, 1} voltage level. When input
I2p enables the transistor 814 to enter an activated conducting
state, while the input I1p enables the transistor 812 to enter an
deactivated nonconducting state, the corresponding voltage as
measured at the Out signal label may correspond to a level 1 signal
level, in an exemplary embodiment of the invention. This voltage
may be referred to as a {1, 0} voltage level. The {0, 1} voltage
level may not be equal to the {1, 0} voltage level. The voltage
difference may comprise mismatch noise.
FIG. 9 is a flowchart illustrating exemplary steps for digital to
analog conversion for power amplifier driver amplitude modulation,
in accordance with an embodiment of the invention. Referring to
FIG. 9, in step 902 a pseudo random bit sequence may be generated
by the pseudo random bit sequence generator 202. In step 904, the
delta sigma modulation quantizer 204 may receive a digital baseband
signal. In step 906, the delta sigma modulation quantizer 204 may
oversample the received digital baseband signal and generate an
output signal. In step 908, the delta sigma modulation quantizer
204 may convert a sample from the digital baseband signal to a
signal level. In step 910, the binary encoder 206 may convert the
signal level to a binary representation. In step 912, the dynamic
element matching block 208 may generate switching control bits for
the DAC 210 based on the signal level. In step 914, the DAC 210 may
generate an analog signal based on the switching control bits.
Aspects of a system for digital to analog conversion for power
amplifier driver amplitude modulation may comprise a Cartesian to
polar converter 158 that enables oversampling, within a single
integrated circuit device 200, of each of a plurality of samples in
a digital baseband signal. A delta sigma modulation quantizer 204
may enable reduction of a number of bits in each of the oversampled
plurality of samples so as to cause displacement of quantization
noise that occurred as a result of the oversampling. A subsequent
signal may be generated based on the oversampling. A digital to
analog converter (DAC) 210 may enable the subsequent signal to be
low-pass filtered utilizing filter circuitry in the single
integrated circuit device 200. Each of the plurality of samples
within the digital baseband signal may comprise an M-bit binary
data word, that is converted to at least one of a plurality of K
signal levels, where K and M represent numbers, wherein the value
of K may be less than 2.sup.M.
The DAC 210 may enable generation of the subsequent signal
comprising a current one of the plurality of K signal levels,
followed by a zero signal level, followed by a subsequent one of
the plurality of K signal levels. A binary encoder 206 may enable
conversion of each of the plurality of K signal levels to a
corresponding N bit binary word, wherein N may represent a number.
A lowest value or highest value for the N bit binary data word may
correspond to a highest or lowest value for one of the plurality of
K signal levels. The value for the N bit binary data word that may
be greater than a lowest value, and less than a highest value, for
the N bit binary data word may correspond to one of the plurality
of K signal levels that may be greater than a lowest level, and
less than a highest level for the one of the plurality of K signal
levels. A dynamic element matching block 208 may enable generation
of switching control bits based on the N bit binary data word and
on bits generated based on a pseudo random bit sequence. The
switching control bits may be differentially encoded.
The DAC 210 may enable selection of one of the plurality of K
signal levels based on positive polarity bits in the differentially
encoded switching control bits. The DAC 210 may enable generation
of a zero signal level based on values for negative polarity bits
in the differentially encoded switching control bits. The DAC 210
may enable generation of the subsequent signal comprising one of
the plurality of K signal levels based on current values for
positive polarity bits, followed by a zero signal level based on
values for negative polarity bits, followed by a subsequent one of
the plurality of K signal levels based on subsequent values for the
positive polarity bits. The current values for the positive
polarity bits may be binary complements to corresponding subsequent
values for the positive polarity bits based on the pseudo random
bit sequence.
Accordingly, the present invention may be realized in hardware,
software, or a combination of hardware and software. The present
invention may be realized in a centralized fashion in at least one
computer system, or in a distributed fashion where different
elements are spread across several interconnected computer systems.
Any kind of computer system or other apparatus adapted for carrying
out the methods described herein is suited. A typical combination
of hardware and software may be a general-purpose computer system
with a computer program that, when being loaded and executed,
controls the computer system such that it carries out the methods
described herein.
The present invention may also be embedded in a computer program
product, which comprises all the features enabling the
implementation of the methods described herein, and which when
loaded in a computer system is able to carry out these methods.
Computer program in the present context means any expression, in
any language, code or notation, of a set of instructions intended
to cause a system having an information processing capability to
perform a particular function either directly or after either or
both of the following: a) conversion to another language, code or
notation; b) reproduction in a different material form.
While the present invention has been described with reference to
certain embodiments, it will be understood by those skilled in the
art that various changes may be made and equivalents may be
substituted without departing from the scope of the present
invention. In addition, many modifications may be made to adapt a
particular situation or material to the teachings of the present
invention without departing from its scope. Therefore, it is
intended that the present invention not be limited to the
particular embodiment disclosed, but that the present invention
will include all embodiments falling within the scope of the
appended claims.
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