U.S. patent number 7,139,338 [Application Number 10/036,623] was granted by the patent office on 2006-11-21 for receiver.
This patent grant is currently assigned to Sony United Kingdom Limited. Invention is credited to Samuel Asangbeng Atungsiri, John Nicholas Wilson.
United States Patent |
7,139,338 |
Wilson , et al. |
November 21, 2006 |
Receiver
Abstract
A receiver is operable to detect and recover data from at least
one set of received signal samples. The signal samples comprise a
plurality of data bearing signal samples and a plurality of guard
signal samples before or after the data bearing signal sample, the
guard signal samples being formed by repeating a plurality of the
data bearing signal samples. The receiver comprises a matched
filter having a matched impulse response, a controller operable to
adapt the impulse response of the matched filter to the signal
samples of the guard signal samples, the matched filter being
operable to produce an output signal which is representative of the
convolution of the guard signal samples of the set with the
received signal samples. A synchronization detector is operable to
estimate the location of a sync position, from a distribution of
energy of the matched filter output signal with respect to the
received samples, the sync position providing the position of a
window of the received signal samples from which the data may be
recovered from the data bearing signal samples. The receiver can
provide an improvement in the detection of the data bearing signal
samples within the set of signal samples, by providing a more
reliable estimate of the synchronization position. The receiver can
be used for recovering data from signals modulated in accordance
with Orthogonal Frequency Division Multiplexing (OFDM) and finds
application as a receiver for Digital Video Broadcast (DVB)
signals.
Inventors: |
Wilson; John Nicholas (Hook,
GB), Atungsiri; Samuel Asangbeng (Basingstoke,
GB) |
Assignee: |
Sony United Kingdom Limited
(Weybridge, GB)
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Family
ID: |
9902901 |
Appl.
No.: |
10/036,623 |
Filed: |
November 7, 2001 |
Prior Publication Data
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Document
Identifier |
Publication Date |
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US 20020110202 A1 |
Aug 15, 2002 |
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Foreign Application Priority Data
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Nov 9, 2000 [GB] |
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0027423.3 |
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Current U.S.
Class: |
375/343 |
Current CPC
Class: |
H04L
25/0216 (20130101); H04L 27/2666 (20130101); H04L
27/2663 (20130101); H04L 27/2665 (20130101); H04L
27/2678 (20130101); H04L 27/2695 (20130101) |
Current International
Class: |
H04L
27/06 (20060101) |
Field of
Search: |
;375/150,152,340,343,355,362,356 ;370/208,210,503,345 |
References Cited
[Referenced By]
U.S. Patent Documents
Foreign Patent Documents
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0 615 352 |
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Sep 1994 |
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EP |
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2 307 155 |
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May 1997 |
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GB |
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Other References
Wahlqvist M et al: "Time Synchronization in the uplink of an OFDM
system" Vehicular Technology Conference, 1996. Mobile Technology
for the Human Race., IEEE 46th Atlanta, GA, USA Apr. 28-May 1,
1996, New York, NY, USA, IEEE, US, Apr. 28, 1996, pp. 1569-1573,
XP010162657 ISBN: 0-7803-3157-5. cited by other.
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Primary Examiner: Tran; Khanh
Attorney, Agent or Firm: Oblon, Spivak, McClelland, Maier
& Neustadt, P.C.
Claims
We claim:
1. A receiver for detecting and recovering data from at least one
set of received signal samples, said signal samples comprising a
plurality of data bearing signal samples and a plurality of guard
signal samples before or after the data bearing signal samples,
said guard signal samples being formed by repeating a plurality of
said data bearing signal samples, and said data is modulated onto
said data bearing signal samples in the frequency domain and
transformed to said data bearing signal samples into the time
domain to form the data bearing signal samples of said set of
received signal samples, said receiver comprising: a matched filter
having an impulse response that is controllably adaptable; a
controller configured to change the impulse response of said
matched filter to correspond with the signal samples of the guard
signal samples, said matched filter being configured to produce an
output signal which is representative of a convolution of the guard
signal samples of said set with said set of received signal
samples; a synchronisation detector operable to estimate the
location of a sync position, consequent upon a distribution of
energy with respect to time of said matched filter output signal
with respect to said received samples, said sync position providing
a position of a window of said received signal samples from which
said data may be recovered from said data bearing signal samples;
and a forward fourier transformer operable to recover the data by
performing a forward fourier transform on the signal samples within
said window.
2. A receiver as claimed in claim 1, wherein said controller is
operable to control said matched filter to convolve said received
signal samples with said matched filter impulse response starting
from a coarse estimate of said sync position providing a temporal
location of said guard signal samples, receiver comprising: a
correlator operable to correlate two samples from said set of
received signal samples separated by a temporal displacement
corresponding to the temporal separation of the samples of the
guard period and the data bearing signal samples from which the
guard signal samples have been formed, said controller being
operable to determine said correlation between said two samples at
each of a plurality of relatively displaced positions along said
received signal samples, and said synchronisation detector is
operable to determine an output value of said correlator for each
of said relatively displaced positions, and to estimate said coarse
sync position estimate in accordance with the displaced position
which produces the greatest output value from the correlator.
3. A receiver as claimed in claim 2, wherein said synchronisation
detector is operable to estimate said coarse sync position by
determining an amount of energy within a shortened averaging window
having a number of samples equal to the number of guard signal
samples divided by an integer number, the energy within said
shortened averaging window being determined for each output value
from said correlator produced for each of said relatively displaced
positions falling within said shortened averaging window, said
coarse estimate of said sync position being determined in
accordance with the relative position of said shortened averaging
window having the most energy.
4. A receiver as claimed in claim 1, wherein said receiver is
operable to process a plurality of said sets of received signal
samples, said synchronisation detector being operable to combine
said output signal from said synchronisation detector for each of a
plurality of sets of received signal samples, and to estimate said
sync position from a peak value of said combined output signal.
5. A receiver as claimed in claim 1, wherein said output signal has
a plurality of temporally separated peaks, said synchronisation
detector being operable to pre-process said output signal by
identifying the temporal position of said peaks within said output
signal which have an amplitude which is less than a predetermined
threshold, and setting the value of said output signal to a
predetermined default value at said identified temporal positions,
said sync position being determined from said pre-processed output
signal.
6. A receiver as claimed in claim 5, wherein said default value is
zero.
7. A receiver as claimed in claim 1, wherein said synchronisation
detector is operable to process said output signal by generating a
representation of the amount of energy in the said output signal
within a period corresponding to the temporal length of said guard
signal samples, for each of a plurality of relative displacements
of said guard period with respect to said output signal, and
determining the relative displacement of said period having the
most energy, and identifying the start of an analysis window of
said output signal from the temporal position of the beginning of
said guard period at said relative displacement of most energy,
said sync position being determined from within said analysis
window of said processed output signal.
8. A receiver as claimed in claim 7, wherein said synchronisation
detector is operable to determine the end of said analysis window
by reversing said output signal in time, generating a
representation of the amount of energy in the said reversed output
signal within said guard period, for each of a plurality of
relative displacements of said guard period with respect to said
reversed output signal, determining the relative displacement of
said period having the most energy, and identifying the end of said
analysis window of said output signal form the temporal position of
the beginning of said guard period at said relative temporal
displacement of most energy.
9. A receiver as claimed in claim 1, wherein said synchronisation
detector is operable to pre-process said output signal by locating
the relative temporal position of the maximum peak within said
output signal, identifying for each other peak sample of said
output signal another sample of said output signal at an opposite
corresponding temporal displacement with respect to said relative
temporal position of said maximum peak, comparing these two samples
and replacing the lower of the two samples with zero.
10. A receiver as claimed in claim 1, wherein said data is
modulated in accordance with Orthogonal Frequency Division
Multiplexing or Coded Orthogonal Frequency Division Multiplexing or
the like.
11. A receiver as claimed in claim 1, wherein said set of received
signal samples are complex samples having real and imaginary parts,
said impulse response having complex samples, and said controller
being operable to represent the real and imaginary components of
each of said received signal samples as a positive or negative
constant in dependence upon the relative sign of said real and
imaginary components, and to represent the real and imaginary
components of the samples of said matched filter impulse response
as a positive or negative constant in dependence upon the relative
sign of said real and imaginary components, said matched filter
being operable to convolve said impulse response with said received
signal samples by logically combining the representation of said
received signal samples and said impulse response.
12. A receiver as claimed in claim 11, wherein said logical
combining of said received signal samples and said impulse response
is summing the XOR compliment of the combination of the
representation of said received signal samples and said impulse
response.
13. A receiver as claimed in claim 12, wherein said matched filter
is operable to perform the convolution of said impulse response
with said received signal samples in accordance with the following
equation: .function..times..times..function..function..function.
##EQU00009## .function..times..times..function..function..function.
##EQU00009.2## where {overscore (XOR)} (a, b) is the compliment of
XOR(a, b), h.sub.m(n).I is the real part and h.sub.m(n).Q the
imaginary part of the complex samples of said output signal.
14. A storage medium for storing computer program product including
computer executable instructions, which when loaded onto a computer
configures the computer to operate as a receiver as claimed in
claim 1.
15. A method of detecting and recovering data from at least one set
of received signal samples, said set of signal samples comprising a
plurality of data bearing signal samples and a plurality of guard
signal samples before or after the data bearing signal samples,
said guard signal samples being formed by repeating a plurality of
said data bearing signal samples, and said data is modulated onto
said data bearing signal samples in the frequency domain and
transformed to said data bearing signal samples into the time
domain to form the data bearing signal samples of said set of
received signal samples said method comprising: controllably
changing an impulse response of a matched filter from a first
impulse response to a second impulse response that corresponds to
the signal samples of the guard signal samples; producing an output
signal which is representative of a convolution of the guard signal
samples of said set with said received signal samples; estimating a
location of a sync position, consequent upon a distribution of
energy with respect to time of said matched filter output signal
with respect to a relative convolution position in said received
samples, said sync position providing a position of a window of
said received signal samples from which said data may be recovered
from said data bearing signal samples, and performing a forward
fourier transform on the signal samples within said window.
16. A method as claimed in claim 15, further comprising controlling
said matched filter to convolve said received signal samples with
said matched filter starting from a coarse estimate of said sync
position, correlating two samples from said set of received signal
samples, separated by a temporal displacement corresponding to the
temporal separation of the samples of the guard period and the data
bearing signal samples from which the guard signal samples have
been formed, at each of a plurality of relatively displaced
positions along said received signal samples, determining an output
value of said correlator for each of said relatively displaced
positions, and generating said coarse estimate of said sync
position in accordance with the displace position which produces
the greatest output from the correlator.
17. A method as claimed in claim 16, wherein said generating said
coarse estimate of said sync position comprises determining an
amount of energy within a shortened averaging window having a
number of samples equal to the number of guard signal samples
divided by an integer number, the energy within said shortened
averaging window being determined for each output value from said
correlator produced for each of said relatively displaced positions
falling within said shortened averaging window, and determining
said coarse estimate of said sync position in accordance with the
relative position of said shortened averaging window having the
most energy.
18. A method as claimed in claim 15, comprising processing a
plurality of said sets of received signal samples, combining said
output signal produced for each of a plurality of sets of received
signal samples, and estimating said sync position from a peak value
of said combined output signal.
19. A method as claimed in claim 15, wherein said output signal has
a plurality of temporally separated peaks, said method comprising
identifying the temporal position of said peaks within said output
signal which have an amplitude which is less than a pre determined
threshold, setting the value of said output signal to a
predetermined default value at said identified temporal positions,
said sync position being determined from said pre-processed output
signal.
20. A method as claimed in claim 19, wherein said default value is
zero.
21. A method as claimed in claim 15, comprising generating a
representation of the amount of energy of said output signal within
a period corresponding to the temporal length of said guard signal
samples, for each of a plurality of relative displacements of said
guard period with respect to said output signal, determining the
relative displacement of said period having the most energy, and
identifying an analysis window of said output signal, said analysis
window starting from the temporal position of the beginning of said
guard period at said relative displacement of most energy, said
sync position being determined from within said analysis window of
said output signal.
22. A method as claimed in claim 21, comprising reversing said
output signal in time, generating a representation of the amount of
energy of said reversed output signal within said guard period, for
each of a plurality of relative displacements of said guard period
with respect to said reversed output signal, determining the
relative displacement of said period having the most energy, and
identifying the end of said analysis window of said output signal
from the temporal position in said reversed output signal
corresponding to the start of said guard period at said determined
relative displacement of most energy.
23. A method as claimed in claim 15, comprising locating the
relative temporal position of the maximum peak within said output
signal, identifying for each other peak of said output signal the
value of said output signal at an opposite corresponding temporal
displacement with respect to said relative temporal position of
said maximum peak, and if said output signal value at said
corresponding displacement is less than said peak value, setting
said output signal value to zero.
24. A method as claimed in claim 15, wherein said set of received
signal samples are complex samples having real and imaginary parts,
said impulse response having complex samples, said method
comprising representing the real and imaginary components of each
of said received signal samples as a positive or negative constant
in dependence upon the relative sign of said real and imaginary
components, and representing the real and imaginary components of
each of the samples of said matched filter impulse response as a
positive or negative constant in dependence upon the relative sign
of said real and imaginary components, said matched filter being
operable to convolve said impulse response with said received
signal samples by logically combining the representation of said
received signal samples and said impulse response.
25. A storage medium for storing computer program product including
computer executable instructions, which when loaded on to a
computer causes the computer to perform the method according to
claim 15.
Description
FIELD OF THE INVENTION
The present invention relates to receivers operable to detect and
recover data from received signal samples. The present invention
also relates to methods of detecting and recovering data from
received signal samples.
BACKGROUND OF INVENTION
Generally data is communicated using radio signals by modulating
the data onto the radio signals in some way, and transmitting the
radio signals to a receiver. At the receiver, the radio signals are
detected and the data recovered from the received radio signals.
Typically this is performed digitally, so that at the receiver, the
detected radio signals are down converted to a base band
representation and converted from analogue form to digital form. In
the digital form the base band signals are processed to recover the
data. However in order to recover the data, the receiver must be
synchronised to the received digital signal samples to the effect
that the relative temporal position of the recovered data symbols
corresponds with the temporal position of the data when
transmitted. This is particularly true for radio communications
systems in which the data is transmitted as bursts or packets of
data.
An example of a radio communications system in which data is
communicated in bursts or blocks of data is the Digital Video
Broadcasting (DVB) system. The DVB system utilises a modulation
scheme known as Coded Orthogonal Frequency Division Multiplexing
(COFDM) which can be generally described as providing K narrow band
carriers (where K is an integer) and modulating the data in
parallel, each carrier communicating a Quadrature Amplitude
Modulated (QAM) symbol. Since the data is communicated in parallel
on the carriers, the same symbol may be communicated on each
carrier for an extended period. Generally, this period is arranged
to be greater than a coherence time of the radio channel so that by
averaging over the extended period, the data symbol modulated onto
each carrier may be recovered in spite of time and frequency
selective fading effects which typically occur on radio
channels.
To facilitate detection and recovery of the data at the receiver,
the QAM data symbols are modulated onto each of the parallel
carriers contemporaneously, so that in combination the modulated
carriers form a COFDM symbol. The COFDM symbol therefore comprises
a plurality of carriers each of which has been modulated
contemporaneously with different QAM data symbols.
In the time domain, each COFDM symbol is separated by a guard
period which is formed by repeating data bearing samples of the
COFDM symbol. Therefore, at a receiver, to detect and recover the
data, the receiver should be synchronised to each COFDM symbol and
the data demodulated from the data bearing signal samples of the
COFDM symbol. A previously proposed technique for acquiring
synchronisation with the data bearing signal samples of a COFDM
symbol is to cross correlate two samples which are temporally
separated by the period over which the data bearing samples are
modulated. A relative temporal position of the two samples is then
shifted within the COFDM symbol, until a position is found at which
the cross-correlation produces maximum energy.
Although the previously proposed synchronisation technique works
adequately in the presence of additive white gaussian noise, in
some situations such as where the signal is received in the
presence of multi-path propagation, this technique produces a
sub-optimum synchronisation point, which can cause the data bearing
signal samples to be corrupted with energy from adjacent signal
samples. This is known as inter-symbol interference (ISI).
SUMMARY OF INVENTION
According to the present invention there is provided a receiver for
detecting and recovering data from at least one set of received
signal samples, the signal samples comprising a plurality of data
bearing signal samples and a plurality of guard signal samples
before or after the data bearing signal samples, the guard signal
samples being formed by repeating a plurality of the data bearing
signal samples, the receiver comprising a matched filter having an
impulse response, a controller operable to adapt the impulse
response of the matched filter to the signal samples of the guard
signal samples, the matched filter being operable to produce an
output signal which is representative of the convolution of the
guard signal samples of the set with the set of received signal
samples, and a synchronisation detector operable to estimate the
location of a sync position, consequent upon a distribution of
energy with respect to time of the filter output signal with
respect to the received samples, the sync position providing the
position of a window of the received signal samples from which the
data may be recovered from the data bearing signal samples.
As will be explained in more detail shortly, a receiver embodying
the present invention provides an improvement in detecting and
recovering of data from data bearing samples forming a set of
received signal samples which include guard signal samples
temporally positioned either before or after the data bearing
signal samples. The guard signal samples are formed by repeating
the data bearing signal samples over a period which is equivalent
to the guard interval. By providing the receiver with a filter and
a controller which adapts the impulse response of the filter to the
signal samples of the guard interval an improved estimate of the
sync position is provided. A synchronisation detector may be
operable to detect a synchronisation (sync) position in accordance
with the distribution of energy with respect to time of the matched
filter output response produced as the received signal passes
through the filter. The peaks of the filter output signal generally
correspond to replicas of the transmitted signal generated by
propagation over the radio path. The receiver can therefore derive
an improved estimate of the sync position by locating the earliest
significant component of the radio channel and locating the window
from which data can be recovered from the data bearing signal
samples starting from this earliest significant component. This is
particularly advantageous in the case of where the received signal
is detected in the presence of multi-path propagation.
Although the sync position can be acquired by filtering all the
signal samples within the set of received signal samples with the
filter, in preferred embodiments the controller may be operable to
control the filter to convolve the received signal samples with the
filter starting from a coarse estimate of the sync position.
Accordingly, the receiver may comprise a correlator operable to
correlate two samples from the set of received signal samples
separated by a temporal displacement corresponding to the temporal
separation of the guard period and the data bearing signal samples
from which the guard signal samples have been formed, the
correlation between the two samples being determined at each of a
plurality of relatively displaced positions, and the
synchronisation detector may be operable to determine an output
value of the correlator for each of the displaced positions, and to
estimate the sync position in accordance with the displaced
position which produces the greatest output from the correlator. A
coarse estimate of the sync position is determined at each of a
plurality of relatively displaced positions, the synchronisation
detector can therefore determine the fine estimate of the sync
position more quickly starting from the coarse estimate. Thus
preferably, the number of positions at which the displaced samples
are cross correlated is determined from the number of guard signal
samples divided by an integer number.
Although the receiver could be arranged to detect the sync position
of only one set of received signal samples, in one application of
the receiver according to the present invention, the receiver is
operable to process a plurality of sets of received signal samples.
Advantageously, therefore the synchronisation detector may be
operable to combine the peaks of the output signal from the
synchronisation detector for each of a plurality of sets of
received signal samples and to estimate the sync position from a
peak value of the combined output signal. Effectively therefore the
synchronisation detector integrates the output signal over
successive sets of received signal samples and therefore provides a
more accurate determination of the sync position from a peak value
of the combined output signal.
Although the example embodiment of the present invention has been
described with reference to Coded Orthogonal Frequency Division
Multiplexing (COFDM), it will be appreciated that the present
invention is not limited to this modulation scheme, but finds
application with any other COFDM variant such as Orthogonal
Frequency Division Multiplexing (OFDM) or indeed any other
modulation and communication scheme could be used. The present
invention finds application with any receiver which is arranged to
detect data from a set of received signal samples comprising data
bearing signal samples and guard signal samples which are
reproduced by replicating the data bearing signal samples.
For some applications, the number of received signal samples in the
set of received signal samples may be relatively large. For
example, for an application with digital video broadcasting, the
COFDM symbols are comprise either 2048 signal samples (2K version)
or 8192 signal samples (8K mode). Furthermore the set of received
signal samples and the impulse response comprise complex samples
having real and imaginary parts. As a result the filter which is
required to perform the convolution of the guard signal samples
with the set of received signal samples would be required to have
either 1536 taps or 6144 taps, for the 2K and 8K modes
respectively, which represents a prohibitive number for
implementation and for real time operation. However in preferred
embodiments the controller may be operable to represent the real
and imaginary components of each of the received signal samples as
a positive or negative constant in dependence upon the relative
sign of the real and imaginary parts and to logically combine the
impulse response with the received signal samples to produce the
output signal. As such by representing the impulse response and the
received signal samples as the constant for each of the real and
imaginary parts, an approximation of the convolution of the
received signal samples and the filter impulse response is
generated from which the sync position can be determined. Thus the
filter is implemented with a considerably reduced complexity.
Various further aspects and features of the present invention are
defined in the appended claims.
BRIEF DESCRIPTION OF THE DRAWINGS
One embodiment of the present invention will now be described by
way of example only with reference to the accompanying drawings
wherein:
FIG. 1 is a schematic representation of two successive COFDM
symbols;
FIG. 2 is a schematic block diagram of a previously proposed
synchronisation detector;
FIG. 3 is a schematic representation of the two COFDM symbols of
FIG. 1 being received at a receiver via a main path and an echo
path;
FIG. 4 is a schematic block diagram of a receiver according to an
embodiment of the present invention;
FIG. 5 is a schematic block diagram of a synchronisation detector
forming part of the receiver shown in FIG. 4;
FIG. 6 is a schematic representation illustrating the processing of
OFDM symbols by a matched filter forming part of the receiver of
FIG. 5;
FIG. 7 is a table providing an indication of the meaning of symbols
which are used throughout the description and drawings.
DESCRIPTION OF PREFERRED EMBODIMENTS
An example embodiment of the present invention will now be
described with reference to detecting and recovering data from a
COFDM symbol produced for example in accordance with the Digital
Video Broadcasting (DVB) standard. The DVB standard is disclosed in
a publication by the European telecommunications standards
institute number EN300744 version 1.1.2 (1997-08) and entitled
"Digital Video Broadcasting (DVB); Frame Structure Channel Coding
And Modulation For Digital Terrestrial Television".
As already explained, a COFDM symbol which is modulated in
accordance with DVB standard is generated by modulating K narrow
band carriers in parallel with the data to be communicated.
Generally as disclosed in the above referenced ETSI publication,
the COFDM symbols are formed in the frequency domain and then
converted in the time domain using an Inverse Fourier Transform. A
diagram representing the form of the COFDM symbols is shown in FIG.
1. In the following description, symbols which are used to
represent various quantities are summarised in a table shown in
FIG. 7.
In FIG. 1 two COFDM symbols represented as blocks 1, 2 are shown as
they would be transmitted by a DVB transmitter with time
progressing from left to right across the page. As shown in FIG. 1,
each COFDM symbol 1, 2 has a useful part of the symbol during which
the data is transmitted. This part of the symbol has duration of
T.sub.u seconds and has N.sub.u samples. A guard interval G.1, G.2
of duration T.sub.g seconds separates the current symbol from the
previous one. The guard interval has Ng samples. For each symbol 1,
2 the guard interval G.1, G.2 therefore precedes the useful part of
the symbol and is formed, as indicated by an arrow 4, by
replicating the samples in the last T.sub.g seconds of the useful
part of the symbol. Each COFDM symbol of N.sub.s samples therefore
has duration T.sub.s=T.sub.g+T.sub.u seconds.
In order to recover the data within the COFDM symbols, the receiver
must detect the data bearing signal samples from within the set of
received signal samples corresponding to each COFDM symbol. Symbol
acquisition entails the location of the optimum point at which the
window for FFT processing should start. The FFT forms the core of
the COFDM demodulator.
The replicated samples during the guard interval G.1, G.2 can be
used to locate the start of each symbol at the receiver. This is
what is referred to above as the location of the FFT window since
the FFT must be performed over a segment of duration T.sub.u that
preferably covers only the useful part of the symbol. However, FFT
windows that start elsewhere within the guard interval can also be
tolerated. Such FFT windows result in a phase slope at the output
of the FFT that can be corrected if the FFT window location is to
within T.sub.g seconds before the correct location. If however the
window location error is excessive, the resultant phase slope wraps
around .+-..pi./2 radians and so cannot be resolved and corrected.
This results in inter-symbol interference (ISI) which degrades the
receiver performance.
FIG. 2 provides an illustration of a previously proposed
synchronisation detector for detecting the FFT window. Once the FFT
window of the data bearing signal samples has been located, the
data is recovered from these data bearing signal samples by
applying an FFT. The FFT therefore converts the signal samples back
into the frequency domain from which the data can be recovered from
the K carriers. In FIG. 2 the two COFDM symbols 1, 2 are shown to
be received by a correlator 10. A delay line 12 is arranged to
delay the signal samples of the received symbols 1, 2 by an amount
corresponding to the length of the signal samples T.sub.u. The
correlator 10 has a first input 14 which receives signal samples
from the delay line 12 and a second input 16 which receives
contemporaneously signal samples from the set of samples forming
the received COFDM symbol. An output of the correlator c(n) is then
fed to an adder 18 on a first input 20. The adder 18 is arranged to
receive the output from the correlator 10 on a second input 22
delayed by a delay 24 by a period equal to that of the guard
interval. The adder 18 also receives on a third input 26 samples
from the output of the adder 18 a(n) fed back via a one sample
delay 28. The output from the adder a(n) is then received on a
first input of a second adder 30 which also has second and third
inputs. The second input receives a version of the output from the
first adder a(n) via a delay producing a delay equal to the guard
interval 32 and the third input of the second adder 30 receives
samples fed back from the output of the second adder via a further
one sample delay 34. The output of the second adder e(n) provides a
signal from which the FFT window for recovering the data from the
COFDM symbols can be determined.
The first adder 18 in combination with the delays 24, 28
effectively form a moving averaging window filter having a length
equal to the period of the guard interval. The second adder 30 and
the delays 32, 34 form a second moving averaging filter having a
window equal to the period of the guard interval. The outputs of
the correlator, the first adder and second adder are also shown in
FIG. 2 plotted with a respect to time in relationship with the time
axis of the COFDM symbols 1, 2. As shown the output of the
correlator c(n) produces a square pulse which corresponds
effectively with the auto-correlation of the guard interval. This
is of course in the absence of noise. The output of the first adder
a(n) produces an integration of the output of the correlator c(n)
and the output e(n) of the second adder 30 produces a further
integration of the output of the first moving averaging window. For
each symbol, the samples in the guard interval also occur in the
last T.sub.g seconds of the symbol. The moving window also has
duration of T.sub.g seconds. Equation (1) illustrates the
computation of the correlation for each received sample r(n)=r(nT)
where T is the sample time of the received sequence at the
receiver: c(n)=r(n)r*(n-N.sub.u)
.function..times..times..function. ##EQU00001## for n=0,1, . . .
N.sub.s-1 where r*(n) is the complex conjugate of r(n).
The value of n which produces the maximum of a(n) within a given
sequence of N.sub.s received samples therefore provides the optimum
coarse sync position. The moving correlation sequence a(n) can be
averaged with similar sequences computed over a number of
successive N.sub.s length windows in order to improve acquisition
performance during low signal to noise ratio (SNR). Furthermore,
the correlation sequence a(n) can be also integrated over a window
of T.sub.g seconds in order to enhance performance in dispersive or
multi-path channels. This integration which can be done prior to
the averaging over successive T.sub.s seconds windows is described
by the equation (2).
.function..times..times..function. ##EQU00002##
Effectively then a peak of the second moving averaging window
provides an indication of the start of the data bearing signal
samples. The FFT window 35 of duration T.sub.u corresponding to the
period of the data bearing signal samples is therefore
determined.
Although the synchronisation detector shown in FIG. 2 can provide
an indication for the FFT window for recovering the data from the
COFDM symbol, in some situations for example where the received
signal is detected in the presence of multi-path propagation, the
sync position from which the FFT window is positioned can be offset
if the received signal is detected in the presence of multi-path,
which can cause inter-symbol interference between COFDM symbols to
the effect that the FFT window includes energy from the guard
signal samples from the next COFDM symbol. This causes errors in
the detected data. In FIG. 3 the COFDM symbols 1, 2 are shown with
a second version of the COFDM symbols 1.sup.1, 2.sup.1 which
provide a schematic illustration of multi-path propagation via a
first main path 50 and a second echo path 52. A representation of
the output of the correlator 10 c(n) due to the echo path 52 and
the main path 50 are represented graphically in FIG. 3 by the lines
54 for the echo signal and for the main signal 56. As detected at
the receiver, the correlator output signal will be effectively a
combination of the contributions from the main signal 50 and the
echo signal 52. The output of the correlator as experienced in the
presence of multi path comprising a main and an echo component is
represented graphically by a third line 58 which effectively forms
the combination of the correlator for the main and echo paths. A
result of the first and second moving averaging filters is
represented graphically on the fourth and fifth plots 59, 61 of
amplitude with respective time for the output of the first adder 18
a(n) and the second adder 30 e(n).
As illustrated by the fourth and fifth graphical representations a
result of the pre-cursor echo path 52 is to shift the
synchronisation point detected by the correlator so that the FFT
window is now biased to a point later in time then the ideal
position which is illustrated in FIG. 3 by an arrow 63. In the
presence of a low power pre-cursive echo, the window location will
be skewed towards the higher energy echo that occurs later. As the
separation between the echoes increases, it becomes more likely
that the window location derived from the above algorithm will fall
outside the guard interval of the low power path and so result in
failure to correctly equalise the low power echo. When this
happens, the low power echo becomes the source of ISI thereby
degrading the performance of the demodulator.
As shown in FIG. 3, a window 60 which defines the signal samples
from which data is recovered now includes signal samples which are
influenced by the guard interval for an un-related COFDM symbol for
the pre-cursive echo path. As a result the samples from within a
region 62 will be affected with inter-symbol interference and
therefore will be more likely to cause errors in all carriers after
the FFT when the data is recovered from the carriers in the
frequency domain.
A receiver for detecting and recovering data from for example a
COFDM symbol is shown in FIG. 4. The receiver shown in FIG. 4 is
operable to correlate the guard signal samples with the set of
received signal samples which provides a representation of the
multi-path components, thereby allowing the FFT window positioning
decision to be taken on the basis of the time occurrence of
significant echoes rather than merely the energy of these echoes.
In implementation, a compact architecture is also proposed which
results in a lower gate count and improved speed. Furthermore, the
synchronisation detector is also used for tracking and time
adjustment of the FFT window position as echoes are born, change in
energy and/or finally disappear.
In FIG. 4 an analogue to digital converter 100 is arranged to
receive an intermediate frequency (IF) signal representative of the
detected radio signal on which the COFDM symbol has been modulated.
The receiver also includes down conversion means and detection
means in order to convert the radio frequency signal into an
intermediate frequency signal which is fed to the analogue to
digital converter 100 via an input 102. Thus it will be appreciated
that the receiver may also include radio frequency receiving and
down converting means which are not shown in FIG. 4. After being
analogue to digitally converted the received signal is processed by
an intermediate frequency to base band version means 104 before
being processed by a re-sampling and carrier offset correction
processor 106. The re-sampling and carrier offset correction
processor is arranged to track in the frequency domain the K
carriers of the COFDM modulation. The base band received signal
samples are then fed to a Fast Fourier transform processor 108
which serves to convert the time domain received signal samples
into the frequency domain. The data is then recovered from the
frequency domain signal samples by a post FFT processor 110. The
data is then fed to a forward error correction processor 112 which
operates to decode the error correction encoded data to finally
produce the recovered data at an output 114.
The receiver according to this example embodiment provides a
synchronisation detector which locates the FFT window from which
the data bearing signal samples are processed by the FFT processor
108. The FFT window position is adjusted in order that the window
includes the maximum energy representative of the data bearing
signal samples. To this end an FFT symbol timing recovery processor
116 is arranged to generate a signal indicative of a sync position
which is fed to the FFT processor 108 via a connecting channel 118.
The FFT symbol timing recovery processor 116 is arranged to detect
the sync position from the received set of signal samples which
represent each COFDM symbol. These are received from the
re-sampling and carrier offset correction processor 106 via a
connecting channel 120. The operation of the FFT symbol timing
recovery processor 116 will now be described with reference to FIG.
5 where parts also appearing in FIG. 4 bear the same numerical
designations.
The symbol time recovery processor shown in FIG. 5 is comprised
generally of a coarse acquisition processor 200 and a fine
acquisition processor 202. The coarse acquisition processor
operates generally in accordance with the previously proposed
synchronisation detector shown in FIG. 2.
Coarse Synchronisation Detector
As shown in FIG. 5 the coarse acquisition processor comprises a
correlator 204 which is arranged to receive the set of received
signal samples corresponding to the COFDM symbol via a first input
206. The set of received signal samples are also received via a
second input 208 but delayed by a period T.sub.u corresponding to
the temporal length of the data bearing signal samples of the COFDM
symbol. The correlator 204 is arranged to cross correlate the two
signal samples from the received signal as previously explained
with reference to the previously proposed detector shown in FIG. 2.
The correlator then feeds the result of the correlation to a first
moving averaging filter 206 which integrates the output of the
correlation. This is in turn fed to a second moving averaging
filter 208 which integrates the output of the first moving
averaging filter. The output of the second moving averaging filter
208 is then integrated on a symbol by symbol basis by an
integration processor 210. The integration processor 210 serves to
integrate the output signal from the second moving averaging signal
208 over successive COFDM symbols so that a combined output is
produced for these successive symbols. The output of the integrator
210 is then fed to a peak detector 212. The peak detector 212 is
arranged to generate a peak value of the symbol integrator. However
unlike the previously proposed arrangement shown in FIG. 2, the
length of the integration over e(n) is reduced to only N.sub.g/4 to
make the energy peak even more biased towards the highest echo as
the fine synchronisation detector depends on accurate location of
the dominant path. A peak detector 212 then determines the relative
displacement which corresponds to the peak of the integrated output
signal from the integrator 210 therefore providing a coarse trigger
point to the fine synchronisation detector 202.
Fine Synchronisation Detector
The fine synchronisation detector provides an improved estimated of
the sync position by utilising a transversal filter which is
adaptively matched to the guard interval of successive COFDM
symbols. An end of symbol marker is obtained for the dominant
multi-path component from the coarse synchronisation detector. This
is used to locate the start of the guard interval on each symbol.
For symbol m, the received signal either side and including its
guard interval, which comprises 3N.sub.g of samples, are used to
set the taps f.sub.m(i) of the transversal filter. In effect,
therefore the received signal is correlated with respect to
3N.sub.g worth of samples. This therefore allows for some error in
the coarse estimated location of the guard signal samples. Once the
filter taps have been set the block r.sub.m(n) of the last N.sub.g
samples of the symbol, which were copied to form the guard interval
are filtered by the matched filter to produce an output signal. As
the filter is excited with these samples, a pulse train h.sub.m(n)
representing an approximation to the channel impulse response (CIR)
during symbol m is produced at the output since the filter is
nominally matched to its excitation. This is represented in
equation (3).
.function..times..times..function..times..function.
##EQU00003##
The fine synchronisation detector 202 is also arranged to receive
the set of received signal samples and the delayed set of received
signal samples from the first and second inputs 206, 208. The
received signal samples from the first and second inputs 206, 208
are fed respectively to first and second binary converters 230,
232. The output from the binary converter is fed to a first input
of an adaptive matched filter 234. A second input to the adaptive
matched filter is fed with samples from the output of the binary
converter 232 via a delay line 236 which serves to delay each
sample by a period corresponding to the number of samples within
the guard period. The output of the adaptive matched filter 234 is
received at an integrator 238 forming part of a synchronisation
detection processor 235. The integrator 238 serves to integrate the
output of the matched filter, the integrated output being presented
on first and second outputs 240, 242 to a centre clip processor 244
and a centre clip level calculator 246. As will be explained
shortly, the centre clip processor and the centre clip level
calculator 244, 246 are arranged to pre-process the output of the
adaptive matched filter which has been integrated by the
integration processor 238 to the effect of cancelling various peaks
of the adaptive filter output which could otherwise give a false
indication of the synchronisation point. As such the performance of
the synchronisation detector is improved particularly in the
presence of noise.
The pre-processed output from the centre clip processor is then fed
to a channel impulse response windowing processor 248. The
windowing processor 248 provides a further pre-processing operation
to the effect of isolating an analysis window within which the
pre-processed output of the adaptive filter produces the maximum
energy. It is within this analysis window that a peak output of the
adaptive matched filter is determined by an error detection
processor 250 with respect to the coarse synchronisation estimate
provided by the coarse synchronisation detector 200. The operation
of the fine synchronisation detector is controlled by a controller
260.
The error detector 250 produces a corrected synchronisation
position at an output 252 which is combined with the coarse
synchronisation estimate provided at an output 220 by a combiner
254 to produce the start point of the FFT window generated at an
output 256. The operation of the fine synchronisation detector and
in particular the operation of the adaptive filter 234 will now be
described with reference to FIG. 6 where parts also appearing in
FIG. 5 have the same numerical references.
In FIG. 6 a representation of each of the guard signal samples for
successive COFDM signals is represented graphically by sections 300
which are arranged to feed a shift register 302. The taps 304 of
the shift register are arranged to provide the samples from the
guard period. For each successive COFDM symbol, the shift register
taps 304 are arranged to represent three guard signal samples worth
3N.sub.g of the received signal samples around, and including the
guard signal samples for the particular COFDM symbol. As such, the
controller 260 within the fine synchronisation detector is arranged
in operation to adapt the taps of the matched filter 234 to
correspond to the 3N.sub.g signal samples of the guard period and
surrounding samples for each of the successive COFDM symbol.
Therefore, as shown in FIG. 6 for the m-th symbol, the matched
filter is adapted to have the 3N.sub.g signal samples around and
including the m-th guard interval. As illustrated by an arrow 306
shown in FIG. 6, the received signal samples from which the guard
signals samples were formed are fed into and excite the adapted
matched filter 234. As a result of the excitation of the matched
filter by the received signal sample copied to form the guard
signal samples which is indicated by the direction of shift with
respect to time 308, an output of the adder 310 when the position
of the copy received signal samples coincides with the guard
samples is to produce a peak output 320 at the output of the adder
316. As each successive symbol is received, the filter is adapted
by replacing its taps with the samples around the guard interval
and then waiting for the arrival of the excitation samples. The
output of the filter for each symbol is an array of 2N.sub.g+1
complex numbers. The output arrays from N.sub.fs successive symbols
as shown in equation (4) are averaged together to filter out noise
from the CIR. Then, by computing the absolute value for each
averaged impulse, the magnitude of the CIR is derived. As such the
output of the matched filter generates effectively the convolution
of the guard signal samples with itself which produces effectively
a representation of the channel impulse response. By analysing the
peaks from the output of the matched filter, an improved
synchronisation position can be formed by detecting the earliest of
the peaks in order to mitigate the ISI. From this sync position the
FFT window can be positioned and from this the data recovered from
the COFDM symbol.
.function..times..times..function. ##EQU00004##
As already explained, coarse acquisition only gives the dominant
multi-path component. Components located within .+-.T.sub.g seconds
of this dominant component can be equalised. Therefore, the
transversal filter has to have a length of at least 3N.sub.g to
cover both pre-cursive and post-cursive components.
As already explained, the synchronisation detector 250 detects the
sync position for the start of the FFT window, from the output of
the matched filter. However, in order to improve the probability of
correctly detecting the optimal sync position, the fine
synchronisation detector 202 is arranged to pre-process the output
signal from the matched filter before the sync position is detected
from the pre-processed output signal. The pre-processing is
performed by the integrator 238, the centre clip processor 244, the
centre clip level calculator 246, and the CIR windowing processor
248. The pre-processing operations performed by each of these
processors will now be explained in the following paragraphs.
Noise Elimination from Magnitude CIR
The signal-to-noise ratio of the output of the matched filter h(n)
can be improved by increasing the number of symbols N.sub.fs over
which the output of the adaptive matched filter is averaged. The
averaging is performed by the integrator 238. This is represented
by equation (4). However, a large value for N.sub.fs also implies
longer acquisition times (during acquisition) and longer update
times (during tracking). The choice of N.sub.fs is therefore of
necessity, a compromise. This means that the magnitude CIR h(n) is
often quite noisy. The noise is reduced by the centre clip
processor 244.
The output signal would generally comprise a plurality of
temporally separated peaks from which the synchronisation position
is detected. However in order to provide an improved estimate of
the sync position particularly for example in the presence of
noise, the synchronisation detector may be processed with a centre
clip processor 244 operable to pre-process the output signal by
identifying the temporal position of the peaks within the output
signal which have an amplitude which is less than a predetermined
threshold and setting the value of the output signal to a
predetermined default value at the identified temporal positions.
Therefore, effectively any peak having a value which is less than a
predetermined threshold is set to a predetermined value, such as
zero so that particularly in the presence of noise the sync
position can be detected from amongst the largest peak.
Ghost Echo Elimination
The output of the matched filter exhibits some peaks which do not
represent real echoes, that is to say, echoes produced by
propagation paths of the radio channel. This is because the matched
filter impulse response and its excitation have both travelled
through the same channel and so have each been convolved with the
channel impulse response. During the matched filtering process,
these channel impulse response components autocorrelate and produce
unwanted peaks at the output of the filter. These unwanted peaks
are referred to as ghost echoes. Preferably, the ghost echoes
should be identified and cancelled in order to improve the
likelihood of correctly detecting the optimum synchronisation
position. The centre clip processor 244 is also arranged to cancel
ghost echoes from the output signal. A further improvement is
provided in the detection of the sync position by pre-processing
the output signal by locating the relative temporal position of the
maximum peak within the output signal, identifying for each other
peak sample of the output signal another sample of the output
signal at an opposite corresponding temporal displacement with
respect to the relative temporal position of the maximum peak,
comparing these two samples and replacing the lower of the two
samples with zero.
Embodiments of the present invention can therefore at least in part
cancel echoes so that the peak value corresponding to the sync
position can be more effectively identified from only the peaks of
the output signal which correspond to the actual channel impulse
response. This is done using the centre-clipper processor 244 in
combination with the centre clip level calculator 246. Significant
echoes in h(n) are determined by comparing their magnitude to a
pre-determined threshold, the level of which is based on the
minimum level of power an un-equalised component or echo needs to
have to cause noticeable degradation on the performance of the
demodulator. Only echoes that lie above this threshold are
retained. All other samples of h(n) are set to a default value such
as zero.
Pulses indicating real echoes are often higher in amplitude than
their corresponding ghosts. For any real echo located N samples
before the main path, a ghost is located N samples after the main
echo in the output signal and vice versa. To eliminate ghost
echoes, the centre clip processor 244 of the synchronisation
detector 235 tests the output signal at equal distances on either
side of the main path, retaining the higher amplitude impulse and
setting the lower to zero.
Determination of the Channel Impulse Response Length
At the receiver, the maximum length of the channel impulse response
of the radio channel which can be resolved corresponds to the guard
period. However, the magnitude of the output of the matched filter
h(n) according to equation (3) will be of length 2N.sub.g+1
samples. From this only an analysis period of length N.sub.g+1 that
forms the actual channel impulse response is relevant to finding
the sync position. This means that only echoes which lie within one
guard interval need to be processed. Therefore in order to further
improve the estimate of the sync position, in preferred embodiments
the synchronisation detector 235 may be provided with a CIR
processor 248 operable to pre-process the output signal by
generating a representation of the amount of energy of the output
signal within a period corresponding to the temporal length of the
guard signal samples, for each of a plurality of relative
displacements of the guard period with respect to the output
signal. The relative displacement of the period having the most
energy is then determined, this corresponding to the start of a
window of the output signal which contains a representation of the
channel impulse response.
In operation, the channel impulse response processor 248 passes the
output h(n) of the matched filter through a moving average filter
of order N.sub.g+1 to produce an output signal y(n) corresponding
to the energy in the signal at the respective displacements
according to equation (5):
.function..times..times..function..times..times..times..times..times..tim-
es. ##EQU00005## where h(n) is the output of the matched
filter.
The index I.sub.y for which y(n) is maximum is therefore the start
of the window which contains a representation of the impulse
response of the channel, the start of the window corresponding to
the optimum sync position being sought
Advantageously, the CIR processor 248 may have a second moving
average filter, the controller 260 being operable to also excite
this second moving average filter with the output samples of the
matched filter but in reverse order. The moving average filter is
operable to produce a second output signal which is representative
of the energy in the reversed matched filter output.
The output of the second moving average filter can be expressed by
equation (6), where z(n) is the output of the second moving average
filter. As in the first moving average filter, the index I.sub.z
for which z(n) is maximum is also identified. The length of the
channel impulse response can then be computed as the difference
(I.sub.z+N.sub.g+I.times.y) between the two indices from equations
(5) and (6).
.function..times..times..function..times..times..times..times..times..tim-
es..times. ##EQU00006##
Therefore a further improvement in the probability of correctly
recovering the data is provided because the channel impulse
response can be more accurately determined within the analysis
window set between the start and end positions determined as a
result of this pre-processing.
Tracking
During normal operation, new multi-path components might emerge
whilst existing ones might change in amplitude. This happens
because of movement of objects around the transmitter and/or
receiver or across the propagation path etc. In such operation a
pre-cursive channel profile might change into a post-cursive
profile and vice versa. To maintain performance, in preferred
embodiments, the controller 260 the FFT window location is arranged
to track these changes in the channel profile. The techniques
described above are also used to track changes in channel profile.
The location of the dominant multi-path component nominally
coincides with the middle tap of the transversal filter. Since the
filter has a length of 3N.sub.g samples, we can see multi-path
components that are located within one guard interval can be
identified either side of the dominant path.
The last output sample of the transversal filter occurs nominally
at 2N.sub.g samples after the start of the dominant path symbol.
From the centre of the filter multi-path components can be
identified to within .+-.N.sub.g. Therefore the location of the FFT
window can be advanced or retarded by adjusting the FFT window
point for the earliest echo. In tracking mode, the complex output
from the matched filter is averaged over N.sub.t symbols so as to
filter out any noise in the estimates. The ideal FFT window start
location is then calculated at T.sub.g seconds later than the start
of the optimum CIR. This is compared with the current FFT window
start position and adjusted accordingly if the two are
different.
Implementation of the Matched Filter
In order to filter the received signal samples with an impulse
response corresponding to the guard signal samples, the filter must
perform a convolution of complex signal samples of the COFDM
symbol. In 2K mode and 1/4 guard for example, the filter therefore
has at least 3*512=1536 taps. Similarly, in 8K mode and 1/4 guard,
the filter has 3*2048=6144 taps. Since each tap is complex and the
excitation samples are also complex, such filters could be
prohibitively costly in gates and/or processing delay to implement.
Preferably, a compact architecture with manageable gate count
and/or processing delay should be used. For reasonable performance
in the presence of interference most COFDM demodulators adopt at
least 10 bits per quadrature sample. This means that for the filter
described above, each tap would be represented with two 10-bit
numbers for the real and imaginary parts. Since however, only the
positions and relative amplitudes of the multi-path components are
required to determine the sync position, instead of the absolute
amplitude, in preferred embodiments, each quadrature tap can be
represented by only the sign of its respective I and Q components.
For each tap, this saves, for this example, eighteen bits. Each
sample of the received signal is also represented similarly.
Therefore, both the dynamic range and bit widths of the arithmetic
used in computing the filter output signal can be now significantly
reduced as outlined below.
Instead of representing the I and Q components of the taps and
excitation as .+-.1, these values are represented with logic 1 for
+/-1 and logic 0 for -/+1. Then the filter equation becomes:
.function..times..times..function..function..function..times..times..func-
tion..times..times..function..function..function. ##EQU00007##
where {overscore (XOR)} (a, b) is the compliment of XOR(a, b). This
avoids a requirement to calculate multiplication and, since the
accumulator is only summing .+-.1, a more compact arrangement can
be made, which can use a slower ripple adder.
Further savings in arithmetic and gates can be achieved by reducing
the number of taps over which the filter equation is computed for
each output. This can be achieved by either taking only a fraction
of the taps e.g. only the middle N.sub.g/2 taps or by decimation of
the taps, for example, for a decimation by 4, the above equations
become:
.function..times..times..function..function..times..function..times..time-
s..times..function..times..times..function..function..times..function..tim-
es. ##EQU00008##
The decimation factor affects the purity of the filter output. In
practice therefore, different optimum decimation factors for each
combination of mode and guard interval duration can be chosen.
As will be appreciated, a receiver having a matched filter
implementation employing the simplified arrangement for calculating
the convolution by logically combining the received signal samples
and the filter impulse response may be used to detect a
synchronisation position in any received signal having a
predetermined characteristic. For the example embodiment described
above, this predetermined characteristic is that the samples of the
guard period are generated by copying data conveyed in another part
of the received signal samples. However in other embodiments the
predetermined characteristic may be any predetermined signal
format, so that the impulse response of the filter is not limited
to being adapted to the guard signal samples. For example, the
filter may be matched to a known data sequence which may be either
a pre-amble to the data to be detected or a mid-amble or a
post-amble.
Various modifications may be made to the example embodiments herein
before described without departing from the scope of the present
invention. In particular, it will be appreciated that the
synchronisation detector can be applied to any signal in which the
guard interval is produced from repeating data bearing signal
samples or repeating any other part of the transmitted signal.
* * * * *