U.S. patent number 6,943,563 [Application Number 10/138,989] was granted by the patent office on 2005-09-13 for probe tone s-parameter measurements.
This patent grant is currently assigned to Anritsu Company. Invention is credited to Jon S. Martens.
United States Patent |
6,943,563 |
Martens |
September 13, 2005 |
Probe tone S-parameter measurements
Abstract
An S-parameter measurement technique allows measurement of
devices under test (DUTs), such as power amplifiers, which require
a modulated power tone drive signal for proper biasing, in
combination with a probe tone test signal, wherein both the
modulated and probe tone signals operate in the same frequency
range. The technique uses a stochastic drive signal, such as a CDMA
or WCDMA modulated signal in combination with a low power probe
tone signal. A receiver in a VNA having a significantly narrower
bandwidth than the modulated signal bandwidth enables separation of
the modulated and probe tone signals. VNA calibration further
improves the measurement accuracy. For modulated signals with a
significant power level in the frequency range of the probe tone
signal, ensemble averaging of the composite probe tone and power
tone signals is used to enable separation of the probe tone signal
for measurement.
Inventors: |
Martens; Jon S. (San Jose,
CA) |
Assignee: |
Anritsu Company (Morgan Hill,
CA)
|
Family
ID: |
26836756 |
Appl.
No.: |
10/138,989 |
Filed: |
May 2, 2002 |
Current U.S.
Class: |
324/638;
324/601 |
Current CPC
Class: |
G01R
27/28 (20130101); H04B 17/20 (20150115) |
Current International
Class: |
G01R
27/00 (20060101); G01R 27/28 (20060101); H04B
17/00 (20060101); G01R 027/28 (); G01R
035/00 () |
Field of
Search: |
;324/601,612,614,623,638,76.19,76.21,76.22,76.45 ;702/85 |
References Cited
[Referenced By]
U.S. Patent Documents
Other References
Cacciola, J., "Direct Vector Analyzer Measurements of Class B and C
Amplifiers," presented at Anritsu Technical Review Series, Jul.
1999. .
Donecker, B., "Determining the Measurement Accuracy of the HP85 10
Microwave Network Analyzer," RF & Microwave Measurement
Symposium, Oct. 1984, pp. 4-70. .
Feher, K., Telecommunications Measurements, Analysis, and
Instrumentation, Prentice-Hall, month missing 1987, pp. 318-321.
.
Gard, K. G., Gutierrez, H. M., and Steer, M. B., "Characterization
of Spectral Regrowth in Microwave Amplifiers Based on the Nonlinear
Transformation of a Complex Gaussian Process," IEEE Trans. On MTT,
vol. 47, Jul. 1999, pp. 1059-1069. .
Marks, R. B., "A Multiline Method of Network Analyzer Calibration,"
IEEE Trans. On MTT, vol. 39, Jul. 1991, pp. 1205-1215. .
Mazumder, S. R., and van der Puije, P.D., "Two-Signal Method of
Measuring the Large Signal S-Parameters of Transistors," IEEE
Trans. On MTT, vol. 26, Jun. 1978, pp. 417-420..
|
Primary Examiner: Lamarre; Guy J.
Assistant Examiner: Kerveros; James C
Attorney, Agent or Firm: Fliesler Meyer LLP
Parent Case Text
CROSS REFERENCE TO PROVISIONAL APPLICATION
This application claims priority to provisional application SC/Ser.
No. 60/288,305, entitled "Probe Tone S-Parameter Measurements"
filed May 2, 2001.
Claims
What is claimed is:
1. A method for measuring scattering parameters of a device under
test (DUT) comprising the steps of: providing a stochastic
modulated signal as a power tone signal; providing a probe tone
signal with a power level substantially less than the power tone
signal; combining the power tone signal and the probe tone signal
to provide a composite signal; providing the composite signal to
the DUT and as a first input to a receiver, the receiver having a
narrow measurement bandwidth relative to a bandwidth of the power
tone signal; providing a signal from the DUT to a second input of
the receiver; and determining an S-parameter measurement for the
DUT using the first and second signals input to the receiver,
wherein the probe tone signal does not increase power from the DUT
more than 0.2 dB relative to power provided when the power tone
signal is provided to the DUT without the probe tone signal.
2. A method for measuring scattering parameters of a device under
test (DUT) comprising the steps of: providing a stochastic
modulated signal as a power tone signal; providing a probe tone
signal with a power level substantially less than the power tone
signal; combining the power tone signal and the probe tone signal
to provide a composite signal; providing the composite signal to
the DUT and as a first input to a receiver, the receiver having a
narrow measurement bandwidth relative to a bandwidth of the power
tone signal; providing a signal from the DUT to a second input of
the receiver; and determining an S-parameter measurement for the
DUT using the first and second signals input to the receiver,
wherein the probe tone signal has a power level no greater than
13.3 dB below the power level of the composite signal.
3. A method for measuring scattering parameters of a device under
test (DUT) comprising the steps of: providing a stochastic
modulated signal as a power tone signal; providing a probe tone
signal with a power level substantially less than the power tone
signal; combining the power tone signal and the probe tone signal
to provide a composite signal; providing the composite signal to
the DUT and as a first input to a receiver, the receiver having a
narrow measurement bandwidth relative to a bandwidth of the power
tone signal; providing a signal from the DUT to a second input of
the receiver; determining an S-parameter measurement for the DUT
using the first and second signals input to the receiver; and
applying ensemble averaging to the first and second signals input
to the receiver to statistically enable the power tone signal to be
removed from measurement of the composite signals.
4. A method for measuring scattering parameters of the device under
test (DUT) comprising the steps of: providing a stochastic
modulated signal as a power tone signal; providing a probe tone
signal with a power level substantially less than the power tone
signal; combining the power tone signal and the probe tone signal
to provide a composite signal; providing the composite signal to
the DUT and as a first input to a receiver, the receiver having a
narrow measurement bandwidth relative to a bandwidth of the power
tone signal; providing a signal from the DUT to a second input of
the receiver; and determining an S-parameter measurement for the
DUT using the first and second signals input to the receiver,
wherein the power tone signal within the receiver range as averaged
from sample measurements has substantially a zero mean.
Description
BACKGROUND OF THE INVENTION
1. Field of the Invention
The present invention relates to measuring scattering parameters
for a device under test (DUT), such as a power amplifier, which
require that the DUT be operating with a modulated drive signal
before accurate measurements can be made.
2. Description of the Related Art
The measurement of a device behavior under complex actual operating
conditions has become increasingly desirable. In particular, it can
be difficult to accurately measure gain and some reflection
coefficients of a power amplifier operating under a realistic
modulated signal drive. A small signal measurement alone of a power
amplifier is generally incorrect since the DUT will not be biased
correctly. A fully modulated measurement, however, may require very
dedicated equipment, long measurement times for adequate stability,
and special calibration techniques.
It has long been known that it is sometimes advantageous to make
S-parameter measurements in the presence of other signals, as
evidenced in S. R. Mazunder and P. D van der Puije, "Two-signal
Method of Measuring the Large Signal S-parameters of Transistors,"
IEEE Trans. On MTT, vol. 26, June 1978, pp. 417-420, incorporated
herein by reference. Another method for taking measurements in the
presence of other signals is the measurement technique termed the
"Hot S22" technique. A typical test system setup for the Hot S22
technique is shown in FIG. 1. In the Hot S22 technique, the DUT 1
(usually a power amplifier) is driven to its normal operating point
by a power tone signal provided from a power tone signal generator
2 to the input of the DUT 1, while a second smaller probe tone
signal is provided to the output of the DUT 1 from a probe tone
signal generator 3. An isolator 4 is sometimes provided between the
probe tone signal generator 2 and the DUT 1 to keep DUT output
power from affecting the probe tone signal generator 3.
Measurements are made from a coupler 4 connected between the second
signal generator 3 and the output of the DUT 1. The coupler 4
provides a signal to a receiver 6 which downconverts the signal
from the coupler to an intermediate frequency (IF) for measurement.
The probe tone signal from generator 3, as reflected from the
output of the DUT 1, is coupled by the coupler 4 to the receiver 6.
The reflected signal is compared with the signal from the probe
tone signal generator 3 to provide an output reflection coefficient
measurement S22. Typically the signals from both the power tone
signal generator 2 and the probe tone signal generator 3 are both
sinusoids and are offset in frequency by at least several IF
bandwidths to avoid effects on measurement due to interference
between the power tone signal and the probe tone signal in the
receiver, although this may not be necessary with specialized
instrumentation to separate the signals based on phase
behavior.
The Hot S22 measurements technique described are not typical load
pull measurements, since the port impedances remain fixed. Without
measurements for different loading, useful information about output
reflection behavior and stability of the DUT, however, will still
be provided for DUTs operating with 50 ohm loads in a 50 ohm
environment. As such, the conventional Hot S22 measurement
technique described is often used to characterize amplifier
subassemblies rather than amplifiers alone.
The S-parameters of a two port device such as DUT 1 characterize
how the device interacts with signals presented to the various
ports of the device. The measurement for Hot S22 is, of course the
S22 S-parameter. The first number following the S in "S22"
indicates the number of the port the signal is leaving, while the
second number is the port that the signal is being injected into.
S12, therefore, is the signal leaving port 1 relative to the signal
being injected into port 2. The four S-parameters associated with
an exemplary two-port DUT are represented in FIG. 2, where: S11 is
referred to as the "forward reflection" coefficient, which is the
signal leaving port 1 relative to the signal being injected into
port 1; S21 is referred to as the "forward transmission
coefficient, which is the signal leaving port 2 relative to the
signal being injected into port 1; S22 is referred to as the
"reverse reflection" coefficient, which is the signal leaving port
2 relative to the signal being injected into port 2; and S12 is
referred to as the "reverse transmission" coefficient, which is the
signal leaving port 1 relative to the signal being injected into
port 2.
An important point about the Hot S22 measurements is that they can
be made with a calibrated vector network analyzer (VNA) 7, as
illustrated in FIG. 3. The VNA is calibrated to remove
uncertainties and provide traceability. With straight power
measurements using a scalar test setup, such as with the signal
generator 2, coupler 5 and receiver 6, rather than the VNA 7 used
to make measurements, uncertainties will not be removed.
In a modem measurement environment with a variety of wide
modulation formats and highly optimized power amplifiers, the need
for performance data increases. For instance, other S-parameter
measurements than S22 might be desirable, since all results under
large signal sinusoidal drive or some other type of drive may not
be the same. As an example, forward parameters may be affected as
well as reverse parameters. Furthermore, a modulated power drive
other than large sinusoidal signals might be desirable since
average compression behavior of components in the receiver varies
as a function of the statistical distribution of an input signal.
More common modulation signals such as code division multiple
access (CDMA) and wideband CDMA (WCDMA) may be desirable.
SUMMARY
In accordance with the present invention, a method and apparatus is
provided to test (DUTs), such as power amplifiers, which require a
modulated drive signal for proper biasing, in combination with a
probe tone test signal, wherein both the modulated and probe tone
signals operate in the same frequency range.
In accordance with the present invention, a stochastic power tone
modulated drive signal, such as a CDMA or WCDMA signal, is provided
in combination with a low power probe tone signal. The probe tone
signal has a power significantly less than the modulated power tone
signal to avoid further compression of the DUT, and so that a
statistical average of a combined probe tone signal and power tone
signal does not change significantly from the average of the power
tone signal alone. A narrowband receiver having an IF measurement
bandwidth significantly less than the overall power tone signal
bandwidth enables separation of the modulated and probe tone
signals. Test signals are provided using a VNA, and calibration is
used to further improve the measurement accuracy. For modulated
power tone signals with a significant power level in the frequency
range of the probe tone signal, ensemble averaging of the modulated
signal over a wide signal frequency range is used to enable
separation of the probe tone signal for measurement.
BRIEF DESCRIPTION OF THE DRAWINGS
Further details of the present invention are explained with the
help of the attached drawings in which:
FIG. 1 a conventional test setup for hot S22 measurements of a
DUT;
FIG. 2 illustrates S-parameters for a two port DUT;
FIG. 3 illustrates measurement of parameters of a two port DUT
using a two port VNA;
FIG. 4 shows test system configuration for making probe tone
S-parameter measurements using the method of the present
invention;
FIG. 5 plots a composite signal power distribution vs. frequency
probability with (dotted) and without (solid) a probe tone signal
component;
FIG. 6 shows VNA measurements of S21 for a thru line for both a
probe tone signal composite signal (probe tone and power tone);
FIG. 7 shows VNA measurements of S21 for a small signal amplifier
for both a probe tone signal and a composite signal;
FIG. 8 shows S21 measurements of a small signal amplifier similar
to FIG. 7, but with power levels increased where the DUT is in
compression;
FIG. 9 shows S21 measurement from an amplifier as probe tone power
is increased and the modulated power signal remains fixed;
FIG. 10 shows a comparison between measurements made using the
improved probe tone technique of the present invention and a more
conventional scalar approach;
FIG. 11 shows conventional vector measurement techniques made with
and without the DUT in compression;
FIG. 12 shows a comparison between scalar and vector
measurements;
FIG. 13 illustrates the effects of an impedance mismatch between
the test signal port and the DUT;
FIG. 14 shows a plot of gain for a power amplifier with and without
modulation power to illustrate profiling; and
FIG. 15 illustrates the group delay and magnitude measurements for
the same test conditions as in FIG. 14.
DETAILED DESCRIPTION
A. Test Method
Without specialized measurement instrumentation, previous
measurement techniques required that the multiple signals provided
to the DUT be separate in frequency space. For more stochastic
natured modem modulation power tone drive signals, such as CDMA and
WCDMA, measurements can be made with the separation restriction
removed if:
a. The probe tone is sufficiently small that the statistics of the
composite signal including the probe tone and power tone signals,
are not significantly different from the modulated power tone
signal alone. The probe signal must, or course, be large enough
that the signal-to-noise (S/N) ratio in the measurement device
receiver is adequate to distinguish the probe tone signal from
noise; and
b. The use of a narrowband receiver along with the ensemble
averaging allows a measurement of the probe tone while excluding
the modulated signal from the measurement receiver.
To make these improved probe tone measurements of value, vector
network analyzer (VNA) calibrations are used to make the
measurements more accurate than conventional probe tone measurement
techniques made with scalar devices rather than a VNA. Further,
with a receiver having a large dynamic range, power levels
manipulated correctly, and a standard VNA calibration, a
significant improvement in measurements can be made. Indeed dynamic
range is somewhat easier to achieve with a narrowband receiver in
that large amounts of the composite signal power distributed over a
wide range are automatically excluded from the receive path.
The improved probe tone measurement techniques will be described
generally herein since the technique has broad applicability.
Examples using the improved measurement technique will largely
focus on a special measurement case: two port devices where the
composite stimulus is composed of a continuous-wave (CW) probe tone
signal plus some modulated power tone signal of significant
bandwidth relative to that of the measurement system.
The improved probe tone S-parameter measurement is characterized by
the following:
1. Any number of ports of the DUT may be driven by a signal and
there may be any number of signals driving a given port.
2. One of these signals, designated the probe tone signal, will be
the basis of a direct S-parameter measurement. It will generally be
CW.
3. The probe tone signal power will be small relative to the total
power level at the port being measured (port in question). Small is
defined as the probe tone having only a quasi-linear impact on DUT
performance and a negligible effect (relative to other
uncertainties) on the total signal statistically at the port in
question.
4. If any signals being combined with the probe tone are also CW,
they may not be at the same frequency as the probe tone and
generally must be separated by at least a bandwidth of the test
measurement receiver. If the other power tone signals are modulated
(and sufficiently stochastic), they may coexist in the same
frequency range.
5. The composite power delivered to the receiver must not alter its
linearity state (although the DUT state can, or course, be a
function of composite power).
How the improved measurement is made is next defined. Generally,
the receiver channels will get a combination of the probe signal
and some other power tone signals. The general S-parameter is
defined as an output wave (b.sub.n, transmitted or reflected)
ratioed against the incident wave (a.sub.m, a.k.a. the reference).
Historically, these waves have been assumed to be measured at a
single frequency, but with receiver bandwidth affecting this
measurement a new concept will be generalized by defining an
S-parameter term S.sub.mn as follows: ##EQU1##
Where g represents the complete receiver behavior which is defined
by a convolution with the change in frequency .DELTA.. The extent
of .DELTA. is limited by a narrow IF filter. f.sub.p is the probe
tone center frequency (to which the receiver is tuned), and state 1
describes the power state that the device/system is operating under
(the dependence on power state is significant). The waves b.sub.n
and a.sub.m represent the composite test signal (b.sub.n received
from the DUT at port n) and the composite reference signal (a.sub.m
generated by the power tone source at port m). If one can assume
that b and a do not change over the frequency span defined by
.DELTA. (which can practically be as small as a few Hz), then the
definition reduces simply to the familiar: ##EQU2##
If the combined signal includes the probe tone signal plus
additional CW signals, then the measurement process is complete
since the IF filter will presumably remove all of the other
signals. If the combined signal includes a modulated signal that
has no significant energy at the frequency of the probe signal,
then the measurement process is complete for the same reason.
Alternatively, the case of the modulated power tone signal having
significant energy at the probe tone frequency with the signal
a.sub.m changing rapidly over the scale of .DELTA. due to a
modulated power tone signal (a.sub.m is a sum of the probe tone,
p.sub.m, and a modulated power tone signal, d.sub.m). In this case,
some additional averaging must be considered to make the
measurement tractable. Presumably .DELTA. can be made narrower
(often digitally) using a mathematical process termed averaging
beyond a certain point (say 1 Hz wide .DELTA. as an example,
although .DELTA. can be other values as in subsequent examples
presented herein). This averaging may be sweep-to-sweep or
point-to-point as long as the sampling process is statistically
independent of the modulated power tone signal d.sub.m. If d.sub.m
has zero mean, it is independent of the sampling process, and hence
can be removed by sufficient ensemble averaging, then
Note that this does not say d.sub.m has zero power, just that the
complex valued function has zero mean in the frequency range of
interest. The `approximately equal` is used to denote a trade-off
between resulting data jitter and the amount of ensemble averaging
performed; it converges to an equality in the limit of high
averaging. Note that standard IS-95 CDMA and 3GPP (among others)
WCDMA waveforms fit this criteria; it is expected that many other
standards do as well. If a variant of equation 3 holds for b.sub.n
(a and b will generally be of the same type), and we define b.sub.n
as the sum of a received probe tone portion, q.sub.n, and received
power tone portion, e.sub.n, then: ##EQU3##
Since p and q, probe tone values, are defined to be small enough to
not affect DUT state (quasi-linearity assumption), then equation 4
still represents a small-signal quantity although it is restricted
to a defined operating state of the DUT. This allows full
S-parameter analysis to proceed and is a key point. Such analysis
is conceptually more difficult with the more traditional large
signal S-parameters (in which the probe tone is the only signal and
is large) since any quasi-linearity claims would be difficult to
justify.
The practical uncertainties introduced by this process are
primarily limited to the residual quasi-random nature of a and b
after a finite amount of ensemble averaging. Preferrably,
sufficient ensemble averaging is performed relative to the base
measurement uncertainty as described in B. Donecker, "Determining
the measurement accuracy of the HP8510 Microwave Network Analyzer,"
R F & Microwave Measurement Symposium, October 1984, pp. 4-71),
incorporated herein by reference. Practical uncertainties may be
elevated compared to traditional measurements if the probe tone
powers are low enough to meet the statistical insignificance
criteria.
For this improved measurement technique, standard VNA 12-term
calibrations can be performed at the probe tone center frequency
f.sub.p and the actual calibration steps can be performed with only
the probe tone present. As such, the standard calibration-related
uncertainty terms (corrected port match, directivity, and tracking)
will be unaffected. This measurement is not specific to a given
calibration technique (Short-Open-Load-Thru, variations on
Thru-Reflect-Line, etc.).
When applying the calibration, the coefficients still apply as long
as equation 4 holds and one assumes the summed signal does not
generate significant non-linearities in the receiver. Subject to
receiver compression levels and sufficient ensemble averaging,
there should be no increase in calibrated uncertanties over that in
the base measurement. As stated previously, this compression risk
is perhaps not as great as might be imagined since the bulk of the
incident power is outside the bandwidth of the receiver function
and hence will not contribute as much to compression as in a
wideband receiver. Note that it is the S/N established by the probe
tone in the receiver that is the relevant component to be used in
the uncertainty computations. The key point is that since the
standard calibration procedures, the uncertainty calculations, and
the small signal measurement characteristics all hold, measurement
integrity and traceability should be maintained.
B. Test System Description
FIG. 4 shows test system configuration for making measurements in
accordance with the improved probe tone S-parameter measurement
technique of the present invention. The system includes a VNA 7
with some internal components shown to illustrate the preferred
components for use in making measurements according to the present
invention. An example of a VNA which includes the components shown
is the Anritsu MS4623CVNA, manufactured by Anritsu company of
Morgan Hill, Calif. The VNA 7 can be calibrated using a calibration
kit, such as the Anritsu 3750LF. Although these specific
instruments are shown and described, other VNAs and calibration
kits might be used.
Attached to an external port of the VNA 7 is a signal source 2 for
generating the power tone signal. Also, connected to the external
test ports, Port 1 and Port 2, is a two port DUT which requires a
modulated drive signal for accurate measurements to be made.
Internal to the VNA is a signal source 3 for generating a
measurement test signal, and which will be used to generate the
probe tone signal required for measurements in accordance with the
present invention. The probe tone signal source 3 is connected by a
switch to connection units 10 and 11 to provide the probe tone
signal to one of two signal connection terminals providing signals
labeled RF1 and RF2. An external signal can be applied to the VNA 7
to a connection unit 9 to create a signal RF3. Although shown as an
external connection, a separate signal source could be included in
the VNA 7 and be connected to unit 9 to provide the RF3 signal.
A receiver 6 is connected to receive signals incident and reflected
from the test ports Port 1 and Port 2 to enable S-parameters to be
calculated for the DUT 1 connected between the test ports. Coupler
13 couples the RF2 signal incident to Port 2 to the receiver port
a2. Coupler 14 couples a reflected signal from port 2 to the
receiver port b2. The RF1 and RF3 signals are combined in power
combiner 15 to provide a test signal to Port 1. Coupler 17 couples
the RF1 and/or RF3 signal incident to Port 1 from combiner 15 to
test port al of the receiver 6. Coupler 18 couples a reflected
signal from Port 1 to test port b1 of the receiver 6.
A directional coupler 20 couples the power tone signal from the
external power tone signal source 2 to be combined with the probe
tone RF1 signal. The diagram in FIG. 4 uses a directional coupler
20, but as an alternative, the power tone signal generator 2 could
be connected to the connection unit 9 to form the signal RF3 and be
combined in the Wilkinson-class combiner 15 with the probe tone
signal RF1 from generator 3 with relatively low insertion loss and
some isolation. The choice of a coupler or Wilkinson-class combiner
for combining the probe tone and power tone signals will be
dictated by the amount of reverse power from the power tone signal
source 2 into the probe tone signal source 3 that the VNA can
tolerate, as well as by the drive requirements of the DUT. If the
isolation of the coupling device 20 is poor, enough energy may be
returned to the VNA port to disturb the function of its automatic
leveling circuitry. This is balanced against the amount of
modulated power required to bias the DUT 1 into its desired
operating state.
Note in FIG. 4 that the combining of the power tone signal with the
probe tone signal accomplished by coupler 20 is done prior to the
reference couplers 17 and 18. This implies that the modulated
signal will be present in both numerator and denominator (a.sub.1
and b.sub.1) of an S-parameter calculation. An important advantage
of this is that the stochastic nature of the data will at least
partially cancel (one of the general benefits of the ratioed
measurement although it is usually used to reduce the effects of
thermal noise, not noise-like modulation), thus reducing the amount
of averaging required in the method of the present invention. The
degree of cancellation will be dependent on the coherence between
the reference and test channels and, hence, on the path length
difference in the VNA 7 between these two channels (receiver
channel a.sub.1 vs. b.sub.1). In most configurations, this path
length difference is less than 10 ns. With current bit times of at
least 100 ns (for 10 Mbps) in personal communication systems, one
can normally expect a high degree of coherence and hence
cancellation. If higher bit rates are required and coherence falls,
equalizing line lengths can be used to decrease the path length
difference.
The IF bandwidth (IFBW) of the receiver 6 will normally be set to a
low value although, depending on receiver architecture, it can be
widened to improve speed. The trade-off for the increased speed is
that the amount of required ensemble averaging for the present
invention will then increase. Preferrably, the sampling and
averaging process of the present invention must be statistically
independent from the modulated signal. Normally this is not an
issue when uncommon clocks are used to drive the power tone
generator 2 and probe tone generator 3, but it can interfere with
the measurement in unusual cases. Smoothing, or boxcar averaging,
can be an acceptable substitute for ensemble averaging if the macro
DUT response is not changing in frequency over the range of
interest. If it is changing, then the frequency resolution will
decrease.
In order to accommodate power requirements for the probe tone
signal generator 3, the probe tone source power of approximately
-15 dBm may be employed for a target DUT that requires 0 dBm to be
biased into the desired state. Assuming classical coupling
coefficients of about 20 dB, one can compute an uncertainty of
about 0.03-0.05 dB for .vertline.S21.vertline. above about -40 dB
(ignoring compression). Since the target DUT is most likely a power
amplifier, this constraint is of no relevance. The main effect of
lowering the probe tone power is to bring up the
.vertline.S21.vertline. level at which uncertainty starts to
degrade.
The next issue is setting the power level for the probe tone signal
relative to the power tone signal. A critical point is that the
behavior of the power tone signal in the presence of the probe tone
signal must not change in the sense of S-parameters, which are
fairly macro-level measurements of a DUTs performance (the criteria
would be much tighter if, for example, the measurement was a bit
error rate or error vector magnitude as opposed to S-parameters).
The criteria for S-parameter measurements can be described as
follows:
1. The average power resulting from application of the probe tone
signal to the modulated signal should not substantively increase.
This is important so that the DUT does not experience compression,
or does not experience further compression if compression in the
current bias condition is occurring. There are many possible
thresholds for composite power level change, but a 0.2 dB maximum
is preferable since it is primarily S-parameters that are measured
with the present invention and not more hyper-sensitive parameters.
This 0.2 dB level is also smaller than the typical measurement
uncertainty of average power.
2. The composite statistics of the waveform with the probe tone
added further should not substantively change. Again the intent
here is to avoid altering the compression behavior of the DUT.
Since at a given frequency a constant power is added, the
statistics themselves are not changing; just the mean is shifting
as illustrated in FIG. 5. FIG. 5 plots the composite signal power
distribution vs. frequency probability with (dotted) and without
(solid) the probe tone signal component. Note that the probe tone
signal does not alter the overall statistics, except it moves the
mean at a given frequency. Thus the peak power and the amount of
time the device spends above a certain level is not changing except
that level is shifted to a higher power. For this argument, the
conditions above are resorted to--that if this shift is small
enough, it will not affect the parameters being analyzed. In an
older measurement system, one may want to make sure that the peak
to average ratio does not change substantively, but more modern
components assure this is the case. It is easy to show that if the
average power has changed less than x dB for a probe tone of
P.sub..DELTA., then the peak to average ratio will have changed
less than x dB as well. Let P.sub.a be the initial average power,
P.sub..DELTA. be the probe tone power, .epsilon.(>1) be the
allowed ratio of new average power to old average power. By
definition ##EQU4##
Let C be the old peak to average ratio (crest factor), then the new
crest factor is ##EQU5##
Since C>1 and P.sub..DELTA. >0, this ratio will always be
smaller than C. Thus R, the ratio of the old to the new crest
factor (to keep R>1 without loss of generality), is given by
##EQU6##
The inequality approaches an equality as C and/or P.sub.a gets
large. Thus if one meets the average power criteria, one
automatically meets that same criteria for crest factor. For a 0.2
dB maximum variation with the probe signal added, this requires
that the probe tone power level be at least 13.3 dB below the
starting average power. If other levels than 0.2 dB maximum are
desired, the probe tone power level will likewise change from 13.3
dB. For signal to noise in the receiver and to minimize measurement
time, it is generally desirable to get the probe tone signal power
relatively close to this level.
C. Test Measurement Examples
Example test measurements will next be discussed, where the
measurements use the test setup of in FIG. 4. The probe tone signal
used will be a sinusoid generated internally by signal source 3 of
the VNA 7, while the modulated signal will be an IS-95 CDMA signal
(chip rate 1.2288 Mcps, 9 forward channels). The modulated signal
is injected from power tone source 2 using the coupler 20. The
power levels vary in the different examples but a 15 dB offset
between modulated power tone signal and probe tone signal power is
maintained (with the exception of one example to show the effect of
high probe powers). Receiver linearity is discussed only in the
first example but it was verified in all examples.
In the first test measurement shown in FIG. 6, an
.vertline.S21.vertline. measurement of a thru line is provided with
the probe tone and with an IS-95 modulated power tone signal with a
center frequency at 1800 MHz. The bandwidth of the modulating
signal is about 1.25 MHz, so a sweep range from 1799 MHz to 1801
MHz is shown. Two test measurements are shown, one with the
composite (probe tone and power tone) signal, and the other with
the probe tone signal separated out. The jitter on the data is a
little higher than what one would normally expect since the
resulting reference power is quite low. This could be improved by
adding reference amplification or increasing averaging (although
the latter will have an attached speed penalty). Since the amount
of jitter is roughly equivalent with and without the modulating
signal, one can conclude that the receiver is effectively filtering
the modulated energy, even with the composite signal (where the
probe tone is still embedded), illustrating that the IS-95 easily
meets the conditions of being sufficiently stochastic. Since the
mean did not move, we can be reasonably sure that the receiver is
not being compressed.
In another example shown in FIG. 7, .vertline.S21.vertline.
measurements are again shown, but this time with the DUT being a
small signal amplifier. As in FIG. 6, in FIG. 7 one measurement is
made with the composite signal, and another with the probe tone
signal separated out. Further, VNA calibration is used for both
measurements. The composite power level required for the small
signal amplifier is low, so the same signal grouping and powers
used in FIG. 6 are used in FIG. 7 so that the DUT amplifier will
not be heavily compressed. As one might expect, the traces nearly
overlay. The slight suppression in the mean with the modulated
signal suggests the onset of compression, but by an amount less
than 0.1 dB. Note that for many larger signal power amplifiers,
this display would be quite different since the gain is a function
of drive level over a large range. This particular DUT has flat
gain up until its compression behavior starts at about a -3 dBm
input.
In another example shown in FIG. 8, .vertline.S21.vertline.
measurements are again shown with the DUT being a small signal
amplifier, as in FIG. 7, but with increased input signal power so
that significant DUT compression is occurring. The same power delta
of 15 dB was maintained but the modulated signal power is
increased. The lower trace in FIG. 8 represents the gain of the
amplifier with this level of modulated drive signal. It is quire
clear that .vertline.S21.vertline., as expressed by the probe tone
measurement, is suppressed upon application of the higher modulated
signal. While this should not be surprising it is a key point of
this measurement.
The 0.4 dB of compression from the small power signal behavior of
FIG. 7 is quite clear in FIG. 8, and the level of jitter again does
not differ much with and without the modulating signal. It is
interesting to note that the level of compression is the same
whether the modulated energy and probe tone overlap in frequency or
not (recall the IS-95 signal has about 1.25 MHz of bandwidth in
this sweep from 1799 MHz through 1801 MHz). Partially this is a
result of the use of a wideband amplifier DUT for this test (4 GHz)
and the fact the modulated power signal is globally changing the
bias state of the device.
FIG. 9 illustrates the effects of amplifier compression on the
measurements in accordance with the present invention to examine
the effect of applying probe tone power in excess of what is
recommended. FIG. 9 shows .vertline.S21.vertline. of an amplifier
in about 0.2 dB compression from the applied IS-95 signal as the
probe tone power is increased and the modulated power signal
remains fixed. The measurement changes composite power levels from
-5.0 dBm to 15 dBm while measuring S21 power. The input probe tone
power varies here from 25 dB below the modulated power (-5 on the
x-axis) to about 5 dB below the modulated power (+15 on the
x-axis). The recommended maximum operating point of 13.3 dBm occurs
at the center of the x-axis between the -5 dBm and 15 dBm points.
As one can see, the degradation of data is not obvious until well
above this threshold. Note that the error is on the order of 0.15
db at the worst case point of 15 dBm on the x-axis. The worst case
shown in FIG. 9 is demanding since the DUT has entered compression
and helps justify the power levels used.
FIG. 10 illustrates a comparison of measurements made using the
improved probe tone technique of the present invention, with a more
conventional scalar approach of modulated gain measurements, as
illustrated in FIG. 1. For the scalar technique, measurements were
made using a spectrum analyzer with an integrating function. The
spectrum analyzer was used to measure input and output power (or
Gain) (integrating over 2 MHz with a 30 kHz bandwidth) of an
amplifier as a function of drive signal power level (Pin) level.
The same measurement was performed using the probe signal approach
of the present invention while maintaining a 15 dB power delta
between the probe and power tone signals, and sweeping the
composite power levels. The same modulated IS-95 signal was used
for both measurements but, of course, the probe tone was not
present for the integrated spectrum analyzer measurement. The
results span the range of small signal to heavily compressed and
hence should represent a reasonable cross section of DUT behaviors.
The results show good agreement between the conventional scalar
measurement, and the vector measurement of the present
invention.
Having established a certain level of agreement with scalar
results, conventional vector measurement techniques are compared.
The simplest of the class of vector probe tone measurements is Hot
S22 in which a full 1 port calibration is used, as illustrated in
FIGS. 1 and 2. In this measurement, unlike measurements with the
present invention, an offset sinusoid is applied to the DUT input
and a probe tone is bounced off the DUT output. Further, as in
conventional measurements, there is a frequency offset between the
probe and power town signal of several receiver bandwidths (300 kHz
>>1 kHz IFBW for FIG. 11). The vector calibrations provide
correction for port matches, directivity and tracking to reduce the
measurement uncertainty in this case to about 0.2 dB, considerably
better than that of a scalar measurement (uncertainty >0.5 dB
typically) and more stable.
An example measurement is shown in FIG. 11 at small signal and
modest compression drive levels. The difference between the
measurements is quite noticeable. The important point about this
measurement is that it uses full vector correction of the
reflection measurement. The uncertainties are much lower (<0.1
dB) and more stable than what one could obtain with a scalar
measurement.
Comparison of vector and scalar measurements are shown in FIG. 12
to illustrate the benefits of vector measurements. On a gain
measurement, for example, the obvious benefit of vector
measurements is correcting for raw port matches. In other
measurements, directivity and vector tracking errors can also be
effectively corrected which will not happen in a scalar
measurement. This issue becomes more important as the DUT
approaches compression since its impedance levels will be changing.
In FIG. 12, the differences are under 0.1 dB at low power levels
(Pin), but at higher power levels, the differences increase, such
as to 0.3 dB at a 7 dBm power level. The difference at up to about
+4 dBm input power, being under 0.1 dB is likely due to source
match-S11 interactions possibly together with a better tracking
term characterization. The difference of 0.3 dB at 7 dBm, however,
is well beyond the measurement uncertainty of about 0.07 dB (for
the vector corrected data) and can be considered significant from a
measurement point of view.
FIG. 13 illustrates the effects of an impedance mismatch between
the test signal ports and the DUT. Effects are shown for an input
mismatch (S11) and an output mismatch (S22). As shown, the
measurement deviation increases as a function of power level. These
parameters were measured using a full 12 term probe tone
calibration. While the input match is actually improving with
drive, the output match is definitely not. The effects of a
mismatch between measurement system and DUT will increase at these
higher power levels and lead to increased scalar measurement
errors.
In a final example, measurements are made using a 900 MHz handset
variety power amplifier as a DUT. This amplifier exhibits gain
expansion for a wide range of input powers, a gain flattening near
the desired operating point and finally gain compression. As such,
a measurement like that in FIG. 7 requires careful interpretation
since the DUT gain will be lower with probe tone alone compared to
the composite signal. The probe tone measurement is still quite
valid but the DUT operating point will be strictly established by
the modulated signal and must be carefully considered.
FIG. 14 shows a plot of gain with and without modulation power
present (again IS-95 with a 15 dB power delta). In both cases, the
amplifier is being operated somewhat below its normal operating
point (13 dBm output power vs. 27 dBm) so that it is still in gain
expansion. The point of this example is that it shows some of the
power of S-parameter profiling possible with this measurement
technique. Since the modulated power is fixed in frequency while
the probe tone sweeps, it is possible to see what behavior an
interfering signal would experience at a different frequency. In
this case, one can see two gain drop outs at 15 MHz away from the
carrier frequency 900 MHz (note that the frequency scale is 10
MHz/div and the modulated power is only .apprxeq.1.25 MHz wide).
This type of S-parameter profiling could conceivably be useful in
studying and/or tailoring out-of-band and band-edge responses.
While FIG. 14 illustrates the difference with and without modulated
power applied, FIG. 15 illustrates the group delay
(-d.phi./d.omega. interpreted as deviation from linear phase) as
well as magnitude of the same amplifier under slightly higher drive
levels (+17 dBm output power). The group delay deviation (of about
20 ns) can be seen to coincide with the magnitude variation. This
type of data is included to illustrate additional profiling power
as well as the group delay using the measurement technique of the
present invention.
Although the present invention has been described above with
particularity, this was merely to teach one of ordinary skill in
the art how to make and use the invention. Many additional
modifications will fall within the scope of the invention, as that
scope is defined by the claims which follow.
* * * * *