U.S. patent number 6,882,680 [Application Number 09/591,196] was granted by the patent office on 2005-04-19 for quadrature phase modulation receiver for spread spectrum communications system.
This patent grant is currently assigned to Umbrella Capital, LLC. Invention is credited to Vladislav A. Oleynik.
United States Patent |
6,882,680 |
Oleynik |
April 19, 2005 |
Quadrature phase modulation receiver for spread spectrum
communications system
Abstract
A quadrature phase modulation receiver for a spread spectrum
communications system includes a mixer for mixing a received spread
spectrum signal with a heterodyne signal to convert the frequency
of the received signal to an intermediate frequency. A regulated
oscillators module is coupled to the receiver for producing the
heterodyne signal and signal equal to the intermediate frequency
signal. A frequency multiplier circuit is coupled to the mixer for
receiving the intermediate frequency signal and multiplying the
frequency of the signal by a predetermined multiplication factor.
An oscillator control signal is produced based on the frequency
multiplied signal to maintain synchronization between the receiver
and the transmitter.
Inventors: |
Oleynik; Vladislav A. (Chapel
Hill, NC) |
Assignee: |
Umbrella Capital, LLC (Miami,
FL)
|
Family
ID: |
34435249 |
Appl.
No.: |
09/591,196 |
Filed: |
June 9, 2000 |
Current U.S.
Class: |
375/147; 375/141;
375/267; 375/329; 375/376; 375/E1.002 |
Current CPC
Class: |
H04B
1/707 (20130101); H04L 27/227 (20130101); H04L
2027/0016 (20130101); H04L 2027/0048 (20130101); H04L
2027/0055 (20130101) |
Current International
Class: |
H04B
1/69 (20060101); H04B 001/69 () |
Field of
Search: |
;375/141,147,267,329,376
;327/47 ;332/103 ;329/315 ;455/86 |
References Cited
[Referenced By]
U.S. Patent Documents
Primary Examiner: Tse; Young T.
Assistant Examiner: Chang; Edith
Attorney, Agent or Firm: Jenkins, Wilson & Taylor,
P.A.
Claims
What is claimed is:
1. A quadrature phase modulation receiver for a spread spectrum
communications system, the receiver comprising: (a) a mixer for
mixing a received spread spectrum signal with a heterodyne signal
to convert the frequency of the received spread spectrum signal to
an intermediate frequency and thereby produce a received signal
having the intermediate frequency, the received spread spectrum
signal including a carrier signal modulated by a data signal and
being spread by a spreading code; (b) a regulated oscillators
module coupled to the mixer for producing the heterodyne signal and
an intermediate frequency signal; (c) a frequency multiplier
coupled to the mixer for receiving the received signal having the
intermediate frequency and multiplying the frequency of the
received signal having the intermediate frequency by a fixed
multiplication factor to produce a frequency multiplied signal,
wherein the frequency-multiplied signal has a phase that does not
depend on a phase of the data signal, wherein the frequency
multiplied signal produced by the frequency multiplier is
characterized by the following expression:
where .omega. is the frequency of the received signal having the
intermediate frequency and .phi. is the initial phase of the
received signal having the intermediate frequency; and (d) means
for producing an oscillator control signal based on the frequency
multiplied signal output from the frequency multiplier, wherein the
regulated oscillators module produces the intermediate frequency
signal based on the oscillator control signal.
2. The quadrature phase modulation receiver of claim 1 comprising:
(e) a phase discriminator for receiving the signal output from the
mixer and the intermediate frequency signal and producing a signal
indicative of transmitted data and a spreading code; and (f) a
demodulator for receiving the signal output from the phase
discriminator and removing the spreading code.
3. The quadrature phase modulation receiver of claim 2 wherein the
demodulator comprises a frequency hopping spread spectrum
demodulator.
4. The quadrature phase modulation receiver of claims 2 wherein the
demodulator comprises a direct sequence spread spectrum
demodulator.
5. The quadrature phase modulation receiver of claim 1 wherein the
regulated oscillators module comprises: a heterodyne signal
oscillator for producing the heterodyne signal; a voltage
controlled oscillator (VCO) for producing an output signal having a
predetermined relationship with the intermediate frequency; and a
frequency synthesizer for receiving the output signal from the VCO
and for producing the intermediate frequency signal.
6. The quadrature phase modulation receiver of claim 5 wherein the
voltage controlled oscillator is adapted to receive the oscillator
control signal and produce the output signal based on the
oscillator control signal.
7. A quadrature phase modulation receiver for a spread spectrum
communications system, the receiver comprising: (a) a mixer for
mixing a received spread spectrum signal with a heterodyne signal
to convert the frequency of the received spread spectrum signal to
an intermediate frequency and thereby produce a received signal
having the intermediate frequency; (b) a regulated oscillators
module coupled to the mixer for producing the heterodyne signal and
an intermediate frequency signal; (c) a frequency multiplier
coupled to the mixer for receiving the received signal having the
intermediate frequency and multiplying the frequency of the
received signal having the intermediate frequency by a
predetermined multiplication factor to produce a frequency
multiplied signal; and (d) means for producing an oscillator
control signal based on the frequency multiplied signal output from
the frequency multiplier, wherein the regulated oscillators module
produces the intermediate frequency signal based on the oscillator
control signal, wherein the means for producing the oscillator
control signal comprises: (i) a phase shifter coupled to the
frequency multiplier for receiving the frequency multiplied signal
and shifting the phase of the frequency multiplied signal by a
predetermined amount to produce an output signal; (ii) a comparison
signal formation circuit for receiving the intermediate frequency
signal output from the regulated oscillators module and for
producing an output signal having a predetermined relationship with
the intermediate frequency signal; and (iii) a frequency and phase
discriminator for receiving the output signals from the phase
shifter and the comparison signal formation circuit and for
producing the oscillator control signal based on the output signals
from the phase shifter and the comparison signal formation
circuit.
8. The quadrature phase modulation receiver of claim 7 wherein the
comparison signal formation circuit comprises first and second
squaring circuits and a multiplier circuit connected in series for
receiving and processing the intermediate frequency signal output
from the regulated oscillators module.
9. The quadrature phase modulation receiver of claim 8 wherein: the
first squaring circuit comprise a first voltage multiplier-having
first and second inputs connected to each other and having a first
output; the second squaring circuit comprises a second voltage
multiplier having third and fourth inputs connected to each other
and to the first output and having a second output; and the
multiplier circuit comprises a three-input multiplier having a
first input coupled to the first output, a second input coupled to
the second output, and a third input coupled to the regulated
oscillators module.
10. A quadrature phase modulation receiver for a spread spectrum
communications system, the receiver comprising: (a) a mixer for
mixing a received spread spectrum signal with a heterodyne signal
to convert the frequency of the received spread spectrum signal to
an intermediate frequency and thereby produce a received signal
having the intermediate frequency; (b) a regulated oscillators
module coupled to the mixer for producing the heterodyne signal and
an intermediate frequency signal; (c) a frequency multiplier
coupled to the mixer for receiving the received signal having the
intermediate frequency and multiplying the frequency of received
signal having the intermediate frequency by a predetermined
multiplication factor to produce a frequency multiplied signal; and
(d) means for producing an oscillator control signal based on the
frequency multiplied signal output from the frequency multiplier,
wherein the regulated oscillators module produces the intermediate
frequency signal based on the oscillator control signal, wherein
the received signal is a quadrature phase modulated signal and the
frequency multiplier is a 4.times. frequency multiplier, and
wherein the 4.times. frequency multiplier comprises: (i) a first
multiplier for squaring the received signal output from the mixer
to produce a squared signal; (ii) a first DC blocking capacitor for
removing DC offset components from the squared signal; (iii) a
second multiplier for squaring the squared signal to produce a
third output signal having a frequency that is four times the
frequency of the intermediate frequency signal; and (iv) a second
DC blocking capacitor coupled to the second multiplier for removing
DC components from the third output signal to produce the frequency
multiplied signal.
11. A method for maintaining synchronization between a quadrature
phase modulation spread spectrum transmitter and a quadrature phase
modulation spread spectrum receiver, the method comprising: at the
quadrature phase modulation spread spectrum receiver: (a) receiving
a quadrature phase modulated spread spectrum signal; (b) mixing the
quadrature phase modulated spread spectrum signal with a heterodyne
signal to produce an intermediate frequency signal; (c) removing
the influence of data changes in the quadrature phase modulated
spread spectrum signal from the intermediate frequency signal to
produce an oscillator control signal, wherein removing the
influence of data changes in the quadrature phase modulated spread
spectrum signal from the intermediate frequency signal includes
multiplying a frequency of the intermediate frequency signal by a
fixed multiplication factor and wherein removing the influence of
data changes in the quadrature phase modulated spread spectrum
signal from the intermediate frequency signal includes producing
the intermediate frequency signal characterized by the following
equation:
where .omega. is the frequency of the intermediate frequency signal
and .phi. is the initial phase of the intermediate frequency
signal; (d) generating a synchronization signal based on the
oscillator control signal; and (e) demodulating the quadrature
phase modulated spread spectrum signal using the synchronization
signal.
12. The method of claim 11 wherein demodulating the quadrature
phase modulated spread spectrum signal includes outputting the
quadrature phase modulated spread spectrum signal to a frequency
hopping spread spectrum demodulator.
13. The method of claim 11 wherein demodulating the quadrature
phase modulated spread spectrum signal includes outputting the
quadrature phase modulated spread spectrum signal to a direct
sequence spread spectrum demodulator.
14. A method for maintaining synchronization between a quadrature
phase modulation spread spectrum transmitter and a quadrature phase
modulation spread spectrum receiver, the method comprising: at the
quadrature phase modulation spread spectrum receiver: (a) receiving
a quadrature phase modulated spread spectrum signal; (b) mixing the
quadrature phase modulated spread spectrum signal with a heterodyne
signal to produce an intermediate frequency signal; (c) removing
the influence of data changes in the quadrature phase modulated
spread spectrum signal from the intermediate frequency signal to
produce an oscillator control signal, wherein removing influence of
data changes in the quadrature phase modulated spread spectrum
signal from the intermediate frequency signal includes multiplying
the frequency of the intermediate frequency signal by a
predetermined multiplication factor, wherein multiplying the
frequency of the intermediate frequency signal by the predetermined
multiplication factor includes multiplying the intermediate
frequency signal by a factor of four to produce a frequency
multiplied signal, wherein producing the oscillator control signal
includes: (i) shifting the phase of the frequency multiplied signal
by a predetermined amount to produce a phase-shifted signal; and
(ii) producing the oscillator control signal based on the
phase-shifted signal; (d) generating a synchronization signal based
on the oscillator control signal; and (e) demodulating the
quadrature phase modulated spread spectrum signal using the
synchronization signal.
15. The method of claim 11 wherein generating the synchronization
signal comprises generating a signal having a frequency equal to
the frequency of the intermediate frequency signal based on the
oscillator control signal.
16. A method for maintaining synchronization between a quadrature
phase modulation spread spectrum transmitter and a quadrature phase
modulation spread spectrum receiver, the method comprising: at the
quadrature phase modulation spread spectrum receiver: (a) receiving
a quadrature phase modulated spread spectrum signal; (b) mixing the
quadrature phase modulated spread spectrum signal with a heterodyne
signal to produce an intermediate frequency signal; (c) removing
the influence of data changes in the quadrature phase modulated
spread spectrum signal from the intermediate frequency signal to
produce an oscillator control signal, wherein removing influence of
data changes in the quadrature phase modulated spread spectrum
signal from the intermediate frequency signal includes multiplying
the frequency of the intermediate frequency signal by a
predetermined multiplication factor, wherein multiplying the
frequency of the intermediate frequency signal by the predetermined
multiplication factor includes multiplying the intermediate
frequency signal by a factor of four to produce a frequency
multiplied signal, wherein multiplying the frequency of the
intermediate frequency signal by the factor of four includes: (i)
squaring the intermediate frequency signal to produce a squared
signal; (ii) filtering out constant components from the squared
signal; (iii) squaring the squared signal to produce a signal
having a frequency equal to four times the frequency of the
intermediate frequency signal; and (iv) filtering out constant
components from the signal having the frequency equal to four times
the frequency of the intermediate frequency signal; (d) generating
a synchronization signal based on the oscillator control signal;
and (e) demodulating the quadrature phase modulated spread spectrum
signal using the synchronization signal.
17. A computer program product comprising computer-executable
instructions embodied in a computer-readable medium for performing
steps comprising: (a) receiving a quadrature phase modulated spread
spectrum signal; (b) mixing the quadrature phase modulated spread
spectrum signal with a heterodyne signal to produce an intermediate
frequency signal; (c) removing the influence of data changes in the
quadrature phase modulated spread spectrum signal from the
intermediate frequency signal to produce an oscillator control
signal, wherein removing the influence of data changes in the
quadrature phase modulated spread spectrum signal from the
intermediate frequency signal includes multiplying a frequency of
the intermediate frequency signal by a fixed multiplication factor
and wherein removing the influence of data changes in the
quadrature phase modulated spread spectrum signal from the
intermediate frequency signal includes producing the intermediate
frequency signal characterized by the following equation:
where .omega. is the frequency of the intermediate frequency signal
and .phi. is the initial phase of the intermediate frequency
signal; (d) generating a synchronization signal based on the
oscillator control signal; and (e) demodulating the quadrature
phase modulated spread spectrum signal using the synchronization
signal.
18. The computer program product of claim 17 wherein demodulating
the quadrature phase modulated spread spectrum signal includes
outputting the quadrature phase modulation spread spectrum signal
to a frequency hopping spread spectrum demodulator.
19. The computer program product of claim 17 wherein demodulating
the quadrature phase modulated spread spectrum signal includes
outputting the quadrature phase modulation spread spectrum signal
to a direct sequence spread spectrum demodulator.
20. A computer program product comprising computer-executable
instructions embodied in a computer-readable medium for performing
steps comprising: at a quadrature phase modulation spread spectrum
receiver: (a) receiving a quadrature phase modulated spread
spectrum signal; (b) mixing the quadrature phase modulated spread
spectrum signal with a heterodyne signal to produce an intermediate
frequency signal; (c) removing the influence of data changes in the
quadrature phase modulated spread spectrum signal from the
intermediate frequency signal to produce an oscillator control
signal, wherein removing the influence of data changes in the
quadrature phase modulated spread spectrum signal from the
intermediate frequency signal includes multiplying the frequency of
the intermediate frequency signal by a predetermined multiplication
factor, wherein multiplying the frequency of the intermediate
frequency signal by the predetermined multiplication factor
includes multiplying the intermediate frequency signal by a factor
of four to produce a frequency multiplied signal; wherein producing
the oscillator control signal includes: (i) shifting the phase of
the frequency multiplied signal by a predetermined amount to
produce a phase-shifted signal; and (ii) producing the oscillator
control signal based on the phase-shifted signal; (d) generating a
synchronization signal based on the oscillator control signal; and
(e) demodulating the quadrature phase modulated spread spectrum
signal using the synchronization signal.
21. The computer program product of claim 20 wherein generating the
synchronization signal comprises generating a signal having a
frequency equal to the frequency of the intermediate frequency
signal based on the oscillator control signal.
22. A computer program product comprising computer-executable
instructions embodied in a computer-readable medium for performing
steps comprising: at a quadrature phase modulation spread spectrum
receiver: (a) receiving a quadrature phase modulated spread
spectrum signal; (b) mixing the quadrature phase modulated spread
spectrum signal with a heterodyne signal to produce an intermediate
frequency signal; (c) removing the influence of data changes in the
quadrature phase modulated spread spectrum signal from the
intermediate frequency signal to produce an oscillator control
signal, wherein removing the influence of data changes in the
quadrature phase modulated spread spectrum signal from the
intermediate frequency signal includes multiplying the frequency of
the intermediate frequency signal by a predetermined multiplication
factor, wherein multiplying the frequency of the intermediate
frequency signal by the predetermined multiplication factor
includes multiplying the intermediate frequency signal by a factor
of four to produce a frequency multiplied signal; wherein
multiplying the frequency of the intermediate frequency signal by
the factor of four includes: (i) squaring the intermediate
frequency signal to produce a squared signal; (ii) filtering out
constant components from the squared signal; (iii) squaring the
squared signal to produce a signal having a frequency equal to four
times the frequency of the intermediate frequency signal, and (iv)
filtering out constant components from the signal having a
frequency equal to four times the frequency of the intermediate
frequency signal; (d) generating a synchronization signal based on
the oscillator control signal; and (e) demodulating the quadrature
phase modulated spread spectrum signal using the synchronization
signal.
Description
The invention relates to spread spectrum (SS) communications
systems. More particularly, the present invention relates to a
receiver for use in spread spectrum communications systems.
RELATED ART
The communication technique known as spread spectrum was developed
during World War II with the primary intent of protecting military
and diplomatic communications. Spread spectrum communication
techniques differ from conventional narrow-band communication
techniques because they spread, rather than concentrate,
transmitted signals over a wide frequency range. In other words,
spread spectrum communication systems effectively spread a
narrow-band information signal into a corresponding wide-band
signal that closely resembles background radio frequency (RF)
noise. Such noise-like characteristics are one of the great
advantages of spread spectrum communication systems. That is,
because spread spectrum signals are noise-like, they are difficult
to detect and hence, there is an inherently high degree of security
with SS type communication techniques. Consequently, SS has been
and remains the communication technique of choice for many military
applications. Without going into great detail, it should also be
appreciated that, for similar reasons, SS signals are difficult to
intercept and even more difficult to jam or interfere with than
conventional narrow-band signals. Again, such exceptional low
probability of intercept (LPI) and anti-jam (AJ) characteristics
are the reasons that the military has used SS based communication
systems for so many years.
Because spread spectrum signals have a high spectral width, the
power spectral density (Watts per Hertz) of such signals is lower
than that of conventional narrow-band signals. This lower
transmitted power density characteristic is another significant
advantage of SS communication systems, as SS and narrow band
signals can occupy the same band, with little or no interference.
Consequently, SS communication systems exhibit a high degree of
immunity to interference generated by other equipment. As a result
of this interference immunity, the Federal Communications
Commission (FCC) and other international regulatory agencies allow
RF equipment to transmit at higher power levels (i.e., longer range
transmission) if spread spectrum transmission techniques are
employed. Hence, all the commercial interest in SS communication
systems today.
The expansion or widening of bandwidth in SS type communication
systems is accomplished through the implementation of a
pseudo-random sequence of binary information, known as a spreading
code. The random quality of the spreading code is ultimately
responsible for the noise-like appearance of the transmitted
broadband SS signal. In reality, the binary sequence that comprises
the spreading code is predictable, and consequently does repeat
(hence the "pseudo" term). However, the randomness of the code is
sufficient to minimize the possibility of accidental duplication or
discovery, and as such the spreading code functions much like a
security encryption key.
Spread Spectrum Modulation Techniques
With regard to spread spectrum type communication systems, the two
most commonly employed signal-spreading techniques are direct
sequencing and frequency-hopping. Both modulation techniques are
characterized by wide frequency spectra and modulated output
signals that occupy a much greater bandwidth than the information
or baseband signal component. In general, to qualify as a spread
spectrum signal, the transmitted signal bandwidth must be much
greater than the information bandwidth, and a function, dependent
on information other than the information being transmitted, must
be employed to determine the transmitted bandwidth. Many commercial
spread spectrum communication systems transmit an RF signal
bandwidth as wide as 20 to 254 times the bandwidth of the
information being sent. Some spread spectrum communications systems
have employed RF bandwidths 1000 times their information
bandwidth.
Direct sequence is perhaps one of the most widely known and
utilized spread spectrum communications systems and it is
relatively simple to implement, in that a narrow band carrier is
modulated by a code sequence. More particularly, direct sequence
spread spectrum (DSSS) systems are so called because they employ a
high speed spreading code sequence, along with the basic
information being sent, to modulate their RF carrier. The high
speed spreading code sequence is used directly to modulate the
carrier, thereby directly setting the transmitted RF bandwidth.
Binary spreading code sequences as short as 2.sup.4 bits or as long
as 2.sup.89 have been employed for this purpose, at code rates from
under one bit per second to several hundred megabits per second.
Direct sequence spectra vary somewhat in spectral shape depending
upon the actual carrier and data modulation used. The most common
signal modulation technique used in DSSS systems is known as binary
phase shift keyed (BPSK) modulation. Using such a BPSK modulation
scheme, the carrier phase of the transmitted signal is abruptly
changed in accordance with the code sequence. Once again, as
discussed above, it will be appreciated that the spreading code
sequence is generated by a pseudo-random noise (PSN) generator and
has a fixed length (i.e., after a given number of bits the code
repeats itself exactly). The speed of the code sequence is called
the chipping rate, measured in chips per second (cps). For direct
sequence, the amount of spreading is dependent upon the ratio of
chips per bit of information. At the receiver, the information is
recovered by multiplying the incoming signal with a locally
generated replica of the spreading code sequence. The result is a
signal that is a maximum when the two signals exactly equal one
another or are "correlated." The correlated signal is then filtered
and sent to a BPSK demodulator.
Signals generated using this DSSS technique appear as noise in the
frequency domain. The wide bandwidth provided by the spreading code
allows the transmitted signal power to drop below the noise
threshold without loss of information.
In another spread spectrum modulation scheme, known as frequency
hopping (FH), the desired wide-band frequency spectrum is generated
in a different manner. In the FH scheme, the carrier frequency of
the transmitter abruptly changes (or hops) in accordance with a
pseudo random spreading code sequence. The specific order in which
frequencies are occupied is a function of the spreading code
sequence, and the rate of hopping from one frequency to another is
a function of the information rate. A frequency hopping spread
spectrum receiver is capable of tracking these frequency changes
and reproduces the original information signal.
Shown in FIG. 1 is a greatly simplified example of a typical direct
sequence spread spectrum (DSSS) transmitter, generally indicated by
the numeral 100. DSSS transmitter 100 includes an information
signal 102 in the form of a serial sequence of binary digits. Also
included is a pseudo-random noise (PSN) generator 104 that is
responsible for reliably generating a fixed-length pseudo-random
binary sequence known as the spreading code. A logic gate 106 that
implements an exclusive-OR logic function is adapted to receive and
logically process the information signal 102 and the spreading code
produced by PSN generator 104. The output of the logic gate 106 is
directed to a binary phase shift key (BPSK) modulator 108. BPSK
modulator 108 also receives a signal from carrier oscillator 110.
The modulated output of BPSK modulator 108 is amplified via RF
power amplifier 112 and subsequently broadcast by antenna 114. As
such, it will be appreciated that input information signal 102 is
logically combined with a spreading code produced by PSN generator
104, and the resulting composite signal is provided as input to
BPSK modulator 108. Modulator 108, with the aid of a carrier signal
produced by carrier oscillator 110, encodes and modulates the
composite signal using a BPSK modulating algorithm. The resulting
BPSK modulated output signal is subsequently amplified via RF power
amplifier 112 and broadcast from transmitter antenna 114.
Shown in FIG. 2 are samples of the initial, intermediate, and final
waveforms generated by the DSSS transmitter 100 shown in FIG. 1.
More particularly, FIG. 2 includes a sample waveform diagram 150
corresponding to a portion of the input information signal 102.
Also included is another sample waveform diagram 152 corresponding
to a portion of a spreading code sequence produced by PSN generator
104. Waveform diagram 154 illustrates a sample output of the XOR
logic gate 106 corresponding to logical processing of the
information waveform 150 and the spreading code waveform 152.
Finally, waveform diagram 156 illustrates the broadband, BPSK
modulated output signal produced by the BPSK modulator 108.
Shown in FIG. 3 is a simplified example of a typical direct
sequence spread spectrum (DSSS) receiver, generally indicated by
the numeral 120. DSSS receiver 120 includes a receiving antenna
122, a broadband RF amplifier 124, and a first signal mixer 126.
Signal mixer 126 is adapted to receive an amplified broadband
signal from amplifier 124 as well as a signal generated by a local
oscillator 128 having a frequency equal to f.sub.c -f.sub.IF, where
f.sub.c is the carrier frequency and f.sub.IF is the intermediate
frequency. The output signal produced by mixer 126 is then compared
at a second mixer 130 to a signal that is generated by a third
mixer 132. The signal generated by mixer 132 is produced using the
same spreading code sequence as that used by the corresponding DSSS
transmitter 100 shown in FIG. 1. This spreading code sequence is
generated in much the same manner as described above for
transmitter 100. That is, a PSN generator 134 and associated clock
function 136 are used to create the binary spreading code sequence.
More particularly, the binary spreading code sequence produced by
PSN generator 134 is combined with an IF carrier signal that is
produced by an IF oscillator 138. It will be further appreciated
that the signal output by the correlating mixer 130 is used to
drive a synchronization circuit 140, which in turn is responsible
for ensuring that the IF carrier signal generated by oscillator 138
is synchronized with carrier oscillator 110 of transmitter 100.
That is, synchronization circuit 140 must reproduce the exact phase
and frequency of the signal output from carrier oscillator 110.
Synchronization circuit 140 performs this function, in part, by
altering the frequency of clock source 136 such that the PSN or
spreading code chip rate matches that of the incoming modulated
broadband signal. Since the spreading code produced by the PSN
generator is the same as that contained within the received signal,
adjusting the clock in the manner described above will eventually
allow the two signals to be brought into a synchronized state.
Again, it will be appreciated that the spreading code produced by
PSN generator 134 is used to modulate the IF carrier produced by
oscillator 138 at mixer 132. This spreading code modulated IF
carrier output of mixer 132 is subsequently provided as one input
to the correlating mixer 130. Again, it will be appreciated that
the output of mixer 132 is a BPSK modulated signal that is similar
to the received broadband signal. This BPSK modulated signal
produced by mixer 132 is compared to the received broadband signal
in mixer 130, which effectively acts as a correlator. The output of
the correlating mixer 130 is then filtered via a low pass filter
(LPF) 142 so as to recover the original sequence of binary
information sent by transmitter 100.
Synchronization Problems of Conventional Spread Spectrum
Receivers
In both direct sequence and frequency hopping spread spectrum
communications systems, the receiver must extract the information
signal from the received RF signal. In order to exact the
information signal, the IF oscillator at the receiver must be
synchronized with the carrier oscillator at the transmitter. In
FIG. 3, this synchronization is performed based on the BPSK signal.
Because the BPSK signal phase changes based on the transmitted
information, the control signals output from synchronization
circuits 140 also vary with the transmitted information. As a
result, synchronization between the oscillators at the transmitter
and receiver cannot be reliably achieved.
FIG. 4 illustrates an example of another type of conventional
spread spectrum receiver in which there is no feedback to the local
oscillator at the receiver. In FIG. 4, the spread spectrum receiver
includes an antenna 400, a pre-selector 402, a mixer 404, and a
phase discriminator 406 connected in series. An oscillator module
408 is connected to mixer 404 and phase discriminator 406.
Antenna 400 receives the signal transmitted from the transmitter
(not shown in FIG. 4). Pre-selector 402 is a bandpass amplifier
that amplifies the received signal over a predetermined frequency
band. Mixer 404 mixes the received signal with one signal output
from oscillator module 408. Phase discriminator 406 outputs a
signal that is proportional to the phase difference between the IF
signal output from mixer 404 and the IF signal output from the
second output from oscillator module 408. Additional modules that
are not shown in FIG. 1 may be used to remove the spreading code
from the signal output from phase discriminator 406.
FIG. 5 illustrates yet another conventional receiver that includes
all of the modules illustrated in FIG. 4 and, in addition, a
frequency and phase adjustment module 500. Frequency and phase
adjustment module 500 receives the output signal from mixer 404.
Frequency and phase adjustment module 500 produces a control signal
that is fed back to an oscillator module 408a. The purpose of the
control signal is to correct the phase and frequency of the IF
signal output from oscillator 408 so that the IF signal is
synchronized with the carrier oscillator of the transmitter.
In the spread spectrum receiver illustrated in FIG. 5, phase
adjustment cannot be made fine enough because of the influence of
the phase changes in the received signal due to the transmitted
information. Phase modulation, e.g., BPSK, QPSK, etc., on the
transmitter side result in nonconherence of the received signal.
Existing methods for generating the oscillator module control
signal are not able to avoid the influence of phase changes in the
received signal caused by the transmitted information. As a result,
the signal output from local oscillator 408a does not remain in
phase with the received signal and demodulation becomes
impossible.
FIG. 6 illustrates yet another conventional spread spectrum
receiver. In FIG. 6, the spread spectrum receiver includes an
antenna 400, a pre-selector 402, a mixer 404, and a phase
discriminator 406 connected in series. An oscillator module 408a is
connected to mixer 404 and phase discriminator 406. These
components and connections are the same as the corresponding
components and connections described above with respect to FIGS. 4
and 5. Hence, a description thereof is not repeated herein.
The spread spectrum receiver illustrated in FIG. 6 also includes a
demodulated signal processing module 600. Demodulated signal
processing module 600 is connected to the output of phase
discriminator 406 to remove the spreading code from the received
signal. Since phase discriminator 406 effectively removes the
carrier from the received signal, the output of demodulated signal
processing module 600 is the information signal.
In addition to removing the spreading code, demodulated signal
processing module 600 also produces a control signal for adjusting
the IF signal output from regulated oscillators module 408a.
However, like the conventional receiver illustrated in FIG. 3, this
control signal is susceptible to changes based on the transmitted
information. As a result, maintaining synchronization is
difficult.
An example of a spread spectrum signal receiver in which the
control signal for adjusting the receiver oscillator is produced by
the demodulated signal processing module 600 is the PRISM.RTM. II
receiver available from Intersil Corporation. The PRISM.RTM. II
receiver is a quadrature phase modulation receiver marketed for use
in wireless local area networks (LANs). However, the Intersil
receiver has the following disadvantages: When the frequency and
phase difference between the local oscillators of the transmitter
and the receiver varies, there is high probability of losing
connection. In addition, restoration of synchronization after
losing the connection is also difficult when the IF signal output
from local oscillator is unstable. A long uninterrupted connection
using the Intersil receiver is problematic due to the large
adjustment time constant using the selected method of phase
adjustment using the demodulated signal output from demodulated
signal processing module 600.
Accordingly, there exists a need for a novel quadrature phase
modulation receiver that avoids at least some of the difficulties
associated with conventional receivers.
DISCLOSURE OF THE INVENTION
A phase modulation spread spectrum receiver maintains
synchronization with a phase modulation spread spectrum transmitter
by removing the influence of the transmitted information from the
intermediate frequency produced by the local oscillator at the
receiver. In order to remove this influence, a frequency multiplier
receives the IF signal produced by the down converter at the
receiver and multiplies the frequency of the signal by a
predetermined multiplication factor, which is preferably equal to
four. Once the IF signal has been frequency-multiplied, the
influence of the transmitted information is removed. This signal
can then be used to adjust the frequency of the receiver
oscillator.
A phase modulation spread spectrum receiver according to the
invention can be implemented in hardware, software, or a
combination of hardware and software. Accordingly, some aspects of
the invention may be implemented as computer program products
comprising computer executable instructions in a computer readable
medium.
Accordingly, it is an object of the invention to provide a spread
spectrum receiver capable of reliably maintaining synchronization
with the transmitter.
It is yet another object of the present invention to provide a
spread spectrum receiver in which phase fluctuations in the
received signal due to the data content of the received signal do
not affect the reference frequency of the phase discriminator which
is equal to the intermediate frequency.
It is yet another object of the invention to reduce the time
constant for adjusting the intermediate frequency signal.
Some of the objects of the invention having been stated
hereinabove, other objects will be evident as the description
proceeds, when taken in connection with the accompanying drawings
as best described hereinbelow.
BRIEF DESCRIPTION OF THE DRAWINGS
A description of preferred embodiments of the invention will now
proceed with reference to the accompanying drawings of which:
FIG. 1 is a block diagram of a conventional spread spectrum
transmitter;
FIG. 2 is a timing diagram illustrating conventional spread
spectrum modulation techniques;
FIG. 3 is a block diagram of a conventional spread spectrum
receiver;
FIG. 4 is a block diagram of another conventional spread spectrum
receiver;
FIG. 5 is a block diagram of yet another conventional spread
spectrum receiver with a frequency and phase adjustment module for
adjusting the phase and frequency of the intermediate signal output
from the local oscillator;
FIG. 6 is a block diagram of yet another conventional spread
spectrum receiver in which phase adjustment of the received signal
is based on the demodulated signal output from a demodulated signal
processing module;
FIG. 7 is a block diagram of a phase modulation spread spectrum
receiver according to an embodiment of the present invention;
FIG. 8 is a block diagram of a phase discriminator suitable for use
in the phase modulation spread spectrum receiver illustrated in
FIG. 7;
FIG. 9 is a schematic diagram of a frequency multiplier circuit
suitable for use with the phase modulation spread spectrum receiver
illustrated in FIG. 7;
FIG. 10 is a schematic diagram of a phase shifter circuit suitable
for use with the phase modulation spread spectrum receiver
illustrated in FIG. 7;
FIG. 11 is a partial schematic/partial block diagram illustrating a
comparison signal generation circuit suitable for use with the
phase modulation spread spectrum receiver illustrated in FIG.
7;
FIG. 12 is a graph illustrating the signal output from the
comparison signal generation circuit illustrated in FIG. 11;
FIG. 13 is a schematic diagram of a frequency and phase
discriminator suitable for use with the phase modulation spread
spectrum receiver illustrated in FIG. 7; and
FIG. 14 is a block diagram of a regulated oscillators module
suitable for use with the phase modulation spread spectrum receiver
illustrated in FIG. 7.
DETAILED DESCRIPTION OF THE INVENTION
FIG. 7 is a block diagram of a spread spectrum receiver according
to an embodiment of the present invention. In FIG. 7, spread
spectrum signal receiver 700 includes an antenna 400, a
pre-selector 402, a mixer 404, a phase discriminator 406, and a
demodulated signal processing module 600. These components are the
same as the correspondingly-numbered components described above
with respect to FIGS. 1-6. However, in order to provide a complete
description of the invention, these components and the connections
between these components will now be discussed in detail.
Antenna 400 is connected to the input of pre-selector 402. Antenna
400 may be a Yagi antenna or any other type of antenna capable of
receiving a spread spectrum signal. The structure of the antenna
depends on the range of frequencies that are desired to be
received. In the United States, the Federal Communications
Commission (FCC) has allocated three frequency bands for spread
spectrum communications: 902-928 MHz, 2.4000-2.4835 GHz, and
5.725-5.850 GHz. Accordingly, antenna 400 may be adapted to receive
signals in one or more of these frequency bands. However, the
present invention is not limited to receiving signals only in these
frequency bands. Receiving spread spectrum signals in any frequency
band allocated for such signals is within the scope of the
invention.
Pre-selector 402 may be a bandpass filter for performing initial
frequency selection of the received signal. Bandpass filters
suitable for use with embodiments of the present invention include
active filters and Butterworth bandpass filters having upper and
lower cutoff frequencies that correspond to the desired spread
spectrum frequency band.
The output of pre-selector 402 is connected to the first input of
mixer 404. Mixer 404 may be any type of mixer capable of frequency
mixing two or more signals. For example, a mixer suitable for use
with embodiments of the present invention may include a
continuously variable transconductance device, such as a dual gate
field effect transistor (FET). In such a device, the output voltage
may be the frequency-converted input signal. The input voltage at
one of the gates may be the receiver input signal. The input
voltage at the other gate may be the frequency of the heterodyne
signal output from a regulated oscillators module 408a. Such a
multiplier is preferably of the four quadrant type so that
multiplier action is obtained regardless of the sign of the
received input signal and the frequency of the heterodyne
signal.
The output of mixer 404 is connected to the input of phase
discriminator 406 and frequency multiplier 702. Phase discriminator
406 also receives the IF signal output from a frequency
synthesizer, generally designated 710.
FIG. 8 illustrates an exemplary phase discriminator 406 suitable
for use with embodiments of the present invention. In FIG. 8, phase
discriminator 406 includes mixers 800 and 802 and phase shifter
804. Mixers 800 and 802 mix the signal output from mixer 404
illustrated in FIG. 7 with a frequency multiplied signal output
from frequency synthesizer 710. More particularly, mixer 802 mixes
a phase-shifted version the frequency multiplied signal with the
signal output from mixer 404, and mixer 800 mixes a
non-phase-shifted version of the frequency multiplied signal with
the signal output from mixer 404. The reason that the signal having
a frequency output from mixer 404 is mixed with the
frequency-multiplied signal having a frequency equal to the
intermediate frequency is to provide quadrature demodulation. The
outputs from mixers 800 and 802 are input to demodulated signal
processing module 600 illustrated in FIG. 7.
The reason that the signal output from mixer 404 is mixed with both
a phase-shifted and a non-phase-shifted version of the
frequency-multiplied signal output from frequency synthesizer 710
is that in the module 406, quadrature de-modulation is being
performed, which requires two frequencies as references for
demodulation. The signal outputs from mixers 800 and 802 are
combined in the bus connecting phase discriminator 406 and
demodulated signal processing module 600. The signal produced by
phase discriminator 406 is output to demodulated signal processing
module 600 for de-spreading. The signal output from phase
discriminator 406 is a PSN code similar to PSN code 154 illustrated
in FIG. 2.
Demodulated signal processing module 600 illustrated in FIG. 7 can
be adapted to demodulate a frequency hopping spread spectrum
signal, a direct sequence spread spectrum signal, as well as other
types of spread spectrum signals. Thus, the present invention is
not limited to any particular type of spread spectrum demodulation.
Any type of spread spectrum demodulator is intended to be within
the scope of the invention. Exemplary spread spectrum demodulators
suitable for use with embodiments of the present invention include
model numbers HFA3824 and HFA3860 available from Intersil
Corporation.
Referring again to FIG. 7, components 400, 402, 404, 408a, 406, and
600 are conventional components of a spread spectrum receiver and
may cause loss of synchronization between the sender and the
receiver due to variations in the intermediate frequency signal
output from regulated oscillators module 408a. However, according
to the present embodiment, frequency multiplier 702 removes the
influence of data changes in the received signal from the
synchronization signal input to phase discriminator 406.
Frequency multiplier 702 receives the signal output from mixer 404
to multiply it by a predetermined multiplication factor. For a
quadrature phase modulation receiver, the multiplication factor of
frequency multiplier 702 is preferably set to four. Setting the
multiplication factor to four removes the influence of data changes
in the received signal from the synchronization signal input to
phase discriminator 406. However, the present invention is not
limited to a multiplication factor of four. Any other
multiplication factor that reduces or removes the influence of data
changes in the received signal from the synchronization signal is
within the scope of the invention.
Multiplying the signal output from mixer 404 by a factor of four
may be accomplished, for example, by squaring the signal power of
the IF signal twice using standard power multipliers. Squaring the
signal power may be accomplished by simultaneously applying the
signal to both inputs of a first power multiplier, filtering the
constant part of the resulting signal, passing the signal through a
DC blocking capacitor, squaring the filtered signal by
simultaneously applying the signal to both inputs of a second power
multiplier, and filtering the constant part of the resulting
signal.
FIG. 9 is a schematic diagram of a frequency multiplier suitable
for use with embodiments of the present invention. In FIG. 9
frequency multiplier 702 comprises first and second power
multipliers 900 and 902 and first and second DC blocking capacitors
904 and 906. Power multipliers 900 and 902 are standard circuits
that multiply the signals at the input terminals. The internal
details of such circuits are generally known to one of ordinary
skill in the power electronics art and need not be described
herein. What is important for purposes of the present invention is
that the signal at the output terminals of each multiplier is the
product of the voltages at the input terminals. In other words:
The signal v.sub.o2 applied to the input terminals of power
multiplier 902 is equal to v.sub.o1 without the DC component, due
to the action of DC blocking capacitor 904. In other words,
Power multiplier 902 squares the signal v.sub.o2 to produce the
signal v.sub.o3. The relationship between v.sub.o2 and v.sub.o3 is
as follows:
Capacitor 906 removes the DC component k.sub.2 from the signal
v.sub.o3 to produce the signal v.sub.o4. v.sub.o4 is thus equal to
the square of v.sub.o2, which is equal to the square of the input
signal. In other words,
Squaring an input signal, filtering the constant component,
squaring the resulting signal and filtering the constant component
from the resulting signal has the effect of multiplying the
frequency of the input signal by four. This effect will be
discussed in more detail below.
The present invention is not limited to using a power multiplier to
increase the frequency of the signal output from mixer 404. Other
schematics achieving the same results or equivalent results are
within the scope of the invention. The 4.sup.th harmonic must be
obtained from the combinations frequencies, in particular by
multiplying signals by themselves.
Once the frequency of the signal output from the frequency
multiplier 702 is multiplied by four, the signal will have no
components dependent on the modulating signal, which, in this
example, changes the phase of the received signal in multiples of
PI/2, i.e., (3.1416 . . . )/2, since quadrature phase modulation is
used. The proof of the independence of the phase of the resulting
signal from the modulating signal is illustrated by the following
mathematical equations.
A signal with quadrature phase modulation after passing the
pre-selector 402 and the mixer 404 can be represented by the
following expression:
where .omega. is the frequency of the signal, which is equal to the
immediate frequency, .phi. is the initial phase of the signal,
which does not depend on the modulating signal, and A, B are
amplitude coefficients with possible values of 1 and -1. These
coefficients determine the dependency of the phase of the received
signal on the modulating signal.
After squaring the signal, the following expression is
obtained:
In light of the following trigonometric identities:
expression (2) can be rewritten as
As stated above, the constants A and B have values of .+-.1 in
accordance with the modulating signal. Thus, A.sup.2 =B.sup.2 =1.
Expression (3) reduces to:
After filtering out the constant component, the signal output from
the first stage of frequency multiplier 702 can be represented
as:
After the second squaring, the following expression is obtained:
##EQU1##
After filtering out the constant component the signal can be
represented by the following expression:
The signal represented by equation (7) is an output signal of the
frequency multiplier 702. From equation (7), the phase of the
signal does not depend on the modulating signal, and the frequency
is equal to the intermediate frequency multiplied by four. Because
the influence of the modulating signal is removed, the frequency
multiplied signal can be used to demodulate the received signal
with a reduced likelihood of losing synchronization with the
received signal.
Referring back to the block diagram illustrated in FIG. 7, output
from the frequency multiplier 702 is input to the first input of
the frequency and phase discriminator 708 through the phase shifter
704. The output signal f.sub.IF from the frequency synthesizer 710
is input to comparison signal formation circuit 706. The second
input of the frequency and phase discriminator 708 receives the
output signal from the comparison signal formation circuit 706. The
frequency and phase discriminator 708 produces a control signal for
regulating the intermediate frequency signal output from regulated
oscillators module 408a.
As stated above, phase shifter 704 receives the signal output from
the frequency multiplier 702. Phase shifter 704 shifts the phase of
the signal by a factor of PI/2. The reason for shifting the signal
output from mixer 404 by a factor of PI/2 is to provide the
required phase shift for signal synchronization in phase
autocorrelation of the signal performed by the phase and frequency
discriminator 708 and to compensate for the phase shift between the
input synchronization signal from the output of mixer 404 and the
VCO signal output from regulated oscillators module 408a.
FIG. 10 is a block diagram of an exemplary phase shifter suitable
for use with embodiments of the present invention. In the
illustrated embodiment, phase shifter 704 comprises a tuned circuit
that shifts the phase of the input signal by ninety degrees. The
tuned circuit includes both capacitive and inductive elements,
which may be distributed or discrete components. The values of the
inductive and capacitive elements are preferably chosen such that
phase shifter 704 functions as a quarter-wave transformer of the
input signal. A quarter-wave transformer produces a PI/2 phase
shift of the signal having the intermediate frequency multiplied by
four between its input and output terminals and can be readily
formed using microstrip line or discrete components.
Comparison signal formation circuit 706 receives a signal having
the frequency f.sub.IF from the frequency synthesizer 710 and
produces a control signal to be input to frequency and phase
discriminator 708. FIG. 11 illustrates an example of a comparison
signal formation circuit suitable for use with embodiments of the
present invention. In the illustrated embodiment, comparison signal
formation circuit 706 comprises first and second signal squaring
circuits 1100 and 1102, a three-input multiplier 1104, and a
half-wave amplitude detector 1106. Squaring circuits 1100 and 1102
each comprise power multipliers in which the inputs receive the
same signal. Module 1104 also comprises a three input power
multiplier. Half-wave amplitude detector 1106 is a rectifier
circuit that only produces an output signal when the input signal
is greater than zero in magnitude.
Squaring circuit 1100 receives and squares the signal of the
intermediate frequency f.sub.IF output from frequency synthesizer
710. Squaring circuit 1102 squares the signal output from
multiplier 1100. The outputs from squaring circuits 1100 and 1102
and the output from frequency synthesizer 710 are multiplied by
multiplier 1104. The purpose of squaring the signal twice and then
multiplying the results of squaring and also multiplying the result
by the input signal is to get the signal of the special wave (or
shape) illustrated in FIG. 12. The signal is formed so that there
is a part that is a positive impulse with a porosity or porousness
of 8, i.e., with period equal to the period of the IF signal and a
duration close to half of the period of the 4.sup.th harmonic of
the IF signal from which it is obtained. In other words, the ratio
of the time duration of the positive portion of the signal output
from multiplier 1104 to the period of the signal is preferably
equal to 8.
FIG. 12 illustrates the signal output from multiplier 1104. The
resulting signal can be represented by the expression:
Such signal is necessary for the phase and frequency discriminator
708, which receives the output from both phase shifter 704 and
comparison signal formation circuit 706.
The signal illustrated in FIG. 12 is then half-wave rectified by
half wave amplitude detector circuit 1106 to remove all negative
portions of the signal. In this manner, the required signal is
formed. The signal is then input to the control input of the
frequency and phase discriminator 708.
FIG. 13 is a schematic diagram of an exemplary frequency and phase
discrimination circuit suitable for use with the receiver
illustrated in FIG. 7. In the illustrated embodiment, frequency and
phase discriminator circuit 708 receives the signals output from
phase shifter 704 and comparison signal formation circuit 706 and
compares the signal input from module 704, which is the result of
processing of the input quadrature phase modulated signal at the
intermediate frequency, with the reference signal from the output
of module 706. As a result of the comparison, an error signal is
obtained at the output of frequency and phase discrimination
circuit 708. The error signal is used for frequency adjustment of
the signal output from the VCO of regulated oscillators module
408a, so that the signal output from frequency synthesizer 710 has
the intermediate frequency and is in synchronization with the
signal output from mixer 404.
In the illustrated example, frequency and phase discrimination
circuit 708 can be built according to the frequency-phase
discriminator diagram based on the Foster--Sili frequency
discriminator with reference frequency oscillations as an input.
The operation of such a frequency and phase discriminator is
described in V. Manassevitsch, Frequency Synthesizers: Theory &
Design, New York, London, Sydney, Toronto (1979), the disclosure of
which is incorporated herein by reference in its entirety.
FIG. 14 is a block diagram illustrating an exemplary regulated
oscillators module suitable for use with a phase modulation spread
spectrum receiver according to an embodiment of the present
invention. In FIG. 14, regulated oscillators module 408a includes a
heterodyne signal oscillator 1400, a voltage controlled oscillator
1402, and a frequency synthesizer 710. Heterodyne signal generator
1400 generates an output signal having a frequency equal to the
difference between the carrier frequency of the received signal and
the intermediate frequency. This signal is output to mixer 404 to
be mixed with the received signal and produce a signal having the
intermediate frequency. Voltage controlled oscillator 1402 receives
the control signal from frequency and phase discriminator 708 and
outputs a signal having the intermediate frequency f.sub.IF divided
by n (f.sub.IF /n). This signal having a frequency of f.sub.IF /n
is output to the frequency synthesizer 710 and demodulated signal
processing module 600. The frequency f.sub.IF from the frequency
synthesizer 710 is output to the phase discriminator 406 to be
mixed with the IF-converted signal output from mixer 404, as
discussed above.
Frequency synthesizer 710 receives the frequency signal f.sub.IF /n
output from VCO 1402 of regulated oscillators module 408a and
produces a signal having the intermediate frequency. The
synchronization frequency f.sub.IF /n of demodulated signal
processing module 600, which is output from VCO 1402, is a multiple
of the frequency of the information elements in the received signal
and depends on the manner of signal processing in module 600 and on
its type. The frequency f.sub.IF /n can be equal to the chipping
rate or a multiple of the chipping rate. Frequency synthesizer 710
can be, for example, a typical multiplier based on a
voltage-controlled oscillator with a phase lock loop (PLL) system,
where the comparison frequencies are the oscillator frequency
divided using counters and the frequency of the output signal from
frequency synthesizer 710.
Because the intermediate frequency signal output from frequency
synthesizer 710 of regulated oscillators module 408a is controlled
based on a frequency multiplied version of the input signal, the
intermediate frequency signal remains stable and in sync with the
transmitter carrier oscillator. Accordingly, reliable demodulation
can occur.
It will be understood that various details of the invention may be
changed without departing from the scope of the invention.
Furthermore, the foregoing description is for the purpose of
illustration only, and not for the purpose of limitation--the
invention being defined by the claims.
* * * * *