U.S. patent number 6,483,480 [Application Number 09/589,859] was granted by the patent office on 2002-11-19 for tunable impedance surface.
This patent grant is currently assigned to HRL Laboratories, LLC. Invention is credited to Robert Y. Loo, James H. Schaffner, Daniel Sievenpiper, Greg Tangonan.
United States Patent |
6,483,480 |
Sievenpiper , et
al. |
November 19, 2002 |
Tunable impedance surface
Abstract
A tuneable impedance surface for steering and/or focusing a
radio frequency beam. The tunable surface comprises a ground plane;
a first plurality of elements disposed in an array a first distance
from the ground plane, the distance being less than a wavelength of
the radio frequency beam; and a second plurality of elements
disposed in an array a second distance from the ground plane, the
second plurality of elements be moveable relative to the first
plurality of elements.
Inventors: |
Sievenpiper; Daniel (Los
Angeles, CA), Tangonan; Greg (Oxnard, CA), Loo; Robert
Y. (Agoura Hills, CA), Schaffner; James H. (Chatsworth,
CA) |
Assignee: |
HRL Laboratories, LLC (Malibu,
CA)
|
Family
ID: |
27065645 |
Appl.
No.: |
09/589,859 |
Filed: |
June 8, 2000 |
Related U.S. Patent Documents
|
|
|
|
|
|
|
Application
Number |
Filing Date |
Patent Number |
Issue Date |
|
|
537923 |
Mar 29, 2000 |
|
|
|
|
537922 |
Mar 29, 2000 |
|
|
|
|
Current U.S.
Class: |
343/909;
343/700MS; 343/754; 343/853 |
Current CPC
Class: |
H01Q
3/44 (20130101); H01Q 9/0442 (20130101); H01Q
15/0066 (20130101); H01Q 15/008 (20130101); H01H
59/0009 (20130101) |
Current International
Class: |
H01Q
15/00 (20060101); H01Q 3/44 (20060101); H01Q
9/04 (20060101); H01Q 3/00 (20060101); H01H
59/00 (20060101); H01Q 015/02 () |
Field of
Search: |
;343/7MS,754,756,778,850,853,909,910 |
References Cited
[Referenced By]
U.S. Patent Documents
Foreign Patent Documents
|
|
|
|
|
|
|
196 00 609 |
|
Apr 1997 |
|
DE |
|
WO 98/21734 |
|
May 1998 |
|
WO |
|
WO 99/50929 |
|
Oct 1999 |
|
WO |
|
WO 00/44012 |
|
Jul 2000 |
|
WO |
|
Other References
Balanis, C., "Aperture Antennas", Antenna Theory, Analysis and
Design, 2nd Edition, (New York, John Wiley & Sons, 1997), Chap.
12, pp. 575-597. .
Balanis, C., "Microstrip Antennas", Antenna Theory, Analysis and
Design, 2nd Edition, (New York, John Wiley & Sons, 1997), Chap.
14, pp. 722-736. .
Cognard, J., "Alignment of Nematic Liquid Crystals and Their
Mixtures" Mol. Cryst. Liq, Cryst. Suppl. 1, 1 (1982) pp. 1-74.
.
Doane, J.W., et al., "Field Controlled Light Scattering from
Nematic Microdroplets", Appl. Phys. Lett., vol. 48 (Jan. 1986) pp.
269-271. .
Jensen, M.A., et al., "EM Interaction of Handset Antennas and a
Human in Personal Communications", Proceedings of the IEEE, vol.
83, No. 1 (Jan. 1995) pp. 7-17. .
Jensen, M.A., et al., "Performance Analysis of Antennas for
Hand-held Transceivers using FDTD", IEEE Transactions on Antennas
and Propagation, vol. 42, No. 8 (Aug. 1994) pp. 1106-1113. .
Ramo, S., et al., Fields and Waves in Communication Electronics,
3rd Edition (New York, John Wiley & Sons, 1994) Section
9.8-9.11, pp. 476-487. .
Sievenpiper, D., et. al., "High-Impedance Electromagnetic Surfaces
with a Forbidden Frequency Band", IEEE Transactions on Microwave
Theory and Techniques, vol. 47, No. 11, (Nov. 1999) pp. 2059-2074.
.
Sievenpiper, D., "High-Impedance Electromagnetic Surfaces", Ph.D.
Dissertation, Dept. of Electrical Engineering, University of
California, Los Angeles, CA, 1999. .
Wu, S.T., et al., "High Birefringence and Wide Nematic Range
Bis-tolane Liquid Crystals", Appl. Phys. Lett. vol. 74, No. 5,
(Jan. 1999) pp. 344-346. .
Bradley, T.W., et al., "Development of a Voltage-Variable
Dielectric (VVD), Electronic Scan Antenna," Radar 97, Publication
No. 449, pp. 383-385 (Oct. 1997)..
|
Primary Examiner: Phan; Tho G.
Attorney, Agent or Firm: Ladas & Parry
Parent Case Text
CROSS REFERENCES TO RELATED APPLICATIONS
This application is a continuation in part of U.S. patent
application Ser. No. 09/537,923, filed Mar. 29, 2000 and entitled
"A Tunable Impedance Surface", the disclosure of which is hereby
incorporated herein by reference. This application is also a
continuation in part of U.S. patent application Ser. No.
09/537,922, filed Mar. 29, 2000 and entitled "An Electronically
Tunable Reflector", the disclosure of which is also hereby
incorporated herein by reference.
The present application is also related to U.S. patent application
Ser. No. 09/537,921, filed Mar. 29, 2000 and entitled "An End-Fire
Antenna or Array on Surface with Tunable Impedance" the disclosures
of which is hereby incorporated herein by reference.
Claims
What is claimed is:
1. A tuneable impedance surface for reflecting a radio frequency
beam, the tunable surface comprising: (a) a ground plane; (b) a
first plurality of elements disposed in an array a first distance
from the ground plane, the distance being less than a wavelength of
the radio frequency beam; and (c) a second plurality of elements
disposed in an array a-second distance from the ground plane, the
second plurality of elements being moveable relative to the first
plurality of elements.
2. The tuneable impedance surface of claim 1 further including a
first substrate having first and second major surfaces, said
substrate supporting said ground plane on the first major surface
thereof and supporting said first plurality of elements on the
second major surface thereof.
3. The tuneable impedance surface of claim 1 further including a
second substrate having first and second major surfaces, said
substrate supporting said second plurality of elements on the
second major surface thereof.
4. The tuneable impedance surface of claim 3 wherein each element
of the first and second pluralities of elements has an outside
dimension which is less than the wavelength of the radio frequency
beam.
5. The tuneable impedance surface of claim 1 wherein the first
plurality of elements is coupled to the ground plane by
electrically conductive vias in a substrate supporting said ground
plane and said first plurality of elements.
6. The tuneable impedance surface of claim 1 wherein the first
plurality of elements is arranged in a planar array.
7. The tuneable impedance surface of claim 1 wherein the second
plurality of elements is arranged in a planar array.
8. The tuneable impedance surface of claim 1 wherein the first
plurality of elements and the second plurality of elements are
separated by a dielectric layer.
9. The tuneable impedance surface of claim 8 wherein the first
plurality of elements and the second plurality of elements abut
said dielectric layer.
10. The tuneable impedance surface of claim 8 wherein the first
plurality of elements is fixed relative to said dielectric layer
and the second plurality of elements is moveable relative to said
dielectric layer.
11. The tunable tunable impedance surface of claim 1 wherein the
first plurality of elements are disposed in a two dimensional
array, wherein each of the first plurality of elements are spaced
from one another, wherein the second plurality of elements are
disposed in a two dimensional array, wherein each of the second
plurality of elements are spaced from one another and wherein the
second plurality of elements are disposed between the first
plurality of elements and the ground plane.
12. A method of tuning a high impedance surface for reflecting a
radio frequency signal comprising: arranging a first plurality of
spaced-apart conductive surfaces in an array disposed essentially
parallel to and spaced from a conductive back plane, arranging a
second plurality of spaced-apart conductive surfaces in an array
disposed essentially parallel to and spaced from said conductive
back plane by a distance greater than the distance said first
plurality of spaced-apart conductive surfaces is spaced from said
conductive back plane, and moving the second plurality of
spaced-apart conductive surfaces relative to the first plurality of
spaced-apart conductive surfaces.
13. The method of claim 12 wherein said pluralities of spaced-apart
conductive surfaces are arranged on a printed circuit board.
14. The method of claim 12 wherein the step of moving the second
plurality of spaced-apart conductive surfaces relative to the first
plurality of spaced-apart conductive, surfaces comprises rotational
movement in a plane essentially parallel to said arrays.
15. The method of claim 12 wherein the size of each conductive
surface along a major axis thereof is less than a wavelength of the
radio frequency signal, and preferably less than one tenth of a
wavelength of the radio frequency signal, and the spacing of each
conductive surface of the first plurality from the back plane is
less than a wavelength of the radio frequency signal.
16. The method of claim 12 wherein the high impedance surface is
tuned so that a generally linear reflection phase function is
impressed on the high impedance surface.
17. The method of claim 16 wherein the linear phase function has
discontinuities of 2.pi. therein.
18. The method of claim 12 wherein the conductive surfaces are
generally planar and wherein the array is generally planar.
19. The method of claim 12 wherein the conductive surfaces are
metallic and wherein the conductive back plane is metallic.
20. The method of claim 12 wherein the size of each conductive
surface along a major axis thereof is less than one tenth of a
wavelength of the radio frequency signal and the spacing of each
conductive surface of the first plurality from the back plane is
less than a wavelength of the radio frequency signal.
21. A tuneable impedance surface for reflecting a radio frequency
beam, the tunable surface comprising: (a) a first substrate formed
of a dielectric material having a thickness which is less than a
wavelength of the radio frequency beam; (b) a conductive plane
disposed on a major surface of said first substrate; (c) a first
plurality of conductive elements disposed in an array on another
major surface of said first substrate, wherein each element of the
first plurality of elements has an outside dimension which is less
than the wavelength of the radio frequency beam; (d) a second
substrate disposed (i) in a confronting relationship to said first
substrate and (ii) relatively moveable to said first substrate; and
(e) a second plurality of conductive elements disposed in an array
on said second substrate wherein each element of the second
plurality of elements has an outside dimension which is less than
the wavelength of the radio frequency beam.
22. The tuneable impedance surface of claim 21 wherein the first
plurality of elements are coupled to the conductive plane by
electrically conductive vias arranged in said first substrate.
23. A tuneable impedance surface for reflecting a radio frequency
beam impinging the tuneable impedance surface, the tunable surface
comprising: (a) a ground plane; (b) a first plurality of elements
disposed in a two dimensional array a first distance from the
ground plane, the distance being less than a wavelength of the
radio frequency beam; and (c) a second plurality of elements
disposed in a two dimensional array a second distance from the
ground plane, the second plurality of elements being disposed
adjacent to and moveable relative to the first plurality of
elements for changing a direction by which the radio frequency
signal reflects from the high impedance surface.
24. The tunable tunable impedance surface of claim 23 wherein each
of the first plurality of elements are spaced from one another,
wherein each of the second plurality of elements are spaced from
one another and wherein the second plurality of elements are
disposed between the first plurality of elements and the ground
plane.
25. The tunable tunable impedance surface of claim 24 wherein the
first plurality of elements are arranged on a first substrate,
wherein the first plurality of elements are ohmically isolated from
one another on the first substrate, wherein the second plurality of
elements are arranged on a second substrate, and wherein the second
plurality of elements are ohmically isolated from one another on
the second substrate.
26. A method of tuning a high impedance surface for reflecting a
radio frequency signal impinging the high impedance surface,
comprising: arranging a first plurality of spaced-apart, isolated
conductive surfaces in a two dimensional array disposed essentially
parallel to and spaced from a conductive back plane, arranging a
second plurality of spaced-apart, isolated conductive surfaces in a
two dimensional array disposed essentially parallel to and spaced
from said conductive back plane by a distance greater than the
distance said first plurality of spaced-apart conductive surfaces
is spaced from said conductive back plane, and moving the second
plurality of spaced-apart conductive surfaces relative to the first
plurality of spaced-apart conductive surfaces in order to change a
direction by which the radio frequency signal reflects from the
high impedance surface.
Description
TECHNICAL FIELD
This invention relates to a surface having a tunable
electromagnetic impedance which acts as a reconfigurable beam
steering reflector.
BACKGROUND OF THE INVENTION
Steerable antennas today are found in two common configurations:
those with a single feed or reflector that is mechanically steered
using a gimbal, and those with a stationary array of electronically
phased radiating elements. Both have shortcomings, and the choice
of system used is often a tradeoff between cost, speed,
reliability, and RF (radio frequency) performance. Mechanically
steered antennas are inexpensive, but moving parts can be slow and
unreliable, and they can require an unnecessarily large volume of
unobstructed free space for movement. Active phased arrays are
faster and more reliable, but they are much more expensive, and can
suffer from significant losses due to the complex feed structure
required to supply the RF signal to and/or receive the RF signal
from each active element of the phased array. Losses can be
mitigated if an amplifier is included in each element or subarray,
but this solution contributes to noise and power consumption and
further increases the cost of the antenna.
One alternative is to use a reflectarray geometry, and replace the
lossy corporate feed network with a free space feed. The actively
phased elements operate in reflection mode, and are illuminated by
a single feed antenna. The array steers the RF beam by forming an
effective reflection surface defined by the gradient of the
reflection phase across the array. Using current techniques, such a
system still requires a large number of expensive phase
shifters.
There is a need for a reflective surface, in which the reflection
phase could be arbitrarily defined, and easily varied as a function
of position. The surface should be less expensive than a comparably
sized array of conventional phase shifters, yet hopefully offer
similar RF performance. Such a surface could behave as a generic
reconfigurable reflector, with the ability to perform a variety of
important functions including steering or focusing of one or more
RF beams. It is the object of this invention to fulfill this
need.
The reconfigurable reflector disclosed herein is based a resonant
textured ground plane, often known as the high-impedance surface or
simply the Hi-Z surface. This electromagnetic structure has two
important RF properties that are applicable to low profile
antennas. It suppresses propagating surface currents, which
improves the radiation pattern of antennas on finite ground planes
and it provides a high-impedance boundary condition, acting as an
artificial magnetic conductor, which allows radiating elements to
lie in close proximity to the ground plane without being shorted
out. It has origins in other well-known electromagnetic structures
such as the corrugated surface and the photonic band gap surface. A
prior art high-impedance surface is disclosed in a pending US
patent application of D. Sievenpiper, E. Yablonovitch, "Circuit and
Method for Eliminating Surface Currents on Metals", U.S.
provisional patent application Ser. No. 60/079,953, filed on Mar.
30, 1998.
A prior art high-impedance surface is shown in FIG. 1. It consists
of an array of metal top plates or elements 10 on a flat metal
sheet 12. It can be fabricated using printed circuit board
technology with the metal plates or elements 10 formed on a top or
first surface of a printed circuit board and a solid conducting
ground or back plane 12 formed on a bottom or second surface of the
printed circuit board. Vertical connections are formed as metal
plated vias 14 in the printed circuit board, which connect the
elements 10 with the underlying ground plane 12. The metal members,
comprising the top plates 10 and the vias 14, are arranged in a
two-dimensional lattice of cells or cavities, and can be visualized
as mushroom-shaped or thumbtack-shaped members protruding from the
flat metal surface 12. The thickness of the structure, which is
controlled by the thickness of the printed circuit board, is much
less than one wavelength for the frequencies of interest. The sizes
of the elements 10 are also kept less than one wavelength for the
frequencies of interest. The printed circuit board is not shown for
ease of illustration.
Turning to FIG. 2, the properties of this surface can be explained
using an effective circuit model or cavity which is assigned a
surface impedance equal to that of a parallel resonant LC circuit.
The use of lumped cavities to describe electromagnetic structures
is valid when the wavelength is much longer than the size of the
individual features, as is the case here. When an electromagnetic
wave interacts with the surface of FIG. 1, it causes charges to
build up on the ends of the top metal plates 10. This process can
be described as governed by an effective capacitance C. As the
charges slosh back and forth, in response to a radio-frequency
field, they flow around a long path P through the vias 14 and the
bottom metal surface 12. Associated with these currents is a
magnetic field, and thus an inductance L. The capacitance C is
controlled by the proximity of the adjacent metal plates 10 while
the inductance L is controlled by the thickness of the
structure.
The structure is inductive below the resonance and capacitive above
resonance. Near its resonance frequency, ##EQU1##
the structure exhibits high electromagnetic surface impedance. The
tangential electric field at the surface is finite, while the
tangential magnetic field is zero. Thus, electromagnetic waves are
reflected without the phase reversal that occurs on a flat metal
sheet. In general, the reflection phase can be 0, .pi., or anything
in between, depending on the relationship between the test
frequency and the resonance frequency of the structure. The
reflection phase as a function of frequency, calculated using the
effective medium model, is shown in FIG. 3. Far below resonance, it
behaves like an ordinary metal surface, and reflects with a .pi.
phase shift. Near resonance, where the surface impedance is high,
the reflection phase crosses through zero. At higher frequencies,
the phase approaches -.pi.. The calculated model of FIG. 3 is
supported by the measured reflection phase, shown for an example
structure in FIG. 4.
A large number of structures of the type shown in FIG. 1 have been
fabricated with a wide range of resonance frequencies, including
various geometries and substrate materials. Some of the structures
were designed with overlapping capacitor plates, to increase the
capacitance and lower the frequency. The measured and calculated
resonance frequencies for twenty three structures with various
capacitance values are compared in FIG. 5. Clearly, the resonance
frequency is a predictable function of the capacitance. The dotted
line in FIG. 5 has a slope of unity, and indicates perfect
agreement. The bars indicate the instantaneous bandwidth of the
surface, defined by the frequencies where the phase is between
.pi./2 and -.pi.2.
For a more detailed description and analysis of the high-impedance
surface, see D. Sievenpiper, L. Zhang, R. Broas, N. Alexopolous, E.
Yablonovitch, "High-Impedance Electromagnetic Surfaces with a
Forbidden Frequency Band", IEEE Transactions on Microwave Theory
and Techniques, vol. 47, pp. 2059-2074, 1999 and D. Sievenpiper,
"High-Impedance Electromagnetic Surfaces", Ph.D. dissertation,
Department of Electrical Engineering, University of California, Los
Angeles, Calif., 1999.
When the resonant cavities are much smaller than the wavelength of
interest, the electromagnetic analysis can be simplified by
considering them as lumped LC circuits. The proximity of the
neighboring metal plates provides capacitance, while the conductive
path that connects them provides inductance. The textured ground
plane supports an electromagnetic boundary condition that can be
characterized by the impedance of an effective parallel LC circuit,
given by ##EQU2##
The sheet inductance is L=.mu.t, where .mu. is the magnetic
permeability of the circuit board material, and t is its thickness.
For a structure with parallel plate capacitors arranged on a square
lattice, the sheet capacitance is C=.di-elect cons.A/d, where e is
the electric permitivity of the dielectric insulator, and A and d
are the overlap area and separation, respectively, of the metal
plates.
The surface has a frequency-dependent reflection phase given by
##EQU3##
where .eta. is the impedance of free space. Far from the resonance
frequency, the surface behaves as an ordinary electric conductor,
and reflects with a .pi. phase shift.
Near the resonance frequency, the cavities interact strongly with
the incoming waves. The surface supports a finite tangential
electric field across the lattice of capacitors, and the structure
has high, yet reactive surface impedance. At resonance, it reflects
with zero phase shift, providing the effective boundary condition
of an artificial magnetic conductor. Scanning through the resonance
condition from low to high frequencies, the reflection phase varies
from .pi., to zero, to -.pi.. Thus, by tuning the resonance
frequency of the cavities, one can tune the reflection phase of the
surface for a fixed frequency.
This tunable reflection phase is the basis of the reconfigurable
beam steering reflector disclosed herein. By varying the reflection
phase as a function of position across the surface, one can perform
a variety of functions. For example, a linear phase gradient is
equivalent to a virtual tilt of the reflector. A saw-tooth phase
function transforms the surface into a virtual grating. A parabolic
phase function can focus a plane wave onto a small feed horn,
allowing the flat surface to replace a parabolic dish.
BRIEF DESCRIPTION OF THE INVENTION
Features of the present invention include: 1. A device with tunable
surface impedance; 2. A method for focusing an electromagnetic wave
using the tunable surface; and 3 . A method for steering an
electromagnetic wave using the tunable surface.
This invention provides a reconfigurable electromagnetic surface
which is capable of performing a variety of functions, such as
focusing or steering a beam. It improves upon the high-impedance
surface, which is the subject of U.S. Provisional Patent Serial No.
60/079,953, to include the important aspect of tunability.
The present invention provides, in one aspect, a tuneable impedance
surface for steering and/or focusing a radio frequency beam, the
tunable surface comprising: a ground plane; a first plurality of
top plates disposed a distance from the ground plane, the distance
being less than a wavelength of the radio frequency beam; and a
second plurality of top plates disposed a different distance from
the ground plane, the second plurality being moveable relative to
the first plurality.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 depicts a conventional high-impedance surface fabricated
using printed circuit board technology of the type disclosed in
U.S. Provisional Patent Serial No. 60/079,953 and having metal
plates on the top side connect through metal plated vias to a solid
metal ground plan on the bottom side;
FIG. 2 is a circuit equivalent of a pair of adjacent metal top
plates and associated vias;
FIG. 3 depicts the calculated reflection phase of the
high-impedance surface, obtained from the effective medium model
and shows that the phase crosses through zero at the resonance
frequency of the structure;
FIG. 4 shows that the measured reflection phase agrees well with
the calculated reflection phase;
FIG. 5 depicts the measured resonance frequency compared to the
calculated resonance frequency, using the effective circuit model
of FIG. 2, for twenty three examples of the surface shown in FIG.
1;
FIGS. 6(a) and 6(b) depict a pair of printed circuit boards, in
side elevation and plan views, one board of which is a
high-impedance surface while the second board is slidable relative
to the high-impedance surface and includes an array of conductive
plates or patches which overlap the plates or patches of the
high-impedance surface;
FIG. 7 depicts a circuit topology corresponding to FIGS. 6(a) and
6(b) showing how the change in capacitance depends on the
polarization of an incoming wave;
FIG. 8 is a somewhat more detailed version of FIG. 6(a), showing
the two boards contacting each other and showing the effect of
movement of one board relative to the other in terms of capacitance
changes;
FIG. 9 is a graph of the measured reflection phase of the
experimental structure shown in FIGS. 6(a) and (b) as a function of
frequency for ten different positions of the one board, displaced
in the direction of the applied electric field relative to the
other board;
FIG. 10 shows rotation of one board relative to the other in order
to vary the resonance frequency and thus the reflection phase, as a
function of position, of the tunable surface so that it can be used
to steer a reflected beam;
FIG. 11 is a graph of the measured reflection magnitude as a
function of incidence angle with the two boards aligned with each
other;
FIGS. 12(a) and 12(b) are graphs of the measured reflection
magnitude as a function of incidence angle with for two different
relative orientations of the two boards;
FIG. 13 demonstrates a test of the microwave grating having two
periods in which the movable board of the experimental structure
was physically divided down its center into two portions were
offset as shown in this figure;
FIGS. 14(a) and 14(b) are graphs of the measured reflection
magnitude as a function of incidence angle with for two different
relative orientations of the two boards when set up to have two
periods as shown in FIG. 13;
FIG. 15 is a graph of phase discontinuities which can occur with
movement or rotation of the one of the board relative to the other
board; and
FIG. 16 depicts two boards, one with conductive patches of a
uniform size and arrangement and the other of a uniform size but a
non-uniform arrangement.
DETAILED DESCRIPTION
The Tunable Impedance Surface
FIGS. 6(a) and 6(b) depict a tunable impedance surface in
accordance with the present invention. FIG. 6(b) is a plan view
thereof while FIG. 6(a) provides a side elevation view thereof. The
tunable impedance surface includes a pair of printed circuit boards
16, 18. The first board 16 has a lattice of conductive structures
10, 14 resembling the conventional high-impedance surface
previously described. The back of this first board has a ground
plane 12, preferably made of a thin, but solid, metal, and the
front is covered with an array of conductive plates or patches 10
preferably made of metal, which are connected to the ground plane
by conductive vias 14 preferably formed by plated metal. The
conductive patches 10 and their associated conductive vias 14 form
the conductive thumbtack-like structures. This structure can be
easily fabricated, for example, on FR4, a standard fiberglass-based
printed circuit material.
The second board 18 includes an array of conductive tuning plates
or patches 20, preferably made of metal, which are designed to
overlap the conductive patches 10 on the first board 16. The tuning
patches 20 are supported on a sheet of FR4, and are preferably
covered by an insulating layer 22 such as Kapton polyirnide. The
two boards may be pressed together with the conductive plates or
patches 10, 20 separated by the polyimide insulator, forming a
lattice of parallel plate capacitors. The confronting surfaces are
designed to slide against each other, to allow adjustment of the
overlap area between the matching sets of metal plates 10, 20, and
thus allow the capacitors to be tuned. Indeed the confronting
surfaces are preferably brought into close contact with each other
as is even better depicted in FIG. 8.
The two boards 16, 18 typically have a large number of conductive
plates or patches 10, 20 formed thereon and the figures only show a
small number of the plates or patches which would typically be
formed for clarity of representation. In the experimental
structure, which is discussed below, each board has approximately
1600 patches disposed thereon. The number of patches utilized is a
matter of design choice.
An Experimental Structure
An experimental structure has been made and tested. In the
experimental structure, the plates 10, 20 were provided by square
metal patches 10, 20 formed on both boards 16, 18 which measured
6.10 mm on each side and they were distributed on a 6.35 mm
lattice. The fixed board 16 was 6.35 mm thick, and the conducting
vias 14 were 500 .mu.m in diameter, centered on the square metal
plates 10. The movable board 18 was 1.57 mm thick, and the
polyimide insulator 22 that covered the tuning plate was 50 .mu.m
thick. Both boards measured 25.4 cm on each edge. As such each
board had an array of approximately 40 by 40 conducive patches 10,
20 thereon. To ensure uniform, intimate contact between the two
matching surfaces, a vacuum pump was attached to the back of the
fixed board. This evacuated the space between the boards by way of
the hollow openings 15 preferably provided in the vias 14 and
forced the two together.
By sliding the upper board 18 relative to the lower board 16, the
overlap area of the capacitors is changed, tuning the resonance
frequency of the small cavities on the surface. However, only
movement that is parallel to the applied electric field contributes
to a change in resonance frequency. This can be understood from the
following discussion: The resonance frequency of the cavities is
given by ##EQU4##
where C is the effective capacitance produced by a combination of
four separate capacitors C.sub.1 -C.sub.4 indicated in FIG. 6(b).
The mode that is excited in the cavities, and the circuit topology
that produces the effective capacitance, depends on the
polarization of the incoming wave. The circuit topology for two
cases is shown in FIG. 7.
For example, consider an incoming wave polarized along direction Y,
referring to FIG. 6(b) for orientation. The effective capacitance
is (C.sub.1 +C.sub.2) in series with (C.sub.3 +C.sub.4). If the top
board 18 is moved in the +Y direction, parallel to the applied
field, then C.sub.1 and C.sub.2 are increased while C.sub.3 and
C.sub.4 are decreased by the same amount, as shown in FIG. 8. Since
the motion occurs along the direction of pairs of capacitors that
are in series, the result is a net change in capacitance, and thus
a change in resonance frequency. Conversely, if the top plate 18 is
moved in the +X direction, perpendicular to the applied field, then
C.sub.2 and C.sub.4 are increased while C.sub.1 and C.sub.3 are
decreased by the same amount. Since the motion occurs along the
direction of pairs that are in parallel, there is no net change in
capacitance, and no change in resonance frequency. The maximum
effective capacitance, and thus the lowest resonance frequency,
occurs when the upper plate is centered such that capacitors that
are in series have equal value. Those skilled in the art will
appreciate that this justification, of why the square shapes work
when one set is rotated with respect to the other set, does not
limit the invention to square shaped top plates 18 and square
shaped lower plates 14. These same sort of effect is obtained if
(i) non-square shapes are used, (ii) non-uniform shapes are used
with relative translation movement and (iii) shapes based on a
polar coordinate system (like segmented rings of metal plates) are
used with rotational movement.
The resonance frequency of the high impedance surface defines the
frequency where the reflection phase crosses through zero. For a
fixed test frequency, a change in the resonance frequency of the
surface appears as a change in reflection phase. To measure the
reflection phase of the experimental structure, a network analyzer
was used and a pair of horn antennas, one for transmitting and the
other for receiving, were also used. The horns were placed next to
each other, both aimed at the tunable surface, and separated by a
sheet of microwave absorber. Microwave energy was transmitted from
one horn, reflected by the surface, and received with the other
horn, while the reflection phase was monitored for various
positions of the movable board. The use of separate transmitting
and receiving horns was used for this experiment because it
eliminates interference from internal reflections within the
antennas. The data was compared to a reference scan taken using a
flat metal surface, which is known to have a reflection phase of
.pi..
The reflection phase of the experimental structure is shown in FIG.
9 as a function of frequency for ten different positions of the
upper board, displaced in the direction of the applied electric
field. By varying the overlap area of the capacitor plates, the
resonance frequency is tuned from roughly 1.7 GHz to 3.3 GHz. The
series of scans shown corresponds to a total translation of
one-half period of the textured surface, or 3.2 mm. The tuning
range is limited by the maximum and minimum achievable capacitance,
which depend on the area of the plates, the thickness of the
insulator, and the fringing field in the surrounding medium.
Reflective Beam Steering
By varying the resonance frequency, and thus the reflection phase,
as a function of position, the tunable surface can be used to steer
a reflected beam. The simplest approach to beam steering is to
create a monotonic, preferably linear phase gradient across the
surface. For a mechanically tuned reflector, this can be
accomplished by a rotation of one printed circuit board with
respect to the other one, as shown in FIG. 10. From the discussion
set forth above, the reflection phase is only affected by
translation of the capacitor plates in the direction parallel to
the applied electric field. For a wave polarized along Y, only the
component of translation in the Y direction is relevant, and the
translation along X has no effect. For each individual capacitor
plate, a small rotation of one board relative to the other produces
a translation in Y that is roughly a linear function of X, but is
largely independent of Y. Thus, rotation generates a monotonic
phase gradient in the direction perpendicular to the applied
electric field, which is equivalent to a virtual tilt of the
surface. Only a small mechanical motion is required, since the
maximum displacement needed at the edge of the board is only
one-half of the lattice period.
To measure the beam steering properties of a tunable reflector
afforded by the previously discussed experimental structure, the
experimental structure was mounted vertically on a rotating
pedestal and the reflection magnitude was measured as a function of
incidence angle using two stationary horn antennas. Adjustment
screws placed at two corners of the surface allowed independent
control of both the relative orientation and the relative vertical
displacement of the two boards. Repeated measurements of the
reflection pattern were taken for various positions of the movable
board. The measurements described below were performed at 3.1
GHz.
With the plates 10, 20 of two boards 16, 18 of the experimental
structure aligned with each other, the surface has no phase
gradient, and the angle of reflection is equal to the angle of
incidence. The reflection magnitude as a function of incidence
angle is shown in FIG. 11. As expected from the foregoing
discussion, the reflection is strongest at 0 and 180 degrees when
the front and back surfaces of the reflector are directly facing
the horns. The lobes at other angles are due to reflections from
the rotating stage, the edges of the boards, the adjustment screws,
the walls of our anechoic chamber, and other objects. The asymmetry
in the reflection magnitude and angular profile between the front
and back sides of the pattern is due to an acrylic vacuum plate
which was attached to the back of the reflector to hold the two
printed circuit boards making up the experimental structure
together. The difference in reflection phase between the two
surfaces also contributes to this asymmetry, because it affects the
way the reflected waves interfere with other reflections from the
surroundings.
When one board of the experimental structure is rotated against the
other, the resulting phase gradient causes a normally incident wave
to be reflected at an angle given by ##EQU5##
where g is the phase gradient in radians per meter and .lambda. is
the wavelength. The reflection patterns for two different relative
orientations of the plates 10, 20 of the two boards 16,18 are shown
in FIGS. 12(a) and 12(b). FIGS. 12(a) and 12 (b) are graphs of the
measured reflection magnitude as a function of incidence angle with
for two different relative orientations of the two boards. In FIG.
12(a) the graph is for the orientation shown by FIG. 10, while FIG.
12(b) is for rotation of the upper board 18 in a direction opposite
to that shown by FIG. 10. The main lobes can be seen at angles of
about +/-8 degrees, indicating that the surface no longer reflects
in the specular direction, but rather in a direction determined by
magnitude and direction of the phase gradient. By rotating the
upper surface between these extremes, the reflection angle can be
tuned in an analog fashion. Of course, the lobe in the backward
direction still appears at 180 degrees, because the back of the
surface is untextured. It should be noted that because the
transmitting and receiving horns are stationary and mounted next to
each other, the main lobes of the reflection pattern indicate
angles at which a plane wave is reflected directly back towards its
source. This means that a normally incident plane wave would be
reflected to twice the angle measured in this experiment, and could
be steered over a range of +/-16 degrees.
Because the resonance frequency is not a linear function of the
displacement, as seen from FIG. 9, the maximum useful range of
motion is actually less than one-half period. For the results
described above, the difference in displacement between the two
edges of the structure was roughly 1 mm, or 0.01 wavelength. The
higher-frequency region is preferred between 2.5 GHz and 3.3 GHz,
where the resonance frequency is roughly a linear function of
displacement. This region also defines the bandwidth over which the
surface can effectively steer a beam.
Microwave Grating
Using a monotonic phase function, the maximum reflection angle is
achieved when the phase varies by 2.pi. across the width of the
surface. This limits the beam steering capabilities of a surface
with a width w to ##EQU6##
In order to steer to larger angles, a larger phase gradient must be
used. Since phase can only be defined modulo 2.pi., periodic
discontinuities of 2.pi. must be included in the phase function.
Such a surface can effectively be considered a grating. Generally
speaking, gratings are physical structures. In this embodiment the
present invention mimics a grating.
In order to test a microwave grating with two periods using the
experimental structure, the movable board 18 was physically divided
down its center into two portions 18a and 18b, and the two portions
were offset as shown in FIG. 13. This provided the phase
discontinuity used to produce a two-period grating, which has twice
the phase gradient as the monotonic surface previously described.
FIGS. 14(a) and 14 (b) are graphs of the measured reflection
magnitude as a function of incidence angle with for two different
relative orientations of the two boards when set up to have two
periods as shown in FIG. 13. In FIG. 14(a) the graph is for the
orientation shown by FIG. 13, while FIG. 14(b) is for rotation of
the upper board 18 in a direction opposite to that shown by FIG.
13. The maximum reflection angle now occurs at +/-19 degrees. For a
normally incident plane wave this corresponds to beam steering of
+/-38 degrees. As before, the beam could be steered to any angle
within this range by adjusting the phase gradient, while
maintaining the 2.pi. phase discontinuity. For larger angles, or
for larger surfaces, multiple discontinuities can of course be
used.
The patterns shown for this experiment exhibit scattering at other
angles. This is because rotation of the upper board of the
experimental structure does not produce a perfectly linear phase
function, as dictated by the functional dependence of the resonance
frequency on the displacement of the capacitor plates. The problem
is most severe at the phase discontinuities, as shown in FIG. 15.
With more accurate control over the resonance frequency of each
individual cavity, the pattern could be improved.
While the phase function produced by this rotational motion tends
to be nonlinear, it can be close enough to linear to produce a
well-formed beam, as seen in the data. Moreover, it may well be
possible to compensate for this non-linearity, and one way of doing
this could be to adjust the spacing of the cells C.sub.1 -C.sub.4
formed by plates 10, 20. Another approach would be to adjust the
size of the cells C.sub.1 -C.sub.4, while keeping the spacing of
the plates uniform. The main objective of this approach would be to
provide a surface in which the capacitance is decreased more slowly
near the edge on which it is being decreased the most--in other
words, to cancel the non-linearity of the phase function. One
example of a structure that could do this is shown by FIG. 16. The
plates 20 are made longer and narrower on one side, but shorter and
wider on the other side. The total capacitance is the same, and but
the side with the longer and narrower squares will be slightly less
sensitive to translation in the vertical direction. Rotation, as
represented by arrow 27, around pivot point 25 should produce a
more linear phase function than a uniform lattice would produce.
This technique could be used to make any other phase function
desired.
In the embodiments shown by the drawings the tunable impedance
surface is depicted as being planar. However, the invention is not
limited to planar tunable impedance surfaces. Indeed, those skilled
in the art will appreciate the fact that the printed circuit board
technology preferably used to provide substrates 16, 18 for the
tunable impedance surface can provide a very flexible substrate.
Thus, the tunable impedance surface can be mounted on any
convenient surface and conform to the shape of that surface.
However, a planar configuration is preferred since that should make
it easier to move board 18 relative to board 16 when the surface it
tuned. However other shapes of surfaces can easily slide one
relative to another, such as spherical surfaces having slightly
different diameters.
The top plate elements 10 and the ground or back plane element 12
are preferably formed from a metal such as copper or a copper alloy
conveniently used in printed circuit board technologies. However,
non-metallic, conductive materials may be used instead of metals
for the top plate elements 10 and/or the ground or back plane
element 12, if desired. This is also true for plates 20 formed on
board 18. The use of conductors 14 to connect the patches 10, 20 on
the two plates 16, 18 is optional, particularly if the RF waves
impinging the surface do so at a relatively high angle of
incidence. The use of conductors 14 is preferable if the RF waves
impinging the surface do so at a relatively low angle of
incidence.
Having described the invention in connection with certain
embodiments thereof, modification will now certainly suggest itself
to those skilled in the art. As such, the invention is not to be
limited to the disclosed embodiments except as required by the
appended claims.
* * * * *