U.S. patent number 6,141,541 [Application Number 09/001,762] was granted by the patent office on 2000-10-31 for method, device, phone and base station for providing envelope-following for variable envelope radio frequency signals.
This patent grant is currently assigned to Motorola, Inc.. Invention is credited to Lawrence E. Connell, Steven F. Gillig, John Grosspietsch, Andrew Merritt Khan, Pallab Midya, George Francis Opas, Robert Louis Palandech.
United States Patent |
6,141,541 |
Midya , et al. |
October 31, 2000 |
Method, device, phone and base station for providing
envelope-following for variable envelope radio frequency
signals
Abstract
A method (200) and device (100) provide an efficient linear
power amplifier that generates a variable-envelope radio frequency
RF signal. The method includes the steps of: A) using an efficient
envelope-following unit to output a supply voltage in accordance
with a variable envelope of an input baseband signal, wherein using
the efficient envelope-following unit includes: 1) using a
bandwidth-limiting mapping unit to determine a reference signal
based on the baseband signal; and 2) using an envelope-tracking
power converter to output a supply voltage, responsive to the
reference signal, to the linear RF power amplifier; B) providing an
RF input signal with amplitude and phase information to a linear RF
power amplifier; and C) using the linear RF power amplifier to
output a power-efficient amplified variable-envelope RF signal with
substantially a same amplitude and phase information as the RF
input signal.
Inventors: |
Midya; Pallab (Schaumburg,
IL), Connell; Lawrence E. (Naperville, IL), Gillig;
Steven F. (Roselle, IL), Grosspietsch; John
(Libertyville, IL), Khan; Andrew Merritt (Schaumburg,
IL), Opas; George Francis (Park Ridge, IL), Palandech;
Robert Louis (Sunrise, FL) |
Assignee: |
Motorola, Inc. (Schaumburg,
IL)
|
Family
ID: |
21697721 |
Appl.
No.: |
09/001,762 |
Filed: |
December 31, 1997 |
Current U.S.
Class: |
455/91; 455/108;
455/115.1; 455/127.1 |
Current CPC
Class: |
H03F
1/0227 (20130101); H04W 52/52 (20130101); H03F
2200/102 (20130101) |
Current International
Class: |
H03F
1/02 (20060101); H04B 7/005 (20060101); H04B
001/02 () |
Field of
Search: |
;455/126,127,91,115,108
;330/129,149 |
References Cited
[Referenced By]
U.S. Patent Documents
Other References
Leonard R. Kahn, "Single-Sideband Transmission by Envelope
Elimination and Restoration", Proceedings of the I.R.E. Jul. 1952,
pp. 803-806. .
Fred Raab and Daniel J. Rupp "Class-S High-Efficiency Amplitude
Modulator" GMRR TP93-1: RF Design vol. 17. No. 5 pp. 70-74, May
1994. .
F. H. Rabb and D. J. Rupp, "High-Efficiency Single-Sideband HF/VHF
Transmitter based upon Envelope Elimination and Restoration", Green
Mountain Radio Research Company, USA, pp. 21-25. .
PhD Thesis of Pallab Midya, 1995 at University of Illinois at
Champaign-Urbana, IL (not enclosed--to be sent upon receipt of a
copy)..
|
Primary Examiner: Bost; Dwayne D.
Assistant Examiner: Redmon; Joy
Claims
We claim:
1. A device for providing an efficient linear power amplifier that
generates a variable-envelope radio frequency RF signal,
comprising:
A) an efficient envelope-following unit, coupled to receive a
baseband signal, for outputting a supply voltage in accordance with
a variable envelope of the baseband signal, wherein the efficient
envelope-following unit includes:
1) a bandwidth-limiting mapping unit, coupled to receive the
baseband signal, for determining a reference signal; and
2) an envelope-tracking power converter, coupled to receive the
reference signal and to a power source, for outputting a supply
voltage to the linear RF power amplifier responsive to the
reference signal;
B) an RF signal generator, coupled to receive the baseband signal
and an RF carrier signal, for providing an RF input signal with
amplitude and phase information to a linear RF power amplifier;
and
C) the linear RF power amplifier, coupled to the efficient
envelope-following unit and to the RF signal generator, for
outputting a power efficient amplified variable-envelope RF signal
with substantially a same amplitude and phase information as the RF
input signal.
2. The device of claim 1 wherein the bandwidth-limiting mapping
unit utilizes a polynomial mapping function of the baseband signal
to provide an implicitly bandlimited signal.
3. The device of claim 2 wherein coefficients of the polynomial
mapping function are programmable.
4. The device of claim 2 wherein coefficients of the polynomial
mapping function are selected to maintain a substantially constant
gain for the linear RF power amplifier.
5. The device of claim 2 wherein coefficients of the polynomial
mapping function are selected to minimize the phase shift through
the linear RF power amplifier.
6. The device of claim 2 wherein coefficients of the polynomial
mapping function are selected to limit a minimum value of the
supply voltage for the linear RF power amplifier.
7. The device of claim 2 wherein coefficients of the polynomial
mapping function are selected to maximize efficiency of the linear
RF power amplifier.
8. The device of claim 1 wherein the bandwidth-limiting mapping
unit uses a polynomial mapping function of a square of the variable
envelope of the baseband signal to provide an implicitly
bandlimited signal.
9. The device of claim 1 wherein the bandwidth-limiting mapping
unit utilizes a transcendental function of the variable envelope of
the baseband signal to provide a bandlimited signal, wherein the
transcendental function is bounded and is characterized by even
symmetry.
10. The device of claim 1 wherein the linear RF power amplifier is
a class AB amplifier.
11. The device of claim 1 wherein the linear RF power amplifier is
a class B amplifier.
12. The device of claim 1 further including an envelope detector
coupled to the linear RF power amplifier output for providing a
feedback signal to the envelope tracking power converter.
13. The device of claim 1 further including:
A) a demodulator, coupled to the RF output and to the RF carrier
signal, for demodulating the amplified variable envelope RF
signal;
B) a feedback linearizer, coupled to the demodulator and to the
baseband signal, for linearizing the RF power amplifier.
14. A method for providing an efficient linear power amplifier that
generates a variable envelope radio frequency RF signal, comprising
the steps of:
A) using an efficient envelope-following unit to output a supply
voltage in accordance with a variable envelope of an input baseband
signal, wherein using the efficient envelope-following unit
includes:
1) using a bandwidth-limiting mapping unit to determine a reference
signal based on the baseband signal; and
2) using an envelope tracking power converter to output a supply
voltage, responsive to the reference signal, to the linear RF power
amplifier;
B) providing a RF input signal with amplitude and phase information
to a linear RF power amplifier; and
C) using the linear RF power amplifier to output a power efficient
amplified variable envelope RF signal with substantially a same
amplitude and phase information as the RF input signal.
15. The method of claim 14 wherein the linear RF power amplifier is
a class AB amplifier.
16. The method of claim 14 wherein the linear RF power amplifier is
a class B amplifier.
17. The method of claim 14 wherein using a linear RF power
amplifier further includes providing an envelope detector coupled
to the linear RF power amplifier output to provide a feedback
signal to the envelope tracking power converter.
18. The method of claim 14 wherein using the bandwidth-limiting
mapping unit includes utilizing a polynomial mapping function of
the baseband signal to provide an implicitly bandlimited
signal.
19. The method of claim 14 wherein using the bandwidth-limiting
mapping unit includes using a polynomial mapping function of a
square of the variable envelope of the baseband signal to provide
an implicitly bandlimited signal.
20. The method of claim 14 wherein using the bandwidth-limiting
mapping unit includes utilizing a transcendental function of the
variable envelope of the baseband signal to provide a bandlimited
signal.
21. The method of claim 20 wherein the transcendental function is
one of: a cosine function and a series consisting of powers of a
cosine function.
22. The method of claim 20 wherein the transcendental function is
approximated by an nth order polynomial, where n is an integer, and
which lacks a linear/first order term.
23. The method of claim 20 wherein the nth order polynomial is
limited to even-order terms and an offset constant.
Description
RELATED INVENTION
The present invention is related to another invention, CR00151 M,
METHOD, DEVICE, PHONE, AND BASE STATION FOR PROVIDING AN EFFICIENT
TRACKING POWER CONVERTER FOR VARIABLE SIGNALS, by Pallab Midya,
Lawrence Connell, John Grosspietsch and Ronald Gene Meyers, which
is being filed concurrently and
is also assigned to Motorola, Inc.
FIELD OF THE INVENTION
The present invention relates to power amplifiers and more
particularly to power amplifiers for variable envelope radio
frequency signals.
BACKGROUND OF THE INVENTION
Switched power converters are widely used to convert between DC
(direct current) source and load as well as to interface to slowly
moving AC (alternating current) inputs and outputs. These uses
represent one mode of operation of switched power converters. For
the purpose of improving efficiency of RF (radio frequency) power
amplifiers, a variable supply is used to power the amplifier. The
switched power converter providing the variable supply must have a
high efficiency, very low switching noise, high bandwidth and slew
rate. This represents a different mode of operation for a switched
power converter. Using conventional control schemes which have been
developed for essentially DC sources and loads, these objectives
can be met only by switching at a rate much higher than the
envelope bandwidth, resulting in lower efficiency and EMI
(electromagnetic interference) problems. A new method and device
are needed specifically for providing an efficient linear power
amplifier that generates a variable envelope radio frequency RF
signal that alleviates such problems.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a block diagram of one embodiment of a
device/phone/mobile/radio telephone/base station for using a
variable envelope RF signal in accordance with the present
invention to provide an efficient linear power amplifier.
FIG. 2 is a flow chart of one embodiment of steps of a method for
using a variable envelope RF signal in accordance with the present
invention to provide an efficient linear power amplifier.
FIG. 3 is a flow chart showing, with greater particularity, the
step from FIG. 2 of using a bandwidth-limiting mapping unit in
accordance with the present invention.
FIG. 4 is a flow chart showing, with greater particularity, the
step from FIG. 2 of using an envelope tracking power converter in
accordance with the present invention.
FIG. 5 is a graphical representation comparing the approximate
square root used for the envelope supply mapping to the ideal
square root function.
FIG. 6 is a graphical representation showing the ideal envelope
spectrum for a 25 kHz QPSK signal.
FIG. 7 is a graphical representation showing the spectrum of the
polynomial mapped function for the 25 kHz QPSK signal.
FIG. 8 shows a graphical representation of a plot of modulator
output voltage vs. input envelope voltage utilizing polynomial
shaping in accordance with the present invention.
FIG. 9 shows a graphical representation of one embodiment of supply
voltage and RF input voltage with respect to time where supply
voltage is shaped by a polynomial function in accordance with the
present invention.
FIG. 10 shows a graphic representation of a comparison of the
modulator output spectrum as known in the prior art and with
polynomial mapping in accordance with the present invention.
FIG. 11 shows a graphic representation of a comparison of the RF
power amplifier output spectrum as known in the prior art and with
polynomial mapping in accordance with the present invention.
FIG. 12 shows a graphical representation of the envelope to supply
mapping incorporating a voltage limit for a boost converter and
constant gain for the RF power amplifier MHW920.
FIG. 13 shows a graphical representation of the IM
(intermodulation) performance of the MHW913 for a 10 kHz two-tone
signal with envelope-following in accordance with the present
invention.
FIG. 14 shows a graphical representation of the IM
(intermodulation) performance of the MHW913 for a 10 kHz two-tone
signal with envelope-following with a fixed supply.
DETAILED DESCRIPTION OF A PREFERRED EMBODIMENT
FIG. 1, numeral 100, is a block diagram of one embodiment of a
device/phone/mobile/radio telephone/base station for using a
variable envelope RF signal in accordance with the present
invention to provide an efficient linear RF power amplifier (106).
An efficient envelope-following unit (102) is coupled to receive a
baseband signal and outputs a supply voltage in accordance with a
variable envelope of the baseband signal. The efficient
envelope-following unit (102) includes: 1) a bandwidth-limiting
mapping unit (108), coupled to receive the baseband signal, for
determining a reference signal; and 2) an envelope-tracking power
converter (110), coupled to receive the reference signal and to a
power source, for outputting a supply voltage to the linear RF
power amplifier (106) responsive to the reference signal. An RF
signal generator (104) is coupled to receive the baseband signal
and an RF carrier signal and provides an RF input signal with
amplitude and phase information to the linear RF power amplifier
(106). The linear RF power amplifier (106) is coupled to the
efficient envelope-following unit (102) and to the RF signal
generator (104) and outputs a power efficient amplified variable
envelope RF signal with substantially a same amplitude and phase
information as the RF input signal.
The bandwidth-limiting mapping unit (108) typically utilizes a
polynomial mapping function of the baseband signal to provide an
implicitly bandlimited signal. Coefficients of the polynomial
mapping function are generally selected to at least: maintain a
substantially constant gain for the linear RF power amplifier
(106), to reduce the phase shift imposed by the linear RF power
amplifier (106), to limit a minimum value of the supply voltage for
the linear RF power amplifier (106), or to maximize efficiency of
the linear RF power amplifier (106).
The bandwidth-limiting mapping unit (108) may use a polynomial
mapping function of a square of the variable envelope of the
baseband signal to provide an implicitly bandlimited signal.
Alternatively, the bandwidth-limiting mapping unit (108) may
utilize a transcendental function of the variable envelope of the
baseband signal to provide a bandlimited signal--the function
characterized by even mathematical symmetry.
In one embodiment, a demodulator (126) may be coupled to the RF
output and to the RF carrier signal, for demodulating the amplified
variable envelope RF signal, and a feedback linearizer (124) may be
coupled to the demodulator (126) and to the baseband signal for
linearizing the RF power amplifier (106).
The envelope tracking power converter (110) typically includes a
feedforward feedback control unit (112), a pulse width modulation
unit (114), and a power converter (116). The feedforward feedback
control unit (112) is coupled to the bandwidth-limiting mapping
unit (108) and, where selected, to receive at least one feedback
signal. The feedforward feedback control unit (112) determines an
optimal control signal. The pulse width modulation unit (114) is
coupled to the feedforward feedback control unit (112) and is used
for modifying a pulse width to provide a constant frequency
switching signal. The power converter (116) is coupled to the pulse
width modulation unit (114) and to the power source (120) and
provides a variable output signal to the linear RF power amplifier
(106).
Generally, the linear RF power amplifier (106) is a class AB
amplifier or a class B amplifier.
Where selected, at least one feedback signal is sent to the
feedforward feedback control unit (112). The power converter (116)
may provide a feedback signal and/or an envelope detector (118)
coupled to the linear RF power amplifier (106) output may provide a
feedback signal to the feedforward feedback control unit (112). The
bandwidth-limiting mapping unit (108) may use a polynomial mapping
function of I.sup.2 plus Q.sup.2, which is a square of the envelope
of the baseband signal to provide an implicitly bandlimited
signal.
The device (122) is typically implemented in a cellular phone, a
satellite phone, a mobile radio, a radio telephone, a base station
or the like.
FIG. 2, numeral 200, is a flow chart of one embodiment of steps of
a method for using a variable envelope RF signal in accordance with
the present invention to provide an efficient linear power
amplifier. The method includes the steps of: A) using (202) an
efficient envelope-following unit to output a supply voltage in
accordance with a variable envelope of an input baseband signal,
wherein using the efficient envelope-following unit includes: 1)
using a bandwidth-limiting mapping unit to determine a reference
signal based on the baseband signal; and 2) using an envelope
tracking power converter to output a supply voltage, responsive to
the reference signal, to the linear RF power amplifier; B)
providing (204) a RF input signal with amplitude and phase
information to a linear RF power amplifier; and C) using (206) the
linear RF power amplifier to output a power efficient amplified
variable envelope RF signal with substantially a same amplitude and
phase information as the RF input signal.
Where selected, using a linear RF power amplifier further may
include providing (208) an envelope detector coupled to the linear
RF power amplifier output to provide a feedback signal to the
envelope tracking power converter.
Again, the linear RF power amplifier is typically a class AB
amplifier or a class B amplifier.
As shown in FIG. 3, numeral 300, the step from FIG. 2 of using a
bandwidth-limiting mapping unit in accordance with the present
invention may include one of: utilizing (302) a polynomial mapping
function of the baseband signal to provide an implicitly
bandlimited signal, using (304) a polynomial mapping function of a
square of the variable envelope of the baseband signal to provide
an implicitly bandlimited signal or utilizing (306) a
mathematically even transcendental function of the variable
envelope of the baseband signal to provide a bandlimited
signal.
As shown in FIG. 4, numeral 400, the step from FIG. 2 of using an
envelope tracking power converter in accordance with the present
invention typically includes: A) utilizing (402) a feedforward
feedback control unit, where the feedforward feedback control unit
is coupled to the bandwidth-limiting mapping unit and, where
selected, to receive at least one feedback signal, for determining
an optimal control signal in accordance with a predetermined
scheme; B) utilizing (404) a pulse width modulation unit, where the
pulse width modulation unit is coupled to the feedforward feedback
control unit, for modifying a pulse width to provide a constant
frequency switching signal; and C) utilizing (406) a power
converter, where the power converter is coupled to the pulse width
modulation unit and to the power source, for providing the supply
voltage to the linear RF power amplifier. Generally, the linear RF
power amplifier is a class AB amplifier or a class B amplifier.
A feedback signal may be provided to the feedforward feedback
control unit by the power converter, or alternatively, an envelope
detector coupled to the linear RF power amplifier output may
provide a feedback signal to the feedforward feedback control unit.
In another embodiment both feedback signals may be utilized.
In a preferred embodiment, the step of utilizing the
bandwidth-limiting mapping unit generally includes using a
polynomial mapping function of I.sup.2 plus Q.sup.2, which is a
square of an envelope of the baseband signal to provide an
implicitly bandlimited signal.
Envelope-following is characterized by a variable supply to the RF
power amplifier, and an RF input signal that is unmodified and has
phase as well as amplitude information. Since the supply does not
directly determine the output amplitude, the supply voltage can be
chosen to optimize power efficiency and/or RF distortion
performance.
To achieve improved spectral performance while maintaining
efficiency, a mapping function or transformation is implemented.
The function is derived by considering the complex composite input
signal in Cartesian format:
where I(t) and Q(t) are bandlimited baseband quadrature signals.
The envelope of V.sub.i (t) is given by: ##EQU1## Since I(t) and
Q(t) are bandlimited, I.sup.2 (t) and Q.sup.2 (t) are also
bandlimited. But, due to the square root operation associated with
envelope generation, the function v.sub.e (t) has an extremely wide
spectrum in general. Even-ordered polynomial functions of the
variable v.sub.e (t)--which generate bandlimited powers of (I.sup.2
(t)+Q.sup.2 (t))--may be applied to the envelope function to reduce
the spectral expansion effects of the square root operation.
Odd-order terms of order greater than one also provide improvement
over the pure envelope signal, but not to the extent that the
even-order terms do, and consequently produce a wider spectrum than
even-order terms. In addition to the spectrum occupancy issues, the
calculation of the square root function is computationally
intensive. Therefore, alternate methods to create the envelope
signal are required.
In FIG. 5, numeral 500, the approximate square root (504) used for
the envelope supply mapping is compared to the ideal square root
function (502). The following section shows an analysis of a
polynomial mapping function that maps the ideal square root
function to the approximate square root
Define the supply to the RF PA to be y. Computing a square root is
a computationally intensive operation. Thus, the supply may be
computed as a function of the envelope squared.
Consider a fourth order polynomial mapping function of the envelope
squared. Since the baseband (I,Q) signals are band limited to
F.sub.channel the supply bandwidth in this case would be limited to
8 (F.sub.channel).
The error in the supply compared to the ideal square-root is as
follows: The average error squared is defined to be the cost
function J. The weighting function is chosen to be the power of the
error signal since the out of band power is to be minimized.
This cost function is minimized by choosing the appropriate
polynomial coefficients. ##EQU2## Though the base band signals
(I,Q) are time varying, the associated statistics are not time
varying. In other words the condition of ergodicity is imposed on
this analysis. Under this condition, a large sample of the data may
be used to predict all statistical information about the process. A
large section of the data is typically used to optimize the
polynomial coefficients. ##EQU3## Differentiating under the
integral sign, the following equations are obtained: ##EQU4## The
higher order terms that follow may be computed similarly. The
baseband data (I,Q) is usually available in sampled format. Thus,
the integrals are reduced to summation. Using a large number of
discrete data points, the values for the polynomial coefficients
are obtained by solving a set of simultaneous equations. Though
this analysis is being done here for a fourth order polynomial of
the envelope squared, the method extends to a polynomial of
arbitrary order. The results may be combined as follows. ##EQU5##
The values for the elements of the matrix C are as follows.
The values of the elements of the vector D are as follows.
After the polynomial coefficients are computed, the supply voltage
is computed in terms of the envelope squared. The number of
addition and
multiplication operations may further be reduced by factoring the
polynomial. However, it is possible for some of the polynomial
coefficients to be complex.
The example case here is a 25 ksps QPSK system where the baseband
data is sampled at 400 ksps. The amplitudes are normalized from
zero to unity. In this case, the roots of the polynomial are
1.5035, 0.4931+0.7802i, 0.4931-0.7802i and -0.0272. The computation
of the envelope to supply mapping is as follows.
The polynomial approximates the square root very closely. However,
the small difference between the two signals results in a large
difference in the bandwidth of the signals. This difference is
shown clearly in the spectral representations of the ideal input
envelope and the polynomial-mapped envelope (602) in FIG. 6,
numeral 600 and FIG. 7, numeral 700, respectively. The signal used
here is a 25 ksps QPSK signal used in a satellite subscriber phone
application.
Here the error power has been minimized. Other functions may be
minimized using the same algorithm. Obviously this would result in
different polynomial coefficients. However, the algorithm taught
here may be used without any changes. It is interesting to note
that the linear term proportional to the envelope has been
completely eliminated. This is a significant departure from all
prior art that uses the linear term only for supply and bias
modulation.
A practical application of this technique is illustrated for a base
station power amplifier where improved power efficiency through the
application of supply modulation is desired but another important
requirement is the maintenance of very low levels of transmitter
noise in adjacent channels that are at relatively wide offsets from
the carrier frequency. The basic operational requirement in
achieving low noise levels while employing supply modulation is
avoidance of a wide spectrum in the supply modulation signal. The
reason for this is the imperfect multiplier action provided by a
typical real-world RF power amplifier when its supply is
modulated--whether the amplifier is operated in an EER mode with a
phase-modulated constant-envelope RF input or in an
envelope-following class-AB mode with a full composite input having
both amplitude and phase variations. This leads to a mapping of the
input envelope signal as has been previously discussed in order to
reverse the spectral expansion created by the envelope generation
or detection process. The mapping may take place in a circuit,
software or a look-up table.
A polynomial series function of powers of the envelope squared--or
alternately, even-ordered powers of the envelope itself--will be
used for the mapping. As has been previously shown, a fourth-order
polynomial (envelope to the eighth power) can be made to closely
fit the square root response of an ideal envelope detector. There
are tradeoffs that can be made, however, in the number of terms in
the series and how close the fit is in specific portions of the
response. It will be assumed in this example that a linear response
over a significant portion of the dynamic range of the input
envelope is preferred because it provides an essentially constant
level of gain compression in this region. If a high-order
polynomial were employed, a very close fit could be achieved down
to nearly zero input and output. If this were done, however, the
baseband spectrum of the mapped envelope would be n/2 times wider
than the spectrum of the envelope squared, where n is even and also
the order of the polynomial. An excessively wide spectrum generated
by the mapping would have negative implications for the required
bandwidth and slew-rate of the modulator's power converter as well
as the bandwidth of any type of linearization system that might be
employed concurrently with modulator. A significant reduction in
the required order of the polynomial consistent with adequate
tracking of the linear response--along with a corresponding
reduction in the mapped spectrum--can be achieved by recognizing
where close tracking of the linear response can be relaxed.
The most critical portion of the envelope signal with respect to
power efficiency is approximately the upper three quarters of the
total envelope dynamic range from 0 volts to peak supply. At one
quarter of the peak supply, for example, a maximum of only 6.25% of
the peak power is supplied to the PA. Averaged over the statistics
of typical modulating signals, power inefficiencies at these levels
impose little impact on the overall PA efficiency. The polynomial
can therefore deviate from the ideal linear mapping in the lower
quarter of the dynamic range to facilitate the representation. In
the proposed polynomial function, a linear or first-order term
would represent a direct translation of the wide envelope spectrum
to the mapped envelope and is therefore excluded. Coefficients of
higher order terms are selected to cause the function to track a
linear response above a threshold level (e.g., one quarter of the
peak supply) for optimum efficiency and RF distortion, although a
limited number of even-order terms is preferred for reasons already
stated.
To continue the illustrative example, the envelope is mapped
through a sixth-order polynomial of the form:
where k represents the maximum negative excursion from the peak
supply level of 1.0 and a.sub.2 through a.sub.6 are coefficients of
the polynomial. The coefficients can be determined in a simplified
quasi-empirical manner by setting three criteria for matching the
classic linear response as follows:
where x is an intermediate value within the dynamic range of
v.sub.e (t) (0.0 to 1.0) that will provide an acceptable fit of the
polynomial to the linear transfer function. These criteria result
in the following three simultaneous equations which permit solution
for the three coefficients:
An example is shown in FIG. 8, numeral 800, where v.sub.m (t) is
plotted vs. v.sub.e (t) for k=0.8 and x=0.68. Above a normalized
input envelope voltage of approximately 0.32 volts, the polynomial
response (802) tracks the classic linear envelope modulator
response (804) very closely, thereby providing the same efficiency
performance and supply-induced RF distortion effects as the linear
response in this region. Below 0.32 volts, the polynomial response
(802) provides higher voltage to the power amplifier compared to
the linear response (804). Although there is an efficiency
degradation in this region, the additional power consumed is very
small compared to the total power consumed by the power amplifier
over the modulation's full dynamic range. As a result, the
efficiency of a power amplifier modulated by the transformed
envelope will degrade very little compared to modulation through
the classic linear modulator response.
Although a sixth-order polynomial realization of the transfer
function is described, it is clear that the approach is general.
For example, a fourth-order response could be employed, although
the it would have greater degradation in tracking accuracy than the
sixth-order response and a consequent degradation in efficiency and
RF distortion. Certain bounded transcendental functions
characterized by even symmetry with respect to their
argument--e.g., cosine, hyperbolic secant functions, certain
elliptic and Bessel functions, and series' of powers of these
functions--could also be employed. Even symmetry is a requirement
because it corresponds to absence of a linear term. Ultimately, all
of these approaches to the description of the mapping function
could be expressed as an even-order polynomial series. Although not
as desirable because of the additional spectrum introduced,
odd-order terms could also be beneficially employed in the
polynomial. Clearly, the first-order or linear term--which is
commonly depicted in the prior art-is not used in the mapping
function since it provides a direct translation of the wide
envelope spectrum to the power amplifier modulation signal.
In a preferred embodiment, the polynomial form of the envelope
modulation mapping function is characterized by:
1) the function lacks a linear or first-order term;
2) the function contains only even-order terms and an offset
constant; and
3) the order of the function is minimal, but is selected to provide
a predetermined level of efficiency and RF distortion.
FIG. 9, numeral 900, shows a graphical representation of one
embodiment where the envelope of the RF input voltage (902) for a
two-tone signal and the resulting supply voltage (904) with respect
to time where supply voltage is mapped by a sixth-order polynomial
function in accordance with the present invention.
FIG. 10, numeral 1000, shows the dramatic reduction in the
frequency content of a conventional two-tone envelope (1002) as
compared to the same envelope processed through polynomial-mapping
(1004) in accordance with the present invention.
FIG. 11, numeral 1100, shows the corresponding spectra at the RF
carrier frequency for a computer simulation made using a Motorola
MRF839 RF power device model operating in a class-AB
envelope-following mode. Application of a conventional two-tone
envelope to the supply terminal of circuit results in a very wide
spectrum at the carrier frequency (1102). The collector efficiency
for this case was 69.2%. There is a very pronounced reduction in
wide-offset products beyond .+-.30 kHz (1104) when the same
envelope signal is processed through polynomial-mapping in
accordance with the present invention. The collector efficiency in
this case decreases slightly to 67.5%. This example illustrates
that modulation mapping provides supply modulation with a favorable
off-channel spectrum while imposing minimal degradation in
efficiency. While the mapping results in significant improvements
in wide-offset spectrum, the closer-in products are essentially
unaffected. However, conventional linearization techniques such as
feedback or predistortion can be implemented to reduce these
products as well. Envelope mapping provides an additional benefit
in this scenario as well by reducing the bandwidth of the PA
distortion products and thereby reducing the bandwidth requirements
of the linearization system. Yet another benefit of modulation
mapping including a departure from tracking at low inputs is a
reduction not only in the modulator bandwidth but also in slew-rate
requirements.
The example above which illustrates the viability of departure from
tracking of the envelope response down to zero volts has benefits
beyond reduction of wide-offset emissions and reduction of
modulator bandwidth and slew-rate requirements. Certain classes of
power converters are limited in the range of output voltages that
they can produce. For example, a boost converter can produce
voltages greater than the DC input voltage. Therefore, the envelope
to supply mapping used with this type of converter must have a
lower limit equal to the input voltage. If a hard lower limit is
imposed on the mapped outputs, abrupt transitions are created in
the envelope to supply mapping output. This results in high
frequency components that are undesirable.
This problem may be resolved by using polynomial mapping as
follows. An idealized linear mapping of envelope to supply is
chosen. The lower bound is then introduced into this mapping
function. This results in a piecewise linear function. A
least-squares estimator can be used to approximate the piecewise
linear function by an even-order polynomial of the baseband
envelope. Thus we obtain a polynomial of (I.sup.2 +Q.sup.2).
This method allows the use of boost converters for
envelope-following applications. There has been a migration of
cellular phones and other portable communication systems to lower
voltage batteries. In the future further lowering of this voltage
is anticipated as a single cell battery system is approached. In
this context, the ability to use a simple boost converter for
envelope-following is of considerable advantage. The RF power
amplifier could then be operated at higher supply voltage than the
battery. This would allow the use of higher voltage RF power
amplifiers that have lower cost and higher efficiency and which are
readily available at this time.
The linearity of a linear RF power amplifier may be characterized
by its gain and phase variation in response to amplitude
variations. Thus, if an amplifier has little gain and phase
variation in response to the entire range of amplitudes applied, it
has high linearity. This results in a spectrum that produces
minimal coupled power (or splatter) into the adjacent channels. In
most radio systems, there are strict limits on the amount of
splatter interfering with the adjacent channel, resulting in strict
limits on the gain and phase variations allowed in a RF power
amplifier.
For any RF power amplifier, linearity is a critical issue. Once the
modulation format is chosen, the way to improve linearity is by
improving the RF power amplifier. Alternate methods to improve
linearity include predistortion and feedback linearization.
Unfortunately, all these methods increase power consumption and
lower efficiency. In the context of an envelope-following system,
the choice of the envelope-to-supply mapping can affect the amount
of gain and phase variation. For example, the phase variation for
typical RF power amplifiers increases dramatically if the supply
voltage approaches zero. Unfortunately, this is the case in many
modulation schemes which utilize the envelope of the RF signal to
generate a proportional supply modulation signal . . . . If the
supply is simply proportional to the envelope as is typical of
prior art, large phase variations through the RF power amplifier
will result. The consequences include increased nonlinearity and
adjacent-channel splatter. To avoid this scenario, zero avoidance
techniques have been developed that prevent the RF envelope from
approaching zero. This solves the linearity problem at the cost of
significant increase in system complexity.
The improved envelope-to-supply mapping as described above for a
boost converter can accomplish zero avoidance without changing the
envelope of the RF signal. Here, the envelope of the RF signal is
allowed to go to zero, but the supply is prevented from following
it. This is effectively a zero avoidance strategy for
envelope-following systems. As was shown in the base station
example, there is negligible difference in power consumption by the
RF power amplifier associated with the zero avoidance. Thus, the
linearity improvement obtained is not at the cost of power
consumption.
It is also possible to limit gain variations by using other
appropriate envelope to supply mappings. The gain of the RF power
amplifier is a function of the input RF signal envelope and the
supply voltage. Consider a Cartesian coordinate system with the
signal envelope and the supply voltage as the axes. In this plane a
locus of points is plotted for constant gain. This plot can be
interpreted as a piecewise linear function for the supply to
envelope mapping. Again, this piecewise linear function may be
approximated as a polynomial of the envelope squared.
In FIG. 12, numeral 1200, the envelope-to-supply mapping (1202)
incorporating both a voltage limit to facilitate a boost converter
and constant gain for the RF power amplifier MHW920 is shown
graphically. The mapping is obtained as a curve-fit to constant
gain data points (1204). This is an RF power amplifier designed to
operate at a nominal supply of 6 V. A boost converter using the
envelope-to-supply mapping shown may be used to operate this power
amplifier from a 3.6 V battery. This would result in significant
power savings while allowing the migration to a lower voltage
battery. Here, a low-order polynomial has been chosen for the
mapping further limiting the supply bandwidth requirements.
The preferred embodiment of the polynomial mapping function is in
digital hardware. In this context, the mapping function is easily
changed by changing the polynomial coefficients. For a given RF
power amplifier and modulation scheme there exists a specific
polynomial that would result in the most linear operation. It would
be advantageous to operate the radio with the polynomial
coefficients for this specific polynomial.
The coefficients are programmable and can be tuned manually at the
factory as part of a procedure called phasing of the radio. This is
a common practice for tuning certain critical features in a radio.
Also, the
polynomial coefficients can be programmed in the field to
compensate for changes in the RF power amplifier characteristics
with temperature. This could be extended to tuning the polynomials
for the best linearity.
If there is significant variation in the radio due to aging,
temperature, load mismatch and other parameter variations, it may
be necessary to adjust the polynomial coefficients dynamically. A
gradient algorithm is suggested here for this tuning process.
One advantage of dynamic tuning is minimization of power
consumption. The current practice is to design RF power amplifiers
with linearity well above specification at nominal conditions such
that linearity specification is met in the worst case situation.
This results in higher power consumption at nominal conditions. Any
dynamic tuning of the supply voltage would allow optimization of
both linearity and power consumption over the entire range of
operating conditions.
In the prior art, the system may introduce a feedback linearization
method to linearize a RF power amplifier to meet stringent splatter
requirements. In these radios it would be advantageous to employ
supply modulation to improve efficiency. However, the bandwidth of
the linearization scheme is limited. Any disturbance introduced
above the bandwidth of the linearization loop cannot be corrected.
The use of the polynomial mapping function strictly limits the
bandwidth of the supply voltage. The order of the polynomial is
chosen such that the supply bandwidth is less than the loop
bandwidth of the linearization scheme.
Predistortion is another method for linearization of RF power
amplifiers. Predistortion may be combined with envelope-following
for systems requiring high efficiency and high linearity. The
ability of predistortion to correct for wide-band distortion is
limited. This limitation arises due to sampling associated with the
digital implementation of the predistortion process. Sampling
fundamentally limits any response above the Nyquist rate, which is
half the sampling frequency. Thus, any spectral content in the
supply above half the sampling rate of the predistortion scheme
would not be corrected. Again an appropriate choice of the order of
the polynomial mapping function limits the bandwidth of the supply
voltage and maintains the linearization obtained from the
predistortion scheme.
The mapping relation between the envelope and the supply voltage is
determined by efficiency and linearity criteria. Two kinds of
nonlinearities in RF power amplifiers are described as
amplitude-to-amplitude distortion (AM-to-AM) and amplitude-to-phase
distortion (AM-to-PM). The latter is increased when the PA supply
is allowed to approach zero volts. This often defines the allowable
lower limit of the supply variation.
AM to AM distortion can be reduced by introducing an appropriate
amount of envelope signal in a manner consistent with improving
efficiency. The power amplifier is operated at a constant
saturation in the upper range of envelope amplitudes. This
determines a portion of the mapping between the envelope and the
supply voltage which is approximately linear. For lower envelope
amplitude levels, the supply voltage is held constant. The
combination of the two mappings create a piece-wise linear
relationship between the envelope and supply voltage. This may be
approximated such that the supply voltage is a polynomial function
of the envelope similar to the base station modulator and boost
converter implementations.
Other implementations of envelope-following are possible, employing
envelope detection of the RF input as opposed to using a baseband
input. However, the approach described above allows the precise
measurement of the envelope from the input data. This results in a
system whose linearity is not compromised by imperfect detection of
the envelope signal. Further, it also allows compensation for the
time delay in the switcher.
Envelope-following is expected to affect the linearity of the RF
power amplifier. The data in the following tables, which is well
within acceptable limits, was taken to illustrate that there is
little degradation in signal quality. TABLE 1, shown below, lists
the adjacent- and alternate-channel power levels with and without
supply modulation. Note that this data was taken with a switching
converter providing the supply modulation. The supply modulation
was derived from the detected RF input signal, and this introduced
a time delay of 2-4 .mu.s between the supply and the envelope. This
much delay in an EER (envelope elimination and restoration) system
would be quite unacceptable. This illustrates the robustness of the
envelope-following approach:
TABLE 1 ______________________________________ ADJACENT-CHANNEL
COUPLED POWER DATA Envelope-Following Fixed Supply Condi- Upper
Lower Upper Lower tion ______________________________________
adjacent- -31.6 dB -32.5 dB -34.7 dB -33.5 dB Modula- channel tion
power Only alternate- -47.5 dB -49.1 dB -50.5 dB -50.9 dB channel
power adjacent- -31.6 dB -32.5 dB -34.7 dB -33.5 dB Modula- channel
tion & power Transient alternate- -47.5 dB -48.4 dB -50.0 dB
-50.0 dB channel power adjacent- -27.4 dB -27.4 dB -31.6 dB -31.3
dB Worst- channel Case power Load alternate- -48.6 dB -43.8 dB
-47.7 dB -48.2 dB Mismatch channel power
______________________________________
There was also some concern that envelope-following would affect
the demodulated signal. The experimental data in TABLE 2 (shown
below) indicates that there is little signal degradation. It was
possible to use a very aggressive envelope that maximized the power
savings, and the in band signal degradation was still
insignificant. However, this degraded adjacent-channel splatter
which may be unacceptable in certain radio systems.
TABLE 2 ______________________________________ ERROR VECTOR
MAGNITUDE DATA FOR 25 ksps OFFSET QPSK SYSTEM Envelope Following
Fixed Supply EVM Mag Phase EVM Mag Phase Condition (rms) error err.
(rms) error err. ______________________________________ Steady
State 4.0% 3.0% 1.3 deg. 2.3% 1.9% 1.0 deg. First 10 symbols 4.8%
3.5% 1.8 deg. 4.0% 3.1% 1.4 deg. of burst Aggressive 4.3% 3.0% 1.5
deg. Envelope-Following ______________________________________
Another envelope-following circuit was built using the Motorola
MHW913 LDMOS RF module that was capable of handling higher power
(.about.10 W RF average output power compared to .about.1 W for the
MHW920 bipolar RF module). FIG. 13, numeral 1300, shows a graphical
representation of the third (1302) and fifth-order (1304) IM
(intermodulation) performance of the MHW913 for a 10 kHz-spaced
two-tone signal using envelope-following in accordance with the
present invention. This data compares favorably with FIG. 14,
numeral 1400, which shows a graphical representation of the IM
performance of the third (1402) and fifth-order(1404) IM of the
MHW913 with a fixed supply. Note that envelope-following resulted
in worse IM at lower power levels, but the IM was not significantly
affected at the highest power level. Thus, the worst-case IM is not
affected, and for a given IM specification the envelope-following
system performs just as well as the system without supply
modulation. These results indicate that the envelope-following
scheme is applicable over a range of output powers and for
different semiconductor technologies used in RF power
amplifiers.
In one embodiment, the device of the present invention may be
utilized in a multi-mode radio-telephone having a system for
providing an efficient linear power amplifier that generates a
variable-envelope RF signal among other possible signals. The
system includes: A) an efficient envelope-following unit, coupled
to receive a baseband signal, for outputting a supply voltage in
accordance with either the variable envelope of the baseband signal
or a constant envelope; B) an RF signal generator, coupled to
receive the baseband signal and an RF carrier signal, for providing
an RF input signal with amplitude and phase information to a linear
RF power amplifier; and C) the linear RF power amplifier, coupled
to the efficient envelope-following unit and to the RF signal
generator, for outputting either: a power-efficient amplified
variable-envelope RF signal with substantially the same amplitude
and phase information as the RF input signal based on the variable
envelope of the baseband signal or a power-efficient amplified
constant-envelope RF signal with substantially the same phase
information as the constant-envelope RF input signal. Clearly, this
embodiment of the invention provides a flexibility for utilizing
the multi-mode radio telephone in any one of a plurality of modes,
depending on the choice of the user. The choice may be based on
availability of a particular mode, or alternatively, on a provider
rate differential in selection of modes.
The present invention may be embodied in other specific forms
without departing from its spirit or essential characteristics. The
described embodiments are to be considered in all respects only as
illustrative and not restrictive. The scope of the invention is,
therefore, indicated by the appended claims rather than by the
foregoing description. All changes which come within the meaning
and range of equivalency of the claims are to be embraced within
their scope.
* * * * *