U.S. patent number 5,844,949 [Application Number 08/728,020] was granted by the patent office on 1998-12-01 for power line communication system.
This patent grant is currently assigned to General Electric Company. Invention is credited to Richard Charles Gaus, Jr., John Erik Hershey, Richard August Korkosz, Gary Jude Saulnier, Kenneth Brakeley Welles, II.
United States Patent |
5,844,949 |
Hershey , et al. |
December 1, 1998 |
Power line communication system
Abstract
A system for communicating digital information over wires having
a great deal of harmonic distortion, such as a power line, employs
a transmitter which transmitter which creates a carrier wave for
each of a plurality of signals to be sent. This carrier wave has
frequency lobes positioned between the frequency lobes of the
harmonic distortion. Each of the lobes of a single carrier signal
is encoded with the same bit value during a given bit period. This
signal is then mixed with any existing signal on the wire. At a
remote receiver coupled to the wires, the signal is sensed,
filtered, and Fourier transformed into coefficients. The
signal-to-noise (S/N) ratio of each Fourier coefficient is
determined by a novel S/N estimation technique. The coefficients
are weighted based upon the S/N ratio estimation, and decoded,
preferably by an inner product of the weighted Fourier
coefficients. Additionally, the S/N ratio estimates could be time
averaged before being used in the weighting and bit decoding.
Inventors: |
Hershey; John Erik (Ballston
Lake, NY), Korkosz; Richard August (Rotterdam Junction,
NY), Saulnier; Gary Jude (Rexford, NY), Gaus, Jr.;
Richard Charles (Burnt Hills, NY), Welles, II; Kenneth
Brakeley (Scotia, NY) |
Assignee: |
General Electric Company
(Schenectady, NY)
|
Family
ID: |
26147676 |
Appl.
No.: |
08/728,020 |
Filed: |
October 9, 1996 |
Current U.S.
Class: |
375/346; 375/259;
370/482; 340/310.12; 340/12.33 |
Current CPC
Class: |
H04L
5/06 (20130101); H04L 1/04 (20130101); H04B
3/54 (20130101); H04B 2203/5416 (20130101); H04B
2203/5495 (20130101) |
Current International
Class: |
H04B
3/54 (20060101); H04L 5/06 (20060101); H04L
1/02 (20060101); H04L 5/02 (20060101); H04L
1/04 (20060101); H04B 001/10 (); H04L 025/08 ();
H03D 001/04 () |
Field of
Search: |
;340/310.02,310.03,310.01 ;375/260,200,205,296,295,259 ;385/27 |
References Cited
[Referenced By]
U.S. Patent Documents
Other References
"Multitone Signals With Low Crest Factor" by S. Boyd, IEEE Trans.
on Circuits and Systems, vol. CAS-33, No. 10, pp. 1018-1022. .
"Substation Noise at Distribution Line Communications Frequencies",
J. B. O'Neil, Jr., IEEE Trans. on Electromagnetic Compatibility,
vol. 30, No. 1, Feb. 1988..
|
Primary Examiner: Chin; Stephen
Assistant Examiner: Park; Albert
Attorney, Agent or Firm: Zale; Lawrence P. Snyder;
Marvin
Claims
What we claim is:
1. A system for communicating digital information over a wire
having a harmonic interference frequency lobes comprising:
a) a transmitter for transmitting message bits of a signal s.sub.1
having:
i. a carrier wave synthesis device for creating a carrier wave
having a plurality of frequency lobes between interference lobes
described by: ##EQU4## where t is time, f.sub.ik are frequencies
between said harmonic interference lobes, M is the number of signal
frequency lobes desired, .alpha..sub.k, .beta..sub.k, are
predetermined signal amplitudes, and .phi.k,.theta.k are
predetermined phase offsets;
ii. a bit encoder coupled to a timing device, the synthesis device,
and said wire, which inverts a carrier waveform for a single bit
period for one bit value of said message bit, and leaves the
carrier wave unchanged for a second message bit value to create an
encoded message;
iii. a summation device coupled to the bit encoder and said wire,
for summing the encoded message with existing signals on said
wire;
b) a receiver coupled to said wire which decodes the encoded
message into message bits.
2. The system for communicating digital information over a wire of
claim 1 wherein the carrier wave synthesis device comprises:
a) a timing device to determine bit periods, and bit rate
timing;
b) a storage device having a prestored carrier waveform;
c) a playback device coupled to the timing device and the storage
device for playing back the prestored waveform according to bit
rate timing.
3. The system for communicating digital information over a wire of
claim 1 wherein the receiver comprises:
a) a sampler which converts a continuous signal into a series of
discrete time samples;
b) a quantizer coupled to the sampler which reduces the precision
of the discrete time samples;
c) a Fourier Transform device which performs a Fourier transform on
the samples obtained at time t to result in Fourier coefficients
X.sub.k (t);
d) a decoder which performs an inner product of the Fourier
coefficients X.sub.k (t), and determine if the inner product is
above or below a predetermined threshold to produce decoded message
bits.
4. The system for communicating digital information over a wire of
claim 3 further comprising:
a) a signal-to-noise (S/N) estimation device which estimates S/N
based upon a reliability indication estimate
.vertline..delta..sub.k .vertline. for each of the Fourier
coefficients;
b) a combiner which weights each of the Fourier coefficients
X.sub.k (t) according to .vertline..delta..sub.k .vertline. to
result in weighted Fourier coefficients X.sub.k (t) passed to the
decoder used in place of X.sub.k (t) in determining bit
decisions.
5. The system for communicating digital information over a wire of
claim 3, further comprising:
a) a signal-to-noise (S/N) estimation device which estimates
.vertline..delta..sub.k .vertline. from Fourier coefficients
acquired at times t and t-1, X.sub.k (t), X.sub.k (t-1),
respectively according to the equation: ##EQU5## where X.sub.k
(t-1)=x.sub.1 +jy.sub.1 and X.sub.k (t)=X.sub.2 +jy.sub.2 for each
of the Fourier coefficients;
b) a combiner which weights each of the Fourier coefficients
X.sub.k (t) according to .vertline..delta..sub.k .vertline. to
result in weighted Fourier coefficients X.sub.k (t) passed to the
decoder used in place of X.sub.k (t) in determining bit
decisions.
6. The system for communicating digital information over a wire of
claim 4 further comprising:
an averager coupled between S/N estimation device and the combiner
for receiving a plurality of .vertline..delta..sub.k .vertline.
estimates for Fourier coefficients X.sub.k (t) over time, and
performing an average of .vertline..delta..sub.k .vertline. to
result in an .vertline..delta..sub.k .vertline. for each lobe,
provided to the combiner, and used in place of
.vertline..delta..sub.k .vertline. to result in the modified
Fourier coefficients X.sub.k (t).
7. The system for communicating digital information over a wire of
claim 1 further comprising a plurality of transmitters each
transmitting digital signals s.sub.i, where i .noteq.1 at
frequencies between harmonic interference lobes, but at frequencies
different than those of other transmitters of the system.
Description
BACKGROUND OF THE INVENTION
1. Field of the Invention
The present invention relates to communications systems and more
particularly to systems communicating a signal over existing power
lines.
2. Description of Related Art
Electric power companies typically have a service person visit
customers and read a power meter to determine the amount of power
consumed for billing purposes. This can become very cumbersome and
time consuming with a large number of customers.
It would be beneficial for power companies to read these electric
power meters remotely. Since there are existing power wires to each
customer's power meter, the existing power wires may possibly be
used to communicate with the power meters, however, existing
modulation techniques may not operate easily in the high-power,
high-noise environment of a power line.
An additional problem involves voltage transformers, which are
inherently inductively coupled and thereby introduce non-linear
phase shifts in a signal passing through the transformer.
Complicated communications systems are required to perform reliably
in the high-power, high-noise, phase distorted power line channels.
For example, some of these systems monitor different frequency
channels, select an appropriate channel, and then indicate the
appropriate frequency to other communicating units.
Some, such as U. S. Pat. No. 5,185,591 Shuey issued Feb. 9, 1993,
employ a plurality of signals which are not harmonically related.
This requires complicated filtering and signal extraction.
Currently there is a need for a less complicated communication
system which can employ existing power lines to remotely read
multiple power meters simultaneously.
BRIEF DESCRIPTION OF THE DRAWINGS
The features of the invention believed to be novel are set forth
with particularity in the appended claims. The invention itself,
however, both as to organization and method of operation, together
with further objects and advantages thereof, may be best understood
by reference to the following description taken in conjunction with
the accompanying drawing in which:
FIG. 1 is a graph of a Harmonic Modulation (HM) signaling power
spectra of the present invention intended to be interleaved with
power line interference.
FIG. 2 is a simplified block diagram of a power line communication
transmitter employing modified GHM signaling waveform according to
the present invention.
FIG. 3 is a simplified block diagram of an embodiment of a power
line communication receiver for decoding harmonic modulation (HM)
signaling according to the present invention.
FIG. 4 is an graph of a signal-to-noise (S/N) ratio estimation
curve.
FIG. 5 is graph of estimated vs. actual S/N ratios after
averaging.
OBJECTS OF THE INVENTION
It is an object of the present invention to provide a system to
communicate information over a power line.
It is another object of the present invention to provide a
modulation scheme which can reliably communicate a message over a
channel having a great deal of harmonic interference, e.g.,
consisting of many lobes centered at frequencies that are multiples
of the power fundamental frequency.
SUMMARY OF THE INVENTION
A communication system transmits message bits over a wire having
harmonic interference lobes, such as a power line.
A carrier wave synthesis device creates a carrier wave having a
plurality of frequency lobes selected to be positioned between
interference lobes.
In a preferred embodiment, the a calculation device preprocesses a
carrier waveform and stores it in a storage device. The calculation
device may be a digital device which synthesizes samples of the
waveform, or be a device which produces a continuous waveform which
is sampled by a sampler. In its most general form, the calculation
device produces a waveform with frequencies between the harmonic
interference lobes, and in a preferred embodiment, is defined
according to a Shapiro-Rudin sequence, described in "Multitone
Signals with Low Crest Factor" by S. Boyd, IEEE Trans. on Circuits
and Systems, Vol. CAS-33, No. 10, pp. 1018-1022.
The samples of this waveform are stored in a storage device, such
as a ROM. This set-up portion need not be repeated unless a
different carrier waveform is desired.
A timing device in the transmitter monitors the power signal
passing through the power line for timing reference and control
purposes.
A playback device is coupled to the timing device and the storage
device and plays back the prestored waveform according to a
predetermined rate.
A bit encoder is provided with digital information to be
transmitted. This information is converted from a series of ones
and zeros to ones and negative ones. The bit encoder inverts the
carrier waveform for a single bit period for one message bit value,
and leaves the carrier wave unchanged for the other message bit
value to create an encoded message.
The encoded message is mixed with existing signals on the power
line by a summation device. A remote receiver coupled to the power
line, decodes the received encoded message into message bits.
In the receiver, a sampler converts a continuous signal into a
series of discrete time samples which are truncated by a
quantizer.
A Fourier Transform device performs a Fourier transform on the
samples to result in Fourier coefficients X.sub.k (t) at time
t.
A decoder performs an inner product of the Fourier coefficients
X.sub.k (t) and X.sub.k *(t-1), and tests if the real part of the
inner product is above or below a predetermined threshold. Above or
at the threshold is one bit value, while other values produce the
other bit value. In this way the decoder produces decoded message
bits.
A signal-to-noise (S/N) estimation device may be employed which
computes a reliability indicator estimate .vertline..delta..sub.k
.vertline. for each of the Fourier coefficients X.sub.k.
A combiner weights each of the frequency coefficient X.sub.k (t)
according to its .vertline..delta..sup.k .vertline. to result in
weighted Fourier coefficients X.sub.k (t) used by the decoder in
place of X.sub.k (t) in determining bit decisions.
An averager may also be employed between the S/N estimation device
and the combiner. It receives a plurality of
.vertline..delta..sub.k .vertline., and performs a time average for
each of the .vertline..delta..sub.k .vertline. to result in an
average S/N estimate .vertline..delta..sub.k .vertline. for each
Fourier coefficient. This .vertline..delta..sub.k .vertline. is
then provided to the combiner, and used in place of
.vertline..delta..sub.k .vertline. to result in the modified
Fourier coefficients X.sub.k (t), having greater accuracy.
DETAILED DESCRIPTION OF THE INVENTION
Basic GHM
Geometric Harmonic Modulation (GHM) for communications systems has
been described for radiowave communication in U.S. Pat. No.
5,519,725 issued May 21, 1996, "Geometric Harmonic Modulation (GHM)
for Combined Analog/Digital Transmissions", Hershey, Saulnier,
Hassan, assigned to the present assignee and hereby incorporated by
reference.
GHM allocates signaling energy into lobes, or tones, at different
frequencies being evenly spaced at geometrically increasing
multiples of a base frequency. The GHM signaling waveforms {W.sub.n
(.PHI.,R;t)} are true spread spectrum signals in that the signal
bandwidth, the bandwidth from the lowest frequency tone to the
highest, vastly exceeds the information bandwidth conveyed by the
GHM transmission.
Binary GHM signals convey binary data by inverting or not inverting
the GHM waveform {W.sub.n (.PHI.,R;t) } during a bit duration
interval.
It is not necessary for the GHM transmitter and GHM receiver to
"agree" on the best frequency on which to send data as the same
data is sent on each of the GHM tones which are spread throughout
the 5-10 kHz band. Thus the operation or signaling protocol of the
GHM system is less complex than a single tone system for which the
transmitter and receiver must jointly search for and agree upon a
frequency which is not attenuated by a fading phenomenon such as
standing waves for example.
It was found that by modifying the GHM signaling waveform {W.sub.n
(.PHI.,R;t) }, that it can have spectral properties well suited to
the synchronous noise environment encountered on much of the power
line network.
By taking a more general look at GHM and the power line noise, it
is determined that lobes of the modulated signal should be
positioned on a frequency spectrum interleaved with the power line
noise. Since power line noise has a great deal of interference at
the basic oscillation frequency, and at overtones, multiples of the
basis frequency, the signal should not reside in these frequency
regions. By modulating the signal to be between the interference
lobes, the signal will be defined by: ##EQU1## where M is the
number of lobes of the signal s.sub.i (t), and i indicates one of a
plurality of different signals which may be transmitted
simultaneously. The phases .PHI..sub.k, .theta..sub.k of Eq. (1)
may be assigned specific values for each signaling set or, in its
most general form, may take on random values for any desired
signaling set. A particular choice of phases, along with amplitudes
and frequencies, determines a unique signaling set that may be used
for a particular power line communication application depending on
considerations of information capacity, multiplicity of users, line
coupling response, and communication channel characteristics.
By selecting frequencies f.sub.ik correctly, the signals s.sub.1
may be interleaved with harmonic power line noise and will be
referenced to as Harmonic Modulation (HM).
In FIG. 1, a frequency v. amplitude spectrum graph shows harmonic
power line noise lobes, marked "N" at the basic frequency (here 60
Hz), and integer multiples of the basic frequency. An HM signal to
be sent s.sub.1 has frequency components f.sub.11, f.sub.12, . . .
f.sub.1M. Similarly, HM signal S.sub.2 has components f.sub.21,
f.sub.22, . . . f.sub.2M. Again, each frequency component of signal
i carries the same information.
Also, a dotted trace depicts an example of frequency-selective
fading. The example shows the first frequency lobes of signals
s.sub.1, and S.sub.2 being f.sub.11, and f.sub.21, will be
attenuated, (having a lower S/N ratio) while other frequencies will
be unaffected.
Modifying GHM Frequencies
The HM signal of FIG. 1 has no appreciable frequency content at 60
Hz and its harmonics. Thus, such a modified GHM signal may be
received over a power line communication link without much
interference from the 60 Hz synchronous signal.
Note also that the parameter M controls the number of local maxima
("main lobes") of the power spectral density and the signaling rate
controls the spectral width of the main lobes. For a meter reading
or other power line communications application, a larger value of M
may be used for greater spectrum spreading.
The spectrum of the modified HM signaling waveform must be in a
frequency range which will be sufficiently above the 60 Hz
fundamental frequency and still carry through a distribution
transformer. The modulation of the HM waveform must not be
significantly affected or corrupted by non-linear phase shifts
which occur as a signal passes through an inductively coupled
transformer.
Phase Shift Keying
Phase Shift keying signaling may be Differential or Coherent. If
the signaling is Differential (DPSK), then the receiver need not
correct for frequency selective phase rotations. The receiver will
incur a non coherent combining loss as a trade for the relative
simplicity of the demodulation algorithm. If coherent signaling is
used, the receiver may be able to achieve a higher signal-to-noise
ratio at the expense of greater demodulation complexity.
By selecting DPSK in the present invention the system is less
sensitive to phase distortion introduced by non-linear transformers
and results in a less complex system.
S/N Estimates
Frequency selective fading typically occurs in received signals due
to standing wave phenomena and there is therefore a need to devise
a low complexity algorithm to estimate the signal-to-noise ratio
(S/N) of each of the tones in the multi-tone signal. Such
information is required in order to better merge information from
received tones to make a better decision in demodulating an
information bit. Conventional methods, such as maximal ratio
combining, may be used. Tones that are highly corrupted by noise
may not be used at all.
The receiver can estimate the S/N of each tone in additive white
Gaussian noise (AWGN) by computing, for each tone k, the absolute
value of the normalized dot product, .vertline..delta..sup.k
.vertline., between X.sub.k (t-1) and X*.sub.k (t) where X.sub.k
(.multidot.) is the Fourier coefficient, and X*.sub. (.multidot.)
is the complex conjugate of X.sub.k (.multidot.). This value of
.vertline..delta..sub.k can be computed by the following. Assume
X.sub.k (t-1)=x.sub.1 +jy.sub.1 and X.sub.k (t)=X.sub.2 +jy.sub.2
then ##EQU2##
If AWGN is assumed for the bandwidth associated with a single tone,
then .vertline..delta..sub.k .vertline. may be used to determine
S/N for each of the tones per the graph in FIG. 4.
Bit Determination
If Differential Phase Shift Keying were used and if the GHM signal
were unaffected by frequency selective fading on the channel, and
if the noise were AWGN, then near optimal demodulation could be
effected by simply computing the inner product: ##EQU3## If Re(b)
.gtoreq.0, the decoded bit value being "0", and Re(b) <0 the
decoded bit value being "1", where Re(b) is the real part of the
complex number b.
Transmitter Implementation
FIG. 2 shows a simplified block diagram of a transmitter 10 coupled
to a power line. This is one of many transmitters which may be
coupled to the same power line. A given number of tones M to be
transmitted is selected. Signals according to Eq. (1) are then
synthesized by a calculation device 3, which may be a general
purpose computer, or may be a device which creates a continuous
waveform which is sampled by sampler 5. The samples are saved in
storage device 13. Storage device 13, in its preferred embodiment
is a ROM. This may all be performed prior to transmission, and need
not be performed before each use of transmitter 10.
Preferably, a transient protection device 11 which limits power
surges, protecting equipment downline, is coupled to the power
line, and passes the signal to a timing device 17.
Timing device 17 determines master timing information from the
fundamental power line frequency, for example by counting
zero-crossings.
A playback device 15 coupled to storage device 13, reads out the
samples at a rate synchronous with the power line voltage.
A multiplier 19, also coupled to timing device 17, multiplies the
waveform created by playback device 15 by a single bit during a
single bit period. The bit sequence is modified prior to
transmission changing bit values to a series of ones and negative
ones, instead of ones and zeros. Therefore, when the waveform and
signal bits are provided to multiplier 19, a bit value of one will
not change the waveform for a bit period, while a negative one will
invert the waveform. This results in an HM modulated signal which
is summed with the power line signal at a summer 18, and
communicated over the power line to a receiver.
Receiver Implementation
FIG. 3 shows a simplified HM receiver 20 according to the present
invention. Again, several receivers may be coupled to the same
power line. Receiver 20 is connected to the power line through a
transient protection device 21 which limits power surges,
protecting equipment downline. Transient protection device 21 is
coupled to a timing controller 27 and a sampler 25.
Timing controller 27 monitors the power line signal r(t) passed
from transient protection device 21 and determines fundamental
power line frequency. Timing controller 27 provides a timing signal
to other elements of receiver 20.
Sampler 25, clocked by the timing controller 27, samples continuous
signal x(t) provided to it from transient protection device 21 to
produce a series of discrete samples {x(n)}.
Optionally, a bandpass filter 23 is connected between transient
protection device 21 and sampler 25 which removes noise outside the
useful spectrum of the HM signaling.
The discrete samples, {x(n) } are provided to a quantizer 29 which
reduces the precision of the samples to a set of 2.sup.B values,
where B is the number of bits allocated per sample.
The quantized samples, {x(n)}, are passed to a Fast Fourier
Transform (FFT) module 31 which determines the inverse Fourier
transform coefficients {X.sub.k (t)} for signal {x(n)}. In one
embodiment, the Fourier Transform coefficients from FFT module 31
may be passed directly to a decoder 33, which recovers the message
bits from the HM signal, as described in Eq. (3) in "Bit
Determination" above.
Optionally, a signal-to-noise (S/N) estimation device receives the
Fourier coefficients from FFT 31 and determines an estimate for
determining the S/N, .vertline..delta..sub.k .vertline. from Eq.
(2) above for each Fourier coefficient.
A combiner circuit 34 connected between FFT 31 and decoder 33
receives S/N estimate, .vertline..delta..sub.k .vertline., and the
Fourier coefficients X.sub.k (t), and provides weighting to the
coefficients based upon the S/N estimate for that frequency band.
This may be any conventional weighting technique, and may be as
simple as throwing out coefficients which do not meet a
predetermined threshold.
The S/N estimation curve related to Eq. (2) is shown in FIG. 4. The
dotted lines indicate a region of .+-. one sigma standard
deviation. There is a high variance to .vertline..delta..sub.k
.vertline. at the lower S/N ratios.
Averaging
Because the data rate at which the automatic meter reading
communications will take place is so low, it will be possible to do
a significant amount of post processing. A better estimate of the
S/N ratio may be made by averaging .vertline..delta..sub.k
.vertline. over the number of symbols in a particular message. This
averaging will reduce the standard deviation of the estimate of
.vertline..delta..sub.k .vertline. by the square root of the number
of symbols.
Therefore, in an optional embodiment, an averager 36 is placed
between S/N estim. unit 35 and combiner 34 which averages
.vertline..delta..sub.k .vertline. over a plurality of estimations.
In FIG. 5, the estimate of the signal-to-noise ratio, produced by
averaging .vertline..delta..sub.k .vertline. over 100 symbols, is
graphed as a function of the true signal-to-noise ratio. Dotted
lines are shown bounding the estimate region by one sigma.
Reduced Peak/RMS Power
By limiting the peak to RMS power transmitted by transmitter 10,
the dynamic range of the transmitted signal will be narrowed,
easing the requirements of amplifier linearity. It was determined
that for a certain M, e.g. M=8, certain forms of Eq. (1) would
result in near optimum Crest factors by selecting signs of the
terms of Eq. (1) according to a Shapiro-Rudin sequence, (See Boyd
above). For example, this results in an 8-tone composite signal
within the 5-10 kHz band with each tone having equal energy. The
form of the ith signal is:
The table of frequencies {f.sub.i,k } for the ten simultaneous
waveforms is given in Table 1.
TABLE 1 ______________________________________ {f.sub.i, k } in Hz
i k = 1 k = 2 k = 3 k = 4 k = 5 k = 6 k = 7 k = 8
______________________________________ 1 5010 5610 6210 6810 7410
8010 8610 9210 2 5070 5670 6270 6870 7470 8070 8670 9270 3 5130
5730 6330 6930 7530 8130 8730 9330 4 5190 5790 6390 6990 7590 8190
8790 9390 5 5250 5850 6450 7050 7650 8250 8850 9450 6 5310 5910
6510 7110 7710 8310 8910 9510 7 5370 5970 6570 7170 7770 8370 8970
9570 8 5430 6030 6630 7230 7830 8430 9030 9630 9 5490 6090 6690
7290 7890 8490 9090 9690 10 5550 6150 6750 7350 7950 8550 9150 9750
______________________________________
For these signals {s.sub.i (t) }, the Crest Factor is 6 dB. By
employing the signaling waveform of a Shapiro-Rudin sequence, the
Crest Factor is significantly reduced.
While several presently preferred embodiments of the novel
invention have been described in detail herein, many modifications
and variations will now become apparent to those skilled in the
art. It is, therefore, to be understood that the appended claims
are intended to cover all such modifications and variations as fall
within the true spirit of the invention.
* * * * *