U.S. patent number 5,789,974 [Application Number 08/682,102] was granted by the patent office on 1998-08-04 for calibrating the dc-offset of amplifiers.
This patent grant is currently assigned to Analog Devices, Inc.. Invention is credited to Gangadhar Burra, Paul F. Ferguson, Jr., Michael Mueck.
United States Patent |
5,789,974 |
Ferguson, Jr. , et
al. |
August 4, 1998 |
Calibrating the DC-offset of amplifiers
Abstract
The dc-offset voltage of an amplifier is calibrated by: (1)
configuring the amplifier as a comparator, (2) using the output of
the comparator to drive the up/down select input of an up/down
counter, and (3) using the output count of the up/down counter to
control: (a) a dc-offset correction voltage being: (i) applied
across the inputs of the amplifier, or (ii) being used to adjust a
voltage which controls an operating parameter of a device in the
amplifier, or (b) switches which selectively adjust the effective
size or operating conditions of a transistor or other device such
that the dc-offset voltage of the amplifier is adjusted
corresponding to the value of the output count. At the end of a
calibration cycle, the output count of the up/down counter is
maintained and is used to: (a) control a voltage which is applied
permanently in series with one of the inputs of the amplifier or to
an operating parameter control lead of a device in the amplifier,
or (b) maintain the position of switches used to adjust the
effective size or operating conditions of the transistor or other
device in the amplifier to compensate for the offset voltage
thereof.
Inventors: |
Ferguson, Jr.; Paul F. (Dracut,
MA), Burra; Gangadhar (Acton, MA), Mueck; Michael
(Andover, MA) |
Assignee: |
Analog Devices, Inc. (Norwood,
MA)
|
Family
ID: |
24738210 |
Appl.
No.: |
08/682,102 |
Filed: |
July 17, 1996 |
Current U.S.
Class: |
330/2; 330/9 |
Current CPC
Class: |
H03F
1/304 (20130101); H03F 3/45475 (20130101); H03F
3/45968 (20130101); H03F 2203/45048 (20130101); H03F
2200/375 (20130101) |
Current International
Class: |
H03F
1/30 (20060101); H03F 001/02 () |
Field of
Search: |
;330/2,9,51,253,257 |
References Cited
[Referenced By]
U.S. Patent Documents
Other References
Yu and Geiger, An Automatic Compensation Scheme with Ping-Pong
Control of CMOS Operational Amplifiers, May, 1994 Jrnl. of Solid
State Circuits, vol. 29, pp. 601-610..
|
Primary Examiner: Mottola; Steven
Attorney, Agent or Firm: Wolf, Greenfield & Sacks,
P.C.
Claims
What is claimed is:
1. An amplifier circuit, comprising:
an amplifier element having a first input and a second input, the
amplifier element being arranged to be configured as an amplifier
during a period of normal operation and to be configured as a
comparator during a calibration period of the amplifier
element;
a logic circuit coupled to an output of the amplifier element at
least during the calibration period to receive an output signal
from the output of the amplifier element and to output an offset
compensation control signal representing an offset compensation
voltage for the amplifier element; and
an offset compensation circuit including:
a positive voltage lead and a negative voltage lead, coupled,
respectively, to the first input and the second input during the
calibration period, and
an offset compensation control lead coupled to the logic circuit to
receive the offset compensation control signal from the logic
circuit, the offset compensation circuit being configured to
provide, responsive to the offset compensation control signal, only
the offset compensation voltage between the positive voltage lead
and the negative voltage lead of the offset compensation circuit
during all times when the amplifier element is receiving power.
2. The amplifier circuit as claimed in claim 1, wherein the logic
circuit is configured to provide a digital output signal which is
altered in response to the output signal from the amplifier element
such that, responsive to a clock signal, the offset compensation
voltage is increased if the output signal from the amplifier
element is in a first logic state and decreased if the output
signal from the amplifier element is in a second logic state.
3. The amplifier circuit as claimed in claim 2, wherein:
the logic circuit includes an up/down counter having an up/down
select input and a count output, the output of the amplifier
element being coupled to the up/down select input of the up/down
counter at least during the calibration period; and
the offset compensation circuit includes a digital-to-analog
converter having a digital input coupled to the count output of the
up/down counter to receive an output count of the up/down counter
as the offset compensation control signal.
4. An amplifier circuit, comprising:
an amplifier including:
an input to receive an input signal,
an output to provide an output signal, and
a plurality of primary devices coupled between the input and the
output;
at least one secondary device, the at least one secondary device
being substantially smaller than each of the plurality of primary
devices; and
at least one switch coupled to at least one of the amplifier and
the at least one secondary device to selectively couple the at
least one secondary device to the amplifier to alter the effective
size of at least one of the plurality of primary devices to
compensate for an offset voltage of the amplifier.
5. The amplifier circuit as claimed in claim 4, wherein the at
least one of the plurality of primary devices is a transistor in a
current mirror of the amplifier.
6. The amplifier circuit as claimed in claim 4, wherein the
amplifier is connected to be configured as a comparator during a
calibration period, and wherein the amplifier circuit further
comprises a logic circuit coupled to the output of the amplifier at
least during the calibration period to receive a logic signal from
the output of the amplifier, and the logic circuit is coupled to
the at least one switch to provide the at least one switch with a
control signal which, responsive to the logic signal from the
output of the amplifier, selectively activates the at least one
switch.
7. The amplifier circuit as claimed in claim 6, wherein the logic
circuit is configured such that the control signal includes a
digital count that, responsive to a clock signal, is increased if
the logic signal from the output of the amplifier is in a first
logic state and is decreased if the logic signal from the output of
the amplifier is in a second logic state.
8. The amplifier circuit as claimed in claim 4, wherein the
plurality of primary devices includes a differential pair of the
primary transistors and the amplifier further includes a shorting
switch arranged to couple together a pair of outputs of the
differential pair during a portion of the calibration period.
9. An amplifier circuit, comprising:
an amplifier element arranged to be configured as an amplifier
during a period of normal operation and to be configured as a
comparator during a calibration period, the amplifier element
including at least one device coupled to an internal node of the
amplifier element, the at least one device having an operating
parameter control lead to control an operating parameter of the at
least one device;
a logic circuit coupled to an output of the amplifier element at
least during the calibration period to receive an output signal
from the output of the amplifier element and to output an offset
compensation control signal; and
an offset compensation control circuit coupled to the logic circuit
and to the operating parameter control lead of the at least one
device to receive the offset compensation control signal from the
logic circuit and provide a parameter control signal to the
operating parameter control lead to control a particular operating
parameter of the at least one device to compensate for an offset
voltage of the amplifier element.
10. An amplifier circuit as claimed in claim 9, wherein the at
least one device includes a transistor and the operating parameter
control lead is coupled to a substrate biasing node of the
transistor.
11. An amplifier circuit as claimed in claim 9, wherein the at
least one device includes a transistor included in a differential
pair of input transistors of the amplifier element.
12. An amplifier circuit, comprising:
an amplifier connected to be configured as a comparator during a
calibration period, the amplifier including:
an input to receive an input signal,
an output to provide an output signal, and
at least one transistor included in a current mirror of the
amplifier, the at least one transistor being coupled to an internal
node of the amplifier and having an operating parameter control
lead to control an operating parameter of the at least one
transistor;
a logic circuit coupled to the output of the amplifier at least
during the calibration period to receive a signal from the output
of the amplifier and to output an offset compensation control
signal; and
an offset compensation control circuit coupled to the logic circuit
and to the operating parameter control lead of the at least one
transistor to receive the offset compensation control signal from
the logic circuit and provide a parameter control signal to the
operating parameter control lead to control a particular operating
parameter of the at least one transistor to compensate for an
offset voltage of the amplifier.
13. An amplifier circuit as claimed in claim 9, wherein the logic
circuit is configured such that the offset compensation control
signal, responsive to a clock signal, is increased if the signal
from the output of the amplifier element is in a first logic state
and is decreased if the signal from the output of the amplifier
element is in a second logic state.
14. An amplifier circuit, comprising:
an amplifier element arranged to be configured as an amplifier
during a period of normal operation and to be configured as a
comparator during a calibration period;
a logic circuit coupled to an output of the amplifier element at
least during the calibration period to receive an output signal
from the output of the amplifier element and to output an offset
compensation control signal, the logic circuit being configured to,
responsive to a clock signal, increase the offset compensation
control signal if the signal from the output of the amplifier
element is in a first logic state and decrease the offset
compensation control signal if the signal from the output of the
amplifier element is in a second logic state; and
an offset compensation control circuit coupled to the logic circuit
and the amplifier element to receive the offset compensation
control signal from the logic circuit and, responsive to the offset
compensation control signal, to compensate for an offset voltage of
the amplifier element.
15. An amplifier circuit, comprising:
means for detecting an offset voltage of an amplifier element
during an offset calibration period in which the detecting means
includes means for configuring the amplifier element as an
amplifier during a period of normal operation and configuring the
amplifier element as a comparator during a calibration period;
and
means for compensating for the offset voltage of the amplifier
element, the compensating means being coupled to the detecting
means at least during the calibration period.
16. A method for controlling an offset voltage of an amplifier
element, comprising the steps of:
configuring the amplifier element as a comparator during a
calibration period;
coupling a controllable voltage source that generates only an
offset compensation voltage for the amplifier element between a
pair of inputs of the amplifier element;
responsive to an output of the amplifier element, generating an
offset compensation control signal that controls the offset
compensation voltage;
responsive to the offset compensation control signal, adjusting the
offset compensation voltage until the offset compensation voltage
approximates an inherent offset voltage of the amplifier element;
and
configuring the amplifier element as an amplifier during a period
of normal operation.
17. The method as claimed in claim 16, wherein the step of
generating an offset compensation control signal responsive to the
output of the amplifier element includes the steps of:
responsive to a first logic state at the output of the amplifier
element, increasing a value of the offset compensation control
signal; and
responsive to a second logic state at the output of the amplifier
element, decreasing the value of the offset compensation control
signal.
18. A method for compensating for an offset voltage of an amplifier
including a primary device having a first size and at least one
secondary device having a second size, the second size being
substantially smaller than the first size, the method comprising
the steps of:
detecting an offset voltage of the amplifier; and
responsive to a detected offset voltage, selectively coupling the
at least one secondary device to an internal node of the amplifier
such that the first size of the primary device is effectively
increased or decreased by the second size of the secondary device
so as to alter the offset voltage of the amplifier circuit.
19. The method as claimed in claim 18, wherein:
the step of detecting the offset voltage of the amplifier includes
the step of configuring the amplifier as a comparator; and
the step of selectively coupling the at least one secondary device
to the internal node of the amplifier responsive to the detected
offset voltage includes the steps of:
responsive to a first logic state at an output of the amplifier,
increasing an offset compensation control signal which controls the
selective coupling of the at least one secondary device to the
amplifier circuit, and
responsive to a second logic state at the output of the amplifier,
decreasing the offset compensation control signal which controls
the selective coupling of the at least one secondary device to the
amplifier circuit.
20. A method for controlling an offset voltage of an amplifier
element, comprising the steps of:
configuring the amplifier element as a comparator during a
calibration period;
connecting a pair of inputs of the amplifier element to one of a
common node and two nodes having known voltages thereon;
responsive to an output signal of the amplifier element,
selectively adjusting a level of an operating parameter of at least
one device included in or coupled to the amplifier element to
compensate for the offset voltage of the amplifier element; and
configuring the amplifier element as an amplifier during a period
of normal operation.
21. The method as claimed in claim 20, wherein the step of
selectively adjusting the voltage of the operating parameter of the
at least one device responsive to the output signal of the
amplifier element includes the steps of:
responsive to a first logic state at the output of the amplifier
element, increasing the level of the operating parameter of the at
least one device; and
responsive to a second logic state at the output of the amplifier
element, decreasing the level of the operating parameter of the at
least one device.
22. A method for compensating for an offset voltage of an amplifier
element including the septs of:
configuring the amplifier element as a comparator during a
calibration period;
connecting a pair of inputs of the amplifier element to one of a
common node and two nodes having known voltages thereon; and
responsive to a first logic state at an output of the amplifier
element, increasing a value of the compensation signal that
controls the offset compensation of the amplifier element;
responsive to a second logic state at the output of the amplifier
element, decreasing the value of the compensation signal that
controls the offset compensation of the amplifier element; and
configuring the amplifier element as an amplifier during a period
of normal operation.
23. The amplifier circuit as claimed in claim 9, wherein the offset
compensation control circuit is configured such that the parameter
control signal provided to the operating parameter control lead
includes a parameter control voltage.
24. The amplifier circuit as claimed in claim 12, wherein the
offset compensation control circuit is configured such that the
parameter control signal provided to the operating parameter
control lead includes a parameter control voltage.
25. The method of claim 20, wherein the step of selectively
adjusting the level of the operating parameter of the at least one
device includes selectively adjusting a voltage of the operating
parameter of the at least one device.
Description
FIELD OF THE INVENTION
The present invention relates generally to electronic amplifiers
and, more particularly, to methods and circuity for calibrating the
dc-offset voltage of an amplifier.
BACKGROUND OF THE INVENTION
Operational amplifiers (op-amps) commonly are used to attenuate,
amplify, filter or otherwise condition electronic signals (e.g.,
audio signals) prior to transmitting the signals to output devices
(e.g., audio reproduction devices such as speakers). Typically,
low-impedance speakers, e.g., eight ohms, are used to reproduce
such audio signals. To attain a power output from these
low-impedance speakers that is satisfactory for audio applications,
high-power, high-gain amplifiers are required to drive them.
Although modem device fabrication technology has resulted in
significant improvements in the performance characteristics of
op-amps, most op-amps still have an inherent flaw which commonly is
referred to as dc-offset. The inherent dc-offset voltage of an
op-amp may be explained as follows. When a common voltage is
applied to both inputs of an op-amp, the op-amp typically does not
perform as though it has received a common voltage at its inputs
but, rather, it performs as if it has received a voltage at a first
input that is offset by a particular value from the voltage
received at a second input. Therefore, it is common practice to
represent an op-amp as having a dc-offset voltage (VOS) located
between its inputs or between one of its inputs and a reference
(e.g., ground).
A typical amplifier circuit employing an op-amp is shown generally
at 20 in FIG. 1.
Referring to FIG. 1, voltage source 25, which is connected between
reference node GND and non-inverting input 30 of op-amp 22,
represents the inherent dc-offset voltage (VOS) of op-amp 22.
Additionally, voltage source 26 represents a dc-offset voltage
(VINOFF) present at input 28 of amplifier circuit 20, which has
been passed to amplifier circuit 20 from previous stages of the
circuit (not shown) in which op-amp 22 is used (i.e., the driving
stage(s)). The gain (-N) between input 28 and output 32 of
amplifier circuit 20 is selected by adjusting the position of wiper
34 of potentiometer 24, which alters the ratio between the
resistance values of its two resistor portions 24A and 24B. If
wiper 34 is positioned so that the resistance value of resistor
portion 24B is nearly zero, the amplifier circuit 20 is in a
maximum gain configuration for the given resistance values.
Conversely, if wiper 34 is positioned so that the resistance value
of resistor portion 24A is zero, the amplifier circuit 20 is in a
muting configuration (i.e., the gain (-N) of the amplifier circuit
is equal to zero).
Because of the presence of the dc-offset voltages (VOS and VINOFF),
the dc output voltage (VDCOUT) at output 32 of amplifier circuit 20
is determined by the following equation: VDCOUT =(VOS *
(1+N)-(VINOFF * N)). It follows, then, that due to the dc-offset
voltages, the dc voltage at output 32 will be affected by changes
in the gain (-N) of amplifier circuit 20. Consequently, in
applications where the value of the gain (-N) is changed by
discrete values (rather than by continuous time changes), voltage
steps will result at the output of the op-amp circuit. In audio
applications, when these voltage steps are large, for example, when
an amplifier circuit experiences a discrete change from a circuit
configuration having a large value of N to a muting configuration,
or vice versa, an audible "click" or "pop" may result when the
resultant voltage step is output by a speaker.
Many modern audio producing devices have a muting function, e.g.,
mute buttons on remote controls of televisions and stereos.
Therefore, amplifier circuits are required for such devices that
somehow prevent the annoying clicking or popping described above.
One way to reduce the adverse effects of dc-offset voltages in
amplifier circuits is to calibrate the circuits so as to compensate
for the dc-offset voltages thereof.
For example, a known method for calibrating the dc-offset of a
digital-to-analog converter (DAC) circuit includes the steps of:
(1) forcing the digital input signal of the DAC to zero, (2)
converting the digital input signal to an analog signal, (3)
temporarily converting an amplifier in the DAC into a comparator,
(3) using this comparator to compare the analog conversion of the
digital input signal to an analog ground, and (4) adding or
subtracting a digital value (representing a dc-offset compensation
voltage) to or from the digital input signal at a digital summation
node until the output of the comparator switches states. In this
manner, a digital representation of a dc-offset voltage of the
entire DAC circuit is obtained and thereafter is added continuously
to the digital input signal. Such DAC circuits successfully achieve
their intended purpose, i.e., to convert a digital input signal
into an analog output signal while, at the same time, compensating
for the dc-offset voltage of the entire DAC circuit. However, such
circuits require a digital input signal and a high-performance DAC
in order to function properly.
Frequently, op-amps are employed in circuits having only an analog
input signal. Thus, in order to use the known dc-offset voltage
compensation technique (applicable to DAC circuits) to compensate
for the dc-offset voltage of op-amps in such analog circuits, a
user would first have to use an analog-to-digital converter (ADC)
to convert the analog input signal into a digital format and then
convert the resulting digital signal back into an analog format
(using an-offset compensated DAC circuit). Such analog-to-digital
and subsequent digital-to-analog conversion would: (1) introduce
any dc-offset voltage caused by the ADC into the circuit, which
dc-offset voltage would not be compensated for by the
offset-compensated DAC circuit, (2) require a large amount of die
area to accommodate the high-performance ADC and the DAC circuit,
(3) significantly increase the power consumption of the circuit,
and (4) increase the probability of signal distortion due to the
inherent inaccuracies of ADCs and DACs. Instead, a simple circuit
is desired which will compensate for the dc-offset voltage of an
op-amp itself, rather than compensating for the dc-offset voltage
of an entire DAC circuit, and which does not require a digital
input signal and a highperformance DAC in order to function
properly.
Dc-offset calibration of op-amp circuits having analog input
signals has been accomplished by a circuit which performs the steps
of: (1) temporarily converting an amplifier into a comparator, (2)
connecting both inputs of the amplifier to a common node, (3) using
a counter to control the stepwise injection of a bias current into
a node of the amplifier until the output of the amplifier changes
states. To control the stepwise injection of bias currents, the
circuit employs a counter as follows: (a) the circuit resets the
counter to zero (with a zero count of the counter causing a maximum
amount of available current from a current DAC to be injected into
a node of a first transistor of a differential pair of input
transistors of the amplifier), (b) the counter increments its
output count until it reaches one-half the value of its maximum
count (with the current flowing from the current DAC being stepwise
decreased with every increment of the counter), wherein a count
equal to one-half of the maximum count causes a minimum amount of
current from the current DAC to be injected into the node of the
first transistor, (c) when the counter passes the count that is
onehalf of its maximum count, the maximum current output of the
current DAC then is caused to be injected into a node of a second
transistor of the differential pair of input transistors, thus
causing the output of the amplifier to switch states, and (d) the
counter continues counting until it reaches its maximum count (with
the current injected into the node of the second transistor by the
current DAC decreasing stepwise in conjunction with the stepwise
increase of the output count of the counter), wherein a maximum
count of the counter causes a minimum current from the current DAC
to be injected into the node of the second transistor. Logic is
employed during this stepwise count/current injection process to
determine at what current level (and corresponding count) the
output of the amplifier (operating as a comparator) was caused to
change states due to the current injection (not including the point
at which the current was caused to switch from a minimum current in
the first transistor to a maximum available current in the second
transistor). After a calibration cycle, the bias current level
which caused the output of the amplifier to change states then is
injected permanently into the node of the amplifier in order to
compensate for the dc-offset thereof.
This stepwise current injection (based on an output count of a
counter) has several drawbacks. One drawback is that the die area
required to implement a seven-bit current DAC may be unacceptably
large for many applications. Another drawback is that such a
current injection scheme requires a moderate amount of power to be
implemented, both during the calibration cycle and after the
circuit is calibrated. That is, even after the circuit is
calibrated, it constantly will require a biasing current to be
injected into a node of a transistor, which results in a static
power consumption that may be considered unreasonable under certain
circumstances. Additionally, the current injection scheme is not
efficient in terms of the time it takes to calibrate an amplifier.
That is, the counter merely begins at zero and increments through
all of its possible outputs until it happens upon a count which
causes a current to be injected into a node of the amplifier which
is sufficient to override the inherent dc-offset of the
amplifier.
It therefore is a general object of the present invention to
provide an improved dc-offset compensation circuit which reduces or
eliminates the adverse effects of the dc-offset voltage of an
op-amp.
SUMMARY OF THE INVENTION
According to one aspect of the present invention, a circuit and
method for calibrating the dc-offset voltage of an amplifier is
disclosed. The circuit includes: (a) an amplifier which is
configured as a comparator during an offset calibration sequence,
(b) a logic circuit coupled to an output of the amplifier to
receive a signal from the output of the amplifier and to output an
offset compensation control signal representing an offset
compensation voltage for the amplifier, and (c) an offset
compensation circuit having an input to receive the offset
compensation control signal from the logic circuit and a pair of
voltage outputs to provide only an offset compensation voltage
across its outputs, which outputs are connected to the inputs of
the amplifier during an offset calibration sequence. The method
includes the steps of: (a) configuring the amplifier as a
comparator, (b) coupling a controllable voltage source that
generates an offset compensation voltage for the amplifier between
a pair of inputs of the amplifier, (c) responsive to an output of
the amplifier, generating an offset compensation control signal
that controls the offset compensation voltage, and (d) responsive
to the offset compensation control signal, adjusting the offset
compensation voltage until the offset compensation voltage
approximates an inherent offset voltage of the amplifier.
According to an additional aspect of the invention, another circuit
and method for calibrating the dc-offset voltage of an amplifier is
disclosed. The circuit includes: (a) an amplifier including several
primary devices coupled between the amplifier's input and output,
(b) at least one secondary device, which is substantially smaller
than each of the primary devices, and (c) a switch coupled to
either the amplifier or the secondary device to selectively couple
the secondary device to the amplifier to alter the effective size
of the primary device to compensate for an offset voltage of the
amplifier. The method includes the steps of: (a) detecting an
offset voltage of the amplifier, and (b) responsive to a detected
offset voltage, selectively coupling the secondary device to an
internal node of the amplifier to adjust the effective size of the
primary device to compensate for the offset voltage of the
amplifier.
According to yet another aspect of the invention, another circuit
and method for compensating for the offset voltage of an amplifier
is disclosed. The circuit includes: (a) an amplifier configurable
as a comparator during an offset calibration period, the amplifier
having a device internal thereto which has a control lead to
control an operating parameter of the device, (b) a logic circuit
coupled to the output of the amplifier at least during the
calibration period to receive an output from the amplifier and to
output an offset compensation control signal, and (c) an offset
compensation control circuit coupled to the logic circuit to
receive the offset compensation control signal and to provide a
control voltage to the operating parameter control lead of the
device internal to the amplifier. The method includes the steps of:
(a) configuring the amplifier as a comparator, (b) connecting a
pair of inputs of the amplifier to a common node or to nodes having
known voltages thereon, and (c) responsive to an output signal of
the amplifier, selectively adjusting the voltage on the operating
parameter control lead of the device internal to the amplifier to
compensate for the offset voltage of the amplifier.
According to another aspect, yet another circuit and method for
compensating for the offset voltage of an amplifier is disclosed.
The circuit includes: (a) an amplifier configurable as a comparator
during an offset calibration control period, (b) a logic circuit
that, responsive to a clock signal, increases its output when the
output signal of the amplifier is in a first logic state and
decrease its output when the output signal of the amplifier is in a
second logic state, and (c) an offset compensation control circuit
that receives an output from the logic circuit and responsive to
the output of the logic circuit, compensates for an offset of the
amplifier. The method includes the steps of: (a) configuring the
amplifier as a comparator, (b) connecting a pair of inputs of the
amplifier to a common node or to two nodes having known voltages
thereon, and (c) responsive to a first logic state at an output of
the amplifier, increasing a value of a compensation control signal
that controls the offset compensation of the amplifier, and (d)
responsive to a second logic state at an output of the amplifier,
increasing a value of a compensation control signal that controls
the offset compensation of the amplifier.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a partial schematic/partial block diagram of a prior art
amplifier circuit in a gain or attenuation configuration.
FIG. 2 is a partial schematic/partial block diagram of an offset
calibration circuit according to the invention.
FIG. 3 is a partial schematic/partial block diagram of another
offset calibration circuit according to the invention.
FIG. 4 is a more detailed partial schematic/partial block diagram
of the offset calibration circuit shown in FIG. 3.
FIG. 5 is a partial schematic/partial block diagram of yet another
offset calibration circuit according to the invention.
FIG. 6 is a partial schematic/partial block diagram of an amplifier
circuit including a main amplifier, which is calibrated according
to the invention, and a catching amplifier, which is used to
maintain a relatively constant voltage level at the output of the
amplifier circuit when the main amplifier is disconnected from
and/or reconnected to the amplifier circuit.
FIG. 7 is a partial schematic/partial block diagram of another
amplifier circuit including a main amplifier having a first stage
and a second stage, which first stage is calibrated according to
the invention, and a catching amplifier, which is used to maintain
a relatively constant voltage level at the output of the amplifier
circuit when the first stage of the main amplifier is disconnected
from and/or reconnected to the amplifier circuit.
FIG. 8 is a timing diagram of several signals used in conjunction
with the offset calibration circuits shown FIGS. 2-5 and the
amplifier circuits shown in FIGS. 6 and 7 when those circuits enter
an amplifier offset calibration cycle.
FIG. 9. is a timing diagram of several signals used in conjunction
with the offset calibration circuits shown FIGS. 2-5 and the
amplifier circuits shown in FIGS. 6 and 7 when those circuits exit
an amplifier offset calibration cycle.
DETAILED DESCRIPTION OF THE INVENTION
FIG. 2 is a block diagram of an offset calibration circuit 36
according to one aspect of the invention. Referring to FIG. 2, when
switch S12 is closed and switches S17 and S18 are open, circuit 36
is configured in a gain or attenuation configuration. That is, an
input signal V.sub.IN presented at node 28 will be output at node
90 after being amplified or attenuated by amplifier circuit 36 (by
a factor determined by the ratio of the resistances of resistor
portions 24A and 24B of potentiometer 24, which ratio is selected
by adjusting the position of wiper 34). Conversely, when switch S12
is open, and switches S17 and S18 are closed, amplifier 40 in
circuit 36 is configured as a comparator and, as such, may be
calibrated according to one aspect of the present invention. While
this configuration of switches is suitable for switching circuit 36
between a gain/attenuation and a comparator configuration, it
should be readily apparent to those skilled in the art that any
similar switch arrangement could accomplish the desired result
(i.e., temporarily configuring amplifier 40 as a comparator).
Circuit 36 normally operates in its gain or attenuation
configuration and is configured as a comparator only during a
particular time interval (i.e., a calibration cycle). It is during
this calibration cycle that the dc-offset voltage of amplifier 40
is calibrated.
Such a calibration cycle may be initiated upon power-up of a device
including circuit 36 or may be initiated responsive to a user
command. During an offset calibration cycle, a voltage is placed
across inverting and non-inverting inputs 52 and 54, respectively,
of amplifier 40 via outputs V- and V+, respectively, of offset
compensation circuit 38. Output 50 of amplifier 40 (which is
operating as a comparator) provides a high or low logic signal to
input 56 of logic circuit 42, depending on whether amplifier 40
senses a greater voltage at input 52 or at input 54. The voltage
that is sensed by amplifier 40 at inputs 52 and 54, and that thus
determines the voltage at output 50 of amplifier 40, includes: (1)
the voltage across outputs V+ and V- of offset compensation circuit
38, and (2) the inherent dc-offset voltage of amplifier 40. Logic
circuit 42 provides an n-bit digital output signal (representing an
offset compensation voltage) at output 58 that is received by
offset compensation circuit 38 at input 48. During a calibration
cycle, the digital output signal at output 58, in response to a
clock signal (CLK) at input 46 and depending of the logic level of
input 56, is increased or decreased. This increase or decrease in
the digital input signal at input 48 of offset compensation circuit
38 causes a corresponding increase or decrease in the offset
compensation voltage across outputs V+ and V- of offset
compensation circuit 38.
In one embodiment of the circuit shown in FIG. 2, offset
compensation circuit 38 and logic circuit 42 can be implemented
with a DAC and an up/down counter, respectively. Any circuitry,
however, which performs a substantially equivalent function to that
performed by these devices may alternatively be used without
departing from the scope of the invention. The digital "count"
output of up/down counter 42 at output 58 controls the voltage
level across outputs V+ and V- of DAC 38. When a signal LOGIC
ENABLE 122 (at counter enable input 44 of up/down counter 42) is
asserted, e.g., is in a logic high state, up/down counter 42,
responsive to a clock signal (CLK) at input 46, will cause up/down
counter 42 to increment or decrement its count at output 58.
Whether up/down counter 42 increments or decrements its count
depends on the state (i.e., logic high or logic low) of the signal
received by up/down selection input 56 of up/down counter 42 from
amplifier 40. Thus, if the input voltage at up/down selection input
56 is in a first state (e.g., logic high), up/down counter 42 will
increment its output count by one every clock cycle so long as
up/down counter 42 is enabled, and if the input voltage at up/down
selection input 56 is in a second state (e.g., logic low) up/down
counter 42 will decrement its output count by one every clock cycle
so long as up/down counter 42 is enabled.
During a calibration cycle, i.e., when amplifier 40 is configured
as a comparator and up/down counter 42 is enabled, the voltage
across outputs V+ and V- of DAC 38 will increase or decrease every
clock cycle corresponding to changes in the digital signal, i.e.,
the count, at input 48. For example, if the voltage across outputs
V+ and V- of DAC 38 is less than the inherent dc-offset voltage of
amplifier 40, output 50 of amplifier 40 will provide a logic high
voltage, up/down counter 42 will increment its count by one, and
the voltage across outputs V+ and V- of DAC 38 will increase by a
discrete voltage level corresponding to the change in the digital
count at input 48. In this manner, the voltage across outputs V+
and V- of DAC 38 will increase or decrease every clock cycle until
this voltage approximates the inherent dc-offset voltage of
amplifier 40.
When the inherent dc-offset voltage of amplifier 40 is reached, the
output of amplifier 40 will switch states. This switching of states
will cause up/down counter 42 to reverse the direction of its
count, causing the voltage across outputs V+ and V- to change
accordingly, and thus causing the output of amplifier 40 to switch
states once again. Therefore, once the dc-offset voltage of
amplifier 40 is approximated by the voltage across outputs V+ and
V-, the output of amplifier 40 will switch states every clock cycle
until the signal LOGIC ENABLE 122 at input 44 of up/down counter 42
is deasserted, e.g., placed in a logic low state. After the signal
LOGIC ENABLE 122 is deasserted, the final output count of up/down
counter 42 is maintained at output 58, and is used as a permanent
input to DAC 38. At the end of the calibration cycle, switch S12 is
closed and switches S17 and S18 are opened to reconfigure circuit
36 in a gain or attenuation configuration. Due to the dc-offset
calibration cycle, however, the voltage between outputs V+ and V-
of DAC 38 now is applied, respectively, between non-inverting input
54 of amplifier 40 and a reference node, e.g., ground node GND,
thereby compensating for the inherent dc-offset voltage of
amplifier 40.
For this calibration circuit to function properly, the maximum
positive and negative offset compensation voltages which can be
produced by DAC 38 (based on the output of up/down counter 42)
should be larger than the expected inherent dc-offset voltage of
amplifier 40. Also, the calibration cycle should be long enough
with respect to a clock (CLK) period such that a sufficient number
of clock cycles occur during the calibration period to ensure that
up/down counter 42 will reach its maximum or minimum count, if
required, during a single calibration period. Additionally, if
possible, up/down counter 42 should be configured such that once
its minimum or maximum count is reached, it maintains that count
rather than wrapping to the maximum or minimum count,
respectively.
FIG. 3 is a block diagram of a circuit 76 according to another
aspect of the invention. As shown, circuit 76 is similar to circuit
36, except that the dc-offset voltage calibration of the amplifier
in circuit 76 is accomplished by using several switches to alter
the effective size of a primary device, e.g., a transistor, which
is internal to amplifier 40. Primary amplifier device 66 in FIG. 3
could represent, for example, a transistor in a current mirror of
the amplifier or one of a differential pair of input transistors.
Secondary amplifier devices 29, 31 and 33 represent amplifier
devices, e.g., transistors, which are substantially smaller than
primary amplifier device 66. For example, if the primary and
secondary devices are transistors, then the secondary transistor
could have a channel length that is substantially (e.g., a factor
of 100) shorter than the channel length of the primary transistor
(or a channel width that is substantially (e.g., a factor of 100)
wider than the channel width of the primary transistor).
In FIG. 3, (conceptual) amplifier 60 represents an ideal op-amp,
i.e., an op-amp with no inherent dc-offset voltage. Amplifier 40
represents an actual amplifier and includes ideal amplifier 60,
primary amplifier device 66 and voltage source 62 (representing the
inherent dc-offset of amplifier 40). Op-amp 64, which includes
secondary amplifier devices 27, 29 and 33 and switches S19, S20 and
S21 (which are configured to switchably connect secondary devices
29, 31 and 33 in parallel with primary amplifier device 66)
represents a dc-offset voltage compensating amplifier according to
this aspect of the invention.
In a similar manner as with circuit 36, when switches S16 and S18
are open and switch S12 is closed, circuit 76 is configured as an
amplifier or attenutator and an input signal V.sub.IN presented at
node 28 will be amplified or attenuated (by a factor depending on
the ratio of the resistances of resistor portions 24A and 24B of
potentiometer 24, which ratio is determined by the position of
wiper 34) and output at node 90. Conversely, when switches S16 and
S18 are closed and switch S12 is open, amplifier 64 will be
configured as a comparator and, as such, may be used to calibrate
the inherent dc-offset voltage of amplifier 40 (which actually is a
part of amplifier 64). Switch S16 is employed to electrically short
together inputs 52 and 54 of amplifier 64 (when it is configured as
a comparator during the calibration cycle). Again, the
configuration of switches S12, S16 and S18 shown in circuit 76 is
merely exemplary and any similar switching configuration that
achieves a similar result alternatively may be employed.
A calibration cycle to calibrate amplifier 64 using circuit 76 may
be initiated upon power up of a device including circuit 76 or may
be initiated responsive to a user command. As mentioned previously,
amplifier 64 is configured as a comparator during a particular time
interval, i.e., the calibration cycle. Logic circuit 42 (which
receives an output of the comparator) outputs a logic signal that
controls several switches, e.g., switches S19, S20 and S21, which
switches selectively connect corresponding secondary amplifier
devices, e.g., secondary amplifier devices 29, 31 and 33, in
parallel with a primary amplifier device, e.g., primary amplifier
device 66.
At the end of a calibration cycle, when the intentionally created
offset voltage introduced by the secondary amplifier devices should
approximate the inherent dc-offset voltage (represented by voltage
source 62) of amplifier 40, the final digital output of logic
circuit 42 is held by logic circuit 42 and is used to permanently
maintain the positions of the several switches, e.g., switches S19,
S20 and S21, when amplifier 64 returns to normal operation (e.g., a
gain or attenuation configuration). In this manner, switches S19,
S20 and S21 (by controlling which of secondary amplifier devices
29, 31 and 33 are connected in parallel with primary amplifier
device 66) may be used to calibrate the inherent dc-offset voltage
of amplifier 40 by optimizing the effective size, for example, of a
current mirror transistor within amplifier 40, as is explained in
more detail below.
FIG. 4 is a more detailed diagram of an exemplary embodiment of
circuit 76. As shown, a differential pair of (p-channel) input
transistors M1 and M2 share a common source node 106. The common
source node 106 is connected to and driven by current source 74,
which also is connected to positive supply voltage node VDD.
Complementary input nodes IN- and IN+ are connected to the gates
(i.e., the control electrodes) of transistors M1 and M2,
respectively. The output (i.e., the drain) of transistor M2 is
connected to the drain of transistor M3, which drain is an input to
a current mirror including (n-channel) transistors M3 and M4. The
drain of transistor M3 is connected to the gate of transistor M3 as
well as to the gate of transistor M4, and the sources of
transistors M3 and M4 are connected to ground node GND. The output
of the current mirror comprising transistors M3 and M4, i.e., the
drain of transistor M4, is connected to the output, i.e., the
drain, of input transistor M1 at output node 50.
Since amplifier 64 in circuit 76 is configured as a comparator
during its calibration cycle, the voltage at output node 50 during
this cycle will represent either logic high or logic low, depending
on the value of the voltages applied to input nodes IN+ and IN-.
Logic circuit 42 includes buffer inverters 71 and 72, up/down
counter 68, and decode logic circuit 70. Output 50 is connected via
switch S18 (which is closed during the calibration cycle) and
buffer inverters 71 and 72 to the up/down selection input 56 of
up/down counter 68. Up/down counter 68 also receives a signal LOGIC
ENABLE 122 and a clock (CLK) signal at inputs 44 and 46,
respectively. When the signal LOGIC ENABLE 122 at input 44 is
asserted, e.g., during a calibration cycle, up/down counter 68,
responsive to a rising edge of the clock signal at input 46, will
count up or down, depending on the state of the up/down selection
signal at input 56.
The four-bit (for example) output count (i.e., bits DO, D1, D2 and
D3) of up/down counter 68 is passed to decode logic circuit 70,
which circuit decodes the output count and outputs seven individual
control lines at output 58. Each of these individual control lines
carries a logic high or a logic low signal which is used to control
one of seven switches S3-S9 (three of which could correspond with
switches S19, S20 and S21 in FIG. 3) within amplifier 64, as is
explained in more detail below. In addition, the most significant
bit (D3) and the complement of the most significant bit (D3') of
the output count are used to control switches S1 and S2,
respectively, within amplifier 64.
Each of (n-channel) transistors M5-M11 in circuit 76 has a channel
length that is similar to the channel length of transistor M4, but
has a channel width that is substantially smaller than the channel
width of transistor M4. In an exemplary embodiment, the channel
width of transistor M4 is approximately one hundred times the
channel width of each of transistors M5-M11. The sources and gates
of each of transistors M5-M11 are connected, respectively, to
ground node GND and to the gate of transistor M4. The drain node of
each of transistors M5, M6, M7, M8, M9, M10, and M11 is connected
to one pole of a corresponding one of switches S3, S4, S5, S6, S7,
S8 and S9. By using switches S1-S9 to selectively connect or
disconnect the drain nodes of transistors M5-M11 to or from either
the output of transistor M1 or the output of transistor M2, the
effective size of current mirror transistor M4 is increased or
decreased. This trimming of the effective size of current mirror
transistor M4 will, in turn, change the current flowing through the
current mirror (comprised of transistors M3 and M4, as well as any
of transistors M5-M11 that are switched in parallel with transistor
M4) by discrete amounts. These discrete changes in current mirror
output current will, in turn, alter the dc-offset voltage of the
amplifier by discrete voltage values. Transistor M12 has its drain
connected to the common node between switches S1-S9, its gate
connected to the gate of transistor M4 and its source connected to
ground node GND. Transistor M12 can have a channel length that is
approximately twice the channel length of each of transistors
M5-M11 and is used to add a slight offset to the amplifier when
either switch S1 or S2 is activated, thus stabilizing the output
signal at output 50.
As explained previously, up/down counter 68 and decode logic
circuit 70 cooperate to control switches S1-S9 so as to increment
or decrement the dc-offset correction voltage (created by
transistors M5-M11) in conjunction with the counting up or counting
down, respectively, of up/down counter 68. Without departing from
the scope of the invention, switches analogous to switches S1-S9,
by controlling corresponding devices which are appropriately
connected, also could be used to alter the effective size of any
device in the amplifier so as to compensate for the amplifier's
inherent dc-offset voltage. For example, the effective size of the
other current mirror transistor M3 could be trimmed using switches
comparable to switches S1-S9.
During a calibration cycle, i.e., when amplifier 64 is configured
as a comparator and the signal LOGIC ENABLE 122 at input 44 of
up/down counter 68 is asserted, inputs IN+ and IN- are shorted
together by closing shorting switch S116. Thus, any inherent
dc-offset voltage of amplifier 40 will result in a corresponding
logic state appearing at output 50 of amplifier 64. Up/down counter
68 will count either up or down, responsive to a clock signal at
input 46, depending on the state of the signal at input 56. As the
current in the current mirror (comprised of transistors M3 and M4,
and any of transistors M5-M11 that are switched in parallel
therewith) increases or decreases incrementally along with changes
in the output count of up/down counter 68, at some point in the
counting, the dc-offset voltage created by transistors M5-M11 will
approximate the otherwise inherent dc-offset voltage of amplifier
40, and the signal at output 50 will change states. This change in
state of the signal at output 50 will cause up/down counter 68 to
count in the opposite direction, which, in turn, will cause the
signal at output 50 to change states once again.
Thus, when the inherent dc-offset voltage (represented by voltage
source 62) of amplifier 40 is approximated by the dc-offset voltage
caused by transistors M5-M11, the signal at output 50 will change
states every clock cycle, causing the counter to toggle its least
significant bit until the end of the calibration cycle. At the end
of the calibration cycle, the final digital count of up/down
counter 68 is maintained by up/down counter 68 after the signal
LOGIC ENABLE 122 is deasserted. This maintained data then is used
to engage or disengage each of switches S1-S9 permanently, as
determined by the maintained data, after amplifier 64 is
reconfigured in its normal operational mode (e.g., in a gain or
attenuation configuration). For this calibration circuit to
function properly, the maximum positive and negative offset
compensation voltages which can be produced trimming transistor
M5-M11 (based on the output of logic circuit 42) should be larger
than the expected inherent dc-offset voltage of amplifier 40. Also,
the calibration cycle should be long enough with respect to the
clock (CLK) period such that a sufficient number of clock cycles
occur during the calibration period to ensure that up/down counter
68 will, if necessary, reach its maximum or minimum count during a
single calibration period. In addition, if possible, up/down
counter 68 should be configured such that once its minimum or
maximum count is reached, it maintains that count rather than
wrapping to the maximum or minimum count, respectively.
At the end of a calibration cycle, switch S18 is opened and output
50 of amplifier 64 is reconnected to the circuit in which the
amplifier is used. If output 50 has a logic high voltage present at
the end of a calibration cycle, it might cause a voltage spike at
an output of the circuit when it is reconnected to the circuit.
Therefore, switch S10, which is connected between the outputs of
the differential pair M1 and M2, is used to short output 50 to the
output of transistor M2 just before the calibration cycle
terminates. Because transistor M3 is diode-connected to ground node
GND, the voltage at the output of transistor M2 remains relatively
stable. Thus, shorting output 50 to the output of transistor M2
will reduce the voltage at output 50 to the voltage level at the
output of transistor M2, thereby substantially reducing the
amplitude of the voltage spike which might occur during the
transition out of the calibration cycle.
FIG. 5 shows another circuit 21 according to yet another aspect of
the invention. Circuit 21 is similar to circuit 76 (shown in FIG.
3) except for the manner in which an intentionally created offset
voltage is created with amplifier device 66. Again, amplifier 40 is
converted into a comparator (during a calibration cycle) by opening
switch 12. Switches S16 and S18 also are closed during a
calibration cycle. The closing of switch S16 (i.e., electrically
shorting the inputs of amplifier 40) causes any inherent dc-offset
voltage of amplifier 40 (e.g., the dc-offset voltage represented by
voltage source 62) to force an output of amplifier 40 (operating as
a comparator) to a particular state (e.g., logic high or logic
low). Logic circuit 42 receives the logic high or logic low output
of amplifier 40 at input 56 and, responsive to a clock signal,
increases or decreases the value of a multi-bit output signal at
output 58. Logic circuit 42 could be, for example, an up/down
counter having its up/down select input connected to the output of
amplifier 64.
The multi-bit output of logic circuit 42 is provided to input 48 of
offset compensation circuit 38. Offset compensation circuit 38
could be, for example, a simple DAC that has an input 48 to receive
the multi-bit output signal from output 58 of logic circuit 42 as
well as positive and negative voltage outputs V+ and V-,
respectively, across which an analog conversion of the digital
input signal is provided. Negative voltage output V- is connected
to a reference node having a particular voltage thereon. Positive
voltage output V+ is connected to operating characteristic input
lead 25 of amplifier device 23. Amplifier device 23 could
represent, for example, a transistor in a current mirror of
amplifier 40 and operating parameter control lead 25 could
represent a lead which is connected to the substrate of the
transistor. The output of offset compensation circuit 48, then,
would bias the substrate of a transistor in amplifier 40. Such a
biasing of the substrate of a transistor in amplifier 40 would
cause a distinct alteration of the dc-offset voltage of amplifier
40.
Therefore, if logic circuit 42 is an up/down counter, offset
compensation circuit 38 is a DAC and amplifier device 23 is a
transistor in a current mirror of amplifier 40, then the counting
up or down of logic circuit 42 would cause a stepwise increase or
decrease in the dc-offset voltage introduced by the biasing of the
substrate of the transistor. The calibration sequence used to
calibrate the dc-offset of amplifier 40 in circuit 21 is similar to
that used to calibrate the amplifiers in the circuits shown in
FIGS. 2, 3 and 4. That is, logic circuit 42, e.g., an up/down
counter, would count up or down in response to the logic level at
its up/down select input until the inherent dc-offset voltage of
amplifier 40 is approximated by the offset voltage intentionally
introduced by the biasing of an operating characteristic of
amplifier device 23. After a calibration cycle, then, the final
output of logic circuit 42 may be held so as to maintain the
voltage across outputs V+ and V- of offset compensation circuit 48,
thereby permanently compensating for the dc-offset voltage of
amplifier 40. Switches S16 and S1 8 may thereafter be opened and
switch 12 may be closed so as to reconfigure amplifier circuit 21
in a gain or attenuation configuration.
When an op-amp is configured as a comparator during a calibration
cycle, the output of the op-amp must be switched out of the circuit
in which it is used to avoid having the comparator's output signal,
i.e., a high/low switching signal, influence the output of the
circuit. Such a switching signal would be highly undesirable, for
example, in an audio reproduction application as it would cause
noise to emanate through an output device (such as a speaker).
Instead, it is desirable generally to switch the
amplifier/comparator out of the circuit and to maintain a constant
signal level at the output of the circuit throughout the duration
of the calibration sequence. The output signal level of the circuit
should remain constant: (1) during the time period that the
amplifier/comparator is switched out of the circuit, (2) during the
time period the amplifier/comparator is being used as a comparator
to calibrate the dc-offset, and (3) during the time period that the
amplifier/comparator is switched back into the circuit.
FIG. 6 is a circuit 78 that is designed to reduce voltage glitches
at an output of a circuit when an output of an amplifier is
switched out of and back into the circuit in order to undergo, for
example, offset calibration. As shown, main amplifier 40 in circuit
78, which also could represent amplifier 40 in circuit 36 (shown in
FIG. 2), amplifier 40 (as a part of amplifier 64) in circuit 76
(shown in FIGS. 3 and 4), or amplifier 40 in circuit 21 (shown in
FIG. 5), receives an input signal at inverting input 52 (through
resistor portion 24B of potentiometer 24) from input node 28.
Potentiometer 24 (comprising resistor portions 24A and 24B) is
connected between input node 28 and output node 90 of amplifier
circuit 78 so as to allow the gain or attenuation of amplifier
circuit 78 to be altered by adjusting the position of wiper 34.
Non-inverting input 54 of main amplifier 40 is connected to a
constant voltage reference node, e.g., ground node GND. Catching
amplifier 80 in circuit 78 should be (proportionately)
substantially identical to (although it may be substantially
smaller than) main amplifier 40, as voltage glitches are reduced
most effectively when these two amplifiers match one another (at
least proportionately). Catching amplifier 80 could be, for
example, two hundred times smaller than main amplifier 40 in order
to reduce the power consumption of the circuit when catching
amplifier 80 is switched into the circuit in place of main
amplifier 40, as explained below.
Circuit 78 includes three switches S11, S12 and S13, which are
controlled by switch control signals represented by the symbols
.PHI.1, .PHI.2 and .PHI.3, respectively, shown next to the
switches. Switch S11 is connected between output 96 of catching
amplifier 80 and output node 90 of circuit 78, and is used to
connect catching amplifier 80 to output node 90 during a time
interval when the signal .PHI.1 is asserted. Switch S12 is
connected between output 50 of main amplifier 40 and output node 90
of circuit 78, and is used to disconnect output 50 from output node
90 during a time interval when the signal .PHI.2 is asserted.
Switch S13 is connected between output node 90 and inverting input
52 of main amplifier 40, and is used to place main amplifier 40 in
a muting configuration during a time interval when the signal
.PHI.3 is asserted and the signal .PHI.2 is deasserted. That is,
when switches S12 and S13 are closed, amplifier 40 has its
inverting input connected to its output and, thus, is in a
unity-gain feedback configuration. In such a unity-gain
configuration the voltage at output 50 will follow the voltage at
non-inverting input 54. Since non-inverting input 54 is connected
to a constant voltage reference node, e.g., ground node GND, when
switches S12 and S13 are closed, the voltage at output 50 will
remain permanently at a voltage near the reference voltage, e.g.,
ground potential, regardless of the voltage placed at input node
28. Therefore, circuit 78 has a gain of approximately zero between
input node 28 and output node 90. As used herein, an amplifier
circuit in a muting configuration means any amplifier circuit
configured to have a gain of approximately zero between its input
and its output.
Catching amplifier 80 has a conductor connected between its
inverting input 100 and its output 96 to place it permanently in a
unity-gain feedback configuration. Catching amplifier 80 also has
its non-inverting input 102 connected to a constant voltage
reference node, e.g., ground node GND. Thus, when switch S11 is
closed and switch S12 is open, output 90 of circuit 78 will be
driven by catching amplifier 80 and, as such, circuit 78 will be
configured to have a gain of approximately zero between input node
28 and output node 90. Therefore, amplifier circuit 78 may be
placed in a muting configuration either by closing switch S13 (when
switch S12 is closed) or by closing switch S11 (when switch S12 is
open).
The signals .PHI.1, .PHI.2 and .PHI.3 are shown in the timing
diagrams of FIGS. 8 and 9, along with several other signals. These
signals are shown on the same time scale (horizontal axis) for
comparison. The physical placement of one signal above another does
not denote that one attains a higher magnitude than the other
signals. The signal SHORT 120 in FIGS. 8 and 9 represents the
signal controlling the operation of switch S10 in FIG. 4, the
signal LOGIC ENABLE 122 represents the signal at input 44 of logic
circuit 42 (shown in FIGS. 2, 3, 4 and 5), and the signal VOUT 124
represents the output voltage at the output node 90 of circuit 78
(or of circuit 108 (shown in FIG. 7)). Although the signal VOUT 124
(shown in FIGS. 8 and 9) is the output voltage resulting from a
circuit simulation of circuit 108, a simulation of circuit 78
should yield a similar result. Each of switches S11, S13 and S10 is
closed when its respective signal .PHI.1, .PHI.3 and SHORT 120 is
asserted, while switch S12 is opened when the signal .PHI.2 is
asserted.
The timing diagram of FIG. 8, in conjunction with FIG. 6, shows how
circuit 78 enters its calibration cycle. All of the signals are
deasserted prior to entering a calibration cycle. First, the signal
.PHI.3 is asserted so as to place amplifier circuit 78 in a muting
configuration (by closing switch S13). Secondly, the rising edge of
the signal .PHI.1 causes switch S11 to close, thereby connecting
output 96 of catching amplifier 80 to output node 90. This closing
of switch S11 causes a slight glitch in the output signal VOUT 124.
After catching amplifier 80 is connected to output 90, the
assertion of the signal .PHI.2 causes switch S12 to open, thereby
disconnecting output 50 from output node 90 and from the catching
amplifier. Since switch S11 is now closed, however, amplifier
circuit 78 is maintained in a muting configuration. A slight glitch
may appear in the output signal VOUT 124 when output 50 is
disconnected from the circuit, as shown. At approximately the same
time that the signal .PHI.2 is asserted, the signal .PHI.3 is
deasserted (causing switch S13 to open), thereby removing the
unity-gain feedback loop from main amplifier 40 and causing
amplifier 40 to run open-loop. Thus, with switches S12 and S13 both
being open, main amplifier 40 is configured as a comparator and may
be used as such to calibrate the dc-offset voltage of main
amplifier 40, as explained above in connection with the
descriptions of circuits 36 (shown in FIG. 2), 76 (shown in FIGS. 3
and 4) and 21 (shown in FIG. 5). Meanwhile, the signal LOGIC ENABLE
122 is asserted so as to enable the up/down counter and/or
associated logic to determine a proper (digitally represented)
offset calibration level for the circuit.
Referring now to FIG. 9, in conjunction with FIG. 6, the previously
discussed signals are depicted as they appear when circuit 78 exits
its calibration cycle. First, the signal LOGIC ENABLE 122 is
deasserted to disable the logic circuit and to permanently register
the output of the up/down counter and/or related circuitry after
calibration. Next, the signal .PHI.2 is deasserted and the signal
.PHI.3 is asserted causing switches S12 and S13, respectively, to
close. Thus, output 50 of main amplifier 40 is reconnected to
circuit node 90 and main amplifier 40 is connected back into a
unity gain configuration, i.e., a configuration having a gain of
one between non-inverting input 54 and output 50 of main amplifier
40. A slight glitch appears in the signal VOUT 124 due to this
reconnection. Meanwhile, after the signal LOGIC ENABLE 122 is
deasserted, but before the signal .PHI.1 is deasserted, the signal
SHORT 120 is asserted, causing the outputs of differential input
transistors M1 and M2 (shown in FIG. 4) of main amplifier 40 to be
shorted together. Next, just before the signal .PHI.1 is deasserted
(causing the catching amplifier to be disconnected from output node
90) the signal SHORT 120 is deasserted (reopening switch S10). The
signal .PHI.1 then is deasserted, causing switch S11 to open and
disconnecting catching amplifier 80 from output node 90. Since
switch S13 is closed, however, amplifier circuit 78 remains in a
muting configuration. Once again, a slight glitch occurs in the
signal VOUT 124 due to the disconnection of catching amplifier 80.
Now, after main amplifier 40 has its dc-offset voltage calibrated,
the signal .PHI.3 may be deasserted and amplifier circuit 78 may
resume normal operation as, for example, a gain or attenuation
circuit.
FIG. 7 is another circuit 108 according to the invention. Circuit
108 is similar to circuit 78 except for the presence of main
amplifier output stage 82 and catching amplifier output stage 84,
and the configuration of the switches to service both stages of the
amplifiers. Catching amplifier output stage 84 should be
substantially identical to (i.e., sized proportionately to) main
amplifier output stage 82, but can be, for example, two hundred
times smaller than main amplifier output stage 82 to reduce the
power consumption of the circuit. Potentiometer 24 (comprising
resistor portions 24A and 24B) is connected between input node 28
and output node 90 of amplifier circuit 108 so as to allow the gain
or attenuation of amplifier circuit 108 to be altered by adjusting
the position of wiper 34. Switch S12 is connected between output 50
of main amplifier 40 and input 92 of main amplifier output stage
82. Switch S12 serves to disconnect output 50 from input 92 during
a calibration cycle. Switch S14 is connected between input 52 of
main amplifier 40 and output 94 of main amplifier output stage 82
to connect amplifier circuit 108 in a muting configuration when
switch S14 is closed, i.e., to configure circuit 108 to have a gain
of approximately zero between input node 28 and output node 90.
Switch S11 in connected between input 98 of catching amplifier
output stage 84 and input 92 of main amplifier output stage 82, and
switch S15 is connected between output 88 of catching amplifier
output stage 84 and output 94 of main amplifier output stage 82.
Output 96 of catching amplifier 80 is connected to input 98 of
catching amplifier output stage 84, a conductor is connected
between inverting input 100 of catching amplifier 80 and output 88
of catching amplifier output stage 84, and non-inverting input 102
of catching amplifier 80 is connected to a constant voltage
reference node, e.g., ground node GND. Since the two-stage catching
amplifier is so configured, when switches S11 and S15 are closed
and switch S12 is opened, amplifier circuit 108 will be held in a
muting configuration by the two-stage catching amplifier.
As in circuit 78, the symbols .PHI.1, .PHI.2 and .PHI.3 beside the
switches in amplifier circuit 108 represent signals that control
the operation of the switches. These signals are shown in FIGS. 8
and 9, along with the other signals discussed in conjunction with
FIG. 6. The operation of circuit 108 is nearly identical to that of
circuit 78. First, switch S14 is closed responsive to the assertion
of the signal .PHI.3, placing amplifier circuit 108 in a muting
configuration. Next, the assertion of the signal .PHI.1 causes
switches S11 and S15 to close, thereby engaging the catching
amplifier. The signal .PHI.2 then is asserted (opening switch S12),
the signal .PHI.3 is deasserted (opening switch S14), and the
signal LOGIC ENABLE 122 is asserted. In this manner, the catching
amplifier holds amplifier circuit 108 in a muting configuration as
the main amplifier 40 is disengaged from the circuit, converted
into a comparator, and calibrated. After the calibration sequence
is complete, the signal LOGIC ENABLE 122 is deasserted, causing the
final count of the up/down counter to be maintained at the output
thereof. Just after the signal LOGIC ENABLE 122 is deasserted, the
signal SHORT 120 is asserted (closing switch S10 in FIG. 4). Next,
the signal .PHI.2 is deasserted (closing switch S12) and the signal
.PHI.3 is asserted (closing switch S14), deconfiguring the main
amplifier from its comparator mode. Finally, the signal SHORT 120
is deasserted (opening switch S10 in FIG. 4) just before the signal
.PHI.1 is deasserted (opening switches S11 and S15), disconnecting
the catching amplifier from the main amplifier. The signal .PHI.3
may subsequently be deasserted, allowing the main amplifier
comprising main amplifier 40 and main amplifier output stage 82 to
operate in a gain or attenuation configuration.
It should be understood that while the embodiments of the present
invention described herein employ up/down counters to produce an
increasing or decreasing digital signal, any logic circuitry
producing a similar result may be used without departing from the
scope of the invention. Also, the dc-offset of an amplifier may be
calibrated with a finer or coarser resolution by using a counter
with a greater or lesser number of bits and/or employing additional
or fewer switches and trimming devices in the circuit. The size of
the trimming devices also may be adjusted to suit a particular
application. In addition, while the amplifier stages shown and
described each comprise a single stage, each stage could likewise
comprise several stages, also without departing from the scope of
the invention.
Having thus described at least one illustrative embodiment of the
invention, various alterations, modifications and improvements will
readily occur to those skilled in the art. Such alterations,
modifications and improvements are intended to be within the spirit
and scope of the invention. Accordingly, the foregoing description
is by way of example only and is not intended as limiting. The
invention is limited only as defined in the following claims and
the equivalents thereto.
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