U.S. patent number 5,361,072 [Application Number 07/843,183] was granted by the patent office on 1994-11-01 for gated fmcw df radar and signal processing for range/doppler/angle determination.
This patent grant is currently assigned to Codar Ocean Sensors, Ltd.. Invention is credited to Donald E. Barrick, Jimmy Isaacson, Peter M. Lilleboe, Belinda J. Lipa.
United States Patent |
5,361,072 |
Barrick , et al. |
November 1, 1994 |
Gated FMCW DF radar and signal processing for range/doppler/angle
determination
Abstract
A lower-frequency compact radar system for wide-angle
surveillance. Direction-finding receive antennas consisting of
colocated orthogonal electric and magnetic dipoles provide target
angles from the radar. The size of this antenna unit is reduced to
the point where internal noise is comparable to external to achieve
maximum compactness. High sensitivity is achieved with an efficient
class of pulsed/gated, linearly swept-frequency waveforms that are
generated and processed digitally. For backscatter radars, close to
50% duty factors are realized. The rules for waveform design and
processing overcome problems of range/Doppler aliasing and/or blind
zones. After mixing in the receiver, processing bandwidths are much
less than RF signal bandwidths, so that simple, inexpensive
personal computers are used for real-time signal processing.
Digital FFT algorithms determine target range and Doppler, and DF
algorithms determine its angles. Frequency hopping can be
incorporated in the waveform design by synchronizing all timing and
sampling functions, allowing spread-spectrum advantages while still
achieving the high sensitivity afforded by coherent processing.
Inventors: |
Barrick; Donald E. (Redwood
City, CA), Lipa; Belinda J. (Portola Valley, CA),
Lilleboe; Peter M. (Los Altos, CA), Isaacson; Jimmy
(Mountain View, CA) |
Assignee: |
Codar Ocean Sensors, Ltd. (Los
Altos, CA)
|
Family
ID: |
25289274 |
Appl.
No.: |
07/843,183 |
Filed: |
February 28, 1992 |
Current U.S.
Class: |
342/133; 342/175;
342/196; 343/793 |
Current CPC
Class: |
G01S
7/35 (20130101); G01S 13/0218 (20130101); G01S
13/42 (20130101) |
Current International
Class: |
G01S
13/00 (20060101); G01S 13/02 (20060101); G01S
13/42 (20060101); G01S 7/35 (20060101); G01S
7/02 (20060101); G01S 013/08 (); H01Q 009/16 () |
Field of
Search: |
;342/432,13,107,133,175,196 ;343/793 |
References Cited
[Referenced By]
U.S. Patent Documents
Other References
Barrick, D. E., "FM/CW radar signals and digital processing," NOAA
Tech. Report ERL 283--WPL 26, U.S. Dept. of Commerce, Boulder,
Colo. 1973. .
Prandle and Ryder, "Measurement of surface currents in Liverpool
Bay by high frequency radar," Nature, vol. 315, pp. 128-131, 1985.
.
Lipa and Barrick, "Least-squares methods for extraction of surface
currents from CODAR crossed-loop data," IEEE J. Oceanic Engr., vol.
OE-8, pp. 226-253, 1983. .
Lipa and Barrick, "Extraction of sea state from HF radar sea echo,"
Radio Science, vol. 21, pp. 81-100, 1986..
|
Primary Examiner: Hellner; Mark
Attorney, Agent or Firm: Codar Ocean Sensors, Ltd.
Claims
We claim:
1. A compact antenna for reception of radar signals, the antenna
being part of a radar signal receiving system which includes a
receiver and a transmission line, the antenna having an electrical
efficiency which is a function of frequency and which causes an
external source to contribute an external noise signal during
detection of a desired signal, the antenna comprising:
two magnetic dipoles with each dipole having a phase center, the
magnetic dipoles being contained in orthogonal planes and having
co-linear phase centers; and
an electric dipole having a phase center, the electric dipole being
contained in a plane which is mutually orthogonal to the planes of
the magnetic dipoles, the phase center of the electric dipole being
co-linear with the phase centers of the magnetic dipoles, wherein
the antenna has a desired electrical efficiency which results in
the external noise signal being substantially equivalent to but of
greater magnitude than an internal noise signal resulting from
sources internal to the receiver, transmission line, and
antenna.
2. The compact radar antenna of claim 1, further comprising:
a third magnetic dipole having a phase center, wherein the phase
centers of the three magnetic dipoles and the electric dipole are
co-located.
3. The compact radar antenna of claim 1, wherein the desired
efficiency of the antenna is achieved by having:
where
F.sub.e is the amount by which internal noise falls below external
noise at the frequency corresponding to the received radar signal
and F.sub.A is proportional to the antenna efficiency,
where ##EQU6## R.sub.R is a radiation resistance of the antenna,
R.sub.L is a remaining resistance of the antenna, transmission line
and/or load, and X.sub.L is a remaining reactance of the antenna
and/or load.
4. The compact radar antenna of claim 1, wherein the magnetic
dipoles are constructed from multiple turns of wire around a
non-permeable core, and further wherein, the total length of the
multiple turns of wire is less than one tenth of the wavelength of
the received radar signal.
5. The compact radar antenna of claim 1, wherein the magnetic
dipoles are constructed from multiple turns of wire around a
permeable core, and further wherein, the total length of the
multiple turns of wire is less than one tenth of the wavelength of
the received radar signal.
6. The compact radar antenna of claim 1, wherein the magnetic
dipoles are constructed from capacitor-tuned loopsticks.
7. A method for determining the azimuth angle to an object, the
method comprising:
receiving a vertically polarized signal from the object by means of
a compact radar antenna, the antenna being part of a radar signal
receiving system which includes a receiver and a transmission line,
the antenna having an electrical efficiency which is a function of
frequency and which causes an external source to contribute an
external noise signal during detection of a desired signal, wherein
the antenna comprises:
two magnetic dipoles with each dipole having a phase center, the
magnetic dipoles being contained in orthogonal planes and having
co-linear phase centers;
an electric dipole having a phase center, the electric dipole being
contained in a plane which is mutually orthogonal to the planes of
the magnetic dipoles, the phase center of the electric dipole being
co-linear with the phase centers of the magnetic dipoles, wherein
the antenna has a desired electrical efficiency which results in
the external noise signal being substantially equivalent to but of
greater magnitude than an internal noise signal resulting from
sources internal to the receiver, transmission line, and antenna;
and processing the received signal to determine the azimuth
angle.
8. The method of claim 7, wherein the processing of the received
signal to obtain the azimuth angle comprises the following
steps:
(1) measuring the voltages produced on each of the magnetic and
electric dipoles upon reception of the vertically polarized
signal;
(2) representing the voltage measured on one of the magnetic
dipoles as S cos .PHI., the voltage measured on the second magnetic
dipole as S sin .PHI. and the voltage measured on the electric
dipole as S, where S is the complex amplitude of the received
signal; and
(3) using the voltages measured on the two magnetic dipoles and the
electric dipole to solve for .PHI., where .PHI. represents the
azimuth angle.
9. The method of claim 7, wherein the desired efficiency of the
antenna is achieved by having:
where
F.sub.E is the amount by which internal noise falls below external
noise at the frequency corresponding to the received radar signal
and F.sub.A is proportional to the antenna efficiency,
where ##EQU7## R.sub.R is a radiation resistance of the antenna,
R.sub.L is a remaining resistance of the antenna, transmission line
and/or load, and X.sub.L is a remaining reactance of the antenna
and/or load.
10. A method for determining the azimuth and elevation angles to an
object, the method comprising:
receiving an arbitrarily polarized signal from the object by means
of a compact radar antenna, the antenna being part of a radar
signal receiving system which includes a receiver and a
transmission line, the antenna having an electrical efficiency
which is a function of frequency and which causes an external
source to contribute an external noise signal during detection of a
desired signal, wherein the antenna comprises:
three magnetic dipoles with each dipole having a phase center, the
magnetic dipoles being contained in orthogonal planes and including
two orthogonal horizontal magnetic dipoles and one mutually
orthogonal vertical magnetic dipole; and
an electric dipole having a phase center, wherein the phase centers
of the magnetic dipoles and the phase center of the electric dipole
are co-located, and further, wherein the antenna has a desired
electrical efficiency which results in the external noise signal
being substantially equivalent to but of greater magnitude than an
internal noise signal resulting from sources internal to the
receiver, transmission line, and antenna; and
processing the received signal to determine the azimuth and
elevation angles.
11. The method of claim 10, wherein the processing of the received
signal to obtain the azimuth and elevation angles comprises the
following steps:
(1) measuring the voltages produced on each of the magnetic and
electric dipoles upon reception of the arbitrarily polarized
signal;
(2) representing the voltage measured on the vertical magnetic
dipole as F.sub.h cos .mu., the voltage measured on the electric
dipole as F.sub.v cos .mu., the dipoles as F.sub.v cos
.PHI.+F.sub.h sin .mu. sin .PHI. and the voltage measured on the
other horizontal magnetic dipole as-Fv sin .PHI.+F.sub.h sin .mu.
cos .PHI., where F.sub.v is the field strength of the vertically
polarized component of the received signal and F.sub.h is the field
strength of the horizontally polarized component of the received
signal; and
(3) using the voltages measured on the three magnetic dipoles and
the electric dipole to solve for and .mu., where .PHI. represents
the azimuth angle and .mu. represents the elevation angle.
12. The method of claim 10, wherein the desired efficiency of the
antenna is achieved by having:
where
F.sub.E is the amount by which internal noise falls below external
noise at the frequency corresponding to the received radar signal
and F.sub.A is proportional to the antenna efficiency,
where ##EQU8## R.sub.R is a radiation resistance of the antenna,
R.sub.L is a remaining resistance of the antenna, transmission line
and/or load, and X.sub.L is a remaining reactance of the antenna
and/or load.
13. A method of generating a radar waveform and processing a
received radar signal which is the result of the waveform being
reflected by an object to obtain direction, range, and doppler or
radial velocity data for the object, the method comprising:
(1) generating a waveform having a substantially linear increase in
frequency during a sweep repetition time interval, the waveform
having a sweep width or increase in frequency dependent upon a
desired range resolution, wherein the waveform is repeated during
each subsequent sweep repetition time interval, the sweep
repetition interval being equal to the reciprocal of a sweep
repetition frequency;
(2) transmitting the waveform by an antenna which is connected to a
transmitter operating in a pulsed mode, wherein the pulse period is
less than the sweep repetition interval, and further, wherein the
transmitter is operated so that it has a duty factor of up to fifty
percent;
(3) receiving the reflected waveform with an antenna which is
connected to a receiver, the receiver being gated so that it is
turned off during the transmission of the waveform by the
transmitter;
(4) mixing the received waveform with the waveform generated in
(1), thereby producing a baseband radar echo signal having a center
near zero frequency, the radar echo having an offset from zero
frequency which depends on a range to the object and the object's
radial velocity;
(5) sampling the mixed waveform at a sampling rate which is less
than a signal bandwidth corresponding to a desired target range
resolution, wherein the sampling rate is higher than two times a
bandwidth of a modulation of the mixed waveform corresponding to
data which is the target signal information; and
(6) processing the sampled waveform to obtain range, doppler, and
directional information for the object.
14. The method of claim 13, wherein the processing of the sampled
waveform further comprises:
(1) performing a first Fast-Fourier-Transform on the sampled data
to obtain the range of the object; and
(2) performing a second Fast-Fourier-Transform on a subset of the
data obtained from performing the first Fast-Fourier-Transform of
the sampled data to obtain the doppler or radial velocity of the
target.
15. The method of claim 13, wherein the sweep width, B, is related
to the desired range resolution, .DELTA.R, by
where c is equal to the speed of light.
16. The method of claim 13, wherein the sweep repetition frequency,
SRF, is substantially equal to twice a maximum doppler value
expected for the object reflecting the transmitted waveform, the
maximum doppler value being equal to
where v is a maximum radial velocity of the object and .lambda. is
a wavelength of the signal transmitted by the transmitter.
17. The method of claim 13, wherein the pulsing of the transmitter
and gating of the receiver are performed in a manner so that a
first spectral sideband at the pulse repetition frequency is
greater in magnitude than a baseband information signal of all
possible received object signals, thereby providing a method of
generating a radar waveform and processing a received radar signal
which significantly reduces range aliasing.
18. The method of claim 13 wherein the processing of the sampled
waveform further comprises:
sampling the previously sampled waveform to produce a time series
of data, the sampling being performed at a sampling rate equal to M
times the pulse repetition frequency, wherein M is a power of
two;
segmenting the time series into M separate sub-series, wherein each
sub-series has a length substantially equal to 1/M of the
unsegmented time series, and further wherein, each sub-series
consists of every M-th consecutive point of the unsegmented time
series;
performing a first Fast-Fourier-Transform on each of the M separate
sub-series to obtain the range of the object;
performing a second Fast-Fourier-Transform on the data obtained
from performing the first Fast-Fourier-Transform on each of the M
separate sub-series to obtain the doppler or radial velocity of the
target; and
recombining the data obtained from performing the second
Fast-Fourier-Transform on each of the M separate sub-series into a
single series, thereby significantly reducing range or doppler
aliasing when the pulsing of the transmitter and gating of the
receiver cannot be repeated at a rate so that a first spectral
sideband at the pulse repetition frequency is greater in magnitude
than a baseband information signal of all possible received object
signals.
19. The method of claim 13, wherein the receiving antenna further
comprises:
two magnetic dipoles with each dipole having a phase center, the
magnetic dipoles being contained in orthogonal planes and having
co-linear phase centers; and
an electric dipole having a phase center, the electric dipole being
contained in a plane which is mutually orthogonal to the planes of
the magnetic dipoles, the phase center of the electric dipole being
co-linear with the phase centers of the magnetic dipoles, wherein
the antenna has a desired electrical efficiency which results in an
external noise signal being substantially equivalent to but of
greater magnitude than an internal noise signal.
20. The method of claim 19, wherein the desired efficiency of the
antenna is achieved by having:
where
F.sub.E is the amount by which internal noise falls below external
noise at the frequency corresponding to the received radar signal
and F.sub.A is proportional to the antenna efficiency,
where ##EQU9## where, R.sub.R is a radiation resistance of the
antenna, R.sub.L is a remaining resistance of the antenna,
transmission line and/or load, and X.sub.L is a remaining reactance
of the antenna and/or load.
21. The method of claim 13, wherein the receiving antenna further
comprises:
three magnetic dipoles with each dipole having a phase center, the
magnetic dipoles being contained in orthogonal planes and including
two orthogonal horizontal magnetic dipoles and one mutually
orthogonal vertical magnetic dipole; and
an electric dipole having a phase center, wherein the phase centers
of the magnetic dipoles and the phase center of the electric dipole
are co-located, and further, wherein the antenna has a desired
electrical efficiency which results in the external noise signal
being substantially equivalent to but of greater magnitude than an
internal noise signal.
22. The method of claim 21, wherein the desired efficiency of the
antenna is achieved by having:
where
F.sub.E is the amount by which internal noise falls below external
noise at the frequency corresponding to the received radar signal
and F.sub.A is proportional to the antenna efficiency,
where ##EQU10## where, R.sub.R is a radiation resistance of the
antenna, R.sub.L is a remaining resistance of the antenna,
transmission line and/or load, and X.sub.L is a remaining reactance
of the antenna and/or load.
23. The method of claim 19, wherein the processing of the sampled
waveform to obtain the directional information to the object
further comprises:
(1) measuring the voltages produced on each of the magnetic and
electric dipoles upon reception of a vertically polarized
signal;
(2) representing the voltage measured on one of the magnetic
dipoles as S cos .PHI., the voltage measured on the second magnetic
dipole as S sin .PHI. and the voltage measured on the electric
dipole as S, where S is the complex amplitude of the received
signal; and
(3) using the voltages measured on the two magnetic dipoles and the
electric dipole to solve for .PHI., where .PHI. represents the
azimuth angle.
24. The method of claim 21, wherein the processing of the sampled
waveform to obtain the directional information to the object
further comprises:
(1) measuring the voltages produced on each of the magnetic and
electric dipoles upon reception of an arbitrarily polarized
signal;
(2) representing the voltage measured on the vertical magnetic
dipole as F.sub.h cos .mu., the voltage measured on the electric
dipole as F.sub.v cos .mu., the voltage measured on one of the
horizontal magnetic dipoles as F.sub.v cos .PHI.+F.sub.h sin .mu.
sin .PHI. and the voltage measured on the other horizontal magnetic
dipole as-Fv sin .PHI.+F.sub.h sin .mu. cos .PHI., where F.sub.v is
the field strength of the vertically polarized component of the
received signal and F.sub.h is the field strength of the
horizontally polarized component of the received signal; and
(3) using the voltages measured on the three magnetic dipoles and
the electric dipole to solve for .PHI. and .mu., where .PHI.
represents the azimuth angle and .mu. represents the elevation
angle.
25. The method of claim 13, wherein the generated waveform further
comprises:
a substantially linear decrease in frequency during the sweep
repetition time interval, the linear decrease being performed
consecutive to the linear increase in frequency, wherein the linear
increase and decrease in frequency have the same sweep width.
26. The method of claim 25, wherein the processing of the sampled
waveform further comprises:
performing a Fast-Fourier-Transform on the sampled data to obtain
the range and doppler or radial velocity of the object.
Description
BACKGROUND 1. Field of Invention
This invention relates to lower-frequency radars (below microwave),
and embodies improvements by way of compact electronics and
antennas, efficient signal waveforms and their digital
generation/processing, and direction-finding (DF) angle
measurements.
2. Cross-Reference to Related Applications
______________________________________ Field of Search 432/107,
432/131, 432/132, 432/139, 432/195, 432/196 343/726, 342/728,
343/742 U.S. Pat. No. 3,882,506 1975 Mori et al. 343/728 4,053,884
1977 Cantrell and Lewis 432/132 4,172,255 1979 Barrick and Evans
432/107 4,309,703 1982 Blahut 432/132 4,433,336 1984 Carr 343/728
4,896,159 1990 Sabatini et al. 432/131 5,023,618 1991 Reits 432/196
______________________________________
3. Other Publications
Barrick, D. E. (1973), FM/CW radar signals and digital processing,
NOAA Tech. Report ERL 283-WPL 26, U.S. Dept. of Commerce, Boulder,
Col.
Prandle, D. & D. K. Ryder (1985), Measurement of surface
currents in Liverpool Bay by high frequency radar, Nature, vol.
315, pp. 128-131.
Lipa, B. J. & D. E. Bartick (1983), Least-squares methods for
the extraction of surface currents from CODAR crossed-loop data:
Application at ARSLOE, IEEE J. Oceanic Engr., vol. OE-8, pp.
226-253.
Lipa, B. J., & D. E. Barrick (1986), Extraction of sea state
from HF radar sea echo: Mathematical theory and modeling, Radio
Sci., vol 21, pp. 81-100.
4. Description of Prior Art
Lower-frequency radars operating in the MF, HF, and VHF bands, are
useful for a number of applications. Among them are ocean wave and
surface current monitoring, as well as detection of discrete
targets, e.g., aircraft, ships, misstics, etc. The advantages are:
(i) their ability to see beyond the horizon, in both skywave and
surface-wave propagation modes; (ii) the comparable size of their
wavelength with scattering target dimensions, allowing resonance
with the target; and, (iii) lower data rates (resulting from the
low frequency) permit easy digital signal generation and
processing. The radars considered here operate typically three
orders of magnitude lower in frequency than the much more more
common microwave radars. Their consequent disadvantages compared to
microwave radars have to do primarily with larger antenna sizes
required for antenna gains comparable to microwave; their sizes can
be larger by as much as three orders of magnitude. Penalties of
this antenna size are: (i) they become prohibitively costly or too
impractical for most applications; or (ii) if antenna size is
reduced, inadequate target detection sensitivity may result when
standard radar signal formats are used. In addition, the narrower
bandwidth of such radar signals makes them more susceptible to
intercept and jamming.
The normal way a microwave radar determines target direction is to
form a narrow beam. This is done with an aperture many wavelengths
in extent. The beamwidth (in radians) is nearly the wavelength
divided by the antenna length. When beam forming is used with HF
skywave radars, for example, phased array antennas several
kilometers in length are required. Narrow-beam surface-wave radars,
such as the British OSCR for ocean current mapping [Prandle and
Ryder, 1985] use phased array antennas that require more than 100
meters of lineal coastal access, a frequently impractical
constraint. Antennas at HF with sizes the order of a wavelength in
extent (e.g., 10-20 meters) have nearly omni-directional patterns,
and are considered inadequate for accurate radar angle
determination if beam forming and scanning are employed. An
alternate way to determine angle is to employ direction-finding
(DF) principles, which has not commonly been used with radars. U.S.
Pat. Nos. 3,882,506 and 4,433,336 describe hardware implementations
of two crossed single-turn air-loops and a monopole all mounted
along the same axis. However, these loop antennas are still quite
large, e.g., 1-4 meters across at mid-HF; it was believed that the
antennas had to be highly efficient to provide adequate sensitivity
and angle accuracy. The point being missed was that a receive
antenna at lower frequencies does not have to be highly efficient,
and therefore be large, in order to provide maximum possible radar
sensitivity and accuracy. The reason is that external noise
dominates, and the antenna need only possess an efficiency so that
external and internal noise are comparable. Any size or cost
increase to improve efficiency of the receive antennas beyond this
point is wasted. The present invention goes beyond the
hardware-only inventions of the above patents by giving algorithms
for extracting angles, and allowing for more than three colocated
orthogonal elements for radar signal DF.
Since both transmit and receive antenna gains of lower-frequency
radars are less, target detection and location accuracy are worse
if the same waveforms are used as for microwave radars. Microwave
radars use pulse waveforms having low duty factors (the ratio of
pulse width to pulse repetition interval), usually 1% or less. To
gain back the sensitivity and accuracy lost by antenna size,
lower-frequency radars have typically gone to high duty-factor
signals. These radars are usually operated against moving targets,
and Doppler processing is part of the waveform design and
utilization. HF skywave radars, where the transmit and receive
sites are separated by tens of kilometers, use 100% duty factor
signals, i.e., transmitter and receiver are on all the time. Here
the favored waveform has been the simple linearly swept
frequency-modulated continuous-wave (FMCW) signal as described by
Bartick [1973].
When the transmitter and receiver are colocated, as are in compact
backscatter radars at HF, one cannot transmit and receive at the
same time because the strong transmit signal overwhelms the
receiver. Then the highest possible duty factor is 50%; i.e., the
transmitter is turned off while receiving and vice versa. The
design of efficient, non-ambiguity-producing waveforms and digital
signal processing that combine high (e.g., 50%) duty factor
pulsing/gating with modulation formats that give target range, such
as linear FMCW, has not yet been successfully implemented. Three
periodic processes are happening simultaneously: (i) the modulation
used for target range determination, e.g., linear FMCW; (ii) the
pulsing/gating process; (iii) the digital sampling occurring in the
analog-m-digital (A/D) convertor. Each of these three periodic
processes replicates the target echo in the frequency domain, and
when all three happen at once, severe aliasing and ambiguities can
result. For example, the simple linearly swept FMCW signal starts
with a potential range-Doppler ambiguity. Attempting to mitigate
this problem for high speed targets can cause Doppler aliasing,
i.e., two or more possible choices for Doppler. To overcome this,
one would increase the linear sweep repetition frequency, but then
range aliasing can occur (two or more possible choices for target
range). And the pulse/gate repetition frequency itself--if chosen
improperly--can cause both range and Doppler aliasing. Attempts to
eliminate the latter by shortening the pulse will either: (i) lower
the duty factor, or (ii) result in blind zones, where targets will
not be illuminated or seen. Although jittering of any of these
repetition rates--as well as the frequency itself--can unravel the
ambiguities and eliminate blind spots, these inelegant solutions
add complexity and can make Doppler processing very difficult.
Examples are U.S. Patent Nos. 4,896,159; 4,309,703; and
4,053,884.
Another disadvantage of conventional pulse-Doppler or chirp
waveforms with time-domain pulse compression is the high digitizing
and data-processing rates required. The A/D must generally sample
at least twice the RF bandwidth, this latter bandwidth being
dictated by the range resolution desired. Even for lower-frequency
radars, the rates required are typically higher than 200 kHz per
receiver channel. This precludes use of 16-bit convertors, with
resultant limitations on signal dynamic range, and thence clutter
and interference rejection. It also rules out use of widely
available, commercial, inexpensive DSP (digital-signal-processing)
boards.
A third problem with existing pulse radar design is the requirement
for STC (sensitivity-time control) circuitry. These active circuits
change the gain of the receiver circuits rapidly with time after
transmission of each pulse, in order to flatten the dynamic range
between very strong close-in clutter echoes and the most distant
target echoes. The difference between these echoes can exceed 140
dB, far beyond the range of practical linear receiver operation.
STC circuitry increases the cost and complexity of radars
considerably, and hence incentive exists to eliminate this
function.
It is often desirable to spread the radar signal's spectral energy
over a very wide bandwidth. There will be less interference to
others, and in military radars, it makes the signal less
susceptable to detection/intercept and jamming. In the range-only
(no Doppler) 100% duty-factor FMCW radar described in U.S. Pat. No.
5,023,618, this was a primary objective of their signal design and
processing. Nevertheless, such FMCW signals, sweeping slowly and
repetitively over a moderately wide bandwidth--as well as
repetitive pulse signals--are both becoming easier to detect and
jam with modem sophisticated military systems. Hence, a signal that
can be used with a lower-frequency backscatter radar, provides high
sensitivity, range, and Doppler information, avoids aliasing and
ambiguities, has much lower digitizing rates than the RF signal
bandwidth, and is difficult to intercept and jam, has not yet been
implemented. The present invention reveals a waveform design
methodology and its digital generation/processing that accomplishes
these goals.
OBJECTS AND ADVANTAGES
Accordingly, advantages of the present invention have to do with
size and cost reductions, accompanied by sensitivity increases, for
lower-frequency radars that determine target range, Doppler, and
angles. Several of the objects listed below have to do with the
antenna unit, while the remainder relate to the unique waveform and
its digital generation and processing. These objects are:
(a) To provide an efficient, nearly omnidirectional transmit
antenna element.
(b) To provide a compact set of receive antenna elements whose
phase centers are colocated, the elements being orthogonally
oriented electrically small loops and monopoles that have patterns
of electric and magnetic dipoles, with adequate efficiencies so
that external noise is comparable to internal noise.
(c) To provide good receive magnetic dipole efficiency at
ultra-small size by employing ferrite-loaded loopstick designs that
are electrically isolated from each other and from the other
orthogonally oriented elements, as well as from all feedlines.
(d) To provide software methods for DF, i.e., using the antenna
outputs to determine the angles to the target.
(e) To provide efficient operation by digitally generating,
radiating, and processing a signal whose linearly swept FMCW
waveform has a nearly 50% duty factor, but has the ability to:
(i) provide both target range and Doppler with a matched-filter
receiver/processor;
(ii) permit much lower data bandwidth and digital processing rate
than contained in the RF bandwidth required to achieve the desired
range resolution;
(iii) provide the desired range coverage without blind zones;
(iv) taper the energy incident upon and received from targets at
different ranges so as to offset the normal rapid echo signal
falloff with range, or to achieve any other application-specific
distribution of energy with distance;
(v) cover the span of anticipated target ranges and Dopplers
without aliasing, ghosting, or other echo ambiguities;
(vi) provide the coherent range and Doppler processing described
above, but when desired, employ rapid random frequency hopping to
avoid signal intercept or interference to others.
(f) To provide recursive, running averaging of received signal
cross spectra in order to identify and/or remove ship echoes or
other interference from the sea scatter background.
(g) To provide means for using the echo signals to calibrate for
inter-element antenna and receive channel amplitude and phase
drifts with time.
(h) To provide a means for determining target angles after
detection in range-Doppler space using the signals from the various
receive antenna elements, while allowing correction for distorted
antenna patterns and coupling among elements that are inevitable in
practical situations.
A principal obstacle overcome by the present invention is the
signal processing complexities resulting from the gating/pulsing
required in the backscatter mode, since the receiver cannot be
turned on while the transmit signal is being radiated. The objects
here are to reveal a design and implementation methodology that
overcomes the difficulties caused by three periodic processes:
modulation sweeping that provides range information; gating and
pulsing; and digital sampling. Robustness against interference to
others and intercept/jamming is provided by inclusion of spread
spectrum techniques in the signal design. Another obstacle
surmounted here is the requirement for large antennas and/or phased
arrays (because of the long wavelength) in order to provide
accurate target angles. Still further objects and advantages will
become apparent from a consideration of the ensuing description and
drawings.
BRIEF DESCRIPTION OF DRAWINGS
The invention will be explained in detail with reference to the
accompanying drawings.
FIG. 1 is block diagram of an embodiment of the present invention
including the hardware (at the left, with light lines) and software
functions (at the right, with heavy lines). Three receive antenna
channels are indicated here, but only one is shown following the
gate switch, 28; the other two are identical. Software processing
functions are shown through the second FFT, 50, after which both
range and Doppler of the target echoes have been obtained.
FIG. 2 shows the-additional software processing functions that
embody the present invention. These begin with the output of the
second FFT from FIG. 1 for all three receive antenna channels.
Therefore, the functions shown in FIG. 2 are repeated for all three
channels (either simultaneously of sequentially), down to the
second to last boxes (64 and 78 ), where signals from the three
channels are combined to obtain the angle directions required. The
processing channels to the left are representative of embodiments
we have used for sea-surface parameter extraction (e.g., current
and wave information). The processing channels to the right are
embodiments used for hard target detection (e.g., aircraft,
missiles, etc.). The diamond boxes in the middle are stored data
that are required in the processing.
FIG. 3 shows one embodiment of the three-element colocated
crossed-loopstick / monopole receive antenna unit. The box
containing the loopstick elements is weatherproof plastic. On the
board beneath the loopsticks is mounted a printed-circuit
implementation of a preamplifier that makes these two elements
"active antennas." The monopole is a fibreglass whip that screws
onto the top 90. If used also as the transmit antenna, this would
normally be a quarter wavelength at the lower frequencies of
intended operation. Four radial whip elements 106 screw into the
four corners of the base, serving as an electrical counterpoise to
the monopole whip.
FIG. 4 illustrates examples of 50% duty-factor pulsing-gating
arrangements that provide different target echo energy
distributions with distance from the radar. The top strip of each
of the three examples represents the transmit signal pulsing
pattern; a white square indicates a pulse is present at that time
interval, and a black indicates no pulse. Immediately below is the
required gate pattern in the receiver, ensuring that the receiver
is turned off when transmitting, and is turned on when not
transmitting; a white square indicates the receiver is on at that
particular time interval, and a black that the receiver is switched
off. These patterns repeat themselves indefinitely at the end of
the twenty shown here. Twenty intervals in a pattern are not
required, and are an example used here only for illustrative
purposes. The graph shows the target echo energy vs range from the
radar for the three pulse/gate patterns above it, along with the
reference from a conventional shortpulse radar (or alternately, one
with 100% duty factor).
DESCRIPTION OF PREFERRED EMBODIMENTS
System and Hardware Implementation
Radar System
FIG. 1. A high-duty-factor (approaching 50%) gated FMCW signal is
employed. A stored map of the signal parameters to be transmitted
and used in the receiver is downloaded to the digital control unit
18 when the radar is started. This map contains all required
timing, frequency, gating/pulsing, and A/D sampling information.
The actual timing for all of these processes is obtained from the
master clock or oscillator 22 through normal digital dividedown
counters. The direct digital synthesizer (DDS) 20 then generates
the discrete frequencies required of the linear FMCW sweep to be
transmitted, as well as that to be fed to the receiver mixer 32
(the latter may be different from that transmitted in order to
provide an IF offset in the receiver, if desired).
The linear frequency ramp signals to be transmitted and mixed in
the receiver are therefore stair-step frequency-vs-time functions.
This means that at given time increments, the frequency of the
sinusoid being generated changes, but in a way so that the phase
remains continuous at the jump. (This phase continuity is the
natural characteristic of DDS chips.) The frequency jump must be
less than the frequency increment corresponding to each range
cell's spectral spacing; the rules defining the latter are
described later. On the transmit side, the RF signal whose
frequency is thusly being digitally controlled and swept, is passed
to the pulse switch 16. Here, pulsing signals (shown dashed) turn
on and off the RF signal before transmission, usually in
square-wave fashion, i.e., close to 50% duty factor. The duration
for the square wave on/off time is much less than the time required
for the FMCW sweep. This is a departure from the normal "chirp"
radar signals, where the sweep is completed and repeated every
pulse. Power amplifiers 14 (either Class A or C) amplify this
signal to its desired level for radiation. The harmonic filter 12
has an adequate low-pass characteristic so as to remove all
harmonics of the RF signal frequency, eliminating the possibility
of out-of-band interference to others. Finally, the efficient but
(nearly) omnidirectional transmit antenna 10 radiates the signal.
For example, its pattern might be omni-directional in bearing, as
would be obtained from a resonant quarter-wave monopole. Or, it
might illuminate a bearing sector 120.degree., obtainable from a
YAGI two-element monopole array. The pattern depends on the radar
application, i.e., where one expects to find target echoes.
Three or more colocated receive antennas 24 might pass their
signals to preamplifiers 26, which can include some RF bandpass
filtering. The purposes of the preamplifiers are to equalize
approximately the signals among the various receive channels, and
to fix the noise factor for that channel. Often, in the case of the
loopstick, the antenna will be capacitively tuned to the frequency
band of operation, providing both good matching to the line as well
as bandpass filtering that rejects out-of-band signals. Loops are
kept electrically small so as to produce "cosine" or figure-8
antenna patterns, as this is useful in the subsequent DF processes.
The receive monopole element can also serve as the transmit
element, if omni-directional illumination is desired. If this is
the case, a preamplifier is normally not needed because of the
antenna's higher efficiency; the resonant monopole serves as a
natural bandpass filter for out-of-band signal rejection. Next,
each receiver channel is gated, i.e., the channel is turned off by
the gate switch 28 during transmit pulsing to prevent subsequent
receiver overload. Finally, the RF amplifier 30 (which can also
include some bandpass filtering) increases all receive signals to
the desired level; these amplifiers can include some coarse
automatic-gain control (AGC) feedback, so as to keep the signals
within desired dynamic range limits.
In a typical embodiment, the receive signals at this point are
mixed 32 with a local oscillator signal obtained from the DDS 20.
Both contain the linear frequency ramp, and the mixing process
removes the ramp and converts to a coherent baseband signal
(near-zero IF). This embodiment is called a homodyne process.
Because of the time delay from radar targets, the receive signal's
frequency is proportional to target range, resulting from the FMCW
linear frequency sweep employed. As a result of the sweep, separate
I and Q (in-phase and quadrature) channels are not required for
coherent Doppler processing. Very strong, close-in clutter signals
will appear at or near zero frequency after mixing, while distant
target echoes will be offset the most in frequency. This has the
advantage that a properly designed, simple high-pass filter 34 can
suppress the strong clutter signals significantly, thereby reducing
the subsequent dynamic range required. The low-pass filter portion
of 34 is meant to cut off at frequencies beyond the limit expected
of the most distant target echoes, so as to reject out-of-band
noise and interference, thereby ensuring the desired
"match-faltering" receiver function. Also, this low-pass filter
rejects the sidebands of the periodic pulsing/gating process; the
signal beyond this point therefore appears smooth, i.e., its pulsed
nature has been removed. Additional baseband amplifiers 36,
accompanied in some cases with AGC feedback gain control, serve to
stretch the signal span so as to use optimally the dynamic range of
the A/D convertor 38. Usually a 12-bit A/D is adequate to handle
the signal dynamic range spans encountered (especially when
followed by the spectral processing that further extends dynamic
range). The bandwidth and data rate through the A/D of this
properly designed FMCW receiver are typically orders of magnitude
lower than the RF sweep bandwidth required to supply the desired
range resolution; this simplifies and reduces costs of the A/D and
subsequent digital processing.
Signal levels are checked at the output of the A/D. If any of the
three channel's signals (which should be of the same order in
magnitude) are too high (tending to saturate the system) or too low
(generating unnecessary quantizing noise), AGC feedback signals 40
adjust the gain of both RF and baseband, amplifiers accordingly.
Only coarse adjustments (e.g., in 20 dB steps) are required. The
output of the A/D for each antenna channel is a real voltage time
series, which digitally appears as an array of numbers. This
digital time series may require digital mixing 42 and/or digital
low-pass filtering followed by decimation, depending on the
application and corresponding signal design. At step 44, a
multiplicative window/weighting may be applied to the time series
array to reduce subsequent spectral sidelobes; the Hamming is one
example of a window we use that guarantees -43 dB sidelobes or
lower. The length of this time series array is made equal to the
frequency sweep repetition interval, and the number of points is
selected (by choice of the A/D sampling rate) to be a power of two
(take this number to be N), as required by the FFT algorithm. The
output of the first FFT 46 is a complex array, of which we retain
only the first N/2 points. The signals at each of these points then
correspond to radar target echo signals that come from
progressively greater ranges. These N/2 complex array points are
collected as rows of a matrix every sweep repetition interval until
a total of M rows are obtained, over a period of time that
corresponds to the reciprocal of the desired Doppler frequency
resolution. This time is selected so M is a power of two also. Then
another window/weighting vector multiplies each column (or range
cell) of this matrix 48, and a second FFT is performed over each
column 50. The latter provides Doppler processing for each range
cell, and since it operates on a complex input array, the output
preserves the sense of Doppler (positive or negative), just as
though in-phase and quadrature channels had been used. Normally, we
do all of the digital processing steps from 42 to 50 on
digital-signal-processing (DSP) boards that are commercially
available for use in personal computers or their microprocessor
chips; the data rates are generally less than 4096 kHz (or words
per channel), so these boards are more than adequate to handle
several such antenna channels simultaneously, providing real-time
processing. Furthermore, the required FFT and windowing algorithms
are generally available from the DSP-board vendors as library
call-up functions.
Subsequent Processing
FIG. 2. The top of FIG. 2 is the output of the second FFT. Further
digital mixing and/or digital filtering of portions of the
range-Doppler cells may be appropriate, and we allow for this
possibility in 52. Beyond this point two signal-flow algorithmic
paths are shown: (i) to the fight, representing typical embodiments
for hard-target detection (e.g., aircraft, missiles, etc.), and
(ii) to the left, representing ocean-surface parameter extraction
(e.g., currents and waves). Consider the flow to the right first,
for hard targets. In 54 and 56, we allow for additional FFF
processing to provide multiple coherent integration times. For
example, assume the output of the second FFT represents an
integration time of 2 seconds (giving a system noise bandwidth of
0.5 Hz). A four-point FFT done on the same range-Doppler cell of
four consecutive 2-second runs results in an effective 8-second
coherent integration time for that cell, thereby increasing its
signal-to-noise (S/N) ratio by 6 dB. Multiple integration times are
thereby realized; the shorter provides more rapid updates, while
the longer provides higher S/N.
In 58, we allow for the option of measuring antenna channel
amplitude and phase calibration factors directly from the (sea)
clutter echo. Such methods are described in Lipa and Bartick [1983
and 1986]. These factors correct for mismatches in the channels due
to system unknowns and/or gain drifts with time; they are stored 84
for later use or diagnostic messages. The next step involves
detection of the target. A standard CFAR threshholding is employed
60, where signals in given range-Doppler cells are flagged as
potential targets if they lie a predetermined level (in dB) above
the flat noise floor. At this point, the stored amplitude/phase
factors--as well as other antenna calibration parameters including
mutual impedances 86 and patterns 88--are applied in 64 to
determine the angles (e.g., bearing and elevation) to the target,
based on the complex signals from the three antenna channels for
the signals in the cell containing the flagged target echo. One
implementation of this DF algorithm will be discussed subsequently.
It is important to note that--unlike normal microwave
radars--target bearing is determined here after target detection
rather than before. Finally in 66, additional target
identification, classification, sorting, and/or tracking algorithms
may be employed.
The left-side algorithmic flow of FIG. 2 relates to ocean-surface
(e.g., current or wave) measurements made with HF/VHF radars, with
the physical principles described in Lipa and Barrick [1983 and
1986]. The first step 68, however, differs from that presented
earlier, in that two or more running averages are continuously
calculated, rather than one set of averages formed by simply adding
J cross spectra and dividing by J. The latter process was done in
the past when computers could not acquire and process radar data
simultaneously; hence, it was necessary to stop the radar data
acquisition while processing a segment of data. Our present
invention involves continuous radar data acquisition and,
simultaneous processing in a multi-tasking environment, and hence,
continuously updated averages are desirable. The algorithm
accomplishing this is a simple recursive, IIR filter, 68 that takes
a new cross spectral sample, x.sub.i , and adds it to the value of
the running average up to that point, Y.sub.i-1, to get the newest
running average estimate, Y.sub.i, by using the following rule:
Y.sub.i =(1-w).times.Y.sub.i-1 +W.times.X.sub.i. where the weight
"w" is given by 2/(1+J), with "J" being the effective number of
samples included in the running average. For example, a one-hour
running average, where new samples are acquired every 256 seconds,
would have approximately J=14. The first place where these multiple
running averages are used is in t he ship-removal algorithm 70
(which can also be used to identify and track ships if desired).
Ship echoes appear as spectral spikes that can mask and contaminate
portions of the sea echo required for current or wave extraction.
They have the property, however, that they are present in a given
range cell for only a few minutes, their residence time being equal
to the range cell width divided by the ship's radial velocity.
Their echo spikes appear and disappear in short time scales
compared to typical changes in sea-echo cross spectra. The latter
have time constants of hours. Hence, all spectral points in each
newly computed cross spectrum are compared to a long-term average
(e.g., 3-hour); if an echo increase is observed of more than 8-9
dB, that region is suspected as being spurious (a ship or other
noise/interference). In this case, spectral points which fail this
stability test are not included in that particular running average.
Thus, slow changes typical of sea-surface conditions are allowed,
but rapid changes due to a ship traversing the range cell (or a
noise burst) are detected and excluded.
In 72, amplitude/phase correction factors are calculated from the
sea echo, as already described above in 58. In 74, the parts of the
sea-echo used for current extraction (the first-order Bragg echo)
and wave data (the second-order echo) are identified and separated
from each other and from the noise; this process is described in
the above references to Lipa and Barrick. Next in 76, the amplitude
and phase correction factors stored and/or determined from 72 are
applied to the three antenna signals at each spectral point to be
processed. At 78 and 80, we employ algorithms to determine currents
and waves, along with their directions of arrival; the mathematics
and physics behind these algorithms are described in Lipa and
Barrick, cited earlier.
Receive Antenna Unit
FIG. 3. One preferred embodiment of a three-element receive
antenna, contained in a plastic case and covered with a
weatherproof lid 92, is shown in FIG. 3. Ferrite rods 94 of Type
#61 material (supplied by Amidon, Fair-Rite, and other companies)
provide adequate sensitivity below 25 MHz so that external noise is
always higher than internal. The rods are arranged as shown, with a
space in the middle, through which the monopole element passes 90;
this ensures that all three antennas have the same phase center for
vertical polarization. The rods on each side (two or more rods, as
shown, are preferred because their efficiency is higher than one)
are wound with wire 96, sometimes coaxial cable in which the outer
shield carries the radiating current. In the embodiment shown in
FIG. 3, the isolation between loops and monopole is increased
because of the symmetrical splitting of each loop into two halves
arranged on either side of the monopole; also, the orthogonal loops
are layed out in the same plane, rather than in the over/under
configurations of our former designs. Each half of a complete
loop--split on each side of the center--is wound and counterwound
with the same number of turns as shown in 96; this counterwinding
cancels any unsymmetric longitudinal E-field component along the
loop axis that can distort the pattern and/or couple to feedlines.
Balance is achieved by connecting the loop halves in series, with a
tuning capacitor 98 between the two halves; the other ends of each
loop half 100 are fed into differential preamplifiers (a standard
push-pull arrangement with respect to ground) layed out on the
board beneath the rods 102. The preamplifiers have 50-ohm inputs
and noise figures typically better than 4 dB; these are available
from several suppliers. In some embodiments, mismatching into
high-impedance preamplifiers is preferred, where broader bandwidth
or more stability are desired. The vertical whip monopole is fed
against short radial elements 106. Because the monopole is usually
more efficient than the loopsticks, normally it does not require
preamplifiers. If the monopole is also used for transmit (as is the
case in some applications), a special T/R (transmit-receive) switch
is employed to keep the strong transmit signal out of the receiver
front end.
Although ferrite rods have been used for decades as the heart of
lower-frequency loop antennas, a design methodology has not yet
appeared in the literature. We outline a tunedloopstick design
approach of one embodiment for radar applications. The number of
turns is kept sufficiently small that the loop is operating below
its first anti-resonance. Its resulting pattern is a cosine
function (figure-8) vs beating angle. The exact number of turns is
increased until the input resistance is about 50 ohms to provide an
impedance match to the lines. At this point, the input reactance
(measured with a network analyzer) will typically lie between 800
and 2000 ohms, and will be inductive. A series capacitance is
inserted and adjusted until this inductance is canceled by tuning.
The equivalent circuit for the antenna is therefore a loss
resistance (nearly 50 ohms) in series with a very small radiation
resistance, in series with an inductor followed by the series
capacitor of equal and opposite reactance at the center of the band
of desired operation. Loopstick antennas we have thusly designed
and tested have 3-dB bandwidths spanning 7-10% of the band center.
Most of the input resistance is due to losses, mainly ferrite core
dissipation.
The radiation resistance part can be estimated from the
formula:
where A is the cross-sectional area of the ferrite rod core; N is
the number of turns; .lambda., is the wavelength of the radio
signal; and .mu..sub.e is the effective relative permeability of
the ferrite core. Typical 7.5-inch-long rods of #61 material with
dispersed windings have an effective relative permeability of about
70. Efficiency (or power gain) of the loops is then approximately
R.sub.r /R.sub.i , where R.sub.i (=50 ohms) is the input resistance
of the loopstick. In units we have built and tested using #61
ferrite rod material, gains we typically achieve are: -55 dB at 6
MHz; -35 dB at 12 MHz; and -22 dB at 25 MHz. At these frequencies,
external noise typically exceeds internal by at least 60 dB; 40 dB;
and 28 dB, respectively. Hence, the efficiencies of these miniature
receiving antennas provide optimum radar sensitivity, i.e., loop
antennas designed with greater efficiency result in no radar
performance improvement. We have found that increasing the number
of rods in the winding center from one to two typically increases
the gain by 3 dB, when using the 50-ohm criterion to select the
number of turns. A further increase from two to four rods increases
gain by 1-2 dB, indicating diminishing return.
Above 25 MHz, where external noise drops further, we find that
small air-core loops are more effective in keeping internal noise
below external. These are usually single or two-turn capacitively
tuned loops, whose number of turns and size are adjusted to achieve
the best compromise between required bandwidth and gain across this
band. The bandwidth is typically broadened to 10% by reducing their
size so the input resistance (which in this case is nearly the
radiation resistance) is a fraction of the 50-ohm line value,
purposely forcing a mismatch to the line and also broadens the
bandwidth. At 35 MHz, for example, a 2-turn air loop only 30 cm in
diameter has a bandwidth of 3 MHz and a gain of -10 dB, keeping
internal noise levels below external while maintaining compact
size.
DESCRIPTION OF OPERATION
Signal Processing and Waveform Design
Antenna Signal Processing for Angle Determination
Software methodologies are described here, as examples, for both
three and four-element crossed-element receive antennas. Consider
first the three-element configuration of FIG. 3, with ideal
patterns. The voltages received on the two horizontal loopsticks
(which are magnetic dipoles) and the vertical whip (which is an
electric dipole) responding to an incoming, vertically polarized
signal of complex amplitude S from bearing angle .PHI. with respect
to Loopstick #1 axis are v.sub.1 =S cos.PHI., v.sub.2 =S sin.PHI.
for the orthogonal loops; and v.sub.3 =S for the monopole. (Bold
lettering denotes complex quantities.) After target detection,
software divides the two loop signals by the monopole signal so
that signal strength drops out, and the ratios: r.sub.1 3 =v.sub.1
/v.sub.3 =cos.PHI.; and r.sub.23 =v.sub.2 /v.sub.3 =sin.PHI. are
obtained. A simple call to the computer library's ATAN2 function
with these ratios as inputs then gives the desired unambiguous
bearing angle, .PHI.. This assumes that the amplitudes and phases
of the signals into the loops have been adjusted to be matched,
which is accomplished by algorithms 58 and 72 described earlier.
This ratio process is also used as a robust basis for bearing
determination when the patterns are distorted from their idealized
dipole cosine-functions. In this case, the actual bearing patterns
are measured during the calibration phase. These ratios then
comprise the software look-up library for bearing DF, in place of
the idealized "ATAN2" function.
The above process works only for vertically polarized signals,
where only bearing (azimuth) of the target signal is required.
Another embodiment of a radar receive antenna that can obtain
bearing and elevation angles for signals of arbitrary polarization
is described here. In addition to the two horizontal crossed
magnetic dipoles (loopsticks) and vertical electric dipole, a
vertical magnetic dipole (loopsticks) is included and configured
around the whip, to form a four-element unit. The relevant
definitions here are:
The received antenna voltages are then:
(voltage received on vertical magnetic dipole)
(voltage received on vertical electric dipole)
(voltage received on horizontal magnetic dipole #1)
(voltage received on horizontal magnetic dipole #2)
Substituting Equations (1) and (2) into (3) and (4) to eliminate
the unknown field strengths in the latter two yields:
The complex voltages are the quantities observed by the four
antennas. These two complex equations become four real equations in
two real unknowns, beating, .PHI. and elevation angle, .mu.. The
method employed in algorithm 64 to solve this set is maximum
likelihood, which provides the optimum solution for these angles in
the presence of noise and other errors. In order to show here that
these equations do indeed have unique solutions, we derive a
closed-form expression for the elevation angle, and thence the
beating angle. First, form quadrature sums of the voltages from the
two crossed horizontal dipoles, i.e., Equations (5) and (6). (This
can be done either in hardware using 90.degree. directional
couplers, or digitally in software.) Denote by subscripts "R" and
"L" right and left-handed quadrature sums.
Now multiply Equations (7) and (8) each by their complex
conjugates. This gets rid of the complex exponentials, eliminating
.PHI., and allows solution for elevation angle, .mu.. Then add the
resulting equations together and use the identity sec.sup.2
.mu.=1+tan.sup.2 .mu. to obtain the following solution for
elevation angle in terms of the measured voltages: ##EQU1## The
above squares are taken on the absolute values (amplitudes) of the
voltage signals. Using this solution for elevation angle and the
measured voltages, one solves directly for bearing angle, .PHI..
This is done by dividing Equation (7) by (8) and taking the
principal complex square root: ##EQU2##
The above analysis demonstrates that both elevation and bearing
angles can be determined from the four-element antenna unit
described, even when the signal is arbitrarily polarized, and its
polarization state is unknown.
FMCW Waveform Design and Digital Processing Rules: Double FFT
Method
We describe here the rules for the waveform design and its digital
processing that underly the present invention. In the fundamental
embodiment, the frequency of the waveform is linearly swept over an
interval of time called the sweep repetition interval; its
reciprocal is referred to as the sweep repetition frequency, SRF.
The objective of all signal designs herein is to avoid creating
blind zones in radar coverage, and to avoid aliasing, ambiguities,
and/or ghosting in range and Doppler space. In the "double-FFT"
processing to be described now, the SRF, if possible, is taken to
be at least twice the maximum Dopplers expected from the class of
targets under consideration. As an example, assume the radar is to
observe aircraft targets, with maximum speeds of Mach 0.5
(.about.150 m/s). At 20 MHz radar frequency, the maximum Doppler is
therefore 2v/.lambda.=20 Hz, where wavelength, .lambda., is 15 m.
Setting the SRF at 40 Hz meets this criterion. The sweep width, B,
depends on the desired range resolution according to the following
role: B=c/(2.DELTA.R), where c is the speed of light
(3.times.10.sup.5 km/s) and .DELTA.R is the desired range
resolution. Assume for the example that .DELTA.R is to be 3 km;
then B=50 kHz. In the simple embodiment being illustrated, the
frequency is therefore swept in one direction (e.g., upward) from
20 MHz over 50 kHz in 1/40 second; this sweep process is then
repeated every 1/40 second. If no pulsing or gating were employed
(i.e., 100% duty factor or pure CW), the radar's baseband signals
(created after mixing the received target echo signals with the
transmitted linear frequency ramp 32 ) represents consecutive range
cells every 40 Hz, and the spectral region about each 40 Hz
position is the target's Doppler shift of .+-.20 Hz from the range
cell center at the multiple of 40 Hz. Each range cell is spaced at
3 km from the radar, e.g., 0 Hz.fwdarw.0 km; 40 Hz=.fwdarw.3 km; 80
Hz.fwdarw.6 km; . . . ; 600 Hz.fwdarw.45 km. This frequency spread
is referred to here as the baseband information spectrum. The
embodiment of the processing shown in FIG. 1 employs a double-FFT
processing sequence to separate first the target echo space into
range cells 46, and then to process these range cells in a second
FFT for Doppler 50. If all ranges from 0-45 km are desired, then
the first FFT is performed on the time series taken over a single
sweep of 1/40 second, with 32 real points sampled at a
32.times.40=1280 Hz rate. The first 16 complex output points from
this FFT represent 16 range cells every 3 km. The length of the
second FFT depends on the reciprocal of the desired Doppler
resolution. For our example, 512-point second FFT processing will
require output accumulation from the first FFF for each range cell
512.times.(1/40)=12.8 seconds, providing a Doppler resolution
1/12.8=0.078125 Hz. The linear FMCW signal design and processing to
this point for pure CW signals are not new; the principles are
described in the first inventor's reference cited earlier [Barrick,
1973]. It is the addition of pulsing/gating, and the rules and
methods for avoiding ambiguities, that are new and are described
next.
The pulse/gate widths are set according to the following role,
recognizing that this pulse/gate modulation of the signal
replicates the baseband information spectrum at multiples of the
gate repetition frequency (GRF). In simple square-wave on/off
sequencing of transmitter and receiver, the pulse width is made
sufficiently long that targets from the radar to maximum desired
range are seen without interruption. If possible, it is kept short
enough that the GRF is at least twice the information spectral span
near DC containing the desired range cells, in order to avoid
aliasing or ghosting. For example, assume 15 range cells are
desired, from 0 out to 45 km, that occupy the region out to 620 Hz.
If the target at maximum range, 45 km, is to be illuminated 50% of
the time, then the pulse width must be 2.times.45/c=300 .mu.s.
Since the off-time (gate width) is also 300 .mu.s, then the total
repetition interval for the square wave is 600 .mu.s, and its
reciprocal, GRF=1667 Hz. Therefore, the first sideband center of
the pulse/gate modulation lies at 1667 Hz, and the 15 range-cell
span is 1667+620 Hz about this sideband center. The sidebands,
therefore, do not overlap the baseband information spectrum from DC
to .+-.600 Hz, and there is no range-cell aliasing caused by the
gate-modulation sidebands. The low-pass filter 34 in the receiver
baseband section cuts off slightly beyond 620 Hz, to eliminate the
undesired modulation sidebands and noise, making the receiver
implementation a true matched filter. This signal, after low-pass
filtering, can be sampled by the A/D 38 at a rate as low as 1280 Hz
(the Nyquist rate of twice the information bandwidth, 640 Hz).
Therefore, we achieve: (i) the desired optimal match-filter
receiver processing; (ii) no aliasing or ambiguities in range or
Doppler information; (iii) a much lower baseband bandwidth and data
rate (1280 Hz) than the 50 kHz RF bandwidth required for the
specified range resolution; the bandwidth reduction is identically
the ratio of the SRF to the GRF. In conventional "chirp" radar
signals, the linear sweep occurs over a single pulse; here, the
sweep repeats much less frequently than the pulsing. Also, in the
present scheme, the match-filter processing takes place digitally,
in the frequency domain, whereas with chirp, the compression is
produced in the time domain by dispersive delay lines.
Overcoming Range/Doppler Aliasing
Described next is an improvement that overcomes aliasing and
ambiguity problems when the above design goals cannot be realized.
The rules behind this invention are best illustrated with an
example, taken as a variation on that of the above two paragraphs.
Suppose that the maximum target velocity to be encountered is now
Mach 1 (300 m/s) instead of Mach 0.5. The maximum Doppler shifts
for a 20 MHz radar frequency are now .+-.40 Hz. Following the
preceding rules, the SRF must be 80 Hz, and each range cell spans
.+-.40 Hz, centered every 80 Hz from DC. Thus, 15 range cells now
extend to 1200 Hz instead of to 600 Hz. If the pulse/gate width
were designed for a maximum range of 45 km, as before, the
sidebands of the GRF at 1667 Hz would cause overlapping, or
aliasing: the baseband information spectrum extends from DC to 1240
Hz, while its first sideband spans 1667 Hz+1240 Hz. Therefore,
several outer range cells of the 16 are aliased, and recognition of
the proper target ranges is ambiguous.
After mixing to baseband, the echo signals in each channel have
frequency f.sub.R proportional to the range to the target,
resulting from the frequency sweep. An idealized square-wave
modulation, representing the pulse/gate process, gives a signal
representable as: ##EQU3## where A and .PHI. are the signals
arbitrary amplitude and phase, and f.sub.G =GRF is the gate
repetition frequency. The series represents replication of the
baseband signal, A sin (2.pi.f.sub.R t+.PHI.), at the GRF
harmonics. To illustrate the interplay of gating and sampling,
reconsider the prior example of the Mach 0.5 situation, where we
take the GRF to be exactly 1280 Hz (twice the baseband bandwidth
that contains the 16 range cells each 40 Hz wide). This GRF, which
is slightly lower than our previous 1667 Hz, gives maximum target
illumination slightly further out, i.e., at 58.6 km instead of 45
km. Let us sample the signal at exactly this 1280-Hz rate, this
being also the Nyquist rate. Then time "t" is discretized as
t.sub.j =j.DELTA.t, where .DELTA.t=1/f.sub.G is the sampling
interval. The sampled signal is: ##EQU4## Therefore, we have
recovered our desired baseband signal, A sin (2.pi.f.sub.R t.sub.j
+.PHI.), multiplied by a fixed constant, C.sub.s , representing the
summation. No aliasing is encountered.
Now come back to the Mach 1 target-speed example, where the gate
frequency is kept the same, f.sub.G =1280 Hz, to allow desirable
range coverage. The SRF is increased to 80 Hz in order to give
unambiguous Doppler. The baseband information for 16 range cells 3
km in extent spanning 0-45 km thus extends to 1280 Hz. Aliasing now
occurs, since the lower 16 range-cell frequency bins from the first
sideband overlay the 16 that extend upward from DC. Make the A/D
sampling frequency exactly twice this; i.e., 2.times.16.times.80
Hz=2560 Hz, with sampling interval .DELTA.t=1/(2f.sub.G). The
equation for the sampled signal is now: ##EQU5##
Sampling at twice the GRF replaces the 2.pi.n phase factor in the
previous equation by .pi.n. This produces different weightings on
every other time-series point, C.sub.s or C.sub.d, as defined
above. Since these two weighting constants are known (calculable or
measurable), one can divide the even time-series points by C.sub.s
and the odd points by C.sub.d, thereby removing the unequal
weightings that cause the aliasing. Thereby, the original
unweighted time series is recovered; i.e., A sin (2.pi. f.sub.R
t.sub.j +.PHI.), as though no gating had been used, and the
range-gate aliasing is eliminated. This time series is sampled at
2560 Hz, adequate to contain the 16 range cells 80 Hz each,
spanning a total of 1280 Hz. This time series for the single sweep
over 1/80-second, is now ready for the first 32-point FFF.
Processing from here on remains the same as described earlier. The
constants C.sub.s and C.sub.d, needed to remove the aliasing, could
also be measured by injecting test signals into the receiver,
rather than being calculated from the series above. The pure
square-wave modulation, which is what the summations represent, are
distorted by the receiver filters and have their higher harmonics
(terms for larger "n") suppressed/removed by the low-pass filter
34.
Another, often more suitable method for removing these constants
(and thereby the aliasing) is now described. Here, each of the two
sets of time series (for the odd and even time points) is processed
separately. For the above example, each set contains 16 points,
which after the first FFT, yields 8 aliased range bins; i.e., the
0-th and 15-th are aliased; the 1-st and 14-th; the 2-nd and 13-th;
etc. The eight complex output bins--for each of the two separate
odd/even arrays--are then processed as they would have been through
the second FFT, giving Doppler. At this point, suppose a target
echo is found in a given range/Doppler cell. Since range is
aliased, there are two possibilities. This target echo will appear
at the same range/Doppler cell for both of the two sets of odd/even
outputs. This target echo in each will have constants C.sub.d and
C.sub.s as multiplicative factors. (In fact, this is a convenient
way to determine the relative factors, their ratio being all that
is needed.) Then, if the (relative) ratio is divided out of the
target echo, equalizing the echoes, a third FFT 56 (consisting of
two points for our example here) will then "de-alias" them. Suppose
the target appeared in the third bin of 8 in each of the two sets
(at the same Doppler point); this could correspond to the target
lying in either range cell 2 or 13. After the third FFT, if the
output shows up as the first (of two) points, the true range cell
is 2; if the second (of two), the range cell is 13. The result is
the same as though the constants had been divided out first, as
described in the preceding paragraph, and the second FFT taken. The
advantages of the present approach are: (i) the required constants
are more easily determined, as described here; (ii) fewer
operations are necessary, because the de-aliasing described need be
done only on spectral peaks representing targets; (iii) the
aliasing constants C.sub.d or C.sub.s are generally
range-dependent, and this gives a way of determining and applying
the correct constants to the proper bins.
Although this method of removing aliasing was illustrated with an
overlap/aliasing factor of two, the method is general and works for
any integral multiple (e.g., factors of two) of the sampling
frequency. If the Doppler window were 160 Hz instead of 80 Hz, then
the 16 range cells would occupy 2560 Hz, and one would sample at
5120 Hz; there would then be a set of four weights representing to
be determined and removed. Again, either dividing each out
initially, or the third-FFF method (being now a 4-point FFT) would
be employed. And so on.
FMCW Waveform Design and Digital Processing Rules: Single FFT
Method
A second range-Doppler processing method is now described that can
often be used advantageously with certain classes of very
high-speed targets. The double-FFT procedure heretofore described,
applied to this case, would lead to both: (i) very high baseband
information spectral content; as well as (ii) extensive
range-Doppler aliasing for which the de-aliasing methods discussed
above become cumbersome. The method described here uses single-FFT
processing applied to linear FMCW signals having two (consecutive
or simultaneous) sweeps of different rates. In this method, Doppler
becomes the large baseband frequency offset, and range is the
smaller offset, contrary to the prior technique. We illustrate the
design rules with an example. Assume the targets under
consideration have velocities lying between Mach 4 and Mach 10
(e.g., corresponding to ballistic missiles). At 20 MHz radar
frequency, the corresponding span of Doppler frequencies is 160 Hz
to 400 Hz. Examine first a constant-frequency radar signal, with no
sweep applied. After mixing to baseband, this signal is low-pass
filtered 34 to 500 Hz and sampled at a 1024-Hz rate for one second.
An FFT of this signal produces a target spike that is 1 Hz wide,
falling between 160 and 400 Hz, corresponding to the target's
Doppler. No range information is conveyed. Next, assume the signal
is swept linearly over 10 kHz during this one-second period, and
this swept replica is mixed 32 with the received signals, as
before. In the no-sweep case, a Mach-6 target velocity produced an
echo peak (or Doppler) at 240 Hz. The sweep adds or subtracts an
offset (corresponding to whether the sweep was down or up) that is
proportional to the target's range. For example, if the target is
600 km from the radar, this range offset is 40 Hz. Since frequency
resolution in this example is 1 Hz, each spectral resolution bin
corresponds to a 15 km target range shift. If two consecutive
1-second sweeps are applied (up and down), they will produce echoes
with their peaks at 200 Hz and 280 Hz. The mean position of these
two peaks (i.e., 240 Hz)is therefore the target's Doppler. Their
difference (i.e., 80 Hz) gives the target's range. Since there are
80 bins between the two peaks, the range resolution for this
example is 600 km/80=7.5 km.
The sweep design and processing procedures now become clear. If the
up/down sweep embodiment described above is employed, the linear
variation in each sweep is c/(4.DELTA.R) Hz, where "c" is the radio
wave propagation speed and .DELTA.R is the desired range
resolution. The sweep time, "T", determines the frequency shift
offset from the target's pure CW Doppler. A spectral bin of width
.DELTA.F=1/T corresponds to one range cell .DELTA.R wide. This time
is usually determined by the nature of the target; for example: (i)
how long is one willing to wait between target information updates;
and (ii) is the target accelerating sufficiently over time T that
target energy is spread over several bins, negating the
signal-to-noise advantage of longer times? The maximum target
spectral frequency at baseband is the sum of: (i) the maximum
target Doppler shift expected; (ii) the added offset due to maximum
target range, R.sub.m, anticipated, which is F.sub.m
=.DELTA.F.times.R.sub.m /.DELTA.R; and (iii) any intentional IF
offset introduced at the mixer 32. The signal is then sampled at a
rate exceeding twice this sum, according to the Nyquist
criterion.
This single-FFT method works well against one or a few high speed
targets, where clutter and multiple lower speed targets can be
separated by their lower Dopplers. Even with their range offsets
added to Doppler, the undesired, low-speed targets are all
segregated to a well-defined sector of the spectrum after digital
processing. As with the preceding double-FFT methods, aliasing or
ghosting occurs if the GRF is lower than twice the baseband
information spectrum, represented by the sum defined above. In this
case, the de-aliasing methodology described earlier may be
applicable. In other cases this aliasing can be totally
circumvented by using a very high GRF. Since a "blind zone" occurs
at range multiples of c/(2.times.GRF), these are mitigated by the
fact that the high-speed target passes out of a blind region during
the FFT processing period, T. For example, consider square-wave
gating of 10 .mu.s on/10 .mu.s off, corresponding to GRF=50 kHz.
Blind zones are now spaced 3 km apart, and the distance between a
blind zone and an illumination maximum is 1.5 km in range. A Mach 6
target flying at 1.8 km/s will therefore always pass through one
blind zone and through an illumination maximum during each
processing interval, and hence the radar will never be blind to
targets with these higher speeds. Designing Target Echo Strength
Dependence on Range: FIG. 4. We describe in greater detail how the
target echo energy depends on its range, R, using the signal design
and processing rules described above and embodied in the present
invention. This target echo energy seen by the receiver does not
follow the R.sup.-4 law, as it does with conventional radars. In
fact, we describe how the target energy can be purposely tapered
with range in order to provide an illumination profile tailored to
a specific application. Examine first the simple pulsing/gating
example discussed up to now. To achieve uninterrupted illumination
of target-space up to 58.6 kin, wc modulated with a square wave
having on and off times each of 390.625 .mu.s, with a GRF of 1280
Hz. With the aid of FIG. 4, this is depicted by the pulse/gate
sequencing shown in 112. The upper panel of 112 represents the
transmit pulsing, and the lower panel of 112 represents the
receiver gating (with white meaning "on" and black meaning "off").
As required, the receiver is always off while the transmitter is
on, and vice versa. Ten time units therefore represent 390.625 Its
for this example, with one time unit being 39.0625 .mu.s. A target
located at the maximum range for which this square-wave is designed
(i.e., 58.6 kin) sees and radiates the voltage signal 50% of the
time. However, a target closer to the radar will see and radiate
the signal less than this amount. The resulting target-echo
baseband information spectral energy is proportional to R.sup.2
from the radar out to 58.6 km. This partially offsets the normal
R.sup.-4 two-way space loss, leaving an overall target energy
dependence on range at the receiver output of R.sup.-2. The
relative plots of target-echo energy (both in absolute power units
and also in decibels) is shown in the lower part of FIG. 4, with
the symbols (circles) given by 110 being R.sup.-4 (the falloff of a
conventional radar), and the solid dots 112 being the R.sup.-2
falloff of the square-wave gating. Both achieve the same levels for
the same average power at maximum range (10 units=58.6 km). The
slower target-energy decrease with range thereby achieved is
desirable, because conventional radars usually require complex,
active STC (sensitivity-time control) circuitry to clamp down the
receiver gain for targets from close-in ranges, so as not to
overdrive the receiver and to flatten the signals' dynamic range.
We have found that the present invention therefore eliminates the
need for STC circuitry. Further target-energy flattening with
range, if required, is conveniently realized using a simple,
passive high-pass filter 34 at baseband; a properly tailored filter
characteristic will suppress signals at lower frequencies that come
from shorter ranges.
In certain applications it may be desirable to have greater target
energy at shorter ranges than is afforded by the square-wave
pattern of 112 with its R.sup.-2 dependence. This can be done with
different pulsing/gating patterns than the square wave. These still
maintain the 50% duty factor for maximum signal efficiency, and
ensure that transmitter and receiver are never on at the same time;
furthermore, the signals are periodic. Two examples of pulse/gate
patterns that accomplish this are shown in FIG. 4 as 114 and 116.
In 116, energy at short ranges is greater than the square wave by
12 dB, while at maximum range it suffers only a 2 dB drop; it is
worse at range unit 7 by about 10 dB, however. In 114, the signal
at maximum range is 8 dB worse, but has comparable or higher energy
than the square-wave 112 up to range unit 7. The energy for these
on-off patterns is the square of the convolution of the upper and
lower strips, and this is used therefore to design range energy
tapers in a methodical way. In this mathematical convolution, white
pulses are ones and black are zeros. The zero-range position output
from the convolution corresponds to the overunder registration
shown; maximum range occurs when the lower strip is slid ten units
with respect to the upper (for the ten-bit code exemplified
here).
Incorporating Spread-Spectrum Method into Waveform Design and
Processing
A final invention is now described that allows spread-spectrum
radar operation along with any of the waveforms discussed above.
The embodiment examined here effectively causes the frequency that
is downloaded from the digital control unit 18, generated in the
direct digital synthesizer 20, radiated, and mixed with the
received signals to be jumped in a completely random manner at
short time intervals. These time intervals are much shorter than
the overall processing times required for range-Doppler
determination, and could typically be as small as the gate interval
(e.g., 300 .mu.s). If such rapid, random hopping is done over a
wide band, intercept by others is difficult. Yet the invention
achieves the advantages of the same coherent processing gain--and
the increased S/N sensitivity implied therefrom--as from the
un-hopped versions discussed earlier.
The essence of the method (before sweeping is considered) varies
the gating and sampling frequencies in direct proportion to the
random radar frequency selected. Consider first an unswept but
gated 20 MHz CW radar signal, which before hopping, would have been
square-wave gated at a 1280 Hz rate (390.625 .mu.s on/off periods)
and sampled also at a 1280 Hz rate. Assume for now a single FFT is
applied, 1024 points long, over data collected for a time interval
(1024/1280 Hz)=0.8 seconds. Before processing, a Mach 0.5 target
echo will be a sine wave with a 20 Hz Doppler offset (from DC), and
will be sampled exactly 1280/20=64 times per cycle. After
processing, a spectral peak with 1/0.8=1.25 Hz frequency resolution
will appear in bin number 20/1.25=16 from DC. Now after ten
pulse/gate periods, let the radiated frequency jump to 24 MHz. (Up
to this time, ten samples have been acquired during the ten gate
periods.) The target's Doppler shift will now also jump in direct
proportion: i.e., become 24 Hz instead of 20 Hz. If samples are
collected at the same 1280 Hz rate, the jump in target Doppler
frequency would destroy the advantage of coherent processing, since
the target would appear at a different frequency. Coherent gain is
achieved only if the target stays in one frequency bin during the
processing period. However, let the signal sampling also change at
the instant of the frequency jump, at an exactly proportional
higher rate, to 1280.times.24/20=1536 Hz. There are now exactly
1536/24=64 samples per sine wave for our Doppler shifted target
echo, same as before. Hence, there will be no longer be any
apparent change to the coherent processing after this frequency
jump, because the sampling rate was adjusted in direct proportion.
The GRF should be increased proportionately, to 1536 Hz from 1280
Hz, meaning the on/off times drop from 390.625 .mu.s to
390.625.times.20/24 .mu.s; thus during ten pulses, exactly ten
samples are still collected, like before. Thus, every ten pulses
the frequency can jump randomly, and if the sampling and gating
frequencies are changed in direct proportion, the appearance will
be the same: a continuous sine-wave signal is being sampled and
processed, as though the radar frequency were kept constant.
Although it is important that the frequency be changed at each jump
in a phase-continuous manner, this is also the nature of
direct-digital synthesizers 20, so the system of FIG. 1 meets this
condition automatically.
Additional changes required to the linear frequency sweep (that was
imposed to obtain range) are: (i) the sweep-repetition frequency
(SRF) must be changed; and (ii) the sweep rate (bandwidth change
per sweep interval) must changed; both in exact direct proportion
to the random frequency jump. These ensure that the frequency
offset due to range (after mixing to baseband) is increased so
that--after the proportional sampling increase--the same number of
samples per target-echo sine wave are being collected. Thusly,
coherent processing is maintained after any number of random
frequency jumps by keeping all waveform and processing timing
locked to the jump size. It matters not whether double or single
FFT processing is employed. With this procedure, the actual
processing time may vary (since timing jumps when the random
frequency jumps), but the number of samples in the FFTs always
remains fixed. The one effect that limits the performance of this
(or any spread-spectrum method) is the frequency response of
antennas, the system, and the target echo radar cross section. When
the frequency band across which hopping occurs is so broad that any
of these factors changes significantly, some coherent processing
gain will be lost, if correction is not made. Although the system
factors can be calibrated and corrections for them applied in
software, such is not true for the radar target, whose radar
response must be considered unknown. However, target responses vs
frequency vary much less for low-frequency radars than for
microwave radars, mitigating this issue to some extent.
Nonetheless, this is a tradeoff that must be considered in
selecting the frequency-hopping bandwidth.
SUMMARY, RAMIFICATIONS, AND SCOPE OF INVENTION
Thus the reader will see that the gated FMCW DF lower-frequency
radars of the present invention represent compact, efficient but
simple alternatives to conventional microwave radars for many
commercial and military applications. By employing DF principles on
receive rather than beam forming, target angles are determined with
compact antennas and with high system sensitivity for wide-angle
surveillance missions. Doppler processing, along with range
determination, provides another dimension to the target's
observables. The unique waveform with very high duty factor, along
with its digital generation and processing, permits very low data
rates that allow real-time processing to be done with simple,
inexpensive personal computers. Methods for avoiding and overcoming
range-Doppler ambiguities and blind zones against moving targets
are detailed herein. A technique is included that allows
spread-spectrum radar operation (when this feature is needed), to
avoid signal intercept as well as interference to others.
Although the description above contains many specifities, these
should not be construed as limiting the scope of the invention, but
rather as exemplifications of presently preferred embodiments
thereof. For example, although three and four-element DF
crossed-dipole receive antennas were discussed, two or five
elements employing the same principles can also be used to
determine bearing in certain situations. Or, the spread-spectrum
frequency-hopping can be used with 100%--as well as 50% or
lower--duty-factor waveforms.
Accordingly, the scope of the invention should be determined by the
appended claims and their legal equivalents, rather than by the
examples given.
* * * * *