U.S. patent number 5,173,702 [Application Number 06/234,046] was granted by the patent office on 1992-12-22 for all weather tactical strike system (awtss) and method of operation.
This patent grant is currently assigned to Raytheon Company. Invention is credited to Rosario Mangiapane, Jerome H. Pozgay, Benjamin L. Young.
United States Patent |
5,173,702 |
Young , et al. |
December 22, 1992 |
All weather tactical strike system (AWTSS) and method of
operation
Abstract
An AWTSS is shown to be made up of an improved synthetic
aperture radar (SAR) for generating radar maps with various degrees
or resolution required for navigation of an aircraft and detection
of ground targets in the presence of electronic countermeasures and
clutter. The SAR consists, in effect, of four frequency-agile
radars sharing quadrants of a single array antenna mounted within a
radome on a "four axis" gimbal with a sidelobe cancelling subarray
mounted at the phase center of each quadrant. Motion sensors are
also mounted on the single array antenna to provide signals for
compensating for vibration and stored compensating signals are used
to compensate for radome-induced errors. In addition, a single
processor is shown which is selectively operable to generate radar
maps of any one of a number of desired degrees of resolution, such
processor being adapted to operate in the presence of clutter or
jamming signals.
Inventors: |
Young; Benjamin L. (Westford,
MA), Mangiapane; Rosario (Burlington, MA), Pozgay; Jerome
H. (Needham, MA) |
Assignee: |
Raytheon Company (Lexington,
MA)
|
Family
ID: |
22879661 |
Appl.
No.: |
06/234,046 |
Filed: |
December 29, 1980 |
Current U.S.
Class: |
342/17;
342/379 |
Current CPC
Class: |
G01S
13/4472 (20130101); G01S 13/9092 (20190501); G01S
13/4463 (20130101); G01S 13/90 (20130101); G01S
7/36 (20130101); G01S 13/5242 (20130101); G01S
7/4065 (20210501); G01S 7/403 (20210501); G01S
7/4069 (20210501); G01S 7/4034 (20210501); G01S
7/4078 (20210501) |
Current International
Class: |
G01S
7/36 (20060101); G01S 13/00 (20060101); G01S
13/44 (20060101); G01S 13/90 (20060101); G01S
13/524 (20060101); G01S 007/36 () |
Field of
Search: |
;343/18E,1LE,379-382
;342/16,17 |
References Cited
[Referenced By]
U.S. Patent Documents
Primary Examiner: Tobbesing; T. H.
Attorney, Agent or Firm: Mofford; Donald F. Sharkansky;
Richard M.
Claims
What is claimed is:
1. In a monopulse radar system utilizing a four quadrant phased
array antenna and a multi-channel radio frequency receiver having a
separate channel fed by signals from each antenna quadrant and an
auxiliary channel, a sidelobe canceller for nulling jamming signals
received in the sidelobes of the four quadrant phased array
antenna, such sidelobe canceller comprising:
(a) a sidelobe canceling antenna array disposed at the center of
each of the antenna quadrants;
(b) means for forming monopulse sum and monopulse difference
signals from signals received by the sidelobe canceling antenna
array in each quadrant;
(c) switching means for selecting, for further processing, either
the monopulse sum or monopulse difference signals out of the
monopulse sum and monopulse difference forming means;
(d) means, following said switching means, for coupling the
selected monopulse sum or monopulse difference signals to the
auxiliary channel; and
(e) means, following said coupling means, for modifying the
selected monopulse sum or monopulse difference signals and applying
such modified sum or difference signal to each channel in the radio
frequency receiver.
Description
BACKGROUND OF THE INVENTION
This invention pertains generally to air-to-ground guided missile
systems and particularly to systems of such type wherein a
synthetic aperture radar is used to provide a map of optical
quality of the terrain underlying an aircraft so that the position
of ground targets may be determined with precision and an
air-to-ground missile may be accurately guided to impact on any
selected one of such ground targets.
It has been postulated for some years that, in the field of
guidance systems for air-to-ground guided missiles, advantage could
very well be taken of the speed and precision with which radar maps
may be generated by processing echo signals to a synthetic aperture
radar to provide guidance systems having "all-weather"
capability.
Any satisfactory "All Weather Tactical Strike System" (hereinafter
referred to by the acronym "AWTSS") must, in addition to possessing
a guidance capability for air-to-ground guided missiles, have a
capability of functioning as a navigational aid for the aircraft
carrying the AWTSS and as a target acquisition and tracking means.
All of the required functions, further, must be carried out in a
hostile environment, meaning when the aircraft is maneuvering
violently to avoid interdicting fire and when electronic
countermeasures are being taken by the enemy. The fact that
satisfactory operation in a hostile environment is a requisite has,
to date, made it impossible to provide an AWTSS which may be used
with any reasonable confidence of success.
SUMMARY OF THE INVENTION
With the foregoing background of this invention in mind, it is a
primary object of this invention to provide an AWTSS which is
capable of performing, in a hostile environment, all of the
requisite actions of such a system.
A further object of this invention is to provide an AWTSS which
uses a single radar to carry out the desired functions.
The foregoing and other objects of this invention are generally
attained by providing, in a tactical aircraft carrying
air-to-ground missiles, an improved pulse Doppler radar and signal
processor, which radar and processor operate in conjunction with
inertial measuring devices also carried on the aircraft to operate
as desired in any of the modes of operation required for an AWTSS.
Specifically, the contemplated radar and signal processor are
arranged to be substantially immune to electronic countermeasures
and to be operative even when the aircraft equipped with the
contemplated AWTSS is violently maneuvered.
BRIEF DESCRIPTION OF THE DRAWINGS
For a more complete understanding of this invention, reference is
now made to the following description of the accompanying drawings,
wherein:
FIG. 1 is a block diagram showing generally the interconnections
between the major components of the contemplated AWTSS;
FIGS. 2 A and 2B, taken together, constitute a block diagram of the
exciter shown in FIG. 1 to be used;
FIG. 3A is a sketch illustrating how the antenna elements of a
planar array are arranged to provide the quadrants and segments
here contemplated;
FIGS. 3B and 3C are views showing details of construction of the
planar array of FIG. 3A;
FIGS. 4A and 4B, taken together, constitute a block diagram of the
radar synchronizer shown in FIG. 1, somewhat simplified to show how
an exemplary receiver channel is controlled;
FIGS. 5 and 5A are diagrams of an exemplary channel of the radio
frequency receiver showing radome calibration;
FIG. 6 is a block diagram of an exemplary channel of the
intermediate frequency receiver shown in FIG. 1;
FIG. 7A is a sketch illustrating how movement of an aircraft
carrying the contemplated AWTSS imposes limits on the extent of the
field which may be satisfactorily mapped;
FIG. 7B is a sketch illustrating how the antenna of FIG. 1 may be
rotated about an axis to enlarge the field which may be
satisfactorily mapped;
FIG. 7C is a block diagram showing how various elements of the
contemplated AWTSS may be interconnected to compensate for movement
of the aircraft during a dwell;
FIGS. 8A and 8B are sketches showing how the gimbal axes of the
antenna shown in FIG. 1 are arranged and an outline drawing of the
gimbal assembly supporting the antenna of FIG. 1;
FIG. 9A is a block diagram of the radar synchronizer of FIG. 1;
FIGS. 9B through 9F are block diagrams of various portions of the
radar synchronizer shown in FIGS. 1 and 9A;
FIG. 10 is a sketch showing, in a general manner, the geometry of a
tactical situation when the contemplated AWTSS is used to aid in
the navigation of an aircraft over a terrain;
FIGS. 11 and 11A are flow charts showing how signals received from,
respectively, stationary and moving targets are contemplated to be
processed;
FIGS. 12A and 12B are sketches showing how the flight path of an
aircraft and the pulse repetition interval are contemplated to be
controlled during the navigation mode;
FIG. 13 is a sketch showing how the received signal strength from
moving targets varies with spacing between phase centers on the
antenna of FIGS. 3A, 3B and 3C; and
FIG. 14 is a flow chart showing how a high resolution map is
generated.
DESCRIPTION OF THE PREFERRED EMBODIMENT
General (FIG. 1)
Referring now to FIG. 1, the herein contemplated synthetic aperture
pulse Doppler radar system 10 is shown generally to include an
antenna 14, a power distribution network 16, a multichannel radio
frequency receiver (sometimes hereinafter referred to simply as
R.F. receiver 18) and an inertial measuring unit 24, all of which
are mounted on a four-axis gimbal assembly 12 and are to be
described hereinbelow. Suffice it to say here that the antenna 14
has dual polarization and is divided into four separate quadrants
and four segments, as shown in FIG. 3A, with a subarray of sidelobe
cancelling elements in each separate quadrant. Radio frequency
(R.F.) power from a transmitter 22 is fed to the antenna 14 via the
power distribution network 16 which, in the receive mode of
operation, directs the signals received by each antenna quadrant to
the R.F. receiver 18. It is noted here in passing that the output
signals from the subarrays are multiplexed into a single receiver
channel in a manner which will be described in detail hereinbelow.
The R.F. receiver 18 is effective to heterodyne the received
signals to suitable intermediate frequency (I.F.) signals which are
subsequently passed to an I.F. receiver 20 mounted off the
four-axis gimbal assembly 12.
The inertial measuring unit 24 (sometimes hereinafter referred to
as the head mounted IMU 24) comprises a pair of cross
accelerometers, a boresight accelerator located on the back center
of the antenna and three mutually orthogonal rate gyroes. In
addition, a total of five vibration-sensitive piezoelectric
accelerometers (not shown) are provided directly on the rear face
of the antenna 14, one at the geometric center of such face and one
at the center of each antenna quadrant. The output signals from the
piezoelectric accelerometers (not shown) are passed to the R.F.
receiver 18 wherein they are processed in a manner to be described
in detail hereinbelow ultimately to phase-compensate the received
signals to eliminate vibration-induced sidebands.
The output signals from the I.F. receiver 20, converted to digital
format, are sent to a signal processor 26 wherein motion
compensation and spectrum analysis are performed in accordance with
a Fast Fourier Transform (FFT) algorithm. The resulting data are
passed to a digital data processor 30 which serves as the command
and control computer for the radar system 10 and generates, inter
alia, angle pointing commands for the four-axis gimbal assembly 12
and timing commands for a radar synchronizer 34. It is noted here
in passing that the requisite tracking loops (not shown) to
maintain frequency control and to control the antenna 14 are
implemented within the digital data processor 30. Completing the
radar system 10 are an exciter 32 (which generates the transmitted
waveforms as well as the local oscillator signals for the R.F.
receiver 18 and the I.F. receiver 20), a utilization device 40
(which may, for example, be a cathode ray display tube), an
inertial platform 38, and a boresight sensor assembly 36. The
latter two units provide phase rotation multipliers to the signal
processor 26 which are used for motion compensation of the received
signals.
Exciter (FIGS. 2A, 2B)
The transmit, receive and ECCM waveforms required to support the
radar system 10 (FIG. 1) are generated in the exciter 32. The
coherent transmitted signals are either CW waveforms, frequency
modulated (FM) chirp or binary phase modulated waveforms, sometimes
referred to hereinafter as a Burst A waveform. In addition, a
noncoherent waveform, referred to hereinafter as a Burst B waveform
or a "spoof" waveform, may be transmitted as a counter
counter-measure against jammers. Finally, the requisite local
oscillator signals for coherently processing Burst A waveforms are
generated in the exciter 32.
Each kind of Burst A waveforms contains signals which are
frequency-agile from pulse-to-pulse following an ordered set of
four frequencies. Each Burst B waveform is made up of a total of
one to four uncoded, frequency-agile, but noncoherent, pulses which
are interlaced between pulses of a Burst A waveform.
Referring now to FIGS. 2A and 2B, control signals from the radar
synchronizer 34 (FIG. 1) are shown applied to a digital interface
control unit 101. Such control signals are applied to the digital
interface control unit 101 in serial format, non-real time. Control
signals applied to the digital interface control unit 101 are
stored until required by the various subassemblies of the exciter
32. A DATA STROBE pulse, also provided by the synchronizer 34 (FIG.
1), is then effective to cause control signals to be taken out of
the digital interface control unit after having been converted in a
known manner to analog signals whenever required.
Control signals from the digital interface control unit 101 are
applied to a frequency-agile generator 103, a waveform generator
105 and a second local oscillator (designated L.O. 2 generator
107). The output signals from the frequency-agile generator 103 and
the waveform generator 105 are provided to an unconverter 109
(shown in greater detail in FIG. 2B) wherein the RF INPUT signal
for the transmitter 22 (FIG. 1), the first local oscillator signal
(L.O. 1) for the RF receiver 18 (FIG. 1) and the PILOT PULSE signal
are developed.
A 100 MHz COHO signal (the coherent local oscillator signal) is
passed directly from the synchronizer 34 (FIG. 1) to a local
oscillator (L.O. 3 generator 111) to "phase lock" a 100 MHz
voltage-controlled oscillator (not shown) in a conventional manner
to provide a low noise COHO signal having both good long-term and
short-term stability. The narrow band characteristics of the phase
lock loop (not shown) eliminate interference picked up in the cable
(not shown) between the synchronizer 34 and the exciter 32 which
would otherwise appear as jitter on the generated waveforms. The
low noise COHO signal is provided as a reference signal to the
waveform generator 105 and the L.O. 2 generator 107 and, after
appropriate multiplication to 500 MHz, is applied to the
frequency-agile generator 103 and also to phase shifters 351U, 351L
(FIG. 6).
The frequency-agile generator 103 is of conventional design, here
being made up of a total of four voltage-controlled oscillators
(not shown) arranged so that a total of four tunable bandwidths of
from 1 to 1.6 GHz may be achieved. The output frequency of each of
the voltage-controlled oscillators is determined in a known manner
by an associated phase-lock loop (PLL) with the 500 MHz COHO signal
from the L.O. 3 generator 111 as the reference signal. The control
signals to the frequency-agile generator 103 from the digital
interface control unit 101 are converted to equivalent offset
voltages for each of the voltage-controlled oscillators. The
resulting signals out of the voltage-controlled oscillators are
then selectively heterodyned in a conventional manner with a
reference signal at C-band (such signal being generated within the
frequency-agile generator 103 in a conventional manner). The
resulting C-band signals are filtered and amplified to provide a
set of frequency-agile signals to the upconverter 109.
The waveform generator 105, which is also of conventional design,
generated modulating waveforms which are impressed on the output
signals from the frequency-agile generator 103 to produce the input
signals to the transmitter 22 (FIG. 1). Here the waveform generator
105 provides either a binary phase-coded waveform, a FM chirp
waveform or a CW waveform. To produce the binary phase-coded
waveform a first portion of the 100 MHz COHO input signal to the
waveform generator 105 is multiplied by a factor of 11 in a
frequency multiplier to produce a 1.1 GHz signal. Such signal is
band-pass filtered and applied to a PIN diode modulator wherein
phase is switched between 0.degree. and 180.degree. in response to
control signals from the digital interface control unit 101. The
output signals from the PIN diode modulator are passed, via a
single-pole double-throw switch and an amplifier, as an input
signal to the upconverter 109. The single-pole double-throw switch,
in response to control signals supplied by the digital interface
control unit 101, is effective to select the waveform applied to
the upconverter 109. It will be appreciated by those of skill in
the art that a CW waveform for application to the upconverter 109
may be derived from the output of the waveform generator 105 by not
actuating the pin diode modulator. It will also be appreciated
that, as is done here for reasons to be explained, a CW waveform
may be applied to the upconverter 109 by tapping the output of the
bandpass filter.
In order to produce an FM chirp waveform, a portion of the 100 MHz
COHO input signal to the waveform generator 105 is first multiplied
by a factor of 72 in a frequency multiplier to produce a signal at
7.2 GHz. A voltage-controlled oscillator is operated periodically
to sweep the frequency of its output signal from 8.3 to 8.9 GHz
which is mixed with a signal at 7.2 GHz to produce, inter alia, a
signal (a chirp waveform) which periodically varies from 1.1 to 1.7
GHz. After appropriate filtering, only the latter chirp waveform is
passed through the single-pole double-throw switch to the
upconverter 109.
At the upconverter 109 the 8.3 to 8.9 GHz output signal from the
frequency-agile frequency generator 103 is applied as an input
signal to each one of a pair of mixers 113 and 115. The second
input signal to mixer 113 is a CW signal at 1.1 GHz from the
waveform generator 105, such DW signal being taken from the output
of the bandpass filter (not shown) in the waveform generator 105.
The output signals from the mixer 113 are passed, via a single-pole
single throw switch 117, to a low pass filter 119. That filter
allows only the lower sideband signal from the mixer 113 to be
passed so that a set of first local oscillator (L.O.) signals with
frequencies 7.2 to 7.8 GHz appear at the output of the low pass
filter 119 corresponding with the signals out of the
frequency-agile frequency generator 103. The first L.O. signals are
passed, via an amplifier 121 and an isolator 123, to the R.F.
receiver 18 (FIG. 1).
The second input signal to the mixer 115 is the coded signal from
the waveform generator 105 (a CW, phase-coded or chirp waveform).
The output signals from the mixer 115 are passed, via a single-pole
single-throw switch 125, to a high pass filter 127 which is
effective to pass only the upper sideband signals from the mixer
115. Such filtered signals are passed, via directional couplers
133, 139, an amplifier 129 and an isolator 131, to the RF
transmitter 22 (FIG. 1). It will now be appreciated that the
single-pole single-throw switches 117, 125 are actuated (by a
control signal from the digital interface control unit 101) so that
the drive signal to the transmitter 22 (FIG. 1) is "off" when the
RF receiver 18 (FIG. 1) is "on".
The directional coupler 133 and a single-pole single-throw switch
137 are provided to allow Burst "B" waveforms from a Burst "B"
V.C.O. 135 to be passed to the transmitter 22 (FIG. 1). The Burst
"B" generator 135 incorporates a voltage-controlled oscillator
whose frequency differs from any passing through the high pass
filter 127. The actuation of the single-pole single-throw switch
137 is determined by control signals provided by the digital
interface control unit 101.
The second directional coupler 139 is positioned to couple a
portion of the signal to the transmitter 22 (FIG. 1) to a pilot
pulse generator 141. The latter, which is here of conventional
design, is provided to generate a simulated target return signal
for the purpose of calibrating the radar system 10 (FIG. 1). Thus,
the pilot pulse generator 141 is shown to receive from the digital
interface control unit 101, GAIN, FREQUENCY and RANGE commands. The
GAIN command controls the gain of an amplifier (not shown) to
provide simulated target return signals with different amplitudes.
The FREQUENCY command controls a voltage-controlled oscillator (not
shown) whose output signal is mixed with the coupled input signal
in order to provide simulated target return signals with different
Doppler frequencies. The RANGE command controls a variable delay
line (also not shown) which is used to simulate target return
signals from targets at different ranges.
The L.O. 2 generator 107 is also of conventional design. This unit
develops any of the requisite L.O. signals to down-convert received
signals in the I.F. receiver 20 (FIG. 1) to a second I.F.
frequency. To this end, the 100 MHz COHO signal from the L.O. 3
generator 111 is multiplied by a factor of 66 in a frequency
multiplier to produce a 6.6 GHz signal which is applied as a
reference signal to a mixer (not shown). The second input signal to
the latter is obtained from a voltage-controlled oscillator which
is tunable from 8.3 to 8.9 GHz. The control signals to control that
oscillator (as well as a ramp signal required to "dechirp" a
received chirp waveform) are obtained from the digital interface
control unit 101. The signal from the mixer is passed through a 1.7
to 2.3 GHz low pass filter to reject the upper sideband signal from
the mixer, amplified and then passed (via an isolator) to a mixer
315 (FIG. 6) in the F.F. receiver 20 (FIG. 1).
Transmitter (FIG. 1)
The transmitter 22 (FIG. 1) is of conventional design and comprises
a liquid-cooled traveling wave tube (TWT) amplifier. A variable
attenuator (not shown) and a drive amplifier (also not shown), both
of which are controlled by the radar synchronizer 34 (FIG. 1), are
provided on the input to the TWT to provide a power management
capability as well as to ensure that the power from the TWT remains
constant with frequency and the TWT (not shown) does not saturate.
The output power from the TWT is passed via an isolator (not shown)
and a harmonic filter (also not shown) to the power distribution
network 16 (FIG. 1).
Dual Polarization Antenna (FIGS. 3A, 3B, 3C)
Referring now to FIGS. 3A to 3C, the dual polarization antenna 14
comprises an array of 560 dual-polarized, stripline fed annular
slot radiating elements 140 (sometimes hereinafter referred to
simply as elements 140), which are capable of simultaneously
supporting orthogonal linear polarizations. That is to say, in the
transmit mode the elements 140 radiate circularly polarized signals
of either sense, while in the receive mode the received circularly
polarized signals are resolved into orthogonal linearly polarized
signals. The antenna aperture is subdivided into four identical
quadrants 14A, 14B, 14C and 14D, each containing 100 elements, and
four identical segments 14AS, 14BS, 14CS and 14DS, each containing
40 elements. The center four elements of each of the quadrants 14A,
14B, 14C and 14D are combined to form a sidelobe cancelling
subarray (not numbered) for their respective quadrants. It should
be noted here that while the sidelobe cancelling subarrays (not
numbered) are utilized primarily in the receive mode, provision is
made to allow the former to be utilized in the transmit mode. It
should also be noted here that the antenna aperture (not numbered)
is divided into square quadrants rather than serrated quadrants
even though the latter design would produce low quadrant sidelobes
in the 0.degree., 45.degree. and 90.degree. planes of the antenna
14. If the quadrants 14A, 14B, 14C and 14D were serrated the
grating lobes would be randomly distributed to either side of the
principal and diagonal planes of the antenna 14. The square
quadrant configuration was here chosen because, in such a
configuration, the sidelobes are known to lie along the diagonal
planes so that compensation for such sidelobes may be effected.
The dual polarization antenna 14 and its requisite low and medium
power feed networks (not shown) are contained entirely in two
stripline packages 143, 145, respectively. The elements 140 are
annular slots, one free-space wavelength in circumference, with an
inter-element spacing of 0.707 inches by 0.707 inches on a
45.degree. grid. The elements 140 are etched in the upper ground
plane 147 of the first stripline package 143 which also contains
the low power feed network. Mode suppression pins 149 are provided
around each of the radiating elements 140.
The low power feed network (not shown) comprises a plurality of
reactive type, binary 3 dB power dividers of conventional design,
arranged to provide each radiating element 140 with a pair
(referred to as the vertically and horizontally polarized feed
signals) of linearly polarized feed signals in space and phase
quadrature. Each of the reactive type power dividers feeds a total
of eight radiating elements 140 with the power for each such
divider being derived from a medium power feed network contained in
the second stripline package 145. The requisite interconnections
between the low and medium power feed network (not shown) are
provided by means of feedthrough pins 151. Dielectric plugs 153 are
inserted over the feedthrough pins 151 and metallic plates 155 are
provided over the dielectric plugs 153. Mode suppression pins 157
are provided around each of the feedthrough pins 151. Coaxial
connectors 159 are attached in any convenient manner, as by means
of screws (not numbered), to the stripline package 143 containing
the low power feed network (not shown). Coaxial connectors 161 are
similarly attached to stripline package 145 containing the medium
power feed network (also not shown). It should be noted here that
the latter network provides a total of sixteen output signals (two
vertically and two horizontally polarized signals from each one of
the four antenna quadrants 14A, 14B, 14C and 14D) which are passed,
via coaxial cables (not shown), to the power distribution network
16 (FIG. 1). A total of 16 outputs, corresponding to two vertically
and two horizontally polarized signals from the side-lobe
cancelling elements in each of the four quadrants of the antenna
14, are provided by the low power feed network (not shown) and are
passed, via coaxial connectors 159, and coaxial cables (not shown),
to the R.F. receiver 18 (FIG. 1).
The stripline packages 143, 145 are bonded together to form a
composite assembly mounted to an aluminum honeycomb support
structure 163. Such structure then renders the stripline packages
143, 145 impervious to the effects of vibration and, at the same
time, provides a support member for the power distribution network
16 (FIG. 1) and the R.F. receiver 18 (FIG. 1).
A small monopole antenna (not shown) is disposed at the geometric
center of the antenna 145 and is excited by the PILOT PULSE signals
provided by the power distribution network 16 when the antenna 14
is to be calibrated.
POWER DISTRIBUTION NETWORK (FIGS. 4A and 4B)
Before proceeding with a detailed description of the power
distribution network 16 it is noted that, for the sake of
simplicity, only one portion of that network (the portion required
to feed a single antenna quadrant, here quadrant 14A), is shown in
detail. Similarly, only the sidelobe cancelling networks for
antenna quadrant 14A are shown in detail. It should be appreciated
that the sidelobe cancelling networks for the remaining quadrants,
as well as the requisite hardware within the power distribution
network 16 for feeding the remaining antenna quadrants, are
identical to the portion shown and to be described. Finally, before
proceeding, it should be noted that antenna segments 14CS and 14DS
are used only on transmit and, therefore, there are some minor
differences (to be described) between the segment feeds.
Referring now to FIGS. 4A, 4B, power from the transmitter 22 (FIG.
1) is fed to the power distribution network 16 via a four axis
waveguide rotary joint (not shown). A waveguide coupler 231 couples
off a portion of the input power to the power distribution network
16 for antenna segment illumination. It is noted here in passing
that the high power feed portion (not numbered) of the power
distribution network 16 is implemented using half-height aluminum
waveguide networks. Also, because of the high power airborne
environment the waveguide is pressurized. Following the coupler 231
in the antenna segment feed network (not numbered) are a series of
power dividers 233A, 233B and 233C which are here reactive H-plane
tees and which serve to divide the input power and deliver it to a
series of single-pole single-throw waveguide switches 235A, 235B,
235C and 235D. The latter are controlled by control signals from
the radar synchronizer 34 (FIG. 1). It should be noted here that,
for the sake of drawing clarity, all control signals from the radar
synchronizer 34 will be represented simply as small arrows on the
units being controlled and the requisite digital-to-analog (D/A)
converters will not be shown.
The switches 235A . . . 235D provide the radar system 10 (FIG. 1)
with a power management and beam shaping feature which is required
in some modes (to be described in detail hereinafter) of the
system. Suffice it to say here that by controlling the position of
switches 237A . . . 237D in the antenna quadrant feed network (not
numbered) and switches 235A . . . 235D in the antenna segment feed
network (also not numbered), it is possible to transmit through any
portion, or portions, of the antenna 14 (FIG. 3A) required for a
particular mode. The output signals from switches 235C and 235D are
passed, respectively, to waveguide power dividers 239C, 239D which
are also reactive H-plane tees. The output arms (not numbered) of
the latter are terminated in coaxial connectors (not shown) and the
interconnections between antenna segments 14CS, 14DS and power
dividers 239C, 239D are via semi-rigid coaxial cables (not
shown).
As mentioned hereinabove, in certain system operating modes, it is
desirable to transmit through only one or two of the antenna
segments 14AS . . . 14DS. In these modes the waveguide switches
237A . . . 237D in the quadrant feed network (not numbered) as well
as the requisite switches in the segment feed network (also not
numbered) are synchronously switched to their open position so that
the power incident thereon may be reflected back to an isolator
(not shown) within the transmitter 22 (FIG. 1) where adequate
cooling is provided. The switches are synchronously switched to
prevent the dissipation of large amounts of power in the loads (not
numbered) provided on the power dividers 233A . . . 233C and 241A .
. . 241C. The quadrant switching function is not provided by a
single switch preceding power dividers 241A . . . 241C because a
single switch having the requisite power handling capability could
not be packaged within the allocated volume.
The output signals from switches 235A, 235B are split in waveguide
power dividers 239A, 239B and passed via waveguide, three-port
circulators 243A . . . 243D as input signals to antenna segments
14AS and 14BS. The interconnections between antenna segments 14AS,
14BS and the three-port waveguide circulators 243A . . . 243D are
made by semi-rigid coaxial cables (not shown). It should be noted
here that all the segment feed networks (not numbered) are made to
be of equal electrical path length and that two inputs,
corresponding to two orthogonal linear polarizations, are provided
to each of the antenna segments 14AS, 14BS, 14CS and 14DS.
The three port waveguide circulators 243A . . . 243D serve as
duplexers to pass, in the receive mode, the vertically and
horizontally polarized signals from antenna segments 14AS, 14BS to
switch networks 245V, 245H (where the letters V and H refer to the
vertical and horizontal components of polarization as resolved by
the element feed networks (not shown)). It should be noted here
that each of the switch networks 245V, 245H includes a 180.degree.
hybrid (not shown) and, therefore, each such network may, in
response to a control signal provided by the radar synchronizer 34
(FIG. 1) produce an output signal corresponding to: (a) the input
signal received from antenna segment 14AS; (b) the input signal
received from antenna segment 14BS; or (c), the difference between
the input signals received from antenna segments 14AS and and 14BS.
The signals out of switch networks 245V, 245H are passed as input
signals to switching networks 247V, 247H, respectively.
The quadrant feed network (not numbered) includes a series of
high-power waveguide power dividers 241A . . . 241C which are here
"magic-tees" whose E-ports are terminated by loads (not numbered)
and which are effective in splitting the input power into four
equal phase, equal amplitude output signals which are passed to the
waveguide switches 237A . . . 237D. As mentioned hereinabove, only
the hardware required for feeding antenna quadrant 14A will be
described in detail. Thus, the output signal from switch 237A is
passed to a waveguide "magic-tee" power divider 249 wherein it is
split into a pair of equal amplitude and equal phase output signals
which are passed to a pair of three port waveguide circulators
251V, 251H. The remaining two ports on each of the latter are
terminated in waveguide-to-coax adaptors (not shown). Thus, the
output signals from the circulators 251V, 251H are passed to a pair
of stripline power dividers 253V, 253H via coaxial cables (not
shown). The latter dividers provide two pairs of input signals to
the element feed networks (not shown) in antenna quadrant 14A (FIG.
3A).
It is noted here in passing that the transmission line networks
(not numbered) between power dividers 253V, 253H and circulators
251V, 251H contain two pairs of directional couplers 255V, 255H and
257V, 257H. The directional couplers 255V, 255H are used in the
receive mode of operation to inject pilot pulse signals during
calibration of the radar system 10 (FIG. 1). That is to say, since
the herein-contemplated radar system 10 (FIG. 1) utilizes a
so-called "pipeline" receiver wherein the signals received by each
of the antenna quadrants 14A . . . 14D (FIG. 3A) are brought down
to the signal processor 26 (FIG. 1) in separate channels rather
than being combined in a monopulse arithmetic network into sum
(.SIGMA.) and difference (.DELTA.) channels, each of the separate
channels must be calibrated to ensure that phase and amplitude
tracking is maintained within very precise tolerances. Such
calibration is here provided through the use of pilot pulses. The
pilot pulses are obtained from the exciter 32 (FIG. 1) and are
split in a power splitting network 261. The pilot pulses then are
of equal amplitude and phase and, therefore, any detected
differences between the channels may then be compensated for within
the signal processor 26 (FIG. 1).
It should be noted here that one of the output signals from the
power splitting network 261 is passed, via a circulator 262, to the
monopole antenna at the center of the antenna 14 (FIG. 1). As
mentioned briefly hereinabove, that monopole antenna (not shown) is
excited by the applied pilot pulse signal to calibrate the antenna
14 (FIG. 1). In order to minimize the effect of the antenna radome
(not shown) on the antenna 14 (FIG. 1) during the calibration
procedure, the latter is always gimballed to the identical position
for purposes of calibration. In this manner the antenna 14 (FIG. 1)
will always receive the identical reflections from the radome (not
shown) and the effect of the latter may, therefore, be
neglected.
As mentioned hereinbefore, the center four elements of each of
antenna quadrants 14A . . . 14D (FIG. 3A) are combined to form a
sidelobe cancelling subarray for their respective quadrants. These
sidelobe cancelling subarrays are utilized only in the receive mode
and their operation in each of the quadrants is identical with the
exception of an added feature (to be described) provided between
the subarrays in antenna quadrants 14A and 14B. As also mentioned
hereinabove, only the sidelobe cancelling networks (not numbered)
for antenna quadrant 14A will be described in detail. Thus, the
vertically and horizontally polarized signals from the sidelobe
cancelling elements of quadrant 14A are combined in monopulse
arithmetic networks 259V, 259H, respectively. The monopulse sum
(.SIGMA.) and difference (.DELTA.) signals then formed are passed
to switches 261V, 261H which are responsive to control signals
provided by the radar synchronizer 34 (FIG. 1) to gate either the
sum or the difference signals for further processing. The signals
out of the switches 261V, 261H are passed through a pair of
directional couplers 263V, 263H which couple off a portion of the
signals from the switches 261V, 261H. Such coupled signals are
passed, via the switching networks 247V, 247H, through the
auxiliary channel (not shown) of the R.F. receiver 18 (FIG. 1) to
the signal processor 26 (FIG. 1). The .SIGMA. and .DELTA. signals
from the switches 261V, 261H are also passed to sidelobe cancelling
circuits (not numbered) comprising variable attenuators 265V, 265H
and phase shifters 267V, 267H, all of which are of conventional
design and are controlled by control signals supplied by the radar
synchronizer 34 (FIG. 1). The output signals from the sidelobe
cancelling circuits (not numbered) are injected, via the
directional couplers 257V, 257H, into the quadrant receiver
channels (not numbered).
It will now be appreciated by those of skill in the art that
sidelobe cancellers are utilized to reduce the level of jamming
power entering the main channel sidelobes of radar systems. That is
to say, the magnitude and phase of the jamming signals received by
the sidelobe cancelling circuits (not numbered) are modified by the
variable attenuators 265V, 265H and the phase shifters 267V, 267H
in response to the applied control signals so that, when such
modified signals are injected into the quadrant receiver channels
(not numbered) they are of equal magnitude but 180.degree.
out-of-phase with the jamming signals received by quadrant 14A. The
requisite control signals are developed within the digital data
processor 30 (FIG. 1) from analysis of the .SIGMA. channel signals
received by the R.F. receiver 18 (FIG. 1) from the directional
couplers 277V, 277H.
Digressing now for a moment, it will also be appreciated by those
of skill in the art that, in general, the sidelobe antennas also
receive returns from the target being tracked and inject such
target return signals, together with the jammer cancellation
signals, into the channel being nulled. Since the phase of the
injected target return signals is rotated by the sidelobe canceller
phase angle, a bias error will develop. However, because the
herein-contemplated radar system 10 (FIG. 1) is continuously
self-calibrated by pilot pulses to the signal processor 26 (FIG.
1), and because the commanded amplitude and phase of the sidelobe
cancelling circuits (not numbered) are known a priori by the
digital data processor 30 (FIG. 1), the bias error due to sidelobe
cancelling may be neutralized. Thus, the digital commands from the
signal processor 26 (FIG. 1) to the sidelobe cancelling circuits
(not numbered) are also sent to the signal processor 26 (FIG. 1)
for compensation purposes, thereby ensuring that the bias error on
target signals is corrected.
It should be noted here that the herein-contemplated sidelobe
cancelling circuits (not numbered) provide an advantage in terms of
wide instantaneous bandwidth. That is to say, conventional sidelobe
cancellers incorporate cancelling antennas which are physically
displaced from the aperture to be cancelled. Such displacement
results in a time delay which induces a phase dispersion across a
finite bandwidth, thereby limiting the degree of nulling attainable
in such cancelling systems. Because the sidelobe cancelling
subarrays are here centered in the antenna aperture and because the
cancelling is realized at microwave frequencies rather than at I.F.
frequencies as in standard sidelobe cancellers, a unique solution
to the problem of phase dispersion is provided. Further, not only
are the centered sidelobe cancelling subarrays positioned to reduce
phase dispersion, each needs to operate on a smaller aperture (a
quadrant only as opposed to the entire antenna aperture). The
single antenna quadrants 14A . . . 14D (FIG. 3A) have twice the
sidelobe frequency stability as that of the full aperture sum beam
and therefore amplitude dispersal effects are minimized. That is,
the null produced on a small quadrant by a centered sidelobe
canceller is twice as wide in angle as that normally obtained for
full aperture cancelling and, therefore, cancelling over a greater
area is realized. This feature is advantageous for handling
multiple spatially displaced jammers and/or multipath effects.
The herein-contemplated radar system 10 (FIG. 1) also includes a
sidelobe monitor (to be described to prevent false target repeater
type jammers from overloading the target tracking capability of the
system by injecting false target returns through the antenna
sidelobes, thereby preventing true target identification and
tracking. Briefly, when the sidelobe monitor (not numbered) is
utilized, two simultaneous maps of a given area are generated
within the digital data processor 30 (FIG. 1), a first one of such
maps being a conventional map generated from the data received by
the four antenna quadrants 14A through 14D (FIG. 3A), and the
second being generated from the data received through the sidelobe
monitors (not numbered). The two maps are compared within the
digital data processor 30 (Fig. 1) and nay potential targets whose
amplitude is greater in the sidelobe monitor generated map than in
the conventional map are rejected as false targets. A moment's
thought here should make it apparent that, as the false target
repeater type jammers enter the conventional map via the antenna
sidelobes, their amplitude in that map vis-a-vis their amplitude in
the sidelobe monitor generated map will be decreased by the ratio
of the gain of the sidelobe monitor antenna pattern to the gain of
the antenna sidelobes. That is to say, the gain of the sidelobe
monitor pattern is designed to be greater than that of the antenna
sidelobes in the region of the antenna sidelobes.
The output signals from the sidelobe monitor (not numbered) are
passed through the auxiliary channel of the R.F. receiver 18 (FIG.
1). Directional couplers 269V, 269H are included in the
transmission line sections (not numbered) carrying the signals from
the sidelobe cancelling subarrays to the switching networks 247V,
247H. Preceding the directional coupler 269V are a variable
attenuator 271V, a switch 273SV and a phase shifter 273V, while the
directional coupler 269H is preceded by a variable attenuator 271H,
a switch 273SH and a phase shifter 273H. The input signals to the
variable attenuators 271V, 271H are obtained from directional
couplers (not shown) disposed, respectively, in the vertically and
horizontally polarized sidelobe cancelling channels (also not
shown) carrying the sidelobe cancelling signals from antenna
quadrant 14B to the switching networks 247V, 247H. The variable
attenuators 271V, 271H, the switches 273SV, 273SH and the phase
shifters 273V, 273H are all controlled by control signals supplied
by the digital data processor 30 (FIG. 1). It should now be
appreciated by those of skill in the art that the variable
attenuators 271V, 271H and the phase shifters 273V, 273H provide
the sidelobe monitor (not numbered) with a null forming capability
which is useful in cancelling noise jammers which might otherwise
render the latter ineffective.
Digressing now for a moment, it will be appreciated by those of
skill in the art that synthetic aperture radar maps are
particularly vulnerable to dedicated tracking standoff spot
jammers. That is to say, since most synthetic aperture radars
employ waveforms having necessarily long coherent dwell times to
obtain azimuth angle resolution, they normally cannot shift
frequency during a dwell in order to avoid spot jammer noise. Thus,
the herein-contemplated synthetic aperture radar system 10 (FIG. 1)
employs a multiple frequency waveform comprising four coherent
frequencies, as was mentioned hereinabove. Although the use of
multiple frequency waveforms can defeat a single spot jammer,
additional immunity is required to counter the threat of multiple
spot jammers.
The sidelobe cancelling circuits (not numbered) will step in
frequency with the multiple frequency waveform and will form
time-multiplexed spatial nulls in each resolved carrier frequency
band. Thus, the sidelobe cancelling circuits (not numbered) will
self-adapt and neutralize jamming in each carrier band; however,
they will only be successful when the carrier channel is not being
jammed by more than one jammer. Thus, two triple frequency jammers
can double cover a maximum of three frequencies and essentially
desirable the sidelobe cancelling in these channels, but the fourth
carrier channel would still be clear.
Additional immunity from spot jammers is provided in the
herein-contemplated system through the transmission of four proof
frequencies. As explained hereinabove, the exciter 32 (FIG. 1) may
generate interlaced bursts of four frequency agile coherent pulses
and four frequency agile noncoherent pulses which are used as spoof
frequency waveforms. The four noncoherent spoofing frequency
waveforms are narrow pulses to conserve on power and to fit within
the coherent pulse repetition interval. The spoof waveforms are
radiated through antenna segments 14AS and 14BS so that the spoof
peak power will appear larger than that of the coherent synthetic
aperture radar waveforms. The spoof transmissions precede a
synthetic aperture radar dwell time in order to capture a multiple
frequency spot jammer throughout the dwell time.
As was explained above, the switching networks 247V, 247H are
effective to gate selected ones of the signals incident thereon to
the fifth or auxiliary channel of the R.F. receiver 18 (FIG. 1).
Directional couplers 277V, 277H are provided in the output arms
(not numbered) of the former for the purpose of injecting pilot
pulse signals into the auxiliary channel of the R.F. receiver 18
(FIG. 1).
Finally, before proceeding it should be noted that a third
waveguide directional coupler 279 is provided in the input channel
of the power distribution network 16. The former couples off a
portion of the input power to a power splitting network 281 which
divides said portion into five equal amplitude and equal phase
output signals, referred to here as REPLICA signals. The REPLICA
signals are provided to compensate for the effects of vibration on
the waveguide components in the power distribution network 16 which
otherwise would be manifested in the form of false targets or
paired echoes appearing in the Doppler band on each side of the
return signals. That is to say, vibrations of the R.F. components
impress an amplitude and phase modulation on a chirp signal,
thereby producing deviations from ideal quadratic phase
characteristics which distinguish a chirp waveform. A second
problem, existing with any type of transmitted waveform, is that
the phases of successive pulses vary randomly. Because each one of
the REPLICA signals corresponds with a transmitted signal,
successively derived REPLICA signals may be demodulated, converted
to digital numbers and processed to form (for each receiver
channel) a running "nominal" complex conjugate of the transmitted
pulses. When a received signal is multiplied by the nominal complex
conjugate existing as the received signal is being processed in the
I.F. receiver 20 (FIG. 1), the resulting signal is compensated for
deviations in phase or amplitude of the corresponding transmitted
signal.
R. F. Receiver (FIG. 5)
Before proceeding with a detailed description of the R.F. receiver
18 it is noted that while that receiver actually has a total of ten
identical channels (a separate channel for each of the signals from
each of the four antenna quadrants 14A, 14B, 14C, 14D (FIG. 3A) and
two auxiliary channels which are time shared between the sidelobe
cancelling subarrays (not shown) and the antenna segments 14AS,
14BS, only a pair of channels, corresponding to those passing the
signals from antenna quadrant 14A, will be described in detail. It
should be appreciated that the remaining channels are identical to
those to be described.
Referring now to FIG. 5, the vertically and horizontally polarized
input signals from circulators 251V, 251H (FIG. 4A), respectively,
are applied to a pair of bandpass filters 283V, 283H. Such filters,
have waveguide devices, are provided to reject undesired,
out-of-band signals of any nature. Following the bandpass filters
283V, 283H are a pair of waveguide limiters 285V, 285H which are of
conventional design and may, for example, comprise self-activated
TR tube limiters. The latter are provided to limit the input power
to the R.F. receiver 18, thereby preventing burnout of the latter.
Following the limiters 285V, 285H are: (a) a pair of variable PIN
diode attenuators 287V, 287H; (b) a pair of isolators 289V, 289H;
(c) a pair of field effect transistor (FET) amplifiers 291V, 291H;
(d) a second pair of isolators 293V, 293H; and (e) a second pair of
PIN diode variable attenuators 295V, 295H, all of which are of
conventional design and are here fabricated in stripline. Each of
the variable attenuators 287V, 287H, 295V, 295H (all of which are
controlled by control signals provided by the radar synchronizer 34
(FIG. 1), has a 40 dB dynamic range. The inclusion of the variable
attenuators 287V, 287H prior to, and the variable attenuators 295V,
295H after the FET amplifiers 291V, 291H allows spatial filtering,
i.e. three phase center monopulse filtering to be described
hereinafter, to be employed as a counter-countermeasure against an
escort jammer and a delta jammer. The attenuators in such a
tactical situation prevent saturation so that the requisite
linearity of response is maintained to allow spatial filtering to
be carried out in a proper manner. It is noted here that the
bandwidth of the R.F. receiver being described is one-half the
first I.F. frequency. This relationship then inhibits the
appearance of a false I.F. signal from a delta jammer. A bandpass
filter 303 is provided in front of the mixer 305 to protect the
latter from spurious cross products which may be generated when the
FET amplifiers 291V, 291H are forced into saturation.
The output signals from the variable attenuators 295V, 295H are
passed, via phase shifters 297V, 297H, respectively, to a power
combiner 299 wherein they are combined to provide a single output
signal. The variable attenuators 295V, 295H and phase shifters
297V, 297H are, as will be explained hereinbelow, included to
provide the radar system 10 (FIG. 1) with a randome compensation
capability. The output signals from the power combiner 299 are
passed, via a directional coupler 301 and the bandpass filter 303,
to the mixer 305 wherein they are heterodyned with the first L.O.
signal from the exciter 32 (FIG. 2), ultimately to produce an
S-band first I.F. signal. A preamplifier 307 is provided to amplify
the S-band signals from the mixer 305 as such signals are passed to
the I.F. receiver 20 (FIG. 1). The directional coupler 301 is
included to allow injection of the REPLICA of the transmitted
signal (obtained form the power distribution network 281 (FIG. 4B)
into the receiver channel (not numbered). The REPLICA is processed
and stored in the signal processor 26 (FIG. 1) prior to the
reception of any echo signals.
It is here noted that the REPLICAs of the transmitted pulses are
passed through A/D converters after correlation in the I.F.
receiver 20 (FIG. 1) with the second L.O. signal from the exciter
32 (FIG. 1). The resultant digital signals therefore represent the
phase of the L.O. at the time of transmission and the phase errors
due to differential transmission line perturbations between the
antenna 14 (FIG. 1) and the second correlation mixer. After T.B
samples (where B is the A/D sampling rate and T is the pulse
duration) are taken of the REPLICAS, the resulting number is stored
within the signal processor 26 (FIG. 1) and is used to represent a
phase reference for all received signals resulting from that
particular pulse transmission. Therefore, the signal processor 26
(FIG. 1) may cause return signals to be corrected for the effects
of phase modulation and phase distortion as mentioned
hereinbefore.
As also mentioned hereinabove, phase shifters 297V, 297H are
included to provide the radar system 10 (FIG. 1) with a radome
compensation capability. Before proceeding with a detailed
description of this capability the need for a radome compensation
technique will first be explained. Thus, the radar system 10 (FIG.
1) employs a so-called "R.F. Cueing Mode" for the detection,
location and subsequent attack of ground-based threat radars. In
this mode the radar system 10 (FIG. 1) must passively detect a
source of R.F. energy (a threat radar) and locate the source both
in angle and range so that a high resolution synthetic aperture
radar map may be made of the surrounding area. After mapping of
such surrounding area, the radar system 10 (FIG. 1) must then track
the threat radar so that an attack may be mounted and carried out
even when the threat radar ceases to transmit or to scan.
In order to initiate tracking, accurate directional information of
the threat radar must be obtained (for example, by using the
antenna 14 (FIG. 1) as a passive sensor) and the Doppler shift of
return signals from the cell in which such radar is located
derived. The requisite directional information (in elevation and
azimuth relative to the aircraft carrying the radar system 10 (FIG.
1)) determines the initial position to which the antenna 14 (FIG.
1) is to be slued to commence map making of the proper area of the
ground. The Doppler shift of the signals from the cell at such
initial position is, of course, a function of aircraft velocity and
the directional information. With the antenna 14 (FIG. 1) slued to
the initial position, the radar system 10 (FIG. 1) is actuated to
transmit. Range strobes are then used to sweep in range until the
range to the center of the elevation monopulse null is measured.
The radar system 10 (FIG. 1) then commences mapping the area (here
to a 20 square foot resolution), measuring the azimuth and
elevation angular location of each cell in the vicinity f the map
center as the mapping is effected. For each cell the azimuth and
elevation angles, the range and Doppler shift are recorded.
Consequently, a file of data is built up, defining ground
reflectors in the cells about the line-of-sight (LOS) to the threat
radar. With knowledge that mapping was initiated when the antenna
14 (FIG. 1) pointed directly at the threat radar, the cell at the
intersection of the azimuth elevation nulls is first assumed to be
the cell in which the threat radar is located. It will be
appreciated, however, that situations may exist when return signals
from adjacent cells may be so similar to the returns from the first
assumed cell that the threat radar could very well be located in
one of such cells. In such a situation, the cell with the return
signals having the largest signal-to-noise ratio is selected. In
any event, the range and Doppler coordinates of the selected cell
are used to initiate a Doppler tracking loop (not shown) and the
azimuth and elevation tracking loops (also not shown).
It should now be apparent that the accuracy of the foregoing threat
radar tracking technique is dependent upon knowing the precise
angular location of a ground target. To put it another way, the
azimuth and elevation of the threat radar must be determined to
within 0.50 milliradians (mrad) to provide a cross-range accuracy
of 15 meters at a 30 Km range. Therefore, the effects of radome
induced errors must be considered because radome boresight errors
resulting from cross-polarization and refraction effects can
introduce up to .+-.20 and .+-.8 mrad of error, respectively.
Digressing here now for a moment, it will be appreciated by those
of skill in the art that, in general, in a phase monopulse radar
system the 93 channel signal is formed by vectorially adding the
received signals in the four quadrants while the .DELTA. channel
signals are formed by vectorially subtracting the received signals
in opposite halves of the aperture of the antenna. After limiting,
or otherwise equalizing, the amplitude of the received signals
prior to forming the .SIGMA. and .DELTA. signals, the quotient of
.DELTA./.SIGMA. is a measure of boresight error which is
independent of signal amplitude. However, any phase error
introduced into the received signals prior to forming the .SIGMA.
and .DELTA. signals will cause a boresight error.
In order to understand the effect of randome cross-polarization on
boresight error, a short discourse on the nature of polarized
signals is in order. In general, a polarized microwave ray passing
through a radome at an incident angle (i.e., an angle not
perpendicular to the surface of the radome where the ray strikes),
can be resolved into a polarization component parallel to the plane
of incidence (the plane between the incident ray and the normal to
the radome surface) and a polarization component perpendicular to
the plane of incidence. The radome passes these two components with
different degrees of attenuation and different microwave phase
delays with the result that the ray emerging from the inside of the
radome on the antenna has had its polarization altered. If the
receiving antenna was originally co-linearly polarized with the
impinging ray and the polarization of that ray was shifted by the
radome, a component of the cross-polarized energy will be accepted
by the antenna. Since the cross-polarized energy has a different
microwave phase than the co-polarized energy, the resulting vector
will be phase-shifted and will create a bore-sight error when
processed.
For purposes of explanation, vizualize two microwave rays of the
same polarization impinging on a radome at equal and opposite
angles of curvature (as they would when entering the two halves of
a monopulse antenna). It is also assumed that the polarization of
the two rays is neither parallel to, or orthogonal to, the plane of
incidence where they enter the radome. For radome transmission
purposes, the polarization of each of the rays may be resolved into
a component parallel to the plane of incidence and a component
orthogonal to the plane of incidence. The transfer function through
the radome for each resolved component then results in differing
signal vectors which, when combined, define the microwave signal
after passage through the radome. The apparent origin of such a
signal ordinarily differs from the actual origin of the microwave
signal being processed, i.e. there ordinarily is a boresight
error.
The two prime methods of compensating for radome induced boresight
errors are either: (a) to apply a correction factor to the
boresight error, or (b) to compensate each monopulse quadrant
signal with an inverse radome transfer function. To correct the
boresight error requires a knowledge of the signal polarization
which cannot be determined until the signal is processed. The
contemplated inverse transfer function method of compensation was
chosen because it can be applied to signals of any polarization
without a priori knowledge of that polarization.
The inverse transform method of radome calibration requires a
monopulse antenna in which each quadrant is sensitive to two
orthogonal polarizations. A microwave attenuation and phase
shifting network is connected to each of the orthogonal ports of
each quadrant. The transfer functions of these networks are
digitally set to the mathematical inverse of the radome/antenna
transfer function for the associated polarization and quadrant.
Since this transfer function changes with, inter alia, the incident
angle of the signal upon the radome, a multiplicity of commands
must be stored in order to cover all possible incident angles so
that the residual boresight error after compensation never exceeds
0.5 milliradians.
Referring briefly now to FIG. 5A, a flow path in mathematical form
for a particular quadrant is illustrated to show how compensation
is to be effected in the contemplated way. The radome/antenna
transfer function can be represented as a two-by-two matrix which
multiplies the orthogonally polarized signals passing through the
radome to reach the antenna quadrant. The method of compensation
illustrated in FIG. 5A then performs an inverse matrix to the one
representing the effect of the radome to produce two compensated
signals which are identical to the ones which entered the radome
for that particular antenna quadrant. It should be noted here that
in the radar system 10 (FIG. 1) complete polarization information
is not maintained after the compensation process is completed since
this would require ten receiver channels instead of five. A
selection is made of a particular or desired polarization (say for
purposes of rain rejection) by adding together the horizontal and
vertical components as illustrated in FIG. 5A.
Referring briefly now to both FIGS. 5 and 5A, the contemplated
radome calibration technique will be described. It will be
understood, of course, that the contemplated technique may be
carried out on any convenient type of radar test range where the
desired polarizations and frequencies of microwave energy may be
directed toward an assembly to be calibrated and the orientation
(relative to the source of the microwave energy) of the radome in
such assembly may be adjusted as required. Before proceeding,
however, it should be noted that the calibration procedure will be
described with reference to the receiver channel (not numbered)
dedicated to antenna quadrant 14A and that the calibration
procedure would be repeated for each one of the remaining channels
(not shown).
As mentioned hereinabove, the degree of radome cross-coupling is
dependent, inter alia, on the sense of the incident polarization,
the operating frequency and the antenna position. Consequently, a
vertically polarized signal is first transmitted toward the antenna
14 (FIG. 1) with the radome being calibrated in place so that the
response of the associated receiver may be measured and recorded.
Next, keeping the operating frequency and antenna position
constant, a horizontally polarized signal is transmitted and the
response of the associated receiver may again be measured and
recorded. The process is systematically repeated, changing the
orientation of the radome being calibrated and the frequency of the
microwave energy so that the response of the associated receiver to
any microwave energy under different operational conditions may be
measured and recorded.
It will be instructive at this point to recall that a polarized
microwave ray passing through a radome at an incident angle (i.e.,
an angle not orthogonal to the surface where the ray strikes) may
be resolved into a polarization component parallel to the plane of
incidence (the plane defined by the incident ray and the normal to
the radome surface) and a polarization component perpendicular to
the plane of incidence. The two components are passed through the
radome with different degrees of attenuation and different
microwave phase delays, with the result that the polarization and
amplitude of a ray emerging from the inside of the radome differs
from an incident ray.
Referring now to FIG. 5A, a radome compensation flow path in
mathematical form for a particular antenna quadrant is illustrated.
Thus, the effect of the radome may be mathematically represented as
a two-by-two matrix which multiplies orthogonally polarized signals
which pass through the radome to reach the antenna quadrant. The
compensation circuitry (not numbered) performs an inverse matrix to
the one representing the effect of the radome and, therefore, two
compensated signals are produced which correspond with the incident
orthogonally polarized signals.
With the foregoing in mind, it is here contemplated that each
radome and antenna assembly be calibrated on a range where the
orientation of such assembly to a linearly polarized test signal
may be measured and the operating frequency of such a signal may be
controlled to derive appropriate compensation signals for any
orientation of the radome and antenna assembly with respect to any
incident signal at any frequency within the operating bandwidth of
such assembly. Thus a radome and antenna assembly to be calibrated
is positioned in any convenient manner on a range to receive a
linearly polarized test signal, E, of the form (simplified)
where E.sub.i is the amplitude of the test signal and .theta. is
the polarization angle. It will be recognized that such a signal is
the equivalent of two orthogonally polarized signals, one (the
cosine component) being horizontally polarized and the other (the
sine component) being vertically polarized. Each such component,
upon passing through the radome and antenna assembly under test is,
in turn, subjected to differing degrees of attenuation and phase
delays. Thus, after the test signal passes through the radome and
antenna assembly the original horizontally polarized component may
be expressed as
where G.sub.XX and G.sub.XY are attenuation factors and D.sub.1 and
D.sub.2 are phase shifts.
Similarly, for the original vertically polarized component.
The attenuation factors and phase shifts describe the transfer
function of the radome and antenna assembly under the given
conditions.
The calibration procedure then is directed to the calculation of
the inverse radome and antenna transfer function for the given
conditions.
The determinant G [ ] associated with the radome compensation
network (not numbered) is of the form:
From the foregoing it should now be apparent that if G.sub.H =cos
.theta. and G.sub.V =sin .theta., the resultant signal from a
radome compensation and polarization selection network should be
A.sub.i.spsp.j(wt).
The contemplated calibration technique will now be described in
terms of the mathematical model of FIG. 5A. Thus, as mentioned
above, the calibration procedure for a given antenna look angle,
antenna roll angle, and operating frequency involves first
illuminating the radome with a horizontally polarized signal and
measuring the response, and next, keeping everything else constant,
illuminating the radome with a vertically polarized signal and
measuring the response to this input. The detailed procedure is as
follows: With a horizontally polarized input signal variable
attenuators 287V and 295V (FIG. 5) are set to provide maximum (80
dB) attenuation (in effect turning OFF the vertically polarized
channel (not numbered)) and the response as seen by the signal
processor 26 (FIG. 1) is recorded. This response corresponds to the
G.sub.XX .sup.j(D.sbsp.1) term in mathematical model of FIG. 5A.
Next, still with the horizontally polarized input signal, variable
attenuators 287H and 295H (FIG. 5) are set to provide maximum
attenuation (thereby turning OFF the horizontally polarized
channel) while variable attenuators 287V, 295V are set to provide
minimum attenuation and the response as seen by the signal
processor 26 (FIG. 1) is recorded. This response corresponds to the
G.sub.YX e.sup.j(D.sbsp.3) term. The foregoing procedure is
repeated with a vertically polarized input signal to yield the
G.sub.YY e.sup.j(D.sbsp.4) and G.sub.XY e.sup.j(D.sbsp.2) terms. On
the basis of these measurements, the determinant G [ ] and the
terms ##EQU1## are computed and stored in memory.
Digressing briefly here now for a moment, it should be noted that
the variables considered here to be important to radome
compensation are antenna look angle, antenna roll and operating
frequency. Given the variables selected, it may be estimated that a
solid angle of 0.03 steradians (10.degree..times.10.degree.) would
encompass a volume within which the boresight error would be
permissible and that a total of eight roll positions would cover
the .+-.75.degree. of roll required by a 4 g acceleration
specification. Twenty frequencies were selected to cover the
operating bandwidth of the system. Based on the foregoing, a total
of approximately 4.times.10.sup.6 bits of memory are required for
the requisite data for radome calibration. Further, it should also
be noted here that the transfer function through the radome (not
shown), the antenna 14 (FIG. 1) and the R.F. receiver 18 (FIG. 1)
may be represented solely by a network comprising a pair of
attenuators and a pair of phase shifters. Thus, the radome
compensation is provided through the use of a pair of variable
attenuators and a pair of phase shifters in each channel of the
R.F. receiver 18.
Proceeding now with the description of the calibration technique,
once all the constants are calculated and stored, the procedure is
to select a desired polarization (for a given operating frequency,
antenna look angle and antenna roll angle), calculate the requisite
transfer function, and convert the calculated transfer function
into equivalent attenuator and phase shifter values.
It is noted here that the tolerances of the attenuators 295V, 295H
and the phase shifters 297V, 297H are relaxed because the pilot
pulse is used to set them to the correct value.
It should be noted here that the contemplated radome calibration
technique also provides an advantage in terms of two way antenna
sidelobe control with a nonreciprocal antenna. That is to say, on
traversing the radome a transmitted beam will have impressed on it
a sidelobe pattern dependent on the attenuation and phase delay
through the radome. Without compensation, the corresponding
received beam would, upon traversing the radome, have an entirely
different sidelobe pattern impressed on it, depending upon the
motion of the aircraft, between the pulse transmission and
reception times. The contemplated radome compensation technique
would, however, remove the effect of the radome on the received
beam with a concomitant improvement in operation.
I. F. Receiver (FIG. 6)
Referring now to FIG. 6, a single channel of the I.F. receiver 20,
corresponding to that associated with antenna quadrant 14A, will be
described in detail. It should be appreciated that the remaining
channels (not shown) are identical to that to be described. The
S-band I.F. signals from the R.F. receiver 18 (FIG. 5) are passed,
via an isolator 311, through a bandpass filter 313 to a correlation
mixer 315, which also is fed by the second L.O. signal from the
exciter 32 (FIG. 2A) to down-convert the S-band I.F. signals to a
second I.F. frequency of 100 MHz. It should be noted here that the
correlation mixer 315 also performs the correlation or "stretch"
processing for the chirped waveforms. The second I.F. signal from
the correlation mixer 315 is amplified by an amplifier 317 and
split into two channels (not numbered), the upper channel being
used in the short range tracking mode and the lower channel being
used in the mapping and long range tracking modes. It should be
noted here that two channels are provided because of the different
dynamic range requirements which are imposed on the two tracking
modes.
The lower channel (sometimes hereinafter referred to as the mapping
and long range tracking channel) comprises an amplifier 319, a
limiter 321L, an amplifier 323L and a bandpass filter 325L, all of
which are of conventional design. The upper channel (sometimes
hereinafter referred to as the short range tracking channel)
includes a limiter 321U, an amplifier 323U and a bandpass filter
325U, all of which are also of conventional design. The bandpass
filters 325U, 325L are provided to remove any harmonics or spurious
signals generated within the limiters 321U, 321L and the amplifiers
323U, 323L.
The filtered output signals from the bandpass filters 325U, 325L
are split and applied to mixers 327UI, 327UQ, 327LI, 327LQ wherein
they are downconverted to baseband quadrature video signals by
being heterodyned with COHO L.O. signals obtained from the exciter
32 (FIG. 2A). It should be noted that the requisite quadrature
relationships are obtained by passing the COHO reference signals to
mixers 327UQ, 327LQ through 90.degree. phase shifters 329U, 329L,
respectively. The output signals from mixers 327UI, 327UQ, 327LI,
327LQ are filtered by low pass filters 331UI, 331UQ, 331LI, 331LQ
to remove undesired sidebands and are amplified by amplifiers
333UI, 333UQ, 333LI, 333LQ prior to being passed to switches 335I,
335Q. The latter, which are controlled by control signals provided
by the radar synchronizer 34 (FIG. 1), are effective to gate either
the long or short range tracking waveforms through a video receiver
(not numbered). The long range tracking waveforms are obtained from
the mapping and long range tracking channel wherein they are
subjected to an additional stage of amplification by virtue of the
amplifier 319.
The selected waveforms from the switches 335I, 335Q are sampled and
gain-scaled in sample/hold circuits 337I, 337Q, of conventional
design, by mode control signals from the radar synchronizer 34
(FIG. 1). The sampled output signals from the sample/hold circuits
337I, 337Q are digitized by ten bit A/Ds 339I, 339Q and are passed,
via buffers 341I, 341Q, to the signal processor 26 (FIG. 1).
The mapping signals in the long range track and mapping channel
(not numbered) are amplified in buffer amplifiers 343I, 343Q prior
to being applied to eight bit A/Ds 345I, 345Q. The latter have a
throughput rate of 50 MHz which is faster than the throughput rate
of the signal processor 26 (FIG. 1). In consequence, then, "first
in, first out" memories (FIFO 349I, 349Q) are provided to store a
series of real time samples from the A/Ds 345I, 345Q and provide
output signals to the signal processor 26 (FIG. 1) at a slower date
rate. To reduce the storage requirements of the FIFO memories 349I,
349Q, digital filters 347I, 347Q are provided between the A/Ds
345I, 345Q and the FIFO memories 349I, 349Q. The digital filters
347I, 347Q perform an integration (or summation) of several
contiguous samples which appear at the output of the A/Ds 345I,
345Q. The number of outputs to be summed before being dumped into
the FIFO memories 349I, 349Q is determined by a control signal
provided by the radar synchronizer 34 (FIG. 1).
Completing the I.F. receiver 20 are a pair of phase shifters 351U,
351L and a phase shifter control unit 353, which is shown to
receive input signals from the radar synchronizer 34 (FIG. 1) as
well as from the five vibration sensitive piezoelectric
accelerometers (not shown) on the rear face of the antenna 14 (FIG.
1). The phase shifter control unit 353, which here is simply a
double integrating filter, is effective to develop control signals
for the phase shifters 351U, 351L, depending upon the vibration
experienced at each antenna quadrant phase center at time of pulse
transmission as well as the frequency of the transmitted pulse. The
piezoelectric accelerometers (not shown) are provided to measure
the differential vibratory motion between the antenna phase
centers, corresponding to the centers of antenna quadrants 14A . .
. 14D (FIG. 3A), which, unless compensated for, would result in
phase tracking errors between the latter. Thus, the phase shifters
351U, 351L are provided to compensate for any differential
vibratory motion between the antenna quadrants 14A . . . 14D (FIG.
3A) by rotating the COHO signal before it is applied to the
quadrature downconversion mixers 327UI, 327UQ, 327LI, 327LQ.
It is anticipated that the vibratory motion of the antenna will
introduce approximately .+-.6 degrees of phase error in the
received signals and, therefore, the design of the phase shifters
351U, 351L becomes less critical. Thus the latter may, for example,
be fabricated by terminating the output ports of a quadrature
hybrid with varactor diodes. As the impedance of the diodes is
changed in response to an applied voltage the phase of the signal
reflected from the output ports will be correspondingly
altered.
Radome Compensation Electronics
As hereinbefore described, the compensation for radome error
requires the adjustment of two compensation elements (variable
attenuators 295V, 295H and phase shifters 297V, 297H being shown in
FIG. 5) in each channel of the R.F. receiver 18 (FIG. 1). This
means that a total of thirty-two compensation elements must be
digitally controlled. The bit length is a maximum of 10 bits for
any one element. Further, because each control word must be updated
whenever frequency or antenna pointing direction is changed and
because the calculation of each control word is performed in the
radar synchronizer 34 (FIG. 1), an interface between each
compensation element and the radar synchronizer 34 (FIG. 1) is
needed. The requisite interface is provided by an interface in the
radar synchronizer 34 (FIG. 1), which interface is similar to the
one shown in FIG. 9B. Because the construction of the interfaces
are so similar, only one will be illustrated.
To allow asynchronous operation, two memories are required, one
storing control signals from the radar synchronizer 34 and the
other providing control signals to a set of thirty-two buffer
registers. The memories perform the so-called "corner turning"
function to convert each control signal to a serial form for entry
in the proper buffer register. Each one of such registers converts
the applied control signal back to a parallel form and holds that
signal for the associated compensation element.
The requisite control signals are partially calculated in the radar
synchronizer 34 (FIG. 1). Common inputs for the calculation of all
of the control signals include frequency and antenna pointing
direction which together determine the address of a block of radome
compensation data stored on a dedicated disk. In addition, signals
indicative of temperature of the antenna 14 (FIG. 1), radar mode
and weather conditions are supplied to digital data processor 30
(FIG. 1) for completing the calculation of the radome compensation
values. Combinations of the addressed block in the dedicated disk
and the outputs of the digital data processor 30 (FIG. 1) then make
up the control signals.
A complete data set for the radome compensation process consists of
thirty-two control signals for each of the four frequencies
transmitted after each PRI. Assuming a 10 bit digital word for each
control signal, there are 1280 bits in a complete data set.
Although updating here of the control signals in only one quarter
(8) of the buffer registers need be done when frequency is changed,
it is preferred here to update all of such buffers when frequency
is changed. The use of two memories makes such complete updating
possible without interfering with calculation of new control
signals and eliminates the necessity for differing updating modes
when, for example, updating is required when the pointing direction
of the antenna is changed with respect to the radome.
Because compensation must be done in real time, double buffering at
each compensation element is dictated to allow each one of the
thirty-two control signals to be changed four times in each PRI
(with the maximum allowable time for change being one microsecond).
Double buffering at each compensation element lengthens the maximum
allowable time for change to twenty-five microseconds without
affecting performance. Just after a pulse is transmitted and before
the R.F. receiver 18 (FIG. 1) is enabled, each newly assembled set
of control signals then is clocked into a corresponding double set
of double buffer registers 298 (FIG. 5).
The provision of two memories and the sets of double buffer
registers 298 (FIG. 5) relieves the digital data processor 30 (FIG.
1) of very high overhead real-time refresh chores.
Within the radar synchronizer 34 (FIG. 1) the memory used for
storing the control signals is arranged so that one bucket of such
memory is loaded while the corresponding bucket of the other is
clocked out. The requisite bucket switching (which is conventional)
is performed when needed under control of the digital data
processor 30 (FIG. 1), at most every 10 ms (during a worst case
roll).
It is noted here that the bandwidths and center frequencies of the
I.F. receiver 20 are designed to be immune to delta jamming by
using the same technique as was described in connection with the
description of the R.F. receiver 18. That is to say, the bandwidth
of each section of the I.F. receiver 20 is designed to be one-half
the frequency of each succeeding downconverter so no false I.F.
signal may be engendered from the difference signal of a delta
jammer.
Motion Compensation (FIGS. 7A and 7B)
Before proceeding with a detailed description of the radar
synchronizer 34 (FIG. 1), the signal processor 26 (FIG. 1) and the
digital data processor 30 (FIG. 1), it will be instructive at this
point to discuss the motion compensation requirements of the radar
system 10 (FIG. 1).
As is known, any synthetic aperture radar used to map a terrain
with high resolution utilizes coherent processing of signals from
each individual ground reflector in the terrain being mapped for a
dwell (or coherent integration time) of sufficient length to
resolve such signals in a set of range gates and Doppler filters.
Such processing makes direct use of the motion of an airborne radar
during the dwell period to provide different Doppler frequencies
for ground reflectors at different angles from the velocity vector
of an aircraft. However, significant movement of the range gates
and Doppler filters during the dwell induce quadratic and cubic
phase components on the terrain and cause degradation of the
resolution of the map finally produced. Compensation for such
movements includes "slipping" range gates and the Doppler filter
bank with sufficient accuracy in a known manner to reduce the
unwanted motion during each dwell to acceptable bounds.
A further source of error in radar mapping is phase modulation,
induced by random or vibrational motions of the phase center of the
radar antenna, during each dwell. With such motions, if the
resulting phase modulation is sufficiently large, modulation
sidebands of signals associated with each particular ground
reflector may overlap with signals from nearby ground reflectors.
Compensation for such motions therefore consists of impressing a
time-varying phase shift on each signal being processed, such phase
shift being determined by a sensor monitoring the phase center
motion of the radar antenna along boresight and being opposite in
sense to the phase modulation introduced by the unwanted motion of
the radar antenna.
The Doppler frequency of any ground reflector will change during
each dwell as a result of the slightly changing angle to each
ground reflector caused by the motion of the radar antenna. If such
change is sufficiently large during each dwell so that the Doppler
resolution width is exceeded, then a spreading of the energy from
ground reflectors into adjacent Doppler filters results. To prevent
such spreading, compensation in the form of an appropriate
time-varying Doppler shift of each received signal is required. In
this case, however, inertial instrumentation is used to sense the
motion causing the unwanted change in Doppler frequency.
In addition to the foregoing, other perturbations of the antenna
cause phase modulation of the received signals. If these modulation
terms are in a small bandwidth, i.e., small with respect to the
Doppler resolution bandwidth (which is approximately equal to the
inverse of the time of dwell), then the effect is apparently the
same as a Doppler modulation caused by a slowly moving ground
reflector, with a maximum Doppler rate given by:
where a(t) is the acceleration of the antenna phase center in the
direction of the line-of-sight and .lambda. is the wavelength of
the transmitted signals. When modulation containing significant
energy at frequencies higher than the coherent resolution bandwidth
are caused, a phase error that varies during each dwell results. If
such error changes sinusoidally during a dwell, a spurious pair of
sidelobes in the Doppler filter output is created. The Doppler
offset of such sidelobes (relative to the mainlobe filter response)
is equal to the frequency of the perturbation causing the unwanted
phase modulation. The amplitude of each such sidelobe (relative to
the mainlobe filter response) is directly proportional to the peak
excursion of the phase center of the radar antenna along
boresight.
Conventional motion compensation techniques cancel the effects of
radial acceleration from the radar return signals. However, under
dynamic flight conditions as here contemplated, cancellation of
such effects is not of prime concern. Rather, the most important
object is to attain width and depth of focus of any map when
violent maneuvers are required in a tactical situation without
exceeding the computational time constraints. It can be shown that
the horizontal component of aircraft acceleration perpendicular to
the antenna boresight direction (usually denoted simply as the
cross-acceleration) has a devastating effect on the width of focus
when any conventional motion compensation technique is
followed.
In order to maintain the width, depth and "height" of focus of the
requisite synthetic aperture maps under severe aircraft
acceleration, the pulse repetition interval (PRI) of the herein
contemplated radar system 10 (FIG. 1) is varied to hold constant
the angular increment swept out by the horizontal component of the
line-of-sight between the antenna and the map reference point. The
width of focus then is a function of the horizontal component of
radial acceleration as opposed to the cross-acceleration. The depth
of focus, on the other hand, is a function of both vertical and
horizontal radial acceleration components. The depth of focus is,
however, not as strongly dependent on aircraft acceleration as is
the width of focus, and the depth of focus is relatively
independent of PRI variation.
"Height" of focus refers to the height (above or below) that a
given ground reflector may be (with respect to that of a map
reference point) without undue defocusing. The height of focus is
limited by the horizontal (radial) and vertical components of
aircraft acceleration and centripetal acceleration effects, with
the vertical component of vehicle acceleration being the most
important. Since the height of focus is proportional to the square
of the azimuth resolution, it imposes a severe restriction when a
map with very high resolution is to be made. The height of focus
cannot be increased in any direct manner by more processing
capability since it is jointly dependent upon vehicle acceleration
and terrain variation.
Referring now to FIGS. 7A to 7C, the effects of aircraft
accelerations on the problem of focusing a synthetic aperture map
will be explained. Thus, an aircraft 302 (FIG. 10) employing a
synthetic aperture radar (not shown) has its radar antenna (also
not shown) located at point a, while a synthetic aperture map
reference point is located at point o. A ground scatterer is
located at a point i with cartesian coordinates X.sub.i, Y.sub.i,
Z.sub.i with respect to the map reference point o. A reference
coordinate frame (the X.sub.o, Y.sub.o and Z.sub.o coordinate
frame) is a cartesian coordinate frame centered at the map
reference point o, with the Z axis being vertical and Y axis being
along the horizontal component of the position vector (not
numbered) from point a to point o at time t.sub.o. The X axis
completes the orthogonal triad, here in the right-handed sense. The
X.sub.1, Y.sub.1 Z.sub.1 coordinate frame is another local vertical
frame centered at point a, with the Z.sub.I axis being vertical and
the Y.sub.I axis disposed along the horizontal projection of the
velocity vector of the aircraft 302 (FIG. 10). A.sub.o is the angle
between the line-of-sight vector to the map reference point o and
the velocity vector. The angles B.sub.o and C.sub.o are,
respectively, the squint and elevation angles.
It will be noted that, as the aircraft 302 (FIG. 10) moves, the
boresight line of the antenna remains fixed on point o, which means
that the x.sub.o, y.sub.o coordinates of the point o rotate about
the Z.sub.o axis and that (for small changes of position of the
aircraft 302 (FIG. 10)) the X.sub.i, Y.sub.i coordinates of the
point i similarly rotate about the Z.sub.o axis.
It will be apparent that no cross-acceleration, i.e. acceleration
along the X axis, resulting from movement of the aircraft 302 (FIG.
10) is experienced at the point o. It follows then that any
compensation technique for nonlinear motion between the aircraft
302 (FIG. 10) and the point o need only be directed to compensation
for radial acceleration, i.e. acceleration along the Y axis. On the
other hand, because of the displacement of the point i from the
point o, motion of the aircraft 302 (FIG. 10) causes (in addition
to a radial movement along the Y axis), cross-acceleration along
the X axis and a concomitant nonlinear change in phase of the
returned signals from the point i. The magnitude and sense of the
cross-acceleration (and also of the radial acceleration) at the
point i is a function of its direction and distance, measured in
the X-Y plane, from the point o. As mentioned hereinbefore,
compensation for the displacement of the point i along the Z axis
may not be directly calculated; however, if vertical acceleration
of the aircraft 302 (FIG. 10) is reduced to a minimum during a
dwell, the compensation for cross-acceleration and radial
acceleration will also provide a modicum of compensation for "Z"
axis displacement. In this connection it will be recognized that an
inherent ambiguity exists between return signals from a point above
the plane of the selected range-Doppler matrix, i.e. the X-Y plane,
and return signals from a point below such plane. A moment's
thought will make it clear that, although the selected
range-Doppler matrix is defined in the X-Y plane by appropriate
pairs of isorange and isodop lines, such pairs of lines are also
lines which lie on curved surfaces. The isorange lines lie on
spherical sectors of "range" spheres which are centered at point a
and the isodop lines lie on the surfaces of halves of "Doppler"
cones having vertices along the velocity vector and intersecting
the terrain being mapped to define the isodops as hyperbolas.
Return signals from any point within the space between selected
spherical sectors and the "Doppler" cones have the same
"range-Doppler" signature and are therefore undistinguishable from
each other. Further, all such signals would have the same
"range-Doppler" signature as return signals from a point on the Y
coordinate. The result is that a three dimensional field mapped by
radar appears as a two dimensional image in which it appears that
there is no difference in elevation across such field. Fortunately,
however, there is little chance for confusion (when a terrain is
being mapped) as to whether a given target is "higher than" or
"lower than" the point o. On any terrain being mapped, all targets
of interest are ground targets approximately on the same level as
the center of the map, here point o. Further, because of the long
experience people have had with conventional photography, the
interpretation of images in two dimensions of features which are
actually three dimensional is easily accomplished.
It will now be appreciated that, if the differences between the
radial acceleration at the points o and i and the
cross-acceleration at the point i are determinable, then
compensation for motion may be effected for points on the map other
than point o. As a result, then, a radar map of higher quality may
be made. It will also be appreciated that the degree of
improvement, in a practical case, is the result of trade-off
between the size and quality of the desired map and the necessity
of providing a computer to calculate and to apply the requisite
compensation factors in real time. Obviously, it would be ideal to
calculate and apply a proper compensation factor to each
range-Doppler cell in order to increase the resolution of each cell
making up a radar map. It is equally obvious, however, that the
amount of data to be processed to obtain the large number of
correction factors required would make such an approach
impractical, considering the present state of the art of airborne
computers. It is, however, here contemplated, using presently known
computers, to calculate compensation factors to improve resolution
by at least an order of magnitude over the resolution usually
attained, making it possible to generate a radar map on which small
targets, such as tanks, may be distinguished.
With the foregoing in mind it is here contemplated to derive
compensation factors which improve the resolution of point i
without allowing the nonlinear change in phase of signals from
point o to exceed a given tolerable amount, say 45.degree.. Because
the phase of any received signal is a function of the range of the
reflector giving rise to such signal, the requisite compensation
factors, i.e. phase shifts, may be calculated by first determining
how the ranges to the points of interest (points o and i) change
during a dwell.
Referring now to FIG. 7A, the geometry of an exemplary situation
wherein an aircraft moves from point a to point b during a dwell
(time interval (t.sub.1 -t.sub.o)) while the boresight of an
antenna (not shown) is kept on point o to measure the depression
angle (which is the same as the elevation angle C when the air
craft is in level flight), the angle A and the angle B and the
range to point o. The aircraft here, in its inertial coordinate
system (X.sub.I, Y.sub.I, Z.sub.I), has a velocity V, and is
subject to acceleration which may be resolved into accelerations
A.sub.X(I), A.sub.Y(I), A.sub.Z(I) along the inertial coordinates
X.sub.I, Y.sub.I, Z.sub.I. In addition, it should be noted that
relative movements between the antenna 14 (FIG. 3A) and the
aircraft 302(FIG. 10) add to the accelerations. It will be apparent
from inspection of FIG. 7A that the range R.sub.i to any selected
points, here points o and i, on the X-Y plane may be determined and
that the amount and direction of rotation of the X-Y axes may also
be determined. To put it another way, using point o as the
reference point during a dwell, the position and orientation of the
isorange lines and isodops in the "range-Doppler" matrix relative
to the inertial coordinates X.sub.I, Y.sub.I may be determined. As
a result, the signals returned from different points on the terrain
being mapped may be appropriately modified to allow the finally
generated radar map to be presented in inertial coordinates with
greater resolution than would be possible if only point o were to
be compensated.
It will now be appreciated that the rationale of the method of
operating a pulse radar disclosed by Nathan Slawsby in U.S. Pat.
No. 4,084,158 (which patent is assigned to the same assignee as
this application) is applicable here. That is to say, because the
Doppler shift of the signals returned from each point on the
terrain being mapped changes in a nonlinear manner, i.e. exhibit a
Doppler acceleration as well as a Doppler velocity, compensation
for such nonlinear changes may be effected here by changing the
pulse repetition frequency of the radar so that the received
signals from each point on the terrain are properly phased with
respect to each other to minimize the effect of Doppler
acceleration.
Although the method described by Slawsby in the cited patent is
effective in compensating for movement of the radar to improve the
resolution of a radar map, it is not completely satisfactory by
itself in all situations. In particular, when a moving target (such
as a tank) is to be detected in the presence of "clutter", the
signal-to-clutter ratio obviously should be optimized. Here
"clutter" is made up of the features on the mapped terrain which
allow the radar map to be made, such features being stationary.
Optimization of the signal-to-clutter ratio may be effected here by
following an iterative procedure whereby: (a) the antenna 14 (FIG.
3A) is oriented so that, with the boresight line intersecting the
point o, the phase centers A', B' are in the plane defined by the
boresight line to the point o and the velocity vector of the
aircraft 302 (FIG. 10); (b) the pulse repetition interval between
successive pulses is adjusted (as described hereinbefore) to reduce
the effect of acceleration on the received signals from point o to
a minimum; (c) the difference between the signals received in the
quadrants 14A, 14B, integrated during a dwell, are compared to a
predetermined level; and the process is repeated (with different
pulse repetition intervals) until only the integrated returns from
moving targets are above the predetermined level. As indicated in
FIG. 7B, the optimum pulse repetition interval occurs when, in the
interval between successive pulses, the phase center B' moves into
coincidence with the line between the phase center A' and the point
o existing when a preceding pulse was transmitted.
It will be appreciated on inspection of FIG. 7B that the
just-described movement of the phase center B' (to the point marked
B") may be accomplished only if the antenna is rotated about both
the Z.sub.I and the Y.sub.I axes as the antenna is moved. It will
also be appreciated that, because the transmitted signals are
coherently modulated and the distance dR in FIG. 7B may be
measured, the phase shift of received signals due to the
propagation delay suffered by any signal moving through the
distance dR is calculable so that compensation of such phase shift
may be effected. To put it another way, the phase centers A' and B'
may (after movement of the latter to point B" and compensation) be
considered to be at the same range from any stationary object at or
near the point o. It follows then that subtraction of one of the
demodulated received signals from the other will effect
cancellation in the same manner as a conventional monopulse
receiver would be processing received signals in two opposite parts
of a monopulse antenna.
A moment's thought will now make it clear that compensation for the
distance dR to make it seem that the phase centers A', B" are
coincident (as far as return signals from a stationary object
adjacent the point o are concerned) may, or may not, have a similar
effect on return signals from a moving target. Thus, the distance
dR may be defined in terms of a number of wavelengths of the
transmitted signal; this in turn means that the phase shift of any
received signal moving through the distance dR may be expressed as
2.pi. n/.lambda., where "n" equals the distance dR expressed in
terms of the number of wavelengths of the transmitted signal. It
will be recognized that the expression just made is the same as the
expression for Doppler "blind" speeds (where "n" is any integer
greater than unity). Therefore, if return signals from a moving
target are being processed, the range rate of such target (which,
of course, determines the phase shift impressed on such return
signals over a period of time) controls the degree of cancellation
suffered when the signals from the phase centers A', B" are
processed. That is to say, whenever a moving target has a Doppler
"blind" speed, the return signals from such target will appear to
be the same as received signals from a stationary object, with the
result that processing cancels such return signal.
To reduce the number of Doppler "blind" speeds, it is necessary to
reduce the distance, in wavelengths, between the phase centers,
i.e., reduce the size of "n". Here, because the phase centers
A.varies. and B' are fixed in position relative to one another, it
is contemplated to process the return signals at phase centers
C.varies., D' to produce an equivalent phase center E', midway
between phase centers A', B'. Thus, the return signals at phase
centers C', D' are added and the resultant sum signal is divided in
half to produce the equivalent phase center E. When the equivalent
phase center E is coincident with the original line from A' to the
point o (again by controlling the pulse repetition intervals and
rotating the antenna) the intervals between Doppler "blind" speeds
are increased. It follows then that "low speed" targets may be
detected using received signals from phase centers A' and B' and
"high speed" targets may be tracked using received signals from
phase centers A' (or B') and E'.
It will now be noted that, although FIG. 7B and the discussion in
connection with FIG. 7B have been specifically directed to how the
signal/clutter ratio adjacent the point o would be enhanced, the
signal/clutter ratio in other areas could be enhanced simply by
having the phase center B" fall on the original line of sight to a
point other than point o. It will also be noted that considering
FIGS. 7A and 7B together, in a particular tactical situation with
known dimensions of the antenna, there are a sufficient number of
measured or known angles and distances to allow a complete solution
of the problem to be made.
Integrated Radar/Inertial System (FIG. 7C)
Referring now to FIG. 7C, a simplified block diagram of the radar
system 10 (FIG. 1) illustrating the interplay between the various
subassemblies for the purposes of motion compensation is
illustrated. Thus, the line labeled "PIEZO" out of the antenna 14
represents the output signals from the piezoelectric accelerometers
(not shown) which are passed, here via the synchronizer 34, to the
I.F. receiver 20 (FIG. 1) wherein they are used, in a manner to be
described in detail hereinbelow, to compensate for any vibratory
motion of the antenna phase centers (not shown). The output signal
from a radial accelerometer 171 (FIG. 8B) as well as the output
signals from the cross accelerometers (not shown) and the rate
gyros (also not shown) within the head mounted IMU 24 are passed in
analog form to the boresight sensor assembly 36 wherein they are
processed ultimately to form motion compensation signals for the
four quadrant phase centers (not shown). Thus, in order to
calculate the antenna axis components of the gravity vector a
processor, referred to as the strapdown processor 173, within the
boresight sensor assembly 36 computes a direction cosine
transformation which defines the angular orientation of the antenna
axes relative to the north, east and local vertical axes determined
by the inertial platform 38. The angular motion of the radar
antenna axes is sensed by the three orthogonal rate gyros (not
shown) within the head mounted IMU and their output signals are
used in the strapdown processor 173 for high frequency (200 Hz)
updating of the direction cosine matrix. Low frequency updating of
the direction cosine is, as will be described in detail
hereinbelow, provided via a transfer alignment Kalman filter (not
shown but within the inertial platform 38) which utilizes the
coordinate frame instrumented by the inertial platform 38 as a
reference for the strapdown direction cosine transformation
matrix.
The strapdown processor 173 rotates the three accelerometer input
signals, via the direction cosine matrix, into locally level
coordinates and integrates the equations of motion to obtain
velocity output signals. The latter, after conventional lever arm
corrections in the signal processor 26 to account for displacement
vector changes between the inertial platform 38 and the antenna 14,
are passed to the inertial platform 38. The latter compares, in a
transfer alignment Kalman filter (not shown), the velocity
estimates from the strapdown processor 173 with its own velocity
estimates. The velocity differences are the result of errors in the
direction cosine matrix within the strapdown processor 173 as well
as in accelerometer and gyro measurement errors. The Kalman filter
(not shown) is designed to estimate these errors which are
subsequently passed back, as corrections, to the strapdown
processor 173.
The output signals from the radial accelerometer 171 (FIG. 8B) are
accumulated in high speed motion compensation buffer registers (not
shown, but each having twenty-one bits) within the boresight sensor
assembly 36. The buffer registers are bumped at a rate
corresponding to the PRF to provide signals for the synchronizer
34, to be digitally integrated along with feedback signals from the
signal processor 26 to form incremented motion compensation (phase
correction multipliers) for the radar signal processing. The
horizontal cross-accelerometer data is also accumulated in the
boresight sensor assembly 36 and sent to the synchronizer 34 to be
used for PRI control. The component of vertical acceleration due to
antenna motion is also passed from the boresight sensor assembly 36
to the synchronizer 34 for use, in a manner to be described in
detail hereinbelow, for ultra high resolution A/D sampling rate
variation.
The signal processor 26 performs the following high speed motion
compensation functions: Phase rotation of the PRF samples from each
of the four antenna phase centers, phase rotation required for
clutter cancellation, and differential phase corrections for map
range sectors. The digital data processor 30, on the other hand,
performs low frequency centripetal motion compensation and supplies
tracking error commands to the boresight sensor assembly 36.
Gimbal Structure and Antenna IMU (FIG. 8A)
Referring now to FIGS. 8A and 8B, the antenna gimbal structure 200
is shown to comprise a four axis gimbal system. The yaw gimbal axis
201 and the pitch gimbal axis 203 are conventional, being required
to point the antenna boresight axis 205 in a given direction
relative to the aircraft centerline 207. In order to meet the
requirement that the plane (not shown) containing the antenna
azimuthal phase centers be maintained parallel to the slant plane
(the plane defined by the velocity vector and the line-of-sight to
the map reference point) a tilt (or elevation) gimbal axis 209 is
required. It should also be noted that the slant plane is also the
plane which cuts the isodop-isoran lines (not shown) at 90.degree.
and that, in consequence, the plane for the azimuth monopulse axis,
which is coincident with the plane (not shown) containing the
antenna azimuthal phase centers, is also coincident to the slant
plane. Finally, a roll bulkhead gimbal axis 211 is provided to
prevent gimbal lock and to maintain the requisite tracking accuracy
in the presence of high angular accelerations.
It is noted than an inertial measuring unit 24 (sometimes
hereinafter referred to simply as the head mounted IMU 24) is
provided on the gimbal structure 200. The herein contemplated
system utilizes both the head mounted IMU 24 and the inertial
platform 38 for the purposes of motion compensation and navigation.
If both functions were to be performed by the head mounted IMU 24,
a precision strap-down system with its attendant performance
degradations and computational difficulties would be required. On
the other hand, if the inertial platform 38 were to be used for
both the motion compensation and navigation functions, its location
within the aircraft 302 presents a problem. That is to say, for
navigation purposes it is desirable to locate the inertial platform
38 at, or within the proximity of, the center of gravity of the
aircraft 302. This location here is about eighteen feet from the
nose of the aircraft 302 where the antenna 14 is located. Since the
instantaneous inertial acceleration at the antenna phase centers is
not well correlated with the inertial acceleration occurring at the
remote inertial platform 38, the accelerometers (not shown) within
the inertial platform 38 are essentially useless for high frequency
motion compensation purposes. Thus, the head mounted IMU 24 is
utilized primarily for compensation of motion of the antenna 14,
while the inertial platform 38 is used for navigation and as a
vertical reference for gravity compensation of the IMU 24, although
both units mutually aid each other, via a coordinate transformation
process to be described in detail hereinbelow.
The uncompensated motion of the antenna 14 in a given radial
direction must be constrained to satisfy the requirements imposed
by the various mapping and tracking modes. The constituent sources
of error in any estimate of the radial acceleration fall into three
categories: (1) the magnitude of the velocity error; (2) the
uncertainty in the angular orientation of the head coordinates with
respect to the coordinates instrumented by the inertial platform 38
for the purpose of gravity compensation; and (3), the error in
acceleration measured by each accelerometer along its input axis.
For map focusing purposes, the angular uncertainty between local
vertical (as measured by the inertial platform 38) and the input
axis of the boresight or radial accelerometer 171 (FIG. 8B) must be
held within a specified tolerance.
As mentioned above, the herein contemplated motion compensation
system employs head mounted gravity compensated accelerometers (two
of which are not shown but are within the IMU 24). The requisite
gravity compensation is a two part process. First, the change with
altitude of the magnitude of gravity must be estimated, and
secondly, the component of gravity along the input axes of the
accelerometers must be calculated. The variation of the magnitude
of gravity relative to some ground reference is given by:
where g.sub.e is the magnitude of the gravity vector at the ground
reference point, H is the altitude of the aircraft 302, and R.sub.e
is the radius of the earth at the ground reference point. The
second part of compensation for gravity is somewhat more complex,
requiring a special version of the standard acceleration matching
technique for determining the direction cosine matrix between the
inertial platform 38 and the coordinates of the accelerometers. The
IMU 24 and the inertial platform 38 then are used in combination,
via the matrix of direction cosines defined by the coordinate axes
of each instrument, to resolve the components of gravity as
required.
Before proceeding, it will be instructive at this point to briefly
describe the antenna gimbal structure 200. Basically, the antenna
gimbal structure 200 allows the beam to be transmitted along the
requisite line-of-sight and isolates the antenna 14 from the
motions of the aircraft 302. The antenna gimbal structure 200
comprises a four-axis gimbal arranged to decouple and stabilize the
antenna 14 in the presence of aircraft motion (roll, pitch and
yaw). The contemplated configuration further allows all of the
cardinal axes to pass through a common point, thereby permitting
servo control about all the axes by means of the head mounted IMU
24. It should be noted here that, as mentioned above, a four-axis
gimbal assembly is provided to prevent a gimbal lock situation
which otherwise might occur with a conventional three-axis gimbal
system. Thus, for example, a conventional three-axis gimbal system
would experience yaw gimbal lock if the antenna 14 were looking
forward and down 45.degree. and the aircraft 302 were to pitch
upward by 45.degree.. Furthermore, even in the absence of gimbal
lock, servo performance degrades as the pointing angle approaches
within 30.degree. of gimbal lock, thereby compromising the
requisite pointing accuracy.
Given that a four-axis gimbal structure 200 is required to meet the
target tracking requirements, a gimbal order of outer roll, pitch,
azimuth and elevation was selected as the optimum configuration
from the point of view of, inter alia, stiffness, damping of
resonances, ease of implementing the waveguide connection to the
antenna 14 and gimbal travel requirements. In such a configuration,
the roll axis 211 and the pitch axis 203 form a stable platform for
a conventional azimuth/elevation mount. The yaw axis 201 and the
elevation axis 209 then form the part of gimbal structure 200
required in the target tracking modes.
Referring now also to FIG. 8B, the roll gimbal 213 and the pitch
gimbal 215 are stabilized by means of rate gyros 217, 219,
respectively, which are mounted on their respective axes. The
antenna 14 is rate stabilized about the yaw (azimuth) gimbal axis
201 and the elevation gimbal axis 209 by means of an azimuth rate
gyro 221 and an elevation rate gyro 222. The roll gimbal 213 and
the pitch gimbal 215 effectively isolate the azimuth gimbal axis
201 and the elevation gimbal axis 209 from the dynamic environment
of the aircraft 302.
The head mounted IMU 24, which, as explained, is provided for
compensation of motion of the antenna 14 relative to the aircraft
302, comprises three mutually orthogonal rate gyros, the radial
accelerometer 171 and a pair of cross accelerometers, such elements
(with the exception of the radial accelerometer 171) being mounted
on the elevation gimbal 223 to serve also as a counterweight to the
antenna 14.
The roll gimbal 213, the pitch gimbal 215, the elevation gimbal 223
and the yaw gimbal 224 are driven by conventional torque motors of
which only the roll torquers 225 and the pitch torquers 226 are
shown. Such motors here are direct current motors with direct drive
to avoid the backlash associated with the use of gear drives.
The roll and pitch axes 211, 203, respectively, are rate controlled
by operation of a roll rate gyro 217 and the pitch rate gyro 219 in
conjunction with rate control commands received from the digital
data processor 30 (FIG. 1). The inner gimbal axes, the yaw gimbal
axis 201 and the elevation gimbal axis 209 are all controlled by
means of coordinate transformation resolvers (not shown) mounted on
their respective gimbals. The rate integrating gyros (not shown)
within the IMU 24 are used to sense rotation about the yaw gimbal
axis 201 and the elevation gimbal axis 209. The output signals from
the former are fed back to both the coordinate transformation
resolvers (not shown) and the digital data processor 30 (FIG. 1).
It should now be appreciated by those of skill in the art that the
coordinate transformation resolvers (not shown) are utilized to
control the jaw and elevation gimbal axes 201, 209, respectively,
to effect a higher control rate and thereby prevent undesired
resonances. That is to say, the outer roll and pitch gimbal axes
211, 203, respectively, are controlled by feeding data directly
back from their respective rate gyros 217, 219 to the digital data
processor 30 (FIG. 1) at a relatively low data rate, here 64 Hz.
Such a data rate is, however, too low to provide the requisite
degree of rate control of the jaw and elevation gimbal axes 201,
209, respectively, and, therefore, the coordinate transformation
resolvers (not shown) are provided to effect a higher data rate
while maintaining an identical interface with the digital data
processor 30 (FIG. 1).
It should be noted here that the antenna gimbal structure 200 is
unique and provides the following advantages vis-a-vis conventional
two or three axis gimbal structures: (a) the four axis gimbal
provides the greatest angle coverage for the antenna 14; (b) it
reduces the dynamic environment seen by the yaw axis 201 and the
elevation axis 209, thereby achieving the requisite stabilization;
(c) it permits placing the jaw axis 201 perpendicular to the slant
plane; and (d) the four axis gimbal structure 200 allows target
tracking without rotation of the antenna 14, thereby providing a
signal processing advantage by easing clutter cancellation
requirements. Furthermore, the inverted roll gimbal 211 design
permits the placement of the head mounted IMU 24 and a
counterweight (not numbered) directly in line behind the antenna
14, making the horizontal and vertical balance about the pitch
gimbal axis 203 a symmetrical balance operation. This approach
serves to produce the lightest possible system weight for a given
antenna size.
Transfer Alignment (FIG. 8B)
As was mentioned above, the herein contemplated coordinate
transformation technique involves resolving the direction cosine
matrix between the instrumented axes of the inertial platform 38
and the head mounted IMU 24. To this end the strapdown processor
173 (FIG. 7C) (sometimes hereinafter referred to simply as
processor 173 and which is here a Texas Instruments Model 9900
microprocessor) is shown to receive the output signals from the
head mounted IMU 24 and from a line-of-sight sensor assembly (not
numbered) mounted at the rear center of the antenna 14. The
line-of-sight sensor assembly (not numbered) comprises the radial
accelerometer 171 and the elevation rate gyro 222, both of which
are aligned with the antenna boresight axis 205. The processor 173
is physically located within a boresight sensor assembly 36 (FIG.
1). It should be noted here in passing that the output signals from
the IMU 24 comprise the output signals from three mutually
orthogonal rate gyros (not shown), a pair of cross accelerometers
(also not shown) and the line-of-sight sensor assembly (not
numbered). It should also be noted that the process of transfer
alignment in inertial navigation systems is well known to those of
skill in the art (e.g. "The Kalman Filter in Transfer Alignment of
Airborne Inertial Guidance Systems" by A. Sutherland and A. Geib,
TASC Report TR-134-2, Jan. 15, 1968; "Strapdown Inertial System
Alignment Using Statistical Filters: A Simplified Formulation"0 by
J. Deyst and A. Sutherland, AIAA Journal, Apr. 1973, pp. 452-6; and
"A New Concept in Strapdown Inertial Navigation" by J. E. Bortz,
NASA Technical Report R-329, Mar. 1970) and will therefore not be
recounted in detail here. Suffice it to say that within the
processor 173 the signals from the boresight or radial
accelerometer 171 and the accelerometers (not shown) within the IMU
are rotated, via a direction cosine matrix, into local vertical
coordinates and integrated to obtain horizontal velocity signals.
High frequency changes in he direction cosine matrix are provided
within the processor 173 by means of the input signals from the
three orthogonal rate gyros which sense the angular motion of the
antenna axes. The horizontal velocity signals are corrected for
errors due to the displacement vector changes between the antenna
14 and the inertial platform 38 by being modified in a so-called
"lever arm correction process" by the output signals from three
rate gyros (not shown) within the boresight sensor assembly 36
which is disposed on the antenna bulkhead (not shown). The actual
lever arm corrections are performed within the digital data
processor 30 (FIG. 1). The thus corrected velocity signals from the
digital data processor 30 are passed to a microcomputer (not shown)
associated with the inertial platform 38 wherein they are compared,
in a Kalman filter process, with the horizontal velocity outputs
obtained from the inertial platform 38. The difference in the
velocities as determined by the inertial platform 38 and the
processor 173 are due to vertical and azimuth errors in the
direction cosine matrix as well as errors in the accelerometer and
gyro input signals to the processor 173. The Kalman filter process
estimates the errors of the processor 173 and supplies these
estimates to the latter as correction signals.
Antenna Phase Center Motion Compensation (FIG. 8B)
Completing the contemplated motion compensation system are a
plurality of piezoelectric accelerometers (not shown, but which are
here Model 606 devices from Columbia Research Laboratory, Woodlin,
Pa.) which are provided to compensate for the effects of
aircraft-induced vibrations on the antenna 14. Thus, although the
signals out of the line-of-sight sensor assembly (not numbered) may
be used to compensate for the effects of translational vibration,
such signals are not useful in compensation for rotational or
differential (between quadrants) vibrations. Compensation for the
effects of differential antenna vibration is especially important
in the ground moving target identification mode wherein
vibration-induced sidebands can appear as moving targets and thus
seriously affect the reliability of the moving target detection
process. In such a mode the differential vibration of the antenna
14 will phase modulate the radar return signals and cause sidebands
(or false targets) to appear, thereby resulting in ambiguous
Doppler angle measurements. Further, in the target tracking mode
specular sidebands can appear in the Doppler cell of a target being
tracked, thereby producing significant angle tracking errors. In
addition, in the target tracking mode the degree of clutter
cancellation between the antenna phase centers is directly affected
by differential vibratory motion which places an upper limit on the
cancellation obtainable, thereby also degrading angle tracking
performance.
As was mentioned above, the phase centers of the antenna 14 are
displaced with respect to each other and can rotate about axes in
the plane of the antenna 14 passing through its geometric center.
If the vibration-induced phase center displacements are not
symmetric for corresponding phase centers, the sum (.SIGMA.) and
and difference (.DELTA.) signals derived will contain uncompensated
vibratory phase modulation even though a radial accelerometer has
been provided to compensate for the pure translatory motion.
Piezoelectric accelerometers (not shown) are used to measure
aircraft-induced vibrations of the antenna 14. A total of five such
devices are provided, one at each of the four antenna phase centers
and a fifth located at the center of the antenna 14. The radial
accelerometer 171 is disposed adjacent to the piezoelectric
accelerometer (not shown) located at the center of the antenna 14.
It should be noted here that the radial accelerometer 171 serves as
a reference for the piezoelectric accelerometers (not shown) which
differ from conventional accelerometers in that their frequency
range is from 2 Hz to 40 KHz.
It should now be appreciated that since the piezoelectric
accelerometers (not shown) are high frequency devices, they do not
measure the low frequency (linear or rotational) components of the
inertial accelerations at the antenna phase centers and the
geometric center of the antenna 14. It can therefore be shown that,
assuming a linear chirp waveform is transmitted, the compensation
signal for a given subaperture phase center would be of the form
##EQU2## where y.sub.i is the differential displacement estimate,
which is given by ##EQU3##
The P.sub.im term in equation (4) corresponds to the piezoelectric
accelerometer signal from each phase center and the P.sub.am term
is the signal from the piezoelectric accelerometer disposed at the
center of the antenna 14. Thus, the signals from the piezoelectric
accelerometers (not shown) disposed at the antenna phase centers
are differenced with that from the piezoelectric accelerometer
disposed at the geometric center of the antenna 14, and the
resultant signal is integrated, in analog fashion, over the
coherent integration time, T. It should be noted here in passing
that the requisite integration is performed in the I.F. receiver 20
(FIG. 6) wherein compensation signals are developed to control
phase shifters in respective channels of such receiver. It should
also be noted here that the actual displacements given by equation
(4) are formulated each PRI. These displacement values are
subsequently modified by four different constants, corresponding to
wavelength values for each of the four frequencies utilized during
a given dwell, to develop the control signals for the phase
shifters 351L, 351U (FIG. 6).
RADAR SYNCHRONIZER (FIGS. 9A-9F)
Before proceeding with a detailed description of the radar
synchronizer 34 (FIG. 1), a brief discussion of its function and
design requirements is in order. Thus, referring briefly back now
to FIG. 1, the primary function of the synchronizer 34 is real time
control of the radar system 10. As such, data and control commands
from the digital data processor 30 (FIG. 1) in the form of digital
words rae interpreted by the synchronizer 34 and translated into
timing and control signals for operating such units as the I.F.
receiver 20 (FIG. 1) and the exciter 32 (FIG. 1). Among the key
timing functions provided by the synchronizer 34 are waveform gate
generation for the exciter 32, analog-to-digital (A/D) sampling
strobes for the I.F receiver 20 and mode control switching for the
power distribution network 16.
Another function of the synchronizer 34 is high speed calculation.
That is to say, the radar system timing depends upon controlling
the pulse repetition interval of the radar system 10 in order to
maintain equal angular increments between the pulse transmission
paths to the map reference point as the aircraft 302 (FIG. 10)
flies along during a mapping action thereby to provide the optimum
focal width for the generated maps, as will be explained in greater
detail hereinbelow. Suffice it to say here that the forementioned
angle equality must be maintained in the presence of aircraft
acceleration of up to 4 g's in magnitude in any direction and
requires that the instantaneous range to the map reference point or
target be known. Range and range rate are supplied to the
synchronizer 34 by the signal processor 26 (FIG. 1) every 7.7 m
sec. However, a range integration at a rate much higher than the
digital data processor update rate is required for smooth
interpolation within the synchronizer 34. The PRI variation
computations also require cross-acceleration data which is provided
by the strapdown processor 173 (FIG. 7C) from cross-acceleration
measurements made at the antenna 14, as was explained
hereinabove.
Within the synchronizer 34 the time spent performing calculations
as well as the basic timing resolution and accuracy are determined
by the requisite map resolution and tracking accuracy. Thus, for
example, in the short pulse tracking mode the quantization noise is
specified to be consistent with a 66 dB receiver dynamic range. In
consequence, then, the A/D strobes sent by the synchronizer 34 to
the signal processor 26 (FIG. 1) must be positioned consistently,
with no more than thirteen picoseconds (ps) root means square (rms)
jitter with respect to the transmitted waveform, to achieve this
dynamic range.
The actual timing of the A/D strobes for the signal processor 26 is
dependent on the range to the target or map reference point. The
foregoing jitter constraint imposes a parallel requirement on how
often the range word must be updated, for the reason that smooth
transitions should occur in the range word to achieve stable
mapping and tracking. The requisite range granularity considered
together with the maximum specified aircraft velocities here define
an update rate of at least 40 KHz for the range word.
As mentioned hereinabove, the digital data processor 30 (FIG. 1)
calculates and delivers to the synchronizer 34 the range, R.sub.o,
and range rate, R.sub.o, every 7.7 msec. The necessity for
instantaneous range every 25 .mu.s means that a range calculation
must be made within the synchronizer 34 at a 40 KHz rate. The range
calculation is defined by the following equation: ##EQU4## where
R.sub.EST is the instantaneous estimated range;
R.sub.o is the range calculated in the digital data processor 30
(FIG. 1);
R.sub.o is the range rate calculated in the digital data processor
30 (FIG. 1);
R.sub.o is the radial acceleration calculated in the strapdown
processor 173 (FIG. 7C); and
.DELTA.t equals 25 microseconds.
The range calculation is reset and restarted in synchronism with
the radar dwell rate timing as soon after a signal processor update
as possible.
The other major computational requirement is PRI variation. The PRI
control processing within the synchronizer 34 will be described in
detail hereinbelow. Suffice it to say here that the basic or
minimum PRI is set by the digital data processor 30 (FIG. 1) for
each dwell or group of dwells by sending a distance increment,
.DELTA.d, to the synchronizer 34 at a 128 Hz rate. Every 25 .mu.s
the strapdown processor 173 (FIG. 7C) sends to the synchronizer 34
a velocity word based on accelerometer measurements. The
synchronizer must predict the next velocity using a GH fading
memory predictive filter and interpolate "instantaneous" velocities
in real time. The rapidly updated (100 KHz) velocity is integrated
by the synchronizer 34 at a 100 KHz rate to develop a travel word,
X.sub..perp.M. During a dwell, an accumulating sum of .DELTA.d's
(i.e. the desired amount of travel, X.sub..perp.D) is compared to
X.sub..perp.M. When X.sub..perp.M equals X.sub..perp.D, a PRI
timing sequence is initiated. At the same time, another .DELTA.d is
added to the previously accumulated X.sub..perp.D. The process
repeats when X.sub..perp.M reaches the new X.sub..perp.D.
The expression for X.sub..perp.M is given by: ##EQU5## It should
now be appreciated that the terms on the right hand side of Eq. (6)
are the "MOTION COMPENSATION" signals of FIG. 7C. The synchronizer
34 then is operative to compare X.sub..perp.M and X.sub..perp.D
every 10 microseconds and to command a pulse to be transmitted
whenever equality occurs. Sufficient accuracy in PRI variation is
achieved if X.sub..perp.M is calculated at a 100 KHz rate. After
each calculation X.sub..perp.M is compared to X.sub..perp.D to
determine if the quantities are equal. A time granularity of 10
.mu.s is adequate for satisfaction of the clutter nulling
requirement.
Digressing briefly here now for a moment and referring to FIGS. 12A
and 12B, the effect of PRI control is illustrated. Thus, during a
given mapping interval starting at time t.sub.o and proceeding to
time t.sub.n, the flight path of the aircraft 302 (FIG. 7A) is
shown to gradually approach the map center point, O. It is apparent
that between pulse transmission times t.sub.o and t.sub.l the
aircraft 302 (FIG. 10) must traverse a greater distance than that
required between pulse transmission times t.sub.n-1 and t.sub.n. It
should also be apparent that the angular increment, .delta..theta.,
swept out between pulse transmission times t.sub.o and t.sub.l, is
identical to that swept out between times t.sub.n-1 and t.sub.n.
Thus, it can be seen that the effect of PRI control is to insure
that, despite the motion of the aircraft 302 (FIG. 10) equal angle
increments are swept out between the pulse transmission times. For
the flight path illustrated the PRI will decrease exponentially
during the dwell period.
As mentioned above, a PRI variation is needed to attain a
satisfactory width of a generated map when the aircraft 302 is
maneuvered during a dwell. A greater depth-of-focus of a generated
map is realized by varying the A/D strobe frequency slightly during
each dwell as a function of change in the depression angle, C.sub.o
(FIG. 7A). As the A/D strobe frequency need only be changed at
relatively infrequent intervals, the 128 Hz update rate of the
synchronizer 34 by the digital data processor 30 is adequate for
satisfactory depth-of-focus and there is therefore no requirement
for real-time calculation in the synchronizer 34 to perform this
function.
Referring briefly now to both FIGS. 1 and 9A, the radar
synchronizer 34 is shown to include a data input/output (I/O)
controller 371, a microcomputer 375, a timing and control logic
unit 377, a clock generator 379, and a 100 MHz coherent oscillator
(COHO) 381. The design of the radar synchronizer 34 is driven by
the requirement to communicate with several subsystems other than
the digital data processor 30. Thus, for example, the digital data
processor 30 sends, either at a 128 Hz rate or once per radar
dwell, approximately 256 digital words (in 16 bit serial form) to
the radar synchronizer 34 and the strapdown processor 173 (FIG. 7C)
delivers a digital word (18 bits) every 25 .mu.s. Further, before
every dwell the radar synchronizer 34 must reprogram the exciter 32
(FIG. 1) with a 96-bit serial word, and the transmitter 22 requires
an 8-bit word for power level control. Still further, before every
PRI the signal processor 26 requires a PRI normalization factor
which is the length of the previous PRI. The data links which are
provided by the data I/O controller 371 are, therefore, data
outputs to the exciter 32, the transmitter 22, the R.F. receiver
18, the I.F. receiver 20 and the power distribution network 16 and
a data input from the strapdown processor 173 (FIG. 7C) and a
bi-directional data link to the digital data processor 30 and the
signal processor 26.
Referring now to FIG. 9B, the data I/O controller 371 is shown to
include a data I/O module 373, a microprocessor 383 and a shared
memory module 385. The data I/O module 373, which here may be
comprised of a plurality of shift registers (not shown), provides
the interface between the radar synchronizer 34 and the other major
systems of the radar system 10 (FIG. 1). An exemplary data
interface, here that with the digital data processor 30 (FIG. 1),
is shown to include both data input and output lines. The data
output lines include a request output (REQO) line, a data output
(DATO) line, a clock output (CLKO) line, and an acknowledge (ACKO)
line. The data input lines include a request input (REQI) line, a
data input (DATI) line, a clock input (CLKI) line and an
acknowledge (ACKI) line. A data output transfer from the data I/O
module 373 is initiated by changing the REQO line from a logic 0
level to a logic 1 level. If the digital data processor 30 is ready
to accept data the ACKI line is switched from a logic 0 to a logic
1 level. The data I/O controller 371 then gates ON (changes to a
logic 1 level) the CLKO line and outputs, synchronously with the
clock signals, a stream of data over the DATO line. The falling
edge of the first clock pulse causes the digital data processor 30
to reset its ACKI output signal back to a logic 0 level. Data
transfer form the digital data processor 30 to the data I/O module
373 is similarly accomplished with the digital data processor 30
initiating the REQ signals for data input and output transfers.
Within the radar synchronizer 34 all the high speed time-critical
computations are performed within the microcomputer 375. The data
I/O controller 371 acts as a buffer between the microcomputer 375
and the various subsystems of the radar system 10 (FIG. 1) to
prevent interrupts of such microcomputer when data transfers are
required. Such buffering is necessary because the microcomputer 375
may have to perform several concurrent calculations at both a 40
KHz and 100 KHz rate, with consistency in the timing of such
calculations determining the accuracy of both the PRI variation and
the A/D strobe frequency for the IF receiver 20 (FIG 1). It
follows, then, that interrupts to the microcomputer 375 are
inconsistent with the task of timing the PRI. Backing up incoming
data from the digital data processor 30 (FIG. 1) in first-in,
first-out memory and sampling it on a polled basis is feasible only
for small amounts of data. The only architecture general enough to
handle the requisite data transfers is one that features an
intelligent data controller. Thus, the data I/O controller 371,
although it is a peripheral of the microcomputer 375, contains
local intelligence provided by the microprocessor 383, which is
here a 2900 Series microprocessor from Advanced Micro Devices, Inc.
Sunnyvale, Calif., 94086. The microprocessor 383 peels off
synchronizer control words from the data stream provided by the
digital data processor 30 (FIG. 1) and places them, together with
an updated status word, in a small scratch-pad memory compartment
of the shared memory module 385. The microcomputer 375 thus has a
small group of data words to process and is protected from the
necessity of directly servicing the radar subsystems.
The data I/O controller 371 is able to respond to data transfer
request on an interrupt basis without affecting radar timing since
it processes data in parallel. At initialization of the radar
system 10 (FIG. 1), the data I/O controller 371 takes priority over
the microcomputer 375 by halting the latter and performing direct
memory address (DMA) data transfers into the shared memory module
385 and the control memory module 387 within the microcomputer
375.
The shared memory module 385 comprises a direct memory address
(DMA) logic section (not shown) and a dual port random access
memory (RAM) (also not shown) which is shared by the microcomputer
375 and the microprocessor 383. The dual port RAM serves as a
macroprogram memory for the microcomputer 375 and, in addition,
stores: (a) mode data for the timing and control unit 377 (FIG.
9A); and (b) numerical data for the microcomputer 375 and the
timing and control logic unit 377 (FIG. 9A). It should be noted
here that all the data transfers within the radar synchronizer 34
are accomplished by data busses (not numbered) comprising exposed
registers (exposed register module 395), which will be described in
detail hereinbelow.
It should be recalled here that all of the high speed,
time-critical algorithms (as, for example, the PRI variation
computations) are performed within the microcomputer 375. The
update rate and the precision required for these calculations as
well as the desire to avoid utilizing double precision arithmetic
lead to the choice of a 24-bit bipolar bit-slice implementation for
the microcomputer 375. The bit-slice family chosen for implementing
the latter is the 2900 series available from Advanced Micro Devices
(AMD). Thus, the sequencer 389 is here a 2910 series microprogram
controller and the arithmetic logic unit (ALU) module 391 is
comprised of six 4-bit 2903 microprocessor slices to achieve the
requisite 24-bit machine width. It should be noted here that each
of such slices includes sixteen randomly accessible dual-port
general purpose internal registers with the necessary "hooks" to
expand the register file by combining external (off-chip) and
internal registers. The multiplier module 393 is here comprised of
a plurality of MPY24HJM multipliers from the LSI Products Div. of
TRW, Redondo Beach, Calif. and the memory module 397 is here a
random access memory because of the versatility of such a
device.
Completing the microcomputer 375 is an exposed register module 395
which handles all the data transfers form the microcomputer 375 to
the data I/O controller 371 and the timing and control logic unit
377. Such unit is implemented using emitter-coupled logic (ECL)
and, therefore, a TTL/ECL interface exists between the
microcomputer 375 and the timing and control logic unit 377 (FIG.
9A). Because the length of each PRI must be controlled to within 10
.mu.s, a very high premium is put on the rapid execution of the PRI
algorithm within the microcomputer 375. Part of that execution time
is expended in outputting the PRI start command to the timing and
control logic unit 377 (FIG. 9A). Performing input/output I/O
instructions with any conventional parallel output technique is too
wasteful of the very limited execution time available to the
microcomputer 375 within a PRI. The exposed register module 395
solves the data transfers problem by taking advantage of the
register expansion capability within the ALU module 391. Thus, it
should be recalled that each 2903 chip in the ALU module 391
contains sixteen general purpose internal registers as well as
"hooks" permitting the connection of additional off-chip registers.
A plurality of such off-chip registers (not shown) which are
assigned dedicated functions are provided in the exposed register
module 395. These off-chip registers are controlled in the same
fashion as the on-chip registers, and TTL to ECL level shifters
(also not shown) present the contents of these off-chip registers
directly and in parallel to the ECL timing logic within the timing
and control logic unit 377 (FIG. 9A). Thus, programming the ECL
logic from the exposed register module 395 requires no processing
time overhead specifically for output operations. That is to say,
as soon as a control word is calculated in the microcomputer 375,
control of the ECL logic within the timing and control unit 377
(FIG. 9A) takes effect because the result of each calculation
always resides in the dedicated exposed register (not shown) within
the exposed register module 395. Eight control words, some of which
change four to six times per PRI, are handled in this fashion
thereby saving significant amounts of processing time during each
PRI.
Before proceeding with a detailed description of the timing and
control logic unit 377 it should be noted here that the timing of
the radar system 10 (FIG. 1) is performed in two different sections
of the synchronizer 34. Thus, PRI variation and scheduling is
performed within the microcomputer 375, while the timing within a
PRI is determined by ECL logic within the timing and control logic
unit 377 which is controlled by, but is much faster than, the
microcomputer 375. It should also be noted that the logic circuitry
within the timing and control logic unit 377 is divided into two
halves corresponding, respectively, to transmit and receive mode
timing. The transmit and receive mode circuitry are similar,
differing only in that the receive mode circuitry does not include
either a phase lock loop module or a range counter module because
the receive mode clock is here generated in the phase lock loop
module 401 (FIG 9E). Thus, for the sake of drawing clarity, only
that portion of the circuitry corresponding to the transmit mode
timing will be shown and described in detail.
Referring now to FIGS. 9D to 9F, the transmit mode timing logic
within the timing and control logic unit 377 is shown to include a
"mode transparent, real time" control module 399, a phase lock loop
module 401, and a range counter module 403, all of which receive
control signals from the microcomputer 375 via exposed registers
(not shown) within either the exposed register module 395 (FIG. 9C)
or the multiplier module 393 (FIG. 9C). The basic clock for the
transmit mode timing is the 50 MHz master digital clock (MDC) which
is obtained from the clock generator 379 (FIG. 9A), and which is
developed within the latter by dividing down the 100 MHz signal
received from the 100 MHz COHO 381 (FIG. 9A).
The mode transparent real time control module 399 is required to
generate, within a given PRI, control signals for the various radar
subsystems as, for example, spoof frequency control signals for the
exciter 32 (FIG. 1), power level control signals for the
transmitter 22 (FIG. 1), and receiver gain levels for the R.F.
receiver 18 (FIG. 1). Prior to the start of a given radar dwell
(which could total 1,000 PRI's) the microcomputer 375 (FIG. 9C)
clears the memory address counter 407 and then loads that counter
(and the random access memory (RAM) 405) with real-time control
states interleaved with state-change times. At the start of each
PRI, the microcomputer 375 (FIG. 9C) resets a PRI timer 409, which
is here a sixteen bit counter, with a word corresponding to the
length of the desired PRI, which could range from 100 .mu.s to 1.3
ms. The memory address counter 407 and the PRI timer 409 are then
synchronously clocked by the 50 MHz MDC. The memory address counter
407 addresses the first event time-stored in the RAM 405 (expressed
as a sixteen bit binary number), which event is then clocked into a
next event time register 411. A comparator 413 detects when the PRI
timer 409 reaches such number stored in the next event time
register 411. When such detection occurs, the comparator 413
provides a control signal to an event control mode logic unit 415,
which is here of conventional design, and which, in turn, causes
the memory address counter 407 to be bumped twice, once to clock a
control word from the RAM 405 to an output register 417 and a
second time to clock a new event time word into the next event time
register 411. It should be noted here that as only changes of state
(pulse edge scheduled times) are stored in the RAM 405, each data
line leaving the output register 417 is a discrete control
line.
Before proceeding with a detailed description of the phase lock
loop module 401, the rationale for providing such a unit will be
described. Thus, in order for the radar system 10 (FIG. 1) to
reliably track a target with the precision desired here, a time
resolution of 140 ps in 266 .mu.s is required. Further, in the
short pulse tracking mode the quantization noise is here specified
to be consistent with a 66 dB dynamic range of a receiver. The A/D
strobes sent by the synchronizer 34 (FIG. 1) to the I.F. receiver
20 (FIG. 1) must, therefore, be positioned consistently with no
more than 13 ps RMS jitter (with respect to the transmitted
waveform) to achieve such a dynamic range. The timing of the A/D
strobes for the I.F. receiver 20 (FIG. 1) depends on the range to
the target or map reference. The range and therefore the time
resolution (140 ps) required relative to the maximum specified
system range dictate that the range be known to eighteen binary
places. The jitter requirement adds three more least significant
bits (LSB's), bringing the total number of bits needed up to 21.
Another result of the jitter constraint is a parallel requirement
on how often the range word must be updated within the
microcomputer 375 (FIG. 9C). Finally, from update to update smooth
LSB transitions should occur in the range word to accomplish
mapping and tracking without danger of the occurrence of
excessively large transients.
The foregoing considerations dictated the requirement that the
transmit-to-receive time interval be timed with 13 ps quantization
and 39 ps resolution. The direct digital implementation of a range
counter having 39 ps time increments requires a 25.6 GHz clock
frequency. Because no commercially available logic family supports
clock frequencies of that order of magnitude, a mixed analog and
digital approach is used here to achieve the desired time
resolution.
Thus, it will be appreciated by those of skill in the art that, in
general, by introducing an analog offset voltage into the control
loop of a phase-locked voltage-controlled oscillator (VCO), the
phase of the VCO can be moved away from that of the phase detector
reference frequency in a very controlled manner. Very precise
departures in phase can be achieved by using an accurate
digital-to-analog (D/A) converter to generate the analog offset
voltage. Thus, if the phase of a VCO were to be locked to the 50
MHz MDC, which, as explained, controls the transmit mode timing,
the phase of such VCO could be controlled over a 0 to 2.pi. radian
phase shift interval (corresponding to a 0 to 20 ns time interval).
In consequence, then, a very fine time resolution can be achieved
using a nine bit D/A converter to attain phase adjustments in
0.70.degree. increments (or 39 ps time increments). If a twelve bit
D/A is utilized with the 3 LSB's grounded, time uncertainty (or
jitter) is controlled down to less than 5 ps, thereby satisfying
the 13 ps jitter specification mentioned hereinabove.
It should now be appreciated that the phase-lock loop module 401
may be provided to develop clock signals for receive mode timing.
Thus, such module is shown to include a first and second phase-lock
loop (not numbered) which develop, respectively, a phase slipped
clock (PSC) signal and a frequency slipped clock (FSC) signal. The
PSC is used within the range counter module 403 (FIG. 9F) for the
receive mode timing, while the FSC signal is used to provide the
requisite A/D strobe frequency variation which, as mentioned
hereinabove, is required to control the depth-of-focus of the
generated maps. The first phase lock loop (not numbered) comprises
a digital phase detector 419, the reference signal to which is the
50 MHz MDC, a loop control amplifier 421, which here may be an
operational amplifier, a loop filter 423 and a voltage-controlled
crystal oscillator VCXO 425. A second control signal is applied to
the loop control amplifier 421, via a range register 427 (a 24 bit
device) and a converter 429 (a 12 bit device with the three least
significant bits grounded). The range register 427 is, at the start
of a given dwell, loaded with a 24 bit estimated range word,
R.sub.EST (which is also a measure of the transmit-to-receive
timing) and updated by the microcomputer 375 (FIG. 9C) at a 40 KHz
rate. The nine LSB's of the estimated range word, R.sub.EST, from
the range register 427 are translated to an analog voltage with 12
bit accuracy by the D/A 429 wherein the 3 LSB's are grounded. Thus,
nine LSB's of the estimated range word, R.sub.EST, are used to
develop the PSC for the receive mode timing. The thus described
phase lock loop (not numbered) is effective to divide one 20 ns 50
MHz clock cycle into 512 parts, each of 39 ps duration. The nine
LSB's of the estimated range word, R.sub.EST, set a phase delay for
the PSC in terms of these 39 ps time units.
A practical problem arises when very little phase difference exists
between the master digital clock (MDC) and the phase slipped clock
(PSC). In this event, the range counter module 403 (FIG. 9F) could
be enabled just before the rising edge of the PSC. The requisite
setup time for the range counter module 403 would not be met in
this instance. This problem is dealt with through the use of a
comparator 439 to detect marginal timing situations in the range
word, R.sub.EST. The output of the comparator 439 is combined in an
exclusive OR gate 441 with the 50 MHz MDC signal. Reference
numbers, corresponding to the ranges occurring 5 ns before and
after R.sub.EST are stored in the comparator 439. When the nine
LSB's of the range word, R.sub.EST, from the range register 427
fall between these reference numbers the comparator 439 causes the
phase of the 50 MHz MDC out of the exclusive OR gate 441 to be
inverted. Such reversal, as will be explained in detail
hereinbelow, ensures that the range counters within the range
counter module 403 (FIG. 9F) are enabled well in advance of the
first PSC signal to which they must respond.
As mentioned hereinabove, an A/D strobe frequency variation must be
included to provide for the depth of focus of the generated maps.
This A/D strobe frequency control is provided by means of the
second phase lock loop (not numbered), which here comprises a phase
comparator 431, the reference signal to which is the PSC signal, a
loop control amplifier 433, which may be an operational amplifier
used as a summing amplifier, a loop filter 435 and a VCXO 437. The
output signal from the phase lock loop (not numbered) is the
frequency slipped clock (FSC) signal which is passed ultimately to
the I.F. receiver 20 (FIG. 1). The loop control amplifier 433 is
shown to receive a pair of control signals, a first one of which is
received from a phase register 443 (12 bits) via a D/A 445, (12
bits with the 3 LSB's grounded). The second control signal to the
loop control amplifier 433 is received from a range register 447
(12 bits), via a D/A 449 (12 bits with the 3 LSB's grounded), and
an integrator 451, comprising a switch S1, a capacitor C1 and an
amplifier A1. The microcomputer 375 (FIG. 9C) provides the phase
register 443 with a phase offset word and the range register 447
with a range offset word. The phase offset word and the range
offset words are effective to produce the FSC having a phase ramp
(frequency offset) with both an adjustable starting phase and phase
slope which are dependent on measured, real-time motion
compensation parameters provided to the microcomputer 375 (FIG.
9C). It should now be appreciated by those of skill in the art that
the FSC signal is a coherent transform of the PSC signal and that
the starting phase and phase slope of the FSC signal are real time
functions of aircraft motion. Finally, it should be noted that the
integrator enable signal to the integrator 451 is provided a signal
from the timing and control logic unit for the receiver (which unit
has not been illustrated but is, as noted above, similar to the
timing and control logic unit being discussed).
Referring now to FIG. 9F, the range counter module 403 is shown to
include four J-K flip-flops (f/f) 453.sub.1, 453.sub.2, 453.sub.3
and 453.sub.4, each of which controls a corresponding counter
455.sub.1, 455.sub.2, 455.sub.3 and 455.sub.4 to make up four
counter paris (not numbered). it should be noted here that four
counter pairs are required to control the four transmitted
frequencies and that a different number of such pairs would be used
if the number of possible transmitted frequencies were changed. The
R.F. receiver 18 (FIG. 1) and the I.F. receiver 20 (FIG. 1) are
activated and strobed by the synchronizer 34 (FIG. 1) to pass the
desired return signal from any transmitted pulse, even though such
signal could occur at any time (even after the transmission of a
third transmitted pulse). The requisite range timing and gating
must be done concurrently for all four pulses because of the
interleaved nature of system timing and, therefore, four
independent range counters are required. Thus, the flip-flops
453.sub.1 . . . 453.sub.4 are shown to be clocked by the 50 MHz
noncoincidence MDC obtained from the exclusive OR gate 441 within
the phase lock loop module 401, while the counters 455.sub.1 . . .
455.sub.4, which are all sixteen bit devices, are clocked by the 50
MHz PSC. The data input to the J data terminals of each of the
flip-flops 453.sub.1 . . . 453.sub.4 are provided by the transmit
sequence control lines (not numbered) from the output register 417
within the mode transparent real time control module 399 (FIG.
9D).
The J data terminals of the flip-flops 453.sub.1 . . . 453.sub.4,
in conjunction with the Q output terminals, provide enabling
signals to the counters 455.sub.1 . . . 455.sub.4. The latter
receive via an exposed register bus (not numbered) the fifteen
MSB's of the range word, R.sub.EST, from the range register 427.
Once enabled, each of the counters 455.sub.1 . . . 455.sub.4 begin
to count the range word, R.sub.EST, and upon completion of the
count, each provides an output signal at the logic 1 level. The
output signals from the counters 455.sub.1 . . . 455.sub.4 are
provided as input signals to the K data terminals of their
respective flip-flops. It will be appreciated by those of skill in
the art that when both the J and K terminals of the flip-flops
455.sub.1 . . . 455.sub.4 are at the logic 1 level and a clock
pulse is received, the Q output will change state. Advantage is
taken of this property of the flip-flops 453.sub.1 . . . 453.sub.4
to provide enabling or disabling signals to the counters 455.sub.1
. . . 455.sub.4.
Synchronously with the terminal count word provided to the
flip-flops 453.sub.1 . . . 453.sub.4, receive sequence start
signals are sent, via an OR gate 457, to the "receive" mode, real
time control module (not shown, but corresponding to that shown in
FIG. 9D) to initiate the receive mode timing.
At this point in time a brief review of the PRI timing is in order.
The PRI is a cycle of four transmitted active pulses, each at a
different frequency. Preceding the transmission of each active
pulse is a panoramic analysis receive period and an additional
transmitted pulse (spoof frequency). The spoof frequency pulses
have unique frequencies different from that of the active pulse
frequencies. The active pulses could be either uncoded, frequency
ramped, or phase coded. The timing of the just described features
(i.e., the panoramic analysis, the spoof frequency transmissions
and the active pulses) is synchronized to the 50 MHz MDC.
Navigation and Position Update Modes (FIG. 10)
Before proceeding with a detailed description of the various
operating modes of the radar system 10 (FIG. 1) together with the
associated signal flow diagrams through the signal processor 26
(FIG. 1) and the digital data processor 30 (FIG. 1), it will be
beneficial at this point to remember that, in addition to being
used for weapon guidance, that radar must also be used to solve the
navigational problem involved. Part of such problem is that of
navigating the aircraft 302 (FIG. 10) to a position in space,
relative to the target area, with sufficient accuracy (both in
position and velocity) to make weapon delivery possible. The
problem is compounded by the fact that the aircraft 302 (FIG. 10)
must operate in a tactical environment, meaning that it will be
exposed to a hostile ECM environment, adverse weather conditions
and interdicting fire.
It will be recalled that the aircraft 302 (FIG. 10) is equipped
with an inertial platform 38 (FIG. 1) which is utilized, inter
alia, for navigation of the aircraft 302 (FIG. 10). Because the
inertial platform 38 (FIG. 1) is subject to long term drift errors
which effect the navigational accuracy, it is periodically updated
with radar derived velocity and position data. The aircraft 302
(FIG. 10), in the navigation mode, may be operating at an altitude
of from 500 to 30,000 feet which imposes a rather severe dynamic
range requirement, as will be explained in detail hereinbelow. To
minimize the risk of detection, the radar system 10 (FIG. 1) will
be operated only periodically and then for the minimum amount of
time. Furthermore, while penetrating enemy territory the aircraft
302 will operate at the minimum altitude and will rise (pop up) to
a high altitude only when required. In consequence, then, the
system of synthetic aperture measurements taken for position and
velocity updates must reflect a great degree of invention to the
effects of altitude.
As was explained hereinabove, synthetic aperture radar systems
require precise measurements of antenna acceleration for motion
compensation of the radar return signals. In order to satisfy
operational requirements and mounting constraints within the
aircraft 302 (FIG. 10), the antenna 14 (FIG. 1) and the inertial
platform 38 (FIG. 1) are physically separated. This physical
separation causes relative motion between the antenna 14 (FIG. 1)
and the inertial platform 38 (FIG. 1) due both to aircraft
maneuvers and to the flexure of the airframe. As was also explained
hereinabove, the motion of the antenna 14 (FIG. 1) can be measured
and therefore compensated by the high resolution inertial measuring
unit 24 (FIG. 1), which is mounted in close proximity to the
former. The long term alignment of the latter is maintained by
slaving it to the inertial platform 38 (FIG. 1).
Referring briefly back now to FIGS. 1, 8A and 8B, the method of
transfer alignment will be reviewed. Thus, the boresight sensor
assembly 36, which comprises three orthogonal ate gyros (not
shown), is mounted near the base of the antenna 14. The rate gyros
(not shown) are aligned to the aircraft roll, pitch and yaw axes
211, 203, 201, respectively. The sensed angular rates of such
gyros, together with the orientation and position of the antenna 14
relative to the inertial platform 38 (as measured by the IMU 24),
are used to determine the north and east velocity differences
between such antenna and platform during aircraft dynamic
maneuvers. The IMU 24 is mounted on the elevation gimbal 223 (FIG.
8B) and comprises two accelerometers (not shown) and three
orthogonal rate gyros (also not shown). The third accelerometer
required to complete the IMU 24 triad is the radial accelerometer
171 which is mounted at the rear center of the antenna 14 (FIG. 1).
The roll rate gyro (not shown) is aligned with the antenna
line-of-sight (LOS) vector (not numbered), while the pitch rate
gyro (not shown) and the Y-axis accelerometer (also not shown) and
the Z-axis accelerometer (also not shown) are mounted orthogonal to
the roll and pitch rate gyros (not shown). The rate gyro outputs
and the accelerometer outputs from the IMU 24, as well as the
accelerometer signal output from the radial accelerometer 171 (FIG.
8B) are passed to the strapdown processor 173 (FIG. 7C) wherein
processing is carried out to generate earth-referenced (north, east
and local vertical) mathematical gimbal axes as well as north and
east velocities. Such velocities are compared within a Kalman
filter (not shown but within the inertial platform 38) with
inertial velocities obtained from the inertial platform 38 to
determine both misalignment angles and sensor errors within the
head mounted IMU 24. These errors are then used to correct the
output signals from the head mounted IMU 24. Antenna axis
accelerations are sent from the strapdown processor 173 to the
digital data processor 30 for motion compensation purposes. This
acceleration data corresponds to the output signals from the head
mounted IMU 24 which have been gravity compensated as well as
corrected for bias and scale factor errors.
Referring now to FIG. 10, the aircraft 302 is shown to be flying
along a path such that its velocity vector, V, is coincident with
Y.sub.i axis of the aircraft inertial reference frame X.sub.i,
Y.sub.i, Z.sub.i. In the velocity update mode, the radar system 10
(FIG. 1) illuminates successive portions of the ground terrain (not
shown) and processes the return data from at least three of such
portions to derive the three components of aircraft velocity. The
theory of operation of Doppler radar navigators is well known to
those of skill in the art and will therefore not be recounted.
Suffice it to say here that the radar system 10 (FIG. 1) may be
thought of as transmitting a cone of constant Doppler frequency
centered about the aircraft velocity vector, V. The cone angle
corresponds to the beam squint angle, A, which is the angle between
the estimated position of the velocity vector, V, and the antenna
boresight axis 205 (FIG. 8B). The azimuth angle, B, is defined as
the angle between the horizontal projections of the antenna
boresight axis 205 and the velocity vector, V. The elevation angle,
C, is the angle of the antenna boresight axis 205 with respect to
the horizontal ground terrain.
The intersection of a constant doppler cone (not shown) with a
terrain produces a contour line of constant Doppler frequency
referred to as an isodop 501, which is here in the form of a
hyperbola when such terrain is horizontal. The particular shape of
the isodop 501 is dependent on the squint angle, A, and the height
of the aircraft 302 above the terrain. If, therefore, the altitude
of the aircraft 302 and the beam squint angle, A, are held constant
as the antenna 14 (FIG. 1) is moved from dwell to dwell, the
antenna boresight axis 205 will be forced to travel along the
isodop 501. It can be shown that the squint angle, A, the azimuth
angle, B, and the elevation angle, C, are related as follows:
Holding the squint angle, A, constant as the antenna 14 (FIG. 1) is
scanned will therefore require changes in both the azimuth angle,
B, and the elevation angle, C. As a given squint angle, A, and
aircraft altitude will define a particular hyperbola, the requisite
gimbal control commands to cause the antenna boresight axis 205 to
follow the isodop 501 may be computed from an a priori knowledge of
the equation of the hyperbola that corresponds to the isodop
501.
It should be recalled here that the widest possible focal width
maps under the conditions of aircraft acceleration are obtained by
controlling the PRI of the radar system 10 (FIG. 1). It can be
shown, however, that the PRI is also dependent on the squint angle,
A. That is, as the antenna 14 (FIG. 1) is squinted from broadside
toward the nose of the aircraft 302, the range to the illuminated
terrain continually increases. Thus, in order to avoid range
ambiguity problems and to prevent pulse eclipsing, the PRF of the
radar system 10 must decrease as the antenna 14 (FIG. 1) is
squinted toward the nose of the aircraft 302. Decreasing the PRF
will result in a concomitant increase in the PRI. The PRI control
problem is further aggravated by the fact that the depth, width and
height of focus of the illuminated terrain are all dependent on the
cell resolution width, .DELTA.a, which, in turn, is dependent,
inter alia, on the height of the aircraft 302 above the
terrain.
Under PRI variation the depth, height and width of focus are given,
respectively, by ##EQU6## where .DELTA.a=(.lambda./2) (h/d) (n/N)
(1/sin C) and n=number of dwells. Using the criteria given by the
foregoing equations ensures that the azimuth cell resolution is
sufficiently large for depth of focus to equal the transmitted
beamwidth. Thus, as the antenna 14 (FIG. 1) is slued to successive
points on the isodop 501, the number of pulses, N, for each
successive dwell is computed from Equation (12). ##EQU7##
As was mentioned above, the radar system 10 (FIG. 1) is equipped
with both an azimuth and elevation monopulse capability. Thus,
range and range rate (Doppler) measurements can be associated with
a specific direction in space. In order to use this capability to
estimate the velocity of the aircraft 302 the estimate of aircraft
velocity obtained from the inertial platform 38 (FIG. 1) is used to
first estimate the component of velocity along the antenna
boresight axis 205. As was mentioned hereinabove, the velocity data
estimates from the inertial platform 38 (FIG. 1) are compensated by
data from the boresight sensor assembly 36 (FIG. 1) and are passed
to the digital data processor 30 (FIG. 1) wherein they are used in
conjunction with estimates of the direction cosine matrix obtained
from the head mounted IMU 24 (FIG. 1) to develop estimates of the
range and range rate to the point on the terrain in the antenna
boresight direction. Radar measurements about these estimated
values of range and range rate are then made and the resulting data
is used to determine the error in the range rate (Doppler)
estimate.
Each beam or dwell measurement consists of data derived from each
of four illumination frequencies. It is noted here in passing that
the number of pulses for each of the four frequencies is determined
from Equation (12) prior to the start of each dwell. The radar
return data from each antenna phase center (quadrant) and for each
illumination frequency are passed through the R.F. receiver 18
(FIG. 1) and the I.F. receiver 20 (FIG. 1) wherein they are
downconverted to in-phase (I) and quadrature-phase (Q) video
signals. The I and Q signals are ultimately digitized and passed to
the signal processor 26 (FIG. 1). Within the latter the data
samples are motion compensated in a manner to be described in
detail hereinbelow. The resulting motion compensated data samples
are pulse compressed and processed by means of a fast Fourier
transform (FFT) algorithm ultimately to form range-Doppler data
arrays (N range cells by M Doppler cells) for the four antenna
phase centers and for each illumination frequency. It should be
noted here that for the purpose of this discussion it is presumed
that the transmitted pulses are binary phase encoded although,
depending on the mapping range, they could just as well be chirped
or uncoded. The data from each of the range-Doppler arrays are
processed, in a known manner such as is described in U.S. patent
application Ser. No. 047,957 filed June 11, 1979 by Ogar et al and
assigned to the assignee of the present application, to generate
normalized azimuth and elevation monopulse data on a cell-by-cell
basis for the N by M array for each illumination frequency. The
normalized azimuth and elevation monopulse data arrays are
processed, in a manner described in the just recited application,
to determine the Doppler frequency associated with the
zero-crossing point of the monopulse discriminator curve which
corresponds to the center of the radar beam. The difference between
the predicted and measured velocity, as determined from the Doppler
frequency associated with the zero crossing point, is the velocity
error for the beam data being processed.
The just described process is repeated for each beam and the
resulting velocity errors are combined with their associated beam
direction cosine matrix to produce the components of velocity error
in inertial reference coordinates for updating the inertial
platform 38 (FIG. 1).
POSITION UPDATE MODE (FIG 10)
As mentioned previously, the inertial platform 38 (FIG. 1) is
periodically updated in order to maintain the long term position
errors so that the aircraft 302 (FIG. 10) can be navigated to a
point (say point p, FIG. 10) in space under varying environmental
conditions and aircraft dynamics with, here, an error of no more
than 0.8 nautical miles. The position update mode entails mapping
the terrain surrounding a predetermined checkpoint whose position
is known and then determining the position errors of the inertial
platform 38 (FIG 1) by measuring the difference between the
position of such check point and the position of the center of the
radar beam in the map.
The position update mode is initiated by the radar operator, in
practice, at successive fifteen minute intervals during flight of
the aircraft 302 (FIG. 10). In the position update mode transmitter
power is radiated from a single antenna segment (say segment 14AS,
FIG. 3A) in order to produce a relatively broad antenna beam,
thereby to illuminate as large an area as possible around the
predetermined checkpoint. Return signals are, however, processed in
a manner similar to that described hereinabove for the velocity
update mode. That is, the map center which corresponds to the
antenna boresight axis 205 is computed both in azimuth and
elevation. In the position update mode, the map produced by
processing return signals is displayed to the radar operator. Using
a cursor (not shown) the operator designates the predetermined
checkpoint on the map. The offset of the predetermined checkpoint
with respect to the map center is then computed in antenna
reference frame coordinates. The cursor address of the
predetermined checkpoint in range/Doppler coordinates provides the
basis for determining an offset error for such checkpoint with
respect to the center of the beam. That offset error, in antenna
space coordinates, is then combined in a known manner with the beam
direction cosine matrix to provide the components of position error
in inertial reference frame coordinates for updating the inertial
platform 38 (FIG. 1).
FIXED TARGET DETECTION (FIG. 11)
The radar system 10 (FIG. 1) is required to automatically detect
trackable targets (meaning targets of significant military value
as, for example, armored columns, aircraft hangars, runways,
bridges, fuel supply depots or air defense radars), both fixed and
moving in the presence of ground clutter, large point targets and
ECM. The detection of fixed trackable targets requires a screening
process to sort the map data while the detection of moving targets
and their relocation to the proper map position requires
discrimination techniques between targets and ground clutter. It
should be noted here that the trackable target concept is important
in that it determines the requisite map resolution size which, in
turn, determines whether or not there will be a range slip problem
during any dwell. That is to say, the amount of range slip during a
given dwell may be expressed as:
where N.sub.RG is the number of range gates processed, .lambda.is
the wavelength of the transmitted signal and .DELTA.X is the map
resolution size. Substituting into the foregoing expression, it may
be seen that for a total of 256 range gates a wavelength of 0.1 ft.
and a resolution of 25 ft., the total range slip during a dwell is
approximately one-half of a cell.
The detection of fixed targets depends on the capability of the
radar system 10 (FIG. 1) to make sufficient quality high resolution
maps and then, by signal processing techniques, to highlight
potential targets on these maps. It should be noted here that a 12
m. (40 ft.) resolution area search map will have sufficient image
quality (even in a severe rain environment) to make both natural
and man-made features such as road intersections, mountains and
rivers clearly visible on the display. A 6 m. resolution map, on
the other hand, is needed to ensure recognition of targets such as
bridges, hangars and water tanks. In a higher resolution map a
bridge contrasts markedly from background (made up of usually of
water, railroad tracks or ravines over which the bridges pass);
hangars are detectable for a like reason because of their proximity
to runways and aprons; and water tanks are readily detectable
because they generally produce a regular geometrical high intensity
map image.
Stationary vehicles, however, present a different type of problem
because the dimensions of such vehicles are usually in the same
order as, or smaller than, the dimensions of a cell when a 6 or a
12 meter resolution is used. Although an ultra-high resolution map,
say one with a resolution of 1 to 2 meters, could be generated to
allow vehicles to be recognized, such a map would be too detailed
to be used by an operator in a tactical situation. That is to say,
there would be so many objects displayed, each of which could be a
vehicle but probably is not, that an operator would take too much
time to identify a vehicle, or vehicles.
The herein-contemplated radar system 10 (FIG. 1) employs a
screening process to winnow out, from a large number of displayed
objects in an ultra-high resolution map, those objects which most
probably are vehicles. The screening process then, obviously, aids
the operator by reducing the number of objects to be recognized and
classified. The screening process will be described in detail
hereinbelow. Suffice it to say here that the contemplated screening
process subjects all the processed radar return signals (which have
been converted to digital numbers representing magnitude in a
range-Doppler matrix) to a sequence of logical tests including: (a)
an amplitude threshold test, (b) a local maxima test, (c) a clear
region test, and (d) a scintillation test. The logical tests are
such that only radar return signals which are probably from
vehicles will pass predetermined set limits. Before testing a
preliminary process is carried on to determine that the digital
numbers being tested are not derived from jamming signals.
Referring now to FIG. 11, a block diagram representation of the
signal processing flow in the contemplated fixed target search mode
is presented. As may be seen the raw (unprocessed) radar return
data corresponding to each of the four transmitted carrier
frequencies from each of the antenna quadrants 14A, 14B, 14C and
14D, (FIG. 3A), as well as the data from the sidelobe channel (not
numbered) are stored in memory (not numbered). It should be noted
here that the data stored in memory (not numbered) have been
previously motion compensated by being complex multiplied by a
series of phase rotation multipliers as, for example, the REPLICA
lock data. The first step in the contemplated signal processing
procedure is to form the monopulse sum (.SIGMA.) signal for each of
the four transmitted frequencies from the antenna quadrant data.
The resulting .SIGMA. channel data for each of the four frequencies
are separately processed by means of an FFT algorithm to produce
four separate range/Doppler matrices (maps). The resulting maps are
then processed through two different signal processing paths, both
of which involve an editing process.
The first or ECM editing process examines each of the four maps to
see if any portions thereof are obscured or obliterated from the
effects of ECM as, for example, by means of active repeater jammers
and/or spot noise jammers. Each of the maps is also examined on a
cell by cell basis against data from the sidelobe monitor (not
numbered) which is provided in the slidelobe channel and which has
been separately FFT processed. The sidelobe monitor data editing
will be described in detail hereinbelow. Suffice it to say here
that the sidelobe monitor editing is effective to reduce the
sensitivity of the radar system 10 (FIG. 1) to the effects of
repeater type jammers.
After ECM editing the data corresponding to the four separate maps
are adaptively compressed from twelve bit magnitude words to four
bit magnitude words in order that they may be displayed in an
optimum manner on the video display unit (not shown). The adaptive
compression process involves rearranging the data samples in each
of the four maps in accordance with their amplitude distribution.
That is, the samples are rearranged in a new matrix wherein
amplitude is the abscissa and the number of samples is the
ordinate. The amplitude corresponding to the greatest number of
samples is then used to form the mean and four bit magnitude words
formed about such mean. The resulting four bit magnitude maps are
then incoherently added to form a single display map. The
incoherent addition tends to fill in the gaps in the individual
maps identified in the ECM editing process.
The second or fixed target detection editing process submits the
data in each of the four maps to: (a) an amplitude threshold test,
(b) a local maxima test, (c) a clear region test, and (d) a
scintillation test. The amplitude threshold test removes most of
the elements within the range-Doppler matrix from further
consideration. The surviving members are then subjected to a local
maxima test wherein it is required that the element of the map
under consideration be greater than its immediate neighbors since
it is desirable that a selected target stand out from its adjacent
neighbors to ensure that a point target is not counted more than
once.
Each of the elements that survive the local maxima test are then
subjected to a clear region test which requires that the local
maximum be greater by a fixed ratio than each of the targets
falling in a ring around the given target. This screening test
favors point targets over spread targets and also tends to select
the largest target in a given neighborhood.
Those elements which survive the clear region test are then
subjected to a scintillation test by comparing the four separate
maps. The scintillation test requires that the three immediately
preceding maps be investigated to determine the presence or absence
of a target which has passed the clear region test. The question to
be determined is whether or not the target under test is also
present in one, two or three of the preceding maps at an amplitude
which is within a given number of dB of its magnitude in the fourth
map. In the scintillation test a lower and upper threshold are
computed which are above and below the magnitude of the target
under test. This maximum is then tested to determine if it lies
within the adaptive bounds. The location of the maxima furnishes
the center point around which the next preceding map is tested.
This process is iterated for the three maps preceding the map under
test. It should be noted here that all the cells tested in the
scintillation test are in the same range bin as the target under
test and that to pass the scintillation test the target must fall
within the thresholds on three out of four maps.
An acceleration drift test is also employed to examine the Doppler
drift between time sequential source maps. Interpolated Doppler
frequencies for the cells surviving the prior tests are determined
for each of the time sequential maps. The individual cell
acceleration consistency is tested and the cell drift rate is
tested against the composite mean. Those cells that fluctuate
excessively or that exhibit drift rates which deviate sufficiently
from the average values will also be deleted.
The foregoing logic testing can be extended to include some
additional ECM editing features, as, for example, discrimination
against corner reflector decoys. As is known, corner reflectors are
utilized to simulate large fixed targets in an attempt to prevent
the detection of actual targets. However, corner reflector decoys
may be discriminated by virtue of their small scintillation
vis-a-vis actual tactical targets as, for example, tanks or trucks.
Further, the clear region ratio testing criteria can be modified to
detect extended or shaped targets. Geometric pattern shape
groupings of cell-sized targets can also be further tested in a
correlation process to evaluate the grouping as a possible convoy
or surface-to-air missile (SAM) site.
Each of the four maps are also edited with respect to the sidelobe
monitor data to remove jammer-induced false targets as was
described above. Following editing with the sidelobe monitor data,
the four maps are incoherently added to form a composite map. In
order for a potential target in the composite map to be
automatically designated to an operator, that target must meet the
following criteria in the composite map: (a) the magnitude of the
eight adjacent cells must be less than or equal to the magnitude of
the test cell; (b) at least five out of eight of the adjacent cells
must be less than half the magnitude of the test cell; and (c) the
magnitude of the nine cells contiguous to the cells adjacent to the
test cell must be no greater than one-quarter the magnitude of the
test cell. Further, a test cell that passes the foregoing criteria
must not have a magnitude in any of the four maps making up the
composite map which is less than half that of its integrated scaled
value on the composite map. The target search in the composite map
is initiated at the center of the map and proceeds to the map
extremities using a spiral search pattern.
Targets passing the foregoing tests are cataloged by range, angle,
Doppler and magnitude and are subsequently stored within the
digital data processor 30 (FIG. 1). Such targets are also
automatically displayed to the operator who may cursor a particular
target for display on a higher resolution map. The cursor addresses
the location of the selected target within the digital data
processor 30 (FIG. 1).
Moving Target Detection (FIG. 11A)
The radar system 10 (FIG. 1) must also detect ground moving targets
in an environment of extended ground clutter, large discrete
scatterers, wind-driven rain and chaff and active jamming. In such
an environment, targets with one percent relative velocity must be
detectable in the forward (.+-.20.degree.) sector, while targets
with radial velocities as low as 1 m/s must be detectable in the
squint (20.degree. to 90.degree.) sector. The latter case involves
the detection of moving targets which are competing with clutter
scatterers well inside the radar antenna sum beam.
Referring now to FIG. 11A, the contemplated moving target detection
technique will be described. Before proceeding, however, it will be
instructive to note that the effects of ground clutter and discrete
scatterer interference may be appreciated by recognizing that
coherent pulse Doppler radars provide high resolution synthetic
aperture mapping by virtue of the fact that each fixed ground point
scatterer that reflects energy has a Doppler phase history
corresponding to its mean angular position relative to the aircraft
velocity vector. Moving targets impart an additional Doppler phase
history to the radar signal; thus, the Doppler history from a
moving target does not correspond to the moving target's angular
position with respect to the velocity vector, V. In many cases
there will exist a ground clutter scatterer which has a Doppler
history due only to the motion of the aircraft 302 (FIG. 10) which
essentially corresponds with the Doppler history of a moving
target. The angle, as viewed from the aircraft, between the moving
target and such a ground clutter scatterer is referred to as the
displacement angle. The displacement angle is, as previously
explained, attributable to the Doppler ambiguities resulting from
the radial velocity of the moving target and the aircraft
line-of-sight cross velocity in the slant plane, and may be
expressed as:
where V.sub.T is the radial velocity of the target and V.sub.AC sin
.alpha. is the cross velocity of the aircraft 302 (FIG. 10). If a
displacement angle of significant magnitude exists, antenna
weighting can be used to attenuate the competing clutter scatterers
such that the moving target becomes detectable.
If the displacement angle is large relative to the transmitted
antenna beamwidth, then conventional .SIGMA. channel detection
techniques can be employed. That is, the competing clutter cell
will be attenuated relative to the target by virtue of the fact
that it is on the far main beam skirts or in the sidelobe region
when the target is within the main lobe. Sum channel detection
methods also apply when the total Doppler shift of the ground
moving target is beyond foldover (i.e., greater than the maximum
Doppler clutter frequency). Strictly speaking, there is no
competing ground clutter when this occurs although either noise
modulated clutter residues or ambiguously sampled clutter may be
detected along with wind-driven rain and chaff. Thus, when
required, rain rejection is provided by means of an adaptive
polarization technique which was described hereinabove.
Before proceeding, it should be noted here that during a given
radar dwell the antenna 14 (FIG. 1) will be slued to maintain the
line-of-sight vector on a single area target focal point. As the
antenna 14 (FIG. 1) is slued, the PRF of the radar system 10 (FIG.
1) must be changed accordingly. It should also be recalled that
during a given radar dwell the PRI of the latter is controlled, for
motion compensation purposes, such that equal angle increments are
swept out between pulse transmission times. Finally, it should be
noted that for long range applications a relatively low radar PRF
is utilized which may result in Doppler ambiguities in the case of
high speed targets. From the foregoing it should now be appreciated
by those of skill in the art that the contemplated moving target
detection technique must be adaptive in order to enable the radar
system 10 (FIG. 1) to detect both low and high speed targets in a
relatively complex clutter background.
Illustrated in FIG. 11A is a block diagram representative of the
contemplated signal processing technique for the ground moving
target detection mode. Similar to the fixed target detection mode,
the raw target return data (from each of the four antenna quadrants
corresponding to each of the four transmitted frequencies) are
complex multiplied by a series of phase rotation multipliers as,
for example, the REPLICA signal, and the phase shifts required as a
result of slueing or rotating the antenna 14 (FIG. 1) during the
radar dwell, as well as other motion compensation phase
multipliers, and are ultimately stored in memory.
Following the phase correction multipliers the data are passed to
an adaptive clutter canceller wherein a displaced phase center
clutter cancelling technique is utilized to cancel background
clutter. At this point it will be instructive to refer back for a
moment to FIG. 3A where the arrangement of the antenna quadrants is
illustrated. The phase centers of antenna quadrants 14A and 14B are
disposed along the horizontal axis (not numbered) of the antenna
14, while the phase centers of antenna quadrants 14C and 14D are
disposed along the vertical axis (also not numbered) of the antenna
14. The phase centers of antenna quadrants 14C and 14D must be
phase compensated to account for their difference in elevation
vis-a-vis the horizontal axis (not numbered) of the antenna 14.
One-half the sum of the two latter is designated "E".
Referring briefly now to FIG. 7A, the contemplated clutter
cancelling technique for the case of a two phase center antenna
will be explained. It should be recalled here that the following
technique was described in detail hereinabove with reference to the
motion compensation section where the technique was expanded to
include a three phase center antenna. Suffice it to say here that
if the phase centers are originally at A.sub.o and B.sub.o, then
according to FIG. 7A the best time to take the B.sub.1 sample for
subtraction from A.sub.o is when B crosses R.sub.Ao. The PRI of the
radar system 10 (FIG. 1) is controlled such that equal angle
increments are swept out by the antenna 14 (FIG. 1) between pulse
transmission times. This ensures that when the second pulse is
transmitted at time t.sub.1, the B phase center will be on the line
between the phase center A at time t.sub.o and the reference point,
o; i.e., phase center B is somewhere along R.sub.Ao. The exact
distance, L.sub.1, that phase center B at time t.sub.1 is from
phase center A at time t.sub.o must be known so that the phase of
the samples obtained from phase center B at time t.sub.1 can be
shifted by 4.pi. L.sub.1 /.lambda. in order to match those obtained
from phase center A at time t.sub.o. The requisite phase shifts are
obtained by generating the range, R.sub.Bo (t), between the B phase
center and the map reference point, 0, from the antenna velocity,
radial acceleration and squint angle and sampling it each time a
pulse is transmitted.
Once the radar return data has been properly phase compensated, the
characterization of clutter rejection is dependent, inter alia, on
the physical separation of the phase centers being cancelled. As a
general rule, the greater the separation between the phase centers
being cancelled, the greater the degree of slow target detection.
It will also be shown in detail hereinbelow that multiple antenna
phase centers may be utilized to distinguish between fast and slow
moving targets.
If the returns from phase center A are to be cancelled with the
returns from phase center B, the procedure involves cancelling
(subtracting) a present sample from phase center A with a previous
sample obtained from phase center B. The particular phase center B
sample chosen for cancellation (meaning the sample that corresponds
to a previously transmitted pulse) is dependent on the distance
between the A and B phase centers and here is the second previously
transmitted pulse. Thus, samples obtained from phase center A at
time t.sub.n are cancelled with phase center E samples obtained at
time t.sub.n-1. In like manner, if phase center E samples are to be
cancelled with phase center B samples, then phase center E samples
obtained at time t.sub.n are cancelled with phase center B samples
obtained at time t.sub.n-1.
Digressing here now for a moment, it should be recalled that each
fixed ground point scatterer that reflects energy has a Doppler
phase history corresponding to its mean angular position relative
to the velocity vector of the aircraft 302 (FIG. 10), and that
moving targets impart an additional Doppler phase history to the
radar signal. The relative angle as viewed from the aircraft 302
(FIG. 10) between a moving target and a ground scatterer having an
identical Doppler frequency as the moving target is known as the
displacement angle. As mentioned hereinbefore, if a sufficient
displacement angle exists, antenna weighting can be used to
attenuate the competing clutter scatterers such that the moving
target becomes detectable.
Referring now to FIG. 13, the effectiveness of the contemplated
antenna weighting technique in detecting both fast and slow moving
ground targets will be explained. Thus, the solid curve of FIG. 13
represents the difference pattern obtained by subtracting the
returns from phase center B from those received by phase center A.
The ordinate of FIG. 13 represents received signal strength, while
the abscissa represents Doppler frequency or, equivalently, angular
extent from a given null position. The returns from any ground
clutter scatterer located in the nulls will be completely
cancelled. As explained hereinabove, any ground moving target will
impart an additional Doppler frequency to the radar signal and,
therefore, the doppler frequency from a moving target will not
correspond to its angular position with respect to the null axis.
Thus, if the total Doppler frequency of a moving target is
sufficient to place the target return in the Doppler filter
corresponding to the null axis, that target will be detected. This
is the so-called moving target anomaly detection technique which is
described in co-pending patent application Ser. No. 582,965
entitled "Moving Target Indicator (MTI) Radar System", filed May
30, 1975, inventors Hiller et al., now U.S. Pat. No. 4,217,583
issued Aug. 8, 1980 It is important to note here that although the
anomaly detection technique permits the detection of moving
targets, further processing is required to accurately position the
target in angle. It should also be noted that if a moving target is
actually located in the reference null or is displaced in angle
from the reference null axis or Doppler filter by an amount equal
to the width of the difference pattern lobe, (i.e., it falls in the
second difference pattern null), it will not be detected. It can be
shown that that angular displacement here corresponds to the
displacement angle. Furthermore, as the displacement angle is
related to the relative velocity of the target, antenna weighting
can be utilized to distinguish between various speed ground
targets. The spacing between the nulls of the A-B pattern is
determined by dividing the operating wavelength, .lambda., by the
distance between phase centers A and B, which is here 12 .lambda..
As the displacement angle is equal to the difference between the
nulls, it can be seen that ground moving targets having a velocity
of 8 percent relative to the cross velocity of the aircraft 302
(FIG. 10) are likely not to be detected by means of the A-B
pattern. Thus, ground moving targets having a velocity of up to 8
percent of the aircraft cross velocity will be detectable by means
of the A-B pattern. This corresponds to the so-called "slow moving
target" detection mode.
If, on the other hand, the returns from phase center E (formed by
combining the returns from phase centers C and D in a manner
described in detail hereinabove) are subtracted from those of phase
center A to form a new null axis, the difference pattern indicated
by the dashed line of FIG. 13 is obtained. As may be seen, the
difference pattern lobe of the a-E pattern is twice as wide as that
of the A-B pattern. This follows from the fact that the distance
between the A and E phase centers is 6 .lambda.. Ground moving
targets having a relative velocity of up to 16 percent of the cross
velocity of the aircraft 302 (FIG. 10) will therefore be detectable
with A-E pattern. That is to say, the angular extent or Doppler
frequency excursion between the nulls of the A-E difference pattern
corresponds to a target having a velocity equal to 16 percent of
the cross velocity of the aircraft 302 (FIG. 10). This is the
so-called "fast moving target" detection mode. It should again be
noted here that the three phase center clutter cancellation
techniques just described simply allow for the detection of either
slow or fast moving ground targets, and does not provide for the
location in angle of the detected targets. Additional processing,
to be described in detail hereinbelow, is required to accurately
position the detected targets in angle.
Referring back now to FIG. 11A, following the adaptive clutter
cancelling phase processing described hereinbelow, the data
corresponding to each of the four transmitted frequencies for each
of the antenna phase centers A, B and E are processed by an FFT
algorithm, the net effect of which is to arrange the data in a
range/Doppler matrix. Following the FFT processing, the data
undergo a thresholding process wherein the difference patterns A-B,
A-E and E-B are formed. Those returns that exceed a predetermined
threshold level in the A-E and E-B patterns are classified as fast
moving target candidates, while those exceeding a similar threshold
in the A-B pattern are classified as slow moving target candidates.
The moving target candidates identified by the thresholding process
are next subjected to a series of editing processes designed to
filter out false targets induced by ground based ECM equipment.
The first of the editing processes is a so-called "SKIRT EDITOR"
designed to identify and remove those false targets introduced into
the radar system 10 (FIG. 1) by repeater type jammers. As is known,
the latter attempt to introduce false targets into the radar system
10 (FIG. 1) via the sidelobes of the antenna 14 (FIG. 1). If the
returns from the repeater jammers occur at frequencies
corresponding to the Doppler frequency of the returns from the main
lobe of the antenna 14 (FIG. 1) or at frequencies corresponding to
the Doppler ambiguities of the main lobe, then the false returns
could be mistaken for real targets. The skirt editing process is
designed to examine the sidelobes of the various antenna patterns
formed to test for the presence of false target returns. Thus, the
returns as seen by a single antenna quadrant, here quadrant A, are
compared with the returns as seen by the sum of antenna quadrants A
and B, (A+B), as well as the sum of all four antenna quadrants,
(A+B+C+D). It will be appreciated by those of skill in the art that
the pattern from the single antenna quadrant A is sufficiently
broad to cover the null in the A+B pattern, which results from the
physical arrangement of antenna quadrants A and B, and the sidelobe
region of the A+B+C+D pattern. In the contemplated skirt editing
process an arbitrary threshold is established based on the
magnitude of the return signals as seen by the single antenna
quadrant A. That threshold is here two times the magnitude of the
returns seen by antenna quadrant A. Hence, if twice the magnitude
of the returns seen by antenna quadrant A in any given region is
greater than the magnitude of the returns seen by either the A+B or
A+B+C+D patterns in the same region, then the returns from the
latter (sum patterns not exceeding the 2A threshold are rejected as
false targets.
It should be noted here that the just described skirt editing
process may be expanded, depending on the threshold level selected,
to provide some degree of protection against main lobe repeaters or
break-lock repeaters which initially repeat the radar signal
faithfully to capture the tracking circuits of the radar system 10
(FIG. 1) and then move off the true target return, either in range
or frequency, thereby pulling the radar system 10 (FIG. 1) off the
true target return signal.
The next step in the editing process is also an ECM editor referred
to as the sidelobe monitor. The sidelobe monitor editing is similar
to the skirt editing just described in that both are designed to
reject the returns from repeater type jammers which attempt to
enter the radar system 10 (FIG. 1) via the antenna sidelobes. It
should be noted here that the intent of these repeater type jammers
is to produce multiple false targets in the radar system 10 (FIG.
1) in an attempt to exceed the data handling capability of the
latter and thereby prevent true targets from being identified. In
the skirt editing process just described the returns detected in a
single antenna quadrant pattern were compared against the returns
seen by the patterns resulting from the sum of antenna quadrants A
and B as well as antenna quadrants A, B, C and D. The sidelobe
monitor editing provides not only protection against repeater type
jammers from entering the sidelobes of the single quadrant antenna
pattern, but also a degree of protection against spot noise
jammers. Thus, it will be recalled that the sidelobe monitor (not
numbered) described hereinabove with reference to FIGS. 4A and 4B
has an adaptive null forming capability which may be used to cancel
the returns from spot jammers.
It should be noted here that the sidelobe monitor data are received
through the auxiliary channel of the RF receiver 18 (FIG. 1) and
are separately FFT processed into a range-Doppler matrix. The
resulting map (range-Doppler matrix) is compared against similar
maps generated from the data received by each of the antenna
quadrants. The sidelobe monitor map is then compared with each of
the quadrant maps and any potential target whose amplitude in the
former is greater than in any of the latter is rejected as a false
target. This editing is dependent on the fact that the gain of the
sidelobe monitor pattern is greater than that of the antenna
quadrant patterns in the sidelobe region of the latter.
Following the sidelobe monitor editing the surviving target
candidates are subjected to an additional ECM editing stage
designed to nullify the effects of swept, frequency modulated (FM)
barrage type jammers on the radar system 10 (FIG. 1). As is known,
the effect of this type of jammer on the latter is to reduce the
signal-to-noise ratio to an extent determined by the effective
radiated power of the jammer, the range to the jammer, and the gain
of the antenna sidelobes in the direction of the jammer. One known
method to counter the effect of such a jammer is to turn off
(blank) the R.F. receiver 18 (FIG. 1) whenever the jammer sweeps
through the radar band. An alternate technique which takes full
advantage of the adaptive null forming capability of the herein
contemplated radar system 10 (FIG. 1) to discriminate against a
number of spatially separated slow sweeping barrage jammers would
be to form a spatial null in the direction of each jammer as they
randomly sequence through the radar spectrum. This technique
presupposes the existence of a means of cataloging the location of
the slow sweeping jammers which here may be provided by means of
the sidelobe cancelling circuits (not numbered) described
hereinabove with reference to FIGS. 4A and 4B. The contemplated
null forming technique will be described in detail hereinbelow with
reference to the target angle estimation section. Suffice it to say
here that that technique takes advantage of the multiple antenna
phase centers to form a pair of steerable monopulse nulls.
If, however, blanking must be utilized for some reason, as for
example, the sheer number of swept jammers that must be nullified,
then the herein contemplated radar system 10 (FIG. 1) still
provides an advantage. That is to say, systematic blanking disturbs
the continuity of the received waveform which will result in the
introduction of Doppler sidelobes in the return signal. However,
since the blanking commands are developed within the radar system
10 (FIG. 1), the signature (time of occurrence and duration) of the
blanks may be stored within the latter and upon completion of the
radar dwell time this blanking sequence (signature) may be
transformed into weights and used to reduce the generated Doppler
sidelobes to an acceptable level.
The potential moving target candidates that survive the foregoing
ECM editing processes are next subjected to an angle gate
thresholding process, sometimes referred to as a spread moving
target gate, which is designed to remove those potential moving
target candidates which are attributable to intrinsic clutter. The
effect of intrinsic clutter which may, for example, be due to the
motion of wind driven trees in leaf, is to cause an increase of
energy in a particular Doppler filter or a spread of energy between
a group of Doppler filters. The angle gate thresholding technique
therefore subjects each moving target candidate, which will be
recalled is positioned in a range Doppler matrix, to a monopulse
angle measurement. That measurement takes full advantage of the
multiple phase center antenna in forming the monopulse angle
estimate. Thus, the estimate is formed by normalizing the
difference of the E-A and B-E difference patterns to the sum of
those difference patterns and may be expressed as:
The procedure now is to form the foregoing ratio for every
candidate target in the range Doppler matrix. As is known,
normalization will assign each moving target candidate in the
range/Doppler matrix (map) to a particular Doppler filter or
angular location. If two or more target candidates are assigned to
the same Doppler filter or angular location, then the angle gate
editor will reject all of these candidates as being intrinsic
clutter due, for example, to the motion of a single tree. In like
manner, if some number n of target candidates, where n>4, are
assigned to adjacent Doppler filters or angular locations, the
angle gate editor will reject all of these candidates as being
intrinsic clutter due, for example, to the motion of a group of
wind driven trees.
Before proceeding, it should be recalled here that the radar system
10 (FIG. 1) is required not only to detect ground moving targets,
but also to accurately position those targets on a displayed map.
Thus, those moving target candidates that survive all of the
foregoing editing processes are subjected to a so-called "three
phase center" monopulse target detection technique. As previously
explained, the radar return data may be combined to provide a
minimum of three colinear phase centers A, B and E which lie in the
slant plane determined by the aircraft velocity vector and the line
of sight vector to a ground reference point. In the three phase
center monopulse target detection technique the returns from the
three phase centers are processed such that the angles to both a
target and its competing clutter patch can be resolved. The
solution is directly analogous to multiple sidelobe jammers in
which more than one jammer is cancelled in the sidelobes of a
tracking radar by using multiple small apertures, one for each
jammer. The three phase centers form a pair of independently
controllable difference patterns exhibiting two nulls, one for the
target and one for clutter. Error signals can then be obtained for
either or both the target or clutter by using the appropriate
source in the denominator of the three phase center monopulse
normalization routine.
As the moving target candidates that survive the foregoing editing
processes are arranged in a range-Doppler matrix (map), an estimate
of the Doppler frequency associated with each target cell is
readily obtained. With knowledge of the Doppler frequency
associated with the potential target cell, the angular location of
that cell with respect to the antenna boresight axis may be
determined by means of the following expression:
where f.sub.D is the Doppler frequency associated with the target
cell relative to boresight, V is the aircraft velocity, B is the
antenna aspect angle with respect to the aircraft velocity vector,
.lambda. is the wavelength of the transmitted signal and .DELTA.B
is the angular location of the target cell.
The first step in the three phase center target detection process
is to place the nulls obtained from both the A-E and E-B patterns
at the angular location of the target cell. It should now be
appreciated by those of skill in the art that the contemplated null
steering may be accomplished by shifting (rotating in angle) the
data from each of the antenna phase centers A, B and E prior to
forming the difference pattern beams. The requisite amount of phase
rotation is given by:
where d is the distance between the antenna phase centers and
.DELTA.B is the angular location of the target cell. It should also
now be appreciated that when both difference pattern nulls are
placed on the given target cell (Doppler filter) any clutter within
that cell (Doppler filter) will, for all practical purposes, be
completely nulled. Therefore, any residual energy within that cell
(Doppler filter) will be due to the motion of an off axis moving
target. The angular position of the moving target relative to the
null axis may be estimated by means of a conventional monopulse
(.DELTA./.SIGMA.) measurement made on the reference cell (Doppler
filter). That is to say, once the monopulse discriminator curve is
formed the off axis location of the target may be estimated from
the magnitude of the residual energy within the target or reference
cell. The procedure now is to phase shift (rotate in angle) one of
the difference pattern nulls by an amount given by:
where .DELTA.B.sub.T is the estimated off axis angular location of
the moving target. Next, with one of the difference pattern nulls
located on the cell (Doppler filter) corresponding to the estimated
position of the moving target, the reference cell (Doppler filter)
is re-examined. If the residual energy within the latter falls to
the level of the null or is otherwise indiscernible, then the
moving target is unambiguously located in angle. That is to say, if
the second difference pattern null is accurately positioned on the
moving target, then the latter will be unable to contribute any
energy to the reference cell or Doppler filter. If, on the other
hand, when the second difference pattern null is placed on the
estimated moving target cell there is little or no change in the
energy level within the reference cell (Doppler filter), then the
moving target is not accurately located and the process is
repeated.
Once the correct angular location of the moving target is
determined that location is stored in memory (not shown) for
subsequent display. The three phase center monopulse detection
technique is then repeated for each of the moving target candidates
that survived the editing processes. When all the moving target
candidates have been accurately located in angle a composite,
incoherent map is formed from the returns corresponding to each of
the transmitted frequencies. The composite map is displayed to an
operator who may then select a particular target or group of
targets for attack.
PPI Mode (FIG. 1)
The radar system 10 (FIG. 1) is required to provide a ground
mapping mode utilized principally for course navigation and in the
designation of sectors for high resolution mapping. In such a mode,
sometimes hereinafter referred to as the PPI mode, a non-coherent
frequency agile sector map is generated. The requisite maps must
cover a 60.degree. azimuth sector and must be displayed in any one
of three operator-designated range scales (i.e. 15 to 100 km, 5 to
40 km, or 5 to 20 km). The range resolution varies with the range
scales being respectively 150, 60 and 30 m. It will be appreciated
by those of skill in the art that in the long range (15 to 100 km)
application the standard antenna .SIGMA. beam will be sufficiently
broadened (due simply to antenna beam dispersion with range) to
provide the requisite elevation coverage. In consequence, then, in
the long range PPI mode the antenna 14 need only be stabilized in
elevation while it is slued in azimuth to provide the requisite
terrain coverage.
In the short range (5 to 20 km) PPI mode a different situation
arises. That is, the .SIGMA. beam will not be sufficiently
dispersed with range to provide the requisite elevation coverage.
In this instance, advantage is taken of one of the antenna
segments, here segment 14AS, to form a fan beam. As is known by
those of skill in the art, a fan beam will be formed from an
antenna where one of the principal dimensions of such antenna
exceeds the other by a ratio in the range of from 2 to about 5. The
length of the antenna segments 14AS, 14BS, 14CS and 14DS exceeds
their width by the required amount and, therefore, each of such
segments will form a fan beam. Thus, in the short range PPI mode
the antenna 14 (FIG. 1) is rotated to align antenna segment 14AS in
a vertical position and the four axis gimbal assembly 12 is used to
stabilize the antenna 14 at a constant elevation angle irrespective
of aircraft motion. Elevation monopulse calculations are then
performed on selected range bins and these monopulse estimates, as
a function of range, are employed to estimate and track the height
of the aircraft 302 (FIG. 10) above the mapped terrain. This
information is employed in controlling the antenna elevation fan
beam pointing. It should be noted here that, as mentioned briefly
hereinabove, the nominal PRF of the radar 10 will be varied as a
cosine of the antenna pointing angle to compensate for the change
in range with antenna scan angle.
Antenna Segment Null Patterns (FIGS. 3A, 4A, 4B)
It should be recalled here that the antenna aperture is subdivided
into four identical quadrants 14A, 14B, 14C and 14D and four
identical segments 14AS, 14BS, 14CS and 14DS. Further, as mentioned
hereinabove, the antenna aperture (not numbered) is divided into
square quadrants rather than serrated quadrants even though
serrated quadrants would provide low quadrant sidelobes in the
0.degree., 45.degree. and 90.degree. planes of the antenna 14. As
also noted, if the quadrants 14A, 14B, 14C and 14D were serrated
the grating lobes would be randomly distributed to either side of
the principal and diagonal planes of the antenna 14. The square
quadrant was here chosen because, in such a configuration, the
sidelobes are known to be along the diagonal planes and, therefore,
compensation for such sidelobes may be effected.
Referring now to FIG. 4B, it should be recalled that the output
signals from antenna segments 14AS, 14BS are passed to switch
networks 245V, 245H and that such switch networks may, in response
to a control signal provided by the radar synchronizer 34 (FIG. 1),
produce an output signal corresponding to: (a) the input signal
received from antenna segment 14AS; (b) the input signal received
from antenna segment 14BS; or (c) the difference between the input
signals received from antenna segments 14AS and 14BS. The output
signals from the switch networks 245V, 245H are ultimately passed
through the auxiliary channel (not shown) of the R.F. receiver 18
(FIG. 1) to the signal processor 26 (FIG. 1) and the digital data
processor 30 (FIG. 1).
It should now be appreciated by those of skill in the art that the
main lobes (fan beam) from antenna segments 14AS, 14BS will be of
greater magnitude vis-a-vis the antenna quadrant side lobes that
reside in their respective diagonal planes. In consequence, then,
the returns from antenna segments 14AS, 14BS may be utilized to
cancel jamming signals which would otherwise enter the radar system
10 via such diagonal plane sidelobes.
It should also be appreciated that the pattern resulting from
taking the difference of antenna segments 14AS, 14BS will have a
null located at the geometric center of the antenna 14
(corresponding to the location of the .SIGMA. beam peak) with lobes
extending radially outward in each of the diagonal planes. Such a
pattern is effective in cancelling jammers that would normally fall
beyond the range of the quadrant sidelobe cancellers. The segment
difference pattern offers a second advantage in that it provides
jammer cancellation without increasing the clutter level as the
clutter is along the range axis.
Difference Channel Sidelobe Cancelling
Referring back now for a moment to FIGS. 4A and 4B, it should be
recalled that the arithmetic networks 259V, 259H are included in
the sidelobe cancelling channels (not numbered) to form both the
monopulse sum and difference signals from the quadrant sidelobe
cancelling elements (not numbered). It should also be recalled that
switches 261V, 261H are provided to selectively apply either the
monopulse sum or difference signals through the R.F. receiver 18
(FIG. 1) to the digital data processor 30 (FIG. 1) for further
processing. The reason for selectively applying either the
monopulse sum or difference signals from the sidelobe cancelling
elements to the digital data processor 30 (FIG. 1) will be
appreciated when it is recognized that conventional sidelobe
cancellers utilize only a single beam to cancel the returns from
the sidelobes of both the sum and difference beams of the antenna
that they are intended to protect. Thus, while such single beam
sidelobe cancellers are effective in cancelling undesired returns
from the sidelobes of the main antenna sum beam, they introduce a
tracking error when they are utilized to cancel undesired returns
from the main antenna difference beam sidelobes. This tracking
error results from the fact that the beam from such conventional
sidelobe cancellers is designed to be of greater magnitude than the
sum beam sidelobes in the region of the sum beam sidelobes. In
consequence, then, the peak of the sidelobe canceller pattern will
be coincident with the null axis of the main antenna difference
beam and, therefore, interference signals from the sidelobe
canceller beam will be injected into the main antenna difference
channels. The radar system will then cause its difference channel
null to be shifted in an attempt to null these error signals,
thereby introducing tracking errors into the antenna system.
The problem in the herein contemplated radar system (wherein the
returns from each of the antenna quadrants 14A, 14B, 14C and 14D
(FIG. 3A) are modified by their sidelobe cancellers before being
combined to form monopulse sum and difference beams) is analogous
to the foregoing problem. Thus, while each of the quadrant
cancellers is controlled to provide the optimum degree of
cancellation for each of the antenna quadrants, the phase and
amplitude settings of each of the quadrant sidelobe cancellers will
not be identical due principally to the actual physical separation
between each of the quadrant sidelobe cancellers. As a result,
then, when the returns from, say, antenna quadrants 14A and 14B are
subtracted to form the monopulse azimuth difference beam it is
possible for the sidelobe cancellation components in each of the
channels to combine in such a manner as to introduce an
interference signal in the difference channel. In light of the
foregoing, the monopulse arithmetic networks 259V, 259H are
provided in the sidelobe cancelling channels (not numbered) so that
a monopulse difference pattern null may be formed from each of the
quadrant sidelobe cancellers and steered in the direction of the
interference. When the quadrant difference pattern is then formed,
the effect of the sidelobe cancellers in the resulting difference
channel will be minimized.
Ultrahigh Resolution Mapping (FIG. 14)
The herein-contemplated system is required to generate a high
resolution (2 ft.) map around an area designated by the radar
operator. This mode is utilized for target classification whereby
vehicles may be distinguished from each other. A chirp waveform
must be used to achieve the wide bandwidth required to achieve the
desired degree of resolution, along with equalization of the
waveform time sidelobes rising the replica lock equalization
technique which was mentioned before hereinabove. Briefly, that
technique involves taking a sample of the transmitted waveform and
using that sample to correct errors in the returned pulse, thereby
to reduce the time sidelobes of the compressed pulse and to correct
random pulse-to-pulse phase errors.
It will be appreciated by those of skill in the art that the focus
zone of the generated map would be an unacceptably small area
around the center of focus of the map if of aircraft motion effects
are not properly compensated. Thus, the techniques of equiangular
sampling, which was described hereinbefore, and polar format
interpolation, which will be described in detail hereinbelow, are
used to expand the focus zone into a usable size. The polar format
technique inherently implies the use of a two-dimensioned FFT for
processing since corrections must be made in the complex plane
before resolving the map into individual cells. In order to be able
to process the two dimensional map data in a reasonable amount of
time, the technique of A/D strobe frequency control, which was
described briefly hereinabove, is used to align the data samples
taken in such a way that they can be processed with a minimum
amount of interpolation and therefore in a minimum amount of
time.
Even with the compensations which are made for aircraft motion,
errors can cause areas of the map to be improperly focused when
there are large accelerations. Thus, a technique of self-focusing,
wherein point targets detected in the map are used to determine the
extent of the acceleration error so that the error may be removed
before the final FFT process which resolves all the cells, is used
to correct for acceleration errors.
Digressing here now for a moment, it should be recalled that the
replica-lock technique is used to solve two problems produced by
distortion of the wideband chirp signal in the high power microwave
portion of the radar system 10. This distortion takes the form of
amplitude and phase modulations which are impressed on the
transmitted waveform producing deviations from the ideal quadratic
phase characteristic of a chirp signal. These deviations, if not
compensated for in some manner, generate time (range) sidelobes and
Doppler sidelobes which degrade the small area contrast ratio of
the map. The second problem which is solved by the replica lock
technique is the removal of pulse-to-pulse random phase changes.
The average phase of the replica is measured for each transmitted
pulse and the deviation from nominal phase is removed in the signal
processor 26 (FIG. 1). The replica lock technique takes samples of
each transmitted pulse for the purpose of measuring the average
phase of the pulse. For the purpose of compensating for the
waveform transmission characteristics, the signal processing
necessary to derive the correction factor is only done once per
dwell as the phase transmission characteristics of the transmitter
and receiver components are slowly changing functions.
When the replica is used to compensate the received signal for time
(range) sidelobes, the compensation consists of multiplying the
digitized radar data by the reciprocal of a stored replica of the
transmitted pulse. Since both quantities are complex, this
operation to a first order is a series of phase rotations which
correct for phase irregularities in the transmitted signal.
As is known, defocusing of the map is caused by quadratic phase
terms in the recorded data for points displaced from the map center
which remain after the focus compensation for the center point has
been performed. While equiangular sampling does, in fact, linearize
the phase history for points displaced from the map center, its
effectiveness is limited in very high resolution applications by
differential range shifts of points within the map during the
mapping dwell. The differential range slip problem may be
understood by considering that as the synthetic aperture map is
developed the radar range gate is fixed on a central point. Points
at the same range but different azimuth positions have varying
ranges with respect to this point and, over the dwell, they will
move into and out of the central gate. If a simple range gated
system is used these targets will smear through several range
gates, and a defocused map will result. The relative target motion
at different azimuth angles is the parameter used to obtain azimuth
resolution and it is implicit in all synthetic aperture maps. For
ultra high resolution maps the differential motion of .lambda./2
(where .lambda. is the wavelength) required to resolve two adjacent
cells is about 1/40 of the cell size and since several hundred
azimuth cells are desired, a differential range slip of several
cells for points at the edges of the map is inherent and
unavoidable.
Polar formating is a signal processing technique used to solve the
differential range slip problem. The concept of polar formating
evolved out of optical processing where initially the dechirped,
uncompressed returns for each transmitted pulse are recorded
directly on film and the returns from successive pulses are laid
down in parallel strips. The result is a hologram of the ground
image. Coherent processing of the hologram (equivalent to a
two-dimensional FFT) recovered the ground image. When ultra high
resolution was attempted it was found that laying the strips down
with a slight angle between successive strips (instead of parallel)
would compensate for differential range slip and give a correct
hologram of the ground image. This slight angle between successive
strips was equivalent to using a polar coordinate system for
recording and became known as "polar formating". The herein
contemplated system uses a discrete digital equivalent of an
optical polar formating technique which has been optimized for the
real time operational requirements of the system. The technique
employs the following sequence of operations. First, a long
frequency ramped signal is transmitted and the received signal is
heterodyned against a similarly ramped local oscillation. In
digitizing the return data the timing of the A/D converter strobe
pulses is varied by means of the frequency slipped clock signal
developed within the synchronizer 34 (FIG. 1) to compensate for
changes in the line of sight depression angle during the dwell. The
PRI being varied, data samples equally spaced in angle around the
map center are produced. The digitized return data are subsequently
thinned in a presummer/roughing filter to lower the sample rate.
Next, interpolation of the data is performed to produce a modified
data set in which the nonlinearity of the phase history of any
point in the map is minimized. Finally, a two-dimensional FFT is
performed on the modified data set to generate the ultra high
resolution maps.
It should be noted here in passing that prior to interpolation the
return data consists of a fan-shaped grid of points in which the
data arranged in vertical columns are separated by a constant angle
as the result of PRI control and the data arranged in horizontal
rows are separated by a constant distance as the result of A/D
strobe control. Interpolation across horizontal points is then used
to effectively arrange the data in a horizontal grid.
The effect of the foregoing sequence of operations is to generate a
sampled data phase plane across which each resolvable point in the
map produces a unique, nearly linear phase gradient. Since the
effect of interpolation is to rearrange the data into the form of
an orthogonal grid, a two-dimensional FFT performed along the two
principal axes will produce a map in which the two dimensions are
also orthogonal. That is, the cross-coupling between the range and
the cross-range dimensions produced by differential range slip is
eliminated by the interpolation process.
As previously mentioned, the times at which A/D strobe pulses and
radar transmitter pulses are produced are both controlled to
simplify the computations involved in producing polar formatted
data. The equation which governs the timing of the A/D strobe
pulses is: ##EQU8## where f.sub.o =transmitter frequency
K=chirp slope
C=velocity of light
.DELTA.=map resolution
.theta.=line-of-sight azimuth angle to the map reference point
.phi.=line-of-sight elevation angle to the map reference point
r.sub.y =range to the map reference point
The strobe timing consists of a constant term and a linear term.
The constant term gives the time delay to the first data sample
taken after any transmitter pulse, while the linear term gives the
repetition interval of the strobe pulse train. Since both .theta.
and .phi. can be considered constant during any radar interpulse
interval, the strobe pulse frequency is constant during any receive
interval. The dependence of t upon cos .phi. is such as to maximize
the extent of the focused region in the presence of vertical
maneuvers.
Before proceeding with a detailed description of the contemplated
interpolation process, it will be instructive at this point to
recall that the result of stretch processing of the chirp waveform
from a point target located within the mapped region is a
relatively long fixed frequency tone whose frequency is a function
of the range to the target. The received tones are sampled with an
A/D at a fixed rate with the result that the point target is
thereby converted to a phase gradient in the sampled domain for
that pulse. This process is repeated over many pulses and a
two-dimensional sampled phase plane results.
Since the PRF is significantly higher than the frequency extent of
the ultra high resolution (UHR) map, the initial returns are over
sampled in the azimuth direction. Consequently, these data are
thinned in the presummer/roughing filter to obtain a lower sample
rate. In order to suppress aliasing around the filter output sample
lines, the roughing filter is sampled at approximately twice the
bandwidth occupied by the mapped region which restricts the phase
quadrant (the phase difference between successive data points)
to.+-.90.degree. for points at the edges of the map.
The basic equation for the phase, P, of a given sample is: ##EQU9##
where: N=number of samples in the range dimension
N.sub.y =y position of point target expressed as an output index
number
n=input index number in range
M=number of samples in the azimuth dimension
M.sub.x =x position of point target expressed as an output index
number
m=input index number in azimuth
B=resolution factor (resolution.div..lambda./2)
If the middle term in the foregoing equation were zero, the phase
would be the linear superposition of two independent functions and
two orthogonal FFTs could be performed to obtain the map. The
center term is not zero except, of course, for M.sub.x =0 at the
map center. This is due to differential range slip and is, as
explained hereinabove, inherent in all synthetic aperture maps.
The function P.sub.n,m can be converted to a function P.sub.n,k of
the form ##EQU10## Since, strictly speaking, the radar system 10
(FIG. 1) operates on complex voltages, the index, k, may be treated
as a noninteger with different gradient values depending on n. A
fractional counter (not shown) may then be used to increment k.
Samples for interpolation may then be selected by the integer
portion of the fractional counter and the weights to be applied by
the fractional part of the counter. For a given n, the
interpolation process will generate harmonics. The nature of the
harmonics, which have frequencies that are a function of n, is that
when the range FFT is performed, they tend to smear into several
Doppler filters. This smearing tends to reduce the harmonic power
in a single Doppler filter.
As mentioned briefly hereinabove, the contemplated UHR signal
processing mode utilizes a self-focus technique to correct for
quadratic phase errors in the map data introduced by inertial
instrument drift. If uncompensated, these errors would degrade the
focus quality of the map in the cross-range dimension thereby
increasing the amplitude of the near-in sidelobes and ultimately
broadening the cell width.
The self-focus technique consists of dividing the data record
obtained from a mapping dwell into subrecords in which the
quadratic phase error is small enough to be ignored. Each of these
subrecords is subsequently used to generate a map in which strong
point targets can be identified. Although the resolution of these
maps will be poorer than that specified for the UHR mode, it will
still be sufficient to permit an accurate determination of target
location in each of them. The principal effect of the quadratic
phase error produced by instrument drift is to cause the images of
fixed point targets to change their locations systematically in
successive maps. Thus, it is possible to estimate the magnitude of
the error by tracking the apparent motion of these targets. The
contemplated self-focus technique extracts acceleration correction
information in this manner and applies these corrections to the map
data gathered during the entire dwell to form a well focused, high
resolution map from the corrected data.
Having described a preferred embodiment of the contemplated system,
it will now be apparent to one of skill in the art that many
changes may be made without departing from our inventive concepts.
That is to say, the actual structural elements making up a system
according to our inventive concepts may be changed from those
illustrated so long as the resulting system is operative as
described. In particular, the provisions here made for: (a)
compensating for the error due to the presence of a radome and for
vibration of the antenna; (b) the assembly, in effect, of a number
of radars using a single array antenna assembly with a "four-axis"
gimbal to make up a monopulse radar which may be operated as a
synthetic aperture radar on a maneuvering aircraft; (c) the
"three-phase" monopulse arrangement to improve operation in the
presence of clutter; (d) the arrangement of a sidelobe cancelling
antenna located at the phase center of each quadrant of the single
array antenna assembly to render the system almost impervious to
electronic countermeasures which would incapacitate any known
system; and (e) the methods used to generate and to process signals
so that maps with various degrees of resolution may be generated
even in the presence of jamming signals, broadly distinguish our
inventive concepts. It is felt, therefore, that this invention
should not be restricted to its disclosed embodiment, but rather
should be limited only by the spirit and scope of the appended
claims.
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