U.S. patent number 4,794,506 [Application Number 07/023,239] was granted by the patent office on 1988-12-27 for resonant dc-dc converter.
This patent grant is currently assigned to Hitachi Medical Corporation. Invention is credited to Takanobu Hatakeyama, Hirofumi Hino.
United States Patent |
4,794,506 |
Hino , et al. |
December 27, 1988 |
Resonant DC-DC converter
Abstract
Disclosed is a resonant DC-DC converter which comprises an
inverter for converting DC power into AC power, a rectifier
connected to the AC output of the inverter through a resonant
transformer for rectifying AC output power of the transformer to
obtain DC power to be applied to a load, means for determining
respective operation phases of a first, a second, a third and a
fourth switching elements constituting the inverter in accordance
with set signals indicating a voltage and a current to be supplied
to the load, and a phase control circuit for controlling the
respective operation phases of the first, second, third and fourth
switching elements on the basis of an output signal of the phase
determining means in a manner so that the first and second
switching elements are alternately turned on with a phase
difference of 180 degrees with respect to an operation frequency of
the inverter, the third and fourth switching elements are
alternately turned on with a phase difference of 180 degrees with
respect to the operation frequency of the inverter, while varying a
phase difference from turn-on of the first switching element to
turn-on of the fourth switching element and a phase difference from
turn-on of the second switching element to turn-on of the third
switching element to thereby control or feed-back control the DC
power to be supplied to the load.
Inventors: |
Hino; Hirofumi (Noda,
JP), Hatakeyama; Takanobu (Ryugasaki, JP) |
Assignee: |
Hitachi Medical Corporation
(Tokyo, JP)
|
Family
ID: |
11965259 |
Appl.
No.: |
07/023,239 |
Filed: |
March 9, 1987 |
Foreign Application Priority Data
|
|
|
|
|
Jan 30, 1987 [JP] |
|
|
62-18209 |
|
Current U.S.
Class: |
363/25; 363/98;
378/112; 363/17; 363/132 |
Current CPC
Class: |
H05G
1/08 (20130101); H05G 1/46 (20130101); H05G
1/32 (20130101); H02M 3/3376 (20130101); H05G
1/20 (20130101); H02M 7/53878 (20210501); Y02B
70/10 (20130101) |
Current International
Class: |
H02M
3/337 (20060101); H02M 3/24 (20060101); H05G
1/46 (20060101); H05G 1/00 (20060101); H05G
1/08 (20060101); H05G 1/20 (20060101); H05G
1/32 (20060101); H02M 003/335 () |
Field of
Search: |
;363/9,17,25,97,98,132,134 ;323/241 ;378/104,111,112 |
References Cited
[Referenced By]
U.S. Patent Documents
Primary Examiner: Salce; Patrick R.
Assistant Examiner: Jones; Judson H.
Attorney, Agent or Firm: Antonelli, Terry & Wands
Claims
What is claimed is:
1. A resonant DC-DC converter comprising:
a DC power source;
an inverter for receiving DC power from said DC source and for
converting the DC power into AC power, said inverter having a first
series connection of first and second switching elements
respectively connected to a positive and a negative pole of said DC
power source, a second series connection connected in parallel to
said first series connection and composed of third and fourth
switching elements disposed so as to respectively correspond to
said first and second switching elements, and first, second, third
and fourth diodes anti-parallelly connected to said first, second,
third and fourth switching elements, respectively;
a transformer, including primary and secondary windings, connected
to an output of said inverter for boosting an output voltage of
said inverter;
a rectifier for converting an AC output voltage of said transformer
into a DC output voltage;
a load connected to said rectifier;
said transformer including a parasitic leakage inductance and stray
capacitance existing among said primary or secondary windings of
said transformer, said leakage inductance and said stray
capacitance being used as resonance elements so that a resonance
voltage induced at said stray capacitance by the resonance between
said leakage inductance and said stray capacitance and a
transformation ratio is applied to said rectifier;
means for smoothing the DC output voltage of said rectifier and for
applying the smoothed DC output voltage to said load;
a phase determination circuit for determining respective operation
phases of said first, second, third and fourth switching elements
of said inverter and substantially over the entire range of
operation of said inverter in accordance with set signals
indicating a voltage and a current to be supplied to said load;
a phase control circuit for controlling the respective operation
phases of said first, second, third and fourth switching elements
on the basis of an output signal of said phase determination
circuit in a manner so that said first and second switching
elements are alternately turned on with a phase difference of 180
degrees with respect to an operation frequency of said inverter,
said third and fourth switching elements are alternately turned on
with a phase difference of 180 degrees with respect to the
operation frequency of said inverter, while varying a phase
difference from a point in time of turn-on of said first switching
element to a point in time of turn-on of said fourth switching
element and a phase difference from a point in time of turn-on of
said second switching element to a point in time of turn-on of said
third switching element to thereby control the power supplied to
said load in accordance with the phase determination made by said
phase determination circuit.
2. A resonance DC-DC converter according to claim 1, in which said
smoothing means is an electrostatic of a high tension cable for
applying said DC output voltage to said load.
3. A resonant DC-DC converter according to claim 1, in which said
load is an X-ray tube.
4. A resonant DC-DC converter according to claim 2, in which said
load is an X-ray tube.
5. A resonant DC-DC converter comprising:
a DC power source;
an inverter for receiving DC power from said DC source and for
converting the DC power into AC power, said inverter having a first
series connection of first and second switching elements
respectively connected to a positive and a negative pole of said DC
power source, a second series connection connected in parallel to
said first series connection and composed of third and fourth
switching elements disposed so as to respectively correspond to
said first and second switching elements, and first, second, third
and fourth diodes anti-parallelly connected to said first, second,
third and fourth switching elements, respectively;
a transformer, including primary and secondary windings, connected
to an output of said inverter for boosting an output voltage of
said inverter;
a rectifier for converting an AC output voltage of said transformer
into a DC output voltage;
a load connected to said rectifier;
said transformer including a parasitic leakage inductance and stray
capacitance existing amoung said primary or secondary windings of
said transformer, said leakage inductance and said stray
capacitance being used as resonance elements so that a resonance
voltage induced at said stray capacitance by the resonance between
said leakage inductance and said stray capacitance and a
transformation ratio is applied to said rectifier;
means for smoothing the DC output voltage of said rectifier and for
applying the smoothed DC output voltage to said load;
a voltage divider for detecting the voltage applied to said
load;
an error amplification phase determination circuit for receiving a
detection signal from said voltage divider and an externally
applied preset target voltage signal so as to amplify a difference
between said detection signal and said target voltage signal, and
for determining respective operation phases of said first, second,
third and fourth switching elements of said inverter substantially
over the entire range of operation of said inverter, on the basis
of said amplified difference; and
a phase control circuit for controlling the respective operation
phases of said first, second, third and fourth switching elements
on the basis of an output signal of said error amplification phase
determination circuit in a manner so that said first and second
switching elements are alternately turned on with a phase
difference of 180 degrees with respect to an operation frequency of
said inverter, said third and fourth switching elements are
alternately turned on with a phase difference of 180 degrees with
respect to the operation frequency of said inverter, while varying
a phase difference from a point in time of turn-on of said first
switching element to a point in time of turn-on of said fourth
switching element and a phase difference from a point in time of
turn-on of said second switching element to a point in time of
turn-on of said third switching element to thereby effect feed-back
control of power supplied to said load in accordance with the phase
determination made by said error amplification phase determination
circuit.
6. A resonant DC-DC converter according to claim 5, in which said
smoothing means is an electrostatic capacitance of a high tension
cable for applying said DC output voltage to said load.
7. A resonant DC-DC converter according to claim 5, wherein said DC
power source is obtained from AC power received from a commercial
power source, rectified into a DC power and then smoothed.
8. A resonant DC-DC converter according to claim 6, wherein said DC
power source is obtained from AC power received from a commercial
power source, rectified into a DC power and then smoothed.
9. An X-ray generating apparatus having a resonant DC-DC converter
comprising:
a DC power source;
an inverter for receiving DC power from said DC source and for
converting the DC power into AC power, said inverter having a first
series connection of first and second switching elements
respectively connected to a positive and a negative pole of said DC
power source, a wecond series connection connected in parallel to
said first series connection and composed of third and fourth
switching elements disposed so as to respectively correspond to
said first and second switching elements, and first, second, third
and fourth diodes anti-parallelly connected to said first, second,
third and fourth switching elements, respectively;
a transformer, including primary and second windings, connected to
an output of said inverter for boosting an output voltage of said
inverter;
a rectifier for converting an AC output voltage of said transformer
into a DC output voltage;
an X-ray tube applied with a smoothed output voltage of said
rectifier via a high tension cable;
said transformer including a parasitic leakage inductance and stray
capacitance existing among said primary or secondary windings of
said transformer, said leakage inductance and said stray
capacitance being used as resonance elements so that a resonance
voltage induced at said stray capacitance by the resonance between
said leakage inductance and said stray capacitance and a
transformation ratio is applied to said rectifier;
a voltage divider for detecting the voltage applied to said X-ray
tube;
an error amplification phase determination circuit for receiving a
detection signal from said voltage divider and an externally
applied preset target voltage signal so as to amplify a difference
between said detection signal and said target voltage signal, and
for determining respective operation phases of said first, second,
third and fourth switching elements of said inverter substantially
over the entire range of operation of said inverter, on the basis
of said amplified difference; and
a phase control circuit for controlling the respective operation
phases of said first, second, third and fourth switching elements
on the basis of an output signal of said error amplification phase
determination circuit in a manner so that said first and second
switching elements are alternately turned on with a phase
difference of 180 degrees with respect to an operation frequency of
said inverter, said third and fourth switching elements are
alternately turned on with a phase difference of 180 degrees with
respect to the operation frequency of said inverter, while varying
a phase difference from a point in time of turn-on of said first
switching element to a point in time of turn-on of said fourth
switching element and a phase difference from a point in time of
turn-on of said second switching element to a point in time of
turn-on of said third switching element to thereby effect in
feed-back control of power supplied to said X-ray tube in
accordance with the phase determination made by said error
amplification phase determination circuit.
10. A resonant DC-DC converter according to claim 9, wherein said
DC power source is obtained from AC power received from a
commercial power source, rectified into a DC power and then
smoothed.
Description
BACKGROUND OF THE INVENTION
The present invention generally relates to a resonant DC-DC
converter and particularly to a resonance DC-DC converter in which
AC power is applied to a resonant transformer from a suitable AC
power source and an AC output power of the transformer is rectified
is supplied to a given load.
Such a resonant DC-DC converter is disclosed, for example, in U.S.
Pat. No. 4,504,895. FIG. 8 is a diagram showing the resonant DC-DC
converter disclosed in U.S. Pat. No. 4,504,895 but somewhat
summarized for the sake of explanation. In FIG. 8, the DC-DC
converter comprises: a DC power source 1; an inverter 2 for
receiving DC power from the DC source and for converting the DC
power into AC power, the inverter having a first series connection
of first and second transistors Tr.sub.1 and Tr.sub.2 respectively
acting as a first and a second switching element respectively
connected to a positive and a negative pole of the DC power source
1, a second series connection connected in parallel to the first
series connection and composed of third and fourth transistors
Tr.sub.3 and Tr.sub.4 respectively acting as third and fourth
switching elements disposed so as to respectively correspond to the
first and second transistors, and first, second, third and fourth
diodes D.sub.1 -D.sub.4 anti-parallelly connected to the first,
second, third and fourth transistors Tr.sub.1 -Tr.sub.4
respetively; a transformer 3 connected to an output of the inverter
for boosting an output voltage of the inverter; a rectifier 4 for
converting an AC output voltage of the transformer into a DC output
voltage; electrostatic capacitance C for smoothing an output
voltage from the rectifier 4, and a load 5 connected to an output
of the rectifier 4. The transistors Tr.sub.1 -Tr.sub.4 are arranged
to be driven by a frequency determination circuit 6 and a frequency
control circuit 7 through driving circuits 8a to 8d,
respectively.
The transformer 3 is used to isolate the input and output of the
converter from each other, and used to boost or reduce the output
voltage in the case where the input voltage is different from the
output voltage. Particularly, in the case where a high voltage of
several tens KV to 200 KV is generated, for example, in a power
source for generating X-rays, the turn ratio of the transformer 3
is very large and hence the number of turns of the secondary
winding is very large. Accordingly, the secondary windings are
formed in layers which are stacked one on one while being insulated
one from one with insulators such as insulating sheets interposed
between adjacent layers. As a result, stray capacitances C.sub.S1
-C.sub.Sn are formed between the layers of the secondary windings
in the transformer 3 as shown in FIG. 9A. The circuit of FIG. 9A
can be expressed by such an equivalent circuit as shown in FIG. 9B.
That is, the serial capacitances C.sub.S1 -C.sub.Sn form a stray
capacitance C.sub.S of the secondary winding. Further, the
transformer 3 per se may be expressed by leakage inductance L.sub.1
and L.sub.2 and excitation inductance L.sub.ex and therefore the
whole of the transformer 3 may be expressed by those inductance
L.sub.1, L.sub.2 and L.sub.ex together with the stray capacitance
C.sub.S as shown in FIG. 9C. Further, generally, L.sub.1
<<L.sub.ex and L.sub.2 <<L.sub.ex, so that leakage
inductance L.sub.S being parasitic on the transformer 3 is
expressed by L.sub.S =L.sub.1 +L.sub.2 and the equivalent cirucit
of the transformer 3 can be shown as FIG. 9D.
If such a transformer 3 is used, the leakage inductance L.sub.S
being parastic on the transformer and the stray capacitance C.sub.S
of the secondary winding can be used as resonance elements so that
a voltage induced at the stray capacitance C.sub.S by the resonance
between the leakage inductance L.sub.S and the stray capacitance
C.sub.S and the transformation ratio of the transformer 3 is
supplied to the rectifier 4 and the output voltage of the rectifier
4 is applied to the load 5 after being smoothed through the
electrostatic capacitance C. In order to control the output power
to be supplied to the load 5, the ratio F.sub.i /F.sub.o of the
operation fequency F.sub.i to the resonance frequency F.sub.o
determined by the leakage inductance L.sub.S and the stray
capacitacne C.sub.S is varied by the frequency determination
circuit 6 and the frequency control circuit 8 shown in FIG. 8. That
is, in the graph of FIG. 10, the respective curve shows the
relation between the ratio F.sub.i /F.sub.o and the output voltage
V.sub.o of the transformer 3 with the ordinate and abscissa
representing the input voltage from the DC power source 1 and the
output voltage V.sub.o and with load resistance R.sub.1, R.sub.2, .
. . , R.sub.5 (R.sub.1 >R.sub.2 >. . . >R.sub.5) as
parameters. Since the resonance frequency takes a constant value
determined by the leakage inductance L.sub.S and the stray
capacitance C.sub.S of the transformer 3, the output voltage
V.sub.o is controlled by suitably varying the operation frequency
F.sub.i of the inverter 2.
In the thus arranged conventional resonant DC-DC converter, as seen
in FIG. 10, the output voltage V.sub.o becomes maximum when the
value of the ratio F.sub.i /F.sub.o is about "1", and if the
operation frequency F.sub.i of the inverter 2 is made lower or
higher than the resonance frequency F.sub.o, the output voltage
V.sub.o is reduced. However, the output voltage V.sub.o cannot be
reduced to zero. In this case, the output voltage V.sub.o can be
made to approach zero if the operation frequency F.sub.i of the
inverter 2 is made extremely low or extremely large. If the
operation frequency F.sub.i is made lower, however, the time
quadrature of the voltage applied to the transformer 3 becomes
larger and therefore it is necessary to make the sectional area of
the core of the transformer 3 larger, resulting in increase in size
of the transformer 3. Further, there is a limit in making the size
of the transformer 3 and the operation frequency F.sub.i cannot be
made so low to thereby cause a limit in the output voltage control
range. Further, even if the operation frequency F.sub.i of the
inverter 2 is made higher than an audio frequency so as to make
noises low, the operation frequency F.sub.i may reach the audio
frequency range to allow noises to become high when the operation
frequency F.sub.i is made low in order to make the output voltage
low in the case of a light load. Also in this case, accordingly,
the operation frequency F.sub.i cannot be made so low to thereby
cause a limit in the output voltage control range.
SUMMARY OF THE INVENTION
It is therefore an objcct of the present invention to eliminate the
drawback in the prior art.
It is another object of the present invention to provide a resonant
DC-DC converter in which a control range of an output power can be
extended without changing the operation frequency of the inverter
and in which the converter can be reduced in size.
In order to attain the above objects, according to the present
invention, an improvement is made in a resonant DC-DC converter
comprising: a DC power source; an inverter for receiving DC power
from the DC source and for converting the DC power into AC power,
the inverter having a first series connection of a first and a
second switching element respectively connected to a positive and a
negative pole of the DC power source 4, a second series connection
connected in parallel to the first series connection and composed
of third and fourth transistors Tr.sub.3 and Tr.sub.4 respectively
acting as a third and a fourth switching element disposed so as to
respectively correspond to the first and second transistors, and
first, second, third and fourth diodes antiparallelly connected to
the first, second, third and fourth switching elements,
respectively; a transformer connected to an output of the invetter
for boosting an output voltage of the inverter; a rectifier for
converting an AC output voltage of the transformer into a DC output
voltage; and a load connected to the rectifier; the transformer
having a parasitic leakage inductance and a stray capacitance
existing among primary or secondary windings of the transformer; in
which the leakage inductance and the stray capacitance are used as
resonance elements so that a resonance voltage induced at the stray
capacitance by resonance between the leakage inductance and the
stray capacitance and a transformation ratio is applied to the
rectifier, and the DC output voltage of the rectifier is smoothed
and applied to the load.
In an aspect of the invention, the DC-DC converter is further
provided with: a phase determination circuit for deteriining
respective operation phases of the first, second, third and fourth
switching elements of the inverter in accordance with set signals
indicating a voltage and a current to be supplied to the load; and
a phase control circuit for controlling the respective operation
phases of the first, second, third and fourth switching elements on
the basis of an output signal of the phase determination circuit in
a mnnner so that the first and second switching elements are
alternately turned on with a phase difference of 180 degrees with
respect to an operation frequency of the inverter, the third and
fourth switching elements are alternately turned on with a phase
difference of 180 degrees with respect to the operation frequency
of the inverter, while varying a phase difference from a point in
time of turn-on of the first switching element to a point in time
of turn-on of the fourth switching element and a phase difference
from a point in time of turn-on of the second switching element to
a point in time of turn-on of the third switching element to
thereby control the power supplied to the load.
In another aspect of the invention, the DC-DC converter is further
provided with: a voltage divider for detectigg the voltage applied
to the load; an error amplification phase determination circuit for
receiving a detection signal from the voltage divider and an
externally applied preset target voltage signal so as to amplify a
difference between the detection signal and the target voltage
signal, and for determining respective operation phases of the
first, second, third and fourth switching elements of the inverter,
on the basis of the amplified difference; a phase control circuit
for controlling the respective operation phases of the first,
second, third and fourth switching elements on the basis of an
output signal of the error amplification phase determination
circuit in a manner so that the first and second switching elements
are alternately turned on with a phase difference of 180 degrees
with respect to an operation frequency of the inverter, the third
and fourth switching elements are alternately turned on with a
phase difference of 180 degrees with respect to the operation
frequency of the inverter, while varying a phase difference from a
point in time of turn-on of the first switching element to a point
in time of turn-on of the fourth switching element and a phase
difference from a point in time of turn-on of the second switching
element to a point in time of turn-on of the third switching
element to thereby effect feedback-control of the power supplied to
the load.
The above and other objects and features of the invention will
appear more fully hereinafter from a consideration of the following
description taken in connection with the accompanying drawings.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a circuit diagram of an embodiment of the resonance DC-DC
converter according to a first aspect of the invention;
FIG. 2 is an equivalent circuit showing the main circuit portion in
the resonance DC-DC converter shown in FIG. 1,
FIG. 3 is a time chart for explaining the operation in the case
where control is made so as to make the output power maximum;
FIG. 4 is a time chart for explaining the operation in the case
where control is made so as to reduce the output power;
FIG. 5 is a graph showing relation between a phase difference and a
tube voltage with load resistance values as parameter in the phase
determination circuit;
FIG. 6 is a circuit diagram of an embodiment of the resonance DC-DC
converter according to a second aspect of the invention;
FIG. 7 a circuit diagram of another embodiment of the resonance
DC-DC converter according to the second aspect of the
invention;
FIG. 8 is a circuit diagram showing a conventional resonance DC-DC
converter;
FIGS. 9A-9D are diagrams showing various equivalent circuits of the
transformer used in the conventional resonant DC-DC converter of
FIG. 8; and
FIG. 10 is a graph for explaining control of the output voltage in
the conventional resonant DC-DC converter of FIG. 8.
DESCRIPTION OF THE PREFERRED EMBODIMENTS
Embodiments of the resonant DC-DC converter according to the
present invention will be described in detail with reference to the
accompanying drawings hereunder.
FIG. 1 is a circuit diagram showing an embodiment of the resonant
DC-DC converter according to a first aspect of the invention. This
resonant DC-DC converter is provided with DC power source 1 such as
a secondary battery or the like, an inverter 2, a transformer 3, a
rectifier 4, and a load 5.
The inverter 2 is arranged to receive DC power from the DC power
source 1 so as to convert the DC power into AC power. The inverter
2 is constituted by: a first series connection including a first
transistor Tr.sub.1 acting as a first switching element and being
connected at its collector to a positive pole of the DC power
source 1 and a second transistor Tr.sub.2 acting a second switching
element and being connected at its emitter to a negative pole of
the DC power source 1; a second series connection connected in
parallel to the first series connection and composed of a third
transistor Tr.sub.3 acting as a third switching element and being
disposed so as to correspond to the first transistor Tr.sub.1 and a
fourth transistor Tr.sub.4 acting as a fourth switching element and
being disposed so as to correspond to the second transistor
Tr.sub.2 ; and first, second, third, and fourth diodes D.sub.1
-D.sub.4 anti-parallelly connected to the first through fourth
transistors Tr.sub.1 -Tr.sub.4, respect. Each of the transistors
Tr.sub.1 through Tr.sub.4 is arranged to be turned on when supplied
with a base current.
The transformer 3 is connected to the output of the inverter 2 so
as to boost an output voltage of the inverter 2. The transformer 3
has a predetermined turns ratio between the primary and secondary
windings and has a leakage inductance L.sub.S and a stray
capacitance C.sub.S which are the same as shown in FIG. 9D.
The rectifier 4 is constituted by four diodes D.sub.5 through
D.sub.8 and arranged to convert an output voltage of the
transformer 3 into a DC output voltage through full-wave
rectification. The output of the rectifier 4 is connected to the
load 5, for example, an X-ray tube or the like. An electrostatic
capacitance C of a high tension cable for applying an output
voltage of the rectifier 4 to the load 5 such as an X-ray tube or
the like serves to smooth the output voltage of the rectifier
4.
In this embodiment according to the first aspect of the invention,
there are further provided: a phase determination circuit 9 for
determining respective operation phases of the transistors Tr.sub.1
through Tr.sub.4 of the inverter 2 in accordance with set signals
instructing a voltage and a current to be supplied to the load 5
such as an X-ray tube (hereinafter, referred to as a tube voltage
and a tube current respectively); a phase control circuit 10 for
generating control signals for controlling respective operation
phases of the transistors Tr.sub.1 through Tr.sub.4 on the basis of
an output signal of the phase determination circuit 9 and for
sending out the control signals in response to an X ray exposure
signal applied from a controller (not shown); and driving circuits
11a through 11d for driving the transistors Tr.sub.1 through
Tr.sub.4 respectively on the basis of the control signals sent-out
from the phase control circuit 10. Power to be supplied to the
X-ray tube acting as the load 5 is controlled by the phase
determination circuit 9 and the phase control circuit 10 in such a
manner that the respective transistors Tr.sub.1 and Tr.sub.2 acting
as the first and second switching elements of the inverter 2 are
alternately turned on with a phase difference of 180 degrees with
respect to an operation frequency of the inverter 2, and the
respective transistors Tr.sub.3 and Tr.sub.4 acting as the third
and fourth switching elements are alternately turned on with a
phase difference of 180 degrees similarly to the transistors
Tr.sub.1 and Tr.sub.2, while properly varying a phase difference
from a point in time of turn-on of the first transistor Tr.sub.1 to
a point in time of turn-on of the fourth transistor Tr.sub.4 and a
phase difference from a point in time of turn-on of the second
transistor Tr.sub.2 to a point in time of turn-on of the third
transistor Tr.sub.3.
Next, description will be made as to the operation of the thus
arranged resonant DC-DC converter. First, a main circuit portion
constituted by the DC power source 1, the inverter 2, the
transformer 3, the rectifier 4, and the load 5 in the resonant
DC-DC converter of FIG. 1 can be expressed by an equivalent circuit
shown in FIG. 2. That is, the transistors Tr.sub.1, Tr.sub.2,
Tr.sub.3, and Tr.sub.4 constituting the inverter 2 are shown by the
first, the second, the third, and the fourth switching element 12a,
12b, 12c, and 12d respectively, and the transformer 3 is shown by
the leakage inductance L.sub.S and the stray capacity C.sub.S
similarly to FIGS. 9A-9D. The first switching lement 12a and the
diode D.sub.1 anti-parallelly connected to the former constitute a
first arm 13a. Similarly to this, the second switching element 12b
and the diode D.sub.2 constitute a second arm 13b, the third
switching element 12c and the diode D.sub.3 constitute a third arm
13c, and the fourth switching element 12d and the diode D.sub.4
constitute a fourth arm 13c.
Then, referring to timing diagrams of FIGS. 3 and 4, description
will be made as to the operation of the main circuit portion by
using the equivalent circuit of FIG. 2.
First, referring to FIG. 3, description will be made as to the
operation in the case where control is performed so as to maximize
the output power of the inverter 2. At this time, the operation
phases of the switching elements 12a through 12d are controlled in
such a manner that the first and second switching elements 12a and
12b are alternately turned on with a phase difference of 180
degrees, the fourth and third switching elements 12d and 12c are
alternately turned on with a phase difference of 180 degrees, the
first and fourth switching elements 12a and 12d are turned on with
a phase difference of zero, that is, turned on simultaneously with
each other, and the second and third switching elements 12b and 12c
are turned on with a phase difference of zero, that is, turned on
simultaneously with each other, as shown in the diagrams (a)
through (d) of FIG. 3.
First, at a point in time Ta.sub.1 in FIG. 3, a resonance current
i.sub.t is being caused to flow by energy of the leakage inductance
L.sub.S through each of a circuit of the leakage inductance L.sub.S
.fwdarw. the diode D.sub.1 .fwdarw. the DC power source 1.fwdarw.
the diode D.sub.4 .fwdarw. the stray capacity C.sub.S .fwdarw. the
leakage inductance L.sub.S, another circuit of the leakage
inductance L.sub.S .fwdarw. the diode D.sub.1 .fwdarw. the DC power
source 1.fwdarw. the diode D.sub.4 .fwdarw. the rectifier 4.fwdarw.
the electrostatic capacity C.fwdarw. the rectifier 4.fwdarw. the
leakage inductance L.sub.S, a further circuit of the leakage
inductance L.sub.S .fwdarw. the diode D.sub.1 .fwdarw. the DC power
source 1.fwdarw. the diode D.sub.4 .fwdarw. the rectifier 4.fwdarw.
the load 5.fwdarw. the rectifier 4.fwdarw. the leakage inductance
L.sub.s. Hereinafter, the state where the resonance current i.sub.t
flows through the three divisional paths after the fourth diode
D.sub.4 is similarly expressed by D.sub.4 .fwdarw. (C.sub.S, 4, C,
and 5).fwdarw.L.sub.S. As a result, negative currents i.sub.1 and
i.sub.4 flow in the first and fourth arms 13a and 13d respectively
(the diagrams (e) and (f) of FIG. 3). At this time, the first and
fourth switching elements 12a and 12d are turned on, resulting in
no influence on the circuits through which the currents i.sub.1 and
i.sub.4 flow. Thereafter, the resonance current i.sub.t approaches
zero as shown in the diagram (i) of FIG. 3 as the energy of the
leakage inductance L.sub.S decreases.
Next, after the resonance current i.sub.t becomes zero at a point
in time Ta.sub.2 as shown in the diagram (i) of FIG. 3, the
resonance current i.sub.t flows through a circuit of the DC power
source 1.fwdarw. the switching element 12a.fwdarw. the leakage
inductance L.sub.S .fwdarw. the stray capacitance C.sub.S .fwdarw.
the switching element 12d.fwdarw. the DC power sou in FIG. 2 while
increasing along an arcuate curve of a resonance frequency
determined by the leakage inductance L.sub.S and the stray
capacitance C.sub.S. Thereafter, when a voltage across the stray
capacitance C.sub.S becomes equal to that across the electrostatic
capacitance C, the resonance current i.sub.t flows through a
circuit of the DC power source 1.fwdarw. the switching element
12a.fwdarw. the leakage element L.sub.S .fwdarw.(CS, 4, C, and
5).fwdarw. the switching element 12d.fwdarw. the DC power source
1.
At a point in time Ta.sub.3, the first and the fourth switching
elements 12a and 12d are turned off, and the second and the third
switching elements 12b and 12c are turned on, as shown in the
diagrams (a) through (d) of FIG. 3. The resonance current i.sub.t,
however, is caused to flow by the energy of the leakage inductance
L.sub.S through a circuit of the leakage inductance L.sub.S
.fwdarw.(C.sub.S, 4, C,) and 5) .fwdarw. the diode D.sub.3 .fwdarw.
the DC power source 1.fwdarw. the diode D.sub.2 .fwdarw. the
leakage inductance L.sub.S, in FIG. 2. Therefore, negative currents
i.sub.2 and i.sub.3 flow in the second and third arms 13b and 13c
respectively (the diagrams (g) and (h) of FIG. 3). Thereafter, the
resonance current i.sub.t approaches zero as shown in the diagram
(i) of FIG. 3 as the energy of the leakage inductance L.sub.S
decreases.
The resonance current i.sub.t becomes zero at a point in time
Ta.sub.4 as shown in the diagram (i) of FIG. 3. Then, the resonance
current i.sub.t flows through a circuit of the DC power source
1.fwdarw. the switching element 12c.fwdarw. the stray capacitance
C.sub.S .fwdarw. the leakage inductance L.sub.S .fwdarw. the
switching element 12b.fwdarw. the DC power source 1 in FIG. 2 so
that the amplitude of the resonance current i.sub.t negatively
increases along the arcuate curve of the resonance frequency
determined by the leakage inductance L.sub.S and the stray
capacitance C.sub.S. When the voltage across the stray capacitance
C.sub.S becomes equal to that across the electrostatic capacitance
C, the resonance current i.sub.t flows through a circuit of the DC
power source 1.fwdarw. the switching element 12c.fwdarw. (C.sub.S,
4, C, and 5).fwdarw. the leakage inductance L.sub.S .fwdarw.
12b.fwdarw. the DC power source 1.
At a point in time Ta.sub.5, the second and third switching
elements 12b and 12c are turned off, and the first and fourth
switching elements 12a and 12d are turned on, as shown in the
diagrams (a) through (d) of FIG. 3. As a result, the switching
elements 12a through 12d become in the same state as that at the
point in time Ta.sub.1, and thereafter the foregoing operation is
repeated.
In the case where the operation phase of the inverter 2 is
controlled as described above, the first and fourth switching
elements 12a and 12d are turned on/off with a phase difference of
zero, that is, simultaneously turned on/off, and the second and
third switching elements 12b and 12c are turned on/off with a phase
difference of zero, that is, simultaneously turned on/off.
Therefore, the current i.sub.1 flowing in the first arm 13a has the
same waveform as that of the current i.sub.4 flowing in the fourth
arm 13d as shown in the diagrams (e) and (f) of FIG. 3, and the
current i.sub.2 flowing in the second arm 13b has the same waveform
as that of the current i.sub.3 flowing in the third arm 13c as
shown in the diagrams (g) add (h) of FIG. 3. The period in which
both the first and fourth switching elements 12a and 12d are in the
on-state (from the point in time Ta.sub.1 to the point in time
Ta.sub.3) is continued in the following period in which both the
second and third switching elements 12b and 12c are in the on-state
(from the point in time Ta.sub.3 to the point in time Ta.sub.5), so
that the output voltage V.sub.t of the inverter 2 has a continuous
square waveform having a positive and a negative crest value equal
to the output voltage of the DC power source 1 as shown in the
diagram (j) of FIG. 3 and the output power of the inverter 2
becomes maximum.
Next, referring to FIG. 4, description will be made as to the
operation in the case where control is performed so as to reduce
the output power of the inverter 2. In this case, the respective
operation phases of the switching elements 12a through 12d of the
inverter 2 of FIG. 2 are controlled in such a manner that the first
and second switching elements 12a and 12b are alternately turned on
with a phase difference of 180 degrees, the fourth and third
switching elements 12d and 12c are alternately turned on with a
phase difference of 180 degrees, the first and fourth switching
elements 12a and 12d are turned on with a time lag, that is, a
phase difference .alpha. from the turn-on of the former to the
turn-on of the latter, and the second and third switching elements
12b and 12c are turned on with a time lag, that is, a phase
difference from the turn-on of the former to the turn-on of the
latter, as shown in the diagrams (a) through (d) of FIG. 4.
First, at a point in time Tb.sub.1 of FIG. 4, the first and the
third switching elements 12a and 12c are in the on-state in FIG. 2,
so that the resonance current i.sub.t is caused to flow by the
energy of the leakage inductance L.sub.S through a circuit of the
leakage inductance L.sub.S .fwdarw. the diode D.sub.1 .fwdarw. the
switching element 12c.fwdarw. ((C.sub.S, 4, C, and 5) .fwdarw. the
leakage inductance L.sub.S. Therefore, the negative current i.sub.1
flows in the first arm 13a as shown in the diagram (e) of FIG. 4
while the positive current i.sub.3 flows in the third arm 13c as
shown in the diagram (h) of FIG. 4. Thereafter, the resonance
current i.sub.t approaches zero as shown in the diagram (i) of FIG.
4 as the energy of the leakage inductance L.sub.S decreases.
The resonance current i.sub.t becomes zero at a timing Tb.sub.2,
and then the resonance current i.sub.t is caused to flow by energy
of the stray capacitance C.sub.S through a circuit of the stray
capacitance C.sub.S .fwdarw. the diode D.sub.3 .fwdarw. the
switching element 12d.fwdarw. the leakage inductance L.sub.S
.fwdarw. S the stray capacitance C.sub.S, while increasing along an
arcuate curve of a resonance frequency determined by the leakage
inductance L.sub.S and the stray capacitance C.sub.S, as shown in
the diagram (i) of FIG. 4. At this time, the positive current
i.sub.1 begins to flow in the first arm 13a as shown in the diagram
(e) of FIG. 4 while the negative current i.sub.3 begins to flow in
the third arm 13c as shown in the diagram (h) of FIG. 4.
The third switching element 12c is turned off at a point in time
Tb.sub.3 as shown in the diagram (d) of FIG. 4, an at the same time
the fourth switching element 12d is turned on as shown in the
diagram (b) of FIG. 4. As a result, both the first and fourth
switching elements 12a and 12dare in the on-state as shown in the
diagrams (a) and (b) of FIG. 4, and the diode D.sub.3 is reversely
biased to be turned off in response to the turn-on of the fourth
switching element 12d, so that the resonance current i.sub.t flows
through a circuit of the DC power source 1.fwdarw. the switching
element 12a.fwdarw. the leakage inductance L.sub.S .fwdarw. the
stray capacitance C.sub.S .fwdarw. the switching element 12.sub.d
.fwdarw. the DC power source 1. Thereafter, when the voltage across
the stray capacitance C.sub.S becomes equal to that across the
electrostatic capacitance C, the resonance current i.sub.t flows
through a circuit of the DC power source 1.fwdarw. the switching
element 12a.fwdarw. the leakage inductance L.sub.S
.fwdarw.(C.sub.S, 4, C, and 5) .fwdarw. the switching element
12d.fwdarw. the DC power source 1. In this period, the current
i.sub.1 flowing in the first arm 131a has the same waveform as the
current i.sub.4 flowing in the fourth arm 13d as shown in the
diagrams (e) and (f) of FIG. 4.
At a point in time Tb.sub.4, the first switching element 12a is
turned off as shown in the diagram (a) of FIG. 4, and the second
switching element 12b is turned on as shown in the diagram (c) of
FIG. 4. At this time, the resonance current i.sub.t is caused to
flow by the energy of the leakage inductance L.sub.S through a
circuit of the leakage inductance L.sub.S .fwdarw. (C.sub.S, 4, C,
and 5).fwdarw. the switching element 12d.fwdarw. the diode D.sub.2
.fwdarw. the leakage inductance L.sub.S. Therefore, the negative
current i.sub.2 flows in the second arm 13b as shown in the diagram
(g) of FIG. 4 while the positive current i.sub.4 flows in the
fourth arm 13d as shown in the diagram (f) of FIG. 4. Thereafter,
the resonance current i.sub.t approaches zero as shown in the
diagram (i) of FIG. 4 as the energy of the leakage inductance
L.sub.S decreases.
The resonance current i.sub.t becomes zero at a point in time
Tb.sub.4, and then the resonance current i.sub.t flows through a
circuit of the stray capacitance C.sub.S .fwdarw. the leakage
inductance L.sub.S .fwdarw. the switching element 12b.fwdarw. the
diode D.sub.4 .fwdarw. the stray capacitance C.sub.S .fwdarw. in
FIG. 2, while negatively increasing its amplitude an arcuate curve
of the resonance frequency determined by the leakage inductance
L.sub.S and the stray capacitance C.sub.S, as shown in the diagram
(i) of FIG. 4.
At a point in time Tb.sub.6, the fourth switching element 12d is
turned off as shown in the diagram (b) of FIG. 4, and the third
switching element 12c is turned on as shown in the diagram (d) of
FIG. 4. As a result, the second and third switching elements 12b
and 12c are in the on-state as shown in the diagrams (c) and (d) of
FIG. 4 respectively, so that the resonance current i.sub.t flows
through a circuit of the DC power source 1 .fwdarw. the switching
element 12c.fwdarw. the stray capacity C.sub.S .fwdarw. the leakage
inductance L.sub.S .fwdarw. the switching element 12b.fwdarw. the
DC power source 1 in FIG. 2. Thereafter, when the voltage across
the stray capacitance C.sub.S becomes equal to that across the
electrostatic capacitance C, the resonance current i.sub.t flows
through a circuit of the DC power source 1.fwdarw. the switching
element 12c.fwdarw. (CS, 4, C, and 5).fwdarw. the leakage
inductance L.sub.S .fwdarw. the switching element 12b.fwdarw. the
DC power source 1. In this period, the current i.sub.2 flowing in
the second arm 13b has the same waveform as that of the current
i.sub.3 flowing in the third arm 13c as shown in the diagrams (g)
and (h) of FIG. 4.
Then, at a point in time Tb.sub.7, the second switching element 12b
is turned off as shown in the diagram (c) of FIG. 4, and the first
switching element 12a is turned on as shown in the diagram (a) of
FIG. 4, so that the switching elements 12a through 12d are in the
state quite the same as that at the point in time Tb.sub.1, the
foregoing operation being repeated thereafter.
In the case where the operation phase of the inverter 2 is
controlled in such a manner as described above, the first and
fourth switching elements 12a and 12d are turned on/off with a time
lag or a phase difference .alpha., and the second and third
switching elements 12b and 12c are turned on/off with a time lag or
a phase difference .alpha.. Therefore, a first simultaneous
on-period in which both the first and fourth switching elements 12a
and 12d are in the on-state (from the point in time Tb.sub.3 to the
point in time Tb.sub.4) is shorter by .alpha. than a period in
which each of the switching elements 12a and 12d is in the
on-state, and a second simultaneous on-period in which the second
and third switching elements 12b and 12c are in the on-state (from
the point in time Tb.sub.6 to the point in time Tb.sub.7) is
shorter by .alpha. than a period in which each of the switching
elements 12b and 12c is in the on-state. Only in each of the
above-mentioned discontinued or intermittent first and second
simultaneous on-periods, DC power is applied from the DC power
source 1 to the load 5. Therefore, the output voltage V.sub.t of
the inverter 2 has an intermittent square waveform which has a
positive and a negative crest value equal in absolute value to the
output voltage of the DC power source 1 respectively in the
foregoing discontinued or intermittent simultaneous on-periods
(each 180-.alpha.degrees), as shown in the diagram (j) of FIG. 4.
Accordingly, the above-mentioned simultaneous on-periods can be
changed so as to control the power supplied to the load by properly
changing the phase difference .alpha. from turn on of the first
switching element 12a to turn-on of the fourth switching element
12d and the phase difference from turn on of the second switching
element 12b to turn-on of the third switching element 12c is
suitably changed. That is, by increasing the respective phase
difference .alpha., the output power can be gradually reduced to
zero as the phase difference increases up to 180 degrees where
there is no period in which both the switching elements 12a and 12d
as well as both the switching elements 12b and 12c are in the
on-state.
The phase determination circuit 9 and the phase control circuit 10
of FIG. 1 which control the main circuit portion of FIG. 2 are
controlled so as to operate the inverter in such a manner as
described above. Now, description will be made as to the operations
of the phase determination circuit 9 and the phase control circuit
10 for controlling the operation phases of the transistors Tr.sub.1
through Tr.sub.4 constituting the inverter 2 as shown in FIG. 1.
First, when a tube voltage and a tube current to be supplied to the
X-ray tube acting as the load 5 are determined, a tube voltage set
signal S.sub.1 and a tube current set signal S.sub.2 corresponding
to the tube voltage and the tube current, respectively, are applied
from the controller (not shown) to the phase determination circuit
9. The phase determination circuit 9 is constituted by a memory
storing tables representing a graph which expresses the
relationship between the phase difference .alpha. and the tube
voltage V of the X-ray tube acting as the load 5 by means of curves
with the abscissa and the ordinate representing the phase
difference and the tube voltage V respectively and with load
resistance values R.sub.1, R.sub.2, R.sub.3. . . (R.sub.1
>R.sub.2 >R.sub.3. . . ) as parameters, for example, as shown
in FIG. 5. Alternatively, the phase determination circuit 9 may be
constituted by a function generator, an operational amplifier, or
the like. In this phase determination circuit 9, the load
resistance value R.sub.1, R.sub.2, . . . , or the like, for example
R.sub.3, is obtained on the basis of the tube voltage set signal
S.sub.1 and the tube current set signal S.sub.2, and then a point
on the curve concerned with the obtained parameter, that is, the
load resistance value R.sub.3 is obtained corresponding to a tube
voltage V to be set now to thereby determine the necessary phase
difference .alpha. for operations of the transistors Tr.sub.1
through Tr.sub.4 of the inverter 2 corresponding to the obtained
point on the curve, as shown by a dotted in FIG. 5. Then, a phase
signal S.sub.3 corresponding to the thus determined phase
difference .alpha. is produced from the phase determination circuit
9, and applied to the phase control circuit 10. On the basis of the
phase signals S.sub.3, the phase control circuit 10 generates
control signals for turnnng on/off the respective transistors
Tr.sub.1 through Tr.sub.4, and a control signal for controlling the
phase difference .alpha. from turn-on of the transistor Tr.sub.1
acting as the first switching element to turn-on of the transistor
Tr.sub.2 acting as the fourth switching element as wel as the phase
difference from turn-on of the transistor Tr.sub.2 acting as the
second switching element to turn-on of the transistor Tr.sub.3
acting as the third switching element. Upon reception of the X-ray
exposure signal S.sub.4 applied from the controller (not shown),
the phase control circuit 10 supplies the generated control signals
to the driving circuits 11a through 11d, so that the driving
circuits 11a through 11d drive the respective transistors Tr.sub.1
through Tr.sub.4 constituting the inverter 2 on the basis of the
control signals supplied from the phase control circuit 10.
When the transistors Tr.sub.1 through Tr.sub.4 are started to
operate as described above, the resonance current i.sub.t as shown
in FIG. 3 or 4 flows in the transformer 3, so that power in
accordance with the set tube voltage and tube current is supplied
to the X-ray tube acting as the load 5. At this time, the inverter
2 operates at a resonance frequency determined by the leakage
inductance L.sub.S and the stray capacitance C.sub.S of the
transformer 3 or a frequency near the resonance frequency.
FIG. 6 is a circuit diagram showing an embodiment of the resonant
DC-DC converter according to a second aspect of the present
invention. In FIG. 6, for the sake of the simplification of
explanation, parts having the same arrangement and function as
those of the parts used in the embodiment of FIG. 1 according to
the first aspect of the invention are correspondingly referenced,
and the explanation of those parts is omitted. In this embodiment
according to the second aspect of the invention, there are provided
a voltage divider 14 for detecting a tube voltage appleed to a load
5 such as an X-ray tube or the like, an error amplification phase
determination circuit 15, a phase control circuit 10, and a signal
conversion circuit 16 for converting a signal representing the tube
voltage detected by the voltage divider 14 into a tube voltage
detection signal S.sub.5 in the form suitable for use in the error
amplification phase determination circuit 15. The error
amplification phase determination circuit 15 is arranged to receive
the detection signal from the signal converter circuit 16 and a
preset target voltage signal (a tube voltage set signal S.sub.1)
and amplify a difference between the detection signal and the
target voltage signal to thereby determine operation phases of
transistors Tr.sub.1 through Tr.sub.4 constituting an inverter 2 on
the basis of the amplified difference. The phase control circuit 10
is arranged to generate control signals for controlling the
operation phases of the respective transistors Tr.sub.1 through
Tr.sub.4 on the basis of an output signal from the error
amplification phase determination circuit 15 and send out the
control signal in response to an X-ray exposure signal S.sub.4
applied from a controller (not shown). Power to be supplied to the
X-ray tube acting as the load 5 is feed-back controlled through the
voltage divider 14, the signal conversion circuit 16, the error
amplification phase determination circuit 15, and the phase control
circuit 10 in such a manner that the transistors Tr.sub.1 and
Tr.sub.2 acting as a first and a second switching element of the
inverter 2 respectively are alternately turned on with a phase
difference of 180 degrees with respect to an operation frequency of
the inverter 2, the transistors Tr.sub.3 and Tr.sub.4 acting as a
third and a fourth switching elements of the inverter respectively
are alternately turned on with a phase difference of 180 degrees
similarly to the transistors Tr.sub.1 and Tr.sub.2, while properly
varying a phase difference from a point in time of turn-on of the
first transistor Tr.sub.1 to a point in time of turn-on of the
fourth transistor Tr.sub.4 and a phase difference from a point in
time of turn-on of the second transistor Tr.sub.2 to a point in
time of turn-on of the third transistor Tr.sub.3.
Next, description will be made as to the operation of the thus
arranged resonant DC-DC converter according to the second aspect of
the invention. First, when a tube voltage to be supplied to the
X-ray tube acting as the load 5 is determined, the error
amplification phase determination circuit 15 is supplied with the
tube voltage set signal S.sub.1 corresponding to the determied tube
voltage from the controller (not shown). The error amplification
phase determination circuit 15 is further supplied with the
above-mentioned tube voltage detection signal S.sub.5 from the
signal conversion circuit 16. In the error amplification phase
determination circuit 15, an error between the tube voltage set
signal S.sub.1 and the tube voltage detection signal S.sub.5 is
detected, and the thus detected error is processed through
proportional-plus-integral control or the like to thereby determine
a difference .alpha. in operation phase between the transistors
Tr.sub.1 through Tr.sub.4 constituting the inverter 2 in accordance
with the above-mentioned error. Then, a phase signal S.sub.3
corresponding to the thus determined phase difference .alpha. is
produced from the error amplification phase determination circuit
15, and applied to the phaee control circuit 10. At this time, the
error between the tube voltage set signal S.sub.1 and the tube
voltage detection signal S.sub.5 is zero prior to the initiation of
the X-ray exposure, and therefore the phase difference .alpha. is
selected to be zero so as to make it possible to supply maximum
power.
On the basis of the phase signals S.sub.3, the phase control
circuit 10 generates control signals for turning on/off the
respective transistors Tr.sub.1 through Tr.sub.4, and a control
signal for controlling the phase difference .alpha. from turn-on of
the transistor Tr.sub.1 acting as the first switching element to
turn-on of the transistor Tr.sub.4 acting as the fourth switching
element as well as the phase difference from turn-on of the
transistor Tr.sub.2 acting as the second switching element to turn
on of the transistor Tr.sub.3 acting as the third switching
element.
Upon reception of the X-ray exposure signal S.sub.4 applied from
the controller (not shown), the phase control circuit 10 supplies
the generated control signals to the driving circuits 11a through
l11d, so that the driving circuits 11a through 11d drive the
respective transistors Tr.sub.1 through Tr.sub.4 constituting the
inverter 2 on the basis of the control signals supplied from the
phase control circuit 10.
When the transistors Tr.sub.1 through Tr.sub.4 are started to
operate as described above, the resonance current i.sub.t as shown
in FIG. 3 or 4 flows in a transformer 3, and the tube voltage is
begun to be applied to the X-ray tube acting as the load 5, so that
the tube current flows. Thereafter, when the tube voltage applied
to the X-ray tube approaches the set value, the error between the
tube voltage set signal S.sub.1 and the tube voltage detection
signal S.sub.5 becomes small, so that the error amplification phase
determination circuit 15 operates to increase the phase difference
.alpha. to thereby reduce the power supplied from a DC power source
1. When the tube voltage of the X-ray tube becomes substantially
equal to the set value, the inverter 2 operates at such a phase
that power equal to that corresponding to the set tube voltage and
tube current can be supplied from DC power source 1 to the load 5.
At this time, the inverter 2 is operated at a resonance frequency
determined by the leakage inductance L.sub.S and the stray
capacitance C.sub. S of the transformer 3 or at a frequency near
the resonance frequency.
FIG. 7 is a circuit diagram showing another embodiment according to
the second aspect of the invention. In this embodiment, a DC power
source 1 is obtained by AC power received from a commercial power
source that is rectified and smoothed. A rectifier 17 is
constituted by four diodes D.sub.9 through D.sub.12 so as to
convert the AC power received from the commercial power source into
DC power through full-wave rectification. Capacitance C' and
inductance L' are provided to smooth the DC output of the rectifier
17. In this embodiment, output power can be increased in comparison
with the foregoing embodiment of FIG. 6. For example, power of
about several tens Kw - several hundreds Kw can be supplied to the
load.
Although the transistors Tr.sub.1 through Tr.sub.4 are used as the
switching elements of the inverter 2 in the foregoing embodiments
of FIGS. 1, 6, and 7, the present invention is not limited to this,
but, for example, GTOs may be used. Alternatively, for purpose of
operation of the inverter 2 at a higher frequency, MOS FETs, IGBTs,
SI transistors, SI thyristors, or the like may be used. Further,
the load 5 is not limited only to the X-ray tube, but the invention
may be applied to any load similarly to the X-ray tube so long as
the load requires a DC output power of a relatively high voltage.
Although processing is generally performed through the
proportional-plus-integral control in the error amplification phase
determination circuit 15 of FIGS. 6 and 7, the present invention is
not limited to this, but the processing may be carried out through
control using software by using various data to be used in the
processing after converted into digital values.
As described above, according to the present invention, the output
power of the inverter can be changed from zero to the maximum value
through the phase control of the inverter 2, so that the range of
output power control can be further extended in comparison with
that provided by the prior art control system. Further, unlike the
prior art system, the output power can be controlled without
changing the operation frequency F.sub.i of the inverter 2, so that
increase in sectional area of the iron core of the transformer 3
due to reduction in the operation frequency F.sub.i becomes
unnecessary to thereby prevent the transformer 3 from being
increased in size. Consequently, the whole apparatus can be reduced
in size. In the case where the operation frequency F.sub.i of the
inverter 2 with a rated load is selected to be higher than an audio
frequency so as to reduce noises, it is possible to maintain the
low noise state without changing the operation frequency F.sub.i in
controlling the output voltage thereafter. Moreover, according to
the second aspect of the invention, power to be supplied to the
load 5 is controlled through feedback in such a manner that an
actual voltage applied to the load 5 is detected and respective
operation phases of the switching elements of the inverter 2 are
determined on the basis of the error between the detection tube
voltage signal and the target tube voltage signal, so that the
output power can be controlled with higher accuracy.
* * * * *