U.S. patent number 4,554,639 [Application Number 06/482,594] was granted by the patent office on 1985-11-19 for audio dosimeter.
This patent grant is currently assigned to E. I. Du Pont de Nemours and Company. Invention is credited to William B. Baker, Harry E. Betsill.
United States Patent |
4,554,639 |
Baker , et al. |
November 19, 1985 |
Audio dosimeter
Abstract
An audio dosimeter for use by an individual for measuring
exposure to sound which contains the following components: (a) an
electronic microphonic sound sensor means, (b) an amplifier and
weighting circuit, (c) an x-squared detector circuit, (d) a
temperature compensation circuit; (e) analog to digital time
converter, (f) a microcomputer which controls functions of the
temperature compensator circuit and analog to digital time
converter and has unique logarithmic mathematics for converting the
signal from the converter into sound measurement values such as
percent of dose, average and maximum decibel levels and
instantaneous sound level.
Inventors: |
Baker; William B. (Newark,
DE), Betsill; Harry E. (Timonium, MD) |
Assignee: |
E. I. Du Pont de Nemours and
Company (Wilmington, DE)
|
Family
ID: |
23916685 |
Appl.
No.: |
06/482,594 |
Filed: |
April 6, 1983 |
Current U.S.
Class: |
702/1;
73/647 |
Current CPC
Class: |
G01H
3/14 (20130101) |
Current International
Class: |
G01H
3/00 (20060101); G01H 3/14 (20060101); G01H
003/12 (); G08B 021/00 () |
Field of
Search: |
;364/556,557,571
;73/648,645,646,632,647 ;179/1.1 |
References Cited
[Referenced By]
U.S. Patent Documents
Other References
Model 4341 Low Cost True RMS-to -DC Converter, Jan. 1975, 4 pages,
Burr-Brown Publication. .
Industrial Hygiene New, May 1982, p. 57. .
Integrated Circuit True RMS-to-DC Converter, 5 pages, Analog Device
Publication. .
LH0091 True RMS-to-DC Converter, Aug. 1976, 7 pages, National
Semiconductor Publication..
|
Primary Examiner: Wise; Edward J.
Attorney, Agent or Firm: Fricke; Hilmar L.
Claims
We claim:
1. An audio dosimeter for measuring exposure to sound
comprising:
(a) an electronic microphonic sound sensor means which generates
electric input signal upon activation by sound;
(b) an amplifier and weighting circuit electrically connected to
the sound sensor means which amplifies the input signal from the
sound sensor means and provides an "A" weighted signal directly
related to the frequency of the sound which activated the sound
sensor;
(c) an x-squared detector circuit electrically connected to the
amplifier and weighting circuit which receives the input signal and
generates a voltage signal which is proportional to the log of the
square of the input signal;
(d) temperature compensation circuit electrically connected to the
x-squared detector circuit which develops reference signals through
ramping and intergrating means matched to the x-squared detector
circuit and thereby temperature compensating the voltage signal of
the x-squared detector circuit;
(e) analog to digital time converter circuit electrically attached
to the temperature compensation circuit which converts the
temperature compensated voltage signal to digital time signal;
(f) a microcomputer electrically connected to the temperature
compensation circuit and controls the integrating and ramping
functions of the circuit and electrically connected to the analog
to digital time converter and controls the functions of the
converter and receives the digital time signals from the converter
and having a calculating means for converting the time signals into
sound measurement values.
2. The audio dosimeter of claim 1 having a gain circuit
electrically connected to the amplifier and weighting circuit and
microcomputer which allows selection of additional amplifier gain
for audio range.
3. The audio dosimeter of claim 2 having a parameter switch
electrically connected to the microcomputer for selection of
doubling rates, threshold levels and criterion levels.
4. The audio dosimeter of claim 1 in which the x-squared detector
circuit comprises a bilateral logging circuit, a bilateral
antilogging and integrating circuit wherein the input signal is
converted to a bilateral current signal and develops a voltage
which is proportional to two times the log of the input signal and
is fed into the bilateral antilogging and integrating circuit
producing a voltage which is proportional to the log of the square
of the input signal.
5. The audio dosimeter of claim 1 in which the temperature
compensation circuit having constant current source and a capacitor
connected in parallel comprises:
(a) a logging voltage to current converter, and a first antilogging
converter feeding the constant current source thereby generating a
low reference signal;
(b) a second antilogging converter feeding the parallel connected
capacitor and constant current source thereby generating a first
ramping reference signal; wherein both the low reference signal and
the ramping reference signal being compensated for temperature and
where the circuit produces a momentary high reference signal across
the capacitor upon a command from the microcomputer to start the
ramping reference signal ramping down;
(c) a first comparator which receives the low reference signal and
the ramping reference signals and detects end-of-ramp;
(d) a second comparator which receives the signals of (c) above and
the input signal from the x-squared detector and determines when
the ramping signals intersect the input signal;
wherein a signal from the first or second comparator fed to the
logic circuit of the microcomputer pulls down the interrupt line of
the microcomputer and the microcomputer which controls ramping
functions ratios time intervals in the ramping functions and
detects sound pressure levels in decibels.
6. The audio dosimeter of claim 1 having a display electrically
connected to the computer to visually show sound measurement values
determined by the microcomputer.
7. The audio dosimeter of claim 6 in which the microcomputer has a
program for calculating values of percentage dose, average decibel
level of exposure, maximum decibel level of exposure, instantaneous
sound level and time of exposure.
8. The audio dosimeter of claim 7 containing a selector switch
which allows for the display of the individual values and allows
the dosimeter to be used as a sound level meter by display of
instantaneous sound level.
9. The audio dosimeter of claim 1 having a control, interface
connected to the microcomputer for sending data to a second
computer.
10. A true x-squared detector circuit which receives an input
signal and determines the x-squared value of the signal; wherein
the circuit comprises:
(a) an x-squared detector circuit which receives the input signal
and generates a voltage signal which is proportional to the log of
the square of the input signal;
(b) temperature compensation circuit electrically connected to the
x-squared detector circuit which develops a reference signal
through ramping and integrating means matched to the x-squared
detector thereby temperature compensating the voltage signal of the
x-squared detector circuit;
(c) analog to digital time converter circuit electrically attached
to the temperature compensation circuit which converts the
temperature compensated logging voltage signal to digital time
signal;
(d) a microcomputer electrically connected to the temperature
compensation circuit and controls the integrating and ramping
functions of the circuit and electrically connected to the analog
to digital time converter and controls the functions of the
converter and receives the digital time signal from the converter
and having a calculating means for converting the time signal into
x-squared value of the input signal.
Description
BACKGROUND OF THE INVENTION
This invention is related to a device which measures noise to which
an individual is exposed.
In a work environment, the accumulated amount of noise or dose, the
average noise level and the maximum level of noise to which an
individual has been exposed during a work day are important to
occupational safety and health of an individual. Currently,
industry and governmental agencies such as Occupational Safety and
Health Administration (OSHA), Mine Safety and Health Administration
(MSHA) and the Department of Defense (DOD) require accurate
measurements and noise data measurements such as the percent of
total daily dose allowable for an individual, the average and
maximum noise levels of exposure and constant sound level
monitoring.
Audio dosimeters are known in the art as shown in Maddox et al.
U.S. Pat. No. 3,868,857 issued Mar. 4, 1975, Ceci U.S. Pat. No.
4,020,286 issued Apr. 26, 1977 and Sima, Jr., et al., U.S. Pat. No.
4,100,810 issued July 18, 1978. Generally, these dosimeters are not
as accurate as is required and only provide dose values and are not
sufficiently versatile to provide the aforementioned noise data
measurements which are currently required.
The audio dosimeter of this invention is a compact and accurate
instrument that provides the above data which can be visually
displayed or can be fed directly into a computer and recorded to
provide accurate records for an individual.
SUMMARY OF THE INVENTION
An audio dosimeter for use by an individual for measuring exposure
to sound which contains the following components:
(a) an electrical microphonic sound sensor means which generates an
electric input signal upon activation by sound;
(b) an amplifier and weighting circuit electrically connected to
the sound sensor means which amplifies the input signal from the
sound sensor means and provides an "A" weighted signal directly
related to the frequency of the sound which activated the sound
sensor;
(c) an x-squared detector circuit electrically connected to the
amplifier and weighting circuit which receives the input signal and
generates a voltage signal which is proportional to the log of the
square of the input signal;
(d) temperature compensation circuit electrically connected to the
x-squared detector circuit which develops reference signals through
ramping and integrating means matched to the x-squared detector
circuit and thus temperature compensates the voltage signal of the
x-squared detector circuit;
(e) analog to digital time converter electrically attached to the
temperature compensation circuit which converts the temperature
compensated voltage signal to digital time signal;
(f) a microcomputer electrically connected to the temperature
compensation circuit and controls the integrating and ramping
functions of the circuit and electrically connected to the analog
to digital time converter and controls the functions of the
converter and receives the digital time signals from the converter
and having unique logarithmic mathematics to convert the digital
time signals into readings such as percent of dose, average and
maximum decibel levels and instantaneous sound level.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 shows a sketch of the audio dosimeter.
FIG. 2 shows a block diagram of the electrical components of the
audio dosimeter.
FIG. 3 shows a functional schematic diagram of the x-squared
detector, temperature compensation circuit and the analog to
digital time converter circuit used in the audo dosimeter.
FIG. 4 shows the timing diagram of a single cycle of the analog to
digital time signal converter used in the audio dosimeter.
FIG. 5 shows a flow diagram of the timer and interrupt routines
used in the audio dosimeter.
FIG. 6 shows a simplified flow diagram used to compute dose from
the ramp timing data.
DETAILED DESCRIPTION OF THE INVENTION
The audio dosimeter is designed for use by an individual in a work
environment and can be carried by a belt or pocket clip. The size
of the dosimeter is about 12 cm. wide.times.17.8 cm. long.times.2.8
cm. deep and weights about 250 g. A microphone is clipped to a
person's shirt or coat pocket or collar or on a tab near the ear.
The dosimeter need not be worn but may be placed in a work area
where noise is monitored.
Typically, the dosimeter is used to monitor the level of noise a
worker is exposed to during an 8-hour work period. An audio
dosimeter is issued to a worker with all readings at zero. At the
end of the work period, readings can be manually read from the
display of the dosimeter or the dosimeter can be plugged into a
computer and the information directly fed into the computer.
Typical items that are monitored and recorded by the audio
dosimeter are as follows:
Dose in percent of total allowable dose,
Average (numerical) dB (decibel) of exposure known as LAVG and
Maximum dB level of exposure known as LMAX, and
Total time of exposure.
The dosimeter also monitors instantaneous sound pressure levels
known as SPL with the display being updated once per second.
The specification requirements and the essential functions of an
audio dosimeter are specified by the American National Standard
Institute Society document ANSI S1.25--1978 which document is
hereby incorporated by reference.
The dosimeter meets the specification requirements of ANSI
S1.25--1978 and has an operating range of 80-148 dB, criterion
levels of 80, 84, 85 and 90 dB, threshold levels of 0, 80, 85 and
90 dB, exchange rate 3, 4, 5, and 6 dB and has a liquid crystal
display which shows dose 0 to 1999%, average dB level of 77-141 dB,
maximum dB level of 77-141 dB, time up to 19 hours 59 minutes, and
instantaneous sound, i.e., sound level meter within range of 50-141
dB.
FIG. 1 shows a sketch of the audio dosimeter. A microphone 1,
typically a 1.2 SCM ceramic type microphone Knowles BL 1830, is
attached to a long cable 2 which is attached to the circuitry in
the case 4 of the dosimeter and held in place in the case 4 by a
large rubber grommet (not shown). A protective cover 3 shown in the
open position slides in slots in the case and protects switch 7
from damage and can be locked into place with a set screw. The
cover 3 has a viewing window 5 and when the cover 3 is closed a
display not shown in the sketch is visible. When the switch 7 is
engaged, light emitting diode 6 flashes to indicate the batteries
are operative. A push button switch 8 selects the desired display,
i.e., % dose, average decibel level, maximum decibel level and the
like. A set screw 9 locks the cover when the dosimeter is in use in
a closed position to protect the switches from damage. A battery
cover 10 can readily be removed to insert new batteries. Typically,
9-volt alkaline is used.
FIG. 2 shows a block diagram of the electrical components used in
the audio dosimeter. The microphone 1 picks up a sound and converts
the sound into an electrical signal which is fed into the amplifier
and weighting circuit 11. The circuit is constructed from three
operational amplifiers; the first and second configured as
noninverting, selectable gain (0 and 20 dB) stages and the third as
an inverting amplifier with adjustable gain to match varying
microphone sensitivities. Typically, three stages of a TL 064A quad
operational amplifier are used.
The amplifier and weighting circuit 11 matches the impedance of the
microphone 1 and amplifies the signal from the microphone 1 and
using common filter circuits provides an "A" weighted frequency
response in relationship to the input frequency of the signal as
defined in Table 1 of the aforementioned ANSI publication.
The resulting input signal A then is fed into x-squared detector
circuit 12. In the x-squared detector circuit the input signal is
converted to a bilateral current signal which develops a voltage
signal which is proportional to two times the log of the input
signal. This voltage signal is fed to a bilateral antilogging
circuit and then feeds into a parallel combination of an
integrating capacitor and a constant current source which produces
a detected sound signal voltage which is proportional to the log of
the square of the input signal.
The resulting detected sound signal is fed into a temperature
compensation circuit. Since the logging and antilogging circuits
used in the x-squared detector are inherently temperature
sensitive, compensation is required for precision performance. The
temperature compensation circuit containing ramping and integrating
means which are controlled by the microcomputer 16 temperature
compensates the voltage signal from the x-squared detector. The
resulting signal B is then fed into the A to D analog to digital
time signal convert circuit 14 which converts the temperature
compensated voltage signal, B, to digital time signal.
The following describes the combined functions of the temperature
compensation circuit 13 and the A to D circuit 14. Reference
voltage signals are developed using a logging voltage to current
converter and a first antilogging converter feeding a constant
current source thereby generating a low reference signal. A second
antilogging converter feeding a capacitor and constant current
source connected in parallel thereby generating a ramping reference
signal. Both signals have the same temperature characteristics as
the detected sound signal thus compensating for any circuit
temperature affects. Also, the circuit can produce a momentary high
reference signal across the capacitor when directed by the computer
to start the ramping reference signal ramping down. The low
reference signal and ramping reference signals are fed to a first
comparator to detect end-of-ramp. The detected sound signal and
ramping reference signal are fed to a second comparator to
determine when the signals are equal; i.e., when the ramp signal
intersects the detected signal. Either comparator signal combined
with a logic circuit will pull down the computer interrupt line. An
output from the computer to the logic will suppress the interrupt
signal only if a ramp crossing caused the interrupt. Thus the
computer which controls start-of-ramp and senses end-of-ramp and
ramp crossing can by ratioing the two time intervals detect sound
pressure level in dB.
The above components 12, 13 and 14, shown within dashed line in
FIG. 2 and indicated as 15, are a semicustom integrated circuit
developed specifically for the audio dosimeter circuit.
The resulting digital signal from A to D 14 (shown in FIG. 2 as
signal D) is fed into the microcomputer 16. The microcomputer 16
through signals C and E controls the functions of the temperature
compensation circuit 13 and A to D circuit 14.
The digital signal D is processed by the microcomputer 16.
Typically, the microcomputer 16 establishes a 1/16 second sample
time base. Within each sample time, digital signal D is used by the
microcomputer 16 to perform a nonlinear numerical integration. The
integration for each sample time is summed and calculations for all
functions measured are made and stored i.e., calculations for %
dose, total time of exposure, average and maximum decibel level and
instantaneous sound level in decibels. Switch 18 controls the
output of these values to the liquid crystal display 17. Engagement
of switch 18 causes the microcomputer 16 select the function to be
displayed and sends the value to the display 17 the function being
shown and the value of that function. A standby position is also
included which suppresses accumulation of new data but maintains
the data in microcomputer 16.
Gain switches 19 are two miniature sets of switches which allow
changing the gain range of the amplifier and weighting circuit 11.
These switches allow the selection of two 20 dB gain steps and
provides inputs to the microcomputer 16 indicating the gain of the
audio range and increases the useful audio range of the dosimeter.
By selecting the correct gain switch, the dosimeter can read noise
levels of as low as 50 dB when used as a sound level meter.
Parameter switch 20 is electrically connected to the microcomputer
16 and tells the microcomputer the desired doubling rate such as 3,
4, 5, and 6 dB), the aforementioned four criterion levels, and
threshold levels to be used in dose calculations.
Optionally, an external data and control interface 21 may be
attached to the computer 16 for logging of data and test control of
the unit. All parameters fed to the display 17 as well as status
information of the state of the battery and switches 19 and 20 may
be output of the interface.
FIG. 3 shows a functional schematic diagram of the x-squared
detector, temperature compensation, and the A to D circuits. The
circuitry, .circle.R , inside the dashed line represents a
semi-custom integrated circuit chip developed specifically for the
audio dosimeter (identified as 15 on FIG. 2).
The custom integrated circuit chip is constructed from a Monochip H
which is a standard chip made by the Interdesign Corporation
Sunnyvale, Calif. The custom chip is prepared according to the
functional schematic of FIG. 3. The various electrical components
shown in FIG. 3 are formed by connected components on the Monochip
which is a technique well known to those skilled in the art of
making integrated circuits from standard chips.
The audio output voltage A from the amplifier and weighting circuit
is driven through C1 typically a 2.2 .mu.F (microfarad) and R1
typically as 3.57 k.OMEGA. (kilohm) resistor, to the input of
operational amplifier A1, typically a CA 3130, which is configured
as a voltage to current converter. Capacitor C7, typically a 47 pF
(picofarad) capacitor, is connected to the amplifier A1 to provide
phase compensation. The amplifier A1 is connected to +V.sub.B which
is the positive side of 9 volt battery used as a power source. The
voltage at S is proportional to the log of the square of the signal
at A since the current drawn through R1 also flows through two
diodes, either CR1 and CR2 or CR3 and CR4 to develop the voltage at
5. On the positive half cycle at S transistor Q1, charges C4,
typically a 68 .mu.F (microfarad) capacitor and on the negative
half cycle the signal is inverted to charge C4 through Q2. A
constant current sink of 1.6 .mu.A, (microamps) as set by R9,
typically a 2.74M.OMEGA. resistor, in conjunction with C4
establishes the one second integration as defined by
where K=Boltzman's constant, 1.3807.times.10.sup.-23 Joules/Kelvin;
T=temperature in Kelvin; C=capacitance, .mu.F;
q=1.6022.times.10.sup.-19 coulombs, and I=current in mA. Under
steady state input conditions the average current supplied by Q1
and Q2 must be equal to the value established by the constant
current sink. For a time varying input signal the current to C4 is
given by ##EQU1## where k, T and q are identified above V.sub.A is
the voltage at node .circle.A , R.sub.1 is 3.57 k.OMEGA., V.sub.k
is voltage at node .circle.K , I.sub.k is the constant current sink
and V.sub.BE is the diode voltage associated with I.sub.k This
current is integrated to a voltage across C4 which is
mathematically equivalent to logging the output of a perfect
squaring detector driving a 1 second time constant RC network. The
voltage at C4 varies in proportion to the log of the input
power.
The resistor network comprising R2, typically 10 k.OMEGA. resistor,
R3 typically a 24 k.OMEGA. resistor, R4 typically 5 k.OMEGA.
resistor and R5 typically 24 k.OMEGA. resistors, develops a voltage
V.sub.Ref at pin 11 which can be adjusted to be 2.5 V. V.sub.Ref is
used throughout the x-squared detector and analog to digital
converter (A to D) as a common bias voltage. Capacitor C2,
typically a 1.0 .mu.F capacitor, bypasses V.sub.Ref to provide a
low impedance reference in the audio frequency range.
The analog to digital converter (A to D) circuitry provides the
means of relating the input signal level to the operating range of
the device as well as temperature compensating the x-squared
detector and one second time constant integrator. Amplifier A3,
transistor Q3 and diode CR5, generate an upper and lower reference
voltage in a circuit configuration which is a unilateral equivalent
of the x-squared detector. To select the upper limit the
microcomputer asserts A to D START at .circle.C pin 18 of the chip.
This forces an open collector inverter, INV1 to ground pin 13 and
establishes the upper limit reference current through resistor R7,
typically a 3.9 k.OMEGA. resistor and R8, typically a 5 k.OMEGA.
variable resistor. Resistor R8 is adjustable to set the upper limit
to a specific value. For a typical case the input voltage range is
64 dB with an upper limit such that 141 dBA which produces a 4 volt
peak to peak signal at .circle.A . The RMS input current at
.circle.A for a sine wave of this magnitude is then
or 396 .mu.A where V.sub.p =2 volt peak and R.sub.1 =3.57 k.OMEGA..
The total current through Q3 and CR5 must be this value to
establish a 141 dBA upper limit.
The low limit is selected at .circle.C when A to D START is forced
low by the microcomputer. Under this condition the current through
Q3 and CR5 is established by R6 typically in 1 M.OMEGA. (megaohm)
resistor, and the voltage at the arm of R4. For a 64 dB dynamic
range, this bias current must be 250 nA. Resistor R4 is adjustable
to allow calibration of the unit. The reference amplifier A3 is
compensated by capacitor C3, typically a 0.001 .mu.F capacitor.
The actual reference to the analog to digital (A to D) converter
appears at the emitter of Q5 which also drives a second 1.6 .mu.A
current source. During the time interval when A to D START is high,
Q4 charges the ramp capacitor C5 to the upper limit voltage with
the third constant current generator or 1.6 .mu.A as a load. The
voltage across C5 is used as the positive reference to two
comparators CMP1 and CMP2. The negative inputs of CMP1 and CMP2 are
fed from node .circle.K and the emitter of Q5, respectively.
R11, typically 68.1 k.OMEGA., sets the bias current as required by
the operational amplifier and comparators. Comparators CMP3
compares a portion of the battery voltage to V.sub.Ref and provides
an output to the computer to indicate if the battery has sufficient
charge for an additional 8 hours of use. The specific voltage
level, determined by R12, typically 182 k.OMEGA., and R13,
typically 130 k.OMEGA. is 7.2 volts.
FIG. 4 gives a timing diagram of a single cycle used in the analog
to digital converter. In the following discussion, reference is
made to points and components of FIG. 3. At the start of the cycle
A to D START node .circle.C has been held high by the microcomputer
which has forced C5 (FIG. 3) to charge to the upper limit--See FIG.
4(d). To initiate a conversion cycle, the microcomputer lowers both
A to D START Node .circle.C and A to D OVERRIDE Node .circle.E .
This action enables the gated 9.7 .mu.A current sink as set by R10
typically 475 k.OMEGA. resistor (FIG. 3) in parallel with the 1.6
.mu.A current sink at the emitter of Q4 (FIG. 3) and simultaneously
selects the low reference current by disabling the open collector
inverter INV1 associated with pin 13--See FIG. 4(f). The voltage
across C5 (FIG. 3) linearly ramps to the low limit voltage
established at the emitter of Q5--See FIG. 4(d). When the ramp
voltage falls below the voltage of C4 as sensed by comparator CMP1
the interrupt, Node .circle.D , pin 2, to the microcomputer is
forced low as shown by FIG. 4(c). The microcomputer measures and
stores the time interval from the start of the cycle to the falling
edge of the interrupt signal at Node .circle.D , and then clears
the interrupt by raising A to D OVERRIDE Node .circle.F --See FIGS.
4(b) and (c). The interrupt pin will remain high until the CMP2
(FIG. 3) changes state when Node .circle.L (FIG. 3) goes below the
low limit voltage established at the emitter of Q5 (FIG. 3). On
this interrupt the microcomputer measures and stores the total ramp
time. The input sound pressure level is then computed as ##EQU2##
where T.sub.R is the total ramp time and T.sub.D is the time from
start of ramp to the first interrupt, R is the range of the ramp in
DB and P.sub.low is the power level associated with the low limit
of the ramp. The actual ramp time as determined by the current
sources and the value of C5 (FIG. 3) is not critical since the
microcomputer calculates the input level based on the ratio of two
time intervals. The nominal value of ramp time was selected to give
a minimum of 256 counts such that a 64 dB range can be measured to
a resolution of 0.25 dB.
The NOR gates 1 and 2 (FIG. 3) associated with INTR Node .circle.D
and A to D OVERRIDE Node .circle.E allow the microcomputer to
operate on signal levels at the input Node .circle.A beyond the
limits of the ramp. The presence of an immediate interrupt at the
start of the ramp indicates that the signal level is above the
upper limit of the ramp. If the signal level is below the range of
the ramp raising, the A to D override will not clear the interrupt.
Since no other interrupt occured the signal level must be below the
lower limit of the ramp.
FIG. 5 shows a flow diagram of the timer and interrupt routines. In
the timer routine a 1/16 second time base is established, the
analog to digital conversion is started and the interrupts enabled.
When an interrupt occurs, the microcomputer gets the timer count
value and then tests the state of A to D OVERRIDE Node .circle.E .
If this output is low, then this is the first interrupt and the
timer count value corresponds to a data crossing. The A to D
OVERRIDE Node .circle.E is then raised and the count saved. The
state of the interrupt input Node .circle.D is then tested. If the
interrupt input is high, then a return is executed to wait for the
end-of-ramp; but if the interrupt is low, then the input level is
equal to or below the end-of-ramp limit. If on entry of the
interrupt routine the A to D OVERRIDE Node .circle.E is high, then
a data sample has already been taken, and the total timer count
value is then saved as the total ramp time value. To prepare for
the next cycle, interrupts are disabled and A to D START Node
.circle.C is set high to charge C5 at Node .circle.L to the upper
limit voltage prior to execution of the return.
The primary function of the dosimeter is to compute dose based on
the equation, ##EQU3## where D is dose, T.sub.c is the criterion
time (8 hours) L is the input level in dBA, L.sub.c is the
criterion level (typically 90 dBA), L.sub.co is the cutoff level
(typically 80 dBA), and q is a factor determined from the doubling
rate. The quantity q is defined as
where D.sub.r is the doubling rate in dB.
In the processor the calculations are performed using base 2
logarithms and the integration is performed numerically. Every 1/16
second a new dose value is computed based on the equation
The parameters above are:
D--dose in binary % of dose.
L--input level in binary dB (64 dB range to 0.25 dB).
L.sub.C --the criterion level in binary dB.
D.sub.r --doubling rate in binary dB.
L.sub.co --cutoff level in binary dB.
K--binary constant to normalize dose for desired output range and
1/16 second sample time.
To facilitate the calculations, a binary log look up table is used
throughout to allow multiplication and division to be performed
using addition and subtraction. The log of a binary number is
represented by a mantissa and an exponent. The mantissa is derived
from the 9 most significant bits of the binary number to be logged.
The exponent is a binary number representing the number of bit
shifts required to normalize the number. The antilog process
results in a multibyte number with a significance of 9 bits shifted
right or left by the value of the exponent. A left bit shift is
equivalent to multiplying by powers of 2. The 9-bit significance is
sufficient to express a number to an accuracy of 0.4%.
FIG. 6 is a simplified flow diagram used to compute dose from the
ramp timing data derived in FIG. 5. When end-of-ramp occurs a flag
bit is set to indicate that new data is ready for processing. In
the main routine the changing of state of this bit indicates the
start of a new dose increment calculation. TDATA is the time from
start-of-ramp to data crossing. The first step in processing
recalculates TDATA by substracting TDATA from TRAMP to reference
the calculation to the lower limit of the system (See Eq. 4). The
value of L in binary dB (see Eq. 7) is computed by first taking the
log of TDATA. The logging subroutine generates an error flag if
TDATA is zero. If TDATA is zero, the level is assumed to be the
lower limit of the ramp; but if it is not, then the ratio of TDATA
to TRAMP (See Eq. 4) is taken by substracting the log of TRAMP from
the log of TDATA. This quantity is then antilogged and multiplied
by 256. This results in an 8-bit binary representation of L to 0.25
dB as reference to that lower limit of the ramp.
The absolute value of L is then computed by adding L to LMIN. LMIN
is a constant determined by the settings of the gain switches (FIG.
2, 19). The value of L is then compared with the cutoff level,
L.sub.co. If L is less than L.sub.co, then the dose calculation is
bypassed. Otherwise, an intermediate value, X, is calculated by
subtracting the criterion level, L.sub.c, and then taking the log
of X. This quantity is divided by the doubling rate, D.sub.r, by
subtracting log.sub.2 (D.sub.r) and taking the antilog. The dose
increment .DELTA.D for this time interval is computed by summing
the log.sub.2 (K) to the value just computed and taking the
antilog. The dose accumulator is then updated by adding in the new
value .DELTA.D. The constants L.sub.c, L.sub.co and D.sub.r are
stored in the program and are selected by parameter switch (See
FIG. 2, 20).
The constant K has a specific value which was selected to make the
24th bit of a 6 byte wide dose accumulator correspond to 0.1% of
dose. Internal to the microcomputer it is possible to accumulate a
dose of up to 1.68.times.10.sup.6 % of dose with the smallest
increment of dose being 0.6.times.10.sup.-9 % of dose. This
technique of computation results in extremely wide dynamic range
integration of dose whereby small values of dose increments can be
accumulated to large dose values without loss of significance.
Using similar techniques, the microcomputer calculates:
LAVG: The sound pressure level which would give the dose
accumulated in run time.
SPL: A calculation of the average sound pressure level for one
second.
LMAX: The highest SPL recorded by the dosimeter.
In addition, the microcomputer maintains a running time clock,
formats data to the LCD display, reads the parameter switch (FIG.
2, 20) and monitors the optional external computer data
interface.
Another aspect of this invention is a circuit which is a true
x-squared detector circuit which can be used as a perfect RMS
detector which receives an input signal and determines the
x-squared value of the signal. This circuit has the following
components:
(a) an x-squared detector circuit which receives the input signal
and generates a voltage signal which is proportional to the log of
the square of the input signal;
(b) temperature compensation circuit electrically connected to the
x-squared detector circuit which develops a reference signal
through ramping and integrating means matched to the x-squared
detector thereby temperature compensating the voltage signal of the
x-squared detector circuit;
(c) analog to digital time converter circuit electrically attached
to the temperature compensation circuit which converts the
temperature compensated logging voltage signal to digital time
signal;
(d) a microcomputer electrically connected to the temperature
compensation circuit and controls the integrating and ramping
functions of the circuit and electrically connected to the analog
to digital time converter and controls the functions of the
converter and receives the digital time signal from the converter
and having a calculating means for converting the time signal into
x-squared value of the input signal.
* * * * *