U.S. patent number 4,013,937 [Application Number 05/593,813] was granted by the patent office on 1977-03-22 for naturally commutated cycloconverter with controlled input displacement power factor.
This patent grant is currently assigned to Westinghouse Electric Corporation. Invention is credited to Laszlo Gyugyi, Brian R. Pelly.
United States Patent |
4,013,937 |
Pelly , et al. |
March 22, 1977 |
Naturally commutated cycloconverter with controlled input
displacement power factor
Abstract
A naturally commutated cycloconverter having at the input a
source of higher frequency than at its output is used as a static
reactive power generator to correct displacement angle in an
alternating current power system when coupled thereto at the
output. Reactive power correction is obtained with such "high
frequency link" by automatically controlling the output voltage of
the cycloconverter so as to deviate by a required amount from the
AC power system voltage. At the same time, a circulating current is
established between the positive and negative banks of thyristors
of the cycloconverter of such a magnitude as to compensate for the
variations of the lagging quadrature component of the input current
of the cycloconverter caused by output current variations, thereby
permitting optimization of the high frequency link. The concept of
generating controllable reactive power with a naturally commutated
cycloconverter is used (1) with one cycloconverter as a
controllable source of reactive power coupled to a power system;
(2) with two cycloconverters tying a power system with a load, or
two power systems together. In either instance, the output voltage
of the cycloconverter is controlled for automatic compensation in
the line power factor, and concurrently a circulating current is
established in the single or in the two cycloconverters for
automatic correction against variations in the lagging current at
the cycloconverter input when the load conditions at the output
thereof vary.
Inventors: |
Pelly; Brian R. (Lingfield,
EN), Gyugyi; Laszlo (Penn Hills Township, PA) |
Assignee: |
Westinghouse Electric
Corporation (Pittsburgh, PA)
|
Family
ID: |
27250499 |
Appl.
No.: |
05/593,813 |
Filed: |
October 9, 1975 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
Issue Date |
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490640 |
Jul 22, 1974 |
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Foreign Application Priority Data
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Jul 22, 1975 [FR] |
|
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75.22821 |
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Current U.S.
Class: |
363/10; 323/207;
363/161 |
Current CPC
Class: |
H02J
3/1835 (20130101); H02M 5/271 (20130101); Y02E
40/30 (20130101) |
Current International
Class: |
H02J
3/18 (20060101); H02M 5/27 (20060101); H02M
5/02 (20060101); H02M 005/27 (); H02J 003/18 () |
Field of
Search: |
;321/7,66,69R
;323/101,102,105,108,119,121,123,127,128 ;318/171,179 |
References Cited
[Referenced By]
U.S. Patent Documents
Primary Examiner: Beha, Jr.; William H.
Attorney, Agent or Firm: Lorin; C. M.
Parent Case Text
CROSS-REFERENCE TO OTHER PATENT APPLICATIONS
This is a continuation-in-part of application Ser. No. 490,640,
filed July 22, 1974 by the same applicants, and which is now
abandoned.
Claims
We claim:
1. In a static frequency changer apparatus including positive and
negative banks of controllable rectifiers operative between input
and output terminals with at least one alternating current power
system of lower frequency at said output terminals and with an
alternating voltage source of higher frequency at said input
terminals for natural commutation thereof; with said one power
system having variable power load conditions; with said voltage
source having a selected VA rating and frequency and generating a
quadrature input current at said input terminals lagging the input
voltage thereof by an amount of lag in relation to said variable
load conditions; the combination of:
signal means for sensing a condition representative of said
quadrature input current to derive a control signal;
control means responsive to said control signal for controlling
said controllable rectifiers to generate between said positive and
negative banks a circulating current of such amplitude as to
maintain said quadrature input current lag to a predetermined
amount.
2. The apparatus of claim 1, including first and second frequency
changers each having positive and negative banks of controllable
rectifiers operative between corresponding said input output
terminals; with said one power system being connected at the output
terminals of said first frequency changer, with the provision of
another alternating current power system connected at the output
terminals of said second frequency changer, said another power
system having variable power load conditions;
with said voltage source generating a first and a second said
quadrature input current at the respective input terminals of said
first and second frequency changers, each having a lag in relation
to the power load conditions of the associated power system;
with said signal means sensing the respective said quadrature input
currents to derive corresponding control signals;
with said control means being responsive to said control signals
and operative with said first and second frequency changers to
generate between positive and negative banks of rectifiers in each
frequency changer a corresponding said circulating current of such
amplitude as to maintain the lag of the corresonding quadrature
input current to a corresponding predetermined amount.
3. The apparatus of claim 1, including first and second frequency
changers each having positive and negative banks of controllable
rectifiers operative between corresponding said input and output
terminals; with said one power system being connected at the output
terminals of said first frequency changer, with the provision of a
load connected at the output terminals of said second frequency
changer, said load having variable power load conditions;
with said voltage source generating a first and a second said
quadrature input current at the respective input terminals of said
first and second frequency changers, each having a lag in relation
to the power load conditions of the associated one of said power
system and said load;
with said signal means sensing the respective said quadrature input
currents to derive corresponding control signals;
with said control means being responsive to said control signals
and operative with said first and second frequency changers to
generate between positive and negative banks of rectifiers in each
frequency changer a corresponding said ciirculating current of such
amplitude as to maintain the lag of the corresponding quadrature
input current to a corresponding predetermined amount.
4. The apparatus of claim 1 with said representative condition
being derived by sensing the quadrature input current supplied by
said source.
5. The apparatus of claim 1 with said representative condition
being derived by sensing the input frequency supplied to the input
terminals of said apparatus.
6. The apparatus of claim 1 with said source consisting of tuned
inductance-capacitance tank circuit means at said selected VA
rating and frequency.
7. The apparatus of claim 1 further including means responsive to a
representation of said variable power load conditions for timing
the operation of said controllable rectifiers for modifying the
output voltage of said apparatus in relation to the output voltage
of said power system to compensate for power factor variations in
said power system.
8. The apparatus of claim 2 further including means responsive to a
first and a second representation of said reactive power load
conditions in said one and another power systems for timing the
operation of the controllable rectifiers of said first and second
frequency changers, respectively for modifying the output voltage
at the respective output terminals in relation to the corresponding
output voltage of the associated one of said power systems to
compensate for power factor variations in such associated power
system.
9. The apparatus of claim 8 further including means associated with
said first and second frequency changers for maintaining the phase
angle of the output current relative to the output voltage at the
output terminals of one frequency changer equal and opposite to the
phase angle of the output current relative to the output voltage at
the output terminals of the other frequency changer, thereby to
control flow of real power between said power systems through said
first and second frequency changers.
10. The apparatus of claim 3 further including means responsive to
a first and a second representation of said power load conditions
in said power system and said load for timing the operation of the
controllable rectifiers of said first and second frequency
changers, respectively for modifying the output voltage at the
respective output terminals in relation to the corresponding output
voltage of the associated one of said power system and load to
compensate for power factor variations in such associated one of
said power system and load.
11. The apparatus of claim 10 further including means associated
with said first and second frequency changers for maintaining the
phase angle of the real component of the output current relative to
the output voltage at the output terminals of one frequency changer
equal and opposite to the phase angle of the real component of the
output current relative to the output voltage at the output
terminals of the other frequency changer, thereby to control flow
of real power between said power system and said load through said
first and second frequency changers.
12. A reactive power generator for supplying variable reactive
power to an alternating current power system of a given voltage and
frequency comprising:
static frequency converter means having controllable rectifier
means, a converter input current, a converter input voltage, a
converter output current and a converter output voltage, the
frequency of said converter input current being higher than the
frequency of said converter output current, said frequency
converter means being coupled to said power system, said converter
output voltage being substantially equal to said power given
voltage, said converter output frequency being equal to said power
system frequency,
said frequency converter means being naturally commutated by said
power system,
means for generating said input current and said input voltage at
said input frequency; and
means controlling the operation of said rectifier means of said
frequency converter means for concurrently (a) adjusting said
converter output voltage in relation to said power system voltage
to correct the power factor of said power system and (b)
establishing a circulating current within said frequency converter
means of such a magnitude to maintain the lagging quadrature
current in said converter input at a predetermined value to
compensate for said variable reactive power in said power sytem,
whereby natural commutation of said converter means with said
generating means occurs at a selected value of said converter input
frequency.
Description
BACKGROUND OF THE INVENTION
A naturally commutated cycloconverter is a static device for
changing electrical power from one frequency to another. An
inherent characteristic of such a cycloconverter is that the
displacement angle of the current drawn at the input is lagging and
is a function of the displacement angle of the output current. This
characteristic is often acceptable, but for some applications it
would be beneficial if the reactive current at the input could be
controlled independently of the amplitude and phase of the current
at the output.
This has been accomplished in the prior art, such as shown by U.S.
Pat. No. 3,742,336, by connecting an external variable reactive
power device across the input terminals and including back-to-back
thyristors which are phase controlled to provide whatever net
reactive current is required at the input, such as disclosed in the
published article appearing in POWER for April 1973, at pages
69-71, irrespective of the output loading conditions of the
cycloconverter. A fixed capacitor provides a fixed amount of
leading reactive power from which is cancelled a lesser or greater
extent of lagging reactive power drawn by the parallel connected
variable inductor, when the cycloconverter experiences variable
load conditions.
There has been in recent years an increased demand for power factor
correction and control in utility and industrial power systems due
to growing use of electrical machines, the major role of the
electric arc furnace in steel production and the general acceptance
of thyristor drives and power controllers in the industry. As a
result, controlled generation of reactive power for improving the
line power factor has become of major importance. This can be
achieved most successfully through the use of static power
switches, for instance thyristors. Traditionally, rotating
synchronous condensers have been used for this purpose. It has been
established, however, that static VAR generators in most
applications provide superior performance at lower cost than
conventional rotating synchronous condensers.
There are three basic modern methods of generating reactive power
(VAR) all using static control of thyristors: (1) thyristor
controlled shunt capacitors and inductors; (2) AC/DC converters and
inverters; (3) AC/AC frequency changers.
The present invention relates to the third category of static VAR
generators, namely to AC/AC frequency changers used for the
generation of reactive power.
The AC/AC frequency changer is itself divided into several
categories according to the mode of control, the range of control,
and the inherent properties of the apparatus in operation. A basic
distinction is made between frequency changers in which the
thyristors are force commutated and those in which the thyristors
are naturally commutated by the voltages of the input source.
The present invention relates to naturally commutated AC/AC
frequency changers or cycloconverters.
Two inherent characteristics of a frequency changer using
thyristors for conversion are: (1) the frequency relation between
the input alternating current and the output alternating current;
and (2) the phase relation between current and voltage at the input
and at the output, i.e., the relationship between the input and
output displacement power factors. In naturally commutated
cycloconverters the input displacement power factor is lagging and
is a function of the displacement power factor of the output
current. It is possible by force commutation to control the input
displacement power factor and in particular, to bring it
automatically to unity. This is not possible with a naturally
commutated frequency changer. Another particularity of frequency
changers is that an alternating sinusoidal wave of a desired
frequency is generated by controlled conduction of the thyristors.
The time and frequency of conduction of the thyristors is generally
variable along the reference waveform used to build the output
waveform.
Static frequency changers offer a unique mode of generating
reactive power. In that respect they provide an interesting
alternative for AC/DC converters, and for inverters, which also are
capable under proper operative conditions of generating reactive
power. In all such instances, practically the reactive power
generator must be operated in an essentially balanced multi-phase
system. However, these types of generators not only perform
generally as well as any rotating synchronous condensers under
steady state conditions, but also have proved to be superior for
transient response.
In the U.S. Pat. No. 3,858,105 of Laszlo Gyugyi reference has been
made to an original concept by B. R. Pelly of a high frequency link
consisting of a cycloconverter and a high frequency source of
reactive power naturally commutating the cycloconverter for
providing, in relation to a power system connected at the output of
the cycloconverter, power factor correction by control of the
cycloconverter.
In the above-mentioned Gyugyi patent it is also stated that the
cycloconverter has at the input an inherent lagging component of
current, similar to that drawn by an inductance. This lagging input
current varies with the output load, so that the frequency of the
reactive power source provided by a tuned L-C circuit, required for
natural commutation of the cycloconverter, varies. Therefore, a
higher VA rating is required from the L-C tuned circuit reactive
source in order to keep its frequency variations small under
varying load conditions. In the patent, the proposition is made to
use two cycloconverters properly controlled to create a system
equivalent to one cycloconverter in which the load variations do
not affect the lagging input power factor exhibited by the two
cycloconverters. The present invention proposes to achieve a
similar result by establishing and controlling a circulating
current between banks of one or two cycloconverters.
SUMMARY OF THE INVENTION
A direct voltage bias is applied to the firing angle control
circuit of one, or two, naturally commutated cycloconverters so as
to establish a circulating current between the positive and
negative banks thereof, of such magnitude that the lagging input
current is varied in relation to variations in the load at the
output.
The invention may take two aspects depending upon whether two
cycloconverters are used for interlinking two power systems (or one
power system and a load), or a single cycloconverter is connected
to one power system. In either instance, a high frequency source is
connected at the input of the cycloconverter for natural
commutation thereof. The high frequency source is common to both
cycloconverters when two are used.
The present invention enables the use of a high frequency generator
of optimum VA rating by keeping the reactive power demand of the
cycloconverter(s) constant.
When one cycloconverter is used as a controllable source of
reactive power, the invention resides in a static reactive power
generator for supplying variable reactive power to an alternating
current power system, comprising: (1) a frequency converter having
positive and negative banks of controllable rectifiers; (2) a
source of reactive power at the input operative at a frequency
which is higher than the frequency of the AC power system for
naturally commutating the rectifiers; (3) controlling means for the
output voltage of the frequency converter to adjust the amount of
reactive power provided for to the AC power system; and (4) means
for controlling the rectifiers to generate circulating current
between the positive and negative banks so as to maintain the
lagging quadrature current in the input of the frequency converter
at a predetermined value independently of the reactive power
provided in the AC power system.
BRIEF DESCRIPTION OF THE DRAWINGS
In FIG. 1 there is diagrammatically shown a prior art arrangement
for correcting the power factor of a cycloconverter connected
between an input power supply and a load;
In FIG. 2 there is schematically shown the power factor correction
arrangement of FIG. 1;
In FIG. 3 there is shown the well known circuit arrangement of the
positive bank of thyristors and the negative bank of thyristors of
a naturally commutated cycloconverter;
In FIGS. 4A to 4K there are shown waveforms to illustrate the
no-circulating current, the just-continuous circulating current and
the increased circulating current conditions of operation of the
here described apparatus;
In FIG. 5 there is illustrated the typical vector diagram for the
input current for a cycloconverter;
In FIG. 6 there is shown a feedback arrangement for controlling the
amplitude of the quadrature current at the input of a
cycloconverter in accordance with the present invention;
In FIGS. 7A and 7B there are shown control voltage waveforms
determined by the bias voltage error signal provided in FIG. 6;
In FIGS. 8A to 8C there are illustrated a practical application of
the control circuit of FIG. 6 in relation to a gas turbine driven
power supply;
In FIG. 9 there is shown a well known high frequency link power
system tie arrangement, including a feedback arrangement for
controlling the amplitude of reactive current supplied by the high
frequency link apparatus;
In FIGS. 10A to 10D there is shown a second practical application
of the present invention in relation to such a high frequency link
power system tie arrangement; and
In FIG. 11 there is shown in greater detail the operation of a high
frequency link including a tank circuit that can be used in each of
the illustrated applications of the invention.
FIG. 12 shows a typical control circuit for controlling power flow
in tying arrangement of two power systems.
FIG. 13 shows a prior art scheme for power factor correction using
fixed capacitor-variable inductor.
FIG. 14 shows the waveforms illustrating the variation of the
inductor current in the circuit of FIG. 12 with varying firing
angle.
FIG. 15 shows waveforms illustrating the operation of a naturally
commutated AC/DC converter in the case of lagging VAR
generation.
FIG. 16 illustrates the basic scheme for providing reactive power
with a static inverter.
FIG. 17 shows the use of a synchronous machine for power factor
correction.
FIG. 18 shows the typical uses of a cycloconverter naturally
commutated by a high frequency machine for power factor
correction.
FIG. 19 is the cycloconverter of FIG. 18 but with the use of an
oscillating tank circuit rather than a machine as the high
frequency source.
FIG. 20 schematically shows a naturally commutated cycloconverter
between a high frequency input power source and an AC power system
to provide reactive power correction for the AC power system.
FIG. 21 shows a typical power circuit embodiment of the
controllable VAR source according to the invention using a
naturally commutated cycloconverter and a high frequency tank
circuit.
FIG. 22 is a diagrammatic representation of the basic functional
scheme for controlling the output voltage of a cycloconverter in
order to adjust the displacement angle at the output.
FIG. 23 combines the circuitry of FIG. 22 with automatic control of
the frequency of the tank circuit of the high frequency source
through control of the circulating current between positive and
negative banks of the cycloconverter.
THE NOVEL CONCEPT OF CONTROLLING THE INPUT CURRENT LAG OF A
NATURALLY COMMUTATED CYCLOCONVERTER
In FIG. 1 there is shown an example of the prior art approach for
correcting the power factor of a cycloconverter 10 operatively
connected between an input power supply 12 in a load 14. The
cycloconverter 10 can be considered as a variable lagging load, in
effect comprising a varying resistance representing the power
component of the load and a varying inductance in parallel with the
resistance, with the varying resistance and inductance depending
upon the load conditions at the output of the cycloconverter.
Normally, in order to correct the equivalent inductance of the
cycloconverter 10, a fixed capacitive device 16 would be connected
of such capacitance as to balance out the equivalent inductance of
the cycloconverter 10. Thus, a unity power factor operating
condition could be achieved. In practice, because the load 14 at
the output of the cycloconverter 10 normally changes during the
course of operation, the equivalent variable inductance as seen
from the input of the cycloconverter 10 changes in value also. A
variable capacitor would then be required in order to cancel out
the varying inductance of the cycloconverter. One way to obtain an
effectively varying capacitor would be to provide a fixed capacitor
such as 16 in combination with a variable inductance 18, which
would cancel the capacitor 16 to a greater or lesser extent as
required.
In FIG. 2 the systems elements shown in FIG. 1 are shown in greater
detail such that the cycloconverters 10 are shown to include an
equivalent variable resistor 20 and an equivalent variable
inductance 22, with the resistor 20 representing the variable power
component of the load 14 and the variable inductance 22 depending
upon the load condition at the output of the cycloconverter. The
variable inductance 18 is shown to include a fixed inductor 24
operative with thyristors 26 for controlling the effective
inductance of the inductor 24 in relation to the fixed capacitance
16, such that the latter provides fixed leading VARS in relation to
the effective lagging power factor of the current drawn at the
input of the cycloconverter 10. In this way, a fixed input power
factor is provided for the cycloconverter as viewed by the input
power supply 12.
As shown in FIG. 3, a naturally commutated cycloconverter includes
a positive bank of thyristors 30 and a negative bank of thyristors
40, with the particular circuit shown in FIG. 3 providing a
one-phase output from a three-phase input power supply. The
positive bank of thyristors 30 carries the positive output current
and the negative bank 40 carries the negative output current. The
control scheme for the cycloconverter may be arranged such that the
negative bank 40 is blocked when the positive bank 30 carries the
output current, and vice versa. It is also practical to fire both
sets of thyristors 30 and 40 at the same time, in which event each
bank carries a circulating component of current in addition to its
share of the output current. Reactors 32, 34, 36 and 38 are
inserted as shown between the two banks 30 and 40, to support the
ripple voltage which inevitably appears between the positive and
negative banks 30 and 40 to thereby limit the circulating ripple
current to an acceptable level. This latter mode of operation is
sometimes employed to obviate the problems otherwise associated
with obtaining a smooth and substantially distortion-free crossover
of the output current. Since the production of appreciable
circulating current at full load is generally considered to be
objectionable because it constitutes an additional load on the
thyristors, as well as an additional lagging reactive load on the
input lines, the control is usually arranged such that the angle of
overlap between the firing pulses of the two banks is virtually
zero at full output current and this angle is automatically
increased until it eventually reaches a full 360.degree. at reduced
output current when the need to operate with circulating current
becomes greater. With this type of control, the loading of the
thyristors and supply lines at intermediate loads is kept to a
level which is less than that obtained at full load. In the prior
art, the operation of a naturally commutated cycloconverter has
usually been with a circulating current between the positive and
negative banks of the cycloconverter to obviate the difficulties of
obtaining a good quality output wave. However, the circulating
current by itself has been looked upon as being an undesirable side
effect, and thus this current has usually been kept just to the
minimum level that would permit the desired circuit operation.
A naturally commutated cycloconverter has the commutations of the
current between the thyristors effected naturally by virtue of the
alternating polarity of the applied input voltage. In other words,
when the one thyristor is fired to turn it on, at the same time
this turns off the preceding thyristor because the input voltage is
instantaneously of such polarity to switch off the thyristor that
was previously conducting. This relates to the commutation within a
given negative or positive bank. For example, in reference to FIG.
3, if the top left thyristor 42 is now conducting and it is desired
to shift the current conduction from the thyristor 42 to the top
middle thyristor 44, this requires the firing of the thyristor 44
so the current flows in the top middle thyristor 44 and at that
time the polarity of the voltage between the two terminals will be
such that the firing of the thyristor 44 will apply a reverse
voltage across the previously conducting thyristor 42 such that the
current flow through the thyristor 42 commutates naturally. The
operating principle of natural commutation is well known in the art
and is described in a book entitled "Thyristor Phase-Controlled
Converters and Cycloconverters" by B. R. Pelly, published in 1971
by John Wiley and Sons, and more specifically in Chapter 3 with
respect to a very simple circuit on pages 27-32. The basic function
of a naturally commutated cycloconverter is to convert from one
frequency to another frequency, with the output frequency being
less than the input frequency. A basic property of a cycloconverter
is that it always appears to the input power supply to which it is
connected as a lagging load, regardless of whether the particular
load connected to the output of the cycloconverter is resistive,
capacitive or inductive. The equivalent load including the
cycloconverter as seen by the input power supply turns out to be
lagging and inductive. Furthermore, the amount of lagging load
reflected by the cycloconverter 10 to the input power supply 12
depends upon the actual load condition at the output of the
cycloconverter 10, such as the amount of actual load and the
phasing of the actual load which is discussed in detail in the
above-referenced book at pages 160-180. The signal waveforms that
explain this effect, for example, are set forth on pages 169-172 of
the above book for various different actual loads at the output of
the cycloconverter 10. The signal waveforms show that for all types
of loads the input current is lagging the input voltage by a
varying amount depending upon the particular load at the output of
the cycloconverter.
The current which circulates between the positive and negative
thyristor banks manifests itself in the input side of the naturally
commutated cycloconverter as a lagging reactive component, and in
accordance with the teachings of the present invention, the input
displacement factor of the cycloconverter is controlled by
regulating the amplitude of the direct component of the current
which circulates between the two converter banks so as to produce
whatever reactive current is desired at the input of the
cycloconverter. In general, this will call for a greater level of
circulating current than is required for the purpose of producing a
good output waveform and thus the control scheme will generally be
compatible with the waveform control requirements. The amplitude of
the direct component of circulating current can be controlled by
introducing the small direct voltage difference between the
positive and negative converters through application of suitable
controlled direct bias voltages to the firing angle control
circuits. Since the direct component of circulating current is
limited only by the relatively small resistance of the circulating
current reactors plus any stray circuit resistance, only a
relatively small biasing of the converter firing angles is
necessary to provide the required level of circulating current.
This can produce only a variable lagging current at the input of
the cycloconverter, and if it is desired to provide either a net
leading or net lagging power factor, depending upon the particular
application requirements, a fixed static capacitor can be connected
across the input terminals of the cycloconverter and the amplitude
of the circulating current can then be controlled as required to
cancel the fixed leading reactive component consumed by the
capacitor to a lesser or greater extent, as desired.
The present invention is directed to modifying the power factor of
the naturally commutated cycloconverter as seen by the input power
supply so that the input power factor can be controlled at will,
and could, for example, be kept substantially constant. The power
factor can be made to be unity or even leading by the teachings of
the present invention. The basic principle here involved is
illustrated by the waveforms shown in FIG. 4. It is assumed that
the fixed capacitors are connected across the input lines of the
three-phase to one-phase cycloconverter as shown in FIG. 3, and
that the output voltage and current remain fixed in amplitude and
phase relationship as the net input displacement factor is
controlled from leading through unity to lagging. In curve 4A the
waveform is shown of the cycloconverter output voltage when the
cycloconverter is operating with no circulating current between the
respective positive bank and negative bank of the cycloconverter,
with the conduction angles of the positive and negative thyristor
banks not overlapping one another. The fixed capacitor current
overcompensates the lagging reactive component drawn by the
naturally commutated cycloconverter, and the net fundamental
component of current at the input is leading. The curve 4E
illustrates the output voltages of the positive and negative banks
of the naturally commutated cycloconverter with the angle of
conduction overlap of the positive and negative banks being
increased to a full 360.degree. and the current carried by each
bank being just continuous. The positive bank curreng i.sub.p as
illustrated in FIG. 3 is determined by the relationship ##EQU1##
The negative bank i.sub.n is determined by the relationship
##EQU2## The output current i.sub.o is determined by the
relationship
where i.sub.p is the instantaneous current of the positive bank,
i.sub.n is the instantaneous current of the negative bank, i.sub.o
is the instantaneous output current, I.sub.o is the peak output
current and .omega..sub.o is the angular output frequency. The
lagging component of current drawn by the cycloconverter itself is
now greater than the showing of curve 4A, and in fact is now just
sufficient to cancel the fixed leading current drawn by the
capacitors and the net input displacement factor is unity.
For an increased circulating current operating condition, as shown
by curves 4I through 4K, the steady component of circulating
current carried by each bank is double that illustrated in curves
4F through 4H and is determined for the positive bank by equation
##EQU3## and for the negative bank is determined by equation
##EQU4## The lagging component of current drawn at the input
terminals of the cycloconverter due to the circulatory current is
double that shown by the waveform 4H and the resulting net input
displacement factor is lagging.
In general, for continuous conduction in both banks of thyristors,
it can be shown that the rms inphase and quadrature components of
current at the cycloconverter input terminals, I.sub.power and
I.sub.q, respectively, are given by the relationship ##EQU5## where
g is the number of output phases, s is the number of three-phase
groups connected in series in each cycloconverter circuit such that
s = 2 for the bridge circuit shown in FIG. 3, r is the ratio of the
actual output voltage to maximum possible output voltage and
I.sub.o is the rms output current, and the relationship ##EQU6##
where values for the quantity a.sub.10 for different values of r
are tabulated on page 296 of the above reference book by Pelly, and
I.sub.c is the steady component of circulating current. Since
continuous conduction is assumed, ##EQU7## It is evident from the
above relationships that by controlling the amplitude of I.sub.c,
the current I.sub.q can be controlled.
The waveforms of FIG. 4 demonstrate the possibility for controlling
the net input displacement factor in both the leading and lagging
directions, independent of the load at the output of the
cycloconverter with no additional active power circuit elements. It
should be noted that the waveforms shown in FIG. 4, which are in
relation to a naturally commutated cycloconverter having a
one-phase output, show a relatively high degree of distortion of
the input current waves. In practice, for a three- (or multiple-)
phase output, most of the distortion seen in these simple waveforms
would cancel, so that the resulting net input current waveform
would be practically sinusoidal, but still would have the same
adjustable phase shift of the fundamental component illustrated in
FIG. 4.
In the example chosen in relation to the waveforms shown in FIG. 4,
the output load current is fixed and the current carried by the
thyristors increases as the input displacement factor becomes more
lagging. In general, the circulating current carried by a thyristor
would be objectionable if it produced a total heating effect
substantially greater than that produced just by the full load
output current with no circulating current. Thus, in practice, it
would be desirable to operate the cycloconverter so that little or
no current circulates between the positive and negative banks at
full load. Then, as the load is reduced, the circulating current
can be increased to the extent necessary to provide whatever net
reactive current is required at the input for that loading
condition. Preferably, the net loading of the thyristors at reduced
external loads would not exceed that obtained at full load.
A typical cycloconverter will be designed for a given rating of
output current and under some conditions of operation, the
cycloconverter will not be required to deliver that amount of
output current. However, under the extreme operating condition, it
will be delivering its full rated current and simultaneously
drawing a full lagging component of current from the input power
supply, which will be the worst case condition as far as the
lagging current in relation to the input power supply 12. As the
load at the output of the cycloconverter is decreased and
correspondingly the load current in the output of the
cycloconverter decreases, the corresponding lagging component of
current at the input of the cycloconverter will also decrease so
that the equivalent inductor 22 shown in FIG. 2 would
correspondingly increase its value as the output load is decreased.
In accordance with the teachings of the present invention, in order
to keep the net quadrature current consumed by the cycloconverter
at a substantially constant value, it is desired to maintain this
equivalent inductor 22 at its lowest value regardless of the output
current being delivered by the cycloconverter. When the full output
current is being provided by the cycloconverter, this inductor
would have the same equivalent value as it does in the operation of
a conventional cycloconverter. As the output current being supplied
by the cycloconverter decreases, this equivalent inductor stays
fixed at the level which is appropriate to the maximum amount of
current even though the maximum current is no longer being drawn
from the output of the cycloconverter. Thusly, in effect, a
constant equivalent shunt inductance is provided regardless of the
load at the output of the cycloconverter, and therefore this
constant equivalent shunt inductance can be corrected by a provided
fixed capacitor. It should be noted that minimum inductance is
related to maximum inductive current, which means maximum
capacitive current would be provided. The present invention
provides an internal control of the thyristors in the respective
positive bank and negative bank of the cycloconverter such that as
viewed from the input power supply, this equivalent shunt
inductance remains substantially constant.
There are basically two ways for operating a naturally commutated
cycloconverter, which comprises two banks of thyristors, a positive
bank and a negative bank. In the first mode of operation, referred
to as the non-circulating current mode in accordance with waveform
4B shown in FIG. 4, the two banks are controlled such that when the
current flowing into the load is positive, that current is supplied
from the positive bank. Because the converters are unidirectional,
the positive bank can only carry positive current; and when the
output current is positive, the positive bank is made conductive
and the firing signals are removed from the negative bank such that
it is not conductive. When it is desired to supply negative
current, the thyristors of the negative bank are fired and made to
be conductive, and the positive bank of thyristors is not provided
with firing signals such that it is not conductive.
A second mode of operation is the circulating current mode in
accordance with the waveforms 4I and 4F shown in FIG. 4, wherein
both the negative bank and the positive bank converters are fired
all of the time regardless of the direction of the flow of the load
current, and the load current flows in each bank in conjunction
with the circulating current which circulates between the negative
and the positive bank of thyristors. The load current by itself
accounts for a lagging reactive current component as seen by the
input power supply, but the circulating current also accounts for a
second lagging reactive current component as viewed by the input
power supply. Therefore, by controlling the circulating current
independently of the load current, it is practical to control the
net effective inductance of the cycloconverter as seen by the input
power supply to have a substantially constant value. It should be
realized that there is a first component of quadrature or lagging
current due to the load and in addition a second component of
quadrature current which is due to the circulating current flowing
between the negative and the positive banks of thyristors of the
cycloconverter, with both the first component and second component
of quadrature current having the same polarity but they could have
a different amplitude. Thusly, as shown in FIG. 5 of the drawings
for the purpose of illustration, the cycloconverter input current
is made up of an in-phase power component I.sub.power and a first
component of quadrature current I.sub.qL due to the load and a
second component of quadrature current I.sub.qcc which is due to
the circulating current flowing between the positive and negative
banks. The two quadrature currents will add together to be equal to
the total quadrature current I.sub.q flowing at the input of the
cycloconverter. As the load 14 connected to the output of the
cycloconverter 10 changes, the in-phase component of input current
I.sub.power will also change, and correspondingly the quadrature
current component I.sub.qL due to the output load changes. As the
current component I.sub.qL changes in one direction, the current
component I.sub.qcc due to the circulating current is
correspondingly changed in the opposite direction such that the sum
I.sub.q of the two component currents I.sub.qL and I.sub.qcc remain
substantially constant or at a controlled magnitude as may be
desired. This control operation takes care of changing load
conditions at the output terminals of the cycloconverter 10.
For the purpose of illustration, a typical load application would
be to drive an induction motor load or a synchronous motor load. To
vary the speed of the motor in an efficient manner, the frequency
and amplitude of the voltage supplied to the motor are controlled.
One way of doing this is with a cycloconverter operative with the
60 -cycle input power supply for providing at the output of the
cycloconverter an output frequency varying from zero up to about 40
Hz. The magnitude of the output load current is essentially
independent of output frequency, but rather depends on the torque
demand on the motor. To change the speed of the AC motor, the
frequency of the output current is changed, but that in itself will
not change the magnitude of the current flowing in the output. In
fact, if a constant torque operation of the motor were desired
regardless of frequency, then the current in the output of the
cycloconverter would stay substantially fixed at all speeds of the
motor. However, this does not mean that the corresponding input
current will stay fixed, for the reason that as the output
frequency is changed, the output voltage is simultaneously
changed.
In relation to the input current supplied to the cycloconverter,
there are two components of lagging input current. One component,
I.sub.qL, is due to the load and cannot be independently controlled
because it is a function of the load, but the other component,
I.sub.qcc, is controllable and is due to the amount of current that
is allowed to circulate between the two banks of the
cycloconverter. The waveform shown in FIG. 4A is representative of
the cycloconverter output voltage under no circulating current mode
of operation for the full output load condition, when the load
impedance is such that the cycloconverter is supplying full rated
current to the load. Under that condition, no current is
circulating between the negative bank and the positive bank
converters such that I.sub.qL is at a maximum value and I.sub.qcc
is zero and I.sub.power is at its maximum value. The current
waveforms in FIG. 4B show that during the positive half cycle of
output load current, the positive converter is in conduction and
the negative converter is blocked and not providing load current.
During the negative half cycle of output load current, the negative
converter is in conduction and the positive converter is blocked.
The waveform shown in FIG. 4C illustrates the current drawn by the
cycloconverter input line A from the input power supply. The
waveform shown in FIG. 4D illustrates the voltage and net current
of input line A of the cycloconverter, assuming that a fixed
capacitor is connected across the input of the cycloconverter such
as illustrated by FIG. 2 and the net fundamental component of
current at the input of the cycloconverter is shown leading the
voltage because of the presence of that capacitor. Since there is
no circulating current to be controlled, the phase of the input
current would not be kept in the same relationship to the input
voltage when the load is varied.
For a fixed load condition at the output of the cycloconverter, by
varying the circulating current, it is practical to obtain a
varying lagging current at the input. For a varying load condition
at the output of the cycloconverter, by varying the circulating
current, it is practical to obtain a substantially fixed lagging
current at the input. For each of the latter load conditions, a
capacitor can be provided at the input to overcorrect the lagging
current component at the input and make it a leading current
component at the input of the cycloconverter. The amplitude of the
circulating current can be controlled by selected firing of the
thyristors in the respective negative bank and positive bank
converters of the cycloconverter such that the effective inductance
remains the same and fixed as the output load changes. In relation
to the waveforms shown in FIG. 4E to illustrate the output voltages
of the positive and negative bank converters, the respective
converters are fired all of the time so that each produces the same
fundamental component of output voltage as illustrated at page 157
of the above reference book by Pelly. Since both converters are in
conduction all the time, the positive bank current i.sub.p is shown
by the top waveform of FIG. 4F and the negative bank current
i.sub.n is shown by the bottom waveform of FIG. 4F. Each current
wave is just continuous in operation with the upper heavy dotted
line 44 representing zero for the positive bank current and the
lower heavy line 46 representing zero for the negative bank
current. Thusly, both the positive and negative bank currents just
reach zero momentarily during the course of the respective half
cycles, and for this reason, this is referred to as just continuous
operation because there is always current flowing in both banks
although at one specific point in each cycle, the current just
reaches zero. The amplitude of the output current i.sub.o is the
difference between the positive bank current and the negative bank
current as shown in FIG. 4F, with the same outward current i.sub.o
being provided in relation to FIG. 4F as was provided in relation
to the waveforms shown in FIG. 4B. The input current waveform shown
by FIG. 4G is now changed in relation to the input current waveform
shown by FIG. 4C, because the converter current waveforms
themselves have changed. If the dotted fundamental component were
drawn in relation to the waveform of FIG. 4G, it could be shown
that it is more lagging than the dotted fundamental component in
relation to the waveform shown in FIG. 4C. Because it is assumed a
fixed value of capacitance is connected across the input of the
cycloconverter, the net sum of that capacitive current plus the
current shown in FIG. 4G results in the dotted fundamental
component of current shown in FIG. 4H being in phase with the
voltage. For the same amount of output current, the cycloconverter
has moved from a leading current condition at the input to a
current condition in phase with the voltage, and this is done by
operating the negative bank converter and the positive bank
converter in a different mode. In relation to the voltage and net
current waveform shown in FIG. 4D, they are operated with no
circulating current and in relation to the voltage and net current
waveform shown in FIG. 4H, they are operated with a just continuous
circulating current condition which has shifted the phase
relationship angle.
In FIG. 4I, the positive bank output current i.sub.p and the
negative bank output current i.sub.n are shown for an increased
circulating current condition of operation of the cycloconverter.
The difference between the positive and negative converter output
current relationships shown in FIG. 4F and the positive and
negative converter output current relationships shown in FIG. 4I is
that each of the respective output currents has been shifted in
relation to the zero current reference such that the positive bank
output current i.sub.p in FIG. 4I has been shifted upward in
relation to the zero current reference 44 and the negative bank
output current i.sub.n shown in FIG. 4I has been shifted downward
relative to its zero current reference 46. The positive converter
output current i.sub.p is more than just continuous as shown in
FIG. 4I because its minimum value is above the zero reference, and
the same is true in relation to the negative bank output current
i.sub.n. This condition of operation is accomplished by including a
bias voltage in the form of a DC voltage component between the two
converter banks to make the two current waveforms i.sub.p and
i.sub.n as shown in FIG. 4I move apart so that each current
waveform no longer hits its respective zero reference. The current
drawn by the cycloconverter input line A for the increased
circulating current condition of operation is shown in FIG. 4J, and
the net input current waveform for line A in relation to the input
voltage of the cycloconverter, which net input current waveform is
the sum of the capacitor current waveform plus the cycloconverter
current waveform, is now clearly lagging the voltage wave as shown
in FIG. 4K. Thusly, it can be seen that with a fixed output
current, by controlling the circulating current between the
positive bank converter and the negative bank converter, the phase
of the fundamental component of input current can be shifted from a
leading condition, to being in-phase and to a lagging condition as
may be desired. Thusly, the applied bias voltage can be increased
to provide a lagging condition and the applied bias voltage can be
decreased to bring the positive bank current and the negative bank
current closer together such that a leading current condition is
thereby obtained.
In general, to obtain the waveforms shown in FIGS. 4A through 4D,
the firing pulses applied to the positive bank converter are
applied during the positive cycle and the firing pulses are applied
to the negative bank converter during the negative cycle. To obtain
the waveform shown in FIGS. 4E through 4H, the firing pulses are
applied continuously to both the negative bank and the positive
bank all the time. To obtain some intermediate condition of
operation between the condition of continuous firing pulse overlap
and the condition of no firing pulse overlap, the firing pulse
control apparatus can be arranged such that there is some period of
time when the firing pulses overlap and some period of time when
just one bank is in conduction. A reference to the above book by
Pelly at pages 190-198 will illustrate the operational principles
here involved.
For the practical operation of the naturally commutated
cycloconverter in accordance with the present invention, if it is
desired to keep the effective power factor of the cycloconverter as
seen by the input power supply substantially constant or at a
controlled magnitude and whereby the effective inductive component
of the cycloconverter is controlled as the output load changes,
then the waveforms shown in FIGS. 4A to 4D could be provided for a
maximum output load condition of operation, as the load current
decreases, so the positive and negative converter firing pulses can
be made to overlap one another, thus providing a sufficient
circulating current to maintain the desired quadrature component of
current at the input. As the load current further decreases, a
point will be reached at which the firing pulses continuously
overlap one another, and the circulating current is just
continuous. Further decrease of load current would then be
accompanied by introduction of an appropriate direct voltage bias
between the positive and negative converters to further increase
the circulating current and keep the net quadrature input current
at a controlled magnitude. Thus, the cycloconverter operation would
change from a full load and no circulating current condition of
operation to a no load and full circulating current condition of
operation.
As shown in FIG. 6, a circulating current control circuit
arrangement can be provided, whereby the input power supply 12 is
operative with the positive converter bank 30 and the negative
converter bank 40 to supply output current to a load 14 through a
circulating current reactor 50. The positive converter bank 30 is
controlled in its operation by a well-known arrangement of a timing
circuit 52 operative with a firing circuit 54 for determining the
conduction of the positive converter bank 30. Similarly, the
negative converter bank is controlled in its operation by a
well-known arrangement of a timing circuit 56 operative with a
firing circuit 58, such as shown at pages 252 and 253 of the above
reference book by Pelly. A desired output voltage reference, in
relation to the output voltage and frequency desired for the load
14, is supplied to terminal 60 and to each of summing junctions 62
and 64 for determining the operation of the respective positive
converter bank 30 and negative converter bank 40 in relation to the
load 14. A current transformer 66 senses the output current i.sub.o
supplied to the load 14 and applies a control signal in accordance
with the output current i.sub.o to each of a comparator circuit 68
and a comparator circuit 70 for determining a firing pulse release
signal to each of the firing circuits 54 and 58, respectively. A DC
voltage in the form of a quadrature current reference signal
I.sub.q * is supplied to the terminal 72 operative with a summing
junction 74 in conjunction with an actual quadrature current signal
I.sub.q applied to terminal 76 by the quadrature current to DC
voltage converter 78 in accordance with input voltage and input
current signals supplied by suitable and well-known sensing devices
80 in relation to the input power supply 12. It should be
understood that a current transformer and a voltage transformer can
be utilized for the sensing devices 80. The quadrature current to
DC voltage converter 78 is well known, and can be a multiplier
responsive to the voltage signal and responsive to the current
signal after a 90.degree. phase shift, with the output signal
passing through a smoothing filter.
A DC bias voltage error signal e.sub.1 representing the error
between the desired quadrature component input current and the
actual quadrature component of input current is applied to the
error signal amplifier 82 and then applied to a second input of the
comparator 68 and passes through an inverter 84 and supplied to a
second input of the comparator 70 for determining the operation of
the respective firing circuits 54 and 58. In addition, the DC bias
voltage error signal e.sub.1 is applied through a threshold circuit
86 such that when it is above a predetermined threshold level, it
is applied to the respective summing junctions 62 and 64 for
controlling the operation of the respective timing circuits 52 and
56, and thereby the operations of the respective positive converter
bank 30 and negative converter bank 40.
A controlled firing pulse overlap is determined by the operation of
the comparators 68 and 70, until a full overlap of the firing
pulses has been provided. Then, the threshold circuit 86 becomes
operative to introduce a direct voltage bias between the two
converter banks 30 and 40 for additional control of the converter
operation.
A quadrature current reference signal I.sub.q * is applied to
terminal 72 according to the desired quadrature current I.sub.q *.
The summing junction 74 compares the desired quadrature current
reference I.sub.q * with the actual quadrature current I.sub.q
signal from the quadrature current to DC voltage converter 78, for
establishing an error signal e.sub.1. The error signal after
amplification by high gain amplifier 82 is applied to comparator 68
for controlling the negative converter bank 40 and passes through
inverter 84 for application to comparator 78 for controlling the
positive converter bank 30. The threshold circuit 86 provides a
deadband such that no output is obtained from the threshold circuit
86 until after the firing pulse overlap control of comparators 68
and 70 is completed to provide additional quadrature current. The
direct voltage bias from the threshold circuit 86 is applied to the
respective summing junctions 62 and 64 to increase the magnitude of
the circulating current beyond the level of just-continuous
operation and until an equilibrium is obtained with the desired
quadrature current being supplied when the control system operation
becomes stabilized. It should be noted that the gain of the error
amplifier can be made high enough, including if desired some
integration, such that an output signal will be provided from the
threshold circuit 86 even when practically a zero error in the
order of one-half percent or so condition of operation is obtained.
When the cycloconverter is operating in the just-continuous
condition of operation as shown in FIGS. 4E to 4H, with 100%
overlapping firing pulses and both converters conducting all of the
time, and it is desired to provide an additional circulating
current, the amplitude of the circulating current can be increased
by providing the DC bias voltage.
In FIG. 7A, there is shown the output current i.sub.o supplied to
the load 14 as represented by the wave 100. The quadrature current
error signal e.sub.1 is shown by the voltage 102 and the inverted
error signal -e.sub.1 is shown by the voltage 104. The output
signal from the comparator 68 is shown by the waveform 106 and the
output signal from the comparator 70 is shown by the waveform 108.
The overlap angle between the waveform 106 and 108 is variable
depending upon the voltage level of the quadrature current error
signal e.sub.1, and for a large value of quadrature current error
signal e.sub.1 the overlap angle could be a full 180.degree..
In FIG. 7B, there is shown the output current i.sub.o as
represented by the waveform 110. The quadrature current error
signal e.sub.1 is shown by the voltage 112 above the indicated
deadband threshold 114 of the threshold circuit 86. The direct
voltage bias e.sub.o is the voltage difference between the
quadrature current error signal 112 and the threshold 114. The
output signal 116 from the comparator 68 is shown, and the output
signal 118 from the comparator 70 is shown.
THE PREFERRED EMBODIMENT OF THE INVENTION WITH TWO CYCLOCONVERTERS
INTERLINKING A POWER SYSTEM WITH ANOTHER POWER SYSTEM OR A LOAD
An illustrative practical application of the present invention, in
relation to a gas turbine driven transportable ground power supply,
is set forth in FIG. 8. A gas turbine 200 running at a
substantially constant speed, such as 50,000 rpm, directly drives
an electrical generator 202 for producing relatively high frequency
output voltage in the order of 850 Hz. This high frequency voltage
is converted through a naturally commutated cycloconverter 204,
with the equivalent circuit being shown in FIG. 8, to a lower
constant frequency output of 60 Hz. The high speed electrical
generator 202 could be a permanent magnet synchronous machine or an
asynchronous induction generator free from the requirement of
producing a magnetic field which rotates synchronously with the
rotor. It is well known that an induction generator 202 would
require external excitation under all operating conditions and
would have to be supplied with an adjustable reactive component of
current precisely controlled in accordance with the loading
conditions. The controllable input displacement factor of the
naturally commutated cycloconverter 204 could provide the desired
excitation for the induction generator 202 under varying load
conditions. FIG. 8A illustrates the equivalent circuit of the power
supply apparatus with full load current flowing at the output of
the cycloconverter 204. The power factor viewed at the output
terminals of the cycloconverter 204 under this condition is assumed
to be unity, and the full load power factor of the induction
generator 202 is assumed to be 0.8 lagging. Assuming that under
this full load condition no current circulates between the positive
and negative banks of the cycloconverter 204 as illustrated by the
idealized current waveforms shown in FIGS. 4A through 4D, then the
level of lagging reactive current relative to the in-phase
component consumed by the cycloconverter 204 is 0.95 p.u. (r = 0.9
assumed). The corresponding lagging reactive current I.sub.q
consumed by the induction generator 202 is 0.75 p.u., and this
fixes the required magnitude of the capacitor current at 0.95 +
0.75 or 1.7 p.u.
In FIG. 8B, there is illustrated the operation of a similar gas
turbine driven transportable ground power supply, with no external
load at the output of the cycloconverter 204, such that the L-C
filter including the inductor 206 and the capacitor 208 are assumed
to draw a purely leading current of one-third the full load value.
The reactive current I.sub.q of the induction generator 202 at
no-load is assumed to be 0.5 p.u. as compared with 0.75 p.u. at
full load. Since the fixed capacitor current through the capacitor
210 would still be 1.7 p.u., this means that the cycloconverter 204
must now be made to consume the quantity 1.7 - 0.5 or 1.2 p.u.
lagging reactive current. From the above equations (1) and (2), it
can be shown that the desired amplitude of direct circulating
current I.sub.c should be about 0.34 times the peak full load
output current I.sub.oFL. The idealized net current waveforms of
the positive and negative bank converters are illustrated in FIG.
8C such that the average loading of the converters is less than the
loading at full load. The output voltage V.sub.o of the induction
generator 202 would be sensed here for quadrature current magnitude
control in relation to a desired output voltage V.sub.o * for the
generator 202.
As another practical application of the present invention, there is
illustrated in FIG. 9 a high frequency link asynchronous power
system tie using two cycloconverters 250 and 252 as shown in basic
functional form, and operative with a reactive current supplying
high frequency link apparatus 254, connected between a first power
sysem 256 and a second power system 258. One problem with such a
power system arrangement is that as the loading conditions change,
the reactive demand on the link circuit apparatus 254 changes. With
high current flowing at the cycloconverter output terminals, the
high frequency tank circuit 254 must deliver high reactive power to
the cycloconverter input terminals. As the system currents
decrease, the reactive demand of the cycloconverter when controlled
in the conventional manner decreases. The prior art solution
proposed for this problem is to allow the link frequency to adjust
automatically to the loading conditions to provide a means for
controlling the reactive power fed to the cycloconverters from the
tank circuit. This solution, however, requires a relatively large
tank in order to keep the swing of link frequency between the
extremes of loading conditions within acceptable limits.
An improved approach would be to utilize the disclosed principle of
the present invention for controlling the reactive input current of
the cycloconverter. The circulating current of the cycloconverter
would be regulated with changing external load so as to preserve a
constant reactive demand on the link apparatus tank circuit. In
this manner, the link frequency could be kept substantially
constant at all loads, and thus the size of the tank circuit could
be minimized.
A voltage transformer 251 is provided to sense the frequency of the
power flow between the tank circuit 254 and the two cycloconverters
250 and 252. The frequency to DC voltage converter is a well known
apparatus, including logic circuitry to sense the zero crossings of
the voltage signal and provide corresponding fixed width pulses
which can be averaged to give an output signal proportional to the
actual frequency sensed by the voltage transformer 251. This actual
frequency f is then compared with a desired frequency reference f*
for controlling the quadrature current relationships of the
cycloconverters 250 and 252, as previously described in relation to
the current control apparatus shown in FIG. 6. If the frequency of
the tank circuit 254 tends to be too low, it is desired that the
quadrature currents I.sub.q should increase and vice versa.
The schematic illustration of FIG. 10A represents the full load
condition of operation for such an asynchronous power system tie,
with power assumed to be at 0.8 power factor being transmitted from
power system 256 to power system 258. Under this condition of
operation, the cycloconverters 250 and 252 are assumed to operate
with no circulating current as indicated by the idealized waveforms
shown in FIG. 10B. The corresponding reactive component of current
I.sub.q drawn at the input of each of the cycloconverters 250 and
252 is designated as 1 per unit. Fig. 10C illustrates the no-load
condition of operation of such a power system, with no current
flowing into or out of either power system 256 or power system 258.
In order to keep the link tank frequency the same as at full load,
each of the cycloconverters 250 and 252 must still draw 1 per unit
reactive current I.sub.q from the tank circuit 254. From the above
equation (2), it can be shown that the steady circulating current
I.sub.c in each cycloconverter tank should be approximately 0.3
times the peak full load current I.sub.oFL, as illustrated by the
idealized waveforms shown in FIG. 10D, assuming r = 0.9. Again, it
should be clear that the loading of the converters 250 and 252
under no-load conditions is less troublesome than at full load.
Thusly, the proposed approach of keeping the tank frequency fixed
with changing load conditions will be a practical solution to the
problem.
In FIG. 11, there is diagrammatically shown the power system tie of
FIG. 9, with the example being for single-phase power systems to
simplify the illustration. For three-phase power systems, three of
the circuit arrangements shown in FIG. 11 would be required for
operation with the same tank circuit 254.
FIG. 12 shows two naturally commutated cycloconverters 250 and 252
forming an intertie between two power systems A and B and having in
common a tank circuit 254, controlled by respective pulse timing
and firing circuits 330 and 331. Summers 332 and 333 provide the
necessary control signals for the firing circuits 330, 331,
respectively. In contrast to the circuit of FIGS. 9 and 11 the
circuit of FIG. 12 includes circuitry for regulating power flow
between the two power systems A, B through the two cycloconverters,
while the previously described circuitry insures that the power
factor at either side of the tie can be maintained at unity, or any
other desired value.
In order to control power flow between the two power systems, real
and quadrature current transducers 301, 302 are respectively
provided on the power line of the two systems A and B. From each
transducer are derived two signals I.sub.Q,FB.sbsb.1, and
I.sub.R,FB.sbsb.1 representing the actual quadrature and the actual
real component for the transducer 301, in the first feedback loop
and I.sub.Q,FB.sbsb.2, I.sub.R,FB.sbsb.2 the actual quadrature and
actual real components for transducer 302 in the second feedback
loop. These signals are each compared with corresponding reference
signals I.sub.Q.sbsb.1 *, I.sub.R *, I.sub.Q.sbsb.2 * to provide,
after summation in respective summers S.sub.1, S.sub.1 ', S.sub.2,
S.sub.2 ', error signals. These error signals after integration
(through integrator I.sub.1, I.sub.1 ', I.sub.2, I.sub.2 ') gate
sinewave generators G.sub.1, G.sub.1 ', G.sub.2, G.sub.2 ' which,
according to the sinewave control method shown on pages 190-192 and
248-254 of the Book of Pelly, determine under voltage reference
V.sub.S.sbsb.1 * and V.sub.S.sbsb.2 * the timing of the firing
pulses to each cycloconverter.
Thus, the reference voltage for each cycloconverter is made up of
three components:
(V.sub.S *), (V.sub.O,180 *) and (V.sub.+-.sub.90 *). The first
component V.sub.S * produces a component of voltage at the output
of the cycloconverter (250, or 252) which is equal to and in phase
with the voltage of the associated system (A, or B). If the two
other components are both zero, no current flows at the output
terminals of the cycloconverter.
Considering component reference voltage V.sub.O,180 * applied to
summer 332, or 333, this voltage is either in phase or in
anti-phase with V.sub.S *. Thus, variation of the amplitude and
sign of this component produces a corresponding variation in the
reactive component of current drawn from the system. The action of
the feedback loop due to the integrated error between I.sub.Q * and
the actual quadrature component feedback signal is to force the
actual system quadrature current to correspond to the reference
value I.sub.Q *.
The third component is the one involving real current. The feedback
loop involved provides a correction signal in relation to the
integrated error between real current reference signal I.sub.R *
and the feedback signal representing the actual real component
I.sub.R,FB of current flowing in the system. Again, this feedback
loop is such as to force the actual real component of current to
correspond to the reference value I.sub.R *.
All that is necessary in order to make the real power flowing into
one side equal to the real power flowing out of the other is to
apply a common real current reference to both cycloconverters, but
with opposite polarity. If the voltage of the two power systems are
equal at both sides, this will produce the desired equal but
opposite power flow at the two sides.
The quadrature currents at each side, on the other hand, are
adjusted independently of one another, since they do not contribute
any net real power flow through the intertie.
Should the voltages of the power systems not be equal, then it is
understood that some circuitry can be added in order to modify the
real current reference signals against the difference in voltage
between the two sides.
PRIOR ART TECHNIQUES FOR POWER FACTOR GENERATORS IN UTILITY
SYSTEMS
The power factor of utility and industrial power lines is usually
corrected by rotating synchronous condensers and/or constant or
mechanically switched passive capacitor banks. The steady state
performance of a rotating synchronous condenser is, under balanced
load conditions, excellent: it provides practically sinusoidal and
continuously variable leading or lagging three-phase current for
the power lines without causing undue transients. However, the
synchronous condenser also has a number of disadvantages which
hinder its wide application: it is expensive, it has moving parts
which need maintenance, its ability to supply unbalanced VAR demand
is limited and its response time is slow. Fixed capacitor banks, on
the other hand, can provide constant reactive power and, therefore,
are only applicable when relatively constant lagging VAR
consumption is encountered. To decrease the effect of varying VAR
consumption, capacitor banks may be switched mechanically in and
out, either individually or in three-phase sets, by suitable
contactors. However, the compensation will be slow and it will
follow the VAR demand in a step-like fashion. In addition, the
switching will, in general, generate transients on the power lines.
The recent advancements in high power thyristor technology and
electronic circuitry make the concept of solid state power factor
correctors more attractive in many practical applications,
promising superior technical performance at economic cost.
There are essentially three methods of static VAR generation: (1)
shunt capacitors and inductors in conjunction with solid state
"on/off" and phase controlled switches; (2) AC/DC converters and
inverters; and (3) the more recently conceived AC/AC frequency
changers (cycloconverters). An evaluation of these three types of
systems indicates that the first group is probably the simplest and
least expensive; in addition, this approach is well suited to
compensate for unbalanced reactive power consumption. On the other
hand, the system is physically large and it represents a resonant
type load (with multiple and variable self frequencies) on the
power lines. The two other groups are more suitable to compensate
for basically balanced three-phase reactive power consumption. The
AC/DC converters are economically competitive and physically small.
However, they must, in most cases, be complemented with passive
filters to provide current waveforms with acceptable distortion.
Inverters are generally expensive, but their performance can
approximate, or even excel, that of the rotating synchronous
condensers. The direct AC/AC frequency changer schemes appear under
proper conditions to give a performance generally equivalent and in
some respects superior to the rotating counterpart and at a
competitive cost.
For power factor correction with thyristor controlled capacitors
and inductors, two basic schemes are possible: one is to control
the leading VAR by switching stationary capacitor banks to the
lines, and the other achieves the same aim by combining a fixed
capacitor bank and a parallel thyristor controlled "variable"
inductor.
An obvious first method of providing controllable leading VAR for
the power lines is to switch in and out appropriately dimensioned
capacitor banks with anti-parallel connected thyristor switches
associated with each capacitor bank.
Another well known system consists of a fixed capacitor in parallel
with a variable inductor as shown in FIG. 13. The magnitude of the
effective AC impedance of the inductor is controllable between
infinity and the absolute value of the impedance of the fixed
capacitor. At maximum leading VAR demand, the impedance of the
variable inductor is infinity and thus the full capacitive current
is drawn from the AC supply. At zero VAR demand, the impedance of
the variable inductor is opposite to that of the fixed capacitor
and consequently the net current provided is zero. Between these
two extreme points, the impedance of the inductor is set so that
the resultant capacitive current corresponds to the instantaneous
VAR demand. The operation of this basic system may be briefly
described as follows:
At zero leading VAR demand, the thyristor switch is closed and the
inductor cancels the effect of the capacitor so that the net
current provided is zero. At some non-zero leading VAR demand, the
closing of the switch is appropriately delayed by a variable angle
("firing angle"), .alpha., with respect to the peak of the supply
voltage in order to reduce the current in the inductor. With
increasing .alpha. (0.degree. to 90.degree.), the inductive current
decreases, as illustrated in FIG. 14, and, consequently, the
leading VAR provided for the AC supply increases. At maximum
leading VAR demand, the switch is open (.alpha. = 90.degree.), the
current in the inductor is zero and, therefore, the maximum rated
capacitive current is drawn from the AC supply. As stated
previously, the fixed reactor in series with phase controlled
thyristor switch can be considered as a variable inductor having a
response time of one half of a cycle (i.e. angle .alpha. can be set
at every half cycle).
Power factor correction is also performed conventionally with AC/DC
converters and inverters. AC/DC converters may be used for pure
reactive VAR generation and thus they may be employed as power
factor correctors. A distinction between converter and inverter can
be made according to the capability of the static equipment to
maintain its AC terminal voltages independently of the AC system to
which it is connected. Specifically, the static equipment is called
converter if its terminal voltage at the AC side is maintained by
the AC system, and it is called inverter if its AC terminal voltage
can be maintained and controlled independently of the AC system.
With this definition, the direction of the power flow is
immaterial; both equipments are capable of absorbing power at the
DC terminals and supplying power at the AC terminals, or vice
versa. However, it will be seen that in the generation of reactive
VA, both the converter and inverter are controlled to absorb the
real power needed from the AC system to replenish their internal
losses, making it possible to use passive reactive elements
(inductors or capacitors) to establish the DC current or voltage
required at the DC terminals.
The AC/DC converter may be either naturally or force commutated.
The AC/DC inverter, on the other hand, must be force commutated.
The naturally commutated converter can only provide lagging VAR; by
contrast, the force commutated converter and inverter can provide
lagging as well as leading VAR.
A naturally commutated converter, illustrated in FIG. 15A for one
bank, can only operate if the thyristors are fired at phase angles
where conduction of the DC current is "naturally" transferred from
one pair of thyristors to the next pair. Referring to curves (a) in
FIG. 15B, representing load current, commutation can only take
place with the indicated polarity of the thyristor from A.sub.1 to
B.sub.1 when line B is more positive than line A. This limits the
maximum range for firing thyristor B.sub.1 from 0.degree. to
-180.degree., measured from the first point of natural commutation.
Consequently, the phase of the AC line current will lag the line
voltage by an angle equal to the firing delay angle. Thus, in the
application considered, where the converter is to be used to
generate purely reactive power, the firing angle has,
theoretically, to be -90.degree.. The output DC voltage is,
therefore, essentially zero and the converter is terminated by a DC
inductor L.sub.1. In order to establish and maintain the required
DC current level, the firing angle must, of course, be slightly
less than 90.degree. so that there is just enough DC voltage
V.sub.o to overcome the thyristor drops, and the resistance of the
DC inductor. It follows that the magnitude of the DC current and,
consequently, the amplitudes of the resultant AC line currents
I.sub.A, I.sub.B and I.sub.C (see curves (f), (g), (h) in FIG. 15B)
drawn by the converter are controllable by means of a slight
adjustment in the firing angle.
The naturally commutated converter can thus be considered as a
reactive current generator, capable of providing three balanced
lagging currents of controllable amplitude for the AC lines. It can
also be viewed, at the fundamental line frequency, as a
continuously variable balanced three-phase inductor, capable of
compensating for reactive leading power consumption.
Controllable leading VAR can be provided to compensate for lagging
power consumption in a manner similar to that discussed earlier.
That is to say, the converter, acting as a variable three-phase
inductor, is connected in parallel with a three-phase capacitor
bank of equal rating. In this way, by controlling the converter
current (inductive) between maximum rated and zero, the capacitive
current in the lines will vary between zero and maximum.
Force commutated converter schemes are another solution to the
problem. However, commutation techniques and circuits create
problems of their own.
With an inverter (in general, regarded as an AC voltage source) a
high quality output waveform that closely approximates a sinusoid
can be produced. If such an inverter has its output connected
through an inductor to an AC source of the same frequency, as shown
in FIG. 16, the voltage across the inductor will be the vectorial
difference between the voltage produced by the inverter (V.sub.inv)
and that produced by the AC source V.sub.AC. This voltage
difference will cause a current to flow in the inductor (I.sub.inv
= - I.sub.AC). If the voltages of the inverter and the AC source
are in phase, then the current in the inverter (and in the AC
source) will be purely reactive. When the inverter's voltage
(V.sub.inv) is greater than that of the generator V.sub.AC, then
the inverter will effectively "see" an inductive load, while the
source will see a capacitive load. If, on the other hand, the
inverter's voltage is lower, then it will see a capacitive load
while the source sees an inductive load. Thus, by keeping the
inverter's voltage exactly in phase with the source voltage while
varying its magnitude, the current drawn by the inverter from the
AC generator is controllable from full rating inductive to full
rating capacitive. If the voltages are exactly in phase, the
inverter absorbs no real power from the AC source and thus losses
will have to be replenished from a separate DC supply. The DC
supply can, however, be dispensed with if a suitable DC reservoir
capacitor is used and the phase of the inverter voltage is made to
slightly lag that of the AC source. A real power component of
current will then flow from the generator to the inverter, and the
losses will be compensated for.
The reservoir capacitor of an inverter operating without a DC input
will have to carry ripple current of a magnitude and frequency
dependent on the power rating, the circuit configuration, operating
mode, and the number of AC phases. At any rate, assuming a balanced
three-phase system, the amplitude of the ripple current is lower,
while its frequency is higher than that of the output current.
Thus, the rating of the single DC capacitor needed to furnish a
given three-phase reactive current through an inverter, may be a
small fraction of the rating of its three-phase AC equivalent
capacitor, or inductor.
The inverter operated to generate reactive VA, is analogous to a
synchronous capacitor whose internal EMF and leakage inductance
determine the magnitude and phase of the reactive current. The
inverter can, therefore, be regarded as a "solid state synchronous
condensor".
Static frequency changers are the third basic approach to static
reactive power generation. In this class of VAR generators, there
are two basic approaches both of which may employ naturally or
force commutated converters.
In the first scheme, the input and output frequencies are the same
and identical to the supply frequency. The main function of the
frequency changer is to invert the phase angle of the throughput
current with respect to the input and output terminal voltage. This
system requires no passive storage components. This scheme has been
described in copending U.S. patent application Ser. No. 575,888,
filed on May 8, 1975, by L. Gyugyi, J. Rosa and E. J. Stacey,
entitled "Static Reactive Power Generating Apparatus".
The second scheme is the concept of a high frequency link used by
the present invention.
DESCRIPTION OF THE PREFERRED EMBODIMENT IN THE CASE OF A SINGLE
CYCLOCONVERTER USED AS A CONTROLLABLE STATIC REACTIVE POWER
GENERATOR
As a result of the availability of high power thyristors (SCR's),
solid state power conversion equipment has become a practical
reality for electric utility type applications. Due to inherent
limitations of these high power thyristors (no gate-controlled
turn-off ability, relatively long turn-off time), however, utility
type applications are predominantly limited to naturally or input
source commutated converters. In these systems, commutation of the
thyristors is accomplished by sequential gating, properly phased
with respect to the AC source voltages so that the "outgoing"
thyristor always receives a reverse bias voltage for a duration of
time necessary to recover from conduction. Input source commutated
converters are conceptually simple but display some disadvantageous
features which become increasingly bothersome at high power levels.
They reflect their load with a lagging power factor to the AC
source and they draw substantial amounts of relatively low order
number harmonic currents from their source. Their response time is
keyed to the source frequency. The novel approach according to the
present invention preserves the advantages of input source
commutated converters, while eliminating their disadvantages.
The basic principle of operation of the static reactive power
generator according to the present invention is better understood
by referring first to the conventional synchronous condenser, as
illustrated in FIG. 17, when used to provide controllable reactive
power. For purely reactive power flow, the induced EMF E is in
phase with the system voltage V. By controlling the excitation of
the machine, and hence the amplitude of E, the reactive power can
be controlled; increasing E above V causes leading current to be
drawn from the system, whereas decreasing E below V produces a
lagging load on the system.
An alternative method of implementing a controllable VAR supply is
illustrated in FIG. 18. The synchronous condenser of FIG. 16 has
been replaced by a relatively high frequency machine, feeding a
static cycloconverter, which converts the machine frequency down to
the system frequency. The amplitude of the output voltage of the
cycloconverter can be controlled to produce either leading or
lagging VAR's, either by control of the excitation of the machine,
or by internal control of the cycloconverter.
The cycloconverter consists in its basic form simply of a circuit
array of thyristors, which, through appropriate control, serves to
convert the machine frequency to system frequency. The basic
frequency conversion process is illustrated by the waveforms in
FIG. 18. Selected segments of the relatively high frequency
voltages produced by the machine are pieced together, by
appropriate gating of the thyristors within the cycloconverter, to
create an output voltage wave with a fundamental frequency equal to
the system frequency. By suitable control of the switching periods
of the thyristors, the amplitude of the fundamental component of
the output voltage of the cycloconverter, relative to the machine
voltage, can be controlled; and, by a similar process, the output
frequency of the cycloconverter can be held constant at the system
frequency against changes in machine frequency. Thus, it is
unnecessary for the machine frequency to be held rigidly fixed.
As the reactive current supplied to (or drawn from) the system is
varied, through control of the amplitude of the fundamental output
voltage of the cycloconverter, this varying reactive power is
reflected through the cycloconverter to the machine. However, due
to the peculiar transfer properties of the line commutated
cycloconverter, both lagging and leading VAR's at the system side
always appear as lagging VAR's to the machine.
Since the machine in FIG. 18 theoretically handles only reactive
power, it can be replaced by a static oscillating tank circuit, as
shown in FIG. 19. As with the scheme of FIG. 18, control of the
reactive power at the system side can be obtained through control
of the voltage generated at the output terminals of the
cycloconverter, which can be accomplished from within the
cycloconverter. The varying reactive load reflected on the tank
circuit by the cycloconverter, as the reactive power at the system
side is varied, can be accommodated by allowing the oscillating
tank frequency to vary appropriately with the loading conditions,
whilst the cycloconverter output frequency is held rigidly at the
system frequency.
It is clear that the scheme of FIG. 19 constitutes an all-static
approach to a continuous VAR controller. It is a naturally
commutated system in which the AC source from which the converter
commutates is a three-phase high-frequency tank circuit. It should
be noted that this is a "three-pulse" system, but comparable
results are obtained by using a system of higher (6 to 12) pulse
numbers.
In FIG. 20, there is schematically shown a cycloconverter 10
operatively connected for reactive power correction between an AC
power system (of voltage V.sub.1 and frequency .omega..sub.1)
connected at the output of the cycloconverter 10, and a source
having a frequency .omega..sub.2, higher than the frequency
.omega..sub.1 of the AC power system, and a voltage V.sub.2. The
thyristors of the cycloconverter are controlled by reference to the
output voltage V.sub.1 of the cycloconverter with a firing pattern
corresponding to the frequencies W.sub.1 and W.sub.2 for
commutation by the reactive power input source. Whatever the angle
of displacement of the output current with respect to the voltage
of the AC power system, the input current of the cycloconverter is
always lagging the input voltage. The output displacement angle may
be represented by either an inductance (lagging VAR) or a capacitor
(leading VAR) at the output. But whatever the nature of the
quadrature component of current at the output, the quadrature
current component at the input is always lagging. The technique of
power factor correction according to the High Frequency Link
technique consists in varying the inequality V.sub.2 < V.sub.1
or V.sub.1 < V.sub.2.
Referring to FIGS. 21, 22 and 23, a new type of static reactive
power generator will now be described in detail in which the VAR is
automatically supplied to an alternating current power system. This
new generator includes an "HF link" which is automatically
maintained at a substantially constant frequency, so that in the
case of an oscillating tank circuit ("passive" link) the VAR rating
can be optimized, and in the case of an external source ("active"
link), the HF source can be of a smaller VA rating.
Referring to FIG. 21, a typical three-phase naturally commutated
balanced 2-quadrant 6-pulse bridge cycloconverter is shown. Each
output phase includes 12 thyristors distributed between two
opposite banks. Each input phase combines two pairs of thyristors
from each bank and from each of the three dozens of thyristors. The
input side, of higher frequency, includes three oscillating tank
circuits HFT.sub.1, HFT.sub.2 and HFT.sub.3 which are
Wye-connected. The output side, of lower frequency, includes the
alternating current power system to be power factor corrected. The
power system is coupled at the output through a transformer having,
with the three phases I, II, III, a Wye connection at the primary
P.sub.1, P.sub.2, P.sub.3 and split secondaries S.sub.1, S.sub.2,
S.sub.3, connected at each end to the three respective groups of
thyristors through the central tap of a corresponding interphase
reactor R.sub.1, R'.sub.1, R.sub.2, R'.sub.2 or R.sub.3, R'.sub.3
between two opposite banks for each direction of flow of the
current during conduction. Each of the three tank circuits HFT.sub.
1, HFT.sub.2 and HFT.sub.3 is responsive to conduction in either
direction for each bank and for the three lines of the AC power
system. Thus, when selected thyristors are controlled for
conduction, while natural commutation turns off the outgoing
thyristors, energy is either accumulated or restituted by the
particular tank circuit. In this sustained exchange of energy, the
tank circuit operates as an input source for the cycloconverter. At
the output side, e.g. at the secondary of the transformer an output
voltage is generated having a lower frequency than the tank
circuit. The frequency of the tank circuit is such that the
frequency at the output matches the frequency of the AC power
system. Also, the output voltage is so selected that at the primary
of the transformer the voltage is equal to the line voltage of the
AC power system.
As explained, when the voltage between terminals A and B of the
interphase reactor (R.sub.1, R'.sub.1, R.sub.2, R'.sub.2, or
R.sub.3, R'.sub.3) is smaller than the voltage between the winding
terminals of the transformer (S.sub.1, S.sub.2, or S.sub.3),
considering the small inductance L.sub.1, L.sub.2 or L.sub.3
present between the output to the AC power system and the output
from the cycloconverter, inductive reactive currents will flow from
the AC power system to the frequency changer output. The currents
lead the output voltage of the frequency changer and lag the output
voltage of the AC power system. Thus, the frequency changer is
capacitively loaded, while the AC power system is inductively
loaded.
In the reverse situation, e.g. when the voltage between terminals A
and B of the interphase reactor (R.sub.1, R'.sub.1, R.sub.2,
R'.sub.2 or R.sub.3, R'.sub.3) is larger than the voltage between
the winding terminals of the transformer (S.sub.1, S.sub.2 or
S.sub.3), the frequency changer is inductively loaded, while the AC
power system is capacitively loaded.
Also, as stated heretofore, in either situation current at the
input side is lagging the input voltage, e.g. between neutral
terminal N and line 1, 2 or 3.
Indeed FIG. 21 is for the purpose of illustration only. The
cycloconverter could have any of the well known configurations such
as explained in Pelly's book. Certain arrangements may reduce the
number of tank circuits necessary, others may require more such
tank circuits.
Referring now to FIG. 22, the cycloconverter, for instance as
described in FIG. 21, will now be considered with the adjunction of
a voltage controller determining the degree of power factor
correction as just explained by reference to the inductances
L.sub.1, L.sub.2 and L.sub.3 of FIG. 21.
FIG. 22 shows schematically the basic functional organization for
power factor correction in the static reactive power generator
according to the present invention when the "HF source" is a tank
circuit.
The tank circuit 110, schematic for a plurality of tank circuits
which may be necessary depending on the particular polyphase
arrangement of thyristors, is coupled to the input of the
cycloconverter 101 including two banks 111, 112, also schematically
representing any combination of thyristors associated with one or
the other polarity. Similarly, an interphase reactor IR is
schematically represented between the outputs of the positive and
negative banks, the center tap of which is connected to the
secondary of transformer T coupling the output of the
cycloconverter to the AC power system to be corrected. A small
inductance L is present in the cycloconverter output between the
secondary of transformer T and the tapping point of reactor IR,
which inductance, as shown in FIG. 20, separates the point at the
cycloconverter output voltage V.sub.2 from the point at voltage
V.sub.1 of the AC power system. These two potentials have a
difference .DELTA.V.sub.1 = V.sub.1 - V.sub.2 which is positive
when the cycloconverter consumes lagging VAR's from the AC power
system, and which is negative when the cycloconverter delivers
lagging VAR's that is, it consumes leading VAR's from the
system.
The control circuitry shown in FIG. 23 causes the cycloconverter to
generate, in response to a control signal
derived on line 120 and applied to summer 123, output voltage
V.sub.2 which is as required for power factor correction in the AC
power system. Control of the output voltage of a cycloconverter is
well known. See for instance pages 190-192 and 248-254 of the Book
of Pelly. Referring to FIG. 21, in normal operation, three
substantially equal sinusoidal voltages, mutually displaced by
120.degree., are developed across the three tank circuits. The
timing of the firing pulses to each cycloconverter is controlled
with respect to the high frequency tank voltages so that each
cycloconverter fabricates a fundamental component of voltage across
its ouput terminals A, B, which is substantially in-phase with the
voltage applied from the connected system across the associated
transformer secondary winding In order to consume lagging VAR's
from the system, the amplitude of the wanted component of
cycloconverter output voltage is made less than the corresponding
applied transformer secondary voltage, through appropriate control
of the timing of the firing pulses in response to a control signal
i.sub.o derived from a current transformer CT at the output of the
cycloconverter. In order to deliver lagging VAR's to the system,
i.e., to consume leading VAR's, the amplitude of the cycloconverter
output voltage is made appropriately greater than the corresponding
transformer secondary voltage.
Referring again to FIG. 22, the alternating reference voltage
.DELTA.V.sub.1 is synchronized and in phase with the system voltage
appearing across the associated transformer secondary. With
.DELTA.V.sub.1 = 0, the fundamental output voltage V.sub.2 of the
cycloconverter is exactly equal to the corresponding transformer
secondary voltage V.sub.1, the voltage V.sub.1 /k being fed back
via lines 122 and 124 as the only input to the cycloconverter pulse
timing circuits 128, 129, and no fundamental current flows (k =
voltage gain between the inter-phase reactor IR connecting point
and summer 123). When the polarity of .DELTA.V.sub.1 is such as to
aid the feedback signal V.sub.1 /k , the cycloconverter delivers
lagging reactive volt-amperes to the connected system. With the
opposite polarity of .DELTA.V.sub.1, the opposite is true. In this
basic scheme, as the external reactive load changes, the reactive
current demand of the cycloconverters on the tank circuit varies,
and the operating frequency of the tank circuit changes
accordingly.
The operation of the circuitry of FIG. 22 is readily understood
from the general information found in the Book of Pelly. The two
banks of thyristors 111 and 112 are controlled by firing circuit
132 for the positive polarity and by firing circuit 133 for the
negative polarity. From the current sensed at the output of the
interphase reactor IR, bank selector 136 determines which bank is
to be fired at a given time. The instant of firing is determined
for each thyristor by the congruence in time between the reference
signal at the frequency .omega..sub.1 and the sinusoidal curve at
frequency .omega..sub.2 of the particular phase from the HF tank
circuit 110, as received on lines 113, 114 and 113, 115 for the
respective banks. The timing circuits 128, 129 establish the
duration of conduction of the oncoming thyristor in relation to the
control signal .DELTA.V.sub.1 on line 120.
Referring now to FIG. 23, circuitry is shown combining output
voltage control such as shown in FIG. 22, for automatic power
factor correction, with control of the circulating current between
the two banks of the cycloconverter so as to correct automatically
for changes in the equivalent inductance seen from the input of the
cycloconverter and therefore maintain operative oscillation between
the cycloconverter and the HF tank circuit at the selected and
desired frequency .omega..sub.2.
The circulating current control circuit includes a detector device
200 operatively connected to line 113, e.g. where energy is
exchanged between the HF tank circuit 110 and the cycloconverter
banks 111, 112. Detector device 200 senses the quadrature current
I.sub.q at the output of the tank circuit as an indication of the
variations in the equivalent inductance seen from the input of the
cycloconverter.
A quadrature current reference signal I.sub.q * is applied to
terminal 204 according to the desired quadrature current I.sub.q.
The summing junction 205 compares the desired quadrature current
reference I.sub.q * with the actual quadrature current I.sub.q
signal derived on line 203 from the quadrature current to DC
voltage converter 202, for establishing an error signal e.sub.i.
The error signal e.sub.i after amplification by high gain amplifier
207 is applied to comparator 211 for controlling the negative bank
112 and also passes through inverter 210 and comparator 212 for
controlling the positive bank 111. The control technique making use
of comparators 211, 212 and firing circuits 132, 133 in relation to
banks 111 and 112 is in accordance with the technique shown on
pages 250 and 251 of the Book of Pelly.
In addition, the DC bias voltage error signal e.sub.i is gated by a
threshold circuit 214 so that when it is above a predetermined
threshold level, e.sub.i is applied via lines 215, 216, and 215,
217 to the respective summing junctions 230 and 231 for controlling
the operation of the respective timing circuits 128 and 129, and
thereby the operation of the respective positive bank 111 and
negative bank 112.
A controlled firing pulse overlap is provided by the operation of
comparators 211 and 212, until a full overlap of the firing pulses
has been reached. Then, the threshold circuit 214 becomes operative
to introduce a direct voltage bias between the two banks 111 and
112 for additional control of the converter operation. The
threshold circuit 214 provides a deadband such that no ouput is
obtained from the threshold circuit until after the firing pulse
overlap control of comparators 211 and 212 is completed to provide
additional quadrature current. The direct voltage bias from the
threshold circuit 214 when applied to the respective summing
junctions 230 and 231 increases the magnitude of the circulating
current beyond the level of just continuous operation until an
equilibrium is obtained with the desired quadrature current being
supplied as the control system operation becomes stabilized. It
should be noted that the gain of the error amplifier can be made
high enough to include if desired, some integration factor so that
an ouput signal will be provided by the threshold circuit 214 even
when practically a zero error (in the order of one-half percent or
so) condition of operation is obtained. When the cycloconverter is
functioning in the just-continuous condition of operation, as
earlier explained by reference to FIGS. 4E to 4H, e.g. with 100
percent overlapping firing pulses and both converters conducting
all of the time, if it is desired to provide an additional
circulating current the amplitude of the circulating current can be
increased by providing a DC bias voltage.
In accordance with the present invention, there has been disclosed
apparatus including one, two, or more, cycloconverters naturally
commutated which have a circulating current established and
controlled so as to keep the input current lag at a predetermined
value despite affecting load conditions at the output.
* * * * *