U.S. patent number 3,909,755 [Application Number 05/489,869] was granted by the patent office on 1975-09-30 for low pass microwave filter.
This patent grant is currently assigned to The United States of America as represented by the Secretary of the Army. Invention is credited to Helmuth M. Kaunzinger.
United States Patent |
3,909,755 |
Kaunzinger |
September 30, 1975 |
Low pass microwave filter
Abstract
A low pass microwave filter comprised of a two conductor
transmission line ncluding at least two cascaded tapered sections
wherein one of the conductors in each filter section has a tapered
surface, preferably linear, and having an electrical length
substantially equal to one half the wavelength of the center
frequency of operation. Geometrically and performance-wise the
subject filter is unsymmetrical with respect to its input and
output ports. The filter does not exhibit the usual rapid
transition between the pass band and stop band which is a general
characteristic of conventional low pass microwave filters utilizing
reflective harmonic structures.
Inventors: |
Kaunzinger; Helmuth M.
(Neptune, NJ) |
Assignee: |
The United States of America as
represented by the Secretary of the Army (Washington,
DC)
|
Family
ID: |
23945604 |
Appl.
No.: |
05/489,869 |
Filed: |
July 18, 1974 |
Current U.S.
Class: |
333/206;
333/208 |
Current CPC
Class: |
H01P
1/211 (20130101); H01P 1/201 (20130101) |
Current International
Class: |
H01P
1/201 (20060101); H01P 1/211 (20060101); H01P
1/20 (20060101); H01P 001/20 (); H01P 003/06 ();
H01P 003/08 () |
Field of
Search: |
;333/73R,73C,73W,84M,84R,97R,98R,73S |
References Cited
[Referenced By]
U.S. Patent Documents
Primary Examiner: Lieberman; Eli
Assistant Examiner: Nussbaum; Marvin
Attorney, Agent or Firm: Edelberg; Nathan Gibson; Robert P.
Murphy; Kenneth J.
Government Interests
The invention described herein may be manufactured and used by or
for the Government for government purposes without payment of any
royalty thereon or therefor.
Claims
Having thus described what is at present considered to be the
preferred embodiments of the subject invention, I claim:
1. A low pass microwave transmission line filter having input and
output end ports and being unsymmetrical about its ports, for
inclusion in a transmission line having first and second continuous
guiding surfaces, said filter comprising,
spaced conductive means providing third and fourth guiding surfaces
facing toward each other,
the third guiding surface having, at each port of said filter, a
configuration to form a continuation of the first guiding surface
of the transmission line,
the fourth guiding surface having, at each port of said filter, a
configuration to form a continuation of the second guiding surface
of the transmission line,
said third guiding surface being geometrically continuous
longitudinally throughout the extent of the filter,
the fourth guiding surface including more than one tapered segment,
the direction of taper of all segments being the same,
each segment having a transition that is essentially perpendicular
to the length of the filter,
whereby said filter is operable in a frequency band having a center
frequency corresponding to a wavelength that is substantially twice
the length of each tapered segment measured longitudinally of the
filter.
2. The filter as defined by claim 1 wherein said tapered segments
have substantially linear periodic tapers.
3. The filter as defined by claim 2 wherein said guiding surfaces
are on inner and outer coaxial conductors.
4. The filter as defined by claim 3 wherein said tapered segments
are generally frusto-conical in configuration and each segment is
terminated by a short cylindrical integral extension at its end of
maximum departure relative to the fourth guiding surface, said
filter additionally including a dielectric spacer between the inner
surface of said outer conductor and said inner conductor at the
region of said tapered segments and wherein said integral
extensions of said tapered segments contact said spacer for
supporting said tapered segments of inner conductor within said
outer conductor.
5. The filter as defined by claim 2 wherein said inner conductor
provides said third guiding surface and said outer conductor
provides said tapered segments formed along the inner surface of
said outer conductor.
6. The filter as defined by claim 5 wherein said inner conductor is
substantially circular in cross section and additionally including
a dielectric sleeve around said inner conductor in the region of
said tapered segments provided by the inner surface of said outer
conductor.
7. The filter as defined by claim 1 wherein said transmission line
is a strip transmission line and wherein said third guiding surface
is provided by a metallic substrate acting as a ground plane, and
additionally including a dielectric layer on said metallic
substrate, and wherein said fourth guiding surface is provided by a
strip conductor of substantially constant thickness located on said
dielectric layer.
8. The filter as defined by claim 1 wherein said transmission line
is hollow rectangular waveguide with a pair of parallel broadwalls
and a pair of parallel sidewalls, said filter being hollow
rectangular waveguide having one broadwall that is geometrically
continuous and providing said third guiding surface, the other
broadwall of said filter having substantially parallel inclined
inner surface segments formed therealong.
Description
BACKGROUND OF THE INVENTION
1. Field of the Invention
The present invention relates generally to transmission line
systems for transmitting high frequency electrical energy at a
desired frequency while rejecting energy at one or more undesired
frequencies and more particularly to a low pass transmission line
filter for microwave applications where it is desirable, for
example, to couple a pulse source and/or a DC bias injection feed
to an active microwave device.
2. Description of the Prior Art
It is well known that the operating frequency of an active
microwave device depends not only on the device characteristics and
optimum circuit configuration for the fundamental frequency
f.sub.o, but also on the optimum tuning or idling of harmonic
frequencies nf.sub.o, where n is an integer greater than one.
Simple known filter techniques for bias injection or driving pulse
feed such as the use of disc capacitors or one or more quarter-wave
sections of low surge impedance lines are only adequate for a
limited fundamental tuning range. Regarding quarter-wave section
microwave filters, all of the even harmonics are not prevented from
escaping from the resonator to the outside of the microwave
enclosure causing radio frequency interference and loss in DC to
microwave conversion effeciency. Moreover, the shape of pulses fed
to the microwave device through such filters is seriously affected
by the excessive capacitance of low impedance quarter-wave
sections.
Thus in many cases the conventional low pass filter is not
acceptable for pulse and DC feed applications because the reactive
component of the filter impedance at the location of the junction
of the filter to the microwave resonator has an excessive range and
varies at an excessive rate causing excessive frequency pulling at
certain portions of the frequency band of interest. One type of low
pass microwave filter commonly utilized for such applications is a
coaxial filter consisting of several sections of high impedance
line such as a relatively thin conductor rod or wire surrounded by
air dielectric for simulating series inductances alternating in
combination with short sections of very low impedance line
consisting of conductive discs of various axial thicknesses and
axial spacings which simulate shunt capacitances.
At the junction point between the filter and microwave resonator,
i.e. at the feed point to the microwave circuit to which it is
coupled, a filter should have consistently low resistance and
reactive impedance components in the stop band over the entire
range of operating frequencies including all related harmonics with
major contribution to the device efficiency. At the same time, the
upper frequency components of the spectrum of a driving pulse which
are in the pass band below the operating frequency range should be
affected as little as possible to preserve pulse shape.
For a more detailed treatment of typical prior art low pass filters
for microwave applications, reference can be made to the text
entitled "Microwave Filters, Impedance Matching Networks and
Coupling Structures", by G. L. Mattahei, et al., McGraw-Hill
Company, Inc., New York, N.Y., 1964, at pages 355-359,
inclusive.
SUMMARY
It is an object of the present invention therefore to provide a new
and improved low pass filter for the purpose of feeding power
supply voltages and/or modulating signals such as pulses with all
essential harmonics into a microwave resonator without distortion
or loss. Briefly, the filter is comprised of a length of
transmission line such as but not restricted to a coaxial line
including therein at least two cascaded tapered conductor sections
wherein one of the conductors, for example the inner conductor, in
each section has a preferably linear, tapered length wherein the
electrical length is substantially equal to one half the wave
length of the center frequency of the operating frequency band.
Both sections are substantially identical in configuration and
orientation with the first section acting as a matching section
whose smaller end dimension is coupled to the energy source via the
input port while the second section has its smaller dimension
beginning at and tapering outwardly from the larger end of the
first section. The larger end of the second section terminates
abruptly and couples to a microwave resonator. The two sections
thus constitute two periodically tapered sections of a transmission
line connected in cascade. When desirable, a still third such
section can be added for greater effect.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a longitudinal cross sectional view of a typical prior
art low pass microwave filter with parts shown in elevation;
FIG. 2 is a similar longitudinal cross sectional view of a first
embodiment of the present invention;
FIG. 3 is a longitudinal cross sectional view of a second
embodiment of the subject invention;
FIG. 4 is a perspective view of yet a third embodiment of the
subject invention; and
FIG. 5 is a perspective view of still another embodiment of the
subject invention.
DESCRIPTION OF THE PREFERRED EMBODIMENTS
Prior to discussing the preferred embodiments in specific detail,
apparatus of the type related to the subject invention utilizes
guiding means such as a two conductor transmission line system in
the TEM mode of propagation which may be, for example, a concentric
line transmission system such as shown in FIG. 1 for transmitting
energy between a source and a load at a given frequency and
provides several circuit elements effectively in cascade embodied
in one conductor of the transmission line. These filter sections
present a high impedance at the rejection frequency while
simultaneously causing much less disturbance to the transmission of
energy at the desired frequency. The load impedance accordingly
will, in general, reflect a significant portion of the wave energy
at the undesired frequency.
Directing attention now to FIG. 1, which represents a typical coax
filter illustrative of the prior art, it is shown comprising a two
conductor transmission line including an inner conductor 10 and an
outer conductor 12. A plurality of conductive discs or pucks 14,
16, 18, 20 and 22 of selective axial thicknesses and mutual
spacings are symmetrically arranged along the inner conductor 10.
The diameter of the first and last disc 14 and 22 is of a
relatively smaller diameter than the intermediate discs 16, 18 and
20, which are additionally adapted to support the inner conductor
10 by having diameters which contact the inner surface of a tubular
dielectric spacer member 24 fitted interiorally of the outer
conductor 12.
That the structure of the coaxial filter shown in FIG. 1 is readily
known to those skilled in the art is shown by the fact that such a
device is manufactured and marketed by the Microlab Corporation,
being identified as FXR Model LA-60N. Additionally, a similar
configuration is disclosed as one element in the teaching of U.S.
Pat. No. 3,600,711, issued to Richard Z. Gerlack, on Aug. 17, 1971,
said patent being identified as "Coaxial Filter Having Harmonic
Reflective and Absorptive Means".
The filter shown in FIG. 1 is symmetrical in its configuration and
when one end of the device is terminated for example with a 50 ohm
resistive termination and a signal applied to the other end, the
following typical impedance resistive and reactive values results
as a function of frequency for the filter looking back toward the
signal source from the load end: freq. Resistance Reactance Ohms
+=induc. - (GHz) Ohms -=capac.
______________________________________ 1.0 46.3 -2.0 2.0 39.0 +5.7
PASSBAND 4.0 73.3 +14.0 8.0 3.34 +147.9 10.0 0.21 -322.5 12.0 0.002
-113.5 STOPBAND 16.0 <0.001 -54.2 20.0 <0.001 -33.6 24.0
0.005 -17.2 ______________________________________
It has already been noted that in order for a filter to optimally
perform, it must consistently exhibit a low output impedance, both
resistive and reactive in the stop band over the entire range of
operating frequencies. It can be observed with respect to Table I
that although the resistance exhibited in the stop band is
relatively low compared to the pass band, the reactance components
are relatively large and undergo sudden changes in polarity
resulting in excessive frequency pulling effects. Thus a certain
limitation inherently exists in such a device, even though it is
widely used and has performed adequately under predetermined
operating conditions.
Turning attention now to the subject invention, the embodiments
shown in FIGS. 2-5 disclose an improved approach to low pass
filtering at microwave frequencies. In contrast to prior art filter
apparatus, the filter according to the subject invention is
unsymmetrical about its input and output ports insofar as geometry
is concerned and its performance characteristics as will be shown
does not exhibit the usual rapid transition between the pass band
and stop band. Although this peculiarity may be undesirable in
certain applications where a sharp discrimination between the stop
band and pass band is required, the subject invention is
particularly adapted for feeding power supply voltages and/or
modulating signals, e.g. pulses with all essential harmonics into
microwave loads coupled to the filter without distortion or loss.
Additionally, the microwave frequency band whose center frequency
is defined as f.sub.o and all major harmonics thereof defined as
nf.sub.o where n is an integer greater than one, are confined to
the microwave load circuitry without the introduction of
appreciable losses or tuning discontinuities caused by the rapid
changes of the filter reactance as a function of fundamental or
harmonic frequency components as evidenced in Table I with respect
to the noted prior art device shown in FIG. 1.
Referring now to FIG. 2, there is disclosed a coaxial configuration
of a two section filter according to the subject invention
comprised of a length of two conductor transmission line 25
including an outer conductor 26 and an inner conductor 28 separated
by a cylindrical or tubular dielectric spacer 30. The center
conductor 28 contains two cascaded or series connected tapered
sections 32 and 34 which have respective electrical lengths
substantially one half wavelength .lambda..sub.o /2 of the center
frequency f.sub.o. Defining port A as the output port which is
adapted to be connected to a microwave load circuit, not shown,
port B is defined as the input port which is adapted to be
connected to a DC bias supply or pulse generator, not shown, by
means of a suitable interconnecting cable, also not shown. The
section 32 closest to the signal source i.e. port B acts as an
impedance transformer whereas section 34 closer to the load i.e.
port A acts as the filter section. When desirable, one or more
additional sections can be fabricated and inserted intermediate the
section 34 and port A to enhance the filtering effect. Considering
the geometrical shape of the two sections 32 and 34 shown in FIG.
2, the constant relatively small normally circular dimension of the
inner conductor 28 is interrupted, for example, at the point 36 and
a generally conical shaped enlargement of the center conductor
takes place, enlarging towards port A providing a generally linear
increasing tapered surface of the inner conductor to the point 38
whereupon a generally flat surface 40 having a width dimension
equal to or less than .lambda..sub.o /20 exists for supporting the
section against the spacer 30. Section 32 abruptly terminates in a
generally flat planar surface 42 normal to the central axis
whereupon the second section 34 becomes axially contiguous
therewith having an initial dimension equal to the dimension of the
inner conductor 28. As mentioned above, the length of the section
32, i.e. from one terminal end which is at point 36 to the other
terminal end, i.e. surface 42 is substantially .lambda..sub.o /2 in
length. Both sections 32 and 34 as well as any additional sections
are preferably substantially identical in configuration and
dimensions being typically 0.552 inches long with a periodic
28.degree. taper.
Where for example a variable frequency source is coupled to the
coaxial filter as shown in FIG. 2 by means of a 50 ohm cable, the
following typical impedance values are provided at the output port
A as a function of input frequency: + = induc. freq. (GHz)
Resistance (Ohms) Reactance (Ohms - = capac. 2 sections 3 sections
2 sections 3 sections
__________________________________________________________________________
0.1 41.1 34.0 -18.8 -22.7 0.5 8.3 4.7 -16.8 -11.1 1.0 3.1 2.5 - 7.9
- 2.2 PASSBAND 2.0 3.9 25.1 + 3.0 -26.4 4.0 0.15 0.08 - 7.6 -10.4
8.0 0.009 0.00015 - 2.41 - 2.8 10.0 = f.sub.o Fundamental 0.014
0.00018 - 1.47 - 1.9 12.0 Band 0.052 0.001 - 0.20 - 1.15 STOPBAND
16.0 0.014 0.018 - 1.71 - 7.2 20.0 = 2f.sub.o Harmonic 0.008 0.0001
- 0.46 - 1.0 24.0 Band 0.0748 0. - 6.38 - 0.18
__________________________________________________________________________
It can be seen with reference to Table II that both low resistive
and reactive impedance values are obtained in the stop band
frequencies adjacent the transition region which is between 4 and 8
GHz.
Considering the theory of operation of the embodiment shown in FIG.
2, an equivalent short circuit is effectively presented to
microwaves and their harmonics at port A. This equivalent short
circuit can be explained by consideration of the resistive
component of the impedance value along the tapered sections 32 and
34 from port B to port A. With a 50 ohm pulse generator connected
to port B by means of a 50 ohm coaxial cable, port B can be
considered as terminated with a matched load for microwave
frequencies and their harmonics. This match is preserved in the
first tapered section 32 toward output port A until the first
section ends at the planar face 42. The beginning of the succeeding
section 34 starts off with a relatively high surge impedance
Z.sub.OH in the 50 ohm region, but it is now mismatched by the
relatively low value of the transformed input impedance at face 42
provided by the first tapered section 32 which happens to coincide
with the value of the effective surge impedance, e.g. Z.sub.OL = 1
ohm. Typically the surge impedance Z.sub.O on the tapered section
varies from Z.sub.OH .congruent. 50 ohms to Z.sub.OL .congruent. 1
ohm, but it is not restricted to these values at all. This typical
50 to 1 mismatch continues along the second tapered section 34 and
results in an effective gap impedance of 1/50 = 0.02 ohms in the
first approximation at the output port A for frequencies near
f.sub.o. A third section, not shown, would reduce the resistive
component to a value in the region of 1/50.sup.2 = 0.0004 ohms. The
presence of reactive components complicates this transformation.
Thus low resistive components of the gap impedance at port A occur
over a wide frequency range from 0.5 f.sub.o to well above 4f.sub.o
and provide a low loss short circuit effect at the modulation or
feed bias plane at the input of any microwave circuit connected to
the filter, i.e. port A. Specifically for the embodiment shown in
FIG. 2, the surge impedance Z.sub.O variation from Z.sub.OL to
Z.sub.OH as a function of distance x along each coaxial tapered
section is defined by the equation: ##EQU1## where D.sub.x is the
diameter of the tapered center conductor at the location x along
the respective tapered section 32 or 34, D.sub.i is the constant
inner diameter of the outer conductor 26, and .epsilon..sub.eff is
the relative dielectric constant at the location x. The equations
for the effective surge impedances, however, for the configurations
shown in FIGS. 3 and 4 and for the intrinsic impedance for the
waveguide case in FIG. 5 are different.
The reactive component of the filter impedance of the embodiment
shown in FIG. 2 at the output port A dominates and remains
relatively low in magnitude and uniform in sign over the lower
portion of the stop band frequency range (8.0 - 12.0 GHz) and over
the major harmonics up to the fourth harmonic. This behavior is
significant where the microwave load circuit has a tuning
capability in that the filter maintains the tuning relatively
smooth and the unloaded circuit Q as a function of the ratio of
reactive over resistive filter components relatively high and
substantially constant. Furthermore, the injection of lower than
operating frequencies into the microwave load circuit up to 1/10 of
the center frequency f.sub.o presents no significant problem
because each filter section represents only a very small portion of
the wavelength of pass band frequencies and thus the entire filter
resembles a lumped capacitance consisting essentially of the
effective capacitances of the lower surge impedance portions of the
filter.
It should be pointed out that the linear taper of the section shown
in the embodiment in FIG. 2 has been shown for purposes of
illustration and is most economical to produce. It is not meant to
be considered in a limiting sense since a non linear taper can be
utilized when more advantageous; nor is it necessary that taper
sections be identical in length or surge impedance variation. Also,
the length of the tapers are not necessarily restricted to half
wavelength sections as long as they remain in the vicinity of half
wavelength of the center frequency of the microwave frequency of
operation. Moreover, the ports A and B can be reversed for certain
applications.
Referring now to the other embodiments of the subject invention
which constitute equivalent structures to the configuration shown
in FIG. 2, reference is now made to FIG. 3 which constitutes a
reversed modification of the coaxial filter in that whereas in the
first embodiment the tapered sections were fabricated on the inner
conductor, in the configuration shown in FIG. 3, the tapered
sections are included in the outer conductor of a length of coaxial
transmission line 43. More particularly, the inner conductor 44
having a relatively small constant diameter is covered by a
dielectric sheath 46 and the outer conductor 48 is modified from a
constant diameter transmission line to include at least two
inwardly tapered sections 50 and 52 wherein the inner diameter
gradually reduces as the distances increases away from the input
port B to the outer surface of the dielectric sheath 46 wherein a
relatively small flat surface 53 is maintained to a planar face 54
whereupon an abrupt return to a larger inner diameter exists
providing the beginning of the second section 52. Again the
electrical length of the sections 50 and 52 are preferably
identical being in the order of one half wavelength .lambda..sub.o
/2 of the center frequency. The embodiment shown in FIG. 3 thus
constitutes a reversal of parts, so to speak, from the previous
embodiment.
A stripline version of a filter according to the subject invention
is shown in FIG. 4 and is comprised of a two conductor microstrip
transmission line 55, one conductor of which comprises a metallic
substrate 56 acting as the ground plane while the other conductor
comprises the strip conductor 58 fabricated on a dielectric layer
60. The dielectric layer 60 is contiguous with the metallic
substrate 56. In the stripline configuration, the constant
thickness strip conductor 58 is configured to include two tapered
sections 62 and 64. Whereas in the first embodiment shown in FIG.
2, the tapered sections consisted in solid surfaces of revolution
defining a generally conical shape, the stripline conductor strip
58 consists in a flat layer of conductor material which is suitably
etched or otherwise fabricated to include diverging or widening
portions of the conductor as it proceeds away from the input port B
for a distance substantially equal to .lambda..sub.o /2 whereupon
it abruptly returns to the smaller width dimension at point 57
whereupon it again diverges forming the second section 64. After a
second distance of .lambda..sub.o /2, the second section again
abruptly returns to the smaller width of the strip 58 at point 59
and proceeds to the output port A which is adapted to be coupled to
a microwave load circuit, not shown. This causes the effective
surge impedance to change along the tapered section similar to the
configuration in FIGS. 2 and 3. This embodiment can be further
modified such that the tapered section is sandwiched between two
conductive planes for a match to modern microwave circuitry
utilizing the technique of stripline circuits between two ground
planes.
Directing attention now to FIG. 5, there is disclosed a waveguide
embodiment of the subject invention in which a length of waveguide
66, having a pair of broadwalls 68 and a pair of narrow walls 70,
include two wedge shaped sections 72 and 74 shown for purposes of
illustration being located along the lower broadwall providing
inclined faces 71 and 73 towards the input port B thus providing
sections wherein the dimension of the narrow walls periodically
decreases. The sections 72 and 74 can either be solid material
inserted in and secured to the inside of the waveguide or may
consist of flat plate elements which are inclined upwardly toward
the upper broadwall to a predetermined distance therefrom for
permitting the passage of microwave energy without electrical
breakdown between the broadwall surfaces thus described. Here the
mechanical section length is approximately one half of the
effective guide wavelength .lambda..sub.g.
It should be pointed out that the flat portions of each section at
the end of the taper for the coaxial embodiment shown in FIGS. 2
and 3 are primarily for purposes of support of the inner conductor.
The stripline emodiment shown in FIG. 4 and the waveguide
embodiment shown in FIG. 5 also are illustrated as having a flat
portion at the end of the tapered segment. This is shown merely for
sake of disclosing the similarity of equivalent structures for it
should be appreciated that when desirable, the taper of the
sections of the embodiment shown in FIG. 4 can extend to the abrupt
return to the smaller dimension at the end of the section
contiguous with the beginning of the second section (unless there
is a sharp edge arcing problem in high power applications).
Thus what has been shown and described is an improved low pass
microwave filter comprised of at least two cascaded tapered
conductor elements formed in one of the two transmission lines of a
microwave transmission line for coupling energy from a power source
to a microwave circuit. The filter sections describe periodic
tapers whose length preferably comprises one half the wave length
of the center frequency of the frequency band of operation.
* * * * *