U.S. patent number 3,904,997 [Application Number 05/397,156] was granted by the patent office on 1975-09-09 for trapped-radiation microwave transmission line.
This patent grant is currently assigned to Microwave Associates, Inc.. Invention is credited to Harold Eugene Stinehelfer, Sr..
United States Patent |
3,904,997 |
Stinehelfer, Sr. |
September 9, 1975 |
**Please see images for:
( Certificate of Correction ) ** |
Trapped-radiation microwave transmission line
Abstract
A novel microwave transmission line is described, in which a
conductor strip is supported on one side of a high-dielectric
substrate body, the other side of which is not backed by a
ground-plane conductor. The side of the dielectric body with the
conductor strip is faced toward and spaced a distance from a ground
plane conductor. A channel is provided in the ground plane
conductor, and the dielectric body is in contact with the
conductive material of the ground plane conductor along two paths
at the sides of the channel. The conductor strip is enclosed in the
channel, between the side paths, "suspended" or spaced from the
ground plane body. Radiation from the conductor strip is minimized
by trapping in the channel and in the dielectric body.
Inventors: |
Stinehelfer, Sr.; Harold Eugene
(Burlington, MA) |
Assignee: |
Microwave Associates, Inc.
(Burlington, MA)
|
Family
ID: |
23570053 |
Appl.
No.: |
05/397,156 |
Filed: |
September 13, 1973 |
Current U.S.
Class: |
333/116; 333/235;
333/247; 333/33; 333/238; 333/263 |
Current CPC
Class: |
H01P
3/084 (20130101); H05K 1/0237 (20130101) |
Current International
Class: |
H01P
3/08 (20060101); H05K 1/02 (20060101); H01p
003/08 () |
Field of
Search: |
;333/84M,84R,96,12 |
References Cited
[Referenced By]
U.S. Patent Documents
Other References
Brenner, H. E., "Numerical Solution of Tem-Line Problems Involving
Inhomogeneous Media," MTT-15, 1967, pp. 485-487. .
Glance et al., "A Waveguide to Suspended Stripline Transition,"
MTT-21, 2-1973, pp. 117-118. .
Schneider et al., "Microwave & Millimeter Wave Hybrid
Integrated Circuits for Radio Systems," B.S.T.J., Vol. 48, 1969,
pp. 1703-1713..
|
Primary Examiner: Smith; Alfred E.
Assistant Examiner: Punter; Wm. H.
Attorney, Agent or Firm: Rosen; Alfred H. Steinhilper; Frank
A.
Claims
I claim:
1. An electric wave transmission line comprising a dielectric body
and an electrically-conductive body spaced apart a distance (h) in
fixed relation with respective first surfaces of each confronting
each other and forming in part the enclosing walls of an enclosed
elongated channel of width (S), said conductive body having two
side walls bounding said channel extending toward said dielectric
body into contact therewith along two paths each having a width
(y), an electrical conductor strip of width (W) supported on said
first surface of the dielectric body within and extending along the
channel, the dielectric body having a second surface outside the
channel and opposite to said first surface, which second surface
directly confronts the surrounding region free of any intervening
electric conductor, said width (y) being approximately W/2, said
width (S) being less than half a wavelength at the operating
frequency, and being substantially two to three times greater than
said width (W), said distance (h) being less than the quantity
(S-W)/2.
2. A transmission line according to claim 1 including electrically
conductive closure means across said channel in electrical
connection with said conductive body, said conductor strip
terminating short of said closure means and thereby having an open
end confronting said closure means, an electrically conductive
adjustable means supported on said conductive body movably relative
to said open end for confining the microwave radiation from said
open end substantially to said channel and said dielectric
body.
3. A transmission line according to claim 1 composed of a
trough-shaped conductive body having up-standing side parts and a
substantially planar dielectric body, said dielectric body being in
contact along two spaced-apart paths on said first surface thereof
with the ends of said side parts, said conductor strip being
between said paths.
4. In a transmission line according to claim 3, a gap in one of
said paths, an electrical conductor passing through said gap out of
electrical contact with said conductive body and extending to a
point on said conductor strip, for bringing a bias voltage to said
condutor strip from outside said transmission line.
5. In a transmission line according to claiam 1, at least one gap
in said conductor strip, and an electrically-conductive adjustable
means supported on said conductive body movable adjacent to and
movable relative to each such gap for tuning the gap.
6. In a transmission line according to claim 3, a second conductor
strip on said first surface of said dielectric body in one of said
paths, for making electrical contact with said conductive body, and
supported on at least one of said conductor strips an electric wave
modifying member.
7. A transmission line according to claim 1 including
electrically-conductive closure means across said channel in
electrical connection with said conductive body, said conductor
strip extending into electrical contact with said closure
means.
8. A transmission line according to claim 1 having two electrical
conductor strips supported on said first surface of said dielectric
body within and extending in spaced-apart relation along said
channel, in a wave-coupling to each other.
9. A transmission line according to claim 8 in which at least two
bridging conductors connecting said conductor strips are also
supported on said first surface, said bridging conductors being
spaced apart along said channel.
10. A transmission line according to claim 1 in which said
conductive body is made of thin metal, and elongated flat sheets of
metal each having a width Y at least approximately W/2 are fitted
one to each of said ends of said side walls.
11. A transmission line according to claim 1 in which said
conductive body is a rigid block of metal containing said channel,
said side walls having said width Y that is at least approximately
W/2.
12. A transmission line according to claim 1 in which the thickness
d of said dielectric body is at least W/3.
13. In a transmission line according to claim 1, diode means in the
space between said dielectric body and said conductive body, and
means connecting said diode means between said conductor strip and
said conductive body in the vicinity of one of said paths.
14. In a transmission line according to claim 1, a stub
transmission line section connected between said conductor strip
and said conductive body in one of said paths, said section having
a conductor supported on said surface of said dielectric body and
connected at one end to said conductor strip.
15. In a transmission line according to claim 1, passage means
through one of said paths, bias conductor means in said passage
means, a reactive connection from said bias conductor means to said
conductor strip, and a capacitive connection from said bias
conductor means to said conductive body in the vicinity of said one
path.
Description
BACKGROUND OF THE INVENTION
The use of microstrip transmission line, generally in the form of
an electrical conductor formed, as by etching on high dielectric
permittivity substrate material, has become widely recognized in
the design of microwave transmission circuits and integrated
circuits, subassemblies and components. A basic or common
microstrip structure consists of a conductor strip or other circuit
pattern supported on one side of a dielectric substrate backed on
the other side by an electrically-conductive ground plane. The
dielectric substrate is usually alumina ceramic because of its high
dielectric constant and low loss tangent although other dielectric
materials are also used. Often the microstrip circuits are
completely enclosed and sealed in an electrically-conductive
enclosure. Various forms of microstrip, strip line, and other
transmission lines are illustrated in FIG. 1 of the article of E.
G. Cristal et al., "MICROGUIDE - A NEW INTEGRATED CIRCUIT
TRANSMISSION LINE" - GMTT May 1972 pages 212-214.
The features of ceramic-based microstrip which led to its
popularity in microwave integrated circuit (MIC) applications
are:
1. its fairly high effective dielectric constant and the consequent
miniaturization of microwave circuits;
2. the ease, reproducibility and economy of producing circuits by
photo etching methods;
3. the ease of mounting unpackaged semiconductor devices directly
on the deposited conductors; and
4. its "open" construction which permits probing and adjustment of
the circuit while it is operating.
Microstrip does have a couple of drawbacks, however, which limit
its applicability. First, its attenuation is relatively high,
compared to coaxial line or stripline. Second, microstrip has a
tendency to radiate RF energy at discontinuities.
The high loss is a consequence of the high effective dielectric
constant. (The effective dielectric constant is a weighted average
of the air and ceramic dielectric constants, which properly
accounts for the decrease in wave phase velocity and line impedance
from the values in air). The high dielectric constant concentrates
the electric energy within the dielectric, enhancing the dielectric
loss, while the small size of the conductor strip raises the
current density, enhancing the "copper" losses.
Because of its open, unbalanced configuration, microstrip tends to
radiate RF power from discontinuities where higher order modes are
excited. Open circuits and high-Q resonators are particularly
troublesome. The radiation can be reduced by increasing the
dielectric constant, which better confines the fields to the
dielectric, but this is done at the cost of increased attenuation.
Radiation into free space can, of course, be eliminated by
enclosing the microstrip in a shielded box, as is common practice.
Nevertheless, the excitation of higher order modes by
discontinuities is not changed by the shielding, and the tendency
toward radiation becomes manifested as increased coupling or
crosstalk among the various circuit elements in the enclosure.
Furthermore, the transverse dimensions of the enclousure should be
less than a half wavelength long to avoid the excitation of box
resonances. Since most circuit functions require circuits somewhat
greater than a quarter wavelength on a side, it is often difficult
to satisfy this criterion without resorting to elaborately shaped
enclosures and circuit substrates.
Suspended substrate microstrip line was introduced as aa way of
decreasing the effective dielectric constant and, thereby, the
loss. Suspended substrate (SS) has a thin strip of conductor
deposited on a dielectric substrate, as in microstrip, but the
substrate is placed or "suspended" nearly equidistant between two
ground planes. The strip may be on either side of the dielectric.
SS has relatively little electric energy stored in the dielectric
compared to microstrip; it thus has a lower dielectric loss and
wider, lower loss strips for a given impedance level. Also, because
of the reduced effect of the dielectric on the wave propagation in
SS lines, the tolerance on the dielectric material properties --
its thickness, dielectric constant, uniformity and surface finish
-- which are rather critical in microstrip, are considerably
relaxed.
Of course, SS line does not alleviate the radiation and box
resonance problems of microstrip. In fact, because of the lower
effective dielectric constant, SS line has an even greater tendency
to radiate and a shielded, narrow enclosure is essentially
mandatory.
GENERAL NATURE OF THE INVENTION
The present invention provides a novel transmission line having a
channel between a dielectric body and a ground plane conductor body
that are spaced a distance apart, there being a conductor strip
supported on that surface of the dielectric body which faces the
ground plane conductor body, the conductor strip being thereby
suspended a distance from the ground plane conductor. The remaining
surface of the dielectric body, outside the channel, is not backed
by a ground plane conductor. The channel is provided in the ground
plane conductor, and the dielectric body is in contact with the
conductive material of the ground plane conductor along two paths
at the sides of the channel which paths are generally parallel to
and spaced from the conductor strip. The width of each path of
contact is related to the width of the conductor strip such that
radiation from the transmission line, e.g.: from discontinuities,
is minimized by trapping the electric fields in the channel and in
the dielectric material. For convenience, this novel transmission
line may be referred to as trapped-radiation microstrip
transmission line.
The invention improves upon the radiative properties of ordinary
microstrip without paying the loss penalty associated with a higher
effective dielectric constant. A line according to the invention
will have a dielectric constant between those characterizing the
equivalent microstrip and suspended substrate lines. Like a
suspended substrate line, it will have low loss and relaxed
tolerances to dielectric material properties. On the other hand,
the fields will be largely confined to the channel region adjacent
the conductor strip by virtue of the high dielectric constant in
the fringing field zone. This "trapping" of the fields in the
channel reduces the coupling to the free space outside the channel
or "above" the substrate, significantly reducing the excitation of
radiation or box resonances by circuit discontinuities. The fact
that a wave propagating in this line gets trapped in the channel
causes the wave to propagate in a uniform manner over longer
distances than in the case of microstrip or suspended substrate
lines.
In addition to maintaining simultaneously low radiation and low
loss, the transmission line according to the invention possesses
several other advantages and useful features, some of which will be
described in detail. It is an attractive transmission line medium
for use in microwave integrated circuit technology, for
example.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is an isometric sketch of a preferred embodiment of the
invention;
FIG. 2 is a section line 2--2 of FIG. 1;
FIG. 3 is a section of another embodiment of the invention;
FIG. 4a is a representation of a typical electric field pattern in
the embodiment illustrated in FIG. 2;
FIG. 4b is a representation of a typical magnetic field pattern in
the embodiment illustrated in FIG. 2;
FIG. 5 is a set of graphs showing typical characteristic impedance
of microstrip and suspended-substrate transmission lines, and a
trapped-radiation transmission line according to the invention.
FIG. 6 is a graphic illustration of the impedance of a transmission
line according to FIGS. 1 and 2 as a function of eccentricity of
the strip conductor therein;
FIG. 7 illustrates a shorted stub in a transmission line according
to FIGS. 1 and 2;
FIG. 8 illustrates structure for incorporating shuntmounted devices
and bias feed-through conductors in a transmission line according
to FIGS. 1 and 2;
FIG. 8a is an enlarged partial cross-section taken along line
8A--8A in FIG. 8 with the parts 14 and 11 closed on each other;
FIG. 9a illustrates the incorporation of a capacitive tuning screw
in a transmission line according to FIGS. 1 and 2;
FIG. 9b illustrates a series capacitive gap in the strip conductor
of a transmission line of the invention;
FIG. 9c is the equivalent circuit of FIG. 9b;
FIG. 10 illustrates a structure for launching wave energy from a
coaxial line into a trapped-radiation transmission line of the
invention;
FIG. 11 is a longitudinal top-section through an embodiment of the
invention which incorporates in one structure a branch-line hybrid
with off-center conductors;
FIG. 12a is a longitudinal top-section through a coupled-strip
coupler employing trapped-radiation transmission line according to
the invention;
FIG. 12b is a section on line 12b--12b FIG. 12a;
FIG. 13a is a longitudinal side-section through an open stub end of
a transmission line according to FIGS. 1 and 2;
FIG. 13b is a top view on line 13b--13b of FIG. 13a;
FIG. 14a is a longitudinal side-section through a "shorted" stub
end of a transmission line according to FIGS. 1 and 2; and
FIG. 14b is a top view on line 14b--14b of FIG. 14a.
DETAILED DESCRIPTION OF THE DRAWINGS
In FIGS. 1 and 2, a strip 10 of electrical conductor is supported
on a surface 12 of a dielectric substrate body 11, in the same
manner as in a microstrip transmission line. The dielectric body 11
is affixed in paths 11a and 11b along the edges of the surface 12
to confronting surfaces 14a and 14b respectively of an electrical
ground-plane conductor 14. The ground plane conductor 14 has a
channel 13 in it, bounded at its bottom by a bottom wall 15 and at
its sides by walls 15a, 15b extending to the surfaces 14a and 14b,
respectively. The channel 13 is an elongated passage of rectangular
cross-section, bounded on the bottom and two sides by electrical
conductor walls 15, 15a and 15b, and on the top by the surface 12
of the dielectric body 11. The strip conductor 10 is suspended from
the dielectric surface 12 spaced from the bottom wall 15 of the
channel 13.
The relative dimensions of the components in FIGS. 1 and 2 are as
follows:
width of conductor 10 = W width of paths 11a-14a and 11b-14b = W/2
(approx.) width of space between each side-wall (inner) of sides
15a, 15b and the nearer longitudinal edge of conductor 10 = X width
of channel 13 (W + 2X) = S thickness of dielectric body 11 = d
thickness of channel 13 = h
In a practical transmission line having Z.sub.o = 50 ohms d >
W/3; W/h = 3 (approx.); S < .lambda./2 at fmax; and X = 2.2h
(approx.)
FIG. 3 is a modification of FIG. 2 in which the dielectric body 11
and strip conductor 10 are the same, but the ground plane member 18
is structurally different, being made of a thin metal channel
having a bottom wall 19 and up-standing thin sidewalls 19a and 19b.
At the top edge of each sidewall 19a, 19b is a flat electrically
conductive plate 20, 21, respectively, of width Y on each of which
the meeting path surface 14a, 14b, respectively, for the dielectric
member 11 is formed. The portion 20a, 21a of each plate 20, 21,
respectively, that extends inward from the sidewalls 19a, 19b,
respectively, should preferably be less than W/4 in width. The
channel 13' is bounded by the electrically conductive walls 19, 19a
and 19b, and the surface 12 of the dielectric body 11.
The dielectric body and the ground plane body may be joined
together in any fashion known to the art, now or hereafter. For
example, metalized strips (16a, 16b) like the conductor 10 can be
laid down on the dielectric member in the paths 11a, 11b.
FIGS. 4a and 4b illustrate typical field patterns of
trapped-radiation transmission lines according to the invention.
These are, respectively, computer-generator plots of the electric
and magnetic fields in a line according to FIGS. 1 and 2 but, the
lines being symmetrical about a longitudinal centrally-located
plane between the side walls of the channel, these figures show
only one half of the transmission line, each being bisected along
the axis of symmetry. The magnetic field pattern (FIG. 4b) is
equivalent to the electric equipotential pattern, and similarly the
electric field pattern (FIG. 4a) is equivalent to the magnetic
equipotential pattern. The patterns were computed by the finite
element program for solving the two-dimensional LaPlace's equation
developed by Sylvester and coworkers at McGill University, (Z.
Csendes and P. Sylvester, "Dielectric Loaded Waveguide Analysis
Program", IEEE Trans. Microwave Theory and Techniques, Volume
MTT-19, p. 789, Sept. 1971; and Z. Csendes and P. Sylvester, "A
Finite-Element Field-Plotting Program", IEEE Trans. MTT, Vol.
MTT-20, p. 294, April 1972). It will be seen that the electric
field lines in FIG. 4a and the magnetic field lines in FIG. 4b
exhibit a high degree of trapping in the channel 13 and in the
dielectric 11, particularly in the portion adjacent the coupling
path 11b- 14b at the side wall 15b.
Comparison with similarly-derived field patterns for micro-strip
line and suspended substrate line, all three lines being in the
same-size enclosure, and employing dielectrics having the same
dielectric constant and physical thickness, and strip conductor
widths chosen to correspond to substantially the same line
impedance, demonstrates that field trapping in lines according to
the invention is greater, and therefore radiation is more greatly
restricted than in the prior lines mentioned. Thus, it is found
that in all three kinds of transmission line most of the electric
field is concentrated in the dielectric. This proportion is greaten
in micro-strip, less in suspended substrate, and intermediate in
the trapped-radiation line of FIGS. 1 and 2. However, when one
considers the electric field above the dielectric substrate (i.e.:
at the side opposite the side bearing the strip conductor 10) the
trapped-radiation line of the present invention has the weakest
fields. The difference is dramatic relative to micro-strip, and
less marked but still significant relative to suspended substrate.
The suspended substrate field in this region is only slightly
greater than in the line according to FIGS. 1 and 2, but it falls
off less rapidly as one moves off the center line. Thus can be
seen, at least qualitatively, the reduced penetration of electric
fields into the air above the substrate, for example, the substrate
11 in FIG. 2. Lateral confinement of the fields by the channel 13
is quite dramatic, there being no equivalent structure in either of
the microstrip or suspended substrate lines. The invention attains
drastic lateral confinement in the channel 13 even though the
effective dielectric constant may be only about half that of
microstrip line.
In addition to reducing the lateral extension of fields, the
trapped-radiation line channel (13, 13'), together with the
dielectric interface, i.e., outer surface of the body 11, above the
line deflect the fields back to the channel edges, effectively
eliminating or greatly reducing the radiative coupling to the
region above, or outside, the dielectric substrate. It is possible
to increase the effective dielectric constant of the
trapped-radiation lines of the invention, e.g.: to 4 or 5, by
increasing the substrate thickness or its dielectric constant, and
thereby further reduce the radiation while keeping the loss below
that of microstrip.
In FIG. 5, curve 31 shows the measured characteristic impedance
Z.sub.o of a set of trapped-radiation transmission lines according
to FIGS. 1 and 2, in which the strip conductor 10 and widths W as
follows:
0.020 inch
0.040 inch
0.060 inch
0.080 inch, and
0.100 inch
In each case the conductor was on a dielectric substrate 11 of
alumina having thickness d = 0.020 inch and dielectric constant
.epsilon..sub.r = 9.0; and the depth h of the channel 13 was 0.020
inch. Each channel had width S that was 0.090 greater than W.
Except for channel width S, the geometry corresponds to that of
FIGS. 4a and 4b, where W was 0.050, and S was 0.150 inch. In FIG.
5, Z.sub.o is plotted as a function of the ratio W/h, and the
characteristic impedance is seen to fall between 80 ohms and 40
ohms, being 50 ohms when W/h = 3 (approx.).
Curve 32 (in dashed-line) shows for comparison the impedance of
suspended substrate line on the same dielectric, as estimated from
the curves of H. E. Brenner "Use a Computer to Design Suspended
Substrate IC's", Microwaves, Vol. 7, No. 9, p. 38, Sept. 1968. FIG.
5 illustrates that the characteristic impedance for these two lines
is closely similar.
Curve 33 shows for comparison the characteristic impedance of
microstrip line, as calculated from Wheeler's formulas, (See M. V.
Schneider, "Microstrip Lines for Microwave Integrated Circuits",
Bell Sys. Tech. Journal, Vol. 48, p. 1421, May-June 1969).
Trapped-radiation lines according to the invention have four
impedance-determining parameters, as is apparent from FIG. 5. These
are:
W/h; d/h; S/h; and .epsilon..sub.r
By contrast, suspended substrate has only three such parameters,
and microstrip has only two. These additional variable parameters
make it possible to develop transmission lines having a wide
variety of specifications. The channel width S can be increased
until the trapping effect is lost, or decreased until the width W
of the strip conductor 10 must be so narrow that losses are
unacceptably high and mechanical tolerances become unacceptably
close. The dielectric can be made thinner to the point where
leakage flux and radiation become unacceptably high, or thicker to
the point where effective dielectric constant and consequent loss
become unacceptably high and the possibility of box resonance
entirely within the dielectric is introduced or becomes
unacceptable. In general, acceptable ranges of dimensisons have
been indicated above, in connection with FIGS. 1 and 2, and FIG. 5,
for frequencies up to 18 GHz.
Since there is no inherent reason for the strip conductor 10 to run
along the center line of the channel 13, the offset, or
eccentricity, .DELTA. of this conductor is a variable parameter
which offers one more degree of freedom in designing transmission
lines according to the invention. FIG. 6 shows the measured effect
of offsetting a 50 ohm strip conductor up to half the channel
width. The characteristic impedance is plotted as a function of the
offset .DELTA., and curve 35 shows that Z.sub.o can be varied,
essentially linearly, from 50 ohms to less than 47 ohms. FIG. 6
illustrates the comparatively relaxed tolerances that are possible
in trapped-radiation lines of the invention. For typical channels,
that is, those which have width S about 2 to 3 times the width W of
the strip conductor 10, the lateral positioning tolerance of the
conductor is not critical; a 15% offset causes only a one-ohm error
in a 50 ohm line. Similarly, the channel width itself is not
critical. The strip conductor having width W = 0.060 inch was 50
ohms in a channel having width S = 0.150 inch, and approximately 49
ohms in a channel having width S = 0.130 inch.
Trapped-radiation lines according to the invention have many
advantages over microstrip and suspended substrate lines. For
example, short-circuited stubs are rarely used in microstrip
because they are inconvenient to make. In order to return the open
end of a stub to ground, either it must be positioned at the edge
of the dielectric board (or substrate), or a hole must be provided
through the board and a separate grounding connection must be made.
On suspended substrate, one does not have even the option to drill
a hole through the board to the ground plane. By contrast, a short
circuit in lines according to the invention can be simply printed
(e.g.: etched out) with the rest of the circuit, as is illustrated
in FIG. 7, where parts correspond to parts in FIG. 2 have the same
reference characters. The stub conductor 41 runs from the main line
conductor 10 to the edge of the channel 13, which is widened in the
region 13a in the vicinity of the stub to accommodate its length.
The contact paths 11a and 11b of the dielectric body 11 are
metallized in the same manner as the conductor 10, in a common
printing or etching operation, for ease in making fixed electrical
contact to the meeting path surfaces 14a, 14b of the ground plane
member 14. Since shorted stubs in microstrip are usually simulated
by an extra quarter wavelength of open stub, they do not have the
same frequency response of true shorted stubs, and they radiate. By
contrast, a shorted stub as shown in FIG. 7 has none of these
deficiencies, giving it a clear advantage in such circuit
applications as interdigital and comb-line filters. This shorted
stub also offers the possibility of a sliding short by filling the
widened channel region 13a with a movable metal block (not
shown).
The advantage of the invention that makes it an easy matter to
fabricate shorted stubs also makes it easy to shunt-mount discrete
circuit elements, such as semiconductors, resistors, capacitors,
directly on the dielectric substrate 11. FIG. 8 (and FIG. 8a)
illustrates several possible arrangements, which can be used
together or in vairous combinations. The main line conductor 10 and
one of the metallized meeting paths 11a can be fitted with
confronting short contact tabs 43, 44, and a diode 45 can be
mounted on one of these tabs 44. A conductor 46 can then be
connected from the diode to the other confronting tab 43, thereby
connecting the diode in shunt from the main line conductor 10 to
ground. A slot 48 may be provided through the upper surface 14b of
the wall 15b of the ground plane conductor 14, to accommodate a
bias feed-through conductor 49 across the path 11b, the metallizing
of which is interrupted for this purpose. The feed-through
conductor connects to a long thin conductor 51 which connects at
its remote end to the main line conductor 10, and has a length
parallel to that conductor sufficient to provide a choke at the
operating microwave frequency. This is a bias choke, and a by-pass
capacitor 52, which may for example be an RF by-pass semiconductor
chip capacitor, is connected between the feed-through conductor 49
and ground via the ground-contact path 11b metallizing, to complete
a bias network.
FIG. 9 illustrates an arrangement, which is easily possible in the
present invention, that provides the ability to do final tuning of
a microwave circuit with a screwdriver, using a non-radiating screw
61 accessible from outside the finished transmission line
structure. Again, parts in common with FIG. 2 bear the same
reference characters, FIG. 9a being a longitudinal section through
an embodiment like FIG. 2. A gap 62 is provided in the main
conductor 10, and the screw 61 is threaded through the bottom 15 of
the ground plane member 14, where it is located precisely with
reference to the gap. This is a capacitive screw, which is
grounded, and with it the coupling of a series gap 62 can be
varied, as shown. Similarly the effective length of an open stub
(see FIG. 13) can be varied. In either case, the radiation trapping
properties described above will be found to be effective.
The series gap 62 and its equivalent circuit are illustrated in
FIGS. 9b and 9c, respectively. The equivalent circuit consists of a
series capacitive susceptance B.sub.b and two shunt susceptances
B.sub.a, in a .pi. network. Two sets of reference planes P.sub.1
P.sub.2 and T.sub.1 are shown for this circuit. For narrow gaps it
is convenient to use a single reference plane through the center
line T.sub.1 of the gap, and for this reference the shunt
capacitors are negative to account for missing line capacitance.
For wide gaps 62 it is more accurate to consider a pair of
reference planes P.sub.1 and P.sub.2 at the ends of the gap. Here,
while the same equivalent .pi. network applies, the shunt
capacitors are positive to account for fringing at the open ends.
In the limit, of course, the gap may be treated as infinitely wide,
and then it is conventianal to treat the open circuit capacitance
as an equivalent length of line and to define a corresponding line
length correction.
A series of gaps 62, each with its tuning screw 61, appropriately
spaced along the main line conductor 10, will provide a gap
filter.
Transition from a coaxial line to a trapped-radiation line of the
invention may follow the technique used with microstrip, and is
illustrated in FIG. 10. A coaxial line 61 is represented by its
inner and outer conductors 62, 63, respectively, the inner
conductor being connected to the main line conductor 10 of the
trapped-radiation line, and the outer conductor being connected to
the ground plane member 14. Test models made according to FIG. 11,
in which the trapped radiation line of the invention was designed
to have Z.sub.o = 50 ohms, and in which the coaxial line was used
to launch microwave energy into the new line, indicated VSWR per
transition to be about 1.2 at X-Band, and 1.4 at 18.0 GHz, prior to
any development work aimed at reducing the VSWR.
The branch line hybrid coupler is often used for 3dB couplers since
it affords the desired tight coupling. FIG. 11 illustrates a
circuit configuration that uses a pair of off-center main line
conductors 70, 71 mounted on a dielectric substrate 74 and
suspended in a common channel 72, the channel being formed in a
ground-plane member (not shown) which corresponds to the
ground-plane member 14 in FIGS. 1 and 2, and has side walls 73, 73.
In FIG. 11, the main line conductors 70 and 71 are connected by
branch line conductors 75, 76, which are formed with the main line
conductors on the substrate 74. At higher frequencies, when the
diameter of the hybrid ring can become so small that the coupled
circuits may be effectively merge together, the coupler is reduced
to a suspended substrate circuit and undesired coupling across the
ring will degrade performance. The provision of an island-like
conductive element 78, projecting from the bottom of the
ground-plane member (not shown) into the center of the hybrid ring
will reduce such undesired coupling at all higher frequencies.
Additional islands 77 and 79 may be provided between the main lines
70, 71, as shown, to further enhance the reduction of undesired
coupling.
A simple arrangement of a pair of coupled strips (or main line
conductors) 80, 81 is illustrated in FIGS. 12a and 12b. This
arrangement is useful for directional couplers, as illustrated, and
for half wave resonator filter circuits, to cite a few examples.
The coupled strips are located closely spaced in the same channel
13 of FIGS. 1 and 2, for example. Because the lateral confinement
of the electric and magnetic fields by the channel provides
excellent isolation among various circuits that may be formed on
the same substrate 11, the feed lines 82, 83, 84 and 85 to the pair
of coupled strips 80, 81 can be easily, simply and cleanly
separated from each other. Each feed line is located in its own
channel 82', 83', 84' or 85', respectively, thereby helping to
reduce parasitic reactances and undesired coupling between the two
lines. By properly proportioning the thickness of the dielectric 11
and the depth of the channel 13 at the coupled-line pair 80, 81, it
is also possible to equalize the phase velocities of the even-and
odd-mode waves. This has the advantage of making broader-band
higher directivity couplers easier to achieve in lines according to
the invention than in microstrip.
FIGS. 13a and 13b show how to make an open stub in a transmission
line according to the invention. A section of FIG. 13a taken along
line 2--2 will look identical to FIG. 2. The channel 13 is
terminated or blocked at one end 13.5 and the main line conductor
10 terminates short of that end. A non-radiating turning screw 61,
like the same component in FIG. 10a, serves to capacitively
terminate the stub. Trapping lines 101, 102 show how the dielectric
member 11 and the screw 61 (which is grounded) reduce radiation
from the free end of the stub.
FIGS. 14a and 14b show how to make a closed stub. The technique is
similar to that in FIGS. 13a and 13b, except that the top surface
91 of the end wall 13b obstruction is fitted with a groove 90, and
the main line conductor extends into that groove to make electrical
contact with the ground-plane member 14.
The embodiments of the invention which have been illustrated and
described herein are but a few illustrations of the invention.
Other alternative circuit arrangements may be made within the scope
of this invention by those skilled in the art. No attempt has been
made to illustrate all possible embodiments of the invention, but
rather only to illustrate its principles and the best manner
presently known to practice it. Therefore, while certain specific
embodiments have been described as illustrative of the invention,
such other forms as would occur to one skilled in this art on a
reading of the foregoing specification are also within the spirit
and scope of the invention.
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