U.S. patent number 3,890,537 [Application Number 05/430,088] was granted by the patent office on 1975-06-17 for solid state chopper ballast for gaseous discharge lamps.
This patent grant is currently assigned to General Electric Company. Invention is credited to John N. Park, Steven C. Peak, Robert L. Steigerwald.
United States Patent |
3,890,537 |
Park , et al. |
June 17, 1975 |
Solid state chopper ballast for gaseous discharge lamps
Abstract
A single phase, high frequency, transistor or gate turnoff
thyristor chopper ballast circuit especially suited for mercury
vapor lamps preferably operates on a unfiltered full wave rectified
line voltage and electronically shapes the lamp current and
therefore the line current to obtain a high power factor. The
ballast circuit is lightweight with low volume due to elimination
of large low frequency energy storage, filtering, and transformer
components. The forced, high frequency ripple lamp current
waveshape, achieved by comparison of the sensed current with an
appropriate reference signal, provides for good regulation, an
initially high starting current to eliminate glow-to-arc mode,
automatic sweeping of the chopping frequency to avoid acoustic
resonance effects, and a minimum current in the valley regions of
the supply voltage for improved reignition characteristics.
Inventors: |
Park; John N. (Rexford, NY),
Peak; Steven C. (Schenectady, NY), Steigerwald; Robert
L. (Scotia, NY) |
Assignee: |
General Electric Company
(Schenectady, NY)
|
Family
ID: |
23706015 |
Appl.
No.: |
05/430,088 |
Filed: |
January 2, 1974 |
Current U.S.
Class: |
315/208;
315/DIG.7; 315/247; 315/307; 315/209R; 315/224; 315/287 |
Current CPC
Class: |
H02M
1/4208 (20130101); H05B 41/28 (20130101); Y02B
70/10 (20130101); Y10S 315/07 (20130101) |
Current International
Class: |
H02M
1/00 (20060101); H05B 41/28 (20060101); H05b
041/24 (); G05f 001/08 () |
Field of
Search: |
;315/2R,207,208,224,287,283,307,DIG.5,DIG.7,29R |
References Cited
[Referenced By]
U.S. Patent Documents
Primary Examiner: Demeo; Palmer C.
Assistant Examiner: LaRoche; E. R.
Attorney, Agent or Firm: Campbell; Donald R. Cohen; Joseph
T. Squillaro; Jerome C.
Claims
What is claimed is:
1. A solid state ballast circuit for gaseous discharge lamps
comprising
a solid state chopper circuit for energization by low frequency
alternating-current line voltage and line current and including
high frequency filter means for supplying sinusoidal power voltage
between a pair of supply terminals and controlled switching means
and coasting device means effectively coupled to said supply
terminals and coasting inductor means to be conductive alternately
to supply lamp current through said coasting inductor means to a
gaseous discharge lamp,
current sensor means coupled to sense the instantaneous lamp
current and produce a sensor signal indicative thereof,
a control circuit comprising generating means for generating a
reference signal with a preselected waveshape and magnitude to
determine the power level and to effect shaping of said lamp
current and therefore the line current to obtain a high power
factor, comparing means for effectively comparing said sensor and
reference signals and producing an output signal, and means
actuated by said output signal for supplying turn-on and turn-off
signals to operate said controlled switching means at a variable
high frequency chopping rate and shape said lamp current as
determined by said reference signal,
said control circuit further comprising means for temporarily
shaping and increasing the magnitude of said reference signal at
start-up to obtain a high starting lamp current, and
said control and chopper circuits further comprising means for
supplying minimum lamp current for good reignition characteristics
during the low voltage regions of said line voltage in each cycle
when said comparing means is ineffective to shape the lamp
current.
2. A solid state ballast according to claim 1 additionally
including an adaptive control circuit connected with said
generating means to further shape said reference signal according
to a selected control.
3. A solid state ballast according to claim 1 wherein said
generating means is comprised by a transformer energized by the low
frequency line voltage for deriving a control voltage, and control
function generator means for regulating and shaping said control
voltage according to a predetermined control function to produce
said reference signal,
said control function generator means further including said means
for temporarily shaping said reference signal at start-up to obtain
a high starting lamp current.
4. A solid state ballast according to claim 3 wherein said
comparing means includes a comparator with hysteresis, and said
comparator has a series pass transistor regulator low voltage power
supply energized by said transformer which supplies regulated
clipped voltage to said comparator except during the low voltage
regions of the line voltage.
5. A solid state ballast according to claim 1 wherein said
generating means is comprised by means energized by the low
frequency line voltage for deriving a control voltage in phase with
the line voltage, and control function generator means for shaping
said control voltage according to a predetermined control function
to produce said reference signal,
said control function generator means further including said means
for temporarily shaping said reference signal at start-up to obtain
the high starting lamp current.
6. A solid state ballast according to claim 5 wherein said means
for temporarily shaping said reference signal at start-up is
provided by a long time constant resistor-capacitor network for
temporarily modifying operation of said control function generator
means.
7. A solid state ballast according to claim 6 wherein said
comparing means includes a comparator with hysteresis, and said
comparator has low voltage power supply means energized by said
means for deriving a control voltage which supplies regulated
voltage to said comparator except during the low voltage regions of
the line voltage.
8. A solid state ballast circuit for gaseous discharge lamps
comprising
a solid state chopper circuit for energization by low frequency
alternating-current line voltage and line current and including
full wave rectifying means and high frequency filter means for
supplying rectified sinusoidal power voltage between a pair of
unidirectional voltage supply terminals, and further including a
power transistor and coasting diode connected in series between
said supply terminals that conduct alternately and supply lamp
current through a coasting inductor and gaseous discharge lamp that
in turn are connected in series across said coasting diode,
current sensor means coupled to sense the instantaneous lamp
current and produce an instantaneous sensor signal indicative
thereof,
a control circuit comprising a transformer and a first bridge
rectifier energized by the low frequency line voltage for
generating full wave rectified sinusoidal control voltage, a
control function generator circuit for shaping said control voltage
and generating a symmetrically curved reference signal with a
waveshape and magnitude selected to determine the power level and
to effect shaping of the lamp current and therefore the line
current to obtain a high power factor in excess of 90 percent, a
comparator circuit with hysteresis for effectively comparing said
sensor and reference signals and producing an output signal, a
second bridge rectifier connected to said transformer for deriving
full wave rectified sinusoidal base drive power supply voltage, a
positive and negative base drive circuit connected to said second
bridge rectifier for supplying alternate turn-on and turn-off
signals to said power transistor with a base current proportional
to collector current, and means for coupling said comparator output
signal to energize said negative base drive circuit and de-energize
said positive base drive circuit to thereby operate said power
transistor at a variable high frequency chopping rate and effect
shaping of the lamp current as determined by said reference signal
waveshape,
said control circuit further comprising means for temporarily
shaping and increasing the magnitude of said reference signal at
start-up to obtain a high starting lamp current, and
said control and chopper circuits further comprising means for
supplying minimum lamp current for good reignition characteristics
during the valleys of the rectified sinusoidal power voltage when
said comparator circuit is ineffective to shape the lamp
current.
9. A solid state ballast according to claim 8 wherein said
comparator circuit has a low voltage power supply circuit energized
by said transformer that is operative to clip the full wave
rectified control voltage at a selected low voltage level and
supply power to said comparator circuit except during the valleys
of the rectified sinusoidal power voltage.
10. A solid state ballast according to claim 8 wherein said chopper
circuit has a pair of input terminals and said high frequency
filter means includes a shunt capacitor and series inductor
connected between said input terminals and full wave rectifying
means, and a polycrystalline varistor connected between the input
terminals of said full wave rectifying means, said transformer
likewise having a primary winding connected between the input
terminals of said full wave rectifying means, to thereby provide
filtering and protection for both the control circuit and chopper
circuit.
11. A solid state ballast according to claim 8 wherein said
comparator circuit has at least one series pass transistor
regulator low voltage power supply circuit energized by said
transformer that is operative to clip the full wave rectified
control voltage at a selected low voltage level and supply power to
said comparator circuit except during the valleys of the rectified
sinusoidal power voltage.
12. A solid state ballast according to claim 8 wherein said control
function generator circuit is comprised by a resistive voltage
divider circuit having a variable resistance branch including and
controlled by an insulated-gate field effect transistor, and
said means for temporarily shaping said reference signal at
start-up to obtain a high starting lamp current is provided by a
long time constant resistor-capacitor network connected to the gate
of said field effect transistor for temporarily determining the
gate voltage and therefore the resistance of said field effect
transistor. l
13. A solid state ballast according to claim 8 wherein said control
function generator circuit is comprised by a resistive voltage
divider circuit having a variable resistance branch including and
controlled by an insulated-gate field effect transistor, and
further comprises a control voltage peak detector circuit connected
to the gate of said field effect transistor for determining the
gate voltage and therefore the resistance of said field effect
transistor to thereby provide automatic gain control of said
reference signal to regulate the lamp current,
said peak detector circuit having a long time constant to thereby
provide said means for temporarily shaping said reference signal at
start-up to obtain a high starting lamp current.
14. A solid state ballast according to claim 8 wherein said
positive base drive circuit is connected to said second bridge
rectifier to be normally conductive in the absence of said
comparator output signal, and
said means for supplying minimum lamp current during the valleys of
the rectified sinusoidal power voltage includes local energy
storage capacitors coupled to said second bridge rectifier and to a
reference point that discharge to provide base current to maintain
conductivity of said power transistor during the power voltage
valleys, said high freguency filter means including a filter
capacitor connected between said unidirectional voltage supply
terminals which supplies lamp current during the power voltage
valleys.
15. A solid state ballast according to claim 8 wherein said control
function generator circuit is comprised by a resistive voltage
divider circuit having a variable resistance branch including an
insulated-gate field effect transistor, and
an adaptive control circuit connected to the gate of said field
effect transistor to determine the gate voltage and therefore the
resistance of said field effect transistor to further shape said
reference signal according to a selected control.
Description
BACKGROUND OF THE INVENTION
This invention relates to a solid state ballast circuit for gaseous
discharge lamps, and more particularly to a high frequency chopper
ballast for mercury vapor lamps which utilizes electronic
techniques to shape the line current for high power factor and to
obtain good regulation.
The majority of mercury lamps presently in use employ
electromagnetic ballasts with bulky low frequency transformers,
inductors and large power factor correcting capacitors. Although a
number of circuits using solid state devices have been developed
for ballasting high intensity discharge mercury lamps and similar
lamps, those circuits which operate on 60 Hz alternating-current or
full wave rectified voltage incorporate bulky and expensive
components. More sophisticated high frequency circuit approaches do
not achieve an economic solution to the problem and ignore some of
the major problem areas such as acoustic resonance effects and
electrode degradation due to arc initiation.
The combination of features an electronic ballast desirably should
have are to provide high power factor, high efficiency, low
acoustic and radio frequency interference noise, and good
regulation in a single phase circuit without requiring heavy power
frequency magnetics and large correction and energy storage
capacitors. Further, the ballast circuit should be relatively
insensitive to normal line transients, the lamp should not
extinguish upon rapid excursion to 65 percent of rated line
voltage, lamp operation should avoid visible flicker or acoustic
resonance effects caused by continuous operation at a constant high
frequency, and circuit operation should be stable for very long
periods of time. The circuit should operate over an ambient
temperature range of -30.degree.C to +85.degree.C and provide
negligible electrical interference to its surroundings.
In the concurrently filed application Ser. No. 429,914 by Robert L.
Steigerwald and John N. Park, entitled "Power Circuits for
Obtaining a High Power Factor Electronically," and assigned to the
same assignee, a number of single phase chopper circuits for
alternating-current and direct-current loads are described which
use only high frequency filtering and electronically shape the line
current to obtain a high power factor. As a typical application, a
mercury lamp ballast circuit having many of the foregoing desirable
features is disclosed. The present application relates to an
improvement on this ballast circuit with emphasis on obtaining good
lamp operation in a more satisfactory circuit configuration.
SUMMARY OF THE INVENTION
The new solid state, high frequency chopper ballast is suitable for
energization by unfiltered low frequency alternating-current line
voltage, preferably full wave rectified with only high frequency
filtering, and broadly includes a controlled switching means, such
as a power transistor, and coasting device means, such as a power
diode, that conduct alternately and supply lamp current through a
coasting inductor to a mercury vapor lamp or other gaseous
discharge lamp. A current sensor is coupled to sense the
instantaneous, high frequency ripple lamp current. The control
circuit has provision for generating a preselected reference signal
waveshape to determine the power level, optionally regulate the
lamp current, and to effect shaping of the lamp current and
therefore the line current to obtain a high power factor. l By
effectively comparing the sensor and reference signals, an output
signal is produced for controlling the application of alternate
turn-on and turn-off signals to operate the controlled switching
means at a variable high frequency chopping rate to shape the lamp
current as determined by the reference signal waveshape. As a
result of automatic sweeping of the chopping frequency and as a
result of the low ripple amplitude, acoustic resonance effects are
avoided.
In accordance with the invention, an improved lamp current
waveshape is obtained at initial start-up of the lamp. Also, the
lamp current is improved by supplying in a more satisfactory manner
a minimum lamp current in each cycle when the comparing means is
ineffective to shape the lamp current, i.e., during the valleys or
low voltage regions of the pulsating or sinusoidal power voltage.
To avoid the undesirable glow-to-arc mode, the control circuit has
provision for temporarily shaping the reference signal at initial
start-up to obtain a high starting current, as by using a long time
constant network to modify the action of the control function
generator in the reference signal generating means. Minimum lamp
current in the valley or low voltage regions for improved
reignition characteristics is supplied by the high frequency filter
and, in the preferred transistor d-c chopper ballast, by using
local energy storage capacitors in the improved transistor drive
curcuit power supply to provide base current to the normal
conducting positive base drive circuit to maintain power transistor
conductivity. Other control circuit improvements include a low
voltage power supply for the comparator which supplies clipped,
regulated voltage except during the valley regions when it is not
needed, thereby eliminating the need for electrolytic capacitors.
An improved transistor base drive circuit and power supply therefor
are also disclosed. The new high frequency chopper ballast for
mercury lamps incorporates the desirable features previously
mentioned, is highly efficient with low volume and light weight,
and does not employ low frequency energy storage and correction
capacitors, inductors, and power transformers.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a simplified schematic circuit diagram partly in block
diagram form of a d-c chopper ballast for a mercury vapor lamp and
is used to explain the principles of the invention;
FIG. 2 is a waveform diagram of a sinusoidal reference signal with
closely adjacent control band limits for controlling the intervals
of conduction and nonconduction of the power transistor in FIG.
1;
FIG. 3 is a schematic power circuit diagram with control circuit
connections according to the preferred embodiment of the mercury
lamp solid state ballast circuit;
FIG. 4 shows ideal waveform diagrams of the line current and
voltage, lamp current, and reference signal for the preferred
ballast circuit;
FIG. 5 is an enlarged diagram of the flattened sinusoidal reference
signal and control band limits for the control circuit logic
signals;
FIG. 6 are typical oscilloscope waveforms of the lamp voltage and
lamp current illustrating, at an enlarged scale, the high frequency
ripple produced by operation of the chopper ballast;
FIG. 7 is a detailed control circuit schematic diagram for the
mercury lamp ballast circuit;
FIG. 8 is a diagrammatic side view of a transformer with a pair of
secondary windings for supplying power to the logic and power
transistor base drive circuits in FIG. 7; and
FIG. 9 is a sketch of a portion of the control function generator
in FIG. 7 modified to obtain auxiliary adaptive control of the
mercury lamp chopper ballast, for example, in response to sensing
the ambient light level.
DESCRIPTION OF THE PREFERRED EMBODIMENT
The high frequency, single phase, direct current chopper circuit
shown in FIG. 1 supplies a controlled current waveshape and
controlled power to a mercury vapor lamp or other appropriate
gaseous discharge lamp, and the line current is accordingly
electronically shaped to obtain a high power factor. The power
circuit is relatively simple and economical, and uses no bulky
supply frequency transformers, inductors, or large energy storage
or power factor correcting capacitors. The control circuit operates
on the basis of continuously comparing the sensed lamp current with
a preselected reference signal waveshape to thereby determine the
high frequency switching rate of the power transistor and generate
the desired lamp current waveshape. In the preferred chopper
ballast of FIGS. 3-9, other desirable operating characteristics
such as good regulation, a good starting current waveform, etc.,
are provided as will be explained.
The single phase power circuit (FIG. 1) has a pair of input
terminals 20 and 21 connected, by way of illustration, to a 60 Hz,
277 volt source of alternating current, but other power frequencies
and voltages can be used depending on the application. A diode
bridge rectifier 22 connected to the a-c input terminals produces a
full wave rectified sinusoidal voltage which is supplied
essentially unfiltered to the chopper circuit. A high frequency
filter provided for example by a series inductor 23 and a shunt
capacitor 24 is connected across the output terminals of the bridge
rectifier 22, but these high frequency filter components
essentially are provided to isolate the high frequency chopping
from the 60 Hz line. It may be preferable to further include a
second shunt filter capacitor connected between the input lines,
and other variations are possible depending upon the amount of line
filtering required. In the chopper circuit, a power transistor 25
and power coasting diode 26 are connected in series between the
high voltage, 120 Hz, pulsating d-c supply terminals 27 and 28, and
a coasting inductor 29 is connected in series with the mercury lamp
30 across the coasting diode 26. A suitable load current sensor 31,
such as a small current transformer or sensing resistor, is coupled
in series with the lamp 30, and continuously supplies an input
signal to the control circuit which is indicative of the magnitude
of the instantaneous lamp current. In operation, in the same manner
as a time ratio control circuit, the power transistor 25 is turned
on and off at a high frequency switching rate. During conducting
intervals of the transistor 25 power is supplied to the load 30
through the coasting inductor 29, and during nonconducting
intervals of the transistor 25 the coasting diode 26 becomes
forward biased and provides a path for load current as the stored
energy in coasting inductor 29 discharges. The circuit is
preferably operated in the tens of kilohertz frequency range, in
the range of about 10 kHz to 40 kHz for this application. With this
power circuit configuration, it is noted, there is inherently a
small high frequency ripple in the load current.
The coasting diode 26 and power transistor 25 are preferably
matched devices in order to eliminate additional power circuit
components in the coasting path. In each high frequency cycle when
the power transistor is rendered conductive, the coasting diode
does not immediately block due to stored charges and higher than
normal currents flow in the power transistor. The peak current
generated during this transient is limited by employing a fast
recovery coasting diode and by making a reasonably close match of
the turn-on time of transistor 25 to the recovery time of the
coasting diode. A controlled recovery diode is used rather than a
"snap off" diode to prevent large transient voltages from
developing across the diode and to prevent generation of high
frequency disturbances.
The control circuit generates a reference signal which is basically
in phase with the applied line voltage and has a predetermined
waveshape and magnitude to achieve high power factor and deliver a
selected amount of power to the load. As has been pointed out, in
this power circuit the reference signal determines the load current
waveshape and thus the line current waveshape and input power for a
given lamp. The reference signal waveshape can also be selected to
achieve additional desirable features such as good regulation and
suitable load current waveshapes to meet the range of load
operation conditions. Accordingly, the exact reference signal
waveshape that is selected depends upon the combination of features
that are required or the best compromise, depending upon the
particular circumstances. In order to eliminate the need for
special signal generating equipment such as low frequency
oscillators, the control signal is derived directly from the a-c
input lines and then shaped according to a selected control
function to obtain the desired reference signal waveshape. The
reference signal is then also in phase with the line voltage. To
this end, a step-down transformer 32 is connected across the input
lines and, for the power circuit configuration, feeds a diode
bridge rectifier 33 so that the input to a control function
generator 34 is a full wave rectified d-c voltage. Generally
speaking, the control function is selected as previously described
and can be a constant gain, an electronically variable gain, a
squaring circuit, a square root circuit, etc., depending upon the
type of load and control desired. Referring also to FIG. 2, there
are closely adjacent control band limits associated with the
reference signal that effectively determine the limits of excursion
of the lamp current as shaped by the controlled switching action of
the power transistor 25. The control band is effectively placed
about the reference signal, or can be entirely at one side of the
reference signal or closely spaced from it. In any case, the
control band limits are close to or coincide with the reference
signal and conform to its waveshape. Although other circuitry can
be employed to obtain the control band limits, a simple and
effective implementation is by the use of a comparator 35 with
hysteresis. The hysteresis characteristic may be obtained by a
feedback connection from the output of the comparator to the
positive input of the comparator, as is further explained with
regard to FIG. 7. The reference signal is applied to the positive
input of comparator 35, while the negative input is a sensor signal
indicative of the instantaneous lamp current generated by the
current sensor 31.
An output from the comparator 35 is amplified by amplifier 36 and
is effective to apply a base drive signal to the power transistor
25 to drive it into saturation and render it conductive. Assuming
that lamp current is circulating in the coasting path and is
decreasing, and that there is a low output from the comparator 35
so that power transistor 25 is turned off, the lamp current
continues to decrease until the current sensor signal at the
negative input of the comparator is equal to and about to go below
the reference signal control band limit at the positive input of
the comparator (i.e., the reference signal minus hysteresis). A
comparator output is now produced, turning on the power transistor
25 and causing an increase in the lamp current as current is drawn
from the supply. The reference signal now switches to its upper
control band limit value (i.e., the reference signal plus
hysteresis), and the comparator output remains high and the power
transistor 25 remains conductive until the lamp current increases
and the current sensor signal becomes equal to the value of the
other reference signal control band limit. The comparator output
then goes low, thereby turning off the power transistor 25 and
switching the value of the reference signal at the positive input
of the comparator to its lower control band limit. The lamp current
therefore has a small amount of ripple about the nominal value
determined by the reference signal hysteresis. The chopping
frequency of the circuit is not constant during each cycle of the
rectified sinusoidal voltage supplied to the chopper circuit. The
chopping frequency is determined primarily by the value of the
coasting inductor 29, the instantaneous voltage difference between
the rectified sinusoidal voltage feeding the chopper and the actual
lamp voltage, the storage time of power transistor 25, and the
comparator hysteresis. For the circuit shown in FIG. 1, the
chopping frequency is considerably higher at the middle of the half
cycle than at either end where the supply voltage is low. This
periodically variable chopping frequency is desirable for some
loads, for example as a factor in eliminating acoustic resonance
problems in mercury vapor lamps which can occur under certain
constant high frequency conditions.
Having discussed the underlying principles of the chopper circuit
and control technique, the preferred embodiment of the invention
will be described with regard to FIGS. 3-9. The power circuit (FIG.
3) is similar to FIG. 1 and is suitable for HID mercury lamps in
the 74 to 1000 watt range and also, without modification, for other
gaseous discharge lamps that are operable on unfiltered full wave
rectified 120 Hz (or 100 Hz) voltage which cyclically drops to zero
in the valley regions. The preferred circuit is discussed with
regard to ballasting a 250 watt mercury vapor lamp, giving typical
values of the voltages, currents and other parameters to clarify
the presentation. When appropriately modified, the ballast circuit
may be used with still different types of gaseous discharge lamps
that require a supply of lamp current in the valleys to maintain
sufficient lamp ionization until the 120 Hz wave rises to a usable
level. Within the broader scope of the invention, the chopper
ballast can be constructed in a-c versions without a full wave
rectifier using a pair of inverse-parallel power switches and
coasting devices to provide bidirectional conducting capability.
This is further explained in the aforementioned concurrently filed
application Ser. No. 429,914, to which the reader may refer for
further information. Instead of the power transistor, a gate
turn-off thyristor can be used, both of these being described
generically as a controlled solid state switch with a single
electrode for turn-on and turn-off. The power circuit is preferably
fabricated by power module techniques while the control curcuit is
fabricated using integrated circuit and microelectronic
techniques.
In FIG. 3, high frequency filtering and transient voltage
protection is provided at the input to the chopper circuit and is
effective as to both the power circuit and the control circuit. The
input high frequency filter serves primarily to limit the amount of
radio frequency interference which appears across the line due to
the operation of the chopper, and includes a second shunt capacitor
24' as well as the series filter inductor 23 which now has a small
parallel resistor 40 to prevent ringing in the filter circuit due
to transient excitation. The filter inductor also provides
sufficient series impedance to permit effective line transient
voltage suppression for all the power and control circuit
components by means of a single polycrystalline varistor 41
effectively connected between the input terminals of diode bridge
rectifier 22. The input voltage for the control circuit 42 is taken
between these same two lines. By way of example, varistor 41 is a
GE-MOV varistor (trademark of the General Electric Company), type
V275LA 20. The high frequency filter also includes the shunt
capacitor 24 (now provided with a parallel bleeder resistor 43) to
provide a circulating path for the high frequency current
components of the transistor chopper circuit. Thus, the voltage
feeding the chopper is essentially a full wave rectified 60 Hz line
voltage.
The power transistor 25, for instance, is a Toshiba 2SC 1172A
transistor and a suitable matched coasting diode 26 is a MR856
power diode manufactured by Motorola, Inc. The current sensor is a
small sensing resistor 31', such as a one-half ohm resistor,
connected in series with coasting inductor 29 and mercury lamp 30,
the coasting diode 26 being connected across all these elements.
The voltage across the sensing resistor 31' is supplied to the
control circuit 42 and is indicative of the instantaneous lamp
current. This is a negative-going signal voltage in this circuit
arrangement. The input voltage derived from the line supplies power
to the control circuit 42 and also provides a control signal that
is modified by the selected control function to provide the
reference signals. The control circuit 42 further includes dual
base drive circuitry for the power transistor 25 which is effective
to turn on, hold on, positively turn off, and hold off the power
transistor. The base drive current and voltage supplied by control
circuit 42 provides the proper conditions for chopper operation
with a full wave rectified supply voltage. As will be further
explained, the base current is proportional to the collector
current in the power transistor, and electrolytic capacitors are
not needed in the base drive circuitry, nor also in the control
function generator and comparator circuitry.
Referring to the waveform diagrams in FIG. 4, it is realized in
practice that the line voltage varies under normal conditions. The
reference signal e.sub.ref is a full wave rectified, flattened
sinusoidal signal, and the control function additionally provides
an electronically variable gain characteristic so that the lamp
current remains approximately constant despite line voltage
variations. This provides good lamp current regulation for a
reasonable range of line voltage variations. A mercury lamp load is
a non-linear load with a negative resistance characteristic at low
frequencies, and further has some of the characteristics of a back
emf load. There is some lamp current at the beginning of each cycle
before ignition and at the end of each cycle, when the line voltage
is low. To further explain the concept of the back emf load, if it
is assumed that the load is a battery being charged, it is readily
seen that power is transferred to the battery only in those
portions of the cycle when the instantaneous applied voltage is
greater than the battery voltage. For instance, for a battery of
100 volts and a peak full wave rectified sinusoidal voltage of 400
volts, no power is transferred to the battery at the beginning and
end of the cycle when the instantaneous voltage is below 100 volts.
The lamp voltage between the terminals of ordinary mercury vapor
lamps is typically about 130 volts. It will be further understood
that there is an impedance transformation by virtue of the
operation of the chopper circuit, so that the lamp current and the
line current do not necessarily have the same waveshape or
magnitude. From the foregoing example, it is seen that there is a
voltage transformation, and in like manner, there is also a current
transformation. Based on the foregoing analysis, there is some lamp
current at the beginning and end of each cycle when the supply
voltage is low, and in the intermediate portion of each cycle the
lamp current is forced to follow the flattened sinusoidal reference
signal. The shaped line current draws increased current due to
ignition of the lamp near the beginning of the cycle, but can be
described as being roughly constant in the intermediate portion of
each cycle, dropping at the end of the cycle in the valley regions
of the rectified supply voltage. This line current waveshape is in
phase with the line voltage and provides high power factor, easily
in excess of 90 percent, with good regulation of the lamp current
and input power.
The flattened and regulated sinusoidal reference signal is actually
negative-going as shown in FIG. 5. The lamp current has a high
frequency ripple about a nominal value, and in the reproductions of
typical oscilloscope waveforms of the lamp current given in FIG. 6,
the ripple in the shaped, flattened sinusoidal lamp current is
illustrated diagrammatically at enlarged scale. The lamp voltage
waveform also exhibits a high frequency ripple and shows the
momentarily higher voltage drawn at reignition at the beginning of
each cycle. In the valleys of the full wave rectified supply
voltage, the lamp plasma actually deionizes to a certain extent,
such that it can be said that the lamp reignites in each cycle. The
minimum lamp current in the valleys maintains sufficient ionization
for good reignition characteristics. For a 277 volt to 208 volt
source, either 60 Hz or 50 Hz, the supply voltage rises to a
sufficiently high level near the beginning of a cycle to permit
reignition.
FIG. 7 is a detailed schematic circuit diagram of the improved
control circuit 42. The step-down transformer 45 is energized by
the high frequency filtered, varistor-protected line voltage and
has a pair of center-tapped secondary windings, one of which
supplies low voltage, high current power (typically 12 volts peak,
1 amp peak) for the dual base drive circuitry of power transistor
25, while the other pair of center-tapped secondary windings
supplies high voltage, low current power (typically 50 volts peak,
30 milliamps peak) for the logic portions of the control circuit. A
suitable transformer construction that provides a low capacitance
between the primary winding and each secondary winding is shown in
FIG. 8. The bobbin 47 is disposed about the central leg of the
magnetic core 48. Bobbin 47 has a multiple wall structure that
provides a series of axially spaced compartments for the winding of
the separate transformer windings in the axial sequence of S1, S3,
P, S4, and S2. The secondary winding designations correspond to
those in FIG. 7. This low capacitance "wafer wound" design is
effective to prevent the coupling of high frequency current
components between the secondary windings and between the secondary
and primary windings, i.e., it provides a low rfi coupling. The
centertapped secondary windings S1 and S2 (FIG. 7) are connected to
a first full wave diode bridge rectifier 49 and generates a
positive-going rectified sinusoidal voltage at one output junction
50 and a negative-going rectified sinusoidal voltage at the other
output junction 51. This negative rectified sinusoidal voltage,
with a typical peak value of about 50 volts, is fed to the control
function generator 34 which produces the flattened sinusoidal,
automatic gain controlled reference signal.
Control function generator 34 is comprised by a resistive voltage
divider connected between the junction 51 and a reference or common
bus 52 which includes the resistors 53-56. The generated reference
signal is taken at the junction of resistors 55 and 56 and supplied
to the positive input of comparator 35. Flattening of the
sinusoidal control signal is accomplished by a small resistor 57
and a small Zener diode 58 connected in series between the junction
of resistors 54 and 55 and the common bus. A small amount of
current is diverted through this network. The automatic gain
control feature is obtained by means of a MOS or insulated-gate
field effect transistor (FET) which is connected in series with a
potentiometer 60 across the resistor 56 and acts as a variable
resistance in the shunt path. The peak voltage of the sinusoidal
control voltage is detected by a peak detector circuit 61 and
determines the gate voltage of FET 59. To this end, a high
resistance value potentiometer 62 is connected between the junction
of resistors 53 and 54 and the common bus 52, and the voltage at
the potentiometer pointer is supplied through a blocking diode 63
and a very large resistor 64 to the peak detector 61, which is
comprised by a large resistor and a capacitor connected in parallel
between the gate of FET 59 and bus 52. In this arrangement, diode
63 prevents the capacitor from discharging rapidly. In operation,
peak detector 61 changes the gate voltage and hence the resistance
of FET 59, and therefore the value of the shunt resistance path in
the resistive voltage divider, so that the reference voltage at the
junction of resistors 55 and 56 is approximately constant despite
variations in the peak value of the control voltage due to line
voltage variations.
An additional important function of the peak detector circuit 61
for controlling the gate voltage of FET 59 is to provide an
improved starting current waveform for the lamp to minimize
electrode degradation during arc initiation. The time constant of
the series RC network (primarily resistor 64 and the capacitor) is
relatively long and is effective to delay the divider action a few
seconds. That is, the capacitor at the gate of FET 59 charges
slowly upon exciting the ballast circuit, with the result that the
FET resistance is initially high as is the value of the generated
reference voltage. The starting lamp current therefore is
momentarily relatively high to quickly heat the cathode and avoid
the undesirable glow-to-arc mode. By way of illustration, for a 250
watt mercury lamp, the starting current ramps from 5 amps peak to
the normal 3 amps peak in approximately 8 seconds. Also, it is
possible to provide the circuit with sophisticated adaptive control
by controlling the voltage at the gate of FET 59. For example,
referring to FIG. 9, the ambient light level may be sensed by a
phototransistor 65 or other photosemiconductor and used to actuate
an auxiliary adaptive control circuit 66 which in turn can
determine the voltage at the gate of the field effect transistor
and consequently the magnitude of the reference voltage.
Alternatively, the output from the adaptive control circuit can be
connected directly to a comparator input. In this manner, a lamp
can automatically be turned on and off in response to the ambient
light level, or timing circuits may be employed to control the
power level of operation over any given time duration, and there
are other possibilities. Adaptive control circuit 66 can be
employed to control the power delivered to the lamp. That is, by
sensing the lamp voltage as well as the lamp current, the sensed
lamp voltage is used to control the output of auxiliary adaptive
control circuit 66. The reference signal is then modified by
control of the gate voltage of FET 59 to keep the lamp power
approximately constant.
The comparator 35 is preferably an integrated circuit component
such as the LM-311 device manufactured by the National
Semiconductor Corp. A positive low voltage power supply and a
negative low voltage power supply produces the respective voltages
+V.sub.dc and -V.sub. dc for supplying power to the comparator.
These power supplies are produced by clipping the high positive and
negative voltages at the outputs of bridge rectifier 49, and
obviate the need for electrolytic capacitors. Although there is no
power for comparator 35 during the valleys of the full wave
rectified 120 Hz voltage, power is not needed at these times since
lamp current during the valley regions is not determined by action
of the comparator. Some lamp current is provided by another
mechanism, as will be explained. The positive low voltage power
supply is comprised by a resistor 67 and a small Zener diode 68
conneceted in series between the bridge output junction 50 and the
center-tap between the secondary transformer windings S1 and S2,
and a transistor 69 having its collector-base connected across the
resistor 67 while the emitter is connected to the appropriate pins
in comparator 35. This is recognized as being a series pass
transistor regulator. As the voltage at point 5. rises, current is
supplied through resistor 67 to Zener diode 68 and to the
base-emitter junction of transistor 69. After the Zener diode
clamps the base voltage, the circuit functions as a series voltage
regulator with the voltage at the emitter of transistor 69 being
approximately +5 volts. The negative low voltage power supply at
the other side of bridge rectifier 49 is similar with corresponding
components designated by corresponding primed numerals. In
addition, it is noted that a high frequency filter capacitor 70 is
connected between the junction of resistors 53 and 54 and the
common bus 52 to filter undesirable high frequency transients in
the control voltage. Also, to provide filtering of the comparator
power supply, small capacitors 71 and 71' are respectively
connected between the +V.sub.dc and -V.sub.dc buses and the common
bus 52.
The minus terminal of sensing resistor 31', as previously
mentioned, is coupled to the negative input of comparator 35, while
the plus terminal is referenced to the center-tap between the pair
of secondary windings S1 and S2 of transformer 45, which is the
common point. To eliminate fast switching transients which could
give a false peak lamp current identification, an RC filter
comprised by resistor 72 and capacitor 73 is effectively connected
across the sensing resistor. The positive input is connected
through a resistor 74 to the junction of resistors 55 and 56 at
which the flattened sinusoidal reference signal is generated. To
provide a comparator hysteresis characteristic, a relatively large
resistor 75 is connected in a feedback path between the output and
the positive input of the comparator, and functions with resistor
74 as a voltage divider. The amount of feedback voltage or feedback
current at the positive input has two values depending upon whether
the comparator output is high or low. The net instantaneous voltage
at the positive input is thus determined by the instantaneous value
of the negative-going flattened sinusoidal reference signal and by
the amount of feedback voltage. As a result of the normal operation
of the chopper circuit, as previously explained, the changing
current sensor signal in the logic circuit at the negative input of
the comparator is alternately compared with the two reference
signal control band limits. This will be reviewed again later.
The output of comparator 35, which typically has a low output of -5
volts and a high output of +5 volts, is coupled to an output
transistor 76 which provides the interface between the logic
circuit and the power transistor drive circuit. In particular, the
comparator output is connected to the junction of a pair of
resistors 77 and 78 that are connected between the base and an
emitter resistor 93 for transistor 76, the resistor 93 further
being connected to the -V.sub.dc bus. Upon the occurrence of a high
comparator output, current is supplied to transistor 76 thereby
rendering it conductive. The net effect of this action, it will be
recalled, is to turn off the power transistor 25.
Relatively low voltage, high current, full wave rectified 120 Hz
unidirectional voltage is supplied to the power transistor base
drive circuit by means of a second diode bridge rectifier 80 that
is energized by the second pair of secondary windings S3 and S4 of
transformer 45. A pair of relatively small, local energy storage
capacitors 81 and 82 are respectively coupled between the
center-tap of the transformer's secondary windings S3 and S4 and
the positive and negative d-c supply terminals 83 and 84 of bridge
rectifier 80. In each cycle, these capacitors store energy which is
available for discharge in the valley regions of the rectified 120
Hz wave, thereby providing a source of base current for the power
transistor 25 in the valley regions when the control logic does not
function. These capacitors also provide low impedance sources so
that fast rising current waves can be developed to properly drive
power transistor 25. The power transistor base drive circuitry is
divided into alternately operating positive and negative base drive
circuits 85 and 86 that serve to turn on, hold on, positively turn
off, and hold off the power transistor 25. The magnitude of the
positive base current varies as a chopped half sinusoid since only
high frequency filtering is provided by the capacitors 81 and 82,
and therefore the collector current in power transistor 25 is
proportional to the base current in transistor 25. Thus, the peak
base current (1 amp) is supplied only when it is absolutely needed
at the point of highest collector current, while at other times
base current is reduced, thereby obtaining high efficiency. In the
positive base drive circuit 85, the collectors of a pair of
transistors in a Darlington amplifier 87 are connected through a
pair of parallel resistors 88 to the positive bridge output
terminal 83, while the emitter of the Darlington amplifier is
coupled to the base electrode 89 of the power transistor. The base
of the first transistor is coupled through a biasing resistor 90 to
the terminal 83, with the result that the transistor Darlington
amplifier 87 is normally conducting and supplies base current to
the base electrode 89. The negative base drive circuit includes a
second Darlington amplifier 91 comprised by a pair of opposite type
transistors whose emitter and collector are respectively connected
together and tied to the base electrode 89. The emitter of the
Darlington amplifier 91 is coupled through a resistor 92 to the
negative bridge supply terminal 84, and the base of the Darlington
amplifier is coupled directly to the collector of the transistor 76
and is also coupled directly to the base of the other Darlington
amplifier 87. With this arrangement, a positive output from
comparator 35 turns on the transistor 76, which is effective in
turn to turn off the Darlington amplifier 87 in the positive base
drive circuit while simultaneously rendering conductive the
Darlington amplifier 91 in the negative base drive circuit.
Excitation of the negative base drive circuit 86, of course,
renders the power transistor 25 nonconductive. Upon application of
the negative base drive current to the base electrode, stored
charge in the base of power transistor 25 is extracted and it turns
off. During the remainder of the off-time, transistor 76 remains
conductive since there is current through resistor 90, and the
small negative voltage at the junction of resistor 90 and the
collector of transistor 76, to which the base of Darlington
amplifier 91 is connected, is effective to maintain the
conductivity of Darlington amplifier 91 and apply a negative bias
to the base electrode 89 which positively holds off the power
transistor 25. With this arrangement, clamping diodes between the
base and emitter of the power transistor 25 are not needed since
base electrode 89 is effectively clamped to the -V.sub.dc supply
through transistor 76 and resistor 93. In the valley regions of the
pulsating 120 Hz unidirectional voltage, the local energy storage
capacitor 81 discharges to provide a small amount of current
through resistors 88 and 90 and Darlington amplifier 87 to maintain
the conductivity of power transistor 25 in the valley regions. The
value of high frequency filter capacitor 24 in the power circuit as
shown in FIG. 3 is sufficiently large (such as 3 microfarads for
the circuit being described) to maintain some lamp current in the
valleys of the energizing power voltage, a condition which is
desirable for good reignition characteristics.
The operation of the solid state mercury lamp chopper ballast will
be reviewed only briefly with reference to FIGS. 3-7. Since only
high frequency filtering of the line voltage is provided in the
power circuit, the voltage supplied to the transistor chopper
circuit is essentially a pulsating, full wave rectified, 120 Hz
sinusoidal voltage. The line voltage is also supplied by means of
stepdown transformer 45 to the control circuit 42. In the control
circuit (FIG. 7), the negative full wave rectified, relatively high
voltage, low current (50 volts, 30 milliamperes) sinusoidal voltage
at the output junction 51 of bridge rectifier 49 is used as a
control voltage for the control function generator 34. In this
sub-circuit, a voltage divider comprised by resistors 53-56 has a
variable resistance component provided by FET 59, the channel of
which is connected in series with variable resistor 60 across the
resistor 56. The automatic gain control feature is obtained since
the gate voltage as determined by the peak detector 61 is
proportional to the peak of the rectified control voltage. When the
magnitude of this voltage drops, for example, FET 59 tends to turn
off and increases the variable resistance in the voltage divider so
that the reference signal taken at the junction between resistors
55 and 56 remains approximately constant with line voltage
variations. The sinusoidal control voltage is further flattened
somewhat by means of a small current drain through the resistor 57
and Zener diode 58. The regulated, flattened sinusoidal reference
signal supplied to the positive input of comparator 35 results in
good lamp current regulation and a slight reduction in the peak
current which the power transistor 25 conducts (as compared to the
unflattened sinusoidal case).
Upon energizing the ballast circuit, the positive base drive
circuit 85 automatically conducts and supplies base current to the
power transistor 25, thus applying line voltage to the lamp. For a
208-277 line, the peak voltage is sufficient to start a mercury
lamp. The starting lamp current is momentarily relatively high
because the reference signal is initially high due to the long RC
circuit time constant of the resistor 64 and the resistor and
capacitor in peak detector circuit 61. The high starting current
quickly heats the cathode of the lamp to avoid the undesirable
glow-to-arc mode, and the current typically ramps down from a 5
amps peak to the normal 3 amps peak in about 8 seconds. The buildup
in lamp current is sensed by the sensing resistor 31' and supplied
as a negative-going sensor signal to the negative input of
comparator 35. The RC filter 72, 73 prevents fast switching
transients from giving a false peak lamp current signal. Assuming
that the line voltage is high enough to cause ignition of the
mercury lamp (see FIGS. 4 and 6), the lamp current is thereafter
shaped in accordance with the flattened sinusoidal reference
voltage until near the end of the cycle in the valley region of the
pulsating 120 Hz d-c voltage. In the steady state, the base current
of the power transistor 25 is at all times proportional to the
collector current whose envelope varies approximately as half
sinusoid. It will be recalled that the comparator 35 has a
hysteresis characteristic and that there is a polarity inversion
since the reference signal is negative-going while the lamp current
is positive. Assuming that power transistor 25 is conducting and
the lamp current is increasing, with a low output from comparator
35, the lamp current increases until the current sensor signal is
equal to the reference signal control band limit corresponding to
maximum current (see FIG. 5). The output of comparator 35 now
changes to the high output and renders conductive the transistor
76, which is the interface between the logic circuitry and the
power transistor base drive circuit, thereby causing the negative
base drive circuit 86 to conduct while simultaneously turning off
the positive base drive circuit 85. Power transistor 25 is now
nonconductive, and load current now circulates through the coasting
path provided by the forward biased power diode 26 and begins to
decrease. In the meantime, the amount of feedback voltage from the
output of comparator 35 to the positive input has changed, thereby
switching the basis for comparison to the other reference signal
control band limit corresponding to the minimum current value. As
previously explained, power transistor 25 turns off before the end
of the high frequency cycle and is held in the nonconducting
condition by the fact that transistor 76 and Darlington amplifier
91 remain conducting to apply a negative potential to the base
electrode 89 of power transistor 25. When the decreasing current
sensor signal becomes equal to the other control band limit, the
output of comparator 35 goes low, thereby turning off the interface
transistor 76 and the negative base drive circuit 86 while
rendering conductive the positive base drive circuit 85.
Primarily because the rate of rise of load current is variable
since it is determined primarily by the difference between the
instantaneous sinusoidal supply voltage (about 400 volts peak) and
the lamp voltage (about 130 volts constant, except for the rapid
increase and decrease at ignition), the switching frequency of
power transistor 25 automatically varies from approximately 10 kHz
to 30 kHz and back to 10 kHz over a complete cycle. In a mercury
lamp ballast, this sweeping of frequency, which occurs
automatically and is inherent in the operation of the circuit,
helps eliminate acoustic resonance problems. As was previously
explained, the other pair of center-tapped secondary windings S3
and S4 of transformer 45 supplies, via the second bridge rectifier
80, relatively low voltage, high current (12 volts peak, 1 amp
peak), full wave rectified, pulsating 120 Hz unidirectional voltage
to the transistor base drive circuit. In the valley regions when
the power circuit supply voltage goes low, the comparator 35 and
associated control logic does not function, and the local energy
storage capacitor 81 discharges to supply base current through the
resistors 88 and 90 and Darlington amplifier 87 to the base
electrode 89 of power transistor 25. This maintenance of lamp
current in the valley regions is desirable for good lamp
maintenance. The positive and negative low voltage power supply for
comparator 35, provided by the series pass transistor regulators
including elements 67-69 and 67'-69', does not function and thus
the comparator does not function in the valley regions when the
control voltage provided by bridge rectifier 49 goes low, however
this makes no difference since the power voltage is low and the
power transistor is maintained in the conducting condition.
By shaping and forcing the lamp current to a flattened sinusoidal
waveshape as shown in FIG. 4, the line current is in phase with the
line voltage and is electronically shaped to obtain a high power
factor exceeding 90 percent. By properly selecting the magnitude of
the reference signal according to the power level desired and by
using a control function to obtain an electronically variable gain,
lamp current is regulated for a nominal line voltage of 277 volts
to less than one-half percent for a plus or minus 10 percent line
voltage variation. The magnitude of the high frequency ripple in
the lamp current (see FIG. 6) is preselected and can be variable,
and for this circuit has approximately 0.25 ampere ripple about the
nominal value. The basic chopper thus can be used for mercury vapor
lamps having different wattage values by properly tailoring the
resistors 88 and 92 in the base drive circuitry, changing the value
of sensing resistor 31', and by adjusting the values of the
appropriate resistors in control function generator 34 to change
the magnitude of the reference signal according to the size of the
lamp being powered. With this solid state ballast circuit, lamp
operation is sustained down to 65 percent of rated line voltage.
Other advantages previously mentioned are that the chopping
frequency is automatically variable to help avoid acoustic
resonance effects and lamp flicker.
By sustaining a minimum lamp current in the valley regions of the
120 Hz lamp current waveform, the resulting lamp voltage waveform
is more suitable for lamp reignition in each cycle and promotes
prolonged lamp life. The provision of a momentarily high starting
current for the mercury lamp minimizes electrode degradation during
arc initiation and eliminates the undesirable cathode glow-to-arc
mode. The glow-to-arc mode puts a high voltage and high current on
the cathode. The chopper ballast operates over a -30.degree.C to
+85.degree.C ambient temperature range. In this regard, and of
importance to the potential commercial attractiveness of the
ballast, is the fact that the high frequency circuit operation is
achieved with minimum capacitive energy storage so as to eliminate
electrolytic capacitance and their associated problems.
Furthermore, this circuit operates reliably under either short
circuit or open circuit lamp load conditions. In the event that a
short circuit in the lamp occurs, the circuit operates inherently
to keep the current in power transistor 25 within its control
limits, and in the event of an open circuit, voltage is
continuously applied to the lamp terminals so that the circuit
restarts automatically, assuming that the mercury lamp is cold or
has cooled down enough so that it will restart immediately.
In summary, an improved chopper ballast is particularly suitable
for operation of mercury vapor lamps from commercially available 60
Hz single phase line voltage in an advantageous transistor d-c
chopper configuration that eliminates the need for bulky
transformers, inductors, large correction and energy storage
capacitors, undesirable electrolytic capacitors, and power
frequency filtering. In addition to the fundamental requirement of
high power factor and good regulation, the circuit supplies a lamp
current waveshape especially suited for mercury and other gaseous
discharge lamps operated on high frequency ripple current, with
provision for a good starting current waveform, automatic sweeping
of the chopping frequency to eliminate acoustic resonance problems,
and a minimum lamp current in the valley regions of the pulsating
energizing voltage for improved reignition. The new chopper ballast
is economical, light-weight, has low volume, and can be built with
state-of-the-art solid state devices.
While the invention has been particularly shown and described with
reference to a preferred embodiment thereof, it will be understood
by those skilled in the art that the foregoing and other changes in
form and detail may be made therein without departing from the
spirit and scope of the invention.
* * * * *