U.S. patent number 3,860,928 [Application Number 05/411,934] was granted by the patent office on 1975-01-14 for super-directive system.
This patent grant is currently assigned to Raytheon Company. Invention is credited to Stanley L. Ehrlich.
United States Patent |
3,860,928 |
Ehrlich |
January 14, 1975 |
SUPER-DIRECTIVE SYSTEM
Abstract
A system of radiating elements arranged for forming one or more
beams of radiation having radiation patterns such as a monopole,
dipole, quadrupole, other multipoles or combination thereof. The
individual radiating elements of the array are interconnected by
circuitry providing for the summing and differencing of signals
provided by adjacent radiating elements in response to incident
radiation. In one embodiment the signal of one radiating element is
delayed relative to the signal of an adjacent radiating element.
The differencing of the signals provides for the deep nulls found
in radiation patterns such as the dipole and quadrupole radiation
patterns, while the delay between signals of adjacent radiating
elements is adjusted for varying the shape of the directivity
pattern. The system provides for varying the direction and shape of
beams of the radiation pattern to provide for the detection,
classification and/or tracking of a distant source of radiation as
well as for illuminating a distant object. The system is responsive
to the intensity of radiation received along one or more beams of
the radiation pattern, and in response thereto, varies the delay
between the signals of adjacent radiating elements and also
provides for the selective coupling of specific radiating elements
of the system.
Inventors: |
Ehrlich; Stanley L.
(Middletown, RI) |
Assignee: |
Raytheon Company (Lexington,
MA)
|
Family
ID: |
26953069 |
Appl.
No.: |
05/411,934 |
Filed: |
November 1, 1973 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
Issue Date |
|
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268416 |
Jul 3, 1972 |
3821740 |
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Current U.S.
Class: |
342/375; 367/97;
367/100; 367/126; 342/377; 367/103 |
Current CPC
Class: |
G10K
11/346 (20130101) |
Current International
Class: |
G10K
11/34 (20060101); G10K 11/00 (20060101); H04b
007/04 () |
Field of
Search: |
;343/113R,119,16R,12R,1SA ;340/6R,16R |
References Cited
[Referenced By]
U.S. Patent Documents
Primary Examiner: Hubler; Malcolm F.
Assistant Examiner: Montone; G. E.
Attorney, Agent or Firm: Warren; David M. Pannone; Joseph D.
Bartlett; Milton D.
Parent Case Text
This is a division of application Ser. No. 268,416 filed July 3,
1972, now U.S. Pat. No. 3,821,740.
Claims
What is claimed is:
1. In combination:
an array of radiating elements, said array having dimensions less
than a wavelength of energy radiated therefrom;
a plurality of signal carrying channels;
means for coupling individual ones of said channels with respective
ones of said radiating elements, each of said channels being
coupled to all of said radiating elements to provide combinations
of signals of said radiating elements, one of said channels having
means for varying the gain of its signal combination relative to
that of another of said channels, at least one of said combinations
having at least a portion thereof which is the signal of a
quadrupole radiation pattern, at least one of said combinations
having at least a portion thereof which is the signal of a dipole
radiation pattern and at least one of said combinations having at
least a portion thereof which is the signal of a monopole radiation
pattern;
means for rotating said dipole pattern relative to said quadrupole
pattern; and
means for arithmetically combining each of said radiation patterns
to form a multiple mode type of radiation pattern, said varying of
gain and said rotating serving to shape said multiple mode
pattern.
2. A combination according to claim 1 further comprising means
coupled between said arithmetic means and said gain varying means
for varying the shape of said multiple mode type of radiation
pattern in response to signals thereof.
3. A combination according to claim 1 further comprising means
coupled between said arithmetic means and said rotating means for
varying the shape of said multiple mode type of radiation pattern
in response to signals thereof.
4. In combination:
an array of radiating elements;
a plurality of signal carrying channels;
means for coupling individual ones of said channel with respective
ones of said radiating elements, said channels being coupled to
provide combinations of signals of said radiating elements, one of
said combinations being the summation of signals of alternate pairs
of said radiating elements subtracted from the summation of the
signals of the remaining radiating elements to provide a first
quadrupole pattern, a second of said combinations being composed of
the summation of signals of pairs of elements subtracted from the
sum of the signals of the remaining elements wherein each pair of
signals of said pair of elements of said second combination
comprises only one signal of the corresponding pair of signals of
said first combination to form a second quadrupole pattern having
an axis inclined at an angle with respect to the axis of said first
quadrupole pattern; and
means for combining at least one of said quadrupole patterns with
signals of said channels to provide a polypole pattern.
5. In combination:
an array of radiating elements;
means for algebraically combining signals from individual ones of
said radiating elements, such algebraic combination comprising the
summing and differencing of paired groups of said signals of said
radiating elements, the number of signals in one of said groups of
one of said combinations being double the number of signals in one
of said groups of a second of said combinations;
means coupled to said combining means for varying the gain of one
of said combinations relative to a second of said combinations;
means for determining which of said combinations provide a desired
signal; and
means coupled to said determination means for displaying a
direction of radiation corresponding to said determination.
6. The combination according to claim 5 further comprising
multiplexing means and signal generation means, said array being
coupled via said multiplexing means to said algebraic combining
means, said array being coupled to said signal generation means via
said multiplexing means, said signal generation means providing a
signal for transmission by said array to be reflected from a
distant object back towards said array to provide information with
respect to the direction of said object from said array.
7. The combination according to claim 6 further comprising timing
means and gating means, said timing means synchronizing said signal
generation means with said gating means, said gating means
inhibiting said algebraic combining means during periods of time
different from a time delay corresponding to the anticipated range
of said object for providing information relative to said
range.
8. A system for combining signals of a radiator array
comprising:
an array of radiating elements;
first combining means coupled to a first and a second group of said
radiating elements for subtracting the sum of the signals of said
second group from the sum of the signals of said first group of
radiating elements;
second combining means coupled to a third group and a fourth group
of said radiating elements, said third group comprising the
radiating elements of a portion of said first group and the
radiating elements of a portion of said second group, said fourth
group comprising the radiating elements of a portion of said second
group and a portion of said first group of radiating elements, said
second combining means subtracting the sum of the signals of said
fourth group from the sum of the signals of said third group of
radiating elements;
means coupled to said first and said second combining means for
comparing an output signal of said first combining means with an
output signal of said second combining means; and
means coupled to said comparing means for varying the gain of an
output signal of said first combining means relative to an output
signal of said second combining means, said varying of gain
providing a variation in the shape of a radiation pattern of said
radiator array.
9. A system for combining signals of a radiator capable of
vibrating in multiple modes comprising:
a ring of acoustically active material;
a plurality of electrodes positioned along a surface of said ring,
each of said electrodes receiving a signal according to its
position on said ring and in response to a mode of vibration of
said ring, said ring being capable of vibrating in a mode of
vibration corresponding to a monopole, a dipole and a quadrupole
radiation pattern;
first combining means coupled to a first and a second group of said
electrodes for subtracting the sum of the signals of said second
group from the sum of the signals of said first group of
electrodes;
second combining means coupled to a third group and a fourth group
of said electrodes, said third group comprising the electrodes of a
portion of said first group and the electrodes of a portion of said
second group, said fourth group comprising the electrodes of a
portion of said second group and a portion of said first group of
radiating elements, said second combining means subtracting the sum
of the signals of said fourth group from the sum of the signals of
said third group of electrodes; and
means coupled to said first and said second combining means for
comparing an output signal of said first combining means with an
output signal of said second combining means.
10. A system according to claim 9 further comprising means coupled
to said comparing means for varying the gain of an output signal of
said first combining means relative to an output signal of said
second combining means, said varying of gain providing a variation
in the shape of a radiation pattern of said radiator.
Description
BACKGROUND OF THE INVENTION
This invention relates to super-directive arrays and more
particularly to a system including such an array wherein signals
from adjacent elements of the array are delayed with respect to
each other by preselectable amounts of delay.
Dipole and quadrupole radiators have been well known for many years
in the fields of electromagnetic and acoustic radiators. These
radiation patterns are obtained at all frequencies for which the
interelement spacing and the size of an element are substantially
smaller than a wavelength of the radiation transmitted or received
by the radiator or array of radiators. For example, in the case of
the electromagnetic dipole radiator, the two active regions of the
radiator, namely, the positively and negatively charged termini of
the radiating element are spaced apart, preferably less than
one-quarter wavelength. In the case of the acoustic dipole
radiator, a pair of monopole or omnidirectional radiating elements
are placed close together, preferably less than one-quarter
wavelength spacing, and the effect of a positive and negative
pulsation is provided by subtracting the signal received by one of
the elements from that of the other element. A directional sonar
system employing omnidirectional and dipole radiation patterns is
disclosed in U.S. Pat. No. 3,176,262 which issued to S. L. Ehrlich,
et al., on Mar. 30, 1965. Since the invention to be described
hereinafter is particularly useful for acoustic radiators, the
ensuing description will be directed towards acoustic systems while
it is understood that these teachings are also applicable to
electromagnetic systems.
It is evident that an array of closely spaced radiating elements is
essential for the directional transmission and reception of
acoustic energies at very low frequencies since a typical phased
array antenna in which the radiating aperture is many wavelengths
long would be prohibitively large in many applications utilizing
frequencies as low as, for example, 10 hertz. However, even with
higher frequencies, an array of closely spaced radiating elements
may be useful since the directivity pattern of such an array is
substantially invariant with the frequency of radiation for all
radiations having a wavelength substantially larger than the
overall dimensions of the array. At still higher frequencies where
the wavelength becomes smaller than the array, the array may still
be useful for specific applications such as where a multiple lobed
beam pattern is desired, however, such a system tends to be of very
narrow band width. It is also apparent that an array of closely
spaced radiating elements is inherently useful in situations
requiring small size and expense.
A problem arises in that such an array produces beam widths which
are excessively wide for the standard techniques utilized in sonar
tracking, for example, the tracking of an object submerged in the
ocean. In addition, due to the subtraction of signals of adjacent
elements as compared with the summation of such signals in the
standard phased array antenna, the sensitivity of such an array,
namely, the amplitude of the resultant signal produced by the array
in response to an incident amplitude of sound pressure, is
significantly reduced thereby increasing the unwanted effects of
noise.
SUMMARY OF THE INVENTION
In accordance with the invention there is provided a system
incorporating an array of radiating elements which overcome the
aforementioned problems in the prior art to permit the use of such
an array for communicating information by acoustic signals. In one
embodiment providing for a polypole radiation pattern, the
invention comprises a plurality of radiating elements each of which
provides signals in response to incident acoustic energy, the
radiating elements being intercoupled via delay lines and circuitry
for combining the signals to permit the summing of such signals,
the differencing of such signals and the scaling of such signals.
In another embodiment utilizing three radiating elements, a
simplification is presented which permits the use of a single delay
line by connecting the radiating elements via summing resistors
having values which provide for a scaling of the signal voltage of
one radiating element relative to the signal voltage of the other
radiating element; this provides, in the case of small phase
angles, (less than approximately 20.degree.), the equivalent of
three delay lines. There is also disclosed a multiple element array
including a switching circuit for selectively coupling groups of
three radiating elements whereby a beam of acoustic radiation can
be directed in any one of a plurality of directions by
appropriately selecting the group of three elements. In addition
there is disclosed the use of an angle tracking circuit
interconnected with the switching circuit and the delay lines for
operating the switching circuit and the delay lines to rotate a
beam of radiation by means of the switching circuit and to alter
the shape of the array directivity pattern by varying the amount of
delay provided by each of the delay lines. The combination of
signals from the radiating elements provides for polypole radiation
patterns described mathematically by a power series and polymode
radiation patterns described mathematically by a Fourier series.
Means are also disclosed for digitizing the signal as by means of a
sampling circuit which provides voltage states corresponding to
logical 1's for the positive portions of the sinusoidal waveform of
the acoustic wave and logical 0's for the negative portions of the
wave. A digital delay line in the form of a shift register with
multiple outputs is utilized in the digital embodiment as a
variable delay line.
BRIEF DESCRIPTION OF THE DRAWINGS
The aforementioned features and other advantages of the invention
are explained in the following description and taken in connection
with the accompanying drawings wherein:
FIG. 1 is a view, partially isometric and partially diagramatic, of
the radiating system in accordance with the invention;
FIG. 2 shows, by way of example, one form of directivity pattern
obtainable with the array of radiating elements of FIG. 1;
FIG. 3 shows the underside of a boat in the ocean with a radiating
array of the invention mounted thereon, the figure further showing
a cardioid radiating pattern with the notch in the direction of the
propeller for inhibiting propeller noise;
FIG. 4 shows a block diagram of a beam former of the system of FIG.
1 for received signals;
FIG. 5 shows a block diagram of a delay unit of the beam former of
FIG. 4;
FIG. 6 shows a block diagram of an arithmetic unit of the beam
former of FIG. 4;
FIG. 7 shows a block diagram of a beam former of FIG. 1 for
transmitted signals;
FIG. 8 is a block diagram of an alternative embodiment of the beam
former of FIG. 7;
FIG. 9 is a diagrammatic presentation of a generalized beam forming
system for a multiple element array;
FIG. 10 is a block diagram, partially schematic, of a simplified
embodiment of the beam former of FIG. 4;
FIG. 11 is a diagram of a portion of an array illuminated by a wave
of radiant energy for use in a mathematical description of the beam
formers of FIGS. 4 and 9;
FIGS. 12, 13 and 14 represent successive mathematical
transformations utilized in developing the simplified beam forming
system of FIG. 10;
FIG. 15 is a block diagram of an angle tracking unit of the system
of FIG. 1;
FIG. 16 is a schematic diagram of a polymode transducer and
receiver of the invention;
FIG. 17 is a pictorial equation showing the combination of various
receiving directivity patterns provided by the receiver of FIG. 16
to form a polymode directivity pattern;
FIG. 18 is a block diagram of a transmitting and receiving system
incorporating the polymode array and receiver of FIG. 16; and
FIG. 19 is a block diagram of an alternate embodiment of the
receiver of FIG. 16.
DESCRIPTION OF THE PREFERRED EMBODIMENT
Referring now to FIG. 1, there is shown a system 20 comprising, in
accordance with the invention, an array 22 of radiating elements
which is coupled to a transceiver 24 via multiplexing circuitry 26.
Each of the radiating elements takes the form of a transducer 28
suitable for converting electrical energy signals on wires 30 to
sonic energy signals indicated by waves 32, and for converting from
sonic energy to electrical energy.
The transducers 28 are of a well-known construction typically
comprising an electrostrictive ceramic material such as barium
titanate and are suitably mounted upon a base plate 34 to serve as
monopole radiators. The surface 34 of each transducer 28 pulsates
to provide a directivity pattern having axial symmetry. Six
transducers 28 have been arranged, in this embodiment of the
invention, in a circle about a seventh transducer 28 at the center
of the array 22. As will be described hereinafter, this arrangement
of the transducers 28 in the array 22 is particularly useful for
providing a variety of radiation patterns as well as a capability
for steering beams of radiation about a central axis 36 which is
normal to the base plate 34.
The transceiver 24 comprises a beam former 38 for forming the
receiving beam for radiation incident upon the array 22 and a beam
former 40 for forming the transmitted beam of radiation. In the
transceiver 24 a signal generator 42 provides a signal suitable for
transmission via the array 22 to a distant object such as a boat
44, and is coupled to the array 22 via the beam former 40 and the
multiplexing circuitry 26. Detection of a signal such as an echo
from the boat 44, indicated by waves 46, is accomplished by
processing the output of the beam former 38 on line 48 by
conventional means. Accordingly, the transceiver 24 utilizes a
correlator 50 which correlates the received signal on line 48 with
a replica of the transmitted signal provided on line 52 from the
signal generator 42. Such a correlator may employ recirculating
delay lines which may be implemented digitally by shift registers
as is disclosed in U.S. Pat. No. 3,594,718 which issued to Charles
I. Black, et al., on July 20, 1971.
The transceiver 24 further comprises an angle tracker 54 which, in
response to the correlator output on line 56, provides signals on
lines 58 and 60 to the beam formers 38 and 40, as well as along
line 62 to a beam switcher 64 for providing a desired shape to the
directivity pattern of the array 22 and for switching the direction
of a beam of radiation provided by the array 22. The beam switcher
64 comprises a well-known switching circuitry which may be either
electrical or mechanical, such as a stepping switch for coupling
individual ones of the transducers 28 to the beam formers 38 and
40, the specific coupling arrangement being in accordance with the
signal provided on line 62. This switching of the beam, analogous
to sequential lobing in the radar art, enables the angle tracker 54
to obtain information on the direction of radiation, such as the
waves 46, incident upon the array 22. Angle data provided by the
angle tracker 54 on line 56 and range data provided by the
correlator 50 on line 68 are presented on a display 70 which may be
of conventional form such as a cathode ray tube or a graphical
display. The signal generator 42, correlator 50, angle tracker 54,
display 70 and the beam formers 38 and 40 are coordinated by timing
signals from the timing unit 72. The timing signals applied to the
beam formers 38 and 40 and the angle tracker 54 are clock pulses
indicated by the symbol C.
The multiplexing circuitry 26 comprises in addition to the beam
switcher 64, a transmit-receive circuit 74 and amplifiers 76 and 78
for amplifying respectively the received and transmitted signals.
The transmit-receive circuit 74 is of conventional form and may
comprise, for example, a diode circuit such as is frequently
utilized in sonar applications.
Referring now to FIG. 2, there is seen a typical directivity
pattern achievable with a linear array of three radiating elements
80 which may be, for example, the transducers 28 of FIG. 1. In FIG.
2 the radiation pattern is that of a quadrupole radiator modified
such that one lobe is substantially larger than any other lobe. The
particular spacings between the radiating elements 80 and the
signal delays to be applied between signals of these elements 80
for producing a specific radiation pattern will be disclosed
hereinafter. Comparing the array 22 of FIG. 1 with the arrangement
of radiating elements 80 of FIG. 2, it is apparent that any set of
three transducers 28 arranged along a diameter of the array 22
correspond to the linear array of three radiating elements 80 of
FIG. 2. Accordingly, the radiation pattern of FIG. 2 can be
obtained with any three transducers 28 arranged along a diameter of
the array 22. Furthermore, by selectively switching in any such set
of three transducers 28 by means of the beam switcher 64, the
radiation pattern of FIG. 2 can be rotated about the axis 36 of the
array 22 of FIG. 1.
Referring now to FIG. 3 there is shown a bottom view of a boat 82
sailing through the ocean waters 84 and having an enclosure 86
mounted on the boat 82 in acoustic contact with the water, the
enclosure housing the array 22 of FIG. 1, not seen in FIG. 3. The
multiplexing circuitry 26 and the transceiver 24 of the system 20
are not shown in FIG. 3 but are carried within the boat 82. The
system 20 is seen providing a directivity pattern 88 having the
general cardioid-type form with a notch 90 in the direction of the
stern of the boat 82 so that noise caused by the movement of the
ocean water 84 past the propeller shaft 92, the propeller 94 and
the rudder 96 is greatly attenuated. It is noted that the
directivity pattern 88 differs from that of the typical phased
array antenna in that there are no reduced level side lobes in the
direction of the propeller 94 but rather the notch 90 which can be
accurately positioned to exclude the propeller noise. This feature
permits the use of the invention in a relatively high noise
environment even though the sensitivity of the array 22 of FIG. 1
is substantially decreased when the signal from the transducers 28
are combined in a manner, to be described hereinafter, wherein the
signals of adjacent transducers 28 are subtracted from each other
to produce cardioid-type or polypole-type radiation patterns.
Referring now to FIG. 4, there is seen a block diagram of the
receiving beam former 38. The beam former 38 may utilize
conventional analog delay lines for delaying the signals between
adjacent radiating elements of the array 22 of FIG. 1 or,
alternatively, may employ digital techniques in the form of shift
registers for providing this delay. In the preferred embodiment of
the invention the beam former 38 utilizes digital techniques, and,
accordingly, the beam former 38 comprises samplers 98 for sampling
the signals of the transducers 28 indicated diagrammatically in
FIG. 4, arithmetic units 100 for combining the signals of the
transducers 28 and delay units 102 for delaying the signals between
the transducers 28. The delay unit 102 and the arithmetic unit 100
will be more fully described with reference to FIGS. 5 and 6.
The beam former 38 operates as follows. Signals received by the
transducers 28 pass via the multiplexing circuitry 26 to respective
samplers 98. Each sampler is of conventional design and is driven
by clock pulses received from the timing unit 72 of FIG. 2. Each
sampler 98 preferably comprises a switch or gate driven by the
clock pulses for obtaining samples of the incoming signal, and a
comparator (not seen in the figures) which produces an output pulse
on line 104 when the sample of the incoming signal is above a
preset threshold. To simplify this explanation, it is assumed that
the incoming acoustic signal is a sinusoid and, accordingly, the
value of this threshold is selected so that for each positive half
cycle of the incoming sinusoidal signal, a succession of pulses
having a logic state of 1 is provided along line 104, and that
during each negative portion of the sinusoid there is an absence of
the pulses on line 104 or, equivalently, the 0 logic state is
present on line 104. The clock pulse frequency is greater than the
frequency of the incoming signal by a factor of, for example, 1,024
so that a total of 1,024 samples can be taken during each cycle of
the incoming signal. The 1,024 samples permit a 10-bit resolution
of the phase of the incoming signal, namely, increments of
approximately 1/3 degree.
Referring now to the block diagram of FIG. 5, the delay unit 102
comprises a shift register 106 and a digital selector 108 (also
known as a decoder or multiplexer switch). The shift register 106
has a single terminal in which the input signal is applied, and a
plurality of output terminals on lines 110, each of the lines 110
corresponding to one stage of the shift register 106. The shift
register 106 is driven by clock pulses, indicated by the letter C
in the figure, with the result that each pulse of the input signal
appears after successive increments of delay at successive ones of
the output terminals on lines 110, with each of these delay
increments being equal to the interval between clock pulses. The
digital selector 108 has multiple input terminals corresponding to
each of the lines 110 and a single output terminal. A specific
connection is made between one of the input terminals and the
output terminal in accordance with a binary number applied at
terminal D and provided by the angle tracker 54 as seen in FIG. 4.
Thus, the angle tracker 54 is able to select, via the digital
selector 108, a specific output terminal of the shift register 106
so that the delay unit 102 provides a desired amount of delay
between the input and output signals of the delay unit 102.
Referring now to the block diagram of FIG. 6, the arithmetic unit
100 comprises digital selectors 112 and 114, each of which is
similar to and functions in a fashion analogous to the digital
selector 108 of FIG. 5. The arithmetic unit 100 further comprises a
scaling unit 116, an operational amplifier 118, a band-pass filter
120 and a sampler 122 similar to the sampler 98 of FIG. 4. The
arithmetic unit 100 has two inputs, one providing a signal from the
sampler 98 and the other providing a signal from the delay unit
102, and combines these two signals to provide a single output
signal. The two input signals may be combined by summing the two
together by taking their difference or by first scaling the signals
to combine a fraction of one with the other. This is readily
accomplished by summing resistors (not seen in the figures) applied
to the input terminals of the operational amplifier 118; in this
case, a multiplicity of such summing resistors of differing values
are provided by the scaling unit 116, with the digital selectors
112 and 114 interconnecting each input signal with any desired
summing resistor. The digital selectors 112 and 114 provide the
inverse function of the digital selector 108, that is, each digital
selector 112 and 114 has a single input and a multiplicity of
outputs on lines 124 with a specific interconnection of one of the
outputs to the input being provided in accordance with a binary
number appearing at terminal A and provided by the angle tracker 54
as seen in FIG. 4. The two input signals to the arithmetic unit 100
are in binary digital format and, after being combined by the
operational amplifier 118, are converted by the band-pass filter
120 to an analog signal on line 126. The pass band of the filter
120 is selected to pass the frequency of the acoustic radiation
incident upon the array 22 of FIG. 1 while excluding the sampling
frequencies introduced by the samplers 122. The analog signal on
line 126 is then converted by the sampler 122 to a binary digital
format to appear at the output terminal of the arithmetic unit
100.
Referring now to FIG. 7, there is shown a block diagram of the
transmitting beam former 40 of FIG. 1. The beam former 40 accepts a
signal in sampled format on line 128 from the signal generator 42
of FIG. 1 and converts this signal into analog signals suitable for
being transmitted via the transducers 28. The beam former 40
comprises inverters 130, delay units 102 previously described with
reference to the receiving beam former 38, a summer 132 and
band-pass filters 134. The beam former 40 may utilize analog
signals and analog delay lines, however, the preferred embodiment
of the invention, as shown in FIG. 7, utilizes digital delay units
102 with the aforementioned sampled or digital format of the signal
on line 128. Each of the three transducers 28, namely, the left,
center and the right transducers 28, receive separate signals in
order to radiate energy having a desired directivity pattern. The
amplitude of the signals for each of the transducers 28 is
controlled by amplifiers 136 and in accordance with a signal
provided at terminal A from the angle tracker 54 of FIG. 1. To
provide the signal for the left transducer 28, the digital format
signal on line 128 is passed through the band-pass filter 134
having a pass band centered at the frequency of the acoustic signal
radiated by the transducers 28; the pass band is sufficiently
narrow to attenuate components of the sampling frequency so that
only the baseband signal passes via the amplifier 136 and the
multiplexing circuitry 26 to the transducer 28. To provide the
signal for the right transducer 28, the digital format signal on
line 128 is first applied to inverter 130, then delayed by delay
unit 102, again inverted by a second inverter 130 and again delayed
by a delay unit 102, and is finally filtered by a band-pass filter
134 to remove all of the sampling frequency components thereby
converting it to the baseband signal suitable for radiation from
the right transducer 28. Each of the inverters 130 is a digital
inverter and converts a logic state of 1 to a logic state of 0 and
vice versa. The reason for using two inverters as well as two
delays will become apparent in a mathematical description of the
operation to be provided hereinafter. Each of the delays 102
provides a suitable phase shift between the signals emanating from
the right and left transducers 28 as is required for producing the
desired directivity pattern. The center transducer 28 obtains its
signal via the summer 132 which sums together the delayed signal
provided by a pair of delay units 102 and the inverters 130. It is
evident that the diagram of FIG. 7 can be simplified, as shown in
FIG. 8, however, the diagram of FIG. 7 parallels that of FIG. 4 and
thus shows an arrangement suitable for forming a polypole radiation
pattern just as the diagram of FIG. 4 is suitable for recieving a
polypole radiation pattern.
Referring now to FIG. 8 there is shown an alternative transmitting
beam former 137 which may be substituted in place of the
transmitting beam former 40. The beam former 137 is simpler than
the beam former 40 in that the three delay units 102 of FIG. 7 have
been replaced in FIG. 8 with two delay units 138 similar in
operation to the delay units 102 but having different delays. In
addition two selector switches 140 are provided to select either
the signal on line 128 or the inverted form of the signal and
thereby provide a more general synthesis of directivity
patterns.
Referring now to FIG. 9 there is shown a schematic diagram of the
arithmetic units and delay units of a beam former 142 which is
similar to the receiving beam former 38 of FIG. 1 except that the
beam former 142 is shown in a generalized form which is suitable
for any number of radiating elements such as the transducers 28.
Arithmetic units 144, some of which are further identified by the
letters A-C, are shown providing the differences between signals of
neighboring transducers 28 while delay units 146 are shown
providing the appropriate delays. In the beam former 38 of FIG. 4,
the arithmetic units 100 are shown as being capable of forming
either the sum or the difference of signals from neighboring
transducers 28, however, in FIG. 9 the arithmetic units are
presumed to be forming only the difference and this is adequate for
demonstrating one form of higher order system in which any number
of elements may be utilized for multiple radiation patterns. It is
also noted that the delay units 146 in the first row of the diagram
of FIG. 9 are of equal value as is indicated by the symbol
.tau..sub.1 ; similarly the values of the delays provided by the
delay units 146 in the second row of the diagram are of equal value
as is indicated by the symbol .tau..sub.2, and that furthermore,
there is one less delay in the second row than in the first row of
the diagram. Similar comments apply to the third and subsequent
rows of the diagram. Control lines 148 select the desired amount of
delay, each control line 148 being energized from a suitable source
similar to the angle tracker 54 of FIG. 1.
Referring now to FIG. 10 there is shown a simplified embodiment of
a beam former 150 which may be used in lieu of the receiving beam
former 38 of FIG. 1 in the situation wherein a polypole radiation
pattern is to be received. The beam former 150 comprises four
resistors 151-154, a delay unit 156 and summer 158. The appropriate
values for the resistors and the value of delay of the delay unit
156 will be derived in terms of the values of delay utilized in the
beam former 38 of FIG. 4 in the ensuing mathematical analysis.
The directivity patterns obtainable with a pair of radiating
elements, such as the elements A and B seen diagrammatically in
FIG. 11, may readily be expressed mathematically. A plane wave of
radiation having an angular frequency .omega. is impinging upon the
array of the two elements A and B at an angle .theta. to an axis of
the array. Each radiating element is an omnidirectional hydrophone
and has s sensitivity of K volts/microbar. If a pressure of 1
microbar is assumed, the voltages provided by each element in
response to the impinging radiation is, in exponential
notation,
v.sub.A =Ke.sup.j.sup..omega.t
e.sup.j.sup..pi.(D/.sup..lambda.)cos.sup..theta. (1)
and
v.sub.B =Ke.sup.j.sup..omega.t
E.sup.-.sup.j.sup..pi.(d/.sup..lambda.)cos.sup..theta. (2)
respectively for the elements A and B where the point of zero phase
is taken for convenience at the midpoint between the two elements
and where d is the interelement spacing. The sum of these two
voltages forms a monopole v.sub.M given by
v.sub.M =2Ke.sup.j.sup..omega.t
cos[.pi.(d/.lambda.)cos.theta.].apprxeq.2Ke.sup.j.sup..omega.t
(3)
the approximation being valid for small values of d/.lambda. such
that
.pi.(D/.lambda.)cos.theta..ltoreq.20.degree.
approximately. The difference of these two voltages is proportional
to the pressure gradient and forms a dipole v.sub.D given by
v.sub.D =2jKe.sup.j.sup..omega.t
sin[.pi.(d/.lambda.)cos.theta.].apprxeq.2jK.pi.(d/.lambda.)e.sup.j.sup..om
ega.t cos.theta. (4)
the approximation being valid similarly for small values of the
argument .pi.(d/.lambda.)cos.theta.. A cardioid response is
obtained by combining the monopole and dipole patterns
(j.pi.d/.lambda.)v.sub.M +v.sub.D wherein the factor
(j.pi.d/.lambda.) is the same as that seen in the expression for
v.sub.D in Equation 4. The cardioid expression contains a term
invariant with .theta. and a term dependent on .theta.. A
quadrupole radiation pattern has a term dependent on cos.sup.2
.theta. or cos2.theta.. The general representation of the voltage
P.sub.N (.theta.) provided by the beam pattern of a polypole line
array of (N+1) elements may be expressed as ##SPC1##
where .theta.=0.degree. and 180.degree. along the endfire
directions of the line, and where the a.sub.n are normalized such
that ##SPC2##
Since reciprocity applies, the same expression may be utilized for
a transmitting array.
Considering the case of the three elements A, B and C in the array
of FIG. 11 and, for convenience, setting the point of zero phase at
the radiating element nearest the impinging wave, namely, element
A, the output voltages are respectively
v.sub.A =Ke.sup.j.sup..omega.t (7)
v.sub.B =Ke.sup.j.sup..omega.t
e.sup.-.sup.j2.sup..pi.(d/.sup..lambda.)cos.sup..theta. (8)
v.sub.C =Ke.sup.j.sup..omega.t
e.sup.-.sup.j4.sup..pi.(d/.sup..lambda.)cos.sup..theta. (9)
These voltages are readily combined in accordance with the
invention by reference to FIG. 9 where the array is understood to
contain only the first three transducers 28 further identified by
the letters A, B and C. Three arithmetic units 144A-C and three
delay units 146A-C are utilized with the three element array and
the combined output voltage appears on line 160. It is presumed for
the purposes of this calculation that each arithmetic unit 144
forms the difference of two voltages without performing any
scaling. Each of the delay units 146 are presumed for the purposes
of this calculation to be analog delays (for ease of mathematical
representation) with the delays and resultant phase shifts
expressed in terms of the interelement spacing qd and rd where q
and r are constants, typically fractions. Thus, letting .tau..sub.1
and .tau..sub.2 be delays equivalent to the incident wave
propagating through distances qd and rd respectively, the voltage
present on line 162 is given by
V.sub.162 =2jKe.sup.j.sup..omega.t
sin[.pi.(d/.lambda.)(q+cos.theta.)]e.sup.-.sup.j.sup..pi.(D/.sup..lambda.)
(Q.sup.+cos.sup..theta.) (10)
the voltage on line 164 is given by
V.sub.164
=2jKe.sup.j.sup.[.sup..omega.t.sup.-2.sup..pi.(d/.sup..lambda.)cos.sup..th
eta..sup.]
sin[.pi.(d/.lambda.)(q+cos.theta.)]e.sup.-.sup.j.sup..pi.(d/.sup..lambda.)
(q.sup.+cos.sup..theta.) (11)
and the voltage on line 160 is given by
V.sub.160 =-
4Ke.sup.j.sup.[.sup..omega.t.sup.-.sup..pi.(d/.sup..lambda.)(r.sup.+Q.sup.
+2cos.sup..theta.).sup.]
sin[(.pi.d/.lambda.)(q+cos.theta.)]sin[(.pi.d/.lambda.)(r+cos.theta.)].
(12)
For small values of the arguments .pi.(d/.lambda.)(q+cos.theta.)
and .pi.(d/.lambda.)(r+cos.theta.) in the expression for the
voltage on line 160, the peak amplitude of this voltage may be
approximated by
V.sub.p =4K(.pi.d/.lambda.).sup.2 (q+cos.theta.)(r+cos.theta.)
(13)
or
V.sub.p =4K(.pi.d/.lambda.).sup.2
(1/p)[pqr+p(q+r)cos.theta.+pcos.sup.2 .theta.] (14)
where (1/p) serves a normalization factor (1/p=qr+q+r+1) in
Equation 14, and in which the bracketed term is recognized as a
power series such as the polypole representation of Equation 5. It
is also interesting to note that for small values of these
arguments the voltages on the lines 162 and 164 are approximated by
the sum of a monopole and a dipole radiation pattern, the monopole
being provided by the factor q and the dipole being provided by the
factor cos.theta..
The three constants pqr, p(q+r) and p may be given any desired
values for providing a desired directivity pattern to the three
element array. In particular, a maximum directivity, in the sense
of a maximum fraction of the overall radiant energy being found in
the main lobe, is obtained in the three element case for the
following values, namely,
pqr=1/6
p(q+r)=1/3 (15) p=5/6
for which
q=0.29
r=0.69 (16)
The minus sign in front of q simply means that the delay units
146A-B should provide a time advance or equivalently, the delay
units 146A-B need be placed on the opposite input ports of the
arithmetic units 144A-B respectively so that a delayed voltage from
element A is combined with an undelayed voltage from the element B,
and similarly, with respect to elements B and C.
The preceding mathematical development also applies to the beam
former 38 of FIG. 4 in the case where the arithmetic units 100
perform simply a subtraction operation as did the arithmetic unit
144 of FIG. 9. In such a case, the simplified beam former 150 of
FIG. 10 closely approximates the beam former 38 and may be
substituted in its place as will now be seen.
Referring now to the signal flow diagrams of FIGS. 12, 13 and 14,
the simplified configuration of the beam former 150 of FIG. 10 is
now derived. There are provided in FIGS. 12-14 delay units 166-170,
and in each delay unit the exponential phase factor of the
preceding mathematics is shown. The radiating elements are
identified as in FIG. 9 by the letters A, B and C. Inverters 174A-C
are provided in FIG. 12, and each applies a 180.degree. phase shift
to a sinusoidal signal waveform applied thereto from delay units
respectively 166-168. These inverters 174A-C will be replaced by
inverter 176 in FIGS. 13 and 14. Summers 178A-C in FIG. 12 combine
the various signals by adding them together, the subtraction of
FIG. 9 being accomplished by the use of the inverters 174A-C. In
FIGS. 13 and 14, these summers 178A-C will be replaced by summer
180.
The beam former of FIG. 12 is functionally equivalent to that
portion of the generalized beam former of FIG. 9 comprising the
first three radiating elements A-C and the delay units 146A-C
described above with reference to FIG. 9; and the preceding
mathematics applies also to FIG. 12. In FIG. 13 the delays
presented to a signal from radiating element C by the delay units
167 and 168 have been combined into a single delay provided by the
delay unit 169, and the delayed and inverted signals from radiating
element B are now shown by an equivalent representation wherein
inverter 176 replaces the inverters 174A and 174C. Since, in the
FIG. 13 a single delay unit, namely, delay unit 169, provides all
of the delay for the radiating element C, the summing together of
the various signals can now be accomplished with a single summer,
the summer 180.
The two delays provided by delay units 166 and 168 in FIG. 13 can
be combined into the single delay of delay unit 170 of FIG. 14 in
the following manner. In summing together the signals of the delay
units 166 and 168, it is seen that the exponential delay terms add
(since the two signals are otherwise equal and may be factored out)
with the result
e.sup.-.sup.j2.sup..pi.qd/.sup..lambda.
+e.sup.-.sup.j2.sup..pi.rd/.sup..lambda. =
2cos[.pi.(r-q)d/.lambda.]e.sup.-.sup.j.sup..pi.(Q.sup.+r)d/.sup..lambda.
.apprxeq.2e.sup.-.sup.j.sup..pi.(q.sup.+r)d/.sup..lambda. (17)
where .pi.(r-q)d/.lambda. is small. The factor of 2 is provided for
in the diagram of FIG. 14 by scaling the signals of the elements A
and C by a factor of 1/2.
FIG. 10 is readily seen to be equivalent to FIG. 14 since the
resistors 151-154 provide for the scaling of 1/2, these resistors
being of equal value, and the inverter 176 as well as the
amplifiers of the multiplexing circuitry 26 function as voltage
sources. With respect to the signal from element A, the signal
splits between the resistors 152 and 153 so that one-half goes to
the summer 158 and one-half goes to the delay unit 156. Thus, at
the output of the summer 158 there appears the following
combination of delay factors
1/2e.sup.-.sup.j2.sup..pi.(q.sup.+r)d/.sup..lambda.
+1/2=cos[.pi.(q+r)d/.lambda.]e.sup.-.sup.j.sup..pi.(q.sup.+r)d/.sup..lambd
a. .apprxeq.e.sup.-.sup.j.sup..pi.(q.sup.+r)d/.sup..lambda.
(18)
where .pi.(q+r)d/.lambda. is small. The final expression in the
above equation is seen to be the delay factor in the delay unit 170
of FIG. 14.
Referring now to the block diagram of FIG. 15, the angle tracker 54
comprises four programming units, respectively, a search selector
programmer 182, a search pattern programmer 184, a track selector
programmer 186, and a track pattern programmer 188, a switch 190
for selectively coupling the programming units 182 and 186 to a
memory 192, and a switch 194 for selectively coupling the
programming units 184 and 188 to a memory 196. The angle tracker 54
further comprises a detector 198 and a predictor 200 both of which
are responsive to signals on line 56 from the correlator of FIG. 1,
the detector 198 actuating the switches 190 and 194, and the
predictor adjusting the programs of the track selector programmer
186 and the track pattern programmer 188.
In operation, the detector 198 senses the magnitude of the
correlator signal on line 56 to determine the presence of a distant
source of sound such as the reflected waves 46 from the boat 44 in
FIG. 1. The detector 198 has a preset threshold for determining
that such a distant source of sound is or is not present. When no
such source of sound is present, the switch 190 couples the search
selector programmer 182 to the memory 192, and the switch 194
couples the search pattern programmer 184 to the memory 186. The
memory 192 stores data with respect to specific ones of the
transducers 28 of FIG. 1 which are to be coupled by the beam
switcher 64 for orienting the radiation directivity pattern of the
array 22 in a specific direction. The search selector programmer
182 provides a succession of directions, in response to the clock
pulses at terminal C, for redirecting the directivity pattern of
the array 22 for searching for a distant source of acoustic
radiation. These directions are sent from the search selector
programmer 182 via the switch 190 to the memory 192 which, in
response thereto, provides signals along line 62 for operating the
beam switcher 64 to couple the appropriate ones of the transducers
28. In a similar manner, the search pattern programmer 184
sequentially selects specific shapes for the directivity pattern
during each orientation of the directivity pattern as provided by
the beam switcher 64 to aid in the searching. Accordingly, the
search pattern programmer 184 designates a specific shape via the
switch 194 to the memory 196 which, in response thereto, commands
the appropriate set of delays and scaling factors for the beam
formers 38 and 40 of FIG. 1, these commands being transmitted in
the form of digital signals on the lines 58 and 60.
When the detector 198 determines that a suuitable source of
acoustic radiation is present, it actuates the switches 190 and 194
to couple the track selector programmer 186 to the memory 192, and
the track pattern programmer 188 to the memory 196. The track
selector programmer 186 continually reorients the directivity
pattern of the array 22 of FIG. 1 so that the source of acoustic
radiation is present on alternating sides of a null in the
directivity pattern, this providing effectively a tracking error
signal analogous to the tracking error signal developed by a
monopulse feed in a radar antenna. Similarly, the track pattern
programmer 188 commands the bean formers 38 and 40 via the switch
194 and the memory 196 to adjust the directivity pattern to provide
a shape more amenable to tracking the source of radiation.
The predictor 200 monitors the angle coordinates of successive
positions of the source of radiation and instructs the track
selector programmer 186 to reorient the directivity pattern when
the source of radiation tends to move away from the null in the
directivity pattern. When the source of radiation moves an angular
distance smaller than an angular increment that can be provided by
the beam switcher 64, the predictor 200 commands the track pattern
programmer 188 to move the null of the directivity pattern to
accommodate the angular movement of the source of radiation rather
than to command the track selector programmer 186 to update the
orientation of the directivity pattern. The average angular
orientation of the null in the directivity pattern corresponds to
the angular coordinate of the source of radiation, and this
information is transmitted from the track selector programmer 186
along line 66 to the display 70 of FIG. 1. To facilitate
discrimination betweeen targets of differing ranges from the array
22 of FIG. 1, the correlator 50 of FIG. 1 is provided with a range
gate, not seen in the figures, which, in response to a signal from
the timing unit 72, passes only those pulses on line 56
corresponding to a radiation source at the desired range, this
range being selected by a manual control provided on line 201 in
FIG. 1.
It is also apparent that while the preceding discussion has
described the implementation of a directivity pattern formed by a
linear array of three of the transducers 28 of FIG. 1, it is
apparent that four or more of the transducers 28 can be switched
into the circuit to further tailor the directivity pattern to a
desired shape so that it is possible to keep a null of the
directivity pattern oriented towards the stern of the boat 82 to
reduce the effects of propeller noise. It is also apparent that a
cylindrical array of the transducer elements 28 can be provided by
coupling all of the transducers 28, except for the central
transducer 28, to the beam formers 38 and 40. Such an
interconnection of the transducers 28 with the beam formers 38 and
40 can be accomplished, for example, by coupling together the pair
of the transducers 28 labeled A and D, the pair of transducers
labeled F and E, and the pair of transducers labeled E and C with
these three pairs of transducers 28 being connected to the beam
formers 38 and 40 in lieu of the previous connections of the
transducers indicated earlier in FIG. 9 with reference to the
transducers A, B and C.
The description of the preferred embodiment of the invention,
hereinbefore presented, provided for an implementation of the
expression for a polypole radiation pattern as given by Equation 5.
An analogous embodiment of the invention will now be presented with
reference to FIGS. 16-18 for providing a polymode radiation pattern
given by Equation 19 ##SPC3##
which is readily seen to be of the same form as Equation 5 except
that the summation is seen to be a Fourier series rather than a
power series. For the three element array considered in FIG. 2, or
the eight element array to be described in FIG. 16, the Fourier
series of Equation 19 can readily be converted by trigonometric
identities to the power series of Equation 5. Accordingly, the
terms polypole and polymode are seen to be convenient terms for
describing the mathematical series representation of multiple mode
types of radiation patterns.
Referring now to FIG. 16, there is seen a schematic diagram of an
alternative embodiment of the invention showing an array 202 of
radiating elements which are in the form of a cylindrical radiator
having inner and outer concentric metallic electrodes located on
the inner and outer surfaces of a ceramic cylinder 204. The outer
electrode is identified by the numeral 206 while the inner
electrode is segmented into eight segments, each of which are
insulated from each other and are identified generally by the
numeral 208 with specific segments being further identified by the
letters A-H. In the most general form the array is of spherical or
spheroidal shape; however, the preferred practical configuration is
cylindrical when, for example, elevation angles are secondary to
azimuthal angles.
The array 202 is shown coupled to a receiver 210 having eight
preamplifiers which will be identified generally by the numeral 212
with individual ones of these preamplifiers 212 being further
identified by the letters A-H which correspond respectively to the
eight inner electrode segments 208A-H. The inner electrode segments
208 are coupled to their respective preamplifiers 212 via wires
214. Each of the preamplifiers 212 are also identified in the
figures by legends such as NNE meaning north-by-northeast and WSW
meaning west-by-southwest. A coordinate axis indicating the
directions north, east, south and west is also seen overlaid upon
the array 202.
The receiver 210 further comprises a first set of operational
amplifiers 216 and a second set of operational amplifiers 218 with
individual ones of these amplifiers being identified respectively
by 216A-G and 218A-H. A filter 220 is provided at the output of the
operational amplifier 216A and filters 222A-B are provided at the
outputs of the operational amplifiers respectively 216F-G.
Adjustable attenuators 223 are coupled to the outputs of the
filters 220, 222A and 222B as well as to the outputs of the
amplifiers 216B-E for scaling the respective signals, these
attenuators being further identified by the letters A-G
corresponding respectively to the amplifiers 216A-G. The
operational amplifiers 216 are further identified by legends such
as the legend M representing a monopole or an omnidirectional
radiation pattern, NW/SE representing a dipole radiation pattern
oriented with its axis running from northwest to southeast, 000
which identify a quadrupole radiation pattern having its reference
axis at an angle of 0.degree. (along the north-south axis) with
respect to the array 202, and 045 which represents a quadrupole
radiation pattern having its reference axis inclined at an angle of
45.degree. with respect to the array 202. Each of the preamplifiers
212 is connected to all of the operational amplifiers 216, however,
for purposes of simplifying the drawing, these interconnections are
indicated by means of the numerals 1-8 identifying the output
terminals of the preamplifiers 212 and the corresponding numerals
1-8 at each of the eight input terminals to each of the seven
operational amplifiers 216. As is well known, operational
amplifiers are typically provided with a positive input and a
negative input with summing resistors connecting with both the
positive and the negative inputs to provide an algebraic
combination in which a plurality of signals may be summed together
and subtracted from a second plurality of signals which are summed
together.
The filters 220 and 222A-B serve as equalizing networks to equalize
the frequency response of the omnidirectional pattern and the
frequency response of the quadrupole pattern with that of the
dipole pattern. As is well known from the mathematics of dipole
patterns in acoustic systems, the receiving frequency response of
ceramic transducers for frequencies below resonance tends to rise
at a rate of 6 dB/oct (decibels per octave) change in frequency.
Accordingly, the omnidirectional pattern which has a flat frequency
response below resonance is passed through the filter 220 which may
be, for example, a lead network, introducing a phase shift of
+90.degree. and having a frequency response of +6 dB/oct. Since
quadrupole radiating patterns have a frequency response for
frequencies below the second order mode resonance frequency of the
array 202, wherein the amplitude of signals provided by the
amplifiers 216F and 216G rises at a rate of +12 dB/oct, each of the
filters 222A-B which may be, for example, a lag network, provide a
phase shift of -90.degree. and a frequency responsivity of -6
dB/oct. Thus the signals on each of the lines 224 are characterized
by a frequency responsivity which varies at the rate of +6 dB/oct
for all frequencies below the resonance region of the array 202. In
the situation where a source of sound 226 transmits sound waves 228
towards the array 202, the sound waves 228 having become
substantially planar waves in the vicinity of the array 202, the
array 202 is built with a diameter which is sufficiently smaller
than the wavelength of the waves 228 so that the frequency of these
waves is below the resonance region of the array 202.
Each of the lines 224 is connected to at least some of the
operational amplifiers 218 but not necessarily all of these
amplifiers. Here again, in order to simplify the drawings, the
interconnection of individual ones of the lines 224 are indicated
by legends affixed to each of the inputs of the respective
operational amplifiers 218A-H. Thus, with respect to the
operational amplifier 218A, the omnidirectional signal provided by
the filter 220 is supplied to one input, the NE/SW dipole signal
provided by the operational amplifier 216D is applied to a second
input, and the quadrupole signal provided by the filter 222A is
applied to the third input of the operational amplifier 218A. As is
indicated by the plus signs adjacent each input of the operational
amplifier 218A, each of these inputs are summed together to give a
voltage V1 which is the voltage associated with one polymode
directivity pattern for receiving the waves 228, as will now be
described with reference to FIG. 17.
Referring now to FIG. 17, there is shown a schematic representation
of the combination of an omnidirectional radiation pattern 230, a
dipole radiation pattern 232 and a quadrupole radiation pattern 234
to give a polymode pattern 236 represented by the voltage V1 of the
operational amplifier 218A of FIG. 16. The omnidirectional pattern
230 is a graph of the voltage appearing at the output of the filter
220 of FIG. 16 as the source 226 is moved in a large circle around
the array 202. Similarly, the dipole radiation pattern 232 and the
quadrupole radiation pattern 234 represent the voltages provided by
respectively the operational amplifier 216C and the filter 222A as
the source of sound 226 is moved in a large circle around the array
202. The summation of these voltages by the operational amplitude
218A is represented in FIG. 17 by the pictorial equation having two
plus signs and an equal sign. It is noted that the polymode pattern
236 has one main lobe in the north direction corresponding to the
contributions of the northerly portions of the three radiation
patterns 230, 232 and 234, each of which have a positive voltage in
the northerly direction. On the east and west axis the negative
portions of the lobes of the quadrupole radiation pattern 234 are
portrayed as having equal and opposite amplitudes to the amplitude
of the omnidirectional radiation pattern 230 at the east and west
directions and therefore cancel each other so that zero amplitude
appears at the east and west directions of the polymode pattern
236. Similar comments apply to the construction of the polymode
pattern 236 and other directions. Mathematically this construction
is indicated by the algebraic expressions beneath each of the
radiation patterns 230, 232 and 234 in which the maximum amplitudes
of these radiation patterns have all been set equal to one-third so
that the peak amplutude of the polymode pattern 236 is equal to
unity.
It is frequently desirable to scale the respective amplitudes of
the respective radiation patterns 230, 232 and 234 to provide a
maximum directivity for the polymode pattern 236 with the maximum
directivity being understood to be a maximization of the ratio of
energy received from the northerly direction of the radiation
pattern 236 to the total isotropic incident energy received by all
of the lobes of the pattern 236. This can be readily accomplished
mathematically by integrating the energy received in each direction
and forming the standard ratio for the directivity index, with the
result that the maximum directivity is obtained for the scaling
factors indicated in the attenuators 223A-G of FIG. 16. These
scaling factors are, respectively, a value of 3 for the attenuator
223A a value of 4 for each of the attenuators 223B-E, and a value
of 5 for each of the attenuators 223F-G. In FIG. 16, the arrow
adjacent the scaling factor in each of the attenuators 223
indicates that the attenuation of these attenuators 223 may be
varied to provide selective scaling of the monopole, dipole and
quadrupole radiation patterns to produce a polymode pattern in
which the positions of the nulls can be varied as well as for
providing a selective tailoring of the shapes of the various lobes
in the polymode radiation pattern.
Returning momentarily to FIG. 16, the operational amplifier 218B is
seen to provide a polymode pattern having the same shape as the
polymode pattern 236 of FIG. 17 but being oriented at an angle of
45.degree. with respect to the coordinate system of the array 202.
This results because of the summation of the monopole signal of the
filter 220 with the signal of the operational amplifier 216B and
the quadrupole signal of the filter 222B, for which the dipole
radiation pattern and the quadrupole radiation patterns have their
main axis in the 45.degree. or NW/SE direction. With respect to the
voltage V3, provided by the operational amplifier 218C, a polymode
radiation pattern perpendicular to that of the polymode pattern 236
of FIG. 17 is provided since here the quadrupole pattern provided
by the filter 222A is subtracted from the sum of the monopole
pattern of the filter 220 and the east-west dipole pattern of the
operational amplifier 216E; this subtraction is indicated by the
minus sign adjacent the third input of the operational amplifier
218C. Similar comments apply to the remaining voltages V4-V8
wherein it is seen that a family of eight polymode radiation
patterns are simultaneously produced by the receiver 210.
Referring now to FIG. 18, there is seen a system 238 comprising the
array 202 and the receiver 210 of FIG. 16, here shown coupled via
multiplexing circuitry 240 similar to the multiplexing circuitry 26
of FIG. 1. A signal generator 242 similar to the signal generator
42 of FIG. 1 provides signals which are coupled via the
multiplexing circuitry 240 to the array 202 for transmission
therefrom as in an active sonar system. The receiver 210 receives
either echoes, which are reflected from an object such as the boat
44 of FIG. 1, or signals generated by a distant source of sound
such as the source of sound 226 of FIG. 16. The direction of such a
distant source of sound can readily be determined by means of logic
circuitry 244 which performs the function of noting which of the
voltages V1-V8 has a maximum value. A display 246, such as a
cathode ray tube display, is coupled to the logic circuitry 244 for
displaying the direction of the distant source of sound.
Additionally, a timing unit 248, similar to the timing unit 72 of
FIG. 1, and a range gating unit 250 may be utilized in an active
sonar mode for gating out all received signals except those
corresponding to echoes emanating from a preselected range from the
array 202. To implement this range gating, the timing unit 248
supplies timing signals to both the signal generator 242 and the
range gating unit 250 so that a range gate can be set corresponding
to a preselected time interval subsequent to the transmission of a
signal generated by the signal generator 242. The range gating unit
250 is coupled to the receiver 210 at the "gate" terminals of the
preamplifiers 212 (FIG. 16) to gate out all the polymode voltages
of the receiver 210 except for such voltages falling within the
designated range gate.
Referring now to FIG. 19 there is shown a block diagram of a
receiver 250 which is similar to the receiver 210 of FIG. 16 in
that it utilizes the eight preamplifiers 212, five of the
operational amplifiers 216, namely, the operational amplifiers
216A, 216C, and 216E-G, and one of the operational amplifiers 218.
Each of the preamplifiers 212 connect via the wires 214 to the
array 202 as in FIG. 16. The receiver 250 differs from the receiver
210 in that the receiver 250 combines the output from a pair of
dipole operational amplifiers, namely, the north-south and
east-west dipole operational amplifiers 216C and 216E via a
coordinate converter 252 to obtain a dipole, represented by a
voltage on line 254, with an axis having any desired orientation
with respect to the coordinate system of the array 202. The
omnidirectional channel comprises the operational amplifier 216A
and the filter 220 of FIG. 16 but no coordinate converter is
utilized since the omnidirectional pattern is symmetric about the
axis of the array 202. A quadrupole radiation pattern oriented in
any direction about the axis of the array 202 is provided by the
operational amplifiers 216F-G and the filters 222A-B of FIG. 16 in
combination with a coordinate converter 256 providing a voltage on
line 258 representing the magnitude of signals received by the
polypole radiation pattern.
The coordinate converters 252 and 256 comprise multipliers 260A-D,
trigonometric units 262A-B and amplifiers 264A-B each of which has
a variable gain control as indicated by the legend G.sub.1 and
G.sub.2. The output signals of the multipliers 260A and 260B are
summed together by the amplifier 264A, and the output signals of
the multipliers 260C and 260D are summed together by the amplifier
264B. The multipliers 260A-D may operate in an analog or in a
digital fashion. Thus, the multiplier 260A multiplies the
north-south dipole signal by cos.theta..sub.1 and the multiplier
260B multiplies the east-west dipole signal by sin.theta..sub.1
whereupon the two products are summed together to give a resultant
dipole having the same amplitude as the north-south dipole but
having a dipole axis inclined at an angle .theta..sub.1 relative to
the axis of the array 202. Subscripts 1 and 2 are utilized with the
angle .theta. to indicate that the dipole pattern may be rotated at
different amounts from the quadrupole pattern. Similar comments
apply to the operation in the coordinate converter 256 except that
here the multiplying factors are cos(2.theta..sub.2) and
sin(2.theta..sub.2) since the expression for the quadrupole
radiation pattern is in terms of cos(2.theta.), while that of the
dipole is in terms of cos.theta. where .theta. is the angle
relative to the coordinate system of the array 202.
With respect to the capability of rotating the dipole patterns
relative to the quadrupole pattern, as seen with the embodiment of
FIG. 19, it is noted that such a capability exists also in the
embodiments of FIGS. 1 and 16 but to a limited extent. Thus, in
FIG. 1 a rotation of 180.degree. imparted by the beam switcher 64
rotates the dipole pattern 180.degree. while rotating the
quadrupole pattern 0.degree., .+-. 90.degree. or 180.degree. since
each of these rotations are fully equivalent with the quadrupole
pattern which is periodic with each rotation of 180.degree.. With
respect to the embodiment of FIG. 16, a dipole pattern can be
rotated with respect to a quadrupole radiation pattern by simply
selecting the outputs from a different operational amplifier 216
such as the combination of the outputs of the operational amplifier
216C and 216F as is provided by the operational amplifier 218A or
the combination of the outputs of the operational amplifiers 216E
and 216F as is provided by the operational amplifier 218C.
To implement the coordinate conversion digitally, the multipliers
260A-D would comprise an analog-to-digital converter, a digital
multiplier, and a digital-to-analog converter (not seen in the
drawings). The trigonometric units 262A-B in digital form would
comprise, by way of example, a read-only memory in which the value
of cos.theta..sub.1 and of sin.theta..sub.1 would be presented in
response to an address supplied on the line labeled .theta..sub.1 ;
and with respect to the trigonometric unit 262B the values of
cos(2.theta. .sub.2) and sin(2.theta..sub.2) would appear in
response to an address supplied along the line labeled
.theta..sub.2.
To implement the multiplication of the multipliers 260A-D in analog
fashion, the multipliers would comprise, for example, gain control
amplifiers in which the gain would be proportional to
cos.theta..sub.1 in the case of the multiplier 260A, and similarly
for the other multipliers 260B-D. With respect to the trigonometric
units 262A-B, in analog form they would comprise, for example, a
nonlinear diode-resistor circuit in which the output voltage
approximates cos.theta..sub.1 in response to input voltage of
.theta..sub.1 with a second such circuit being provided to produce
sin.theta..sub.1 in response to .theta..sub.1.
The omnidirectional channel also comprises an amplifier 266 coupled
to the output of the filter 220 and having a variable gain in
accordance with a voltage applied at terminal G.sub.0 for varying
the gain of the omnidirectional pattern relative to the gains of
the dipole and quadrupole patterns. The relative gains for the
three channels, namely, G.sub.0, G.sub.1 and G.sub.2, as well as
the angles of rotation of the dipole, .theta..sub.1, and of the
quadrupole .theta..sub.2, are provided by a beam former 268 in
response to signals provided by a logic circuit 270. The logic
circuit 270 is similar to the logic circuit 244 of FIG. 18 and is
useful for tracking a source of sound such as the source 226 of
FIG. 16. Thus, the logic circuit 270 in response to the polymode
voltage, V, at the output of the operational amplifier 218 provides
signals to the beam former 268 for rotating the polymode
directivity pattern, and furthermore, for varying the shape of the
directivity pattern by varying the gain G.sub.0, G.sub.1, and
G.sub.2. The shape of the directivity pattern can also be varied by
making .theta..sub.1 different from .theta..sub.2, this providing a
flexibility not found in the receiver 210 of FIG. 16. The receiver
250 and the logic circuit 270 may be substituted in the system 238
of FIG. 18 in place of the receiver 210 and the logic circuit 244
to provide a tracking system with improved tracking capability.
It is also evident with respect to the receivers 210 and 250 of
FIGS. 16 and 19 that an octopole radiation pattern could be formed
by summing and differencing alternate outputs of the amplifiers
212. In this respect it is noted that alternate pairs of inputs to
the operational amplifiers 216F-G are summed and differenced to
provide the quadrupole radiation pattern, there being two such
patterns spatially offset by 45.degree.. In the case of the
octopole pattern, only one fixed pattern could be produced with the
eight inner electrode segments 208, it being necessary to employ 16
such segments for providing a second octopole radiation pattern,
and similarly for providing a combination of these patterns to give
a resultant octopole radiation pattern in any arbitrary direction.
Via the same reasoning, it is also evident that if only four inner
electrode segments 208 were employed in the array 202, then one
fixed quadrupole radiation pattern could also be produced; however,
there would be no facilities for altering the directional quality
of such a pattern.
it is also apparent from a comparison of the array 202 of FIG. 16
and the array 22 of FIG. 1 that the two arrays may be combined into
a single composite array (not seen in the figures) in which one or
more transducers 28 of the array 22 are replaced with an array
having the form of the array 202. In view of the fact that the
diameter of the array 202 is substantially smaller than a
wavelength, such a radiator may readily be substituted for a
transducer 28. With respect to connecting this composite or hybrid
array to the multiplexing circuitry 26 and the transceiver 24, the
multiplexing circuitry 26 must be enlarged to accommodate a
transmit-receive circuit for each of the wires 214 of FIG. 16 for
each one of the arrays 202 utilized in the composite array. A
plurality of the receivers 210 of FIG. 16 are utilized, one for
each of the arrays 202 with the omnidirectional channel in each of
these receivers being connected to the receiving beam former 38 of
FIG. 1. This provides a composite receiving system having both
features of the system 20 of FIG. 1 and features of the system 238
of FIG. 18, and may be referred to as a polypole, polymode
receiving system; the composite array may similarly be referred to
as a polypole-polymode array. Transmission of signals generated by
the signal generator 42 of FIG. 1 may still be accomplished in the
polypole fashion by energizing each of the inner electrode segments
208 of an array 202 in parallel circuit from a single output of the
transmitting beam former 40 of FIG. 1.
A polypole or polymode or hybrid array may also be constructed in
three dimensions, such as a four element array in which each of the
radiating elements are located at the vertices of a tetrahedron.
Such an array is readily implemented for polypole operation with
radiating elements having a spherical radiating surface so as to
radiate an omnidirectional pattern in three dimensions. In the case
of the polymode operation, a radiating element must be segmented in
three dimensions as by utilizing sectors of a sphere. The resultant
radiating patterns of the tetrahedronal array are significantly
more complex than those presented hereinbefore for the planar
array, as may be understood by noting that the projections of the
interelement spacing upon an axis of the radiation pattern is
dependent on the orientation of the axis relative to the
tetrahedron configuration. In particular, there are six major axes
each of which passes through the center of the tetrahedron and is
parallel to an edge of the tetrahedron. It may also be desirable to
place a radiating element at the center of the tetrahedron to
provide greater flexibility in forming the radiation pattern in any
desired direction. The same form of circuitry as has been disclosed
hereinbefore, may be utilized for the tetrahedronal array, however,
the beam forming circuitry and the logic control for the beam
forming circuitry is extended to encompass all three dimensions in
a manner which may be seen by extending the preceding mathematics
to three dimensions.
It is understood that the above described embodiments of the
invention are illustrative only and that modifications thereof will
occur to those skilled in the art. Accordingly, it is desired that
this invention is not to be limited to the embodiments disclosed
herein but is to be limited only as defined by the appended
claims.
* * * * *