U.S. patent number 3,750,175 [Application Number 04/879,775] was granted by the patent office on 1973-07-31 for modular electronics communication system.
This patent grant is currently assigned to Texas Instruments Incorporated. Invention is credited to Ray E. Cooper, George E. Goode, Robert M. Lockerd, Mark W. Smith.
United States Patent |
3,750,175 |
Lockerd , et al. |
* July 31, 1973 |
MODULAR ELECTRONICS COMMUNICATION SYSTEM
Abstract
A communication system including a plurality of radiating
elements formed into an antenna array for transmitting and
receiving communication frequency signals and employing a central
processor to generate the transmitted signals and process the
received frequencies through a manifold arrangement. Each radiating
element connects to the manifold through a module made up of
integrated microwave circuitry including a mixer coupled to a local
oscillator and a phase shifter coupled to a beam steering computer.
By means of the beam steering computer the antenna can be made to
scan various preselected areas.
Inventors: |
Lockerd; Robert M. (Dallas,
TX), Smith; Mark W. (Dallas, TX), Cooper; Ray E.
(Richardson, TX), Goode; George E. (Richardson, TX) |
Assignee: |
Texas Instruments Incorporated
(Dallas, TX)
|
[*] Notice: |
The portion of the term of this patent
subsequent to January 5, 1988 has been disclaimed. |
Family
ID: |
27104631 |
Appl.
No.: |
04/879,775 |
Filed: |
November 25, 1969 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
Issue Date |
|
|
690638 |
Dec 14, 1967 |
3553693 |
|
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Current U.S.
Class: |
342/371;
342/377 |
Current CPC
Class: |
H01Q
3/34 (20130101); H01Q 3/38 (20130101); H01Q
21/0025 (20130101) |
Current International
Class: |
H01Q
21/00 (20060101); H01Q 3/30 (20060101); H01Q
3/34 (20060101); H01Q 3/38 (20060101); H01q
003/26 () |
Field of
Search: |
;343/1SA |
References Cited
[Referenced By]
U.S. Patent Documents
Primary Examiner: Borchelt; Benjamin A.
Assistant Examiner: Berger; Richard E.
Parent Case Text
This application is a continuation-in-part application of U.S. Pat.
No. 3,553,693, and assigned to the same assignee as this patent
application.
Claims
What is claimed is:
1. A communication system having a central processor for generating
a transmitted signal and processing a received signal
comprising:
an antenna array of a plurality of transmitting/receiving elements,
and
a transmit/receive module for each element of said antenna array,
said module including phase shift means for shifting the phase of a
transmitted signal generated by said central processor independent
of signals received at the antenna array, power amplification means
for increasing the power of said transmitted signal, frequency
converting means for converting a low frequency signal to a high
frequency transmit signal, a transmit channel coupling said phase
shift means and said power amplification means to said frequency
converting means to convey phase shifted transmitted signals of
increased power to said element, said module also including phase
shift means for selecting the phase of the received signal
processed by the central processor independent of the transmitted
signal, a filter for separating a received signal frequency from
other signals, and a receive channel being coupled to said filter
and said phase shift means for conveying a received signal to said
central processor, said receive channel including a mixer coupled
to said phase shift means.
2. A communication system as set forth in claim 1 wherein said
transmit/receive module includes in the receive portion a
multiplier to change the frequency of the output of the phase shift
means prior to coupling to said mixer.
3. A communication system as set forth in claim 2 wherein said
transmit/receive module includes in said receive channel an IF
filter-amplifier coupled to said mixer.
4. A communication system as set forth in claim 1 wherein said
frequency converting means is a multiplier.
Description
This invention relates to a communication system involving solid
state microwave modules, and more particularly to a modular
electronic communication system having a multi-element phased array
antenna.
While this invention is immediately advantageous in connection with
construction and operation of satellite and airborne communication
systems, it has application to other communication systems such as
those used for mobile ground applications and microwave
transmission.
In comparison to conventional communication systems, a modular
electronics system provides the advantages of less weight per watt
of effective radiating power, noninertial electronic beam steering
(where required), and increased system reliability because of the
use of integrated circuits and distributed-function, building block
construction. These advantages are of particular importance for
satellite communication and airborne communication systems.
However, they are also important to ground based systems especially
those requiring high reliability.
Satellite and airborne communication systems have been faced with
the problems of minimizing weight and increasing reliability while
generating high power microwave energy. Other major problems of
conventional mobile communication systems have been concerned with
auxiliary equipment such as rotary joints, servo motors for the
antennas, and the like. Restrictions imposed by such components on
reliability exist in the most modern of transistorized systems
produced for mobile service. Further, the magnetrons for
transmitting, and klystrons for local oscillator service all have
been found to restrict the reliability of the system.
In accordance with the present invention, there is provided an
improved communications system in which solid state circuitry is so
constructed and arranged as to be capable of overcoming the major
obstacle heretofore encountered in the development of such a
system, namely, the generation of high power microwave energy. This
problem is overcome by the use of solid state functional electronic
blocks or modules, so constructed as to operate as a modular
antenna array which may be responsive to beam steering control and
which may be operated at an adequate power level. The electronic
blocks may be designed to operate at almost any communications
frequencies. At the same time, such construction lends itself to a
lightweight multi-element antenna array using electronic beam
scanning thereby eliminating wave guides, rotary joints, motors,
synchros, gears, and other servo components normally essential to a
moving antenna communication system. As a result, a substantial
reduction in total volume and weight over known or existing systems
is achieved. This is accompanied by a substantial increase in the
reliability of the system.
Another advantage of the phase array integrated circuit
communication system is the ability to restrict the transmit and
receive beam width thereby increasing the antenna efficiency. In
conventional satellite or aircraft communication systems, the beam
width must be made sufficiently large to completely cover a desired
section plus an area outside the desired section necessitated by
tolerances in the satellite's or aircraft's attitude control
system. Because of the much wider beam width in the conventional
communication system, a transmitter must have a greater transmitter
power to achieve effective communication.
Beam pointing and control in a phased array is accomplished by the
setting of phase shifters in series with each radiating element. A
beam scanning computer calculates the necessary phase for each
element in order to collimate the beam at some position in space
and generates an individualized digital code for each element.
Typically, the scan computer is a digital device and the variable
phase shifter is either a digital device or a digitally controlled
analog device. Phased array antennas are formed by radiating
elements arranged in any desired pattern, for example, a planar
array of rows and columns. In order to control the direction of the
antenna beam, an individualized digital code is provided to the
phase shifter for each of the radiating elements in the antenna
array. In operation, the digital computer generates individual
codes for each radiating element depending on its position in the
array. This code is applied to the phase shifter to impart a
predetermined amount of phase shift to the signal received or
transmitted from the element. If each phase shifting code imparts a
particular degree of phase shift, the entire beam can be pointed in
any desired direction. The phase shifting code may be applied to
all the elements simultaneously or on a sequential basis. In the
sequential mode, one phase shifter at a time would be updated,
while all other shifters would remain unchanged.
Each radiating element in the antenna array is part of an antenna
module with individual power generation and phase control
circuitry. The use of solid state modules, together with microwave
transistors, permits operation in the X-band and higher. Production
of transmitting and receiving power at the desired frequency
involves the use of frequency multipliers and up-converters of
integrated circuit construction. Thus, each transmit module
includes the necessary circuitry for amplifying relatively low IF
energy applied simultaneously to all of the modules, and
multiplying the frequency to a higher frequency for transmission
from the transmit radiating elements. Each receive module includes
circuitry for processing one or more high frequency signals
received by the receive radiating elements to produce low frequency
IF signals, which are also preamplified. In addition, each module
includes phase shifting means for the transmitted or received
energy so that the beams to or from the fixed antenna array can be
electronically scanned.
The phased array integrated circuit communication system of this
invention as set forth in the appended claims includes a multiple
radiating element antenna coupled to a signal processor through
amplification and phase shifting devices. For X-band frequency
operation, a local oscillator output frequency is multiplied and
mixed with an IF signal for both the transmit and receive
operation.
A more complete understanding of the invention and its advantages
will be apparent from the specification and claims and from the
accompanying drawings illustrative of the invention.
Referring to the drawings:
FIG. 1 is a pictorial of a satellite mounted modular electronic
communication system employing independent transmit and receive
antenna arrays;
FIG. 2 is a basic block diagram of a complete phased array
integrated circuit communication system with independent transmit
and receive antennas;
FIG. 3 is a block diagram of a multiple signal low frequency
modular electronics receiving system;
FIG. 4 is a block diagram of a low frequency modular electronics
transmitting system;
FIG. 5 is a block diagram of a X-band modular electronics receive
and transmit communication system;
FIG. 6 is an integrated circuit mixer employing a surface-oriented
diode wafer and ceramic substrate;
FIG. 7 illustrates a video amplifier construction for the IF
preamplifier of FIG. 5;
FIG. 8 is a top view of an intergrated circuit embodiment of a
multiplier of FIG. 5;
FIG. 9 is an illustration of beam scanning; and
FIG. 10 is a block diagram of an X-band modular electronic
communications system using a single phased array antenna for
transmit and receive.
FIG. 11 is another embodiment of the modular electronic
communication system according to the present invention.
This invention will be described as it is employed as a satellite
communication system contained within a satellite 10, referring to
FIG. 1, having a phased array transmitting antenna 11 and a similar
phased array receiving antenna 12. A plurality of booms 13-18
provide a means for instructing the satellite 10 to maintain a
predetermined position relative to the earth. This orientation is
such that the antenna arrays 11 and 12 are always facing toward the
earth's surface. Electrical power to operate the communication
system of the present invention is provided by means of solar cells
19 covering all outer surfaces of the satellite 10 except for areas
covered by the antennas 11 and 12. Solar cell power supplies are
now well known in the art and additional description is not deemed
necessary.
By using separate transmitting and receive antennas 11 and 12
separated by several feet, greater isolation between transmit and
receive modes is possible. Each of the antennas 11 and 12 comprises
an array of individual radiating building blocks terminating in a
radiating element, such as the crossed dipole 21. Other radiating
element configurations that can be used include the slotted dipole
configuration, the orthogonal slot configuration, and various open
ended wave guides. The antennas 11 and 12 consist of no less than
two radiating elements with the maximum number limited only by
practical considerations. An antenna array for a 3-degree beam
width with 0.75.lambda. element-to-element spacing requires
approximately 550 radiating elements.
Each radiating element 21a, 21b, . . . 21n of the antenna 11,
referring to FIG. 2, is coupled to a manifold 22 through a modular
electronic transmitting building block 23a, 23b, . . . 23n,
respectively. A central processor 24 composed of a limiter and
power amplification section provides a signal to the manifold 22
for distribution to the radiating elements of the antenna 11. For
the system shown, the central processor 24 could be either a
frequency translation repeater or demod-remod system, either of
which to be a limiting or linear system. The relative amplitude and
phase of the signal supplied to each of the radiating elements are
controlled to obtain the desired radiation pattern from the
combined action of all the elements in the antenna 11. A local
oscillator 26 coupled to the manifold 22 provides power to the
transmitting building blocks for use in the frequency conversion
process where required. A beam steering computer 27, also coupled
to the manifold 22, provides the phasing information for beam
steering purposes.
A system similar to that described in the preceding paragraph is
also provided to process communication signals received by the
antenna 12 made up of radiating elements 28a, 28b, . . . , 28n.
Each signal received by elements of the antenna 12 is transmitted
to a manifold 29 through a modular electronic receiving block 31a,
31b, . . . , 31n. The beam steering computer 27 again provides
phasing information to the signals received by the antenna 12 prior
to processing by the central signal processor 24. Where required,
the local oscillator 26, coupled to the manifold 29, converts the
transmitted frequency into an IF signal for handling by the central
signal processor 24. Thus, the basic modular electronic
communication system contains six major elements: the receive
antenna array 12, the central processor 24, the transmit antenna
array 11, the manifolds 22 and 29, the beam steering computer 27,
and the local oscillator 26.
Referring to FIG. 3, there is shown a simple low frequency (2.0
GHz) receiving module including a filter and pre-amplifier 32 and
four phase shifters 33, 34, 36, and 37, one for each of four
possible signals received by a radiating element. The radiating
element 28 connects to the input of the pre-amplifier and filter
32. Each of the phase networks 33, 34, 36, and 37 receives a
control signal from and transmits a signal to an associated section
of the manifold 29. The various sections of the manifold 29 in turn
transmit and receive signals from the signal processor 24.
In an exemplary operation of a multiple phase shifter and manifold
receive system, the four receive signals from the element 28 are
coupled to each phase shifter 33, 34, 36 and 37. A given phase
displacement is applied to each signal in each phase shifter in
accordance with the control code connected thereto. The signals
from all the phase shifters coupled to one manifold are then summed
in such a manner that only one of the received signals will be
coherent. The remaining three will be non-coherent resulting in an
unintelligent signal (noise), which is rejected by the central
processor 24. Each phase shifter and manifold operates in a similar
manner to coherently add one received signal and non-coherently add
the other three.
A typical 2-GHz preamplifier includes thin film resistors and
capacitors, and several amplification stages employing chip
transistors in a microstrip circuit on a ceramic substrate. Such a
device will provide low noise amplification of the signal received
at the radiating element 28. Phase shifting with a 4-bit phase
shifter, such as phase shifters 33, 34, 36 and 37, is obtained by
switching two PIN diodes, one at each end of a quarter wave section
of transmission line. Typically, a shifter uses two parallel sets
of diodes on each quarter wave section. Thus, two bits of phase
shift are obtained by one section of line. With a receiving system
as shown in FIG. 3, various amounts of phase shift can be applied
to a signal received by the antenna element 28.
For a low frequency communication system, a transmit building block
for one radiating element is shown in FIG. 4 and includes an
amplifier 38 coupled to a 4-bit phase shifter 39. An antenna
element 21 is coupled to the amplifier 38 and a control network 41
connects to the phase shifter 39. A control code from the beam
steering computer 27 is coupled to the control network 41 through
the manifold 22. The phase shifter 39 receives a signal through the
manifold 22 from the signal processor 24. The amplifier 38 and the
phase shifter 39 are again microstrip circuits with discrete
components on a ceramic substrate as described with respect to the
receive building block of FIG. 3.
While the receive and transmit building blocks of a low frequency
system are relatively simple, including only an amplifier and a
phase shifter, the same building blocks for an X-band communication
system requires additional components as shown in FIG. 5. An
antenna element 28 receives a circularly polarized wave of about
8.3 GHz and converts it into the input of an unblanced strip line
42 leading to a mixer 43. Insofar as possible, the antenna element
28 rejects a transmit signal emitting from the antenna element 21
to provide a limited amount of isolation. As discussed previously,
a crossed dipole is one of many configurations for the antenna
element 28 and provides good isolation between transmit and receive
signals. Typically, the element 28 will be mounted on a 1 .times. 1
inch block face followed by a tapered coax balun.
Conversion of the RF received signal into an IF processing signal
takes place in the mixer 43 by means of a signal from a multiplier
44 connected to the output of a 2.235 MHz local oscillator through
a receive manifold 46. The mixer 43 utilizes a Schottky-barrier,
GaAs diode of the type described in the U.S. Pat. No. 3,388,000,
issued to Warren P. Waters. Metal semiconductor diodes, which are
known in the art as Schottky barriers, are commonly used in high
frequency circuits such as, for example, the mixer 43. The Schottky
barrier of the above United States Patent is fabricated on a
semiconductor substrate to conform to the total integrated circuit
technique of a modular electronics communication system. An example
of an X-band mixer circuit is shown in FIG. 6; it is a thin film
circuit using metallization on a ceramic substrate 50. The Schottky
barrier mixer diodes 47 and 48 are mounted as chips on either side
of an open one quarter wavelength stub 49 providing a short circuit
at the input signal frequency. A complete description of a
microwave integrated circuit mixer is given in the U.S. Pat. No.
3,416,042 issued to Philip R. Thomas, et al.
The fundamental operation of a mixer 43 is to convert a microwave
frequency of a lower frequency with a minimum of added noise. The
conversion for optimum operation should be with minimum loss.
Generally speaking, the received microwave signal and a signal from
a local oscillator are applied to a semiconductor junction from
which the difference in frequency or an IF output is extracted. To
optimize the noise level for the receiver system, both the signal
to noise ratio of the mixer and the conversion loss in the mixer
must be as low as possible.
The frequency multiplier 44 can be of a varactor type with a single
idler circuit. In the design of an integrated circuit 3X
multiplier, careful attention must be given to harmonic noise
problems arising from spurious emission from the multiplier. For
example, if the 3X multiplier produces a 4X component, the 4X
component would mix with the transmitted signal to produce noise at
the IF frequency. However, a 3X multiplier circuit using varactor
diodes should not directly contain a fourth harmonic if designed
with a single idler circuit at twice the input frequency.
Output signals at an IF frequency appear on channel 51 and are
applied from the mixer 43 to an IF filter and preamplifier 52. The
IF preamplifier is a three or four stage hybrid structure to
provide, for example, a 30-dB gain with a 6dB noise figure.
Filtering of the IF signal on channel 51 is required at the input
of the preamplifier to reduce amplifier loading due to the noise
outside the IF band. The principal noise source will be the
transmit signal which is converted by the mixer 43. Since this
frequency is about one and one-half octaves below the IF signal, a
multipole band pass microstrip filter should achieve at least 40-dB
of attenuation. A completely monolithic IF preamplifier circuit can
be constructed using thin film resistors, capacitors, and inductors
and epitaxial transistors on a high resistivity silicon
substrate.
FIG. 7 illustrates an integrated circuit 53 of a construction
suitable for the IF amplifier 52 of FIG. 5. The circuit 53 is
comprised of a substrate 54 of single crystal, high resistivity
silicon or other semi-insulating or high resistance semiconductor
material having first and second surfaces 56 and 57. The resistance
required between the surfaces 56 and 57 will vary with the
frequency at which the circuit is operated, the lower the frequency
the greater the resistance required. However, for high frequency
applications, high resistivity semiconductor material is adequate.
The components for the IF amplifier are formed at the surface of
the semiconductor substrate 54 using any conventional technique.
For example, in addition to epitaxial techniques, a transistor 58
may be formed in the surface by sequentially diffusing N-type,
P-type, and N-type regions into the surface 57 of the substrate
through openings etched in an oxide film 59. The circuit may also
include interconnecting strip conductors such as 61, 62, and 63
which may be placed directly on the high resistivity substrate 54
or on the oxide film 59. The conductors may also form inductors
such as indicated by the dotted outline at 64.
An insulating layer 66, such as glass, is deposited over and
inherently bonded to the portion of the second surface of the
substrate 54 which is exposed, and to the components of the
circuit. The insulating layer 66 is therefore integral with the
substrate. Metallized films 67 and 68 are inherently bonded to the
insulating layer 66 and to the first side 56 of the substrate. When
the metallized film 67 and 68 are connected to ground, as
represented by the conductors 71 and 72, the entire integrated
circuit is disposed between two closely spaced ground planes. The
ground planes and the circuit are interconnected so as to provide a
rugged, sealed package. For high frequency transmission lines, the
dielectric properties between the circuit components and each of
the ground planes may be made approximately equal for improved
performance.
A phase shift network 73 receives the amplified IF signal by way of
a channel 74 and delivers output signals of IF frequency by way of
channel 76 to the receive manifold 46. A beam steering or phase
control voltage is applied to a control network 77 from the
manifold 46 by way of channel 78 and produces a 3-bit control
signal to the phase shifter 73.
The control network 77 is a register which accepts a
one-out-of-eight phase step command through the manifold 46 and
stores it until another command is received. The register is
composed of three flip-flops and an input logic gate, all in low
power integrated circuit form, similar to Texas Instruments Series
54L Logic Modules.
The output signal of the central register of the network 77 drives
the phase shifter 73, which may be implemented as 3-series
shunt-loaded microstrip transmission line quarter wave sections.
Phase shifting is obtained by switching two PIN diodes, one at each
end of a quarter wave section transmission line. A PIN diode is
primarily capacitive under reverse bias and resistive under forward
bias and provides a low power loss phase shifting mechanism. The
three phase shift sections provide phase shifts in 45-degree steps
from 0 to 360 degrees. That is, a 45-degree section, a 90-degree
section, and a 180-degree section is provided in the phase shift
network 73.
Output signals at an IF frequency appear on channel 76 and are
applied through the manifold 46 for processing in a central
processor 79 by means of a channal 81. A distributive manifold for
a modular microwave system made up a large number of building block
modules, such as the one shown coupled to the manifold 46 and
including the radiating element 28, is described in the U.S. Pat.
No. 3,438,029 issued to Troy D. Fuchser et al. As described in the
referenced patent, the manifold 46 is a submanifold of a complete
distributive manifold system. In addition to the module associated
with the radiating element 28, as shown in FIG. 5, the manifold 46
would be coupled to three other radiating elements of the antenna
array 12 of FIG. 1. One quarter of the total number of said
submanifolds are coupled to a main manifold which in turn are
coupled to a four-way divider. In FIG. 5, the channel 81 is
intended to represent the main manifold and four-way divider system
coupling the radiating element 28 to the central processor 79.
For the transmit section of a modular electronics communication
system, an IF frequency signal is trnasmitted from the central
processor 79 through a transmit manifold 82 by means of a channel
83 to a phase shift network 84 by means of a channel 86. Again, the
channel 83 is intended to represent a complete manifold system for
dividing a transmit signal into a plurality of transmit signals to
each radiating element of the antenna array 11. A beam steering or
phase control code is applied to a control network 87 from the
manifold 82 by way of a channel 88 and supplies a 3-bit control
signal to the phase shift network 84. The phase shift network 84
and the control network 87 are similar in construction and
operation to the corresponding phase shift network 73 and control
network 77 of the received module as described previously.
An IF signal from the phase shift network 84 is coupled to an IF
amplifier 89 by means of a channel 91. The IF amplifier 89 is
similar to the preamplifier 52 described with reference to FIG. 7,
except that it will be designed to operate at higher power
levels.
The upward frequency conversion process performed on the output of
the IF amplifier 89 is basically the same as the downward shifting
done in the mixer 43, except that a lower side band up-converter is
employed for the actual translation process. A 4X multiplier 92
supplies the carrier frequency to an up-converter 93. The 4X
multiplier 92 is similar in construction to the 3X varactor type
multiplier 44 in the receive module.
A 4X multiplier circuit is shown in FIG. 8 and employs a varactor
diode 94 operating as a quadrupler with idlers at second and third
harmonics. More particularly a tuned circuit 96 and 97 may be
considered to be resonant at the second harmonic and the tuned
circuit 98 and 94 at the third harmonic. The varactor diode 94 and
the strip line transmission circuits forming inductance and
capacitance are formed on a semiconductor substrate. The substrate
101 has about one-half of its area covered by a highly conductive
surface layer 102. The layer 102 is then covered by a thin
dielectric layer 103 so that layer 102 serves as a common plate for
all but two condensors in the multiplier.
The input L section is formed by the strip transmission line 104
which extends over the thick dielectric portion of the substrate
101 to the plate 106 of the input capacitor. The capacitor 106
overlays the relatively thin dielectric layer 103 to form a
condensor with the common conductive layer 102. A loop 107 forms an
inductance over the thick dielectric layer and leads to a capacitor
plate 108 over the thin dielectric layer. Similarly, a loop 109
leads to the capacitor plate 111. A transmission line filter system
will thus be characterized by long thin transmission lines over a
thick dielectric section to provide primarily inductance
characteristics. Wide transmission line sections overlaying thin
high dielectric layers form zones in the transmission line system
primarily capacitive in nature. A loop 112 extends from plate 111
over the thick dielectric to the juncture with a loop 96 which
leads to a capacitor plate 97. Loop 112 also leads to one terminal
of the varactor diode 94. A strip extending from the juncture 98
and loop 116 then leads to a capacitor plate 117. The capacitor
plate 117 is positioned on top of a conductive layer 118 which
overlays one-half of a condensor plate 119. Condensor plate 124
similarly overlays the plate 119. The transmission line loop 121
then extends to the output capacitance plate 122 with the matching
conductance 123 extending from the plate 122. The plate 119 is
capacitively coupled to the capacitor plates 117 and 124 and to the
high conductive layer 102. The process of forming an integrated
circuit multiplier is thoroughly described in the U.S. Pat. No.
3,386,092 of Tom M. Hyltin.
The 4X multiplier 92 quadruples the 2.235 GHz output of a local
oscillator as amplified by a LO amplifier 126 coupled to the
transmit manifold 82 by means of a channel 127. The up-converter 93
is a varactor type similar in some respects to the varactor
multiplier circuits 44 and 92. It is of the lower side band type
rather than the more conventional upper side band type to reduce
harmonic noise production in the receiver IF band pass and reduce
the factor by which the local oscillator must be multiplied prior
to the conversion process. The converted IF signal from the
amplifier 89 is then transmitted as an RF frequency signal from the
radiating element 21 coupled to the up-converter 93 by means of an
unbalanced strip line 128.
By way of example, the operation of the system of FIG. 5 is such
that the 8.325 GHz .+-. 50 MHz circularly polarized wave applied to
the antenna element 28 will be converted to a 1.620 GHz .+-. 50 MHz
signal in the mixer 43. The frequency conversion function by the 3X
multiplier 44 changes the 2.235 GHz local oscillator signal into a
6.705 GHz signal coupled to the mixer 43. The input power level to
the multiplier 44 is 4 mW, to provide the required 2 mW input to
the mixer 43. The IF signal of 1.620 GHz .+-. 50 MHz is amplified
in the IF filter preamplifier 52 having three or more stages of
transistor amplification to generate a gain of 30 dB.
Again, by way of example, the transmit section emits a circularly
polarized 7.320 GHz .+-. 50 MHz signal from the radiating element
21 from the up-converter 93. The up-converter 93 receives a 1.620
GHz .+-. 50 MHz, 15 mW signal from the IF amplifier 89 and an 8.940
GHz, 100 mW signal from the 4X multiplier 92. The LO amplifier 126
has a gain of 20 dB at 2.235 GHz, with an output power of 250 mW,
while the IF amplifier 89 has a gain of 20 dB. Thus, the power
generation chain of the transmit section consists of one stage of
amplification at 2.235 GHz followed by a 4X varactor multiplier
with an output of 8.940 GHz.
As explained previously, phased array antennas such as 11 and 12
are formed by radiating elements arranged in any desired geometry,
for example, triangular spacing. They may be planar or non-planar
(conformal). In order to control the phase of the antenna transmit
or receive beam, phase control codes are provided for each of the
radiating elements in the antenna array. These codes are generated
by the beam steering computer 27 such as shown in FIG. 2. Thus, the
beam pointing and control in a phased array is accomplished by the
phase setting of a phase shift network in series with each
radiating element such as network 73 and 84 of FIG. 5. A beam
scanning computer calculates the necessary phase for each element
in order to collimate the beam at some position in space. The scan
computer can be a digital device and the variable phase shift
networks either a digital device or a digitally controlled analog
device. Because of their digital nature, phase shift networks may
be classified as follows:
Phase Shifter Minimum Discrete Phase Shift 1-bit 180.degree. 2-bit
90.degree. 3-bit 45.degree. 4-bit 22.5.degree. 5-bit
11.25.degree.
Thus, the larger number of "bits" available to control a phase
shifter, the more accurately the array beam may be positioned. It
is believed that a 3-bit phase shift network will provide
sufficient accuracy for beam steering for the system of this
invention.
Referring to FIG. 9, there is illustrated the relationship between
beam pointing angle and the required phase shift network setting
which is given by:
.PSI.n = -(d/.lambda.) (2.pi.) sin .theta.
where
.PSI.n = phase shift setting (radians)
d = spacing between radiators
.lambda. = wavelength (same units as d)
.theta. = scan angle
For the radiating elements shown in the lower portion of FIG. 9,
each separated a distance d, the phase shifter setting for a scan
angle of .theta. is as follows:
.PSI..sub.0 = 0
.PSI..sub.1 = - (d/.lambda.) (2.pi.) sin .theta.
.PSI..sub.2 = 2 .PSI..sub.1
.PSI..sub.3 = 3 .PSI..sub.1
.PSI..sub.n = n .psi..sub.1
Modern electronically scanned arrays using digital means for
controlling the phase of each antenna element frequently use binary
devices, such as diodes, for the actual phase control. The two
state characteristic of these devices naturally leads to binary
phase quantization. In FIG. 9, the actual quantized phase front for
each of the seven elements shown is given by the stair-step curve
131. For a control network such as 77 and 87 of FIG. 5, the minimum
step size, Q, for the curve 131 is given by:
Q =2.pi./2.sup.n radians (2)
where n = the number of control bits. Increasing the number of
control bits reduces the quantization error at the expense of
increased complexity, insertion loss, driving power, cost, and
weight of the overall system. Therefore, there is usually some
tradeoff between the accuracy with which the array beam may be
positioned and the quantization error.
Radar phased array antenna beam steering, which is similar in many
respects to the beam steering required in the system of this
invention, is adequately described in the literature. For example,
the work of Merrill I. Skolink entitled "Introduction to Radar
Systems," McGraw-Hill, contains a section on phased array antennas
and beam steering. A phased array antenna scan control system is
also described in the U.S. Pat. No. 3,345,631 issued to Leo A.
Chamberlain, Jr.
Referring to FIG. 10, there is shown a duplexed operation employing
a single antenna 132 for the received and transmitted signals which
are separated by a circulator 133, sometimes known as a diplexer.
The circulator 133 is s three-port device that has the property
that a wave incident in port one is coupled into port two only, a
wave incident in port two is coupled into port three only, and so
on. Ideal circulators are matched devices, that is, all ports
except one terminate in matched loads, the input impedance of the
remaining port is equal to the characteristic impedance of its
input line, and hence presents a matched load. H. J. Carlin in an
article entitled "Principles of Gyrator Networks," Polytech
Institute Brooklyn, Vol. 4, 1955, describes how a lossless,
matched, nonreciprocal three-port microwave junction as an ideal
three port circulator. A practical realization of a three-port
circulator involves a symmetrical Y-junction of three identical
"strip-line" type transmission lines with an axial magnetized
ferrite rod or disk at the center of the Y-junction. Thus, if an
8.3 GHz signal transmitted to port one of the circulator 133 is
incident in one leg of the Y-junction, it is coupled to the antenna
element 132 through only the second leg of the junction. The 7.3
GHz signal received by the antenna element 132 will be incident in
the second leg of the Y-junction and coupled to only the third leg.
Typical characteristics that can be obtained from a circulator are
insertion loss of less than 1 dB, and isolation between the
transmit and receive signal of from 30 to 40 dB.
Additional separation between the 8.3 GHz, 500 mW transmit signal
output of an up-converter 134 from the received signal is performed
by a notch filter 136 coupled to the circulator 133 by means of a
channel 137. The notch filter 136 is of a microstrip configuration
providing on the order of 80 to 130 dB isolation between the
transmit signal and the 7.3 GHz receive signal. A conversion of the
7.3 GHz signal from the filter 136 to a 1.6 GHz signal of IF
frequency is performed in a mixer 138 coupled to receive a 5.7 GHz,
2 mW signal from a 3X multiplier 139. The 3X multiplier 139
receives a 1.9 GHz local oscillator signal on channel 141 through a
manifold 142.
Additional filtering of the IF signal from the mixer 138 is
performed in a notch filter 143 coupled to an IF filter and
preamplifier 144 by means of a channel 146. A phase shift network
147 receiving control signals over a channel 148 couples the 1.6
GHz signal to a central processor (not shown) through the manifold
142.
The transmit channel of the system of FIG. 10 is similar to the
transmit channel of the system of FIG. 5. In addition to the
up-converter 134, it includes a local oscillator amplifier 149 for
power amplification of a 2.235 GHz local oscillator signal and a 3X
multiplier 151 for changing the local oscillator signal into a 6.7
GHz signal connected to the up-converter 134. A phase shift network
152 applies the appropriate phasing information to a 1.6 GHz signal
on channel 153 in accordance with control information on channel
154. An IF amplifier 156 couples the transmit signal from the phase
shifter network 152 to the up-converter 134 wherein a conversion
process takes place to the 8.3 GHz transmit signal.
Again, a module containing the components shown in FIG. 10 is
required for each radiating element of the antenna array 132. The
operation and construction of the various components of FIG. 10
have been described previously with respect to FIG. 5, except for
the notch filters 136 and 143, and the diplexer 133.
Referring to FIG. 11, there is shown another embodiment of a
duplexed operation employing a single antenna 160 for the received
and transmitted signals. To obtain acceptable signal separation,
different transmit and receive frequencies are employed as in the
previously described systems. In addition, about 10 dB signal
separation is obtained by circularly polarizing the transmit
received signal and an attenuated transmit signal. This output is
one sense and circularly polarizing the receive signal with the
opposite sense. For additional signal separation, a filter 162 is
designed to pass the frequency of the received signal and attenuate
the frequency of a transmitted signal. For example, the filter 162
may be designed to attenuate a transmitted frequency between 7.725
to 7.850 GHz by approximately 40 dB and pass a 7.125 GHz signal
received by the antenna 160 with a loss of about 1dB. Thus, the
output of the filter 162 will consist of both the output is
processed through a receive channel similar to that described with
respect to FIG. 10.
The output of the filter 162 is converted into an IF frequency by a
mixer 164 which also receives a signal from a multiplier 166
connected to a phase shift network 168. The phase shift network 168
is similar to that described previously. In the embodiment of FIG.
11, the phase of the received signal is controlled substantially
simultaneously with the conversion of the RF signal received at the
antenna 160 into an IF processing signal by varying the phase of
the local oscillator signal applied to the mixer 164. Phase shift
instructions from a central processor (not shown) are connected to
a 4-bit logic network 170 which in turn connects to the phase shift
network 168. One third of the phase shift desired for the signal
received at the antenna 160 is applied to the output of
preamplifier 172. The subsequent multiplication of this signal in
the 3X multiplier 166 produces a mixer input signal with the
desired phase shift. The input to the preamplifier 172 is the
output of a local oscillator (not shown). It should be noted that
if preamplifier 172 is capable of providing the desired local
oscillator frequency via phase shifter 168 to mixer 164, the 3X
multiplier 166 would be unnecessary and the actual desired amount
of phase shift for the signal received at the antenna 160 would
have to be provided by phase shifter 168.
Typically, the local oscillator may have an output of approximately
2.208 GHz at 2mw. This signal is amplified in the amplifier 172 to
a power level of 20 mw. As a result of passing through the phase
shift network 168, the power level at the input of the multiplier
166 is 10 mw. In the multiplier 166, the 2.208 GHz oscillator
output is increased in frequency to 6.625 GHz at a power level of
4mw. Considering a received signal of 7.125 GHz, the output of the
mixer 164 will be an IF signal at 500 MHz .+-. 62.5 GHz plus some
level of unwanted transmitter component at 1162.5.+-.62.5 MHz.
Additional filtering of the IF signal from the mixer 164 is carried
out in an IF filter 174 coupled to an IF amplifier 176, reducing
the unwanted transmitter component another 50 dB. Output signals at
an IF frequency from the amplifier 176 are applied through a
manifold (not shown) for processing in a central processor of the
type described previously.
The transmit channel of the system of FIG. 11 includes a
preamplifier 178 coupled to the output of the signal processor. A
phase shift network 180 applies the appropriate phasing information
to the output of the amplifier 178 in accordance with control
information coupled to a 4-bit logic network 182. Instructions to
the 4-bit logic network 182 are received from the central
processor. The output of the preamplifier 178, with the appropriate
phase shift information applied thereto, is connected to the input
of a power amplifier 184 which in turn connects to a multiplier
186. The multiplier 186 performs the same function as the
up-converter described previously. It converts the low frequency
output of the central processor into a high frequency transmit
signal.
Considering that the transmitted frequency is in the range of from
7.725 to 7.850 GHz, then with a 3X multiplier, the output of the
central processor is a signal at a frequency in the range from
2.575 to 2.617 GHz. Typical power levels for a system of the type
illustrated are 2mw to the amplifier 178 and 20 mw to the phase
shift network 180. The power level of the signal transmitted from
the phase shift network 180 to the power amplifier 184 is on the
order of 10 mw with the output of the power amplifier at a 250 mw
level. It should be understood, that the values discussed with
regard to FIG. 11 are given only by way of example.
In addition to the use of modular electronics technology to
satellite communcation systems, there are several other areas for
its application; the most appealing of which is the implementation
of skin-mounted conformal arrays for aircraft. A modular electronic
communication system, capable of high antenna gains and coupled to
a beam steering compunter, allows relative small tactical aircraft
to communicate freely through a satellite system. Presently, only
large, equipment filled aircraft are capable of communicating
through such a satellite system. Possibilities exist in this
context for the implementation of a combined navigation and
communication system employing exclusively satellite terminals. The
modular electronics array also is compatible with implementation as
an electronically adaptive system or a retrodirective system which
permits automatic tracking of satellites from aircraft
terminals.
The frequency flexibility of a modular electronics communication
system with its high speed, inertialess beam steering also lends
itself for adaptation to other applications. For example,
spacecraft systems with a high gain, steerable antenna are
envisioned as becoming important as data rates and communication
distances increase. These same characterisitcs, combined with the
reliability and relative ruggedness of an integrated circuit
system, make it equally suitable for portable mobile or shipboard
applications.
Another area contemplated as within the scope of the present
invention includes a system for generating several radiation
patterns from one antenna array simultaneously. This, for example,
permits the simultaneous, narrow-beam tracking of several
satellites by one earth based array. A multibeam function may be
achieved in several ways. The simplest of which is to use
designated sections of the antenna array for each beam. Another
approach would be to use multiple phase shifters and manifolds in
the transmit section in a manner similar to the receive section of
FIG. 3. With this approach, the entire antenna array is used for
each transmit beam. The phase shift networks for each group of
radiating elements be independently controlled by the beam steering
computer.
While several embodiments of the invention, together with
modifications thereof, have been described in detail herein and
shown in the accompanying drawings, it will be evident that various
further modifications are possible in the arrangement and
construction of its components without departing from the scope of
the invention.
* * * * *