U.S. patent number 3,678,414 [Application Number 05/081,943] was granted by the patent office on 1972-07-18 for microstrip diode high isolation switch.
This patent grant is currently assigned to Collins Radio Company. Invention is credited to Ben R. Hallford.
United States Patent |
3,678,414 |
Hallford |
July 18, 1972 |
**Please see images for:
( Certificate of Correction ) ** |
MICROSTRIP DIODE HIGH ISOLATION SWITCH
Abstract
A shield enclosed microstrip circuit microwave multi-diode
switch with undesired waveguide signal transmission minimized.
Diodes are spaced and connective microstrip sections dimensioned
for optimum signal isolation in the OFF state. Further, diode dc
bias isolation is provided for multi-diode sections of such
switches by signal path coupling capacitors, along with dc bias
supply RF rejection filtering minimizing loss of signal power and
that minimizes disturbance of signal line impedance over a
relatively broad band range of operation.
Inventors: |
Hallford; Ben R. (Dallas,
TX) |
Assignee: |
Collins Radio Company (Dallas,
TX)
|
Family
ID: |
22167396 |
Appl.
No.: |
05/081,943 |
Filed: |
October 19, 1970 |
Current U.S.
Class: |
333/104; 333/103;
333/238; 333/128 |
Current CPC
Class: |
H01P
1/15 (20130101) |
Current International
Class: |
H01P
1/10 (20060101); H01P 1/15 (20060101); H01p
005/12 () |
Field of
Search: |
;333/7,8,84M,84 |
References Cited
[Referenced By]
U.S. Patent Documents
Primary Examiner: Saalbach; Herman Karl
Assistant Examiner: Chatmon, Jr.; Saxfield
Claims
I claim:
1. In a shield enclosed microstrip diode high isolation switch, a
circuit board with a rigid electrically conductive ground plane
plate mounting a dielectric material layer and microstrip circuitry
in laminate relation thereon; electrically conductive shield means
substantially completely enclosing the microstrip circuitry above
said ground plane plate; signal input through wall electric circuit
connective means for interconnecting a signal source and said
microstrip circuitry; signal output through wall electric circuit
connective means for interconnecting signal output utilizing means
and said microstrip circuitry; a microstrip section with signal
coupling dc blocking means at opposite ends in the microstrip
circuitry between said input and said signal output connective
means; diode means in the form of a plurality of PIN diodes and
assembly structures connected between said microstrip section and
said ground plane plate with common diode electrodes connected to
said microstrip section and opposite electrodes connected to said
ground plane plate; each diode assembly structure including a
highly electrically and thermally conductive mounting metal base
seated at locations in the switch structure in close metal to metal
engagement in said ground plane plate, said diodes spaced at
substantially one-quarter wavelength of the switch operational
designed center frequency apart along said microstrip section and
controlled switch voltage bias circuit means connected to said
microstrip section for controlled biasing said diode means to
conduction for a switched OFF state of said switch between said
input and said output connective means from a switched ON state
diode means not biased to conduction.
2. The shield enclosed microstrip diode high isolation switch of
claim 1, with a plurality of said microstrip sections in
microcircuitry interconnecting a plurality of said signal input
through wall connective means and said signal output through wall
electric circuit connective means.
3. The shield enclosed microstrip diode high isolation switch of
claim 2, wherein a plurality of said controlled switch voltage bias
circuit means is provided one for each of said plurality of
microstrip sections.
4. The shield enclosed microstrip diode high isolation switch of
claim 3, wherein transformer means interconnects the plurality of
said microstrip sections and said signal output through wall
electric circuit connective means.
5. The shield enclosed microstrip diode high isolation switch of
claim 3, wherein said transformer is a microstrip transformer; and
RF rejection filter means is included in each of the plurality of
said controlled switch voltage bias circuit means.
6. The shield enclosed microstrip diode high isolation switch of
claim 5, wherein two of said signal input through wall connective
means are located at opposite signal input ends of said
electrically conductive shield means in the form of a rectangular
elongate microwave microstrip switch box; and said signal output
through wall electric circuit connective means is in the form of a
signal output connection located substantially midway of said
switch box opposite signal input ends.
7. The shield enclosed microstrip diode high isolation switch of
claim 7, wherein diodes of a first microstrip section are connected
cathodes to the first microstrip section and anodes to said ground
plane plate; and diodes of a second microstrip section are
connected anodes to the second microstrip section and cathodes to
said ground plane plate.
8. The shield enclosed microstrip diode high isolation switch of
claim 6, wherein said microstrip transformer is a T-shaped
transformer with T-opposite side legs approaching one-fourth
wavelength of the switch operational designed center frequency
length, and having a common T-shank substantially one-fourth
wavelength of the switch operational designed center frequency long
that is output end connected to said signal output through wall
electric circuit connective means.
9. The shield enclosed microstrip diode high isolation switch of
claim 8, wherein said signal coupling dc blocking means are each
capacitors; a gap is provided in each of the transformer T-opposite
side legs; and a capacitor signal couple spans each of the
transformer T-opposite side leg gaps.
10. The shield enclosed microstrip diode high isolation switch of
claim 9, wherein the apparent signal electrical length of said
transformer T-opposite side legs are substantially one-fourth
wavelength of the switch operational designed center frequence long
with a gap spanning capacitor in place on each transformer side
leg.
11. The shield enclosed microstrip diode high isolation switch of
claim 10, wherein said dc blocking signal coupling capacitors are
relatively small discreet multiplate capacitors mounted with plates
substantially parallel to the longitudinal direction of the
microstrip circuit and across the gaps and with the plates
substantially vertically perpendicular to the plane of the
microstrip circuitry in the area of gap ends.
12. The shield enclosed microstrip diode high isolation switch of
claim 6, wherein electrically conductive material baffle plate
means is provided within said rectangular elongate microwave
microstrip switch box extended substantially through the
longitudinal length of the box from end to end and generally
spanning the interior of the shield box from said ground plane
plate to intimate close engagement with the inside of a shield box
lid.
13. The shield enclosed microstrip diode high isolation switch of
claim 12, wherein the plurality of said microstrip sections are
located to one side of said baffle plate means; and said RF
rejection filter means are located to the other side of said baffle
plate means from the side of said microstrip sections.
14. The shield enclosed microstrip diode high isolation switch of
claim 13, wherein each of the plurality of said controlled switch
voltage bias circuit means includes a through wall bias connector;
and a substantially one-fourth wavelength of the switch operational
designed center frequency long relatively high impedance line
interconnecting each RF rejection filter and the respective
microstrip section.
15. The shield enclosed microstrip diode high isolation switch of
claim 14, wherein said baffle plate means is in the form of two
baffle plates extending from opposite side switch box ends to
relatively closely spaced inner ends approaching the transformer
T-shank of the microstrip circuitry within the box.
16. The shield enclosed microstrip diode high isolation switch of
claim 8, wherein length and impedance parameters of transformer
T-opposite side legs are design selected to optimize the attainment
of microstrip circuit matched characteristic impedances.
17. The shield enclosed microstrip diode high isolation switch of
claim 16, wherein length and impedance parameters of the common
T-shank of said microstrip transformer are designs determined to
optimize the attainment of microstrip circuit matched
characteristic impedances.
Description
This invention relates in general to RF switches, and in
particular, to microstrip multi-diode high isolation microwave
signal path switches.
Heretofore, high isolation switch systems in microwave signal paths
have been waveguide switching devices as the only such microwave
frequency signal switch giving as much or more than 90 db
isolation. Furthermore, such microwave waveguide switches have been
very narrow-band in the attainment of desired isolation levels and
are rather expensive. Additional factors such as VSWR and insertion
loss with such approaches are higher than desired, although lived
with for some time since better solutions did not appear to be
available. A stripline approach has been tried but, with this
approach, a prototype 2 GHz 90 db isolation switch, the VSWR and
insertion loss were prohibitively high. Furthermore, with
preexisting systems, a junction mismatch has been a significant
problem where two diode switches are joined to a common line where
a mismatch would exist at the common tie point if no corrections
are made such as would be the case where two transmitters are
operating, one switched to deliver to a common single antenna and
the other switched off in the hot standby state. The mismatch under
these operational conditions exists since the presence of the "OFF"
switch transmission line alters the impedance of the "ON" switch
transmission line section. An approach to this problem in
minimizing such mismatch problems and disturbances arising
therefrom is to locate a short (or low impedance) of the "OFF"
switch transmission line substantially one quarter wavelength from
where it joins the transmission line of the "ON" side. This quarter
wavelength long line makes the "OFF " switch short appear as,
effectively, an open circuit where it joins the "ON" side
transmission line.
There is a problem with matching of a switched "ON" transmission
line connected to a common junction with respect to another
transmission line connected to the same junction in the switched
"OFF" state with respect to quarter wavelengths being a determinate
factor since such a length exists only at one frequency with
special matching techniques called for when a rather broad band
range of frequencies is to be utilized. Such an idealistic matching
technique has been achieved through an optimization routine in a
computer program actually trying different line lengths and
impedances at such a common junction and calculating the total
switch VSWR. The computer program is such that differences in the
VSWR between changes are compared with the next calculation being
made in a direction tending to improve VSWR. Such a programming
approach has been employed utilizing good equivalent circuits for
PIN diodes, or the functional equivalent thereof, with the only
component elements ignored being connectors and capacitors since
their impedance versus frequency relationship is not known. With
this approach, the matching stub on a final optimized model is
adjusted to compensate for the contribution of these two elements
to the overall switch VSWR inherently with the optimization
programming approach.
It is, therefore, a principle object of this invention to provide a
microstrip microwave multi-diode switch capable of relatively fast
switching speeds, relatively high RF power handling capabilities up
to, for example, 10 watts CW, with a low VSWR factor, low insertion
loss, and high isolation.
A further object is to provide such a microwave multi-diode switch
in the form of a shield enclosed microstrip circuit single pole,
double throw diode switch.
Another object is to provide such a microstrip multi-diode switch
that lends itself to economical fabrication techniques resulting in
high reliability and consistently predictable performance
characteristics along with performance reliability with minimal
service maintenance requirements quite adequately satisfying
commercial market requirements.
Still another object with such a microstrip multi-diode switch is
to provide selective low loss connection to one transmitter while
providing a high isolation to the signal of another transmitter
connected to the switch circuit.
A further object with such a microstrip multi-diode switch is the
minimizing of microwave RF current leakage between input and output
connectors and to prevent any significant signal coupling between
an RF rejection filter connected to the transmitter ON section of
the switch and a transmitter connector of an OFF portion of the
switch.
Still another object is the obtainment with such a microstrip
multidiode switch of a relatively wide frequency operational
range.
Features of the invention useful in accomplishing the above objects
include in a shield enclosed microstrip circuit single pole double
throw diode switch, a plurality of diodes in each of two
transmitters legs of a switch with each leg of the switch as a
switch activated or deactivated transmission line interconnected by
a microstrip transformer section to a single antenna connective leg
thereof. The diodes employed are PIN semiconductor diodes having a
P layer, an intrinsic or I layer, and an N layer. These PIN diodes
are mounted in shunt with the line and with no bias current
appearing in the diodes, they are in a high resistance state.
However, as bias current appears and is increased, the shunt
resistance rapidly decreases to a very low value. With no bias
applied to the PIN diodes of a switch leg, each of the diodes
appears as a 50 ohm impedance with a low loss and represent thereby
a good impedance match to a 50 ohm circuit line with application of
such voltage bias to the diodes of a switch leg resulting in, for
example, a 100 milliamperes bias current flow through each diode of
that leg. The impedance therethrough becomes very low, in the
neighborhood of one ohm thereby resulting in high insertion loss to
RF of the transmitter connected to that leg of the diode switch to
thereby accomplish the switched-off state for the RF transmitter of
that switch leg. DC blocking capacitors are positioned in series
with the transmission line circuit path at each end of the diode
section of each switch leg to prevent other than desired bias
current from flowing into the respective diode switch legs of the
switch. Voltage bias connection is provided to each of the
respective switch legs from different level voltage supplies via a
bias voltage control switch and a connection to and through an
individual respective RF rejection filter to each of the diode
switch sections of the respective legs between the dc blocking
capacitors of the respective switch legs. End-line fed type of coax
connectors are provided through walls of the shielded box
containing the microstrip diode switch for connection to the three
pairs of terminals of the switch with connectors being matched to
50 ohm impedance in accord with design practices well known to
those skilled in the art. The switch is formed with a common
T-shaped transformer section. The attainment of desired operational
results through optimized impedance matching entails spacing
between diodes at an optimum as well as spacing through transformer
legs in accord with the desired frequency range objectives for the
switch and with the spacing distance being approximately a quarter
wavelength between diodes at the center operating frequency. This
is with the exact spacing distance being determined empirically
through measuring the frequency where the isolation is the highest
when the diodes are mounted a quarter wavelength apart, at the
center operating frequency, and then correcting the distance
thereof by scaling the measured peak isolation frequency to the
desired center operating frequency. The two diode switch legs are
joining to the common T-transformer output leg while maintaining a
low insertion loss and a good match to the ON side with impedances
and lengths of the T-shaped transformer being adjusted for least
VSWR over the operating frequency range; for example, in a specific
switch 1,700 to 2,300 MH.sub.z. Baffle plates are provided within
the shield box of the switch that prevent undesired leakage of RF
currents between the input and output connectors and to suppress
undesired coupling of RF rejection filters from the ON side to the
transmitter connector on the OFF side thereby insuring maximum
isolation possible.
High power handling capabilities of the switch are in large measure
the result of large breakdown voltage characteristics of the PIN
diodes used along with good heat sink action attained between the
gold plated copper bars of the diode semiconductor assemblies that
are mounted in intimate contact with the relatively thick aluminum
plate that forms the backing plate of the polyolefin microstrip
board employed for the switch. This is with a slot milled through
the polyolefin substrate and directly into the aluminum of the
ground plane plate for each individual diode in providing the
setforth good metal to metal contact for each of the diodes. The dc
blocking capacitors employed in the switch are chip style resting
above gaps in the microstrip conductors of each switch leg with
each capacitor being, for example, a 100 picofarad capacitor as
required for a 2 GH.sub.z center frequency switch for the series
impedance to be negligibly small. The dc blocking capacitors have
many plates, separated by thin dielectric with a high dielectric
constant, and the capacitors are mounted with the plates
perpendicular to the microstrip's conductor. The chip capacitors
electrically lengthen the transmission line paths at 2 GH.sub.z
approximately 0.040 inches per capacitor with this length being a
significant factor subtracted from the length of T-shaped
transformer legs in satisfying functional operational design
requirements. The operational control switching waveform to the
diodes is a negative voltage that supplies the necessary bias when
turning a switch leg off. A positive back voltage is applied to
quickly remove the stored charges in the individual PIN diode
junctions of a switch leg when switching to the ON state. Further,
this positive voltage also reduces the insertion loss factor in the
diodes by depleting the diode junctions.
A specific embodiment representing what is presently regarded as
the best mode of carrying out the invention is illustrated in the
accompanying drawings.
In the drawings:
FIG. 1 represents a top plan view looking down at the top of a
microstrip diode high isolation switch contained in a shield box
with the lid removed showing microstrip isolation switch circuitry
for a two transmission line alternate switch to common output
switch circuit;
FIG. 2, a planar view of the shield box lid;
FIG. 3, a cut away and sectioned view taken along line 3--3 of FIG.
1 of the microstrip diode high isolation switch in its box shielded
environment;
FIG. 4, a circuit schematic showing of the microstrip diode high
isolation switch system of FIGS. 1 and 3;
FIGS. 4A and 4B, transmitter "ON" and "OFF" diode equivalent
circuit operational states;
FIG. 5, a plan view of microstrip art work circuit board results
prior to further machining and the mounting of circuit components
thereon for the two transmission line microstrip diode high
isolation switch of FIGS. 1, 3, and 4;
FIG. 6, a partial T-junction detail view showing electrical
reference measurement detail supplemental to the showing of FIG.
5;
FIGS. 7 and 8, side and top views respectively illustrating dc
diode switch section bias isolating microwave signal coupling
capacitors;
FIG. 9, a VSWR to frequency in GH.sub.z performance characteristic
curve for the same embodiment;
FIG. 10, an isolation in db to frequency in GH.sub.z performance
characteristic curve for the same embodiment;
FIG. 11, an insertion loss in db to frequency in GH.sub.z
performance characteristic curve for a microstrip diode high
isolation switch system constructed in accord with the embodiment
of FIGS. 1, 3, 4, and 5; and
FIG. 12, a two transmission line embodiment of a microstrip diode
high isolation switch with the switching diodes of one switch
section inverted from those of the other switch section, and with
the diode biasing switch control system different from the diode
switch bias control shown with the FIGS. 1, 3, 4, and 5
embodiment.
Referring to the drawings:
The diode high isolation switch 20 of FIG. 1 is shown to be in
microstrip circuit form on a microstrip circuit board 21 contained
in a shield conductive metal box 22 with the lid 23, shown in FIG.
2, removed that is normally fixed in place on box ledge 24 by
screws extended through lid openings 25 and threaded into shield
box threaded openings 26. The shield box 22 is provided with
through wall inserted bias voltage coax connectors 27a and 27b, one
of which appears in cross section in FIG. 3, along with the coax
connector 28 for connection to an antenna 29, schematically
indicated in FIG. 4. The shield box 22 is also provided with
through wall coax connectors 29a and 29 b for connection to RF
transmitters 20a and 20 b for selective switch connection via the
diode switch 20 for signal connecting one or the other RF
transmitter 20a or 20b to feed the antenna 29 as diode switch
control selected.
The microstrip board 21 is constructed of a relatively thick rigid
ground plane plate 30 with a relatively uniformly thick plastic
dielectric laminate sheet 31 bonded to the upper surface thereof,
and with RF conductive microstrip circuitry 32 formed on the upper
side of the plastic dielectric laminate sheet 31, such as having
been formed through patterned etching of a copper cladded side of
the plastic dielectric laminate sheet 31. The microstrip art work
of a microstrip board 21 is shown in FIG. 5 whereon microstrip
circuit section elements are produced including a center microstrip
T-shaped transformer 33 having a transmitter output connective leg
34 and top opposite side RF transmitter switch connective arms 35a
and 35b, each with a nonconductive gap spacing 36a and 36b that are
bridged, respectively, by signal coupling chip capacitors 37a and
37b. The microstrip board 21 is also equipped, successively
outwardly from the respective transformer arms 35a and 35b, with
microstrip transmission line switch sections 38a, 39a, 40a, and
41a, and 38a, 39b, 40b, and 41b, respectively. This is with
transmission line sections 38a and 38b aligned with and spaced from
transformer arms 35a and 35b, respectively, for insertion therein
of individual diode assemblies 42. Furthermore, additional diode
assemblies 42 are inserted in the gaps between the transmission
line switch sections 38 a and 39a, 39a and 40a, 38b and 39b, and
39b and 40b. Diode assemblies 42 are PIN type diodes having a
through conductive element bridging the respective gaps between
adjacent transmission line elements and provide a connection
through the diode structures in shunt with the respective
transmission lines to ground at each diode location. This is with,
as shown in the schematic of FIG. 4, the cathodes connected to the
respective transmission lines and anodes connected to ground, that
is, the ground plane plate 30 of the microstrip board 21. The PIN
diodes 42 are each in the form of a semiconductor having a P layer,
an intrinsic or I layer, and an N layer. As has been pointed out,
these diodes are mounted in shunt with the respective transmission
lines and each appear as a high resistance when there is no bias
current through the respective diodes 42 and, as bias current
through the diodes is developed, there is a rapid decrease in
resistance therethrough to a very low value. Relatively high power
handling capabilities of the switch are provided as a result of the
relatively large breakdown voltage characteristics of the PIN
diodes 42 coupled with good heat sink action between gold plated
copper bars 43 of the diode assemblies 42 for cooling, particularly
with the bars 43 being tightly mounted as by screws 44 to ground
plane plate 30. This is with 43 receiving slots 45 milled through
the dielectric layer 31, in switches that have been built being in
the form of a laminate layer of polyolefin, and into the aluminum
of the ground plane plate 30 to provide good metal to metal
contact. This is with the grooves in the ground plane plate being
of such depth that the through ribbon conductive leads of the
diodes 42 conveniently rest on the adjacent microstrip transmission
line conductor ends thereby achieving well matched impedance
connections from the respective microstrip transmission line
conductors to the diodes 42.
The nonconductive gap spacings 46a and 46b between microstrip
transmission line switch sections 40a and 41a and 40b and 41b are
bridged by signal coupling chip capacitors 47a and 47b,
respectively, that are substantially the same as signal coupling
chip capacitors 37a and 37b. These dc blocking capacitors are
placed in series with the respective circuit transmission lines to
dc isolate the diode switch portions of the RF transmitter switch
connective transmission lines signal coupled to the microstrip
transformer 33 (shown with additional detail in FIG. 6) to prevent
bias current in the respective switch sections from flowing into
the circuits joined to the switch 20. The particular capacitors
37a, 37b, 47a, and 47b employed, as shown in FIGS. 7 and 8, are
ceramic chip style having metalized ends 48 and 49 for connection
to strip conductors by soldering or brazing 50 or through the use
of other highly conductive fastening media such as silver filled
epoxy with the capacitors relatively small cubes of approximately
50 mils cube. The four dc blocking capacitors so employed are chip
style components resting above respective gaps in the microstrip
circuit conductors providing approximately 100 picofarads at a
center operational frequency of 2 GH.sub.z, with a switch that has
been built, such as to cause the series impedance to be negligibly
small. These capacitors have many small plates which are narrowly
separated by relatively very thin dielectric layers having a high
dielectric constant and the capacitors are mounted with the plates
perpendicular to the microstrip conductor since that it has been
found that when the capacitors are mounted with the plates parallel
to the RF conductors that an absorption type of resonance occurs
causing a sharp rise in insertion loss. It appears that the
multipath propagation through the high dielectric constant
capacitor dielectric causes sufficient phase change between the
energy along the lower and upper plates when the coupling
capacitors are mounted with the plates parallel to the microstrip
conductors to form a resonant circuit. It appears that such
multipath condition exists since the alternate plates are joined
together at the two ends of the capacitor of each capacitor
structure with the electrical distance through each plate
combination of the capacitor structure being different when so
mounted. It should be noted further that while relatively small,
these chip capacitors do electrically lengthen the microstrip
conductor with such additional length at, for example, 2 GH.sub.z
being effectively measured to be approximately 0.040 inches per
capacitor chip. Such length factors must be subtracted from the
length of the T-shaped transformer 33 where the capacitors are
mounted to satisfy the conditions called for in optimized switch
design. It should be noted further, at this point, that actual
location of the chip capacitors along the respective transmission
lines varies the effective electrical length through the respective
capacitor chips, that is, those capacitor chips within the
microstrip T-transformer.
The two transmitter fed diode switch 20 is joined to the common
output connector 28 for feeding a single antenna 29 via leg 34 of
the T-shaped transformer 33 while maintaining a low insertion loss
and good match to the ON side of the diode switch. The complete
switch 20 is analyzed with dimensions optimized in the T-shaped
transformer 33 to attain impedances and lengths for least VSWR over
the operating frequency range, in a particular instance for a
particular switch 1,700 to 2,300 MH.sub.z with the optimized
dimensions and factors as set forth with FIG. 6. Location of
electrical reference planes such as illustrated in FIG. 6 is
important in determining the measurements for the legs of the
T-transformer with the electrical reference planes defining the
location of the respective conductor ends from the junction to
determine the total physical lengths involved. It should be noted
again that the chip capacitors 37a and 37b are significant in
effectively electrically lengthening the microstrip conductors of
the opposite side diode switch legs of the transformer 33 with, for
the specific 2 GH.sub.z center frequency diode switch, the
additional length to be computed for each leg being 0.040 inches
per capacitor. This additional 0.040 inch length must be subtracted
from the length of the diode switch arms of the T-shaped
transformer in satisfying the design conditions imposed. With a
T-head microstrip conductor width of .108 inches and a stem width
of 0.1045 for the antenna output connection, the electrical
reference plane with respect to the ON side diode switch leg
activated is displaced 0.009 inches from the center line of the
common stem leg 34 in the direction of the ON side, and the
electrical reference plane for the common antenna output connective
stem leg 34 is displaced 0.048 inches in the direction of the
common stem leg 34. The additional resulting dimensions are 0.856
inches from the T vertical center line to a switching diode side
leg end resulting in a 0.206 wavelength from the electrical
reference plane to the switched ON end, and an effective impedance
for the diode ON arm of 39.87 ohms. Please note that the diode
switching arm ends of the transformer 33 are beveled from a width
of 0.045 inches at 45.degree. in order to provide an improved and
proper impedance match with the through conductive ribbon of the
respective switching diodes 42 connected thereto. The vertical
dimension of the common antenna connective leg 34 is 1.058 inches
from the adjacent edge of the switching diode legs of the
T-transformer to the connective end, other than for a relatively
narrow connective extension tab 51 of the leg 34, with this length
resulting in the leg 34 being 0.2415 wavelength effectively long
from the electrical reference plane therefor and having a
characteristic impedance of 40.83 ohms. Further, capacitor 37a and
37b location gaps of 0.020 inches are located at a distance of
0.262 inches from the adjacent edge of the common transformer leg
34. At this spacing the chip capacitors 37a and 37b look like
capacitive reactances to effectively lower to some degree the
operational frequency range of the diode switch 20.
In addition to critical dimensions set forth for the T-transformer
33 of the diode switch 20, it is of interest to note that in
optimum spacing the equivalent of quarter wavelength distance
spacings of the switch center frequency is provided for the three
PIN diodes 42 in each of the diode switch legs of the diode switch
20. This is with the exact distance determined empirically through
measuring the frequency where the isolation is highest when the
diodes are mounted a quarter wavelength apart at the center
operating frequency, and then correcting through scaling the
measured peak isolation frequency to the desired center operating
frequency. Bias line connections are provided from a minus voltage
supply 52 and a positive voltage supply 53 through a switch 54
that, while illustrated in FIG. 4 as being a mechanical switch, has
in at least one instance assumed the form of a fast acting
electronic switch for alternately connecting bias line leads to the
minus voltage supply 52 and the positive voltage supply 53 such as
in accord with the switching waveform illustrated therewith. One
switch output is connected through a lead line 55a that includes
the center conductor 56a of through wall coaxial connector 27a to a
RF rejection filter 57a, such as is the subject matter of copending
application entitled "RF Rejection Filter" filed by applicant on
July 24, 1970 and assigned to the common assignee hereof, and
through a quarter wavelength, of the center operational frequency
of the switch, line 58a from the rejection filter 57a to connection
with the diode switch side section 35a of a transformer diode
switch leg for bias switching ON-OFF control of the diodes 42 of
that leg. In like manner, the minus voltage supply 52 and the
positive voltage supply 53 are connected through switch 54 to and
through line 55b including the center conductor 56b of through wall
coaxial connector 27b to RF rejection filter 57b and through a
quarter wavelength, of the center operational frequency of the
switch, line 58b from the rejection filter 57b to connection with
the diode switch side section 35b of a transformer diode switch leg
for bias switching ON-OFF control of the diodes 42 of that leg.
Each of the lines 55a and 55b also include jumper line sections 59a
and 59b, respectively, extended from the center conductors 56a and
56b to a microstrip connective pad 60a and 60b, respectively, and
also interconnective lines 61a and 61b therefrom to the respective
rejection filters 57a and 57b, respectively.
A switching control voltage waveform, such as that illustrated with
the voltage supplies 52 and 53 in FIG. 4, with switching from a
minus voltage supply of -0.95 volts to a positive voltage supply of
+24 volts with the negative voltage the necessary bias to bias the
diodes of one switch side to conduction and provide switch OFF of
the signal of the RF transmitter of that side, and the positive
back voltage of +24 volts is quite adequate to quickly remove
stored charges in the PIN diode junctions of the diodes of that
side to thereby accomplish turnoff of the diodes and turn ON switch
connection of the RF transmitter of that side. Furthermore, the
positive voltage is of such level to reduce insertion loss of the
diodes through quickly depleting the diode junctions of the side
being positively voltage biased. The particular diodes 42 that have
been used in working diode switches to date are Hewlett Pachard
Associate PIN diodes, part number 5082-3040. FIG. 4a illustrates
the PIN diode equivalent circuit for a diode 42 biased OFF and
transmitter biased switched ON state of operation with two series
connected coils of 0.25.times.10.sup.-.sup.9 Henry with a common
junction thereof connected through a 1170 ohm resistor and a
0.01.times.10.sup.-.sup.12 farad capacitor in parallel to ground.
Then in the diode biased to conduction state with the diodes
conducting approximately a 100 milliamperes current equivalent to
the transmitter switched OFF state for the transmitter of that
side, the equivalent circuit of FIG. 4b shows, again, the
0.25.times.10.sup.-.sup.9 Henry value series connected coils with
the common junction thereof connected serially through a 1 ohm
resistor and a 17.5.times.10.sup.-.sup.12 Henry coil to ground.
With this being the case, when the positive voltage is applied to a
diode switching side, the PIN diodes thereof appear as 50 ohm
impedances with a low loss thereby presented in a good impedance
match in a 50 ohm circuit line. Conversely, when a negative voltage
is applied generating approximately 100 milliamperes of current
through the diodes, the impedance therethrough to ground becomes
very low, in the neighborhood of 1 ohm, resulting in, relatively
speaking, a very high insertion loss to RF signals.
In order to attain performance objectives desired, particularly
VSWR of the assembled switch, matching stubs 62a and 62b are
provided approximately midway in transmission line sections 39a and
39b, respectively, approximately midway between the outermost PIN
diodes 42 of the respective PIN diode switching sections of the
diode switch 20. These matching stubs 62a and 62b are size and
location empirically determined in looking at a swept response of
the return loss of the switch at the antenna connector output to
compensate for a mismatch departure from calculated performance
contributed by coaxial connectors in the dc blocking capacitors in
series with the RF conductors.
In addition to all the features hereinbefore presented, conductive
baffle plates 63a and 63b are required within the diode switch
enclosing shield box 22 in the attainment of the desired isolation
of the transmitter switched OFF side of the switch. These baffle
plates 63a and 63b are seated in milled grooves 64a and 64b, that
are milled through the substrate dielectric 31 and into the ground
plane backing plate 30 of microstrip board 21, wherein they are
securely seated as by screws (detail not shown). The baffle plates
also span the vertical interior dimension of the shield box 22 from
the ground plane plate 30 to intimate close engagement with the
inside of the shield box lid 23 that may be further biased into
tight engagement contact therewith by screws extended through the
lid into the baffle plates 63a and 63b (detail not shown). Further,
the baffle plates 63a and 63b extend from opposite side end
engagement with the outer RF transmitter connective ends of the
diode switch shield box 22 to inner spaced ends 65a and 65b that
are relatively closely spaced from closely adjacent bias lines 58a
and 58b, respectively, and much closer toward the center of the
switch than the rejection filters 57a and 57b, respectively. This
baffle plate structure arrangement within the switch shield case 22
substantially prevents leakage of RF currents between the input and
output connectors that would otherwise occur between the shield box
floor and the metal backing plate of the polyolefin laminated board
that, were such loss to be present, would be manifest as a loss in
the isolation of the OFF side. Coupling of the RF rejection filter
of the transmitter switched ON side of the diode switch is
substantially eliminated from RF coupling to the transmitter
connector of the OFF side of the switch, an RF coupling factor that
would otherwise reduce the maximum isolation possible. Another
problem that is substantially eliminated through the use of the
baffle plates 63a and 63b is that without such plates within the
switch shield box, the interior box size is sufficiently large to
support the dominate TE.sub.10 waveguide mode, such as is well
known to those skilled in the art, with RF propagation in this mode
being through the box interior, not depending on microstrip
conductors, with in such mode, low isolation between the antenna
conductor and the OFF transmitter conductor resulting from such
propagation mode.
The dominant waveguide TE.sub.10 mode cutoff wavelength is
.lambda.c = 2a, where a is the inside width, or opposite end to end
length, of the waveguide. For the example cited, a = 8.25
inches.
Thus, .lambda.c = 2(8.25) = 17.50 inches
The corresponding cutoff frequency is:
fc = V/.lambda.c = 11.8/8.25 = 1.43 GH.sub.z
where V is the free space velocity in inches per nanosecond. The
operating frequency range is 1.7 to 2.3 GH.sub.z , so propagation
would occur. With the baffle plates present, no waveguide path is
possible between the antenna connector and the transmitter
connectors.
The bias lines to the PIN diodes account for no appreciable loss of
the RF signal and are of such high impedance where they are joined
to the main circuit conductor that no change to RF match was
observable. The effectiveness of this bias line decoupling results
from the very low impedance over a broad frequency range that is
possible with the special RF suppression filters acting in
combination with a quarter wavelength of high impedance line which
serves to invert the low filter impedance to a very high impedance
where the bias line is joined to the RF circuit. Other types of
decoupling schemes caused a noticeable and undesirable coupling to
the main circuit conductor.
FIGS. 9, 10, and 11 illustrate actual measured performance
characteristics as compared to theoretical optimum characteristics
with FIG. 9 illustrating a VSWR measured waveform, FIG. 10
isolation in db, and FIG. 11 insertion loss in db and with these
waveforms taken through the 1.7 to 2.3 GH.sub.z frequency range for
a particular switch using the specific PIN diodes identified and
other circuit component values and parameters set forth. The
performance characteristics portrayed by these waveform curves are
together very highly idealistically optimized operational results.
With diode switches designed for operation above about 3 GH.sub.z
RF absorbing material may be used under the shield box lid in the
continued attainment of high isolation with the move to higher
operational frequencies.
With the alternate embodiment of FIG. 12, illustrated in circuit
schematic form, those portions duplicating those of the embodiment
of FIGS. 1 through 8, providing the operational characteristic
curves of FIGS. 9, 10, and 11, are numbered the same and those
portions that include some variance therefrom carry primed numbers.
With this embodiment, the diodes 42 and 42', while they may be PIN
diodes such as used with the other embodiment, the switching diodes
42' of one side are reversed from those of the previously described
embodiment with the cathodes thereof connected to the ground plane
plate and the anodes connected to the switching transmission line
section of that side that interconnects RF transmitter 20a and the
microstrip T-transformer 33. Consistent with these changes, a minus
voltage supply 52' and a positive voltage supply 53' are connected
to respective poles of single throw, double pole switch 54' that,
in turn, is connected through dropping resisters 66a and 66b to
respective side bias connection line systems 55a' and 55b', that
are otherwise the duplicates of lines 55a and 55b with the
components recited therefore with the other described embodiment.
With the embodiment of FIG. 12, however, the minus voltage supply
52' and the positive voltage supply 53' are of substantially
equivalent equal voltage magnitudes, with only one supply at a time
simultaneously applying voltage biasing to conduction voltage to
one side and the reverse bias to the diodes of the other switch
side, interchangeably in each instance as it is switched from one
pole switched position to the other. With this particular
embodiment, all other operational actions are substantially the
same as with the previously described embodiment and a further
description with respect thereto is not again repeated.
Whereas this invention is herein illustrated and described with
respect to a specific embodiment hereof, it should be realized that
various changes may be made without departing from the essential
contributions to the art made by the teachings hereof.
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