Microwave Horn-paraboloidal Antenna

Abstract

A horn-paraboloidal antenna in which a discontinuity is introduced into the feed horn so as to excite higher-order modes from the dominant mode. The various modes are phased in order to radiate a main lobe and side lobes from the horn. The latter lobes reflect from the paraboloid out-of-phase with the main lobe so as to produce a far field radiation pattern having a more uniform signal strength for a given beam angle.

Description

FIELD OF THE INVENTION

This invention relates to a horn-paraboloidal antenna which may be used in communication satellites to provide a radiation pattern having a more uniform signal strength for a given beam angle.

DESCRIPTION OF THE PRIOR ART

Satellites used for communications over a large segment of the earth ideally require an antenna which gives a uniform signal strength over the entire coverage area. Such satellites may use a despun antenna that is carefully aligned to a particular point in its coverage area. Various tolerances in the mechanical design of the satellite result in perturbations causing the antenna boresight to wander about a nominal position. If the antenna pattern is not uniform, the signal strength within the coverage area will fluctuate. The amount of the fluctuation will depend upon the degree of wander and the slope of the radiation pattern at a prescribed point. With a conventional horn-paraboloidal antenna (comprising a parabolic reflector driven from a feed horn), the radiation pattern is ellipse shaped with maximum signal strength on the boresight and dropping significantly near the edges, thus resulting in large signal variations. These variations are undesirable, detracting from the overall system performance.

Various designs utilizing antenna arrays, such as multi-horn antennas, have been developed in order to achieve a more uniform signal strength. However, mechanical difficulties and increased weight resulting from the required complex feeder system have limited the success of such approaches for satellite applications.

In a horn-paraboloidal reflector antenna, the overall radiation pattern can be altered by varying the illumination of the reflector. The ideal situation from the standpoint of highest gain and narrowest beamwidth occurs when the signal distribution over the reflector is uniform in magnitude and phase. However, with such an illumination, a significant proportion of the primary energy from the horn will be lost due to spillover at the edges of the reflector. In addition, the reduction in beamwidth is accompanied by the generation of unwanted secondary or side lobes.

The illumination of the reflector is controlled by varying the dimensional parameters of the horn. An increase in the waveguide cross-section of the feed horn reduces its beam angle but is also accompanied by the appearance of secondary lobes particularly in the E-plane. One cause of the secondary lobes is the presence of higher-order waves which are generated at the mouth and at the throat discontinuity of the horn. In the H-plane the throat reflection is usually small in comparison with the mouth reflection and hence the horn has much lower side lobes. In previous systems, the approach has been to illuminate the whole reflector with in-phase energy from the main lobe and to provide a flare angle from the horn which will cause its side lobes to miss the reflector.

SUMMARY OF THE INVENTION

It has been discovered that if the outer portion of the parabolic reflector is illuminated with energy which will reflect .pi. radians out-of-phase with respect to its center, the out-of-phase energy will subtract and reduce the boresight gain of the far field radiation pattern. However, at a point off the axis, an additional phase difference is introduce due to the differential path length between the two regions. At some angle, this phase difference will be .pi. radians with the net effect that all the aperture energy will appear in phase and add. By a proper balance of energy levels and aperture size, a nearly constant flat-topped radiation pattern can be achieved. As a result, there is less deviation in the gain for a given beamwidth and the overall gain at the beam edges is higher.

The generation of this out-of-phase energy is obtained by permitting the primary set of side lobes to strike the edges of the reflector. In the H-plane however, the amplitude of the side lobes is normally insufficient to achieve the desired radiation pattern. It has also been discovered that by introducing an H-plane discontinuity and by phasing the horn so that the resulting higher-order waves arrive at the mouth of the feed horn .pi. radians out-of-phase with those of the dominant mode, significant side lobes can be generated in the H-plane. As a result, a substantially flat-topped radiation pattern can be produced in the H-plane as well as the E-plane.

Thus, in accordance with the present invention there is provided a microwave antenna comprising a paraboloidal metallic reflector and a metallic feed horn located at the former's focal point. The feed horn has a throat section and a mouth aperture. Included in the throat section in a waveguide feed section which propagates the dominant mode but is beyond cut-off for high-order modes. In addition, there is a discontinuity between the throat section and the mouth aperture for excitation of a higher-order mode, and a differential phasing section for reversing the phase of the higher-order mode with respect to the dominant mode. As a result, the radiation pattern from the feed horn has a main lobe, and also has side lobes which radiate from the reflector out-of-phase with the main lobe.

In a preferred embodiment of the invention, the antenna comprises a pyramidal feed horn having a rectangular waveguide feed section which propagates only TE.sub.10 waves. An H-plane step in the feed horn between the throat section and the mouth aperture excites TE.sub.30 waves while a differential phasing section is used to reverse the phase of the TE.sub.30 waves with respect to the TE.sub.10 waves.

In a still more preferred embodiment, the antenna is used for two waves of orthogonal polarization. In this embodiment the H-plane step for one wave will act as an E-plane step for the other thus resulting in the generation of undesired higher-order waves. These latter waves are suppressed by having a portion of the differential phasing section which is beyond cut-off for the higher-order waves generated by the E-plane step.

BRIEF DESCRIPTION OF THE DRAWINGS

An example embodiment of the invention will now be described with reference to the accompanying drawings in which:

FIG. 1 is a perspective view of a microwave antenna in accordance with the present invention;

FIG. 2 is a substantially side-elevational view of the microwave antenna illustrated in FIG. 1;

FIG. 3 is a cross-sectional view, taken along the line III--III of FIG. 2;

FIG. 4 is a cross-sectional view, taken along the line IV--IV of FIG. 1;

FIG. 5 is a graph of relative power and phase vs normalized horizontal reflector aperture of the microwave antenna illustrated in FIG. 1;

FIG. 6 is a graph of relative power vs the far field directional pattern of a conventional horn-paraboloidal antenna and the microwave antenna illustrated in FIG. 1.

DESCRIPTION OF THE PREFERRED EMBODIMENT

Referring to FIG.'s 1 and 2, the microwave antenna basically comprises a paraboloidal metallic reflector 10 having a metallic pyramidal feed horn 11 located at its focus. The feed horn 11 is driven from a rectangular waveguide feed section 12 which is driven from an ortho-coupler 13 in a well known manner. The whole antenna assembly is mounted on a rotary joint 14.

FIG.'s 3 and 4 show horizontal and vertical cross-sections of the metallic pyramidal feed horn 11 respectively. The feed horn 11 comprises a throat section generally 20 and a mouth aperture 21. The throat section 20 includes at least a portion of the rectangular waveguide feed section 12. In addition, a discontinuity, shown as a H-plane step 22, is located between the throat section 20 and the mouth aperture 21. Following the H-plane step 22 is a differential phasing section 23 which has a uniform cross section. Also shown are a plurality of tapered vanes 24 which intercept the E-plane field for one polarization and a further plurality of tapered vanes 25 which intercept the E-plane filed for the other polarization.

The microwave antenna of the present invention may be used in, although not necessarily limited to, a communications satellite. In a typical such application, the received band would be at 6 GHz. while the transmit band would be at 4 GHz. In addition, the 6 GHz. signals are vertically polarized while the 4 GHz. signals are horizontally polarized.

In the present embodiment, the primary requisite is to provide a more uniform signal strength of the horizontal radiation pattern of the antenna shown in FIG.'s 1 and 2, at both the 4 GHz. and 6 GHz. bands. It necessary, the technique could be extended to provide a more uniform radiation pattern of the vertical pattern. Unless otherwise stated, the following description will be directed towards the two horizontal radiation patterns of the antenna.

Initially, the horizontal length and flare angle of the horn 11, as shown in FIG. 3, are selected to generate the main lobe and a primary set of side lobes for a horizontally polarized E-plane radiation pattern at 4 GHz. As is well known, utilizing a large cross-section at the mouth aperture 21 results in a narrow beam with high side lobes in the E-plane. However, at 6 GHz. the horn would have very low side lobes since it is an H-plane pattern, and would be ineffectual for the purpose of beam shaping. To create the side lobes in this plane, the symmetrical H-plane step 22 (with respect to the 6 GHz. field) is introduced which generates the TE.sub.30 mode from an incident TE.sub.10 mode. Thus, as shown in FIG.'s 1, 2 and 3, the discontinuity 22 is an H-plane step at 6 GHz. while an E-plane step at 4 GHz.

Being an E-plane step at 4 GHz., the discontinuity 22 will tend to generate the TE.sub.12 and TM.sub.12 mode pair. These latter modes would have undesirable effects at 4 GHz. and thus cannot be allowed to propagate. In addition, the TE.sub.30 mode must arrive at the mouth aperture 21 .pi. radians out-of-phase with respect to the TE.sub.10 mode in order to generate the necessary side lobes. However, the horizontal length and flare dimensions have already been set by the 4 GHz. requirements. Therefore, a differential phasing section 23 is added in order to provide the necessary phase shift of .pi. radians between the TE.sub.10 and TE.sub.30 modes at 6 GHz. This can be done by assuming the two modes (TE.sub.10, TE.sub.30) to have an initial phase difference of 180.degree. at the mouth aperture 21. The differential phase velocity between the two modes can be integrated along the length of the horn 11 until a point is reached where the two modes are in phase. This is the first possible position for the H-plane discontinuity 22. However, at this point, the cross-section of the horn 11 is of sufficient size to propagate the TE.sub.12 and TM.sub.12 modes at 4 GHz. and therefore cannot be used. The throat section 20 is therefore extended to a cross-section where the latter mode pair are beyond cutoff at 4 GHz. but will still propagate the TE.sub.30 mode at 6 GHz. At this point, it is necessary to add the differential phasing section 23 of uniform cross-section to bring the TE.sub.10 and TE.sub.30 modes in phase again. The H-plane step 22 is then inserted at that point. The step size is selected to produce sufficient TE.sub.30 mode to obtain the desired side lobe levels at 6 GHz. In order to assure that all the energy of the TE.sub.30 mode propagates toward the mouth aperture 21, the throat section 20 is further reduced to beyond cut-off for all modes except the dominant TE.sub.10 mode. Hence, only 6 GHz. signals of the TE.sub.10 along the waveguide feed section 12 in the vertical plane.

Thus, by utilizing the flare angle and length of the horn 11 to control the E-plane pattern at 4 GHz. and by utilizing the H-plane step 22 and the phasing section 23 to control the generation of the TE.sub.30 mode at 6 GHz., the required side lobes can be generated at the mouth aperture 21 for both horizontal signal patterns.

Since the beamwidth of a given aperture decreases in portion to the signal wavelength, it is necessary to widen the horizontal beamwidth at 6 GHz. relative to that at 4 GHz., so that the far field radiation patterns will be approximately the same. This requires concentrating more of the 6 GHz. energy at the center of the reflector 10 by decreasing the horizontal beamwidth of the feed horn 11 at that frequency. To achieve this, the width of the horn 11, as shown in FIG. 3, is increased. However, such an increase would likewise affect the beamwidth at 4 GHz. To overcome this, the tapered vanes 24 are inserted parallel to the E-plane at 4 GHz. The vanes 24 are spaced closer than one-half wavelength apart at this frequency so that the region between the vanes 24 is in cutoff and therefore the 4 GHz. fields parallel to the vanes 24 cannot propagate between them. The 6 GHz. electric fields perpendicular to the vanes 24 are not affected and are only restricted by the sidewalls of the horn 11 as shown in FIG. 3. By using the vanes 24 to define the 4 GHz. E-plane horn dimensions, the size of the horn 11 is effectively increased by the depth of the vanes 24 at 6 GHz. so as to narrow the radiation pattern from the horn 11 at that frequency.

In the vertical direction, the flare angle and length of the horn 11 are selected to optimize the 4 GHz. H-plane illumination of the reflector 10. However, at 6 GHz., the horn 11 is an E-plane radiator thus producing sides lobes. Because of the difference in frequencies between the two bands the 6 GHz. vertical pattern from the feed horn 11 is narrower than that at 4 GHz. The pattern is further narrowed due to the difference in efficiencies in the E-plane and H-plane. Utilizing only the walls of the horn 11 as shown in FIG. 4, would result in side lobes from the 6 GHz. E-plane signal stricking the reflector 10. This would result in a shaped beam at 6 GHz. in this plane. However, since the width of the reflector 10 was selected for a conventional radiation pattern at 4 GHz., the reflector 10 is not large enough to give the correct shaped beamwidth. As a result, the shaped beam at 6 GHz. would be excessively wide resulting in a loss of gain. To overcome this, the vanes 25 are inserted parallel to the 6 GHz. E-plane, thereby widening this pattern from the feed horn 11 sufficiently to divert most of the side lobe energy off the edge of the reflector 10. This provides a more uniform illumination of the reflector 10 thereby narrowing the 6 GHz. far field vertical radiation pattern of the antenna.

The width of the reflector 10, and the flare angle and length of the horn 11 are selected so that both the main lobe and the primary side lobes of both horizontal radiation patterns strike the surface of the reflector 10. As stated above, the side lobes which radiate from the reflector 10 must be .pi. radians out-of-phase with the main lobe. Minor variations to the shape of the paraboloidal reflector 10 can be made in order to optimize this phase relationship. In addition, the edges of the reflector 10 can be shaped as shown in order to prevent unwanted signal energy from striking the corners of the reflector 10 thus improving the radiation patterns. This also saves weight, an important design aspect of a satellite antenna.

FIG. 5 illustrates the relative power and the phase of signals at both 4 GHz. and 6 GHz. vs the normalized horizontal reflector aperture. The side lobes at 4 GHz. are controlled by the flare angle and length of the horn 11, as shown in FIG. 3, which alter the E-plane illumination and phase distribution from the mouth aperture 21. The side lobes at 6 GHz. are generated primarily by the H-plane step 22. As is apparent from the bottom portion of FIG. 5, the phase of the two side lobes at both 4 GHz. and 6 GHz. is substantially .pi. radians with respect to that of the main lobe.

FIG. 6 illustrates a typical far field horizontal directional pattern of the microwave antenna described above together with a typical pattern of a conventional horn-paraboloid. With a beamwidth of .+-.4.2.degree. the boresight gain is reduced by about 1 db. over the conventional horn-paraboloid. However, the signal strength at the .+-.4.2.degree. points is about 1.3 db. greater, thereby yielding an overall reduction in signal variation of 2.3 db.

By utilization of the present invention, an improved horizontal directional pattern is achieved at both the 4 GHz. and 6 GHz. bands. While not illustrated in the present embodiment, the invention could be readily extended to provide the same advantages for both the vertical radiation patterns.

Claims

What is claimed is:

1. A microwave antenna comprising:

a paraboloidal metallic reflector for illumination by a main lobe and a primary set of side lobes at its center and edge portions respectively;

a metallic feed horn disposed at the focal point of the reflector and having a throat section and a mouth aperture;

the throat section including a waveguide feed section which propagates the dominant mode and is beyond cutoff for high-order modes;

a discontinuity in the feed horn between the throat section and the mouth aperture for excitation of a higher-order mode; and

a differential phasing section in the feed horn between the discontinuity and the mouth aperture for reversing the phase of said higher-order mode with respect to said dominant mode;

radiated signals resulting from the propagation of said modes produce the main lobe, and the primary set of side lobes which radiate from the reflector out-of-phase with the main lobe.

2. A microwave antenna comprising:

a paraboloidal metallic reflector for illumination by a main lobe and a primary set of side lobes at its center and edge portions respectively;

a metallic pyramidal feed horn disposed at the focal point of the reflector and having a throat section and a mouth aperture;

a throat section including a rectangular waveguide feed section which propagates TE.sub.10 waves and is beyond cutoff for higher-order waves;

an H-plane step in the feed horn between the throat section and the mouth aperture for excitation of TE.sub.30 waves from the TE.sub.10 waves; and

a differential phasing section in the feed horn between the H-plane step and the mouth aperture for reversing the phase of the TE.sub.30 waves with respect to the TE.sub.10 waves;

radiated signals resulting from the propagation of said modes produce the main lobe, and the primary set of side lobes which radiate from the reflector out-of-phase with the main lobe.

3. A microwave antenna for first and second electromagnetic waves of orthogonal polarization;

the antenna comprising:

a paraboloidal metallic reflector for illumination by a main lobe and a primary set of side lobes at its center and edge portions respectively;

a metallic pyramidal feed horn disposed at the focal point of the reflector and having a throat section and a mouth aperture;

the throat section including a rectangular waveguide feed section which propagates the TE.sub.10 mode and is beyond cutoff for higher-order modes of both the first and second electromagnetic waves;

an H-plane step in the feed horn between the throat section and the mouth aperture for excitation of the TE.sub.30 mode from the TE.sub.10 mode of the first electromagnetic waves; and

a differential phasing section in the feed horn between the H-plane step and the mouth aperture for reversing the phase of the TE.sub.30 mode with respect to the TE.sub.10 mode of the first electromagnetic waves, said phasing section having at least a portion thereof adjacent said step which is beyond cutoff for higher-order modes of the second electromagnetic wave;

radiated signals resulting from the propagation of said mode produce the main lobe, and the primary set of side lobes which radiate from the reflector out-of-phase with the main lobe.

4. A microwave antenna as defined in claim 3 in which said portion has a uniform cross-section.

5. A microwave antenna as defined in claim 3 in which the frequency of said first electromagnetic waves is higher than that of said second electromagnetic waves, and

in which the feed horn additionally comprises a plurality of tapered vanes parallel to the E-plane of the second electromagnetic waves, said vanes being less than one-half wavelength apart so as to decrease the effective mouth aperture of the horn for the second electromagnetic waves.