U.S. patent number 3,631,528 [Application Number 05/063,727] was granted by the patent office on 1971-12-28 for low-power consumption complementary driver and complementary bipolar buffer circuits.
Invention is credited to Robert S. Green.
United States Patent |
3,631,528 |
Green |
December 28, 1971 |
**Please see images for:
( Certificate of Correction ) ** |
LOW-POWER CONSUMPTION COMPLEMENTARY DRIVER AND COMPLEMENTARY
BIPOLAR BUFFER CIRCUITS
Abstract
A low-power complementary driver comprising a first
complementary inverter is provided with a special circuit for
turning the N-channel device off before the P-channel device is
turned on and vice versa to greatly reduce power consumption. This
is accomplished by two additional complementary inverters having
different transition voltages connected between the input signal
and the gates of the N and P-channel devices, respectively, of the
first complementary inverter. A complementary-bipolar inverter is
provided using the same technique with one of said additional
inverters being replaced by a double inverter.
Inventors: |
Green; Robert S. (Murray,
UT) |
Family
ID: |
22051092 |
Appl.
No.: |
05/063,727 |
Filed: |
August 14, 1970 |
Current U.S.
Class: |
326/84; 326/103;
326/17 |
Current CPC
Class: |
H03K
19/0948 (20130101); H03K 19/0013 (20130101) |
Current International
Class: |
H03K
19/00 (20060101); H03K 19/0948 (20060101); H03k
017/60 () |
Field of
Search: |
;307/205,237,238,251,279,288,303,304,313 |
References Cited
[Referenced By]
U.S. Patent Documents
Primary Examiner: Krawczewicz; Stanley T.
Claims
I claim:
1. A low power consumption complementary transistor circuit
comprising an input node adapted to receive an input signal having
a transition between first and second signal levels and an output
node, signal generating means operatively connected to said input
node for generating, in response to the transition of said input
signal, first and second signals each having a transition between
third and fourth signal levels, said transition of said first
signal preceding in time said transition of said second signal, a
driving stage comprising first and second semiconductor switching
devices having their output circuits connected across a reference
voltage source, said first and second switching devices receiving
said first and second signals respectively at their control
terminals and having their output circuit terminals connected to
said output node, said transitions of said first and second
signals, respectively, shifting said first switching device from a
conductive to a nonconductive state and shifting said second
switching device from a nonconductive to a conductive state.
2. The circuit of claim 1 wherein said driving stage comprises a
complementary inverter, and wherein said first and second switching
devices are complementary devices.
3. The circuit of claim 1, wherein said signal generating means
comprises first and second complementary inverter stages having
different transition voltages and adapted to generate said first
and second signals, respectively.
4. The circuit of claim 1, wherein said signal generating means
comprises a first complementary inverter stage adapted to generate
said first signal and a double inverter comprising second and third
complementary inverter stages adapted to generate said second
signal, said first and second inverter stages having different
transition voltages.
5. The circuit of claim 2, wherein said signal generating means
comprises first and second complementary inverter stages having
different transition voltages and adapted to generate said first
and second signals respectively.
6. The circuit of claim 2, wherein said complementary devices are
N- and P-channel field effect transistors respectively.
7. The circuit of claim 5, wherein said complementary devices are
N- and P-channel field effect transistors respectively.
8. The circuit of claim 4 wherein said switching devices comprise a
bipolar transistor and a field effect transistor respectively.
9. The circuit of claim 1, wherein said third and fourth signal
levels are equal to said first and second signal levels,
respectively.
10. The circuit of claim 8, wherein the control terminal of said
field effect transistor switching device is connected to the output
of said double inverter, and the base of said bipolar transistor is
connected to the output of said first complementary inverter
stage.
11. The circuit of claim 3, wherein said first and second
complementary inverter stages each comprise a P-channel and
N-channel field effect transistor connected in series across a
reference voltage source, the ratio of the channel widths of said
P- and N-channel field effect transistors of said first inverter
stage being different than the ratio of the channel widths of said
P- and N-channel field effect transistors of said second inverter
stage.
12. The circuit of claim 4, wherein said first and second
complementary inverter stages each comprise a P-channel and
N-channel field effect transistor connected in series across a
reference voltage source, the ratio of the channel widths of said
P- and N-channel field effect transistors of said first inverter
stage being different than the ratio of the channel widths of said
P- and N-channel field effect transistors of said second inverter
stage.
Description
This invention relates to transistor circuits and more particularly
to MOS and MOS-bipolar complementary circuits.
The metal-oxide semiconductor (MOS) field effect transistor (FET)
is a voltage controlled device which exhibits an extremely high
input resistance, allowing direct coupling in digital circuits.
MOSFETS may be P-channel or N-channel. In a typical P-channel
device, two highly doped P-type areas defining the source and the
drain are diffused into an N-type silicon substrate. (In practice
they are separated by about 0.2 mils for a driver device.) A thin
insulating material, commonly silicon dioxide, is grown or
deposited on the silicon substrate between the source and the drain
over which a metal electrode is deposited to form the insulated
gate electrode. If the gate is zero potential with respect to the
source, no current flows from source to drain because the
PN-junctions are reverse biased. As the gate is made negative,
positively charged holes are induced into a thin channel region in
the silicon substrate adjacent the oxide interface. When enough
holes are induced in the channel to overcompensate for the N-type
doping, the channel changes from N-type to P-type and is said to be
inverted; ohmic conduction then occurs between source and drain.
The gate voltage at which conduction begins is known as the
threshold voltage, V.sub.T. As the gate is made more negative the
inversion layer is driven deeper and conduction continues to
increase, approaching a steady state "on" or saturation value. An
N-channel device operates on the same principles with the
polarities reversed--that is, the channel is inverted from P-type
to N-type and the device is normally on (conductive) at zero gate
bias as opposed to the P-channel device which is normally off
(nonconductive) at zero gate bias.
Since it is desirable to use an initially off device for switching
in digital circuits, commercial MOS single polarity integrated
circuits (which operate on negative supply voltages) commonly
utilize the P-channel device. In the common mode of operation of
such devices, the source is usually connected to the substrate and
grounded and the drain is connected to a negative potential via a
load resistor. The operative node capacitances at the junction
between the output circuits of the switching device and the load
resistor are charged through the resistor or discharged through the
switching device, depending upon the control signal at the gate
terminal of the switching device.
Single polarity MOSFET circuits of this type have a number of
disadvantages. It will be apparent that if the device is turned on
by the control signal, current and power is drawn continuously to
discharge the node capacitance. Another disadvantage of this
structure is that because the gate of the load resistor is tied to
its drain, there is always an offset voltage equal to V.sub.T
across the load resistor. This can be avoided by providing a
separate supply to the gate. However, the provision for a separate
supply voltage is cumbersome and generally undesirable.
Complementary MOS technology has long been regarded as the ideal
solution to many of these difficulties. In effect, a complementary
MOS structure replaces the load resistor with an N-channel device.
With this arrangement a positive supply voltage is used and the
N-channel device is normally off at zero (ground) gate bias while
the P-channel device is normally on when its gate is at 0 volts.
The node capacitances are thus charged or discharged only during
switching. As a result, a complementary gate draws approximately
zero power while in the quiescent state and circuit voltages swing
the full supply voltage, making two supplies unnecessary. During
switching enough current must be supplied to charge the stray
capacitance associated with each node that changes state. Since
this is done through the saturation resistance of an active device
instead of through a load resistor, the charging time is very short
and operating frequency very high. Complementary structures are
therefore extremely useful for low-power high-speed applications,
such as large computer memories.
While complementary circuits have extremely low standby (DC) power
characteristics, they do consume considerable power during
switching since both N- and P-type devices are conducting current
during this period. It has been found that the major portion of
this current does not go to charging the load capacitor but instead
effectively shorts out the power supply. As a result, the power
consumed during switching is largely dissipated and the capacitive
driving capability and switching speeds of such circuits are
limited.
It is a primary object of the present invention to design a
complementary MOS driver circuit with significantly reduced power
consumption during switching.
It is a further object of the present invention to provide a
complementary MOS circuit which is adapted to drive a large
capacitive load at a high speed and still dissipate a small amount
of power.
It is yet another object of the present invention to provide a
complementary MOS-bipolar inverter circuit in which substantially
all the current is used to charge the load capacitor.
It is still another object of the present invention to provide
circuits of the type described in which the output rise and fall
time is reduced and power dissipation is minimized at all
frequencies.
To these ends there is provided a complementary MOS driver circuit
having three inverter stages. Two of these stages comprise small
complementary MOS devices and are both adapted to invert the input
signal with a slight time delay between their outputs. The inverted
output which is first in time is applied to the initially on device
and the subsequent inverted signal is applied to the initially off
device of the third inverter stage so that the initially on device
is rendered fully nonconductive prior to the initiation of
conduction through the initially off device.
The same technique can be used when a bipolar output is used in
place of the P-channel device by using a double inverter to drive
the N-channel device.
To the accomplishment of the above, and to such other objects as
may hereinafter appear, the present invention relates to MOS and
MOS-bipolar complementary circuits, as defined in the appended
claims and as described in this specification, taken together with
the accompanying drawings, in which:
FIG. 1 is a circuit diagram schematically illustrating a typical
prior art complementary MOS inverter circuit and showing a typical
pulse-type input signal;
FIG. 2 is a circuit diagram schematically illustrating the
complementary MOS driver circuit of the present invention and
showing a typical pulse-type input signal;
FIG. 3 is a circuit diagram schematically illustrating a prior art
complementary MOS-bipolar inverter circuit and showing a typical
pulse-type input signal;
FIG. 4 is a circuit diagram schematically illustrating the
complementary MOS-bipolar inverter circuit of the present
invention, and showing a typical pulse-type input signal;
FIGS. 5 and 6 are graphical illustrations of the input and output
signal and the current through both the P- and N-channel devices of
the circuit of FIG. 1 versus time for the positive and negative
going edges, respectively, of the input pulse shown in FIG. 1;
FIGS. 7 and 8 are graphical illustrations of the output signals
from the three inverter stages of the circuit of FIG. 2 versus time
for the positive and negative going edges, respectively, of the
input pulse shown in FIG. 2; and
FIG. 9 illustrates the time relationship of the input and output
signals and the signals generated by the three intermediate
inverter stages of the circuit of FIG. 4.
A typical prior art complementary inverter circuit is illustrated
in FIG. 1. P- and N-channel field effect devices designated P1 and
N1 respectively (both of the enhancement type) are connected in
series between a positive supply V.sub.DD and ground. An input
signal in the form of a positive going pulse generally designated 6
is applied to the gate terminals 2 and 4 of FETs P1 and N1,
respectively, the output being defined at the junction 8 between
the output circuits of the two devices. Load capacitor C1
represents the capacitive load driven by the circuit and may
comprise the stray or interelectrode capacitances of the devices
driven or a combination of such stray capacitance and discrete
capacitors. In the present illustration C1 represents approximately
8 pf. In operation when the input is low (at ground or 0 volts),
the N-channel device N1 is rendered nonconductive (off) as its gate
is returned to its source voltage (ground). The P-channel device
P1, however, has its gate at the most negative potential in the
circuit and is thus rendered conductive (on). Under these
conditions the output at 8 goes high (positive) and is inverted
relative to the low-input signal, C1 being charged positive. When
the input goes high, FET N1 is turned on and FET P1 is turned off,
resulting in a low (ground) output. It should be noted that in
either quiescent state, one device is on (presenting a low driving
impedance to C1) and the other device is off (limiting the current
drain, and thus power, to the leakage value). However, as
previously noted, because the output current characteristics of MOS
devices during switching are gradual, both devices are conducting
during a portion of the time that switching is taking place.
FIGS. 5 and 6 show the current characteristics of P1 and N1 (in
milliamperes) superimposed on the input and output signals (in
volts) during switching. For purposes of illustration the V.sub.DD
supply and the input pulse amplitude will be assumed to be +12
volts. It should be noted, however, that the pulse level need not
be equal to V.sub.DD. The only requirement is that both the input
pulse level and the V.sub.DD supply (output pulse level) be
sufficiently positive to produce effective switching. It will be
recalled that MOSFET conduction begins when the gate bias reaches
the threshold voltage of the device V.sub.T. Thus, as shown in FIG.
5, current begins to flow through N1, as the input rises to V.sub.T
(which in the present case is approximately 2.5 volts). Since FET
P1 is still fully "on" at this point (its gate bias being 12-2.5 or
9.5 volts) current also begins to flow through FET P1. The current
in both devices rises as FET N1 becomes more conductive. When the
input reaches approximately 5 volts, P1 begins to turn off and
current in both devices begins to drop off reaching zero when the
input reaches approximately 9.5 volts (within 2.5 volts or V.sub.T
of V.sub.DD). The magnitudes of the peak currents are proportional
to the P1 and N1 device widths. The difference between the currents
through P1 and N1 is the current discharging the load capacitor to
0 volts (as represented by the output voltage curve), and is
proportional to the load capacitance (8 pf. for the present
case).
As illustrated in FIG. 6, the reverse process occurs at the
negative going edge of the input pulse. It will be noted that in
this case the current through P1 exceeds the current through N1 by
an amount necessary to charge C1 to the V.sub.DD supply level. In
both cases it will be noted that N1 and P1 are conducting while the
input is between V.sub.DD - V.sub.TP and V.sub.TN (about 400 ns.),
where V.sub.TP and V.sub.TN represent the threshold voltages (in
this case -2.5 and +2.5 volts) of P1 and N1 respectively. Thus it
can be seen that current and power consumption is a function of the
input rise and fall time, which in most cases it is not possible to
reduce. It will be apparent, however, that if the P-channel device
can be turned fully off before the N-channel device is turned fully
on (for the positive going edge of input pulse 6) and vice versa
(for the negative going edge of the input pulse 6), current and
power consumption can be significantly reduced.
The circuit of FIG. 2 is adapted to accomplish this result. As
there shown a complementary inverter generally designated 10
comprises N- and P-channel devices N2 and P2 respectively connected
in series between the V.sub.DD supply and ground. Inverter 10 is
adapted to drive a large capacitive load C2 at its output node 12
at the junction of the output circuits of FETs P2 and N2. The gate
terminal 14 of FET P2 is driven by the output A of a second
complementary inverter stage generally designated 16 comprising
complementary FETs P3 and N3 connected in series between the
V.sub.DD supply and ground and having an output node 18 at the
junction of their output circuits. Similarly the gate 20 of FET N2
is driven by the output B of a third inverter stage 22 comprising
FETs P4 and N4 connected in series between V.sub.DD and ground and
having an output node 24 at the junction of their output circuits.
The gates of P3, N3, P4 and N4 are all driven by the input signal
shown in the form of a positive pulse 26 (having an amplitude of
V.sub.DD). When the input is low (0 volts) as at 28 the outputs of
inverter stages 16 and 22 are both high (at the V.sub.DD level) and
thus FET N2 is on and FET P2 is off, making the output at 12 low.
In order to turn N2 off before P2 turns on as the input goes high
along the positive going edge 30 and vice versa along the negative
going edge 32, the output A from inverter 16 must be slow going
negative and the output B from inverter 24 must be slow going
positive, i.e., the transition voltage of inverter 16 must be less
than that of inverter 22. The transition voltage of a complementary
inverter is defined as the point at which the input and output
voltages are equal and can be calculated by the equation:
##SPC1##
where K is a function of the channel width to length ratio W/L of
the device, the subscripts representing the P and N devices,
respectively.
By making the ratio of the channel widths of the P3 and N3 devices
relatively high and the ratio of the widths of the P4 and N4
devices relatively low, the transition voltage of inverter 16 may
be made lower than that of inverter 24. This is illustrated
graphically in FIGS. 7 and 8. As shown in FIG. 7, as the input
rises output B begins to fall first, having a transition voltage
V.sub.TR (B) of approximately 5.5 volts. Output A, however, does
not begin to fall steeply until output B has almost reached 0
volts, the transition of A occurring at approximately 7 volts. It
will be apparent from FIG. 7 that output A does not drop to a level
sufficient to begin turning on FET P2 (9.5 volts) until the level
of output B has reached a level sufficient to fully turn off FET N2
(2.5 volts).
As illustrated in FIG. 8, the situation is just the reverse at the
negative going edge of the input pulse. Thus, output B does not
reach the turn-on level of N2 (2.5 volts) until output A has
reached the level sufficient to turn FET P2 fully off (9.5 volts).
It will thus be apparent that current never flows through devices
P2 and N2 simultaneously so that the power supply is never shorted
through inverter 10 and only the small amount of current necessary
to charge and discharge load capacitor C1 flows through P2 and N2,
respectively, during switching. As a result the rise and fall time
of the output signal is considerably reduced from about 400 ns.
(FIGS. 5 and 6) to about 50 ns. (FIGS. 7 and 8) and power
consumption is kept to a minimum. Moreover, because current and
power consumption is limited to that necessary to charge the load
capacitor, the P2 and N2 devices may be designed with an "on"
resistance low enough to drive a capacitive load of up to 200 pf.
without a large power consumption. While power is of course
dissipated by the concurrent conduction of the P and N devices in
inverter stages 16 and 24, this power consumption can generally be
kept relatively low by reducing the size of these devices since
these stages need only drive the gates 14 and 20 of FETs P2 and N2,
respectively.
The foregoing technique may also be adapted to circuits having a
complementary MOS-bipolar output. FIG. 3 shows a typical prior art
complementary MOS-bipolar inverter circuit, having a first inverter
stage 36 comprising complementary FETs P5 and N5 having their
output circuits connected in series between a positive voltage
supply V.sub.DD and ground. A second inverter stage 38 comprises an
NPN-bipolar transistor B1 having its collector connected to the
V.sub.DD supply and its emitter connected to the source of an
N-channel FET N6 having its drain connected to ground. The gate
terminals of FETs P5, N5 and N6 all receive the input signal shown
in the form of a positive pulse 40 (of amplitude V.sub.DD). The
base of transistor B1 receives the output of inverter 36 at the
junction 42 of the output circuits of FETs P5 and N5. The output of
circuit 38 at the junction 44 of the emitter of B1 and the source
of N6 drives a capacitive load C3.
It will be apparent that when the input is low P5 is on, N5 and N6
are off and the output of inverter 36 at node 42 is high.
Accordingly, B1 is on and capacitor C3 is charged to the V.sub.DD
supply level. As the input 40 goes high P5 is turned off, N5 and N6
are turned on and the output of inverter 36 at node 42 goes low.
Thus B1 is turned off and capacitor C3 is discharged to ground via
FET N6. The NPN-bipolar transistor is desirable since it reduces
the capacitance seen at node 42 and thus increases switching speed.
However, here again considerable power is consumed as both B1 and
N6 are conducting for some appreciable time during switching.
FIG. 4 illustrates a complementary MOS-bipolar inverter utilizing
the same technique as that used in the driver circuit of FIG. 2. As
there shown, the circuit comprises a complementary bipolar circuit
44, a complementary MOS inverter stage 46 and a double inverter
stage 48 comprising two MOS complementary inverters 50 and 52 in
cascade. Complementary bipolar circuit 44 comprises an NPN-bipolar
transistor B2 having its collector connected to the V.sub.DD supply
and its emitter connected to the source of an N-channel FET N7
having its drain terminal connected to ground. Inverter stages 46,
50 and 52 comprise complementary devices P8, N8, P9, N9, P10, and
N10, respectively, connected in the usual manner in series between
the V.sub.DD supply and ground. The gate terminals of FETs P8, N8,
P9, and N9 all receive the input signal shown in the form of a
positive pulse 53. FETs P10 and N10 have their gate terminals
connected to the output Y from inverter stage 50 at the junction
between the output circuits of FETs P9 and N9. The gate terminal of
FET N7 receives the output Z from inverter stage 52 at the junction
of the output circuits of FETs P10 and N10. The base of transistor
B2 receives the output X from inverter stage 46 at the junction
between the output circuits of FETs P8 and N8. The output of
circuit 44 at the junction between the emitter of transistor B2 and
the source of FET N7 drives a capacitive load designated C4. As
shown an additional P-channel FET P11 receiving the output Z at its
gate terminal may be connected in parallel with bipolar transistor
B2 between the V.sub.DD supply and the output to provide a full
voltage swing at the output. However, FET P11 is not essential to
the operation of the circuit.
The operation of the circuit of FIG. 4 will now be apparent. When
the input 53 is low outputs X and Y are high and output Z is low.
Accordingly, FET N7 is off (nonconductive) and transistor B2 is on
(conductive) and load capacitor C4 is charged to the V.sub.DD level
through transistor B2. When the input goes high outputs X and Y go
low and output Z goes high. Transistor B2 is turned off and FET N7
is turned on to discharge C4 to ground. By making the ratio of the
widths of FETs P8 and N8 relatively low and the ratio of widths of
FETs P9 and N9 relatively high the transition voltage of inverter
46 may be made higher than that of inverter 50. Thus output X is
slow going positive and output Y is slow going negative. The
transition voltage of inverter 52 is unimportant since the rise and
fall of output Z must of necessity occur within the fall and rise
times respectively of output Y. Thus if output Y is slow going
negative, output Z is slow going positive. This is illustrated
schematically in FIG. 9. As there shown, as the input rises output
X falls first, its transition voltage being indicated at 54. Output
Y does not begin falling until output X has reached a level
sufficient to turn off transistor B2. The transition voltage of
output Y is indicated at 55. Output Z, of course, cannot begin its
rise until output Y begins to fall. Thus FET N7 remains fully off
until transistor B2 is turned fully off to prevent the shorting out
of the power supply via transistor B2 and FET N7. While some
current will flow through FET P11 during switching, this can be
kept to a minimum by making this FET small since it is not the
driving device. When the input goes low, first output Z goes low to
turn FET N7 off and FET P11 on, after which output X goes high to
turn transistor B2 on to charge load capacitor C4 positive. In
practice the voltage drop across the collector-emitter terminals of
transistor B2 would prevent capacitor C4 from being charged to the
full supply voltage. Accordingly, FET P11 is effective to complete
the swing to the V.sub.DD level.
Again it will be noted that the circuit of FIG. 4 is adapted to
drive high capacitive loads (up to 200 pf.) without significant
power dissipation. The devices of inverter stages 46, 50 and 52 are
small enough to limit current and power in those stages during
switching to a minimum.
It will thus be seen that the complementary circuits of both FIGS.
2 and 4 are adapted to drive extremely high capacitive loads with
very little power dissipation. This is made possible by the ability
to control the transition voltage of an inverter stage by varying
the relative widths of the P- and N-channel devices. By utilizing
inverter stages having different transition voltages between the
input signal source and the complementary drive devices, the
initially on device may be switched off before the initially off
device is switched on to completely eliminate unnecessary current
drain through both devices. As a result virtually all current goes
to driving the capacitive load and the output signal rise and fall
time is considerably reduced thereby increasing the switching speed
of the driven circuit.
While only two of the embodiments of the present invention are here
specifically described, it will be apparent that many variations
may be made thereto without departing from the scope of the instant
invention as defined in the following claims.
* * * * *