U.S. patent number 3,579,088 [Application Number 04/814,393] was granted by the patent office on 1971-05-18 for ferroresonant transformer with controllable flux.
Invention is credited to Taylor C. Fletcher, Lawrence M. Silva, Bruce L. Wilkinson.
United States Patent |
3,579,088 |
Fletcher , et al. |
May 18, 1971 |
**Please see images for:
( Certificate of Correction ) ** |
FERRORESONANT TRANSFORMER WITH CONTROLLABLE FLUX
Abstract
A ferroresonant transformer with means for varying saturation
flux capacity for controlling the transformer output. A
ferroresonant transformer with at least a portion of the core
structure carrying the secondary winding having two separate
sections providing parallel magnetic paths for the secondary and
resonant flux, with a control winding on one of the sections and a
switching circuit for opening and closing the control winding. A
low frequency version utilizing E and I laminations with the
control winding on one of the outer legs of the E. Another low
frequency version utilizing E and I laminations with the control
winding encircling a portion of the secondary magnetic circuit. A
high frequency version utilizing a plurality of toroid cores with
the control winding on one of the cores.
Inventors: |
Fletcher; Taylor C. (Fullerton,
CA), Silva; Lawrence M. (Portuguese Bend, CA), Wilkinson;
Bruce L. (Torrance, CA) |
Family
ID: |
25214929 |
Appl.
No.: |
04/814,393 |
Filed: |
April 8, 1969 |
Current U.S.
Class: |
323/248 |
Current CPC
Class: |
G05F
1/13 (20130101); G05F 3/06 (20130101); H01F
29/146 (20130101); H01F 2029/143 (20130101) |
Current International
Class: |
H01F
29/00 (20060101); H01F 29/14 (20060101); G05F
1/13 (20060101); G05F 3/04 (20060101); G05F
1/10 (20060101); G05F 3/06 (20060101); G05f
001/38 (); G05f 007/00 () |
Field of
Search: |
;323/6,48,50,56,57--61,8,81,85 ;321/57,68 |
References Cited
[Referenced By]
U.S. Patent Documents
Primary Examiner: Miller; J. D.
Assistant Examiner: Goldberg; Gerald
Claims
We claim:
1. A ferroresonant transformer including a magnetic core structure,
a primary winding and a secondary winding,
wherein the core structure with the secondary winding includes two
separate sections providing parallel magnetic paths for the
secondary and resonant flux;
a control winding on one of said sections; and
switch means for opening and closing a circuit across said control
winding, with no current source connected to said control winding
whereby the only current in said control winding is self induced
current resulting from flux in said one section.
2. 2. A ferroresonant transformer as defined in claim 1 wherein the
core structure includes an opening therethrough, with said two
sections forming opposite walls thereof.
3. A ferroresonant transformer as defined in claim 1 wherein said
core structure includes stacked E and I laminations, with said two
sections comprising the outer legs of the E laminations.
4. A ferroresonant transformer as defined in claim 1 wherein said
core structure includes at least two groups of stacked E and I
laminations, with said two sections comprising the center legs of
two different groups of said stacked laminations.
5. A ferroresonant transformer as defined in claim 1 wherein said
core structure is formed of a plurality of toroids, with said two
sections comprising separate toroids.
6. A ferroresonant transformer as defined in claim 1 wherein said
core structure includes three sets of stacked E and I laminations
assembled side by side, with the center set inverted with respect
to the outer sets, and with said two sections comprising the center
legs of the outer sets, respectively.
7. A ferroresonant transformer including a magnetic core structure,
a primary winding and a secondary winding,
wherein the core structure with the secondary winding includes two
separate sections providing parallel magnetic paths for the
secondary and resonant flux;
a control winding on one of said sections; and
switch means for opening and closing a circuit across said control
winding as a function of the secondary voltage, with no current
source connected to said control winding whereby the only current
in said control winding is self induced current resulting from flux
in said one section.
8. A ferroresonant transformer as defined in claim 7 wherein said
switch means includes a synchronized oscillator.
9. A ferroresonant transformer as defined in claim 7 wherein said
switch means includes a phase shifter.
10. A ferroresonant transformer as defined in claim 7 wherein said
switch means includes a saturable reactor.
11. A ferroresonant transformer as defined in claim 7 wherein said
switch means provides for switching operation every half-cycle of
the supply voltage.
12. A ferroresonant transformer as defined in claim 7 wherein said
switch means provides for switching operation every cycle of the
supply voltage.
13. A ferroresonant transformer as defined in claim 7 including
rectifier means connected across the secondary winding for
producing a DC control signal voltage; and
said switch means includes a switch element and a control circuit
having said control signal voltage as an input for closing said
switch element when said control signal voltage exceeds a
predetermined magnitude.
14. A ferroresonant transformer as defined in claim 13 in which
said control circuit includes an integrator connected for charging
from said control winding, and a trigger unit having the integrator
output and said control signal voltage as inputs with said trigger
unit actuating said switch element at least once each cycle of the
supply voltage connected to the primary winding.
15. A ferroresonant transformer as defined in claim 14 in which
said integrator comprises a resistor and capacitor circuit.
16. A ferroresonant transformer as defined in claim 14 in which
said integrator comprises an operational amplifier connected as an
integrator.
17. A ferroresonant transformer including a magnetic core
structure, a primary winding and a secondary winding,
wherein the core structure with the secondary winding includes two
separate sections providing parallel magnetic paths for the
secondary and resonant flux; and
means for varying the saturation flux capacity in the core
structure by switching the secondary and resonant flux between a
first condition with flux change occurring in both of said sections
and a second condition with substantially all of the flux change
occurring in one of said sections,
said means for varying including a control winding on the other of
said sections, and means for closing and opening a circuit across
said control winding, with no current source connected to said
control winding whereby the only current in said control winding is
self induced current resulting from flux in said other section.
Description
This invention relates to ferroresonant transformers and in
particular, to a new and improved ferroresonant transformer
incorporating means for varying the saturation flux capacity in the
core structure for controlling the transformer output.
Ferroresonant transformers are widely used today for a variety of
regulating and control purposes and the basic design is shown in
the U.S. Pat. to Sola, No. 2,143,745. A conventional ferroresonant
transformer utilizing toroid cores is shown in the U.S. Pat. to
Sola, No. 2,753,513. Various modifications and improvements are
shown in other patents to the same patentee. It is an object of the
present invention to provide a new and improved ferroresonant
transformer construction which may be utilized with any of the
presently known ferroresonant transformers and which provides for
saturation flux capacity variation and transformer output
control.
The conventional ferroresonant transformer has a magnetic core
structure with a primary winding, a secondary winding and a
resonant winding thereon. A capacitance is connected across the
resonant winding. The resonant winding may be a separate winding or
may actually be the secondary winding, with the capacitance
connected across the secondary winding. A leakage flux path is
provided in the core structure. In a transformer utilizing E and I
laminations for the core structure, a quantity of core material is
installed between the primary and secondary windings to provide a
magnetic shunt for the leakage flux path. In transformers utilizing
a toroid for the core structure, at least two toroids are utilized
with the primary winding linking all of the toroids and with the
secondary winding not linking all of the toroids. The ferroresonant
type of operation may also be achieved with a core structure having
two or more separate core units, and several such devices utilizing
a saturating transformer, without a shunt, and a series choke are
shown in the U.S. Pats. to Schmutz et al., No. 2,179,353, John et
al., No. 2,505,620, Buie No. 2,764,725, and Kohn No. 2,967,271. The
theory of operation and the details of construction of these
conventional regulating devices may be obtained from various prior
art publications, including the aforementioned patents. The term
"ferroresonant transformer" as used herein is intended to include
all such regulating devices.
In all ferroresonant transformers, the saturation flux capacity of
the secondary magnetic circuit is predominant in determining the
output voltage for a given number of secondary turns and a given
frequency.
Because of external drops such as rectifiers and wiring resistance
and because of other parameter changes such as frequency and
saturation drifts with temperature, means for controlling the
saturation flux capacity and hence the output voltage is desirable.
Attempts have been made to achieve such control in the past by
using windings with DC bias which affect the effective saturation
flux density of the core material. These techniques require a
sizeable mmf. in order to achieve control and the cost and
complexity of the required driving circuits offset the advantages
of using a ferroresonant approach.
The present invention provides for changing the saturation flux
capacity of the secondary magnetic circuit in a ferroresonant
transformer by isolating a portion of the secondary core material
and controlling the flux in this section. In the present invention
the secondary magnetic cross-sectional area is divided into two
sections, which may be referred to as an uncontrolled section and a
controlled section. Means are provided in the controlled section to
limit the maximum value of the instantaneous flux passing through
this section. Said means comprises a control winding encircling the
controlled section and a switch element for shorting or opening
said winding. By varying the time in the cycle at which the switch
is operated, the maximum instantaneous flux in the controlled
section can be varied from zero to a maximum value equal to the
saturation flux capacity of the controlled section. As used herein,
the term saturation flux capacity means the sum of (1) the product
of saturation flux density of the magnetic material multiplied by
the cross-sectional area of the core at the uncontrolled section
and (2) one-half the total change of flux in the core at the
controlled section.
Accordingly, it is an object of the invention to provide a new and
improved ferroresonant transformer with a core construction
incorporating two separate sections in the secondary path, with a
control winding on one of the sections, and switch means for
opening and closing a circuit across the control winding. A further
object is to provide transformers incorporating the invention for
operation at low frequencies and for operation at high frequencies.
An additional object is to provide such a transformer which may
incorporate the various features of conventional ferroresonant
transformers and which may be constructed utilizing present day
manufacturing techniques, including E and I laminations and
toroidal cores.
Other objects, advantages, features and results will more fully
appear in the course of the following description. The drawings
merely show and the description merely describes the preferred
embodiments of the present invention which are given by way of
illustration or example. Various configurations for the magnetic
core structure may be utilized including simple rectangular cores,
E and I lamination cores, toroids, C-cores, and separate choke and
transformer elements, and several specific forms are illustrated.
The switch means for shorting and opening the control winding
typically may include a switch element and a control circuit for
actuating the switch element. Various devices may be used as the
switch element, including a simple mechanical switch or relay, a
transistor, an SCR, a Triac, a thyratron, an ignitron, a gas
triode, a magnetic amplifier, and a saturable reactor. Various
circuitry arrangements may be used as the control circuit,
including a synchronized oscillator, a phase shifter, a magnetic
amplifier, and a saturable reactor. Several specific examples of
the switch means are set out herein. The switching operation may be
performed every cycle or every half-cycle, and circuits for both
modes are illustrated.
In the drawings:
FIG. 1 is a diagram of a ferroresonant transformer incorporating an
embodiment of the present invention;
FIG. 2 is a view of an alternative form of construction of a
ferroresonant transformer of the present invention suitable for use
at power frequencies;
FIG. 3 is a diagram indicating typical flux waveforms in the
controlled and uncontrolled sections and the total secondary flux
waveform;
FIG. 4 is a schematic diagram of a circuit providing a synchronized
oscillator with half-wave control for use with the transformers of
FIG. 2 and FIG. 11;
FIG. 5 is a view of an alternative form of construction of a
ferroresonant transformer incorporating another embodiment of the
present invention;
FIG. 6 is a view of another alternative form of construction of a
ferroresonant transformer incorporating a preferred embodiment of
the present invention for use at higher frequencies;
FIG. 7 is a schematic diagram of a circuit providing a phase
shifter control for use with the transformer of FIG. 6;
FIG. 8 is a block diagram illustrating the circuit of FIG. 7;
FIG. 9 is a schematic diagram of an alternate circuit providing a
synchronized oscillator with full-wave control for use with
transformers described herein;
FIG. 10 is a diagram of a ferroresonant transformer of this
invention used in conjunction with a saturable reactor to obtain DC
load compensation;
FIG. 11 is an isometric view of a form of construction of a
ferroresonant transformer incorporating a preferred embodiment of
the present invention suitable for use at power frequencies and for
frequencies up through the audio range;
FIG. 12 is a skeletonized view of the core of the transformer of
FIG. 11, indicating the placement of the various laminations;
FIG. 13 is a sectional view taken along the line 13-13 of FIG. 11;
and
FIGS. 14, 15 and 16 are sectional views taken along the lines
14-14, 15-15, and 16-16, respectively, of FIG. 13.
The transformer of FIG. 1 includes a magnetic core 10 which may be
constructed in the usual manner of interleaved or butt stacked
laminations. The various conventional transformer manufacturing
processes may be utilized. Typical core constructions for use with
E and I laminations are illustrated in FIGS. 2 and 11--16. Core
constructions for use with toroids are illustrated in FIGS. 5 and 6
and will be described later. Referring to FIG. 1, an input or
primary winding 11 is provided on a leg portion 12 of the core 10.
An output or secondary winding 13 is provided on another leg
portion 14. In the embodiment illustrated, the secondary winding
also serves as the resonant winding, and a capacitor 16 is
connected across the secondary winding 13. In an alternative
arrangement, a separate resonant winding could be provided on the
leg portion 14, with the capacitor 16 connected thereacross.
A leakage flux path is provided in the core 10 between the primary
winding 11 and the secondary winding 13, and comprises a magnetic
shunt of core sections 20, 21 with a nonmagnetic gap 22, usually an
air gap, therebetween. The elements described thus far are found in
the conventional ferroresonant transformers such as described in
the aforementioned Sola patents and reference may be made to the
prior art for a study of the theory of ferroresonance. A power
source is connected to the primary winding and a load is connected
to the secondary winding, and the transformer functions to maintain
the output voltage of the secondary winding constant within
predetermined limits for variations in load and variations in
voltage of the power source.
In the core of the invention, as illustrated in FIG. 1, the
secondary leg portion 14 is divided into two sections 23, 24, which
may be identified as the uncontrolled section 23 and the controlled
section 24. The uncontrolled section is designed to saturate. A
control winding 25 is provided on the section 24. A control circuit
26 operates a switch 27 connected across the control winding 25.
The control circuit serves to open and close the switch at a
particular phase angle of the output waveform. This phase angle is
a prescribed function of a controlling signal. If a closed loop
regulating device is desired, the controlling signal is derived
from the output voltage. While a variety of devices may be utilized
as the switch, contemporary solid state devices are presently
preferred, such as a silicon-controlled rectifier, and two examples
of control circuits and switches are described herein below.
When the switch 27 is open, the control winding 25 is open
circuited and the secondary and resonant flux varies in both the
uncontrolled section 23 and the controlled section 24. With the
switch 27 closed, the control winding 25 is short circuited and
further variation of flux in the controlled section is prevented.
Since the flux in the controlled section is maintained constant by
the action of switch 27, the total saturation flux capacity is
reduced as compared to the total saturation flux capacity when the
switch is closed for a minimum duration; the time required for the
capacitor ring-over cycle.
In order for the device to operate in the ferroresonant mode the
uncontrolled section must saturate. During periods when the mmf.
across the controlled section exceeds the mmf. required to maintain
the desired flux in this section the switch must be shorted or
closed. The maximum switch closure time is a half-cycle and this
condition corresponds to the the minimum total secondary saturation
flux capacity. Since increasing the switch closure time causes a
reduction in total secondary saturation flux capacity, a
corresponding reduction in the output voltage is obtained.
Therefore by opening or closing the circuit across the control
winding, the saturation flux capacity and hence the output voltage
may be changed.
Prior to closing the switch and shorting the control winding the
flux variation in the controlled section follows the instantaneous
secondary voltage. The magnitude of the flux rate of change in the
controlled section is determined by the instantaneous secondary
voltage and the ratio of the reluctances of the two sections. After
the switch in the control winding closes, and thereby shorts the
control winding, the flux in the controlled section remains
constant because any further changes in flux are prevented by the
induced mmf. in the short-circuited control winding.
As a consequence of shorting the control winding and forcing the
flux in the controlled section to remain constant, the rate of
change of flux in the uncontrolled section must then increase to a
level sufficient to maintain the voltage that existed in the
secondary winding at the time of switch closure.
The secondary voltage prior to and after switch closure at time
t.sub.2, (FIG. 3), must be identical because the resonant capacitor
across the secondary magnetic circuit prevents any instantaneous
change of voltage. The instantaneous variation of fluxes in the two
sections and the control action of the switch and control winding
is indicated in FIG. 3. In the drawing, t.sub.1 and t.sub.5 are the
times when the flux waveforms pass through zero, t.sub.2 is the
time when the switch is closed, and t.sub.4 is the time when the
switch is opened.
After the switch closes, the flux in the uncontrolled section
continues to increase until the magnetic material in the
uncontrolled section saturates. When this occurs, at time t.sub.3,
the secondary resonant winding reflects a low impedance to the
resonant capacitor and initiates a ring-over cycle that inverts the
voltage on the secondary and resonant circuits. As a result of the
inversion of the secondary voltage by the capacitor ring-over cycle
the rate of change of fluxes reverses and the flux in the
uncontrolled section begins to decrease. At time t.sub.4, when the
flux in the uncontrolled section decreases to a value approximately
equal to the magnitude of flux that existed at the time of switch
closure, t.sub.2, the switch is opened. With the switch open the
flux in both sections can now change. The total rate of change of
flux in both sections is again determined by the instantaneous
value of the secondary voltage, and the flux division between the
two sections is again given by the ratio of the reluctances. After
switch opening the magnitude of the rate of change of flux in the
uncontrolled section decreases since the rate of change of total
flux before and after switch opening must be identical.
The process then repeats itself in the following half-cycle with
the polarity of fluxes and voltages being reversed.
The peak value of the total flux is equal to the peak value of the
flux in the controlled section plus the peak value of flux in the
uncontrolled section.
By varying the time of switch closure the peak flux in the
controlled section is varied, since the flux in this section
increases up the time the switch is closed. If the switch is closed
at the time of flux zero crossing the peak flux in the controlled
section will be approximately zero. If the switch is closed at the
time the secondary flux waveform has a maximum, the peak flux in
the controlled section will be a maximum and will be equal to the
saturation flux capacity of the controlled section.
The peak value of the flux in the uncontrolled section is not
affected by varying the switch closing point since the uncontrolled
section is driven into saturation each half-cycle.
As a consequence, the peak value of the total secondary flux will
vary as the peak value of the controlled flux.
Since the average secondary output voltage is directly proportional
to the peak value of the total flux, the magnitude of the average
secondary output voltage can be varied by varying the length of the
interval when the switch is closed.
Since the device of this invention is operating in the
ferroresonant mode and hence the secondary voltage does not ring
over until the uncontrolled section saturates, the shorting or
opening of the control winding does not introduce a discontinuity
in the output voltage waveform. The only effect of shorting the
control winding is to cause a change in the amplitude of the
waveform over the entire half-cycle.
For operation of the transformer as a voltage regulator, a smooth
control between the two conditions is desirable. This may be
achieved by shorting the control winding for a certain portion of
each cycle or each half-cycle of the primary supply frequency. By
varying the duty cycle between open circuit and closed circuit
conditions, the output may be smoothly adjusted from minimum to
maximum voltage. Suitable circuits for effecting this control are
illustrated in FIGS. 4 and 7 and 9. FIGS. 4 and 7 are half-wave
control circuits and operate once per cycle. FIG. 9 is a full wave
control that closes the switch each half-cycle.
The ferroresonant transformer of the invention circulates sizeable
mmf. in the control winding. One advantage of the transformer of
the present invention is that this mmf. is generated by the
transformer in the form of induced current and the control circuit
must merely contain a switch capable of carrying this current. No
separate power supply is required to supply the mmf. in the control
winding.
When the controllable ferroresonant transformer of the invention is
used as a closed loop regulator, a method for determining when in
the cycle to close the control winding is desired. Most existing
phase control methods are troubled with the fact that the
instantaneous phase angle of the ferroresonant transformer varies
with load changes and varies during dynamic voltage variations in
the output. Such phase variations can cause severe instabilities in
a phase controlled loop.
A more satisfactory mode of control is the type embodied in the
circuit of FIG. 7. In this circuit the control winding voltage is
integrated with respect to time and the switch across the control
winding is closed when this integrated value reaches the required
level. This level is determined by comparing a signal proportional
to the output voltage with a reference voltage and amplifying the
error signal. By varying the proportionality relationship between
the output voltage and the reference voltage, the output voltage
amplitude can be adjusted, as by resistors 109 and 110. The
integrator may be reset at the time the short circuit is applied to
the control winding so that the integrator will be ready for the
next half-cycle.
In some applications, it is only necessary to short the control
winding once every cycle instead of every half-cycle because the
magnetic core will automatically saturate at the desired time
during the other half-cycle. This occurs when a core with square
loop material is used in the controlled section of the transformer.
Under these conditions, the integrator of the control circuit may
be reset at any time between the application of the short circuit
and the beginning of the next cycle. A variety of integrators may
be utilized, such as a resistor and capacitor circuit, a Miller
integrator, an operational amplifier connected as an integrator, a
magnetic amplifier, and the like.
Turning now to the embodiment of FIGS. 2 and 4, a core 30 is formed
of butt stacked E and I laminations 31, 32, respectively. A primary
winding 33 is positioned about the center leg 34 of the stack of E
laminations. A shunt 35 is positioned between the center leg 34 and
the outer leg 36 and a similar shunt 37 is positioned between the
center leg 34 and the outer leg 38. These shunts may be
conventional in construction and typically each comprises a stack
of laminations.
A secondary winding 41 is positioned about the center leg 34 and a
control winding 42 is positioned about portion 44 of the outer leg
38 so that the portion 44 functions as the controlled section of
the core and the portion 43 of the outer leg 36 functions as the
uncontrolled or saturating section of the core. In the embodiment
illustrated, the leg 36 is reduced in cross-sectional area at 43
for the purpose of assuring saturation in this section and the leg
38 is reduced in cross-sectional area at 44 for the purpose of
minimizing the induced voltage in the control winding.
In the diagram of FIG. 4, the resonant capacitor 48 is connected
across the secondary winding 41. The AC output voltage on the
secondary 41 is converted to a DC control signal voltage at points
49, 50 by a full-wave rectifier 51 and a filter capacitor 52. A
silicon-control rectifier 53 is connected across the control
winding 42 and functions as the switch for opening and closing the
circuit across the control winding. An integrator comprising a
resistor 54 and a capacitor 55 is connected across the points 49,
50. A resistor 56 is connected between the point 49 and the control
winding for the purpose of supplying holding current to SCR 53. A
trigger diode 57 is connected between the junction point 58 of the
integrator and the control element of the silicon-control rectifier
53. A resistor 59 is connector from the silicon-control rectifier
control element to the point 50 to provide a load to ground for the
trigger diode. The trigger diode 57 passes substantially zero
current until voltage thereacross builds up to a given level, after
which the diode conducts with a relatively low impedance. A typical
diode would be an MPT 28.
In operation, the control rectifier 53 and the diode 57 are
initially not conducting. As the output voltage of the transformer
builds up, the capacitor 55 is charged through the resistor 54.
When the voltage level at point 58 reaches a predetermined value,
the diode 57 conducts, discharging the capacitor 55 into the
control rectifier 53 and turning the control rectifier on. The
control rectifier remains conducting until the end of the
half-cycle when it automatically turns off. The operation is
repeated for every alternate half-cycle.
The resistor 54, the capacitor 55 and the diode 57 function as a
relaxation oscillator which normally runs in synchronism with the
line frequency under equilibrium conditions. If the output voltage
of the transformer increases, the DC voltage at points 49, 50
increases and the frequency of the relaxation oscillator increases,
resulting in an earlier shorting of the control winding 42 and a
reduction in the output of the transformer which in turn causes the
oscillator to return to synchronism. Similarly, a reduction in the
transformer output causes a reduction in frequency of the
oscillator and a later closing of the switch 53 with a subsequent
increase in transformer output voltage.
A preferred embodiment of this invention is illustrated in FIGS.
11--16, and may be used with the control circuit of FIG. 9. FIG. 11
is an overall view of the complete transformer assembly which
includes a magnetic core 300, a primary coil 301, a secondary coil
assembly 302, shunts 305 and 306 and gaps 303 and 304 (FIG. 15).
The secondary coil assembly includes a resonant winding 307, a
secondary winding 308 and control winding 309 (FIGS. 13 and 16).
The secondary magnetic circuit includes a controlled section 310,
and uncontrolled sections made up of 311 and 312 (FIG. 16). To
assure saturation in the uncontrolled section 311 and 312, a window
315 is cut in the tongue of the uncontrolled section (FIGS. 12, 13
and 16). This window 315 reduces the cross-sectional area of the
uncontrolled secondary magnetic circuit relative to the
cross-sectional area of the outer legs 314 and 313 and the total
primary core cross-sectional area of core tongues 316 and 316a and
310.
The control winding 309 encircles the controlled section 310. The
resonant winding 307 and the secondary winding 308 encircle both
the controlled core section 310 and the uncontrolled core section
311 and 312. The control winding 309 is encircled by secondary coil
308 and the resonant coil 307 encircles both the secondary winding
308 and the control winding 309 (FIG. 16).
FIG. 15 is a view through the magnetic shunt structure. Shunts 305
and 306 appear in plan view and have an L-shape to provide magnetic
flux shunting of the primary flux existing in the primary magnetic
circuit 316 and 316a and the controlled section core tongue 310.
The primary coil 301 encircles both core tongues 310 and 316, 316a.
If the cross-sectional area of tongue 310 is small relative to the
cross-sectional area 316, 316a, then the shunts 306 and 305 may be
straight sections that only extend the length of the primary core
tongue 316, 316a.
FIG. 14 is a view showing the primary coil structure. The primary
coil 301 encircles both the controlled section core tongue 310 and
the primary core tongue 316, 316a.
The core 300 of the transformer structure is assembled from
modified standard E and I laminations. The laminations in stack 320
are only encircled by the resonant winding 307 and secondary 308
winding and consist of standard E's and I's with a window 315 cut
at the back of the E. In the stack 321, space for the control
winding 309 is obtained by cutting off a portion of the tongue of a
standard E lamination. The stack 322 is made from the same standard
E and I laminations with the center leg 310 of the E reduced in
width to receive the control winding 309.
A particularly simple form of full-wave control for the transformer
of FIGS. 11--16 is shown in FIG. 9. The circuit of FIG. 9 utilizes
a Triac for the switch element. Since the Triac operates each
half-cycle, the nominal frequency of the relaxation oscillator must
be twice the transformer operating frequency.
Diodes 200, 201, 202, and 203 form a rectifier bridge which
rectifies the output of the transformer. This rectified AC is
applied to resistor 204 and the anode of rectifier 205. The cathode
of rectifier 205 is connected to capacitor 206. This capacitor
filters the rectified signal to DC. This DC in turn supplies
current through resistor 208 to the Zener diode 211 which results
in a constant potential appearing on the cathode of the Zener 211.
The DC appearing on capacitor 206 also causes resistor 207 to
supply current to capacitor 209 which charges until the firing
point of the unijunction 212 is reached. At this point, the
unijunction switches to a conducting state which causes the charge
on capacitor 209 to be delivered to the control electrode of the
Triac 213 which in turn causes the Triac to conduct, shorting the
control winding.
The function of resistor 204 and Zener 210 is to supply a
synchronizing pulse to the B2 electrode of unijunction 212. This is
accomplished as follows. The rectified voltage out of the bridge is
not filtered because rectifier 205 isolates this point from the
filter capacitor 206. As a result, when the AC voltage into the
bridge crosses through zero, the voltage out of the bridge drops to
zero which stops the current flow in resistor 204. At this time,
the voltage on the B2 electrode of the unijunction 212 is given by
the Zener voltage of Zener 211 minus the Zener voltage of Zener
210. During the remainder of the half-cycle, when the voltage is
high, current flows in resistor 204 which causes the Zener 204 to
become forward biased. The voltage on the B2 electrode of
unijunction 212 is then given by the Zener voltage of Zener 211
plus the forward voltage of Zener 210. This latter voltage appears
during most of the half-cycle but during the zero-crossing period
of the AC, the voltage momentarily drops to the former level
resulting in a negative pulse on the B2 electrode of unijunction
212.
Since the firing point of the E electrode of unijunction 212 is a
fixed percentage of the voltage on the B2 electrode, reducing the
B2 electrode potential reduces the firing point of the E electrode.
This will cause the unijunction to fire at this time if capacitor
209 is charged to a potential near the firing level when the
voltage on the B2 electrode is at its high level.
In operation, resistor 207 is adjusted such that capacitor 209 will
charge to the firing level of the E electrode of unijunction 212 in
one-half cycle of the AC period when the DC voltage on capacitor
206 is at the desired potential. If the voltage is too high, the
capacitor 209 charges faster, causing the firing angle of
unijunction 212 and Triac 213 to advance which in turn reduces the
voltage on capacitor 206 until equilibrium is established with the
voltage on capacitor 206 at the proper potential for capacitor 209
to charge to the firing level in one-half cycle. Likewise, if the
voltage on capacitor 206 is too low, the charging time of capacitor
209 lengthens which retards the firing angle until equilibrium is
again established as before.
In the event the transformer is unable to supply the desired
voltage, the firing angle is delayed until the following zero
crossing of the AC waveform is reached. At this point the
previously mentioned synchronizing pulse causes the unijunction to
fire which prevents any further delay in the firing angle and the
resultant loss of synchronization. A ferroresonant transformer
constructed as illustrated in FIGS. 11--16 and operated with the
circuit of FIG. 9 at 60 Hz. as a voltage regulator provided
substantially constant output voltage (i.e., .+-.0.3-volts
variation at 166.6-volts output) over the range of no load to full
load for input voltage variations in the range of 100 to 135
volts.
FIGS. 5 and 6 illustrate alternative constructions for the
transformer of the invention particularly suitable for operation at
higher frequencies, such as 20 kHz., and utilizing a plurality of
toroids for the magnetic core structure. In FIG. 5, a primary
winding 65 is linked through toroids 66, 67 and 68. A secondary
winding 69 is linked through the toroids 67 and 68. A control
winding 70 is linked through the toroid 68. The resonant capacitor
71 is connected across the secondary winding, and the control
circuit 72 provides for closing the switch 73 across the control
winding. The cores 67, 68, comprise the secondary portion of the
core structure, with the core 67 corresponding to the uncontrolled
section 23 of FIG. 1 and with the core 68 corresponding to the
controlled section 24 of FIG. 1. The operation of the transformer
will be the same as described in conjunction with the transformer
of FIG. 1 and the transformer of FIGS. 2--4.
FIG. 6 illustrates an alternative form for the transformer of FIG.
5. The toroid 67 is replaced by two toroids 67a, 67b to provide the
desired amount of magnetic material. A compensation winding 74 may
be provided on one of the toroids, here the toroid 66. The
compensation winding is used in the same manner as in the
conventional ferroresonant transformer as described in some of the
aforementioned patents. The toroids of FIGS. 5 and 6 may be formed
as unitary molded elements, or pairs of C-cores, or wound ribbons,
or in other forms as desired.
FIG. 7 illustrates the use of a ferroresonant transformer as shown
in FIGS. 5 and 6, as the output transformer in a transistor
inverter circuit. A typical inverter utilizes a pair of transistors
connected in series opposition across a primary winding of an
output transformer, with the AC output appearing at a secondary or
load winding of the transformer. The DC power source is connected
to the junction point of the transistors and to a center tap of the
primary winding, and a drive circuit provides a drive current to
the transistors for turning the transistors off and on. The drive
circuit may be energized from a winding on the transformer
providing a self oscillating inverter or the drive circuit may be
energized from an external source. The two transistors operate as
switches with first one and then the other being closed to connect
the DC source current through the transformer primary winding
alternating in the positive and negative directions to provide the
AC voltage on the transformer. This basic inverter circuit is well
known and several variations thereof are shown in U.S. Letters Pat.
Nos. 2,990, 519; 3,256,495; 3,317,856; and 3,405,342.
The output of an inverter 80 is connected to the primary winding
65. The secondary winding 69 is connected to a full-wave rectifier
comprising diodes 81, 82 and a filter comprising a capacitor 83 and
an inductance 84, to provide a DC output at the terminals 85, 86.
The resonant capacitor 71 is connected across a portion of the
secondary winding 69.
The DC output voltage is connected to a control circuit 90 via
lines 91, 92. The control circuit 90 includes an integrated circuit
component 94 which provides a reference voltage and a DC amplifier,
a threshold or trigger device in the form of an unijunction
transistor 95, a switch in the form of a silicon-controlled
rectifier 96, and an integrator comprising a resistor 97 and a
capacitor 98.
The function of the control circuit is to vary the area under the
voltage vs. time waveform of the control winding 70. The reset
period occurs in the following half-cycle and will exhibit the same
area as the controlled area because a magnetic device will not
support a DC unbalance. As a result, it is only necessary to
control during one-half cycle out of every cycle. Since the prime
objective is to control the area, a means of controlling the firing
angle as a direct function of area is used in this control circuit.
The basic operation of the circuit is shown in the block diagram of
FIG. 8.
The voltage on the control winding 70 is integrated with time by
the integrator 97, 98. Since the value of the integral is
proportional to the area to be controlled, it is only necessary to
close the switch 96 when the integral reaches a predetermined
level. This triggering function is accomplished by the threshold
device 95 which supplies a trigger or drive signal to the switch 96
when the integral exceeds the preset threshold value. In order to
obtain control however, it is necessary to vary the area and hence
the threshold value. The threshold device has another input which
controls the threshold level in proportion to the voltage applied
to that input. To provide a control loop, the voltage applied to
the other input of the threshold device is derived by comparing the
output voltage of the transformer at terminals 85, 86 with a
reference voltage and amplifying the resulting error signal with a
DC amplifier 94.
One circuit arrangement suitable for use as the control circuit 90
is illustrated in FIG. 7. A diode 100 is connected in series with
the rectifier switch 96 to enhance the reverse blocking of the
switch. Diode 101 is connected across the capacitor 94 of the
integrator to limit the swing during the reset period of the
control winding to essentially zero so that the integrator will
start at zero for the next control interval.
When the voltage on the electrode E of the unijunction transistor
95 reaches a certain percentage of the voltage on the electrode B2,
usually 60 to 80 percent, the device switches from essentially an
open circuit between the electrodes E and B1 to a very low
impedance. This action causes the charge on capacitor 98 to be
delivered to the gate or control electrode of the rectifier 96,
firing the rectifier into conduction. The impedance between
electrode B2 and electrode B1 is substantially reduced during this
period and a resistor 102 is included as a current limiting
resistor.
The integrated circuit 94 contains a voltage regulator which
supplies a fixed reference voltage and contains a DC amplifier. A
typical integrated circuit may be a Fairchild .mu. A723C. Terminals
7 and 8 are the positive input terminals, terminal 5 is the
negative terminal, terminal 4 is the reference voltage output and
is connected to terminal 3 which is the noninverting or reference
input for the DC amplifier. Terminal 4 is used as the return point
for diode 101, capacitor 98, diode 103, and rectifier 96, enabling
a reverse bias to be developed on the gate of rectifier 96 and the
leakage current of transistor 95 to be bled off by resistor 104.
Diode 103 serves to limit the reverse bias on the rectifier 96.
Terminal 6 of the integrated circuit 94 is the DC amplifier output
and provides the varying control signal voltage for the electrode
B2 of the threshold device 95. Resistor 108 functions to prevent
the voltage on electrode B2 from becoming so low that the charge
from the capacitor 98 is insufficient to fire the rectifier 96.
Terminal 2 of the integrated circuit 94 is the inverting input to
the DC amplifier and resistors 109, 110 form a voltage divider
which divides the output voltage down to an appropriate level for
the amplifier. Terminal 9 is a terminal provided for frequency
compensation of the DC amplifier to prevent high frequency
instability. Capacitor 111 provides this compensation. Additional
frequency compensation is provided by resistor 112, capacitor 113,
resistor 114, and capacitor 115. Capacitor 116 serves to prevent
noise from disturbing the reference voltage on terminal 4.
A ferroresonant transformer connected as illustrated in FIGS. 6--8
operated at 20 kHz. as a voltage regulator provided substantially
constant output voltage (i.e., 0.06-volts variation at 30.03-volts
output) at no load and at full load for input voltage variations in
the range of 20 to 36 volts for a 28-volt rated input.
Thus, it is seen that the objects of the invention are achieved in
providing very close control of a ferroresonant transformer
utilizing flux switching and without requiring separate power
supplies.
The control winding can be used in other ways. By way of example, a
mechanical switch may be used to change the output voltage or the
operating frequency for a fixed output voltage. By connecting a
suitable frequency sensitive impedance, such as a series resonant
circuit, across the control winding, frequency compensation can be
obtained. The ferroresonant transformer of the invention can be
used with any device having an impedance that is dependent on load,
line, frequency or other desired parameter, in conjunction with the
control winding to obtain close loop control or open loop
compensation.
FIG. 10 illustrates the utilization of the transformer of this
invention in combination with a saturable reactor to obtain DC load
compensation.
The circuit of FIG. 10 utilizes the transformer of FIG. 1 and
corresponding elements are identified by the same reference
numerals. A full-wave rectifier 225 is connected across the
secondary and resonant winding 13. The rectified output from the
rectifier 224 is connected through one winding of a saturable
reactor 226 to the load 227. The other winding of the saturable
reactor 226 is connected across the winding 25.
The saturable reactor 226 offers a low impedance to the control
winding 25 when it is saturated and a high impedance during
unsaturated periods. Essentially, the reactor functions as a
switch. To obtain load compensation, the reactor windings are
interconnected in a manner that will cause the conduction period of
the saturable reactor to be a maximum at no load and a minimum at
full load. With this configuration of elements it is possible to
obtain a variety of load compensation characteristics.
* * * * *