U.S. patent number 10,476,454 [Application Number 16/280,180] was granted by the patent office on 2019-11-12 for power amplifier module.
This patent grant is currently assigned to MURATA MANUFACTURING CO., LTD.. The grantee listed for this patent is Murata Manufacturing Co., Ltd.. Invention is credited to Yuri Honda, Shota Ishihara, Kazuhiro Koshio, Fuminori Morisawa, Shizuki Nakajima, Seiko Ono, Masato Sato, Yusuke Shimamune.
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United States Patent |
10,476,454 |
Ishihara , et al. |
November 12, 2019 |
**Please see images for:
( Certificate of Correction ) ** |
Power amplifier module
Abstract
A power amplifier module includes a first current source that
outputs a first current corresponding to a level control voltage
for controlling a signal level of an amplified signal, a second
current source that outputs a second current corresponding to the
level control voltage, a first transistor in which an input signal
and a first bias current are supplied to a base and an emitter is
grounded, a second transistor in which an emitter is connected to a
collector of the first transistor, the second current is supplied
to a base, and a first amplified signal obtained by amplifying the
input signal is output from a collector, and a third transistor in
which the first current is supplied to a collector, a bias control
current or voltage is supplied to a base, and the first bias
current is supplied from an emitter to the base of the first
transistor.
Inventors: |
Ishihara; Shota (Kyoto,
JP), Ono; Seiko (Kyoto, JP), Shimamune;
Yusuke (Kyoto, JP), Morisawa; Fuminori (Kyoto,
JP), Nakajima; Shizuki (Kyoto, JP), Honda;
Yuri (Kyoto, JP), Koshio; Kazuhiro (Kyoto,
JP), Sato; Masato (Kyoto, JP) |
Applicant: |
Name |
City |
State |
Country |
Type |
Murata Manufacturing Co., Ltd. |
Kyoto |
N/A |
JP |
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Assignee: |
MURATA MANUFACTURING CO., LTD.
(Kyoto, JP)
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Family
ID: |
66696499 |
Appl.
No.: |
16/280,180 |
Filed: |
February 20, 2019 |
Prior Publication Data
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Document
Identifier |
Publication Date |
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US 20190181816 A1 |
Jun 13, 2019 |
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Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
Issue Date |
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16013039 |
Jun 20, 2018 |
10256778 |
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15710426 |
Aug 7, 2018 |
10044330 |
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Foreign Application Priority Data
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Sep 21, 2016 [JP] |
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2016-184752 |
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Current U.S.
Class: |
1/1 |
Current CPC
Class: |
H03F
3/21 (20130101); H03F 1/0261 (20130101); H03F
3/245 (20130101); H03F 1/34 (20130101); H03F
3/191 (20130101); H03F 1/22 (20130101); H03F
2200/408 (20130101); H03F 2200/451 (20130101); H03F
2200/18 (20130101); H03F 2200/318 (20130101); H03F
2200/144 (20130101) |
Current International
Class: |
H03F
3/21 (20060101); H03F 1/22 (20060101); H03F
1/34 (20060101); H03F 3/191 (20060101); H03F
3/24 (20060101); H03F 1/02 (20060101) |
References Cited
[Referenced By]
U.S. Patent Documents
Foreign Patent Documents
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105009447 |
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Oct 2015 |
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CN |
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2009-100197 |
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May 2009 |
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JP |
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2014-068358 |
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Apr 2014 |
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JP |
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I508432 |
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Nov 2015 |
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TW |
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Primary Examiner: Nguyen; Patricia T
Attorney, Agent or Firm: Pearne & Gordon LLP
Parent Case Text
This application is a continuation-in-part of U.S. patent
application Ser. No. 16/013,039 filed on Jun. 20, 2018, which is a
continuation from Ser. No. 15/710,426 filed on Sep. 20, 2017, which
claims priority from Japanese Patent Application No. 2016-184752
filed on Sep. 21, 2016. The content of these applications are
incorporated herein by reference in its entirety.
Claims
What is claimed is:
1. A power amplifier module comprising: a first current source that
outputs a first current; a first transistor, wherein an input
signal and a first bias current are supplied to a base of the first
transistor; a second transistor, wherein a first amplified signal
obtained by amplifying the input signal is output from a collector
of the second transistor; a third transistor, wherein the first
current is supplied to the collector of the third transistor, a
bias control current or voltage is supplied to a base of the third
transistor, and the first bias current is supplied from an emitter
of the third transistor to the base of the first transistor; and an
impedance element connected between a collector of the third
transistor and the first current source.
2. The power amplifier module according to claim 1, wherein the
impedance element comprises a resistor.
3. The power amplifier module according to claim 1, wherein the
impedance element comprises a diode.
4. The power amplifier module according to claim 3, wherein a
cathode of the diode is connected to the collector of the third
transistor, and an anode of the diode is connected to the first
current source.
5. The power amplifier module according to claim 1, further
comprising: a second current source that outputs a second current;
and a second transistor, wherein an emitter of the second
transistor is connected to a collector of the first transistor, the
second current is supplied to a base of the second transistor, and
a first amplified signal obtained by amplifying the input signal is
output from a collector of the second transistor, wherein the first
current is greater than the second current.
6. The power amplifier module according to claim 1, further
comprising: a fourth transistor, wherein a power supply voltage is
supplied to a collector of the fourth transistor, the first
amplified signal and a second bias current are supplied to a base
of the fourth transistor, an emitter of the fourth transistor is
grounded, and a second amplified signal obtained by amplifying the
first amplified signal is output from the collector of the fourth
transistor; and a fifth transistor, wherein the first current is
supplied to a collector of the fifth transistor, the bias control
current or voltage is supplied to a base of the fifth transistor,
and the second bias current is supplied from an emitter of the
fifth transistor to the base of the fourth transistor.
7. The power amplifier module according to claim 1, further
comprising: a second transistor, wherein an emitter of the second
transistor is connected to a collector of the first transistor, and
a first amplified signal obtained by amplifying the input signal is
output from a collector of the second transistor; and a negative
feedback circuit disposed between the collector of the second
transistor and the base of the first transistor.
8. The power amplifier module according to claim 7, wherein the
negative feedback circuit comprises a first resistor and a
capacitor, the first resistor and the capacitor being connected in
series.
9. The power amplifier module according to claim 1, further
comprising a second resistor connected between the collector of the
third transistor and ground.
10. The power amplifier module according to claim 1, wherein the
first current corresponds to a level control voltage for
controlling a signal level of the first amplified signal.
Description
BACKGROUND
The present disclosure relates to a power amplifier module.
A mobile communication device such as a cellular phone includes a
power amplifier module that is used to amplify the power of a radio
frequency (RF) signal to be transmitted to a base station. For
example, in GSM (registered trademark) (Global System for Mobile
Communications), the gain of a power amplifier module is controlled
to implement slope control (ramp-up and ramp-down) of a
transmission signal from a mobile communication device to a base
station.
Japanese Unexamined Patent Application Publication No. 2009-100197
discloses a configuration for supplying a voltage V.sub.LDO output
from a low dropout (LDO) regulator to a collector terminal of each
of three-stage amplifiers in a power amplifier module. In the
disclosed configuration, the level of the voltage V.sub.LDO is
adjusted on the basis of a level control voltage Vramp to control
the gain of the power amplifier module.
U.S. Pat. No. 7,605,651 discloses a configuration of a power
amplifier module that includes three-stage amplifiers, in which a
voltage V.sub.REG output from an LDO regulator is supplied to the
first- and second-stage amplifiers and a constant power supply
voltage is supplied to the three-stage amplifier.
As described above, in the configurations disclosed in Japanese
Unexamined Patent Application Publication No. 2009-100197 and U.S.
Pat. No. 7,605,651, an LDO regulator is used to control the gain of
a power amplifier module. In general, an LDO regulator leads to a
large circuit size because of large current flow.
BRIEF SUMMARY
Accordingly, the present disclosure provides a power amplifier
module for performing slope control of a transmission signal, in
which an increase in circuit size is prevented or reduced.
According to embodiments of the present disclosure, a power
amplifier module includes a first current source that outputs a
first current corresponding to a level control voltage for
controlling a signal level of an amplified signal, a second current
source that outputs a second current corresponding to the level
control voltage, a first transistor having a collector, a base, and
an emitter, wherein an input signal and a first bias current are
supplied to the base and the emitter is grounded, a second
transistor having a collector, a base, and an emitter, wherein the
emitter is connected to the collector of the first transistor, the
second current is supplied to the base, and a first amplified
signal obtained by amplifying the input signal is output from the
collector, and a third transistor having a collector, a base, and
an emitter, wherein the first current is supplied to the collector,
a bias control current or voltage is supplied to the base, and the
first bias current is supplied from the emitter to the base of the
first transistor.
According to embodiments of the present disclosure, in a power
amplifier module for performing slope control of a transmission
signal, it may be possible to prevent or reduce an increase in
circuit size and it may also be possible to satisfy desired
characteristics.
Other features, elements, characteristics and advantages of the
present disclosure will become more apparent from the following
detailed description of embodiments of the present disclosure with
reference to the attached drawings.
BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS
FIG. 1 is a diagram illustrating an example configuration of a
transmitting unit that includes a power amplifier module according
to an embodiment of the present disclosure;
FIG. 2 is a diagram illustrating an example of slope control of a
transmission signal;
FIG. 3 is a diagram illustrating an example of a voltage (level
control voltage) input to the power amplifier module;
FIG. 4 is a diagram illustrating the configuration of a power
amplifier module that is an example of the power amplifier module
illustrated in FIG. 1;
FIG. 5 is a diagram illustrating an example configuration of an
amplifier;
FIG. 6 is a diagram illustrating an example configuration of a bias
circuit;
FIG. 7 is a diagram illustrating an example configuration of a
voltage-current conversion circuit and current sources;
FIG. 8 is a graph illustrating an example relationship between the
voltage (level control voltage) and currents;
FIG. 9A is a schematic diagram of an ideal output waveform of an RF
signal;
FIG. 9B is a schematic diagram of an output waveform of an RF
signal in a comparative example;
FIG. 9C is a schematic diagram of an output waveform of an RF
signal in the amplifier;
FIG. 10 is a graph illustrating results of simulation of adjacent
channel leakage ratio (ACLR) characteristics in the amplifier and
the comparative example;
FIG. 11 is a diagram illustrating the configuration of a power
amplifier module that is another example of the power amplifier
module illustrated in FIG. 1; and
FIG. 12 is a diagram illustrating the configuration of a power
amplifier module that is still another example of the power
amplifier module illustrated in FIG. 1.
DETAILED DESCRIPTION
An embodiment of the present disclosure will be described
hereinafter with reference to the drawings. FIG. 1 is a diagram
illustrating an example configuration of a transmitting unit that
includes a power amplifier module according to an embodiment of the
present disclosure. A transmitting unit 100 is included in a mobile
communication device such as a cellular phone, for example, and is
used to transmit various signals such as audio and data signals to
a base station. The mobile communication device also includes a
receiving unit to receive a signal from the base station, which is
not described herein.
As illustrated in FIG. 1, the transmitting unit 100 includes a
modulation unit 110, a power amplifier module 120, a front end unit
130, and an antenna 140.
The modulation unit 110 modulates an input signal in accordance
with the GSM (registered trademark) or Enhanced Data Rates for GSM
Evolution (EDGE) modulation scheme and generates a radio frequency
(RF) signal for wireless transmission. The RF signal lies in a
range of about several hundreds of megahertz (MHz) to about several
gigahertz (GHz), for example.
The power amplifier module 120 amplifies the power of an input
signal RFin to a level necessary for transmission to a base station
and outputs an amplified signal RFout. Further, the power amplifier
module 120 performs slope control of the amplified signal by
controlling the gain on the basis of a voltage Vramp (level control
voltage) for controlling a signal level.
FIG. 2 is a diagram illustrating an example of the slope control of
a transmission signal. In the slope control, as illustrated in FIG.
2, it is necessary to control the signal level of a transmission
signal so that the signal level falls within a range between a
lower limit DL and an upper limit UL. In a portion (an area
including a rising edge) represented by A in FIG. 2, the signal
level of the transmission signal needs to be controlled so that a
predetermined rate of change (slope) is maintained in such a manner
that the signal level does not exceed the upper limit UL or fall
below the lower limit DL. In a portion (an area including a falling
edge) represented by B in FIG. 2, the signal level of the
transmission signal needs to be controlled so that a predetermined
rate of change is maintained in such a manner that the signal level
does not exceed the upper limit UL or fall below the lower limit
DL.
FIG. 3 is a diagram illustrating an example of the voltage Vramp
input to the power amplifier module 120. In the power amplifier
module 120, the gain is controlled on the basis of the voltage
Vramp, which changes in the manner illustrated in FIG. 3, to
control the signal level of the transmission signal in the manner
illustrated in FIG. 2.
Referring back to FIG. 1, the front end unit 130 performs
operations such as filtering the amplified signal RFout and
switching between the amplified signal RFout and a reception signal
that is received from the base station. The amplified signal output
from the front end unit 130 is transmitted to the base station via
the antenna 140.
FIG. 4 is a diagram illustrating the configuration of a power
amplifier module 120A-1. The power amplifier module 120A-1 is an
example of the power amplifier module 120. The power amplifier
module 120A-1 includes amplifiers 400, 401, and 402, inductors 410,
411, and 412, matching networks (MNs) 420, 421, 422, and 423, bias
circuits 430, 431, and 432, a voltage-current conversion circuit
440, and current sources 450, 451, and 452.
The amplifiers 400 to 402 constitute three-stage amplifiers. The
amplifier 400 amplifies an input signal RFin and outputs an
amplified signal (first amplified signal). The amplifier 401
amplifies the amplified signal (first amplified signal) output from
the amplifier 400 and outputs an amplified signal (second amplified
signal). The amplifier 402 amplifies the amplified signal (second
amplified signal) output from the amplifier 401 and outputs an
amplified signal. A constant power supply voltage Vcc is supplied
to the amplifier 400. A bias current Ibias1 (first bias current) is
also supplied to the amplifier 400 from the bias circuit 430.
Likewise, the power supply voltage Vcc and a bias current Ibias2
(second bias current) are supplied to the amplifier 401. The power
supply voltage Vcc and a bias current Ibias3 are supplied to the
amplifier 402. The number of stages of amplifiers is not limited to
three; the number of stages of amplifiers may be two or less or may
be four or more.
FIG. 5 is a diagram illustrating an example configuration of the
amplifier 400. As illustrated in FIG. 5, the amplifier 400 in the
initial stage includes a transistor 500 (first transistor) and a
transistor 501 (second transistor), and the transistor 501 is
connected in cascode with the transistor 500. The transistors 500
and 501 are each a transistor such as a heterojunction bipolar
transistor (HBT). Specifically, the lower-stage transistor 500 has
a collector connected to an emitter of the transistor 501, a base
to which the input signal RFin is supplied, and an emitter
grounded. The upper-stage transistor 501 has a collector to which
the power supply voltage Vcc is supplied via the inductor 410, a
base to which a current Icas corresponding to the voltage Vramp is
supplied, and an emitter connected to the collector of the
transistor 500. Thus, the amount of current flowing through the
lower-stage transistor 500 is limited by the amount of current
flowing through the upper-stage transistor 501. Further, the
collector voltage of the lower-stage transistor 500 is limited by
the emitter voltage of the upper-stage transistor 501. The bias
current Ibias1, which corresponds to the voltage Vramp, is supplied
to the base of the transistor 500. Thus, an amplified signal RFout1
(first amplified signal) corresponding to the amount of the bias
current Ibias1 is output from the collector of the transistor 501.
The transistors 500 and 501 can have the same size, for
example.
As illustrated in FIG. 5, in the amplifier 400, the gain is
controlled by the bias current Ibias1. The gain may be controlled
by a bias voltage instead of a bias current; however, a
configuration in which the gain is controlled by a bias current has
higher controllability. This will now be described. It is assumed
that the collector current is represented by Ic, the base voltage
by Vb, the base current by Ib, the current amplification factor by
hFE, the saturation current by Is, the Boltzmann coefficient by k,
the absolute temperature by T, the elemental charge of an electron
by q, and the thermal voltage by Vt=k.times.T/q. In the case of
control using the base voltage,
.DELTA.Ic.apprxeq.Is.times.exp(.DELTA.Vb/Vt) holds. Accordingly,
the collector current Ic rapidly increases when the base voltage Vb
exceeds a threshold voltage. This results in low controllability.
In the case of control using the base current, in contrast,
.DELTA.Ic=.DELTA.Ib.times.hFE holds. Accordingly, the collector
current Ic changes with respect to the base current Ib at a
constant rate. This results in high controllability.
In this embodiment, the amplifiers 401 and 402 in the second and
subsequent stages have a configuration in which, compared with the
configuration illustrated in FIG. 5 (hereinafter referred to also
as a "cascode-connection configuration"), the cascode-connected
transistor 501 is not included (this configuration is hereinafter
referred to also as a "single-transistor configuration"). Instead
of the amplifier 400 in the initial stage, the amplifier 401 or 402
in any other stage may have a cascode-connection configuration, or
a plurality of or all of the amplifiers 400, 401, and 402 may have
a cascode-connection configuration. The amplifier 400 in the
initial stage provides less output power of a signal than the
amplifiers 401 and 402 in the second and three stages. Thus, when
only the amplifier 400 in the initial stage has a
cascode-connection configuration, signal attenuation caused by the
cascode-connection configuration is prevented or reduced, compared
with when any other amplifier has a cascode-connection
configuration. The effect of the cascode-connection configuration
will be described below.
Referring back to FIG. 4, the inductors 410 to 412 are each a choke
inductor for preventing or reducing leakage of an RF signal to a
power supply circuit.
The matching network 420 is disposed before the amplifier 400, the
matching networks 421 and 422 are respectively disposed before and
after the amplifiers 401 and 402, and the matching network 423 is
disposed after the amplifier 402. The matching networks 420 to 423
are circuits for matching impedances between the preceding and
subsequent circuits. The matching networks 420 to 423 are formed
using capacitors and inductors, for example.
The bias circuits 430 to 432 supply the bias currents Ibias1 to
Ibias3 to the amplifiers 400 to 402, respectively. The bias
currents Ibias1 to Ibias3 are adjusted on the basis of a current
Ibb output from the current source 452 and a current Ibc output
from the current source 450.
FIG. 6 is a diagram illustrating an example configuration of the
bias circuit 430. The bias circuit 430 includes a transistor 600,
diodes 610 and 611, and a resistor 620. The transistor 600 (third
transistor) is, for example, an HBT. The diodes 610 and 611 are
connected in series. The anode of the diode 610 is connected to a
base of the transistor 600, and the cathode of the diode 611 is
grounded. The resistor 620 has an end connected to an emitter of
the transistor 600, and another end connected to the base of the
transistor 500 included in the amplifier 400. The predetermined
current Ibb (bias control current) is supplied to the base of the
transistor 600. The current Ibc, which corresponds to the voltage
Vramp, is supplied to the collector of the transistor 600.
Accordingly, the bias current Ibias1, which corresponds to the
voltage Vramp, is output from the emitter of the transistor 600 via
the resistor 620. The configurations of the bias circuits 431 and
432 are similar to the configuration of the bias circuit 430 and
are not described in detail herein. Instead of the diodes 610 and
611, transistors (diode-connected transistors), each having a
collector and a base connected to each other, may be used.
Referring back to FIG. 4, the voltage-current conversion circuit
440 converts the voltage Vramp into a current Iramp and outputs the
current Iramp. The current source 450 (first current source)
outputs the current Ibc (first current) on the basis of the current
Iramp. The current Ibc is supplied to the bias circuits 430 to 432.
The current source 451 (second current source) outputs the current
Icas (second current) on the basis of the current Iramp. The
current Icas is supplied to only the amplifier 400 in the initial
stage. The current source 452 outputs the current Ibb. The current
Ibb remains constant during the slope control of the transmission
signal.
FIG. 7 is a diagram illustrating an example configuration of the
voltage-current conversion circuit 440 and the current sources 450
and 451.
The voltage-current conversion circuit 440 includes an operational
amplifier OP1, a p-channel MOSFET (MP1), an n-channel MOSFET (MN1),
and a resistor R1.
The operational amplifier OP1 has a non-inverting input terminal to
which the voltage Vramp is supplied, an inverting input terminal
connected to a source of the n-channel MOSFET (MN1), and an output
terminal connected to a gate of the re-channel MOSFET (MN1). The
p-channel MOSFET (MP1) has a source to which the power supply
voltage Vcc is supplied, a gate connected to a drain thereof, and
the drain connected to the drain of the n-channel MOSFET (MN1). The
source of the re-channel MOSFET (MN1) is connected to one end of
the resistor R1. The other end of the resistor R1 is grounded.
An imaginary short established between the non-inverting input
terminal and inverting input terminal of the operational amplifier
OP1 allows the voltage at the one end of the resistor R1 to be
equal to Vramp. If the resistor R1 has a resistance value r1, the
current flowing through the p-channel MOSFET (MP1) and the
n-channel MOSFET (MN1) is given by Vramp/r1. Accordingly, the
current Iramp that flows through the p-channel MOSFET (MP1) has a
value corresponding to the voltage Vramp.
P-channel MOSFETs (MP2a and MP2b) have sources to which the power
supply voltage Vcc is supplied, gates connected to the gate of the
p-channel MOSFET (MP1), and drains connected to drains of n-channel
MOSFETs (MN2a and MN2b). The n-channel MOSFETs (MN2a and MN2b) have
drains connected to the drains of the p-channel MOSFETs (MP2a and
MP2b), gates connected to the drains thereof, and sources grounded.
The p-channel MOSFETs (MP2a and MP2b) are each connected to the
p-channel MOSFET (MP1) to form a current mirror. Thus, a current
Ipa that flows through the p-channel MOSFET (MP2a) and the
n-channel MOSFET (MN2a) and a current Ipb that flows through the
p-channel MOSFET (MP2b) and the n-channel MOSFET (MN2b) have values
corresponding to the current Iramp (that is, values corresponding
to the voltage Vramp).
The current source 450 includes an operational amplifier OP2a,
p-channel MOSFETs (MP3a, MP4a, MP5a, and MP6a), n-channel MOSFETs
(MN3a, MN4a, MN5a, MN6a, and MN7a), and a current source Ja. The
current source 451 includes an operational amplifier OP2b,
p-channel MOSFETs (MP3b, MP4b, MP5b, and MP6b), n-channel MOSFETs
(MN3b, MN4b, MN5b, MN6b, and MN7b), and a current source Jb. The
following description will be given taking the current source 450
as an example. The configuration of the current source 451 is
similar to the configuration of the current source 450 and is not
described in detail herein.
The n-channel MOSFETs (MN3a and MN4a) have drains connected to
drains of the p-channel MOSFETs (MP5a and MP6a), gates connected to
the gate of the n-channel MOSFET (MN2a), and sources grounded. That
is, the n-channel MOSFETs (MN3a and MN4a) are each connected to the
n-channel MOSFET (MN2a) to form a current mirror. Accordingly,
currents In3 and In4 that flow through the n-channel MOSFETs (MN3a
and MN4a) are each proportional to the current Ipa that flows
through the re-channel MOSFET (MN2a) (In3.varies.Ipa and
In4.varies.Ipa).
The p-channel MOSFETs (MP3a and MP4a) have sources to which the
power supply voltage Vcc is supplied, gates connected to the output
terminal of the operational amplifier OP2a, and drains connected to
sources of the p-channel MOSFETs (MP5a and MP6a). A current that is
the sum of the current In3 and a current In6 flowing through the
n-channel MOSFET (MN6a) flows through the p-channel MOSFET (MP3a).
Thus, a current Ip3 that flows through the p-channel MOSFET (MP3a)
has a value corresponding to the current Ipa that flows through the
re-channel MOSFET (MN2a) {Ip3.apprxeq.(In3 (.varies.Ipa)+In6}).
Since the p-channel MOSFET (MP4a) is mirror-connected with the
p-channel MOSFET (MP3a), a current Ip4 that flows through the
p-channel MOSFET (MP4a) is proportional to the current Ip3 that
flows through the p-channel MOSFET (MP3a) (Ip4.varies.Ip3).
Accordingly, the current Ip4 that flows through the p-channel
MOSFET (MP4a) has a value corresponding to the current Ipa that
flows through the n-channel MOSFET (MN2a)
(Ip4.varies.Ip3.apprxeq.{In3 (.varies.Ipa)+In6}). Hence, the
current Ibc (i.e., the output current of the current source 450),
which is obtained by subtracting a current Ip6 that flows through
the p-channel MOSFET (MP6a) and a current In7 that flows through
the n-channel MOSFET (MN7a) from the current Ip4, also has a value
corresponding to the current Ipa (i.e., an amount of current
corresponding to the voltage Vramp).
The operational amplifier OP2a has an inverting input terminal to
which a reference voltage Vset is supplied, a non-inverting input
terminal to which a voltage Vdet is supplied, and an output
terminal connected to the gates of the p-channel MOSFETs (MP3a and
MP4a). The operational amplifier OP2a controls the gate voltages of
the p-channel MOSFETs (MP3a and MP4a) so that, as a result of an
imaginary short between the non-inverting input terminal and the
inverting input terminal, the voltage Vdet is equal to the
reference voltage Vset.
The p-channel MOSFET (MP5a) has a source connected to the drain of
the p-channel MOSFET (MP3a), a gate connected to a gate of the
p-channel MOSFET (MP6a), and a drain connected to the drain of the
n-channel MOSFET (MN3a). The p-channel MOSFET (MP6a) has a source
connected to the drain of the p-channel MOSFET (MP4a), a gate
connected to a drain thereof, and the drain connected to the drain
of the n-channel MOSFET (MN4a). Accordingly, the p-channel MOSFET
(MP5a) is connected to the p-channel MOSFET (MP6a) to form a
current mirror. The p-channel MOSFET (MP5a) can have the same
current density as the p-channel MOSFET (MP6a).
The p-channel MOSFETs (MP5a and MP6a) operate so that the drain
voltages of the p-channel MOSFETs (MP3a and MP4a) have the same
potential. Specifically, if the drain voltages of the p-channel
MOSFETs (MP3a and MP4a) are represented by Vd3 and Vd4 and the
gate-source voltages of the p-channel MOSFETs (MP5a and MP6a) are
represented by Vgs5 and Vgs6, then, Vd3-Vgs5=Vd4-Vgs6 holds since
the gates of the p-channel MOSFETs (MP5a and MP6a) are connected to
each other. Here, when the p-channel MOSFETs (MP5a and MP6a) have
an equal current density, the gate-source voltages Vgs5 and Vgs6 of
the p-channel MOSFETs (MP5a and MP6a) are equal. Thus, Vd3=Vd4
holds. Therefore, the drain voltages Vd3 and Vd4 of the p-channel
MOSFETs (MP3a and MP4a) can be maintained so as to have the same
potential, and current can be stably output.
The n-channel MOSFET (MN5a) has a drain to which a reference
current Isa is supplied from the current source Ja, a gate
connected to the drain thereof, and a source grounded. The
n-channel MOSFET (MN6a) has a drain connected to the drain of the
p-channel MOSFET (MP3a), a gate connected to the gate of the
n-channel MOSFET (MN5a), and a source grounded. The re-channel
MOSFET (MN7a) has a drain connected to the drain of the p-channel
MOSFET (MP4a), a gate connected to the gate of the n-channel MOSFET
(MN5a), and a source grounded. That is, the n-channel MOSFETs (MN6a
and MN7a) are each connected to the n-channel MOSFET (MN5a) to form
a current mirror. Hence, the current In6 that flows through the
n-channel MOSFET (MN6a) and the current In7 that flows through the
n-channel MOSFET (MN7a) have values corresponding to the reference
current Isa output from the current source Ja. Thus, the amounts of
the currents In6 and In7 are adjusted in accordance with a change
in the amount of the reference current Isa.
Even when the current In3 has a very small value (i.e., the voltage
Vramp is very low), the n-channel MOSFET (MN6a) causes a constant
amount of the current In6 to flow through the p-channel MOSFET
(MP3a), thereby maintaining the effect of the imaginary short of
the operational amplifier OP2a. This prevents or reduces
deterioration of the frequency characteristics of the feedback of
the operational amplifier OP2a or prevents or reduces the
occurrence of oscillation of the operational amplifier OP2a. In
addition, the amount of the current Ibc output from the current
source 450 is given by subtracting the current Ip6 and the current
In7 from the current Ip4 that flows through the p-channel MOSFET
(MP4a). Accordingly, the amount of the current Ibc changes in
accordance with a change in the amount of the reference current
Isa, and thus the responsiveness of the current Ibc to the
reference current Isa is improved.
As described above, the current source 450 generates the current
Ibc, which has a target level corresponding to the current Iramp
(i.e., corresponding to the voltage Vramp) and outputs the current
Ibc. The target level of the current Ibc is a level necessary to
achieve the current to be supplied to the collectors of the
respective transistors in the bias circuits 430 to 432.
Specifically, the current Ibc increases in accordance with an
increase in the voltage Vramp and decreases in accordance with a
reduction in the voltage Vramp. The configuration of the current
source 451 is similar to that of the current source 450.
Accordingly, the current sources 450 and 451 can output the
currents Ibc and Icas, which change in accordance with the voltage
Vramp. For example, the configuration, element sizes, etc. of the
current source 451 may be made different from those of the current
source 450 to generate a smaller amount of current Icas than the
amount of the current Ibc.
FIG. 8 is a graph illustrating an example relationship between the
voltage Vramp and the currents Ibc and Icas. As illustrated in FIG.
8, it is found that the current Ibc and the current Icas increase
linearly as the voltage Vramp increases. Furthermore, the current
Ibc changes by a larger amount with a change in the voltage Vramp
than the current Icas. In this embodiment, therefore, the amount of
the current Ibc is larger than the amount of the current Icas.
With the configuration described above, in the power amplifier
module 120A-1, the gain of the amplifier 400 is controlled using
both the currents Ibc and Icas, and the gains of the amplifiers 401
and 402 are controlled using the current Ibc. Specifically, since
the current Ibc increases in accordance with an increase in the
voltage Vramp, the bias currents Ibias1 to Ibias3 supplied from the
bias circuits 430 to 432 increase. Accordingly, the gains of the
amplifiers 400 to 402 increase in accordance with an increase in
the voltage Vramp. In addition, since the current Icas increases in
accordance with an increase in the voltage Vramp, the base current
of the transistor 501 in the amplifier 400 in the initial stage
increases. Accordingly, the gain of the amplifier 400 in the
initial stage increases in accordance with an increase in the
voltage Vramp. That is, in the power amplifier module 120, during a
low-output-power period, the bias currents Ibias1 to Ibias3 are
decreased to reduce the gains of the amplifiers 400 to 402, and
during a high-output-power period, the bias currents Ibias1 to
Ibias3 are increased to increase the gains of the amplifiers 400 to
402. The gains of the amplifiers 400 to 402 are controlled in the
way described above, and thereby slope control of the transmission
signal is performed. This configuration therefore enables slope
control of a transmission signal while preventing or reducing an
increase in circuit size, compared with the configuration in which
an LDO regulator is used. Next, the operation of a
cascode-connected transistor in the amplifier 400 in the initial
stage will be described in detail.
FIG. 9A is a schematic diagram of an ideal output waveform of an RF
signal, FIG. 9B is a schematic diagram of an output waveform of an
RF signal in a comparative example, and FIG. 9C is a schematic
diagram of an output waveform of an RF signal in the amplifier 400.
In the comparative example, a configuration that does not include a
cascode-connected transistor (single-transistor configuration) is
used.
As illustrated in FIG. 9A, a transistor typically outputs a signal
with a waveform w1 having no distortion when supplied with a bias
voltage VbiasA so that the base-emitter voltage exceeds a threshold
voltage Vth even when the voltage decreases in accordance with the
waveform of an input signal. In the comparative example, however,
the bias current is reduced in accordance with a reduction in the
voltage Vramp during a low-output-power period. As a result, as
illustrated in FIG. 9B, a bias voltage VbiasB excessively
decreases, and no signal is output in an area where the
base-emitter voltage is less than or equal to the threshold voltage
Vth. Thus, a signal of a waveform w2 that is greatly distorted on
one side (low-voltage side) is output. In the comparative example,
accordingly, distortion characteristics deteriorate, causing a
problem in that the linearity of the gain is impaired.
In the amplifier 400, in contrast, during a low-output-power
period, the amount of the current Icas also decreases in accordance
with a decrease in the voltage Vramp. As a result, the amount of
current flowing through the transistor 501 decreases. Since the
amount of current flowing through the transistor 500 is limited by
the amount of current flowing through the transistor 501, the
amount of current flowing through the transistor 500 also
decreases. Accordingly, the gain of the amplifier 400 can be
reduced, compared with the comparative example, when the amount of
bias current is the same as that in the comparative example. In
other words, even if the amount of bias current is larger than that
in the comparative example, the gain of the amplifier 400 can be
reduced to substantially the same level as that of the comparative
example. Thus, during a low-output-power period, deterioration of
distortion characteristics caused by an excessive reduction in bias
voltage can be prevented or reduced in an area where the voltage
Vramp is low.
In this case, as illustrated in FIG. 9C, the signal output from the
amplifier 400 has a waveform w3 in which, for example, no signal is
output in both an area where the base-emitter voltage of the
transistor 500 is less than or equal to the threshold voltage Vth
and an area where the collector voltage of the transistor 500 is
greater than or equal to an upper limit value Vc that is limited by
the emitter voltage of the transistor 501. In the waveform w3, no
signal is output on both the high- and low-voltage sides. Thus, an
output waveform is exhibited which is more symmetrical with respect
to a bias voltage VbiasC than the waveform w2 in which no signal is
output only on the low-voltage side. Accordingly, the amplifier 400
has improved distortion characteristics, compared with the
comparative example, resulting in improved linearity of the
gain.
FIG. 10 is a graph illustrating results of simulation of adjacent
channel leakage ratio (ACLR) characteristics in the amplifier 400
and the comparative example. In the graph illustrated in FIG. 10,
the vertical axis represents the ACLR in dBc, and the horizontal
axis represents the frequency in Hz when the center frequency of
the transmission signal is set to 0.
As illustrated in FIG. 10, it is found that in the amplifier 400,
compared with the comparative example, the ACLR is reduced at
frequencies other than the target frequency band of the
transmission signal. It is therefore found that the amplifier 400
provided with a cascode-connection configuration has improved
distortion characteristics, compared with the comparative example,
and that, as a result, the linearity of the gain is improved.
FIG. 11 is a diagram illustrating the configuration of a power
amplifier module 120A-2. The power amplifier module 120A-2 is an
example of the power amplifier module 120. The same or
substantially the same elements as those of the power amplifier
module 120A-1 illustrated in FIG. 4 are assigned the same numerals
and are not described herein. In FIG. 11, for convenience of
illustration, elements included in the amplifiers 400 to 402 are
represented by numerals that are similar to those in FIG. 5, and
elements included in the bias circuits 430 to 432 are represented
by numerals that are similar to those in FIG. 6.
The power amplifier module 120A-2 includes negative feedback
circuits 700 and 701 and a resistor 710, in addition to the
elements of the power amplifier module 120A-1.
The negative feedback circuit 700 includes a resistor 720 (first
resistor) and a capacitor 721. The resistor 720 and the capacitor
721 are connected in series and are provided between the base of
the transistor 500a and the collector of the transistor 501. The
resistor 720 and the capacitor 721 stabilize the amplification
operation of the amplifier 400 in the initial stage, which is
constituted by the transistors 500a and 501. It is assumed here
that the amplifier 400 in the initial stage has a typical
configuration in which the configuration (single-transistor
configuration) adopted in the comparative example includes the
negative feedback circuit 700. When such a typical
single-transistor configuration is modified into a
cascode-connection configuration, the negative feedback circuit 700
disposed between the base of the transistor 500a and the collector
of the transistor 501, rather than between the base and collector
of the transistor 500a, can prevent or reduce a change in the
output impedance of the amplifier in the initial stage.
Accordingly, the typical single-transistor configuration described
above can be easily modified in design.
The negative feedback circuit 701 includes a resistor 730. The
resistor 730 is disposed between the base and collector of the
transistor 500b (fourth transistor). The resistor 730 stabilizes
the amplification operation of the amplifier 401 in the second
stage, which is constituted by the transistor 500b. As described
above, the negative feedback circuit may not include one of a
resistor and a capacitor. The number of stages of amplifiers that
include a negative feedback circuit is not limited to that in the
illustrated example, and any one amplifier may include a negative
feedback circuit or every amplifier may include a negative feedback
circuit.
The resistor 710 (second resistor) has an end connected to the
collectors of the transistors 600a (third transistor), 600b (fifth
transistor), and 600c respectively included in the bias circuits
430 to 432, and another end grounded. The resistor 710 is, for
example, a resistor having a comparatively small resistance value
(e.g., about 250 to 700 .OMEGA.). The use of the resistor 710
allows the current Ibb supplied from the current source 452 to the
bias circuits 430 to 432 to flow through the resistor 710 via the
bases and collectors of the transistors 600a, 600b, and 600c. This
prevents or reduces the flow of the current Ibb into the amplifiers
400 to 402 via the bases and emitters of the transistors 600a,
600b, and 600c.
With the configuration described above, the power amplifier module
120A-2 can also achieve advantages similar to those of the power
amplifier module 120A-1. Since the power amplifier module 120A-2
includes the negative feedback circuits 700 and 701, a stable
amplification operation of amplifiers is achieved, compared with
the power amplifier module 120A-1. In addition, since the power
amplifier module 120A-2 includes the resistor 710, the flow of the
current Ibb into the amplifiers 400 to 402 is prevented or reduced,
compared with the power amplifier module 120A-1.
In this embodiment, the currents Ibc and Icas output from the
current sources 450 and 451 change linearly with respect to the
voltage Vramp. Alternatively, the current sources 450 and 451 may
be configured to output a current Ibc and a current Icas that
change non-linearly (e.g., with substantially the square) with
respect to the voltage Vramp. In this case, for example, the rates
of change of the currents Ibc and Icas when the voltage Vramp has a
comparatively high level are higher than the rates of change of the
currents Ibc and Icas when the voltage Vramp has a comparatively
low level. Accordingly, the rates of change of the bias currents
Ibias1 to Ibias3 to be supplied to the amplifiers 400 to 402 can be
increased (the slopes can be made steeper) in an area where the
voltage Vramp is high. Therefore, as illustrated in FIG. 2, slope
control of the transmission signal can be performed so as not to
change the signal level too gently at or near a falling edge (i.e.,
in an area with a high signal level).
The current sources 450 and 451 may be configured such that one of
the current Ibc or the current Icas (e.g., the current Ibc) changes
non-linearly with respect to the voltage Vramp and the other
current (e.g., the current Icas) changes linearly with respect to
the voltage Vramp.
The current source 450 may be a voltage source (first voltage
source) that outputs a voltage (first voltage) corresponding to the
voltage Vramp, instead of a current source. Likewise, the current
source 451 may be a voltage source (second voltage source) that
outputs a voltage (second voltage) corresponding to the voltage
Vramp, instead of a current source. The first voltage may have a
higher voltage value than the second voltage. The current source
452 may use a voltage source, instead of a current source, to
supply a voltage (bias control voltage) to the bases of transistors
in bias circuits.
FIG. 12 is a diagram illustrating the configuration of a power
amplifier module 120A-3. The power amplifier module 120A-3 is an
example of the power amplifier module 120. The same or
substantially the same elements as those of the power amplifier
module 120A-2 illustrated in FIG. 11 are assigned the same numerals
and are not described herein. In FIG. 12, for convenience of
illustration, elements included in the amplifiers 400 to 402 are
represented by numerals that are similar to those in FIG. 5, and
elements included in the bias circuits 430 to 432 are represented
by numerals that are similar to those in FIG. 6.
The power amplifier module 120A-3 includes an impedance element 800
in addition to the elements of the power amplifier module
120A-2.
The impedance element 800 is disposed between the collector of the
transistor 600a and the current source 450. The impedance element
800 can adjust the amount of the current Ibc flowing from the
current source 450 to the collector of the transistor 600a. In the
configuration illustrated in FIG. 12, the amount of current flowing
through the transistor 600a can be reduced compared with that
through the transistors 600b and 600c.
Suitable examples of the impedance element 800 include a resistor
and a diode. The impedance element 800 may be an element having a
resistance component. The resistor preferably has a resistance
value of, for example, about 800 .OMEGA. to about 1500 .OMEGA. in
terms of the size of the resistor and the amount of current
Ibc.
The effect of the arrangement of a resistor or a diode will be
described. First, the operation mechanism without the impedance
element 800 will be described. In an area where the output of an
output signal RFout is presumably small (about 0 to 10 dBm), the
current Ibc supplied from the current source 450 is small, and the
collector voltage of the transistor 600a is low. A portion of the
current Ibb supplied to the base of the transistor 600a flows to
the collector thereof. In this situation, if the input signal RFin
having a high signal level (e.g., 6 dBm) is input to the base of
the transistor 500a, the average voltage at the base of the
transistor 500a is lower than that when the input signal RFin
having a low signal level is input to the base of the transistor
500a. A decrease in the average voltage at the base of the
transistor 500a causes a decrease in the potential at the emitter
of the transistor 600a, and the collector voltage of the transistor
600a also decreases accordingly. As a result, the amount of current
flowing from the base to the collector of the transistor 600a
increases. In this case, the amount of emitter current decreases,
which leads to a reduction in the gain of the transistor 500a and a
reduction in the output of the output signal RFout.
The case where the impedance element 800 is connected between the
current source 450 and the collector of the transistor 600a as
illustrated in FIG. 12 will be described. For simplicity of
illustration, the impedance element 800 is assumed to be a
resistor. In FIG. 12, the operations in the area where the output
of the output signal RFout is presumably small (about 0 to 10 dBm)
are substantially the same as those in FIG. 11, except that the
voltage at the collector of the transistor 600a is increased by an
amount corresponding to the potential difference across the
resistor. The increase in the voltage at the collector of the
transistor 600a reduces the potential between the base and
collector of the transistor 600a. Thus, in the configuration
illustrated in FIG. 11, current flowing from the base to the
collector of the transistor 600a is present, whereas in the
configuration illustrated in FIG. 12, the current flowing from the
base to the collector of the transistor 600a can be decreased. As a
result, even if the output of the output signal RFout is small and
the signal level of the input signal RFin is high, the gain of the
transistor 500a can be maintained.
When the impedance element 800 is implemented as a diode, it is
desirable to connect the cathode of the diode to the collector of
the transistor 600a and to connect the anode of the diode to the
current source 450. This configuration allows the diode to block
the flow of DC current from the base to the collector of the
transistor 600a. Thus, a diode is more effective than a
resistor.
An illustrative embodiment of the present disclosure has been
described. In the power amplifier modules 120A-1, 120A-2, and
120A-3, the current Ibc output from the current source 450 is
supplied to the collectors of transistors in the bias circuits 430
to 432. The current Ibc is controlled in accordance with the
voltage Vramp, and thereby slope control of a transmission signal
is performed. Thus, the power amplifier modules 120A-1, 120A-2, and
120A-3 can perform slope control of a transmission signal without
necessarily using an LDO regulator. Accordingly, it may be possible
to prevent or reduce an increase in circuit size, compared with the
case where an LDO regulator is used. The amplifier 400 includes the
transistor 501, which is connected in cascode with the transistor
500. The current Icas, which is controlled in accordance with the
voltage Vramp, is supplied to the base of the transistor 501.
Accordingly, it may be possible to reduce the gain of the amplifier
400, compared with a typical single-transistor configuration, even
when a large amount of bias current is supplied during a
low-output-power period. Thus, deterioration of distortion
characteristics caused by an excessive reduction in bias voltage
can be prevented or reduced in an area where the voltage Vramp is
low.
The relationship between the current Ibc and the current Icas is
not limited to any particular one. For example, the amount of the
current Ibc may be larger than the amount of the current Icas.
The power amplifier modules 120A-1, 120A-2, and 120A-3 include a
plurality of amplifiers 400 to 402, and the transistors in the
amplifier 400 in the initial stage are connected in cascode. The
amplifier 400 in the initial stage provides less output power of a
signal than the amplifiers 401 and 402 in the subsequent stages.
Thus, the amplifier 400 in the initial stage, which is provided
with a cascode-connection configuration, prevents or reduces signal
attenuation caused by the cascode-connection configuration.
The power amplifier module 120 may also be configured such that a
voltage output from a voltage source is supplied to the collectors
of the transistors in the bias circuits 430 to 432 and a voltage
output from a voltage source is supplied to the base of the
transistor 501, which is connected in cascode with the transistor
500 in the amplifier 400. This configuration allows the power
amplifier module 120 to perform slope control of a transmission
signal while preventing or reducing an increase in circuit size. In
addition, since the transistors in the amplifier 400 are connected
in cascode, the distortion characteristics of an amplifier can be
improved, compared with a configuration having no cascode
connection.
The relationship between a voltage (first voltage) supplied to a
transistor included in a bias circuit and a voltage (second
voltage) supplied to a transistor included in an amplifier is not
limited to any particular one. For example, the first voltage may
have a higher voltage value than the second voltage.
The power amplifier module 120A-2 includes the negative feedback
circuit 700 between the collector of the transistor 501 and the
base of the transistor 500a, and the negative feedback circuit 701
between the collector and base of the transistor 500b. Thus, the
amplification operation of the amplifier 400 and the amplifier 401
is stabilized.
The configuration of the negative feedback circuit 700 is not
limited to any particular one. For example, the negative feedback
circuit 700 may include the resistor 720 and the capacitor 721,
which are connected in series.
The power amplifier module 120A-2 further includes the resistor 710
between ground and collectors of the transistor 600a to 600c
respectively included in the bias circuits 430 to 432. This
configuration allows the current Ibb supplied from the current
source 452 to the bias circuits 430 to 432 to flow through the
resistor 710 via the bases and collectors of the transistors 600a,
600b, and 600c. Accordingly, the flow of the current Ibb into the
amplifiers 400 to 402 via the emitters of the transistors 600a,
600b, and 600c is prevented or reduced.
The embodiment described above is intended for easy understanding
of the present invention, and it is not intended to construe the
present invention in a limiting fashion. Various modifications or
improvements can be made to the present invention without departing
from the gist of the present invention, and equivalents thereof are
also included in the present invention. That is, the embodiment may
be appropriately modified in design by those skilled in the art,
and such modifications also fall within the scope of the present
invention so long as the modifications include the features of the
present invention. For example, the elements included in the
embodiment and the arrangement, materials, conditions, shapes,
sizes, and the like thereof are not limited to those described in
the illustrated examples but can be modified as appropriate. In
addition, the elements included in the embodiment can be combined
as much as technically possible, and such combinations of elements
also fall within the scope of the present invention so long as the
combinations of elements include the features of the present
invention.
While preferred embodiments of the invention have been described
above, it is to be understood that variations and modifications
will be apparent to those skilled in the art without departing from
the scope and spirit of the invention. The scope of the invention,
therefore, is to be determined solely by the following claims.
* * * * *