U.S. patent application number 17/695106 was filed with the patent office on 2022-06-30 for aperture-fed, stacked-patch antenna assembly.
This patent application is currently assigned to SKYLINE PARTNERS TECHNOLOGY LLC. The applicant listed for this patent is SKYLINE PARTNERS TECHNOLOGY LLC. Invention is credited to David Andrew G. LEA, Kevin J. NEGUS.
Application Number | 20220209427 17/695106 |
Document ID | / |
Family ID | |
Filed Date | 2022-06-30 |
United States Patent
Application |
20220209427 |
Kind Code |
A1 |
LEA; David Andrew G. ; et
al. |
June 30, 2022 |
APERTURE-FED, STACKED-PATCH ANTENNA ASSEMBLY
Abstract
Directive gain antenna elements implemented with an aperture-fed
patch array antenna assembly are described. A feed network for the
aperture-fed patch array may include offset apertures and may also
include meandering feed lines. Scalable aperture shapes and
orientations that can be used with antennas operating at any
frequency and with dual orthogonal polarizations are also
disclosed. Directive gain antenna elements implemented with arrays
of orthogonal reflected dipoles are also described with optimal
feed networks and parasitic elements to achieve desired directive
gain characteristics. Such arrayed dipole antennas feature dual
orthogonal polarizations with assembly tabs that lower cost and
improve reliability. Backhaul radios that incorporate said antennas
are also disclosed.
Inventors: |
LEA; David Andrew G.;
(Vancouver, B.C., CA) ; NEGUS; Kevin J.;
(Philipsburg, MT) |
|
Applicant: |
Name |
City |
State |
Country |
Type |
SKYLINE PARTNERS TECHNOLOGY LLC |
Boulder |
CO |
US |
|
|
Assignee: |
SKYLINE PARTNERS TECHNOLOGY
LLC
Boulder
CO
|
Appl. No.: |
17/695106 |
Filed: |
March 15, 2022 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
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16393560 |
Apr 24, 2019 |
11283192 |
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17695106 |
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15470080 |
Mar 27, 2017 |
10313898 |
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16393560 |
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14559859 |
Dec 3, 2014 |
9609530 |
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15470080 |
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14197158 |
Mar 4, 2014 |
8928542 |
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14559859 |
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13645472 |
Oct 4, 2012 |
8811365 |
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14197158 |
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13371366 |
Feb 10, 2012 |
8311023 |
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13645472 |
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13212036 |
Aug 17, 2011 |
8238318 |
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13371366 |
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International
Class: |
H01Q 21/24 20060101
H01Q021/24; H01Q 25/00 20060101 H01Q025/00; H01Q 1/48 20060101
H01Q001/48; H01Q 1/50 20060101 H01Q001/50; H01Q 9/04 20060101
H01Q009/04; H01Q 9/28 20060101 H01Q009/28; H01Q 21/00 20060101
H01Q021/00; H01Q 21/06 20060101 H01Q021/06; H01Q 21/08 20060101
H01Q021/08; H01Q 21/26 20060101 H01Q021/26; H01Q 21/29 20060101
H01Q021/29; H01Q 1/24 20060101 H01Q001/24 |
Claims
1. An antenna assembly comprising: a plurality of first substrates,
each of the plurality of first substrates comprising a unitary
dipole antenna element and a slot; a second substrate comprising a
plurality of coplanar dipole antenna elements and a plurality of
slots; and a third substrate, wherein the plurality of first
substrates are coupled to the third substrate, and wherein the
plurality of slots in the second substrate are configured to engage
with the slot in each of the plurality of first substrates to form
a two-port, orthogonally polarized dipole antenna array.
Description
CROSS-REFERENCE TO RELATED APPLICATIONS
[0001] The present application is a continuation of U.S. patent
application Ser. No. 16/393,560, filed Apr. 24, 2019, currently
pending, which is a continuation of U.S. patent application Ser.
No. 15/470,080, filed Mar. 27, 2017, now U.S. Pat. No. 10,313,898,
which is a continuation of U.S. patent application Ser. No.
14/559,859, filed on Dec. 3, 2014, now U.S. Pat. No. 9,609,530,
which is a continuation of U.S. patent application Ser. No.
14/197,158, filed on Mar. 4, 2014, now U.S. Pat. No. 8,928,542,
which is a continuation-in-part of U.S. patent application Ser. No.
13/645,472, filed on Oct. 4, 2012, now U.S. Pat. No. 8,811,365,
which is a continuation of U.S. patent application Ser. No.
13/371,366, filed on Feb. 10, 2012, now U.S. Pat. No. 8,311,023,
which is a continuation of U.S. patent application Ser. No.
13/212,036, filed on Aug. 17, 2011, now U.S. Pat. No. 8,238,318,
the disclosures of which are hereby incorporated herein by
reference in their entireties.
[0002] The present application is also related to U.S. patent
application Ser. No. 13/898,429, filed May 20, 2013 and U.S. Pat.
No. 8,467,363, the disclosures of which are hereby incorporated
herein by reference in their entirety.
[0003] The present application is also related to U.S. patent
application Ser. No. 13/271,051, filed Oct. 11, 2011 and U.S. Pat.
No. 8,300,590, the disclosures of which are hereby incorporated
herein by reference in their entirety.
[0004] The present application is also related to U.S. patent
application Ser. No. 14/108,200, filed Dec. 16, 2013 and U.S. Pat.
Nos. 8,638,839 and 8,422,540, the disclosures of which are hereby
incorporated herein by reference in their entirety.
BACKGROUND
1. Field
[0005] The present disclosure relates generally to data networking
and in particular to a backhaul radio for connecting remote edge
access networks to core networks.
2. Related Art
[0006] Data networking traffic has grown at approximately 100% per
year for over 20 years and continues to grow at this pace. Only
transport over optical fiber has shown the ability to keep pace
with this ever-increasing data networking demand for core data
networks. While deployment of optical fiber to an edge of the core
data network would be advantageous from a network performance
perspective, it is often impractical to connect all high bandwidth
data networking points with optical fiber at all times. Instead,
connections to remote edge access networks from core networks are
often achieved with wireless radio, wireless infrared, and/or
copper wireline technologies.
[0007] Radio, especially in the form of cellular or wireless local
area network (WLAN) technologies, is particularly advantageous for
supporting mobility of data networking devices. However, cellular
base stations or WLAN access points inevitably become very high
data bandwidth demand points that require continuous connectivity
to an optical fiber core network.
[0008] When data aggregation points, such as cellular base station
sites, WLAN access points, or other local area network (LAN)
gateways, cannot be directly connected to a core optical fiber
network, then an alternative connection, using, for example,
wireless radio or copper wireline technologies, must be used. Such
connections are commonly referred to as "backhaul."
[0009] Many cellular base stations deployed to date have used
copper wireline backhaul technologies such as T1, E1, DSL, etc.
when optical fiber is not available at a given site. However, the
recent generations of HSPA+ and LTE cellular base stations have
backhaul requirements of 100 Mb/s or more, especially when multiple
sectors and/or multiple mobile network operators per cell site are
considered. WLAN access points commonly have similar data backhaul
requirements. These backhaul requirements cannot be practically
satisfied at ranges of 300m or more by existing copper wireline
technologies. Even if LAN technologies such as Ethernet over
multiple dedicated twisted pair wiring or hybrid fiber/coax
technologies such as cable modems are considered, it is impractical
to backhaul at such data rates at these ranges (or at least without
adding intermediate repeater equipment). Moreover, to the extent
that such special wiring (i.e., CAT 5/6 or coax) is not presently
available at a remote edge access network location; a new high
capacity optical fiber is advantageously installed instead of a new
copper connection.
[0010] Rather than incur the large initial expense and time delay
associated with bringing optical fiber to every new location, it
has been common to backhaul cell sites, WLAN hotspots, or LAN
gateways from offices, campuses, etc. using microwave radios. An
exemplary backhaul connection using the microwave radios 132 is
shown in FIG. 1. Traditionally, such microwave radios 132 for
backhaul have been mounted on high towers 112 (or high rooftops of
multi-story buildings) as shown in FIG. 1, such that each microwave
radio 132 has an unobstructed line of sight (LOS) 136 to the other.
These microwave radios 132 can have data rates of 100 Mb/s or
higher at unobstructed LOS ranges of 300 m or longer with latencies
of 5 ms or less (to minimize overall network latency).
[0011] Traditional microwave backhaul radios 132 operate in a Point
to Point (PTP) configuration using a single "high gain" (typically
>30 dBi or even >40 dBi) antenna at each end of the link 136,
such as, for example, antennas constructed using a parabolic dish.
Such high gain antennas mitigate the effects of unwanted multipath
self-interference or unwanted co-channel interference from other
radio systems such that high data rates, long range and low latency
can be achieved. These high gain antennas however have narrow
radiation patterns.
[0012] Furthermore, high gain antennas in traditional microwave
backhaul radios 132 require very precise, and usually manual,
physical alignment of their narrow radiation patterns in order to
achieve such high performance results. Such alignment is almost
impossible to maintain over extended periods of time unless the two
radios have a clear unobstructed line of sight (LOS) between them
over the entire range of separation. Furthermore, such precise
alignment makes it impractical for any one such microwave backhaul
radio to communicate effectively with multiple other radios
simultaneously (i.e., a "point to multipoint" (PMP)
configuration).
[0013] In wireless edge access applications, such as cellular or
WLAN, advanced protocols, modulation, encoding and spatial
processing across multiple radio antennas have enabled increased
data rates and ranges for numerous simultaneous users compared to
analogous systems deployed 5 or 10 years ago for obstructed LOS
propagation environments where multipath and co-channel
interference were present. In such systems, "low gain" (usually
<6 dBi) antennas are generally used at one or both ends of the
radio link both to advantageously exploit multipath signals in the
obstructed LOS environment and allow operation in different
physical orientations as would be encountered with mobile devices.
Although impressive performance results have been achieved for edge
access, such results are generally inadequate for emerging backhaul
requirements of data rates of 100 Mb/s or higher, ranges of 300 m
or longer in obstructed LOS conditions, and latencies of 5 ms or
less.
[0014] In particular, "street level" deployment of cellular base
stations, WLAN access points or LAN gateways (e.g., deployment at
street lamps, traffic lights, sides or rooftops of single or
low-multiple story buildings) suffers from problems because there
are significant obstructions for LOS in urban environments (e.g.,
tall buildings, or any environments where tall trees or uneven
topography are present).
[0015] FIG. 1 illustrates edge access using conventional
unobstructed LOS PTP microwave radios 132. The scenario depicted in
FIG. 1 is common for many 2.sup.nd Generation (2G) and 3.sup.rd
Generation (3G) cellular network deployments using "macrocells". In
FIG. 1, a Cellular Base Transceiver Station (BTS) 104 is shown
housed within a small building 108 adjacent to a large tower 112.
The cellular antennas 116 that communicate with various cellular
subscriber devices 120 are mounted on the towers 112. The PTP
microwave radios 132 are mounted on the towers 112 and are
connected to the BTSs 104 via an nT1 interface. As shown in FIG. 1
by line 136, the radios 132 require unobstructed LOS.
[0016] The BTS on the right 104a has either an nT1 copper interface
or an optical fiber interface 124 to connect the BTS 104a to the
Base Station Controller (BSC) 128. The BSC 128 either is part of or
communicates with the core network of the cellular network
operator. The BTS on the left 104b is identical to the BTS on the
right 104a in FIG. 1 except that the BTS on the left 104b has no
local wireline nT1 (or optical fiber equivalent) so the nT1
interface is instead connected to a conventional PTP microwave
radio 132 with unobstructed LOS to the tower on the right 112a. The
nT1 interfaces for both BTSs 104a, 104b can then be backhauled to
the BSC 128 as shown in FIG. 1.
[0017] In the conventional PTP radios 132, as described in greater
detail in U.S. patent application Ser. No. 13/645,472 and
incorporated herein, the antenna is typically of very high gain
such as can be achieved by a parabolic dish so that gains of
typically >30 dBi (or even sometimes >40 dBi), can be
realized. Such an antenna usually has a narrow radiation pattern in
both the elevation and azimuth directions. The use of such a highly
directive antenna in a conventional PTP radio link with
unobstructed LOS propagation conditions ensures that a modem within
such radios has insignificant impairments at the receiver due to
multipath self-interference and further substantially reduces the
likelihood of unwanted co-channel interference due to other nearby
radio links. However, the conventional PTP radio on a whole is
completely unsuitable for obstructed LOS or PMP operation.
[0018] In U.S. patent application Ser. No. 13/645,472 and the
related applications and patents summarized above, a novel
Intelligent Backhaul Radio (or "IBR") suitable for obstructed LOS
and PMP or PTP operation is described in great detail in various
embodiments of those inventions. Additionally, in U.S. patent
application Ser. No. 13/898,429, certain exemplary antenna
assemblies were described. Applicants have identified herein
additional improvements to antenna assembly designs for both
patch-based and dipole-based radiating element structures.
[0019] Aperture-fed antennas have been previously known in the art.
For example, in D. M. Pozar, "A microstrip antenna aperture-coupled
to a microstripline," Electron. Lett., vol. 21, no. 2, pp. 49-50,
1985, and in D. M. Pozar and S. D. Targonski, "Improved coupling
for aperture-coupled microstrip antennas," Electron. Lett., vol.
27, no. 13, pp. 1129-1131, 1991, an aperture-fed patch antenna was
disclosed. Additionally, in S. C. Gao et al., "Dual-polarized
slot-coupled planar antenna with wide bandwidth," IEEE Trans.
Antennas and Propagation, vol. 51, no. 3, pp. 441-448, 2003, a
dual-polarization aperture-fed antenna was disclosed. However, the
conventional art is completely unsuitable for application in an
IBR. For example, the conventional aperture fed antennas have
insufficient antenna gain for IBR directive gain antenna elements,
have unacceptable coupling efficiencies, have unacceptable
backwards facing radiation and are impractical to manufacture
cost-effectively and reliably.
SUMMARY
[0020] The following summary of the invention is included in order
to provide a basic understanding of some aspects and features of
the invention. This summary is not an extensive overview of the
invention and as such it is not intended to particularly identify
key or critical elements of the invention or to delineate the scope
of the invention. Its sole purpose is to present some concepts of
the invention in a simplified form as a prelude to the more
detailed description that is presented below.
[0021] Some embodiments of the claimed inventions are directed to
an improved antenna assembly including an array of resonant
radiating patch antenna elements and transmission line feed
networks that are electromagnetically coupled using apertures.
Other embodiments of the claimed inventions are directed to an
improved antenna assembly including an array of dipole antenna
elements and transmission line feed networks that are conductively
connected at junctions formed with substrate tabs and cutouts.
Backhaul radios that include the improved antenna assemblies are
also disclosed.
[0022] According to an aspect of the invention, an antenna assembly
is provided that includes a first substrate comprising a plurality
of conductive patch elements; a second substrate comprising a first
layer with at least a conductive ground plane and a plurality of
pairs of apertures, wherein the number of pairs of apertures is
equal to the number of conductive patch elements, and a second
layer with at least a first transmission line feed network coupled
to a first feed point and a second transmission line feed network
coupled to a second feed point; and a spacer interposed between the
first substrate and the second substrate, the spacer comprising a
dielectric material and at least one spacer opening in the
dielectric material, wherein the dielectric material is absent
within the at least one spacer opening; wherein the first
transmission line feed network overlaps a first aperture of each
pair of the plurality of pairs of apertures and the second
transmission line feed network overlaps a second aperture of each
pair of the plurality of pairs of apertures; wherein the first
aperture of each pair of the plurality of pairs of apertures
electromagnetically couples the first transmission line feed
network and the second aperture of each pair of the plurality of
pairs of apertures electromagnetically couples the second
transmission line feed network to a respective one of the plurality
of conductive patch elements; and wherein the first aperture of
each pair of the plurality of pairs of apertures is orthogonal to
the second aperture of each pair of the plurality of pairs of
apertures.
[0023] The first substrate may be a printed circuit board. The
second substrate may be a printed circuit board. The second
substrate may be a printed circuit board having more than two
layers.
[0024] The first transmission line feed network and the second
transmission line feed network each may include striplines. The
first transmission line feed network and the second transmission
line feed network each may include microstrip lines. The first feed
point and the second feed point may each be coupled to respective
components on an outside layer of the second substrate. The
respective components may be at least one of an RF bandpass filter
or a low noise amplifier within a receiver.
[0025] The at least one spacer opening may extend beyond a
projected area of one or more of the plurality of conductive patch
elements by at least a distance equal to a thickness of the
spacer.
[0026] The first aperture of each pair of the plurality of pairs of
apertures may excite a respective resonant radiating cavity formed
between each respective one of the plurality of conductive patch
elements and the conductive ground plane in an electromagnetic mode
corresponding to a vertical polarization far-field pattern, and
wherein the second aperture of each pair of the plurality of pairs
of apertures may excite said respective resonant radiating cavity
in an electromagnetic mode corresponding to a horizontal
polarization far-field pattern.
[0027] The antenna assembly may further include a plurality of
plastic fasteners to hold the first substrate, the second substrate
and the spacer together.
[0028] The first aperture of each respective pair of the plurality
of pairs of apertures may be oriented relative to the second
aperture of each respective pair of the plurality of pairs of
apertures in a T-shape. Each of the first aperture and the second
aperture of each respective pair of the plurality of pairs of
apertures may include a rectangular aperture body with an aperture
body width and a pair of aperture ends with an aperture end width.
Each aperture end may include a rectangular end and a semi-circular
end with a radius equal to one half of the aperture end width. The
aperture end width may be at least five times greater than the
aperture body width. Each aperture end may be tapered or rounded.
The rectangular end may have a width equal to the aperture end
width and a thickness equal to one sixth of the aperture end width.
The aperture end width may be equal to one third of an aperture
length.
[0029] The first transmission line feed network may be terminated
by a first via to the conductive ground plane after a feedline
portion of the first transmission line feed network crosses over
the rectangular aperture body of the first aperture of each pair of
the plurality of pairs of apertures, and the second transmission
line feed network may be terminated by a second via to the
conductive ground plane after a feedline portion of the second
transmission line feed network crosses over the rectangular
aperture body of the second aperture of each pair of the plurality
of pairs of apertures.
[0030] The plurality of conductive patch elements may be arranged
in an array with a plurality of rows wherein each row comprises at
least one conductive patch element. The plurality of conductive
patch elements may be arranged in an array with a plurality of rows
and a plurality of columns wherein each row comprises a number of
conductive patch elements equal to the number of columns. The
number of columns may be equal to two.
[0031] A first feedline portion of the first transmission line feed
network may cross over a rectangular aperture body of the first
aperture of each pair of the plurality of pairs of apertures in a
first direction for each first aperture that excites each
respective resonant radiating cavity formed between each respective
one of the plurality of conductive patch elements and the
conductive ground plane for conductive patch elements may be
arranged in a first column and a second feedline portion of the
first transmission line feed network may cross over the rectangular
aperture body of the first aperture of each pair of the plurality
of pairs of apertures in a second direction for each first aperture
that excites each respective resonant radiating cavity formed
between each respective one of the plurality of conductive patch
elements and the conductive ground plane for conductive patch
elements arranged in a second column, and the second direction may
be opposite to the first direction.
[0032] The second feedline portion may be electrically longer than
the first feedline portion by a distance equivalent to 180 degrees
in phase at a target operating frequency for the antenna
assembly.
[0033] A third feedline portion of the second transmission line
feed network may cross over a rectangular aperture body of the
second aperture of each pair of the plurality of pairs of apertures
in a third direction for each second aperture that excites each
respective resonant radiating cavity formed between each respective
one of the plurality of conductive patch elements and the
conductive ground plane for conductive patch elements arranged in
the first column and a fourth feedline portion of the second
transmission line feed network may cross over the rectangular
aperture body of the second aperture of each pair of the plurality
of pairs of apertures in a fourth direction for each second
aperture that excites each respective resonant radiating cavity
formed between each respective one of the plurality of conductive
patch elements and the conductive ground plane for conductive patch
elements arranged in the second column, and the third direction may
be the same as the fourth direction.
[0034] The third feedline portion may be equivalent in electrical
length to the fourth feedline portion. Each of the first
transmission line feed network and the second transmission line
feed network may include at least one meandering line portion. Each
meandering line portion may include one or more bends, and wherein
an electrical length of each meandering line portion may match a
group delay from the respective first or second feed point to at
least one of the respective first or second apertures with that of
another group delay from the respective first or second feed point
to at least one other of the respective first or second
apertures.
[0035] Each of the first transmission line feed network and the
second transmission line feed network may include at least one
tunable element. An input signal applied to at least one tunable
element may adjust at least one characteristic of the antenna
assembly, said characteristic being at least one selected from the
group consisting of a far-field radiation pattern, a coupling
between the first feed point and the second feed point, and a
coupling to one or more nearby antennas.
[0036] According to another aspect of the invention, an antenna
assembly is provided that includes a plurality of first substrates
each comprising a unitary dipole antenna element, wherein each
unitary dipole antenna element comprises a first pair of dipole
branches, a first coplanar feed line pair and a first conductor
connection substrate tab; a second substrate comprising a plurality
of coplanar dipole antenna elements, wherein each coplanar dipole
antenna element comprises a second pair of dipole branches, a
second coplanar feed line pair and a second conductor connection
substrate tab; and a third substrate comprising a plurality of
conductor connection cutouts, a first layer and a second layer,
wherein the first layer comprises a conductive plane with a
plurality of conductor connection clearances and wherein the second
layer comprises a first transmission line feed network and a second
transmission line feed network; wherein the second substrate is
orthogonal to each of the plurality of first substrates and wherein
the third substrate is orthogonal to the second substrate and each
of the plurality of first substrates; wherein the first
transmission line feed network conductively connects to each
respective unitary dipole antenna element via its respective first
coplanar feed line pair at a respective one of a plurality of first
conductive junctions, each said first conductive junction
comprising the respective first conductor connection substrate tab,
a first corresponding one of the plurality of conductor connection
cutouts, and a first corresponding one of the plurality of
conductor connection clearances; and wherein the second
transmission line feed network conductively connects to each
respective coplanar dipole antenna element via its respective
second coplanar feed line pair at a respective one of a plurality
of second conductive junctions, each said second conductive
junction comprising the respective second conductor connection
substrate tab, a second corresponding one of the plurality of
conductor connection cutouts, and a second corresponding one of the
plurality of conductor connection clearances.
[0037] The first pair of dipole branches of each unitary dipole
antenna element may be located on a same surface as the first
coplanar feed line pair. Each unitary dipole antenna element may
further include a first pair of parasitic elements. The first pair
of parasitic elements of each unitary dipole antenna element may be
located on the same surface as the first pair of dipole
branches.
[0038] The first pair of parasitic elements may broaden a radiation
pattern of each unitary dipole antenna element in a plane of the
same surface as the first pair of dipole branches. The first pair
of parasitic elements may include half-wavelength resonant dipole
elements at a target operating frequency of the antenna assembly.
The first pair of parasitic elements may be asymmetrically offset
relative to an axis of the respective first pair of dipole branches
towards an end of the respective first substrate having the
respective first conductor connection substrate tab.
[0039] Each of the plurality of first substrates may further
include a first assembly slot and the second substrate may further
include a plurality of second assembly slots. A respective one of
the plurality of second assembly slots may align with a respective
first assembly slot within each respective first substrate.
[0040] Each of the plurality of first substrates further include
one or more first mechanical tabs. The third substrate may further
include additional cutouts, each additional cutout corresponding to
a respective first mechanical tab amongst the plurality of first
substrates.
[0041] Each of the plurality of first substrates may further
include one or more first metalized pads corresponding to
respective ones of each first mechanical tab. The second layer of
the third substrate may further include a plurality of third
metalized pads corresponding to respective ones of each first
mechanical tab. Each first metalized pad may adjoin a respective
third metalized pad.
[0042] The second substrate may further include one or more second
mechanical tabs. The third substrate may further include additional
cutouts, each additional cutout corresponding to a respective
second mechanical tab.
[0043] The second substrate may further include one or more second
metalized pads corresponding to respective ones of each second
mechanical tab. The second layer of the third substrate may further
include a plurality of third metalized pads corresponding to
respective ones of each second mechanical tab. Each second
metalized pad may adjoin a respective third metalized pad.
[0044] Each of the plurality of conductor connection clearances may
be asymmetrically offset relative to a respective one of the
plurality of conductor connection cutouts. The asymmetric offset
may center each of the plurality of conductor connection clearances
relative to a projected intersection with the third substrate for a
respective one of first coplanar feed line pairs or second coplanar
feed line pairs.
[0045] The second substrate may be oriented such that each of the
plurality of coplanar dipole antenna elements radiates in a
vertical polarization far-field pattern and the plurality of first
substrates may be oriented such that each unitary dipole antenna
element radiates in a horizontal polarization far-field
pattern.
[0046] The first transmission line feed network may include a first
feed point, a first microstrip distribution portion, and a
plurality of first microstrip feed structure portions and the
second transmission line feed network may include a second feed
point, a second microstrip distribution portion, and a plurality of
second microstrip feed structure portions.
[0047] Each first microstrip feed structure portion may include a
first balun structure that couples a first pair of balanced
microstrip lines at a respective one of the plurality of first
conductive junctions to a first unbalanced microstrip line within
the first microstrip distribution portion and each second
microstrip feed structure portion may include a second balun
structure that couples a second pair of balanced microstrip lines
at a respective one of the plurality of second conductive junctions
to a second unbalanced microstrip line within the second microstrip
distribution portion.
[0048] Each of the first and second balun structures may include a
first microstrip line, a second microstrip line, and a T-junction,
and the second microstrip line may be electrically longer than the
first microstrip line by one half wavelength at a target operating
frequency of the antenna assembly and the second microstrip line
may include at least one additional bend than the first microstrip
line.
[0049] Each of the first and second microstrip lines may function
as an impedance transformer of an electrical length that is an
integer multiple of one quarter wavelength at a target operating
frequency of the antenna assembly.
[0050] Each of the first microstrip feed structure portion and the
second microstrip feed structure portion may further include an
impedance transformer from the T-junction within its respective
first or second balun structure to its respective first or second
unbalanced microstrip line within the respective first or second
microstrip distribution portion. The impedance transformer may
include an unbalanced microstrip line of an electrical length that
is an integer multiple of one quarter wavelength at a target
operating frequency of the antenna assembly.
[0051] The first feed point and the second feed point may each be
coupled to respective components on the second layer of the third
substrate. The respective components may be at least one of an RF
filter or a power amplifier within a transmitter.
[0052] The first microstrip distribution portion may equally divide
a first power and matches a first group delay from the first feed
point to each of the plurality of first microstrip feed structure
portions and the second microstrip distribution portion may equally
divide a second power and matches a second group delay from the
second feed point to each of the plurality of second microstrip
feed structure portions. Each of the first microstrip distribution
portion and the second microstrip distribution portion may include
at least one tunable element. An input signal applied to at least
one tunable element may adjust at least one characteristic of the
antenna assembly, said characteristic being one or more of a
far-field radiation pattern, a coupling between the first feed
point and the second feed point, or a coupling to one or more
nearby antennas.
[0053] A numerical count of unitary dipole antenna elements may
exceed that of a numerical count of coplanar dipole antenna
elements.
BRIEF DESCRIPTION OF THE DRAWINGS
[0054] The accompanying drawings, which are incorporated into and
constitute a part of this specification, illustrate one or more
examples of embodiments and, together with the description of
example embodiments, serve to explain the principles and
implementations of the embodiments.
[0055] FIG. 1 is an illustration of conventional point to point
(PTP) radios deployed for cellular base station backhaul with
unobstructed line of sight (LOS).
[0056] FIG. 2 is an illustration of intelligent backhaul radios
(IBRs) deployed for cellular base station backhaul with obstructed
LOS according to one embodiment of the invention.
[0057] FIG. 3 is a block diagram of an IBR according to one
embodiment of the invention.
[0058] FIG. 4 is a block diagram of an IBR antenna array according
to one embodiment of the invention.
[0059] FIG. 5A is an assembly view of an antenna assembly according
to one embodiment of the invention.
[0060] FIG. 5B is a side view of the antenna assembly according to
one embodiment of the invention.
[0061] FIG. 5C is an assembly view of an alternate embodiment of
the invention.
[0062] FIG. 6 is a view of the plurality of conductive patch
elements on the first substrate of the antenna assembly according
to one embodiment of the invention.
[0063] FIG. 7 is a view of the spacer laid over the plurality of
conductive patch elements on the first substrate of the antenna
assembly according to one embodiment of the invention.
[0064] FIG. 8A is a detailed view of both the first layer and the
second layer of the second substrate of the antenna assembly
according to one embodiment of the invention.
[0065] FIG. 8B is a detailed view of the transmission line feed
network portions near the apertures of the second substrate of the
antenna assembly according to one embodiment of the invention.
[0066] FIG. 8C is a detailed view of the first layer of the second
substrate of the antenna assembly according to one embodiment of
the invention.
[0067] FIG. 8D is a detailed view of the second layer of the second
substrate of the antenna assembly according to one embodiment of
the invention.
[0068] FIG. 8E is a detailed view of one of the plurality of
apertures within the first layer of the second substrate of the
antenna assembly according to one embodiment of the invention.
[0069] FIG. 8F is a view of the first and second substrates showing
how the plurality of pairs of apertures on the first layer of the
second substrate align with the plurality of conductive patch
elements on the first substrate according to one embodiment of the
invention.
[0070] FIG. 9 is a view showing the surface current of the second
substrate using the aperture feed arrangement according to one
embodiment of the invention.
[0071] FIG. 10A is a detailed view of a unitary dipole antenna
element for a dipole array antenna assembly according to one
embodiment of the invention.
[0072] FIG. 10B is a detailed view of a plurality of coplanar
dipole antenna elements for a dipole array antenna assembly
according to one embodiment of the invention.
[0073] FIG. 11A is a detailed view of a microstrip feed structure
portion for a dipole array antenna assembly according to one
embodiment of the invention.
[0074] FIG. 11B is a detailed view of an orthogonal interconnection
of substrates for a dipole array antenna assembly according to one
embodiment of the invention.
[0075] FIG. 12 is a schematic diagram of cascade impedances for a
dipole antenna array assembly according to one embodiment of the
invention.
[0076] FIG. 13A is an assembly view of a dipole array antenna
assembly according to one embodiment of the invention.
[0077] FIG. 13B is an alternative assembly view of a dipole array
antenna assembly according to one embodiment of the invention.
[0078] FIG. 14 is a detailed view of first and second layers of the
third substrate of a dipole array antenna assembly according to one
embodiment of the invention.
DETAILED DESCRIPTION
[0079] FIG. 2 illustrates deployment of intelligent backhaul radios
(IBRs) in accordance with an embodiment of the invention. As shown
in FIG. 2, the IBRs 200 are deployable at street level with
obstructions such as trees 204, hills 208, buildings 212, etc.
between them. The IBRs 200 are also deployable in configurations
that include point to multipoint (PMP), as shown in FIG. 2, as well
as point to point (PTP). In other words, each IBR 200 may
communicate with more than one other IBR 200.
[0080] For 3G and especially for 4.sup.th Generation (4G), cellular
network infrastructure is more commonly deployed using "microcells"
or "picocells." In this cellular network infrastructure, compact
base stations (eNodeBs) 216 are situated outdoors at street level.
When such eNodeBs 216 are unable to connect locally to optical
fiber or a copper wireline of sufficient data bandwidth, then a
wireless connection to a fiber "point of presence" (POP) requires
obstructed LOS capabilities, as described herein.
[0081] For example, as shown in FIG. 2, the IBRs 200 include an
Aggregation End IBR (AE-IBR) and Remote End IBRs (RE-IBRs). The
eNodeB 216 associated with the AE-IBR is typically connected
locally to the core network via a fiber POP 220. The RE-IBRs and
their associated eNodeBs 216 are typically not connected to the
core network via a wireline connection; instead, the RE-IBRs are
wirelessly connected to the core network via the AE-IBR. As shown
in FIG. 2, the wireless connections between the IBRs include
obstructions (i.e., there may be an obstructed LOS connection
between the RE-IBRs and the AE-IBR).
[0082] FIG. 3 illustrates an exemplary embodiment of the IBRs 200
shown in FIG. 2. In FIG. 3, the IBRs 200 include interfaces 304,
interface bridge 308, MAC 312, modem 324, channel MUX 328, RF 332,
which includes Tx1 . . . TxM 336 and Rx1 . . . RxN 340, antenna
array 348 (includes multiple antennas 352), a Radio Link Controller
(RLC) 356 and a Radio Resource Controller (RRC) 360. The IBR may
optionally include an Intelligent Backhaul Management System (IBMS)
agent as shown in FIG. 7 of U.S. patent application Ser. No.
13/645,472. It will be appreciated that the components and elements
of the IBRs may vary from that illustrated in FIG. 3. U.S. patent
application Ser. No. 13/645,472 and the related applications and
patents summarized above describe in detail the various elements of
the IBR including their structural and operational features in
numerous different embodiments both as depicted in FIG. 3 and as
depicted with various additional elements not shown in FIG. 3. A
brief summary of certain elements of the IBR is also provided
herein.
[0083] The external interfaces of the IBR (i.e., the IBR Interface
Bridge 308 on the wireline side and the IBR Antenna Array 348
(including antennas 352) on the wireless side) are a starting point
for describing some fundamental differences between the numerous
different embodiments of the IBR 200 and either conventional PTP
radios or other commonly known radio systems, such as those built
to existing standards including 802.11n (WiFi), 802.11ac (WiFi),
802.16e (WiMax) or 4G LTE.
[0084] In some embodiments, the IBR Interface Bridge 308 physically
interfaces to standards-based wired data networking interfaces 304
as Ethernet 1 through Ethernet P. "P" represents a number of
separate Ethernet interfaces over twisted-pair, coax or optical
fiber. The IBR Interface Bridge 308 can multiplex and buffer the P
Ethernet interfaces 304 with the IBR MAC 312. In exemplary
embodiments, the IBR Interface Bridge 308 preserves "Quality of
Service" (QoS) or "Class of Service" (CoS) prioritization as
indicated, for example, in IEEE 802.1q 3-bit Priority Code Point
(PCP) fields within the Ethernet frame headers, such that either
the IBR MAC 312 schedules such frames for transmission according to
policies configured within or communicated to the IBR 200, or the
IBR interface bridge 308 schedules the transfer of such frames to
the IBR MAC 312 such that the same net effect occurs. In other
embodiments, the IBR interface bridge 308 also forwards and
prioritizes the delivery of frames to or from another IBR over an
instant radio link based on Multiprotocol Label Switching (MPLS) or
Multiprotocol Label Switching Transport Profile (MPLS-TP). U.S.
patent application Ser. No. 13/645,472 provides additional
description of exemplary embodiments of the interfaces 304 and the
interface bridge 308 of the IBR 200. U.S. patent application Ser.
No. 13/271,051 provides additional description of exemplary
embodiments of an IBMS that includes an IBMS Agent in communication
with or IBMS components and the IBR Interface Bridge 308 as well as
MAC 312 and/or RRC 360. U.S. patent application Ser. No. 13/271,051
also describes an IBR with an integrated Carrier Ethernet
switch.
[0085] FIG. 4 illustrates an exemplary embodiment of an IBR Antenna
Array 348. FIG. 4 illustrates an antenna array having Q directive
gain antennas 352 (i.e., where the number of antennas is greater
than 1). In FIG. 4, the IBR Antenna Array 348 includes an IBR RF
Switch Fabric 412, RF interconnections 404, a set of Front-ends 408
and the directive gain antennas 352. The RF interconnections 404
can be, for example, circuit board traces and/or coaxial cables.
The RF interconnections 404 connect the IBR RF Switch Fabric 412
and the set of Front-ends 408. Each Front-end 408 is associated
with an individual directive gain antenna 352, numbered
consecutively from 1 to Q.
[0086] U.S. patent application Ser. No. 13/645,472, U.S. patent
application Ser. No. 13/898,429, and U.S. patent application Ser.
No. 14/108,200 provide additional description of the Front-end 408
and various embodiments thereof as applicable to different IBR
duplexing schemes such as Time Division Duplexing (TDD), Frequency
Division Duplexing (FDD) and Zero Division Duplexing (ZDD). For
example, with TDD embodiments where certain directive gain antenna
elements 352 are used for both transmit and receive at different
times, then Front-end 408 may include a transmit/receive switch,
one or more RF low pass and/or bandpass filters, and either a
low-noise amplifier (LNA) in the receive path or a power amplifier
(PA) in the transmit path. Similarly, with FDD embodiments where
certain directive gain antenna elements 352 are used for both
transmit and receive at the same time, then Front-end 408 may
include a duplex filter, one or more additional RF low pass and/or
bandpass filters, and either a low-noise amplifier (LNA) in the
receive path or a power amplifier (PA) in the transmit path.
Another common embodiment for FDD has certain directive gain
antenna elements 352 used only for transmit and then Front-end 408
for such transmit antenna elements would have a PA and one or more
RF filters for a transmit FDD sub-band and has certain directive
gain antenna elements 352 used only for receive and then Front-end
408 for such receive antenna elements would have an LNA and one or
more RF filters for a receive FDD sub-band. In most ZDD
embodiments, certain directive gain antenna elements 352 are used
only for transmit and others only for receive with respective
Front-ends as described for FDD except that the RF filters overlap
in the frequency domain for both transmit and receive (i.e. no
separate transmit and receive sub-bands).
[0087] Note that each antenna 352 has a directivity gain Gq. For
IBRs intended for fixed location street-level deployment with
obstructed LOS between IBRs, whether in PTP or PMP configurations,
each directive gain antenna 352 may use only moderate directivity
compared to antennas in conventional PTP systems at a comparable RF
transmission frequency. As described in greater detail in U.S.
patent application Ser. No. 13/645,472, U.S. patent application
Ser. No. 13/898,429, and U.S. patent application Ser. No.
14/108,200, typical values of Gq are on the order of 10 to 20 dBi
for each antenna at RF transmission frequencies below 10 GHz.
[0088] In the IBR Antenna Array 348, the total number of individual
antenna elements 352, Q, is at least greater than or equal to the
larger of the number of RF transmit chains 336, M, and the number
of RF receive chains 340, N. In some embodiments, some or all of
the antennas 352 may be split into pairs of polarization diverse
antenna elements realized by either two separate feeds to a
nominally single radiating element or by a pair of separate
orthogonally oriented radiating elements. In some embodiments,
certain antenna elements 352 may be configured with different
antenna gain Gq and/or radiation patterns compared to others in the
same IBR. Also, in many embodiments, such as for those employing
FDD or ZDD, U.S. patent application Ser. No. 13/645,472, U.S.
patent application Ser. No. 13/898,429, and U.S. patent application
Ser. No. 14/108,200 provide additional description of advantageous
arrangements of separate transmit and receive antenna subsets with
the total set Q of individual antenna elements 352.
[0089] The IBR RF Switch Fabric 412 provides selectable RF
connections between certain RF-Tx-m and/or certain RF-Rx-n to the
various individual antenna elements 352 via various front-end 408
embodiments. Note specifically that in certain embodiments the
individual antenna elements 352 are coupled via a transmit-only
front-end and/or the IBR RF Switch Fabric 412 to only a transmit
chain output RF-Tx-m or coupled via a receive-only front-end and/or
the IBR RF Switch Fabric 412 to only a receive chain output RF-Rx-n
to advantageously enable separate optimization of the receive
antenna array from that of the transmit antenna array. U.S. patent
application Ser. No. 13/645,472, U.S. patent application Ser. No.
13/898,429, and U.S. patent application Ser. No. 14/108,200 provide
additional description of different embodiments of the IBR RF
Switch Fabric 412 as applicable to TDD, FDD and ZDD in different
product configurations.
[0090] With reference back to FIG. 3, the IBR RF 332 also includes
transmit and receive chains 336, 340. In one embodiment, each
element of transmit chain 336 takes a transmit chain input signal
such as digital baseband quadrature signals I.sub.Tm and Q.sub.Tm
and then converts them to a transmit RF signal RF-Tx-m at an RF
carrier frequency typically below 10 GHz. Similarly, each element
of receive chain 340 converts a receive RF signal RF-Rx-n at an RF
carrier frequency typically below 10 GHz to a receive chain output
signal such as digital baseband quadrature signals I.sub.Rn and
Q.sub.Rn.
[0091] Other IBR elements include the IBR MAC 312, the Radio Link
Control (RLC) 356, the Radio Resource Control (RRC) 360 and the
optional IBMS Agent. Although IBR embodiments are possible wherein
the MAC 312, RLC 356, RRC 360 and the optional IBMS Agent are
distinct structural entities, more commonly IBRs are realized
wherein the MAC 312, RLC 356, RRC 360 and the optional IBMS Agent
as well as portions of the IBR Interface Bridge 308 are software
modules executing on one or more microprocessors. Note also that in
some IBR embodiments that use of a "Software Defined Radio" (SDR)
for the IBR Modem 324 and/or IBR Channel MUX 328 or portions
thereof may also be realized in software executing on one or more
microprocessors. Typically in SDR embodiments, the one or more
microprocessors used for elements of the PHY layer are physically
separate from those used for the MAC 312 or other layers and are
physically connected or connectable to certain hardware cores such
as FFTs, Viterbi decoders, DFEs, etc. As SDR processing power
increases over time, functions traditionally implemented in
hardware cores advantageously migrate to the SDR processor cores as
software modules for greater implementation flexibility.
[0092] The RRC 360 and RLC 356 interact with the IBR MAC 312 and
various elements of the IBR PHY both via "normal" frame transfers
and direct control signals via the conceptual IBR Control plane.
Both the RRC 360 and the RLC 356 may execute concurrent control
loops with the respective goals of optimizing radio resource
allocations and optimizing radio link parameters for current
resources in view of the dynamic propagation environment conditions
(including uncoordinated interference if applicable), IBR loading,
and possibly system-wide performance goals (via the optional IBMS
Agent or other IBR to IBR control communications links). It is
instructive to view the RLC 356 as an "inner loop" optimizing
performance to current policies and radio resource allocations for
each active link and to view the RRC 360 as an "outer loop"
determining if different policies or radio resource allocations are
desirable to meet overall performance goals for all IBRs currently
interacting with each other (intentionally or otherwise). Typically
both the RRC 360 and the RLC 356 are implemented as software
modules executing on one or more processors.
[0093] The primary responsibility of the RLC 356 in exemplary IBRs
is to set or cause to be set the current transmit Modulation and
Coding Scheme (MCS) and output power for each active link. The RLC
356 causes the transmit power control (TPC) of the IBR to be
maintained both in a relative sense amongst active links,
particularly of interest for the AE-IBR in a PMP configuration, and
also in an overall sense across all transmits chains and
antennas.
[0094] In some embodiments, the RLC 356 can determine its MCS and
TPC selections across active links based on information from
various sources within the IBR. For example, the IBR MAC can
deliver RLC control frames from other IBRs with information from
such other IBRs (for example, RSSI, decoder metrics, FCS failure
rates, etc.) that is useful in setting MCS and TPC at the
transmitting IBR. Additionally, such RLC control frames from an
associated IBR may directly request or demand that the RLC in the
instant IBR change its MCS and/or TPC values for transmit directly
on either a relative or absolute basis. U.S. patent application
Ser. No. 13/645,472 and U.S. patent application Ser. No. 14/108,200
provide additional description of different embodiments of the RLC
356 as applicable to TDD, FDD and ZDD in different product
configurations.
[0095] The primary responsibility of the RRC 360 is to set or cause
to be set at least the one or more active RF carrier frequencies,
the one or more active channel bandwidths, the choice of transmit
and receive channel equalization and multiplexing strategies, the
configuration and assignment of one or more modulated streams
amongst one of more modulator cores, the number of active transmit
and receive RF chains, and the selection of certain antenna
elements and their mappings to the various RF chains. Optionally,
the RRC may also set or cause to be set the superframe timing, the
cyclic prefix length, and/or the criteria by which blocks of
Training Pilots are inserted. The RRC 360 allocates portions of the
IBR operational resources, including time multiplexing of currently
selected resources, to the task of testing certain links between an
AE-IBR and one or more RE-IBRs. The RRC 360 evaluates such tests by
monitoring at least the same link quality metrics as used by the
RLC 656. Additionally, in some embodiments, additional RRC-specific
link testing metrics are also used. The RRC 360 can also exchange
control frames with a peer RRC at the other end of an instant link
to, for example, provide certain link testing metrics or request or
direct the peer RRC to obtain link specific testing metrics at the
other end of the instant link for communication back to RRC
360.
[0096] In some embodiments, the RRC 360 causes changes to current
resource assignments in response to tested alternatives based on
policies that are configured in the IBR and/or set by the optional
IBMS Agent. An exemplary policy includes selecting resources based
on link quality metrics predicted to allow the highest throughput
MCS settings at lowest TPC value. Additional exemplary policies may
factor in minimizing interference by the instant link to other
AE-IBR to RE-IBR links (or other radio channel users such as
conventional PTP radios) either detected at the instant IBRs or
known to exist at certain physical locations nearby as set in
configuration tables or communicated by the optional IBMS Agent or
other IBR to IBR control communications links as described, for
example, in co-pending U.S. patent application Ser. No. 14/098,456,
the entirety of which is hereby incorporated by reference. For
example, U.S. patent application Ser. No. 14/098,456 discloses
exemplary systems and methods for control communications links in
the form of inline or embedded signals that may be suitable for
exchange of control information between IBRs that otherwise lack
any IBR to IBR communication path. Such policies may also be
weighted proportionately to reach a blended optimum choice amongst
policy goals or ranked sequentially in importance.
[0097] In some embodiments, for either PTP or PMP deployment
configurations, the selection of either the one or more active RF
carrier frequencies used by the RF chains of the IBR RF, the one or
more active channel bandwidths used by the IBR MAC, IBR Modem, IBR
Channel MUX and IBR RF, the superframe timing, the cyclic prefix
length, or the insertion policy for blocks of Training Pilots is
determined at the AE-IBR for any given link. The RE-IBR in such an
arrangement can request, for example, an RF carrier frequency or
channel bandwidth change by the AE-IBR by sending an RRC control
frame in response to current link conditions at the RE-IBR and its
current RRC policies. Whether in response to such a request from
the RE-IBR or due to its own view of current link conditions and
its own RRC policies, an AE-IBR sends the affected RE-IBRs an RRC
control frame specifying at least the parameters for the new RF
frequency and/or channel bandwidth of the affected links as well as
a proposed time, such as a certain superframe sequence index, at
which the change-over will occur (or alternatively, denies the
request). The AE-IBR then makes the specified change after
receiving confirmation RRC control frames from the affected RE-IBRs
or sends a cancellation RRC control frame if such confirmations are
not received before the scheduled change.
[0098] An RE-IBR typically attempts to utilize all available
modulator and demodulator cores and streams as well as all
available RF chains to maximize the robustness of its link to a
particular AE-IBR. In an RE-IBR embodiment where at least some
redundancy in antenna elements amongst space, directionality,
orientation, polarization and/or RF chain mapping is desirable, the
primary local RRC decision is then to choose amongst these various
antenna options. In other embodiments the AE-IBR and RE-IBR
optimize their resource allocations independently such that there
is little distinction between the RRC strategies at the AE-IBR
versus the RE-IBR. U.S. patent application Ser. No. 13/645,472,
U.S. patent application Ser. No. 13/898,429, and U.S. patent
application Ser. No. 14/108,200 provide additional description of
different embodiments of the RRC 360 as applicable to TDD, FDD and
ZDD in different product configurations.
[0099] The specific details of the IBR Modem 324 and IBR Channel
MUX 328 depend somewhat on the specific modulation format(s)
deployed by the IBR. In general, the IBR requires a modulation
format suitable for a broadband channel subject to
frequency-selective fading and multipath self-interference due to
the desired PHY data rates and ranges in obstructed LOS propagation
environments. Many known modulation formats for such broadband
channels are possible for the IBR. Two such modulation formats for
the IBR are (1) Orthogonal Frequency Division Multiplexing (OFDM)
and (2) Single-Carrier Frequency Domain Equalization (SC-FDE). Both
modulation formats are well known, share common implementation
elements, and have various advantages and disadvantages relative to
each other. U.S. patent application Ser. No. 13/645,472 provides
additional detail regarding OFDM and SC-FDE as applicable to
various IBR embodiments.
[0100] The specific details of the IBR Modem 324 and IBR Channel
MUX 328 also depend somewhat on the specific antenna array signal
processing format(s) deployed by the IBR. In general, the IBR
utilizes multiple antennas and transmit and/or receive chains,
which can be utilized advantageously by several well-known baseband
signal processing techniques that exploit multipath broadband
channel propagation. Such techniques include Multiple-Input,
Multiple-Output (MIMO), MIMO Spatial Multiplexing (MIMO-SM),
beamforming (BF), maximal ratio combining (MRC), and Space Division
Multiple Access (SDMA). U.S. patent application Ser. No. 13/645,472
provides additional detail regarding such techniques as applicable
to various IBR embodiments.
[0101] In many embodiments, the IBR Modem 324 comprises one or
modulator cores each of which comprises such functional elements as
scramblers, encoders, interleavers, stream parsers, symbol groupers
and symbol mappers. At a high level, each modulator core within the
IBR Modem 324 typically transforms a data stream from the IBR MAC
312 into a symbol stream that can be passed to the IBR Channel MUX
328. Similarly, in many embodiments, the IBR Modem 324 also
comprises one or demodulator cores each of which comprises such
functional elements as descramblers, decoders, deinterleavers,
stream multiplexers, and soft decision symbol demappers. At a high
level, each demodulator core within the IBR Modem 324 typically
transforms a stream of estimated receive symbols, such as
represented by a Log-Likelihood Ratio (LLR), from the IBR Channel
MUX 328 into a data stream that can be passed to the IBR MAC 312.
U.S. patent application Ser. No. 13/645,472, U.S. patent
application Ser. No. 13/898,429, and U.S. patent application Ser.
No. 14/108,200 provide additional description of different
embodiments of the IBR Modem 324 as applicable to TDD, FDD and ZDD
in different product configurations.
[0102] In many embodiments, the IBR Channel MUX 328 comprises a
transmit path channel multiplexer that may or may not be frequency
selective and that in turn may comprise such functional elements as
block assemblers, transmit channel equalizers, transmit
multiplexers, cyclic prefix adders, block serializers, transmit
digital front ends, preamble inserters, and pilot inserters. At a
high level, the transmit path of the IBR Channel MUX 328 transforms
one or more symbol streams from the IBR Modem 324 into inputs for
the one or more transmit chains each comprised of baseband symbol
samples. Similarly, in many embodiments, the IBR Channel MUX 328
also comprises a frequency selective receive path channel
multiplexer that in turn may comprise that in turn comprises such
functional elements as synchronizers, receive digital front ends,
cyclic prefix removers, channel equalizer coefficients generators,
receive channel equalizers, receive stream multiplexers and complex
Discrete Fourier Transformers (DFT). At a high level, the receive
path of the IBR Channel MUX 328 transforms the outputs of the one
or more receive chains each comprised of baseband symbol samples
into one or more streams of estimated receive symbols for input
into the IBR Modem 324. U.S. patent application Ser. No.
13/645,472, U.S. patent application Ser. No. 13/898,429, and U.S.
patent application Ser. No. 14/108,200 provide additional
description of different embodiments of the IBR Channel MUX 328 as
applicable to TDD, FDD and ZDD in different product
configurations.
[0103] In exemplary embodiments, the IBR MAC 312 comprises such
functional elements as a management entity, a Tx buffer and
scheduler, a control entity, an Rx buffer, a frame check sum (FCS)
generator, a header generator, a header analyzer and an FCS
analyzer. U.S. patent application Ser. No. 13/645,472, U.S. patent
application Ser. No. 13/898,429, and U.S. patent application Ser.
No. 14/108,200 provide additional description of different
embodiments of the IBR MAC 312 as applicable to TDD, FDD and ZDD in
different product configurations.
[0104] Additional details regarding numerous optional functional
components and regarding additional exemplary embodiments of the
IBR are provided in commonly assigned U.S. patent application Ser.
No. 13/645,472, U.S. Pat. Nos. 8,311,023 and 8,238,318, U.S. patent
application Ser. No. 13/898,429 and U.S. Pat. No. 8,467,363, U.S.
patent application Ser. No. 13/271,051 and U.S. Pat. No. 8,300,590,
and U.S. patent application Ser. No. 14/108,200 and U.S. Pat. Nos.
8,638,839 and 8,422,540, the disclosures of which are hereby
incorporated herein by reference in their entirety.
[0105] Antenna assembles having improved feed mechanisms to address
these problems will now be described. The patch array antenna
assembly includes an array of resonant radiating patch antenna
elements that are aperture-fed (instead of pin or probe-fed). In
this improved antenna design, the feed network is coupled to the
resonant radiating cavity via apertures in the conductive ground
plane. In one embodiment, the feed network is composed of
transmission lines such as, for example, microstrip lines on one
side of a printed circuit board (PCB) wherein the conductive ground
plane is on the other side of the PCB. In another embodiment, the
feed network is composed of transmission lines such as, for
example, striplines within a multi-layer PCB wherein at least one
layer is a conductive ground plane that includes the apertures on
the outside of the PCB. Exemplary advantages of an aperture-fed
antenna array include, for example, lower cost due to reduced
assembly time/complexity and higher reliability due to no solder
joints securing any pins. The shape of the aperture also provides
distinct advantages over the prior art. In particular, by using an
aperture having a modified shape (e.g., rounded or tapered)
compared to a conventional art "H" or "dogbone" slot shape, the
wanted coupling to the resonant radiating cavity is maximized and
unwanted backwards facing radiation from is minimized. The use of a
multi-layer PCB and a stripline transmission line feed network
further minimizes the unwanted backwards facing radiation as well
as enables greater flexibility for placing active electronic
components on the opposite side of the PCB from the side with the
apertures in the conductive ground plane.
[0106] Another improvement is the termination of the feed line. In
conventional aperture-fed patch antennas the magnetic (inductive)
coupling between the transmission line feed and resonant radiating
cavity results in excessive inductive reactance at the antenna
feed. To counter this inductive reactance, the feed line is
commonly left open-stub after crossing the aperture to provide a
series capacitive reactance. This open-stub tuning has the
undesirable side effect of increasing backwards-facing radiation.
By co-optimizing the conductive patch element dimensions and
aperture dimensions, the feed can be tuned for zero-net reactance,
and a desired input resistance (e.g. 50 ohms, or 100 ohms). This
allows the feed line to be terminated in a short circuit
immediately after crossing the aperture, resulting in lower
backwards-facing radiation than the conventional art open circuit
counterpart.
[0107] In some embodiments, a dielectric spacer is provided between
the two PCBs to provide the desired spacing between the conductive
patch elements and conductive ground plane thereby forming a
resonant radiating cavity. In an exemplary embodiment, this spacer
can be a simple injection-molded plastic part. The spacer includes
one or more symmetric openings that remove any dielectric material
from within the resonant radiating cavity between the conductive
patch element and conductive ground plane. Dielectric material
within the resonant radiating cavity is undesirable because any
variation in the material dielectric constant can cause the
resonant frequency of the resonant radiating cavity to shift,
thereby reducing radiation efficiency. Additional details of this
patch array antenna assembly design will now be discussed with
reference to several Figures. The exemplary antenna assembly
embodiments described herein may be used in the IBR embodiments
described above and in the incorporated co-pending applications as
a pair of directive gain antenna elements for a facet comprising a
first directive gain antenna element with a first polarization and
a second directive gain antenna element with a second polarization
that is orthogonal to the first polarization.
[0108] FIG. 5A illustrates a patch array antenna assembly 500 in
accordance with some embodiments of the invention. A detailed view
of the side of the antenna assembly is shown in FIG. 5B.
[0109] As shown in FIG. 5A, the patch array antenna assembly 500
includes a first substrate 504, a spacer 508 and a second substrate
512. The first substrate 504, spacer 508 and second substrate 512
have approximately the same overall width and length and are
approximately aligned with one another.
[0110] In FIG. 5A, a plurality of rivets 516 are inserted into
openings 520 in the first substrate 504, spacer 508 and second
substrate 512 to hold the first substrate 504, spacer 508 and
second substrate 512 together. It will be appreciated that other
methods may be used to secure or hold the first substrate 504,
spacer 508 and second substrate 512 together. For example, rivets
having an alternative shape, different fasteners (e.g., screws,
bolts, clamps, etc.), adhesives or other methods known to those of
skill in the art may be used to hold the first substrate 504,
spacer 508 and second substrate 512 together. It will further be
appreciated that fewer than or more than the number of rivets 516
shown in FIG. 5A may be used to secure or hold the first substrate
504, spacer 508 and second substrate 512 together.
[0111] The first substrate 504 is typically a printed circuit board
(PCB). In some embodiments, the first substrate 504 is at least a
conductive patch element carrier. A number of conductive patch
elements may be located on one side of the first substrate 504. In
some embodiments, the plurality of conductive patch elements is
located on the surface of the first substrate 504 adjacent to the
spacer 508. The conductive patch elements, as will be described in
further detail hereinafter, define a number of resonant radiating
patch antenna elements.
[0112] The second substrate 512 is also typically a printed circuit
board (PCB). The second substrate 512 includes a transmission line
feed network for the patch array antenna assembly. The top outer
layer of the second substrate 512 may also be a conductive ground
plane 528 that, in combination with the conductive patch elements
of the first substrate, forms resonant radiating patch antenna
elements. As will be described in further detail herein, the
conductive ground plane 528 on the second substrate 512 includes a
plurality of apertures (or openings) for coupling the resonant
radiating cavities to the transmission line feed network on the
second substrate 512.
[0113] The spacer 508 is positioned between the first substrate 504
and the second substrate 512. The spacer 508 has a thickness that
is selected to maintain the proper height within the resonant
radiating cavity (said height being the distance between the
conductive ground plane 528 on the second substrate 512 and the
conductive patch elements on the first substrate 504 in some
embodiments) as shown in FIG. 5A. Alternatively, in other
embodiments where the conductive patch elements on the first
substrate 504 are on the opposite side of substrate 504 from the
surface adjacent to spacer 508 or on an inner layer of a
multi-layer PCB, the height within the resonant radiating cavity is
equal to the thickness of the spacer 508 plus the thickness of any
PCB layers between the spacer and the surface with the conductive
patch elements.
[0114] In FIG. 5A, the spacer 508 includes two openings 524A, 524B
where dielectric material is absent. The spacer 508 and the
openings 524, in particular, ensure there is no dielectric material
in the resonant radiating cavity, other than air, making the
structure more robust against variations in the electrical
properties of the spacer material. Although two openings 524A, 524B
are shown in FIG. 5A, it will be appreciated that the spacer 508
may include fewer than two or more than two openings 524. For
example, the spacer 508 may include one opening. In another
example, the spacer 508 may include three or more openings. In
another example, the number of openings in the spacer 508 may be
equal to the number of conductive patch elements on the first
substrate 504. Alternatively, the number of openings in the spacer
508 may be one quarter or one half the number or any other number
of conductive path elements on the first substrate 504. Further,
the shape of the openings may differ from that shown in FIG. 5A.
For example, if one opening is provided for each conductive patch
element of circular shape, then the spacer openings may also be
circular but larger in diameter than that of the circular
conductive patch element. In another example, if the shape of the
conductive patch element is non-circular and there is a one-to-one
correspondence of openings to conductive patch elements, the shape
of the opening may be the same shape as the conductive patch
element but also a larger dimension than the conductive patch
element.
[0115] In order to minimize the effect of the spacer electrical
properties on the antenna performance, the spacer opening should be
larger than the conductive patch element (or alternatively, the
spacer opening should extend beyond the projected area of the
conductive patch element) by at least the spacer thickness and
preferably the spacer opening should be larger than the conductive
patch element by two times the spacer thickness. Additional
differences in shape and size will be understood by one of skill in
the art and additional details regarding the spacer 508 will be
described hereinafter. This approach allows more flexibility in the
choice of spacer material than the conventional-art (i.e., a solid
spacer) because some variation in the dielectric parameters of the
spacer 508 can be tolerated with minimal effect on the patch array
antenna assembly. This also allows the use of less expensive
materials for fabrication of the spacer 508 than the
conventional-art (i.e., a solid spacer).
[0116] FIG. 5C shows an exemplary patch antenna assembly that
shares the same construction features as shown in FIGS. 5A and 5B,
but with the addition of a second instance of spacer 508 and a
third substrate 532. This assembly interleaves two spacers, 508A
and 508B, between a first substrate 504, a third substrate 532, and
a second substrate 512. Third substrate 532 is a conductive patch
element carrier, having a plurality of conductive patch elements
located on either or both of its two surfaces. In some embodiments,
the number of conductive patch elements on the third substrate 532
is equal to the number of conductive patch elements on the first
substrate 504, and the plurality of conductive patch elements on
the third substrate 532 are concentric with the respective ones of
the plurality of conductive patch elements on the first substrate
504 (or alternatively, coincident with a projection of the
plurality of conductive patch elements on the first substrate 504).
In some embodiments, the conductive patch elements on the first
substrate 504 are of a larger area (or diameter if circular) than
the conductive patch elements on the third substrate 532. In an
embodiment with circular conductive patch elements, the diameter of
the conductive patch elements on the first substrate 504 is
typically 10% larger than the diameter of the conductive patch
elements on the third substrate 532.
[0117] The conductive patch elements on the first substrate 504,
the conductive patch elements on the third substrate 532, and
conductive ground plane 528 on the second substrate 512 form
stacked resonant radiating patch antenna elements. A stacked
resonant radiating patch antenna element provides wider bandwidth
than can be achieved with radiating patch antenna elements
comprising a single conductive patch element. Typically, a
conventional, single conductive patch element can achieve a
resonant radiating patch antenna element with an impedance
bandwidth of about 5% of the target operating frequency, whereas a
stacked resonant radiating patch antenna element, as described
herein, can achieve an impedance bandwidth of up to 20% of the
target operating frequency.
[0118] In some embodiments, one or both of first substrate 504 and
third substrate 532 are formed of a dielectric film material, such
as, for example, polyimide. It will be appreciated that other
dielectric film materials may used to form the first substrate 504
and/or third substrate 532. The conductive patch elements may be
formed by a copper deposition process, such as, for example, copper
sputtering. It will be appreciated that alternative methods may be
used to form the conductive patch elements. Dielectric film
materials are advantageous because they can be used to form a very
thin substrate (e.g., 0.1 mm), which, in turn, minimizes
undesirable dielectric loading of the resonant radiating cavities
by the substrate material. In other embodiments, one or both of the
first substrate 504 and third substrate 532 are formed from a
printed circuit board. In yet other embodiments, one of either
first substrate 504 or third substrate 532 is formed from a printed
circuit board while the other is formed from a dielectric film
material. The exemplary aperture feeding techniques disclosed below
are compatible with this combination three-substrate and two-spacer
assembly. In some embodiments, the conductive patch elements 604
are etched in the first substrate 504 using known techniques and
known materials. In one embodiment, the conductive patch elements
604 are formed from a metal such as etched copper elements.
[0119] FIG. 6 shows a bottom view of the first substrate 504. As
shown in FIG. 6, the first substrate 504 includes a surface 600 on
which conductive patch elements 604 are located. When assembled,
the conductive patch elements 604 on the first substrate 504 and
the conductive ground plane 528 on the second substrate 512 form
resonant radiating patch antenna elements that can be excited by
aperture feeds as described herein. These resonant radiating patch
antenna elements support simultaneous excitation in multiple,
orthogonal, electromagnetic modes that can correspond to vertical
polarization and horizontal polarization or dual-slant 45-degree
polarization for the respective far-field directive gain antenna
patterns. With the aperture design as described herein, the
coupling between the orthogonal modes can be very low.
[0120] In some embodiments, the combination of the feed network and
the plurality of resonant radiating patch antenna elements form a
phased array. In a phased array embodiment, the resonant radiating
patch antenna elements are excited in a specific relative phase and
amplitude to attain different performance characteristics than what
is realizable from a single resonant radiating patch antenna
element. Exemplary performance characteristics of a phased array
antenna assembly are higher far-field pattern gain, improved
spatial selectivity, and increased control over coupling to nearby
antennas as also described in greater detail in U.S. patent
application Ser. No. 13/898,429 and U.S. Pat. No. 8,467,363, the
disclosures of which are hereby incorporated herein by reference in
their entirety.
[0121] In the embodiment depicted in FIG. 6, the conductive patch
elements 604 are arranged in an array of four rows with two columns
where each row/column combination corresponds to one conductive
patch element 604 within an array. In the embodiment of FIG. 6, the
four rows of conductive patch elements 604 in the array are
arranged such that the aperture feeds described herein can provide
two orthogonal directive gain antenna elements of a patch array
antenna assembly each with increased directive gain in the
elevation pattern compared to an antenna formed by a single row of
one or more resonant radiating patch antenna elements. Similarly,
the two columns of conductive patch elements 604 in the array are
arranged such that the aperture feeds described herein can provide
two orthogonal directive gain antenna elements of an antenna
assembly each with increased directive gain in the azimuthal
pattern compared to an antenna formed by a single column of one or
more resonant radiating patch antenna elements. Other array
embodiments (not shown) may have only a single column of conductive
patch elements 604 or may have more than two columns of conductive
patch elements 604 or may have either more or less than four rows
of conductive patch elements 604. It will be further appreciated
that the first substrate 504 may include fewer than eight or more
than eight conductive patch elements 604.
[0122] In the embodiment depicted in FIG. 6, the conductive patch
elements 604 are located on the bottom surface (that is, the
surface adjacent to the spacer when assembled) of the first
substrate 504 so that, when assembled, the material that forms the
spacer 508 is not within the resonant radiating cavity formed by
the conductive patch elements 604 and the conductive ground plane
528. Those skilled in the art will recognize that the conductive
patch elements 604 can alternately be located on the top surface of
the first substrate 504. This arrangement is less desirable, since
the dielectric parameters of the first substrate 504 will more
heavily influence the resonant frequency of the resonant radiating
cavity, or hence each resonant radiating patch antenna element.
This dielectric loading can be accounted for in design of the
conductive patch element shape and size, but in practice,
variability in the dielectric parameters of the first substrate
will then cause undesirable variability in resonant frequency of
the resonant radiating patch antenna element.
[0123] FIG. 7 illustrates an exemplary arrangement of the spacer
508 relative to the bottom surface of the first substrate 504. As
shown in FIG. 7, the openings 524 in the spacer 508 are designed to
coordinate with the conductive patch elements 604. In particular,
the opening 524A is aligned with four of the patch elements 604,
and opening 524B is aligned with the other four patch elements 604
on the bottom surface of the first substrate 504. As explained
above, each of the openings 524A, 524B is designed to ensure the
spacer does not significantly impinge into the resonant radiating
cavity formed between the conductive patch element and the
conductive ground plane. For example, in a specific embodiment
optimized for operation at 5.3 GHz, the patch elements have a
radius of 12.42 mm, the spacer thickness is 2.4 mm and the spacer
opening is larger than the projected area of each conductive patch
element by at least 5.08 mm.
[0124] As shown in FIG. 7, the openings are shown having a
rectangular shape having rounded edges. However, it will be
appreciated that alternative shapes may be used. For example, the
openings may be circular, the radius of the curve may be less than
or greater than that shown in FIG. 7, the openings may be
rectangular, etc., as understood by those of skill in the art.
[0125] In one embodiment, the openings 524 in the spacer 508 are
designed for multiple antenna assemblies that operate at differing
target operating frequencies. As understood by those of skill in
the art, the dimensions and relative position of the conductive
patch elements differ depending on the desired target operating
frequency of the antenna assembly. By correctly sizing the openings
in the spacer, one spacer may be used with antenna assemblies for
these differing operating frequencies. For example, the same spacer
508 may be used with a substrate 504 that is configured for a 5.3
GHz target operating frequency, a 5.6 GHz target operating
frequency and a 5.8 GHz target operating frequency. In this
example, setting the spacer opening dimensions sufficiently large
enough to cause negligible antenna performance variation effects to
the 5.3 GHz conductive patch element, as described above, also
causes the spacer opening to be large enough for the 5.6 GHz and
5.8 GHz optimized designs.
[0126] FIG. 8A is a view of the second substrate 512 illustrating
both a first layer or "top surface" that comprises a plurality of
pairs of apertures 820, 824 being repeated across the conductive
ground plane 528 and a second layer or "bottom surface" that
comprises transmission line feed networks such as 804A and 804B. As
shown in FIG. 8A, the second substrate 512 includes microstrip
transmission line feed networks 804A and 804B each with respective
feed points 836A and 836B. As understood by those of skill in the
art, these feed networks might also be realized as stripline
transmission line structures if the second substrate were a
multi-layered printed circuit board with more than two layers and
multiple ground planes. In FIG. 8A, radio transceiver electronics
component placement patterns 808a and 808b are co-located with the
microstrip feed networks 804 on the bottom side of the printed
circuit board substrate 512 which in this embodiment corresponds to
the second layer. The co-location of radio transceiver components
on the same substrate as the feed network provides a very short,
and in turn low-loss, interconnect between the feed points 836A and
836B and the rest of the radio components. For example, co-locating
a low noise amplifier and preferably also at least one RF bandpass
filter on the same substrate as the feed network is advantageous
because this increases the amount of loss in cables that may be
part of the coupling between directive gain antenna elements and
receive RF chains that can be tolerated without any degradation in
the radio link performance.
[0127] In some embodiments, electronic components may be located
within the feed networks 804. The integration of electronic
components for tunable elements such as tunable capacitors in
series or shunt with lumped element or distributed circuit elements
within the feed network allows dynamic adjustment of the
characteristics of the patch array antenna assembly, such as
far-field radiation patterns, cross-coupling between the orthogonal
polarizations as measured at their feed points, or coupling to
nearby antennas, in order to optimize a desired radio link metric,
such as a signal to noise and/or interference ratio or such as a
degree of isolation between an directive gain antenna element used
for transmit and another used for receive under full duplex
operation conditions. In an exemplary IBR embodiment, the RRC 360
(or some other controller such as a ZDD Canceller Loop Coefficients
Generator described in U.S. patent application Ser. No. 14/108,200
and U.S. Pat. Nos. 8,638,839 and 8,422,540, the disclosures of
which are hereby incorporated herein by reference in their
entirety) provides or causes to be provided an input signal to the
tunable element so that the antenna assembly characteristic is
adjusted according to the desired metric.
[0128] FIG. 8A shows a plurality of pairs of apertures 820, 824
being repeated across the conductive ground plane 528 on the top
surface, or first layer, of the second substrate in a pattern that
matches the distribution of conductive patch elements on the first
substrate, as shown within an array in FIG. 7. Preferably, the
apertures are divided into pairs 820, 824. Each pair of apertures
includes a first aperture 820 in a first direction, and a second
aperture 824 in a second direction that is orthogonal to the first
aperture 820. In some embodiments, each pair of apertures 820, 824
corresponds to one of the conductive patch elements 604 on the
first substrate 504. Each aperture 820 or 824 excites its
corresponding resonant radiating patch antenna element in a single
respective electromagnetic mode, and the respective modes are then
orthogonal to each other in an electromagnetic sense. The two
apertures in each pair of apertures are arranged such that the end
point of a first aperture 824 is aligned with the mid-point of a
second aperture 820, in a T-shape. This T-shape arrangement
achieves a very low coupling between the two orthogonal apertures
as compared with the more common L-shape arrangement of the
conventional art where the two apertures are aligned at one
endpoint. Such desirable lower coupling occurs at least because the
relative excitation of the T-shape arrangement is in common mode
which causes a cancellation effect to the other aperture.
[0129] As shown in FIG. 8A, the left hand column of vertical
apertures 824 is fed from the right-hand side. This means that each
respective feedline portions 812A to 812D overlap each aperture by
crossing over a rectangular aperture body for each respective left
hand vertical aperture 824 from a center point on the right of the
rectangular aperture body to a termination point 828 on the left of
the rectangular aperture body, as also illustrated in FIG. 8B.
Similarly, the right hand column of vertical apertures 824 is fed
from the left-hand side. Thus, in some embodiments, the vertical
apertures depicted in FIG. 8A for excitation of the resonant
radiating patch antenna elements set by the array of conductive
patch elements depicted in FIG. 6 are fed in an opposite direction
for the first or left hand column of respective pairs of apertures
and conductive patch elements from the direction for the second or
right hand column of respective pairs of apertures and conductive
patch elements. The feedline portions 812A through 812D must be
offset by an electrical length of 90 degrees towards one column to
phase align the electrical modes excited in the two columns of
conductive patch elements in order to achieve the desired array
properties. The feedline portions 812A to 812D are shown with an
offset of 90 degrees equivalent electrical length at the target
operating frequency to the left-hand side of the "center" of each
of the feedline portions 812A to 812D, but could also be offset by
90 degrees to the right hand side. This feed arrangement is
advantageous because it minimizes the space needed on the substrate
512 so that more feeds, apertures and conductive patch element
combinations can be included in the patch array antenna assembly,
thereby increasing the potential array gain that can be realized
from a given substrate size.
[0130] As shown in FIG. 8A, one or more of the feed lines may also
include a meandering line portion 832. The meandering line portion
832 may include one or more bends so that the physical distance or
electrical length (and hence the group delay) is the same from each
common feed point 836A or 836B to each respective aperture 824 or
820 in the array structure Conventional art approaches, such as
series feeding techniques, match only the relative phase of each
aperture--not the group delay, like the feeding technique
illustrated in FIG. 8A. The matched group delay approach (e.g.,
using the feed line having a meandering line portion shown in FIG.
8A) is advantageous because the resulting feed network maintains
proper phase excitation of the array of resonant radiating patch
antenna elements independently of frequency, thereby resulting in a
very broadband feed structure.
[0131] Additionally, as shown in FIG. 8A, the feed lines 804 are
terminated in a short circuit. By tuning the aperture width as
described herein, the feed lines 804A and 804B can be terminated in
a short circuit--rather than an open stub as in the conventional
art--using the vias 828. The vias 828 extend from the feed network
804A and 804B on one surface of the second substrate 512 (or
optionally, if stripline, from a feed network on an inner layer)
and through to the ground plane to the opposing surface of the
second substrate 512 to form a short circuit termination, thereby
realizing the advantage of lower backwards facing radiation as
described above.
[0132] FIG. 8B further illustrates the offset arrangement of the
feed network. As shown in FIG. 8B, horizontal apertures 820 are fed
by a feedline portion 852, which splits into two feedline portions
856 and 860. In FIG. 8B, feedline portions 856 and 860 are the same
distance from the 3-way T-junction joining feedline portions 852,
856 and 860, and the resulting symmetric structure ensures
horizontal apertures 820 are excited by feedline portions that
overlap the rectangular aperture body by overlapping (via crossing
over) the rectangular aperture body in the same direction with the
same relative phase angle as is desirable for the array performance
in this exemplary embodiment. As shown in FIG. 8B, vertical
apertures 824 are fed by a feedline portion 848 which splits into
feedline portions 840, 844. In FIG. 8B, the feedline portion 844 is
electrically longer than the feedline portion 840 by a distance
equivalent to 180 degrees in phase to correct for the phase offset
of the mirrored feeds (left-hand versus right-hand) as described
above.
[0133] FIG. 8B further illustrates the short circuit termination.
As shown in FIG. 8B, each of the feedline portions 856, 860, 840,
844 terminates with a via 828 that connects to the conductive
ground plane 528. As explained above, the vias 828 result in short
circuit terminations of each of the feedline portions.
[0134] In some embodiments, as shown in FIG. 8C, a first layer of
the second substrate 512 is located at the surface adjacent to
spacer. In FIG. 8C, the second substrate 512 includes a ground
plane with a plurality of apertures 820, 824. Other openings in the
conductive ground plane 528 are locations of drilled holes used for
assembly purposes and are not additional apertures within the
antenna assembly.
[0135] In some embodiments, as shown in FIG. 8D, a second layer is
located at the surface opposite to spacer 508 of the second
substrate 512, including first and second transmission line feed
networks 804A and 804B. In other embodiments that use a multi-layer
PCB, the portion of FIG. 8D showing the first and second
transmission line feed networks 804A and 804B may be located on an
inner layer between two ground planes using stripline structures
instead of microstrip.
[0136] FIG. 8E is a detailed view of the apertures 820, 824 that
are used to feed the resonant radiating patch antenna elements. As
shown in FIG. 8E, the aperture is defined by a rectangular aperture
body 864 and two aperture ends 868A and 868B. The aperture ends
868A and 868B are wider than the width of the rectangular aperture
body 864. In some embodiments, as shown in FIG. 8E, the edges of
the aperture ends 868 are tapered or rounded.
[0137] The rectangular aperture body 864 has an aperture body width
w that is chosen to be as narrow as can be reliably fabricated by
standard printed circuit board construction capabilities for the
selected PCB that forms the second substrate 512. Under current
processing capabilities, an aperture body width w of about as low
as 10 mils (0.38 mm) may be used, although smaller values are
already possible. However, it will be understood that as etching
process capabilities improve, the width of the aperture body w may
decrease. The aperture body width w may also be selected to other
widths as known to those of skill in the art. In order to minimize
rearward facing radiation and maximize coupling to the resonant
radiating cavity, the aperture may have a narrow aperture body
width, w, with wider openings at both aperture ends 868A and 868B.
In one embodiment, the aperture end width is more than five times
wider than the aperture body width. These aperture openings are
described further below.
[0138] In some embodiments, the shape of the aperture ends 868A and
868B consists of a combination of a rectangular end 872 and a
semi-circular end 876. The rectangular end 872 has a width t and a
length equal to the aperture end width W, and the semi-circular end
876 has a radius r. In some embodiments, the radius r of the
semi-circular end 876 is half of the aperture end width W. In some
embodiments, the aperture end width W is selected to be one third
of the aperture length L, and the width t of the rectangular end
872 is one third of the radius r of the semi-circular end 876 (or
hence one sixth of the aperture end width W). Thus, the shape of
the apertures is scalable according to a target operating frequency
for the antenna assembly using the relationships between the
aperture dimensions described in this paragraph (for example,
W=L/3, r=L/6, t=L/18), where a single variable, L, is scaled
proportionally to the desired frequency of operation.
[0139] In one particular embodiment, the aperture length L is 10.42
mm for a patch array antenna assembly operating at 5300 MHz. In
another particular embodiment, the aperture length L is 9.98 mm for
a patch array antenna assembly operating at 5600 MHz. In yet
another particular embodiment, the aperture length L is 9.7 mm for
a patch array antenna assembly operating at 5788 MHz. It will be
appreciated that the aperture may have different aperture lengths L
depending on, for example, the operating frequency, the thickness
of the spacer 508, and the size of the conducting patch element
604, as understood by those of skill in the art. It follows that
ability to dynamically alter the electrical size of the aperture,
whether by electrical or mechanical mechanism, allows dynamic
adjustment of the resonant frequency (or hence, the optimal
operating frequency) of the aperture feed structure and hence
performance of the antenna assembly.
[0140] In some embodiments, as shown in FIG. 8F, the projection of
the respective pairs of apertures 820 and 824 within the first
layer of the second substrate 512 located at the surface adjacent
to spacer 508 upon the outlines of the conductive patch elements
604 in the first substrate 504. As can be seen from FIG. 8F, the
apertures 820 and 824 do not need to be precisely centered relative
to each conductive patch element 604. Each of the apertures 820 and
824 has a long axis parallel to the long dimension of the aperture
and in the plane of the conductive ground plane, and a narrow axis
parallel to the narrow dimension of the aperture and in the plane
of the conductive ground plane. The magnetic fields for the
respective electromagnetic mode excited by an aperture are
approximately constant inside the resonant radiating cavity in the
dimension of the long axis, but vary significantly in the dimension
of the narrow axis with maximum in the center of the conductive
patch element and diminishing to near zero at the edges of the
conductive patch element. The aperture electromagnetic coupling is
primarily magnetic coupling (inductive). Moving the apertures along
the long axis does not significantly affect this magnetic coupling
because the magnetic fields within the resonant radiating cavity
are approximately constant in this direction. Moving the apertures
along the short axis significantly affects the magnetic coupling
mechanism because the magnetic fields vary in this direction.
Hence, the arrangement of apertures illustrated in FIG. 8F
co-optimizes the respective magnetic coupling to the resonant
radiating cavity in both apertures 820 and 824. Aperture 824 is
shifted significantly along its long axis, and aperture 820 is
shifted slightly along its narrow axis.
[0141] FIG. 9 illustrates a diagram showing the surface current
distribution at a pair of apertures 820, 824 showing additional
advantages of the aperture design according to embodiments of the
invention. In FIG. 9, feed lines 960 which terminate in a short
circuit 828 drive the aperture 820. As shown in FIG. 9, the current
distortion owing to the second orthogonal aperture 824 is minimized
by rounded edges 928A and 928B, thereby making the input impedance
of each aperture evenly balanced and improving the radiation
efficiency.
[0142] In an embodiment as described herein with four rows and two
columns of conductive patch elements operating at a target
frequency of 5300 MHz and aperture dimensions as described above, a
patch diameter is 28.5 mm, a conductive patch element thickness is
18 um, a spacer thickness is 2.4 mm, a first substrate thickness is
0.508 mm, a second substrate thickness is 0.762 mm, a center to
center column spacing is 34 mm, a center to center row spacing is
42.45 mm, outer dimensions for both a first substrate and second
substrate are 78 mm.times.185 mm, aperture 824 offset is 5.5 mm
from the center of the respective conductive patch element and
aperture 820 offset is 2.5 mm from the center of the respective
conductive patch element. This embodiment achieves a patch array
antenna assembly with port to port isolation of >35 dB across
the operating band of 5250 MHz to 5350 MHz, a vertically polarized
far-field radiation pattern with 16.3 dB gain, 16.1 degree vertical
beamwidth, 42 degree horizontal beamwidth, -0.8 dB radiation
efficiency, and a second horizontally polarized far-field radiation
pattern with 16.3 dB gain, 16.9 degree vertical beamwidth, 39
degree horizontal beamwidth, and -0.8 dB radiation efficiency.
[0143] FIGS. 10A illustrates an exemplary unitary dipole antenna
element 1000 having a driven coplanar dipole 1032 with dipole
branches 1032A and 1032B and respective parasitic elements 1004A
and 1004B. In one embodiment, each unitary dipole antenna element
1000 will be arranged as an array of such elements to provide a
horizontally polarized far-field directive gain antenna pattern.
The structure of driven coplanar dipole 1032 is referred to as
coplanar because both dipole branches 1032A and 1032B are located
on the same surface of substrate 1012 as coplanar feed line pair
1008, which serve as the feed transmission line. Parasitic elements
1004A and 1004B broaden the radiation pattern in the plane of the
surface of substrate 1012 having the driven coplanar dipole 1032
and the coplanar feed line pair 1008, and may or may not be on the
same side of the substrate as driven coplanar dipole 1032 and
coplanar feed line pair 1008.
[0144] Parasitic elements 1004A and 1004B are approximately
half-wavelength resonant dipole elements at the target operating
frequency. In some embodiments, these parasitic elements 1004A and
1004B are asymmetrically offset towards the conductor connection
end of substrate 1012 relative to the axis of driven coplanar
dipole 1032, as shown in FIG. 10A. This offset enhances the mutual
coupling between the driven coplanar dipole branches 1032A and
1032B and respective parasitic elements 1004A and 1004B. Driven
coplanar dipole 1032 is approximately a one-half wavelength
resonant dipole and features strong electric fields at the ends of
dipole branches 1032A and 1032B where respective parasitic elements
1004A and 1004B are located. As a result, the coupling mechanism
between driven coplanar dipole 1032 and parasitic elements 1004 is
primarily electric (capacitive), as opposed to magnetic
(inductive). The electric fields created by driven coplanar dipole
1032 are symmetric about the axis of dipole branches 1032A and
1032B. This relative offset between driven coplanar dipole 1032 and
parasitic elements 1004 causes the symmetric electrical fields of
dipole branches 1032A and 1032B to couple to a differential
electromagnetic mode in respective parasitic elements 1004A and
1004B. Half-wave dipoles resonate readily when excited via
differential-mode electromagnetic stimulus, but not to a
common-mode electromagnetic stimulus, and so the offset is
necessary to achieve adequate mutual coupling. Adjusting the length
of the parasitic elements 1004A and 1004B controls the relative
phase of the mutual coupling, and in turn the relative phase
between the electric current on driven coplanar dipole 1032 and the
electric current on parasitic elements 1004. Adjusting this
relative phase between these electric currents achieves the desired
far-field pattern. The input impedance of driven coplanar dipole
1032 can be tuned by adjusting the length and shape of dipole
branches 1032A and1032B, whilst maintaining a fixed relative
spacing to respective parasitic elements 1004A and 1004B.
[0145] Coplanar feed line pair 1008 connects the driven unitary
dipole antenna element 1000. Dipoles are a balanced antenna, and as
such are well suited to excitation by balanced transmission lines,
such as coplanar strips arranged as a coplanar feed line pair.
Coplanar feed line pair 1008 extends onto conductor connection
substrate tab 1016. The conductor connection substrate tab 1016 can
be inserted into a slot (or conductor connection cutout) on an
orthogonal backplane substrate, where the increased spacing 1020
between the branches of the coplanar feed line pair 1008
facilitates connection to another balanced transmission line
structure, such as coupled microstrip lines. In some embodiments,
each substrate 1012 having a unitary dipole antenna element 1000 is
repeated as individual elements in a dipole array antenna assembly,
wherein each individual unitary dipole antenna element 1000 is
coupled separately to a feed network on the orthogonal backplane
substrate. Additional features for mechanical fastening to
orthogonal substrates can be included, such as assembly slot 1024
and mechanical tabs 1028A and 1028B. Each mechanical tab 1028A or
1028B can also have one or more metalized pads 1034 as depicted in
FIG. 10A that can be on either or both surfaces of the substrate
1012 such that each mechanical tab aligns with an additional cutout
in an orthogonal substrate and each metalized pad 1034 adjoins
another metalized pad on the orthogonal substrate for soldering.
These orthogonal substrates may include, but are not limited to, an
orthogonal backplane substrate as described herein. Metalized pad
1022, typically located on the opposite surface of substrate 1012
from the surface comprising the coplanar feed line pair 1008, is an
exemplary feature for providing additional mechanical fastening.
The use of these mechanical fastening features will be illustrated
in further detail hereinafter.
[0146] In the embodiment depicted in FIG. 10A, a unitary dipole
antenna element for a target operating frequency of 5.66 GHz
comprises a driven coplanar dipole 1032 with overall length of 18
mm and outer width of 1.5 mm, and parasitic elements 1004 of length
of 16 mm and of outer width of 1.5 mm. The parasitic elements 1004
are located on the opposite side of substrate 1012 from the driven
coplanar dipole 1032 with rearward offset of 1.5 mm relative to the
centerline of the dipole branches 1032 and spacing of 11.05 mm from
the center of coplanar strips to center of parasitic elements.
Coplanar feed line pair 1008 have an inside edge-edge spacing of
0.3 mm and each a width of 1 mm. The distance from the axial
centerline of dipole branches 1032 to the bottom edge of substrate
1012, the plane of the orthogonal backplane substrate, is 20
mm.
[0147] FIG. 10B illustrates four coplanar dipole antenna elements
1060 each fed by a respective coplanar feed line pair 1056, and
arranged as a vertical array on common substrate 1036 to provide a
vertically polarized far-field directive gain antenna pattern. Each
coplanar dipole antenna element 1060 comprises dipole branches
1060A and 1060B. For each of coplanar dipole antenna elements 1060,
respective ones of coplanar feed line pair 1056 and dipole branches
1060A and 1060B are located on the same surface of substrate 1036,
forming a coplanar half-wavelength resonant dipole. The coplanar
feed line pair 1056 extends onto conductor connection substrate tab
1040. The conductor connection substrate tab 1040 can be inserted
into a slot (or conductor connection cutout) on an orthogonal
backplane substrate, where the increased spacing 1048 between the
branches of the coplanar feed line pair 1056 allows connection to
another balanced transmission line structure, such as coupled
microstrip lines. Additional features for mechanical fastening to
orthogonal substrates can be included such as assembly slots 1052A
to 1052E and mechanical tabs 1044A and 1044B. Each mechanical tab
1044A or 1044B can also have one or more metalized pads 1046A and
1046B as depicted in FIG. 10B that can be on either or both
surfaces of the substrate 1036 such that each mechanical tab aligns
with an additional cutout in an orthogonal substrate and each
metalized pad 1034 adjoins another metalized pad on the orthogonal
substrate for soldering. These orthogonal substrates may include,
but are not limited to, an orthogonal backplane substrate as
described herein. Respective metalized pads 1058, typically located
on the opposite surface of substrate 1036 from coplanar feed line
pair 1056, are an exemplary feature for providing additional
mechanical fastening. The use of these mechanical fastening
features will be illustrated in further detail hereinafter.
[0148] Coplanar dipoles antenna elements 1060 of FIG. 10B are all
located on the same surface of substrate 1036; however, in some
embodiments, one or more of coplanar dipole antenna elements 1060
may be located on one side of the substrate 1036 and the remainder
of the coplanar dipole antenna elements 1060 on the other side of
substrate 1036.
[0149] In the embodiment depicted in FIG. 10B, coplanar dipole
antenna elements 1060 for a target operating frequency of 5.66 GHz
have an overall length of 19.75 mm and outer width of 1.6 mm.
Coplanar feed line pair 1056 has an inside edge-to-edge spacing of
0.2 mm and each a width of 1 mm. The center-to-center spacing
between adjacent instances of coplanar dipole antenna elements 1060
is 32.46 mm. The distance from the axial centerline of dipole
branches 1060 to the left side of substrate 1036, the plane of the
orthogonal backplane substrate, is 21 mm.
[0150] FIG. 11A illustrates two instances of exemplary microstrip
feed structure portion 1100 of a transmission feed line network
that facilitates electrical interconnect between a balanced
element, such as one of unitary dipole antenna elements 1000 or
coplanar dipole antenna elements 1060, to a feed network on
orthogonal backplane substrate 1112. Two alternate orientations of
the exemplary microstrip feed structure portion, 1100A and 1100B,
are illustrated in FIG. 11A. In some embodiments, the backplane
substrate 1112 is a printed circuit board. Orthogonal backplane
substrate 1112 features a bottom outer layer (or first layer) that
is a conductive plane 1144, which fulfills several functions
including providing a reflective plane for unitary dipole antenna
elements 1000 and coplanar dipole antenna elements 1060, and also
providing a ground plane for the microstrip feed structure portion
1100.
[0151] Orthogonal backplane substrate 1112 contains a plurality of
conductor connection cutouts 1120 that are sized to accommodate
conductor connection substrate tabs 1040 and 1016. When conductor
connection substrate tab 1016, for example, is inserted into
conductor connection cutout 1120A, substrates 1012 and 1112 are
oriented orthogonally to each other. Three distinct connections can
be made between the substrates 1012 and 1112 using, for example,
solder fillet. These connections include (1) one of the conductors
that forms a branch of the coplanar feed line pair 1008 is
connected to microstrip line 1132A, (2) the other conductor from
the other branch of coplanar feed line pair 1008 is connected to
microstrip line 1108A, and (3) metalized pad 1022 is connected to
metalized pad 1136A for mechanical fastening (see also FIG. 11B).
Likewise, when conductor connection substrate tab 1040 is inserted
into conductor connection cutout 1120B that is rotated 90 degrees
relative to a conductor connection cutout 1120A, substrates 1036
and 1112 are oriented orthogonally to each other. Again, three
distinct connections between substrates 1036 and 1112 can be made
using, for example, solder fillet. These three connections include:
(1) one of the conductors that forms a branch of the coplanar feed
line pair 1056 is connected to microstrip line 1132B, (2) the other
conductor from the other branch of coplanar feed line pair 1056 is
connected to microstrip line 1108B, and (3) metalized pad 1058 is
connected to metalized pad 1136B for mechanical fastening.
[0152] Conductive plane 1144 also has a respective conductor
connection clearance 1116A and 1116B to reduce parasitic
capacitance between coplanar feed line pair 1008 or 1056 and the
conductive plane 1144 in the vicinity of the conductive junction
for each conductor connection cutout 1120. In one embodiment,
conductor connection clearance 1116 is asymmetrically offset from
conductor connection cutout 1120 as shown in FIG. 11A so as to be
centered, both horizontally and vertically in either orientation of
conductor connection cutout 1120, about a projected intersection of
the coplanar feed line pair 1008 or 1056 with the orthogonal
substrate instead of being centered about the conductor connection
substrate tabs 1016 or 1040 (or centered about the conductor
connection cutouts 1120). This centering further minimizes
parasitic capacitance between the coplanar feed line pair 1008 or
1056 and the conductive plane 1144. For example, at a target
operating frequency of 5.66 GHz, the size of the conductor
clearance is equivalent to the distance between conductive plane
1144 and the plane that comprises the transmission line feed
structures such as microstrip feed structure portions 1100.
[0153] In some embodiments, the microstrip feed structure portion
1100 includes balun elements to connect the balanced coplanar feed
line pairs 1008 and 1056 to the unbalanced microstrip lines 1104
and 1148 respectively. Microstrip lines 1104 and 1148 are both
located within respective transmission feed line networks.
Unbalanced microstrip lines are better suited for large parts of
the transmission feed line networks as they have fewer conductors
(and avoid cross-overs) compared to a balanced structure such as
coupled microstrip lines. However, at the actual electrical
connection point between the balanced coplanar feed line pair 1008
or 1056 and its respective transmission feed line network, the
conductive junction is preferably formed by a connection to a
separate balanced microstrip line from each branch of a coplanar
feed line pair as described above. Thus, microstrip feed structure
portion 1100 needs to at least include balanced microstrip lines at
the conductive junction and a balun structure that includes
impedance matching between such balanced microstrip lines and
unbalanced microstrip line 1104 within the transmission feed line
network. In some embodiments, the balun structure includes the
microstrip lines 1132 and 1108 and the T-junction 1124 as shown in
FIG. 11A. In other embodiments (not shown), discrete components may
be used instead to transform balanced microstrip lines at the
conductive junction to an unbalanced microstrip line 1104 within
the transmission feed line network as is known in the conventional
art.
[0154] In the embodiment of FIG. 11A, the electrical length of
microstrip line 1132 is 90 degrees (or 1/4 wavelength) at the
desired operating frequency, and the electrical length of
microstrip line 1108 is 270 degrees (or 3/4 wavelength) at the
desired operating frequency. The bends in 1108 provide for the
additional electrical length required (180 degrees or 1/2
wavelength) in a compact arrangement, and also provides a structure
that minimizes any distance where microstrip line 1108 runs
parallel to other parts of the same trace. Parallel lengths of
microstrip line increase undesired electrical coupling effects,
which reduces the effectiveness of the balun structure. The
T-junction 1124 provides a common connection to both the 90 degree
microstrip line 1132 and 270 degree microstrip line 1108.
[0155] The feed structure 1100 could alternatively be implemented
in a stripline structure as opposed to a microstrip structure shown
here if orthogonal backplane substrate 1112 is implemented as a
multi-layer PCB.
[0156] FIG. 11B illustrates a cross-sectional view of an exemplary
conductive junction between orthogonal substrates 1012 and 1112. In
some embodiments, as illustrated in FIG. 11B, the coplanar feed
line pair 1008, when assembled, can be easily soldered to the
microstrip feed structure portion 1100 to minimize overall feed
losses and the metalized pad 1022 can be easily soldered to the
metalized pad 1136 to provide additional mechanical ruggedness to
the overall assembly.
[0157] FIG. 12 illustrates an equivalent electrical circuit
representation of microstrip feed structure portion 1100 for one
embodiment. The load impedance, Z.sub.D is the differential
impedance of the coplanar feed line pair 1056 or 1008 at the plane
where the connection is made to microstrip lines 1132 and 1108. In
some embodiments, the driven unitary dipole antenna elements 1000
and driven coplanar dipole elements 1060 are tuned for an input
impedance of 100 ohms with the previously described spacing to
reflective plane 1144, and the dimensions of coplanar feed line
pair 1008 and 1056 are similarly chosen to achieve a characteristic
impedance of approximately 100 ohms. This tuning sets load
impedance Z.sub.D to approximately 100 ohms. The differential
impedance is split evenly between the microstrip lines 1132 and
1108 at the target operating frequency, such that each microstrip
line has an effective single-ended load impedance of about 50 ohms.
In this exemplary embodiment, each of the microstrip lines 1132 and
1108 has a characteristic impedance of approximately 65 ohms.
Microstrip lines 1132 and 1108, both provide for quarter-wavelength
impedance transformation and hence transform the 50 ohm load
impedance to approximately 65{circumflex over ( )}2/50=85 ohms,
shown as Z.sub.A in FIG. 12. The T-junction 1124 divides the
impedance Z.sub.A by a factor of 2, owing to parallel impedances,
making Z.sub.B equal to approximately 42 ohms. In some embodiments,
an additional microstrip line 1128 is included as a
quarter-wavelength impedance transformer, with characteristic
impedance of approximately 65 ohms and transforms the load
impedance Z.sub.B back to Z=100 ohms, which is convenient for
realizing transmission feed line networks for phased array
applications. This circuit analysis is reciprocal and applies
whether the antenna is receiving or transmitting a signal.
[0158] Constraining the microstrip lines 1132 and 1108 to integer
lengths of quarter-wavelengths as described above integrates an
impedance matching function within the balun. The characteristic
impedance of microstrip lines 1132 and 1108 can be set to obtain
the desired impedance transformation ratio. In some embodiments,
cascading a second quarter-wavelength microstrip line 1128 allows
the impedance transformation to be spread over multiple elements,
resulting in a more broadband structure than choosing to do the
required transformation in only a single element, and provides
flexibility to accommodate for variations in element impedance
without changing the other elements of the microstrip feed portion
1100.
[0159] When conductor connection substrate tabs 1040 and 1016 are
inserted into respective conductor connection cutouts 1120,
coplanar feed line pair 1008 or 1056 extends a short distance, such
as 1 mm in some embodiments, beyond microstrip lines 1108 and 1132.
This additional length is necessary to provide adequately large
surfaces for reliable solder joints on each branch of coplanar feed
line pair 1008 and 1056. This short length of open circuit
transmission line creates parasitic capacitance at the conductive
junction. The increased spacing 1048 and 1029 results in parasitic
inductance near the conductive junction. In one embodiment, the
length of the increased spacing 1048 and 1020 is optimized such
that the resulting parasitic inductance resonates with the open
stub parasitic capacitance, thereby adding no net additional
reactance at the conductive junction for the target operating
frequency. For example, at a target operating frequency of 5.66
GHz, the length of increased spacing may be 1.25 mm.
[0160] FIGS. 13 and 14 illustrate an exemplary dipole array antenna
assembly 1304 based on an orthogonal assembly of five first
substrates 1012A-E, a second substrate 1036, and a third substrate
1300. Second substrate 1036 contains four instances of coplanar
dipole antenna elements 1060A-1060D. Here, the second substrate is
used for vertical polarization if the entire antenna assembly 1304
is oriented in a backhaul radio, such as the IBR. The longer
dimension of assembly 1304 represents up/down and the smaller
dimension represents left/right. Each of the five first substrates
1012A-E contains a unitary dipole antenna element 1000, used here
for horizontal polarization given the antenna assembly orientation
described above. The third substrate 1300 is the orthogonal
backplane substrate 1112 described above with a plurality of
conductor connection cutouts 1120, as well as various mechanical
connection cutouts. In some embodiments, the third substrate 1300
is a multi-layer substrate having at least two layers. In an
exemplary embodiment, the first layer of third substrate 1300
comprises at least the conductive plane 1144 and the respective
conductor connection clearances 1116, as well as other
slots/clearances associated with mechanical tabs used for
mechanical assembly purposes. Also in an exemplary embodiment, the
second layer of third substrate 1300 comprises at least the
transmission line feed networks including respective microstrip
feed structure portions 1100 for each element in the overall array.
Thus, dipole array antenna assembly 1304 effectively interleaves
vertically and horizontally polarized antenna elements to create a
two-port, orthogonally polarized dipole array antenna assembly.
[0161] Although two-port, orthogonally polarized dipole array
antenna assemblies with crossed dipole elements have been disclosed
previously, these conventional antenna assemblies result in a
crossed dipole assembly that is more complex, costly, and prone to
failure than the novel interleaved array structure described as
well as lower performing in terms of antenna efficiency and
isolation. FIGS. 13A and 13B further illustrate the functionality
of assembly slots 1024 and 1052, which permit the orthogonal,
interleaved assembly of substrates 1012 and 1036. Upon assembly,
second substrate 1036 effectively captures, retains, and provides
additional lateral support for the multiple instances of first
substrate 1012. Likewise, the multiple instances of first substrate
1012 provide additional lateral support for substrate 1036. As
shown in FIG. 13A, each assembly slot 1052 aligns with a
corresponding assembly slot 1024 to set the spacing between
successive first substrates in the array and to set the orientation
of each first substrate 1012 as orthogonal to the second substrate
1036. In some embodiments, the tabs can be soldered, both for
mechanical retention and electrical connectivity, in a single
soldering process, to corresponding pads or conductive feed lines
on the second layer of the third substrate 1300.
[0162] The interleaved arrangement of the dipoles of opposite
polarity also achieves very low mutual coupling between the
elements of opposite polarity. This is because the symmetric
electric field of each element, couples in a common-mode fashion to
the dipole elements that are orthogonally polarized. As previously
discussed, half-wave dipoles to not resonate in response to a
common-mode excitation. There is significant mutual coupling
between the elements of a similar polarization, but this coupling,
which is deterministically known, can be minimized through proper
design of a feed network. It is important to minimize mutual
coupling between orthogonally polarized elements, so that the
resulting orthogonally polarized antenna arrays do not couple
significantly to each other. Mutual coupling between two arrays
will reduce the efficiency of each antenna, and also has been shown
to increase correlation between the two antennas, resulting in
degraded MIMO performance for a backhaul radio that uses such
antenna assemblies.
[0163] FIG. 14 shows exemplary details of the third substrate 1300
(or orthogonal backplane substrate), which can be realized as a
printed circuit board. Third substrate 1300 features a first layer
that is a conductive plane 1144, which fulfills several functions
including providing a reflective plane for unitary dipole antenna
elements 1000 and coplanar dipole antenna elements 1060, and also
providing a ground plane for the plurality of microstrip feed
structure portions 1100 and the microstrip distribution portions
1428A and 1428B of the first transmission line feed network 1408A
and the microstrip distribution portions 1428C of the second
transmission line feed network 1408B. Microstrip-based transmission
line feed networks 1408A and 1408B connect to unitary dipole
antenna elements 1000 and coplanar dipole antenna elements
1060A-1060D, respectively through a plurality of microstrip feed
structure portions 1100 as detailed previously in FIGS. 11A and
11B. Each microstrip-based transmission line feed network 1408A and
1408B also comprises a respective feed point 1404A and 1404B and
respective microstrip distribution portions 1428A through 1428C as
shown in FIG. 14. In some embodiments, additional components such
as filters and transmit power amplifiers or receive low noise
amplifiers may be located near the feed points 1404A and 1404B to
minimize losses and improve isolation performance in view of
interconnects such as cables from these dipole array antenna
assemblies to the rest of the radio.
[0164] FIG. 14 also shows additional cutouts 1416 and metalized
pads 1424 where the mechanical tabs 1028 and 1044 are fastened to
the third substrate preferably in a single soldering step with the
conductor connections to further increase mechanical rigidity and
reliability of the overall antenna assembly. FIG. 14 further
depicts that for one subset of five of the conductor connection
cutouts, each conductor connection cutout 1120 is arranged to align
with the conductor connection substrate tabs 1016 of the unitary
dipole antenna elements, and that for the other subset of four of
the conductor connection cutouts, each conductor connection cutout
1120 is arranged to align with the conductor connection substrate
tabs 1040 of the coplanar dipole antenna elements. It will be
appreciated that third substrate may have fewer than or more than
five conductor connection cutouts in the vertical array subset and
fewer than or more than four conductor connection elements in the
horizontal array subset.
[0165] Horizontally oriented dipoles tend to have a broader pattern
beamwidth in elevation than the vertically oriented dipoles. The
additional number of elements in the horizontal array compared to
the vertical array adds additional array factor gain, such that the
resulting elevation beamwidth of the two polarizations is similar.
FIG. 14 further illustrates the advantage of the compact size of
microstrip feed structure portion 1100, which permits both
horizontal and vertical dipole array element interleaving at
desirable element spacing, such as, for example 0.65 times the
free-space wavelength at the target operating frequency, while
leaving adequate room for the remaining transmission feed line
network interconnects.
[0166] The transmission line feed network 1408B is a corporate feed
network providing for uniform and matched group delay excitation of
the four vertically oriented coplanar dipole antenna elements 1060.
Those skilled in the art will recognize that uniform excitation is
commonly used to denote equal amplitude excitation amongst antenna
elements, and that the matched group delay excites the elements in
the same relative phase. Owing to the 2{circumflex over ( )}N
number of elements (here N=2, or hence 4 elements), matched group
delay from common feed point 1404B to each coplanar dipole antenna
element 1060, and uniform excitation of each coplanar dipole
antenna element 1060 is achieved via the symmetry of microstrip
distribution portions 1428C. In this exemplary embodiment, the
quarter-wavelength microstrip lines 1128 are adjusted differently
depending on whether the associated coplanar dipole element 1060 is
an outer antenna element in the array or an inner antenna element
in the array. This custom tuning is necessary to ensure the desired
uniform, phase-aligned excitation of the plurality of coplanar
dipole antenna elements 1060.
[0167] The transmission line feed network 1408A is also a uniformly
excited, matched group delay corporate feed network, but the uneven
number of unitary dipole elements 1000 necessitates the inclusion
of microstrip distribution portion 1428B which includes additional
bends and length necessary to achieve the desired matched group
delay from common feed point 1404A to each unitary dipole element
1000. The quarter-wavelength microstrip line 1128 that couples to
microstrip distribution portion 1428B is also uniquely shaped to
accommodate the bends of microstrip portion 1428B whilst also
providing tuning for uniform excitation. Open stub tuning feature
1432 corrects for parasitic effects introduced in the feed network
owing to the undesired coupling between parallel lengths of line
and undesired parasitic capacitance at bends in the microstrip
distribution portions 1428A and 1428B. This open stub tuning
feature 1432 further ensures the desired uniform, phase-aligned
excitation of the plurality of unitary dipole antenna elements
1000.
[0168] Uniform and phase-aligned excitation of antenna elements in
an array assembly is known to achieve the maximum realizable
far-field directive antenna gain pattern in the broadside
direction; however, in some embodiments, non-uniform, and/or
non-phase-aligned excitation may be preferable. Both the amplitude
and relative phase of each antenna element can be adjusted to
optimize some desired characteristic of the antenna array assembly,
such as side lobe levels, peak gain orientation, and isolation to
nearby antenna assemblies. These parameters may also be dynamically
adjustable by including tuning elements within microstrip
distribution portions 1428A, 1428B, and 1428C. In particular,
adaptive control of the phase, and or amplitude, of either each
individual antenna element, or a subset of antenna elements, can be
used to dynamically tune both near-field and far-field coupling to
an adjacently located antenna assembly to achieve maximum
port-to-port isolation between the two antenna assemblies.
[0169] In some applications it may be desirable to conform to an
EIRP (effective isotropic radiated power) elevation mask, and in
this case it is beneficial to shape the transmit antenna far-field
radiation pattern, either statically or dynamically, to minimize
the transmitted EIRP in the vertical direction. This shaping can be
achieved by altering the relative phase and/or the amplitude
excitation of each antenna element in the exemplary antenna array
assemblies.
[0170] One exemplary embodiment for achieving far-field radiation
pattern shaping is by tapering the amplitude excitement of the
individual antenna elements within the array assembly by some
pattern as a function of the location of the antenna element in the
array assembly. A tapered amplitude excitation, wherein the inner
antenna elements in an antenna array assembly are driven with
higher relative amplitude than the outer antenna elements, is known
to achieve lower far-field antenna pattern side lobe levels than
the equivalent array assembly with uniform excitation. Microstrip
lines 1140A to 1140B and 1128A to 1128B can be used to control the
input impedances of each element in the dipole array antenna
assembly as seen by transmission line feed networks 1408A and
1408B. This impedance can be adjusted to control the relative
amplitude excitation of the elements, providing the desired tapered
amplitude excitation, and in turn, the desired far-field antenna
radiation pattern side lobe suppression. Similarly, these amplitude
tapers can be applied to the aperture-fed patch element array
described herein.
[0171] Another exemplary embodiment for far-field antenna radiation
pattern shaping is via a progressive relative phase shift in the
relative excitation of each antenna element in the array antenna
assembly. A progressive relative phase shift between the antenna
elements of an antenna array assembly is known to scan the main
beam of the far-field antenna radiation pattern in an angular
sense. The lengths of microstrip line distribution portions 1428
can be adjusted to vary the relative phase excitation of each
element. Alternatively, electronic phase shifters can be inserted
into microstrip line distribution portions 1428 to dynamically vary
the individual antenna element relative phase excitation. The
ability to dynamically scan the main beam in a downward direction
can help conform to the EIRP elevation mask in response to changes
in elevation alignment of the IBR. Similarly, these progressive
relative phase shifts can be applied to the aperture-fed patch
element array described herein.
[0172] For example, if the IBR is installed with a tilt angle
upwards towards the sky then a sensor such as based on a multi-axis
accelerometer can determine the amount of upward tilt and then a
controller, such as the RRC, can provide or cause to be provided
certain control signals to the antenna array assembly, whether
based on aperture-fed patch elements or substrate tab connected
dipole elements or otherwise, so that the main beam is either
adjusted in a downward direction or has additional sidelobe
suppression applied, thereby either optimizing link performance
and/or conforming with a regulatory domain elevation mask EIRP
limit at a particular elevation angle.
[0173] In one embodiment, the use of progressive relative phase
shifts and/or amplitude tapers between either aperture-fed patch
elements or substrate tab connected dipole elements that can be
dynamically altered is applied based on a tilt sensor input to
ensure that the maximum EIRP above an upward elevation such as 30
degrees or higher is at least 13 dB lower than the maximum EIRP at
zero degrees elevation angle, or alternatively to ensure that the
maximum EIRP above an upward elevation such as 30 degrees or higher
is not greater than a prescribed limit such as +23 dBm. In other
embodiments, the IBR uses the tilt sensor input and the known
characteristics of a particular antenna assembly far-field
radiation pattern to limit the maximum conducted power into the
antenna assembly to ensure that the maximum EIRP above an upward
elevation such as 30 degrees or higher is not greater than a
prescribed limit such as +23 dBm.
[0174] In the example described herein with five elements in the
horizontal array and four elements in the vertical array operating
at a target frequency of 5660 MHz and with the dipole element
dimensions described above, the dipole array antenna assembly
achieves port-to-port isolation of more than 35 dB across the
operating band of 5470 MHz to 5850 MHz, a vertically polarized
far-field radiation pattern with 11.1 dB gain, 19.5 degree vertical
beamwidth, 120.1 degree horizontal beamwidth, -0.75 dB radiation
efficiency, and a second horizontally polarized far-field radiation
pattern with 11.5 dB gain, 16.7 degree vertical beamwidth, 122.7
degree horizontal beamwidth, and -0.75 dB radiation efficiency.
[0175] Numerous additional variations of the above-described
elements of the IBR and antennas can also be advantageously
utilized in substitution for or in combination with the exemplary
embodiments described above. For example, in certain embodiments
the aperture-fed patch array antenna assemblies are used as
directive gain antenna elements that can be coupled to receive RF
chains and the dipole array antenna assemblies are used as
directive gain antenna elements that can be coupled to transmit RF
chains. In other exemplary embodiments, the aperture-fed patch
array antenna assemblies are used as directive gain antenna
elements that can be coupled to both receive RF chains and transmit
RF chains, typically wherein a first subset of such antenna
assemblies is configured for receive usage and a second subset is
configured for transmit usage. When an aperture-fed patch array
antenna assembly is configured for transmit usage, active
components such as power amplifiers and filters may also be
integrated into such antenna assemblies preferably with minimal
loss between the feed points and the power amplifiers.
[0176] One or more of the methodologies or functions described
herein may be embodied in a computer-readable medium on which is
stored one or more sets of instructions (e.g., software). The
software may reside, completely or at least partially, within
memory and/or within a processor during execution thereof. The
software may further be transmitted or received over a network.
[0177] The term "computer-readable medium" should be taken to
include a single medium or multiple media that store the one or
more sets of instructions. The term "computer-readable medium"
shall also be taken to include any medium that is capable of
storing, encoding or carrying a set of instructions for execution
by a machine and that cause a machine to perform any one or more of
the methodologies of the present invention. The term
"computer-readable medium" shall accordingly be taken to include,
but not be limited to, solid-state memories, and optical and
magnetic media.
[0178] Embodiments of the invention have been described through
functional modules at times, which are defined by executable
instructions recorded on computer readable media which cause a
computer, microprocessors or chipsets to perform method steps when
executed. The modules have been segregated by function for the sake
of clarity. However, it should be understood that the modules need
not correspond to discrete blocks of code and the described
functions can be carried out by the execution of various code
portions stored on various media and executed at various times.
[0179] It should be understood that processes and techniques
described herein are not inherently related to any particular
apparatus and may be implemented by any suitable combination of
components. Further, various types of general purpose devices may
be used in accordance with the teachings described herein. It may
also prove advantageous to construct specialized apparatus to
perform the method steps described herein. The invention has been
described in relation to particular examples, which are intended in
all respects to be illustrative rather than restrictive. Those
skilled in the art will appreciate that many different combinations
of hardware, software, and firmware will be suitable for practicing
the present invention. Various aspects and/or components of the
described embodiments may be used singly or in any combination. It
is intended that the specification and examples be considered as
exemplary only, with a true scope and spirit of the invention being
indicated by the claims.
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