U.S. patent application number 17/150127 was filed with the patent office on 2021-06-24 for medical device.
The applicant listed for this patent is Ethicon LLC. Invention is credited to Peter Ralph Bonham, Alan Edward Green, Paul Christopher Roberts, Mark David Tuckwell.
Application Number | 20210186554 17/150127 |
Document ID | / |
Family ID | 1000005436470 |
Filed Date | 2021-06-24 |
United States Patent
Application |
20210186554 |
Kind Code |
A1 |
Green; Alan Edward ; et
al. |
June 24, 2021 |
MEDICAL DEVICE
Abstract
A medical device is described having a handle, a shaft coupled
to the handle and an end effector coupled to the shaft. In one
embodiment, the device includes an ultrasonic transducer and is
arranged so that ultrasonic or electrical energy can be delivered
to a vessel or tissue to be treated. Various novel sensing circuits
are described to allow a measure of the drive signal to be measured
and fed back to a controller. An active fuse circuit is also
described for protecting one or more batteries of the device from
an over-current situation.
Inventors: |
Green; Alan Edward;
(Cumbria, GB) ; Roberts; Paul Christopher;
(Cambridge, GB) ; Tuckwell; Mark David;
(Cambridge, GB) ; Bonham; Peter Ralph; (Cambridge,
GB) |
|
Applicant: |
Name |
City |
State |
Country |
Type |
Ethicon LLC |
Guaynabo |
PR |
US |
|
|
Family ID: |
1000005436470 |
Appl. No.: |
17/150127 |
Filed: |
January 15, 2021 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
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15105444 |
Jun 16, 2016 |
10912580 |
|
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PCT/US2014/069039 |
Dec 8, 2014 |
|
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17150127 |
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Current U.S.
Class: |
1/1 |
Current CPC
Class: |
A61B 2017/00141
20130101; A61B 2018/00994 20130101; A61B 2017/0003 20130101; A61B
2018/00875 20130101; A61B 18/1206 20130101; A61B 2017/320069
20170801; A61B 2017/320094 20170801; A61B 2018/1226 20130101; A61B
2017/00734 20130101; A61B 2017/320093 20170801; A61B 17/320092
20130101; A61B 2017/00137 20130101; A61B 2017/320095 20170801; A61B
2018/00642 20130101 |
International
Class: |
A61B 17/32 20060101
A61B017/32; A61B 18/12 20060101 A61B018/12 |
Foreign Application Data
Date |
Code |
Application Number |
Dec 16, 2013 |
GB |
1322210.4 |
Claims
1.-22. (canceled)
23. A medical device comprising: an end effector for gripping a
vessel/tissue; one or more batteries for providing a DC voltage
supply; a signal generator coupled to the one or more batteries for
generating a cyclically varying drive signal from the DC voltage
supply for driving energy into the vessel/tissue; a controller
operable to control the signal generator to control the energy
delivered to the vessel/tissue; and an active fuse circuit coupled
between the one or more batteries and the signal generator for
protecting the one or more batteries.
24. A device according to claim 23, wherein the active fuse circuit
comprises a switch that is electrically coupled between a terminal
of the one or more batteries and the signal generator; and control
circuitry configured to switch the switch.
25. A device according to claim 24, wherein the switch is arranged
to disconnect the signal generator from the one or more batteries
or is arranged to connect a large impedance between the signal
generator and the one or more batteries.
26. A device according to claim 24, wherein the control circuitry
of the active fuse comprises circuitry for sensing a measure of the
current being drawn from the one or more batteries and is
configured to switch the switch in the event that the current
measure exceeds a threshold.
27. A device according to claim 26, wherein the control circuitry
of the active fuse comprises a comparator for comparing the current
measure with the threshold and wherein an output of the comparator
controls the opening and closing of the switch.
28.-39. (canceled)
Description
[0001] The present invention relates to the field of medical
devices and in particular, although not exclusively, to medical
cauterization and cutting devices. The invention also relates to
drive circuits and methods for driving such medical devices.
[0002] Many surgical procedures require cutting or ligating blood
vessels or other internal tissue and many procedures are performed
using minimally invasive techniques with a hand-held cauterization
device to perform the cutting or ligating. Some existing hand-held
cauterization devices use an ultrasonic transducer in the
cauterization device to apply ultrasonic energy to the tissue to be
cut or ligated. Other hand-held cauterization devices apply RF
energy directly to the tissue/vessel being cauterized via forceps
of the device.
[0003] The present invention aims to provide an alterative surgical
device that is able to apply ultrasonic energy or RF energy to the
vessel or tissue to be cauterized. Other aspects of the invention
relate to the way in which control circuitry is provided to select
between the different operating modes. Other aspects of the
invention relate to the way in which voltage and current
measurements can be made in the circuit design for reporting to a
controller, such as a microprocessor; and to the way in which
control circuitry can be provided to ensure that too much current
is not drawn from the battery.
[0004] According to one aspect, the present invention provides a
medical device comprising: an end effector for gripping a
vessel/tissue; an ultrasonic transducer coupled to the end
effector; a drive circuit coupled to the end effector and to the
ultrasonic transducer and operable to generate a periodic drive
signal and to provide the drive signal either to the ultrasonic
transducer or to the end effector; and a controller operable to
control the drive circuit so that the drive signal is applied to a
desired one of the ultrasonic transducer and the end effector.
[0005] In one embodiment, the drive circuit comprises a first
resonant circuit having a first resonant frequency and a second
resonant circuit having a second resonant frequency that is
different to the first resonant frequency, wherein the first
resonant frequency corresponds to a resonant characteristic of the
ultrasonic transducer and wherein the controller is operable to
control the drive circuit so that the drive circuit generates a
drive signal having a frequency corresponding to the first resonant
frequency when the drive signal is to be applied to the ultrasonic
transducer and so that the drive circuit generates a drive signal
having a frequency corresponding to the second resonant frequency
when the drive signal is to be applied to the end effector.
[0006] A signal generator may also be provided that is coupled
between the controller and the drive circuit for generating a
cyclically varying voltage from a DC voltage supply in dependence
upon control signals from the controller and for supplying the
cyclically varying voltage to the first and second resonant
circuits of the drive circuit.
[0007] The controller may be arranged to vary the period of the
drive signal about the first resonant frequency or the second
resonant frequency to vary the energy supplied to the vessel or
tissue gripped by the end effector. The controller may vary the
period of the drive signal so that the frequency of the drive
signal varies around the first resonant frequency within 0.1% to 1%
of the first resonant frequency; or so that the frequency of the
drive signal varies around the second resonant frequency within 40%
to 60% of the second resonant frequency.
[0008] Typically, the resonant characteristics of the first and
second resonant circuits vary with the tissue or vessel gripped by
the forceps and in one embodiment, the controller is configured to
vary the period of the drive signal to track changes in the
respective resonant characteristic.
[0009] An ultrasonic waveguide may be provided that is coupled to
the ultrasonic transducer for guiding ultrasonic energy generated
by the ultrasonic transducer towards the end effector. The end
effector may comprise first and second jaws and the second resonant
circuit may be electrically coupled to the first and second jaws of
the end effector. For example, the first jaw of the end effector
may be electrically coupled to the waveguide and the second
resonant circuit may be electrically coupled to the first jaw of
the end effector via the ultrasonic waveguide. In some embodiments,
the first resonant circuit is electrically coupled to the
ultrasonic transducer and to the waveguide.
[0010] Sensing circuitry may be provided for sensing a drive signal
applied to the ultrasonic transducer or to the end effector. In one
embodiment, one or both of the first and second resonant circuits
may comprise at least one of an inductor coil, a capacitor and a
resistor and wherein the sensing circuitry may comprise an op-amp
circuit for sensing the voltage across the inductor coil or the
capacitor or the resistor and for converting the sensed voltage to
a sensor signal suitable for inputting to the controller. In an
alternative embodiment, one or both of the first and second
resonant circuits may comprise an impedance element that is
connected between the resonant circuit and a reference potential
and wherein the sensing circuitry comprises a divider circuit for
obtaining a measure of the voltage across the impedance element and
a bias signal generator for applying a DC bias signal to the
voltage measure. In this case, the impedance element may comprise a
capacitor or a resistor. Typically, the sensing circuitry comprises
DC blocking circuitry for preventing the DC bias signal from the
bias signal generator from coupling with the drive circuit. The
bias signal generator may comprises a voltage divider circuit
connected between a reference voltage and a supply voltage of the
controller.
[0011] The device is preferably a battery operated device and
comprises one or more batteries for powering the device and further
comprising an active fuse circuit for protecting the one or more
batteries. The active fuse circuit may comprise a switch
electrically coupled between a terminal of the one or more
batteries and the drive circuit and control circuitry configured to
open the switch to isolate the supply terminal from the drive
circuit.
[0012] The present invention also provides a medical device
comprising: an end effector for gripping a vessel/tissue; a drive
circuit for generating a cyclically varying drive signal for
driving energy into the vessel/tissue; sensing circuitry for
sensing a drive signal generated by the drive circuit; and a
controller responsive to the sensing circuitry and operable to
control the drive circuit to control the energy delivered to the
vessel/tissue; wherein the drive circuit comprises an impedance
element that is coupled to a reference potential and wherein the
sensing circuitry comprises a divider circuit for obtaining a
measure of the voltage across the impedance element and a bias
signal generator for applying a DC bias signal to the voltage
measure.
[0013] The impedance element may be a capacitor or a resistor. The
sensing circuitry may also comprise DC blocking circuitry for
preventing the DC bias signal from the bias signal generator from
coupling with the drive circuit. The bias signal generator may
comprise a divider circuit connected between a reference voltage
and a supply voltage of the controller. The divider circuit of the
bias signal generator may be connected to the supply voltage of the
controller via a switch and wherein the controller is configured to
open the switch when the controller does not require signals from
the sensing circuitry.
[0014] The present invention also provides a medical device
comprising: an end effector for gripping a vessel/tissue; one or
more batteries for providing a DC voltage supply; a signal
generator coupled to the one or more batteries for generating a
cyclically varying drive signal from the DC voltage supply for
driving energy into the vessel/tissue; a controller operable to
control the signal generator to control the energy delivered to the
vessel/tissue; and an active fuse circuit coupled between the one
or more batteries and the signal generator for protecting the one
or more batteries.
[0015] The active fuse circuit may comprise a switch that is
electrically coupled between a terminal of the one or more
batteries and the signal generator; and control circuitry
configured to switch the switch. The switch may be arranged to
disconnect the signal generator from the one or more batteries or
may connect a large impedance between the signal generator and the
one or more batteries.
[0016] The control circuitry of the active fuse may comprise
circuitry for sensing a measure of the current being drawn from the
one or more batteries and is configured to switch the switch in the
event that the current measure exceeds a threshold. The control
circuitry of the active fuse may comprise a comparator for
comparing the current measure with the threshold and wherein an
output of the comparator controls the opening and closing of the
switch.
[0017] The present invention also provides a method of operating a
medical device comprising generating a periodic drive signal and
applying the drive signal to an ultrasonic transducer or to an end
effector of the medical device and controlling the drive circuit so
that the drive signal is applied to a desired one of the ultrasonic
transducer and the end effector.
[0018] The present invention also provides a method of cauterising
or cutting a vessel or tissue, the method comprising: gripping the
vessel or tissue with an end effector of a medical device; using a
drive circuit to apply a periodic drive signal either to an
ultrasonic transducer or to the end effector; and controlling the
drive circuit so that the drive signal is applied to a desired one
of the ultrasonic transducer and the end effector. The method may
use the above described medical device.
[0019] The present invention also provides electronic apparatus for
use in a medical device having an ultrasonic transducer and an end
effector, the electronic apparatus comprising: a drive circuit for
generating a periodic drive signal; and a controller operable to
control the drive circuit so that the drive signal is applied to a
desired one of the ultrasonic transducer and the end effector;
wherein the drive circuit comprise a first resonant circuit having
a first resonant frequency and a second resonant circuit having a
second resonant frequency that is different to the first resonant
frequency, and wherein the controller is operable to control the
drive circuit so that the drive circuit generates a drive signal
having a frequency corresponding to the first resonant frequency
when the drive signal is to be applied to the ultrasonic transducer
and so that the drive circuit generates a drive signal having a
frequency corresponding to the second resonant frequency when the
drive signal is to be applied to the end effector.
[0020] The present invention also provides a medical device
comprising: an end effector for gripping a vessel/tissue; a drive
circuit coupled to the end effector and operable to generate a
drive signal and to provide the drive signal to the end effector; a
controller operable to generate and output control signals to the
drive circuit to control the drive signal generated by the drive
circuit; wherein the drive circuit and a load formed by the
vessel/tissue gripped by the end effector define a resonant circuit
whose resonant frequency varies as the impedance of the load formed
by the vessel/tissue gripped by the end effector changes; wherein
the controller is arranged to generate control signals which cause
the drive circuit to generate a drive signal having a frequency
that tracks said resonant frequency as it changes; and wherein said
controller is further arranged to reduce one or more of the power,
current or voltage delivered to the load formed by the
vessel/tissue gripped by the end effector.
[0021] Sensor circuitry may be provided for sensing signals applied
to the load formed by the vessel/tissue gripped by the end effector
and measurement circuitry for processing the signals from the
sensor circuitry to determine a measure of the impedance of the
load formed by the vessel/tissue gripped by the end effector. In
this case, the controller can generate said control signals in
dependence upon said measure of the impedance of the load formed by
the vessel/tissue gripped by the end effector.
[0022] In one embodiment, the controller generates control signals
having sequences of pulses and the controller skips one or more
pulses from the control signals in order to reduce one or more of
the power, current or voltage delivered to the load formed by the
vessel/tissue gripped by the end effector.
[0023] Typically, in this case, the controller comprises a pulse
signal generator that generates pulses at a desired frequency that
depends on said resonant frequency and the controller skips pulses
generated by said pulse signal generator by suppressing pulses
generated by the pulse signal generator.
[0024] These and various other features and aspects of the
invention will become apparent from the following detailed
description of embodiments which are described with reference to
the accompanying Figures in which:
[0025] FIG. 1 illustrates a hand-held cauterization device that has
batteries and drive and control circuitry mounted in a handle
portion of the device;
[0026] FIG. 2 is a part block diagram illustrating the main
components of the cauterization device used in one embodiment of
the invention;
[0027] FIG. 3 is a circuit diagram illustrating the main electrical
components of the cauterization device shown in FIG. 2;
[0028] FIG. 4 schematically illustrates the way in which the
ultrasonic transducer is coupled to a waveguide for delivering the
generated ultrasonic energy to the forceps and illustrating the way
in which the circuitry shown in FIG. 3 can deliver electrical
energy to the forceps;
[0029] FIG. 5 is a block diagram that schematically illustrates
processing modules that form part of the microprocessor shown in
FIG. 2;
[0030] FIG. 6 illustrates the form of control signals generated by
the microprocessor to control the drive circuit whilst minimising
3.sup.rd harmonic content;
[0031] FIG. 7 is a contour plot illustrating the delivered power
versus the load resistance and the drive frequency;
[0032] FIG. 8a is a circuit diagram illustrating one way in which a
measure of the load current can be determined and supplied to the
microprocessor;
[0033] FIG. 8b is a circuit diagram illustrating another way in
which a measure of the load current can be determined and supplied
to the microprocessor;
[0034] FIG. 8c is a circuit diagram illustrating one way in which a
measure of the load voltage can be determined and supplied to the
microprocessor;
[0035] FIG. 9a is a circuit diagram illustrating one way in which a
measure of the load current can be determined and supplied to the
microprocessor without using an op-amp circuit;
[0036] FIG. 9b is a circuit diagram illustrating one way in which a
measure of the load current can be determined and supplied to the
microprocessor and illustrating one way in which a measure of the
load voltage can be determined and supplied to the microprocessor
without using op-amp circuits; and
[0037] FIG. 10 is a circuit diagram illustrating an active fuse
circuit used to protect the batteries shown in FIG. 2 from
excessive current demand.
MEDICAL DEVICE
[0038] Many surgical procedures require cutting or ligating blood
vessels or other vascular tissue. With minimally invasive surgery,
surgeons perform surgical operations through a small incision in
the patient's body. As a result of the limited space, surgeons
often have difficulty controlling bleeding by clamping and/or
tying-off transected blood vessels. By utilizing
ultrasonic-surgical forceps or electro-surgical forceps, a surgeon
can cauterize, coagulate/desiccate, and/or simply reduce bleeding
by controlling the ultrasonic energy applied to the tissue/vessel
by an ultrasonic transducer or by controlling the RF energy applied
to the tissue/vessel via the forceps.
[0039] FIG. 1 illustrates the form of an ultrasonic/RF-surgical
medical device 1 that is designed for minimally invasive medical
procedures, according to one embodiment of the present invention.
As shown, the device 1 is a self contained device, having an
elongate shaft 3 that has a handle 5 connected to the proximal end
of the shaft 3 and an end effector 7 connected to the distal end of
the shaft 3. In this embodiment, the end effector 7 comprises
medical forceps 9 that are controlled by the user manipulating
control levers 11 and 13 of the handle 5.
[0040] During a surgical procedure, the shaft 3 is inserted through
a trocar to gain access to the patient's interior and the operating
site. The surgeon will manipulate the forceps 9 using the handle 5
and the control levers 11 and 13 until the forceps 9 are located
around the vessel to be cut or cauterised. Electrical energy is
then applied, in a controlled manner, either to the tissue directly
via the forceps 9 (as RF energy) or to an ultrasonic transducer 8
that is mounted within the handle 5 and coupled to the forceps 9
via a waveguide (not shown) within the shaft 3, in order to perform
the desired cutting/cauterisation using ultrasonic energy. As shown
in FIG. 1, in this embodiment, the handle 5 also houses batteries
15 and control electronics 17 for generating and controlling the
electrical energy required to perform the cauterisation. In this
way, the device 1 is self contained in the sense that it does not
need a separate control box and supply wire to provide the
electrical energy to the forceps 9. However, such a separate
control box may be provided if desired.
[0041] System Circuitry
[0042] FIG. 2 is a schematic block diagram illustrating the main
electrical circuitry of the cauterization/cutting device 1 used in
this embodiment to generate and control the electrical energy
supplied to the ultrasonic transducer or to the forceps 9. As will
be explained in more detail below, in this embodiment, the
circuitry is designed to control the period of an electrical drive
waveform that is generated in order to control the amount of power
delivered to the tissue/vessel being cauterized.
[0043] As shown in FIG. 2, the cauterization/cutting device 1
comprises a user interface 21--via which the user is provided with
information (such as an indication that energy is being applied to
the gripped tissue/vessel by electrical energy or ultrasonic
energy) and through which the user controls the operation of the
cauterization/cutting device 1, including selection of ultrasonic
operation or RF operation. As shown, the user interface 21 is
coupled to a microprocessor 23 that controls the
cutting/cauterisation procedure by generating control signals that
it outputs to gate drive circuitry 25. In response to the control
signals from the microprocessor 23, the gate drive circuitry 25
generates gate control signals that cause a bridge signal generator
27 to generate a desired drive waveform that is applied either to
the ultrasonic transducer 8 or to the forceps 9 via a drive circuit
29. Voltage sensing circuitry 31 and current sensing circuitry 33
generate measures of the current and voltage applied to the
ultrasonic transducer 8 or to the forceps 9, which they feed back
to the microprocessor 23 for control purposes. FIG. 2 also shows
the batteries 15 that provide the power for powering the electrical
circuitry shown in FIG. 2. In this embodiment, the batteries 15 are
arranged to supply 0V and 14V rails.
[0044] FIG. 3 illustrates in more detail the components of the gate
drive circuitry 25, the bridge signal generator 27 and the drive
circuit 29. FIG. 3 also shows an electrical equivalent circuit 30
of the piezo-electric ultrasonic transducer 8 and the load
(R.sub.load) formed by the tissue/vessel to be treated. As shown in
FIG. 3, the gate drive circuitry 25 includes two FET gate drives
37--FET gate drive 37-1 and FET gate drive 37-2. A first set of
control signals (CTRL.sub.1) from the microprocessor 23 is supplied
to FET gate drive 37-1 and a second set of control signals
(CTRL.sub.2) from the microprocessor 23 is supplied to FET gate
drive 37-2. FET gate drive 37-1 uses the first set of control
signals (CTRL.sub.1) to generate two drive signals--one for driving
each of the two FETs 41-1 and 41-2 of the bridge signal generator
27. The FET gate drive 37-1 generates drive signals that causes the
upper FET (41-1) to be on when the lower FET (41-2) is off and vice
versa. This causes the node A to be alternately connected to the
14V rail (when FET 41-1 is switched on) and the 0V rail (when the
FET 41-2 is switched on). Similarly, FET gate drive 37-2 uses the
second set of control signals (CTRL.sub.2) to generate two drive
signals--one for driving each of the two FETs 41-3 and 41-4 of the
bridge signal generator 27. The FET gate drive 37-2 generates drive
signals that causes the upper FET (41-3) to be on when the lower
FET (41-4) is off and vice versa. This causes the node B to be
alternately connected to the 14V rail (when FET 41-3 is switched
on) and the OV rail (when the FET 41-4 is switched on). Thus the
two sets of control signals (CTRL.sub.1 and CTRL.sub.2) output by
the microprocessor 23 control the digital waveform that is
generated and applied between nodes A and B. Each set of control
signals (CTRL.sub.1 and CTRL.sub.2) comprises of a pair of signal
lines, one to indicate when the high side FET is on and the other
to indicate when the low side FET is on. Thus the microprocessor
23, either through software or through a dedicated hardware
function can ensure that the undesirable condition when both high
and low side FETs are simultaneously turned on does not occur. In
practice this requires leaving a dead time when both high and low
side FETs are turned off to ensure that, even when allowing for
variable switching delays, there is no possibility that both FETs
can be simultaneously on. In the present embodiment a dead time of
about 100 ns was used.
[0045] As shown in FIG. 3, the nodes A and B are connected to the
drive circuit 29, thus the digital voltage generated by the bridge
signal generator 27 is applied to the drive circuit 29. This
applied voltage will cause current to flow in the drive circuit 29.
As shown in FIG. 3, the drive circuit 29 includes two transformer
circuits 42-1 and 42-2. The first transformer circuit 42-1 is
designed for efficient driving of the ultrasonic transducer 8 and
includes a capacitor-inductor-inductor resonant circuit 43-1 formed
by capacitor C.sup.US.sub.s 45, inductor L.sup.US.sub.s 47 and
inductor L.sup.US.sub.m 49. When driving the ultrasonic transducer
8, the microprocessor 23 is arranged to generate control signals
for the gate drive circuitry 25 so that the fundamental frequency
(f.sub.d) of the digital voltage applied across nodes A and B is
around the resonant frequency of the resonant circuit 43-1, which
in this embodiment is about 50 kHz. As a result of the resonant
characteristic of the resonant circuit 43-1, the digital voltage
applied across nodes A and B will cause a substantially sinusoidal
current at the fundamental frequency (f.sub.d) to flow within the
resonant circuit 43-1. This is because higher harmonic content of
the drive voltage will be attenuated by the resonant circuit 43-1
and the impedance of L.sub.t and C.sub.t1 referred to the
transformer primary.
[0046] As illustrated in FIG. 3, the inductor L.sup.US.sub.m 49
forms the primary of the transformer circuit 42-1, the secondary of
which is formed by inductor L.sup.US.sub.sec 53. The transformer
up-converts the drive voltage (V.sup.US.sub.d) across the inductor
L.sub.m 49 to a load voltage (V.sub.L; typically about 120 volts)
that is applied to the ultrasonic transducer 8. The electrical
characteristics of the ultrasonic transducer 8 change with the
impedance of the forceps' jaws and any tissue or vessel gripped by
the forceps 9; and FIG. 3 models the ultrasonic transducer 8 and
the impedance of the forceps' jaws and any tissue or vessel gripped
by the forceps 9 by the inductor L.sub.t 57, the parallel
capacitors C.sub.t1 59 and C.sub.t2 61 and the resistance
R.sub.load.
[0047] The inductor L.sup.US.sub.s and capacitor C.sup.US.sub.s of
the drive circuit 29 are designed to have a matching LC product to
that of inductor L.sub.t and capacitor C.sub.t1 of the ultrasonic
transducer 8. Matching the LC product of a series LC network
ensures that the resonant frequency of the network is maintained.
Similarly, the magnetic reactance of the inductor L.sup.US.sub.m is
chosen so that at resonance it matches with the capacitive
reactance of the capacitor C.sub.t2 of the ultrasonic transducer 8.
For example, if the transducer 8 is defined such that capacitor
C.sub.t2 has a capacitance of about 3.3 nF, then the inductor
L.sup.US.sub.m should have an inductance of about 3 mH (at a
resonant frequency of about 50 kHz). Designing the drive circuit 29
in this way provides for the optimum drive efficiency in terms of
energy delivery to the tissue/vessel gripped by the forceps 9. The
efficiency improvement is realised because the current flowing in
C.sup.US.sub.s and consequently the FET bridge (27) is reduced,
because the transformer magnetising current cancels out the current
flowing in C.sub.t2 In addition, because of this current
cancellation, the current flowing in C.sup.US.sub.s is proportional
to the current flowing in Rload, which allows the load current to
be determined by measuring the current flowing in
C.sup.US.sub.s.
[0048] The second transformer circuit 42-2 is designed for
efficient driving of electrical RF energy directly to the
tissue/vessel via the forceps 9 and includes a
capacitor-inductor-inductor resonant circuit 43-2 formed by
capacitor C.sup.F.sub.s 46, inductor L.sup.F.sub.s 48 and inductor
L.sup.F.sub.m 50. When driving the forceps 9 directly with
electrical energy, the microprocessor 23 is arranged to generate
control signals for the gate drive circuitry 25 so that the
fundamental frequency (f.sub.d) of the digital voltage applied
across nodes A and B is around the resonant frequency of the
resonant circuit 43-2, which in this embodiment is about 500 kHz.
As a result of the resonant characteristic of the resonant circuit
43-2, the digital voltage applied across nodes A and B will cause a
substantially sinusoidal current at the fundamental frequency
(f.sub.d) to flow within the resonant circuit 43-2. This is because
higher harmonic content of the drive voltage will be attenuated by
the resonant circuit 43-2.
[0049] As illustrated in FIG. 3, the inductor L.sup.F.sub.m 50
forms the primary of the transformer circuit 42-1, the secondary of
which is formed by inductor L.sup.US.sub.sec 54. The transformer
up-converts the drive voltage (V.sup.F.sub.d) across inductor
L.sup.F.sub.m 50 to the load voltage (V.sup.F.sub.L; typically
about 120 volts) that is applied to the forceps 9. The tissue or
vessel gripped by the jaws of the forceps 9 is represented as the
resistive load R.sub.load in the box labelled 9 in FIG. 3. In
practice, this will be the same resistive load that is illustrated
in the electrical equivalent circuit 30 of the ultrasonic
transducer 8.
[0050] FIG. 4 is a schematic diagram illustrating the way in which
the ultrasonic transducer 8 couples to the tissue/vessel to be
cauterized and the way in which the circuit components illustrated
in FIG. 3 connect to the ultrasonic transducer 8 and to the forceps
9. In particular, FIG. 4 shows the shaft 3, the forceps 9 and the
ultrasonic transducer 8. FIG. 4 also shows the waveguide 72 along
which the ultrasonic signal that is generated by the ultrasonic
transducer 8 is guided. The waveguide 72 is connected to the node
"BB" show in FIG. 3, whilst the input supply to the ultrasonic
transducer 8 is connected to node "AA" shown in FIG. 3. The output
node "CC" of the second transformer circuit 42-2 is connected to a
conductive inner wall of the sheath 3, which is electrically
connected to the upper jaw 74 of the forceps 9. The return path is
through the tissue/vessel to be cauterized and the lower jaw 76,
which is electrically connected to the node "BB".
[0051] When the drive signal has a drive frequency of about 50 kHz,
very little current will flow within the second transformer circuit
42-2 because the drive frequency is far away from the resonant
frequency of the resonant circuit 43-2 such that the input
impedance of the second transformer circuit 42-2 will be very high
for this drive signal. Therefore, the power will be delivered
almost entirely via the first transformer circuit 42-1. Similarly,
when the drive signal has a drive frequency of about 500 kHz, very
little current will flow within the first transformer circuit 42-1
because the drive frequency is far away from the resonant frequency
of the resonant circuit 43-1 such that the input impedance of the
first transformer circuit 42-1 will be very high for this drive
signal. Therefore, the power will be delivered almost entirely via
the second transformer circuit 42-2. In this way, the two
transformer circuits 42-1 and 42-2 can be driven by a common bridge
signal generator 27; although it is also feasible to drive each
transformer circuit with separate bridge signal generators.
[0052] It is not always desired to apply full power to the
tissue/vessel to be treated. Therefore, in this embodiment in the
ultrasonic mode of operation, the amount of ultrasonic energy
supplied to the vessel/tissue is controlled by varying the period
of the digital waveform applied across nodes A and B so that the
drive frequency (f.sub.d) moves away from the resonant frequency of
the ultrasonic transducer 8. This works because the ultrasonic
transducer 8 acts as a frequency dependent (lossless) attenuator.
The closer the drive signal is to the resonant frequency of the
ultrasonic transducer 8, the more ultrasonic energy the ultrasonic
transducer 8 will generate. Similarly, as the frequency of the
drive signal is moved away from the resonant frequency of the
ultrasonic transducer 8, less and less ultrasonic energy is
generated by the ultrasonic transducer 8. In addition or instead,
the duration of the pulses of the drive signals may be varied to
control the amount of ultrasonic energy delivered to the
tissue/vessel.
[0053] Similarly, in the electrical mode of operation, the amount
of electrical power supplied to the forceps 9 is controlled by
varying the period of the digital waveform applied across nodes A
and B so that the drive frequency (f.sub.d) moves away from the
resonant frequency of the resonant circuit 43-2. This works because
the resonant circuit 43-1 acts as a frequency dependent (lossless)
attenuator. The closer the drive signal is to the resonant
frequency of the resonant circuit 43-1, the less the drive signal
is attenuated. Conversely, as the frequency of the drive signal is
moved away from the resonant frequency of the circuit 43-1, the
more the drive signal is attenuated and so the electrical energy
supplied to the tissue/vessel reduces. The drive frequency needs to
move away from the resonant frequency by about 50% of the resonant
frequency to achieve the desired range of power variation. An
alternative approach to controlling the power, current or voltage
applied during the electrical mode of operation is to continuously
tune the frequency of the excitation signal to keep it matched with
the resonant frequency of the drive circuit (as it changes with the
changing load impedance) and thereby maintain efficient operation
and to skip some of the pulses of the drive control signals until
the average power, current and/or voltage is below the relevant
limit. A further alternative, which is most effective when using
the pulse skipping technique, is to remove the inductor 48 shown in
FIG. 3 and therefore the drive circuit for the electrical mode of
operation becomes a substantially parallel LC resonant circuit (it
is not a pure parallel LC resonant circuit because the transformer
leakage inductance appears in series with inductor 48 and cannot be
entirely removed). The advantage of removing the inductor 48 is
that the overall efficiency can be increased, because there are no
longer any losses in the inductor. A further advantage is that the
physical size of the circuit can be reduced, because often the
inductor is a physically large component relative to the FETs,
microprocessor, capacitors and other system components.
[0054] The microprocessor 23 controls the power delivery based on a
desired power to be delivered to the circuitry 30 (which models the
ultrasonic transducer 8 and the tissue/vessel gripped by the
forceps 9) or to the forceps 9 and based on measurements of the
load voltage (V.sub.L) and of the load current (i.sub.L) obtained
from the voltage sensing circuitry 31 and the current sensing
circuitry 33. The microprocessor 23 also selects the frequency of
the drive signal (around 50 kHz or around 500 kHz) based on a user
input received via the user interface 21 that selects either
electrical operation or ultrasonic operation.
[0055] Microprocessor
[0056] FIG. 5 is a block diagram illustrating the main components
of the microprocessor 23 that is used in this embodiment. As shown,
the microprocessor 23 includes synchronous I,Q sampling circuitry
81 that receives the sensed voltage and current signals from the
sensing circuitry 31 and 33 and obtains corresponding samples which
are passed to a measured voltage and current processing module 83.
The measured voltage and current processing module 83 uses the
received samples to calculate the impedance of, and the RMS voltage
applied to and the RMS current flowing through, the ultrasonic
transducer 8 and/or directly to the tissue/vessel gripped by the
forceps 9; and from them the power that is presently being supplied
to the circuitry 30 or directly to the tissue/vessel gripped by the
forceps 9. The determined values are then passed to a power
controller 85 for further processing. The measured voltage and
current processing module 83 can also process the received I and Q
samples to calculate the phase difference between the load voltage
(V.sub.L) and the load current (i.sub.L). During the ultrasonic
mode of operation, at resonance, this phase difference should be
around zero and so this phase measure can be used as a feedback
parameter for the power controller 85.
[0057] The power controller 85 uses the received impedance value
and the delivered power value to determine, in accordance with a
predefined algorithm and a power set point value and a mode
indication signal (received from a medical device control module 89
and indicating ultrasonic operation or electrical operation), a
desired period/frequency (.DELTA.t.sub.new) of the control signals
(CTRL.sub.1 and CTRL.sub.2) that are used to control the gate drive
circuit 25. This desired period/frequency is passed from the power
controller 85 to the control signal generator 95, which changes the
control signals CTRL.sub.1 and CTRL.sub.2 in order to change the
waveform period to match the desired period. The CTRL control
signals may comprise square wave signals having the desired period
or they may comprise periodic pulses with the period corresponding
to the desired period (.DELTA.t.sub.new) and with the relative
timing of the pulses of the control signals being set to minimise
harmonic content of the waveform that is generated by the bridge
signal generator 27 (such as to minimise the 3.sup.rd order
harmonic). In this embodiment, the control signals CTRL.sub.1 are
output to the FET gate drive 37-1 (shown in FIG. 2), which
amplifies the control signals and then applies them to the FETs
41-1 and 41-2; and the control signals CTRL.sub.2 are output to the
FET gate drive 37-2 (shown in FIG. 2), which amplifies the control
signals and then applies them to the FETs 41-3 and 41-4, to thereby
generate the desired waveform with the new period
(.DELTA.t.sub.new).
[0058] In order to drive the circuitry with the optimal RMS
waveform, the MOSFETs 41 are driven as complementary, opposing
pairs. Although the maximum output voltage is achieved when the
MOSFET pairs are driven at a phase shift of 180 degrees, the
resulting harmonic content of such a drive waveform, particularly
the 3rd harmonic (which is poorly excluded by the output filter) is
quite high. The inventors have found that the optimal phase shift
between the control signals applied to the two pairs of MOSFETs 41,
for 3rd harmonic reduction, is around 120.degree.. This is
illustrated in FIG. 6 which shows in the upper plot the output from
the first MOSFET pair 41-1 and 41-2; in the middle plot the output
from the second MOSFET pair 41-3 and 41-4 (shifted by 120.degree.
relative to the upper plot); and in the lower plot the resulting
(normalised) output voltage applied across inputs A and B. The
shape of this normalised output voltage has very low 3.sup.rd order
harmonic content.
[0059] I & Q Signal Sampling
[0060] Both the load voltage and the load current will be
substantially sinusoidal waveforms, although they may be out of
phase, depending on the impedance of the load represented by the
transducer 8 and/or the vessel/tissue gripped by the forceps 9. The
load current and the load voltage will be at the same drive
frequency (f.sub.d) corresponding to the presently defined waveform
period (.DELTA.t.sub.new). Normally, when sampling a signal, the
sampling circuitry operates asynchronously with respect to the
frequency of the signal that is being sampled. However, as the
microprocessor 23 knows the frequency and phase of the switching
signals, the synchronous sampling circuit 81 can sample the
measured voltage/current signal at predefined points in time during
the drive period. In this embodiment, during the ultrasonic mode of
operation, the synchronous sampling circuit 81 oversamples the
measured signal eight times per period to obtain four I samples and
four Q samples. Oversampling allows for a reduction of errors
caused by harmonic distortion and therefore allows for the more
accurate determination of the measured current and voltage values.
However, oversampling is not essential and indeed under sampling
(less than two samples per period) is performed when the device is
operating in the electrical mode of operation and is possible due
to the synchronous nature of the sampling operation. The timing
that the synchronous sampling circuit 81 makes these samples is
controlled, in this embodiment, by the control signals CTRL.sub.1
and CTRL.sub.2. Thus when the period of these control signals is
changed, the period of the sampling control signals CTRL.sub.1 and
CTRL.sub.2 also changes (whilst their relative phases stay the
same). In this way, the sampling circuitry 81 continuously changes
the timing at which it samples the sensed voltage and current
signals as the period of the drive waveform is changed so that the
samples are always taken at the same time points within the period
of the drive waveform. Therefore, the sampling circuit 81 is
performing a "synchronous" sampling operation instead of a more
conventional sampling operation that just samples the input signal
at a fixed sampling rate defined by a fixed sampling clock. Of
course, such a conventional sampling operation could be used
instead.
[0061] Measurements
[0062] The samples obtained by the synchronous sampling circuitry
51 are passed to the measured voltage and current processing module
83 which can determine the magnitude and phase of the measured
signal from just one "I" sample and one "Q" sample of the load
current and load voltage. However, in this embodiment, to achieve
some averaging, the processing module 83 averages consecutive "I"
samples to provide an average "I" value and consecutive "Q" samples
to provide an average "Q" value; and then uses the average I and Q
values to determine the magnitude and phase of the measured signal.
Of course, it should be recognised that some pre-processing of the
data may be required to convert the actual measured I and Q samples
into I and Q samples of the load voltage or the load current, for
example, scaling, integration or differentiation of the sample
values may be performed to convert the sampled values into true
samples of the load voltage (V.sub.L) and the load current
(i.sub.L). Where integration or differentiation is required, this
can be achieved simply by swapping the order of the I and Q
samples--as integrating/differentiating a sinusoidal signal simply
involves a scaling and a 90 degree phase shift.
[0063] The RMS load voltage, the RMS load current and the delivered
power, P.sub.delivered, can then be determined from:
V RMS = 1 2 ( V I 2 + V Q 2 ) ##EQU00001## I RMS = 1 2 ( I I 2 + I
Q 2 ) ##EQU00001.2## Power = V I * = 1 2 ( V I + j V Q ) ( I I - j
I Q ) = P delivered + j P reactive ##EQU00001.3## P delivered = 1 2
( V I I I + V Q I Q ) ##EQU00001.4## P reactive = 1 2 ( V Q I I - V
I I Q ) ##EQU00001.5## Power = V RMS I RMS = P delivered + j P
reactive ##EQU00001.6##
[0064] The impedance of the load represented by the ultrasonic
transducer 8 and the vessel/tissue gripped by the forceps 9 (or
just the impedance of the forceps 9 and the vessel/tissue gripped
by the forceps 9 if the electrical energy is directly applied to
the forceps 9) can be determined from:
Z Load = ( V I + j V Q ) ( I I + jI Q ) = ( V I + j V Q ) ( I I - j
I Q ) ( I I + jI Q ) ( I I - j I Q ) = ( V I I I + V Q I Q + j V Q
I I - jV I I Q ) 2 I RMS 2 = R Load + jX Load ##EQU00002##
[0065] An alternative way of computing R.sub.Load and X.sub.Load is
as follows:
R Load = P delivered 2 I RMS 2 X Load = P reactive 2 I RMS 2
##EQU00003##
and the phase difference between the load voltage and the load
current can be determined from:
Phase.sub.measured=a tan 2(P.sub.reactive, P.sub.delivered)
[0066] A computationally efficient, approximation to the atan2
function can be made using look up tables and interpolation in
fixed point arithmetic, or using a `CORDIC` like algorithm,
[0067] Limits
[0068] As with any system, there are certain limits that can be
placed on the power, current and voltage that can be delivered
either to the ultrasonic transducer 8 or to the forceps 9. The
limits used in this embodiment and how they are controlled will now
be described.
[0069] In this embodiment, the drive circuitry 29 is designed to
deliver ultrasonic energy into tissue or to deliver electrical
energy into tissue with the following requirements:
[0070] 1) Supplied with a nominally 14V DC supply
[0071] 2) Substantially sinusoidal output waveform at approximately
50 kHz in the case of ultrasonic operation
[0072] 3) Substantially sinusoidal output waveform at approximately
500 kHz in the case of RF electrical operation
[0073] 4) Power limited output of 90W in the case of ultrasonic
operation
[0074] 5) Power limited output of 100W in the case of electrical
operation
[0075] 6) Current limited to 1.4 A.sub.rms and voltage limited to
130V.sub.rms in the case of ultrasonic operation
[0076] 7) Current limited to 1.4 A.sub.rms and voltage limited to
100V.sub.rms in the case of electrical operation
[0077] 8) In the case of ultrasonic operation, the measured phase
is greater than a system defined phase limit
[0078] The power controller 85 maintains data defining these limits
and uses them to control the decision about whether to increase or
decrease the waveform period or whether to skip pulses of the
control signals given the latest measured power, load impedance
and/or measured phase. In this embodiment, when operating in the
ultrasonic mode of operation, the phase limit that is used depends
on the measured load impedance. In particular, the power controller
85 maintains a look up table (not shown) relating load impedance to
the phase limit; and the values in this table limit the phase so
that when the measured load impedance is low (indicating that the
jaws of the forceps 9 are open and not gripping tissue or a
vessel), the delivered power is reduced (preferably to zero).
[0079] As discussed above, one of the ways to control the operation
of the device (when operating in the electrical mode of operation)
is to maximise the drive efficiency. When controlling the device in
this way, the power controller 85 tracks a maximum power delivery
condition as the load changes. The way that this can be done will
now be described.
[0080] Maximum Power Delivery Tracking Condition
[0081] The complex impedance of the circuitry shown in FIG. 3 (when
operating in the electrical mode of operation and with inductor 48
removed) can be approximated by the following equation:
Z = j 2 .pi. f L S F + 1 j 2 .pi. f C S F + j 2 .pi. f L M F R load
_ ref j 2 .pi. f L M F + R load _ ref + R s ##EQU00004##
[0082] Where:
[0083] Road.sub.load_ref is the load resistance referred to the
primary (by the square of the turns ratio); and R.sub.s represents
the equivalent series resistance of the inductor, transformer
capacitor and switching devices. This complex impedance may be
rewritten as:
Z = j 2 .pi. f L S F + 1 j 2 .pi. f C S F + 4 .pi. 2 f 2 L M F 2 R
load _ ref 4 .pi. 2 f 2 L M F 2 + R load _ ref 2 + j 2 .pi. f L M F
R load _ ref 2 4 .pi. 2 f 2 L M F 2 + R load _ ref 2 + R s
##EQU00005##
[0084] Therefore, the real part of this complex impedance is:
( Z ) = 4 .pi. 2 f 2 L M F 2 R load _ ref 4 .pi. 2 f 2 L M F 2 + R
load _ ref 2 + R s ##EQU00006##
[0085] And the imaginary part of this complex impedance is:
( Z ) = 2 .pi. fL S F - 1 2 .pi. fC S F + 2 .pi. fL M F R load _
ref 2 4 .pi. 2 f 2 L M F 2 + R load _ ref 2 ##EQU00007##
[0086] When the drive frequency (f) corresponds to the resonant
frequency of this complex impedance, the imaginary part (Z)=0.
Therefore, the power controller 85 can vary the drive frequency (t)
to keep the imaginary part (Z) at or around zero using a phase
locked loop. Indeed, it can be shown that when (Z)=0 the maximum
power (for a given supply voltage) is delivered to the load.
[0087] FIG. 7 is a contour plot showing the power contours than can
be delivered to the load versus the drive frequency and the load
resistance (Rload). As shown in FIG. 7, the power that can be
delivered varies with the load resistance and the drive frequency.
FIG. 7 also shows the line 92 of maximum power delivery that can be
achieved as the load resistance and drive frequency change.
Therefore, the power controller 85 can use the measured value of
Rload together with stored data defining the line 92 shown in FIG.
7 (which may be a look-up-table) to determine the corresponding
drive frequency to be used. In this way, the microprocessor 23 will
track along the line 92 shown in FIG. 7 as the load resistance
changes during the cutting/cauterisation process.
[0088] One of the advantages of this approach is that it enables a
useful operating condition at low values of Rload, in particular
for values of Rload less than the critical value (i.e. when
R.sub.load_ref<2.pi.fL.sup.F.sub.M), in which a maximum power
will be delivered even if this is below the desired power level.
However, operating along the line 92 of maximum power delivery can
result in some of the above system limits being breached unless
further control action is taken. In the preferred embodiment, this
further control action is to use pulse skipping techniques until
the average power, current and/or voltage is below the relevant
limit. For example, as can be seen from the measurements described
above, the measured voltage and current processing module 83 can
determine the delivered power, the RMS voltage and the RMS current.
The power controller 85 can therefore use these values to skip one
or more pulses of the CTRL control signals until the measured
voltage and current values are below the relevant system limits and
the delivered power is at or below the power set-point defined by
the medical device control module 89.
[0089] Pulses may be skipped, for example, by passing the pulses
generated by the control signal generator 87 through a logic gate
(not shown) and selectively suppressing pulses that are generated
by the control signal generator 87 by controlling the logic level
of another input to the logic gate. For example, the pulses of each
control signal that is generated by the control signal generator 87
may be passed through an AND gate, with another input of the AND
gate being generated by the power controller 85 and being a logic
"1" when the pulses are to be output to the FET gate drives 37 as
normal and being a logic "0" when the pulses are to be skipped or
suppressed. Other pulse skipping techniques could of course be
used.
[0090] Medical Device Control Module
[0091] As mentioned above, the medical device control module 89
controls the general operation of the cauterisation/cutting device
1. It receives user inputs via the user input module 91. These
inputs may specify that the jaws of the forceps 9 are now gripping
a vessel or tissue and that the user wishes to begin
cutting/cauterisation and specify whether ultrasonic energy or
electrical energy is to be applied to the vessel/tissue. In
response, in this embodiment, the medical device control module 89
initiates a cutting/cauterisation control procedure. Initially, the
medical device control module 89 sends an initiation signal to the
power controller 85 and obtains the load impedance measurements
determined by the measured voltage and current processing module
83. The medical device control module 89 then checks the obtained
load impedance to make sure that the load is not open circuit or
short circuit. If it is not, then the medical device control module
89 starts to vary the power set point to perform the desired
cutting/cauterisation and sets the initial period/frequency of the
drive signal to be generated. As discussed above, for ultrasonic
operation, the initial frequency of the drive signal will be set
around 50 kHz and for RF electrical operation, the initial
frequency will be set around 500 kHz.
[0092] Voltage/Current Sensing Circuitry As shown in FIG. 2,
voltage sensing circuitry 31 is provided to sense the load voltage
applied to the load and current sensing circuitry 33 is provided to
sense the current applied to the load. The sensed signals are
supplied to the microprocessor 23 for use in controlling the
operation of the medical device. There are various ways of sensing
the load voltage and the load current and some of these will now be
described.
[0093] FIG. 8a illustrates the primary side of the first
transformer circuit 42-1 and one way in which the current sensing
circuitry 33 obtains a measure of the load current. As shown, the
current sensing circuitry 33 comprises an additional inductor
turn(s) 67 which link the flux present in inductor 47 (or inductor
49) and that consequently outputs a voltage across inductor 67 that
varies with the rate of change of load current. The voltage across
the inductor 67 is a bipolar voltage whose amplitude is directly
proportional to the rate of change of load current and the number
of turns in 67. This bipolar voltage is scaled and converted into a
unipolar voltage suitable for input to the microprocessor 23 by the
op-amp circuit 69-1 which outputs a measured voltage (V.sup.meas).
This measured voltage will also depend on the current flowing in
the inductor 47 and so will also depend on the current flowing on
the secondary side of the transformer circuit 42-1 and thus the
current flowing through the load. As the ratio of the number of
turns of inductor 47 to inductor 67 is known, the measured voltage
and current processing module 83 can use V.sup.meas to determine
the voltage across inductor 47. The voltage across inductor 47 is
related to the current flowing through the inductor 47 by V=Ldi/dt.
As the inductance of inductor 47 is known, the measured voltage and
current processing module 83 can determine the current flowing in
the primary side of the transformer circuit 42-1 by integrating the
voltage across the inductor 47 and by scaling the result to account
for the inductance of the inductor 47 (and the scaling of the
op-amp-circuit 69-1). This current measure can then be converted
into a suitable measure of the load current (i.sub.L) by a further
scaling to take into account the number of turns between inductor
49 and inductor 53. Of course, the measured voltage and current
processing module 83 does not need to integrate the voltage across
the inductor 47--as the measured signals are sinusoidal and so
integration can be achieved by applying a suitable scaling factor
and a 90 degree phase shift. Thus, the measured voltage and current
processing module 83 can determine the load current by applying a
suitable (pre-stored) scale factor to the measured voltage
(V.sup.meas) and by applying a suitable 90 degree phase shift
(which can be achieved simply by swapping the order of the I and Q
samples as discussed above).
[0094] FIG. 8b illustrates the primary side of the first
transformer circuit 42-1 and another way in which the current
sensing circuitry 33 can obtain a measure of the load current. As
shown, in this case, the current sensing circuitry 33 measures the
voltage across the capacitor 45. The voltage across the capacitor
45 is a bipolar voltage. This bipolar voltage is scaled and
converted into a unipolar voltage suitable for input to the
microprocessor 23 by the op-amp circuit 69-2 which outputs a
measured voltage (V.sup.meas)This voltage is related to the current
flowing in the primary side of the transformer circuit 42 by
I=CdV.sup.meas/dt and thus the current flowing through the load. As
the capacitance of the capacitor 45 is known, the measured voltage
and current processing module 83 can determine the current flowing
in the primary side of the transformer circuit 42-1 by
differentiating the voltage across the capacitor 45 and by scaling
the result to account for the capacitance of the capacitor 45 (and
the scaling of the op-amp-circuit 69-2). This current measure can
then be converted into a suitable measure of the load current
(i.sub.L) by a further scaling to take into account the number of
turns between inductor 49 and inductor 53. Of course, the measured
voltage and current processing module 83 does not need to
differentiate the voltage across the capacitor 45--as the measured
signals are sinusoidal and so differentiation can be achieved by
applying a suitable scaling factor and a 90 degree phase shift.
Thus, the measured voltage and current processing module 83 can
determine the load current by applying a suitable (pre-stored)
scale factor to the measured voltage (V.sup.meas) and by applying a
suitable 90 degree phase shift (which can be achieved simply by
swapping the order of the I and Q samples as discussed above).
[0095] FIG. 8c schematically illustrates how a measure of the load
voltage can be determined. FIG. 8c shows the secondary side of the
first transformer circuit 42 and illustrates the use of a voltage
divider circuit (in this case formed by resistors R1 and R2), with
the voltage across resistor R2 being input to the op-amp circuit
69-3. Therefore, by applying an appropriate scale on the measured
voltage from op-amp 69-3, the measured voltage and current
processing module 83 can determine the load voltage.
[0096] The sensing circuits described above use op-amp circuits 69
to convert the bipolar drive signals into unipolar voltages that
are suitable for input to the microprocessor 23. The use of such
op-amp circuits has a number of disadvantages, including that they
are costly, they consume power and they requires space within the
electronics. These are important factors when the device is
designed to be battery powered and the electronics are housed
within the handle 5 of the device. FIG. 9 illustrates various
sensing circuits that can be used without an op-amp. The circuit of
FIG. 9a is suitable when the AC drive signal is unipolar. This may
be achieved by replacing the full bridge signal generator 27 with a
half bridge signal generator. This would mean removing, for
example, the FETs 41-3 and 41-4 and connecting node B to ground. In
this case, the microprocessor 23 only needs to generate one control
signal (CTRL.sub.1) to control FETs 41-1 and 41-2. The circuit of
FIG. 9b is suitable for both unipolar and bipolar drive
signals.
[0097] In FIG. 9a, the capacitor 45 has been moved to be between
the inductor 49 that forms the primary of the transformer circuit
42-1 and ground (GND). The current sensing circuitry 33 is arranged
to measure the voltage across the capacitor 45 via a potential
divider formed by resistors R1//R2 and R3. The resistor R1 connects
the output of a DC blocking capacitor C.sub.B to a supply voltage
rail of the microprocessor 23 (in this case at 3.3V) through the
switch 121; and resistor R2 connects the output of the DC blocking
capacitor C.sub.B to a reference potential (in this case ground).
The resistors R1 and R2 therefore provide a divider circuit that
applies a DC bias to the measured AC signal. The DC blocking
capacitor prevents this DC bias from coupling to the drive circuit.
Typically, the resistors R1 and R2 are equal so that the DC bias
will be at 1.65V. Thus, the voltage output from the sensing circuit
33 will be an AC voltage having a mid-rail value of about 1.65 V
and whose peak voltage will be a proportion of the voltage across
the capacitor 45. The potential divider formed by resistors R1//R2
and R3 is such that the peak to peak amplitude of the AC signal
passed to the microprocessor 23 is less than the 3.3V input range
of the microprocessor 23. If the microprocessor 23 operates at a
different voltage rail (for example 5V), then the values of the
resistors R1, R2 and R3 can be adjusted accordingly. To minimise
current drawn either from the 3.3V rail or from the transformer
circuit 42-1, the resistors R1, R2 and R3 can have relatively large
values and typical values for these resistors are: R1=R2=200.OMEGA.
and R3=1000.OMEGA.. The values of R1 and R2 should be chosen to
meet the input impedance requirements of the Analogue to digital
converter used to sample the signals. The switch 121 allows the
microprocessor 23 to disconnect the sensing circuit 33 from the
3.3V rail-so that when sensing is not required, the circuit 33 does
not consume any power.
[0098] FIG. 9b illustrates the way in which similar circuits can be
provided on the secondary side of the transformer circuit 42-1. In
particular, FIG. 9b shows the voltage sensing circuit 31 used to
obtain a measure of the load voltage V.sub.L via a voltage divider
formed by capacitors C1 and C2. If the capacitors C1 and C2 were
replaced with resistors, then a blocking capacitor C.sub.B should
be provided before the divider circuit formed by resistors R1 and
R2 in the manner shown in FIG. 9a. FIG. 9b also shows the current
sensing circuit 33, which senses the load current by sensing the
voltage across capacitor C3 via a potential divider formed by
resistors R4//R5 and R6. As shown, a DC blocking capacitor C.sub.B
is provided to enable a DC bias signal to be added to the output to
the microprocessor via the divider circuit formed from resistors R4
and R5. As before, the switches 121-1 and 121-2 allow the
microprocessor to switch off the sensing circuits 31 and 33 when
sensor signals are not required. As those skilled in the art will
appreciate, the "op-ampless" sensing circuits 31 and 33 illustrated
in FIG. 9 are cheaper to manufacture, can be made to consume less
power and take up less space on the circuit board than the op-amp
circuits illustrated in FIG. 8.
[0099] The circuits illustrated in FIGS. 8 and 9 were used to
obtain measurements from the first transformer circuit 42-1 of the
drive circuit 29. As those skilled in the art will appreciate, the
same or similar sensing circuits would be provided for sensing
signals in the second transformer circuit 42-2. Further, although
the sensing circuits illustrated in FIG. 9 sense the voltage across
a capacitor, the circuits could also sense the voltage across
another impedance element, such as across a resistor of the
transformer circuit 42.
[0100] Active Battery Protection
[0101] Typically, with battery operated devices such as the medical
device 1 described above, a fuse is provided between the batteries
and the electric circuitry, to protect the battery from damage
caused by short circuits and the like. However, standard fuses have
a resistance of about 10 mOhms. With such a standard fuse, when 10
A is drawn from the batteries, approximately 1 W is dissipated
through the fuse. An active fuse circuit 130 that is used in this
embodiment will now be described with reference to FIG. 10 that
reduces the power dissipation associated with such standard
fuses.
[0102] FIG. 10 shows the batteries 15, which supply the 14V rail
and the GND rail to the circuitry shown in FIG. 3. The active fuse
circuit 130 comprises a differential amplifier 131 that measures
the voltage drop across part of a PCB conductor trace 133 that is
connected between the 14V rail and the positive terminal of the
batteries (V.sub.bat+). The conductor trace 133 has a resistance of
approximately 1-2 mOhms and so the voltage drop is proportional to
the current that is being drawn from the batteries 15. The measured
voltage drop is low pass filtered by the low pass filter 135 to
avoid transient spikes triggering the fuse circuit--so that it is
just the voltage corresponding to the DC current drawn from the
batteries 15 that will pass through the filter 135. In this
embodiment, the low pass filter 135 has a cut-off frequency of
about 10 Hz. The output from the low pass filter 135 is then
compared with a reference voltage (V.sub.ref) using a latching
comparator 137. The reference voltage is set in advance to
correspond to a desired limit on the current drawn from the
batteries. In this embodiment, V.sub.ref is set to correspond to a
current limit of 15 A. When the voltage drop across the trace 133
is less than the reference voltage the output of the comparator 137
remains at a high value--which maintains the FET switch 139 on and
so current can be drawn from the batteries 15 by the bridge signal
generator 27. However, when the voltage drop across the trace 133
is greater than the reference voltage, then the output of the
comparator 137 goes low and it is maintained low even if the
current drawn from the batteries drops below the defined limit.
When the comparator output is low the FET 139 is switched off,
thereby disconnecting the batteries 15 from at least the bridge
signal generator 27.
[0103] In this embodiment, the FET 139 is an N-channel enhanced
mode switch having an on resistance of just 2 mOhms. This means
that when the switch 139 is switched on and 10 A is being drawn
from the batteries, just 0.2 W is dissipated through the switch
139.
[0104] In this embodiment, when the comparator 137 is triggered and
the switch 139 is switched off (open circuit), the batteries 15
have to be removed to reset the active fuse circuitry 130--as the
opening of the switch 139 disconnects all the electronics except
the circuit components of the active fuse 130 from the batteries.
Alternatively, if the microprocessor 23 (or some other control
circuitry) is powered directly from the batteries, then the
comparator 137 could be reset either in response to a user input
(for example in response to the user pressing a reset button or the
like) or in response to some other trigger event (such as after a
predetermined time-out period).
Modifications and Alternatives
[0105] A medical cauterisation/cutting device has been described
above. As those skilled in the art will appreciate, various
modifications can be made and some of these will now be described.
Other modifications will be apparent to those skilled in the
art.
[0106] In the above embodiment, various operating frequencies,
currents, voltages etc were described. As those skilled in the art
will appreciate, the exact currents, voltages, frequencies,
capacitor values, inductor values etc. can all be varied depending
on the application and any values described above should not be
considered as limiting in any way. However, in general terms, the
circuit described above has been designed to provide a drive signal
to a medical device, where the delivered power is desired to be at
least 10 W and preferably between 10 W and 200 W; the delivered
voltage is desired to be at least 20 V.sub.Rms and preferably
between 30 V.sub.Rms and 120 V.sub.Rms; the delivered current is
designed to be at least 0.5 A.sub.RMS and preferably between 1
A.sub.RMS and 2 A.sub.RMS; and the drive frequency for ultrasonic
operation is desired to be at least 20 kHz and preferably between
30 kHz and 80 kHz; and the drive frequency for RF operation is
desired to be at least 100 kHz and preferably between 250 kHz and 1
MHz.
[0107] In the above embodiment, the resonant circuits 43-1 and 43-2
were formed from capacitor-inductor-inductor elements. As those
skilled in the art will appreciate, other resonant circuit designs
with multiple capacitors and inductors in various series and
parallel configurations or simpler LC resonant circuits may also be
used. Also, in some applications there is no need for a transformer
to step-up the drive voltage, as the FETs can deliver the required
drive voltage.
[0108] FIG. 1 illustrates one way in which the batteries and the
control electronics can be mounted within the handle of the medical
device. As those skilled in the art will appreciate, the form
factor of the handle may take many different designs. Indeed, it is
not essential for the device to be battery powered, although this
is preferred for some applications to avoid the need for power
cords and the like.
[0109] The embodiment described above included a description of
various novel features, including the novel ability to selectively
apply ultrasonic energy or RF energy to the tissue gripped by the
forceps, the novel way in which the microprocessor controlled the
operation of the device in the electrical mode of operation; the
way in which load current/voltage is measured and the way in which
the batteries are protected using an active fuse circuit. As those
skilled in the art will appreciate, these novel features do not
need to be employed together. For example, the current/voltage
sensing techniques described above can be used with other devices
as can the active fuse circuit. Similarly, the way in which the
electrical mode of operation is controlled by tracking the maximum
power delivery condition and by using pulse skipping techniques can
be used in a device that does not have an ultrasonic
transducer.
[0110] In the above embodiment, an exemplary control algorithm for
performing the cutting/cauterisation of the vessel or tissue
gripped by the forceps was described. As those skilled in the art
will appreciate, various different procedures may be used and the
reader is referred to the literature describing the operation of
such cutting/cauterisation devices for further details.
[0111] In the above embodiment, four FET switches were used to
convert the DC voltage provided by the batteries into an
alternating signal at the desired frequency. As those skilled in
the art will appreciate, it is not necessary to use four
switches--two switches may be used instead (using a half bridge
circuit). Additionally, although FET switches were used, other
switching devices, such as bipolar transistor switches may be used
instead. However, MOSFETs are preferred due to their superior
performance in terms of low losses when operating at the above
described frequencies and current levels.
[0112] In the above embodiment, the I & Q sampling circuitry 81
oversampled the sensed voltage/current signal in the ultrasonic
mode of operation and undersampled the sensed voltage/current
signal in the electrical mode of operation. As those skilled in the
art will appreciate, this is not essential. Because of the
synchronous nature of the sampling, samples may be taken more than
once per period or once every n.sup.th period if desired. The
sampling rate used in the above embodiment was chosen to maximise
the rate at which measurements were made available to the power
controller 85 and the medical device control module 89 as this
allows for better control of the applied power during the
cauterisation process.
[0113] In the above embodiment, a 14V DC supply was provided. In
other embodiments, lower (or higher) DC voltage sources may be
provided. In this case, a larger (or smaller) transformer turns
ratio may be provided to increase the load voltage to the desired
level or lower operating voltages may be used.
[0114] In the above embodiment, the medical device was arranged to
deliver a desired power (in the form of ultrasonic energy or
electrical energy) to the tissue/vessel gripped by the forceps. In
an alternative embodiment, the device may be arranged to deliver a
desired current or a desired voltage level to the ultrasonic
transducer or the forceps.
[0115] In the above embodiment the battery is shown integral to the
medical device. In an alternative embodiment the battery may be
packaged so as to clip on a belt on the surgeon or simply be placed
on the Mayo stand. In this embodiment a relatively small two
conductor cable would connect the battery pack to the medical
device.
[0116] In the above embodiment, a microprocessor based control
circuitry was provided. This is preferred due to the ease with
which the microprocessor can be programmed to perform the above
control actions using appropriate computer software. Such software
can be provided on a tangible carrier, such as a CD-ROM or the
like. Alternatively, hardware control circuitry can be used in
place of the microprocessor based circuitry described above.
[0117] In the above embodiment, the user controlled whether the
energy delivered to the vessel/tissue was ultrasonic energy or RF
electrical energy. In alternative embodiments, the microprocessor
may control the selection based on an internally generated control
signal or in response to a control signal received from another
device.
[0118] In the above embodiment, the active fuse circuit opened a
switch that disconnected the bridge signal generator from the
batteries. In an alternative embodiment, the active fuse circuit
could instead switch in a large impedance between the batteries and
the bridge signal generator to limit the current drawn from the
batteries. Also, the switch could be used to disconnect the
positive voltage supply from the signal generator instead of
disconnecting the negative terminal of the batteries.
* * * * *