U.S. patent application number 16/731432 was filed with the patent office on 2020-07-02 for pulse-shaped orthogonal frequency division multiplexing.
This patent application is currently assigned to IDAC Holdings, Inc.. The applicant listed for this patent is IDAC Holdings, Inc.. Invention is credited to Erdem BALA, Daniel R. COHEN, Jialing LI, Rui YANG.
Application Number | 20200213168 16/731432 |
Document ID | / |
Family ID | 50116187 |
Filed Date | 2020-07-02 |
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United States Patent
Application |
20200213168 |
Kind Code |
A1 |
BALA; Erdem ; et
al. |
July 2, 2020 |
PULSE-SHAPED ORTHOGONAL FREQUENCY DIVISION MULTIPLEXING
Abstract
A method and apparatus for performing pulse shaping using
different windowing functions for different sub-bands of a
transmission is disclosed. A method for use in a wireless
transmit/receive unit (WTRU) may include the WTRU receiving data
symbols. The WTRU may assign the data symbols to a plurality of
subcarriers in different sub-bands and map the data symbols on each
of the plurality of subcarriers in the different sub-bands to a
plurality of corresponding subcarriers of an inverse fast Fourier
transform (IFFT) block. The WTRU may take an IFFT of the block for
each sub-band and pad an output of the IFFT block with a prefix and
a postfix for each sub-band. The WTRU may apply a windowing
function to an output of the padding for each sub-band and form a
composite signal for transmission by adding an output of the
windowing of each sub-band. The WTRU may transmit the signal.
Inventors: |
BALA; Erdem; (East Meadow,
NY) ; YANG; Rui; (Greenlawn, NY) ; LI;
Jialing; (San Diego, CA) ; COHEN; Daniel R.;
(Huntington, NY) |
|
Applicant: |
Name |
City |
State |
Country |
Type |
IDAC Holdings, Inc. |
Wilmington |
DE |
US |
|
|
Assignee: |
IDAC Holdings, Inc.
Wilmington
DE
|
Family ID: |
50116187 |
Appl. No.: |
16/731432 |
Filed: |
December 31, 2019 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
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14766040 |
Aug 5, 2015 |
10523475 |
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PCT/US2014/014717 |
Feb 4, 2014 |
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16731432 |
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61871461 |
Aug 29, 2013 |
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61760938 |
Feb 5, 2013 |
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Current U.S.
Class: |
1/1 |
Current CPC
Class: |
H04L 27/2649 20130101;
H04L 27/2607 20130101; H04L 27/2634 20130101; H04L 27/2602
20130101; H04L 25/03834 20130101 |
International
Class: |
H04L 25/03 20060101
H04L025/03; H04L 27/26 20060101 H04L027/26 |
Claims
1. (canceled)
2. A method implemented by an 802.11 device for transmitting a
signal, the method comprising: mapping a plurality of data symbols
on each of a plurality of subcarriers in a plurality of sub-bands;
performing an IFFT on each of the plurality of sub-bands to
generate an output for each of the plurality of sub-bands; padding
the output with a prefix or a postfix for each sub-band; applying a
windowing function to the padded output for each sub-band; forming
a composite signal for transmission from each of the windowed
padded output; and transmitting the composite signal.
3. The method of claim 2, wherein different windowing functions are
applied to different sub-bands.
4. The method of claim 2, wherein the prefix or the postfix is a
guard interval.
5. The method of claim 2, wherein the subcarriers in the sub-bands
are non-contiguous.
6. The method of claim 2, wherein the prefix for each sub-band is
of a different length.
7. An 802.11 device configured to transmit a signal, the WTRU
comprising: a processor configured to: map a plurality of data
symbols on each of a plurality of subcarriers in a plurality of
sub-bands; perform an IFFT on each of the plurality of sub-bands to
generate an output signal for each of the plurality of sub-bands;
padding an output of the IFFT block with a prefix or a postfix for
each sub-band; apply a windowing function to the padded output for
each sub-band; form a composite signal for transmission from each
of the windowed padded output; and a transmitter, operatively
coupled to the processor, configured to transmit the composite
signal.
8. The 802.11 device of claim 7, wherein different windowing
functions are applied to different sub-bands.
9. The 802.11 device of claim 7, wherein the prefix or the postfix
is a guard interval.
10. The 802.11 device of claim 7, wherein the subcarriers in the
sub-bands are non-contiguous.
11. The 802.11 device of claim 7, wherein the prefix for each
sub-band is of a different length.
Description
CROSS REFERENCE TO RELATED APPLICATIONS
[0001] This application is a continuation of U.S. patent
application Ser. No. 14/766,040 filed on Aug. 5, 2015 which issued
as U.S. Pat. No. 10,523,475 on Dec. 31, 2019, which is the U.S.
National Stage, under 35 U.S.C. .sctn. 371, of International Patent
Application No. PCT/US2014/014717, filed on Feb. 4, 2014, which
claims the benefit of U.S. Provisional Application No. 61/871,461
filed on Aug. 29, 2013 and U.S. Provisional Application No.
61/760,938 filed on Feb. 5, 2013, the contents of which are hereby
incorporated by reference herein.
BACKGROUND
[0002] Multicarrier modulation (MCM) is based on the idea of
splitting a high-rate wideband signal into multiple lower-rate
signals, where each signal occupies a narrower band. Orthogonal
frequency division multiplexing (OFDM) has proved itself as one of
the most popular MCM techniques and is currently used in many
wireless communication systems such as 3rd Generation Partnership
Project (3GPP) Long Term Evolution (LTE), 802.11, etc. OFDM offers
many advantages such as robustness to multipath propagation, simple
equalization, a simple transceiver architecture and efficient use
of the available bandwidth through overlapping subchannels. On the
other hand, OFDM has several disadvantages such as spectral leakage
due to high sidelobes, and high peak-to-average power ratio
(PAPR).
[0003] The demand for higher data rates has been increasing
significantly. Several techniques have been studied and proposed to
meet this demand, such as overlaying small cells over macro cells
to allow spectral reuse, opening new bands to wireless
communication, and utilizing the bandwidth more efficiently by
spectrum sharing via cognitive radio. Since wireless systems are
evolving towards a "network of networks" architecture where many
networks are expected to share the spectrum, spectrally agile
waveforms with small out-of-band leakage are important. To that
end, the adjacent channel interference created by the spectral
leakage of OFDM makes this waveform unsuitable for these
networks.
[0004] As an alternative to OFDM, filter bank multicarrier (FBMC)
modulation schemes, specifically OFDM-Offset quadrature amplitude
modulation (QAM), have recently taken interest. OFDM-OQAM is
another MCM technique where data on each sub-carrier is shaped with
an appropriately designed pulse so that sidelobes are lower. A real
data symbol is transmitted in each subchannel and on each OFDM-OQAM
symbol. Consecutive OFDM-OQAM symbols are staggered. Adjacent
subchannels overlap to maximize the spectral efficiency, creating
inter-carrier interference (ICI); and consecutive OFDM-OQAM symbols
interfere with each other due to the long pulse, creating
intersymbol interference (ISI). In an ideal single path Additive
White Gaussian Noise (AWGN) channel, perfect orthogonality may be
achieved and ISI/intercarrier interference (ICI) may be cancelled.
The OFDM-OQAM transmitter and receiver may be implemented in an
efficient manner by using the polyphase filterbanks. Although
OFDM-OQAM offers less spectral leakage, its implementation in
practical systems poses several challenges due to its complexity,
latency, and more complex channel estimation and equalization
algorithms in doubly dispersive channels. Therefore, it is
desirable to design an OFDM-like, but spectral contained waveform
with improved out-of-band emission characteristics.
[0005] Therefore, there is a need for an advanced waveform for
spectral agile systems that is capable of sharing opportunistically
available and non-contiguous spectrum resources with other users.
The characteristics of such a waveform should include low
out-of-band emission (OOBE), low in-band distortion, low
complexity, low latency, low PAPR, robustness to frequency and
timing asynchronous, and robustness to power amplifier (PA)
nonlinearity. The existing baseband waveforms in those systems
possess very large OOBE, which may make it difficult for the
existing baseband waveforms to be used in spectral agile
systems.
SUMMARY
[0006] Methods and apparatus for performing transmitter and
receiver side pulse shaping using different windowing functions for
different sub-bands of a transmission are disclosed. A method for
use in a wireless transmit/receive unit (WTRU) for performing
transmitter side pulse shaping may include the WTRU receiving data
symbols. The WTRU may assign the data symbols to a plurality of
subcarriers in the different sub-bands and map the data symbols on
each of the plurality of subcarriers in the different sub-bands to
a plurality of corresponding subcarriers of an inverse fast Fourier
transform (IFFT) block. The WTRU may take an IFFT of the block for
each sub-band and pad an output of the IFFT block with a cyclic
prefix (CP) and a postfix for each sub-band. The WTRU may apply a
windowing function to an output of the padding for each sub-band
and form a composite signal for transmission by adding an output of
the windowing of each sub-band. The WTRU may transmit the
signal.
[0007] A method for use in a wireless transmit/receive unit (WTRU)
for performing receiver side pulse shaping may include a WTRU
receiving a signal comprising data symbols and assigning the data
symbols to a plurality of subcarriers in the different sub-bands.
The WTRU may apply a receive windowing function to each sub-band
and map the data symbols on each of the plurality of subcarriers in
the different sub-bands to a plurality of corresponding subcarriers
of a fast Fourier transform (FFT) block. The WTRU may take an FFT
of the block for each sub-band and apply further processing to an
output of the FFT block for each sub-band.
[0008] Methods and apparatus for performing transmitter and
receiver side pulse shaping with zero-padded OFDM instead of OFDM
with cyclic prefix (CP) are also disclosed. Methods and apparatus
for improving performance of receive windowing, including
interference cancellation, CP overhead reduction, and the
utilization of CP samples, are also disclosed.
BRIEF DESCRIPTION OF THE DRAWINGS
[0009] A more detailed understanding may be had from the following
description, given by way of example in conjunction with the
accompanying drawings wherein:
[0010] FIG. 1A is a system diagram of an example communications
system in which one or more disclosed embodiments may be
implemented;
[0011] FIG. 1B is a system diagram of an example wireless
transmit/receive unit (WTRU) that may be used within the
communications system illustrated in FIG. 1A;
[0012] FIG. 1C is a system diagram of an example radio access
network and an example core network that may be used within the
communications system illustrated in FIG. 1A;
[0013] FIG. 2 is a flow chart of an example transmit windowing
implementation procedure using cyclic prefix (CP);
[0014] FIG. 3 is a block diagram of an example transmitter module
configured to implement transmit windowing with cyclic prefix
(CP);
[0015] FIG. 4 is a diagram of an example application of transmit
windowing in accordance with the method described in FIG. 2;
[0016] FIG. 5 is a flow chart of an example transmitter windowing
procedure with zero-padding;
[0017] FIG. 6 is a diagram of an example application of transmit
windowing with zero-padding in accordance with the method described
in FIG. 5.
[0018] FIG. 7 is an example diagram of receiver side procedures for
windowing with zero-padding;
[0019] FIG. 8 is a diagram of an example method of applying
different windowing functions to different sub-bands of the
transmission band;
[0020] FIG. 9 is an example diagram of the boundaries of receive
windowing;
[0021] FIG. 10 is a block diagram of a receiver module configured
to implement an example type of receive windowing;
[0022] FIG. 11 is a diagram of an example windowing interval in a
practical system;
[0023] FIG. 12 is a diagram of an example windowing interval in a
practical system;
[0024] FIG. 13 is a diagram of a receiver module configured to
implement an example type of receive windowing by point-wise
multiplying;
[0025] FIG. 14 is an example case of transmit windowing;
[0026] FIG. 15 is an example case of transmit windowing;
[0027] FIG. 16 is an example case of transmit windowing;
[0028] FIG. 17 is a diagram of an example implementation of
receiver windowing for non-contiguous sub-bands;
[0029] FIG. 18 is a graph depicting the improvement in BER in an
AWGN channel due to adding back CP samples when IEEE 802.11af is
the underlying radio access medium;
[0030] FIG. 19 is a diagram of an example transmitter capable of
transmitting symbols with varying CP lengths for different
sub-bands;
[0031] FIG. 20 is a graph illustrating the received interference
power in dB at the first receiver;
[0032] FIG. 21 is a graph showing a close-up view of FIG. 20 that
depicts the first half of the spectrum allocated to the first
user.
[0033] FIG. 22 is a diagram of an example transmitter that uses
transmitter windowing to reduce the ICI when using variable CP
lengths;
[0034] FIG. 23 is a diagram of example receivers that use receiver
windowing to reject the interference when using variable CP
lengths;
[0035] FIG. 24 is a diagram of an example transmitter using
transmitter side filtering when using variable CP lengths;
[0036] FIG. 25 is a diagram of example receivers that uses receiver
side filtering to reject the interference when using variable CP
lengths;
[0037] FIG. 26 is a diagram of an example RB-F-OFDM based
transmitter;
[0038] FIG. 27 is a diagram of an example Type-I per-RB F OFDM
transmit module (F-OFDM Tx.sub.k);
[0039] FIG. 28 is a diagram of an example RB F-OFDM Receiver
(RB-F-OFDM Rx) corresponding to the RB F-OFDM transmitter in FIG.
26;
[0040] FIG. 29 is a diagram of an example Type-I per-RB F-OFDM
receive module (F-OFDM Rx.sub.k);
[0041] FIG. 30 is a diagram of an example frame structure
corresponding to two signals; and
[0042] FIG. 31 is a diagram of an example frame structure
corresponding to two signals.
DETAILED DESCRIPTION
[0043] FIG. 1A is a diagram of an example communications system 100
in which one or more disclosed embodiments may be implemented. The
communications system 100 may be a multiple access system that
provides content, such as voice, data, video, messaging, broadcast,
etc., to multiple wireless users. The communications system 100 may
enable multiple wireless users to access such content through the
sharing of system resources, including wireless bandwidth. For
example, the communications systems 100 may employ one or more
channel access methods, such as code division multiple access
(CDMA), time division multiple access (TDMA), frequency division
multiple access (FDMA), orthogonal FDMA (OFDMA), single-carrier
FDMA (SC-FDMA), and the like.
[0044] As shown in FIG. 1A, the communications system 100 may
include wireless transmit/receive units (WTRUs) 102a, 102b, 102c,
102d, a radio access network (RAN) 104, a core network 106, a
public switched telephone network (PSTN) 108, the Internet 110, and
other networks 112, though it will be appreciated that the
disclosed embodiments contemplate any number of WTRUs, base
stations, networks, and/or network elements. Each of the WTRUs
102a, 102b, 102c, 102d may be any type of device configured to
operate and/or communicate in a wireless environment. By way of
example, the WTRUs 102a, 102b, 102c, 102d may be configured to
transmit and/or receive wireless signals and may include user
equipment (UE), a mobile station, a fixed or mobile subscriber
unit, a pager, a cellular telephone, a personal digital assistant
(PDA), a smartphone, a laptop, a netbook, a personal computer, a
wireless sensor, consumer electronics, and the like.
[0045] The communications systems 100 may also include a base
station 114a and a base station 114b. Each of the base stations
114a, 114b may be any type of device configured to wirelessly
interface with at least one of the WTRUs 102a, 102b, 102c, 102d to
facilitate access to one or more communication networks, such as
the core network 106, the Internet 110, and/or the other networks
112. By way of example, the base stations 114a, 114b may be a base
transceiver station (BTS), a Node-B, an eNode B, a Home Node B, a
Home eNode B, a site controller, an access point (AP), a wireless
router, and the like. While the base stations 114a, 114b are each
depicted as a single element, it will be appreciated that the base
stations 114a, 114b may include any number of interconnected base
stations and/or network elements.
[0046] The base station 114a may be part of the RAN 104, which may
also include other base stations and/or network elements (not
shown), such as a base station controller (BSC), a radio network
controller (RNC), relay nodes, etc. The base station 114a and/or
the base station 114b may be configured to transmit and/or receive
wireless signals within a particular geographic region, which may
be referred to as a cell (not shown). The cell may further be
divided into cell sectors. For example, the cell associated with
the base station 114a may be divided into three sectors. Thus, in
one embodiment, the base station 114a may include three
transceivers, i.e., one for each sector of the cell. In another
embodiment, the base station 114a may employ multiple-input
multiple-output (MIMO) technology and, therefore, may utilize
multiple transceivers for each sector of the cell.
[0047] The base stations 114a, 114b may communicate with one or
more of the WTRUs 102a, 102b, 102c, 102d over an air interface 116,
which may be any suitable wireless communication link (e.g., radio
frequency (RF), microwave, infrared (IR), ultraviolet (UV), visible
light, etc.). The air interface 116 may be established using any
suitable radio access technology (RAT).
[0048] More specifically, as noted above, the communications system
100 may be a multiple access system and may employ one or more
channel access schemes, such as CDMA, TDMA, FDMA, OFDMA, SC-FDMA,
and the like. For example, the base station 114a in the RAN 104 and
the WTRUs 102a, 102b, 102c may implement a radio technology such as
Universal Mobile Telecommunications System (UMTS) Terrestrial Radio
Access (UTRA), which may establish the air interface 116 using
wideband CDMA (WCDMA). WCDMA may include communication protocols
such as High-Speed Packet Access (HSPA) and/or Evolved HSPA
(HSPA+). HSPA may include High-Speed Downlink Packet Access (HSDPA)
and/or High-Speed Uplink Packet Access (HSUPA).
[0049] In another embodiment, the base station 114a and the WTRUs
102a, 102b, 102c may implement a radio technology such as Evolved
UMTS Terrestrial Radio Access (E-UTRA), which may establish the air
interface 116 using Long Term Evolution (LTE) and/or LTE-Advanced
(LTE-A).
[0050] In other embodiments, the base station 114a and the WTRUs
102a, 102b, 102c may implement radio technologies such as IEEE
802.16 (i.e., Worldwide Interoperability for Microwave Access
(WiMAX)), CDMA2000, CDMA2000 1.times., CDMA2000 EV-DO, Interim
Standard 2000 (IS-2000), Interim Standard 95 (IS-95), Interim
Standard 856 (IS-856), Global System for Mobile communications
(GSM), Enhanced Data rates for GSM Evolution (EDGE), GSM EDGE
(GERAN), and the like.
[0051] The base station 114b in FIG. 1A may be a wireless router,
Home Node B, Home eNode B, or access point, for example, and may
utilize any suitable RAT for facilitating wireless connectivity in
a localized area, such as a place of business, a home, a vehicle, a
campus, and the like. In one embodiment, the base station 114b and
the WTRUs 102c, 102d may implement a radio technology such as IEEE
802.11 to establish a wireless local area network (WLAN). In
another embodiment, the base station 114b and the WTRUs 102c, 102d
may implement a radio technology such as IEEE 802.15 to establish a
wireless personal area network (WPAN). In yet another embodiment,
the base station 114b and the WTRUs 102c, 102d may utilize a
cellular-based RAT (e.g., WCDMA, CDMA2000, GSM, LTE, LTE-A, etc.)
to establish a picocell or femtocell. As shown in FIG. 1A, the base
station 114b may have a direct connection to the Internet 110.
Thus, the base station 114b may not be required to access the
Internet 110 via the core network 106.
[0052] The RAN 104 may be in communication with the core network
106, which may be any type of network configured to provide voice,
data, applications, and/or voice over internet protocol (VoIP)
services to one or more of the WTRUs 102a, 102b, 102c, 102d. For
example, the core network 106 may provide call control, billing
services, mobile location-based services, pre-paid calling,
Internet connectivity, video distribution, etc., and/or perform
high-level security functions, such as user authentication.
Although not shown in FIG. 1A, it will be appreciated that the RAN
104 and/or the core network 106 may be in direct or indirect
communication with other RANs that employ the same RAT as the RAN
104 or a different RAT. For example, in addition to being connected
to the RAN 104, which may be utilizing an E-UTRA radio technology,
the core network 106 may also be in communication with another RAN
(not shown) employing a GSM radio technology.
[0053] The core network 106 may also serve as a gateway for the
WTRUs 102a, 102b, 102c, 102d to access the PSTN 108, the Internet
110, and/or other networks 112. The PSTN 108 may include
circuit-switched telephone networks that provide plain old
telephone service (POTS). The Internet 110 may include a global
system of interconnected computer networks and devices that use
common communication protocols, such as the transmission control
protocol (TCP), user datagram protocol (UDP) and the internet
protocol (IP) in the TCP/IP internet protocol suite. The networks
112 may include wired or wireless communications networks owned
and/or operated by other service providers. For example, the
networks 112 may include another core network connected to one or
more RANs, which may employ the same RAT as the RAN 104 or a
different RAT.
[0054] Some or all of the WTRUs 102a, 102b, 102c, 102d in the
communications system 100 may include multi-mode capabilities,
i.e., the WTRUs 102a, 102b, 102c, 102d may include multiple
transceivers for communicating with different wireless networks
over different wireless links. For example, the WTRU 102c shown in
FIG. 1A may be configured to communicate with the base station
114a, which may employ a cellular-based radio technology, and with
the base station 114b, which may employ an IEEE 802 radio
technology.
[0055] FIG. 1B is a system diagram of an example WTRU 102. As shown
in FIG. 1B, the WTRU 102 may include a processor 118, a transceiver
120, a transmit/receive element 122, a speaker/microphone 124, a
keypad 126, a display/touchpad 128, non-removable memory 130,
removable memory 132, a power source 134, a global positioning
system (GPS) chipset 136, and other peripherals 138. It will be
appreciated that the WTRU 102 may include any sub-combination of
the foregoing elements while remaining consistent with an
embodiment.
[0056] The processor 118 may be a general purpose processor, a
special purpose processor, a conventional processor, a digital
signal processor (DSP), a plurality of microprocessors, one or more
microprocessors in association with a DSP core, a controller, a
microcontroller, Application Specific Integrated Circuits (ASICs),
Field Programmable Gate Array (FPGAs) circuits, any other type of
integrated circuit (IC), a state machine, and the like. The
processor 118 may perform signal coding, data processing, power
control, input/output processing, and/or any other functionality
that enables the WTRU 102 to operate in a wireless environment. The
processor 118 may be coupled to the transceiver 120, which may be
coupled to the transmit/receive element 122. While FIG. 1B depicts
the processor 118 and the transceiver 120 as separate components,
it will be appreciated that the processor 118 and the transceiver
120 may be integrated together in an electronic package or
chip.
[0057] The transmit/receive element 122 may be configured to
transmit signals to, or receive signals from, a base station (e.g.,
the base station 114a) over the air interface 116. For example, in
one embodiment, the transmit/receive element 122 may be an antenna
configured to transmit and/or receive RF signals. In another
embodiment, the transmit/receive element 122 may be an
emitter/detector configured to transmit and/or receive IR, UV, or
visible light signals, for example. In yet another embodiment, the
transmit/receive element 122 may be configured to transmit and
receive both RF and light signals. It will be appreciated that the
transmit/receive element 122 may be configured to transmit and/or
receive any combination of wireless signals.
[0058] In addition, although the transmit/receive element 122 is
depicted in FIG. 1B as a single element, the WTRU 102 may include
any number of transmit/receive elements 122. More specifically, the
WTRU 102 may employ MIMO technology. Thus, in one embodiment, the
WTRU 102 may include two or more transmit/receive elements 122
(e.g., multiple antennas) for transmitting and receiving wireless
signals over the air interface 116.
[0059] The transceiver 120 may be configured to modulate the
signals that are to be transmitted by the transmit/receive element
122 and to demodulate the signals that are received by the
transmit/receive element 122. As noted above, the WTRU 102 may have
multi-mode capabilities. Thus, the transceiver 120 may include
multiple transceivers for enabling the WTRU 102 to communicate via
multiple RATs, such as UTRA and IEEE 802.11, for example.
[0060] The processor 118 of the WTRU 102 may be coupled to, and may
receive user input data from, the speaker/microphone 124, the
keypad 126, and/or the display/touchpad 128 (e.g., a liquid crystal
display (LCD) display unit or organic light-emitting diode (OLED)
display unit). The processor 118 may also output user data to the
speaker/microphone 124, the keypad 126, and/or the display/touchpad
128. In addition, the processor 118 may access information from,
and store data in, any type of suitable memory, such as the
non-removable memory 130 and/or the removable memory 132. The
non-removable memory 130 may include random-access memory (RAM),
read-only memory (ROM), a hard disk, or any other type of memory
storage device. The removable memory 132 may include a subscriber
identity module (SIM) card, a memory stick, a secure digital (SD)
memory card, and the like. In other embodiments, the processor 118
may access information from, and store data in, memory that is not
physically located on the WTRU 102, such as on a server or a home
computer (not shown).
[0061] The processor 118 may receive power from the power source
134, and may be configured to distribute and/or control the power
to the other components in the WTRU 102. The power source 134 may
be any suitable device for powering the WTRU 102. For example, the
power source 134 may include one or more dry cell batteries (e.g.,
nickel-cadmium (NiCd), nickel-zinc (NiZn), nickel metal hydride
(NiMH), lithium-ion (Li-ion), etc.), solar cells, fuel cells, and
the like.
[0062] The processor 118 may also be coupled to the GPS chipset
136, which may be configured to provide location information (e.g.,
longitude and latitude) regarding the current location of the WTRU
102. In addition to, or in lieu of, the information from the GPS
chipset 136, the WTRU 102 may receive location information over the
air interface 116 from a base station (e.g., base stations 114a,
114b) and/or determine its location based on the timing of the
signals being received from two or more nearby base stations. It
will be appreciated that the WTRU 102 may acquire location
information by way of any suitable location-determination method
while remaining consistent with an embodiment.
[0063] The processor 118 may further be coupled to other
peripherals 138, which may include one or more software and/or
hardware modules that provide additional features, functionality
and/or wired or wireless connectivity. For example, the peripherals
138 may include an accelerometer, an e-compass, a satellite
transceiver, a digital camera (for photographs or video), a
universal serial bus (USB) port, a vibration device, a television
transceiver, a hands free headset, a Bluetooth.RTM. module, a
frequency modulated (FM) radio unit, a digital music player, a
media player, a video game player module, an Internet browser, and
the like.
[0064] FIG. 1C is a system diagram of the RAN 104 and the core
network 106 according to an embodiment. As noted above, the RAN 104
may employ an E-UTRA radio technology to communicate with the WTRUs
102a, 102b, 102c over the air interface 116. The RAN 104 may also
be in communication with the core network 106.
[0065] The RAN 104 may include eNode-Bs 140a, 140b, 140c, though it
will be appreciated that the RAN 104 may include any number of
eNode-Bs while remaining consistent with an embodiment. The
eNode-Bs 140a, 140b, 140c may each include one or more transceivers
for communicating with the WTRUs 102a, 102b, 102c over the air
interface 116. In one embodiment, the eNode-Bs 140a, 140b, 140c may
implement MIMO technology. Thus, the eNode-B 140a, for example, may
use multiple antennas to transmit wireless signals to, and receive
wireless signals from, the WTRU 102a.
[0066] Each of the eNode-Bs 140a, 140b, 140c may be associated with
a particular cell (not shown) and may be configured to handle radio
resource management decisions, handover decisions, scheduling of
users in the uplink and/or downlink, and the like. As shown in FIG.
1C, the eNode-Bs 140a, 140b, 140c may communicate with one another
over an X2 interface.
[0067] The core network 106 shown in FIG. 1C may include a mobility
management entity gateway (MME) 142, a serving gateway 144, and a
packet data network (PDN) gateway 146. While each of the foregoing
elements are depicted as part of the core network 106, it will be
appreciated that any one of these elements may be owned and/or
operated by an entity other than the core network operator.
[0068] The MME 142 may be connected to each of the eNode-Bs 140a,
140b, 140c in the RAN 104 via an Si interface and may serve as a
control node. For example, the MME 142 may be responsible for
authenticating users of the WTRUs 102a, 102b, 102c, bearer
activation/deactivation, selecting a particular serving gateway
during an initial attach of the WTRUs 102a, 102b, 102c, and the
like. The MME 142 may also provide a control plane function for
switching between the RAN 104 and other RANs (not shown) that
employ other radio technologies, such as GSM or WCDMA.
[0069] The serving gateway 144 may be connected to each of the
eNode Bs 140a, 140b, 140c in the RAN 104 via the Si interface. The
serving gateway 144 may generally route and forward user data
packets to/from the WTRUs 102a, 102b, 102c. The serving gateway 144
may also perform other functions, such as anchoring user planes
during inter-eNode B handovers, triggering paging when downlink
data is available for the WTRUs 102a, 102b, 102c, managing and
storing contexts of the WTRUs 102a, 102b, 102c, and the like.
[0070] The serving gateway 144 may also be connected to the PDN
gateway 146, which may provide the WTRUs 102a, 102b, 102c with
access to packet-switched networks, such as the Internet 110, to
facilitate communications between the WTRUs 102a, 102b, 102c and
IP-enabled devices.
[0071] The core network 106 may facilitate communications with
other networks. For example, the core network 106 may provide the
WTRUs 102a, 102b, 102c with access to circuit-switched networks,
such as the PSTN 108, to facilitate communications between the
WTRUs 102a, 102b, 102c and traditional land-line communications
devices. For example, the core network 106 may include, or may
communicate with, an IP gateway (e.g., an IP multimedia subsystem
(IMS) server) that serves as an interface between the core network
106 and the PSTN 108. In addition, the core network 106 may provide
the WTRUs 102a, 102b, 102c with access to the networks 112, which
may include other wired or wireless networks that are owned and/or
operated by other service providers.
[0072] One way to improve the spectral containment of OFDM may be
by filtering the time domain signal at the output of the OFDM
modulator. In a fragmented spectrum where available sub-bands are
not contiguous, filtering becomes challenging since a separate
filter may need to be designed and used for each fragment.
[0073] Another method used to improve the spectral containment of
OFDM is pulse shaping, also known as windowing. It should be noted
that the terms pulse shaping and windowing may be used
interchangeably throughout this description and are meant to have
the same connotation. Pulse shaping is a method used to reduce the
spectral leakage at the transmitter. Pulse shaping may also be used
to reject adjacent channel interference at the receiver. In this
technique, the rectangular pulse shape of the OFDM symbol is
smoothed to prevent sharp transitions between consecutive OFDM
symbols, resulting in lower sidelobes. A mechanism is deployed at
the receiver to reject the adjacent channel interference leakage.
This is because even if the interfering signal in the adjacent band
has low out-of-band emission, the spectral leakage from the
interfering signal increases after cyclic prefix (CP) removal if
the received filter covers the whole accessible band. Therefore,
before CP is removed, the received signal should be filtered for
individual sub-bands.
[0074] Similar to transmitter filtering, receive filtering imposes
challenges in fragmented spectrum. Receive windowing has been used
to reduce the impact of ICI due to carrier frequency offset or
Doppler and to suppress radio frequency interference (RFI) in
discrete multitone (DMT) systems.
[0075] In current OFDM-based communications systems, e.g., LTE, a
CP may be used in OFDM to mitigate ISI due to multipath channel or
timing offset distortion. CP may be prepended at the output of the
inverse fast Fourier transform (IFFT) at the transmitter side and
discarded at the receiver side before the fast Fourier transform
(FFT). The overhead due to the CP may be significant. Therefore,
reducing this overhead while not degrading the system performance
is beneficial. This is because some WTRUs may be in different
locations in a cell and experience different delay spreads, and
some WTRUs may need shorter CP than others. This is true for both
downlink and uplink transmission.
[0076] In addition, CP in OFDM carries useful information since it
is a replica of the time domain samples at the tail of an OFDM
symbol. CP may also be used at single carrier systems and again
consists of the time domain samples at the tail of the symbol. As
noted previously, CP is discarded at the receiver since it is
contaminated by ISI. However, most of the time, the channel delay
spread is smaller than the length of the CP, resulting in some of
the CP samples being free of ISI. These samples may be used at the
receiver to improve performance.
[0077] A transceiver architecture based on transmit and receive
pulse shaping to reduce spectral leakage and reject adjacent
channel interference in multicarrier modulation systems; methods
and apparatus for transmitter side implementation of pulse shaping
on time domain samples of each symbol for OFDM based systems;
methods and apparatus for receiver side implementation of windowing
on time domain samples of each symbol for OFDM based systems; and
methods for applying windowing over a plurality of received OFDM
data block are described herein.
[0078] A general MCM scheme will now be described. For a general
MCM scheme, the input data sequence to be transmitted on the k'th
subcarrier and 'th symbol may be denoted as S.sub.k[], where k
denotes the subcarrier index and denotes the symbol index. Then,
the input data symbols for the k'th subcarrier may be written
as
x.sub.k(t)=S.sub.k[].delta.[t-T'], Equation (1)
where T' is the symbol interval. The data on each subcarrier may be
convolved by a filter p(t) that is modulated to the frequency of
that subcarrier. The aggregate transmitted signal may be written
as
x(t)=.SIGMA..sub.k=0.sup.M-1[{S.sub.k[].delta.[t-T']}*{p(t)e.sup.j2.pi.k-
F.sup.s.sup.t}], Equation (2)
where M is the total number of subcarriers and F.sub.s is the
spacing between the subcarriers.
T = det 1 F s ##EQU00001##
is typically equal to or smaller than T'.
[0079] After expanding the convolution in Equation 2, the following
may be obtained:
x(t)=.SIGMA..sub.k=0.sup.M-1.SIGMA..sub..tau.=-.infin..sup..infin.S.sub.-
k[].delta.(.tau.-T')p(t-.tau.)e.sup.j2.pi.kF.sup.s.sup.(t-.tau.).
Equation (3)
[0080] Since .delta.(.tau.-T')=0, .tau..noteq.T', the following may
be obtained:
x(t)=.SIGMA..sub.k=0.sup.M-1S.sub.k[]p(t--T'). Equation (4)
Equation (4) may be viewed as the general multicarrier modulation
scheme.
[0081] An example of the OFDM structure will now be described. With
respect to OFDM, it may be assumed that the signal in Equation (4)
is sampled at a sampling rate of T.sub.s=T/N and that
T'=.lamda.T.sub.s. Here, the discrete-time of Equation (4) may be
written as:
x [ n ] = det x ( nT s ) = k = 0 M - 1 = - .infin. .infin. S k [ ]
p ( nT s - .lamda. T s ) e j 2 .pi. knT s NT s e - j 2 .pi. k (
.lamda. T s ) NT s = k = 0 M - 1 = - .infin. .infin. S k [ ] p [ n
- .lamda. ] e j 2 .pi. kn N e - j 2 .pi. k .lamda. N , Equation ( 5
) ##EQU00002##
where n denotes the time sample index.
[0082] For OFDM without CP with critical sampling, the parameters
in Equation (5) are as follows:
N = M = .lamda. ##EQU00003## p [ n ] = { 1 , n = 0 , 1 , N - 1 0 ,
otherwise . ##EQU00003.2##
With these parameters, Equation (5) may be written as
x [ n ] = k = 0 N - 1 = - .infin. .infin. S k [ ] p [ n - .lamda. ]
e j 2 .pi. k n N . Equation ( 6 ) ##EQU00004##
Since the consecutive symbols do not overlap, only a single 'th
[0083] OFDM symbol may be considered:
x [ n ] = k = 0 N - 1 S k [ ] e j 2 .pi. k n N , n = N , N + N - 1.
Equation ( 7 ) ##EQU00005##
[0084] From Equation (7), the 'th OFDM symbol may be computed by
taking the inverse fast Fourier transform (IFFT) of the input data
symbols.
[0085] When a CP is appended, the pulse shape may be defined
as:
p [ n ] = { 1 , n = - N G , , 0 , 1 , N - 1 0 , otherwise ,
##EQU00006##
where N.sub.G is the number of samples in the guard interval, e.g.,
cyclic prefix. Again, the consecutive OFDM symbols may not overlap,
so it may be sufficient to consider a single OFDM symbol. The
parameters in Equation (5) may be as follows:
N = M , .lamda. = N + N G , Equation ( 8 ) x [ n ] = k = 0 M - 1 S
k [ ] p [ n - ( N + N G ) ] e j 2 .pi. k ( n - ( N + N G ) ) N , n
= ( N + N G ) - N G , , ( N + N G ) + N - 1. ##EQU00007##
[0086] Defining n=(N+N.sub.G)+m, where m=-N.sub.G, . . . , 0, 1, .
. . N-1, then:
x [ ( N + N G ) + m ] = k = 0 M - 1 S k [ ] p [ m ] e j 2 .pi. k (
m ) N , Equation ( 9 ) m = - N G , , 0 , 1 , N - 1 Since p [ m ] =
1 , and e j 2 .pi. k ( m ) N = e j 2 .pi. k ( m + N ) N , x [ ( N +
N G ) + m ] = k = 0 M - 1 S k [ ] p [ m ] e j 2 .pi. k ( m ) N , m
= - N G , , - 1 and x [ ( N + N G ) + m ] = k = 0 M - 1 S k [ ] p [
m ] e j 2 .pi. k ( m + N ) N , m = N - N G , , N - 1
##EQU00008##
are equal. Therefore, Equation (8) may be implemented by taking the
IFFT of the input data symbols, and padding the last N.sub.G
samples of the IFFT output to the front of the IFFT output.
[0087] As discussed above, pulse shaping, also known as windowing,
may be used at the transmitter side and receiver side to improve
the spectral containment of OFDM.
[0088] Methods and apparatus for transmitter side windowing will
now be described. As an example, the pulse shaping function for
windowing in Equation (5) may be defined as
Equation ( 10 ) ##EQU00009## p [ n ] = { 0.5 ( 1 + cos { .pi. ( 1 +
n .beta. N T ) } 0 .ltoreq. n < .beta. N T 1 .beta. N T .ltoreq.
n < ( .beta. + 1 ) N T 0.5 ( 1 + cos { .pi. n - ( .beta. + 1 ) N
T .beta. N T } ( .beta. + 1 ) N T .ltoreq. n .ltoreq. ( 2 .beta. +
1 ) N T - 1 } ##EQU00009.2##
where N.sub.T=N+N.sub.G, and .lamda.=(1+.beta.)N.sub.T. Other pulse
shaping functions are also possible. Pulse shaping functions should
generally create a smooth transition at the boundary of two
consecutive symbols. In this case, the new guard interval is
generally larger than the cyclic prefix. N'.sub.G=N.sub.G+N.sub.EGI
may be defined, where N.sub.EGI is the extended guard interval.
However, from a signal processing point of view, is nothing but a
longer cyclic prefix. It should be noted that windowing functions
other than the one in Equation (10) are also possible, but the
following approach will be similar.
[0089] Defining n=i(1+.beta.)N.sub.T+m, where m=0, . . . ,
(1+.beta.)N.sub.T-1, Equation (4) may be written as
x [ i ( 1 + .beta. ) N T + m ] = k = 0 M - 1 = - .infin. .infin. S
k [ ] p [ i ( 1 + .beta. ) N T + m - ( 1 + .beta. ) N T ] .times. e
j 2 .pi. kF 0 ( i ( .beta. + 1 ) N T + m ) e - j 2 .pi. kF 0 ( ( 1
+ .beta. ) N T ) , Equation ( 11 ) ##EQU00010##
which may be written as
x.sub.i'[m]=E.sub.k=0.sup.M-1S.sub.k[]p[(i-)(1-.beta.)N.sub.T+m]e.sup.j2-
.pi.kF.sup.0.sup.(i(.beta.+1)N.sup.T.sup.+m)e.sup.-j2.pi.kF.sup.0.sup.(.su-
p.(1+.beta.)N.sup.T.sup.), Equation (12)
defining x'[m]=x[i(1+.beta.)N.sub.T+m].
[0090] Since for the i'th block, only two symbols overlap due to
the pulse shape design where the pulse shape should not be much
longer than N, the terms corresponding to =i and =i-1 in the
summation over are retained. Then,
x i ' [ m ] = k = 0 M - 1 S k [ i - 1 ] p [ ( 1 + .beta. ) N T + m
] e j 2 .pi. k ( i ( .beta. + 1 ) N T + m ) N e - j 2 .pi. k ( ( i
- 1 ) ( 1 + .beta. ) N T ) N + k = 0 M - 1 S k [ i ] p [ m ] e j 2
.pi. k ( i ( .beta. + 1 ) N T + m ) N e - j 2 .pi. k ( i ( 1 +
.beta. ) N T ) N , Equation ( 13 ) ##EQU00011##
which is equal to
x i ' [ m ] = k = 0 M - 1 S k [ i - 1 ] p [ ( 1 + .beta. ) N T + m
] e j 2 .pi. k ( ( 1 + .beta. ) N T + m ) N + k = 0 M - 1 S k [ i ]
p [ m ] e j 2 .pi. k m N . Equation ( 14 ) ##EQU00012##
[0091] In Equation (14), it can be seen that for the first term is
non-zero for m=0, 1, . . . , .beta.N.sub.T has non-zero values. The
limits where the pulse shape is defined may be seen in Equation
(10). Therefore, the implementation, as shown in FIG. 2, may be
used.
[0092] FIG. 2 is a flow chart of an example transmit windowing
implementation procedure using CP 200. The example transmit
implementation procedure 200 is a direct implementation from
Equation (14). Referring to FIG. 2, the OFDM symbol index is set to
m=0 in step 205. Next, the transmitter takes the IFFT of a block of
frequency domain QAM symbols in step 210. For example, if the IFFT
size is N=8, the samples at the output of the IFFT for the m-th
OFDM symbol are a.sup.(m)=[a.sub.1.sup.(m), a.sub.2.sup.(m),
a.sub.3.sup.(m), a.sub.4.sup.(m), a.sub.5.sup.(m), a.sub.6.sup.(m),
a.sub.7.sup.(m), a.sub.8.sup.(m)]. In step 215, a cyclic prefix
(CP) and a postfix are attached to the output samples to generate a
composite signal. Assuming the CP length is 4, and the postfix
length is 2, after the CP and postfix is attached, the composite
signal becomes s.sup.(m)=[a.sub.5.sup.(m), a.sub.6.sup.(m),
a.sub.7.sup.(m), a.sub.8.sup.(m), a.sub.1.sup.(m), a.sub.2.sup.(m),
a.sub.3.sup.(m), a.sub.4.sup.(m), a.sub.5.sup.(m), a.sub.6.sup.(m),
a.sub.7.sup.(m), a.sub.8.sup.(m), a.sub.1.sup.(m),
a.sub.2.sup.(m)]. In step 220, a windowing function is applied to
the composite signal to generate a windowed composite signal. It
should be noted that the windowing function may be applied in
different ways. For example, if ".beta.N.sub.T"=2, that is,
windowing takes two samples on each side to ramp up and down, then
s.sub.w.sup.(m)=[p.sub.1a.sub.5.sup.(m), p.sub.2a.sub.6.sup.(m),
a.sub.7.sup.(m), a.sub.8.sup.(m), a.sub.1.sup.(m), a.sub.2.sup.(m),
a.sub.3.sup.(m), a.sub.4.sup.(m), a.sub.5.sup.(m), a.sub.6.sup.(m),
a.sub.7.sup.(m), a.sub.8.sup.(m), p.sub.13a.sub.1.sup.(m),
p.sub.14a.sub.2.sup.(m)], where 0=p.sub.1<p.sub.2<1, and
0=p.sub.14<p.sub.13<1. In step 225, the postfix portion of
the previous (m-1).sup.th windowed composite signal is added to the
leftmost part of the CP portion of the current m.sup.th windowed
composite signal. In step 230, the signals generated from step 225
are transmitted, excluding the postfix portion, which will be used
in the next iteration. The transmitted samples for the m.sup.th
symbol is
s.sub.w-TX.sup.(m)=[p.sub.2a.sub.1.sup.(m-1)+p.sub.1a.sub.5.sup.(m),p.sub-
.1a.sub.2.sup.(m-1)+p.sub.2a.sub.6.sup.(m)m
a.sub.7.sup.(m),a.sub.8.sup.(m),a.sub.1.sup.(m),a.sub.2.sup.(m),a.sub.3.s-
up.(m),a.sub.4.sup.(m),a.sub.5.sup.(m),a.sub.6.sup.(m),a.sub.7.sup.(m),a.s-
ub.8.sup.(m)] In step 235, the OFDM symbol index is increased by 1,
m+1=>m, to apply the process to the next consecutive symbol. If
m+1 is not the last symbol, steps 210-240 are repeated for the next
consecutive symbol. If m+1 is the last OFDM symbol, then the
process is complete in step 245.
[0093] FIG. 3 is a block diagram of an example transmitter module
configured to implement transmit windowing with cyclic prefix. The
implementation shown in FIG. 3 is a practical implementation of
Equation (14).
[0094] Referring to FIG. 3, the ' th symbol 305 from an input data
stream that has been spread across a plurality of multiple parallel
sub-carriers 310 of a frequency band is input into an IFFT unit
315. The IFFT unit may be an N-point IFFT unit. The IFFT unit 315
converts the signals in the plurality of sub-carriers 310 from the
frequency domain to corresponding time domain signals 320a-n. The
CP and the postfix are appended to generate a composite signal and
then, the composite signal is multiplied by a windowing function
p[m], m=0, 1, . . . , (2.beta.+1)N.sub.T-1, at the respective
windowing modules 325a-n. The last .beta.N.sub.T samples,
corresponding to the postfix, of the output of the windowing
modules 325n-1, 325n may be kept in buffers 330. The last
.beta.N.sub.T samples, corresponding to the postfix, in the buffers
330 are then added to the first .beta.N.sub.T samples corresponding
to the prefix portion of the output of the next symbol by windowing
modules 325a, 325b. Note that the samples may be added to the
buffer before the contents of the buffer are updated. The
transmitter may then transmit the first (1+.beta.)N.sub.T samples.
A parallel-to-serial converter (P/S) 335 receives the samples and
converts them into an OFDM signal 340 for transmission.
[0095] FIG. 4 is a diagram of an example application of transmit
windowing in accordance with the method described in FIG. 2. It
should be noted that for purposes of illustration and for the
reader's convenience, only two data blocks are shown. It should be
noted that there may be any number of data blocks before and after
the data blocks shown in FIG. 4. As described in FIG. 2, the method
may continue until the last data block in the signal is processed.
Referring to FIG. 4, a first data block 410 and a second data block
420 are shown. The first data block 410 has a CP 412 prepended to
the beginning of the data block 410 and a postfix 414 appended to
the end of the data block 410. The leftmost part 411 of the CP 412
and the postfix 414 of the first data block 410 have been windowed.
Similarly, the second data block 420 has a CP 422 prepended to the
beginning of the data block 420 and a postfix 424 appended to the
end of the data block 420. The leftmost part 426 of the CP 422 and
the postfix 424 of the second data block 420 have been windowed. It
is important to note that windowing is applied independently on
each of the data blocks 410, 420 as in the method described in FIG.
2. The postfix 414 of the first data block 410 overlaps in time
with the windowed part 426 of the CP 422 of the second data block
420. The postfix 414 and the windowed part 426 of the CP 422 are
effectively summed 430.
[0096] A transmitter side windowing method and apparatus using
zero-padding (ZP) OFDM instead of CP will now be described. FIG. 5
is a flow chart of an example transmitter windowing procedure with
zero-padding. Referring to FIG. 5, the OFDM symbol index is set to
m=0 in step 505, Next, the transmitter takes the IFFT of a block of
frequency domain QAM symbols. For example, if the IFFT size is N=8,
then the samples at the output of IFFT for the m.sup.th OFDM symbol
is a.sup.(m)=[a.sub.1.sup.(m), a.sub.2.sup.(m), a.sub.3.sup.(m),
a.sub.4.sup.(m), a.sub.5.sup.(m), a.sub.6.sup.(m), a.sub.7.sup.(m),
a.sub.8.sup.(m)]. In step 515, a zero prefix (ZP) and postfix are
attached to generate the composite signal. For example, if the zero
padding length is 4 and the postfix length is 2, then after ZP and
the postfix are appended, the composite signal becomes
s.sup.(m)=[0, 0, 0, 0, a.sub.1.sup.(m), a.sub.2.sup.(m),
a.sub.3.sup.(m), a.sub.4.sup.(m), a.sub.5.sup.(m), a.sub.6.sup.(m),
a.sub.7.sup.(m), a.sub.8.sup.(m), a.sub.1.sup.(m), a.sub.2.sup.(m)]
In step 520, the windowing function is applied to the composite
signal to generate a windowed composite signal. In the above
example, since the first 4 samples are zero, the first 4 samples of
the windowing function are also zero. For example, if
".beta.N.sub.T"=2, that is, windowing takes 2 samples on each side
to ramp up and down, then s.sub.w.sup.(m)=[0, 0, 0, 0, p.sub.11
a.sub.1.sup.(m), p.sub.12a.sub.2.sup.(m), a.sub.3.sup.(m),
a.sub.4.sup.(m), a.sub.5.sup.(m), a.sub.6.sup.(m), a.sub.7.sup.(m),
a.sub.8.sup.(m), p.sub.13a.sub.1.sup.(m), p.sub.14a.sub.2.sup.(m)]
where 0=p.sub.1<p.sub.2<1, and 0=p.sub.14<P.sub.13<1.
In step 525, the postfix portion of the previous (m-1)th windowed
composite signal is added to the leftmost part of the ZP portion of
the current m.sup.th windowed composite signal. In step 530, the
signals generated from step 525 are transmitted, excluding the
postfix portion, which will be used in the next iteration. The
transmitted samples for the m.sup.th symbol is
s.sub.w-TX.sup.(m)=[p.sub.2a.sub.1.sup.(m-1),
p.sub.1a.sub.2.sup.(m-1), 0, 0, a.sub.1.sup.(m), a.sub.2.sup.(m),
a.sub.3.sup.(m), a.sub.4.sup.(m), a.sub.5.sup.(m), a.sub.6.sup.(m),
a.sub.7.sup.(m), a.sub.8.sup.(m)]. In step 535, the OFDM symbol
index is increased by 1, m+1=>m, to apply the process to the
next consecutive symbol. If m+1 is not the last symbol, steps
510-540 are repeated for the next consecutive symbol. If m+1 is the
last OFDM symbol, then the method is complete in step 545.
[0097] FIG. 6 is a diagram of an example application of transmit
windowing with zero-padding in accordance with the method described
in FIG. 5. It should be noted that for purposes of illustration and
for the reader's convenience, only two data blocks are shown. It
should be noted that there may be any number of data blocks before
and after the data blocks shown in FIG. 6. As described in FIG. 5,
the method may continue until the last data block in the signal is
processed. Referring to FIG. 6, a first data block 610 and a second
data block 620 are shown. The first data block 610 has a ZP 612
consisting of zeros prepended to the beginning of the data block
610 and a postfix 614 appended to the end of the data block 610.
The leftmost part of the data block 610 and the postfix 614 have
been windowed. Similarly, the second data block 620 has a ZP 622
consisting of zeros prepended to the beginning of the data block
620 and a postfix 624 appended to the end of the data block 620.
The leftmost part of the data block 620 and the postfix 624 have
been windowed. It is important to note that windowing is applied
independently on each of the data blocks 610, 620. The overlapping
segment of the windowed postfix 614 of the first data block 610 and
the foremost portion 626 of the ZP 622 of the second data block 620
are effectively summed 630.
[0098] FIG. 7 is an example diagram of receiver side procedures for
windowing with zero-padding. At the receiver, the signal is
restored by overlapping and adding the postfix 715 to the first in
samples of the received data block 710. Referring to FIG. 7, a
block of data 710 with a prepended ZP 705 and an appended postfix
715 is received at a receiver. The ZP 705 is discarded, leaving the
data block 710 and the postfix 715. The postfix 715 is removed from
the tail of the data block 710 and added to the head of the data
block 710. The output is then sent to the FFT block 730 which
converts the data block in in the sub-carriers from the time domain
back to the frequency domain.
[0099] A method and apparatus for applying different window
functions to separate groups of a data signal will now be
described. Different windowing functions may be applied to
different sub-bands of the transmission band. As an example, the
sub-bands next to the edges may be shaped with longer windows to
get better spectral containment of those sub-bands. On the other
hand, the sub-bands away from the edges, or in the middle of the
band, may be shaped with shorter windows. Since windowing may
introduce distortion, the possibly larger distortion introduced by
the longer windows will be limited to the sub-bands on the
edges.
[0100] FIG. 8 is a diagram of an example method of applying
different windowing functions to different sub-bands of the
transmission band. In FIG. 8, the subcarriers in each sub-band are
shown to be contiguous. This is for illustration purposes only, and
in general, a sub-band may consist of contiguous or non-contiguous
group(s) of subcarriers. The sub-bands are non-overlapping, i.e.,
subcarriers in a sub-band are different than the sub-carriers in
the other sub-bands.
[0101] Referring to FIG. 8, incoming modulated symbols s[n] 805 are
input into a serial-to-parallel converter (S/P) 810 which outputs
in groups of modulated output signals, 815a-m to be divided among a
plurality of subcarriers. For purpose of explanation, it is assumed
that in different windowing functions will be applied to in
different sub-bands. For each sub-band, the modulated output
signals 815a-m are mapped to the plurality of subcarriers in the
IFFT blocks 820a-m corresponding to the subcarriers in those
sub-bands. Note that FIG. 8 shows multiple IFFT blocks. This is to
show that conceptually, in IFFTs are taken. However, in a hardware
implementation there may be one IFFT block that may be used in
times. This is also the case for the other units shown in the
diagram. Let .sub.i denote the set of indices of subcarriers in the
i'th sub-band where each element of N.sub.i {0, 1, . . . , N-1}.
Let the input to the IFFT be a N.times.1 vector of zeros. The group
of modulated symbols that are to be transmitted on subcarriers
.sub.i are inserted into the elements of the vector where the
indices of the elements are .sub.i. For each sub-band, the IFFT
output 825a-m is padded with a prefix and a postfix at the padding
attacher module 830a-m. The output 835a-m of each padding attacher
module 830a-m is point-wise multiplied with the appropriate
windowing function at the respective window filters 840a-m. As
shown in FIG. 8, a different windowing function may be used in each
branch (i.e., window type 1, window type k, window type, etc.).
However, the length of the signal in each branch should be equal.
Therefore, the size of the prefix and postfix in each branch is the
same. After the windowing function is applied, the output of each
windowed signal 845a-m is input into a parallel-to-serial converter
(P/S) 850a-m. The output of each branch 855a-m is then added
together to create the composite signal that will be
transmitted.
[0102] Methods and apparatus for receiver side windowing will now
be described. A mechanism may be used at the receiver to reject the
adjacent channel interference leakage. Such a mechanism is used
because even if the interfering signal in the adjacent band has low
out-of-band emission, the spectral leakage from the interfering
signal increases after the CP removal. Therefore, before the CP is
removed, the received signal may be filtered. OFDM achieves this by
using the rectangular windowing, which corresponds to a sinc-type
filter with high tails, and is therefore not satisfactory for
interference rejection capability.
[0103] Similar to the transmitter side filtering, receive filtering
imposes challenges in a fragmented spectrum. An alternative method
is to use windowing at the receiver. In general, if the transmitter
attaches a prefix and postfix as illustrated in FIG. 3, receive
windowing may be applied to the prefix, data, and postfix
samples.
[0104] One way the receive window may be defined is as follows:
p [ n ] = { 0.5 ( 1 + cos { .pi. ( 1 + n .beta. N T } .alpha.
.ltoreq. n < .beta. N T 1 .beta. N T .ltoreq. n < N T -
.beta. N T 0.5 ( 1 + cos { .pi. n - ( .beta. + 1 ) N T .beta. N T }
N T - .beta. N T .ltoreq. n .ltoreq. N T } Equation ( 15 )
##EQU00013##
[0105] In general, the receive window may be defined beyond
N.sub.T. However, this may call for the next symbol to be used,
causing a small delay. The following approach will hold
regardless.
[0106] FIG. 9 is an example diagram of the boundaries of receive
windowing. Referring to FIG. 9, N.sub.pre 910 is the length of the
prefix. N.sub.post 920 is the length of the postfix. Note that the
boundaries shown for the window in FIG. 9 are just one example, and
it should be noted that the window non-zero window coefficients
boundaries may extend from -N.sub.pre to N.sub.post.
[0107] In the case of no transmit windowing, i.e., all 1's, and no
overlapping between consecutive symbols, the transmitted symbol may
be written as
x [ n ] = k = 0 M - 1 = - .infin. .infin. S k [ ] p [ n - .lamda. ]
e j 2 .pi. k ( n - .lamda. ) N , Equation ( 16 ) ##EQU00014##
where .lamda.=N+N.sub.pre+N.sub.post. For n=l.lamda.-N.sub.pre, . .
. , l.lamda.+(N+N.sub.post-1),
x [ n ] = k = 0 M - 1 S k [ ] e j 2 .pi. k ( n - .lamda. ) N
##EQU00015##
where the N.sub.pref may include the guard interval for the CP as
well. Let n=l.lamda.+m for l=-.infin., . . . , 0, . . . , .infin.,
m=-N.sub.pre, . . . , N+N.sub.post. Then,
x l [ m ] = x [ l .lamda. + m ] = k = 0 M - 1 S k [ ] e j 2 .pi. k
( l .lamda. + m - .lamda. ) N = k = 0 M - 1 S k [ ] e j 2 .pi. k (
m ) N . Equation ( 17 ) ##EQU00016##
[0108] The receiver windowing coefficients may be defined as:
{w[m], m=-N.sub.pre, . . . , N+N.sub.post}. Applying windowing and
converting the received signal back to the frequency domain by FFT,
i.e., frequency demodulation at frequency k/N,
m = - N pre N + N post - 1 x l [ m ] w [ m ] e j 2 .pi. mk N = m =
- N pre - 1 x l [ m ] w [ m ] e - j 2 .pi. mk N + k = 0 N - 1 x l [
m ] w [ m ] e - j 2 .pi. mk N + m = N N + N post - 1 x l [ m ] w [
m ] e - j 2 .pi. mk N Equation ( 18 ) ##EQU00017##
[0109] If m' is defined as follows, m'=N+m for m=N.sub.pre, . . . ,
-1, then, the terms in Equation 18 may be written as follows:
The first term = m ' = N - N pre N - 1 x l [ m ' - N ] w [ m ' - N
] e - j 2 .pi. m ' k N Equation ( 19 ) ##EQU00018##
[0110] Expanding w[m] such that w[m]=0 for m=-N, . . . ,
-N.sub.prep-1. Then the first term becomes:
The first term = m ' = 0 N - 1 x l [ m ' - N ] w [ m ' - N ] e - j
2 .pi. mk N Equation ( 20 ) Define m '' = m - N for m = N , , N + N
post The first term = m '' = 0 N post - 1 x l [ m '' + N ] w [ m ''
+ N ] e - j 2 .pi. m '' k N Equation ( 21 ) ##EQU00019##
[0111] Expanding w[m] such that w[m]=0 for m=N+N.sub.post, . . . ,
2N. Then,
The last term = m '' = 0 N - 1 x l [ m '' + N ] w [ m '' + N ] e -
j 2 .pi. m '' k N Equation ( 22 ) ##EQU00020##
[0112] Combining the terms, gives
m = - N pre N + N post - 1 x l [ m ] w [ m ] e - j 2 .pi. mk N = m
= 0 N - 1 i = - 1 1 x l [ m + iN ] w [ m + iN ] e - j 2 .pi. mk N .
Equation ( 23 ) ##EQU00021##
[0113] To recover the transmitted symbols, make
x.sup.l[m]=x.sup.l[m-N] for m=N-N.sub.pre, . . . , N-1 and
x.sup.l[m]=x.sup.l[m+N] for m=0, . . . , N.sub.post-1.
[0114] The transmitted signal may be considered as
x [ n ] = k = 0 N - 1 l = - .infin. .infin. S k [ l ] p [ n - l
.lamda. ] e j 2 .pi. k ( n - l .lamda. ) N Equation ( 24 )
##EQU00022##
where .lamda.=N+N.sub.post+N.sub.pre. Assuming that p[n].noteq.0
for n=-N.sub.pre, . . . , N+N.sub.post, define m=n-l.lamda.. Then,
taking samples from x[n]:
x l [ m ] = x [ m + l .lamda. ] = k = 0 N - 1 S k [ l ] p [ m ] e j
2 .pi. k ( m ) N = p [ m ] k = 0 N - 1 S k [ l ] e j 2 .pi. k ( m )
N = p [ m ] y [ m ] Equation ( 25 ) ##EQU00023##
[0115] For m=-N.sub.pre, . . . , N+N.sub.post. Note that
y[m]=y[m-N] for m=N-N.sub.pre, . . . , N-1 and y[m]=y[m+N] for m=0,
. . . , N.sub.post-1. At the receiver side, the transmitter window
p[m] should be chosen such that x.sup.l[m]=p[m]y[m] also satisfies
this condition.
[0116] Applying the transmitted signal at the receiver side,
m = 0 N - 1 i = - 1 1 x l [ m + iN ] w [ m + iN ] e - j 2 .pi. mk '
N = m = 0 N - 1 i = - 1 1 ( p [ m + iN ] k = 0 N - 1 S k [ l ] e j
2 .pi. k ( m + iN ) N ) w [ m + iN ] e - j 2 .pi. mk ' N = m = 0 N
- 1 ( i = - 1 1 p [ m + iN ] w [ m + iN ] ) k = 0 N - 1 S k [ l ] e
j 2 .pi. m ( k - k ' ) N Equation ( 26 ) If i = - 1 1 p [ m + iN ]
w [ m + iN ] = 1 , Equation ( 27 ) ##EQU00024##
or a constant, for m=0, . . . , N-1, the above expression is
S.sub.k'[l].
[0117] FIG. 10 is a block diagram of a receiver module 1000
configured to implement an example type of receive windowing.
Referring to FIG. 10, the receiver may receive an OFDM signal y[n]
1005. A serial-to-parallel converter (S/P) 1010 samples the
received signal y[n] 1005, which may include a data block spread
across a plurality of sub-carriers of a frequency band, and divides
it into a plurality of sub-carriers. Receive windowing may be
applied at window filters 1015a-n. The prefix samples 1020a-n and
the postfix samples 1025a-n may be added to the data block 1030a-n
and then removed. It should be noted that receive windowing may be
applied to the prefix, datablock, and postfix samples, before the
prefix and postfix is removed. The datablock 1030a-n is input into
an FFT unit 1040, which may be an N-point FFT unit, that converts
the datablock in the sub-carriers from the time domain back to the
frequency domain.
[0118] As an alternative method, the received signal on subcarrier
{circumflex over (k)} after windowing and taking the FFT may be
written as
S [ k ^ ] = k = 0 N - 1 n = - N pre N + N post - 1 S k [ ] p [ n ]
w [ n ] e j 2 .pi. k 1 N n e - j 2 .pi. k ^ 1 N n Equation ( 28 )
##EQU00025##
where the receive window is denoted as w[n].
[0119] Defining n=iN+m, and ignoring p[n] because it is all 1's,
Equation 29 is obtained using only i=-1, 0, 1 because of the extent
of the window.
S [ k ^ ] = k = 0 N - 1 m = 0 N - 1 i = - 1 1 S k [ ] w [ iN + m ]
e j 2 .pi. ( k - k ^ ) 1 N ( iN + m ) Equation ( 29 )
##EQU00026##
[0120] From Equation 29, it can be seen that, to ensure
orthogonality, .SIGMA..sub.i=-1.sup.1w[iN+m]=constant. Under this
condition, orthogonality is preserved, and
[{circumflex over
(k)}]=.SIGMA..sub.k=0.sup.N-1.SIGMA..sub.m=0.sup.N-1s.sub.k[]e.sup.j2.pi.-
(k-{circumflex over (k)})m Equation (30)
[0121] Note that, similar to adding the CP in Equation 8, the
extension of the OFDM signal over duration N is equal to taking the
IFFT and adding the first samples as postfix.
[0122] From Equation (29),
S [ k ^ ] = i = - 1 1 m = 0 N - 1 w [ iN + m ] k = 0 N - 1 S k [ ]
e j 2 .pi. ( k - k ^ ) 1 N m Equation ( 31 ) ##EQU00027##
[0123] The above window is the most general case. In practical
systems, by way of example, the postfix of one symbol serves as the
prefix of the next symbol. Therefore, applying windowing to the
postfix introduces additional ISI from the following symbol and
increases latency. FIGS. 11 and 12 are diagrams of an example
windowing interval in a practical system. Referring to FIG. 11, the
windowing interval 1105 covers the samples of the current symbol
1115, including the data part 1112 and the cyclic prefix part 1110.
Additionally, the window interval 1105 may cover samples of the
extended guard interval parts 1120, if it exists. Similarly, this
may be seen in FIG. 12. Note that, from a system perspective, an
extended guard interval may be assumed as part of the cyclic
prefix.
[0124] For this case,
S [ k ^ ] = k = 0 N - 1 m = 0 N - 1 i = - 1 0 S k [ ] w [ iN + m ]
e j 2 .pi. ( k - k ^ ) 1 N ( iN + m ) . Equation ( 32 )
##EQU00028##
[0125] For orthogonality, .SIGMA..sub.i=-1.sup.0
w[iN+m]=constant.
[0126] With transmit windowing, it can be shown that to maintain
orthogonality,
.SIGMA..sub.i=-1.sup.0w[iN+m]p[m+iN]=constant. Equation (33)
[0127] Receive windowing may also be implemented by point-wise
multiplying the input received block by the receiver windowing
coefficients. FIG. 13 is a diagram of a receiver module 1300
configured to implement an example type of receive windowing by
point-wise multiplying. Referring to FIG. 13, the receiver may
receive an OFDM signal y[n] 1305. A serial-to-parallel converter
(S/P) 1310 samples the received signal y[n] 1305, which may include
a data block spread across a plurality of sub-carriers of a
frequency band, and divides it into a plurality of subcarriers.
Receive windowing is applied by point-wise multiplying the data
block by the receiver windowing coefficients at window filters
1315a-n. The first samples 1320a-n are added to the corresponding
last samples 1325a-n. The first samples 1320a-n are then removed.
The first samples 1320a-n may also be referred to as guard
interval. Then, the data block 1330a-n is input into an FFT unit
1340, which may be an N-point FFT unit that converts the datablock
in the sub-carriers from the time domain back to the frequency
domain.
[0128] Equations (27) and (33) specify the conditions to maintain
orthogonality at the receiver when both transmit and receive
windowing are applied. Given a transmit window function, the
receiver window function may be computed from these equations.
[0129] Intersymbol interference due to receive windowing will now
be described. The following analysis assumes that the second type
of receive windowing, as described in FIG. 11 and Equations
(28)-(33), is used since it is more appropriate from a system
design point of view.
[0130] Assume that the transmitted signal goes through a channel
denoted as h[n]. The received signal, after windowing, may be
written as
[{circumflex over
(k)}]=.SIGMA..sub.i=-1.sup.0.SIGMA..sub.m=0.sup.N-1w[iN+m]r[iN+m]e.sup.-j-
2.pi.{circumflex over (k)}F.sup.0.sup.m Equation (34)
where r[n]=x[n]*h[n]=.SIGMA..sub.u=0.sup.L-1x(n-u)h(u) Then,
[{circumflex over
(k)}]=.SIGMA..sub.i=-1.sup.0.SIGMA..sub.m=0.sup.N-1w[iN+m].SIGMA..sub.u=0-
.sup.L-1x(iN+m-u)h(u)e.sup.-j2.pi.{circumflex over
(k)}F.sup.0.sup.m. Equation (35)
[0131] Note that, the prefix part of the data block may contain
interference from the previous data block. This interference is
multiplied by the window function and added to the desired signal
as shown in Equation (23). The level of interference may depend on
the channel delay spread and the length of the window function. If
the zero-part of the window function is long enough to absorb the
ISI, interference may not occur since the prefix is discarded.
However, if the delay spread is long enough, then interference may
occur. Assuming that the channel delay spread L<N, then ISI is
due to only the previous data block. In this case, interference
contributes from i=-1, due to data transmitted in i=-2. Then, the
interference on the k'th subcarrier may be written as
[{circumflex over
(k)}]=.SIGMA..sub.m=0.sup.N-1w[iN+m].SIGMA..sub.u=0.sup.L-1x(iN+m-u)v(iN+-
m-u)h(u)e.sup.-j2.pi.{circumflex over (k)}F.sup.0m,i=-1, Equation
(36)
where v(iN+m-u)=1 for iN+m-u<-N, else 0.
S [ k ^ ] = m = 0 N - 1 u = 0 L - 1 w [ iN + m ] x ( iN + m - u ) v
( iN + m - u ) h ( u ) e - j 2 .pi. kF 0 m , i = - 1 Equation ( 37
) ##EQU00029##
[0132] From here, the interference power may be computed. Matrix
notation may be used to see the ISI more clearly. The transmitted
signal may be written as
x l = [ 0 .beta. .times. ( N - .beta. ) I .beta. I N ] F H S l ,
Equation ( 38 ) ##EQU00030##
[ 0 .beta. .times. ( N - .beta. ) I .beta. I N ] ##EQU00031##
[0133] where F.sup.H is the IFFT matrix and appends a prefix of
.beta. samples. The received signal after the channel is
y l = H signal [ 0 .beta. .times. ( N - .beta. ) I .beta. I N ] F H
S l + H isi [ 0 .beta. .times. ( N - .beta. ) I .beta. I N ] F H S
l - 1 + n quation ( 39 ) ##EQU00032##
where the first part of Equation (27) is the desired signal, the
second part of Equation (27) is the ISI from the previous block,
and the third part of Equation (27) is the noise. The channel
matrices in Equation (27) may be written as
H signal = [ h 0 0 0 h L - 1 h 0 0 0 h L - 1 h 0 0 0 0 h L - 1 h 0
] , H isi = [ 0 0 h L - 1 h 1 0 h L - 1 0 0 0 ] . Equation ( 40 )
##EQU00033##
[0134] After applying the windowing to the received signal,
r = [ I .beta. .times. .beta. 0 .beta. .times. N 0 .beta. .times.
.beta. 0 N .times. .beta. I N .times. N 0 N .times. .beta. I .beta.
.times. .beta. 0 .beta. .times. N I .beta. .times. .beta. ] Wy l ,
Equation ( 41 ) ##EQU00034##
where W performs the windowing operation and is defined as
follows:
W = diag [ w - .beta. 11111 w N - .beta. w N - 1 ] ##EQU00035## w -
m + w N - m = 1 and [ I .beta. .times. .beta. 0 .beta. .times. N 0
.beta. .times. .beta. 0 N .times. .beta. I N .times. N 0 N .times.
.beta. I .beta. .times. .beta. 0 .beta. .times. N I .beta. .times.
.beta. ] ##EQU00035.2##
takes the first .beta. values of the processed data block and adds
to the last .beta. values. These two matrices together perform the
receive windowing. Equation 29 may be rewritten as
r = [ I .beta. .times. .beta. 0 .beta. .times. N 0 .beta. .times.
.beta. 0 N .times. .beta. I N .times. N 0 N .times. .beta. I .beta.
.times. .beta. 0 .beta. .times. N I .beta. .times. .beta. ] W { H
signal [ 0 .beta. .times. ( N - .beta. ) I .beta. I N ] F H S l + H
isi [ 0 .beta. .times. ( N - .beta. ) I .beta. I N ] F H S l - 1 +
n } . Equation ( 42 ) ##EQU00036##
[0135] After the removal of the guard band, the signal part
preserves the orthogonality. However, ISI from the previous block
is introduced. The amount of the ISI depends on L and .beta.. The
effect of ISI may be more clearly seen in the following:
g = [ 0 0 h L - 1 h 1 0 h L - 1 0 0 0 ] [ 0 .beta. .times. ( N -
.beta. ) I .beta. I N ] F H S l - 1 . Equation ( 43 )
##EQU00037##
[0136] The ISI introduced will be Fy.sub.isi, where
y.sub.isi=[0 0 0 0 . . . w.sub.-.beta.g(0) . . .
w.sub.-1g(.beta.-1)].sup.T. Equation (44)
[0137] The samples multiplied by the window's 0-coefficients will
not contribute to the ISI. If w.sub.-.beta.+.alpha. is the first
non-zero sample of the window, then ISI contribution will be zero
if .alpha..gtoreq.L.
[0138] Methods to improve performance of receive windowing, such as
interference cancellation will now be described. Successive
interference cancellation will now be described. The received
signal may contain interference from the previous transmitted block
due to the multi-path channel. Equation (44) characterizes the
interference in terms of the channel, the previous block, and the
receive windowing coefficients. After windowing, depending on the
CP length and the windowing coefficients, some of this interference
is added to the time samples of the current symbol. One method to
improve the performance is to regenerate and cancel this
interference. Assuming that the channel is known and the previous
symbol has been demodulated, the interference may be regenerated
and subtracted from the current symbol. The subtraction may be done
either in the time domain or the frequency domain.
[0139] If the transmitted signal was also windowed, then even
without multi-path channel, the receive windowing introduces
ISI.
[0140] FIGS. 14, 15 and 16 show various cases of transmit
windowing. Referring to FIG. 14, a first block of data 1400 and a
second block of data 1450 are shown. Extended guard intervals, in
the form of a prefixes 1405, 1455, and postfixes 1415, 1465 are
appended to each respective block of data 1400, 1450. The prefixes
1405, 1455 are appended to the CP 1410, 1460 of each respective
block of data 1400, 1450. The postfixes 1415, 1465 are appended to
the tail of each respective block of data 1400, 1450. The extended
guard intervals are appended in order to perform transmit
windowing. At the receiver side, after receive windowing, the CP
and extended guard intervals are discarded. The receive window may
be applied on the CPs and the extended guard intervals. In this
case, the interference depends on the channel length and the
receive window coefficients as shown in the above analysis. The
difference would be that the interfering samples from the previous
block are now weighted by the transmit window coefficients.
[0141] Referring to FIG. 15, a first block of data 1500 and a
second block of data 1550 are shown. Extended guard intervals, in
the form of prefixes 1505, 1555 and postfixes 1515, 1565 are
appended to each respective block of data 1500, 1550. The prefixes
1505, 1555 are appended to the CP 1510, 1560 of each respective
block of data 1500, 1550. The postfixes 1515, 1565 are appended to
the tail of each respective block of data 1500, 1550. The extended
guard intervals, i.e., prefixes, are appended in order to perform
transmit windowing. In this case, the extended guard intervals,
i.e. the postfix 1515 of the first block of data 1500 and the
prefix 1555 of the second block of data 1550, of consecutive
symbols overlap to reduce spectral loss. At the receiver side,
after receive windowing, the CP and extended guard intervals are
discarded. As in the case shown in FIG. 14, the receive window may
be applied on the CPs and the extended guard intervals. In this
case, the interference due to receive windowing depends on the
extent of the receive window. If the receive window coefficients
are zero on the interval corresponding to the extended guard
interval, then interference would be picked on the cyclic prefix
interval due to multi-path. If some of the receive window
coefficients are non-zero on the interval corresponding to the
extended guard interval, then interference would be picked on the
extended guard interval and cyclic prefix interval even without
multi-path.
[0142] Referring to FIG. 16, a first block of data 1600 and a
second block of data 1650 are shown. In this case, guard intervals,
i.e. cyclic prefixes, are prepended to the respective blocks of
data 1600,1650. Postfixes 1610, 1660 are appended to the end of the
respective blocks of data 1600, 1650. In this case, the CP 1605,
1655 of each respective block of data 1600, 1650 is used to perform
transmit windowing. At the receiver side, after receive windowing,
the CP is discarded. In this case, interference is picked up from
the cyclic prefix interval even without multi-path.
[0143] To understand the cases shown in FIGS. 15 and 16, the
received signal on block may be written as
y l = H siganl { PGF H S l + H signal T .alpha. _ PGF H S l - 1 } +
H isi P [ 0 .beta. .times. ( N - .beta. ) I .beta. I N ] F H S l -
1 + n , where Equation ( 45 ) G = [ 0 .beta. .times. ( N - .beta. )
I .beta. I N I .alpha. x ( N - .alpha. ) 0 .alpha. x ( N - .alpha.
) ] , T .alpha. = [ 0 .alpha. x ( N ^ - .alpha. ) I .alpha. 0 ( N -
.alpha. ) .times. N ^ ] . Equation ( 46 ) ##EQU00038##
[0144] The first term of Equation (45) may be described as follows.
The data block goes through IFFT. To perform transmit windowing,
the last .beta. symbols are copied as the prefix, and the first
.alpha. symbols are copied as the postfix. Note that, since
overlapping of the prefix and postfix of consecutive symbols will
be performed, .alpha. and .beta. are preferred to be equal. Then,
the signal is multiplied by the window coefficients, denoted as the
diagonal matrix P, whose diagonal elements are the window
coefficients. This transmitted signal goes through the channel.
[0145] The second term of Equation (45) is similar to the first
term of Equation (45) and contains the coefficients from the
previous data block. T.sub..alpha. is a matrix that selects the
last .alpha. samples of the windowed signal to create a vector. The
elements of the vector are rearranged from the last coefficient to
the first coefficient. This means that the last .alpha. samples of
the windowed signal is added to the first .alpha. samples of the
current block. {circumflex over (N)}=N+.alpha.+.beta..
[0146] The third term of Equation (45) may be the intersymbol
interference from the previous symbol. The received signal is
processed by the receive windowing. Depending on the window
coefficients, the interference may be estimated and canceled.
[0147] Methods and apparatus for applying different windowing
functions to separate groups of a received signal will now be
described. Different windowing functions may be applied to
different sub-bands of the transmission band. As an example, the
sub-bands next to the edges may be shaped with longer windows to
better reject adjacent channel interference. On the other hand, the
sub-bands away from the edges may be shaped with shorter windows.
Since windowing may introduce distortion, the possibly larger
distortion introduced by the longer windows may be limited to the
sub-bands on the edges. An example of this method is shown in FIG.
17.
[0148] FIG. 17 is a diagram of an example implementation of
receiver windowing for non-contiguous sub-bands. Referring to FIG.
17, an OFDM signal y[n] 1705 containing modulated symbols is
received and input into a serial-to-parallel converter (S/P) 1710
that samples the received signal y[n] 1705, and divides it into a
plurality of subcarriers. It is assumed that in windows will be
applied to in different sub-bands. The signal is copied into in
branches. For each group, or branch, the appropriate receive
windowing is applied at the windowing filters 1715a-n. The receive
windowing applied for each group, or branch may different. For
example, the receive windowing type as shown in FIG. 10 may be
applied to one branch, whereas the receive windowing type as shown
in FIG. 13 may be applied to another branch. It should be noted
that other windowing types, other than those shown in FIGS. 10 and
13 may be used. Then, for each respective branch, the prefix and
postfix are removed, if it exists, at each respective
prefix/postfix removal unit 1720a-n. Then, each branch goes through
an FFT unit 1725a-n, which converts the data in each branch in in
the sub-carriers from the time domain back to the frequency domain.
For each branch, the samples corresponding to the subcarriers of
the sub-band in that branch are selected for further processing at
processing units 1730a-n. Such processing may include equalization
and demodulation.
[0149] The requirements of spectral leakage and interference may
change, therefore, adaptation of the windowing function may be
beneficial. As discussed previously, the window function at the
transmitter shapes the CP and therefore may introduce interference
if the unshaped portion of the CP is not long enough to compensate
for the channel delay spread. Similarly, windowing at the receiver
may introduce interference. There may, however, be a tradeoff
between interference and spectral leakage reduction and adjacent
channel interference rejection. For example, as the roll-off
portion of a window gets longer, e.g., the window gets smoother,
spectral leakage reduction at the transmitter and adjacent channel
interference rejection at the receiver improves. However,
self-created interference, due to ISI and/or ICI may increase.
[0150] Accordingly, it may be beneficial to adaptively change the
window function depending on the requirements on the spectral
leakage and adjacent channel interference rejection. If the
requirements are tight, then a smoother window function may be used
at the transmitter and/or receiver at the expense of more ISI/ICI.
Otherwise, a less smooth window function may be preferred,
resulting in less ISI/ICI. The selection may be done by the
receiver or transmitter based on measurements, such as sensing, or
the receiver may select the receive window function while the
transmitter may select the transmit window function.
[0151] In transmit windowing, the window function may be chosen
such that the samples corresponding to the data block (e.g., those
that are produced by the IFFT before CP attachment) are multiplied
by unity (i.e., the weights of the window function are 1). The
samples that correspond to the CP and postfix may be multiplied by
non-unity weights. If the lengths of the CP, data block, and
postfix are k, n, k, respectively, then the total length of the
window may be n+2k, with the middle samples indexed by k+1 to k+n
being unity. Sometimes, a tight requirement on spectral leakage may
necessitate a very smooth window function to be used at the
transmitter. But since the size of the CP may be fixed, this may
not be possible. One method to overcome this may be to allow some
of the n weights of the window function to have non-unity values.
For example, Equation (47) below may be used. The smoothness of the
window function may be improved significantly without bit error
rate (BER) degradation.
p [ n ] = { 0.5 ( 1 + cos { .pi. ( 1 + n ( .beta. N T + b ) ) } 0
.ltoreq. n < .beta. N T + b 1 .beta. N T + b .ltoreq. n < (
.beta. + 1 ) N T - b 0.5 ( 1 + cos { .pi. n - ( .beta. + 1 ) N T -
b .beta. N T - b } ( .beta. + 1 ) N T - b .ltoreq. n .ltoreq. ( 2
.beta. + 1 ) N T - 1 } Equation ( 47 ) ##EQU00039##
[0152] During receiver windowing, the overall performance of the
receiver may be improved. In one example, the received
signal-interference-to-noise-ratio (SINR) may be improved by adding
the samples corresponding to the CP to the end of the received
symbol. It should be noted that if windowing is not applied, this
operation may be done independently.
[0153] For example, the transmitted signal may be written in matrix
notation as
x l = [ 0 N G .times. ( N - N G ) I N G I N ] F H S l , Equation (
48 ) ##EQU00040##
where F.sup.H is the inverse fast Fourier transform (IFFT) matrix
and
[ 0 N G .times. ( N - N G ) I .beta. I N ] ##EQU00041##
appends a prefix of N.sub.G samples. The received signal after the
channel may be written as
y l = H signal [ 0 N G .times. ( N - N G ) I N G I N ] F H S l + H
isi [ 0 N G .times. ( N - N G ) I N G I N ] F H S l - 1 + w ,
Equation ( 49 ) ##EQU00042##
where the first part of Equation (49) is the desired signal, the
second part is the inter-symbol interference (ISI) from the
previous block, and the third part is the noise. The channel
matrices in Equation (49) may be written as
H signal = [ h 0 0 0 h L - 1 h 0 0 0 h L - 1 h 0 0 0 0 h L - 1 h 0
] , H isi = [ 0 0 h L - 1 h 1 0 h L - 1 0 0 0 ] , Equation ( 50 )
##EQU00043##
where [h.sub.0, h.sub.1, . . . , h.sub.L-1] is the channel response
of the multipath channel.
[0154] In the above example, if the channel order L=N.sub.G, then
all samples of the cyclic prefix may be contaminated. However, when
a wireless communication system is designed, the length of the
cyclic prefix, N.sub.G, may be selected based on worst case
scenarios. Therefore, very often, L<N.sub.G, so N.sub.G-L
samples of the cyclic prefix may be ISI-free. The initial paths of
the channel may have significant power, and the delayed paths may
have much less power. Thus, many samples of the cyclic prefix may
only be contaminated by low-power ISI.
[0155] If sample n of the received OFDM symbol (with the CP) is ISI
free, the received OFDM symbol may be expressed as
y[n]=x[n]+w.sub.1[n]. Due to how the cyclic prefix is formed,
sample (n+N) may be written as y[n+N]=x[n+N]+w.sub.2[n+N], where
x[n]=x[n+N].
[0156] The noise samples w.sub.1 and w.sub.2 may be independent and
may have the same statistics. Assuming they are zero-mean and have
variance .delta..sup.2, if y[n] and y[n+N] are added, the result
may be expressed as
y[n]+y[n+N]=x[n]+x[n+N]+w.sub.1[n]+w.sub.2[n+N]=2x[n]+w.sub.1[n]+w.sub-
.2[n+N].
[0157] The SINR of the added symbols may be
SINR = 4 x n 2 2 .delta. 2 = 2 P o .delta. 2 , ##EQU00044##
where P.sub.0=E{x.sub.n.sup.2}. Accordingly, the power of one
sample is doubled (in the time domain) due to the CP being a copy
of the original samples. In the frequency domain, assuming the
received signal is written as
y [ m ] = k = 0 M - 1 S k [ ] p [ m ] e j 2 .pi. k ( m ) N + w [ n
] , m = - N G , , 0 , 1 , N - 1 , Equation ( 51 ) ##EQU00045##
the estimate of the transmitted signal on subcarrier k may be
written as
S ^ k [ ] = ( 1 N ) m = 0 N - NG - 1 k = 0 N - 1 S k [ ] p [ m ] e
j 2 .pi. k ( m ) N e - j 2 .pi. k ^ ( m ) N + w [ m ] + ( 1 N ) m =
N - NG N - 1 { 0.5 ( k = 0 N - 1 S k [ ] p [ m ] e j 2 .pi. k ( m )
N + w [ n ] ) e - j 2 .pi. k ^ ( m ) N + 0.5 ( k = 0 N - 1 S k [ ]
p [ m ] e j 2 .pi. k ( m ) N + w [ m - N ] ) e - j 2 .pi. k ^ ( m )
N } . Equation ( 52 ) ##EQU00046##
[0158] From this, the SINR on the subcarrier k may be given as
SINR AWGN ( k ) = P 0 { ( N - N G ) + 0.5 N G } .sigma. 2 N = P 0 {
( 1 - N G N ) + 0.5 N G N } .sigma. 2 . Equation ( 53 )
##EQU00047##
[0159] By way of example, if it is assumed that the CP is 25% of
the total symbol duration, and half of the samples are ISI-free,
then the improvement in SINR per sample in the frequency domain is
0.3 dB. Assuming that there is a channel with L.sub.1 paths, where
L.sub.1<N.sub.G, sample n may be written as
y[n]=y[n+N]=.SIGMA..sub.i=0.sup.N.sup.G.sup.L.sup.1.sup.+1h[i]x[n-i].
From this, the SINR improvement per sample in the time domain may
be 2.times. for ISI-free samples. After the IFFT, SINR on the kth
subcarrier before equalization may be SINR.sub.AWGN(k)|H(k)|.sup.2,
where H(k) is the FFT of the channel on the kth subcarrier.
[0160] Even a small increase in SINR may result in a jump in the
rate because the Modulation and Coding Scheme (MCS) works on
discrete channel quality indicator (CQI) values. If a small
increase results in the next MCS, the rate may be doubled in some
scenarios. On the other hand, WTRUs closer to the transmitter may
have most of the samples in the CP ISI-free. Further, even if a
sample is not ISI-free, it may still be useful if the contribution
of the desired signal outweighs the contribution of the ISI. For
example, assuming that the received samples are
y[n]=x[n]+w.sub.1[n]+z[n] and y[n+N]=x[n+N]+w.sub.2[n+N], where
z[n] is the ISI, if y[n] and y[n+N] are added,
y[n]+y[n+N]=x[n]+x[n+N]+w.sub.1[n]+w.sub.2[n+N]+z[n]=2x[n]+w.sub.1[n]+w.s-
ub.2[n+N]+z[n]. The SINR of the added symbols in the time domain
is
SINR = 4 P o 2 .delta. 2 + .sigma. z 2 . ##EQU00048##
If
[0161] SINR = 4 P o 2 .delta. 2 + .sigma. z 2 > P o .delta. 2 ,
##EQU00049##
then it may be beneficial to use that sample from the CP.
[0162] When the samples from the CP are added to the samples at the
tail of the OFDM symbol, each sample may be divided by 2 so that
orthogonality is preserved. This may be due to the orthogonality
condition in receive windowing.
[0163] The result in Equation (53) may be generalized to the case
where a receive window is applied. After receive windowing of the
received samples, the CP may be added to the end of the OFDM
symbol. Assuming an additive white Gaussian noise (AWGN) channel is
used (e.g., ISI has not contaminated the CP samples), the received
signal may be:
y [ n ] = k = 0 N - 1 S k [ ] p [ n ] e j 2 .pi. k ( n ) N + w [ n
] , n = - N G , , 0 , 1 , N - 1 , Equation ( 54 ) ##EQU00050##
where z[n] is the AWGN with zero mean and variance .sigma..sup.2.
The estimate of the transmitted data symbol on subcarrier
{circumflex over (k)} may be written, after windowing with v[n],
as
S ^ k [ ] = ( 1 N ) n = 0 N - N G - 1 k = 0 N - 1 S k [ ] p [ n ] e
j 2 .pi. k ( n ) N e - j 2 .pi. k ^ ( n ) N + z [ n ] + ( 1 N ) n =
N - N G N - 1 { v [ n ] ( k = 0 N - 1 S k [ ] p [ n ] e j 2 .pi. k
( n ) N + z [ n ] ) e - j 2 .pi. k ^ ( n ) N + ( 1 - v [ n ] ) ( k
= 0 N - 1 S k [ ] p [ n ] e j 2 .pi. k ( n ) N + z [ n - N ] ) e -
j 2 .pi. k ^ ( n ) N } Equation ( 55 ) ##EQU00051##
[0164] From Equation (55), the SINR on the subcarrier k may be
given as
( k ) = P 0 { ( N - N G ) + .GAMMA. N G } .sigma. 2 N = P 0 { ( 1 -
N G N ) + .GAMMA. N G N } .sigma. 2 , .GAMMA. = n = N - N G N - 1 1
+ 2 v [ n ] 2 - 2 v [ n ] . Equation ( 56 ) ##EQU00052##
[0165] Similarly, with a multi-path channel (at the absence of
ISI), the SINR becomes
( k ) = SINR H ( k ) 2 { ( 1 - N G N ) + .GAMMA. N G N } - 1 .
##EQU00053##
However, due to ISI, windowing may introduce interference, and the
exact SINR may depend on the channel model.
[0166] With channel estimation, the WTRU may be able to determine
the channel delay spread and distribution of the paths. This would
allow the WTRU to determine if the CP contains samples that are
ISI-free or are contaminated with small enough ISI. Since WTRUs
closer to the transmitter may be most likely to benefit from this
(i.e., CP length is larger than the delay spread), it may be
assumed that they already have better a better signal-to-noise
ratio (SNR) and channel estimation may be reliable.
[0167] FIG. 18 is a graph depicting the improvement in BER in an
AWGN channel due to using CP when IEEE 802.11af is the underlying
radio access medium.
[0168] As discussed previously, the length of the CP in current
wireless systems using CP may be set to one value and may be used
for all users. However, in a given transmission area, some users
may be experiencing channels with smaller delay spread than others.
If that is the case, it may be beneficial to use a shorter CP for
those users experiencing channels with smaller delay spread.
[0169] FIG. 19 is a diagram of an example transmitter 1900 capable
of transmitting symbols with varying CP lengths for different
sub-bands. In the example illustrated in FIG. 19, only two
sub-bands 1905, 1910 are shown. It should be noted that this is for
illustration purposes, and there may be any number of sub-bands,
and sub-bands may include non-contiguous subcarriers or resource
blocks (RBs). The concept of RBs may be generalized as a group of
subcarriers, where the RB size (i.e., the number of subcarriers)
may vary for different RBs. In addition, the transmitter may not
need to be OFDM-based. For example it may be discrete Fourier
transform-spread OFDM (DFT-spread-OFDM), as used in the uplink of
Third Generation Partnership Project (3GPP) LTE, or pure single
carrier (SC). In DFT-spread-OFDM, the data goes through a fast
Fourier transform (FFT) before being mapped to the subcarriers. In
SC, the CP is directly attached to the time domain signal.
[0170] Referring to FIG. 19, for this example, it is assumed that
the transmission for each sub-band 1905, 1910 is similar to
conventional OFDM. That is, symbols from an input data stream are
spread across a plurality of multiple parallel sub-carriers or RBs
of multiple sub-bands, sub-bands 1905, 1910 as illustrated in FIG.
19, which are then input into their respective IFFT units, 1915,
1920. Note that FIG. 19 shows multiple IFFT blocks. This is to show
that conceptually, in IFFTs are taken. However, in a hardware
implementation there may be one IFFT block that may be used in
times for different input streams. The IFFT units 1915, 1920
convert the signals in the plurality of sub-carriers or RBs from
the frequency domain to corresponding time domain signals. CP,
denoted in the illustration as CP-1 and CP-2 is prepended at the
respective CP adder units 1930, 1935. As illustrated in FIG. 19,
CP-1, which is prepended to the signal corresponding to sub-band
1905 is shorter than CP-2, which is prepended to the signal
corresponding to sub-band 1910. A parallel-to-serial converter
(P/S) 1940, 1945 receives the respective samples and converts them
into their respective OFDM signals 1955, 1960 and adds them at an
adder unit 1965 prior to transmission.
[0171] Use of a variable CP may cause, in general, ICI. As in the
example described in FIG. 19, it is assumed that two symbols are
transmitted on two subcarriers and are intended for two different
receivers. If one has a longer CP, the p(t) functions are
different, and maybe written as
x(t)=S.sub.kp.sub.k(t)exp(j2.pi.F.sub.kt)+S.sub.mp.sub.m(t)exp(j2.pi.F.s-
ub.mt). Equation (57)
[0172] At one of the receivers, the data symbol may be estimated as
in Equation (58).
= .intg. x x + T x ( t ) exp ( - j 2 .pi. F k t ) dt = .intg. x x +
T S k p k ( t ) dt + .intg. x x + T S m p m ( t ) exp ( j 2 .pi. (
F m - F k ) t ) dt Equation ( 58 ) ##EQU00054##
The first part of Equation (58) is the desired signal and the
second part is the interference. Integration is performed over an
interval of T, and the CP discarded. But, for the interfering
signal, this T duration may cover an arbitrary portion of its CP
and its data part. Assume that in this T interval, the interfering
signal does not jump from one block to a new one, e.g., there is
only one data symbol. As shown in Equation (59), there will not be
any interference.
.intg. x x + T S m p m ( t ) exp ( j 2 .pi. 1 T ( m - k ) t ) dt =
0 Equation ( 59 ) ##EQU00055##
[0173] However, if the T interval covers two symbols, then 0.
.intg. x y S m 1 p m ( t ) exp ( j 2 .pi. 1 T ( m - k ) t ) dt +
.intg. y x + T S m 2 p m ( t ) exp ( j 2 .pi. 1 T ( m - k ) t ) dt
.noteq. 0. Equation ( 60 ) ##EQU00056##
The same may be true also for multi-path channels since each path
introduces a multiplicative coefficient. If the interval T has only
one data symbol of the interfering signal, then there will be no
interference. So, in general, there may be interference most of the
time. The symbols may drift because they have different lengths.
For some symbols, orthogonality may be preserved.
[0174] A simulation is described to evaluate the interference power
caused by using a variable CP. In this simulation, a transmitter
similar to that described above and in FIG. 19 uses 1024
subcarriers. Half of the 1024 subcarriers are reserved for a first
receiver. The other half of the 1024 subcarriers are reserved for a
second receiver. The CP attached to the signal generated from the
first half of the subcarriers has a length of 32 samples. The CP
attached to the signal generated from the second half of the
subcarriers has a length of 64 samples. FIG. 20 is a graph
illustrating the received interference power in dB at the first
receiver. FIG. 21 is a graph showing a close-up view of FIG. 20
that depicts the first half of the spectrum allocated to the first
user. In the examples provided in FIGS. 20 and 21, the interference
due to ICI/ISI created by different CP timings is higher at the
edges of the spectrum. The effect of the interference may be
reduced by not using several subcarriers between the bands and
using filtering or pulse shaping to reduce the out-of-band emission
at the transmitter and/or improve interference rejection at the
receiver when using variable CP lengths.
[0175] Pulse shaping, or windowing, as described heretofore is one
technique that may be used to reduce the ICI at the transmitter
when using variable CP lengths. FIG. 22 is a diagram of an example
transmitter that uses transmitter windowing to reduce the ICI when
using variable CP lengths. In the example illustrated in FIG. 22,
only two sub-bands 2205, 2210 are shown. It should be noted that
this is for illustration purposes, and there may be any number of
sub-bands, and sub-bands may include non-contiguous subcarriers or
RBs. In addition, the transmitter may not need to be
OFDM-based.
[0176] Referring to FIG. 22, symbols from an input data stream are
spread across a plurality of multiple parallel sub-carriers or RBs
of multiple sub-bands, sub-bands 2205, 2210 as illustrated in FIG.
22, which are then input into their respective IFFT units, 2215,
2220. Note that FIG. 22 shows multiple IFFT blocks. This is to show
that conceptually, in IFFTs are taken. However, in a hardware
implementation there may be one IFFT block that may be used in
times for different input streams. The IFFT units 2215, 2220
convert the signals in the plurality of sub-carriers or RBs from
the frequency domain to corresponding time domain signals. CP,
denoted in the illustration as CP-1 and CP-2 is prepended at the
respective CP adder units 2230, 2235. As illustrated in FIG. 22,
CP-1, which is prepended to the signal corresponding to sub-band
2205 is shorter than CP-2, which is prepended to the signal
corresponding to sub-band 2210. The appropriate windowing functions
are applied at window filters 2240, 2245. It should be noted that a
different windowing function may be used for branch. A
parallel-to-serial converter (P/S) 2250, 2255 receives the
respective windowed samples and converts them into their respective
OFDM signals 2260, 2265. It is important to note that the windowing
function may be applied at window filters positioned after
parallel-to-serial conversion. The placement of the windowing
filters may be implementation specific. The signals 2260, 2265 are
then added together at an adder unit 2275 prior to
transmission.
[0177] FIG. 23 is a diagram of example receivers that use receiver
windowing to reject the interference when using variable CP
lengths. The transmitter structure illustrated in FIG. 23 is the
same as that shown and described in FIG. 19. The transmitter 1900
transmits the added signals over channels 2305, 2310. The signals
are received at the respective receivers 2301, 2302. Each receiver
2301, 2302 corresponds to a different WTRU. Receive windowing may
be applied at window filters 2315, 2320, respectively. Note that
the windows applied by the WTRUs, in general, may be different.
After windowing is applied, the signals are output for additional
processing, as shown in blocks 2330, 2335. These processes may
include discarding each respective CP, e.g., CP-1 and CP-2, taking
the FFT, and equalization. Each WTRU knows the length of CP used
for them. Therefore, at the receiver, when CP is discarded, the
length of the signal discarded is equal to the CP length.
[0178] Another technique that may be used to reduce the ICI when
using variable CP lengths is filtering at the transmitter, at the
receiver, or both. FIG. 24 is a diagram of an example transmitter
using transmitter side filtering when using variable CP lengths.
Referring to FIG. 24, symbols from an input data stream are spread
across a plurality of multiple parallel sub-carriers or RBs of
multiple sub-bands, sub-bands 2405, 2410 as illustrated in FIG. 24,
which are then input into their respective IFFT units, 2415, 2420.
Note that FIG. 24 shows multiple IFFT blocks. This is to show that
conceptually, in IFFTs are taken. However, in a hardware
implementation there may be one IFFT block that may be used in
times for different input streams. The IFFT units 2415, 2240
convert the signals in the plurality of sub-carriers or RBs from
the frequency domain to corresponding time domain signals. CP,
denoted in the illustration as CP-1 and CP-2 is prepended to the
signals at the respective CP adder units 2430, 2435. In the example
shown in FIG. 24, CP-1, which is prepended to the signal
corresponding to sub-band 2405 is shorter than CP-2, which is
prepended to the signal corresponding to sub-band 2410. A
parallel-to-serial converter (P/S) 2440, 2445 receives the
respective samples and converts them into their respective OFDM
signals 2455, 2460. The appropriate filtering operations are
applied at filters 2465, 2470. It should be noted that the filters,
in general, may be different. The signals are then added together
at an adder unit 2475 prior to transmission. FIG. 25 is a diagram
of example receivers that use receiver side filtering to reject the
interference when using variable CP lengths. The transmitter
structure illustrated in FIG. 25 is the same as that shown and
described in FIG. 19. This for illustration purposes only, as
different transmitter structures may be used. The transmitter 1900
transmits the added signals over channels 2505, 2510. The signals
are received at the respective receivers 2501, 2502. Each receiver
2501, 2502 corresponds to a different WTRU. Receive filtering may
be applied at filters 2515, 2520, respectively. Note that the
filters applied by different WTRUs, in general, may be different.
After filtering is applied, the signals are output for additional
processing, as shown in blocks 2530, 2535. These processes may
include discarding each respective CP, (e.g., CP-1 and CP-2),
taking the FFT, and equalization. Each WTRU knows the length of CP
used for them. Therefore, at the receiver, when CP is discarded,
the length of the signal discarded is equal to the CP length. In
addition, as previously noted, the sub-bands used for transmission
may consist of non-contiguous subcarriers or RBs. Here, filtering
may become challenging but may be addressed by either having each
sub-band include only contiguous sub-carriers or dividing each
sub-band into sub-units (e.g., RBs), filtering each RB separately,
and adding the signals from each sub-unit to form the final signal.
This may be similar to resource block filtered OFDM
(RB-F-OFDM).
[0179] FIG. 26 is a diagram of an example RB-F-OFDM based
transmitter 2600. Referring to FIG. 26, the RB-F-OFDM transmitter
2600 comprises multiple filtered-OFDM transmit modules (F-OFDM Tx)
2650a, 2650b, . . . , 2650n, one for each RB, which output per-RB
multicarrier modulated signals 2660a, 2660b, . . . , 2660n for each
RB from the respective symbol vectors 2620a, 2620b, . . . , 2620n.
of each RB. The per-RB multicarrier modulated signals 2660a, 2660b,
. . . , 2660n form the transmit signal 2660 when summed together.
The RB-F-OFDM transmitter 2600 is different than CP-OFDM
transmitters or filtered OFDM transmitters, in that the per-RB
filtered-OFDM transmit modules 2650a, 2650b, . . . , 2650n,
comprised in the RB-F-OFDM transmitter 2600, each only modulate the
subcarriers in one RB and therefore, the low rate of OFDM signal
may be generated and then up-converted to a high rate.
[0180] Each per-RB multicarrier modulated signal 2660a, 2660b, . .
. , 2660n only has a signal overlapping its adjacent RBs but not
the RBs beyond its adjacent RBs. It is assumed that a per-RB
transmit filter brings the signal leakage of a per-RB multicarrier
modulated signal to its non-adjacent RBs to be negligible. The
signal overlap between adjacent RBs may not create inter-subcarrier
interference due to orthogonality between subcarriers in different
RBs.
[0181] FIG. 27 is a diagram of an example Type-I per-RB F OFDM
transmit module (F-OFDM Tx.sub.k) 2700. The F-OFDM Tx.sub.k 2700
may be used as the per-RB filtered-OFDM transmit module 2650 in the
RB-F-OFDM transmitter 2600 as shown in FIG. 26. Referring to FIG.
27, the F-OFDM Tx.sub.k 2700 comprises an IFFT unit 2705, a
parallel-to-serial converter (P/S) 2710, a CP adder unit 2715, an
upsampling unit 2720, a transmit filter 2725, and a RB modulation
unit 2730. The IFFT of the symbol vectors for the kth RB 2620b is
taken at the IFFT unit 2705. CP is prepended at CP adder unit 2715.
Note that CP of length k is prepended. CPs attached in different
RBs do not need to be of the same length. Parallel-to-serial
conversion is performed at P/S 2710. The signal is then upsampled
at the upsampling unit 2720. After upsampling, the signal is
filtered at transmit filter 2725 which outputs the filtered signal
2740. The filtered signal 2740 is modulated into the frequency band
of the kth RB at the RB modulation unit 2730 to form the per-RB
multicarrier modulated signal 2660b. Which is summed with the other
per-RB multicarrier modulated signals 2660a, . . . , 2660n to form
the transmit signal 2660 as shown in FIG. 26.
[0182] FIG. 28 is a diagram of an example RB F-OFDM Receiver
(RB-F-OFDM Rx) corresponding to the RB F-OFDM transmitter 2600 in
FIG. 26. Referring to FIG. 28, the RB-F-OFDM receiver 2800
comprises per-RB F-OFDM receive modules 2850a, 2850b, . . . ,
2850n, that output per-RB demodulated symbol vectors 2890a, 2890b,
. . . , 2890n for each RB from the received multicarrier modulated
signal 2805. The RB-F-OFDM receive modules 2850a, 2850b, . . . ,
2850n each only demodulate the subcarriers in one RB and therefore,
the signal may be down-converted to low rate and then
demodulated.
[0183] FIG. 29 is a diagram of an example Type-I per-RB F-OFDM
receive module (F-OFDM Rx.sub.k) 2900. The F-OFDM Rx.sub.k may be
used as the per-RB-filtered-OFDM receive module 1650 in the
RB-F-OFDM receiver 2800 as shown in FIG. 28. The OFDM Rx.sub.k
shown in FIG. 29 has reverse operations of the F-OFDM Tx.sub.k
shown in FIG. 27. The F-OFDM Rx.sub.k 2900 comprises a RB
demodulation unit 2910, a receive filter 2915, a downsampling unit
2920, a serial-to-parallel converter (S/P) 2925, a CP removal unit
2930, and an FFT unit 2935. Referring to FIG. 29, for the kth RB,
the received signal 2905 is demodulated from the frequency band of
the kth RB to baseband at the RB demodulation unit 2910 to form a
RB demodulated signal 2950. The RB demodulated signal 2950 is then
filtered at filter 2915. The filtered signal is downsampled at the
downsampling unit 2920, the S/P converter 2925, and the CP removal
unit 2930. Note that here, the CP that is discarded may be
different for each RB. The FFT is taken at the FFT unit 2935. The
outputs from the FFT unit 2935 form the demodulated symbol vectors
2990. The demodulated symbol vectors are obtained RB by RB this
way, similar to FIG. 28.
[0184] Since the transmitter will be able to use CPs of different
lengths at a given time, the frame structure may need to be
addressed.
[0185] One option is to keep the subframe length unchanged but to
change the number of OFDM symbols depending on the CP length. This
will result in a different amount of data blocks existing in
different signals. FIG. 30 is a diagram of an example frame
structure corresponding to two signals 3000, 3001. In the example
illustrated in FIG. 30, it is assumed that the transmitter
generates OFDM symbols with two different CPs 3005, 3010. As shown
in FIG. 30, the length of CP 3005 in signal 3000 is shorter than
the length of CP 3010 in signal 3001. The lengths of the data parts
3015a, 3015b of the OFDM symbols in the respective signals 3000,
3001, in the illustrated example are the same. Since the CP 3010 is
longer than CP 3005, the transmitter will fit a smaller number of
OFDM data blocks 3015b when using the longer CP 3010. Thus, signal
3001 may contain less OFDM data blocks 3015b than the OFDM data
blocks 3015a in signal 3000.
[0186] In an example, assume the transmitter is LTE-based and uses
short and long CP simultaneously. The subframe length is 1 ms.
Here, 14 OFDM data blocks may exist in the first signal, and 12
OFDM data blocks may exist in the second signal.
[0187] FIG. 31 is a diagram of another example frame structure
corresponding to two signals 3100, 3101. In the example illustrated
in FIG. 31, different frame formats correspond to different
signals. Referring to FIG. 31, it is assumed that the transmitter
generates OFDM symbols with two different CPs 3105, 3110. As shown
in FIG. 31, the length of CP 3105 in signal 3100 is shorter than
the length of CP 3110 in signal 3101. The lengths of the data parts
3115 of the OFDM symbols in the illustrated example are the same.
Note that the subframe corresponding to signal 3100 is shorter than
the subframe corresponding to signal 3101. Therefore, the same
number of OFDM data blocks may be included in each signal.
Embodiments
[0188] 1. A method for performing pulse shaping at a transmitter,
comprising:
[0189] performing an inverse fast Fourier transform (IFFT) over N
symbols of a current block of data.
[0190] 2. The method of embodiment 1, further comprising:
[0191] padding a prefix of the current block of data.
[0192] 3. The method of embodiment 2, wherein the prefix has a
predetermined length.
[0193] 4. The method of any one of embodiments 1-3, further
comprising: multiplying a result of the IFFT by a windowing
function.
[0194] 5. The method of embodiment 4, wherein the windowing
function is defined by the equation:
p[m],m=0,1, . . . ,(.beta.+1)N.sub.T-1.
[0195] 6. The method of any one of embodiments 1-5, further
comprising: perform a second IFFT on a previous block of data.
[0196] 7. The method of embodiment 6, further comprising:
[0197] adding a prefix and a postfix to a result of the second
IFFT.
[0198] 8. The method of embodiment 7, further comprising:
[0199] multiplying a result of adding the prefix and the postfix to
the result of the second IFFT by a second windowing function.
[0200] 9. The method of embodiment 8, wherein the second windowing
function is defined by the equation:
p[m],m=0,1, . . . ,(2.beta.+1)N.sub.T-1.
[0201] 10. The method of embodiment 9, further comprising:
[0202] taking a last .beta.N.sub.T samples of the windowed signal;
and
[0203] adding the last .beta.N.sub.T samples to a first
.beta.N.sub.T samples of the output of the windowing function.
[0204] 11. The method of embodiment 1, further comprising:
[0205] adding a prefix and a postfix to a result of the IFFT.
[0206] 12. The method of embodiment 11, further comprising:
[0207] multiplying a result of adding the prefix and the postfix to
the result of the IFFT by a windowing function.
[0208] 13. The method of embodiment 12, wherein the windowing
function is defined by the equation:
p[m],m=0,1, . . . ,(2.beta.+1)N.sub.T-1.
[0209] 14. The method of embodiment 13, further comprising:
[0210] retaining a last .beta.N.sub.T samples of the windowed
signal in a buffer.
[0211] 15. The method of embodiment 14, further comprising:
[0212] adding the samples in the buffer to a first .beta.N.sub.T
samples of the windowed signal.
[0213] 16. The method of embodiment 15, further comprising:
[0214] transmitting a first (1+.beta.)N.sub.T samples.
[0215] 17. A method for performing transmitter windowing of a
signal, comprising:
[0216] mapping modulation symbols to corresponding elements of an
inverse fast Fourier transform (IFFT) block.
[0217] 18. The method of embodiment 17, further comprising:
[0218] taking an IFFT of the block.
[0219] 19. The method of embodiment 18, further comprising:
[0220] taking a first M samples with indices; and
[0221] adding the samples to a tail of the block as a postfix.
[0222] 20. The method of embodiment 19, further comprising:
[0223] adding K zeros to a head of the block as a prefix, wherein
the prefix is a zero prefix.
[0224] 21. The method of embodiment 20, further comprising:
[0225] point-wise multiplying the block with a windowing
function.
[0226] 22. The method of embodiment 21, further comprising:
[0227] discarding the zero prefix at a receiver side.
[0228] 23. The method of embodiment 22, further comprising:
[0229] adding a postfix to a first head of the data block; and
[0230] discarding the postfix.
[0231] 24. The method of embodiment 23, further comprising:
[0232] processing a result of the adding the postfix by a fast
Fourier transform.
[0233] 25. A method for performing transmitter windowing for
non-contiguous sub-bands, comprising:
[0234] applying a different windowing function to different
sub-bands.
[0235] 26. The method of embodiment 25, wherein longer windows are
applied to sub-bands adjacent to edges of a transmission band.
[0236] 27. The method of embodiment 25, wherein shorter windows are
applied to sub-bands distant from edges of a transmission band.
[0237] 28. The method of any one of embodiments 25-27, wherein the
sub-bands are non-overlapping, whereby subcarriers in a sub-band
are different than subcarriers in other sub-bands.
[0238] 29. The method of any one of embodiments 25-28, further
comprising:
[0239] passing incoming modulated symbols through a serial to
parallel processor.
[0240] 30. The method of embodiment 29, further comprising:
[0241] mapping the modulated symbols to subcarriers in an inverse
fast Fourier transform (IFFT) block corresponding to the
subcarriers in those sub-bands.
[0242] 31. The method of embodiment 30, further comprising:
[0243] padding an output of the IFFT with a prefix and a postfix
for each sub-band.
[0244] 32. The method of embodiment 31, further comprising:
[0245] point-wise multiplying an output of the padding with a
windowing function.
[0246] 33. The method of embodiment 32, further comprising:
[0247] adding an output of all branches to create a composite
signal to be transmitted.
[0248] 34. A method for performing receive windowing,
comprising:
[0249] rejecting adjacent channel interference leakage.
[0250] 35. The method of embodiment 34, wherein the rejecting
includes
[0251] applying a filter to a received signal.
[0252] 36. The method of embodiment 34, wherein the rejecting
includes
[0253] applying a receive window to a received signal.
[0254] 37. The method of embodiment 36, further comprising:
[0255] converting the received signal back to the frequency domain
via a fast Fourier transform.
[0256] 38. A method for performing successive interference
cancellation, comprising:
[0257] regenerating and canceling interference from a received
signal.
[0258] 39. The method of embodiment 38, wherein the interference is
regenerated from a previous symbol in the received signal and is
subtracted from a current symbol in the received signal.
[0259] 40. The method of embodiment 39, wherein the subtraction is
performed in the time domain.
[0260] 41. The method of embodiment 39, wherein the subtraction is
performed in the frequency domain.
[0261] 42. The method of any one of embodiments 38-41, further
comprising:
[0262] performing receive windowing.
[0263] 43. The method of embodiment 42, further comprising:
[0264] discarding a cyclic prefix.
[0265] 44. The method of embodiments 42 or 43, further
comprising:
[0266] discarding an extended guard interval.
[0267] 45. The method of any one of embodiments 42-44, wherein the
receive window is applied to the cyclic prefix and the extended
guard interval.
[0268] 46. The method of any one of embodiments 38-45, wherein
different windowing functions are applied to different
sub-bands.
[0269] 47. The method of embodiment 46, wherein longer windows are
applied to sub-bands adjacent to edges of a transmission band.
[0270] 48. The method of embodiment 46, wherein shorter windows are
applied to sub-bands distant from edges of a transmission band.
[0271] 49. The method of any one of embodiments 38-48, further
comprising:
[0272] passing modulated symbols of the received signal through a
serial to parallel processor.
[0273] 50. The method of embodiment 49, further comprising:
[0274] copying the received signal to M branches.
[0275] 51. The method of embodiment 50, further comprising:
[0276] applying receive windowing to each of the M branches.
[0277] 52. The method of embodiment 51, further comprising:
[0278] removing the prefix and the postfix from the received
signal.
[0279] 53. The method of embodiment 52, further comprising:
[0280] performing a fast Fourier transform on the received signal
after the prefix and the postfix have been removed.
[0281] 54. The method of embodiment 53, further comprising: