U.S. patent application number 16/468680 was filed with the patent office on 2020-03-12 for flyback switching power supply.
This patent application is currently assigned to MORNSUN GUANGZHOU SCIENCE & TECHNOLOGY CO., LTD.. The applicant listed for this patent is MORNSUN GUANGZHOU SCIENCE & TECHNOLOGY CO., LTD.. Invention is credited to Baojun WANG.
Application Number | 20200083819 16/468680 |
Document ID | / |
Family ID | 59145086 |
Filed Date | 2020-03-12 |
United States Patent
Application |
20200083819 |
Kind Code |
A1 |
WANG; Baojun |
March 12, 2020 |
FLYBACK SWITCHING POWER SUPPLY
Abstract
A flyback switch power supply connects N.sub.P1 heteronymous
terminals in a transformer B to a power source, grounds second
primary side winding N.sub.P2 heteronymous ends, and ensures that
N.sub.P1 and N.sub.P2 are dual-wire parallel windings. Adding a
capacitor C1, one end of C1 is connected to N.sub.P1 homonymous
terminals, and the other end is connected to N.sub.P2 homonymous
terminals. The secondary side winding uses a Q2 connection method
that is the opposite to the prior art, and is controlled by a PWM
signal controlled by another output voltage. The following effect
is realized: when Q1 is connected, N.sub.P1 and N.sub.P2 are both
excited, and there is artificial surplus energy; when Q1 is
disconnected, the secondary side N.sub.S implements a rectified
output voltage via Q2 on the basis of the output load requirements,
and the leakage inductance and excess energy are non-destructively
absorbed by N.sub.P2 via D1.
Inventors: |
WANG; Baojun; (Guangdong,
CN) |
|
Applicant: |
Name |
City |
State |
Country |
Type |
MORNSUN GUANGZHOU SCIENCE & TECHNOLOGY CO., LTD. |
Guangdong |
|
CN |
|
|
Assignee: |
MORNSUN GUANGZHOU SCIENCE &
TECHNOLOGY CO., LTD.
Guangdong
CN
|
Family ID: |
59145086 |
Appl. No.: |
16/468680 |
Filed: |
February 2, 2018 |
PCT Filed: |
February 2, 2018 |
PCT NO: |
PCT/CN2018/075024 |
371 Date: |
June 12, 2019 |
Current U.S.
Class: |
1/1 |
Current CPC
Class: |
H02M 3/33523 20130101;
H02M 3/33507 20130101; H02M 3/33592 20130101; H02M 1/44 20130101;
H02M 3/33576 20130101; H02M 2001/0064 20130101 |
International
Class: |
H02M 3/335 20060101
H02M003/335; H02M 1/44 20060101 H02M001/44 |
Foreign Application Data
Date |
Code |
Application Number |
Mar 10, 2017 |
CN |
201710142831.6 |
Claims
1. A flyback switching power supply comprising: a transformer, a
first switch transistor, a second switch transistor, a second
capacitor, and a first diode, wherein the first switch transistor
and the second switch transistor are both N-channel field-effect
transistors, the transformer comprises a first primary-side
winding, a second primary-side winding, and a secondary-side
winding, an undotted terminal of the secondary-side winding is
connected to a drain of the second switch transistor, a source of
the second switch transistor is connected to one end of the second
capacitor to form a positive output, and a dotted terminal of the
secondary-side winding is connected to the other end of the second
capacitor to form a negative output; a positive terminal of an
input DC power supply is connected to both a dotted terminal of the
first primary-side winding and a cathode of the first diode, and an
undotted terminal of the first primary-side winding is connected to
a drain of the first switch transistor; an anode of the first diode
is connected to an undotted terminal of the second primary-side
winding, a source of the first switch transistor is connected to a
dotted terminal of the second primary-side winding, and a
connection point is also connected to a negative terminal of the
input DC power supply; a gate of the first switch transistor is
connected to a primary-side control signal; wherein the first
primary-side winding and the second primary-side winding are
bifilar-wound, the flyback switching power supply further comprises
a first capacitor, one end of the first capacitor is connected to
the undotted terminal of the first primary-side winding, the other
end of the first capacitor is connected to the undotted terminal of
the second primary-side winding, and a gate of the second switch
transistor is connected to a secondary-side control signal.
2. A flyback switching power supply comprising: a transformer, a
first switch transistor, a second switch transistor, a second
capacitor, and a first diode, wherein the first switch transistor
and the second switch transistor are both N-channel field-effect
transistors, the transformer comprises a first primary-side
winding, a second primary-side winding, and a secondary-side
winding, an undotted terminal of the secondary-side winding is
connected to a drain of the second switch transistor, a source of
the second switch transistor is connected to one end of the second
capacitor to form a positive output, and a dotted terminal of the
secondary-side winding is connected to the other end of the second
capacitor to form a negative output; a positive terminal of an
input DC power supply is connected to both a drain of the first
switch transistor and the undotted terminal of the second
primary-side winding, and a source of the first switch transistor
is connected to a dotted terminal of the first primary-side
winding; the dotted terminal of the second primary-side winding is
connected to a cathode of the first diode, an undotted terminal of
the first primary-side winding is connected to an anode of the
first diode, and a connection point is also connected to a negative
terminal of the input DC power supply; a gate of the first switch
transistor is connected to a primary-side control signal; wherein
the first primary-side winding and the second primary-side winding
are bifilar-wound, the flyback switching power supply further
comprises a first capacitor, one end of the first capacitor is
connected to the dotted terminal of the first primary-side winding,
the other end of the first capacitor is connected to the dotted
terminal of the second primary-side winding, and a gate of the
second switch transistor is connected to a secondary-side control
signal.
3. A flyback switching power supply comprising: a transformer, a
first switch transistor, a second switch transistor, a second
capacitor, and a first diode, wherein the first switch transistor
is a P-channel field-effect transistor and the second switch
transistor is an N-channel field-effect transistor, the transformer
comprises a first primacy-side winding, a second primary-side
winding, and a secondary-side winding, an undotted terminal of the
secondary-side winding is connected to a drain of the second switch
transistor, a source of the second switch transistor is connected
to one end of the second capacitor to form a positive output, and a
dotted terminal of the secondary-side winding is connected to the
other end of the second capacitor to form a negative output; a
negative terminal of an input DC power supply is connected to both
an undotted terminal of the first primary-side winding and an anode
of the first diode, and a dotted terminal of the first primary-side
winding is connected to a drain of the first switch transistor; a
cathode of the first diode is connected to the dotted terminal of
the second primary-side winding, a source of the first switch
transistor is connected to the undotted terminal of the second
primary-side winding, and a connection point is also connected to a
positive terminal of the input DC power supply; a gate of the first
switch transistor is connected to a primary-side control signal;
wherein the first primary-side winding and the second primary-side
winding are bifilar-wound, the flyback switching power supply
further comprises a first capacitor, one end of the first capacitor
is connected to the dotted terminal of the first primary-side
winding, the other end of the first capacitor is connected to the
dotted terminal of the second primary-side winding, and a gate of
the second switch transistor is connected to a secondary-side
control signal.
4. A flyback switching power supply comprising: a transformer, a
first switch transistor, a second switch transistor, a second
capacitor, and a first diode, wherein the first switch transistor
is a P-channel field-effect transistor and the second switch
transistor is an N-channel field-effect transistor, the transformer
comprises a first primary-side winding, a second primary-side
winding, and a secondary-side winding, an undotted terminal of the
secondary-side winding is connected to a drain of the second switch
transistor, a source of the second switch transistor is connected
to one end of the second capacitor to form a positive output, and a
dotted terminal of the secondary-side winding is connected to the
other end of the second capacitor to form a negative output; a
negative terminal of an input DC power supply is connected to both
a drain of the first switch transistor and the dotted terminal of
the second primary-side winding, and a source of the first switch
transistor is connected to an undotted terminal of the first
primary-side winding; the undotted terminal of the second
primary-side winding is connected to an anode of the first diode, a
dotted terminal of the first primary-side winding is connected to a
cathode of the first diode, and a connection point is also
connected to a positive terminal of the input DC power supply; a
gate of the first switch transistor is connected to a primary-side
control signal; wherein the first primary-side winding and the
second primary-side winding are bifilar-wound, the flyback
switching power supply further comprises a first capacitor, one end
of the first capacitor is connected to the undotted terminal of the
first primary-side winding, the other end of the first capacitor is
connected to the undotted terminal of the second primary-side
winding, and a gate of the second switch transistor is connected to
a secondary-side control signal.
5. The flyback switching power supply according to claim 1, wherein
the secondary-side control signal is a PWM signal controlled by a
voltage between the positive output and the negative output.
6. The flyback switching power supply according to claim 1, wherein
a duty cycle of the primary-side control signal is fixed.
7. The flyback switching power supply according to claim 1, wherein
a duty cycle of the primary-side control signal is provided
according to m times of an actual output power of a secondary side,
and the duty cycle does not increase until reaching a maximum duty
cycle.
8. The flyback switching power supply according to claim 1, wherein
directions of physical paths of excitation currents of the first
primary-side winding and the second primary-side winding during PCB
wiring are opposite.
9. The flyback switching power supply according to claim 2, wherein
the secondary-side control signal is a PWM signal controlled by a
voltage between the positive output and the negative output.
10. The flyback switching power supply according to claim 3,
wherein the secondary-side control signal is a PWM signal
controlled by a voltage between the positive output and the
negative output.
11. The flyback switching power supply according to claim 4,
wherein the secondary-side control signal is a PWM signal
controlled by a voltage between the positive output and the
negative output.
12. The flyback switching power supply according to claim 2,
wherein a duty cycle of the primary-side control signal is
fixed.
13. The flyback switching power supply according to claim 3,
wherein a duty cycle of the primary-side control signal is
fixed.
14. The flyback switching power supply according to claim 4,
wherein a duty cycle of the primary-side control signal is
fixed.
15. The flyback switching power supply according to claim 2,
wherein a duty cycle of the primary-side control signal is provided
according to m times of an actual output power of a secondary side,
and the duty cycle does not increase until reaching a maximum duty
cycle.
16. The flyback switching power supply according to claim 3,
wherein a duty cycle of the primary-side control signal is provided
according to m times of an actual output power of a secondary side,
and the duty cycle does not increase until reaching a maximum duty
cycle.
17. The flyback switching power supply according to claim 4,
wherein a duty cycle of the primary-side control signal is provided
according to m times of an actual output power of a secondary side,
and the duty cycle does not increase until reaching a maximum duty
cycle.
18. The flyback switching power supply according to claim 2,
wherein directions of physical paths of excitation currents of the
first primary-side winding and the second primary-side winding
during PCB wiring are opposite.
19. The flyback switching power supply according to claim 3,
wherein directions of physical paths of excitation currents of the
first primary-side winding and the second primary-side winding
during PCB wiring are opposite.
20. The flyback switching power supply according to claim 4,
wherein directions of physical paths of excitation currents of the
first primary-side winding and the second primary-side winding
during PCB wiring are opposite.
Description
TECHNICAL FIELD
[0001] The present invention relates to the field of switching
power supplies, and in particular, to a flyback switching power
supply.
BACKGROUND ART
[0002] At present, switching power supplies have been extensively
used. For applications with input power below 75 W and having no
requirement on power factor (PF), flyback switching power supplies
may have fascinating advantages: a simple circuit topology and a
wide input voltage range. Since the number of components is small,
the reliability of the circuit is relatively high, and the flyback
switching power supply is widely used. For convenience, in many
documents, the flyback switching power supply is also referred to
as a flyback switching power supply, a flyback power supply, and a
flyback converter. In Japan and Taiwan, it is also referred to as a
flyback converter, a flyback switching power supply, and a flyback
power supply. A common topology for AC/DC converters is shown in
FIG. 0. The prototype of the figure is from page 60 of Topology and
Design of Switching Power Supply Power Converters with the book No.
ISBN978-7-5083-9015-4 written by Dr. Zhang Xingzhu. It consists of
a rectifier bridge 101, a filter circuit 200, and a basic flyback
topology unit circuit 300, wherein 300 is also referred to as a
main power stage. A practical circuit is also provided with
pressure-sensitive, NTC thermistor, and electromagnetic
interference (EMI), and other protection circuits in front of the
rectifier bridge, to ensure that the electromagnetic compatibility
of the flyback switching power supply meets a use requirement. In
the flyback switching power supply, a minimum leakage inductance
between primary-side and secondary-side windings is required, so
that conversion efficiency is high. In addition, a withstand
voltage carried by a primary-side main power switch transistor is
also reduced. For the flyback switching power supply that uses an
RCD network for demagnetization and absorption, losses of the RCD
are also reduced. Note: RCD absorption refers to an absorption
circuit consisting of a resistor, a capacitor, and a diode. The
literature in China is the same as the international one. The
letter R is usually used to number of the resistor and represent
the resistor. The letter C is used to number the capacitor and
represent the capacitor. The letter D is used to number the diode
and represent the diode. The resistor and the capacitor are
connected in parallel, and then connected in series with to diode
to form an RCD network.
[0003] When there is no rectifier bridge 101, 200 and 300 can
constitute a DC/DC switching power supply or converter. Because it
is supplied by DC, there is no requirement for the power factor,
and the power may be more than 75 W. In fact, the use of flyback
topology in low-voltage DC/DC switching power supplies is not
mainstream. This is because the input current is discontinuous and
the ripple is relatively large, which has a relatively high
requirement for a previous power supply device. The output current
is also discontinuous, and the ripple is large, which has a
relatively high requirement for capacity of a subsequent filter
capacitor. Especially when the input voltage is relatively low,
since the excitation current becomes large, the primary-side
winding has to be bifilar-wound with a plurality of strands to
reduce the loss of the skin effect. The inductance of the
primary-side winding is also relatively low, and it is often found
that the calculated number of turns is insufficient for winding
full a wire casing of a skeleton from left to right in a tiled
manner. When a working voltage is relatively high, the primary-side
winding may adopt the sandwich series connection scheme. At a low
working voltage, the series connection causes the inductance to be
excessively large, and the sandwich parallel connection scheme has
to be used. Because the two primary-side windings are not in the
same layer, there is leakage inductance between the two
primary-side windings. Consequently, losses are generated, which
makes the efficiency of the switching power supply become low,
causing the following problems:
[0004] During excitation, due to the leakage inductance, the
induced voltage difference has a voltage difference in the leakage
inductance, causing non-negligible loss and it might be easier to
understand in the following manner: for two primary-side windings
that are connected in parallel, if the difference between the
numbers of turns is one, it is equivalent to that inter-turn
short-circuit of this turn exists, but the short-circuit is formed
by using DC internal resistance of the two primary-side windings
that are connected in parallel. Relatively, the loss is not as
large as that of a real inter-turn short-circuit. During
demagnetization, that is, the rectifier diode of the secondary side
is conducted, and the output filter capacitor is continuously
charged. In this case, the primary side induces a reflected
voltage, and the two primary-side windings that are connected in
parallel induce voltages that are not equal. Due to low internal
resistance of the winding, it is induced that current caused by the
voltages that are not equal is not small, causing losses and
relatively large electromagnet interference. If a third winding is
used for demagnetization, which of the two primary-side windings
that are connected in parallel are bifilar-wound with the third
winding? Only two third windings can be used, which are
respectively bifilar-wound with the two primary-side windings that
are connected in parallel, and then are connected in parallel to
form a "third winding". The process is complicated, and the third
winding formed by parallel connection by two windings also induces
voltages that are not equal, causing losses and large
electromagnetic interference. In fact, for the common
demagnetization by the third winding, the advantage is
non-destructive demagnetization, and the efficiency is relatively
high, but the choice of the wire diameter of the third winding is
also a problem: a relatively small wire diameter is selected, and
parallel winding with the primary-side winding is relatively
troublesome, and the thin wire is easily pulled apart. If a wire
diameter the same as that of the primary-side winding is selected,
costs are high. The third winding demagnetizes the flyback
switching power supply is also referred to as "the three-winding
absorbs the flyback switching power supply".
[0005] The flyback switching power supply still has one
shortcoming: the bandwidth of the switching power supply is
insufficient, that is, the loop response is poor. For a common
switching power supply with a working frequency of 65 KHz, the
bandwidth thereof is usually only a few hundred Hz, usually below
400 Hz. To achieve 1 KHz, design engineers need superb design
experience, superb circuit board design level, and superb debugging
skills. For a switching power supply with a working frequency of
280 to 330 kHz, the bandwidth thereof is usually only 1 to 2 KHz.
It is also very difficult to achieve 10 kHz. This is determined by
the inherent working characteristics of the flyback switching power
supply. An optocoupler detects a voltage change on an output end,
to determine the duty cycle of the primary side for the following
implementation: after considering the efficiency loss, in the unit
time, the energy of the primary-side excitation is equal to the
energy output by the secondary side in the same period. However, as
the jump frequency of the load increases, a control loop of the
system cannot keep synchronization. This is also an important
reason why for a flyback switching power supply, such as a printer
or an automatic door, is rarely seen in low-voltage DC/DC switching
power supplies on occasions where a requirement is made on the
bandwidth.
[0006] Two primary-side windings that are connected in parallel are
applied to low-voltage DC/DC switching power supplies. Low-voltage
DC/DC switching power supplies usually refer to switching power
supplies with input voltages below 48V. Low-voltage DC/DC switching
power supplies of some uses can work up to DC 160V, such as railway
power supplies.
SUMMARY OF THE INVENTION
[0007] In view of this, to overcome the deficiencies of the
existing low-voltage flyback switching power supply, the present
invention provides a flyback switching power supply. The
primary-side winding may be used without parallel connection
between two separate primary-side windings. That is, leakage
inductance between primary and secondary-side windings is allowed
to be relatively large, and a third winding is not used for
demagnetization. Moreover, the conversion efficiency is not
reduced, the losses during excitation and demagnetization are
reduced, the bandwidth is increased, and the loop response is
good.
[0008] The objective of the present invention is achieved in the
following manner: a flyback switching power supply, comprising: a
transformer, a first switch transistor, a second switch transistor,
wherein the first and second switch transistors are both N-channel
field-effect transistors, a second capacitor, and a first diode,
wherein the transformer comprises a first primary-side winding, a
second primary-side winding, and a secondary-side winding, an
undotted terminal of the secondary-side winding is connected to a
drain of the second switch transistor, a source of the second
switch transistor is connected to one end of the second capacitor;
to form a positive output, and a dotted terminal of the
secondary-side winding is connected to the other end of the second
capacitor to form a negative output; a positive terminal of an
input DC power supply is connected to both a dotted terminal of the
first primary-side winding and a cathode of the first diode, and an
undotted terminal of the first primary-side winding is connected to
a drain of the first switch transistor; an anode of the first diode
is connected to an undotted terminal of the second primary-side
winding, a source of the first switch transistor is connected to a
dotted terminal of the second primary-side winding, and a
connection point is also connected to a negative terminal of the
input DC power supply; a gate of the first switch transistor is
connected to a primary-side control signal; characterized in that,
the first primary-side winding and the second primary-side winding
are bifilar-wound, and a first capacitor is further comprised; one
end of the first capacitor is connected to the undotted terminal of
the first primary-side winding, and the other end of the first
capacitor is connected to the undotted terminal of the second
primary-side winding, and a gate of the second switch transistor is
connected to a secondary-side control signal.
[0009] Preferably, the secondary-side control signal is a PWM
signal controlled by a voltage between the positive output and the
negative output.
[0010] The present invention further provides a solution equivalent
to solution 1, and the equivalent solution is solution 2: the
objective of the present invention may be further achieved in the
following manner: a flyback switching power supply, comprising: a
transformer, a first switch transistor, a second switch transistor,
wherein the first and second switch transistors are both N-channel
field-effect transistors, a second capacitor, and a first diode,
wherein the transformer comprises a first primary-side winding, a
second primary-side winding, and a secondary-side winding, an
undotted terminal of the secondary-side winding is connected to a
drain of the second switch transistor, a source of the second
switch transistor is connected to one end of the second capacitor
to form a positive output, and a dotted terminal of the
secondary-side winding is connected to the other end of the second
capacitor to form a negative output; a positive terminal of an
input DC power supply is connected to both a drain of the first
switch transistor and the undotted terminal of the second
primary-side winding, and a source of the first switch transistor
is connected to a dotted terminal of the first primary-side
winding; the dotted terminal of the second primary-side winding is
connected to a cathode of the first diode, an undotted terminal of
the first primary-side winding is connected to an anode of the
first diode, and a connection point is also connected to a negative
terminal of the input DC power supply; a gate of the first switch
transistor is connected to a primary-side control signal;
characterized in that, the first primary-side winding and the
second primary-side winding are bifilar-wound, and a first
capacitor is further comprised; one end of the first capacitor is
connected to the dotted terminal of the first primary-side winding,
and the other end of the first capacitor is connected to the dotted
terminal of the second primary-side winding, and a gate of the
second switch transistor is connected to a secondary-side control
signal.
[0011] Preferably, the secondary-side control signal is a PWM
signal controlled by a voltage between the positive output and the
negative output.
[0012] The present invention further provides a technical solution
of using a P-channel field-effect transistor as a first switch
transistor. Based on the solution 1, polarities of the power
supply, the diode, and the dotted terminal need to be reversed, and
the polarity of an output rectifier portion does not need to be
reserved. Then solution 3 is obtained: a flyback switching power
supply, comprising: a transformer, a first switch transistor, a
second switch transistor, wherein the first switch transistor is a
P-channel field-effect transistor and the second switch transistor
is an N-channel field-effect transistor, a second capacitor, and a
first diode, wherein the transformer comprises a first primary-side
winding, a second primary-side winding, and a secondary-side
winding, an undotted terminal of the secondary-side winding is
connected to a drain of the second switch transistor, a source of
the second switch transistor is connected to one end of the second
capacitor to form a positive output, and a dotted terminal of the
secondary-side winding is connected to the other end of the second
capacitor to form a negative output; a negative terminal of an
input DC power supply is connected to both an undotted terminal of
the first primary-side winding and an anode of the first diode, and
a dotted terminal of the first primary-side winding is connected to
a drain of the first switch transistor; a cathode of the first
diode is connected to the dotted terminal of the second
primary-side winding, a source of the first switch transistor is
connected to the undotted terminal of the second primary-side
winding, and a connection point is also connected to a positive
terminal of the input DC power supply; a gate of the first switch
transistor is connected to a primary-side control signal;
characterized in that, the first primary-side winding and the
second primary-side winding are bifilar-wound, and a first
capacitor is further comprised; one end of the first capacitor is
connected to the dotted terminal of the first primary-side winding,
and the other end of the first capacitor is connected to the dotted
terminal of the second primary-side winding, and a gate of the
second switch transistor is connected to a secondary-side control
signal.
[0013] Preferably, the secondary-side control signal is a PWM
signal controlled by a voltage between the positive output and the
negative output.
[0014] The present invention further provides a solution equivalent
to solution 3, and the equivalent solution is a technical solution
of using a P-channel field-effect transistor as a first switch
transistor of solution 2. Based on the solution 2, polarities of
the power supply, the diode, and the dotted terminal need to be
reversed, and the polarity of an output rectifier portion does not
need to be reserved. Then solution 4 is obtained: the objective of
the present invention may be further achieved in the following
manner: a flyback switching power supply, comprising: a
transformer, a first switch transistor, a second switch transistor,
wherein the first switch transistor is a P-channel field-effect
transistor and the second switch transistor is an N-channel
field-effect transistor, a second capacitor, and a first diode,
wherein the transformer comprises a first primary-side winding, a
second primary-side winding, and a secondary-side winding, an
undotted terminal of the secondary-side winding is connected to a
drain of the second switch transistor, a source of the second
switch transistor is connected to one end of the second capacitor
to form a positive output, and a dotted terminal of the
secondary-side winding is connected to the other end of the second
capacitor to form a negative output; a negative terminal of an
input DC power supply is connected to both a drain of the first
switch transistor and the dotted terminal of the second
primary-side winding, and a source of the first switch transistor
is connected to an undotted terminal of the first primary-side
winding; the undotted terminal of the second primary-side winding
is connected to an anode of the first diode, a dotted terminal of
the first primary-side winding is connected to a cathode of the
first diode, and a connection point is also connected to a positive
terminal of the input DC power supply; a gate of the first switch
transistor is connected to a primary-side control signal;
characterized in that, the first primary-side winding and the
second primary-side winding are bifilar-wound, and a first
capacitor is further comprised; one end of the first capacitor is
connected to the undotted terminal of the first primary-side
winding, and the other end of the first capacitor is connected to
the undotted terminal of the second primary-side winding, and a
gate of the second switch transistor is connected to a
secondary-side control signal.
[0015] Preferably, the secondary-side control signal is a PWM
signal controlled by a voltage between the positive output and the
negative output.
[0016] As an improvement to the foregoing four solutions and
preferred solutions thereof, a duty cycle of the primary-side
control signal is fixed.
[0017] Preferably, a duty cycle of the primary-side control signal
is provided according to m times of an actual output power of a
secondary side, and the duty cycle does not increase until reaching
a maximum duty cycle.
[0018] Preferably, directions of physical paths of excitation
currents of the first primary-side winding and the second
primary-side winding during PCB wiring are opposite.
[0019] The working principle will be illustrated in detail with
reference to embodiments. The beneficial effects of the present
invention are: leakage inductance between primary and
secondary-side windings is allowed to be relatively large, and
bifilar winding is still used on the primary side, the conversion
efficiency is high, the EMI performance is good, and the bandwidth
is relatively good.
BRIEF DESCRIPTION OF THE DRAWINGS
[0020] FIG. 0 is a diagram of a principle of using an existing
flyback switching power supply for AC to DC conversion.
[0021] FIG. 1 is a diagram of a principle of a first embodiment
corresponding to solution 1 of a flyback switching power supply of
the present invention.
[0022] FIG. 1-1 is a schematic diagram of charging a capacitor C1
during power-on in the first embodiment.
[0023] FIG. 1-2 is a schematic diagram of a voltage polarity after
charging of the capacitor C1 is completed after power-on in the
first embodiment.
[0024] FIG. 1-3 is a schematic diagram of generating two paths of
excitation currents 41 and 42 during saturation conduction of Q1 in
the first embodiment.
[0025] FIG. 1-4 is a schematic diagram of generating a freewheeling
current 43 and a demagnetizing current 4 during disconnection of Q1
in the first embodiment.
[0026] FIG. 1-5a is a waveform graph of Q2 working in a first mode
in the first embodiment.
[0027] FIG. 1-5b is a waveform graph of Q2 working in a second mode
in the first embodiment.
[0028] FIG. 1-5c is a waveform graph of Q2 working in a third mode
in the first embodiment.
[0029] FIG. 1-6 is an equivalent principle diagram in a
secondary-side rectifier circuit in FIG. 1 in the first
embodiment.
[0030] FIG. 2 is a diagram of a principle of a second embodiment
corresponding to solution 2 of a flyback switching power supply of
the present invention.
[0031] FIG. 2-1 is a diagram of a principle of using a P-channel
field-effect transistor in a secondary-side rectifier circuit in
the present invention.
[0032] FIG. 2-2 is a diagram of an equivalent principle of using a
P-channel field-effect transistor in a secondary-side rectifier
circuit in the present invention.
[0033] FIG. 3 is a diagram of a principle of a third embodiment
corresponding to solution 3 of a flyback switching power supply of
the present invention.
[0034] FIG. 4 is a diagram of a principle of a fourth embodiment
corresponding to solution 4 of a flyback switching power supply of
the present invention.
DETAILED DESCRIPTION OF THE INVENTION
First Embodiment
[0035] FIG. 1 is a diagram of a principle of a flyback switching
power supply according to a first embodiment of the present
invention. The flyback switching power supply includes a
transformer B, a first switch transistor Q1, and a second switch
transistor Q2, where the first switch transistor Q1 and the second
switch transistor Q2 are both N-channel field-effect transistors, a
second capacitor C2, and a first diode D1. The transformer B
includes a first primary-side winding N.sub.P1, a second
primary-side winding N.sub.P2, and a secondary-side winding
N.sub.S. An undotted terminal of the secondary-side winding N.sub.S
is connected to a drain d of the second switch transistor Q2, and a
source s of the second switch transistor Q2 is connected to one end
of the second capacitor C2, to form positive output, which is the
+end of Vout in the figure. A dotted terminal of the secondary-side
winding N.sub.S is connected to the other end of the second
capacitor C2, to form negative output, which is the -end of Vout in
the figure. A positive terminal+ of an input DC power supply
U.sub.DC is connected to both a dotted terminal of the first
primary-side winding N.sub.P1 and a cathode of the first diode D1,
and an undotted terminal of the first primary-side winding N.sub.P1
is connected to a drain d of the first switch transistor Q1. An
anode of the first diode D1 is connected to an undotted terminal of
the second primary-side winding N.sub.P2, a source s of the first
switch transistor Q1 is connected to a dotted terminal of the
second primary-side winding N.sub.P2, and a connection point is
also connected to a negative terminal- of the input DC power supply
U.sub.DC. A gate g of the first switch transistor Q1 is connected
to a primary-side control signal. The first primary-side winding
N.sub.P1 and the second primary-side winding N.sub.P1 are
bifilar-wound, and a first capacitor C1 is further included. One
end of the first capacitor C1 is connected to the undotted terminal
of the first primary-side winding N.sub.P1, and the other end of
the first capacitor C1 is connected to the undotted terminal of the
second primary-side winding N.sub.P2, and a gate g of the second
switch transistor Q2 is connected to a secondary-side control
signal. The secondary-side control signal is a PWM signal
controlled by a voltage between the positive output and the
negative output.
[0036] Dotted terminal: the end marked by a black dot in the
winding in the figure.
[0037] Undotted terminal: the end not marked by a black dot in the
winding in the figure.
[0038] Primary-side control signal: various square waves including
a PWM pulse width modulated signal, a PFM pulse frequency
modulation, and the like.
[0039] Secondary-side control signal: a PWM signal controlled by a
voltage between the positive output and the negative output, and
various square waves including a PWM pulse width modulated signal,
a PFM pulse frequency modulation, and the like are all referred to
as PWM signals.
[0040] Transformer B: magnetic cores of the first primary-side
winding NP1 and the second primary-side winding N.sub.P2 are
connected by a dotted line in the figure, indicating that they are
wound on a transformer, and share a same magnetic core, and the
transformer is not an independent transformer, and the drawing
method in the figure is used only for clear patterns and a simple
connection relationship.
[0041] In FIG. 1, the source of the N-channel field-effect
transistor Q1 is connected to the dotted terminal of the second
primary-side winding N.sub.P2, and the connection point is also
connected to the negative terminal- of the input DC power supply
U.sub.DC. Namely, the source of the field-effect transistor Q1 is
connected to the negative terminal- of the input DC power supply
U.sub.DC, and this is not directly present in actual application,
because in the field of switching power supplies, unnecessary
factors are omitted during analysis of a working principle of basic
topology. In actual application, the source of the field-effect
transistor is connected to a current detection resistor or a
current mutual inductor to detect an average current or a peak
current to implement various control policies. Connection to the
source through the current detection resistor or the current mutual
inductor is equivalent to connection to the source, and this is a
common technology in the art, and this application complies with a
default rule in the industry. If a current mutual inductor is used,
the current mutual inductor may appear anywhere in the exciting
circuit, such as a drain of a field-effect transistor, a dotted
terminal or an undotted terminal of the first primary-side winding,
and in addition to a conventional magnetic core type mutual
inductor whose primary side is one turn, and whose secondary side
is a coil with a plurality of turns, the current mutual inductor
may also be a Hall sensor.
[0042] Working principle: Referring to FIG. 1, when the first
capacitor C1 (for the convenience of analysis, according to the
standard of the textbook, the first capacitor C1 is referred to as
a capacitor C1 or C1 hereinafter; this also applies to other
devices, for example, an input DC power supply is referred to as a
DC power supply or a power supply) does not exist, the primary side
part of the circuit is a flyback switching power supply
demagnetized by a third winding, and the second primary-side
winding N.sub.P2 becomes the "third winding" dedicated for
demagnetization. However, after the capacitor C1 is added to the
present invention, the working principle of the circuit is
completely different from that of the prior art.
[0043] When the circuit of FIG. 1 is powered on, D1 does not work
due to reverse bias, and Q1 is equivalent to an open circuit
because the primary-side control signal is not received. Then the
DC power supply U.sub.DC charges C1 through the first primary-side
winding N.sub.P1. At the same time, the current returns to the
negative terminal of the DC power supply U.sub.DC through the
second primary-side winding N.sub.P2. As shown in FIG. 1-1, the
direction of the current charging C1 is marked by two arrows: the
current flows from the dotted terminal to the undotted terminal in
N.sub.P1; and the current flows from the undotted terminal to the
dotted terminal in N.sub.P2. N.sub.P1 and N.sub.P2 are
bifilar-wound, the two charging currents are equal in magnitude,
and the generated magnetic fluxes are opposite, and are completely
offset. That is, during power-on, the DC power supply U.sub.DC
charges C1 through the two windings of the transformer B. Because
the two windings are offset due to the mutual inductance effect,
and do not work, it is equivalent to that C1 is connected in
parallel to the DC power supply U.sub.DC through DC internal
resistances of N.sub.P1 and N.sub.P2, and C2 stills implements the
function of power supply filtering and decoupling.
[0044] As time passes by, the end voltage of C1 is equal to the
voltage of U.sub.DC, and the left is positive and the right is
negative, as shown in FIG. 1-2.
[0045] When Q1 normally receives the primary-side control signal,
taking one period as an example, when the gate of Q1 is high level,
Q1 is saturated and conducted, and its internal resistance is equal
to the on-state internal resistance R.sub.ds(ON). For the
convenience of analysis, this case is considered as direct
conduction, which is a wire, as shown in FIG. 1-3. D1 is in a
reverse biased state and does not participate in work. In this
case, two excitation currents are generated, as shown by 41 and 42
in FIG. 1-3.
[0046] The current 41: from the positive terminal of the DC power
supply U.sub.DC, enters through the dotted terminal of the first
primary-side winding N.sub.P1 and exits from the undotted terminal
of N.sub.P1, enters through the drain of Q1, and exits through the
source of Q1, and returns to the negative terminal of the DC power
supply U.sub.DC.
[0047] The current 42: from the left positive terminal of the
capacitor C1, enters through the drain of Q1 and exits from the
source of Q1, then enters through the dotted terminal of the second
primary-side winding N.sub.P2, and exits through the undotted
terminal of N.sub.P2, and returns to the right negative terminal of
the capacitor C1.
[0048] For convenience, the negative terminal of the DC power
supply U.sub.DC is referred to as ground. Because the left positive
terminal of C1 is connected to the negative terminal of the DC
power supply U.sub.DC through the saturated and conducted Q1, that
is, grounded, then, the voltage on the right negative terminal of
C1 is approximately -U.sub.DC; in this excitation process, If the
end voltage of C1 has a tendency to decline due to insufficient
capacity, that is, the voltage on the right negative terminal of C1
has a tendency to rise, and its absolute value is less than
U.sub.DC, then, in the excitation process, when Q1 is saturated and
conducted to excite the first primary-side winding N.sub.P1, the
dotted terminal induces a positive voltage, and the undotted
terminal induces a negative voltage, and the size is equal to the
voltages applied to two ends of N.sub.P1, and is equal to U.sub.DC.
In this case, because N.sub.P1 and N.sub.P2 are bifilar-wound, the
two ends of N.sub.P2 also induce: the dotted terminal induces a
positive voltage, and the undotted terminal induces a negative
voltage, and the size is equal to U.sub.DC, and this voltage
directly charges C1. This is a forward process, so that the end
voltage of C1 does not drop due to insufficient capacity. As
mentioned above, the power supply U.sub.DC charges C1 through the
two windings of the transformer B; because the two windings are
offset due to the mutual inductance effect, and do not work, it is
equivalent to that C1 is connected in parallel to the power supply
U.sub.DC through DC internal resistances of N.sub.P1 and N.sub.P2,
and the power supply U.sub.DC directly supplements electric energy
to C1 by using extremely low DC internal resistance, and an end
voltage thereof maintains stable.
[0049] It can be seen that the excitation currents of 41 and 42 are
in parallel relationship. Because N.sub.P1 and N.sub.P2 have the
same inductance and the same excitation voltage, which are both
equal to U.sub.DC, 41 and 42 are completely equal. In the
excitation process, the secondary-side winding N.sub.S also
generates an induced voltage according to a turn ratio. The induced
voltage is: the dotted terminal induces a positive voltage, and the
undotted terminal induces a negative voltage; the size is equal to
U.sub.DC multiplied by the turn ratio n, that is, the N.sub.S
induces a voltage with a positive lower part and a negative upper
part. This voltage is connected in series to the end voltage of C2,
and is applied to two ends of Q2. Q2 is reverse-biased, and is not
conducted. In this case, the secondary side is equivalent to zero
load and has no output.
[0050] In the excitation process, the currents of 41 and 42
increase linearly upward. The current direction is: the currents
flow from the dotted terminal to the undotted terminal in the
inductor.
[0051] To ensure that the electromagnetic compatibility meets the
requirements of use, it is tricky during wiring. Observing 41 and
42 in FIG. 1-3, 41 is in the clockwise current direction, and 42 is
in the counterclockwise direction. If during circuit board wiring,
it is also ensured that the one of the two currents is in the
clockwise direction and the other one is in the counterclockwise
direction, then magnetic fluxes generated during excitation can be
offset when observed from a relatively far place. In this case, the
EMI performance of the flyback switching power supply of the
present invention will be very good.
[0052] When the gate of Q1 changes from high level to low level, Q1
also changes from saturation and conduction to disconnection. The
current in the inductor cannot change abruptly. Even though Q1 is
already disconnected in this case, the currents of 41 and 42 still
flow from the dotted terminal to the undotted terminal. Because the
current loop of the primary side has been cut off, the energy in
the magnetic core flows from the dotted terminal to the undotted
terminal on the secondary side. Referring to FIG. 1-4, the current
flowing from the dotted terminal to the undotted terminal appears
on the secondary-side winding N.sub.S. As shown in 43 in FIG. 1-4,
the initial magnitude of the current=(the sum of 41 and 42 at the
instant of disconnection of Q1)/turn ratio n. Whether the current
exists depends on whether Q2 is in a conducted state synchronously.
Q2 is replaced with a rectifier diode, which is denoted as D2 for
the convenience of description. Then the conduction of D2 is
passive, and the total duration of its conduction is denoted as
T.sub.D. T.sub.D is a variable. According to the law of the
inductance voltage voltage-second balance, when the excitation time
of Q1 changes, T.sub.D is also changing. The total conduction
duration T.sub.D of D2 is a hypothetical variable.
[0053] Because the gate g of Q2 is connected to the secondary-side
control signal, and the secondary-side control signal is a PWM
signal controlled by the voltage between the positive output and
negative output, a plurality of working modes is obtained.
[0054] In a first working mode, Q2 and D2 are synchronously
conducted, and the working duration is equal to T.sub.D. This is
the flyback switching power supply with output synchronous
rectification, and the bandwidth is not changed. This mode is
similar to the conventional synchronous rectification working mode,
but is still different. In the conventional synchronous
rectification, the body diode in the synchronous rectifier tube is
in the same direction as the hypothetical rectifier diode D2, but
in the present invention, referring to FIG. 1, according to the
connection of the technical solution, the body diode of Q2 is in
the direction opposite to the direction of D2.
[0055] In a second working mode, Q2 is conducted in a time shorter
than T.sub.D in the T.sub.D time, and is a switching mode
controlled by a Vout voltage. When the output voltage is lower than
the standard value, after the next period or several periods, the
conduction time of Q2 increases. When the output voltage is higher
than the standard value, after the next period or several periods,
the conduction time of Q2 decreases. That is, in the present
invention, through a special circuit structure, the primary side
implements lossless demagnetization through D1. In this way, it is
possible to make Q1 in an overexcited state. The transformer B
operates as an inductor in the flyback switching power supply. Q1
has sufficient energy stored in transformer B, and the conduction
duty cycle of Q2 can be directly controlled by the secondary-side
output voltage Vout. In this way, secondary side control with
extremely short control delay is implemented. The secondary side
control is almost in real-time mode. When the output voltage is
higher than the standard value, that is, the load becomes lighter,
after the next period or several periods, conduction time of Q2 is
shortened. In this way, the output voltage decreases. When the
output voltage is lower than the standard value, that is, the load
becomes heavier, after the next period or several periods, the
conduction time of Q2 increases, and Q2 releases more energy from
the transformer B to the output end through the N.sub.S, so that
the output voltage rises quickly to reach the standard value.
[0056] That is, the duty cycle of the primary-side control signal
is provided according to m times of the actual output power of the
secondary side. In actual implementation, because the stored energy
is proportional to the square of the excitation current, the
excitation current is linearly proportional to the duty cycle. That
is, when m is 1.1, the actual stored energy is square of 1.1, which
is 1.21. That is, 21% of energy is reserved to deal with the load
abrupt change of the output end. If 100% of the energy is reserved
to deal with the load abrupt change of the output end, that is,
double of the output power, then m only needs to be about 1.41
times, that is, the original duty cycle is 0.1. In this case,
(primary-side excitation energy.times.efficiency)=energy output by
the secondary side in the same period. Then the duty cycle of the
present invention needs to operate according to 0.141. Obviously, m
is a value greater than 1. When m is 2, the square thereof is 4,
the stored energy is 4, and the actual output is only 1, which is
equivalent to reserving 300% energy storage.
[0057] Then, Q2 is conducted in a time shorter than T.sub.D in the
T.sub.D time, and there are also three working modes.
[0058] 1) In a first mode, Q2 and T.sub.D are synchronously
conducted, and the waveform diagram is shown in FIG. 1-5a, that is,
the Q2 switch transistor for output rectification is conducted
synchronously with the hypothetical diode D2, but can be
disconnected in advance. U.sub.gs 1 is the gate control signal of
Q1; i.sub.Q1 is the drain current of Q1, which is also the sum of
the excitation currents of 41 and 42; U.sub.gs 2 is the gate
control signal of Q2; i.sub.Q2 is the drain current of Q2, and is
also the power supply current to the filter capacitor and load; iD1
is the demagnetization current of D1, and the energy is recovered
by U.sub.DC. i.sub.Q2 and i.sub.D1 form a complete demagnetization
current.
[0059] 2) In a second mode, Q2 is conducted in the middle of
T.sub.D, and the waveform diagram is shown in FIG. 1-5b, that is,
the Q2 switch transistor for output rectification is conducted in
the middle of the hypothetical diode D2, is conducted in a delayed
manner, but can be disconnected in advance. U.sub.gs 1 is the gate
control signal of Q1; i.sub.Q1 is the drain current of Q1, which is
also the sum of the excitation currents of 41 and 42; U.sub.gs 2 is
the gate control signal of Q2; i.sub.Q2 is the drain current of Q2,
and is also the power supply current to the filter capacitor and
load; i.sub.D1 is the demagnetization current of D1, is divided
into two segments, and the energy is recovered by U.sub.DC.
i.sub.Q2 and two segments of i.sub.D1 form a complete
demagnetization current.
[0060] 3) In a third mode, Q2 and T.sub.D are synchronously ended,
and the waveform diagram is shown in FIG. 1-5a, that is, the Q2
switch transistor for output rectification is disconnected
synchronously with the hypothetical diode D2. U.sub.gs 1 is the
gate control signal of Q1; i.sub.Q1 is the drain current of Q1,
which is also the sum of the excitation currents of 41 and 42;
U.sub.gs 2 is the gate control signal of Q2; i.sub.Q2 is the drain
current of Q2, and is also the power supply current to the filter
capacitor and load; i.sub.D1 is the demagnetization current of D1,
and the energy is recovered by U.sub.DC. i.sub.Q2 and i.sub.D1 form
a complete demagnetization current. In this mode, because of the
relatively small current of Q2, the output power is small, and this
mode is suitable for a light load mode.
[0061] Certainly, a secondary-side control dedicated IC may be
designed, which is freely switched according to load variation in
the above three modes to achieve optimal control.
[0062] The output end of the conventional flyback switching power
supply obtains energy when the power supply of the primary-side
winding is disconnected, and therefore this name is obtained, and
the output voltage depends on the loop control circuit and is not
related to the turn ratio of the primary side to the secondary side
of the flyback transformer (for example, the transformer in the
series figures such as FIG. 0 and FIG. 1). In the process of energy
transfer, the transformer B does not implement the function of
voltage conversion, but implements the function of the freewheeling
of the magnetic core, and is an isolated version of the Buck-Boost
converter. Therefore, the transformer B is often referred to as a
flyback transformer.
[0063] In the present invention, the output voltage is also
controlled by the secondary-side control signal of Q2, and the
secondary-side control signal is a PWM signal controlled by the
voltage between the positive output and the negative output, which
is actually a very special secondary-side control mode. In most
cases, the conduction of Q2 is less than T.sub.D. Only when the
load current suddenly becomes large, it is possible to have the
completely same working duration as T.sub.D. However, as the load
current continues to stabilize under a large current, the duty
cycle of Q1 of the primary side then increases, and abundant energy
is still supplied to the secondary side, to ensure that energy is
quickly released to the secondary side when the load suddenly
changes again.
[0064] Because the primary-side winding and the secondary-side
winding are generally not possible to be bifilar-wound, there must
be a leakage inductance. The energy stored on the primary-side
winding magnetizing inductance is transmitted to the secondary-side
winding N.sub.S and output end through transformer B after Q1 is
disconnected, but the energy on the leakage inductance is not
transmitted, causing overvoltage on both ends of the Q1 tube and
damaging the Q1 tube. The circuit for demagnetizing the leakage
inductance of the present invention is composed of D1 and the
second primary-side winding N.sub.P2. This circuit simultaneously
demagnetizes Q2 when Q2 is disconnected in advance and is not
conducted in time. The working principle thereof is as follows.
[0065] The first primary-side winding N.sub.P1 and second
primary-side winding N.sub.P2 are bifilar-wound, and the leakage
inductance between the two windings is zero. At the instant of
disconnection of Q1 and thereafter, the energy on the leakage
inductance is not transmitted to secondary side; the current
direction of the electrical energy of the leakage inductance in the
second primary-side winding N.sub.P2 is the same as the direction
during the excitation, and the current flows from the dotted
terminal to the undotted terminal, that is, flows from bottom to
top in FIG. 1-4; D1 is switched on, and the electric energy is
absorbed by the DC power supply U.sub.DC, to form the leakage
inductance demagnetization current shown by 44.
[0066] The electrical energy of the leakage inductance in the first
primary-side winding N.sub.P1 is coupled to the second primary-side
winding N.sub.P2 without leakage inductance. Demagnetization is
implemented by D1, so that the leakage inductance demagnetization
current shown by 44 is also formed. Demagnetization when Q2 is
disconnected in advance and is not conducted in time is similar to
this, and analysis is not made again.
[0067] Obviously, the output voltage V.sub.out is divided by the
turn ratio n to obtain the "reflected voltage" formed on the
primary side when the secondary-side winding N.sub.S is conducted.
For good demagnetization, the reflected voltage cannot be greater
than the value of the DC power supply U.sub.DC. In this way, this
circuit can work well. Because the currents of 41 and 42 are the
same, the first primary-side winding and the second primary-side
winding have the same wire diameter, so that the winding is
convenient. That the wire diameters are the same described herein
includes that they are Litz wires of the same specification, may
have different colors, namely, they are stranded by stranded wires.
To facilitate identification, wire materials of the same
specification including Litz wires may have different colors. As
the working frequency increases, the high frequency current more
tends to flow on the surface of the enameled wire. In this case,
the Litz wire can solve this problem. Certainly, enameled wires of
two different colors are first used to make the Litz wires. Direct
winding is performed, and then the first primary-side winding and
the second primary-side winding are distinguished by color, or the
wire diameters and the number of strands of the two windings are
different, but both achieve the objective of the invention.
[0068] The circuit has an extremely large number of variations, and
it is difficult to completely protect them by claims. For example,
for FIG. 1, because Q2 and N.sub.S are connected in series, when
the positions of them are interchanged, they are completely
operable. That is, the secondary side of FIG. 1 is replaced with
the secondary-side rectifier circuit in FIG. 1-6, and the objective
of the invention is also achieved. The position interchange in the
series circuit is considered as equivalent replacement.
[0069] Referring to FIG. 1-6, another connection relationship of
the secondary-side rectifier circuit is: an undotted terminal of
the secondary-side winding N.sub.S is connected to one end of a
second capacitor C2, to form positive output, which is the +end of
V.sub.out in the figure. A dotted terminal of the secondary-side
winding N.sub.S is connected to the source s of the second switch
transistor Q2, the drain d of the second switch transistor Q2 is
connected to one end of the second capacitor C2, to form negative
output, which is the -end of V.sub.out in the figure. By replacing
the secondary-side rectifier circuit of FIG. 1 with it, the
objective of the invention is also achieved.
[0070] It can be seen that compared with the conventional
three-winding absorption flyback switching power supply, the
present invention has many differences, mainly including: the
"third winding" of the conventional three-winding absorption
flyback switching power supply does not participate in the
excitation, and only participates in demagnetization. In the
present invention, there is no third winding, and both two
primary-side windings participate in the excitation, and in the
demagnetization, the second primary-side winding N.sub.P2
participates in the demagnetization of the leakage inductance,
thereby achieving the lossless absorption of the leakage inductance
energy. Because the lossless absorption of the leakage inductance
energy is realized, the leakage inductance of the primary and
secondary sides is allowed to be large, and the conversion
efficiency of the switching power supply is not affected, so that
high efficiency is achieved, and in the present invention, the
demagnetization winding is the second primary-side winding
N.sub.P2, which also participates in excitation, improves the
current density of the primary-side winding and increases the power
density of the switching power supply. The source of the DC power
supply U.sub.DC can be obtained by electrolytic capacitor filtering
or valley fill circuit filtering after rectification of AC power.
Further, in the present invention, more energy is provided to the
magnetic core of the transformer B, and Q2 of the secondary side is
rectified as needed for output, and the secondary side control is
realized, so that the bandwidth becomes good.
[0071] Therefore, compared with the prior art, the beneficial
effects of the present invention are: leakage inductance between
primary and secondary-side windings is allowed to be relatively
large, and bifilar winding is still used on the primary side, the
conversion efficiency is high, the EMI performance is good, and the
bandwidth is relatively good.
Second Embodiment
[0072] The present invention further provides an equivalent
solution of the first embodiment, corresponding to solution 2.
Referring to FIG. 2, a flyback switching power supply, including a
transformer B, a first switch transistor Q1, a second switch
transistor Q2, where the first switch transistor Q1 and the second
switch transistor Q2 are both N-channel field-effect transistors, a
second capacitor C2, and a first diode D1, where the transformer B
includes a first primary-side winding N.sub.P1, a second
primary-side winding N.sub.P2, and a secondary-side winding
N.sub.S. An undotted terminal of the secondary-side winding N.sub.S
is connected to a drain d of the second switch transistor Q2, and a
source s of the second switch transistor Q2 is connected to one end
of the second capacitor C2, to form positive output, which is the
+end of V.sub.out in the figure. A dotted terminal of the
secondary-side winding N.sub.S is connected to the other end of the
second capacitor C2, to form negative output, which is the -end of
V.sub.out in the figure. A positive terminal+ of an input DC power
supply U.sub.DC is connected to both a drain d of the first switch
transistor Q1 and the undotted terminal of the second primary-side
winding N.sub.P2, and a source s of the first switch transistor Q1
is connected to a dotted terminal of the first primary-side winding
N.sub.P1. The dotted terminal of the second primary-side winding
N.sub.P2 is connected to a cathode of the first diode D1, an
undotted terminal of the first primary-side winding N.sub.P1 is
connected to an anode of the first diode D1, and a connection point
is also connected to a negative terminal- of the input DC power
supply U.sub.DC. A gate g of the first switch transistor Q1 is
connected to a primary-side control signal. The first primary-side
winding N.sub.P1 and the second primary-side winding N.sub.P2 are
bifilar-wound, and a first capacitor C1 is further included; one
end of the first capacitor C1 is connected to the dotted terminal
of the first primary-side winding N.sub.P1, and the other end of
the first capacitor C1 is connected to the dotted terminal of the
second primary-side winding N.sub.P2, and a gate g of the second
switch transistor Q2 is connected to a secondary-side control
signal. The secondary-side control signal is a PWM signal
controlled by a voltage between the positive output and the
negative output.
[0073] In fact, the second embodiment is an equivalent deformation
of the first embodiment: based on FIG. 1 of the first embodiment,
the positions of the devices connected in series of the two
excitation circuits are simultaneously interchanged vertically,
that is, positions of N.sub.P1 and Q1 are interchanged, and
positions of D1 and N.sub.P2 are also interchanged at the same
time, and C1 is still connected between the connection points of
the two devices connected in series, and the circuit of FIG. 2 of
the second embodiment is obtained. Because the source voltage of Q1
is variable, this circuit is floating drive, costs thereof should
be relatively high, and the circuit is not used generally.
[0074] A brief description of its working principle as follows.
[0075] Referring to FIG. 2, when the circuit is powered on, D1 does
not work due to reverse bias, and Q1 does not work because it does
not receive the primary-side control signal, which is equivalent to
an open circuit, then the DC power supply U.sub.DC charges C1
through N.sub.P2. The current simultaneously returns to the
negative terminal of the DC power supply U.sub.DC through N.sub.P1.
Also, during power-on, the DC power supply U.sub.DC charges C1
through the two windings of the transformer B. These two windings
are offset due to the mutual inductance effect, and do not work,
and this is equivalent to that C1 is connected in parallel to the
DC power supply U.sub.DC through DC internal resistances of
N.sub.P2 and N.sub.P1, and C1 still implements the function of
power supply filtering and decoupling.
[0076] As time passes by, the end voltage of C1 is equal to the
voltage of U.sub.DC, and the right is positive and the left is
negative.
[0077] When Q1 is saturated and conducted, the internal resistance
thereof is equal to on-stated internal resistance R.sub.ds (ON),
and is also considered as a wire as stated above. In this case, two
excitation currents are generated.
[0078] The first current: from the positive terminal of the DC
power supply U.sub.DC, enters through the drain of Q1 and exits
from the source of Q1, then enters through the dotted terminal of
the first primary-side winding N.sub.P1, and exits through the
undotted terminal of N.sub.P1, and returns to the negative terminal
of the DC power supply U.sub.DC.
[0079] The second current: from the right positive terminal of the
capacitor C1, enters through the dotted terminal of the second
primary-side winding N.sub.P2 and exits from the undotted terminal
of N.sub.P2, enters through the drain of Q1, and exits through the
source of Q1, and returns to the left negative terminal of the
capacitor C1.
[0080] For convenience, the negative terminal of the DC power
supply U.sub.DC is assumed to be grounded herein, and is referred
to as ground. Because the left negative terminal of C1 is connected
to the positive terminal of the DC power supply U.sub.DC through
the saturated and conducted Q1, then, the voltage of the right
positive terminal of C1 is about 2 U.sub.DC to ground. In this
excitation process, if the end voltage of C1 has insufficient
capacity, that is, the voltage on the right positive terminal of C1
has a tendency to decrease, absolute values of two ends of C1 are
less than U.sub.DC, then, in the excitation process, when Q1 is
saturated and conducted to excite the first primary-side winding
N.sub.P1, the dotted terminal induces a positive voltage, and the
undotted terminal induces a negative voltage, and the size is equal
to the voltages applied to two ends of N.sub.P1, and is equal to
U.sub.DC. In this case, because N.sub.P1 and N.sub.P2 are
bifilar-wound, the two ends of N.sub.P2 also induce: the dotted
terminal induces a positive voltage, and the undotted terminal
induces a negative voltage, and the size is equal to U.sub.DC, and
this voltage directly charges C1. This is a forward process, so
that the end voltage of C1 does not drop due to insufficient
capacity. As mentioned above, the DC power supply U.sub.DC charges
C1 through the two windings of the transformer B; because the two
windings are offset due to the mutual inductance effect, and do not
work, it is equivalent to that C1 is connected in parallel to the
DC power supply U.sub.DC through DC internal resistances of
N.sub.P1 and N.sub.P2, and the DC power supply U.sub.DC directly
supplements electric energy to C1 by using extremely low DC
internal resistance, and an end voltage thereof maintains
stable.
[0081] It can be seen that the first and second excitation currents
are in parallel relationship. Because N.sub.P1 and N.sub.P2 have
the same inductance and the same excitation voltage, which are both
equal to U.sub.DC, the two currents are completely equal. In the
excitation process, the secondary-side winding N.sub.S also
generates an induced voltage according to a turn ratio. The dotted
terminal induces a positive voltage, and the undotted terminal
induces a negative voltage; the size is equal to U.sub.DC
multiplied by the turn ratio n, that is, the N.sub.S induces a
voltage with a positive lower part and a negative upper part. This
voltage is connected in series to the end voltage of C2, and is
applied to two ends of Q2. Q2 is reverse-biased, and is not
conducted. In this case, the secondary side is equivalent to zero
load and has no output.
[0082] In the excitation process, the first and second excitation
currents increase linearly upward. The current direction is: the
currents flow from the dotted terminal to the undotted terminal in
the inductor.
[0083] When Q1 is disconnected, the current in the inductor cannot
abruptly change. The energy in the magnetic core flows from the
dotted terminal to the undotted terminal on the secondary side, and
a current flowing from the dotted terminal to the undotted terminal
appears on the secondary-side winding N.sub.S, and the current
charges the capacitor C2 by using Q2 conducted in some time, and
V.sub.out establishes a voltage or continuously outputs energy, and
this process is also a partial process of demagnetization. When Q2
is not conducted, the demagnetization current implements
demagnetization through D1. The working process thereof is the same
as FIG. 1-5a, FIG. 1-5b, and FIG. 1-5c.
[0084] The second embodiment is a deformation of the first
embodiment, and the working principle is equivalent, and the
objective of the invention is also achieved. The technical solution
of using the N-channel field-effect transistor may also be realized
by using a P-channel field-effect transistor. The P-channel
field-effect transistor has a relatively low cost at a low working
voltage. In this case, based on the first embodiment, polarities of
the power supply, the diode, and the dotted terminal need to be
reversed, and the polarity of the output rectifier part does not
need to be reversed, and then a third embodiment obtained as
follows.
[0085] Herein, implementing the secondary-side rectifier circuit by
using a P-channel field-effect transistor is first presented.
Referring to FIG. 2-1, a secondary-side rectifier circuit: a second
switch transistor Q2 is a P-channel field-effect transistor, an
undotted terminal of a secondary-side winding N.sub.S is connected
to a source s of the second switch transistor Q2, a drain d of the
second switch transistor Q2 is connected to one end of the second
capacitor C2, to form positive output, which is the +end of Vout in
the figure, a dotted terminal of the secondary-side winding N.sub.S
is connected to the other end of the second capacitor C2, to form
negative output, which is the -end of Vout in the figure.
[0086] For example, for FIG. 2-1, because Q2 and N.sub.S are
connected in series, when the positions of them are interchanged,
they are completely operable. The position interchange in the
series circuit is considered as equivalent replacement to obtain
the following secondary-side rectifier circuit.
[0087] Referring to FIG. 2-2, another connection relationship of
the secondary-side rectifier circuit is: a secondary-side rectifier
circuit: the second switch transistor Q2 is a P-channel
field-effect transistor; an undotted terminal of the secondary-side
winding N.sub.S is connected to one end of a second capacitor C2,
to form positive output, which is the +end of Vout in the figure. A
dotted terminal of the secondary-side winding N.sub.S is connected
to the drain d of the second switch transistor Q2, the source s of
the second switch transistor Q2 is connected to one end of the
second capacitor C2, to form negative output, which is the -end of
V.sub.out in the figure. By replacing the secondary-side rectifier
circuit of FIG. 1 and FIG. 2 with it, the objective of the
invention is also achieved.
Third Embodiment
[0088] Referring to FIG. 3, also the foregoing solution 3, a
flyback switching power supply, including a transformer B, a first
switch transistor Q1, a second switch transistor Q2, where the
first switch transistor Q1 is a P-channel field-effect transistor,
and the second switch transistor Q2 is an N-channel field-effect
transistor, a second capacitor C2, and a first diode D1, where the
transformer B includes a first primary-side winding N.sub.P1, a
second primary-side winding N.sub.P2, and a secondary-side winding
N.sub.S. An undotted terminal of the secondary-side winding N.sub.S
is connected to a drain d of the second switch transistor Q2, and a
source s of the second switch transistor Q2 is connected to one end
of the second capacitor C2, to form positive output, which is the
+end of Vout in the figure. A dotted terminal of the secondary-side
winding N.sub.S is connected to the other end of the second
capacitor C2, to form negative output, which is the -end of Vout in
the figure. A negative terminal- of an input DC power supply
U.sub.DC is connected to both an undotted terminal of the first
primary-side winding N.sub.P1 and an anode of the first diode D1,
and a dotted terminal of the first primary-side winding N.sub.P1 is
connected to a drain d of the first switch transistor Q1. A cathode
of the first diode D1 is connected to the dotted terminal of the
second primary-side winding N.sub.P2, a source s of the first
switch transistor Q1 is connected to the undotted terminal of the
second primary-side winding N.sub.P2, and a connection point is
also connected to a positive terminal+ of the input DC power supply
U.sub.DC. A gate g of the first switch transistor Q1 is connected
to a primary-side control signal. The first primary-side winding
N.sub.P1 and the second primary-side winding N.sub.P2 are
bifilar-wound, and a first capacitor C1 is further included; one
end of the first capacitor C1 is connected to the dotted terminal
of the first primary-side winding N.sub.P1, and the other end of
the first capacitor C1 is connected to the dotted terminal of the
second primary-side winding N.sub.P2, and a gate g of the second
switch transistor Q2 is connected to a secondary-side control
signal. The secondary-side control signal is a PWM signal
controlled by a voltage between the positive output and the
negative output.
[0089] Upon comparison between FIG. 1 and FIG. 3, it can be seen
that the third embodiment is obtained by reversing the polarities
of the DC power supply U.sub.DC, the diode D1, the dotted terminals
of the first primary-side winding N.sub.P1 and the second
primary-side winding N.sub.P2, and replacing the N transistor with
a P transistor as Q1 of the first embodiment. It should be noted
that the secondary-side rectifier circuit of the output rectifier
part does not need to be changed and remains as it is. Certainly,
if the secondary-side rectifier circuit uses the solutions of FIG.
1-6, FIG. 2-1, and FIG. 2-2, the objective of invention can be
achieved. In FIG. 3, the positive terminal of the input DC power
supply U.sub.DC is the ground, and belongs to a switching power
supply of negative power supply work, and the P-channel
field-effect transistor itself is also driven by the negative
level, and this is exactly suitable.
[0090] Therefore, the working principle thereof is the same as that
of the first embodiment, and details are not described herein
again, and the objective of the invention is also achieved.
Fourth Embodiment
[0091] The present invention further provides an equivalent
solution of the third embodiment. Referring to FIG. 4, FIG. 4 shows
a technical solution of using a P-channel field-effect transistor
as Q1 of solution 2. Based on the solution 2, polarities of the
power supply, the diode, and the dotted terminal need to be
reversed, and the polarity of an output rectifier portion does not
need to be reserved. Then solution 4 is obtained: a flyback
switching power supply, including a transformer B, a first switch
transistor Q1, and a second switch transistor Q2, where the first
switch transistor Q1 is a P-channel field-effect transistor, and
the second switch transistor Q2 is an N-channel field-effect
transistor, a second capacitor C2, and a first diode D1, where the
transformer B includes a first primary-side winding N.sub.P1, a
second primary-side winding N.sub.P2, and a secondary-side winding
N.sub.S. An undotted terminal of the secondary-side winding N.sub.S
is connected to a drain d of the second switch transistor Q2, and a
source s of the second switch transistor Q2 is connected to one end
of the second capacitor C2, to form positive output, which is the
+end of Vout in the figure. A dotted terminal of the secondary-side
winding N.sub.S is connected to the other end of the second
capacitor C2, to form negative output, which is the -end of
V.sub.out in the figure. A negative terminal- of an input DC power
supply U.sub.DC is connected to both a drain d of the first switch
transistor Q1 and the dotted terminal of the second primary-side
winding N.sub.P2, and a source s of the first switch transistor Q1
is connected to an undotted terminal of the first primary-side
winding N.sub.P1. The undotted terminal of the second primary-side
winding N.sub.P2 is connected to an anode of the first diode D1, a
dotted terminal of the first primary-side winding N.sub.P1 is
connected to a cathode of the first diode D1, and a connection
point is also connected to a positive terminal+ of the input DC
power supply U.sub.DC. A gate g of the first switch transistor Q1
is connected to a primary-side control signal. The first
primary-side winding N.sub.P1 and the second primary-side winding
N.sub.P2 are bifilar-wound, and a first capacitor C1 is further
included; one end of the first capacitor C1 is connected to the
undotted terminal of the first primary-side winding N.sub.P1, and
the other end of the first capacitor C1 is connected to the
undotted terminal of the second primary-side winding N.sub.P2, and
a gate g of the second switch transistor Q2 is connected to a
secondary-side control signal. The secondary-side control signal is
a PWM signal controlled by a voltage between the positive output
and the negative output.
[0092] The fourth embodiment of FIG. 4 is a deformation of the
third embodiment: based on FIG. 3 of the third embodiment, the
positions of the devices connected in series of the two excitation
circuits are interchanged, that is, positions of N.sub.P1 and Q1
are interchanged, and positions of D1 and N.sub.P2 are also
interchanged at the same time, and C1 is still connected between
the two primary-side windings N.sub.P1 and N.sub.P2 connected in
series, and the circuit of FIG. 4 of the fourth embodiment is
obtained. Because the source voltage of Q1 is variable, this
circuit is floating drive, costs thereof should be relatively high,
and the circuit is not used generally.
[0093] Upon comparison between FIG. 2 and FIG. 4, it can be seen
that the fourth embodiment is obtained by reversing the polarities
of the DC power supply U.sub.DC, the diode D1, the dotted terminals
of the first primary-side winding N.sub.P1 and the second
primary-side winding N.sub.P2, and replacing the N transistor with
a P transistor as Q1 of the second embodiment in FIG. 2. It should
be noted that the secondary-side rectifier circuit of the output
rectifier part does not need to be changed and remains as it is.
Certainly, if the secondary-side rectifier circuit uses the
solutions of FIG. 1-6, FIG. 2-1, and FIG. 2-2, the objective of
invention can be achieved. In FIG. 4, the positive terminal of the
input DC power supply U.sub.DC is the ground, and also belongs to a
switching power supply of negative power supply work, and the
P-channel field-effect transistor itself is also driven by the
negative level, and this is exactly suitable.
[0094] Therefore, the working principle thereof is the same as that
of the second embodiment, and details are not described herein
again, and the objective of the invention is also achieved.
[0095] The present invention also has a great advantage. When there
are multiple paths of outputs on the secondary side, since each
path independently controls its rectifier tubes Q2a, Q2b, Q2c, etc.
by using its output voltage, the voltage adjustment rate of each
output does not affect each other, to achieve high-precision output
voltage and good bandwidth of each path.
[0096] It should be noted that the primary-side circuit of the
present invention includes four cases of FIG. 1, FIG. 2, FIG. 3 and
FIG. 4. The secondary-side rectifier circuit includes four cases of
FIG. 1, FIG. 2, FIG. 2-1 and FIG. 2-2. Any one of the primary-side
circuits and any one of the secondary-side rectifier circuits may
be arbitrarily selected for combination, and all can achieve the
objective of the invention. The embodiments that are not mentioned
above are not described in detail one by one in this
specification.
[0097] The foregoing descriptions are only preferred embodiments of
the present invention, and it should be noted that the above
preferred embodiments should not be construed as limiting the
present invention. For a person of ordinary skill in the art,
without departing from the spirit and scope of the present
invention, a number of improvements and modifications, such as
adding a control loop to achieve voltage regulation of the output
may also be made, this is obviously obtained through the prior art.
The improvements and modifications, such as using a switch
transistor Q1 of another symbol, adding multipath outputs to
secondary-side outputs, and using .pi.-type filtering for
filtering, should also be considered as protection scope of the
present invention. The embodiments are not described herein, and
the protection scope of the present invention should be subject to
the appended claims.
* * * * *