U.S. patent application number 16/278394 was filed with the patent office on 2020-03-12 for interleaved llc half-bridge series resonant converter having integrated transformer.
The applicant listed for this patent is JING-YUAN LIN. Invention is credited to HUANG-JEN CHIU, ZONG-SIAN JIANG, SIH-YI LEE, JING-YUAN LIN.
Application Number | 20200083818 16/278394 |
Document ID | / |
Family ID | 68316356 |
Filed Date | 2020-03-12 |
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United States Patent
Application |
20200083818 |
Kind Code |
A1 |
LIN; JING-YUAN ; et
al. |
March 12, 2020 |
INTERLEAVED LLC HALF-BRIDGE SERIES RESONANT CONVERTER HAVING
INTEGRATED TRANSFORMER
Abstract
An interleaved LLC half-bridge series resonant converter having
an integrated transformer includes a power supply, a magnetic core,
a first converter, a second converter and an output load circuit.
The magnetic core has first and second outer columns and a center
column. The first converter includes a first switch circuit, a
first resonant tank, a first transformer, and a first rectifier
circuit. The first transformer is coupled to the first resonant
tank and includes a first primary winding wound on the first outer
column and a first secondary winding wound on the second outer
column. The second converter includes a second switch circuit, a
second resonant tank, a second transformer and a second rectifier
circuit. The second transformer is coupled to the second resonant
tank and includes a second primary winding wound on the first outer
column and a second secondary winding wound on the second outer
column.
Inventors: |
LIN; JING-YUAN; (New Taipei
City, TW) ; JIANG; ZONG-SIAN; (Yunlin County, TW)
; LEE; SIH-YI; (Taipei City, TW) ; CHIU;
HUANG-JEN; (New Taipei City, TW) |
|
Applicant: |
Name |
City |
State |
Country |
Type |
LIN; JING-YUAN |
New Taipei City |
|
TW |
|
|
Family ID: |
68316356 |
Appl. No.: |
16/278394 |
Filed: |
February 18, 2019 |
Current U.S.
Class: |
1/1 |
Current CPC
Class: |
H02M 3/33592 20130101;
H02M 2001/0058 20130101; H01F 30/10 20130101; H02M 2001/0064
20130101; H02M 3/285 20130101; H02M 1/4233 20130101; H02M 3/33576
20130101; H01F 27/00 20130101; H01F 27/40 20130101; H01F 3/14
20130101 |
International
Class: |
H02M 3/335 20060101
H02M003/335; H02M 1/42 20060101 H02M001/42 |
Foreign Application Data
Date |
Code |
Application Number |
Sep 12, 2018 |
TW |
107132062 |
Claims
1. An interleaved LLC half-bridge series resonant converter having
an integrated transformer, comprising: a power supply; a magnetic
core, having a first outer column, a center column and a second
outer column; a first converter, including: a first switch circuit,
configured to control a first input voltage and a first input
current from the power supply; a first resonant tank coupled to the
first switch circuit, including a first resonant inductor, a first
resonant capacitor and a first magnetizing inductor; a first
transformer coupled to the first resonant tank, including: a first
primary winding wound on the first outer column; and a first
secondary winding wound on the second outer column; and a first
rectifier circuit configured to receive and rectify an output
voltage and an output current of the first transformer; a second
converter, including: a second switch circuit, configured to
control a second input voltage and a second input current from the
power supply; a second resonant tank coupled to the second switch
circuit, including a second resonant inductor, a second resonant
capacitor and a second magnetizing inductor; a second transformer
coupled to the second resonant tank, including: a second primary
winding wound on the first outer column; and a second secondary
winding wound on the second outer column; and a second rectifier
circuit configured to receive and rectify an output voltage and an
output current of the second transformer; and an output load
circuit respectively coupled to the first rectifier circuit and the
second rectifier circuit, having an output capacitor and a load,
wherein a ratio of numbers of coils of the first primary winding to
the first secondary winding is determined based on a first gain
value of the first resonant tank and a ratio of the first input
voltage to an output voltage of the output load circuit, a ratio of
numbers of coils of the second primary winding to the second
secondary winding is determined based on a second gain value of the
second resonant tank and a ratio of the second input voltage to the
output voltage of the output load circuit, and the first gain value
and the second gain value are approximately 1.
2. The interleaved LLC half-bridge series resonant converter
according to claim 1, wherein the first switch circuit includes a
first upper bridge switch and a first lower bridge switch, the
second switching circuit includes a second upper bridge switch and
a second lower bridge switch, the first rectifier circuit includes
a first rectifier switch and a second rectifier switch, and the
second rectifier circuit includes a third rectifier switch and a
four rectifier switch.
3. The interleaved LLC half-bridge series resonant converter
according to claim 2, further comprising a control circuit
configured to respectively control the first switching circuit, the
second switching circuit, the first rectifier circuit and the
second rectifier circuit to be switched between multiple switching
states.
4. The interleaved LLC half-bridge series resonant converter
according to claim 3, wherein in a first phase, the first upper
bridge switch is turned on, the first lower bridge switch is turned
off, the second upper bridge switch is turned off, and the second
lower bridge switch is turned on; in a second phase after the first
phase, the second lower bridge switch is turned off; in the third
phase after the second phase, the second upper bridge switch is
turned on; and in a fourth phase after the third phase, the first
upper bridge switch is turned off.
5. The interleaved LLC half-bridge series resonant converter
according to claim 4, wherein in the first phase, the first
rectifier switch is turned on, the second rectifier switch is
turned off, the third rectifier switch is turned off, and the
fourth rectifier switch is switched from ON state to OFF state; in
the second phase after the first phase, the third rectifier switch
is switched from ON state to OFF state; in the third phase after
the second phase, the first rectifier switch is switched from ON
state to OFF state; and in the fourth phase after the third phase,
the second rectifier switch is switched from OFF state to ON
state.
6. The interleaved LLC half-bridge series resonant converter
according to claim 4, wherein in a fifth phase, a sixth phase, a
seventh phase and an eighth phase after the fourth phase, switching
states of the first rectifier circuit and the second rectifier
circuit are opposite to those in the first phase, the second phase,
the third phase and the fourth phase, respectively, in the fifth
phase, the sixth phase and the seventh phase, the switching states
of the first switching circuit are opposite to those in the first
phase, the second stage and the third stage, respectively, and in
the fifth phase, the seventh phase and the eighth phase, the
switching states of the second switch circuit are opposite to those
in the first phase, the third stage and the fourth phase,
respectively.
7. The interleaved LLC half-bridge series resonant converter
according to claim 1, wherein the first outer column and the second
outer column are respectively divided into a first magnetic column
portion and a second magnetic column portion by a first air gap and
a second air gap, respectively.
8. The interleaved LLC half-bridge series resonant converter
according to claim 7, wherein the first primary winding is wound on
the first outer column of the first magnetic column portion, and
the first secondary winding is wound on the second outer column of
the first magnetic column portion; and wherein the second primary
winding is wound on the first outer column of the second magnetic
column portion, and the second secondary winding is wound on the
second outer column of the second magnetic column portion.
9. The interleaved LLC half-bridge series resonant converter
according to claim 7, wherein widths of the first air gap and the
second air gap depend on inductance values of the first magnetizing
inductor and the second magnetizing inductor, respectively, and the
inductance values satisfy soft-switching conditions in which the
first switch circuit and the second switch circuit are
zero-voltage-switched to ON state, and the soft-switching
conditions are designed to be related to operation frequencies,
dead-zone time and switching parasitic capacitances.
10. (canceled)
Description
CROSS-REFERENCE TO RELATED PATENT APPLICATION
[0001] This application claims the benefit of priority to Taiwan
Patent Application No. 107132062, filed on Sep. 12, 2018. The
entire content of the above identified application is incorporated
herein by reference.
[0002] Some references, which may include patents, patent
applications and various publications, may be cited and discussed
in the description of this disclosure. The citation and/or
discussion of such references is provided merely to clarify the
description of the present disclosure and is not an admission that
any such reference is "prior art" to the disclosure described
herein. All references cited and discussed in this specification
are incorporated herein by reference in their entireties and to the
same extent as if each reference was individually incorporated by
reference.
FIELD OF THE DISCLOSURE
[0003] The present disclosure relates to an LLC half-bridge series
resonant converter, and more particularly to an interleaved LLC
half-bridge series resonant converter having an integrated
transformer.
BACKGROUND OF THE DISCLOSURE
[0004] In recent years, with an increasing shortage of energy, the
growth of environmental awareness, the rapid development of science
and technology, the population explosion, and a growing demand for
electricity, it has become an urgent issue to search for
alternative energy and to effectively save energy. With the
continuous progress in semiconductor manufacturing process,
switching power supply has been widely used in all kinds of
electronic products, and most electronic products are developed for
miniaturization and high power density to meet market demand. For
these purposes, the switching frequency of switching power supply
should be increased so as to reduce the volume of magnetic
components.
[0005] Since the switching frequency is increased, the switching
loss during the switching of the power switch assembly between ON
and OFF states is increased. When an LLC Series Resonant Converter
(LLC-SRC) is operated in a low voltage-high current output mode,
since the output current of the secondary side is semi-sinusoidal
current, an excessive ripple current is generated at the output
terminal as the output current rises.
[0006] Therefore, to overcome the above defects structure, it has
become an important issue in the art to improve the converter
structure.
SUMMARY OF THE DISCLOSURE
[0007] In response to the above-referenced technical inadequacies,
the present disclosure provides an interleaved LLC half-bridge
series resonant converter with an integrated transformer.
[0008] In one aspect, the present disclosure provides an
interleaved LLC half-bridge series resonant converter with an
integrated transformer, which includes a power supply, a magnetic
core, a first converter, a second converter and an output load
circuit. The magnetic core has a first outer column, a center
column and a second outer column. The first converter includes a
first switch circuit, a first resonant tank, a first transformer,
and a first rectifier circuit. The first switch circuit is
configured to control a first input voltage and a first input
current from the power supply. The first resonant tank is coupled
to the first switch circuit, which includes a first resonant
inductor, a first resonant capacitor and a first magnetizing
inductor. The first transformer is coupled to the first resonant
tank, which includes a first primary winding wound on the first
outer column and a first secondary winding wound on the second
outer column. The first rectifier circuit is configured to receive
and rectify an output voltage and an output current of the first
transformer. The second converter includes a second switch circuit,
a second resonant tank, a second transformer and a second rectifier
circuit. The second switch circuit is configured to control a
second input voltage and a second input current from the power
supply. The second resonant tank is coupled to the second switch
circuit, which includes a second resonant inductor, a second
resonant capacitor and a second magnetizing inductor. The second
transformer is coupled to the second resonant tank, which includes
a second primary winding wound on the first outer column and a
second secondary winding wound on the second outer column. The
second rectifier circuit is configured to receive and rectify an
output voltage and an output current of the second transformer. The
output load circuit is respectively coupled to the first rectifier
circuit and the second rectifier circuit, which has an output
capacitor and a load.
[0009] One of the advantages of the present disclosure is that the
interleaved LLC half-bridge series resonant converter having an
integrated transformer provided by the present disclosure can
reduce output current ripple and improve efficiency by utilizing
two series-coupled LLC-SRCs combined with a mechanism of phase
shift of 90.degree..
[0010] One of the advantages of the present disclosure is that the
interleaved LLC half-bridge series resonant converter having an
integrated transformer provided by the present disclosure can
reduce errors of two transformers to achieve current-sharing for
the secondary currents by utilizing a technique of magnetic
integration to integrate two transformers into one magnetic core,
and by replacing the conventional winding frame-wound transformer
with a plate transformer.
[0011] In order to further understand the characteristics and
technical contents of the present disclosure, the following
detailed descriptions and drawings related to the present
disclosure are provided. However, drawings are provided for the
purpose of illustration and explanation, and are not intended to
limit the present disclosure.
[0012] These and other aspects of the present disclosure will
become apparent from the following description of the embodiment
taken in conjunction with the following drawings and their
captions, although variations and modifications therein may be
affected without departing from the spirit and scope of the novel
concepts of the disclosure.
BRIEF DESCRIPTION OF THE DRAWINGS
[0013] The present disclosure will become more fully understood
from the following detailed description and accompanying
drawings.
[0014] FIG. 1 is a circuit layout of an interleaved LLC half-bridge
series resonant converter having an integrated transformer of an
embodiment of the present disclosure.
[0015] FIG. 2 is a schematic diagram of a magnetic core, a primary
winding, and a secondary winding of an integrated transformer of an
embodiment of the present disclosure.
[0016] FIG. 3 is a timing chart of an interleaved LLC half-bridge
series resonant converter of an embodiment of the present
disclosure.
[0017] FIG. 4A to FIG. 4H are schematic diagrams showing current
paths of a first phase through an eighth phase of interleaved LLC
half-bridge series resonant converter of an embodiment of the
present disclosure.
[0018] FIG. 5A is a graph showing a comparison of output current
ripples of different structures according to an embodiment of the
present disclosure.
[0019] FIG. 5B is a graph showing a comparison of output current
ripples of a current sharing group and a non-uniform current group
according to an embodiment of the present disclosure.
[0020] FIG. 6 is a schematic diagram of an integrated transformer
core according to an embodiment of the present disclosure.
[0021] FIG. 7A and FIG. 7B are equivalent magnetic circuit diagrams
of an integrated transformer of an embodiment of the present
disclosure.
[0022] FIG. 8 is a timing chart showing operations of the magnetic
flux of the magnetic core of the integrated transformer according
to an embodiment of the present disclosure.
[0023] FIGS. 9A to 9D are magnetic flux path diagrams of the
magnetic core of the integrated transformer according to embodiment
of the present disclosure.
[0024] FIG. 10 is a graph showing a comparison for magnetic flux of
each of columns of the magnetic core of an integrated transformer
of an embodiment of the present disclosure.
[0025] FIG. 11 is a schematic diagram showing magnetic reluctance
blocks of an integrated transformer of an embodiment of the present
disclosure.
DETAILED DESCRIPTION OF THE EXEMPLARY EMBODIMENTS
[0026] The present disclosure is more particularly described in the
following examples that are intended as illustrative only since
numerous modifications and variations therein will be apparent to
those skilled in the art. Like numbers in the drawings indicate
like components throughout the views. As used in the description
herein and throughout the claims that follow, unless the context
clearly dictates otherwise, the meaning of "a", "an", and "the"
includes plural reference, and the meaning of "in" includes "in"
and "on". Titles or subtitles can be used herein for the
convenience of a reader, which shall have no influence on the scope
of the present disclosure.
[0027] The terms used herein generally have their ordinary meanings
in the art. In the case of conflict, the present document,
including any definitions given herein, will prevail. The same
thing can be expressed in more than one way. Alternative language
and synonyms can be used for any term(s) discussed herein, and no
special significance is to be placed upon whether a term is
elaborated or discussed herein. A recital of one or more synonyms
does not exclude the use of other synonyms. The use of examples
anywhere in this specification including examples of any terms is
illustrative only, and in no way limits the scope and meaning of
the present disclosure or of any exemplified term. Likewise, the
present disclosure is not limited to various embodiments given
herein. Numbering terms such as "first", "second" or "third" can be
used to describe various components, signals or the like, which are
for distinguishing one component/signal from another one only, and
are not intended to, nor should be construed to impose any
substantive limitations on the components, signals or the like.
[0028] "INTERLEAVED LLC HALF-BRIDGE SERIES RESONANT CONVERTER
HAVING INTEGRATED TRANSFORMER" of the present disclosure is
described as follows. Those skilled in the art can understand
advantages and effects of the present disclosure from the contents
disclosed in the specification. The present disclosure can be
implemented or applied in various other specific embodiments, and
various modifications and changes can be made without departing
from the spirit and scope of the present disclosure. In addition,
the drawings of the present disclosure are merely illustrative and
are not intended to be in the actual size. The following
embodiments will further explain the related technical content of
the present disclosure, but is not intended to limit the scope of
the present disclosure.
[0029] It should be understood that, although the terms "first",
"second", "third", and the like may be used herein to describe
various elements or signals, however, these elements or signals are
not limited by these terms. These terms are primarily used to
distinguish one element from another or one signal from another. In
addition, the term "or" as used herein may include a combination of
any one or more of the associated listed items, depending on the
actual situation.
[0030] Reference is now made to FIG. 1. An interleaved LLC
half-bridge series resonant converter 1 having an integrated
transformer is provided according to an embodiment of the present
disclosure, and includes a power source Vin, a first converter 11,
a second converter 12, and an output load circuit 13 and a magnetic
core 14.
[0031] The first converter 11 includes a first switch circuit 110,
a first resonant tank 112, a first transformer 114 and a first
rectifier circuit 116. The first switch circuit 110 is configured
to control a first input voltage and a first input current from the
power source Vin.
[0032] The first resonant tank 112 is coupled to the first switch
circuit 110 and includes a first resonant inductor Lr1, a first
resonant capacitor Cr1, and a first magnetizing inductor Lm1. The
first transformer 114 is coupled to the first resonant tank 112,
and includes a first primary side winding L1 wound on a first outer
column OP1 and a first secondary side winding L21 wound on a second
outer column OP2. The first rectifier circuit 116 is used to
receive and rectify an output voltage and output current of the
first transformer 114.
[0033] The second converter 12 includes a second switch circuit
120, a second resonant tank 122, a second transformer 124, and a
second rectifier circuit 126. The second switching circuit 120 is
configured to control a second input voltage and a second input
current from the power source Vin. The second resonant tank 122 is
coupled to the second switch circuit 120, and includes a second
resonant inductor Lr2, a second resonant capacitor Cr2, and a
second magnetizing inductor Lm2. The second transformer 124 is
coupled to the second resonant tank 122, and includes a second
primary side winding L2 wound on the first outer column OP1 and a
second secondary winding L22 wound on the second outer column OP2.
The second rectifier circuit 126 is used to receive and rectify an
output voltage and output current of the second transformer
124.
[0034] The output load circuit 13 is coupled to the first rectifier
circuit 116 and the second rectifier circuit 126, respectively,
having an output capacitor Co and a load RL. The output capacitor
Co is used for filtering and the load RL acts as a load impedance
of an output end.
[0035] The present disclosure utilizes two LLC half-bridge series
resonant converters in parallel, as shown in FIG. 1. In detail, the
half-bridge series resonant circuit operates in a series resonant
converter (SRC) resonance mode and an LLC (LLC-Type Series Resonant
Converter) resonance mode, respectively. Taking the first
transformer 11 as an example, the first magnetizing inductor Lm1
determines whether to participate in resonance according to
different switching operation regions. In the SRC mode, the first
magnetizing inductor Lm1 does not participate in resonance, the
first resonant frequency is determined by the first resonant
inductor Lr1 and the first resonant capacitor Cr1. Since the first
resonant inductor Lr, the first resonant capacitor Cr1 of the
resonant circuit, and the load are in series, the maximum voltage
gain for the SRC resonant mode is generated when a switching
frequency fsw operates with a first resonant frequency FR1, as
shown in the following formula (1):
fsw = fr 1 = 1 2 .pi. LrCr formula ( 1 ) ##EQU00001##
[0036] In the LLC mode, the first magnetizing inductor Lm1
participates in resonance, and forms a resonant network with the
first resonant inductor Lr1 and the first resonant capacitor Cr1.
The first magnetizing inductor Lm1 merges with the first resonant
inductor Lr1, and then produces a second resonant frequency fr2
with the first resonant capacitor Cr1, as shown in the following
formula (2):
fr 2 = 1 2 .pi. ( Lr + Lm ) Cr formula ( 2 ) ##EQU00002##
[0037] Two resonant frequency points can divide three intervals on
the frequency response curve, that is, the two resonant frequency
points are the first resonant frequency fr1 and the second resonant
frequency fr2. The biggest difference between the operations in LLC
mode and in the SRC mode is that the voltage gain of the LLC mode
is greater than 1. In addition, zero voltage switching conditions
of a power transistor in this operation mode is only related to the
first magnetizing inductor Lm1, but not related to the output
current. In other words, the zero voltage switching conditions can
be satisfied as long as the current on the magnetizing inductor is
large enough.
[0038] On the other hand, when the converter switches to the
switching frequency fsw to be operated in the LLC mode, an
operating frequency is smaller than the first resonant frequency
fr1 and greater than the second resonant frequency fr2. The
resonant current is decreased to be equal to a magnetizing current
before the switch is cutoff. When the resonant current is equal to
the magnetizing current, the current does not flow into the primary
side of the first transformer 114, and no energy is transferred to
the load end. The output rectifier switch reaches zero-current
cutoff since there is no current flowing through, and thus the
output voltage cannot clamp the first magnetizing inductor Lm1 with
the first transformer 114 via the output rectifier switch. In this
region, the resonant element includes the first resonant inductor
Lr1, the first resonant capacitor Cr1, and the first magnetizing
inductor Lm1 and enters a second resonant mode. Preferably, the two
LLC half-bridge series resonant converters of the present
disclosure operate in the second region (i.e.,
fr1<fs<fr2).
[0039] In this case, as shown in FIG. 1, the first switch circuit
110 includes a first upper bridge switch Q1 and a first lower
bridge switch Q2, the second switch circuit 120 includes a second
upper bridge switch Q3 and a second lower bridge switch Q4, and
diodes D1, D2, D3 and D4 and capacitors Coss1, Coss2, Coss3 and
Coss4 are respectively body diodes and parasitic capacitors of the
first upper bridge switch Q1, the first lower bridge switch Q2, the
second upper bridge switch Q3 and the second lower bridge switch
Q4. On the other hand, the first rectifier circuit 116 includes a
first rectifier switch Q5 and a second rectifier switch Q6, the
second rectifier circuit 126 includes a third rectifier switch Q7
and a fourth rectifier switch Q8, and diodes D5, D6, D7 and D8 are
respectively body diodes of the first rectifier switch Q5, the
second rectifier switch Q6, the third rectifier switch Q7 and the
fourth rectifier switch Q8. In the following, the rectifier
switches are presented as diodes, such as the diodes D5, D6, D7 and
D8, and the conduction voltage-dropping and resistance effects are
ignored.
[0040] Reference is now made to FIG. 2, which is a schematic
diagram of a magnetic core, a primary winding, and a secondary
winding of the integrated transformer of an embodiment of the
present disclosure. As shown in FIG. 2, the magnetic core 14 has
the first outer column OP1, a center column CP and the second outer
column OP2. In order to prevent the first transformer 114 and the
second transformer 124 from interfering with each other, the
primary winding and the secondary winding of a center tap
transformer are wound on the same magnetic column.
[0041] The first transformer 114 includes the first primary side
winding L1 wound on the first outer column OP1 and the first
secondary winding L21 wound on the second outer column OP2. The
second transformer 124 includes the second primary side winding L2
wound on the outer column OP1 and the second secondary side winding
L22 wound on the second outer side column OP2. The first lateral
column OP1 and the second lateral column OP2 are divided into a
first magnetic column portion MP1 and a second magnetic column
portion MP2 through a first air gap GP1 and a second air gap GP2,
respectively. The first primary side winding L1 is wound on the
first outer column OP1 of the first magnetic column portion MP1,
and the first secondary side winding L21 is wound on the second
outer column OP2 of the first magnetic column portion MP1, the
second primary winding L2 is wound on the first outer column OP1 of
the second magnetic column portion MP2, and the second secondary
side winding L22 is wound on the second outer column OP2 of the
second magnetic column portion MP2.
[0042] The first secondary side winding L21 further includes a
positive half-cycle winding L211 and a negative half-cycle winding
L212, and the second secondary side winding L22 further includes a
positive half cycle winding L221 and a negative half-cycle winding
L222.
[0043] Reference is now made to FIG. 3 and FIGS. 4A to 4H. FIG. 3
is a timing chart of an interleaved LLC half-bridge series resonant
converter of an embodiment of the present disclosure, and FIGS. 4A
to 4H are schematic diagrams showing current paths of a first phase
through an eighth phase of interleaved LLC half-bridge series
resonant converter of an embodiment of the present disclosure. As
shown in FIG. 3, actions in a cycle can be divided into 16 phases,
of which the t0 to t8 phases and the t8 to t16 phases are
pair-states. Therefore, only the t0-t8 phases are described
hereinafter. For the sake of brevity, the synchronous rectifier
switches on the secondary side are presented as diodes, and the
conduction voltage dropping and resistance effect thereof are
ignored. Since the synchronous rectifier switches on the secondary
side have an extremely large output capacitance, the synchronous
rectifier switches on the secondary side can be regarded as
constant voltage sources. In addition, the rest of the components
in the circuit are ideal without any losses, and the integrated
transformer is replaced by a simple transformer model.
[0044] Phase 1 (t0-t1)
[0045] At time t=t0, in the first transformer 11, the first upper
bridge switch Q1 is zero-voltage switched to ON state, the first
lower bridge switch Q2 maintains at OFF state. Since the current
flowing through the first transformer 114 and the rectifier diode
D5 on the primary side transfers energy to the output load RL, the
voltage at the first magnetizing inductor Lm1 is clamped at nVo.
Therefore, the magnetizing current iLm1 is linearly increased. At
time t=t0, in the second transformer 12, the second upper bridge
switch Q4 is turned on, and the second lower bridge switch Q3 is
turned off. Since the current flowing through the second
transformer 124 and the rectifier diode D8 on the primary side
transfers energy to the output load RL, the voltage at the second
magnetizing inductor Lm2 voltage is clamped at -nVo. Therefore, the
magnetizing current iLm2 is linearly decreased. FIG. 4A shows a
current conduction path for Phase 1. Here, Vgs1, Vgs2, Vgs3, Vgs4
are gate voltages, Vds1, Vds2, Vds3, Vds4 are conduction voltages,
iD5, iD6, ID7, ID8, iO1, iO2, iD3 are current, and Vo is an output
voltage.
[0046] Phase 2 (t1-t2)
[0047] In the present phase, the first converter 11 maintains at a
state being the same as Phase 1. At time t is t1, in the second
converter 12, the resonant current iLr2 equals to the magnetizing
current iLm2, and no current flows into the second transformer 124.
The second transformer 124 is considered as being operated in
decoupling state while stopping transmitting energy to the output
end, and the rectifier diode D8 is zero-current cutoff, that is,
the cross voltage on the second magnetizing inductor Lm2 is no
longer clamped. Therefore, the second magnetizing inductor Lm2 is
resonant with the second resonant inductor Lr2 and the second
resonant capacitor Cr2 in this phase, while the output load energy
is provided by the output filter capacitor Co. FIG. 4B shows a
current conduction path for Phase 2.
[0048] Phase 3 (t2-t3)
[0049] In the present phase, the first converter 11 still maintains
at a state being the same as Phase 1. At time t is t2, in the
second converter 12, the second lower bridge switch Q4 is turned
off, while the resonant current iLr2 is still equal to the
magnetizing current iLm2. Therefore, the second transformer 124
continues to maintain the decoupling state. Since the resonant
current ILR2 maintains continuously flowing, the upper bridge
parasitic capacitor Coss4 is charged to the input voltage Vin,
while the lower bridge parasitic capacitor Coss3 is discharged to
zero voltage. In the present phase, the resonant assemblies can be
regarded as the upper and lower bridge parasitic capacitors Coss3,
Coss4, the second resonant inductor Lr2, the second magnetizing
inductor Lm2, and energy of the output load is still provided by
the output filter capacitor Co. FIG. 4C shows a current conduction
path for Phase 3.
[0050] Phase 4 (t3-t4)
[0051] In the present phase, the first converter 11 still maintains
at a state being the same as Phase 1. At time t is t1, in the
second converter 12, the lower bridge parasitic capacitance Coss4
is charged to the input voltage Vin, and the upper bridge parasitic
capacitance Coss3 has been discharged to zero voltage. At this
time, the resonant current iLr2 will turn on the body diode D3 of
the upper bridge switch Q3 so as to maintain continuously flowing.
FIG. 4D shows a current conduction path for Phase 4.
[0052] Phase 5 (t4-t5)
[0053] In the present phase, the first converter 11 still maintains
at a state being the same as Phase 1. At time t is t4, in the
second transformer 12, the upper bridge switch Q3 is turned on, and
the resonant current iLr2 of the primary side flows through the
second upper bridge switch Q3. Since the current flowing through
the second transformer 124 and the rectifier diode D7 on the
primary side transfers energy to the output load RL, the voltage at
the second magnetizing inductor Lm2 voltage is clamped at nVo.
Therefore, the magnetizing current iLm2 is linearly increased. FIG.
4E shows a current conduction path for Phase 5.
[0054] Phase 6 (t5-t6)
[0055] In the present phase, the second converter 12 maintains at a
state being the same as Phase 5. At time t is t5, in the first
converter 11, the resonant current iLr1 equals to the magnetizing
current iLm1, and no current flows into the first transformer 114.
The first transformer 114 is considered as being operated in a
decoupling state while stopping transmitting energy to the output
end, and the rectifier diode D5 is zero-current cutoff, that is,
the cross voltage on the first magnetizing inductor Lm1 is no
longer clamped. Therefore, the first magnetizing inductor Lm1 is
resonant with the first resonant inductor Lr1 and the first
resonant capacitor Cr1 in this phase, while the output load energy
is provided by the output filter capacitor Co. FIG. 4F shows a
current conduction path for Phase 6.
[0056] Phase 7 (t6-t7)
[0057] In the present phase, the second converter 12 still
maintains at a state being the same as Phase 5. At time t is t6, in
the first converter 11, the first upper bridge switch Q4 is turned
off, while the resonant current iLr1 is still equal to the
magnetizing current iLm1. Therefore, the first transformer 124
continues to maintain the decoupling state. Since the resonant
current iLr1 maintains continuously flowing, the upper bridge
parasitic capacitor Coss1 is charged to the input voltage Vin,
while the lower bridge parasitic capacitor Coss2 is discharged to
zero voltage. In the present phase, the resonant assemblies can be
regarded as the upper and lower bridge parasitic capacitors Coss1,
Coss2, the first resonant inductor Lr1, the first magnetizing
inductor Lm1, and energy of the output load is still provided by
the output filter capacitor Co. FIG. 4G shows a current conduction
path for Phase 7.
[0058] Phase 8 (t7-t8)
[0059] In the present phase, the second converter 12 still
maintains at a state being the same as Phase 5. At time t is t7, in
the second converter 11, the lower bridge parasitic capacitance
Coss2 is charged to the input voltage Vin, and the upper bridge
parasitic capacitance Coss1 has been discharged to zero voltage. At
this time, the resonant current iLr2 will make the body diode D3 of
the upper bridge switch Q3 being turned on so as to maintain
continuously flowing. FIG. 4H shows a current conduction path for
Phase 8.
[0060] The switch states of the first upper bridge switch Q1, the
first lower bridge switch Q2, the second upper bridge switch Q3,
the second bridge lower switch Q4, the first rectifier switch Q5,
the second rectifier switch Q6, the third rectifier switch Q7 and
the fourth rectifier switch Q8 are simplified as described in the
following table 1. "o" represents ON state, and X represents OFF
state. Here, Phase 1 to Phase 8 correspond to a first stage to a
fourth stage, and the fifth stage to the eighth stage are dual
states with respect to the first stage to the fourth stage.
TABLE-US-00001 TABLE 1 Stage Q1 Q2 Q3 Q4 Q5 Q6 Q7 Q8 1st Stage
.largecircle. X X .largecircle. .largecircle. X X .largecircle.
(t0-t2) .largecircle. X X .largecircle. .largecircle. X X X 2nd
Stage .largecircle. X X X .largecircle. X X X (t2-t4) .largecircle.
X X X .largecircle. X .largecircle. X 3rd Stage .largecircle. X
.largecircle. X .largecircle. X .largecircle. X (t4-t6)
.largecircle. X .largecircle. X X X .largecircle. X 4th Stage X X
.largecircle. X X X .largecircle. X (t6-t8) X X .largecircle. X X
.largecircle. .largecircle. X 5th Stage X .largecircle.
.largecircle. X X .largecircle. .largecircle. X (t8-t10) X
.largecircle. .largecircle. X X .largecircle. X X 6th Stage X
.largecircle. X X X .largecircle. X X (t10-t12) X .largecircle. X X
X .largecircle. X .largecircle. 7th Stage X .largecircle. X
.largecircle. X .largecircle. X .largecircle. (t12-t14) X
.largecircle. X .largecircle. X X X .largecircle. 8th Stage X X X
.largecircle. X X X .largecircle. (t14-t16) X X X .largecircle.
.largecircle. X X .largecircle.
[0061] In the present embodiment, the interleaved LLC half-bridge
series resonant converter 1 may further include a control circuit
15 for respectively controlling the first switch circuit 110, the
second switch circuit 120, the first rectifier circuit 116, and the
second rectifier circuit 126 to be switched between multiple
switching states. For example, the first upper bridge switch Q1,
the first lower bridge switch Q2, the second upper bridge switch
Q3, the second lower bridge switch Q4, the first rectifier switch
Q5, the second rectifier switch Q6, the third rectifier switch Q7,
and the fourth rectifier switch Q8 are configured to operate in the
manner of Table 1.
[0062] Reference is now made to FIGS. 5A and 5B. FIG. 5A is a graph
showing a comparison of output current ripples of different
structures according to an embodiment of the present disclosure,
and FIG. 5B is a graph showing a comparison of output current
ripples of a current sharing group and a non-uniform current group
according to an embodiment of the present disclosure. FIG. 5A shows
a comparison of the output current ripples between the interleaved
LLC half-bridge series resonant converter of the present disclosure
and a single LLC half-bridge series resonant converter for an
output current of 50 A. It can be seen that the interleaved LLC
half-bridge series resonant converter has advantage in reducing the
output current ripple.
[0063] In the conventional interleaved LLC half-bridge series
resonant converter, two transformers are required so as to transmit
the energy of each primary side, which means that two magnetic
cores are required. However, due to the difference in the material
of each magnetic core, the difference in the winding process and
the difference in the air gap, some errors are existed in the
magnetizing inductance and leakage inductance of the two
transformers, so that the secondary side of each group of
transformers transmits uneven energy, causing the countervail
effect of the output current ripples to deteriorate. As shown in
the drawings, the difference between the first transformer 114 and
the second transformer 124 causes a non-uniform flow condition for
the output current on the secondary side, and thus the output
current ripples thereof become large.
[0064] For this reason, the present disclosure also provides an
integrated transformer for the interleaved LLC half-bridge series
resonant converter. In order to maximize the ability for
suppressing the output current ripples for the interleaved LLC
half-bridge series resonant converter, the integrated transformer
of the present disclosure can greatly reduce the error of the
magnetizing inductance and the leakage inductance, so that the
current on the secondary side can be balanced, thereby maximizing
the ability for suppressing the output current ripples.
[0065] In addition, the integrated transformer of the present
disclosure can reduce the conduction loss of the synchronous
rectifier switches on the secondary side when the converter is
operated with low output voltage and high output current, and can
be provided with the function for dividing the secondary side into
multiple sets of current paths when using the magnetic core.
[0066] Reference is further made to FIG. 6, which is a schematic
diagram of an integrated transformer core according to an
embodiment of the present disclosure. As shown in the drawings, the
magnetic core of the integrated transformer provided by the present
disclosure will be divided into three columns. The magnetic core 14
contains a central column CP, a first outer column OP1 and a second
outer column OP2, coils of the primary side and the secondary side
will be individually wound on the two outer columns. Therefore, one
outer column forms one transformer, such that the two transformers
may be formed by one magnetic core, and the secondary side may also
be divided into two sets of parallel paths. Another advantage of
this structure is that the primary side winding and the secondary
side winding of each of the first transformer 114 and the second
transformer 124 are respectively wound on the same outer column,
and part of magnetic flux can be canceled at the central column CP,
thereby reducing the magnetic core loss.
[0067] Reference is now made to FIGS. 7A and 7B, which are
equivalent magnetic circuit diagrams of an integrated transformer
of an embodiment of the present disclosure. In order to avoid the
interference between the first transformer 114 and the second
transformer 124, the primary winding and the secondary winding are
wound on the same magnetic column in the same group of a center tap
transformer, as shown in FIG. 2.
[0068] Since currents on the primary side of the transformer of the
LLC half-bridge series resonant converter are AC currents, only the
AC flux is analyzed. According to the structure and the winding
type of each magnetic component of FIG. 2, the equivalent magnetic
circuit diagram is shown in FIG. 7A. It should be noted that in the
structure of the integrated transformer of the present disclosure,
the central tap type is selected, and a way for winding is to
replace the general way using the copper wire winding by using the
flat coil layout on the PCB board, so that the coil difference of
two transformers and the parameter error of the transformer are
minimized.
[0069] In addition, in the structure of the LLC half-bridge series
resonant converter, the ratio of the input voltage to the output
voltage varies as a gain value G of the resonant tank and the
numbers of primary side winding and secondary side winding of the
integrated transformer, as shown in the following equation (1):
Vout 1 2 Vin = G .times. Ns Np Eq ( 1 ) ##EQU00003##
[0070] Where Np and Ns are the numbers of the primary side winding
and the secondary side winding, and different winding turn ratios
allow the resonant converters to have different gain values G and
to operate in different region. When the resonant converter
operates in the second region, the gain of the resonant tank will
be greater than 1. Therefore, the relationship between the winding
turn ratio of the transformer and the input voltage Vin and the
output voltage Vout is shown in the following equation (2):
Ns N p .ltoreq. Vout 1 2 Vin Eq ( 2 ) ##EQU00004##
[0071] Based on the above equation (2), the winding turn ratio of
the transformer can be appropriately selected. For example, an
example of circuit specifications is provided in Table 2 below.
TABLE-US-00002 TABLE 2 Items Specification Input voltage (Vin) 380
VDC Output voltage (Vout) 12 VDC Output current (Iout) 50 A Maximum
output power (Pout) 600 W Switching frequency (FS) 100 kHz
Conversion efficiency (.eta.) 95%
[0072] According to this table, Ns/Np.ltoreq.12/190=1/15.833 can be
obtained. Therefore, the number of primary side winding Np is
selected to be 16 turns, and the number of secondary side winding
Ns is selected to be 1 turn.
[0073] Here, Ni is magnetic potential, g, c are magnetic
resistance, and .phi.1, .phi.c, and .phi.2 are flux directions. For
the sake of simplicity, it is assumed that the magnetic material
itself has a very small magnetic reluctance and can be neglected,
and only reluctances of the air gaps are considered. Since the two
transformers have the same structure, the number of primary side
winding is 16 turns, and the number of secondary side winding is 1
turn. Therefore, NP1=NP2=NP, and NS11=NS12=NS21=NS22=NS For the
structural symmetry, the equivalent magnetic reluctances of the air
gaps GP1 and GP2 of the first outer column OP1 and the second outer
column OP2 are both g, and the equivalent reluctance of the central
column CP is c.
[0074] In FIG. 7A, a magnetic flux relationship can be obtained
according to a view from the first outer column OP1 and the second
outer column OP2, as shown in the following equations (3) and
(4):
.PHI. 1 = ( g + C ) [ N P i P 1 - N S ( i S 11 + i S 12 ) ] g ( g +
2 C ) Eq ( 3 ) .PHI. 2 = ( g + C ) [ N P i P 2 - N S ( i S 21 + i S
22 ) ] g ( g + 2 C ) Eq ( 4 ) ##EQU00005##
[0075] The representations of the magnetic flux .phi.1 and the
magnetic flux .phi.2 respectively flowing into the central column
can be the following equations (5) and (6):
.PHI. 1 toC = .PHI. 1 .times. g g + C ; Eq ( 5 ) .PHI. 2 toC =
.PHI. 2 .times. g g + C ; Eq ( 6 ) ##EQU00006##
[0076] Since no air gap is in the central column, g>>c, and
it can be found that the magnetic flux .phi.1 and magnetic flux
.phi.2 of the two outer columns will only flow into the central
column without coupling. Since the left and right sides have
independent transformers, the magnetic circuit of FIG. 7A can be
disassembled into two simplified equivalent magnetic paths as shown
in FIG. 7B, and a new magnetic flux relationship can be obtained,
as shown in the following equations (7) and (8):
.PHI. 1 = N P i P 1 - N S ( i S 11 + i S 12 ) g + C ; Eq ( 7 )
.PHI. 2 = N P i P 2 - N S ( i S 21 + i S 22 ) g + C ; Eq ( 8 )
##EQU00007##
[0077] The flow direction of the magnetic flux in each column and
the equivalent magnetic circuit and variations for the magnetic
flux in one cycle will be illustrated by using the structure of the
magnetic core with the action interval diagrams thereof. Reference
is now made to FIGS. 8, 9A to 9D, which are a timing chart and
magnetic flux path diagrams showing operations of the magnetic flux
of the magnetic core according to an embodiment of the present
disclosure.
[0078] Phase 1 (t0-t1)
[0079] At time t is t0, the first side cross-voltage VP1 of the
first transformer 114 is larger than 0, the first side
cross-voltage VP2 of the second transformer 124 is larger than 0,
and the first side cross-voltage VP1 and the first side
cross-voltage VP2 respectively generate a magnetic flux .phi.1 and
a magnetic flux .phi.2 along the corresponding directions in the
first outer column OP1 and the second outer column OP2. The
magnetic flux .phi.1 will rise linearly with the positive slope,
and the magnetic flux .phi.2 will rise linearly with the negative
slope at the moment, while a magnetic flux .phi.c of the central
column accumulated by the magnetic flux .phi.1 and the magnetic
flux .phi.2 will rise with the positive slope. FIG. 9A shows a
magnetic flux path and an equivalent magnetic circuit for Phase 1.
It can be known from the equivalent magnetic circuit from FIG.
9A:
.PHI. 1 = N P 1 i P 1 - N S 1 i S 1 g + C ; .PHI. 2 = N P 2 i P 2 -
N S 2 i S 2 g + C ; and .PHI. c = .PHI.1 + .PHI.2 .
##EQU00008##
[0080] At time t is t1, the first side cross-voltage VP1 of the
first transformer 114 is smaller than 0, and the first side
cross-voltage VP2 of the second transformer 124 maintains at a
state being the same as Phase 1. Therefore, the magnetic flux
.phi.1 will decline linearly with the negative slope, and the
magnetic flux .phi.2 will rise linearly with the negative slope
while maintaining at the state of Phase 1, the magnetic flux .phi.c
of the central column accumulated by the declined magnetic flux
.phi.1 and the raised magnetic flux .phi.2 at the moment will
remain unchanged at a positive maximum. FIG. 9B shows a magnetic
flux path and an equivalent magnetic circuit for Phase 2. It can be
known from the equivalent magnetic circuit from FIG. 9B:
.PHI. 1 = - ( N P 1 i P 1 - N S 1 i S 1 g + C ) ; .PHI. 2 = N P 2 i
P 2 - N S 2 i S 2 g + C ; and .PHI. c = .PHI.1 + .PHI.2 .
##EQU00009##
[0081] Phase 3 (t2-t3)
[0082] At time t is t2, the first side cross-voltage VP1 of the
first transformer 114 maintains at a state being the same as Phase
2, and the first side cross-voltage VP2 of the second transformer
124 is smaller than 0. Therefore, the magnetic flux .phi.1
maintains declining linearly with the slope of Phase 2, and the
magnetic flux .phi.2 will decline linearly with the negative slope.
The magnetic flux .phi.c of the central column accumulated by the
magnetic flux .phi.1 and the magnetic flux .phi.2 will decline with
a negative slope. FIG. 9C shows a magnetic flux path and an
equivalent magnetic circuit for Phase 3. It can be known from the
equivalent magnetic circuit from FIG. 9C:
.PHI. 1 = - ( N P 1 i P 1 - N S 1 i S 1 g + C ) ; .PHI. 2 = - ( N P
2 i P 2 - N S 2 i S 2 g + C ) ; and .PHI. c = .PHI.1 + .PHI.2 .
##EQU00010##
[0083] Phase 4 (t3-t4)
[0084] At time t is t3, the first side cross-voltage VP1 of the
first transformer 114 is larger than 0, and the first side
cross-voltage VP2 of the second transformer 124 maintains at a
state being the same as Phase 3. Therefore, the magnetic flux
.phi.1 will rise linearly with a positive slope, and the magnetic
flux .phi.2 will maintain declining linearly with the slope of
Phase 3. The magnetic flux .phi.c of the central column accumulated
by the raised magnetic flux .phi.1 and the declined magnetic flux
.phi.2 at the moment will remain unchanged at a negative maximum.
FIG. 9D shows a magnetic flux path and an equivalent magnetic
circuit for Phase 4.
[0085] It can be known from the equivalent magnetic circuit from
FIG. 9D:
.PHI. 1 = N P 1 i P 1 - N S 1 i S 1 g + C ; .PHI. 2 = - ( N P 2 i P
2 - N S 2 i S 2 g + C ) ; and .PHI. c = .PHI.1 + .PHI.2 .
##EQU00011##
[0086] Based on the above, in order to prevent the first
transformer 114 and the second transformer 124 from coupling, the
first air gap GP1 and the second air gap GP2 are respectively added
to the first outer column OP1 and the second outer column OP2 for
dividing into the first magnetic column portion MP1 and the second
magnetic column portion MP2. Here, lengths of the first air gap GP1
and the second air gap GP2 depend on design values of the first
magnetizing inductor Lm1 and the second magnetizing inductor Lm2,
and soft-switching conditions in which the switch circuits on the
primary side being zero-voltage-switched to ON state must be
satisfied, and the design values are related to operation
frequencies, dead-zone time and switching parasitic
capacitance.
[0087] In detail, when the first converter 11 operates in LLC-SRC,
the zero-voltage switching conditions are: in a fixed dead zone
time, the magnetizing current iLm1 can smoothly discharge the
parasitic capacitor Coss1 of the first upper bridge switch Q1 to 0
volts, and charge the parasitic capacitor Coss2 of the first lower
bridge switch Q2 to the input voltage Vin. Here, a curve diagram of
the parasitic capacitance corresponding to the voltage across the
switch can be obtained according to the component manual provided
by the power switch manufacturer, and the total charge for the
parasitic capacitance charged from 0V to the input voltage Vin, for
example, 380V shown in Table 2, can be obtained by using the curve
diagram.
[0088] In this embodiment, the total charge of the parasitic
capacitance Coss1 of the first upper bridge switch Q1 charged from
0 V to 380 V can be, for example, 124585 .rho.C, and the
magnetizing inductance values are obtained by using the following
equations (9) and (10):
Qtotal = Coss .times. Vds = iLm .times. td ; Eq ( 9 ) Lm .ltoreq.
td 16 .times. Coss .times. fs = 381.3 .mu. H ; Eq ( 10 )
##EQU00012##
[0089] By combining the results of equations (9) and (10), the
inductance value of the first magnetizing inductor Lm1 is selected
to be 381.3 .mu.H After obtaining the magnitude of the magnetizing
inductance value from the circuit specification, the value of the
magnetizing inductance depends on the size specification, the
number of coils, the length of the air gap, the magnetic
permeability of the magnetic core, and the like. The magnitudes of
the respective magnetic reluctances can be obtained first, and then
the first magnetizing inductor Lm1 and the second magnetizing
inductor Lm2 are derived. In order to obtain a more accurate
relationship between the magnetizing inductance value and the
magnetic reluctance to facilitate following design procedures, the
magnetic reluctance of the magnet material will be taken into
account. First of all, the magnetic reluctance must be divided into
several blocks, as shown in FIG. 11, which is a schematic diagram
showing magnetic reluctance blocks of the integrated transformer of
an embodiment of the present disclosure.
[0090] As shown in FIG. 11, the specifications of the magnetic core
14 are shown as the lengths a to f in the drawings, and the
equivalent magnetic reluctances of the magnetic reluctance blocks i
to v can be arranged as shown in Table 3 below:
TABLE-US-00003 TABLE 3 Magnetic reluctance Magnetic path length
Magnetic path area block (le) (Ae) i .pi. 8 ( a + c ) ##EQU00013##
f ( a + c ) 2 ##EQU00014## ii .pi. 8 ( e + c ) ##EQU00015## f ( e +
c ) 2 ##EQU00016## iii b c .times. f iv d a .times. f v d - g e
.times. f
[0091] Therefore, according to table 3 above, the relationship
between magnetizing inductance and magnetic reluctance can be
obtained as follows:
L m = N P 2 2 .times. ( i + ii + iii ) + iv + v + g ; ##EQU00017##
L m = N p 2 2 .times. ( ? ? + ? ? + ? ? ) + l iv .mu. 0 .mu. r A iv
+ l v .mu. 0 .mu. r A v + l GAP .mu. 0 A g ; ##EQU00017.2## ?
indicates text missing or illegible when filed ##EQU00017.3##
[0092] It can be seen that the lengths g of the first air gap GP1
and the second air gap GP2 are specially designed for the
inductance values required by the circuit specifications, and the
magnetic flux .phi.1 and the magnetic flux .phi.2 will only flow
into the center column OP1 through the arrangement of the first air
gap GP1 and the second air gap GP2. However, the magnetic flux
.phi.1 and the magnetic flux .phi.2 have the advantage of
offsetting each other due to the phase difference, so that the
maximum value of the magnetic flux .phi.c of the center column OP1
does not become large. Reference is now made to FIG. 10, which is a
graph showing a comparison for magnetic flux of each of columns of
the magnetic core of the integrated transformer of an embodiment of
the present disclosure. Here, .phi.1 and .phi.2 are magnetic flux
of the two outer column, .phi.c is the magnetic flux of the central
column, and the units are WB for the above magnetic flux. From FIG.
10, it can be known that it is feasible to integrate two
conventional transformers into one magnetic core.
[0093] One of the advantages of the present disclosure is that the
interleaved LLC half-bridge series resonant converter having an
integrated transformer provided by the present disclosure can
reduce output current ripple and improve efficiency by utilizing
two series-coupled LLC-SRCs combined with a mechanism in that
90.degree. of phase shift is provided.
[0094] One of the advantages of the present disclosure is that the
interleaved LLC half-bridge series resonant converter having an
integrated transformer provided by the present disclosure can
reduce the error of two transformers to achieve current-sharing for
the secondary currents by utilizing a technique in that two
transformers being integrated into one magnetic core, and by
replacing the conventional winding frame-wound transformer with the
plate transformer.
[0095] The above disclosure is only a preferred embodiment of the
present disclosure, and is not intended to limit the scope of the
present disclosure. Therefore, any equivalent technical changes
made by using the present specification and the contents of the
drawings are included in the scope of the present disclosure.
[0096] The foregoing description of the exemplary embodiments of
the disclosure has been presented only for the purposes of
illustration and description and is not intended to be exhaustive
or to limit the disclosure to the precise forms disclosed. Many
modifications and variations are possible in light of the above
teaching.
[0097] The embodiments were chosen and described in order to
explain the principles of the disclosure and their practical
application so as to enable others skilled in the art to utilize
the disclosure and various embodiments and with various
modifications as are suited to the particular use contemplated.
Alternative embodiments will become apparent to those skilled in
the art to which the present disclosure pertains without departing
from its spirit and scope.
* * * * *