U.S. patent application number 15/463502 was filed with the patent office on 2018-09-20 for synchronization for low-energy long-range communications.
The applicant listed for this patent is SAMSUNG ELECTRONICS CO., LTD.. Invention is credited to JEONGTAEK LEE, JACOB C. SHARPE, MYEONG-CHEOL SHIN, FEI TONG.
Application Number | 20180270091 15/463502 |
Document ID | / |
Family ID | 63491098 |
Filed Date | 2018-09-20 |
United States Patent
Application |
20180270091 |
Kind Code |
A1 |
SHARPE; JACOB C. ; et
al. |
September 20, 2018 |
SYNCHRONIZATION FOR LOW-ENERGY LONG-RANGE COMMUNICATIONS
Abstract
A receiver configured to receive a frequency-modulated
transmission having a preamble and a corresponding method are
provided, the receiver having a buffer coupled to an input terminal
for receiving the transmission, a time-to-frequency transformer
coupled to the buffer, an energy aggregator coupled to the
transformer, a preamble detector coupled to the aggregator, and a
symbol synchronizer coupled to the detector; the method including
receiving a sequence of time-domain frequency-modulated samples,
transforming the sequence of time-domain samples into a spectrum of
frequency-domain data, and matching an actual energy distribution
over a plurality of discrete frequencies in the frequency-domain
data with an expected energy distribution of the preamble to
determine frequency error.
Inventors: |
SHARPE; JACOB C.;
(CAMBRIDGE, GB) ; TONG; FEI; (BASSINGBOURN,
GB) ; SHIN; MYEONG-CHEOL; (SUWON-SI, KR) ;
LEE; JEONGTAEK; (SEONGNAM-SI, KR) |
|
Applicant: |
Name |
City |
State |
Country |
Type |
SAMSUNG ELECTRONICS CO., LTD. |
SUWON-SI |
|
KR |
|
|
Family ID: |
63491098 |
Appl. No.: |
15/463502 |
Filed: |
March 20, 2017 |
Current U.S.
Class: |
1/1 |
Current CPC
Class: |
H04L 27/2692 20130101;
H04W 4/80 20180201; H04L 27/12 20130101; H04L 27/2682 20130101;
H04L 27/265 20130101; H04L 27/2662 20130101; H04L 27/2675 20130101;
H04L 27/2657 20130101; H04W 28/14 20130101; H04L 27/2672
20130101 |
International
Class: |
H04L 27/12 20060101
H04L027/12; H04W 28/14 20060101 H04W028/14; H04L 27/26 20060101
H04L027/26; H04W 4/00 20060101 H04W004/00 |
Claims
1. A method for symbol synchronization in a frequency-modulated
transmission having a preamble, the method comprising: receiving
the frequency-modulated transmission and obtaining a sequence of
time-domain samples of the preamble; transforming the sequence of
time-domain samples into a spectrum of frequency-domain data over
discrete frequencies arranged in a plurality of frequency bins;
comparing combined energies for each of a plurality of subsets of
the frequency bins, each subset corresponding to a relative
frequency distribution of energy expected in the preamble at a
respective carrier frequency; selecting the subset of the frequency
bins with the greatest combined energy, the selected subset being
indicative of a carrier frequency error of the transmission; and
synchronizing receipt of the frequency-modulated transmission in
accordance with the selected subset of the frequency bins.
2. The method of claim 1, wherein the synchronizing receipt of the
frequency-modulated transmission in accordance with the selected
subset of the frequency bins comprises comparing phase
relationships at the plurality of discrete frequencies to determine
symbol timing.
3. The method of claim 1 wherein the frequency-modulated
transmission is a Guassian frequency-shift-keying (FSK) modulated
transmission.
4. The method of claim 1, wherein the receiving of the
frequency-modulated transmission comprises: receiving a plurality
of periodic samples of the frequency-modulated transmission;
buffering the plurality of the received periodic samples; and
wherein the transforming comprises: performing a block
transformation on the plurality of buffered periodic samples into
the plurality of frequency bins.
5. The method of claim 4, further comprising comparing the combined
energies to a threshold, wherein if the combined energies does not
exceed the threshold, receiving the next periodic samples, or
wherein if the combined energies exceeds the threshold,
synchronizing the transmission and determining a fine carrier
frequency error.
6. The method of claim 1, further comprising determining fractional
symbol timing based on relative frequency bin phases.
7. The method of claim 4, wherein: the block transformation is an
n-point time-domain to frequency-domain transform, and fractional
symbol timing is determined in accordance with relative frequency
bin phases of first and second bins separated by n/2 bins for
determining high-precision symbol timing with an ambiguity, a third
bin separated from the first bin by n/8 bins and a fourth bin
separated from the second bin by n/8 bins for resolving the
ambiguity.
8. The method of claim 4, wherein the block transformation is a
64-point Fast Fourier Transform (FFT).
9. The method of claim 1, further comprising: achieving symbol
timing synchronization; and beginning demodulation of a main packet
based on the achieved symbol timing synchronization.
10. The method of claim 4, further comprising: comparing the
greatest combined energy of the subset to a threshold; and if the
greatest combined energy exceeds the threshold, using frequency bin
phases to extract symbol timing.
11. The method of claim 1, wherein the frequency-modulated
transmission is a Bluetooth.TM. Low-Energy Long-Range transmission,
and the preamble includes ten 8 .mu.s repetitions of the 8-bit
pattern 00111100.
12. The method of claim 1 wherein the carrier frequency error is
based on a fractional bin spacing.
13. The method of claim 1, further comprising performing an
alternate FFT with a 1/2 bin frequency offset to obtain a carrier
frequency error within 1/4 bin.
14. A receiving apparatus for symbol synchronization in a
frequency-modulated transmission having a preamble, the receiving
apparatus comprising: a receiver configured to receive the
frequency-modulated transmission and obtain therefrom a sequence of
time-domain samples of the preamble; and a processor configured to:
transform the sequence of time-domain samples into a spectrum of
frequency-domain data over discrete frequencies arranged in a
plurality of frequency bins, compare combined energies for each of
the plurality of subsets of the frequency bins, each subset
corresponding to a relative frequency distribution of energy
expected in the preamble at a respective carrier frequency, select
the subset of the frequency bins with the greatest combined energy,
the selected subset being indicative of a carrier frequency error
of the transmission, and synchronize receipt of the
frequency-modulated transmission in accordance with the selected
subset of the frequency bins.
15. The receiving apparatus of claim 14, wherein the processor is
further configured to compare phase relationships at the plurality
of discrete frequencies to determine symbol timing.
16. The receiving apparatus of claim 14, wherein the
frequency-modulated transmission is a Gaussian FSK modulated
transmission.
17. The receiving apparatus of claim 14, wherein: the receiver is
further configured to receive a plurality of periodic samples of
the frequency-modulated transmission; and the processor is further
configured to: buffer the plurality of the received periodic
samples to a buffer, perform a block transformation on the
plurality of buffered periodic samples into the plurality of
frequency bins.
18. The receiving apparatus of claim 17, wherein the processor is
further configured to compare the combined energies to a threshold,
wherein if the combined energies does not exceed the threshold, the
processor is further configured to receive the next periodic
samples, and wherein if the combined energies exceeds the
threshold, the processor is further configured to synchronize the
transmission, and determine a fine carrier frequency error.
19. The receiving apparatus of claim 14, wherein the processor
further configured to determine fractional symbol timing based on
relative frequency bin phases.
20. A program storage device tangibly embodying a program of
instruction steps executable by a processor for receiver symbol
synchronization in a frequency-modulated transmission having a
preamble, the instruction steps comprising: receiving the
frequency-modulated transmission and obtaining a sequence of time
samples of the preamble; transforming the sequence of time-domain
samples into a spectrum of frequency-domain data over discrete
frequency arranged in a plurality of frequency bins; comparing
combined energies for each of a plurality of subsets of the
frequency bins, each subset corresponding to a relative frequency
distribution of energy expected in the preamble at a respective
carrier frequency; selective the subset of the frequency bins with
the greatest combined energy, the selected subset being indicative
of a carrier frequency error of the transmission; and synchronizing
receipt of the frequency-modulated transmission in accordance with
the selected subset of the frequency bins.
Description
TECHNICAL FIELD
[0001] Embodiments of the present inventive concept relate to
communications, and more particularly, to a system and method for
carrier frequency error determination and symbol synchronization in
low-energy long-range communications.
DISCUSSION OF RELATED ART
[0002] Wireless communications generally require synchronization at
a receiver for accurate decoding. Traditionally, a frame type of
synchronization has been achieved via bit correlation after
frequency demodulation.
SUMMARY
[0003] An exemplary embodiment method is provided for receiver
symbol synchronization in a frequency-modulated transmission having
a preamble, and includes receiving a sequence of time-domain
frequency-modulated samples, transforming the sequence of
time-domain samples into a spectrum of frequency-domain data, and
matching an actual energy distribution over a plurality of discrete
frequencies in the frequency-domain data with an expected energy
distribution of the preamble to determine frequency error.
[0004] The method may include comparing phase relationships at the
plurality of discrete frequencies to determine symbol timing. The
method may be applied where matching the expected energy
distribution includes combining energies over the plurality of
discrete frequencies.
[0005] The method may include receiving periodic samples of the
frequency-modulated transmission, buffering a plurality of the
received samples, performing a block transformation on the
plurality of buffered samples into a plurality of frequency bins,
comparing combined energies for each of a plurality of subsets of
the plurality of frequency bins, each subset corresponding to the
relative frequency distribution of energy expected in the preamble,
selecting the subset with the greatest combined energy to determine
the carrier frequency error of the transmission, and synchronizing
receipt of the frequency-modulated transmission in accordance with
the selected subset. The method may include comparing the combined
energies to a threshold, wherein if the threshold is not exceeded
receiving the next periodic samples, or wherein if the threshold is
exceeded synchronizing the transmission and determining a fine
carrier frequency error.
[0006] The method may include determining fractional symbol timing
based on relative frequency bin phases. The method may be applied
where the block transformation is an n-point time-domain to
frequency-domain transform, and fractional symbol timing is
determined in accordance with relative frequency bin phases of
first and second bins separated by n/2 bins for determining
high-precision symbol timing with an ambiguity, a third bin
separated from the first bin by n/8 bins and a fourth bin separated
from the second bin by n/8 bins for resolving the ambiguity.
[0007] The method may be applied where the block transformation is
a 64-point Fast Fourier Transform (FFT). The method may include
achieving symbol timing synchronization, and beginning demodulation
of a main packet based on the achieved synchronization.
[0008] The method may include comparing the subset with the
greatest combined energy to a threshold, and if the combined energy
exceeds the threshold using frequency bin phases to extract symbol
timing. The method may be applied where the frequency-modulated
transmission is a Bluetooth.TM. Low-Energy Long-Range transmission,
and the preamble includes ten 8 .mu.s as repetitions of the 8-bit
pattern 00111100.
[0009] The method may be applied where the carrier frequency error
is based on a fractional bin spacing. The method may include
performing an alternate FFT with a 1/2 bin frequency offset to
obtain a carrier frequency error within 1/4 bin.
[0010] An exemplary embodiment receiver is provided as configured
to receive a frequency-modulated transmission having a preamble,
the receiver including a buffer coupled to an input terminal for
receiving the transmission, a time-to-frequency transformer coupled
to the buffer, an energy aggregator coupled to the transformer, a
preamble detector coupled to the aggregator, and a symbol
synchronizer coupled to the detector.
[0011] The receiver may include a start-pattern de-mapper coupled
to the synchronizer. The receiver may be configured as an I/Q
receiver. The receiver may be configured as a direct conversion
receiver, a super-heterodyne receiver, or the like.
[0012] The receiver may be applied where the time-to-frequency
transformer is an n-point Fast Fourier Transform (FFT) unit. The
receiver may be applied where the detector is configured to take
squares of absolute values of transformed frequency bins, test
summations of candidate bin sets, and compare to a threshold.
[0013] The receiver may include an amplifier configured to receive
an input signal, an analog-to-digital converter (ADC) connected to
the amplifier, and a filter connected to the ADC, where the
down-sampler is connected to the filter, and the buffer is
connected to the down-sampler and configured for receiving each
sample to simultaneously hold a plurality of data samples so a
time-domain to frequency-domain transform may be periodically
performed on the data within the buffer.
[0014] The receiver may be applied where the down-sampler is
configured to down-samples to symbol rate. The receiver may be
applied where the buffer is a first-in, first-out (FIFO)
buffer.
[0015] The receiver may include a low-noise amplifier (LNA)
configured to receive an input signal, an oscillator configured to
generate a cosine wave and a 90-degree offset sine wave, a first
mixer connected to the LNA and the oscillator and configured to mix
the input with the cosine wave to expose an in-phase (I) portion of
the input signal, a second mixer connected to the LNA and the
oscillator and configured to mix the input with the sine wave to
expose a quadrature-phase (Q) portion of the input signal, a first
analog-to-digital converter (ADC) connected to the first mixer, a
second ADC connected to the second mixer, a first low-pass filter
(LPF) connected to the first ADC, a second LPF connected to the
second ADC, a first down-sampler connected to the first LPF and
configured for down-sampling the in-phase I data, a second
down-sampler connected to the second LPF and configured for
down-sampling the quadrature-phase Q data, and at least one buffer
connected to the down-samplers and configured for receiving each
I/Q sample pair to simultaneously hold a plurality of I/Q data
sample pairs so a time-domain to frequency-domain transform may be
performed on the data within this buffer periodically.
[0016] An exemplary embodiment program storage device is provided
tangibly embodying a program of instructions executable by a
processor for receiver symbol synchronization in a
frequency-modulated transmission having a preamble, the
instructions including receiving a sequence of time-domain
frequency-modulated samples, transforming the sequence of
time-domain samples into a spectrum of frequency-domain data, and
matching an actual energy distribution over a plurality of discrete
frequencies in the frequency-domain data with an expected energy
distribution of the preamble to determine frequency error.
[0017] The instruction steps may include comparing phase
relationships at the plurality of discrete frequencies to determine
symbol timing. The instruction steps may be applied where matching
the expected energy distribution includes combining energies over
the plurality of discrete frequencies.
[0018] The instruction steps may include receiving periodic samples
of the frequency-modulated transmission, buffering a plurality of
the received samples, performing a block transformation on the
plurality of buffered samples into a plurality of frequency bins,
comparing combined energies for each of a plurality of subsets of
the plurality of frequency bins, each subset corresponding to the
relative frequency distribution of energy expected in the preamble,
selecting the subset with the greatest combined energy to determine
the carrier frequency error of the transmission, and synchronizing
receipt of the frequency-modulated transmission in accordance with
the selected subset. The instruction steps may include comparing
the combined energies to a threshold, wherein if the threshold is
not exceeded receiving the next periodic samples, or wherein if the
threshold is exceeded synchronizing the transmission and
determining a fine carrier frequency error.
[0019] The instruction steps may include determining fractional
symbol timing based on relative frequency bin phases. The
instruction steps may be applied where the block transformation is
an n-point time-domain to frequency-domain transform, and
fractional symbol timing is determined in accordance with relative
frequency bin phases of first and second bins separated by n/2 bins
for determining high-precision symbol timing with an ambiguity, a
third bin separated from the first bin by n/8 bins and a fourth bin
separated from the second bin by n/8 bins for resolving the
ambiguity.
[0020] The instruction steps may be applied where the block
transformation is a 64-point Fast Fourier Transform (FFT). The
instruction steps may include achieving symbol timing
synchronization, and beginning demodulation of a main packet based
on the achieved synchronization.
[0021] The instruction steps may include comparing the subset with
the greatest combined energy to a threshold, and if the combined
energy exceeds the threshold using frequency bin phases to extract
symbol timing. The instruction steps may be applied where the
frequency-modulated transmission is a Bluetooth.TM. Low-Energy
Long-Range transmission, and the preamble includes ten 8 .mu.s
repetitions of the 8-bit pattern 00111100.
[0022] The instruction steps may be applied where the carrier
frequency error is based on a fractional bin spacing. The
instruction steps may include performing an alternate FFT with a%
bin frequency offset to obtain a carrier frequency error within 1/4
bin.
BRIEF DESCRIPTION OF THE DRAWINGS
[0023] Aspects of the present inventive concept may become apparent
and appreciated upon consideration of the following description of
exemplary embodiments when taken in conjunction with the
accompanying drawings, in which:
[0024] FIG. 1 is a schematic block diagram of an I/Q receiver in
accordance with an exemplary embodiment of the present inventive
concept;
[0025] FIG. 2 is a schematic block diagram of an I/Q receiver with
FFT detector in accordance with an exemplary embodiment of the
present inventive concept;
[0026] FIG. 3 is a graphical diagram comparing coherent and
incoherent summation of a spoof signal received by an I/Q receiver
with FFT detector in accordance with an exemplary embodiment of the
present inventive concept;
[0027] FIG. 4 is a graphical diagram of FFT bins for the spoof
signal received by an I/Q receiver with FFT detector in accordance
with an exemplary embodiment of the present inventive concept;
[0028] FIG. 5 is a graphical diagram comparing coherent and
incoherent summation of FFT bins for a signal received by an I/Q
receiver with FFT detector in accordance with an exemplary
embodiment of the present inventive concept; and
[0029] FIG. 6 is a schematic flow diagram for a method of operating
an I/Q receiver with FFT detector in accordance with an exemplary
embodiment of the present inventive concept.
DETAILED DESCRIPTION
[0030] Bluetooth.TM. Low-Energy (BLE) Long-Range is a wireless
communications protocol expected to operate at low Signal-to-Noise
Ratios (SNR) such as around zero decibels (dB). Using an 8 .mu.s
repetition period, the BLE Long-Range preamble includes 10
repetitions of the 8-bit pattern 00111100. Various methodologies
for determining carrier frequency error and achieving symbol
synchronization are possible. Here, a coarse carrier frequency
error may be determined through finding the location of a best
matching frequency bin pattern prior to thresholding and
synchronization, while a fine carrier frequency error may be
determined through thresholding and synchronization.
[0031] Symbol synchronization (Sync) may be approached by a
correlation of the received signal with an ideal realization of the
signal. This correlation could be performed in the Cartesian-domain
or in the frequency-domain for each candidate Sync position. A
correlation of sufficient magnitude would indicate Sync, and at the
maximum correlation value provide symbol timing.
[0032] A Cartesian-domain correlation has excellent performance at
low SNR, but the complexity of the correlator increases as n.sup.2
so it quickly becomes an expensive solution for long sequences.
Cartesian correlation is desensitized by frequency errors, which
may necessitate the use of a matched filter bank to consider
multiple frequency offset candidates thereby adding to the
complexity.
[0033] A frequency-domain correlation, which may involve a phase
differentiation (e.g., FM demodulation) of the received I/Q data
and comparison against the expected bit pattern, is efficient at
high SNR and robust to frequency error. It has been used for
standard Bluetooth.TM. (BT) and Bluetooth.TM. Low-Energy (BLE).
Like Cartesian-domain correlation, its complexity rises as n.sup.2.
It is less effective at low SNR as there is significant loss of SNR
in the I/Q to frequency-domain conversion when the operating SNR is
low.
[0034] In Orthogonal Frequency-Division Multiplexing (OFDM) such as
IEEE 802.11 OFDM PHY, the preamble has a frequency structure
resulting in time-domain repetition. As with OFDM modulation, Fast
Fourier Transform (FFT) methods may be used to achieve
time/frequency synchronization in BLE Long-Range. However, the
preamble frequency structure such as tone spacing and tone power
for BLE Long-Range is quite different.
[0035] As shown in FIG. 1, an exemplary embodiment I/Q receiver is
indicated generally by the reference numeral 100. The I/Q receiver
100 includes a low-noise amplifier (LNA) 110 configured to receive
an input signal, an oscillator 111 configured to generate a cosine
wave and a 90-degree offset sine wave, a first mixer 112 connected
to the LNA and the oscillator and configured to mix the input with
the cosine wave to expose the in-phase (I) portion of the input
signal, a second mixer 113 connected to the LNA and the oscillator
and configured to mix the input with the sine wave to expose the
quadrature-phase (Q) portion of the input signal, a first
analog-to-digital converter (ADC) 114 connected to the first mixer,
a second ADC 115 connected to the second mixer, a first low-pass
filter (LPF) 116 connected to the first ADC, a second LPF 117
connected to the second ADC, a first down-sampler 118 connected to
the first LPF and configured for down-sampling the in-phase I data
to 1 MHz, a second down-sampler 119 connected to the second LPF and
configured for down-sampling the quadrature-phase Q data to 1 MHz,
and buffer 120, such as a first-in first-out (FIFO) buffer,
connected to the down-samplers and configured for receiving each
I/Q sample pair to simultaneously hold 64 I/Q data sample pairs so
an FFT may be performed on the data within this buffer at least
every 8 .mu.s.
[0036] The exemplary embodiment receiver 100 uses a direct
conversion receiver architecture capable of implementing the
inventive concept, although the inventive concept is not limited to
this type of architecture. For example, a super-heterodyne type of
receiver or the like may be used in alternate embodiments. At the
LNA 110, the signal is received in the 2.4-2.48 GHz Industrial,
Scientific and Medical (ISM) band, which is next amplified, mixed,
and sampled. The signal is then digitized and at least low-pass
filtered using a filter having sufficiently wide bandwidth, here
1.3 MHz, to allow passage of the BLE Long Range signal including
any frequency offset prior to down-sampling. Thus, the inventive
concept works on the buffered I/Q data, down-sampled in this
exemplary embodiment to 1 million samples per second
(MSample/s).
[0037] Turning to FIG. 2, an exemplary embodiment I/Q receiver is
indicated generally by the reference numeral 200. The I/Q receiver
200 is similar to the I/Q receiver 100 of FIG. 1, so duplicate
description may be omitted. In addition, the I/Q receiver 200
includes a 64-point FFT unit 130 connected to the buffer 120. The
buffer 120 may be a FIFO buffer with serial access, and/or support
simultaneous parallel access to all of its 64 I/Q sample stores.
The receiver 200 further includes a detector 140 configured to take
absolute (ABS) values (and/or squares) of the FFT bins, test
summations (Z) of candidate 4-bin (or alternatively 6-bin) sets,
and compare to a threshold. The detector, in turn, is connected to
a synchronizer 150. If the threshold is not exceeded, the detector
signals the FFT device 130 and awaits the next FFT; but when the
threshold is exceeded, the detector passes a Sync signal to the
synchronizer 150. Upon receipt of the Sync signal, the synchronizer
150 enters a Sync-achieved state, where the 4 (or alternatively 6)
bin set with the largest summation is used to determine the
frequency error, and the bin phases are used to determine symbol
timing. A start-pattern de-mapper 160 is connected to the
synchronizer, in turn, and configured to de-map the synchronized
start-pattern data.
[0038] Turning now to FIG. 3, a graphical comparison of coherent
(e.g., in-phase) and incoherent/non-coherent (e.g., out-of-phase)
summation for a spoof signal received by an exemplary I/Q receiver,
such as the I/Q receiver 200 of FIG. 2, is indicated generally by
the reference numeral 300. Here, the coherent summation 310 has
significantly lower continuous and peak correlated magnitude output
than the incoherent summation 320.
[0039] When summing the result vectors of successive FFT frames
together, one can just add the magnitudes, which is an incoherent
sum, or sum the complex vectors, which is a coherent sum. With the
coherent combination, any stationary sinusoids with a frequency
mid-way between 2 FFT bins across 2 successive FFT frames, for
example, would cancel out instead of summing in spectrum. Here, the
significantly lower peak magnitude of the coherent summation 310
indicates that the frequency bins are not in perfect phase. Here,
coherent indicates that a phase relationship is enforced on the
bins. Thus, if the complex vectors are summed, all of the phases
must be equal to get the maximum sum. The phase relationship of the
bins changes depending on the starting point of the FFT, and only
for one point will all the phases be equal. However, the spoofing
signal chosen has a phase relationship that would never exist for a
real signal, but all phase information is neglected by taking the
absolute value or squaring.
[0040] Use of the inventive concept may be detected by sending such
a spoof signal before the main waveform or normal packet in a
throughput test. A product deploying the inventive concept would
Sync on such an amplitude-modulated (AM) spoof signal but then fail
to decode the access code and packet. The result should be a
detectable loss in throughput. A method not employing the inventive
concept should not Sync to the spoofing signal and thus have no
loss in throughput versus its standard performance. Although this
spoof signal is for six (6) bins, details of how many bins such as
four (4), and which bins are used such as the inner four, and how
the bins are summed such as sum of squares or absolute values, may
lead to different optimal spoof signals in alternate
embodiments.
[0041] As shown in FIG. 4, a graphical representation of normalized
magnitudes for all 6 FFT bins of a real BLE long range signal
received by an exemplary I/Q receiver like that of FIG. 2 is
indicated generally by the reference numeral 400. The normalized
magnitudes of the FFT bins are the same for the real BLE long range
signal compared to the used bins of the spoof signal. Although all
6 frequency bins are shown (i.e., -375 kHz, -250 kHz, -125 kHz,
+125 kHz, +250 kHz, +375 kHz) over the relevant frequency spectrum,
either all of these or fewer, such as the middle 4 frequency bins
(i.e., -250 kHz, -125 kHz, +125 kHz, +250 kHz), may be utilized in
alternate embodiments.
[0042] The BLE Long-Range preamble is 10 repetitions of the bit
sequence 00111100, which is modulated as a Gaussian Frequency Shift
Keying (GFSK) symbol at a symbol rate of 1 million symbols per
second (MSymbol/s) with a modulation index of 0.5 and a
bandwidth-time (BT) product of 0.5. This is substantially the same
BT product as a standard BLE signal. In the frequency-domain, this
BLE Long-Range preamble signal consists of 6 tones at the .+-.125
kHz, .+-.250 kHz, .+-.375 kHz frequency locations. The preamble
repetition helps to concentrate the spectral energy in a subset of
bins. The inventive concept hones in on this concentration of
spectral energy. The exemplary spectrum is for a frequency error of
0 kHz, but the method is robust for frequency error. The effect of
a frequency error is to shift the 6 occupied bins substantially
equally. If the bins are spaced at 15.625 kHz, an error of 46.875
kHz would shift the entire pattern 3 bins to the right (i.e., the
occupied bins would change from 9, 17, 25, 41, 49, and 57 to 12,
20, 28, 44, 52, and 60).
[0043] Turning to FIG. 5, a graphical comparison of coherent and
incoherent summation of the FFT bins for a BLE Long Range preamble
signal received by an exemplary I/Q receiver, such as the I/Q
receiver 200 of FIG. 2, is indicated generally by the reference
numeral 500. Here, the coherent summation 510 has substantially the
same peak correlated magnitude output as the incoherent summation
520.
[0044] A timewise correlation of two signals can be related to the
multiplication of their Fourier transforms through the convolution
theorem. Thus, the correlation of the received I/Q sequence signal
with the ideal I/Q sequence may be obtained by multiplying the FFT
of the received I/Q with the FFT of the ideal signal. As the ideal
signal consists of just 6 bins, one need only consider these bins
in the multiplication for a given frequency error. The timewise
correlation will have strong peaks every 8 .mu.s. With the FFT,
this information is contained in the phases of the 6 tones.
Coherently combining the FFT output with a phase relationship
corresponding to a unique sampling time produces the desired result
for a timewise correlation. Following this methodology, the FFT
could either be performed many times to test all starting times, or
the results of one FFT could be considered with the tone phases of
a series of starting time candidates.
[0045] The present exemplary embodiment combines the squared
magnitudes of the FFT bins. This may reduce the SNR of the
resulting correlation, but greatly simplifies the processing as all
of the symbol time candidates may be considered simultaneously.
Once a BLE Long-Range signal has been detected, the symbol time can
then be calculated from the phases of the FFT bins.
[0046] Coherent combination would enforce a particular phase
relationship on the tones. So, one would multiply the tones from
the FFT by "ideal tones" known from the transmitted sequence and
then sum. Whereas incoherent combination involves summing the
squares of the tones so the phase information is lost. Correlation
would always involve a multiplication, but one could modify the
method to work on the absolute values. Squaring is preferred
because absolute values can be vulnerable to obscure FFT artefacts
such as from ramping the signal.
[0047] A comparison of coherent and incoherent summation may be
achieved by summing the squares or the absolute values, but it is
not limited thereto. All of the frequency error candidates are then
accounted for by considering different sets of summed bins, as
offset by the frequency error.
[0048] Turning now to FIG. 6, a method of operating an I/Q receiver
for frequency error estimation and symbol synchronization in
low-energy long-range communications is indicated generally by the
reference numeral 600. Here, an input block 610 receives I/Q valued
data and passes control to a function block 620. The block 620
down-samples the data to 1 MHz and passes control to a function
block 630. The block 630 buffers 64 samples of the data and passes
control to a function block 640. The block 640 performs a 64-point
FFT on the samples at least every 8 .mu.s into 64 FFT bins and
passes control to a function block 650. The block 650 calculates a
square of the absolute value for all FFT bins and passes control to
a function block 660. The block 660 sums sets of 4 (or 6) bins over
27 combinations to cover all frequency errors and passes control to
a decision block 670. The block 670 takes the maximal combination
and compares it to a threshold, looping control back to block 640
if the threshold is not exceeded, but passing control on to a
function block 680 if the threshold is exceeded. The block 680, in
turn, uses the FFT bin phases to extract Symbol timing.
[0049] Thus, the processing flow of the inventive concept operates
on a buffer of I/Q data that has been down sampled to 1 MHz. 64
samples are stored in the buffer, which is 64 .mu.s of signal while
the total preamble length is 80 .mu.s. Every 8 .mu.s or less, an
FFT is performed on this buffered data; the resulting output will
be the real or in-phase and imaginary or quadrature-phase values of
the 64 FFT bins. The sum of the absolute values squared of all of
these bins is then taken. All bins are used since different
frequency error candidates must typically be considered. The BLE
specification allows for a transmitter to be in error by up to
.+-.150 kHz, while the receiver itself may have a frequency error
of .+-.50 kHz. Such large potential frequency errors must be
accommodated because Bluetooth.TM. was originally specified for
relatively inexpensive hardware devices, and backward compatibility
remains desirable.
[0050] Thus, frequency errors up to .+-.200 kHz are considered, and
a 1.3 MHz filter bandwidth suffices as being 300 kHz greater than
the 1 MHz down-sampled data rate. The bin spacing is 1
MHz/64=15.625 kHz, so this calls for considering bin offsets of
200/15.625=12.8 bins. As only integer bins can be considered, this
is .+-.13 bins. Note that the tone spacing (125 kHz) is a multiple
of the bin spacing (8*15.625 kHz), which allows for a simpler
implementation since all of the bins can be shifted by a fixed
number depending upon the frequency offset.
[0051] Due to the ten 8 .mu.s repetitions of the 00111100 8-bit
pattern in the preamble for BLE Long-Range, the signal energy is
concentrated in 6 tones at .+-.125 kHz, .+-.250 kHz, and .+-.375
kHz. Embodiments of the present inventive concept may use all six
of these tones, or alternately just the center four (i.e., .+-.125
kHz and .+-.250 kHz) since the outer two (i.e., .+-.375 kHz) should
be of relatively lower magnitude. For the 0 Hz offset bin, the
absolute values squared of the bins (17, 25, 41, 49) are combined
when using the central 4 of the 6 bins. Using the absolute values
squared is one way of assuring non-negative results for
complex-valued I/Q samples. For ease of explanation regardless of
the offset frequency error, the 6 bins resulting from the preamble
signal may be thought of as any set of 6 bins (out of 64 in this
embodiment) that could be seen in a frequency-domain plot through a
mask or ruler with 6 equidistant holes spaced at 125 kHz
intervals.
[0052] After the 27 candidates have been calculated, the largest
candidate is compared against a threshold. If the threshold is
exceeded, Sync is detected. The two outer bins are not used in this
exemplary embodiment since they contain only half the signal
strength of the inner bins, and their inclusion may exacerbate the
detection of false maximums, such as one 8 bins offset from the
true frequency error, which due to the presence of noise might have
a slightly larger magnitude than that of the true frequency error.
By removing the 2 outer bins, only 2 of the 4 remaining bins
overlap in this case as compared to 4 of the 6 original bins.
[0053] Setting or adjusting the sync threshold requires care. At
low powers with low SNR, a fixed threshold may be used. As the
signal power increases, the threshold should increase or false Sync
can be detected before the buffer contains 64 samples of preamble.
This is done here by summing the absolute values squared of the
buffered samples to give a signal power (P) estimate, where the
threshold is then set according to the following relationship:
thresh=max(fixed.sub.threshold, grad_threshold*P-threshold_offset)
(Eqn. 1)
[0054] In BLE Long-Range, spacing between preamble tones is
coincidentally fixed by the standard to 125 kHz apart. If a
different FFT length were used, the interval between samples might
not be 8 .mu.s but would preferably be an integer number of bins
apart. That is, FFT length and down-sampling rate are linked.
[0055] Frequency offsets at a half bin spacing, for example, will
reduce the magnitude of the candidates by about 3 dB. To avoid
this, alternate FFT's may be performed with a 1/2 bin frequency
offset. As the preamble length is 10 repetitions and only 8 are
used for the FFT in this exemplary embodiment, one FFT will then
have a frequency error of less than 1/4 bin.
[0056] I/Q data is used because I/Q takes advantage of sine and
cosine signals being mathematically orthogonal to each other, and
therefore separable with minimal effort. With appropriate care, one
signal can be modulated with a cosine wave and the other with a
sine wave that is basically a delayed version of the cosine wave,
where these two signals remain separable at the receiver even
though their frequency spectrums overlap. I/Q pairs are the real
and imaginary components of the complex-valued transmitted baseband
signal. "I/Q data" refers to the real or in-phase (I) and imaginary
or quadrature-phase (Q) samples of the constellation for the
modulation type used. There may be various pairs of I/Q "samples"
that occur during interim processing. Although a complex I/Q data
signal is used in the exemplary embodiments, it shall be understood
that alternate embodiments may use different signals, such as
multiple signals that are separated in time, frequency, phase,
and/or quadrature.
[0057] An FFT is performed on the received I/Q data using an
appropriate sample rate, such as the symbol rate. This places
substantially all of the signal energy into just the 6 bins
allowing a significant increase in SNR. By summing the energy in
various candidate sets of the 6 bins, or of the inner 4 bins in
some embodiments, it is possible at low SNR to determine both the
presence of a BLE Long-Range signal as well as its frequency error.
The relative phases of the bins can then be used to determine
fractional symbol timing with respect to the FFT window. This
information is sufficient to achieve symbol synchronization and
begin demodulation of the main BLE Long-Range packet.
[0058] Exemplary embodiments of the inventive concept use an FFT
per block of samples to achieve time and frequency synchronization.
This exploits the relatively low complexity of FFTs while
maintaining de-noising gain. The relatively low complexity using
FFTs is due to the reduced increase in complexity as nlogn for
sequences of length n, which is significantly less than for other
methods that increase as n.sup.2. In an exemplary embodiment, 64
samples of I/Q data are stored where each I/Q data sample uses a
9-bit word length for I and a 9-bit word length for Q. Each FFT
block uses the n=64 I/Q data samples covering 8 repetitions of the
8-bit BLE preamble pattern 00111100. In alternate embodiments, a
block may include a different number n of I/Q data samples. For
example, n=80 could cover all 10 repetitions of the preamble.
[0059] The inventive concept may further use a combining scheme to
make the detection more robust to frequency offsets. This balances
the difficulties of frequency error estimation with signal
detection. By magnitude combining the candidate bins, SNR may be
somewhat sacrificed to allow the consideration of multiple
frequency error and sampling time candidates in a highly efficient
manner. Nonetheless, the SNR remains sufficient to successfully
detect BLE Long-Range packets with a very low false detection
rate.
[0060] Thus, embodiments of the inventive concept require less
hardware to implement than time-domain correlation schemes.
Time-domain correlation given de-sensitization of the correlation
by frequency error would require a large number of simultaneous
correlations to be performed, such as 20 for example. Time-domain
correlation might otherwise offer better performance at low SNR,
but insufficient benefit would be realized since performance is
limited by the bit error rate (BER) for both Sync methodologies. It
shall be understood that this is the BER of the access code and
payload demodulation, even with ideal Sync.
[0061] The inventive concept has a substantially lower SNR
threshold for a successful detection then frequency-domain
correlation. In theory, frequency-domain correlation is simpler as
it avoids an FFT, but hardware implementation is likely to be at
least as expensive. As an exemplary embodiment of the present
inventive concept can determine symbol timing using the tone
phases, it is only required every 8 .mu.s. For frequency-domain
correlation, the correlation would be determined far more
frequently such as per sample or faster than the 1 MSample/s in the
buffer since, there, the correlation peak or the like is used to
derive the symbol timing.
[0062] The method of the present inventive concept makes use of the
energy in the BLE Long-Range preamble being concentrated at 6 tones
(i.e., -375 kHz, -250 kHz, -125 kHz, +125 kHz, +250 kHz, +375 kHz).
These have a specific magnitude relationship (i.e., 0.47, 1, 0.91,
-0.91, 1, -0.47) at sample offset 1=0. By adjusting the magnitude
relationship, a spoofing signal waveform may be constructed that
while it should be detected by the FFT synchronization algorithm,
should not cause either I/Q correlation or frequency-domain
correlation to detect synchronization. A possible candidate for
this spoofing signal would be [0.5 1 0.9 -0.9 -1 -0.5]. This signal
uses amplitude modulation (AM), so it would not be seen by a
frequency synchronizer and thus the response of an I/Q correlation
should be substantially reduced.
[0063] The next step of the inventive concept is to extract the
symbol timing. This can be done from the phases of the FFT bins.
The phases of the FFT bins vary with starting time as outlined
below. t=0 is exact alignment with the 00111100 * 8 pattern where t
is the offset in input buffer samples from that alignment. t's
range is 0 to 63, but as the FFT is repeated at least every 8
.mu.s, the correction is only needed over an 8 sample (8 ps) range.
The bin numbers (e.g., -8) are given relative to the center bin.
This center bin includes the offset for the selected frequency
error candidate.
Y [ - 16 ] = - t 2 .pi. , Y [ - 8 ] = - t 4 .pi. , Y [ 8 ] = .pi. +
t 4 .pi. , Y [ 16 ] = t 2 .pi. ( Eqn . 2 ) ##EQU00001##
[0064] The difference between Y[-16] and Y[16] might provide the
greatest accuracy, but will change through 8.pi. in 8 samples
providing an ambiguous answer. The phase difference between any two
pairs of bins where the bins of each pair are each separated by 8
bins can be used to resolve the ambiguity. The phase difference
between bins (-16 & -8) and (8 & 16) is used by
calculating:
Coarse Timing = 4 .pi. angle ( Y [ - 8 ] * Y [ - 16 ] _ - Y [ 16 ]
* Y [ 8 ] _ ) ( Eqn . 3 ) = 4 .pi. angle ( e - j t .pi. 4 .times. e
j t .pi. 2 - e j t .pi. 2 .times. e - j ( .pi. + t .pi. 4 ) ) ( Eqn
. 4 ) = 4 .pi. angle ( e j t .pi. 4 + e j t .pi. 4 ) = 4 .pi. angle
( 2 e j t .pi. 4 ) = t ( Eqn . 5 ) ##EQU00002##
[0065] To get a high accuracy estimate for the symbol timing,
.theta. = 1 .pi. angle ( Y [ - 16 ] * Y [ - 16 ] _ ) ( Eqn . 6 )
##EQU00003##
is used to derive 4 timing candidates of 0+.theta., 2+.theta., 4+0
and 6+.theta.. The closest candidate to the coarse timing will then
recover the symbol timing. By retaining fractional accuracy, this
gives an accurate symbol timing even with decimation to 1 MHz.
[0066] Fractional symbol timing may be determined based on relative
frequency bin phases where the block transformation is an n-point
time-domain to frequency-domain transform. Here, the fractional
symbol timing is determined in accordance with relative frequency
bin phases of first and second bins separated by n/2 bins for
determining high-precision symbol timing with an ambiguity, a third
bin separated from the first bin by n/8 bins and a fourth bin
separated from the second bin by n/8 bins for resolving the
ambiguity. This bin spacing is applicable to BLE long range with
the given FFT length.
[0067] The inventive concept is not limited to the direct
conversion receiver architecture illustrated, and could be applied
to other receiver architectures such as super-heterodyne receivers
or the like. The choice of FFT length and buffering rate is
arbitrary. Combinations which make the tone spacing an integer
multiple of the bin spacing will make the processing simpler since
the frequency candidates are integer shifts, but with added
complexity non-integer fractional shifts may be accommodated.
[0068] It is also possible to sum the absolute values of the bins
and compare this to a threshold and to make use of the outer two
bins. If using the outer bins for offsets larger than 7, then a
circular shift ("circshift") operation may be applied to any bins
below 0 or above 63 since the tone may alias. This is feasible
since the filtering before the buffering should not be exactly 1
MHz, but wider (e.g., 1.3 MHz) with allowance for any frequency
error.
[0069] There are numerous algorithm choices for setting the
threshold. Additionally, there are numerous choices for
establishing the sampling time from the FFT bin phases using
different combinations of the tones.
[0070] Accordingly, the present inventive concept reduces the
complexity of the correlator to increasing as nlogn using a FFT per
block of samples to achieve time and frequency synchronization,
rather than increasing as n.sup.2 as in each of the traditional
Cartesian correlation and the traditional frequency-domain
correlation. This facilitates a significant reduction in correlator
complexity for large n. It shall be understood that performing a
single FFT multiple times is still possible, versus performing a
block FFT fewer times.
[0071] Although exemplary embodiments of the present inventive
concept have been shown and described, it shall be understood that
those of ordinary skill in the pertinent art may make changes
therein without departing from the scope, principles, and spirit of
the present inventive concept as defined by the appended claims and
their equivalents.
* * * * *