U.S. patent application number 15/894145 was filed with the patent office on 2018-06-28 for excitation and use of guided surface wave modes on lossy media.
The applicant listed for this patent is CPG Technologies, LLC. Invention is credited to James F. Corum, Kenneth L. Corum.
Application Number | 20180180729 15/894145 |
Document ID | / |
Family ID | 51486974 |
Filed Date | 2018-06-28 |
United States Patent
Application |
20180180729 |
Kind Code |
A1 |
Corum; James F. ; et
al. |
June 28, 2018 |
EXCITATION AND USE OF GUIDED SURFACE WAVE MODES ON LOSSY MEDIA
Abstract
Disclosed are various embodiments systems and methods for
transmission and reception of electrical energy along a surface of
a lossy conducting medium medium. In one example, a receive circuit
is used to receive electrical energy from a guided surface
waveguide probe that transmits the electrical energy in the form of
a Zenneck surface wave along a surface of a a lossy conducting
medium.
Inventors: |
Corum; James F.;
(Morgantown, WV) ; Corum; Kenneth L.; (Plymouth,
NH) |
|
Applicant: |
Name |
City |
State |
Country |
Type |
CPG Technologies, LLC |
Italy |
TX |
US |
|
|
Family ID: |
51486974 |
Appl. No.: |
15/894145 |
Filed: |
February 12, 2018 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
|
|
13789525 |
Mar 7, 2013 |
9910144 |
|
|
15894145 |
|
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Current U.S.
Class: |
1/1 |
Current CPC
Class: |
H01Q 1/36 20130101; G01S
13/88 20130101; G01S 13/0218 20130101; H01Q 1/04 20130101; H01Q
9/30 20130101; H02J 50/12 20160201 |
International
Class: |
G01S 13/02 20060101
G01S013/02; H02J 50/12 20060101 H02J050/12; H01Q 1/04 20060101
H01Q001/04; H01Q 1/36 20060101 H01Q001/36; H01Q 9/30 20060101
H01Q009/30; G01S 13/88 20060101 G01S013/88 |
Claims
1. A method, comprising: positioning a receive circuit relative to
a terrestrial medium; and receiving, via the receive circuit,
energy conveyed in a form of a Zenneck surface wave launched by a
guided surface waveguide probe that generates a plurality of
electromagnetic fields that are substantially mode-matched to a
Zenneck surface wave mode of a surface of a lossy conducting
medium, wherein the electromagnetic fields substantially synthesize
a wave front incident at a complex Brewster angle of the lossy
conducting medium.
2. The method of claim 1, wherein the Zenneck surface wave
propagates along a surface of the terrestrial medium in a guided
transmission line mode.
3. The method of claim 1, wherein the receive circuit loads an
excitation source coupled to the guided surface waveguide
probe.
4. The method of claim 1, wherein the energy further comprises
electrical power, and the method further comprises applying the
electrical power to an electrical load coupled to the receive
circuit, where the electrical power is used as a power source for
the electrical load.
5. The method of claim 1, wherein the receive circuit comprises at
least one of a magnetic coil, a linear probe, or a tuned
resonator.
6. An apparatus, comprising: a receive circuit that receives energy
conveyed by a Zenneck surface wave from a guided surface waveguide
probe that generates a plurality of electromagnetic fields that are
substantially mode-matched to a Zenneck surface wave mode of a
surface of a lossy conducting medium, wherein the electromagnetic
fields substantially synthesize a wave front incident at a complex
Brewster angle of the lossy conducting medium.
7. The apparatus of claim 6, wherein the receive circuit is
configured to load an excitation source coupled to the guided
surface waveguide probe that launches the Zenneck surface wave.
8. The apparatus of claim 6, wherein the energy further comprises
electrical power, and the receive circuit is coupled to an
electrical load, and wherein the electrical power is applied to the
electrical load, the electrical power being employed as a power
source for the electrical load.
9. The apparatus of claim 8, wherein the electrical load is
impedance matched with the receive circuit.
10. The apparatus of claim 6, wherein the receive circuit comprises
a linear probe.
11. The apparatus of claim 6, wherein the receive circuit comprises
a magnetic coil.
12. The apparatus of claim 6, wherein the receive circuit comprises
a tuned resonator.
13. The apparatus of claim 12, wherein the tuned resonator
comprises a series tuned resonator.
14. The apparatus of claim 12, wherein the tuned resonator
comprises a parallel tuned resonator.
15. The apparatus of claim 12, wherein the tuned resonator
comprises a distributed tuned resonator.
16. A power transmission system, comprising: a guided surface
waveguide probe that transmits electrical energy by a plurality of
resultant fields that are substantially mode-matched to a Zenneck
surface wave mode of a surface of a terrestrial medium, where the
guided surface waveguide probe is configured to impose a plurality
of voltage magnitudes and a plurality of phases on a plurality of
charge terminals of the guided surface waveguide probe, wherein the
plurality of voltage magnitudes and a plurality of phases vary
based at least in part on a physical position of a respective one
of the plurality of charge terminals; and a receive circuit that
receives the electrical energy.
17. The power transmission system of claim 16, wherein the receive
circuit loads the guided surface waveguide probe.
18. The power transmission system of claim 16, wherein an
electrical load is coupled to the receive circuit and the
electrical energy is used as a power source for the electrical
load.
19. The power transmission system of claim 18, wherein the
electrical load is impedance matched to the receive circuit.
20. The power transmission system of claim 18, wherein a power
transfer is established from the receive circuit to the electrical
load.
Description
CROSS REFERENCE TO RELATED APPLICATIONS
[0001] This application is a continuation of U.S. patent
application Ser. No. 13/789,525, entitled "EXCITATION AND USE OF
GUIDED SURFACE WAVE MODES ON LOSSY MEDIA," filed on Mar. 7, 2013,
the entire contents of which is hereby incorporated herein by
reference in its entirety.
BACKGROUND
[0002] For over a century, signals transmitted by radio waves
involved radiation fields launched using conventional antenna
structures. In contrast to radio science, electrical power
distribution systems in the last century involved the transmission
of energy guided along electrical conductors. This understanding of
the distinction between radio frequency (RF) and power transmission
has existed since the early 19003 s.
BRIEF DESCRIPTION OF THE DRAWINGS
[0003] Many aspects of the present disclosure can be better
understood with reference to the following drawings. The components
in the drawings are not necessarily to scale, emphasis instead
being placed upon clearly illustrating the principles of the
disclosure. Moreover, in the drawings, like reference numerals
designate corresponding parts throughout the several views.
[0004] FIG. 1 is a chart that depicts a field strength as a
function of distance for a guided electromagnetic field and a
radiated electromagnetic field.
[0005] FIG. 2 is a drawing that illustrates a propagation interface
with two regions employed for transmission of a guided surface wave
according to various embodiments of the present disclosure.
[0006] FIG. 3 is a drawing that illustrates a polyphase waveguide
probe disposed with respect to a propagation interface of FIG. 2
according to an embodiment of the present disclosure.
[0007] FIG. 4 is a drawing that provides one example illustration
of a phase shift in a ground current that facilitates the launching
of a guided surface-waveguide mode on a lossy conducting medium in
the propagation interface of FIG. 3 according to an embodiment of
the present disclosure.
[0008] FIG. 5 is a drawing that illustrates a complex angle of
insertion of an electric field synthesized by the polyphase
waveguide probes according to the various embodiments of the
present disclosure.
[0009] FIG. 6 is a schematic of a polyphase waveguide probe
according to an embodiment of the present disclosure.
[0010] FIGS. 7A-J are schematics of specific examples of the
polyphase waveguide probe of FIG. 6 according to various
embodiments of the present disclosure.
[0011] FIGS. 8A-C are graphs that illustrate field strengths of
guided surface waves at select transmission frequencies generated
by the various embodiments of polyphase waveguide probes according
to the various embodiments of the present disclosure.
[0012] FIG. 9 shows one example of a graph of experimental
measurements of field strength of a guided surface wave at 59
Megahertz as a function of distance generated by a polyphase
waveguide probe according to an embodiment of the present
disclosure.
[0013] FIG. 10 shows a graph of experimental measurements of the
phase as a function of distance of the guided surface wave of FIG.
9 according to an embodiment of the present disclosure.
[0014] FIG. 11 shows another example of a graph of experimental
measurements of field strength as a function of distance of a
guided surface wave generated by a polyphase waveguide probe at
1.85 Megahertz according to an embodiment of the present
disclosure.
[0015] FIGS. 12A-B depict examples of receivers that may be
employed to receive energy transmitted in the form of a guided
surface wave launched by a polyphase waveguide probe according to
the various embodiments of the present disclosure.
[0016] FIG. 13 depicts an example of an additional receiver that
may be employed to receive energy transmitted in the form of a
guided surface wave launched by a polyphase waveguide probe
according to the various embodiments of the present disclosure.
[0017] FIG. 14A depicts a schematic diagram representing the
Thevenin-equivalent of the receivers depicted in FIGS. 12A-B
according to an embodiment of the present disclosure.
[0018] FIG. 14B depicts a schematic diagram representing the
Norton-equivalent of the receiver depicted in FIG. 13 according to
an embodiment of the present disclosure.
DETAILED DESCRIPTION
[0019] Referring to FIG. 1, to begin, some terminology shall be
established to provide clarity in the discussion of concepts to
follow. First, as contemplated herein, a formal distinction is
drawn between radiated electromagnetic fields and guided
electromagnetic fields.
[0020] As contemplated herein, a radiated electromagnetic field
comprises electromagnetic energy that is emitted from a source
structure in the form of waves that are not bound to a waveguide.
For example, a radiated electromagnetic field is generally a field
that leaves an electric structure such as an antenna and propagates
through the atmosphere or other medium and is not bound to any
waveguide structure. Once radiated electromagnetic waves leave an
electric structure such as an antenna, they continue to propagate
in the medium of propagation (such as air) independent of their
source until they dissipate regardless of whether the source
continues to operate. Once electromagnetic waves are radiated, they
are not recoverable unless intercepted, and, if not intercepted,
the energy inherent in radiated electromagnetic waves is lost
forever. Electrical structures such as antennas are designed to
radiate electromagnetic fields by maximizing the ratio of the
radiation resistance to the structure loss resistance. Radiated
energy spreads out in space and is lost regardless of whether a
receiver is present. The energy density of radiated fields is a
function of distance due to geometrical spreading. Accordingly, the
term "radiate" in all its forms as used herein refers to this form
of electromagnetic propagation.
[0021] A guided electromagnetic field is a propagating
electromagnetic wave whose energy is concentrated within or near
boundaries between media having different electromagnetic
properties. In this sense, a guided electromagnetic field is one
that is bound to a waveguide and may be characterized as being
conveyed by the current flowing in the waveguide. If there is no
load to receive and/or dissipate the energy conveyed in a guided
electromagnetic wave, then no energy is lost except for that
dissipated in the conductivity of the guiding medium. Stated
another way, if there is no load for a guided electromagnetic wave,
then no energy is consumed. Thus, a generator or other source
generating a guided electromagnetic field does not deliver real
power unless a resistive load is present. To this end, such a
generator or other source essentially runs idle until a load is
presented. This is akin to running a generator to generate a 60
Hertz electromagnetic wave that is transmitted over power lines
where there is no electrical load. It should be noted that a guided
electromagnetic field or wave is the equivalent to what is termed a
"transmission line mode." This contrasts with radiated
electromagnetic waves in which real power is supplied at all times
in order to generate radiated waves. Unlike radiated
electromagnetic waves, guided electromagnetic energy does not
continue to propagate along a finite length waveguide after the
energy source is turned off. Accordingly, the term "guide" in all
its forms as used herein refers to this transmission mode of
electromagnetic propagation.
[0022] To further illustrate the distinction between radiated and
guided electromagnetic fields, reference is made to FIG. 1 that
depicts graph 100 of field strength in decibels (dB) above an
arbitrary reference in volts per meter as a function of distance in
kilometers on a log-dB plot. The graph 100 of FIG. 1 depicts a
guided field strength curve 103 that shows the field strength of a
guided electromagnetic field as a function of distance. This guided
field strength curve 103 is essentially the same as a transmission
line mode. Also, the graph 100 of FIG. 1 depicts a radiated field
strength curve 106 that shows the field strength of a radiated
electromagnetic field as a function of distance.
[0023] Of interest are the shapes of the curves 103/106 for
radiation and for guided wave propagation. The radiated field
strength curve 106 falls off geometrically (1/d, where d is
distance) and is a straight line on the log-log scale. The guided
field strength curve 103, on the other hand, has the characteristic
exponential decay of e.sup.-ad/ {square root over (d)} and exhibits
a distinctive knee 109. Thus, as shown, the field strength of a
guided electromagnetic field falls off at a rate of e.sup.-ad/
{square root over (d)}, whereas the field strength of a radiated
electromagnetic field falls off at a rate of 1/d, where d is the
distance. Due to the fact that the guided field strength curve 103
falls off exponentially, the guided field strength curve 103
features the knee 109 as mentioned above. The guided field strength
curve 103 and the radiated field strength curve 106 intersect at a
crossover point 113 which occurs at a crossover distance. At
distances less than the crossover distance, the field strength of a
guided electromagnetic field is significantly greater at most
locations than the field strength of a radiated electromagnetic
field. At distances greater than the crossover distance, the
opposite is true. Thus, the guided and radiated field strength
curves 103 and 106 further illustrate the fundamental propagation
difference between guided and radiated electromagnetic fields. For
an informal discussion of the difference between guided and
radiated electromagnetic fields, reference is made to Milligan, T.,
Modern Antenna Design, McGraw-Hill, 1.sup.st Edition, 1985, pp.8-9,
which is incorporated herein by reference in its entirety.
[0024] The distinction between radiated and guided electromagnetic
waves, made above, is readily expressed formally and placed on a
rigorous basis. That two such diverse solutions could emerge from
one and the same linear partial differential equation, the wave
equation, analytically follows from the boundary conditions imposed
on the problem. The Green function for the wave equation, itself,
contains the distinction between the nature of radiation and guided
waves.
[0025] In empty space, the wave equation is a differential operator
whose eigenfunctions possess a continuous spectrum of eigenvalues
on the complex wave-number plane. This transverse electro-magnetic
(TEM) field is called the radiation field, and those propagating
fields are called "Hertzian waves". However, in the presence of a
conducting boundary, the wave equation plus boundary conditions
mathematically lead to a spectral representation of wave-numbers
composed of a continuous spectrum plus a sum of discrete spectra.
To this end, reference is made to Sommerfeld, A., "Uber die
Ausbreitung der Wellen in der Drahtlosen Telegraphie," Annalen der
Physik, Vol. 28, 1909, pp. 665-736. Also see Sommerfeld, A.,
"Problems of Radio," published as Chapter 6 in Partial Differential
Equations in Physics--Lectures on Theoretical Physics: Volume VI,
Academic Press, 1949, pp. 236-289, 295-296; Collin, R. E.,
"Hertzian Dipole Radiating Over a Lossy Earth or Sea: Some Early
and Late 20.sup.th Century Controversies," IEEE Antennas and
Propagation Magazine, Vol. 46, No. 2, April 2004, pp. 64-79; and
Reich, H. J., Ordnung, P. F, Krauss, H. L., and Skalnik, J. G.,
Microwave Theory and Techniques, Van Nostrand, 1953, pp. 291-293,
each of these references being incorporated herein by reference in
their entirety.
[0026] To summarize the above, first, the continuous part of the
wave-number eigenvalue spectrum, corresponding to branch-cut
integrals, produces the radiation field, and second, the discrete
spectra, and corresponding residue sum arising from the poles
enclosed by the contour of integration, result in non-TEM traveling
surface waves that are exponentially damped in the direction
transverse to the propagation. Such surface waves are guided
transmission line modes. For further explanation, reference is made
to Friedman, B., Principles and Techniques of Applied Mathematics,
Wiley, 1956, pp. pp. 214, 283-286, 290, 298-300.
[0027] In free space, antennas excite the continuum eigenvalues of
the wave equation, which is a radiation field, where the outwardly
propagating RF energy with E.sub.z and H.sub..phi. in-phase is lost
forever. On the other hand, waveguide probes excite discrete
eigenvalues, which results in transmission line propagation. See
Collin, R. E., Field Theory of Guided Waves, McGraw-Hill, 1960, pp.
453, 474-477. While such theoretical analyses have held out the
hypothetical possibility of launching open surface guided waves
over planar or spherical surfaces of lossy, homogeneous media, for
more than a century no known structures in the engineering arts
have existed for accomplishing this with any practical efficiency.
Unfortunately, since it emerged in the early 1900's, the
theoretical analysis set forth above has essentially remained a
theory and there have been no known structures for practically
accomplishing the launching of open surface guided waves over
planar or spherical surfaces of lossy, homogeneous media.
[0028] According to the various embodiments of the present
disclosure, various polyphase waveguide probes are described that
are configured to excite radial surface currents having resultant
fields that synthesize the form of surface-waveguide modes along
the surface of a lossy conducting medium. Such guided
electromagnetic fields are substantially mode-matched in magnitude
and phase to a guided surface wave mode on the surface of the lossy
conducting medium. Such a guided surface wave mode may also be
termed a Zenneck surface wave mode. By virtue of the fact that the
resultant fields excited by the polyphase waveguide probes
described herein are substantially mode-matched to a Zenneck
surface wave mode on the surface of the lossy conducting medium, a
guided electromagnetic field in the form of a Zenneck surface wave
is launched along the surface of the lossy conducting medium.
According to one embodiment, the lossy conducting medium comprises
a terrestrial medium such as the Earth.
[0029] Referring to FIG. 2, shown is a propagation interface that
provides for an examination of the boundary value solution to
Maxwell's equations derived in 1907 by Jonathan Zenneck as set
forth in his paper Zenneck, J., "On the Propagation of Plane
Electromagnetic Waves Along a Flat Conducting Surface and their
Relation to Wireless Telegraphy," Annalen der Physik, Serial 4,
Vol. 23, Sep. 20, 1907, pp. 846-866. FIG. 2 depicts cylindrical
coordinates for radially propagating waves along the interface
between a lossy conducting medium specified as Region 1 and an
insulator specified as Region 2. Region 1 may comprise, for
example, any lossy conducting medium. In one example, such a lossy
conducting medium may comprise a terrestrial medium such as the
Earth or other medium. Region 2 is a second medium that shares a
boundary interface with Region 1 and has different constitutive
parameters relative to Region 1. Region 2 may comprise, for
example, any insulator such as the atmosphere or other medium. The
reflection coefficient for such a boundary interface goes to zero
only for incidence at a complex Brewster angle. See Stratton, J.
A., Electromagnetic Theory, McGraw-Hill, 1941, p. 516.
[0030] According to various embodiments, the present disclosure
sets forth various polyphase waveguide probes that generate
electromagnetic fields that are substantially mode-matched to a
Zenneck surface wave mode on the surface of the lossy conducting
medium comprising Region 1. According to various embodiments, such
electromagnetic fields substantially synthesize a wave front
incident at a complex Brewster angle of the lossy conducting medium
that results in zero reflection.
[0031] To explain further, in Region 2, where e.sup.j.omega.t field
variation is assumed and where .rho..noteq.0 and z .gtoreq.0 (z is
a vertical coordinate normal to the surface of Region 1, .rho. is
the radial dimension in cylindrical coordinates), Zenneck's
closed-form exact solution of Maxwell's equations satisfying the
boundary conditions along the interface are expressed by the
following electric field and magnetic field components:
H 2 .phi. = Ae - u 2 z H 1 ( 2 ) ( - j .gamma..rho. ) , ( 1 ) E 2
.rho. = A ( u 2 j .omega. o ) e - u 2 z H 1 ( 2 ) ( - j
.gamma..rho. ) , and ( 2 ) E 2 z = A ( - .gamma. .omega. o ) e - u
2 z H 0 ( 2 ) ( - j .gamma..rho. ) . ( 3 ) ##EQU00001##
[0032] In Region 1, where e.sup.j.omega.t field variation is
assumed and where .rho..noteq.0 and z .ltoreq.0, Zenneck's
closed-form exact solution of Maxwell's equations satisfying the
boundary conditions along the interface are expressed by the
following electric field and magnetic field components:
H 1 .phi. = Ae u 1 z H 1 ( 2 ) ( - j .gamma..rho. ) , ( 4 ) E 1
.rho. = A ( - u 1 .sigma. 1 + j .omega. 1 ) e u 1 z H 1 ( 2 ) ( - j
.gamma..rho. ) , and ( 5 ) E 1 z = A ( - j .gamma. .sigma. 1 + j
.omega. 1 ) e u 1 z H 0 ( 2 ) ( - j .gamma..rho. ) . ( 6 )
##EQU00002##
[0033] In these expressions, H.sub.n.sup.(2)(-j.gamma..rho.) is a
complex argument Hankel function of the second kind and order n,
u.sub.1 is the propagation constant in the positive vertical
direction in Region 1, u.sub.2 is the propagation constant in the
vertical direction in Region 2, .sigma..sub.1 is the conductivity
of Region 1, .omega. is equal to 2.pi.f, where f is a frequency of
excitation, .epsilon..sub.0 is the permittivity of free space,
.epsilon..sub.1 is the permittivity of Region 1, A is a source
constant imposed by the source, z is a vertical coordinate normal
to the surface of Region 1, .gamma. is a surface wave radial
propagation constant, and .rho. is the radial coordinate.
[0034] The propagation constants in the .+-.z directions are
determined by separating the wave equation above and below the
interface between Regions 1 and 2, and imposing the boundary
conditions. This exercise gives, in Region 2,
u 2 = - jk o 1 + ( r - jx ) , ( 7 ) ##EQU00003##
and gives, in Region 1,
u.sub.1=-u.sub.2(.epsilon..sub.r-jx). (8)
The radial propagation constant y is given by
.gamma.=j {square root over (k.sub.0.sup.2+u.sub.2.sup.2)}, (9)
which is a complex expression. In all of the above Equations,
x = .sigma. 1 .omega. o , and ( 10 ) k o = .omega. .mu. o o , ( 11
) ##EQU00004##
where .mu..sub.0 comprises the permeability of free space,
.epsilon..sub.r comprises relative permittivity of Region 1. Thus,
the surface wave generated propagates parallel to the interface and
exponentially decays vertical to it. This is known as
evanescence.
[0035] Thus, Equations (1)-(3) may be considered to be a
cylindrically--symmetric, radially-propagating waveguide mode. See
Barlow, H. M., and Brown, J., Radio Surface Waves, Oxford
University Press, 1962, pp. 10-12, 29-33. The present disclosure
details structures that excite this "open boundary" waveguide mode.
Specifically, according to various embodiments, a polyphase
waveguide probe is provided with charge terminals of appropriate
size that are positioned relative to each other and are fed with
voltages and/or currents so as to excite the relative phasing of
the fields of the surface waveguide mode that is to be launched
along the boundary interface between Region 2 and Region 1.
[0036] To continue further, the Leontovich impedance boundary
condition between Region 1 and Region 2 is stated as
{circumflex over (n)}.times..sub.2(.rho.,.PHI.,0)=.sub.s, (12)
where {circumflex over (n)} is a unit normal in the positive
vertical (+z) direction and .sub.2 is the magnetic field strength
in Region 2 expressed by Equation (1) above. Equation (12) implies
that the fields specified in Equations (1)-(3) may be obtained by
driving a radial surface current density along the boundary
interface, such radial surface current density being specified
by
J.sub..rho.(.rho.')=-AH.sub.1.sup.(2)(-j.gamma..rho.') (13)
where A is a constant yet to be determined. Further, it should be
noted that close-in to the polyphase waveguide probe (for
.rho.<<.lamda.), Equation (13) above has the behavior
J close ( .rho. ' ) = - A ( j 2 ) .pi. ( - j .gamma..rho. ) = - H
.phi. = - I o 2 .pi..rho. ' . ( 14 ) ##EQU00005##
One may wish to note the negative sign. This means that when source
current flows vertically upward, the required "close-in" ground
current flows radially inward. By field matching on
H.sub..phi."close-in" we find that
A = - I o .gamma. 4 ( 15 ) ##EQU00006##
in Equations (1)-(6) and (13). Therefore, Equation (13) may be
restated as
J .rho. ( .rho. ' ) = I o .gamma. 4 H 1 ( 2 ) ( - j .gamma..rho. '
) . ( 16 ) ##EQU00007##
[0037] With reference then to FIG. 3, shown is an example of a
polyphase waveguide probe 200 that includes a charge terminal
T.sub.1 and a charge terminal T.sub.2 that are arranged along a
vertical axis z. The polyphase waveguide probe 200 is disposed
above a lossy conducting medium 203 according to an embodiment of
the present disclosure. The lossy conducting medium 203 makes up
Region 1 (FIG. 2) according to one embodiment. In addition, a
second medium 206 shares a boundary interface with the lossy
conducting medium 203 and makes up Region 2 (FIG. 2). The polyphase
waveguide probe 200 includes a probe coupling circuit 209 that
couples an excitation source 213 to the charge terminals T.sub.1
and T.sub.2 as is discussed in greater detail with reference to
later figures.
[0038] The charge terminals T.sub.1 and T.sub.2 are positioned over
the lossy conducting medium 203. The charge terminal T.sub.1 may be
considered a capacitor, and the charge terminal T.sub.2 may
comprise a counterpoise or lower capacitor as is described herein.
According to one embodiment, the charge terminal T.sub.1 is
positioned at height H.sub.1, and the charge terminal T.sub.2 is
positioned directly below T.sub.1 along the vertical axis z at
height H.sub.2, where H.sub.2 is less than H.sub.1. The height h of
the transmission structure presented by the polyphase waveguide
probe 200 is h=H.sub.1-H.sub.2. Given the foregoing discussion, one
can determine asymptotes of the radial Zenneck surface current on
the surface of the lossy conducting medium J.sub..rho.(.rho.) to be
J.sub.1(.rho.) close-in and J.sub.2(.rho.) far-out, where
Close - in ( .rho. < .lamda. / 8 ) : J .rho. ( .rho. ) .about. J
1 = I 1 + I 2 2 .pi..rho. + E .rho. QS ( Q 1 ) + E .rho. QS ( Q 2 )
Z .rho. , and ( 17 ) F ar - out ( .rho. .lamda. / 8 ) : J .rho. (
.rho. ) .about. J 2 = j .gamma..omega. Q 1 4 + 2 .gamma. .pi.
.times. e - ( .alpha. + j .beta. ) .rho. .rho. ( 18 )
##EQU00008##
where I.sub.1 is the conduction current feeding the charge Q.sub.1
on the first charge terminal T.sub.1, and I.sub.2 is the conduction
current feeding the charge Q.sub.2 on the second charge terminal
T.sub.2. The charge Q.sub.1 on the upper charge terminal T.sub.1 is
determined by Q.sub.1=C.sub.1V.sub.1, where C.sub.1 is the isolated
capacitance of the charge terminal T.sub.1. Note that there is a
third component to J.sub.1 set forth above given by
( E .rho. Q 1 ) Z .rho. , ##EQU00009##
which follows from the Leontovich boundary condition and is the
radial current contribution in the lossy conducting medium 203
pumped by the quasi-static field of the elevated oscillating charge
on the first charge terminal Q.sub.1. The quantity
Z .rho. = j .omega..mu. o .gamma. e ##EQU00010##
is the radial impedance of the lossy conducting medium, where
.gamma..sub.e=(j.omega..mu..sub.1.sigma..sub.1-.omega..sup.2.mu..sub.1.ep-
silon..sub.1).sup.1/2.
[0039] The asymptotes representing the radial current close-in and
far-out as set forth by Equations (17) and (18) are complex
quantities. According to various embodiments, a physical surface
current, J.sub.(r), is synthesized to match as close as possible
the current asymptotes in magnitude and phase. That is to say
close-in, |J.sub.(r)| is to be tangent to |J.sub.1|, and far-out
|J.sub.(r)| is to be tangent to |J.sub.2|. Also, according to the
various embodiments, the phase of J.sub.(r) should transition from
the phase of J.sub.1 close-in to the phase of J.sub.2 far-out.
[0040] According to one embodiment, if any of the various
embodiments of a polyphase waveguide probe described herein are
adjusted properly, this configuration will give at least an
approximate magnitude and phase match to the Zenneck mode and
launch Zenneck surface waves. It should be noted that the phase
far-out, .PHI..sub.2, is proportional to the propagation phase
corresponding to e.sup.-j.beta..rho.plus a fixed "phase boost" due
to the phase of {square root over (.gamma.)}which is arg( {square
root over (.gamma.)}),
j.PHI..sub.2(.rho.)=-j.beta..rho.+arg( {square root over
(.gamma.)}) (19)
where .gamma. is expressed in Equation (9) above, and depending on
the values for .epsilon..sub.r and .sigma. at the site of
transmission on the lossy conducting medium and the operating
frequency f, arg( {square root over (.gamma.)}), which has two
complex roots, is typically on the order of approximately
45.degree. or 225.degree.. Stated another way, in order to match
the Zenneck surface wave mode at the site of transmission to launch
a Zenneck surface wave, the phase of the surface current |J.sub.2|
far-out should differ from the phase of the surface current
close-in by the propagation phase corresponding to
e.sup.-j.beta.(.rho..sup.2.sup.-.rho..sup.1.sup.) plus a constant
of approximately 45 degrees or 225 degrees. This is because there
are two roots for {square root over (.gamma.)}, one near .pi./4 and
one near 5.pi./4. The properly adjusted synthetic radial surface
current is
J .rho. ( .rho. , .phi. , 0 ) = I o .gamma. 4 H 1 ( 2 ) ( - j
.gamma..rho. ) . ( 20 ) ##EQU00011##
By Maxwell's equations, such a J.sub.(.rho.) surface current
automatically creates fields that conform to
H .phi. = - .gamma. I o 4 e - u 2 z H 1 ( 2 ) ( - j .gamma..rho. )
, ( 21 ) E .rho. = - .gamma. I o 4 ( u 2 j .omega. o ) e - u 2 z H
1 ( 2 ) ( - j .gamma..rho. ) , and ( 22 ) E z = - .gamma. I o 4 ( -
.gamma. .omega. o ) e - u 2 z H 0 ( 2 ) ( - j .gamma..rho. ) . ( 23
) ##EQU00012##
Thus, the difference in phase between the surface current |J.sub.2|
far-out and the surface current |J.sub.1| close-in for the Zenneck
surface wave mode that is to be matched is due to the inherent
characteristics of the Hankel functions in Equations (20)-(23) set
forth above. It is of significance to recognize that the fields
expressed by Equations (1)-(6) and (20) have the nature of a
transmission line mode bound to a lossy interface, not radiation
fields such as are associated with groundwave propagation. See
Barlow, H. M. and Brown, J., Radio Surface Waves, Oxford University
Press, 1962, pp. 1-5. These fields automatically satisfy the
complex Brewster angle requirement for zero reflection, which means
that radiation is negligible, while surface guided wave propagation
is dramatically enhanced, as is verified and supported in the
experimental results provided below.
[0041] At this point, a review of the nature of the Hankel
functions used in Equations (20)-(23) is provided with emphasis on
a special property of these solutions of the wave equation. One
might observe that the Hankel functions of the first and second
kind and order n are defined as complex combinations of the
standard Bessel functions of the first and second kinds
H.sub.n.sup.(1)(x)=J.sub.n(x)+jN .sub.n(x) and (24)
H.sub.n.sup.(2)(x)=J.sub.n(x)-jN.sub.n(x). (25)
These functions represent cylindrical waves propagating radially
inward (superscript (1)) and outward (superscript (2)),
respectively. The definition is analogous to the relationship
e.sup..+-.jx=cos x.+-.j sin x. See, for example, Harrington, R. F.,
Time-Harmonic Fields, McGraw-Hill, 1961, pp. 460-463.
[0042] That H.sub.n.sup.(2)(k.sub..rho..rho.) is an outgoing wave
is easily recognized from its large argument asymptotic behavior
that is obtained directly from the series definitions of J.sub.n(x)
and N.sub.n(x),
H n ( 2 ) ( x ) x .fwdarw. .infin. 2 j .pi. x j n e - jx ( 26 )
##EQU00013##
which, when multiplied by e.sup.j.omega.t, is an outward
propagating cylindrical wave of the form e.sup.j(.omega.t-k.rho.)
with a 1/ .rho. spatial variation. The phase of the exponential
component is .psi.=(.omega.t-k.rho.). It is also evident that
H n ( 2 ) ( x ) x .fwdarw. .infin. j n H 0 ( 2 ) ( x ) , ( 27 )
##EQU00014##
and, a further useful property of Hankel functions is expressed
by
.differential. H 0 ( 2 ) ( x ) .differential. x = - H 1 ( 2 ) ( x )
, ( 28 ) ##EQU00015##
which is described by Jahnke, E., and F. Emde, Tables of Functions,
Dover, 1945, p. 145.
[0043] In addition, the small argument and large argument
asymptotes of the outward propagating Hankel function are as
follows:
H 1 ( 2 ) ( x ) x .fwdarw. 0 2 j .pi. x ( 29 ) H 1 ( 2 ) ( x ) x
.fwdarw. .infin. j 2 j .pi. x e - jx = 2 .pi. x e - j ( x - .pi. 2
- .pi. 4 ) . ( 30 ) ##EQU00016##
[0044] Note that these asymptotic expressions are complex
quantities. Also, unlike ordinary sinusoidal functions, the
behavior of complex Hankel functions differs in-close and far-out
from the origin. When x is a real quantity, Equations (29) and (30)
differ in phase by {square root over (j)}, which corresponds to an
extra phase advance or "phase boost" of 45.degree. or, equivalently
.lamda./8.
[0045] With reference to FIG. 4, to further illustrate the phase
transition between J.sub.1 (FIGS. 3) and J.sub.2 (FIG. 3) shown is
an illustration of the phases of the surface currents J.sub.1
close-in and J.sub.2 far-out relative to a position of a polyphase
waveguide probe 200 (FIG. 3). As shown in FIG. 4, there are three
different observation points P.sub.0, P.sub.1, and P.sub.2. A
transition region is located between the observation point P.sub.1
and observation point P.sub.2. The observation point P.sub.0 is
located at the position of the polyphase waveguide probe 200. The
observation point P.sub.1 is positioned "close-in" at a distance Ri
from the observation point P.sub.0 that places the observation
point P.sub.1 between the transition region 216 and the observation
point P.sub.0. The observation point P.sub.2 is positioned
"far-out" at a distance R.sub.2 from the observation point P.sub.0
beyond the transition region 216 as shown.
[0046] At observation point P.sub.0, the magnitude and phase of the
radial current J is expressed as |J.sub.P.sub.2|<.phi..sub.0. At
observation point P.sub.1, the magnitude and phase of the radial
current J is expressed as
|J.sub.P.sub.1|<.phi..sub.0-.beta.R.sub.1, where the phase shift
of .beta.R.sub.1 is attributable to the distance R.sub.1 between
the observation points P.sub.0 and P.sub.1. At observation point
P.sub.2, the magnitude and phase of the radial current J is
expressed as
|J.sub.P.sub.2|<.phi..sub.0-.beta.R.sub.1.phi..sub..DELTA. where
the phase shift of .beta.R.sub.1+.phi..sub..DELTA.is attributable
to the distance R.sub.2 between the observation points P.sub.0 and
P.sub.2 AND an additional phase shift that occurs in the transition
region 216. The additional phase shift .phi..sub..DELTA. occurs as
a property of the Hankel function as mentioned above.
[0047] The foregoing reflects the fact that the polyphase waveguide
probe 200 generates the surface current J.sub.1 close-in and then
transitions to the J.sub.2 current far-out. In the transition
region 216, the phase of the Zenneck surface-waveguide mode
transitions by approximately 45 degrees or 1/8.lamda. This
transition or phase shift may be considered a "phase boost" as the
phase of the Zenneck surface-waveguide mode appears to advance by
45 degrees in the transition region 216. The transition region 216
appears to occur somewhere less than 1/10 of a wavelength of the
operating frequency.
[0048] Referring back to FIG. 3, according to one embodiment, a
polyphase waveguide probe may be created that will launch the
appropriate radial surface current distribution. According to one
embodiment, a Zenneck waveguide mode is created in a radial
direction. If the J.sub.(r) given by Equation (20) can be created,
it will automatically launch Zenneck surface waves.
[0049] In addition, further discussion is provided regarding the
charge images Q.sub.1' and Q.sub.2' of the charges Q.sub.1 and
Q.sub.2 on the charge terminals T.sub.1 and T.sub.2 of one example
polyphase waveguide probe shown in FIG. 3. Analysis with respect to
the lossy conducting medium assumes the presence of induced
effective image charges Q.sub.1' and Q.sub.2' beneath the polyphase
waveguide probes coinciding with the charges Q.sub.1 and Q.sub.2 on
the charge reservoirs T.sub.1 and T.sub.2 as described herein. Such
image charges Q.sub.1' and Q.sub.2' must also be considered in the
analysis. These image charges Q.sub.1' and Q.sub.2' are not merely
180.degree. out of phase with the primary source charges Q.sub.1
and Q.sub.2 on the charge reservoirs T.sub.1 and T.sub.2, as they
would be in the case of a perfect conductor. A lossy conducting
medium such as, for example, a terrestrial medium presents phase
shifted images. That is to say, the image charges Q.sub.1' and
Q.sub.2' are at complex depths. For a discussion of complex images,
reference is made to Wait, J. R., "Complex Image
Theory--Revisited," IEEE Antennas and Propagation Magazine, Vol.
33, No. 4, August 1991, pp. 27-29, which is incorporated herein by
reference in its entirety.
[0050] Instead of the image charges Q.sub.1' and Q.sub.2' being at
a depth that is equal to the height of the charges Q.sub.1 and
Q.sub.2 (i.e. z.sub.n'=-h.sub.n), a conducting mirror 215 is placed
at depth z=-d/2 and the image itself appears at a "complex
distance" (i.e., the "distance" has both magnitude and phase),
given by z.sub.n'=-D.sub.n=-(d+h.sub.n).noteq.-h.sub.n, where n=1,
2, and for vertically polarized sources,
d = 2 .gamma. e 2 + k 0 2 .gamma. e 2 .apprxeq. 2 .gamma. e = d r +
jd i = d .angle..zeta. , ( 31 ) ##EQU00017##
where
.gamma..sub.e.sup.2=j.omega..mu..sub.1.sigma..sub.1-.omega..sup.2.mu..su-
b.1.epsilon..sub.1, and (32)
k.sub.0=.omega. {square root over (.mu..sub.0.epsilon..sub.0)},
(33)
[0051] The complex spacing of image charges Q.sub.1' and Q.sub.2',
in turn, implies that the external fields will experience extra
phase shifts not encountered when the interface is either a
lossless dielectric or a perfect conductor. The essence of the
lossy dielectric image-theory technique is to replace the finitely
conducting Earth (or lossy dielectric) by a perfect conductor
located at a complex depth, z=-d/2. Next, a source image is then
located at a complex depth D.sub.n=d/2+d/2+h.sub.n=d+h.sub.n, where
n=1, 2. Thereafter, one can calculate the fields above ground
(z.gtoreq.0) using a superposition of the physical charge (at z=+h)
plus its image (at z'=-D). The charge images Q.sub.1' and Q.sub.2'
at complex depths actually assist in obtaining the desired current
phases specified in Equations (20) and (21) above.
[0052] From Equations (2) and (3) above, it is noted that the ratio
of E.sub.2z to E.sub.2.rho.in Region 2 is given by
E 2 z E 2 .rho. = A ( - .gamma. .omega. 0 ) e - u 2 z H 0 ( 2 ) ( -
j .gamma. .rho. ) A ( .mu. 2 j .omega. 0 ) e - u 2 z H 1 ( 2 ) ( -
j .gamma. .rho. ) = ( - j .gamma. u 2 ) H 0 ( 2 ) ( - j .gamma.
.rho. ) H 1 ( 2 ) ( - j .gamma. .rho. ) . ( 34 ) ##EQU00018##
Also, it should be noted that asymptotically,
H n ( 2 ) ( x ) .fwdarw. x .fwdarw. .infin. j n H 0 ( 2 ) ( x ) . (
35 ) ##EQU00019##
Consequently, it follows directly from Equations (2) and (3)
that
E 2 z E 2 .rho. = r - jx = n = tan .psi. i , B , ( 36 )
##EQU00020##
where .psi..sub.i,B is the complex Brewster angle. By adjusting
source distributions and synthesizing complex Brewster angle
illumination at the surface of a lossy conducting medium 203,
Zenneck surface waves may be excited.
[0053] With reference to FIG. 5, shown is an incident field E
polarized parallel to a plane of incidence. The electric field
vector E is to be synthesized as an incoming non-uniform plane
wave, polarized parallel to the plane of incidence. The electric
field vector E may be created from independent horizontal and
vertical components as:
(.theta..sub.0)=E.sub..rho.{circumflex over
(.rho.)}+E.sub.z{circumflex over (z)}. (37)
Geometrically, the illustration in FIG. 5 suggests:
E .rho. ( .rho. , z ) = E ( .rho. , z ) cos .psi. 0 , and ( 38 a )
E z ( .rho. , z ) = E ( .rho. , z ) cos ( .pi. 2 - .psi. 0 ) = E (
.rho. , z ) sin .psi. 0 , ( 38 b ) ##EQU00021##
which means that the field ratio is
E z E .rho. = tan .psi. 0 . ( 39 ) ##EQU00022##
However, recall that from Equation (36),
tan .theta..sub.i,B= {square root over (.epsilon..sub.r-jx)}
(40)
so that, for a Zenneck surface wave, we desire
.psi..sub.o=.theta..sub.i,B which results in
E z E .rho. = tan .psi. 0 = r - j .sigma. .omega. 0 . ( 41 )
##EQU00023##
[0054] The Equations mean that if one controls the magnitude of the
complex field ratio and the relative phase between the incident
vertical and horizontal components E.sub.z and E.sub..rho.in a
plane parallel to the plane of incidence, then the synthesized
E-field vector will effectively be made to be incident at a complex
Brewster angle. Such a circumstance will synthetically excite a
Zenneck surface wave over the interface between Region 1 and Region
2.
[0055] With reference to FIG. 6, shown is another view of the
polyphase waveguide probe 200 disposed above a lossy conducting
medium 203 according to an embodiment of the present disclosure.
The lossy conducting medium 203 makes up Region 1 (FIG. 2)
according to one embodiment. In addition, a second medium 206
shares a boundary interface with the lossy conducting medium 203
and makes up Region 2 (FIG. 2).
[0056] According to one embodiment, the lossy conducting medium 203
comprises a terrestrial medium such as the planet Earth. To this
end, such a terrestrial medium comprises all structures or
formations included thereon whether natural or man-made. For
example, such a terrestrial medium may comprise natural elements
such as rock, soil, sand, fresh water, sea water, trees,
vegetation, and all other natural elements that make up our planet.
In addition, such a terrestrial medium may comprise man-made
elements such as concrete, asphalt, building materials, and other
man-made materials. In other embodiments, the lossy conducting
medium 203 may comprise some medium other than the Earth, whether
naturally occurring or man-made. In other embodiments, the lossy
conducting medium 203 may comprise other media such as man-made
surfaces and structures such as automobiles, aircraft, man-made
materials (such as plywood, plastic sheeting, or other materials)
or other media.
[0057] In the case that the lossy conducting medium 203 comprises a
terrestrial medium or Earth, the second medium 206 may comprise the
atmosphere above the ground. As such, the atmosphere may be termed
an "atmospheric medium" that comprises air and other elements that
make up the atmosphere of the Earth. In addition, it is possible
that the second medium 206 may comprise other media relative to the
lossy conducting medium 203.
[0058] The polyphase waveguide probe 200 comprises a pair of charge
terminals T.sub.1 and T.sub.2. Although two charge terminals
T.sub.1 and T.sub.2 are shown, it is understood that there may be
more than two charge terminals T.sub.1 and T.sub.2. According to
one embodiment, the charge terminals T.sub.1 and T.sub.2are
positioned above the lossy conducting medium 203 along a vertical
axis z that is normal to a plane presented by the lossy conducting
medium 203. In this respect, the charge terminal T.sub.1 is placed
directly above the charge terminal T.sub.2 although it is possible
that some other arrangement of two or more charge terminals T.sub.N
may be used. According to various embodiments, charges Q.sub.1 and
Q.sub.2 may be imposed on the respective charge terminals T.sub.1
and T.sub.2.
[0059] The charge terminals T.sub.1 and/or T.sub.2 may comprise any
conductive mass that can hold an electrical charge. The charge
terminal T.sub.1 has a self-capacitance C.sub.1, and the charge
terminal T.sub.2 has a self-capacitance C.sub.2. The charge
terminals T.sub.1 and/or T.sub.2 may comprise any shape such as a
sphere, a disk, a cylinder, a cone, a torus, a randomized shape, or
any other shape. Also note that the charge terminals T.sub.1 and
T.sub.2 need not be identical, but each can have a separate size
and shape, and be comprises of different conducting materials.
According to one embodiment, the shape of the charge terminal
T.sub.1 is specified to hold as much charge as practically
possible. Ultimately, the field strength of a Zenneck surface wave
launched by a polyphase waveguide probe 200 is directly
proportional to the quantity of charge on the terminal T.sub.1.
[0060] If the charge terminals T.sub.1 and/or T.sub.2 are spheres
or disks, the respective self-capacitance C.sub.1 and C.sub.2 can
be calculated. For example, the self-capacitance of an isolated
conductive sphere is C=4.pi..epsilon..sub.0r, where r comprises the
radius of the sphere in meters. The self-capacitance of an isolated
disk is C=8.epsilon..sub.0r, where r comprises the radius of the
disk in meters.
[0061] Thus, the charge Q.sub.1 stored on the charge terminal
T.sub.1 may be calculated as Q.sub.1=C.sub.1V, given the
self-capacitance C.sub.1 of the charge reservoir T.sub.1 and
voltage V that is applied to the charge terminal T.sub.1.
[0062] With further reference to FIG. 6, according to one
embodiment, the polyphase waveguide probe 200 comprises a probe
coupling circuit 209 that is coupled to the charge terminals
T.sub.1 and T.sub.2. The probe coupling circuit 209 facilitates
coupling the excitation source 213 to the charge terminals T.sub.1
and T.sub.2, and facilitates generating respective voltage
magnitudes and phases on the charge terminals T.sub.1 and T.sub.2
for a given frequency of operation. If more than two charge
terminals T.sub.N are employed, then the probe coupling circuit 209
would be configured to facilitate the generation of various voltage
magnitudes and phases on the respective charge terminals T.sub.N
relative to each other. In the embodiment of the polyphase
waveguide probe 200, the probe coupling circuit 209 comprises
various circuit configurations as will be described.
[0063] In one embodiment, the probe coupling circuit 209 is
specified so as to make the polyphase waveguide probe 200
electrically half-wave resonant. This imposes a voltage +V on a
first one of the terminals T.sub.1 or T.sub.2, and a -V on the
second one of the charge terminals T.sub.1 or T.sub.2 at any given
time. In such case, the voltages on the respective charge terminals
T.sub.1 and T.sub.2 are 180 degrees out of phase as can be
appreciated. In the case that the voltages on the respective charge
terminals T.sub.1 and T.sub.2 are 180 degrees out of phase, the
largest voltage magnitude differential is experienced on the charge
terminals T.sub.1 and T.sub.2. Alternatively, the probe coupling
circuit 209 may be configured so that the phase differential
between the charge terminals T.sub.1 and T.sub.2 is other than 180
degrees. To this end, the probe coupling circuit 209 may be
adjusted to alter the voltage magnitudes and phases during
adjustment of the polyphase waveguide probe 200.
[0064] By virtue of the placement of the charge terminal T.sub.1
directly above the charge terminal T.sub.2, a mutual capacitance
C.sub.M is created between the charge terminals T.sub.1 and
T.sub.2. Also, the charge terminal T.sub.1 has self-capacitance
C.sub.1, and the charge terminal T.sub.2 has a self-capacitance
C.sub.2 as mentioned above. There may also be a bound capacitance
between the charge terminal T.sub.1 and the lossy conducting medium
203, and a bound capacitance between the charge terminal T.sub.2
and the lossy conducting medium 203, depending on the respective
heights of the charge terminals T.sub.1 and T.sub.2. The mutual
capacitance C.sub.M depends on the distance between the charge
terminals T.sub.1 and T.sub.2.
[0065] Ultimately, the field strength generated by the polyphase
waveguide probe 200 will be directly proportional to the magnitude
of the charge Q.sub.1 that is imposed on the upper terminal
T.sub.1. The charge Q.sub.1 is, in turn, proportional to the
self-capacitance C.sub.1 associated with the charge terminal
T.sub.1 since Q.sub.1=C.sub.1V, where V is the voltage imposed on
the charge terminal T.sub.1.
[0066] According to one embodiment, an excitation source 213 is
coupled to the probe coupling circuit 209 in order to apply a
signal to the polyphase waveguide probe 200. The excitation source
213 may be any suitable electrical source such as a voltage or
current source capable of generating the voltage or current at the
operating frequency that is applied to the polyphase waveguide
probe 200. To this end, the excitation source 213 may comprise, for
example, a generator, a function generator, transmitter, or other
electrical source.
[0067] In one embodiment, the excitation source 213 may be coupled
to the polyphase waveguide probe 200 by way of magnetic coupling,
capacitive coupling, or conductive (direct tap) coupling as will be
described. In some embodiments, the probe coupling circuit 209 may
be coupled to the lossy conducting medium 203. Also, in various
embodiments, the excitation source 213 maybe coupled to the lossy
conducting medium 203 as will be described.
[0068] In addition, it should be noted that, according to one
embodiment, the polyphase waveguide probe 200 described herein has
the property that its radiation resistance R.sub.r is very small or
even negligible. One should recall that radiation resistance
R.sub.r is the equivalent resistance that would dissipate the same
amount of power that is ultimately radiated from an antenna.
According to the various embodiments, the polyphase waveguide probe
200 launches a Zenneck surface wave that is a guided
electromagnetic wave. According to the various embodiments, the
polyphase waveguide probes described herein have little radiation
resistance R.sub.r because the height of such polyphase waveguide
probes is usually small relative to their operating wavelengths.
Stated another way, according to one embodiment, the polyphase
waveguide probes described herein are "electrically small." As
contemplated herein, the phrase "electrically small" is defined as
a structure such as the various embodiments of polyphase waveguide
probes described herein that can be physically bounded by a sphere
having a radius equal to .lamda./2.pi., where .lamda. is the
free-space wavelength. See Fujimoto, K., A. Henderson, K. Hirasawa,
and J. R. James, Small Antennas, Wiley, 1987, p. 4.
[0069] To discuss further, the radiation resistance R.sub.r for a
short monopole antenna is expressed by
R r = 160 .pi. 2 ( h .lamda. ) 2 ( 42 ) ##EQU00024##
where the short monopole antenna has a height h with a uniform
current distribution, and where A is the wavelength at the
frequency of operation. See Stutzman, W. L. et al., "Antenna Theory
and Design," Wiley & Sons, 1981, p. 93.
[0070] Given that the value of the radiation resistance R.sub.r is
determined as a function of
( h .lamda. ) 2 , ##EQU00025##
it follows that if the height h of the structure is small relative
to the wavelength of the operating signal at the operating
frequency, then the radiation resistance R.sub.r will also be
small. As one example, if the height h of the transmission
structure is 10% of the wavelength of the operating signal at the
operating frequency, then the resulting value of
( h .lamda. ) 2 ##EQU00026##
would be (0.1).sup.2=0.01. It would follow that the radiation
resistance R.sub.r is correspondingly small.
[0071] Thus, according to various embodiments, if the effective
height h of the transmission structure is less than or equal to
.lamda. 2 .pi. , ##EQU00027##
where .lamda. is the wavelength at the operating frequency, then
the radiation resistance R.sub.r will be relatively small. For the
various embodiments of polyphase waveguide probes 200 described
below, the height h of the transmission structure may be calculated
as h=H.sub.1-H.sub.2, where H.sub.1 is the height of the charge
terminal T.sub.1, and H.sub.2 is the height of the charge terminal
T.sub.2. It should be appreciated that the height h of the
transmission structure for each embodiment of the polyphase
waveguide probes 200 described herein can be determined in a
similar manner.
[0072] While
.lamda. 2 .pi. ##EQU00028##
is provided as one benchmark, it is understood that the ratio of
the height h of the transmission structure over the wavelength of
the operating signal at the operating frequency can be any value.
However, it is understood that, at a given frequency of operation,
as the height of a given transmission structure increases, the
radiation resistance R.sub.r will increase accordingly.
[0073] Depending on the actual values for the height h and the
wavelength of the operating signal at the operating frequency, it
is possible that the radiation resistance R.sub.r may be of a value
such that some amount of radiation may occur along with the
launching of a Zenneck surface wave. To this end, the polyphase
waveguide probe 200 can be constructed to have a short height h
relative to the wavelength at the frequency of operation so as to
ensure that little or substantially zero energy is lost in the form
of radiation.
[0074] In addition, the placement of the charge reservoirs T.sub.1
and T.sub.2 along the vertical axis z provides for symmetry in the
Zenneck surface wave that is launched by the polyphase waveguide
probe 200 as described by the Hankel functions in Equations
(20)-(23) set forth above. Although the polyphase waveguide probe
200 is shown with two charge reservoirs T.sub.1 and T.sub.2 along
the vertical axis z that is normal to a plane making up the surface
of the lossy conducting medium 203, it is understood that other
configurations may be employed that will also provide for the
desired symmetry. For example, additional charge reservoirs T.sub.N
may be positioned along the vertical axis z, or some other
arrangement may be employed. In some embodiments, symmetry of
transmission may not be desired. In such case, the charge
reservoirs T.sub.N may be arranged in a configuration other than
along a vertical axis z to provide for an alternative transmission
distribution pattern.
[0075] When properly adjusted to operate at a predefined operating
frequency, the polyphase waveguide probe 200 generates a Zenneck
surface wave along the surface of the lossy conducting medium 203.
To this end, an excitation source 213 may be employed to generate
electrical energy at a predefined frequency that is applied to the
polyphase waveguide probe 200 to excite the structure. The energy
from the excitation source 213 is transmitted in the form of a
Zenneck surface wave by the polyphase waveguide probe 200 to one or
more receivers that are also coupled to the lossy conducting medium
203 or that are located within an effective transmission range of
the polyphase waveguide probe 200. The energy is thus conveyed in
the form of a Zenneck surface wave which is a surface-waveguide
mode or a guided electromagnetic field. In the context of modern
power grids using high voltage lines, a Zenneck surface wave
comprises a transmission line mode.
[0076] Thus, the Zenneck surface wave generated by the polyphase
waveguide probe 200 is not a radiated wave, but a guided wave in
the sense that these terms are described above. The Zenneck surface
wave is launched by virtue of the fact that the polyphase waveguide
probe 200 creates electromagnetic fields that are substantially
mode-matched to a Zenneck surface wave mode on the surface of the
lossy conducting medium 203. When the electromagnetic fields
generated by the polyphase waveguide probe 200 are substantially
mode-matched as such, the electromagnetic fields substantially
synthesize a wave front incident at a complex Brewster angle of the
lossy conducting medium 203 that results in little or no
reflection. Note that if the polyphase waveguide probe 200 is not
substantially mode-matched to the Zenneck surface wave mode, then a
Zenneck surface wave will not be launched since the complex
Brewster angle of the lossy conducting medium 203 would not have
been attained.
[0077] In the case that the lossy conducting medium 203 comprises a
terrestrial medium such as the Earth, the Zenneck surface wave mode
will depend upon the dielectric permittivity Er and conductivity a
of the site at which the polyphase waveguide probe 200 is located
as indicated above in Equations (1)-(11). Thus, the phase of the
Hankel functions in Equations (20)-(23) above depends on these
constitutive parameters at the launch site and on the frequency of
operation.
[0078] In order to excite the fields associated with the Zenneck
surface wave mode, according to one embodiment, the polyphase
waveguide probe 200 substantially synthesizes the radial surface
current density on the lossy conducting medium of the Zenneck
surface wave mode as is expressed by Equation (20) set forth above.
When this occurs, the electromagnetic fields are then substantially
or approximately mode-matched to a Zenneck surface wave mode on the
surface of the lossy conducting medium 203. To this end, the match
should be as close as is practicable. According to one embodiment,
this Zenneck surface wave mode to which the electromagnetic fields
are substantially matched is expressed in Equations (21)-(23) set
forth above.
[0079] In order to synthesize the radial surface current density in
the lossy conducting medium of the Zenneck surface wave mode, the
electrical characteristics of the polyphase waveguide probe 200
should be adjusted to impose appropriate voltage magnitudes and
phases on the charge terminals T.sub.1 and T.sub.2 for a given
frequency of operation and given the electrical properties of the
site of transmission. If more than two charge terminals T.sub.N are
employed, then appropriate voltage magnitudes and phases would need
to be imposed on the respective charge terminals T.sub.N, where N
may even be a very large number effectively comprising a continuum
of charge terminals.
[0080] In order to obtain the appropriate voltage magnitudes and
phases for a given design of a polyphase waveguide probe 200 at a
given location, an iterative approach may be used. Specifically,
analysis may be performed of a given excitation and configuration
of a polyphase waveguide probe 200 taking into account the feed
currents to the terminals T.sub.1 and T.sub.2, the charges on the
charge terminals T.sub.1 and T.sub.2, and their images in the lossy
conducting medium 203 in order to determine the radial surface
current density generated. This process may be performed
iteratively until an optimal configuration and excitation for a
given polyphase waveguide probe 200 is determined based on desired
parameters. To aid in determining whether a given polyphase
waveguide probe 200 is operating at an optimal level, a guided
field strength curve 103 (FIG. 1) may be generated using Equations
(1)-(11) above based on values for the conductivity of Region 1
(.sigma..sub.1) and the permittivity of Region 1 (.epsilon..sub.1)
at the location of the polyphase waveguide probe 200. Such a guided
field strength curve 103 will provide a benchmark for operation
such that measured field strengths can be compared with the
magnitudes indicated by the guided field strength curve 103 to
determine if optimal transmission has been achieved.
[0081] In order to arrive at an optimized polyphase waveguide probe
200, various parameters associated with the polyphase waveguide
probe 200 may be adjusted. Stated another way, the various
parameters associated with the polyphase waveguide probe 200 may be
varied to adjust the polyphase waveguide probe 200 to a desired
operating configuration.
[0082] One parameter that may be varied to adjust the polyphase
waveguide probe 200 is the height of one or both of the charge
terminals T.sub.1 and/or T.sub.2 relative to the surface of the
lossy conducting medium 203. In addition, the distance or spacing
between the charge terminals T.sub.1 and T.sub.2 may also be
adjusted. In doing so, one may minimize or otherwise alter the
mutual capacitance C.sub.M or any bound capacitances between the
charge terminals T.sub.1 and T.sub.2 and the lossy conducting
medium 203 as can be appreciated.
[0083] Alternatively, another parameter that can be adjusted is the
size of the respective charge terminals T.sub.1 and/or T.sub.2. By
changing the size of the charge terminals T.sub.1 and/or T.sub.2,
one will alter the respective self-capacitances C.sub.1 and/or
C.sub.2, and the mutual capacitance C.sub.M as can be appreciated.
Also, any bound capacitances that exist between the charge
terminals T.sub.1 and T.sub.2 and the lossy conducting medium 203
will be altered. In doing so, the voltage magnitudes and phases on
the charge terminals T.sub.1 and T.sub.2 are altered.
[0084] Still further, another parameter that can be adjusted is the
probe coupling circuit 209 associated with the polyphase waveguide
probe 200. This may be accomplished by adjusting the size of the
inductive and/or capacitive reactances that make up the probe
coupling circuit 209. For example, where such inductive reactances
comprise coils, the number of turns on such coils may be adjusted.
Ultimately, the adjustments to the probe coupling circuit 209 can
be made to alter the electrical length of the probe coupling
circuit 209, thereby affecting the voltage magnitudes and phases on
the charge terminals T.sub.1 and T.sub.2.
[0085] It is also the case that one can adjust the frequency of an
excitation source 213 applied to the polyphase waveguide probe 200
to optimize the transmission of a Zenneck surface wave. However, if
one wishes to transmit at a given frequency, other parameters would
need to be adjusted to optimize transmission.
[0086] Note that the iterations of transmission performed by making
the various adjustments may be implemented by using computer models
or by adjusting physical structures as can be appreciated. In one
approach, a field meter tuned to the transmission frequency may be
placed an appropriate distance from the polyphase waveguide probe
200 and adjustments may be made as set forth above until a maximum
or any other desired field strength of a resulting Zenneck surface
wave is detected. To this end, the field strength may be compared
with a guided field strength curve 103 (FIG. 1) generated at the
desired operating frequency and voltages on the terminals T.sub.1
and T.sub.2. According to one approach, the appropriate distance
for placement of such a field meter may be specified as greater
than the transition region 216 (FIG. 4) in the "far-out" region
described above where the surface current J.sub.2 dominates.
[0087] By making the above adjustments, one can create
corresponding "close-in" surface current J.sub.1 and "far-out"
surface current J.sub.2 that approximate the same currents
J.sub.(r) of the Zenneck surface wave mode specified in Equations
(17) and (18) set forth above. In doing so, the resulting
electromagnetic fields would be substantially or approximately
mode-matched to a Zenneck surface wave mode on the surface of the
lossy conducting medium 203.
[0088] Referring next to FIGS. 7A through 7J, shown are additional
examples of polyphase waveguide probes 200, denoted herein as
polyphase waveguide probes 200a-j, according to various embodiments
of the present disclosure. The polyphase waveguide probes 200a-j
each include a different probe coupling circuit 209, denoted herein
as probe coupling circuits 209a-j, according to various
embodiments. While several examples of probe coupling circuits
209a-j are described, it is understood that these embodiments are
merely examples and that there may be many other probe coupling
circuits 209 not set forth herein that may be employed to provide
for the desired voltage magnitudes and phases on the charge
terminals T.sub.1 and T.sub.2 according to the principles set forth
herein in order to facilitate the launching of Zenneck surface
waves.
[0089] In addition, each of the probe coupling circuits 209a-j may
employ, but are not limited to, inductive impedances comprising
coils. Even though coils are used, it is understood that other
circuit elements, both lumped and distributed, may be employed as
reactances. Also, other circuit elements may be included in the
probe coupling circuits 209a-j beyond those illustrated herein. In
addition, it is noted that the various polyphase waveguide probes
200a-j with their respective probe coupling circuits 209a-j are
merely described herein to provide examples. To this end, there may
be many other polyphase waveguide probes 200 that employ various
probe coupling circuits 209 and other circuitry that can be used to
launch Zenneck surface waves according to the various principles
set forth herein.
[0090] Referring now to FIG. 7A, shown is a first example of the
polyphase waveguide probe 200 (FIG. 6), denoted herein as polyphase
waveguide probe 200a, according to one embodiment. The polyphase
waveguide probe 200a includes the charge terminals T.sub.1 and
T.sub.2 that are positioned along a vertical axis z that is
substantially normal to the plane presented by the lossy conducting
medium 203. The second medium 206 is above the lossy conducting
medium 203. The charge terminal T.sub.1 has a self-capacitance
C.sub.1, and the charge terminal T.sub.2 has a self-capacitance
C.sub.2. During operation, charges Q.sub.1 and Q.sub.2 are imposed
on the charge terminals T.sub.1 and T.sub.2, respectively,
depending on the voltages applied to the charge terminals T.sub.1
and T.sub.2 at any given instant. A mutual capacitance C.sub.M may
exist between the charge terminals T.sub.1 and T.sub.2 depending on
the distance there between. In addition, bound capacitances may
exist between the respective charge terminals T.sub.1 and T.sub.2
and the lossy conducting medium 203 depending on the heights of the
respective charge terminals T.sub.1 and T.sub.2 with respect to the
lossy conducting medium 203.
[0091] The polyphase waveguide probe 200a includes a probe coupling
circuit 209a that comprises an inductive impedance comprising a
coil L.sub.1a having a pair of leads that are coupled to respective
ones of the charge terminals T.sub.1 and T.sub.2. In one
embodiment, the coil L.sub.1a is specified to have an electrical
length that is one-half (1/2) of the wavelength at the operating
frequency of the polyphase waveguide probe 200a.
[0092] While the electrical length of the coil L.sub.1a is
specified as approximately one-half (1/2) the wavelength at the
operating frequency, it is understood that the coil L.sub.1a may be
specified with an electrical length at other values. According to
one embodiment, the fact that the coil L.sub.1a has an electrical
length of approximately one-half the wavelength at the operating
frequency provides for an advantage in that a maximum voltage
differential is created on the charge terminals T.sub.1 and
T.sub.2. Nonetheless, the length or diameter of the coil L.sub.1a
may be increased or decreased when adjusting the polyphase
waveguide probe 200a to obtain optimal excitation of a Zenneck
surface wave mode. Alternatively, it may be the case that the
inductive impedance is specified to have an electrical length that
is significantly less than or greater than 1/2 the wavelength at
the operating frequency of the polyphase waveguide probe 200a.
[0093] According to one embodiment, the excitation source 213 is
coupled to the probe coupling circuit 209 by way of magnetic
coupling. Specifically, the excitation source 213 is coupled to a
coil L.sub.P that is inductively coupled to the coil L.sub.1a. This
may be done by link coupling, a tapped coil, a variable reactance,
or other coupling approach as can be appreciated. To this end, the
coil L.sub.P acts as a primary, and the coil L.sub.1a acts as a
secondary as can be appreciated.
[0094] In order to adjust the polyphase waveguide probe 200a for
the transmission of a desired Zenneck surface wave, the heights of
the respective charge terminals T.sub.1 and T.sub.2 may be altered
with respect to the lossy conducting medium 203 and with respect to
each other. Also, the sizes of the charge terminals T.sub.1 and
T.sub.2 may be altered. In addition, the size of the coil L.sub.1a
may be altered by adding or eliminating turns or by changing some
other dimension of the coil L.sub.1a.
[0095] Based on experimentation with the polyphase waveguide probe
200a, this appears to be the easiest of the polyphase waveguide
probes 200a-j to adjust and to operate to achieve a desired
efficiency.
[0096] Referring now to FIG. 7B, shown is an example of the
polyphase waveguide probe 200 (FIG. 6), denoted herein as polyphase
waveguide probe 200b, according to one embodiment. The polyphase
waveguide probe 200b includes the charge terminals T.sub.1 and
T.sub.2 that are positioned along a vertical axis z that is
substantially normal to the plane presented by the lossy conducting
medium 203. The second medium 206 is above the lossy conducting
medium 203. The charge terminals T.sub.1 and T.sub.2 are positioned
along a vertical axis z to provide for cylindrical symmetry in the
resulting Zenneck surface wave as was described above. The charge
terminal T.sub.1 has a self-capacitance C.sub.1, and the charge
terminal T.sub.2 has a self-capacitance C.sub.2. During operation,
charges Q.sub.1 and Q.sub.2 are imposed on the charge terminals
T.sub.1 and T.sub.2, respectively, depending on the voltages
applied to the charge terminals T.sub.1 and T.sub.2 at any given
instant. A mutual capacitance C.sub.M may exist between the charge
terminals T.sub.1 and T.sub.2 depending on the distance there
between. In addition, bound capacitances may exist between the
respective charge terminals T.sub.1 and T.sub.2 and the lossy
conducting medium 203 depending on the heights of the respective
charge terminals T.sub.1 and T.sub.2 with respect to the lossy
conducting medium 203.
[0097] The polyphase waveguide probe 200b also includes a probe
coupling circuit 209b comprising a first coil L.sub.1b and a second
coil L.sub.2b. The first coil L.sub.1b is coupled to each of the
charge terminals T.sub.1 and T.sub.2 as shown. The second coil
L.sub.2b is coupled to the charge terminal T.sub.2 and to the lossy
conducting medium 203.
[0098] The excitation source 213 is magnetically coupled to the
probe coupling circuit 209b in a manner similar as was mentioned
with respect to the polyphase waveguide probe 200a (FIG. 7A) set
forth above. To this end, the excitation source 213 is coupled to a
coil L.sub.P that acts as a primary and the coil L.sub.1b acts as a
secondary. Alternatively, the coil L.sub.2b may act as a secondary
as well.
[0099] In order to adjust the polyphase waveguide probe 200b for
the transmission of a desired Zenneck surface wave, the heights of
the respective charge terminals T.sub.1 and T.sub.2 may be altered
with respect to the lossy conducting medium 203 and with respect to
each other. Also, the sizes of the charge terminals T.sub.1 and
T.sub.2 may be altered. In addition, the size of each of the coils
L.sub.1b and L.sub.2b may be altered by adding or eliminating turns
or by changing some other dimension of the respective coils
L.sub.1b or L.sub.2b.
[0100] Referring now to FIG. 7C, shown is another example of the
polyphase waveguide probe 200 (FIG. 6), denoted herein as polyphase
waveguide probe 200c, according to one embodiment. The polyphase
waveguide probe 200c includes the charge terminals T.sub.1 and
T.sub.2 that are positioned along a vertical axis z that is
substantially normal to the plane presented by the lossy conducting
medium 203. The second medium 206 is above the lossy conducting
medium 203. The charge terminal T.sub.1 has a self-capacitance
C.sub.1, and the charge terminal T.sub.2 has a self-capacitance
C.sub.2. During operation, charges Q.sub.1 and Q.sub.2 are imposed
on the charge terminals T.sub.1 and T.sub.2, respectively,
depending on the voltages applied to the charge terminals T.sub.1
and T.sub.2 at any given instant. A mutual capacitance C.sub.M may
exist between the charge terminals T.sub.1 and T.sub.2 depending on
the distance there between. In addition, bound capacitances may
exist between the respective charge terminals T.sub.1 and T.sub.2
and the lossy conducting medium 203 depending on the heights of the
respective charge terminals T.sub.1 and T.sub.2 with respect to the
lossy conducting medium 203.
[0101] The polyphase waveguide probe 200c also includes a probe
coupling circuit 209c comprising a coil L.sub.1c. One end of the
coil L.sub.1c is coupled to the charge terminal T.sub.1 as shown.
The second end of the coil L.sub.1c is coupled to the lossy
conducting medium 203. A tap that is coupled to the charge terminal
T.sub.2 is positioned along the coil L.sub.1c.
[0102] The excitation source 213 is magnetically coupled to the
probe coupling circuit 209c in a manner similar as was mentioned
with respect to the polyphase waveguide probe 200a (FIG. 7A) set
forth above. To this end, the excitation source 213 is coupled to a
coil L.sub.P that acts as a primary and the coil L.sub.1c acts as a
secondary. The coil L.sub.P can be positioned at any location along
the coil L.sub.1c.
[0103] In order to adjust the polyphase waveguide probe 200c for
the excitation and transmission of a desired Zenneck surface wave,
the heights of the respective charge terminals T.sub.1 and T.sub.2
may be altered with respect to the lossy conducting medium 203 and
with respect to each other. Also, the sizes of the charge terminals
T.sub.1 and T.sub.2 may be altered. In addition, the size of the
coil L.sub.1c may be altered by adding or eliminating turns, or by
changing some other dimension of the coil L.sub.1c. In addition,
the inductance presented by the portions of the coil L.sub.1c above
and below the tap may be adjusted by moving the position of the
tap.
[0104] Referring now to FIG. 7D, shown is yet another example of
the polyphase waveguide probe 200 (FIG. 6), denoted herein as
polyphase waveguide probe 200d, according to one embodiment. The
polyphase waveguide probe 200d includes the charge terminals
T.sub.1 and T.sub.2 that are positioned along a vertical axis z
that is substantially normal to the plane presented by the lossy
conducting medium 203. The second medium 206 is above the lossy
conducting medium 203. The charge terminal T.sub.1 has a
self-capacitance C.sub.1, and the charge terminal T.sub.2 has a
self-capacitance C.sub.2. During operation, charges Q.sub.1 and
Q.sub.2 are imposed on the charge terminals T.sub.1 and T.sub.2,
respectively, depending on the voltages applied to the charge
terminals T.sub.1 and T.sub.2 at any given instant. A mutual
capacitance C.sub.M may exist between the charge terminals T.sub.1
and T.sub.2 depending on the distance there between. In addition,
bound capacitances may exist between the respective charge
terminals T.sub.1 and T.sub.2 and the lossy conducting medium 203
depending on the heights of the respective charge terminals T.sub.1
and T.sub.2 with respect to the lossy conducting medium 203.
[0105] The polyphase waveguide probe 200d also includes a probe
coupling circuit 209d comprising a first coil L.sub.1d and a second
coil L.sub.2d. A first lead of the first coil L.sub.1d is coupled
to the charge terminal T.sub.1, and the second lead of the first
coil L.sub.1d is coupled to the lossy conducting medium 203. A
first lead of the second coil L.sub.2d is coupled to the charge
terminal T.sub.2, and the second lead of the second coil L.sub.2d
is coupled to the lossy conducting medium 203.
[0106] The excitation source 213 is magnetically coupled to the
probe coupling circuit 209d in a manner similar as was mentioned
with respect to the polyphase waveguide probe 200a (FIG. 7A) set
forth above. To this end, the excitation source 213 is coupled to a
coil L.sub.P that acts as a primary and the coil L.sub.2d acts as a
secondary. Alternatively, the coil L.sub.1d may act as a secondary
as well.
[0107] In order to adjust the polyphase waveguide probe 200d for
the excitation and transmission of a desired Zenneck surface wave,
the heights of the respective charge terminals T.sub.1 and T.sub.2
may be altered with respect to the lossy conducting medium 203 and
with respect to each other. Also, the sizes of the charge terminals
T.sub.1 and T.sub.2 may be altered. In addition, the size of each
of the coils L.sub.1d and L.sub.2d may be altered by adding or
eliminating turns or by changing some other dimension of the
respective coils L.sub.1d or L.sub.2d.
[0108] Referring now to FIG. 7E, shown is still another example of
the polyphase waveguide probe 200 (FIG. 6), denoted herein as
polyphase waveguide probe 200e, according to one embodiment. The
polyphase waveguide probe 200e includes the charge terminals
T.sub.1 and T.sub.2 that are positioned along a vertical axis z
that is substantially normal to the plane presented by the lossy
conducting medium 203. The second medium 206 is above the lossy
conducting medium 203. The charge terminals T.sub.1 and T.sub.2 are
positioned along a vertical axis z to provide for cylindrical
symmetry in the resulting Zenneck surface wave as was described
above. The charge terminal T.sub.1 has a self-capacitance C.sub.1,
and the charge terminal T.sub.2 has a self-capacitance C.sub.2.
During operation, charges Q.sub.1 and Q.sub.2 are imposed on the
charge terminals T.sub.1 and T.sub.2, respectively, depending on
the voltages applied to the charge terminals T.sub.1 and T.sub.2 at
any given instant. A mutual capacitance C.sub.M may exist between
the charge terminals T.sub.1 and T.sub.2 depending on the distance
there between. In addition, bound capacitances may exist between
the respective charge terminals T.sub.1 and T.sub.2 and the lossy
conducting medium 203 depending on the heights of the respective
charge terminals T.sub.1 and T.sub.2 with respect to the lossy
conducting medium 203.
[0109] The polyphase waveguide probe 200e also includes a probe
coupling circuit 209e comprising a first coil L.sub.1e and a
resistor R.sub.2. A first lead of the first coil L.sub.1e is
coupled to the charge terminal T.sub.1, and the second lead of the
first coil L.sub.1e is coupled to the lossy conducting medium 203.
A first lead of the resistor R.sub.2 is coupled to the charge
terminal T.sub.2, and the second lead of the resistor R.sub.2 is
coupled to the lossy conducting medium 203.
[0110] The excitation source 213 is magnetically coupled to the
probe coupling circuit 209e in a manner similar as was mentioned
with respect to the polyphase waveguide probe 200a (FIG. 7A) set
forth above. To this end, the excitation source 213 is coupled to a
coil L.sub.P that acts as a primary and the coil L.sub.1e acts as a
secondary.
[0111] In order to adjust the polyphase waveguide probe 200e for
the transmission of a desired Zenneck surface wave, the heights of
the respective charge terminals T.sub.1 and T.sub.2 may be altered
with respect to the lossy conducting medium 203 and with respect to
each other. Also, the sizes of the charge terminals T.sub.1 and
T.sub.2 may be altered. In addition, the size of the coil L.sub.1e
may be altered by adding or eliminating turns or by changing some
other dimension of the respective coils L.sub.1e. In addition, the
magnitude of the resistance R.sub.2 may be adjusted as well.
[0112] Referring now to FIG. 7F, shown is a further example of the
polyphase waveguide probe 200 (FIG. 6), denoted herein as polyphase
waveguide probe 200f, according to one embodiment. The polyphase
waveguide probe 200f includes a charge terminal T.sub.1 and a
ground screen G that acts as a second charge terminal. The charge
terminal T.sub.1 and the ground screen G are positioned along a
vertical axis z that is substantially normal to the plane presented
by the lossy conducting medium 203. The second medium 206 is above
the lossy conducting medium 203. Note that to calculate the height
h of the transmission structure, the height H.sub.2 of the ground
screen G is subtracted from the height H.sub.1 of the charge
terminal T.sub.1.
[0113] The charge terminal T.sub.1 has a self-capacitance C.sub.1,
and the ground screen G has a self-capacitance C.sub.2. During
operation, charges Q.sub.1 and Q.sub.2 are imposed on the charge
terminal T.sub.1 and the ground screen G, respectively, depending
on the voltages applied to the charge terminal T.sub.1 and the
ground screen G at any given instant. A mutual capacitance C.sub.M
may exist between the charge terminal T.sub.1 and the ground screen
G depending on the distance there between. In addition, bound
capacitances may exist between the charge terminal T.sub.1 and/or
the ground screen G and the lossy conducting medium 203 depending
on the heights of the charge terminal T.sub.1 and the ground screen
G with respect to the lossy conducting medium 203. Generally, a
bound capacitance will exist between the ground screen G and the
lossy conducting medium 203 due to its proximity to the lossy
conducting medium 203.
[0114] The polyphase waveguide probe 200f includes a probe coupling
circuit 209f that is made up of an inductive impedance comprising a
coil L.sub.1f having a pair of leads that are coupled to the charge
terminal T.sub.1 and ground screen G. In one embodiment, the coil
L.sub.1f is specified to have an electrical length that is one-half
(1/2) of the wavelength at the operating frequency of the polyphase
waveguide probe 200f.
[0115] While the electrical length of the coil L.sub.1f is
specified as approximately one-half (1/2) the wavelength at the
operating frequency, it is understood that the coil L.sub.1f may be
specified with an electrical length at other values. According to
one embodiment, the fact that the coil L.sub.1f has an electrical
length of approximately one-half the wavelength at the operating
frequency provides for an advantage in that a maximum voltage
differential is created on the charge terminal T.sub.1 and the
ground screen G. Nonetheless, the length or diameter of the coil
L.sub.1f may be increased or decreased when adjusting the polyphase
waveguide probe 200f to obtain optimal transmission of a Zenneck
surface wave. Alternatively, it may be the case that the inductive
impedance is specified to have an electrical length that is
significantly less than or greater than 1/29the wavelength at the
operating frequency of the polyphase waveguide probe 200f.
[0116] According to one embodiment, the excitation source 213 is
coupled to the probe coupling circuit 209f by way of magnetic
coupling. Specifically, the excitation source 213 is coupled to a
coil L.sub.P that is inductively coupled to the coil L.sub.1f. This
may be done by link coupling, a phasor/coupling network, or other
approach as can be appreciated. To this end, the coil L.sub.P acts
as a primary, and the coil L.sub.1f acts as a secondary as can be
appreciated.
[0117] In order to adjust the polyphase waveguide probe 200a for
the launching and transmission of a desired Zenneck surface wave,
the heights of the respective charge terminals T.sub.1 and T.sub.2
may be altered with respect to the lossy conducting medium 203 and
with respect to each other. Also, the sizes of the charge terminals
T.sub.1 and T.sub.2 may be altered. In addition, the size of the
coil L.sub.1f may be altered by adding or eliminating turns or by
changing some other dimension of the coil L.sub.1f.
[0118] Referring now to FIG. 7G, shown is another example of the
polyphase waveguide probe 200 (FIG. 6), denoted herein as polyphase
waveguide probe 200g, according to one embodiment. The polyphase
waveguide probe 200g includes the charge terminals T.sub.1 and
T.sub.2 that are positioned along a vertical axis z that is
substantially normal to the plane presented by the lossy conducting
medium 203. The second medium 206 is above the lossy conducting
medium 203. The charge terminals T.sub.1 and T.sub.2 are positioned
along a vertical axis z to provide for cylindrical symmetry in the
resulting Zenneck surface wave as was described above. The charge
terminal T.sub.1 has a self-capacitance C.sub.1, and the charge
terminal T.sub.2 has a self-capacitance C.sub.2. During operation,
charges Q.sub.1 and Q.sub.2 are imposed on the charge terminals
T.sub.1 and T.sub.2, respectively, depending on the voltages
applied to the charge terminals T.sub.1 and T.sub.2 at any given
instant. A mutual capacitance C.sub.M may exist between the charge
terminals T.sub.1 and T.sub.2 depending on the distance there
between. In addition, bound capacitances may exist between the
respective charge terminals T.sub.1 and T.sub.2 and the lossy
conducting medium 203 depending on the heights of the respective
charge terminals T.sub.1 and T.sub.2 with respect to the lossy
conducting medium 203.
[0119] The polyphase waveguide probe 200g also includes a probe
coupling circuit 209g comprising a first coil L.sub.1g, a second
coil L.sub.2g, and a variable capacitor C.sub.v The first coil
L.sub.1g is coupled to each of the charge terminals T.sub.1 and
T.sub.2 as shown. The second coil L.sub.2g has a first lead that is
coupled to a variable capacitor C.sub.v and a second lead that is
coupled to the lossy conducting medium 203. The variable capacitor
C.sub.v, in turn, is coupled to the charge terminal T.sub.2 and the
first coil L.sub.1g.
[0120] The excitation source 213 is magnetically coupled to the
probe coupling circuit 209g in a manner similar as was mentioned
with respect to the polyphase waveguide probe 200a (FIG. 7A) set
forth above. To this end, the excitation source 213 is coupled to a
coil L.sub.P that acts as a primary and either the coil L.sub.1g or
the coil L.sub.2g may act as a secondary.
[0121] In order to adjust the polyphase waveguide probe 200g for
the launching and transmission of a desired Zenneck surface wave,
the heights of the respective charge terminals T.sub.1 and T.sub.2
may be altered with respect to the lossy conducting medium 203 and
with respect to each other. Also, the sizes of the charge terminals
T.sub.1 and T.sub.2 may be altered. In addition, the size of each
of the coils L.sub.1g and L.sub.2g may be altered by adding or
eliminating turns or by changing some other dimension of the
respective coils L.sub.1g or L.sub.2g. In addition, the variable
capacitance C.sub.v may be adjusted.
[0122] Referring now to FIG. 7H, shown is yet another example of
the polyphase waveguide probe 200 (FIG. 6), denoted herein as
polyphase waveguide probe 200h, according to one embodiment. The
polyphase waveguide probe 200h includes the charge terminals
T.sub.1 and T.sub.2 that are positioned along a vertical axis z
that is substantially normal to the plane presented by the lossy
conducting medium 203. The second medium 206 is above the lossy
conducting medium 203. The charge terminal T.sub.1 has a
self-capacitance C.sub.1, and the charge terminal T.sub.2 has a
self-capacitance C.sub.2. During operation, charges Q.sub.1 and
Q.sub.2 are imposed on the charge terminals T.sub.1 and T.sub.2,
respectively, depending on the voltages applied to the charge
terminals T.sub.1 and T.sub.2 at any given instant. A mutual
capacitance C.sub.M may exist between the charge terminals T.sub.1
and T.sub.2 depending on the distance there between. In addition,
bound capacitances may exist between the respective charge
terminals T.sub.1 and T.sub.2 and the lossy conducting medium 203
depending on the heights of the respective charge terminals T.sub.1
and T.sub.2 with respect to the lossy conducting medium 203.
[0123] The polyphase waveguide probe 200h also includes a probe
coupling circuit 209h comprising a first coil L.sub.1h and a second
coil L.sub.2h. The first lead of the first coil L.sub.1h is coupled
to the charge terminal T.sub.1, and the second lead of the first
coil L.sub.1h is coupled to the second charge terminal T.sub.2. A
first lead of the second coil L.sub.2h is coupled to a terminal
T.sub.T, and the second lead of the second coil L.sub.2h is coupled
to the lossy conducting medium 203. The terminal T.sub.T is
positioned relative to the charge terminal T.sub.2 such that a
coupling capacitance C.sub.c exists between the charge terminal
T.sub.2 and the terminal T.sub.T.
[0124] The excitation source 213 is magnetically coupled to the
probe coupling circuit 209h in a manner similar as was mentioned
with respect to the polyphase waveguide probe 200a (FIG. 7A) set
forth above. To this end, the excitation source 213 is coupled to a
coil L.sub.P that acts as a primary and the coil L.sub.2h acts as a
secondary. Alternatively, the coil L.sub.1h may act as a secondary
as well.
[0125] In order to adjust the polyphase waveguide probe 200h for
the launching and transmission of a desired Zenneck surface wave,
the heights of the respective charge terminals T.sub.1 and T.sub.2
may be altered with respect to the lossy conducting medium 203 and
with respect to each other. Also, the sizes of the charge terminals
T.sub.1 and T.sub.2 may be altered. In addition, the size of each
of the coils L.sub.1h and L.sub.2h may be altered by adding or
eliminating turns or by changing some other dimension of the
respective coils L.sub.1h or L.sub.2h. Also the spacing between the
charge terminal T.sub.2 and the terminal T.sub.T may be altered,
thereby modifying the coupling capacitance C.sub.c as can be
appreciated.
[0126] Referring now to FIG. 71, shown is yet another example of
the polyphase waveguide probe 200 (FIG. 6), denoted herein as
polyphase waveguide probe 200i, according to one embodiment. The
polyphase waveguide probe 200i is very similar to the polyphase
waveguide probe 200h (FIG. 7H) except for the fact that the
excitation source 213 is series-coupled to the probe coupling
circuit 209i as will be described.
[0127] To this end, the polyphase waveguide probe 200i includes the
charge terminals T.sub.1 and T.sub.2 that are positioned along a
vertical axis z that is substantially normal to the plane presented
by the lossy conducting medium 203. The second medium 206 is above
the lossy conducting medium 203. The charge terminal T.sub.1 has a
self-capacitance C.sub.1, and the charge terminal T.sub.2 has a
self-capacitance C.sub.2. During operation, charges Q.sub.1 and
Q.sub.2 are imposed on the charge terminals T.sub.1 and T.sub.2,
respectively, depending on the voltages applied to the charge
terminals T.sub.1 and T.sub.2 at any given instant. A mutual
capacitance C.sub.M may exist between the charge terminals T.sub.1
and T.sub.2 depending on the distance there between. In addition,
bound capacitances may exist between the respective charge
terminals T.sub.1 and T.sub.2 and the lossy conducting medium 203
depending on the heights of the respective charge terminals T.sub.1
and T.sub.2 with respect to the lossy conducting medium 203.
[0128] The polyphase waveguide probe 200i also includes a probe
coupling circuit 209i comprising a first coil L.sub.1i and a second
coil L.sub.2i. The first lead of the first coil L.sub.1c i is
coupled to the charge terminal T.sub.1, and the second lead of the
first coil L.sub.1i is coupled to the second charge terminal
T.sub.2. A first lead of the second coil L.sub.2i is coupled to a
terminal T.sub.T, and the second lead of the second coil L.sub.2i
is coupled to an output of the excitation source 213. Also, a
ground lead of the excitation source 213 is coupled to the lossy
conducting medium 203. The terminal T.sub.T is positioned relative
to the charge terminal T.sub.2 such that a coupling capacitance
C.sub.c exists between the charge terminal T.sub.2 and the terminal
T.sub.T.
[0129] The polyphase waveguide probe 200i provides one example
where the excitation source 213 is series-coupled to the probe
coupling circuit 209i as mentioned above. Specifically, the
excitation source 213 is coupled between the coil L.sub.2i and the
lossy conducting medium 203.
[0130] In order to adjust the polyphase waveguide probe 200i for
the launching and transmission of a desired Zenneck surface wave,
the heights of the respective charge terminals T.sub.1 and T.sub.2
may be altered with respect to the lossy conducting medium 203 and
with respect to each other. Also, the sizes of the charge terminals
T.sub.1 and T.sub.2 may be altered. In addition, the size of each
of the coils L.sub.1i and L.sub.2i may be altered by adding or
eliminating turns or by changing some other dimension of the
respective coils L.sub.1i or L.sub.2i. Also the spacing between The
charge terminal T.sub.2 and the terminal T.sub.T may be altered,
thereby modifying the coupling capacitance C.sub.c as can be
appreciated.
[0131] Referring now to FIG. 7J, shown is an example of the
polyphase waveguide probe 200 (FIG. 6), denoted herein as polyphase
waveguide probe 200j, according to one embodiment. The polyphase
waveguide probe 200j includes the charge terminals T.sub.1 and
T.sub.2 that are positioned along a vertical axis z that is
substantially normal to the plane presented by the lossy conducting
medium 203. The second medium 206 is above the lossy conducting
medium 203. In this embodiment, the charge terminal T.sub.1
comprises a sphere and the charge terminal T.sub.2 comprises a
disk. In this respect, the polyphase waveguide probe 200j provides
an illustration that the charge terminals T.sub.N may comprise any
shape.
[0132] The charge terminal T.sub.1 has a self-capacitance C.sub.1,
and the charge terminal T.sub.2 has a self-capacitance C.sub.2.
During operation, charges Q.sub.1 and Q.sub.2 are imposed on the
charge terminals T.sub.1 and T.sub.2, respectively, depending on
the voltages applied to the charge terminals T.sub.1 and T.sub.2 at
any given instant. A mutual capacitance C.sub.M may exist between
the charge terminals T.sub.1 and T.sub.2 depending on the distance
there between. In addition, bound capacitances may exist between
the respective charge terminals T.sub.1 and T.sub.2 and the lossy
conducting medium 203 depending on the heights of the respective
charge terminals T.sub.1 and T.sub.2 with respect to the lossy
conducting medium 203.
[0133] The polyphase waveguide probe 200j includes a probe coupling
circuit 209j comprising an inductive impedance comprising a coil
L.sub.1j having a pair of leads that are coupled to respective ones
of the charge terminals T.sub.1 and T.sub.2. In one embodiment, the
coil L.sub.1j is specified to have an electrical length that is
one-half (1/2) of the wavelength at the operating frequency of the
polyphase waveguide probe 200j. l While the electrical length of
the coil L.sub.1j is specified as approximately one-half (1/2) the
wavelength at the operating frequency, it is understood that the
coil L.sub.1j may be specified with an electrical length at other
values as was discussed with reference to the polyphase waveguide
probe 200a (FIG. 7A) described above. In addition, the probe
coupling circuit 209j includes a tap 223 on the coil L.sub.1j that
is coupled to the lossy conducting medium 203.
[0134] The excitation source 213 is magnetically coupled to the
probe coupling circuit 209j in a manner similar as was mentioned
with respect to the polyphase waveguide probe 200a (FIG. 7A) set
forth above. To this end, the excitation source 213 is coupled to a
coil L.sub.P that acts as a primary and the coil L.sub.1j acts as a
secondary. The coil L.sub.P can be positioned at any location along
the coil L.sub.1j. Also, the coil L.sub.P can be positioned above
or below the tap 223.
[0135] In order to adjust the polyphase waveguide probe 200j for
the launching and transmission of a desired Zenneck surface wave,
the heights of the respective charge terminals T.sub.1 and T.sub.2
may be altered with respect to the lossy conducting medium 203 and
with respect to each other. Also, the sizes of the charge terminals
T.sub.1 and T.sub.2 may be altered. In addition, the size of the
coil L.sub.1j may be altered by adding or eliminating turns or by
changing some other dimension of the coil L.sub.1j. Further, the
position of the tap 223 on the coil L.sub.1j may be adjusted.
[0136] With reference to the various embodiments of the polyphase
waveguide probes 200a-j in FIGS. 7A-J, each of the polyphase
waveguide probes 200a-j may be excited to transmit energy conveyed
in the form of a guided wave, or waveguide mode along the surface
of the lossy conducting medium 203. To facilitate such
transmission, the elements of each of the polyphase waveguide
probes 200a-j may be adjusted to impose a desired voltage magnitude
and phase on the respective charge terminals T.sub.1 and T.sub.2
when the respective polyphase waveguide probe 200a-j is excited.
Such excitation may occur by applying energy from an excitation
source 213 to the respective polyphase waveguide probe 200a-j as
was described above.
[0137] The voltage magnitudes and phases imposed on the charge
terminals T.sub.1 and T.sub.2 may be adjusted in order to
substantially synthesize the fields that are substantially
mode-matched to the guided or Zenneck surface-waveguide mode of the
lossy conducting medium 203 at the site of transmission given the
local permittivity .epsilon..sub.r, conductivity .sigma., and
potentially other parameters of the lossy conducting medium 203.
The waveguide mode of the surface-guided wave is expressed in
Equations (21), (22), and (23) set forth above. This
surface-waveguide mode has a radial surface current density
expressed in Equation (20) in Amperes per meter.
[0138] It is understood that it may be difficult to synthesize
fields that exactly match the surface-waveguide mode expressed in
Equations (21), (22), and (23) set forth above. However, a guided
surface wave may be launched if such fields at least approximate
the surface-waveguide mode. According to various embodiments, the
fields are synthesized to match the surface-waveguide mode within
an acceptable engineering tolerance so as to launch a guided
surface wave.
[0139] Likewise, it may be difficult to synthesize a radial surface
current density that exactly matches the radial surface current
density of the Zenneck surface-waveguide mode, where the
synthesized radial surface current density results from the
synthesized fields described above. According to various
embodiments, the polyphase waveguide probes 200 may be adjusted to
match the radial surface current density of the guided
surface-waveguide mode within an acceptable engineering tolerance
so as to launch a Zenneck surface wave mode. By creating specific
charge distributions plus their images at complex distances, the
various polyphase waveguide probes 200a-j set forth above excite
surface currents, the fields of which are designed to approximately
match a propagating Zenneck surface wave mode and a Zenneck surface
wave is launched. By virtue of this complex image technique
inherent in the various polyphase waveguide probes 200a-j described
above, one is able to substantially mode-match to the surface
waveguide modes that the guiding interface wants to support at the
location of transmission. The guiding interface is the interface
between Region 1 (FIG. 2) and Region 2 (FIG. 2) as described above.
According to one embodiment, the guiding interface is the interface
between the lossy conducting medium 203 presented by the Earth and
the atmospheric medium as described above.
[0140] When the voltage magnitudes and phases imposed on the charge
terminals T.sub.1 and T.sub.2 are adjusted so that they, plus their
effective images at complex depths, excite complex surface currents
whose fields synthesize the fields that substantially match the
Zenneck surface-waveguide mode of the lossy conducting medium 203
at the site of transmission, by virtue of the Leontovich boundary
condition, such fields will automatically substantially synthesize
a wave front incident at a complex Brewster angle of the lossy
conducting medium 203, resulting in zero reflection. This is the
condition of wave matching at the boundary.
[0141] Referring next to FIGS. 8A, 8B, and 8C, shown are examples
of graphs 300a, 300b, and 300c that depict field strength in Volts
per meter as a function of distance in kilometers for purposes of
comparison between a Zenneck surface wave and conventionally
radiated fields. In addition, the various graphs 300a, 300b, and
300c illustrate how the distance of transmission of a Zenneck
surface wave varies with the frequency of transmission.
[0142] Each graph 300a, 300b, and 300c depicts a corresponding
guided field strength curve 303a, 303b, and 303c and corresponding
radiated field strength curves 306a, 306b, and 306c. The guided
field strength curves 303a, 303b, and 303c were generated assuming
various parameters. Specifically, the graphs 300a, 300b, and 300c
were calculated with a constant charge Q.sub.1 (FIG. 3) applied to
the upper terminal T.sub.1 (FIG. 3) at frequencies of 10 MHz, 1
MHz, and 0.1 MHz, respectively. Constitutive parameters of
.epsilon..sub.r=15 and .sigma.=0.008 mhos/m, which are taken from
the R-3 map for central Ohio set forth by the Federal
Communications Commission (FCC), were assumed for purposes of
calculation. The following table provides the assumed polyphase
waveguide probe operating parameters for the generation of each of
the guided field strength curve 303a, 303b, and 303c.
TABLE-US-00001 Height of Self-Cap of Voltage on Frequency Terminal
T.sub.1 Terminal T.sub.1 Term T.sub.1 Charge Q.sub.1 Curve (MHz)
.lamda. (m) H/.lamda. (pF) (V) (Coulombs) 303a 10 30 0.8 0.027 50
10,000 5 .times. 10.sup.-7 303b 1.0 300 8 0.027 100 5,000 5 .times.
10.sup.-7 303c 0.1 3000 8 0.0027 100 5,000 5 .times. 10.sup.-7
[0143] In order to have physically realizable operation, the height
of terminal T.sub.1 was specified at H.sub.T1=8 meters for f=0.1
MHz and 1.0 MHz, but shortened to 0.8 meters at 10 MHz in order to
keep the current distribution uniform. Also, the self-capacitance
C.sub.1 of the terminal T.sub.1 was set to 100 pF for operation at
f=0.1 MHz and 1.0 MHz. This capacitance is unreasonably large for
use at 10 MHz, so the self-capacitance C.sub.1 was reduced for that
case. However, the resulting terminal charge Q.sub.T1 which is the
controlling parameter for field strength was kept at the same value
for all three guided field strength curve 303a, 303b, and 303c.
[0144] From the graphs it can be seen that the lower the frequency,
the less the propagation attenuation and the fields reach out over
greater distances. However, consistent with conservation of energy,
the energy density decreases with distance. Another way to state
that is that the higher the frequency, the smaller the region over
which the energy is spread, so the greater the energy density.
Thus, the "knee" of the Zenneck surface wave shrinks in range as
the frequency is increased. Alternatively, the lower the frequency,
the less the propagation attenuation and the greater the field
strength of the Zenneck surface wave at very large distances from
the site of transmission using a polyphase waveguide probe 200
(FIG. 6).
[0145] The Zenneck surface wave for each case is identified as
guided field strength curves 303a, 303b, and 303c, respectively.
The Norton ground wave field strength in Volts per meter for a
short vertical monopole antenna, of the same height as the
respective polyphase waveguide probe 200, with an assumed ground
loss of 10 ohms, is represented by the radiated field strength
curves 306a, 306b, and 306c, respectively. It is asserted that this
is a reasonably realistic assumption for monopole antenna
structures operating at these frequencies. The critical point is
that a properly mode-matched polyphase waveguide probe launches a
guided surface wave which dramatically outperforms the radiation
field of any monopole at distances out to just beyond the "knee" in
the guided field strength curves 303a-c of the respective Zenneck
surface waves.
[0146] Given the foregoing, according to one embodiment, the
propagation distance of a guided surface wave varies as a function
of the frequency of transmission. Specifically, the lower the
transmission frequency, the less the exponential attenuation of the
guided surface wave and, therefore, the farther the guided surface
wave will propagate. As mentioned above, the field strength of a
guided surface wave falls off at a rate of e.sup.-ad/ {square root
over (d)}, whereas the field strength of a radiated electromagnetic
field falls off geometrically, proportional to 1/d, where d is the
distance in kilometers. Thus, each of the guided field strength
curves 303a, 303b, and 303c feature a knee as was described above.
As the frequency of transmission of a polyphase waveguide probe
described herein decreases, the knee of the corresponding guided
field strength curve 303a, 303b, and 303c will push to the right in
the graph.
[0147] FIG. 8A shows a guided field strength curve 303a and a
radiated field strength curve 306a generated at a frequency of 10
Megahertz. As shown, the guided surface wave falls off below 10
kilometers. In FIG. 8B, the guided field strength curve 303b and
the radiated field strength curve 306b are generated at a frequency
of 1 Megahertz. The guided field strength curve 303b falls off at
approximately 100 Kilometers. Finally, in FIG. 8C, the guided field
strength curve 303c and the radiated field strength curve 306c are
generated at a frequency of 100 Kilohertz (which is 0.1 Megahertz).
The guided field strength curve 303c falls off at between 4000-7000
Kilometers.
[0148] Note that if the frequency is low enough, it may be possible
to transmit a guided surface wave around the entire Earth. It is
believed that such frequencies may be at or below approximately
20-25 kilohertz. It should be noted that at such low frequencies,
the lossy conducting medium 203 (FIG. 6) ceases to be a plane and
becomes a sphere. Thus, when a lossy conducting medium 203
comprises a terrestrial medium, the calculation of guided field
strength curves will be altered to take into account the spherical
shape at low frequencies where the propagation distances approach
size of the terrestrial medium.
[0149] Given the foregoing, next some general guidance is provided
in constructing a polyphase waveguide probe 200 (FIG. 6) using the
terrestrial medium of Earth as the lossy conducting medium 203
according to the various embodiments. As a practical approach, one
might specify the frequency of operation and identify the desired
field strength of the guided surface wave at a distance of interest
from the respective polyphase waveguide probe 200 to be
constructed.
[0150] Given these parameters, next one may determine the charge
Q.sub.1 (FIG. 6) that is to be imposed on the upper charge terminal
T.sub.1 (FIG. 6) in order to produce the desired field strength at
the specified distance. To determine the charge Q.sub.1 needed, one
would need to obtain the permittivity Er and the conductivity a of
the Earth at the transmission site. These values can be obtained by
measurement or by reference to conductivity charts published, for
example, by the Federal Communications Commission or the Committee
Consultif International Radio (CCIR). When the permittivity Er,
conductivity a, and the desired field strength at the specified
distance are known, the needed charge Q.sub.1 can be determined by
direct calculation of the field strength from Zenneck's exact
expressions set forth in Equations (21)-(23) above.
[0151] Once the needed charge Q.sub.1 is determined, next one would
need to identify what self-capacitance C.sub.1 of the charge
terminal T.sub.1 at what voltage V would produce the needed charge
Q.sub.1 on the charge terminal T.sub.1. A charge Q on any charge
terminal T is calculated as Q=CV. In one approach, one can choose
what is deemed to be an acceptable voltage V that can be placed on
the charge terminal T.sub.1, and then construct the charge terminal
T.sub.1 so as to have the required self-capacitance C.sub.1 to
achieve the needed charge Q.sub.1. Alternatively, in another
approach, one could determine what is an achievable
self-capacitance C.sub.1 by virtue of the specific construction of
the charge terminal T.sub.1, and then raise the resulting charge
terminal T.sub.1 to the required voltage V to achieve the needed
charge Q.sub.1.
[0152] In addition, there is an issue of operational bandwidth that
should be considered when determining the needed self-capacitance
C.sub.1 of the charge terminal T.sub.1 and voltage V to be imposed
on the charge terminal T.sub.1. Specifically, the bandwidth of the
polyphase waveguide probes 200 described herein is relatively
large. This results in a significant degree of flexibility in
specifying the self-capacitance C.sub.1 or the voltage V as
described above. However, it should be understood that as the
self-capacitance C.sub.1 is reduced and the voltage V increased,
the bandwidth of the resulting polyphase waveguide probe 200 will
diminish.
[0153] Experimentally, it should be noted that a smaller
self-capacitance C.sub.1 may make a given polyphase waveguide probe
200 more sensitive to small variations in the permittivity
.epsilon..sub.r or conductivity .sigma. of the Earth at or near the
transmission site. Such variation in the permittivity
.epsilon..sub.r or conductivity .sigma. might occur due to
variation in the climate given the transition between the seasons
or due to changes in local weather conditions such as the onset of
rain, drought, and/or other changes in local weather. Consequently,
according to one embodiment, the charge terminal T.sub.1 may be
specified so as to have a relatively large self-capacitance C.sub.1
as is practicable.
[0154] Once the self-capacitance C.sub.1 of the charge terminal
T.sub.1 and the voltage to be imposed thereon are determined, next
the self-capacitance C.sub.2 and physical location of the second
charge terminal T.sub.2 are to be determined. As a practical
matter, it has been found easiest to specify the self-capacitance
C.sub.2 of the charge terminal T.sub.2 to be the same as the
self-capacitance C.sub.1 of the charge terminal This may be
accomplished by making the size and shape of the charge terminal
T.sub.2 the same as the size and shape of the charge terminal
T.sub.1. This would ensure that symmetry is maintained and will
avoid the possibility of unusual phase shifts between the two
charge terminals T.sub.1 and T.sub.2 that might negatively affect
achieving a match with the complex Brewster angle as described
above. The fact that the self-capacitances C.sub.1 and C.sub.2 are
the same for both charge terminals T.sub.1 and T.sub.2 will result
in the same voltage magnitudes on the charge terminals T.sub.1 and
T.sub.2. However, it is understood that the self-capacitances
C.sub.1 and C.sub.2 may differ, and the shape and size of the
charge terminals T.sub.1 and T.sub.2 may differ.
[0155] To promote symmetry, the charge terminal T.sub.2 may be
positioned directly under the charge terminal T.sub.1 along the
vertical axis z (FIG. 6) as described above. Alternatively, it may
be possible to position the charge terminal T.sub.2 at some other
location with some resulting effect.
[0156] The distance between the charge terminals T.sub.1 and
T.sub.2 should be specified so as to provide for the best match
between the fields generated by the polyphase waveguide probe 200
and the guided surface-waveguide mode at the site of transmission.
As a suggested starting point, this distance may be set so that the
mutual capacitance C.sub.M (FIG. 6) between the charge terminals
T.sub.1 and T.sub.2 is the same or less than the isolated
capacitance C.sub.1 on the charge terminal Ultimately, one should
specify the distance between the charge terminals T.sub.1 and
T.sub.2 to make the mutual capacitance C.sub.M as small as is
practicable. The mutual capacitance C.sub.M may be determined by
measurement, and the charge terminals T.sub.1 and T.sub.2 may be
positioned accordingly.
[0157] Next, the proper height h=H.sub.1-H.sub.2 (FIGS. 7A-J) of
the polyphase waveguide probe 200 is determined. The so-called
"image complex-depth" phenomenon comes into bearing here. This
would entail consideration of the superposed fields on the surface
of the Earth from the charge reservoirs T.sub.1 and T.sub.2 that
have charges Q.sub.1 and Q.sub.2, and from the subsurface images of
the charges Q.sub.1 and Q.sub.2 as height h is varied. Due to the
significant number of variables to consider to ensure that a given
polyphase waveguide probe 200 is mode matched with the guided
surface-waveguide mode of the Earth at the site of transmission, a
practical starting point is a height h at which the bound
capacitance of each of the charge reservoirs T.sub.1 and T.sub.2
with respect to the ground is negligible such that the capacitance
associated with the charge terminals T.sub.1 and T.sub.2 is
essentially their isolated self-capacitance C.sub.1 and C.sub.2,
respectively.
[0158] Another consideration to take into account when determining
the height h associated with a polyphase waveguide probe 200 is
whether radiation is to be avoided. Specifically, as the height h
of the polyphase waveguide probe 200 approaches an appreciable
portion of a wavelength at the frequency of operation, the
radiation resistance R.sub.r will grow quadratically with height h
and radiation will begin to dominate over the generation of a
guided surface wave as described above. One benchmark set forth
above that ensures the Zenneck surface wave will dominate over any
radiation is to make sure the height h is less than 10% of the
wavelength at the frequency of operation, although other benchmarks
may be specified. In some cases, it may be desired to allow some
degree of radiation to occur in addition to launching a guided
surface wave, where the height h may be specified accordingly.
[0159] Next, the probe coupling circuit 209 (FIG. 6) is specified
to provide for the voltage phase between the charge terminals
T.sub.1 and T.sub.2. The voltage phase appears to have a
significant effect on creating fields that mode-match the guided
surface-waveguide mode at the site of transmission. Assuming that
the placement of the charge terminals T.sub.1 and T.sub.2 is along
the vertical z axis to promote symmetry, the probe coupling circuit
209 may be specified to provide for a voltage phase differential of
180 degrees on the charge terminals T.sub.1 and T.sub.2. That is to
say, the probe coupling circuit 209 is specified so that voltage V
on the charge terminal T.sub.1 is 180 degrees out of phase with
respect to the voltage on the charge terminal T.sub.2.
[0160] As was described above, one example approach is to place a
coil L.sub.1a (FIG. 7A) between the charge terminals T.sub.1 and
T.sub.2 as described above with reference to the polyphase
waveguide probe 200a and adjust the coil L.sub.1a until the
resulting system is electrically half-wave resonant. This would
place a voltage V on the charge terminal T.sub.1 and voltage--V on
the charge terminal T.sub.2 such that the largest voltages are
placed on the charge terminals T.sub.1 and T.sub.2 180 degrees out
of phase.
[0161] The excitation source 213 (FIG. 6) may then be coupled to
the probe coupling circuit 209 and the output voltage adjusted to
achieve the required voltage V to provide for the needed charge
Q.sub.1 as described above. The excitation source 213 may be
coupled via magnetic coupling, capacitive coupling, or conductive
coupling (direct) to the probe coupling circuit 209. Note that the
output of the excitation source 213 may be stepped up using a
transformer or via some other approach if necessary. The location
of the coil L.sub.1a can be at any location such as down on the
ground by the excitation source 213. Alternatively, as per best RF
practice, the coil L.sub.1a can be positioned directly between the
charge reservoirs T.sub.1 and T.sub.2. Principles of impedance
matching may be applied when coupling the excitation source 213 to
the probe coupling circuit 209.
[0162] Note that the phase differential does not necessarily have
to be 180 degrees. To this end, one has the option of raising and
lowering one or both of the charge terminals T.sub.1 and/or
T.sub.2, adjusting the voltages V on the charge terminals T.sub.1
and/or T.sub.2, or adjusting the probe coupling circuit 209 to
adjust the voltage magnitudes and phases to create fields that most
closely match the guided surface-waveguide mode in order to
generate a guided surface wave.
Experimental Results
[0163] The above disclosures are supported by experimental
measurements and documentation. With reference to FIG. 9, shown is
a graph that presents the measured field strength of an
electromagnetic field transmitted by one embodiment of an
experimental polyphase waveguide probe measured on Oct. 14, 2012 in
Plymouth, N.H. The frequency of transmission was 59 MHz with a
voltage of 60 mV imposed on the charge terminal T.sub.1 of the
experimental polyphase waveguide probe. The self-capacitance
C.sub.1 of the experimental polyphase waveguide probe was 8.5 pF.
The conductivity a of the ground at the test site is 0.0002 mhos/m,
and the permittivity Er of the ground at the test site was 5. These
values were measured in situ at the frequency in use.
[0164] The graph includes a guided field strength curve 400 that is
labeled a "Zenneck" curve at 80% efficiency and a radiated field
strength curve 403 that is labeled a "Norton" curve at 100%
radiation efficiency, which is the best possible. To this end, the
radiated field strength curve 403 represents the radiated
electromagnetic fields that would be generated by a 1/4 wavelength
monopole antenna operating at a frequency of 59 MHz. The circles
406 on the graph represent measured field strengths produced by the
experimental polyphase waveguide probe. The field strength
measurements were performed with a NIST-traceable Potomac
Instruments FIM-71 commercial VHF field strength meter. As can be
seen, the measured field strengths fall along the theoretical
guided field strength curve 400. These measured field strengths are
consistent with the propagation of a guided or Zenneck surface
wave.
[0165] Referring next to FIG. 10, shown is a graph that presents
the measured phase of the transmitted electromagnetic wave from the
experimental polyphase waveguide probe. The curve J(r) indicates
the phase of the fields incident to the currents J.sub.1 and
J.sub.2 with a transition between the currents J.sub.1 and J.sub.2
as shown. The curve 503 indicates the asymptote depicting the phase
of the current J.sub.1, and the curve 506 indicates the asymptote
depicting the phase of the current J.sub.2. A difference of
approximately 45 degrees generally exists between the phases of the
respective currents J.sub.1 and J.sub.2. The circles 509 indicate
measurements of the phase of the current J.sub.(r) generated by the
experimental polyphase waveguide probe operating at 59 MHz as with
FIG. 9. As shown, the circles 509 fall along the curve J.sub.(r)
indicating that there is a transition of the phase of the current
J.sub.(r) from the curve 503 to the curve 506. This indicates that
the phase of the current J.sub.(r) generated by the experimental
polyphase waveguide probe transitions from the phase generated by
the close-in current J.sub.1 to the far-out current J.sub.2. Thus,
these phase measurements are consistent with the phase with the
presence of a guided or Zenneck surface wave.
[0166] With reference to FIG. 11 shown is a graph of a second set
of measured data that depicts the field strength of an
electromagnetic field transmitted by a second embodiment of an
experimental polyphase waveguide probe measured on Nov. 1, 2003 in
the vicinity of Ashland, N.H. and across the region north of Lake
Winnipesaukee. The frequency of transmission was 1850 kHz with a
voltage of 1250 V imposed on the charge terminal T.sub.1 of the
experimental polyphase waveguide probe. The experimental polyphase
waveguide probe had a physical height of H.sub.1=2 meters. The
self-capacitance C.sub.1 of the experimental polyphase waveguide
probe in this experiment, which was a flat conducting disk of 1
meter radius, was measured to be 70 pF. The polyphase waveguide
probe was arranged as illustrated in FIG. 7J, with spacing h=1
meter and the height of the charge terminal T.sub.2 above the
ground (the lossy conducting medium 203) being H.sub.2=1 meter. The
average conductivity a of the ground in the vicinity of
experimentation was 0.006 mhos/m, and the relative permittivity
.epsilon..sub.r of the ground was on the order of 15. These were
determined at the frequency in use.
[0167] The graph includes a guided field strength curve 600 that is
launched by the experimental polyphase waveguide probe, labeled as
"Zenneck" curve at 85% efficiency, and a radiated field strength
curve 603 that is labeled a "Norton" curve as radiated from a
resonated monopole of the same height, H.sub.2=2 meters, over a
ground screen composed of 20 radial wires equally spaced and of
length 200 feet each. To this end, the radiated field strength
curve 603 represents the conventional Norton ground wave field
radiated from a conventional stub monopole antenna operating at a
frequency of 1850 kHz over the lossy Earth. The circles 606 on the
graph represent measured field strengths produced by the
experimental polyphase waveguide probe.
[0168] As can be seen, the measured field strengths fall closely
along the theoretical Zenneck guided field strength curve 600.
Special mention of the field strength measured at the r=7 mile
point may be made. This field strength data point was measured
adjacent to the shore of a lake, and this may account for the data
departing slightly above the theoretical Zenneck guided field
strength curve 600, i.e. the constitutive parameters,
.epsilon..sub.r and/or a, at that location are likely to have
departed significantly from the path-average constitutive
parameters.
[0169] The field strength measurements were performed with a
NIST-traceable Potomac Instruments FIM-41 MF/HF field strength
meter. The measured field strength data are consistent with the
presence of a guided or Zenneck surface wave. It is apparent from
the experimental data that the measured field strengths observed at
distances less than 15 miles could not possibly be due to
conventional Norton ground wave propagation, and can only be due to
guided surface wave propagation launched by the polyphase probe
operating as disclosed above. Under the given 1.85 MHz experimental
conditions, out at 20 miles it appears that a Norton ground wave
component has finally overtaken the Zenneck surface wave
component.
[0170] A comparison of the measured Zenneck surface wave data shown
in FIG. 9 at 59 MHz with the measured data in FIG. 11 at 1.85 MHz
illustrates the great advantage of employing a polyphase waveguide
probe according to the various embodiments at lower
frequencies.
[0171] These experimental data confirm that the present polyphase
waveguide probes, comprising a plurality of appropriately phased
and adjusted charge terminals, as taught herein, induce a
phase-advanced surface current with a unique phase boost of arg(
{square root over (.gamma.)}), and whose fields synthesize surface
illumination at the complex Brewster angle for the lossy boundary
as disclosed herein. The consequence is the efficient launching of
cylindrical Zenneck-like wave propagation, guided by the boundary
surface as an evanescent, single-conductor radial transmission-line
mode, which attenuates as not as
e - .alpha. d d , ##EQU00029##
not as a radiation field, which would decrease as 1/d due to
geometrical spreading.
[0172] Referring next to FIGS. 12A, 12B, and 13, shown are examples
of generalized receive circuits for using the surface-guided waves
in wireless power delivery systems. FIGS. 12A and 12B include a
linear probe 703 and a tuned resonator 706. FIG. 13 is a magnetic
coil 709 according to various embodiments of the present
disclosure. According to various embodiments, each one of the
linear probe 703, the tuned resonator 706, and the magnetic coil
709 may be employed to receive power transmitted in the form of a
guided surface wave on the surface of a lossy conducting medium 203
(FIG. 6) according to various embodiments. As mentioned above, in
one embodiment the lossy conducting medium 203 comprises a
terrestrial medium.
[0173] With specific reference to FIG. 12A, the open-circuit
terminal voltage at the output terminals 713 of the linear probe
703 depends upon the effective height of the linear probe 703. To
this end, the terminal point voltage may be calculated as
V.sub.T=.intg..sub.0.sup.h.sup.eE.sub.incdl, (43)
where E.sub.inc is the strength of the electric field in vector on
the linear probeb 703 in Volts per meter, dl is an element of
integration along the direction of the linear probe 703, and
h.sub.e is the effective height of the linear probe 703. An
electrical load 716 is coupled to the output terminals 713 through
an impedance matching network 719.
[0174] When the linear probe 703 is subjected to a guided surface
wave as described above, a voltage is developed across the output
terminals 713 that may be applied to the electrical load 716
through a conjugate impedance matching network 719 as the case may
be. In order to facilitate the flow of power to the electrical load
716, the electrical load 716 should be substantially impedance
matched to the linear probe 703 as will be described below.
[0175] Referring to FIG. 12B, the tuned resonator 706 includes a
charge terminal T.sub.R that is elevated above the lossy conducting
medium 203. The charge terminal T.sub.R has a self-capacitance
C.sub.R. In addition, there may also be a bound capacitance (not
shown) between the charge terminal T.sub.R and the lossy conducting
medium 203 depending on the height of the charge terminal T.sub.R
above the lossy conducting medium 203. The bound capacitance should
preferably be minimized as much as is practicable, although this
may not be entirely necessary in every instance of a polyphase
waveguide probe 200.
[0176] The tuned resonator 706 also includes a coil L.sub.R. One
end of the coil L.sub.R is coupled to the charge terminal T.sub.R,
and the other end of the coil L.sub.R is coupled to the lossy
conducting medium 203. To this end, the tuned resonator 706 (which
may also be referred to as tuned resonator L.sub.R-C.sub.R)
comprises a series-tuned resonator as the charge terminal C.sub.R
and the coil L.sub.R are situated in series. The tuned resonator
706 is tuned by adjusting the size and/or height of the charge
terminal T.sub.R, and/or adjusting the size of the coil L.sub.R so
that the reactive impedance of the structure is substantially
eliminated.
[0177] For example, the reactance presented by the self-capacitance
C.sub.R is calculated as
1 j .omega. C R . ##EQU00030##
Note that the total capacitance of the tuned resonator 706 may also
include capacitance between the charge terminal T.sub.R and the
lossy conducting medium 203, where the total capacitance of the
tuned resonator 706 may be calculated from both the
self-capacitance C.sub.R and any bound capacitance as can be
appreciated. According to one embodiment, the charge terminal
T.sub.R may be raised to a height so as to substantially reduce or
eliminate any bound capacitance. The existence of a bound
capacitance may be determined from capacitance measurements between
the charge terminal T.sub.R and the lossy conducting medium
203.
[0178] The inductive reactance presented by a discrete-element coil
L.sub.R may be calculated as j.omega.L, where L is the
lumped-element inductance of the coil L.sub.R. If the coil L.sub.R
is a distributed element, its equivalent terminal-point inductive
reactance may be determined by conventional approaches. To tune the
tuned resonator 706, one would make adjustments so that the
inductive reactance presented by the coil L.sub.R equals the
capacitive reactance presented by the tuned resonator 706 so that
the resulting net reactance of the tuned resonator 706 is
substantially zero at the frequency of operation. An impedance
matching network 723 may be inserted between the probe terminals
721 and the electrical load 726 in order to effect a
conjugate-match condition for maxim power transfer to the
electrical load 726.
[0179] When placed in the presence of a guided surface wave,
generated at 7the frequency of the tuned resonator 706 and the
conjugate matching network 723, as described above, maximum power
will be delivered from the surface guided wave to the electrical
load 726. That is, once conjugate impedance matching is established
between the tuned resonator 706 and the electrical load 726, power
will be delivered from the structure to the electrical load 726. To
this end, an electrical load 726 may be coupled to the tuned
resonator 706 by way of magnetic coupling, capacitive coupling, or
conductive (direct tap) coupling. The elements of the coupling
network may be lumped components or distributed elements as can be
appreciated. In the embodiment shown in FIG. 12B, magnetic coupling
is employed where a coil L.sub.s is positioned as a secondary
relative to the coil L.sub.R that acts as a transformer primary.
The coil L.sub.s may be link coupled to the coil L.sub.R by
geometrically winding it around the same core structure and
adjusting the coupled magnetic flux as can be appreciated. In
addition, while the tuned resonator 706 comprises a series-tuned
resonator, a parallel-tuned resonator or even a distributed-element
resonator may also be used.
[0180] Referring to FIG. 13, the magnetic coil 709 comprises a
receive circuit that is coupled through an impedance coupling
network 733 to an electrical load 736. In order to facilitate
reception and/or extraction of electrical power from a guided
surface wave, the magnetic coil 709 may be positioned so that the
magnetic flux of the guided surface wave, H.sub..phi., passes
through the magnetic coil 709, thereby inducing a current in the
magnetic coil 709 and producing a terminal point voltage at its
output terminals 729. The magnetic flux of the guided surface wave
coupled to a single turn coil is expressed by
.psi.=.intg..intg..sub.A.sub.CS.mu..sub.r.mu..sub.o{circumflex over
(n)}dA (44)
where .psi. is the coupled magnetic flux, .mu..sub.r i s the
effective relative permeability of the core of the magnetic coil
709, .mu..sub.o is the permeability of free space, H is the
incident magnetic field strength vector, n is a unit vector normal
to the cross-sectional area of the turns, and A.sub.cs is the area
enclosed by each loop. For an N-turn magnetic coil 709 oriented for
maximum coupling to an incident magnetic field that is uniform over
the cross-sectional area of the magnetic coil 709, the open-circuit
induced voltage appearing at the output terminals 729 of the
magnetic coil 709 is
V = - N d .PSI. dt .apprxeq. - j .omega. .mu. r .mu. 0 HA CS , ( 45
) ##EQU00031##
where the variables are defined above. The magnetic coil 709 may be
tuned to the guided wave frequency either as a distributed
resonator or with an external capacitor across its output terminals
729, as the case may be, and then impedance-matched to an external
electrical load 736 through a conjugate impedance matching network
733.
[0181] Assuming that the resulting circuit presented by the
magnetic coil 709 and the electrical load 736 are properly adjusted
and conjugate impedance matched, via impedance matching network
733, then the current induced in the magnetic coil 709 may be
employed to optimally power the electrical load 736. The receive
circuit presented by the magnetic coil 709 provides an advantage in
that it does not have to be physically connected to the ground.
[0182] With reference to FIGS. 12A, 12B, and 13, the receive
circuits presented by the linear probe 703, the tuned resonator
706, and the magnetic coil 709 each facilitate receiving electrical
power transmitted from any one of the embodiments of polyphase
waveguide probes 200 described above. To this end, the energy
received may be used to supply power to an electrical load
716/726/736 via a conjugate matching network as can be appreciated.
This contrasts with the signals that may be received in a receiver
that were transmitted in the form of a radiated electromagnetic
field. Such signals have very low available power and receivers of
such signals do not load the transmitters.
[0183] It is also characteristic of the present guided surface
waves generated using the polyphase waveguide probes 200 described
above that the receive circuits presented by the linear probe 703,
the tuned resonator 706, and the magnetic coil 709 will load the
excitation source 213 (FIG. 3) that is applied to the polyphase
waveguide probe 200, thereby generating the guided surface wave to
which such receive circuits are subjected. This reflects the fact
that the guided surface wave generated by a given polyphase
waveguide probe 200 described above comprises a transmission line
mode. By way of contrast, a power source that drives a radiating
antenna that generates a radiated electromagnetic wave is not
loaded by the receivers, regardless of the number of receivers
employed.
[0184] Thus, together a given polyphase waveguide probe 200 and
receive circuits in the form of the linear probe 703, the tuned
resonator 706, and/or the magnetic coil 709 can together make up a
wireless distribution system. Given that the distance of
transmission of a guided surface wave using a polyphase waveguide
probe 200 as set forth above depends upon the frequency, it is
possible that wireless power distribution can be achieved across
wide areas and even globally.
[0185] The conventional wireless-power transmission/distribution
systems extensively investigated today include "energy harvesting"
from radiation fields and also sensor coupling to inductive or
reactive near-fields. In contrast, the present wireless-power
system does not waste power in the form of radiation which, if not
intercepted, is lost forever. Nor is the presently disclosed
wireless-power system limited to extremely short ranges as with
conventional mutual-reactance coupled near-field systems. The
wireless-power system disclosed herein probe-couples to the novel
surface-guided transmission line mode, which is equivalent to
delivering power to a load by a wave-guide or a load directly wired
to the distant power generator. Not counting the power required to
maintain transmission field strength plus that dissipated in the
surface waveguide, which at extremely low frequencies is
insignificant relative to the transmission losses in conventional
high-tension power lines at 60 Hz, all the generator power goes
only to the desired electrical load. When the electrical load
demand is terminated, the source power generation is relatively
idle.
[0186] Referring next to FIG. 14A shown is a schematic that
represents the linear probe 703and the tuned resonator 706. FIG.
14B shows a schematic that represents the magnetic coil 709. The
linear probe 703and the tuned resonator 706 may each be considered
a Thevenin equivalent represented by an open-circuit terminal
voltage source V.sub.s and a dead network terminal point impedance
Z.sub.s. The magnetic coil 709 may be viewed as a Norton equivalent
represented by a short-circuit terminal current source Is and a
dead network terminal point impedance Z.sub.s. Each electrical load
716/726/736(FIGS. 12A-B and FIG. 13) may be represented by a load
impedance Z.sub.L. The source impedance Z.sub.s comprises both real
and imaginary components and takes the form
Z.sub.s=R.sub.s+jX.sub.s.
[0187] According to one embodiment, the electrical load 716/726/736
is impedance matched to each receive circuit, respectively.
Specifically, each electrical load 716/726/736presents through a
respective impedance matching network 719/723/733a load on the
probe network specified as Z.sub.L' expressed as
Z.sub.L'=R.sub.L'+j X.sub.L', which will be equal to
Z.sub.L'=Z.sub.s*=R.sub.s-j X.sub.s, where the presented load
impedance Z.sub.L' is the complex conjugate of the actual source
impedance Z.sub.s. The conjugate match theorem, which states that
if, in a cascaded network, a conjugate match occurs at any terminal
pair then it will occur at all terminal pairs, then asserts that
the actual electrical load 716/726/736will also see a conjugate
match to its impedance, Z.sub.L'. See Everitt, W. L. and G. E.
Tanner, Communication Engineering, McGraw-Hill, 3.sup.rd edition,
1956, p. 407. This ensures that the respective electrical load
716/726/736is impedance matched to the respective receive circuit
and that maximum power transfer is established to the respective
electrical load 716/726/736.
[0188] It should be emphasized that the above-described embodiments
of the present disclosure are merely possible examples of
implementations set forth for a clear understanding of the
principles of the disclosure. Many variations and modifications may
be made to the above-described embodiment(s) without departing
substantially from the spirit and principles of the disclosure. All
such modifications and variations are intended to be included
herein within the scope of this disclosure and protected by the
following claims. In addition, all optional and preferred features
and modifications of the described embodiments and dependent claims
are usable in all aspects of the disclosure taught herein.
Furthermore, the individual features of the dependent claims, as
well as all optional and preferred features and modifications of
the described embodiments are combinable and interchangeable with
one another.
* * * * *