U.S. patent application number 15/365869 was filed with the patent office on 2018-05-31 for resistance-to-frequency converter.
The applicant listed for this patent is Silicon Laboratories Inc.. Invention is credited to Aaron J. Caffee.
Application Number | 20180150031 15/365869 |
Document ID | / |
Family ID | 62193264 |
Filed Date | 2018-05-31 |
United States Patent
Application |
20180150031 |
Kind Code |
A1 |
Caffee; Aaron J. |
May 31, 2018 |
Resistance-to-Frequency Converter
Abstract
A technique for sensing an environmental parameter is disclosed.
The technique generates an oscillating signal using a variable
resistance sensitive to a variable parameter. A frequency of the
oscillating signal is directly dependent on the variable
resistance. A time-to-digital converter generates a digital code
indicative of the variable resistance. The digital code is
generated based on the frequency of the oscillating signal and a
second frequency of a reference clock signal. The second frequency
is insensitive to the variable parameter. The variable resistance
may be a metal resistor and the reference resistance may be
generated using a capacitor that is switched at a particular
frequency. The measured resistance may be used to control a
voltage-controlled oscillator. The oscillating signal frequency may
be converted to a digital signal and post-processed for use as an
indicator of absolute temperature or other environmental
parameter.
Inventors: |
Caffee; Aaron J.;
(Scappoose, OR) |
|
Applicant: |
Name |
City |
State |
Country |
Type |
Silicon Laboratories Inc. |
Austin |
TX |
US |
|
|
Family ID: |
62193264 |
Appl. No.: |
15/365869 |
Filed: |
November 30, 2016 |
Current U.S.
Class: |
1/1 |
Current CPC
Class: |
G01K 7/01 20130101; G04F
10/005 20130101 |
International
Class: |
G04F 10/00 20060101
G04F010/00 |
Claims
1. (canceled)
2. An apparatus comprising: an oscillator circuit configured to
generate an oscillating signal; and a time-to-digital converter
configured to generate a digital code indicative of the variable
resistance, the digital code being generated based on the frequency
of the oscillating signal and a second frequency of a reference
clock signal, the second frequency being insensitive to the
variable parameter, wherein the oscillator circuit comprises: a
variable resistance sensitive to a variable parameter, a frequency
of the oscillating signal being directly dependent on the variable
resistance; a first resistor providing the variable resistance; a
first switched capacitor circuit configured to provide a reference
resistance in response to the oscillating signal; a first node
coupled between the first resistor and the first switched capacitor
circuit; an amplifier configured to generate a control signal based
on a first signal on the first node, the control signal being
indicative of the variable resistance; and a voltage controlled
oscillator circuit configured to generate the oscillating signal in
response to the control signal.
3. The apparatus, as recited in claim 2, wherein the oscillator
circuit further comprises: a second resistor having the variable
resistance; a second switched capacitor circuit configured to
provide the reference resistance in response to the oscillating
signal; and a second node coupled between the second resistor and
the second switched capacitor circuit, wherein the amplifier is
further configured to generate the control signal based on a
differential signal on the first node and the second node.
4. The apparatus, as recited in claim 2, wherein the control signal
is strain-invariant and the variable parameter is absolute
temperature.
5. The apparatus, as recited in claim 2, wherein the first switched
capacitor circuit comprises a plurality of switched capacitors,
each of the plurality of switched capacitors being responsive to a
corresponding phase of a plurality of phases of the oscillating
signal.
6. The apparatus, as recited in claim 2, wherein the variable
resistance has a first dependence on absolute temperature and the
reference resistance has a second dependence on absolute
temperature, the second dependence being less than or having a
polarity opposite to a second polarity of the first dependence.
7. The apparatus, as recited in claim 2, where the apparatus forces
the reference resistance to be equal to the variable resistance
using the oscillating signal.
8. The apparatus, as recited in claim 2, wherein the
time-to-digital converter is a frequency counter and the digital
code indicates the frequency of the oscillating signal relative to
the second frequency.
9. An apparatus comprising: an oscillator circuit configured to
generate an oscillating signal; and a time-to-digital converter
configured to generate a digital code indicative of the variable
resistance, the digital code being generated based on the frequency
of the oscillating signal and a second frequency of a reference
clock signal, the second frequency being insensitive to the
variable parameter; wherein the oscillator circuit comprises: a
variable resistance sensitive to a variable parameter, a frequency
of the oscillating signal being directly dependent on the variable
resistance; and a ring oscillator circuit having a plurality of
stages, each stage of the plurality of stages comprising: an
inverter; and a low-pass filter coupled to an output of the
inverter, the low-pass filter comprising: a resistor providing the
variable resistance; and a capacitor coupled between the resistor
and a voltage reference node.
10. The apparatus, as recited in claim 9, wherein the variable
resistance dominates a resistance at an output of each stage.
11. (canceled)
12. A method comprising: generating an oscillating signal having a
frequency directly dependent on a variable resistance, the variable
resistance being sensitive to a variable parameter; and generating
a digital code indicative of the variable resistance, the digital
code being generated based on the frequency of the oscillating
signal and a second frequency of a reference clock signal, the
second frequency being insensitive to the variable parameter,
wherein generating the oscillating signal comprises: generating a
reference resistance by transferring charge at a switching
frequency in response to the oscillating signal; generating a
control signal based on a first signal on a first node, the control
signal being indicative of the variable resistance of a first
resistor; and generating the oscillating in response to the control
signal.
13. The method, as recited in claim 12, wherein the control signal
is generated further based on a second signal on a second node, the
first node and the second node providing a differential error
signal indicative of a difference between the reference resistance
and the variable resistance.
14. The method, as recited in claim 12, wherein the control signal
is strain-invariant and the variable parameter is absolute
temperature.
15. The method, as recited in claim 14, wherein the digital code is
indicative of the frequency relative to the second frequency.
16. The method, as recited in claim 12, wherein generating the
reference resistance comprises: driving a plurality of switched
capacitors at the switching frequency using corresponding phases of
a plurality of phases of the oscillating signal.
17. The method, as recited in claim 12, wherein the variable
resistance has a first dependence on absolute temperature and the
reference resistance has a second dependence on absolute
temperature, the second dependence being less than or having a
polarity opposite to a second polarity of the first dependence.
18. The method, as recited in claim 12, wherein the oscillating
signal forces the reference resistance to be equal to the variable
resistance.
19. The method, as recited in claim 12, wherein the oscillating
signal has a period directly proportional to the variable
resistance.
20. (canceled)
Description
BACKGROUND
Field of the Invention
[0001] This application relates to integrated circuits and more
particularly to integrated circuit sensors.
Description of the Related Art
[0002] In general, sensing an environmental parameter (e.g.,
temperature or strain) on an integrated circuit includes taking
measurements using integrated circuit devices that have electronic
behavior sensitive to the environmental variable being sensed.
Parameters of the integrated circuit devices may vary as a function
of other environmental parameters (e.g., aging) that cause
electrical nonlinearities, thereby changing the operational
characteristics of the device and that affect the achievable
accuracy of a sensor. Although a resulting parameter shift may have
an expected value, the parameter shift may be unpredictable.
[0003] For example, typical devices used for sensing temperature
have high sensitivity to mechanical strain. In general, strain is a
change in element length .DELTA.L over the original element unit
length L (e.g., S=.DELTA.L/L). Packaging stress may cause strain on
an integrated circuit die, resulting in a shift in electronic
behavior of temperature sensing devices of the integrated circuit
die. If strain on the integrated circuit die is not properly
calibrated, then an apparent shift in temperature may occur in a
temperature sensor of the integrated circuit die. The apparent
shift will reduce the accuracy of a sensed temperature. Although
strain sensors can be included in a system on the integrated
circuit to compensate for these effects, the practical
implementation of such a compensation mechanism can be costly.
Therefore, improved techniques for sensing environmental parameters
by an integrated circuit are desired.
SUMMARY OF EMBODIMENTS OF THE INVENTION
[0004] In at least one embodiment of the invention, an apparatus
includes an oscillator circuit configured to generate an
oscillating signal. The oscillator circuit comprises a variable
resistance sensitive to a variable parameter. A frequency of the
oscillating signal is directly dependent on the variable
resistance. The apparatus includes a time-to-digital converter
configured to generate a digital code indicative of the variable
resistance. The digital code is generated based on the frequency of
the oscillating signal and a second frequency of a reference clock
signal. The second frequency is insensitive to the variable
parameter. The oscillator circuit may include a first resistor
providing the variable resistance, a first switched capacitor
circuit configured to provide a reference resistance in response to
the oscillating signal, a first node coupled between the first
resistor and the first switched capacitor circuit, an amplifier
configured to generate a control signal based on a first signal on
the first node, and a voltage controlled oscillator circuit
configured to generate the oscillating signal in response to the
control signal. The apparatus may further include a second resistor
having the variable resistance, a second switched capacitor circuit
configured to provide the reference resistance in response to the
oscillating signal, and a second node coupled between the second
resistor and the second switched capacitor circuit. The amplifier
may be further configured to generate the control signal based on a
differential signal on the first node and the second node. The
first switched capacitor circuit may include a plurality of
switched capacitors. Each of the plurality of switched capacitors
may be responsive to a corresponding phase of a plurality of phases
of the oscillating signal. The time-to-digital converter may be a
frequency counter and the digital code indicates the frequency of
the oscillating signal relative to the second frequency. The
oscillator circuit may include a ring oscillator circuit having a
plurality of stages. Each stage of the plurality of stages may
include an inverter and a low-pass filter coupled to an output of
the inverter. The low-pass filter may include a resistor providing
the variable resistance and a capacitor coupled between the
resistor and a voltage reference node.
[0005] In at least one embodiment of the invention, a method
includes generating an oscillating signal having a frequency
directly dependent on a variable resistance. The variable
resistance is sensitive to a variable parameter. The method
includes generating a digital code indicative of the variable
resistance. The digital code is generated based on the frequency of
the oscillating signal and a second frequency of a reference clock
signal. The second frequency is insensitive to the variable
parameter. Generating the oscillating signal may include generating
a reference resistance by transferring charge at a switching
frequency in response to an oscillating signal, generating a
control signal based on a first signal on a first node, and
generating the oscillating signal in response to the control
signal. The control signal may be indicative of the variable
resistance of a first resistor. The control signal may be generated
further based on a second signal on a second node. The first node
and the second node may provide a differential error signal
indicative of a difference between the reference resistance and the
variable resistance. The control signal may be strain-invariant and
the variable parameter may be absolute temperature. The digital
code may be indicative of the frequency relative to the second
frequency. Generating the reference resistance may include driving
a plurality of switched capacitors at the switching frequency using
corresponding phases of a plurality of phases of the oscillating
signal.
BRIEF DESCRIPTION OF THE DRAWINGS
[0006] The present invention may be better understood, and its
numerous objects, features, and advantages made apparent to those
skilled in the art by referencing the accompanying drawings.
[0007] FIG. 1 illustrates a functional block diagram of an
integrated circuit sensing system.
[0008] FIG. 2 illustrates a circuit diagram of an exemplary
temperature sensing circuit of the integrated circuit temperature
sensing system of FIG. 1.
[0009] FIG. 3 illustrates a circuit diagram of an exemplary
temperature sensing circuit of the integrated circuit sensing
system of FIG. 1, consistent with at least one embodiment of the
invention.
[0010] FIG. 4 illustrates a circuit diagram of an exemplary
temperature sensing circuit of the integrated circuit sensing
system of FIG. 1, consistent with at least one embodiment of the
invention.
[0011] FIG. 5 illustrates a circuit diagram of an exemplary
temperature sensing circuit of the integrated circuit sensing
system of FIG. 1.
[0012] FIGS. 6A and 6B illustrate a circuit diagram and associated
signal waveforms for an exemplary resistor-switched capacitor
divider that may be included in the sensing circuits of FIGS.
1-5.
[0013] FIG. 7 illustrates a circuit diagram of an exemplary sensing
circuit including a resistance-to-frequency conversion circuit
consistent with at least one embodiment of the invention.
[0014] FIG. 8 illustrates a circuit diagram of an exemplary portion
of the sensor system of FIG. 7 including a multi-phase
switched-capacitor circuit and voltage controlled oscillator
consistent with at least one embodiment of the invention.
[0015] FIG. 9 illustrates a circuit diagram of an exemplary ring
oscillator that may be included in the resistance-to-frequency
conversion circuit consistent with at least one embodiment of the
invention.
[0016] FIG. 10 illustrates a circuit diagram of an exemplary
sensing circuit including a resistance-to-frequency conversion
circuit consistent with at least one embodiment of the
invention.
[0017] The use of the same reference symbols in different drawings
indicates similar or identical items.
DETAILED DESCRIPTION
[0018] A technique for sensing an environmental parameter is
disclosed. The technique generates an oscillating signal using a
variable resistance sensitive to a variable parameter. A frequency
of the oscillating signal is directly dependent on the variable
resistance. A time-to-digital converter generates a digital code
indicative of the variable resistance. The digital code is
generated based on the frequency of the oscillating signal and a
second frequency of a reference clock signal. The second frequency
is insensitive to the variable parameter. The variable resistance
may be a metal resistor and the reference resistance may be
generated using a capacitor that is switched at a particular
frequency. The measured resistance may be used to control a
voltage-controlled oscillator. The oscillating signal frequency may
be converted to a digital signal and post-processed for use as an
indicator of absolute temperature or other environmental
parameter.
[0019] Referring to FIG. 1, exemplary sensor system 100 includes
sensor 102 that generates analog signal S.sub.OUT, which is
indicative of an environmental condition (e.g., absolute
temperature). Analog-to-digital converter 106 converts analog
signal S.sub.OUT to digital value D.sub.SENSE. That digital value
may be used to adjust parameters of an integrated circuit system
(e.g., to compensate for frequency dependence on temperature of a
microelectromechanical systems (MEMS) resonator).
[0020] Typically, a thermoelectric transducer of sensor 102
determines the accuracy of digital value D.sub.SENSE. To maintain a
sufficient level of accuracy for a target application, the
thermoelectric transducer is designed to have less sensitivity to
other environmental variables such as mechanical stress, humidity,
time (i.e. aging), etc. Any non-linearity of the electrical output
with respect to temperature may introduce a need to post-process
analog signal S.sub.OUT or digital value D.sub.SENSE for use in the
target application. Therefore, selection of components for use in
sensor 102 and configuration of those selected components to map
changes in temperature into electrical behavior should be
considered carefully.
[0021] Conventional temperature sensors use differences in
base-emitter voltages of bipolar junction diodes to sense
temperature due to the proportional to absolute temperature (PTAT)
and ratio-metric behavior of those base-emitter voltages. However,
the low transduction gain (e.g., approximately 0.3 mV/C) of bipolar
junction diodes increases noise and accuracy performance
requirements of the associated analog-to-digital converter.
Alternatively, use of the turn-on voltage of a bipolar junction
diode may improve transduction gain to approximately -1.6 mV/C.
However, the absolute accuracy of that technique is more sensitive
to diode biasing. Furthermore, the piezojunction effect, i.e.,
changes in current-voltage characteristics of a p-n junction due to
changes in mechanical stress may degrade accuracy of the resulting
sensor.
[0022] Resistive bridge techniques may improve transduction gain to
approximately 3 mV/C and maintain ratio-metric operation. Referring
to FIG. 2, exemplary sensor 102 includes a resistive bridge circuit
that compares resistance R.sub.T of temperature-sensitive resistors
to resistance R of reference resistors to generate a differential
DC output signal of analog signal S.sub.OUT.sup.- and analog signal
S.sub.OUT.sup.-. The output voltage .DELTA.V is approximately
V.sub.REF(R.sub.T+R/R.sub.T-R) where V.sub.REF is the head voltage
of the bridge circuit. Resistance R is approximately the same as
resistance R.sub.T at a nominal temperature (e.g.,
T.sub.O=25.degree. C.) resulting in .DELTA.V being approximately
zero Volts at the nominal temperature.
[0023] Although it may be desirable to select the reference
resistor with resistance R having a temperature dependence opposite
to a temperature dependence of resistance R.sub.T (e.g., -3600
parts-per million per degree Celsius (i.e., ppm/.degree. C.)), such
resistors are not readily available in a typical integrated circuit
manufacturing process. Instead, a reference resistor is chosen to
have resistance R with a temperature dependence that is as negative
as practicable (e.g., n+ polysilicon resistors having resistance R
with a temperature dependence of approximately -900 ppm/.degree.
C.). Resistance R is more weakly dependent on temperature as
compared to resistance R.sub.T. As a result, resistance R may be
considered to be a reference resistance while the temperature
dependence of resistance R.sub.T dominates the temperature
dependence of differential voltage V. Accordingly, differential
voltage .DELTA.V indicates the change in resistance R.sub.T due to
temperature relative to reference resistance R. Note that since the
first-order temperature effects tend to dominate the temperature
behavior of resistors, the discussion included herein applies only
to first order temperature effects and does not address
higher-order temperature effects. The differential voltage .DELTA.V
will indicate any mismatch between the actual resistance R and the
actual resistance R.sub.T. For example, at a maximum temperature
(e.g., 85.degree. C.), the differential voltage .DELTA.V will be
positive and at a minimum temperature (e.g., -40.degree. C.), the
differential voltage .DELTA.V will be negative. Ideally, the
differential voltage .DELTA.V is a linear function of temperature.
Although R.sub.T-R and R.sub.T+R are linear functions of
temperature, .DELTA.V may not have a perfectly linear temperature
dependence.
[0024] An actual reference resistor having resistance R may have
nonidealities associated with it that reduce the accuracy of the
sensed temperature. In some applications, absolute accuracy of the
sensed temperature is not required, but rather correspondence of a
particular sensed temperature to a particular frequency of
operation of a system is sufficient and any non-idealities may be
calibrated out. For example, a MEMS resonator application has a
temperature coefficient of approximately 10 ppm/.degree. C.
Apparent changes in temperature impact the accuracy of the system.
If an apparent temperature change is 1.degree. C., then the
frequency must be changed by 10 ppm. However, changes in the sensed
temperature due to effects unrelated to temperature may still be
problematic for the system. Even if there is no actual temperature
change, the product specifications for the MEMS resonator must
account for the effects of strain or aging on the reference
resistor of the temperature sensor and the system may only be
accurate to within +/-20 ppm. Accordingly, an ideal reference
resistor has a resistance R that is insensitive to aging and
mechanical strain.
[0025] In at least one embodiment of sensor 102, the bridge circuit
uses polysilicon resistors to implement reference resistance R and
uses temperature-sensitive resistors, i.e., thermistors, are used
to implement PTAT resistance, R.sub.T. Polysilicon resistors
typically have highly linear resistances and are designed to have
small temperature coefficients. However, conventional
polysilicon-based resistors (silicided to form a thermistor, p+/n+
doped to form a reference resistor) may be sensitive to mechanical
stress (e.g., due to their polycrystalline structure) and may age
with time (e.g., due to doping migration). Thus, the output of a
temperature sensor output may change as a function of environmental
factors unrelated to temperature and, thereby reduce overall
accuracy of the temperature sensor.
[0026] In other embodiments of sensor 102, the reference resistors
are implemented using diffusion resistors, which are less commonly
used due to their large voltage and temperature coefficients, but
are less prone to aging. However, diffusion resistors have
resistances with a greater temperature coefficient than the
temperature coefficient of the resistance of a polysilicon
resistor. Thus, use of diffusion resistors as reference resistors
reduces circuit sensitivity and reduces the value of differential
voltage .DELTA.V. This tradeoff may result in higher power
consumption in associated sensing circuits (e.g., an
analog-to-digital converter circuit) for a particular
signal-to-noise ratio. Diffusion and polysilicon resistors are also
considered piezoresistive. Although circuits that use thin film
polysilicon resistors or diffusion resistors are relatively low
cost, the response of those resistors to mechanical strain and/or
aging degrades the accuracy of the temperature measurement. Use of
strain sensors to sense and compensate for the effects of
mechanical strain on diffusion or polysilicon reference resistors
may consume a non-trivial amount of power and area while also
increasing system complexity. To reduce the piezoresistivity and
effects of aging on the resistive bridge, metal resistors may be
used as the thermistors and switched capacitor resistors may be
used as the reference resistors for comparison. FIG. 3 illustrates
an embodiment of sensor 102 configured to provide output current
.DELTA.I using terminals S.sub.OUT+ and S.sub.OUT- coupled to a
virtual ground (illustrated as 0V).
[0027] Referring to FIG. 4, rather than use typical integrated
circuit reference resistors, temperature sensor resistive bridge
302 includes reference resistors 308 and 310, which provide
relatively strain-invariant resistances. Reference resistors 308
and 310 may include current sources derived from temperature
insensitive, strain-invariant circuits that provide currents that
are proportional to the bias voltage, e.g., I=.alpha.V.sub.REF, to
maintain ratio-metric operation. In addition, note that using
reference current sources to realize the reference resistances
increases the sensitivity of the bridge circuit by a factor of two
at the point where the bridge is balanced (i.e., T.sub.O, e.g.,
25.degree. C.) because the current sources are high impedances and
do not load the bridge circuit and eliminate the resistor divider
of the bridge circuits of FIGS. 2 and 3. Referring back to FIGS. 4
and 5, reference resistors 308 and 310 may be implemented using
current sources having a p-type metal-oxide-semiconductor (i.e.,
PMOS) current source for reference resistor 308 coupled to VREF and
an n-type metal-oxide-semiconductor (i.e., NMOS) current source for
reference resistor 310 coupled to a ground node. In other
embodiments, bootstrapped, switched-capacitor based current
sources, or other suitable current sources may be included in
reference resistors 308 and 310.
[0028] Reference resistors 308 and 310 may include a reference
resistor, R.sub.T.sub.o, used in voltage-to-current conversion. If
the current sources used for reference resistances 308 and 310 of
FIG. 4 are a weak function of temperature, output signal .DELTA.V
will have a more linear response with respect to temperature than
.DELTA.V of FIG. 2. Since resistivity is proportional to absolute
temperature in metal resistors, to obtain a linear function for
.DELTA.V, a resistance that has little or no dependence on
temperature in numerator or denominator improves the linearity of
operation. Thus, reference resistances 308 and 310 are designed to
be centered at the temperature where the bridge is balanced (i.e.,
.DELTA.V=0 at temperature T.sub.0 by choosing the reference
resistor R=.alpha..times.V.sub.BIAS=R.sub.T(T.sub.o), where T.sub.0
is, e.g., 25.degree. C.). This may be achieved by choosing constant
.alpha. to be 1/R.sub.T.sub.0. Referring to FIG. 4, a realization
of the reference resistances are coupled to thermistors 304 and
306, in a Wheatstone bridge configuration and coupled to a voltage
reference node and a ground node, respectively. Thus, the output
voltage is approximately .DELTA.V=V.sub.BIAS(2.varies.R.sub.T-1),
where .varies.=1/R.sub.T.sub.o. Accordingly,
.DELTA.V.apprxeq.V.sub.BIAS(R.sub.T-R.sub.T.sub.o/R.sub.T.sub.o).
FIG. 5 illustrates an embodiment of temperature sensor resistive
bridge 302 configured to provide output current .DELTA.I using
terminals S.sub.OUT+ and S.sub.OUT- coupled to a virtual ground
(illustrated as 0V).
[0029] In at least one embodiment, sensor 102 (which may be a
voltage divider or bridge circuit of FIG. 4 or FIG. 5) includes one
or more reference resistors implemented using a switched-capacitor
resistance. Referring to FIGS. 6A and 6B, exemplary reference
resistance 310 includes a switched capacitor driven by a two-phase
clock, having phases .PHI..sub.0 and .PHI..sub.1, or other
non-overlapping clocks having a low sensitivity to environmental
parameters. For example, the two-phase clock may be generated using
a MEMS-based oscillator, an LC-based oscillator, a crystal-based
oscillator, or other suitable oscillator configured to generate a
clock signal having a frequency with a low sensitivity to strain
and aging. When .PHI..sub.1 closes the switch, node 604 charges to
the voltage on node 602, v.sub.o. When the first phase ends,
.PHI..sub.1 opens the switch and the second switch closes in
response to .PHI..sub.0, the capacitor discharges. The reference
resistance conveys current from node 604 to a reference voltage
node (e.g., ground node). The effective reference resistance is
approximately the period of the clock, T.sub.OSC, divided by the
capacitance C, (e.g., R=T.sub.OSC/C) resulting in an average
current through the reference resistance
I.sub.AVG=(v.sub.o)/(T.sub.OSC/C). As long as capacitance C is not
a function of strain or aging and r is not a function of strain or
aging, temperature sensors including bridge circuits 302 of FIGS. 4
and 5 including switched-capacitor circuits as reference resistors
308 and 310 are less sensitive to strain and aging than sensors 102
including typical integrated circuit reference resistors of FIGS. 2
and 3. Referring back to FIGS. 4 and 5, the capacitors included in
the switched-capacitor circuits used by reference resistors 308 and
310 may include a finger capacitor embedded in SiO.sub.2 or a
capacitor formed in a MEMS structural layer, or other suitable
capacitor. The capacitor may be designed to cancel any residual
strain effects by orienting two capacitors rotated 90 degrees from
each other to form a composite capacitor. In other embodiments, the
capacitor may be bootstrapped relative to changing voltage across
it causing the capacitor to behave more like a current source load
on the bridge circuit than a resistor, thus improving bridge
sensitivity to temperature.
[0030] Still referring to FIGS. 4 and 5, variable resistance
R.sub.T of thermistors 304 and 306 is selected to be a strong PTAT
resistance (e.g., metal resistors or silicided-polysilicon
resistors). An exemplary metal resistor has a variable resistance
R.sub.T with a temperature dependence of approximately 3600
ppm/.degree. C. Strain or aging may affect some thermistors.
However, although metal resistors exhibit little or no effects of
aging and may be affected by strain on geometry, the material is
not piezoresistive and therefore the effects of strain on the
resistance are negligible. Since typical metal resistors are fixed
by surrounding SiO.sub.2, the conductivity is substantially
unchanged in response to strain on geometry.
[0031] Metal resistors are not commonly used in conventional analog
circuits since metal layers in typical CMOS processes are intended
to provide low-resistance interconnects and thus have very low
sheet resistance. The low sheet resistance (e.g., 60 milli-Ohms per
square) requires resistors having a large area to implement even
small resistances (e.g., 10-20 kilo-Ohms). However, a stack of
multiple metal layers coupled by conductive via(s) of a CMOS
process may be configured as electrically coupled metal resistors
that have reduced area as compared to a typical CMOS metal
resistor, (e.g., a planar resistor formed using a narrow,
serpentine metal trace implemented using a single CMOS metal
layer).
[0032] In at least one embodiment of a sensor circuit, a thermistor
includes a silicided-polysilicon resistor, which is a polysilicon
resistor without the silicide blocked. Silicide is metal that is
injected into the top of polysilicon or diffusion to decrease the
sheet resistance. Therefore, a thermistor including a
silicided-polysilicon resistor has a combination of polysilicon and
metal resistor properties, which behaves like a PTAT resistor.
Silicided-polysilicon resistors are less sensitive to strain and
aging than conventional CMOS resistors. Typical
silicided-polysilicon resistors have higher sheet resistances than
metal resistors (e.g., 10 times the typical sheet resistance of
metal) and result in metal resistors with higher resistances for
the same area (e.g., 100-200 kilo-Ohms). Although the thermistors
of the bridge circuit of FIGS. 4 and 5 are illustrated as a single
resistor, in other embodiments of a sensor circuit, each thermistor
includes a network of individual thermistor elements and/or
includes one or more silicided-polysilicon resistors.
[0033] Referring to FIGS. 6A and 6B, reference resistor 310, which
includes a switched-capacitor circuit, may generate a ripple in
output signal v.sub.o as a result of loading a resistor (e.g.,
continuous conduction) and resetting a switched capacitor. The
ripple may present challenges to techniques for enhancing the
linearity of the resistive bridge, e.g. by connecting the resistive
bridge to a virtual ground of the analog-to-digital converter
front-end operational amplifier. In a Wheatstone bridge
configuration where the frequency is allowed to vary and an
amplifier controls a voltage-controlled oscillator generated
feedback clock signal, the RC product is equal to the oscillation
period:
v o , avg = V BIAS - I avg .times. R ##EQU00001## I avg = v o , avg
.times. C / T osc , and ##EQU00001.2## v o , avg V BIAS = 1 1 + RC
T osc = 1 2 , R .times. C = T osc . ##EQU00001.3##
If capacitance C is constant, the period of oscillation is directly
proportional to resistance R (e.g., the variable resistance
R.sub.T). Where temperature-induced change (i.e., change in
resistance is only negligibly affected by mechanical strain or
other physical parameters) dominates the change in resistance, the
period of oscillation is directly proportional to temperature.
[0034] The variable resistance, which is represented by the clock
period or clock frequency, may be used as a proxy for temperature
under conditions where the variable resistance is not substantially
sensitive to aging and mechanical strain, as discussed above. Use
of a voltage controlled oscillator analog-to-digital converter
provides a relatively simple interface between a voltage divider or
bridge circuit and the analog-to-digital converter. Referring to
FIG. 7, by forcing a virtual ground at the output of a bridge
circuit, operational amplifier 104 generates control signal v.sub.o
that drives voltage-controlled oscillator 107 to generate output
clock signal CLK.sub.VCO that oscillates with a period that is
proportional to the variable resistance R.sub.T, thereby forcing
the reference resistance R.sub.SC of the switched-capacitor
resistor to be equal to the variable resistance R.sub.T.
Accordingly, system 100 maps changes in resistance, which under
circumstances described above are indicative of, or proportional
to, changes in temperature, to changes in frequency of oscillation
of output clock signal CLK.sub.VCO generated by voltage-controlled
oscillator 107.
[0035] The analog information in output clock signal CLK.sub.VCO
(e.g., period or frequency of oscillation) may then control counter
108 or other time-to-digital converter circuit, which uses the
period of output clock signal CLK.sub.VCO as an interval from which
to count periods of reference clock signal CLK.sub.REF, which is
generated by a stable reference oscillator (e.g., a crystal
oscillator, MEMS oscillator, LC oscillator, etc.) that is
insensitive to environmental parameters as compared to the
sensitivity to environmental parameters of output clock signal
CLK.sub.VCO. For example, reference clock signal CLK.sub.REF may be
at least one order of magnitude less sensitive to the environmental
parameter being sensed than output clock signal CLK.sub.VCO. In
other embodiments, reference clock signal CLK.sub.REF controls
counter 108 to count periods of output clock signal CLK.sub.VCO. In
at least one embodiment, counter 108 is a resettable counter. In
other embodiments, counter 108 is configured to wrap around or
rollover and high-pass filter 110 provides first-order
noise-shaping to generate digitized output Dour from digital count
code D.sub.COUNT. In addition, digital signal processor 112 may be
configured to suppress high frequency noise introduced by the
noise-shaping or filter quantization noise, or to make digital
corrections to the digitized output D.sub.OUT. The resulting
digital output D.sub.SENSE may have higher resolution and accuracy
than digital count code D.sub.COUNT.
[0036] Referring to FIG. 8, in at least one embodiment,
switched-capacitor circuit 114 includes multiple switched capacitor
resistors coupled in parallel to reduce bridge output ripple at the
virtual ground of operational amplifier 104. Switched-capacitor
circuit 114 receives multiple clock phases of output clock signal
CLK.sub.VCO from voltage-controlled oscillator 107 to drive
corresponding switched capacitor resistors of switched-capacitor
circuit 114. Referring to FIG. 7, accuracy of
resistance-to-frequency converter 100 may be further improved by
implementing chopping techniques on the input of the operational
amplifier 104. Exemplary chopping techniques modulate any input
voltage offset or other low frequency noise to a higher frequency
for filter attenuation at a higher frequency.
[0037] Voltage-controlled oscillator 107 may be formed using metal
resistance and metal-based capacitance, which causes the native
frequency of voltage-controlled oscillator 107 to change consistent
with the loop response to temperature changes and requiring only
minor adjustments in response to temperature change, thereby
reducing dynamic range requirements for operational amplifier 104
and improving noise performance of the sensor system. Referring to
FIG. 9, in at least one embodiment, voltage-controlled oscillator
107 includes a ring oscillator, which may have a single-ended or
differential topology and may be formed using a plurality of
stages. Each stage includes an inverter driving an RC filter formed
from metal/silicided resistance and metal capacitance. Different
resistances and capacitances that are multiples of R.sub.T and C
(as indicated by constants .alpha. and .beta. may be used to obtain
a target time constant of the oscillator circuit. The variable
resistance R.sub.T dominates the resistance at an output of each
stage (e.g., variable resistance R.sub.T is much greater than the
resistance of the inverter). Ring oscillator 900 may be
incorporated into voltage-controlled oscillator 107 of FIG. 7 in
high performance applications. In at least one embodiment, ring
oscillator 900 is designed to have a frequency of oscillation at or
near a target frequency that naturally balances the bridge circuit
driving the ring oscillator (i.e., causes R.sub.SC==R.sub.T). The
ring oscillator may be coupled to an analog varactor that provides
relatively small voltage values to tune the frequency of
oscillation, thereby reducing or eliminating any need for feedback
control in simple temperature-to-frequency converter applications
that are more cost sensitive and have relaxed accuracy
requirements. In embodiments where the frequency of oscillation
does not need substantial change, the operation amplifier needs
less dynamic range, a smaller analog varactor may be used, and a
lower-noise design results. In at least one embodiment of a sensor
system, sensor 102 includes a ring oscillator having variable
resistance R.sub.T configured as a stand-alone
resistance-to-frequency converter, without a feedback loop, as
illustrated in FIG. 10. Note that the ring oscillator circuits
illustrated herein are exemplary only, and techniques described
herein may use other types of oscillators (e.g., relaxation
oscillators, Wein bridge oscillators, etc.).
[0038] Referring to FIGS. 1-10, in at least one embodiment of a
sensor, instead of sensing absolute temperature, the sensor
generates a digital signal indicative of mechanical strain on the
integrated circuit. Accordingly, rather than R.sub.T being a
resistance sensitive to absolute temperature, R.sub.T is a
resistance sensitive to mechanical strain. For example, resistances
R.sub.T of FIGS. 2-7, 9, and 10 are formed from strain-sensitive
diffusion or polysilicon resistors, and the reference resistances
are relatively insensitive to mechanical strain. Since resistors
308 and 310 are temperature insensitive and strain-invariant, the
sensors of FIGS. 1-10 can generate D.sub.SENSE that is indicative
mechanical strain on the integrated circuit.
[0039] Thus, various embodiments of a sensor that converts
resistance to frequency for use in providing a digital
representation of absolute temperature. The description of the
invention set forth herein is illustrative, and is not intended to
limit the scope of the invention as set forth in the following
claims. For example, while the invention has been described in an
embodiment in which a Wheatstone bridge circuit configuration of
thermistor resistors and switched-capacitor resistors are used to
detect changes in resistance as a proxy for temperature, one of
skill in the art will appreciate that the teachings herein can be
utilized in other arrangements of thermistors and reference
resistors that produce a signal that is indicative of other
environmental variables. Variations and modifications of the
embodiments disclosed herein, may be made based on the description
set forth herein, without departing from the scope and spirit of
the invention as set forth in the following claims.
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