U.S. patent application number 15/847273 was filed with the patent office on 2018-05-03 for adaptive tracking rail audio amplifier.
The applicant listed for this patent is James K. Waller, JR.. Invention is credited to James K. Waller, JR..
Application Number | 20180123519 15/847273 |
Document ID | / |
Family ID | 59276041 |
Filed Date | 2018-05-03 |
United States Patent
Application |
20180123519 |
Kind Code |
A1 |
Waller, JR.; James K. |
May 3, 2018 |
ADAPTIVE TRACKING RAIL AUDIO AMPLIFIER
Abstract
For use in professional audio amplifier applications, high
current charge pump circuits including charge circuits and dynamic
release circuits are connected between respective storage
capacitors and the audio power amplifier output. The charge
circuits charge their storage capacitors when an output signal
swing of the power amplifier is below a corresponding predetermined
positive and negative threshold. The dynamic release circuits
release power from their charged storage capacitors to dynamically
vary the voltage levels of their positive and negative power supply
rails in direct relation to the output signal swing of the power
amplifier, preferably at unity gain. Multiple positive and negative
power supply rails at different voltage levels can be used with
cascaded multi-stage dynamic release circuits to maximize system
efficiency. The charge and dynamic release circuits can be
connected between their storage capacitors and a buffer amplifier
which also receives the power amplifier audio input.
Inventors: |
Waller, JR.; James K.;
(Clarkston, MI) |
|
Applicant: |
Name |
City |
State |
Country |
Type |
Waller, JR.; James K. |
Clarkston |
MI |
US |
|
|
Family ID: |
59276041 |
Appl. No.: |
15/847273 |
Filed: |
December 19, 2017 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
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15468823 |
Mar 24, 2017 |
9853602 |
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15847273 |
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15218463 |
Jul 25, 2016 |
9641133 |
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15468823 |
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13859889 |
Apr 10, 2013 |
9402128 |
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15218463 |
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61622696 |
Apr 11, 2012 |
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Current U.S.
Class: |
1/1 |
Current CPC
Class: |
H03F 2200/375 20130101;
H03F 3/217 20130101; H03F 1/0233 20130101; H03F 2200/03 20130101;
H03F 3/185 20130101; H03F 3/183 20130101; H04R 3/00 20130101; H03F
3/2173 20130101; H03F 1/0238 20130101 |
International
Class: |
H03F 1/02 20060101
H03F001/02; H03F 3/217 20060101 H03F003/217; H03F 3/183 20060101
H03F003/183; H03F 3/185 20060101 H03F003/185 |
Claims
1. A system for amplifying an audio signal comprising: positive and
negative power supply rails; a power amplifier having an audio
signal input, an audio signal output and positive and negative
adaptive rails; storage capacitors connected in one-to-one
correspondence to respective said positive and negative adaptive
rails; high current charge pump circuits in one-to-one
correspondence with respective said capacitors, each said high
current charge pump circuit being connected between respective said
storage capacitors and said amplifier output and having a charge
circuit and a dynamic release circuit connected in one-to-one
correspondence between respective said storage capacitors and said
amplifier output, said charge circuits and dynamic release circuits
being co-operable to cause is their respective said high current
charge pump circuits to charge respective said storage capacitors
across both said positive and negative power supply rails when an
output signal swing of said power amplifier is below corresponding
respective predetermined positive and negative thresholds and to
cause power to be released from respective said storage capacitors
to dynamically vary voltage levels of respective said positive and
negative power supply rails in a direct relation to said output
signal swing of said power amplifier.
2. A system according to claim 1, said direct relation of said
voltage levels of respective said positive and negative adaptive
rails to said output signal swing of said power amplifier being
unity gain.
3. A system according to claim 1, each said dynamic release circuit
further comprising a sag correction circuit altering its respective
said predetermined threshold to maintain a minimum offset between
said amplifier output swing and dynamically varied voltage levels
of its respective said power supply rail as a voltage of its
respective said storage capacitor discharges.
4. A system according to claim 1 further comprising a buffer
amplifier having a common node audio signal input with said power
amplifier signal output, an audio signal output and common node
positive and negative adaptive rails with said power amplifier
positive and negative adaptive rails, said high current charge pump
circuits, including their respective said charge and dynamic
release circuits, being connected in one to-one correspondence
between respective said storage capacitors and said buffer
amplifier output instead of said power amplifier output.
5. A system according to claim 1, said positive and negative power
supply rails comprising multiple positive and negative power supply
rails each having a different voltage level and each of said
dynamic release circuits comprising multiple stages corresponding
in one-to-one relationship to said multiple positive and negative
power supply rails to cascade operation of said dynamic release
circuit stages in response to corresponding levels of said
amplifier output swing.
6. A system according to claim 5, said direct relation of said
voltage levels of respective said positive and negative power
supply rails to said output signal swing of said power amplifier
being unity gain.
7. A system according to claim 5, each said control circuit further
comprising a sag correction circuit altering its respective said
predetermined threshold to maintain a minimum offset between said
amplifier output swing and the dynamically varied voltage levels of
its respective said power supply rails of said at least two sets of
positive and negative power supply rails as a voltage of its
respective said storage capacitor discharges.
8. A system according to claim 5 further comprising a buffer
amplifier having a common node audio signal input with said power
amplifier signal output, an audio signal output and common node
positive and negative adaptive rails with said power amplifier
positive and negative adaptive rails, said high current charge pump
circuits, including their respective said charge and dynamic
release circuits, being connected in one to-one correspondence
between respective said storage capacitors and said buffer
amplifier output instead of said power amplifier output.
Description
CROSS-REFERENCE TO RELATED APPLICATIONS
[0001] This application is a continuation claiming priority to
pending U.S. patent application Ser. No. 15/468,823, filed Mar. 24,
2017, a continuation-in-part claiming priority to U.S. patent
application Ser. No. 15/218,463, filed Jul. 25, 2016, now U.S. Pat.
No. 9,641,133, which is a continuation of U.S. patent application
Ser. No. 13/859,889, filed Apr. 10, 2013, now U.S. Pat. No.
9,402,128, which claims is priority to provisional U.S. Application
No. 61/622,696, filed Apr. 11, 2012.
BACKGROUND OF THE INVENTION
[0002] Numerous designs have been developed and used over the past
years incorporating various methods of providing a modulated power
supply for an audio amplifier in order to improve efficiency and
reduce dissipation of the power devices. Many prior art designs
have disclosed either dual or multiple rails that switch to a
higher voltage when the output swing of the amplifier is near
clipping. These designs have been termed Class H and Class G where
a secondary or multiple rail voltages are selected as required
based on amplifier output swing. More complex designs have been
realized with continuously variable rails that track the input
signal and adjust the power supply rails so as to maintain a
constant voltage between the output devices output swing and the
power supply rail voltage. Most of these amplifiers are very
complex requiring Pulse Width Modulation of the power supply and
increased manufacturing requirements due to the large associated
circuitry required to provide the tracking supply rails. Countless
Class D designs have also been offered commercially which convert
an input audio signal into a series of output pulses. When the
pulses are averaged over time and low pass filtered to remove
higher order harmonic information the output will be a replica of
the input signal. While Class D offers the highest level of
efficiency it is also one of the most difficult to use in
applications where low EMI/RFI performance is required. While all
of the various topologies have seen varying degrees of success
commercially, the designs that offer the best cost vs. performance
gain the widest market acceptance. Many of the numerous designs
have excellent performance but may also be the most difficult to
manufacture. At the same time high output automotive audio power
amplifiers based on switch mode power supply technology has been
available for years as aftermarket is products but have not been
embraced by the original equipment manufacturers (OEM) due to a
number of undesirable side effects including switching transients
which cause large levels of RFI emissions. In order to deliver high
power automotive audio systems an efficient, high power DC to DC
converter is required, which will convert the 12 volt automotive
battery voltage to a higher supply voltage with high current output
capability. The automotive charging system typically produces
between 13.5 and 14.4 volts when the engine is running. While this
is a slight increase above the 12 volt battery voltage this is not
enough for high power amplification of audio signals. Amplifiers
capable of high output power either need more voltage swing than
the typical 14.4 volts available with the engine running or need to
provide an extremely high current output in order to drive very low
impedance loads. Typically, speakers with lower impedances have
lower efficiencies and therefore a gain in output power with audio
amplifiers that can deliver higher output current may not result in
a large net gain in sound pressure levels. The formula to calculate
output power is given by:
POWER = V rms I rms = V rms 2 R = V peak 2 2 R ##EQU00001##
[0003] An ideal power amplifier that can swing all the way to the
power supply rails with a 14.4 volt supply can deliver a peak
amplitude of 7.2 volts when connected to a 4 ohm load and would
deliver 6.48 watts. Most automotive audio power amplifiers are dual
amplifiers connected in what is termed "Bridge Mode" with one
amplifier channel swinging positive and one swinging negative with
the load connected between the two amplifier outputs and can, as a
result, deliver twice the voltage swing across the load. This means
that an ideal amplifier that can swing to the rails in bridge mode
can deliver 14.4 volts peak which would deliver a total of 25.9
watts into a 4 ohm load. In reality, most power amplifiers are far
from ideal and can typically only swing to within about 1.5 volts
of the positive and negative power supply rails. As a result, the
real world power amplifiers actual power output with the alternator
running and 14.4 volts at the battery is closer to 16 watts. Thus
it becomes obvious that the output of a car audio amplifier is
limited by the voltage of the car battery with the alternator
running In most actual car systems, the amplifiers are connected in
bridge mode configuration as described above, and speaker
impedances are no higher than 4.OMEGA., but it becomes apparent
that the maximum output power per channel is roughly 30 watts even
when driving a 2 ohm load and only about 16 watts with a 4 ohm
load. High-power car amplifiers have been available for many years
in the automotive aftermarket and these amplifiers use a DC-to-DC
converter to generate a higher power supply voltage. In order to
increase the battery voltage to a level capable of producing a
higher power level most aftermarket automotive power amplifiers use
switch mode power supplies SMPS to convert or transfer power from
the 12 volt automotive battery (14.4 volts with the engine running)
to a higher output voltage. While switching power supplies have
seen improvements in terms of output power and efficiency even the
best designs today produce an unacceptable level of radio frequency
interference RFI and as a result have not seen wide acceptance for
use in OEM vehicles. Other improvements in SMPS have been made
offering higher switching frequencies, which allow component sizes
to be reduced but produce even higher levels of RFI emissions. The
common (SMPS) used in automotive aftermarket audio applications
switches the battery voltage at a frequency between 25 kHz and 100
kHz to generate an AC square wave signal at the primary side of a
step-up transformer. The stepped up waveform on the secondary of
the transformer is rectified and filtered back to a DC signal. The
output is typically a symmetrical +/-25 to 35 volts.
[0004] DC-DC converters based on charge pump or flying capacitor
technology is have been widely used in low power DC-DC converters
but have seen limited use in high power applications due to a
number of limitations including high pulse currents that occur at
the switching transients which reduce efficiency and increase RFI
problems. The low power circuits typically switch at higher
frequencies between 20 KHz and 150 KHz which reduce the size of the
capacitors but also contribute to an increase the RFI emissions.
Integrated circuits have been produced for years based on the
concepts of charge pump circuitry and have provided circuits which
offer low current designs capable of delivering only milliamps of
current to an external load. U.S. Pat. No. 5,066,871 is an example
of one such design but many of the integrated circuit manufacturers
offer IC's based on charge pump technology. There are many current
offerings for low power switched capacitor technology in integrated
circuit form from manufacturers including Analog Devices, Linear
Technologies and National Semiconductor, to name just a few.
However, none of these circuits can be used in a higher power
application that can deliver amperes of output power required for
automotive audio power amplification.
[0005] One recent prior art system offers improved switched
capacitor technology by charging a capacitor to the supply voltage
and switching the charged capacitor when additional output swing is
required. In order to keep the amplifier output swing centered, a
reference voltage is added to the input to switch the amplifier
center bias when the additional capacitor voltage is switched on.
One drawback to this system is the adding of undesirable switching
transients in the output signal. While this system will double the
power supply voltage when needed, it is also limited to two times
the power supply voltage, i.e. 28 volts with a 14 volt supply, and
therefore requires relatively low impedance drivers in order to
gain large amounts of output power. This system operates as a class
H or class G amplifier when the additional rail voltage is switched
on and off, which improves dissipated output device heat but does
not gain the full advantage of a tracking rail or adaptive rail
design. A full tracking rail or adaptive rail design will provide
even better reduction of dissipation. The implementation of a
pulsed rail configuration will greatly reduce the heat dissipation
of the power MOSFETs used in switching the supply voltage.
[0006] In another recent prior art system, a power amplifier is
centered between the supply voltage and both the positive and
negative rails are increased as the output of the amplifier
requires more voltage swing. This system basically uses additional
power amplifiers with a gain of 1 to track the audio power
amplifier output at unity gain. The system monitors the audio
amplifiers input or output swing and when a threshold is exceeded
the power boost amplifiers will boost the power supply rails by
driving charged capacitors between the output of the boost
amplifiers and the power supply rails. This system does provide a
tracking rail design which tracks the output swing of the audio
power amplifier. However, the net gain in output power is
relatively small, on the order of a few watts, compared to bridge
mode designs. This is so because it is a single ended design. The
second major drawback to this design is the complexity of the
system, including the power boost amplifiers. In order to provide
boost amplifiers with unity gain, a full amplifier circuit is
implemented with a power MOSFET output stage. A relatively small
gain in total system output power is achieved at the expense of
increased cost and complexity in implementing this system. While
the power boost amplifiers certainly can be fully integrated in IC
form, reducing build complexity, the design requires large output
MOSFETs and will therefore require expensive integrated circuit
packaging that will provide some form of heat sink interface to
keep the output devices cool. The prior art system also teaches
using a gain slightly above unity in order to avoid capacitor
voltage droop or sag due to the capacitor discharging under
amplifier load. The amount of voltage droop or sag is difficult to
measure in operation and, therefore, adding additional gain may be
inadequate in some circumstances or may be excessive in others.
[0007] The low voltage method of controlling tracking rails in an
audio power amplifier disclosed herein is well-suited to automotive
applications, such as using a subwoofer channel at bass
frequencies, where a maximum of 14.4V DC is available with the
alternator running However, because the loop gain of the control
system is extremely high, it can in some applications cause
overshoot of the control resulting in instability, particularly at
frequencies above 3 KHz. Having a larger voltage differential
between the amplifier output and the adaptive rail mitigates the
instability, but may also reduce efficiency.
[0008] There is also a need for improvement of efficiency in
professional audio applications that do not have the voltage
limitations of the automotive applications. The maximum available
power at a typical 120V/20 A AC wall socket is the product of 120V
RMS AC and 20 A or 2400 VA. A professional audio amplifier
connected to a typical AC mains circuit converts this available AC
power into audio power to drive a loudspeaker. If the audio
amplifier were 100% efficient, it could not deliver more than 2400
VA on a continuous basis. But even class D amplifiers do not reach
100 percent efficiency, most designs being closer to 90% and the
best designs not achieving much better than 90% efficiency,
decreasing the final power conversion factor of the system.
Consider known professional audio applications in which audio
amplifiers have positive and negative power supplies and
theoretical 100% efficiency of both the power supplies and the
amplifier. Assume the amplifier can deliver 2500 watts peak power
or 100V peak signal swing into a 4 ohm load at 25 A to power the
rails of the amplifier for both the positive and negative power
supplies. The required peak output power of each power supply is,
therefore, 2500 watts. Such a demand on the power supply is a
severe limitation on the maximum power available from a typical 20
A RMS AC is mains circuit.
[0009] Furthermore, "Watts RMS" is widely used by the audio
community and has become an accepted, though inaccurate, way to
rate audio amplifier output. While "RMS" is an appropriate term
when applied to voltage or current, it is not appropriate to power.
Rating audio amplifier output power in "Watts RMS" could only be
valid if the output power were measured using a pure sine wave (RMS
volts) signal driving a resistive load. Adding the inefficiencies
of both the typical power supply and the audio amplifier circuit,
the actual available audio power is considerably less than a
consumer might expect based on a "Watts RMS" rating.
[0010] Many audio manufacturers promote their audio power in terms
of peak power or "music power" or a number of other power rating
methods. Peak power actually is twice the available RMS power and
has become the most common rating for professional audio power
amplifiers because of the apparently higher power rating. But the
continuous RMS power available simply cannot exceed what is
available from the AC mains circuit. Some manufactures avoid the
issue completely by simply specifying "watts" and not specifying
how the output wattage was measured.
[0011] It is, therefore, an object of the invention to provide an
adaptive rail power amplifier using charge pump circuitry which
overcomes the limitations of the above-mentioned prior art designs
allowing use in OEM automotive vehicles. It is another object to
provide an adaptive rail power amplifier with the ability to
produce an output voltage swing above the input power supply
voltage dynamically and as needed. In particular, it is an object
of the invention to provide a power amplifier with adaptive
tracking power supply rails offering greatly reduced complexity. It
is a further object of the invention to provide an adaptive rail
power amplifier with dynamically controlled charge pump converters
for use in automotive power amplifiers capable of delivering well
in excess of 100 watts of output power. It is a further object of
the invention to provide an increase of up to 3 times the input
supply voltage as required, to increase the amplifier output swing.
It is a yet a further object of the invention to provide an
adaptive rail power amplifier technology capable of high power
output level for use in OEM automotive applications without causing
problems in radio reception. It is still another objective of the
invention to provide an adaptive rail power amplifier technology
with higher efficiency and reduced heat dissipation. It is yet
another objective of the invention to offer improved tracking of
the power supply rails that can automatically adapt to any droop or
sag voltage in the stored charge of the capacitor used to elevate
the power supply rails.
[0012] It is also an object of the invention to provide an
alternate embodiment of the adaptive rail power amplifier which has
the same level of performance as the above described invention with
reduced parts and complexity. It is a further object of the
invention to provide an alternate embodiment with a simplified
design which lends itself to full integration. It is another object
of the alternate embodiment of the invention to require a single
reference voltage for proper operation and signal tracking.
[0013] Furthermore, it is an object of the invention to provide an
adaptive tracking rail audio amplifier which is stable at
frequencies above 3 KHz. Another object of the invention is to
provide an adaptive tracking rail audio amplifier with full
bandwidth operation. Yet another object of the invention is to
provide an adaptive tracking rail audio amplifier which reduces
power consumption. It is also an object of the invention to provide
an adaptive tracking rail audio amplifier which operates over the
entire audio spectrum up to 20 KHz. A further object of the
invention is to provide an adaptive tracking rail audio amplifier
which operates without distortion or collapse of the control rails.
It is a further object of the invention to provide an adaptive
tracking rail audio amplifier which operates without increasing the
difference voltage between the output swing of the amplifier and
the power supply rails. And it is an object of the invention to
provide an adaptive tracking rail audio amplifier with improved
rail control circuit efficiency.
SUMMARY OF THE INVENTION
[0014] With the forgoing and other objects in view there is
provided, in accordance with the invention, an adaptive rail power
amplifier technology that compares the output signal of the power
amplifier with positive and negative adaptive power supply rails
generating a pulsed control signal fed to high current charge pump
circuits for providing an output voltage that is capable of
exceeding the input supply voltage when the output signal swings
beyond the limits of the input voltage.
[0015] The adaptive rail high current charge pump circuit includes
a charge storage device, also called a flying capacitor, with one
of its plates connected to two high current switches. One high
current switch is connected to the supply through a ferrite bead.
The other high current switch is connected to ground through
another ferrite bead. The other plate of the flying capacitor is
connected to two high current diodes. The first high current diode
is connected to the battery. The other high current diode is
connected to the output node and to second charge storage capacitor
to smooth the output voltage and deliver output current to the
load. A dead zone circuit generates the proper timing to control
the high current switches. The dead zone outputs are connected to
the gate control of the high current switches. The charge pump
circuit is controlled by the output of a comparator which compares
the output signal of the power amplifier with the rail voltage,
producing a dynamically varying pulsed output signal in response to
the output swing of the amplifier and the adaptive rail output of
the charge pump circuit.
[0016] In another embodiment of the invention, an adaptive rail
control signal is derived by taking the difference between the
amplifier output signal and the rail voltage. The derived
difference signal is used to control a charge pump circuit in both
saturated and linear mode, producing a linear adaptive rail voltage
which tracks the output signal so as to provide an adaptively
increasing power supply rail voltage.
[0017] In accordance with an added feature of the invention,
multiple sections or blocks of the charge pump circuitry are
combined to provide even higher output voltages, thereby increasing
further the output voltage swing of the amplifier and allowing the
use of higher efficiency higher impedance speaker loads.
[0018] While the invention will be described for use in automotive
sound systems it is understood that many other applications for the
invention are possible including any application where limited
power supply voltage is seen.
[0019] In addition to improving the efficiency of the automotive
system, the automotive application technology is also useful in
professional audio applications that do not have the voltage
limitations of the automotive applications.
[0020] In accordance a single stage above-automobile-voltage
embodiment of the invention, a system for amplifying an audio
signal has positive and negative power is supply rails and a power
amplifier with an audio signal input, an audio signal output and
positive and negative adaptive rails. Storage capacitors are
connected in one-to-one correspondence to respective positive and
negative adaptive rails. Each capacitor has a high current charge
pump circuit, or rail control circuit, which includes a charge
circuit and a dynamic release circuit. The charge and dynamic
release circuits are each connected between their respective
storage capacitor and the amplifier output. The charge and dynamic
release circuits circuit cooperate so that each rail control
circuit charges its respective storage capacitor across both
positive and negative power supply rails when the output signal
swing of the power amplifier is below its respective predetermined
positive and negative thresholds and so that stored power is
released from its respective storage capacitor to dynamically vary
voltage levels of its respective positive and negative power supply
rails in a direct relation to the output signal swing of the power
amplifier.
[0021] In accordance with multi-stage above-automobile-voltage
applications of the invention, multiple positive and negative power
supply rails have different voltage levels and each dynamic release
circuit has multiple stages to cascade its operation in response to
corresponding levels of the amplifier output swing.
[0022] For single stage and multi-stage systems, the direct
relation of the voltage levels of respective positive and negative
power supply rails to the output signal swing of the power
amplifier is preferably unity gain.
[0023] For single stage and multi-stage systems, each dynamic
release circuit may also incorporate a sag correction circuit
altering its respective predetermined threshold to maintain a
minimum offset between the amplifier output swing and the
dynamically varied voltage levels of its respective power supply
rails as the voltage of its respective storage capacitor
discharges.
[0024] For single stage and multi-stage systems, the system may
alternatively use a buffer amplifier with a common node audio
signal input with the power amplifier signal output, an audio
signal output and common node positive and negative adaptive rails
with the power amplifier positive and negative adaptive rails. The
high current charge pump circuits, including the charge and dynamic
release circuits, are then connected between their respective
storage capacitors and the buffer amplifier output instead of
between their respective storage capacitors and the power amplifier
output.
BRIEF DESCRIPTION OF THE DRAWINGS
[0025] Other objects and advantages of the invention will become
apparent upon reading the following detailed description and upon
reference to the drawings in which:
[0026] FIG. 1 is a block diagram of a typical prior art car audio
system;
[0027] FIG. 2 is a simplified schematic of the typical switch mode
power supply used in prior art aftermarket audio systems;
[0028] FIG. 3 is a block diagram of a pulsed mode adaptive rail
power amplifier embodiment of the invention for use in automotive
applications;
[0029] FIG. 4 is a partial block, partial schematic diagram of the
positive charge pump circuit of FIG. 3;
[0030] FIG. 5 is a partial block, partial schematic diagram of the
negative charge pump circuit of FIG. 3;
[0031] FIG. 6 is a graphic representation of the output voltage
swing of the adaptive rail amplifier showing the positive and
negative adaptive rail signals.
[0032] FIG. 7 is a graphic representation of the positive output
signal swing of a 100 Hz sine wave showing the positive adaptive
rail signal and positive pulsed rail control signal of FIG. 3;
[0033] FIG. 8 is a simplified schematic diagram of a typical power
amplifier block 14 of FIG. 3;
[0034] FIG. 9 is a block diagram of a linear mode embodiment of the
invention providing non-pulsed adaptive rails with reduced EMI
emissions;
[0035] FIG. 10 is a schematic diagram of the positive charge pump
circuit of FIG. 9;
[0036] FIG. 11 is a schematic diagram of the negative charge pump
circuit of FIG. 9;
[0037] FIG. 12 is a graphic representation of the positive output
signal swing of a 100 Hz sine wave showing the positive adaptive
rail signal and positive adaptive rail control signal of FIG.
9;
[0038] FIG. 13 is a block diagram of a single comparator charge
pump tracking rail embodiment of the invention;
[0039] FIG. 14 is a block diagram of a switched dual comparator
charge pump tracking rail embodiment of the invention;
[0040] FIG. 15 is a schematic diagram of the positive charge pump
circuit of FIG. 14.
[0041] FIG. 16 is a schematic diagram of the negative charge pump
circuit of FIG. 14;
[0042] FIG. 17 is a schematic diagram of the positive adaptive rail
circuit of FIG. 14;
[0043] FIG. 18 is a schematic diagram of the negative adaptive rail
circuit of FIG. 14; and
[0044] FIG. 19 is a schematic diagram of an alternate or simplified
embodiment of the adaptive rail power amplifier.
[0045] FIG. 20 is a block diagram of an above-automotive-voltage
single stage dynamic release circuit embodiment of the
invention;
[0046] FIG. 21 is a detailed schematic of the positive rail charge
and dynamic release circuits of the embodiment of FIG. 1;
[0047] FIG. 22 is a detailed schematic of the negative rail charge
and dynamic release circuits of the embodiment of FIG. 1;
[0048] FIG. 23 is a block diagram of an above-automotive-voltage
multi-stage dynamic release circuits embodiment of the
invention;
[0049] FIG. 24 is a detailed schematic of the positive rail charge
and dynamic release circuits of the embodiment of FIG. 23;
[0050] FIG. 25 is a detailed schematic of the negative rail charge
and dynamic release circuits of the embodiment of FIG. 23;
[0051] FIG. 26 is a graphical representation comparing the positive
swing of the power amplifier output of FIG. 23 with the positive
adaptive rail voltage of the positive rail charge and dynamic
release circuits of FIG. 24; and
[0052] FIG. 27 is a block diagram of an above-automotive-voltage,
buffer amplifier driven rail control alternative for use with
single stage and multi-stage dynamic release circuit embodiments of
the invention.
[0053] While the invention will be described in connection with
preferred embodiments thereof, it will be understood that it is not
intended to limit the invention to those embodiments or to the
details of the construction or arrangement of parts illustrated in
the accompanying drawings.
DETAILED DESCRIPTION
[0054] In the following description of the Figures, similar
reference symbols designate corresponding structural parts or
functional blocks.
The Prior Art Automotive Aftermarket Power Amplifier
[0055] Turning first to FIG. 1, a block diagram of the typical
aftermarket automotive power amplifier is shown. The switch mode
power supply 11 and the audio power amplifier 14 are contained in
one chassis or unit. A 12 volt battery 10 provides the input power
to the switch mode power supply 11 via a +12 volt and ground
connection. The switch mode power supply 11 converts the 12 volt
battery voltage to a bipolar output voltage referenced to ground,
providing a positive output voltage PS and a negative output
voltage NS, which supplies bipolar power to the amplifier 14. The
audio amplifier 14 receives an audio input signal and amplifies
this audio input signal to drive a speaker 100.
[0056] Looking at FIG. 2, a simplified schematic of the switch mode
power supply 11 of FIG. 1 is shown. The switch mode power supply 11
receives its input power from the 12 volt battery 10, which is
connected to the center tap of a transformer T1. A pulse width
modulator controller 13 controls the switching of high power MOSFET
switching transistors Q1, Q2, Q3 and Q4. The switching topology is
called a "push-pull" converter because the transformer T1 has a
"center-tapped" primary. The center tap is permanently connected to
the 12 volt car battery, typically via an LC filter (not shown), to
reduce switching spikes in the battery voltage. The ends of the
primary side of the transformer T1 are each connected to a
different paralleled pair of MOSFETs that switch them to ground in
each conduction cycle. MOSFETs Q1 and Q2 are parallel connected to
one primary end and MOSFETs Q3 and Q4 are parallel connected to the
other primary end so as to provide higher switching current than
would be possible with a single MOSFET transistor. The secondary
side of the switch mode power supply transformer T1 has its center
tap connected to ground and the two outputs of the transformer T1
are connected to switching diodes in a diode bridge D1. The diodes
of the bridge D1 must be extremely fast diodes and are typically
discrete diodes in a TO220 package with a 10 amp minimum rating and
at a suitable voltage rating based on the output voltage of the
transformer. The output of the diode bridge D1 is filtered by
capacitors C+ and C- at nodes PS and NS, respectively, to provide a
positive and negative 30 volt high current output capable of
providing the required voltage and current to power an audio
amplifier. The typical switching frequency of this type of switch
mode power supply is between 25 KHz and 100 KHz, which reduces the
size of both the transformer T1 and the output filter capacitors C+
and C. Due to the high switching frequency and the large current
pulses generated when switching a high current square wave at high
frequencies, this design suffers from high EMI/RFI emissions. When
the audio power amplifier 14 is delivering large amounts of current
to the speaker load, the RFI emissions can actually reduce FM radio
reception and, in some cases, cause complete loss of certain AM
radio band frequencies. As a result, the automotive OEMs have
avoided using this type of switch mode power supply/power amplifier
technology.
Pulsed Adaptive Rail Power Amplifier for Automotive Use
[0057] Turning now to FIG. 3, a block diagram of a pulsed adaptive
rail power amplifer according to the invention is shown. The power
amplifier 14 receives both positive and negative power from charge
pump circuits 40 and 41, respectively. The input of the power
amplifier 14 is biased at VREF 28, which is 1/2 the auto battery
voltage. The battery voltage will be between 12 volts without the
engine running and 14.4 volts with the alternator charging the
battery. This means that VREF 28 will track at 1/2 the battery
voltage, keeping the output voltage swing of the amplifier 14
centered between the positive battery voltage and ground. A
comparator 24 compares the output voltage swing of the power
amplifier 14 and the positive adaptive rail voltage 21 at the
output of the positive charge pump circuit 40. A resistor 26
provides hysteresis for the comparator 24 which determines in part
the switching frequency of the adaptive pulsed rail 21 and provides
a window of switching between transitions. By increasing the
hysteresis, a larger transition time between switching points
occurs, which reduces the switching transitions of the positive
cycling waveform. When the output voltage swing of the power
amplifier 14 approaches the battery voltage, the output of the
comparator 24 changes from a high output to low. Due to the
positive feedback of the hysteresis resistor 26, the output of the
positive charge pump circuit 40 will increase only a small voltage
based on the amount of hysteresis. This small voltage increase will
cause the comparator 24 to change state again to high. This slight
positive/negative cycle will continue as long as the output signal
swing of the amplifier 14 requires more voltage than the input
battery voltage at B1. This will provide a series of output pulses
at the comparator 24 which change in pulse width and frequency
based on the current demand on the output of the positive charge
pump circuit 40. As the amplifier 14 starts to swing negative with
respect to the input battery voltage, the positive adaptive rail 21
will reach a quiescent point and the amplifier 14 will now draw its
power from the battery voltage available at B1. The voltage
difference between the output of the amplifier 14 and the adaptive
rail voltage 21 is further defined by the rail offset voltage 27
applied to the negative input of the positive rail comparator 24.
This offset is adjusted to a level required to provide sufficient
voltage for the output stage of the amplifier 14 to drive the load,
as shown the speaker 100, but also keeps the voltage across the
output devices of the power amplifier 14 low so as to keep
dissipated heat to a minimum. The output voltage of the positive
adaptive rail 21 of the positive charge pump circuit 40 will track
the output voltage swing of the amplifier 14 within the predefined
offset voltage set by the rail offset 27 regardless of any drop in
stored charge in the capacitors C1P and C2P of the positive charge
pump circuit 40 (seen in FIG. 4) due to the fact that the system
operates within a feedback loop. The system compares the output
signal of the power amplifier 14 plus the offset voltage 27 with
the positive rail voltage 21 and switches the positive charge pump
circuit 40 as required to keep the positive rail voltage 21 within
the specified offset regardless of any loss in the stored charge of
the capacitors C1P and C2P in the charge pump circuit 40 (see FIG.
4). As mentioned above, when the output voltage swing of the
amplifier 14 drops below the battery voltage by an amount equal to
the offset voltage 27, the positive rail voltage 21 returns back to
the voltage of the battery. Thus, the positive charge pump circuit
40 allows an increase above the battery voltage up to two times the
battery voltage as required to track the output voltage swing of
the amplifier 14.
[0058] When the output voltage swing of the amplifier 14 approaches
ground, the is output of the negative rail comparator 25 changes
from high to low. Due to the positive feedback of its hysteresis
resistor 29, the output of the negative charge pump circuit 41 will
decrease only a small voltage, based again on the amount of
hysteresis. This small voltage decrease will cause the negative
rail comparator 25 to change states again to high. This slight
positive/negative cycle will continue, as long as the output signal
swing of the power amplifier 14 requires more voltage than the
ground side of the battery. This will provide a series of output
pulses at the negative rail comparator 25 changing in pulse width
and frequency based on the current demand on the output of the
negative charge pump circuit 41. As the power amplifier 14 starts
to swing positive with respect to the battery ground, the negative
adaptive rail 23 will reach a quiescent point and the power
amplifier 14 will now draw its power from the battery ground. The
voltage difference between the output of the power amplifier 14 and
the negative adaptive rail 23 is also further defined by the rail
offset 27 applied to the negative input of the negative rail
comparator 25. The system compares the output signal of the power
amplifier 14 with the negative rail voltage 23 plus the offset
voltage 27 and switches the negative charge pump circuit 41 as
required to keep the negative rail voltage 23 within the specified
offset regardless of any loss in the stored charge of the
capacitors C1N and C2N in the negative charge pump circuit 41 (see
FIG. 5).
[0059] Both charge pump circuits 40 and 41 receive the battery
voltage applied at B1 and at G1 the battery ground. The charge pump
circuits 40 and 41 are capable of providing large amounts of output
current at the positive adaptive rail 21 and the negative adaptive
rail 23. This allows the amplifier 14 to swing nearly three times
the typical output voltage that would otherwise be available
without the added voltage increase from charge pump circuits 40 and
41.
The Charge Pump Circuits
[0060] Looking now at FIG. 4, a partial block, partial schematic
diagram of the positive charge pump circuit 40 of FIG. 3 is shown.
A positive dead zone circuit 30P receives the positive adaptive
rail control signal 20 from its comparator 24. The dead zone
circuit 30P provides an output signal with a minimum of 10
microseconds dead time between the negative going signal for the
P-channel drive 31P and the negative going signal for the N-channel
drive 32P. This avoids having the P-channel MOSFET 41P and the
N-channel MOSFET 42P turned on at the same time, avoiding
overheating of the switching devices or the device destruction
which would occur if both MOSFETs 41P and 42P were on at the same
time. The dead zone circuit 30P also provides a 10 microseconds
dead time when the positive adaptive rail control signal 20 goes
positive. This will ensure that the N-channel power MOSFET
transistor 42P is completely off before switching on the P-channel
power MOSFET transistor 41P. When the positive charge pump circuit
40 is powered on, the N-channel MOSFET transistor 42P will be
turned on, connecting the negative plate of the first positive
capacitor C1P to ground. This will charge the first positive rail
capacitor C1P through its schottky diode D1P connected to the B1
terminal, which is connected to the 12 volt battery or other power
source. The first positive rail capacitor C1P is the flying
capacitor of the positive charge pump circuit 40, which transfers
its charge through another schottky diode D2P into a second
positive rail capacitor C2P, providing the output voltage 21 of the
positive adaptive pulsed rail 21. Both diodes D1P and D2P are
schottky diodes which provide low forward voltage drop and fast
switching response. The battery voltage at B1 is also fed directly
to the adaptive positive rail 21 through a third diode D3P. When
the positive adaptive rail control signal 20 goes negative, the
N-channel power MOSFET transistor 42P switches off and the
P-channel power MOSFET transistor 41P switches on, connecting the
negative plate of the flying capacitor C1P to the B1 terminal,
allowing forward conduction of the schottky diode D2P and
transferring voltage from the flying capacitor C1P to the positive
is pulsed rail 21. The source connection of the P-channel power
MOSFET transistor 41P and the source connection of the N-channel
power MOSFET transistor 42P connect to the B1 terminal and ground
connection through ferrite beads L1P and L2P, respectively, which
reduce the high speed switching transients reducing RFI emissions
of the circuit. The second positive rail capacitor C2P is typically
between 100 uF and 1000 uF for subwoofer applications which allows
a fast release time for the positive adaptive rail 21. When the
positive adaptive rail control signal 20 goes high, the P-channel
MOSFET transistor 41P turns off and the N-channel MOSFET transistor
42P turns on again, charging the positive rail flying capacitor
C1P.
[0061] FIG. 5 is a partial block, partial schematic diagram of the
negative charge pump circuit 41 of FIG. 3. Its dead zone circuit
30N receives the negative adaptive rail control signal 22 from its
comparator 25. The negative dead zone circuit 30N provides an
output signal with a minimum of 10 microseconds dead time between
the negative going signal for the P-channel drive 31N and the
negative going signal for the N-channel drive 32N. This avoids
having the P-channel MOSFET 41N and the N-channel MOSFET 42N turned
on at the same time, avoiding overheating of the switching devices
and the device destruction which would occur if both negative
charge pump MOSFETs 41N and 42N were on at the same time. The
negative dead zone circuit 30N also provides a 10 microseconds dead
time when the negative adaptive rail control signal 22 goes
negative. This will ensure that the N-channel power MOSFET
transistor 42N is completely off before switching on the P-channel
power MOSFET transistor 41N.
[0062] When the negative charge pump circuit 41 is powered on, the
P-channel MOSFET transistor 41N will be turned on, connecting the
positive plate of its flying capacitor C1N to the terminal B1. This
will cause the negative rail flying capacitor C1N to charge in
cooperation with a first negative rail schottky diode D1N connected
to the ground terminal G1, which is connected to the 12 volt
battery ground or other power source. The flying capacitor C1N of
the negative charge pump circuit 41 transfers its charge through
another schottky diode D2N into a second capacitor C2N, providing
the output voltage 23 of the negative adaptive pulsed rail 23. Both
diodes MN and D2N are schottky diodes which provide low forward
voltage drop and fast switching response. The battery ground is
also fed directly to the adaptive negative rail 23 through a third
diode D3N. When the positive adaptive rail control signal 22 goes
positive, the P-channel power MOSFET transistor 41N switches off
and the N-channel power MOSFET transistor 42N switches on,
connecting the positive plate of the negative rail flying capacitor
C1N to the ground terminal and allowing forward conduction of its
schottky diode D2N, transferring voltage from the flying capacitor
C1N to the negative rail 23. The source connection of the P-channel
power MOSFET transistor 41N and the source connection of the
N-channel power MOSFET transistor 42P connect to the B1 terminal
and ground connection through ferrite beads L1P and L2P and L1N and
L2N, respectively, which reduce the high speed switching transients
and reduce RFI emissions of the circuit. The second negative rail
capacitor C2N is typically between 100 uF and 1000 uF for subwoofer
applications which allows a fast release time for the negative
adaptive rail 23. When the negative adaptive rail control signal 22
goes low, the P-channel MOSFET transistor 41P turns off and the
N-channel MOSFET transistor 42N turns on, again charging the
negative rail flying capacitor C1N.
[0063] FIG. 6 is a graph of the output voltage swing of the
adaptive rail amplifier showing the power amplifier output signal
and the positive and negative adaptive rail signals 21 and 23. The
amplifier output signal starts at VREF 28, which is shown in this
graph at 7 volts and is based on a battery supply voltage of 14
volts. This will allow the amplifier output signal swing to be
centered between the 14 volt battery voltage and battery ground at
0 volts DC. When the output signal swing goes positive by more than
battery voltage minus the preset rail offset voltage 27, the
positive supply rail increases with a series of positive going
pulses, producing the positive adaptive rail 21. The positive
adaptive rail 21 will track the output signal offset positive by
the rail offset 27. When the output signal drops below the 14 volt
battery voltage plus the pre-defined rail offset 27, the positive
rail returns back to the battery voltage at 14 volts. As the output
signal swings negative and approaches the negative rail at 0 volts
DC by more than the predefined rail offset 27, the negative rail
increases negative with a series of negative going pulses,
producing the negative adaptive rail 21.
[0064] FIG. 7 is a graph of the positive output signal swing of a
100 Hz sine wave showing the adaptive rail signal 21 and the
positive pulsed rail control signal 20 of FIG. 3. FIG. 7 shows in
greater detail only the positive going portion of the signal shown
in FIG. 6 and further includes the positive pulsed rail control
signal 20 in FIG. 3. As the output signal swings positive by an
amount equal to the positive battery voltage 14 minus the
predefined rail offset, the positive pulsed rail control starts to
generate a series of pulses, which changes in pulse width depending
on the amount of output current required to source the positive
supply voltage to the power amplifier 14 in FIG. 3. The negative
going pulses get wider as the output signal swings more positive
and the demand for output current increases. When the signal starts
to swing negative from the maximum peak swing, the output pulses
change in both frequency and width until the output signal swings
negative by more than the predefined rail offset 27 with respect to
the 14 volt battery voltage. The positive pulsed rail control
signal 20 stops changing and provides a positive signal allowing
the flying capacitor C1P in the positive charge pump circuit 40 of
FIG. 4 to fully charge. The negative going signal will produce a
similar but negative response from the negative adaptive rail
control signal 22 and the negative charge pump circuit 41 shown in
FIG. 3.
Typical Power Amplifier
[0065] Turning now to FIG. 8, a simplified schematic diagram of a
typical power amplifier 14 of FIG. 3 is shown. The power amplifier
14 includes a preamplifier/pre-drive circuit P1 and a bipolar
output stage including transistors Q5 and Q6. The preamplifier P1
receives a positive input signal via one resistor R3 plus VREF 28
bias via another resistor R4. VREF 28 is also applied via a
resistor R1 to the negative input of the preamp/pre-drive circuit
P1. The preamp circuit P1 receives its power directly from the
battery voltage applied at B1 and battery ground. The pre-drive
output signal drives the base of both transistors Q5 and Q6 which
receive the positive adaptive rail voltage 21 and the negative
adaptive rail voltage 23, respectively. Amplifier negative feedback
is provided via another resistor R2.
[0066] Many possible amplifier configurations will operate with the
disclosed invention, including Class A/B, Class B, Class D and
output stages utilizing bipolar or MOSFET transistors. As shown,
the preamp/pre-drive circuit is connected directly to the battery
voltage, but this stage of the power amplifier 14 can also be
configured to connect to the adaptive rails 21 and 22. If a single
amplifier 14 is used, as in FIG. 8, the output speaker 100 will
need to be decoupled from the amplifier by use of a large value
capacitor (not shown) so as to eliminate the VREF bias from the
output signal applied to the speaker 100. The invention can also be
implemented with dual amplifiers in bridge mode, effectively
doubling the output voltage swing across the speaker load and
eliminating the need for output decoupling capacitors.
[0067] While the above-disclosed embodiments of the invention
provide excellent performance and improved manufacturing costs over
the prior art, it may be desirable to eliminate all of the
switching transients for use in certain automotive applications so
as to eliminate virtually all of the RFI/EMI emissions of the
amplification system. As can be seen with reference to FIG. 3, the
system operates with a large gain in the feedback loop due to the
comparators 24 and 25. The invention can also be implemented with
reduced loop gain, allowing more linear operation of the adaptive
rails and eliminating the high frequency switching transients of
the pulsed rails. The main tradeoff of this embodiment is that the
MOSFETs that drive the positive adaptive rail and negative adaptive
rail will see a slight increase in dissipated heat due to using
them in a linear mode of operation.
Linear Mode Adaptive Rail Power Amplifier
[0068] In FIG. 9, a linear mode embodiment of the invention
provides non-pulsed adaptive rails with reduced EMI emissions. The
FIG. 9 circuit is the same basic circuit as shown in FIG. 3 with
reduced feedback loop gain by changing the comparators 24 and 25
shown in FIG. 3 to differential amplifiers 24 and 25, as shown in
FIG. 9. The power amplifier 14 receives both positive and negative
power directly from the battery due to the schottky diodes D3P and
D3N and also to the positive and negative charge pump circuits 40
and 41, respectively. The amplifier input is biased at VREF 28,
which is 1/2 the auto battery voltage. As described above, the
battery voltage will be between 12 volt s without the engine
running and 14.4 volts with the alternator charging the battery.
This means that
[0069] VREF 28 will track at 1/2 the battery voltage, keeping the
output voltage swing of the amplifier 14 centered between the
positive battery voltage and ground. A differential amplifier 24
provides an output signal which is the difference between the
positive adaptive rail voltage 21 applied to the negative input via
one resistor R6 and the amplifier output signal applied to the
positive input by another resistor R7. A third resistor R5 provides
negative feedback for the differential amplifier 24 and a fourth
resistor R8 provides positive rail offset 27P applied to the
positive input. The rail offset voltage 27P applied to the fourth
resistor R8 is positive, typically 1.5 volts, and provides the
proper offset required for the positive adaptive rail 21 to track
the output signal. The gain of the differential amplifier 24 is
selected to be less than one and typically on the order of 0.3 in
order to avoid clipping the differential amplifier with a full
scale output signal at the output of the amplifier 14. If the
differential amplifier 24 clips prior to detecting the maximum
output signal swing, the positive adaptive rail control signal 20
will not provide proper rail tracking. The differential output
signal 20 is fed to the input of the positive charge pump circuit
40. The linear embodiment of the invention also operates in a
feedback loop by taking the difference between the output
amplifier's signal plus the offset voltage 27P and the positive
rail voltage 21 and controls the output voltage of the positive
charge pump circuit 40 as required to keep the rail voltage 21
within the specified offset regardless of any loss in the stored
charge of the capacitors C1P and C2P in the charge pump circuit 40
(See FIG. 10).
[0070] Another differential amplifier 25 provides an output signal
which is the difference between the negative adaptive rail 23
applied to the negative input via one resistor R11 and the
amplifier output signal applied to the positive input via another
resistor R10. A third resistor R12 provides negative feedback for
the differential amplifier 25 and a fourth resistor R9 provides
negative rail offset 27N applied to the positive input. The rail
offset voltage 27N applied to the fourth resistor R9 is negative,
typically -1.5 volts, and provides the proper offset required for
the negative adaptive rail 23 to track the output signal. The gain
of the differential amplifier 25 is selected to be less than one
and typically on the order of 0.3 in order to avoid clipping the
differential amplifier 25 with a full scale output signal at the
output of the amplifier 14. If the differential amplifier 25 clips
prior to detecting the maximum output signal swing, the negative
adaptive rail control signal 22 will not provide proper rail
tracking. The differential output signal 22 is fed to the input of
the negative charge pump circuit 41.
[0071] FIG. 10 is a schematic diagram of the positive charge pump
circuit 40 of FIG. 9. The positive charge pump circuit 40 receives
the battery voltage at B1 and ground and battery voltage applied to
B1 is fed to the positive adaptive rail 21 via a schottky diode
D3P. The positive adaptive rail control 20 from the differential
amplifier 24 of FIG. 9 is applied to a resistor R14 and to the
positive input of a positive charge pump comparator U1. The
positive charge pump comparator U1 is connected to 1/2 battery
voltage VREF and the output will be low when the output signal from
the power amplifier 14 of FIG. 9 is less than the battery voltage
minus the rail offset 27P applied to the differential amplifier 24,
also seen in FIG. 9. Continuing to look at FIG. 10, a first
transistor Q8 operates as a switch to provide gate drive for the
N-channel power MOSFET transistor 42P. The emitter of the
transistor Q8 is connected to the battery voltage B1 and its
collector is connected to ground through a resistor R16. The output
of the positive charge pump comparator U1 is connected via a 6 volt
zener diode ZD1 to the transistorbase drive resistor R15. When the
positive charge pump comparator U1 is low, the transistor Q8 is
turned on, providing gate drive and turning on the N-channel power
MOSFET transistor 42P, connecting the negative plate of the
positive rail flying capacitor C1P to ground. A ferrite bead L2P is
connected in series with the source connection of the N-channel
MOSFET transistor 42P in order to reduce the switching transient
from the N-channel power MOSFET transistor 42P. This allows the
positive rail flying capacitor C1P to charge through the schottky
diode D1P. When the positive adaptive rail control 20 goes positive
by more than VREF, the positive charge pump comparator U1 will
switch high, turning the transistor Q8 off and turning off the
N-channel MOSFET 42P. When the positive adaptive rail 20 goes high
by more than VREF plus the VBE of the transistor Q7, the transistor
Q7 will start to conduct, providing gate drive to the P-channel
power MOSFET transistor 41P. If the positive rail flying is
capacitor C1P is fully charged, the P-channel power MOSFET
transistor 41P will operate in its linear region pulling the
negative plate of its flying capacitor C1P positive. As the
P-channel MOSFET transistor 41P starts to conduct, the gate drive
signal supplied from the output of the differential amplifier 24 in
FIG. 9 will be reduced and will thus provide a linear voltage
increase that will track the output amplifier signal swing plus the
rail offset voltage 27P. As the signal swing drops below the
battery voltage plus the rail offset voltage, the P-channel power
MOSFET transistor 41P will stop conducting. When the differential
amplifier 24 output signal swings below VREF, the positive charge
pump comparator U1 will switch low, turning on the N-channel power
MOSFET transistor 42P, again providing charging for the positive
rail flying capacitor C1P. The dead zone operation is provided due
to the fact that the comparator U1 switches at VREF and the
transistor Q7 starts to conduct only when the positive adaptive
rail control signal 20 increases above VREF by one VBE. This
ensures that both N-channel and P-channel power MOSFET transistors
41P and 42P cannot conduct at the same time.
[0072] In FIG. 11, the negative charge pump circuit 41 of FIG. 9
receives battery voltage at B1 and ground. Battery ground is fed to
the negative adaptive rail 23 via the schottky diode D3N. The
negative adaptive rail control 22 from the differential amplifier
25 of FIG. 9 is applied to R19 and to the positive input of a
negative charge pump comparator U2. The negative charge pump
comparator U2 negative input is connected to 1/2 battery voltage
VREF and the output will be high when the output signal from the
amplifier 14 of FIG. 9 is greater than the battery ground plus the
rail offset 27N applied to the differential amplifier 25 also of
FIG. 9. The output of the negative charge pump comparator U2 is
connected to a 6 volt zener diode ZD2 and to the base drive
resistor R17. Transistor Q9 operates as a switch and will switch
off when the negative charge pump comparator U2 is high. The
emitter of the transistor Q9 is connected to B1 battery voltage and
its collector is connected to ground through a resistor R18. When
Q9 is switched off, the P-Channel power MOSFET transistor 41N is
turned on due to a resistor R18 pulling the P-Channel gate drive
31N low. When P-Channel power MOSFET transistor 41N is switched on,
the positive plate of the negative rail flying capacitor C1N is
tied to B1 providing ch, transistor Q9 is turned on eliminating
gate drive 31N and turning off the p-channel power MOSFET
transistor 41N. A ferrite bead L1N is connected in series with the
source connection of the MOSFET transistor 41N in order to reduce
the switching transient from the power MOSFET transistor 41N. When
the negative adaptive rail 22 goes negative by more than VREF, the
negative charge pump comparator U2 will switch low, turning the
transistor Q9 on and turning off the P-channel MOSFET 41N. When the
negative adaptive rail control 22 is low by more than VREF plus the
VBE of the other transistor Q10, the transistor Q10 will start to
conduct, providing gate drive to the N-channel power MOSFET
transistor 42N. If the negative rail flying capacitor C1N is fully
charged, the N-channel power MOSFET transistor 42N will operate in
its linear region, pulling the positive plate of the negative rail
flying capacitor C1N negative. As the MOSFET transistor 42N starts
to conduct, the gate drive signal supplied from the output of the
differential amplifier 25 in FIG. 9 will be reduced and will thus
provide a linear negative voltage increase that will track the
output amplifier signal swing plus the rail offset voltage 27N from
FIG. 9. As the signal swing goes above battery ground plus the rail
offset voltage, the N-channel power MOSFET transistor 42N will stop
conducting. When the output signal of the differential amplifier 25
swings above VREF, the negative charge pump comparator U2 will
switch high, turning on the P-channel power MOSFET transistor 41N
again, providing charging for the negative rail flying capacitor
C1N. Dead zone operation is provided due to the fact that the
negative rail comparator is U2 switches at VREF and the transistor
Q9 starts to conduct only when negative adaptive rail signal 22
increases below VREF by one VBE. This ensures that both the
N-channel power MOSFET and the P-channel power MOSFET transistors
cannot conduct at the same time. Both positive and negative
adaptive rails will track the amplifier output voltage swing plus
the predefined positive and negative rail offset voltage up to the
point that the signal swing in either direction exceeds the stored
capacitor voltage available in the flying capacitors C1P and C1N in
the positive and negative charge pump circuits 40 and 41. The
linear embodiment shown in FIG. 9 provides an easy to manufacture
design with reduced parts count and complexity.
[0073] FIG. 12 compares the positive output signal swing of a 100
Hz sine wave with the positive adaptive rail signal 21 and the
positive adaptive rail control signal 20 of FIG. 9. FIG. 12 shows
only the positive going portion of a signal from FIG. 9 and further
includes the positive adaptive rail control 20 in FIG. 9. As the
output signal swings positive by an amount equal to the positive
battery voltage 14 minus the predefined rail offset, the positive
adaptive rail control 20 starts to increase in voltage. The
positive adaptive rail 21 linearly tracks the output signal plus a
rail offset determined by the rail offset 27P in FIG. 9. When the
signal starts to swing negative from the maximum peak swing, the
output voltage continues to track the output signal until the
output signal drops below the battery voltage, shown as 14 volts
plus the rail offset voltage. The output of the difference
amplifier 24 in FIG. 9, which produces positive adaptive rail
control 20 shown in FIG. 12, will remain constant as the adaptive
rail tracks the output signal so as to maintain a constant linear
rail offset which tracks the output signal. The negative going
signal will produce similar negative response from the negative
adaptive rail control 22 and negative charge pump circuit 41 shown
in FIG. 9.
Single Comparator Switched Charge Pump Tracking Rail
[0074] Referring now to FIG. 13, a single comparator switched
charge pump tracking rail embodiment of the invention is shown.
FIG. 13 uses the same reference designations as in FIG. 9 where
similar circuit operation is disclosed. The power amplifier 14
output is connected to the positive input of the comparator 44 and
the negative input of the comparator 44 is connected to VREF, which
is 1/2 supply voltage. The comparator 44 provides a zero crossing
detector and may include positive hysteresis so as to avoid
comparator switching due to noise or very low level signals. When
the output signal from the power amplifier 14 swings positive above
VREF, the output 45 of the comparator 44 switches high and when the
output signal of the power amplifier 14 swings negative below VREF,
the output 45 of the comparator 44 switches low. The comparator 44
generates a control signal 45 to control the positive and negative
charge pump circuits 40 and 41, which generates a positive 2X
voltage at the output 42 of the positive charge pump circuit 40 and
a negative 2X output voltage at the output 43 of the negative
charge pump circuit 41. The charge pump circuits 40 and 41 are
clocked by the audio signal even when the audio signal is low and
does not require any additional output swing beyond the limits of
the supplied battery voltage. Upon power up, the flying capacitors
C1P and C1N in both charge pump circuits 40 and 41 are charged up
to the battery voltage. With each positive swing of the output
signal of the power amplifier 14, the internal flying capacitor C1P
of the positive charge pump circuit 40 will transfer a charge to an
internal output storage capacitor C1P within the positive charge
pump circuit 40, thereby increasing the positive charge pump output
voltage 42. With each negative swing of the output signal of the
power amplifier 14, the internal flying capacitor C1N of the
negative charge pump circuit 41 will transfer a charge to an
internal output storage capacitor C2N within the negative charge
pump circuit 41, thereby increasing the negative charge pump output
voltage 43. The positive 2X voltage 42 is fed to the input of the
positive adaptive rail circuit 50. The positive adaptive rail
circuit 50 is controlled via the control signal 20 from the
differential amplifier 24 and provides a positive adaptive rail
output 21 which increases the positive power supply voltage above
the battery voltage B1 which is connected to a schottky diode D3P.
The nominal rail voltage 21 will be the same as the battery voltage
B1 less the forward drop of the schottky diode D3P and will
increase up to two times the battery voltage less any capacitor
voltage sag that may occur in the positive charge pump circuit 40.
The negative 2X voltage 43 is fed to the input of the negative
adaptive rail circuit 51. The negative adaptive rail circuit 51 is
controlled via the control signal 22 from the differential
amplifier 25 and provides a negative adaptive rail output 23 which
increases the negative power supply voltage below the battery
ground connected to a schottky diode D3N. The nominal rail voltage
23 will be the same as battery ground less the forward drop of the
schottky diode D3N and will increase negative up to two times the
battery voltage less any capacitor voltage sag that may occur in
the negative charge pump circuit 41. When the output voltage swing
of the power amplifier 14 approaches the positive rail voltage 21
minus the rail offset 27P, the positive adaptive rail circuit 50
will become active and increase the positive rail 21 so as to keep
the positive rail above the output voltage swing by an amount equal
to the rail offset 27P. When the output voltage swing of the power
amplifier 14 approaches the negative rail voltage 23 minus the rail
offset 27N, the negative adaptive rail circuit 51 will become
active and increase the negative rail voltage 23 so as to keep the
negative rail below the amplifier 14 output voltage swing by an
amount equal to the rail offset 27N.
[0075] Looking at FIG. 14, a modification to the embodiment of FIG.
13 can be made using two separate comparators 44P and 44N in place
of the single comparator 44. That would allow the flying capacitors
C1P and C1N of both the positive and the negative charge pump
circuits 40 and 41 to charge on power up or idle (no audio signal
present). One comparator 44P would compare the amplifier output
signal swing with a positive reference voltage 28 P to control the
positive charge pump circuit 40 and the second comparator 44N would
compare the amplifier output signal swing with a negative reference
voltage 28N to control the negative charge pump circuit 41. This
modification would allow the flying capacitors C1P and C1N in the
positive and negative charge pump circuits 40 and 41 to both fully
charge when the system is switched on or when no audio is present.
Other modifications will become apparent to the skilled
artisan.
[0076] Referring now to FIG. 15, the positive charge pump circuit
40 of the embodiment of FIG. 14 is the same circuit as shown in
FIG. 10 without the shottky diode D3P and with changes in value for
the capacitors C1P and C2P to optimize performance for this
embodiment. As a result, the detailed description of FIG. 15 is the
same as the description given for FIG. 10. After a few switching
cycles of the comparator 44P of FIG. 14, the second positive charge
pump capacitor C2P will charge up to a voltage equal to two times
the battery voltage minus the forward diode drop of the schottky
diodes D1P and D2P. There is no discharge path for the stored
charge in the capacitor C2P until the positive adaptive rail
circuit 50 of FIG. 14 becomes active, so the stored charge will
remain at this peak until the positive adaptive rail circuit 50
becomes active. When the positive adaptive rail circuit 50 of FIG.
14 becomes active, the positive charge pump comparator 44P will be
switched positive, turning on the P-channel MOSFET 41P of FIG. 15.
This allows the stored charge in both positive charge pump
capacitors C1P and C2P to provide current to increase the positive
adaptive rail 21 of FIG. 14. This provides additional benefits by
increasing the maximum stored charge available to the positive
adaptive rail circuit 50 and decreasing the inrush current required
to charge the single positive rail flying capacitor C1P on startup.
The capacitor C2P and the diode D2P could be omitted, reducing the
maximum available stored charge and providing a switched output
voltage at 42 when the positive charge pump comparator 44P of FIG.
14 switches positive. The output signal 42 would connect to the
cathode side of the schottky diode D1P in this configuration. The
main advantage of this modification is reduced circuit
complexity.
[0077] Referring now to FIG. 16, the negative charge pump circuit
41 of the embodiment of FIG. 14 is shown. FIG. 16 is the same
circuit as shown in FIG. 11 without the schottky diode D3N and with
changes in value for the capacitors C1N and C2N to optimize
performance for this embodiment. As a result, the detailed
description of FIG. 16 is the same as the description given for
FIG. 11. After a few switching cycles of the comparator 44N of FIG.
14, the second negative charge pump capacitor C2N will charge up to
a voltage equal to minus two times the battery voltage minus the
forward diode drop of the schottky diodes D1N and D2N. There is no
discharge path for the stored charge in the capacitor C2N until the
negative adaptive rail circuit 51 of FIG. 14 becomes active, so the
stored charge will remain at this negative peak until the negative
adaptive rail circuit 51 becomes active. When the negative adaptive
rail circuit 51 of FIG. 14 becomes active, the negative charge pump
comparator 44N will be switched negative. Looking again at FIG. 16,
this turns on the N-channel MOSFET 42N and allows the stored charge
in both negative charge pump capacitors C1N and C2N to provide
current to increase the negative adaptive rail 23 of FIG. 14. This
provides additional benefits by increasing the maximum stored
charge available to the negative adaptive rail circuit 51 and
decreasing the inrush current required to charge the single
capacitor C1N on startup. The capacitor C2N and diode D2N could be
omitted, reducing the maximum available stored charge and providing
a switched output voltage at 43 when the negative is charge pump
comparator 44N of FIG. 14 switches negative. The output signal 43
would connect to the anode side of the schottky diode MN in this
configuration. As mentioned above, the main advantage of this
modification is reduced circuit complexity.
[0078] FIGS. 17 and 18 show the positive and negative adaptive rail
circuits 50 and 51 of FIG. 14, respectively. As seen in FIG. 17,
the positive adaptive rail circuit 50 includes the P-channel power
MOSFET transistor 52 which receives the input signal 42 and
provides a variable output voltage 21. The P-channel power MOSFET
52 is controlled by a small signal transistor Q7. The transistor Q7
receives the control signal 20 through a resistor R14 to the base
connection of the transistor Q7. The emitter of the transistor Q7
is tied to VREF 1/2 supply and the collector of the transistor Q7
is connected to the gate of the P-channel MOSFET 52 through a gate
resistor and also to a resistor R13 which is tied to input signal
42. In operation, when a positive going control signal 20 appears,
the transistor Q7 becomes conductive, providing gate drive to the
P-channel MOSFET 52 which in turn increases the output voltage 21,
providing an increase above the battery voltage seen at the power
amplifier 14 of FIG. 14. By connecting the resistor R13 to the
input signal 42, the power P-channel MOSFET 52 does not become
active until the transistor Q7 starts to conduct. The differential
amplifier 24 of FIG. 14 provides the control signal to the base of
the transistor Q7. The output of the differential amplifier 24 is a
voltage equal to the difference between the power amplifier 14
output voltage swing and the positive adaptive rail 21 plus the
rail offset voltage 27P. This will keep the positive adaptive rail
voltage 21 above the output signal swing by an amount equal to the
rail offset 27P at all times up to the point where the output
signal swings beyond the available voltage at 42 and the output
clips of the power amplifier 14.
[0079] Turning to FIG. 18, the negative adaptive rail circuit 51
includes the N-channel power MOSFET transistor 53 which receives
the input signal 43 and provides a variable output voltage 23. The
N-channel power MOSFET 53 is controlled by a small signal
transistor Q10. The transistor Q10 receives the control signal 22
through a resistor R19 to the base connection of the transistor
Q10. The emitter of the transistor Q10 is tied to VREF 1/2 supply
and the collector of the transistor Q10 is connected to the gate of
the N-channel MOSFET 53 through a gate resistor and also to a
resistor R20 which is tied to the input signal 43. In operation,
when a negative going control signal 22 appears, the transistor Q10
becomes conductive, providing gate drive to the N-channel MOSFET 53
which in turn increases the negative output voltage 23, providing
an increase below the battery ground G1 seen at the power amplifier
14 of FIG. 14. By connecting the resistor R20 to the input signal
43, the power N-channel MOSFET 53 does not become active until the
transistor Q10 starts to conduct. The differential amplifier 25 of
FIG. 14 provides the control signal to the base of the transistor
Q10. The output of the differential amplifier 25 is a voltage equal
to the difference between the output voltage swing of the power
amplifier 14 and the negative adaptive rail 23 plus the rail offset
voltage 27N. This will keep the negative adaptive rail voltage 23
below the output signal swing by an amount equal to the rail offset
27N at all times up to the point where the output signal swings
negative beyond the available voltage at 43 and the output clips of
the power amplifier 14.
[0080] In operation, all embodiments of the invention will produce
a signal swing of nearly 40 volts peak into an 8 ohm load which
will deliver 100 watts and nearly 180 watts into a 4 ohm load. As
previously noted, higher impedance 8 ohm and 4 ohm speakers have
higher sensitivity ratings and will therefore provide higher output
sound pressure levels than the lower impedance speakers of 2 or 1
ohms. The invention, therefore, also allows higher sound pressure
levels with reduced current consumption.
Simplified Embodiment of the Adaptive Rail Power Amplifier
[0081] Turning now to FIG. 19, typical Battery voltage of 12 volts
(14.4 volts with the alternator running) is applied at B1 and
Battery Ground is applied at G1. Power amplifier 14 receives a
positive power supply voltage from Positive Adaptive Rail 21 and
receives negative power supply voltage from Negative Adaptive Rail
23. With no input signal, or a low level input signal, the voltage
on Positive Adaptive Rail 21 will equal the Battery voltage less
one diode drop from schottky diode D1P and the Negative Adaptive
Rail 23 will equal the Battery ground less one diode drop from
schottky diode D1N. A reference voltage of 1/2 B1 (battery voltage)
is applied as signal VREF as a bias reference at the input and also
at nodes indicated as VREF. The VREF signal will track the battery
and provide 1/2 battery reference voltage as the battery voltage
fluctuates. The output of amplifier 14 will be centered at VREF
half way between B1, the positive battery voltage, and G1, the
ground side of the battery. The emitter of Q1 is connected to node
21, which is equal to B1 minus the forward diode drop of schottky
diode D1P. The base of Q1 is connected to 5.1K resistor R1 and to
the cathode side of zener diode D2. The anode side of D2 is
connected to the output of amplifier 14. When the output swing of
amplifier 14 is low, base current will flow in Q1, turning Q1 on,
which will in turn pull gate resistor R2 positive ensuring that
P-CHANNEL power MOSFET transistor will be turned off. This keeps
P-CHANNEL power MOSFET transistor 41P turned off until the output
voltage swing of amplifier 14 exceeds the point where Q2 switches
off plus a predefined dead zone voltage. Q2 emitter is tied to VREF
and base resistor R5 is tied to ground which will turn on Q2 with
no input signal thereby turning on N-CHANNEL power MOSFET
transistor 42P by pulling gate resistor R7 positive causing storage
capacitor C1P to charge through schottky diode D1P. When the output
swing of amplifier 14 increases positive above VREF by more than
one volt Q2 will switch off pulling the gate of N-CHANNEL power
MOSFET transistor 42P low due to collector connected pull down
resistor R6. Q2's base drive is a result of the divider voltage of
R5/R4+R5 times VOUT of power amplifier 14. The VBE of Q2 is
effectively cancelled out by the forward drop of diode D3 making
the switch point of Q2 as described above. With the values shown in
FIG. 1 and a battery voltage of 12 volts Q2 will switch off when
the output signal swing exceeds approximately 7 volts positive, one
volt above VREF. As the output signal of amplifier 14 continues to
swing positive approaching B1, Q1 will stop conducting. When power
amplifier 14 output signal approaches B1, minus the diode drop of
schottky diode D1P, minus the VBE of transistor Q1 and the zener
voltage of zener diode D2 transistor Q1 will stop conducting. Zener
diode D2 will typically be a 1N5221 which has a 2.4 volt zener
voltage. This will provide a dead zone of approximately 1.6 volts
between the point where transistor Q2 switches off and the point
where transistor Q1 will start to turn off allowing P-CHANNEL power
MOSFET transistor 41P to become active. The Source connection of
P-CHANNEL power MOSFET transistor 41P is connected directly to the
B1 positive battery connection. As power MOSFET transistor 41P
becomes active it will pull the negative plate of storage capacitor
C1P positive towards the B1 battery potential thereby increasing
the Positive Adaptive Rail 21 supply voltage. As the Positive
Adaptive Rail 21 increases, Q1's emitter voltage increases
positive, which keeps Q1 in a linear or non-saturated condition
providing a positive adaptive power supply rail that will track the
output signal by 3.4 volts positive above the output signal swing.
If the output voltage swing exceeds the battery voltage plus the
available stored charge in capacitor C1P, the positive output swing
will start to clip. The Positive Adaptive Rail voltage will start
to increase above the battery voltage when the output voltage swing
increases positive by more than B1 minus the forward drop of
schottky diode D1P, minus the VBE of Q1 minus the zener voltage of
zener diode D2. With a 2.4 volt zener diode this offset will be
approximately 3.4 volts. The offset voltage can be increased by
changing zener diode D2, thus it will be apparent to the skilled
artisan that increasing the zener voltage will increase the offset
voltage by an equal amount.
[0082] The operation of the Negative Adaptive Rail will now be
described. The emitter of Q4 is connected to node 23, which is
equal to G1 minus the forward diode drop of schottky diode D1N. The
base of Q4 is connected to 5.1K resistor R14 and to the anode side
of zener diode D5. The cathode side of D5 is connected to the
output of amplifier 14. When the output voltage swing of amplifier
14 is low, base current will flow in Q4, turning Q4 on, which will
in turn pull gate resistor R9 negative ensuring that P-CHANNEL
power MOSFET transistor 42N will be turned off. This keeps
P-CHANNEL power MOSFET transistor 42N turned off until the output
voltage swing of amplifier 14 swings negative and exceeds the point
where Q3 switches off plus a predefined dead zone voltage. Q3
emitter is tied to VREF and base resistor R9 is tied to B1 which
will turn on Q3 with no input signal thereby turning on N-CHANNEL
power MOSFET transistor 41N by pulling gate resistor R10 positive
causing storage capacitor C1N to charge through schottky diode D1N.
When the output swing of amplifier 14 increases negative below VREF
by more than one volt Q3 will switch off pulling the gate of
N-CHANNEL power MOSFET transistor 41N high due to collector
connected pull up resistor R8. Q3's base drive is a result of the
divider voltage of R9/R11 +R9 times VOUT of power amplifier 14. The
VBE of Q3 is effectively cancelled out by the forward drop of diode
D4 making the switch point of Q3 as described above. With the
values shown in FIG. 1 and a battery voltage of 12 volt s Q3 will
switch off when the output signal swing exceeds approximately 7
volts negative, one volt below VREF. As the output signal of
amplifier 14 continues to swing negative approaching G1, Q4 will
stop conducting. When power amplifier 14 output signal approaches
G1, minus the diode drop of schottky diode D1N, plus the VBE of
transistor Q4 and the zener voltage of zener diode D5 transistor Q3
will stop conducting. Zener diode D5 will typically be a 1N5221
which has a 2.4 volt zener voltage. This will provide a dead zone
of approximately 1.6 volts between the point where transistor Q3
switches off and the point where transistor Q4 will starts to turn
off allowing N-CHANNEL power MOSFET transistor 42N to become
active. The Source connection of N-CHANNEL power MOSFET transistor
42N is connected directly to the G1 negative battery connection. As
power MOSFET transistor 42N becomes active it will pull the
positive plate of storage capacitor C1N negative towards the G1
battery ground potential thereby increasing the Negative Adaptive
Rail 23 supply voltage. As the Positive Adaptive Rail 23 increases
negative Q4's emitter voltage increases negative, which keeps Q4 in
a linear or non-saturated condition providing a negative adaptive
power supply rail that will track the output signal by 3.4 volts
negative below the output signal swing. If the output voltage swing
exceeds the battery ground potential plus the available stored
charge in capacitor C1N, the negative output swing will start to
clip. The Negative Adaptive Rail voltage will start to increase
below the battery ground potential when the output voltage swing
increases negative by more than G1 minus the forward drop of
schottky diode D1N, minus the VBE of Q4 and the zener voltage of
zener diode D5. With a 2.4 volt zener diode this offset will be
approximately 3.4 volts. The offset voltage can be increased by
changing zener diode D5, thus it will be apparent to the skilled
artisan that increasing the zener voltage will increase the
negative offset voltage by an equal amount.
[0083] The circuit of FIG. 19 will provide a positive and negative
adaptive rail that will track the output signal by a predefined
offset voltage and will charge the storage capacitors C1P and C1N
when the power amplifier 14 output signal is low in amplitude or on
power up when the amplifier output is zero. The circuit of FIG. 19
will also provide accurate positive and negative adaptive tracking
rails regardless of the amount of drop in voltage in the storage
capacitors C1P and C1N. In operation the output signal swing of
amplifier 14 and positive adaptive rail signal 21 will be virtually
identical to that shown graphically in FIG. 12 of the co-pending
application.
[0084] It will be apparent to the skilled artisan that the circuit
of FIG. 19 can be implemented as a bridge mode design with one
amplifier driving one side of the speaker in phase and a second
channel of amplification driving the other side of the speaker out
of phase, thereby providing twice the voltage swing and four times
the output power. It will also be apparent to the skilled artisan
that the circuit shown in FIG. 19 will have advantages and can be
implemented in a bipolar power supply design with a positive
voltage, a ground reference and a negative voltage eliminating the
need for a VREF reference bias voltage.
Above-Automotive-Voltage Audio Amplifiers
[0085] Unlike known professional audio amplifiers, and in
accordance with the invention, assume, for example, that the 120V
power supply rails of a professional audio amplifier were lowered,
perhaps to 33.33V. By charging storage capacitors across both the
positive and negative power supplies, an additional 66.66V stored
in the power supply capacitors is available to lift the rails up to
100V. Therefore, the power supply must deliver 25 A at only 33.3V.
If so, the power supplies need output only 832.5 watts peak instead
of the 2500 watts peak of a conventional power amplifier design.
Thus, a professional power amplifier, in accordance with the
invention, can deliver a higher level of output power with the
available VA from an AC mains circuit.
[0086] In the embodiments disclosed above in relation to FIGS.
1-19, the difference between the amplifier output swing and the
dynamic rail voltage is used to generate a control signal that then
controls the charge pump circuits. In the following embodiments of
FIGS. 20-27, pump circuits are used as low gain source followers
with unity gain directly controlled by the output signal swing of
the audio signal power amplifier.
[0087] The following identification of some of the element numbers
used in FIGS. 20-27 is provided for convenience:
[0088] 200 power amplifier
[0089] 201 positive adaptive rail
[0090] 202 negative adaptive rail
[0091] 210 power supply
[0092] 221 single stage dynamic release positive rail control
[0093] 222 single stage dynamic release negative rail control
[0094] 223 multi-stage dynamic release positive rail control
[0095] 224 multi-stage dynamic release negative rail control
[0096] 231 single stage positive charge circuit
[0097] 232 single stage negative charge circuit
[0098] 233 multi-stage positive charge circuit
[0099] 234 multi-stage negative charge circuit
[0100] 241 single stage positive dynamic release circuit
[0101] 242 single stage negative dynamic release circuit
[0102] 243 multi-stage positive dynamic release circuit
[0103] 244 multi-stage negative dynamic release circuit
[0104] 251 single stage positive sag control circuit
[0105] 252 single stage negative sag control circuit
[0106] 253 multi-stage positive sag control circuit
[0107] 254 multi-stage negative sag control circuit
[0108] 260 power amplifier output
[0109] 270 buffer amplifier
[0110] 280 buffer amplifier output
Single Stage Dynamic Release Adaptive Tracking Rails
[0111] Turning to FIGS. 20-22, an above-automotive-voltage
embodiment of the invention makes positive and negative power
supply voltages available, as shown .+-.18 VDC plus ground, from a
power supply 210 fed by, for example, a 120 VAC mains outlet.
[0112] As seen in FIG. 20, an audio input signal is received at the
audio input of a power amplifier 200 which produces an audio output
signal 260 at the audio amplifier output. The power amplifier 200
has positive and negative adaptive rails 201 and 202 which are
associated with positive and negative rail controls 221 and 222,
respectively.
[0113] As seen in FIG. 21, in the positive rail control 221, a
storage capacitor C1 is connected to the positive adaptive rail
201. A charge circuit 231 is connected between the storage
capacitor C1 and the amplifier output 260. A dynamic release
circuit 241 is also connected between the storage capacitor C1 and
the amplifier output 260. The charge circuit 231 causes the storage
capacitor C1 to charge across both the positive and negative power
supplies when the swing of the output signal 260 of the power
amplifier 200 is below its predetermined positive and negative
thresholds. The dynamic release circuit 241 causes power to be
released from the storage capacitor C1 to dynamically vary the
voltage levels of the positive adaptive rail 201 in direct relation
to the swing of the output signal 260 of the power amplifier 200,
preferably at unity gain.
[0114] Continuing to look at FIG. 21, the dynamic release circuit
241 preferably includes a sag correction circuit 251 altering its
predetermined threshold to maintain a minimum offset between the
swing of the output signal 260 of the power amplifier 200 and the
dynamically varied voltage levels of its positive adaptive rail 201
as the storage capacitor C1 releases its charge.
[0115] Returning to FIG. 20, the positive and negative .+-.18 VDC
power supply voltages are connected to the positive rail control
221 which feeds the positive adaptive rail 201 of the power
amplifier 200. The power amplifier output signal 260 is in turn
connected to the positive rail control 221.
[0116] As best seen in FIG. 21, the positive rail control 221 of
FIG. 20 includes the charge circuit 231 and the dynamic release
circuit 241. The charge circuit 231 charges the storage capacitor
C1 across the positive and negative 18 VDC power supplies to a
total stored voltage of 36 volts, less one diode drop due to a
Schottky diode D1. The dynamic release circuit 241 enables use of
this stored energy to lift the voltage of the positive adaptive
rail 201 above 18 VDC when the power amplifier output 260 swings
within a predetermined offset of the positive adaptive rail 201.
This allows the output voltage swing of the power amplifier 200 to
reach approximately 54 VDC positive from the nominal 18 volt power
supply 210.
[0117] Continuing to look at FIG. 21, the positive rail control 221
receives the positive and negative 18 VDC power supplies and ground
and the power amplifier output signal 260 and outputs the positive
adaptive rail voltage 201 to the power amplifier 200. The power
amplifier output signal 260 is the input of the charge circuit 231.
The charge circuit 231 operates when an activating transistor Q10
is turned on, which occurs when the power amplifier output 260 is
below approximately 1 volt positive. The collector output of the
charge circuit activating transistor Q10 is fed to the bases of
gate drive transistors Q11 and Q12 through a 2 k.OMEGA. build out
resistor R13. This allows a clamp transistor Q13 to clamp the bases
of two gate drive transistors Q11 and Q12 at -18V when a stored
energy release
[0118] MOSFET transistor Q2 is active without being effected by the
collector current of the charge circuit activating transistor Q10.
This ensures that the capacitor storage and release power MOSFETs
Q1 and Q2, respectively, cannot both be active at the same time,
even during an initial power up condition. A resistor R11 provides
a slight negative bias to hold the charge circuit activating
transistor Q10 switched on unless the output swing of the power
amplifier 200 is above 1 volt. The emitter- follower outputs of the
gate drive circuit transistors Q11 and Q12 are connected to a gate
resistor R15 which connects to the gate of the capacitor storage
MOSFET Q1. The capacitor storage MOFSET Q1 is a common Source
configuration and operates like a switch when the gate drive
voltage is approximately 3 volts above the Source voltage, which is
tied to the negative 18 VDC power supply.
[0119] Still looking at FIG. 21, for the operation of the stored
energy dynamic release circuit 241, the power amplifier output
signal 260 is also connected to a small signal diode D21. The small
signal diode D21 provides halve wave rectification, allowing only a
positive voltage above ground to pass due to a cathode-side
resistor R23 tied to ground. The cathode side of the small signal
diode D21 is connected to the cathode side of a Zener diode D22, as
shown a 24 volt Zener diode. The anode side of the Zener diode D22
is connected to a bias resistor R24 which is connected to the
negative 18 volts power supply. This node is also connected to the
bases of emitter follower transistors Q22 and Q23, which provide
gate drive for the capacitor release MOSFET Q2 through a gate
resistor R25. The positive adaptive rail 201 will be at +18 VDC
less the diode drop of the Schottky diode D1 connected between the
storage capacitor C1 and the positive 18 volt power supply. The
Zener diode D22 is selected to allow the power amplifier output
signal 260 to directly drive the gate of the release MOSFET Q2 to
release the stored energy of the capacitor C1 when the output swing
comes within approximately 7 volts of the +18 volt power supply.
The 24 volt Zener voltage plus the forward drop of the small signal
diode D21, the VBE of the emitter follower transistors Q22 and Q23
plus the 3 volt gate threshold of the capacitor C1 release MOSFET
Q2 provides a cumulative offset such that the positive adaptive
rail voltage 201 will begin to lift above +18 volts when the power
amplifier output is 260 swings above 11 volts.
[0120] Unlike the automotive voltage embodiment, the power MOSFET
Q1 that lifts the positive adaptive rail 201 is a Source follower
with a gain of 1, which allows the direct power amplifier output
260 to drive the gate release power MOSFET Q2 with the proper Zener
offset voltage, reducing the loop gain of the control and adding
stability. This allows operation out to 20 Khz without any side
effect, such as an overshoot of the rail control which would cause
high frequency tracking errors.
[0121] Also seen in FIG. 21 is additional circuitry 251 employing
two transistors Q20 and Q21 to provide capacitor voltage sag
correction. One correction transistor Q20 has its base connected to
the power amplifier output 260 through a resistor R20 and a Zener
diode D20, its emitter connected to the positive adaptive rail 201
and its collector connected to -18 volts though a resistor R21.
This Zener diode D20 is selected to provide the minimum desired
offset voltage before sag correction is applied. The diode D20 is a
4.7 volt Zener. When the voltage difference between the power
amplifier output 260 and the positive adaptive rail 201 is more
than 4.7 volts, the first correction transistor Q20 will be turned
on and its collector voltage will be equal to the emitter voltage
holding the gate of the second correction transistor Q1, a JFET
transistor, off through a gate connected resistor R22. As the power
amplifier output 260 delivers current to the load, the lifted rail
voltage 201 will sag due to the capacitor C1 being discharged. When
the voltage difference between power amplifier output 260 and the
positive adaptive rail 201 drops to less than 4.7 volts the first
correction transistor Q20 turns off, pulling its collector voltage
down to -18 VDC due to a bias resistor R21, which turns on the
second correction transistor Q21. When the second correction
transistor Q21 turns on it connects another Zener diode D23 across
the first Zener diode D22, reducing the zener voltage, and the
effective offset, between the power is amplifier output 260 and the
bases of the gate drive transistors Q22 and Q23. The correction
transistors Q20 and Q21 operate in a linear mode and do not
saturate so as to continuously and dynamically vary the offset
threshold to compensate for the continuing voltage drop that occurs
in the storage capacitor C1 as large current is delivered to the
load connected to the output 260 of the power amplifier 200. This
linear operation maintains a minimum difference between the
adaptive tracking rail 201 and the power amplifier output 260 based
on the selected zener voltage of the Zener diode D20 up to the
point of clipping on the output 260 of the power amplifier 200. The
storage capacitor C1 will be fully recharged back to 36 volts less
the diode drop of the Zener diode D1 when the output 260 of the
power amplifier 200 swings below 1 volt.
[0122] The positive rail control 221 disclosed in FIG. 21 provides
reduced loop gain by use of the power MOSFET Q2 configured as a
Source follower, allowing accurate high frequency operation out to
20 KHz. Other Zener voltages can be used to provide different
offset voltages between the power amplifier output 260 and the
positive adaptive rail 201 without changing the basic operation of
the invention. The Zener offset voltages chosen in the above
example are for illustrative purposes. Different power supply
voltages and maximum power amplifier output voltage swings, as well
as various offset tracking levels between the power amplifier
output swing and the adaptive tracking rail, may also be
employed.
[0123] Comparing FIG. 22 to FIG. 21, as well as Schedules I and II
hereinafter presented, the negative rail control 222 of FIG. 22 is
essentially a symmetrically inverted version of the positive rail
control 221 of FIG. 21. Therefore, in FIG. 22, the FIG. 21 supply
voltages and the power MOSFETs Q1 and Q2 are inverted.
[0124] In FIG. 21, the positive rail control element numbers 201,
221, 231, 241 and 251 of FIG. 21 are replaced by the negative rail
control element numbers 202, 222, 232, 242 and 252, respectively.
All other element numbers of FIG. 21 are used to identify
corresponding elements in FIG. 22, which are specifically defined
by reference to Schedule I below. Therefore, the above description
of the positive rail control 221 of FIG. 21 is sufficient to
understand to the negative rail control 22 of FIG. 22.
Multi-Stage Dynamic Release Adaptive Tracking Rails
[0125] Turning to FIGS. 23-25, a multi-stage dynamic release
adaptive tracking rail embodiment of the invention provides
multiple, as shown dual, positive and negative above-automobile
voltage power supply voltages, as shown .+-.15 VDC and .+-.30 VDC
plus ground, from a power supply 260 fed by, for example, an AC
source, such as a 120 VAC mains outlet.
[0126] As seen in FIG. 23, an audio input signal is received by the
power amplifier 200 which produces an audio amplifier output signal
260. The power amplifier 200 has positive and negative adaptive
rails 201 and 202 which are associated with positive and negative
rail controls 223 and 224, respectively.
[0127] Looking at FIG. 24, in the positive rail control 223, the
storage capacitor C1 is connected to the positive adaptive rail
201. A charge circuit 233 is connected between the storage
capacitor C1 and the amplifier output 260. A dynamic release
circuit 243 is also connected between the storage capacitor C1 and
the amplifier output 260. The charge circuit 233 causes the storage
capacitor C1 to charge across the .+-.30 VDC positive and negative
power supply rails when the swing of the output signal 260 of the
power amplifier 200 is below its predetermined positive and
negative thresholds. The dynamic release circuit 243 causes power
to be released from the storage capacitor C1 to dynamically vary
the voltage levels of the .+-.30 VDC positive and negative power
supply rails in direct is relation to the swing of the output
signal 260 of the power amplifier 200, preferably at unity
gain.
[0128] As seen in FIG. 24, each of the dual .+-.15 VDC and .+-.30
VDC positive and negative power supply rails has a different
voltage level and the dynamic release circuit 243 has dual stages
corresponding to the dual positive and negative power supply rails
to cascade operation of the dynamic release circuit 243 in response
to corresponding levels of the swing of the amplifier output
260.
[0129] Returning to FIG. 23, the dual .+-.15 VDC and .+-.30 VDC
positive power supply voltages plus ground are connected to the
positive rail control 243 which feeds the positive adaptive rail
201 of the power amplifier 200. The power amplifier output signal
260 is in turn connected to the positive rail control 223.
[0130] Returning to FIG. 24, the positive rail control 223 is a
high current charge pump including a charge circuit 233 and a
three-stage dynamic release circuit 243. The charge circuit 233
charges the storage capacitor C1 across the higher positive and
negative .+-.30 VDC power supplies to a total stored voltage of 60
VDC less one diode drop due to the Schottky diode D1. The
three-stage dynamic release circuit 243 enables use of this stored
energy to lift the positive adaptive rail voltage 201 above 30 VDC
when the swing of the output signal 260 of the power amplifier 200
swings within a predetermined offset of the positive adaptive rail
201. This allows the swing of the output signal 260 of the power
amplifier 200 to reach approximately 90 VDC positive from the
nominal 30 volt power supply 260. The positive rail control 223
provides a continuous adaptive rail voltage 201 when the output 260
of the amplifier 200 increases above a predetermined threshold, and
does so according the three stages of dynamic release.
[0131] Continuing to look at FIG. 24, in the operation of the
charge circuit 233, the input of the positive rail control 223
receives the dual positive and negative .+-.15 VDC and .+-.30 VDC
power and ground and the power amplifier output signal 260 and
outputs the positive adaptive rail voltage 201 to the power
amplifier 200. The power amplifier output signal 260 is applied to
the input of the charge circuit 233 which is identical to and
operates exactly the same as the charge circuit 231 of FIG. 21, but
with higher power supply voltages of .+-.30 VDC. The storage
capacitor C1 is charged through the Schottky diode D1 when the
charge MOSFET transistor Q1 is turned on. When the power amplifier
output signal 260 increases above 1 volt, the charge MOSFET Q1
turns off. When the power amplifier output signal 260 swings close
to the adaptive rail voltage 201, reaching approximately 7 volts
less than the 30 volt rail, the first stage MOSFET Q4 becomes
active, pulling the negative side of the storage capacitor C1 above
-30 volts by forward biasing the Schottky diode D3 and lifting the
positive adaptive rail 201.
[0132] Still looking at FIG. 24, in the operation of the dynamic
release circuit 243, the first stage diode D41 passes only positive
signals above ground at the power amplifier output 260. A Zener
diode D42, as shown a 39 volt Zener, provides the required voltage
offset needed. The anode side of the Zener diode D42 is biased to
-30 volts by a resistor R44 and also connected to the bases of
emitter follower transistors Q42 and Q43 which provide a gate drive
circuit with their emitters connected to a gate resistor R45 which
is connected to the gate of the first stage power MOSFET Q4. The
collector of one follower transister Q43 is connected to the -30
volt power supply and the collector of the other follower
transistor Q42 is connected to ground, limiting the gate drive
voltage between -30 volts and ground. The first stage MOSFET Q4 is
a Source follower, the same as the release MOSFET Q2 in FIG. 21. As
the output signal 260 of the power amplifier 200 swings above 22
volts, first stage release MOSFET Q4 becomes active and lifts the
positive adaptive rail 201 up to approximately 55 volts. The
operation of the first stage release MOSFET Q4 is graphically
illustrated in FIG. 26 in which the offset of the positive adaptive
rail 201 is shown as being approximately 7 volts.
[0133] Returning to FIG. 24, when the available lift from first
stage release MOSFET Q4 is reached, the difference voltage between
the positive adaptive rail 201 and power amplifier output 260 will
decrease to the point where the second stage of the dynamic release
control circuit 243, including the second stage diode D21, Zener
diode D22, gate drive circuit transistors Q22 and Q23 and the
second stage release MOSFET Q3, become operational and continue to
lift the rail 201. Similar to the first stage, the anode of the
second stage diode D21 is connected to the power amplifier output
260 and the cathode is connected to bias resistor R23 and the anode
of the 43 volt Zener diode D22, which is a higher voltage Zener
than the first stage Zener D42 which allows the second stage rail
lift to operate only when the first stage rail lift reaches its
maximum. The second stage gate drive transistors Q22 and Q23, with
bases connected to a buildout resistor R25, operate the same as
described for the first stage gate drive transistors Q42 and Q43,
except their collectors are connected to the cathode of the second
stage diode D21 and to -30 volts, providing a different operating
gate drive range. Second stage gate resistor R35 has one end
connected to the emitters of the second stage gate drive
transistors Q22 and Q23 and its other end connected to the gate of
the second stage release MOSFET Q3, which is also a Source follower
configuration and provides the advantages previously described for
increased stability and extended high frequency performance. The
drain of the second stage release MOSFET Q3 is connected to +15
volts. The anode of the Schottky diode D2 is connected to the
Source and the cathode to the negative side of the storage
capacitor C1. As described above, when the positive adaptive rail
201 is lifted to the maximum limit from the first stage release
MOSFET Q4, the difference voltage drops to the point where the
second stage Zener diode D22 becomes biased, driving the gate of
the second stage release MOSFET Q3 through the second stage gate
drive transistors Q22 and Q23. Therefore, the second stage release
MOSFET Q3 lifts the positive adaptive rail 201 over the range of 55
volts to approximately 74 volts and will limit at that level
because the second stage release MOSFET's drain is connected to +15
volts. Another Zener diode D30 limits the gate drive voltage to
ensure the maximum gate voltage is not exceeded, which would cause
damage to the second stage release MOSFET Q3.
[0134] The third stage of the dynamic release circuit 243 consists
of a third stage Zener diode D25 which has its cathode connected to
the emitters of the second stage gate transistors Q22 and Q23 and
its anode connected to the third stage gate drive resistor R26
which connects to the gate of the third stage release MOSFET Q2
which is also a Source follower configuration with its drain
connected to the +30 volt power supply. The Source is connected
directly to the negative side of the storage capacitor C1. When the
operating limit of the second stage release MOSFET Q3 is reached,
the difference voltage between power amplifier output 260 and the
positive adaptive rail 201 will decrease again to the point where
the third stage Zener diode D25 will become biased. When the third
stage release MOSFET Q2 becomes active, the voltage level of the
positive adaptive rail 201 continues to increase, as is shown in
FIG. 26. The third stage release MOSFET Q2 provides operation
between approximately 74 volts up to a maximum limit based on the
power supply voltage and the voltage sag of the storage capacitor
C1.
[0135] In sum, the first power MOSFET Q4 lifts the rail 201 by 30
volts but also delivers the lower end of the power output and,
therefore, the lowest portion of current delivered to the load. The
second power MOSFET Q3 and the third power MOSFET Q2 work over a
reduced 15 volt range but at higher power levels and higher current
load.
[0136] Also seen in FIG. 24 is additional circuitry 243 employing
two transistors Q20 and Q21 to provide capacitor voltage sag
correction as previously described with reference to FIG. 21,
except that the first correction transistor Q20 has its collector
connected to -30 volts though the resistor R21.
[0137] While it would be possible to lift the voltage of the
positive adaptive rail 201 with multiple power MOSFETs connected in
parallel, the heat generated due to the increased voltage drop
across the power MOSFETs would be considerably higher than the
three-stage method disclosed in FIGS. 23-25. By lifting the
adaptive rail 201 in three steps, the voltage drop across the power
MOSFETs Q4, Q3 and Q2 is reduced to far less than the 60 volts
between the positive 30 volt and negative 30 volt power supplies,
thus reducing the heat dissipation of the power MOSFETs that lifts
the positive adaptive rail 201 and reducing the required "safe
operating area" (SOA) of the power MOSFETs Q4, Q3 and Q2.
[0138] Comparing FIG. 25 to FIG. 24, as well as Schedules I and II
hereinafter presented, the negative rail control 224 of FIG. 25 is
essentially a symmetrically inverted version of the positive rail
control 223 of FIG. 24. Therefore, in FIG. 25, the FIG. 24 supply
voltages and the power MOSFETs Q1, Q2, Q3 and Q4 are inverted.
[0139] In FIG. 25, the positive rail control element numbers 201,
223, 233, 243 and 253 of FIG. 24 are replaced by the negative rail
control element numbers 202, 224, 234, 244 and 254, respectively.
All other element numbers of FIG. 24 are used to identify
corresponding elements in FIG. 25, which are specifically defined
by reference to Schedule II below. Therefore, the above description
of the positive rail control 223 of FIG. 24 is sufficient to
understand to the negative rail control 224 of FIG. 25.
[0140] While FIGS. 23-25 illustrate a dual power supply, three
stage dynamic release system, using the same principles any number
n of different above-automobile voltage power supply voltages can
be used with any number n+1 of dynamic release stages.
Buffer Amplifier Adaptive Tracking Rails
[0141] Turning to FIG. 27, for both the single stage and
multi-stage dynamic release embodiments of FIGS. 20 and 23, instead
of using the actual output swing 260 of the power amplifier 200,
the positive rail control, high current charge pump circuits 221
and 222 or 223 and 224 are controlled by use of a unity gain low
current buffer amplifier 270. The buffer amplifier 270 has a common
node audio signal input with the power amplifier 200, an audio
signal output 280 and common node positive and negative adaptive
rails 201 and 202 with the power amplifier 200. The high current
charge pump circuits 221 and 222 or 223 and 224, including the
charge circuits 231 and 232 or 233 and 234 and the dynamic release
circuits 241 and 242 or 243 and 244 are connected between the
storage capacitor C1 and the buffer amplifier output 280 instead of
between the storage capacitor C1 and the power amplifier output
260. The unity gain buffer amplifier 270 produces the exact same
output voltage swing 280 as the voltage swing 260 of the power
amplifier 200 and receives the same audio input signal received by
the power amplifier 200.
[0142] The buffer amplifier 270 drives the input of the positive
and negative rail controls 221 and 222 or 223 and 224 which require
very low current drive. Therefore, the output 260 of the buffer
amplifier 200 is designed to deliver only very low current,
simplifying the design of the buffer amplifier 200. Since the
buffer amplifier 200 does not deliver a large output current, it
affords a more stable drive for the inputs of the positive and
negative rail controls 221 and 222 or 223 and 224 when the output
signal 260 of the power amplifier 200 is driving large current to
the output load.
Elements of the Above-Automotive-Voltage Audio Amplifiers
[0143] The following Schedules I and II provide examples of
co-operable combinations of elements of single stage and
multi-stage dynamic release embodiments, respectively, of
above-automotive-voltage audio amplifiers, which includes any
combinations of buffer amplifiers with single stage or multi-stage
amplifiers:
Schedule I
TABLE-US-00001 [0144] Element Single-Stage Positive Rail
Single-Stage Negative Rail Number FIG. 21 Component FIG. 22
Component Q1, Q2 IRFB3207 N-CHANNEL IXTP76P10T P-CHANNEL MOSFET
MOSFET Q11, Q13, BJT_PNP, 2N5551 BJT_NPN, 2N5401 Q22 Q10, Q12,
BJT_PNP, 2N5401 BJT_NPN, 2N5551 Q20, Q23 Q21 JFET_P-CHANNEL J175
JFET_N-CHANNEL J111 D1 SCHOTTKY DIODE SB580 SCHOTTKY DIODE SB580
D20 ZENER, 1N5230B ZENER, 1N5230B D21 DIODE, 1N4148 DIODE, 1N4148
D22 ZENER, 1N5252B ZENER, 1N5252B D23 ZENER, 1N5233B ZENER, 1N5233B
C1 5600UF ELECTROLYTIC 5600UF ELECTROLYTIC CAPACITOR CAPACITOR R10,
R23, RESISTOR, 5.1 k.OMEGA. RESISTOR, 5.1 k.OMEGA. R24 R12, R13
RESISTOR, 2 k.OMEGA. RESISTOR, 2 k.OMEGA. R11 RESISTOR, 47 k.OMEGA.
RESISTOR, 47 k.OMEGA. R14, R20 RESISTOR, 10 k.OMEGA. RESISTOR, 10
k.OMEGA. R21 RESISTOR, 20 k.OMEGA. RESISTOR, 20 k.OMEGA. R22
RESISTOR, 510 k.OMEGA. RESISTOR, 510 k.OMEGA.
Schedule II
TABLE-US-00002 [0145] Element Number Multi-Stage Positive Rail
Multi-Stage Negative Rail Q1, Q2, Q3, Q4 IRFB3207 N-CHANNEL
IXTP76P10T P-CHANNEL MOSFET MOSFET Q11, Q13, Q22, Q42 BJT_NPN,
2N5551 BJT_PNP, 2N5401 Q10, Q12, Q20, Q23, Q43 BJT_PNP, 2N5401
BJT_NPN, 2N5551 Q21 JFET_P, J175 JFET_N, J111 D1, D2, D3 SCHOTTKY
DIODE SB580 SCHOTTKY DIODE SB580 D20 ZENER, 1N5233B ZENER, 1N5233B
D21, D41 DIODE, 1N4148 DIODE, 1N4148 D22 ZENER, 1N5260B ZENER,
1N5260B D23 ZENER, 1N5236B ZENER, 1N5236B D25 ZENER, 1N5221B ZENER,
1N5221B D30 ZENER, 1N5262B ZENER, 1N5262B D42 ZENER, 1N5259B ZENER,
1N5259B C1 CAP_ELECTROLIT, 10000 .mu.F CAP_ELECTROLIT, 10000 .mu.F
R10, R23, R43 RESISTOR, 5.1 k.OMEGA. RESISTOR, 5.1 k.OMEGA. R15,
R26, R35, R45 RESISTOR, 300 .OMEGA. RESISTOR, 300 .OMEGA. R12, R14,
R20, R24, R27, R44 RESISTOR, 10 k.OMEGA. RESISTOR, 10 k.OMEGA. R11
RESISTOR, 47 k.OMEGA. RESISTOR, 47 k.OMEGA. R13, R25 RESISTOR, 2
k.OMEGA. RESISTOR, 2 k.OMEGA. R21 RESISTOR, 20 k.OMEGA. RESISTOR,
20 k.OMEGA. R22 RESISTOR, 510 k.OMEGA. RESISTOR, 510 k.OMEGA.
[0146] Thus, it is apparent that there has been provided, in
accordance with the invention, an adaptive tracking rail audio
amplifier that fully satisfies the objects, aims and advantages set
forth above. While the invention has been described in conjunction
with specific embodiments thereof, it is evident that many
alternatives, modifications and variations will be apparent to
those skilled in the art in light of the foregoing description.
Accordingly, it is intended to embrace all such alternatives,
modifications and variations as fall to the spirit of the appended
claims.
* * * * *