U.S. patent application number 15/609973 was filed with the patent office on 2017-09-14 for resource block based multicarrier modulations for agile spectrum.
This patent application is currently assigned to IDAC Holdings, Inc.. The applicant listed for this patent is IDAC Holdings, Inc.. Invention is credited to Erdem Bala, Juan Fang, Leonid L. Kazakevich, Kenneth Kearney, Jialing Li, I-Tai Lu, Rui Yang, Zihao You.
Application Number | 20170264476 15/609973 |
Document ID | / |
Family ID | 49887209 |
Filed Date | 2017-09-14 |
United States Patent
Application |
20170264476 |
Kind Code |
A1 |
Yang; Rui ; et al. |
September 14, 2017 |
RESOURCE BLOCK BASED MULTICARRIER MODULATIONS FOR AGILE
SPECTRUM
Abstract
A resource block (RB)-based multicarrier modulation (MCM)
transmitter and receiver structure for spectral agile systems are
disclosed. The transmitter and the receiver are capable of sharing
opportunistically available and non-contiguous channels with other
users. The RB-MCM partitions the available spectrum, contiguous or
non-contiguous, into multiple RBs (same or different sizes),
applies a baseband MCM or single carrier modulation, or coded
single carrier or multicarrier schemes in each RB with a type of
spectral leakage reduction technique, and applies RB modulation for
each RB to modulate the signal from baseband to the frequency band
of that RB. At the receiver, the received signal may be filtered
and RB demodulation may be applied to put each RB signal in
baseband and a baseband multicarrier or single carrier or coded
single carrier or coded multicarrier demodulation may be applied to
each RB signal. Different RBs may use different modulation
schemes.
Inventors: |
Yang; Rui; (Greenlawn,
NY) ; Kazakevich; Leonid L.; (Plainview, NY) ;
Li; Jialing; (San Diego, CA) ; Bala; Erdem;
(East Meadow, NY) ; Kearney; Kenneth; (Smithtown,
NY) ; Lu; I-Tai; (Dix Hills, NY) ; Fang;
Juan; (Brooklyn, NY) ; You; Zihao; (Brooklyn,
NY) |
|
Applicant: |
Name |
City |
State |
Country |
Type |
IDAC Holdings, Inc. |
Wilmington |
DE |
US |
|
|
Assignee: |
IDAC Holdings, Inc.
Wilmington
DE
|
Family ID: |
49887209 |
Appl. No.: |
15/609973 |
Filed: |
May 31, 2017 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
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14648160 |
May 28, 2015 |
9692629 |
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PCT/US2013/072410 |
Nov 27, 2013 |
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15609973 |
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61731356 |
Nov 29, 2012 |
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61767556 |
Feb 21, 2013 |
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61774452 |
Mar 7, 2013 |
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Current U.S.
Class: |
1/1 |
Current CPC
Class: |
H04L 27/0008 20130101;
H04W 72/04 20130101; H04L 27/2627 20130101; H04L 27/2631 20130101;
H04L 5/0066 20130101; H04L 27/264 20130101 |
International
Class: |
H04L 27/26 20060101
H04L027/26; H04L 27/00 20060101 H04L027/00; H04L 5/00 20060101
H04L005/00; H04W 72/04 20090101 H04W072/04 |
Claims
1. A method for performing multicarrier modulation in a wireless
communication device, the method comprising: partitioning
non-contiguous available spectrum into a plurality of resource
blocks (RBs), wherein each RB of the plurality of RBs comprises a
plurality of subcarriers; applying a modulation scheme on a block
of input symbols on each of the plurality of RBs separately,
wherein the modulation scheme is different for at least one of the
plurality of RBs; filtering and windowing each modulated block of
input symbols on each of the plurality of RBs separately to form a
plurality of per-RB baseband filtered signals; performing RB
modulation on each of the plurality of per-RB baseband filtered
signals separately to modulate each per-RB baseband filtered signal
into a frequency band of each RB to form per-RB multicarrier
modulated signals; summing the per-RB multicarrier modulated
signals for transmission; and transmitting the summed per-RB
multicarrier modulated signals.
2. The method of claim 1, further comprising: adding a cyclic
prefix (CP) or symbol extension to each modulated block of input
symbols on each of the plurality of RBs separately; and performing
upsampling on each modulated block of input symbols on each of the
plurality of RBs separately.
3. The method of claim 1, wherein the modulation scheme is one of
at least orthogonal frequency division multiplexing (OFDM),
filtered orthogonal frequency division multiplexing (F-OFDM),
single carrier modulation (SCM), or precoded orthogonal frequency
division multiplexing (P-OFDM).
4. The method of claim 1, wherein each RB of the plurality of RBs
is comprised of a different number of subcarriers.
5. The method of claim 1, further comprising: scaling a power of
each of the per-RB multicarrier modulated signals separately.
6. The method of claim 1, further comprising: applying a phase
rotation technique to each per-RB multicarrier modulated signal,
wherein at least one of the phase rotation techniques is
different.
7. The method of claim 6, wherein one of the phase rotation
techniques is selective level mapping (SLM) or partial transmit
sequences (PTS).
8. A wireless communication device for performing multicarrier
modulation, the wireless communication device comprising: a
processor operatively coupled to a transmitter, the processor and
transmitter configured to: partition non-contiguous available
spectrum into a plurality of resource blocks (RBs), wherein each RB
of the plurality of RBs comprises a plurality of subcarriers; apply
a modulation scheme on a block of input symbols on each of the
plurality of RBs separately, wherein the modulation scheme is
different for at least one of the plurality of RBs; filter and
window each modulated block of input symbols on each of the
plurality of RBs separately to form a plurality of per-RB baseband
filtered signals; perform RB modulation on each of the plurality of
per-RB baseband filtered signals separately to modulate each per-RB
baseband filtered signal into a frequency band of each RB to form
per-RB multicarrier modulated signals sum the per-RB multicarrier
modulated signals for transmission; and transmit the summed per-RB
multicarrier modulated signals.
9. The wireless communication device of claim 8, wherein the
processor and transmitter are further configured to: add a cyclic
prefix (CP) or symbol extension to each modulated block of input
symbols on each of the plurality of RBs separately; and perform
upsampling on each modulated block of input symbols on each of the
plurality of RBs separately.
10. The wireless communication device of claim 8, wherein the
modulation scheme is one of at least orthogonal frequency division
multiplexing (OFDM), filtered orthogonal frequency division
multiplexing (F-OFDM), single carrier modulation (SCM), or precoded
orthogonal frequency division multiplexing (P-OFDM).
11. The wireless communication device of claim 8, wherein each RB
of the plurality of RBs is comprised of a different number of
subcarriers.
12. The wireless communication device of claim 8, wherein the
processor and transmitter are further configured to: scale a power
of each of the per-RB multicarrier modulated signals
separately.
13. The wireless communication device of claim 8, wherein the
processor and transmitter are further configured to: apply a phase
rotation technique to each per-RB multicarrier modulated signal,
wherein at least one of the phase rotation techniques is
different.
14. The wireless communication device of claim 13, wherein at least
one of the phase rotation techniques is selective level mapping
(SLM) or partial transmit sequences (PTS).
Description
CROSS REFERENCE TO RELATED APPLICATIONS
[0001] This application is a continuation of U.S. application Ser.
No. 14/648,160, filed May 28, 2015 and claims the benefit of PCT
Application No. PCT/US2013/072410, filed Nov. 27, 2013, U.S.
Provisional Application No. 61/731,356, filed on Nov. 29, 2012,
U.S. Provisional Application No. 61/767,556, filed on Feb. 21,
2013, and U.S. Provisional Application No. 61/774,452, filed on
Mar. 7, 2013, the contents of which are incorporated herein by
reference in their entirety.
BACKGROUND
[0002] Multi-carrier modulation (MCM) techniques enable
transmissions of a set of data over multiple narrow band
sub-carriers. With an advanced wideband modulation and coding
scheme, an MCM system may achieve higher spectral efficiency in
frequency selective channels compared to systems using single
carrier modulation techniques.
[0003] Orthogonal frequency division multiplexing (OFDM) is an
example of a MCM technique used in a wireless communication system.
OFDM divides the total bandwidth into several orthogonal sub-bands
overlapping in frequency and may be efficiently implemented by fast
Fourier transform (FFT). In practical wireless communication
systems, cyclic prefix orthogonal frequency division multiplexing
(CP-OFDM), (i.e., OFDM with CP), is used to combat multipath
channel. With a long enough cyclic prefix (CP), a simple one-tap
frequency domain equalizer (FDE) is sufficient to mitigate the
multipath channel impairment. However, large sidelobes of the
rectangular pulse in OFDM create challenging issues in practice,
including large inter-carrier interference when the system is not
perfectly synchronized and excessive out-of-band emission (OOBE)
exists, which limits the effectiveness of OFDM in networks with
dynamic channel access cap abilities.
SUMMARY
[0004] A resource block based multicarrier modulation (RB-MCM)
transmitter and receiver structure for spectral agile systems are
disclosed. The transmitter and the receiver are capable of sharing
opportunistically available and non-contiguous channels with other
users. The RB-MCM partitions the available spectrum, contiguous or
non-contiguous, into multiple resource blocks (RBs) (same or
different sizes), applies a baseband MCM or single carrier
modulation, or coded single carrier or multicarrier schemes in each
RB with a type of spectral leakage reduction technique, and then
applies frequency shift (RB modulation) for each RB to modulate the
signal from baseband to the frequency band of that RB. At the
receiver, the received signal may be filtered and a frequency shift
(RB demodulation) may be applied to put each RB signal in baseband
and a baseband multicarrier or single carrier or coded single
carrier or coded multicarrier demodulation may be applied to each
RB signal. Different RBs may use different modulation schemes.
[0005] RB-based filtered-OFDM (RB-F-OFDM), RB-based single carrier
modulation (SCM), and, RB-based precoded OFDM (RB-P-OFDM) are
examples under the RB-MCM transmitter and receiver structure.
[0006] The RB-MCM may provide criteria such as low out-of-band
emission (OOBE), low in-band distortion, low complexity, low
latency, low peak-to-average power ratio (PAPR), robustness to
frequency and timing asynchronous, and robustness to PA
nonlinearity.
BRIEF DESCRIPTION OF THE DRAWINGS
[0007] A more detailed understanding may be had from the following
description, given by way of example in conjunction with the
accompanying drawings wherein:
[0008] FIG. 1A is a system diagram of an example communications
system in which one or more disclosed embodiments may be
implemented;
[0009] FIG. 1B is a system diagram of an example wireless
transmit/receive unit (WTRU) that may be used within the
communications system illustrated in FIG. 1A;
[0010] FIG. 1C is a system diagram of an example radio access
network and an example core network that may be used within the
communications system illustrated in FIG. 1A;
[0011] FIG. 2A shows an example of a non-contiguous spectrum
divided into resource blocks of same size;
[0012] FIG. 2B shows an example of a non-contiguous spectrum
divided into resource blocks of different sizes;
[0013] FIG. 3 is a flowchart diagram illustrating procedures for
one embodiment of a RB-based waveform;
[0014] FIG. 4 is a block diagram of an example RB-based MCM
transmitter;
[0015] FIG. 5 is a block diagram of an example RB-based MCM
transmitter that reuses one RB-MCM module;
[0016] FIG. 6 is a block diagram of an example RB-based MCM
receiver in accordance with one embodiment;
[0017] FIG. 7 is a block diagram of an example RB-based MCM
transmitter using selective level mapping (SLM) or partial transmit
sequence (PTS);
[0018] FIG. 8A shows an example of multicarrier modulation (MCM)
within each RB;
[0019] FIG. 8B shows an example of single carrier modulation within
each RB;
[0020] FIG. 8C shows an example of single carrier modulation with
CDMA within each RB;
[0021] FIG. 8D shows an example of coded multicarrier modulation
within each RB;
[0022] FIG. 9 is a representation of an RB based filtered-OFDM
transmitter (RB-F-OFDM Tx);
[0023] FIG. 10 is a representation of an F-OFDM transmit module
(F-OFDM Tx) being used for each RB;
[0024] FIG. 11 shows the PSD of per-RB filtered modulated signals
forming an RB-F-OFDM transmit signal;
[0025] FIG. 12 is an example structure of an RB-F-OFDM transmitter
with the per-RB filtered-OFDM transmit module (F-OFDM Tx);
[0026] FIG. 13 shows power spectrum density (PSD) of per-RB
filtered-OFDM signal components;
[0027] FIG. 14 is an example RB-F-OFDM system with a cyclic prefix
(CP)-OFDM/filtered-OFDM receiver;
[0028] FIG. 15 is a representation of an example RB-F-OFDM
receiver;
[0029] FIG. 16 is a representation of an example F-OFDM receive
module;
[0030] FIG. 17 is a block diagram of an example RB-F-OFDM
receiver;
[0031] FIG. 18 is a block diagram of an example RB-F-OFDM system
that uses a RB-F-OFDM transmitter and a RB-F-OFDM receiver;
[0032] FIG. 19 is a block diagram of an example type-I per-RB
F-OFDM transmit module;
[0033] FIG. 20 is an illustration of the PSD of the upsampled
signal, the per-RB transmit filter, and the per-RB filtered signal
in type-I per-RB F-OFDM Transmitter;
[0034] FIG. 21 is an illustration of the PSD of the passband of the
transmit filter in RB-F-OFDM;
[0035] FIG. 22 is a block diagram of an example type-I
per-RB-F-OFDM receive module;
[0036] FIG. 23 is a block diagram of an example type-II per-RB
F-OFDM transmit module;
[0037] FIG. 24 is a block diagram of an example modified structure
of the type-II per-RB OFDM transmit module of FIG. 23 using the
circular shift via permutation;
[0038] FIG. 25 is a block diagram of an example type-II per-RB
F-OFDM receive module (F-OFDM Rx-II);
[0039] FIG. 26 is a block diagram of an example modified structure
of the type-II per-RB F-OFDM receive module of FIG. 25 using the
circular shift via permutation;
[0040] FIG. 27 is a block diagram of a RB-F-OFDM transmitter using
PAPR reduction techniques;
[0041] FIG. 28 is a block diagram of a RB-F-OFDM receiver using
PAPR reduction techniques;
[0042] FIG. 29 shows an example RB-F-OFDM transmitter using
polyphase implementation of a synthesis filter bank (SFB);
[0043] FIG. 30 is an example block diagram of the OFDM transmit
module used in the RB-F-OFDM transmitter of FIG. 29;
[0044] FIG. 31 is an example RB-F-OFDM receiver using polyphase
implementation of an analysis filter bank (AFB);
[0045] FIG. 32 is an example block diagram of the OFDM receive
module used in the RB-F-OFDM receiver of FIG. 31;
[0046] FIG. 33 is an example block diagram of a time-varying filter
at the AFB;
[0047] FIG. 34 is an example resource block single carrier
modulation (RB-SCM) transmit module;
[0048] FIG. 35 is an example resource block single carrier
modulation (RB SCM) receive module;
[0049] FIG. 36 is a block diagram of an example structure of the
individual precoded OFDM (I-P-OFDM) transceiver in accordance with
one embodiment;
[0050] FIG. 37 is a block diagram of a per-RB precoding transmit
processing module;
[0051] FIG. 38 is a block diagram of a per-RB precoding receive
processing module;
[0052] FIG. 39 is a block diagram of an example uniform resource
block-based precoded OFDM (U-RB-P-OFDM) structure;
[0053] FIG. 40 is a block diagram of an example RB-based MCM
receiver using selective level mapping (SLM) or partial transmit
sequence (PTS) corresponding to the transmitter of FIG. 7;
[0054] FIG. 41 is a block diagram of an example type-III per-RB
F-OFDM transmit module; and
[0055] FIG. 42 is a block diagram of an example type-III per-RB
F-OFDM transmit module.
DETAILED DESCRIPTION
[0056] FIG. 1A is a diagram of an example communications system 100
in which one or more disclosed embodiments may be implemented. The
communications system 100 may be a multiple access system that
provides content, such as voice, data, video, messaging, broadcast,
etc., to multiple wireless users. The communications system 100 may
enable multiple wireless users to access such content through the
sharing of system resources, including wireless bandwidth. For
example, the communications systems 100 may employ one or more
channel access methods, such as code division multiple access
(CDMA), time division multiple access (TDMA), frequency division
multiple access (FDMA), orthogonal FDMA (OFDMA), single-carrier
FDMA (SC-FDMA), and the like.
[0057] As shown in FIG. 1A, the communications system 100 may
include wireless transmit/receive units (WTRUs) 102a, 102b, 102c,
102d, a radio access network (RAN) 104, a core network 106, a
public switched telephone network (PSTN) 108, the Internet 110, and
other networks 112, though it will be appreciated that the
disclosed embodiments contemplate any number of WTRUs, base
stations, networks, and/or network elements. Each of the WTRUs
102a, 102b, 102c, 102d may be any type of device configured to
operate and/or communicate in a wireless environment. By way of
example, the WTRUs 102a, 102b, 102c, 102d may be configured to
transmit and/or receive wireless signals and may include user
equipment (UE), a mobile station, a fixed or mobile subscriber
unit, a pager, a cellular telephone, a personal digital assistant
(PDA), a smartphone, a laptop, a netbook, a personal computer, a
wireless sensor, consumer electronics, and the like.
[0058] The communications systems 100 may also include a base
station 114a and a base station 114b. Each of the base stations
114a, 114b may be any type of device configured to wirelessly
interface with at least one of the WTRUs 102a, 102b, 102c, 102d to
facilitate access to one or more communication networks, such as
the core network 106, the Internet 110, and/or the other networks
112. By way of example, the base stations 114a, 114b may be a base
transceiver station (BTS), a Node-B, an eNode B, a Home Node B, a
Home eNode B, a site controller, an access point (AP), a wireless
router, and the like. While the base stations 114a, 114b are each
depicted as a single element, it will be appreciated that the base
stations 114a, 114b may include any number of interconnected base
stations and/or network elements.
[0059] The base station 114a may be part of the RAN 104, which may
also include other base stations and/or network elements (not
shown), such as a base station controller (BSC), a radio network
controller (RNC), relay nodes, etc. The base station 114a and/or
the base station 114b may be configured to transmit and/or receive
wireless signals within a particular geographic region, which may
be referred to as a cell (not shown). The cell may further be
divided into cell sectors. For example, the cell associated with
the base station 114a may be divided into three sectors. Thus, in
one embodiment, the base station 114a may include three
transceivers, i.e., one for each sector of the cell. In another
embodiment, the base station 114a may employ multiple-input
multiple-output (MIMO) technology and, therefore, may utilize
multiple transceivers for each sector of the cell.
[0060] The base stations 114a, 114b may communicate with one or
more of the WTRUs 102a, 102b, 102c, 102d over an air interface 116,
which may be any suitable wireless communication link (e.g., radio
frequency (RF), microwave, infrared (IR), ultraviolet (UV), visible
light, etc.). The air interface 116 may be established using any
suitable radio access technology (RAT).
[0061] More specifically, as noted above, the communications system
100 may be a multiple access system and may employ one or more
channel access schemes, such as CDMA, TDMA, FDMA, OFDMA, SC-FDMA,
and the like. For example, the base station 114a in the RAN 104 and
the WTRUs 102a, 102b, 102c may implement a radio technology such as
Universal Mobile Telecommunications System (UMTS) Terrestrial Radio
Access (UTRA), which may establish the air interface 116 using
wideband CDMA (WCDMA). WCDMA may include communication protocols
such as High-Speed Packet Access (HSPA) and/or Evolved HSPA
(HSPA+). HSPA may include High-Speed Downlink Packet Access (HSDPA)
and/or High-Speed Uplink Packet Access (HSUPA).
[0062] In another embodiment, the base station 114a and the WTRUs
102a, 102b, 102c may implement a radio technology such as Evolved
UMTS Terrestrial Radio Access (E-UTRA), which may establish the air
interface 116 using Long Term Evolution (LTE) and/or LTE-Advanced
(LTE-A).
[0063] In other embodiments, the base station 114a and the WTRUs
102a, 102b, 102c may implement radio technologies such as IEEE
802.16 (i.e., Worldwide Interoperability for Microwave Access
(WiMAX)), CDMA2000, CDMA2000 1.times., CDMA2000 EV-DO, Interim
Standard 2000 (IS-2000), Interim Standard 95 (IS-95), Interim
Standard 856 (IS-856), Global System for Mobile communications
(GSM), Enhanced Data rates for GSM Evolution (EDGE), GSM EDGE
(GERAN), and the like.
[0064] The base station 114b in FIG. 1A may be a wireless router,
Home Node B, Home eNode B, or access point, for example, and may
utilize any suitable RAT for facilitating wireless connectivity in
a localized area, such as a place of business, a home, a vehicle, a
campus, and the like. In one embodiment, the base station 114b and
the WTRUs 102c, 102d may implement a radio technology such as IEEE
802.11 to establish a wireless local area network (WLAN). In
another embodiment, the base station 114b and the WTRUs 102c, 102d
may implement a radio technology such as IEEE 802.15 to establish a
wireless personal area network (WPAN). In yet another embodiment,
the base station 114b and the WTRUs 102c, 102d may utilize a
cellular-based RAT (e.g., WCDMA, CDMA2000, GSM, LTE, LTE-A, etc.)
to establish a picocell or femtocell. As shown in FIG. 1A, the base
station 114b may have a direct connection to the Internet 110.
Thus, the base station 114b may not be required to access the
Internet 110 via the core network 106.
[0065] The RAN 104 may be in communication with the core network
106, which may be any type of network configured to provide voice,
data, applications, and/or voice over internet protocol (VoIP)
services to one or more of the WTRUs 102a, 102b, 102c, 102d. For
example, the core network 106 may provide call control, billing
services, mobile location-based services, pre-paid calling,
Internet connectivity, video distribution, etc., and/or perform
high-level security functions, such as user authentication.
Although not shown in FIG. 1A, it will be appreciated that the RAN
104 and/or the core network 106 may be in direct or indirect
communication with other RANs that employ the same RAT as the RAN
104 or a different RAT. For example, in addition to being connected
to the RAN 104, which may be utilizing an E-UTRA radio technology,
the core network 106 may also be in communication with another RAN
(not shown) employing a GSM radio technology.
[0066] The core network 106 may also serve as a gateway for the
WTRUs 102a, 102b, 102c, 102d to access the PSTN 108, the Internet
110, and/or other networks 112. The PSTN 108 may include
circuit-switched telephone networks that provide plain old
telephone service (POTS). The Internet 110 may include a global
system of interconnected computer networks and devices that use
common communication protocols, such as the transmission control
protocol (TCP), user datagram protocol (UDP) and the internet
protocol (IP) in the TCP/IP internet protocol suite. The networks
112 may include wired or wireless communications networks owned
and/or operated by other service providers. For example, the
networks 112 may include another core network connected to one or
more RANs, which may employ the same RAT as the RAN 104 or a
different RAT.
[0067] Some or all of the WTRUs 102a, 102b, 102c, 102d in the
communications system 100 may include multi-mode capabilities,
i.e., the WTRUs 102a, 102b, 102c, 102d may include multiple
transceivers for communicating with different wireless networks
over different wireless links. For example, the WTRU 102c shown in
FIG. 1A may be configured to communicate with the base station
114a, which may employ a cellular-based radio technology, and with
the base station 114b, which may employ an IEEE 802 radio
technology.
[0068] FIG. 1B is a system diagram of an example WTRU 102. As shown
in FIG. 1B, the WTRU 102 may include a processor 118, a transceiver
120, a transmit/receive element 122, a speaker/microphone 124, a
keypad 126, a display/touchpad 128, non-removable memory 130,
removable memory 132, a power source 134, a global positioning
system (GPS) chipset 136, and other peripherals 138. It will be
appreciated that the WTRU 102 may include any sub-combination of
the foregoing elements while remaining consistent with an
embodiment.
[0069] The processor 118 may be a general purpose processor, a
special purpose processor, a conventional processor, a digital
signal processor (DSP), a plurality of microprocessors, one or more
microprocessors in association with a DSP core, a controller, a
microcontroller, Application Specific Integrated Circuits (ASICs),
Field Programmable Gate Array (FPGAs) circuits, any other type of
integrated circuit (IC), a state machine, and the like. The
processor 118 may perform signal coding, data processing, power
control, input/output processing, and/or any other functionality
that enables the WTRU 102 to operate in a wireless environment. The
processor 118 may be coupled to the transceiver 120, which may be
coupled to the transmit/receive element 122. While FIG. 1B depicts
the processor 118 and the transceiver 120 as separate components,
it will be appreciated that the processor 118 and the transceiver
120 may be integrated together in an electronic package or
chip.
[0070] The transmit/receive element 122 may be configured to
transmit signals to, or receive signals from, a base station (e.g.,
the base station 114a) over the air interface 116. For example, in
one embodiment, the transmit/receive element 122 may be an antenna
configured to transmit and/or receive RF signals. In another
embodiment, the transmit/receive element 122 may be an
emitter/detector configured to transmit and/or receive IR, UV, or
visible light signals, for example. In yet another embodiment, the
transmit/receive element 122 may be configured to transmit and
receive both RF and light signals. It will be appreciated that the
transmit/receive element 122 may be configured to transmit and/or
receive any combination of wireless signals.
[0071] In addition, although the transmit/receive element 122 is
depicted in FIG. 1B as a single element, the WTRU 102 may include
any number of transmit/receive elements 122. More specifically, the
WTRU 102 may employ MIMO technology. Thus, in one embodiment, the
WTRU 102 may include two or more transmit/receive elements 122
(e.g., multiple antennas) for transmitting and receiving wireless
signals over the air interface 116.
[0072] The transceiver 120 may be configured to modulate the
signals that are to be transmitted by the transmit/receive element
122 and to demodulate the signals that are received by the
transmit/receive element 122. As noted above, the WTRU 102 may have
multi-mode capabilities. Thus, the transceiver 120 may include
multiple transceivers for enabling the WTRU 102 to communicate via
multiple RATs, such as UTRA and IEEE 802.11, for example.
[0073] The processor 118 of the WTRU 102 may be coupled to, and may
receive user input data from, the speaker/microphone 124, the
keypad 126, and/or the display/touchpad 128 (e.g., a liquid crystal
display (LCD) display unit or organic light-emitting diode (OLED)
display unit). The processor 118 may also output user data to the
speaker/microphone 124, the keypad 126, and/or the display/touchpad
128. In addition, the processor 118 may access information from,
and store data in, any type of suitable memory, such as the
non-removable memory 130 and/or the removable memory 132. The
non-removable memory 130 may include random-access memory (RAM),
read-only memory (ROM), a hard disk, or any other type of memory
storage device. The removable memory 132 may include a subscriber
identity module (SIM) card, a memory stick, a secure digital (SD)
memory card, and the like. In other embodiments, the processor 118
may access information from, and store data in, memory that is not
physically located on the WTRU 102, such as on a server or a home
computer (not shown).
[0074] The processor 118 may receive power from the power source
134, and may be configured to distribute and/or control the power
to the other components in the WTRU 102. The power source 134 may
be any suitable device for powering the WTRU 102. For example, the
power source 134 may include one or more dry cell batteries (e.g.,
nickel-cadmium (NiCd), nickel-zinc (NiZn), nickel metal hydride
(NiMH), lithium-ion (Li-ion), etc.), solar cells, fuel cells, and
the like.
[0075] The processor 118 may also be coupled to the GPS chipset
136, which may be configured to provide location information (e.g.,
longitude and latitude) regarding the current location of the WTRU
102. In addition to, or in lieu of, the information from the GPS
chipset 136, the WTRU 102 may receive location information over the
air interface 116 from a base station (e.g., base stations 114a,
114b) and/or determine its location based on the timing of the
signals being received from two or more nearby base stations. It
will be appreciated that the WTRU 102 may acquire location
information by way of any suitable location-determination method
while remaining consistent with an embodiment.
[0076] The processor 118 may further be coupled to other
peripherals 138, which may include one or more software and/or
hardware modules that provide additional features, functionality
and/or wired or wireless connectivity. For example, the peripherals
138 may include an accelerometer, an e-compass, a satellite
transceiver, a digital camera (for photographs or video), a
universal serial bus (USB) port, a vibration device, a television
transceiver, a hands free headset, a Bluetooth.RTM. module, a
frequency modulated (FM) radio unit, a digital music player, a
media player, a video game player module, an Internet browser, and
the like.
[0077] FIG. 1C is a system diagram of the RAN 104 and the core
network 106 according to an embodiment. As noted above, the RAN 104
may employ an E-UTRA radio technology to communicate with the WTRUs
102a, 102b, 102c over the air interface 116. The RAN 104 may also
be in communication with the core network 106.
[0078] The RAN 104 may include eNode-Bs 140a, 140b, 140c, though it
will be appreciated that the RAN 104 may include any number of
eNode-Bs while remaining consistent with an embodiment. The
eNode-Bs 140a, 140b, 140c may each include one or more transceivers
for communicating with the WTRUs 102a, 102b, 102c over the air
interface 116. In one embodiment, the eNode-Bs 140a, 140b, 140c may
implement MIMO technology. Thus, the eNode-B 140a, for example, may
use multiple antennas to transmit wireless signals to, and receive
wireless signals from, the WTRU 102a.
[0079] Each of the eNode-Bs 140a, 140b, 140c may be associated with
a particular cell (not shown) and may be configured to handle radio
resource management decisions, handover decisions, scheduling of
users in the uplink and/or downlink, and the like. As shown in FIG.
1C, the eNode-Bs 140a, 140b, 140c may communicate with one another
over an X2 interface.
[0080] The core network 106 shown in FIG. 1C may include a mobility
management entity gateway (MME) 142, a serving gateway 144, and a
packet data network (PDN) gateway 146. While each of the foregoing
elements are depicted as part of the core network 106, it will be
appreciated that any one of these elements may be owned and/or
operated by an entity other than the core network operator.
[0081] The MME 142 may be connected to each of the eNode-Bs 140a,
140b, 140c in the RAN 104 via an S1 interface and may serve as a
control node. For example, the MME 142 may be responsible for
authenticating users of the WTRUs 102a, 102b, 102c, bearer
activation/deactivation, selecting a particular serving gateway
during an initial attach of the WTRUs 102a, 102b, 102c, and the
like. The MME 142 may also provide a control plane function for
switching between the RAN 104 and other RANs (not shown) that
employ other radio technologies, such as GSM or WCDMA.
[0082] The serving gateway 144 may be connected to each of the
eNode Bs 140a, 140b, 140c in the RAN 104 via the S1 interface. The
serving gateway 144 may generally route and forward user data
packets to/from the WTRUs 102a, 102b, 102c. The serving gateway 144
may also perform other functions, such as anchoring user planes
during inter-eNode B handovers, triggering paging when downlink
data is available for the WTRUs 102a, 102b, 102c, managing and
storing contexts of the WTRUs 102a, 102b, 102c, and the like.
[0083] The serving gateway 144 may also be connected to the PDN
gateway 146, which may provide the WTRUs 102a, 102b, 102c with
access to packet-switched networks, such as the Internet 110, to
facilitate communications between the WTRUs 102a, 102b, 102c and
IP-enabled devices.
[0084] The core network 106 may facilitate communications with
other networks. For example, the core network 106 may provide the
WTRUs 102a, 102b, 102c with access to circuit-switched networks,
such as the PSTN 108, to facilitate communications between the
WTRUs 102a, 102b, 102c and traditional land-line communications
devices. For example, the core network 106 may include, or may
communicate with, an IP gateway (e.g., an IP multimedia subsystem
(IMS) server) that serves as an interface between the core network
106 and the PSTN 108. In addition, the core network 106 may provide
the WTRUs 102a, 102b, 102c with access to the networks 112, which
may include other wired or wireless networks that are owned and/or
operated by other service providers.
[0085] Filtered-OFDM, (F-OFDM) is the use of a digital baseband
filter after the OFDM modulator to combat the OOBE of a CP-OFDM
based system. A matched filter may be added at the receiver side.
In general, F-OFDM may reduce the OOBE of an OFDM signal for
wireless communication systems having fixed spectrum bandwidth. It
may be used for such systems, regardless of the fact that the
sample rate transmit filtering incurs an increase in computational
complexity. However, when it comes to opportunistically available
and non-contiguous channels in a band which is shared by multiple
users in both frequency domains and time domains, a typical
situation for cognitive radio (CR) systems, filtered-OFDM may not
be an effective solution. In such systems, a pre-defined transmit
filter may not adequately reduce the energy leakage of certain
unavailable frequencies (spectrum "holes"), when such frequencies
are in the transmit filter's passband. Therefore, F-OFDM may not be
suitable for wireless communication systems with agile spectrum,
(i.e., when the transmit spectrum is non-contiguous with possible
dynamically changed frequency bands of various sizes).
[0086] In addition to F-OFDM, several other sidelobe suppression
techniques for non-contiguous OFDM (NC-OFDM) (i.e., OFDM with
non-contiguous active subcarrier allocations, aiming for spectral
agile systems) may be used. These sidelobe suppression techniques
include guard subcarriers (GS), windowing, adaptive symbol
transition (AST), constellation expansion (CE), cancellation
carriers (CC), subcarriers weighting (SW), multiple-choice
sequences (MCS), polynomial cancellation coding (PCC), spectral
precoding (SP), extended active interference cancellation (EAIC),
partial response signaling (PRS), N-continuous OFDM (NC), and power
allocation schemes. These techniques may reduce OOBE for spectral
agile systems. However, these techniques may have deficiencies,
such as increased peak-to-average power ratio (PAPR), high
computational complexity, decrease of throughput due to in-band
distortion, etc. Furthermore, these techniques may not be suitable
for high order modulations.
[0087] Filter bank multicarrier (FBMC) modulation techniques may be
used to overcome the drawback of large OOBE in both contiguous and
non-contiguous spectrum. Filter bank multicarrier is a family of
MCM techniques in which a prototype filter, applying to each
subcarrier (or tone), is designed to achieve a certain goal, such
as minimizing inter-symbol interference (ISI), inter-carrier
interference (ICI) and/or stop-band energy. OFDM may be considered
a type of FBMC whose time domain prototype filter is a simple
rectangular pulse. Other FBMC modulation techniques use transmit
and receive prototype filters whose time duration is, in general,
greater than the symbol duration. Therefore, in FBMC, the
successive data symbols overlap in the time domain. By choosing a
prototype filter other than the rectangular pulse used in OFDM, the
peak of the sidelobes may be reduced. As a result, FBMC may provide
better spectral shaping of the sub-bands. In FBMCs, orthogonality
in time and frequency is maintained. One of the FBMCs that have
been considered is filtered multi-tone modulation (FMT). FMT uses a
guard band to maintain orthogonality between subcarriers.
Therefore, FMT reduces OOBE at the cost of spectral efficiency
loss.
[0088] Another FBMC that has been considered is OFDM-offset
quadrature amplitude modulation (OFDM-OQAM). In OFDM-OQAM,
subcarriers of the signal overlap each other to achieve a high
spectral efficiency. Unlike OFDM, the real and imaginary parts of
the quadrature amplitude modulation (QAM) symbols are processed
separately with 2.times. symbol rate. A simple one-tap FDE may be
sufficient for OFDM-OQAM when the multipath channel has short delay
spread, but OFDM-OQAM incurs large throughput degradation when the
channel has long delay spread. A more sophisticated multi-tap
equalizer may be needed for channels with long delay spread,
resulting in a large increase in complexity. Despite this,
OFDM-OQAM may be used for cognitive radio and spectral agile
systems for channels with relatively short delay spread, such as in
an indoor environment. However, OFDM-OQAM has a much higher latency
than other OFDM-based MCMs, which restricts its usage in some
practical systems, such as WiFi.
[0089] Orthogonality in time and frequency may lower the design
freedom of the waveform. Non-orthogonal MCMs have been considered.
Examples of the non-orthogonal MCMs are the non-orthogonal
frequency division multiplexing (NOFDM), and non-contiguous NOFDM
(NC-NOFDM), its version for spectral agile systems. In NOFDM,
overlapping of signal in time and frequency makes the
time-frequency grid denser. By properly designing the
transmitter-receiver pulse pairs, low OOBE may be achieved in
NOFDM. Moreover, due to overlapping of signal in frequency domain,
spectrum may be utilized more efficiently. However, the pulse
shaping in NOFDM significantly increases the PAPR. Additionally,
the filtering and ISI/ICI cancellation in NOFDM results in a high
complexity increase when compared to OFDM.
[0090] Another example of a non-orthogonal MCM is generalized
frequency division multiplexing (GFDM). GFDM is parallel
single-carrier CP systems that are realized in digital domain. CP
is used to enable one-tap FDE. Each subcarrier has a pair of
transmit and receive filters. Between subcarriers, there is ICI.
The ICI cancellation results in high complexity. Therefore, GFDM
may experience in-band performance loss due to loss of
orthogonality between subcarriers.
[0091] The aforementioned MCMs and sidelobe suppression techniques
for better spectral containment share a common drawback of OFDM,
low energy efficiency. These MCMs' large PAPR of the transmitted
signals requires power amplifiers (PA) to have a large linear
range. Otherwise, the nonlinearity may lead to signal distortion,
which may cause larger out-of-band radiation and larger bit error
rate (BER).
[0092] Several PAPR reduction methods have been considered in order
to solve this problem. These PAPR reduction methods may be
generally categorized into two groups, signal scrambling techniques
and signal distortion techniques. Signal scrambling techniques
include all variations of techniques for performing scrambling to
decrease the PAPR. These signal scrambling techniques may include
coding, interleaving, selective level mapping (SLM), partial
transmit sequences (PTS), tone reservation, tone injection, etc.
Side information may be needed for signal scrambling techniques, by
which redundancy is introduced and the effective throughput is
reduced. Signal distortion techniques may reduce high peaks by
directly distorting the signal. These signal distortion techniques
include clipping and filtering, companding, etc. However, OOBE may
increase due to signal distortion, if the MCM signal is sensitive
to PA nonlinearity.
[0093] There is a need for an advanced waveform for spectral agile
systems that is capable of sharing opportunistically available and
non-contiguous spectrum resources with the other users. The
characteristics of such a waveform should include low OOBE, low
in-band distortion, low complexity, low latency, low PAPR,
robustness to frequency and timing asynchronous, robustness to PA
nonlinearity, etc. For different practical systems, such as LTE and
WiFi (802.11x), the advanced waveform may be tuned (by changing
design parameters) to make different tradeoffs between the
aforementioned criteria to satisfy the system requirements.
[0094] Embodiments disclosed herein are applicable to multicarrier
modulation based systems, such as LTE, WiMAX, 802.11x, 802.15x,
etc. and cognitive radio systems.
[0095] In filtered bank multicarrier (FBMC) modulation, the
out-of-subcarrier emission is reduced by filtering at a
per-subcarrier basis. In a multiple access system based on a
multicarrier modulation schemes, each frequency resource may be
partitioned into several groups, or resource blocks (RB), of
several subcarriers (e.g., 12 subcarriers as one resource block in
LTE). The amount of resources allocated to the users is based on
the number of those RBs. The smallest granularity of the resource
is a group of subcarriers, rather than a single subcarrier. The
concept of RBs may be generalized as a group of subcarriers, where
the RB size (i.e., the number of subcarriers) may vary for
different RBs. As shown in FIG. 2A, a non-contiguous spectrum may
be divided into RBs 202a, 202b, 202c, . . . , 202n, 202n+1, of the
same size. Alternatively, as shown in FIG. 2B, a non-contiguous
spectrum may be divided into RBs 204, 206 of different sizes. It is
noted that one skilled in the art would understand that any number
of RBs may be used, and that the RBs depicted in FIGS. 2A and 2B
are for the reader's convenience. In other words, fewer or more RBs
than depicted in FIGS. 2A and 2B may be used. Because the frequency
resource is being partitioned into multiple RBs in a multiple
access system based on a multicarrier modulation schemes, a
waveform may be designed to reduce the out-of-RB emission so as to
minimize the OOBE.
[0096] In one embodiment of such a RB-based waveform, as
represented by the flowchart diagram in FIG. 3, the available
spectrum (contiguous or non-contiguous) is partitioned into
multiple RBs (block 301), a baseband MCM scheme is applied in each
RB with a type of spectral leakage reduction technique (block 302),
and frequency shift (RB modulation) is applied for each RB to
modulate the signal from baseband to the frequency band of that RB
(block 303). The signal power for each RB may be different to
combat the fading of the channel or to control the interference to
the adjacent channel.
[0097] FIG. 4 is a block diagram of an example RB-based MCM
transmitter 400 arrangement in accordance with one embodiment.
Referring to FIG. 4, input symbols 405 are partitioned at a
serial-to-parallel (S/P) converter 410 into N resource blocks 415a,
415b, . . . , 415k. The input symbols 405 may be QAM modulated. A
baseband multicarrier modulation (MCM) scheme is applied to each
resource block 415a, 415b, . . . , 415k with a type of spectral
leakage reduction technique at a RB-MCM unit 420a, 420b, . . . ,
420k creating per-RB signals. Frequency shift (RB modulation) is
then applied for each per-RB signal at an RB modulation unit 425a,
425b, . . . , 425k to modulate the per-RB signals from baseband to
the frequency band of each respective RB, outputting per-RB
modulated signals. The signal power for each RB may be different to
combat the fading of the channel or to control the interference to
the adjacent channel. As a result, the signal power of each RB may
be multiplied by a power factor .lamda..sub.k at mixers 430a, 430b,
. . . , 430k to scale the power of each per-RB signal. f.sub.RB,k
represents the center frequency of k.sup.th RB, and .lamda..sub.k
is the square root of the power for k.sup.th RB. Note that the
multiplication by a power factor .lamda..sub.k at mixers 430a,
430b, . . . , 430k may take place prior to the application of the
baseband multicarrier modulation scheme at RB-MCM units 420a, 420b,
. . . , 420k; may take place after the application of the baseband
multicarrier modulation scheme at RB-MCM units 420a, 420b, . . . ,
420k, but before RB modulation at the RB modulation units 425a,
425b, 425k; or may take place after the RB modulation at the RB
modulation unit 425a, 425b, 425k as shown in FIG. 4. The per-RB
modulated signals may then be summed at a summing unit 435 to form
a transmit signal which may be amplified at power amplifier (PA)
440 for transmission, via antenna 445.
[0098] In a practical implementation of the RB-based MCM
transmitter of FIG. 4, a single RB-MCM module 550 may be used to
generate per-RB modulated signals for all RBs at a high rate. FIG.
5 is a block diagram of a RB-based MCM transmitter 500 that reuses
a single RB-MCM module 550. The RB-MCM module 550 performs the MCM
and RB modulation operations to generate per-RB modulated signals
for all RBs at a high rate. Referring to FIG. 5, input symbols 505,
which may be QAM modulated, are partitioned into N RBs 515a . . . ,
515k at a serial-to-parallel (S/P) converter 510 and are selected
by a multiplexer (MUX) 517 for input into the RB-MCM module 550.
The RB-MCM module comprises a RB-MCM unit 520, that applies a
baseband MCM scheme to each selected resource block 515a, . . . ,
515k with a type of spectral leakage reduction technique; a RB
modulation unit 525 that modulates the per-RB signal from baseband
to the frequency band of each respective RB; and a mixer 530 that
multiplies the signal power of each RB by a power factor
.lamda..sub.k to scale the power of each per-RB signal. f.sub.RB,k
represents the center frequency of k.sup.th RB, and .lamda..sub.k
is the square root of the power for k.sup.th RB. The multiplication
by a power factor .lamda..sub.k at mixer 530 may take place prior
to the application of the baseband multicarrier modulation scheme
at RB-MCM unit 520; may take place after the application of the
baseband multicarrier modulation scheme at RB-MCM unit 520, but
before RB modulation at the RB modulation unit 525; or may take
place after the RB modulation at the RB modulation unit 525, as
shown in FIG. 5. A demultiplexer (DMUX) 532 outputs each per-RB
modulated signal. The process is repeated for each of the N RBs
515a, . . . , 515k. Each per-RB modulated signal output by the DMUX
532 is then summed together at a summing unit 535 to form a
transmit signal which may be amplified at PA 540 for transmission,
via antenna 545.
[0099] FIG. 6 is a block diagram of an example RB-based MCM
receiver 600 corresponding to the RB-based MCM transmitter 400
depicted in FIG. 4. Referring to FIG. 6, a signal is received by an
antenna 605 connected to a wide-band radio-frequency (RF) front-end
unit 610, which outputs total received baseband signals to RB
filters 615a, 615b, . . . , 615k which demodulate each respective
RB from the frequency band of that RB to a baseband frequency. The
RB filter F.sub.RB,k, is centered at f.sub.RB,k and has the
bandwidth equal to N.sub.RB,k subcarriers for k=1, . . . , K. A
multicarrier demodulation process may be applied to the filtered
per-RB signals at de-MCM units 620a, 620b, . . . , 620k. The
channels that all RBs go through are then estimated using the
demodulated per-RB signals and equalized at a channel estimation
(CHEST) and equalizer unit 625a, 625b, 625k, which outputs the
estimated symbols for all K RBs 630a, 630b, . . . , 630k. Those
estimated symbols are then demodulated and decoded at a
demodulating and decoding unit 635 outputting demodulated symbols.
The demodulating unit may be a QAM demodulating unit if the signal
was modulated using QAM at the transmitter side. It should be
appreciated that a high rate operation receiver arrangement may
also be used. In such a high rate receiver, a single module may
contain a single RB filter, a single de-MCM unit, and a single
CHEST and equalizer unit. A demultiplexer (DMUX) may be added to
output per-RB estimated symbols.
[0100] Low in-band distortion, low out-of-RB spectral leakage, low
latency, and low complexity are some of the design goals for the
RB-MCM units 420a, 420b, . . . , 420k, 520, 720a, 720b, . . . ,
720k. To achieve low in-band distortion, the ISI due to multipath
fading may be mitigated by using CP or symbol extension. The ICI
for subcarriers within one RB may be minimized by frequency domain
modulation (e.g., OFDM or other MCMs that have orthogonality
between subcarriers) and advanced equalization techniques. If
non-orthogonal MCM is used for each RB, the ICI for subcarriers
within one RB may also be minimized by de-correlation or ICI
cancellation, as in NOFDM and GFDM. To achieve low out-of-RB
spectral leakage, a spectral leakage reduction technique may be
applied to the per-RB signal, such that the ICI for subcarriers in
different RBs is minimized, and that the OOBE for the entire
transmit signal is minimized. For example, such spectral leakage
reduction techniques may include: time domain filtering (which is
similar to Filtered-OFDM), time domain windowing, spectral
precoding, or specific pulse shaping for OFDM.
[0101] To achieve low latency and complexity, the transmit signal
provided by the spectral leakage reduction technique may be
localized in both time and frequency. For example, if time domain
filtering is used, the filter may be short compared to the symbol
duration. Moreover, overall complexity may be further minimized by
generating a low rate per-RB signal and then up-converting it to
high rate, since the bandwidth of each RB is smaller compared to
the occupied bandwidth of the transmit signal.
[0102] For the RB-MCM with spectral leakage reduction, time domain
filtering may be used. Per-RB complex data streams may go through
an inverse fast Fourier transform (IFFT) operation, followed by CP
insertion and time domain filtering to generate the per-RB
signal.
[0103] Alternatively, time domain windowing may be used. Per-RB
complex data streams may go through an IFFT operation, followed by
time domain windowing and CP insertion to generate the per-RB
signal. Even though windowing is a time domain operation that does
not depend on a RB based structure, the RB based structure allows
some design flexibility to achieve lower PAPR.
[0104] Alternatively, spectral precoding may be used. Per-RB
complex data streams may go through spectral precoding before an
IFFT operation and CP insertion to generate the per-RB signal.
[0105] Alternatively, pulse shaping may be used. Symbol extension
may be applied to per-RB complex data streams before a specific
pulse shaping, followed by subcarrier modulation. Even though the
pulse shaping technique is applied on a per-subcarrier basis and
does not depend on a RB based structure, the RB based structure
allows some design flexibility to achieve lower PAPR.
[0106] Furthermore, since the transmit signal is the sum of the
per-RB signals, the RB based structure provides additional degrees
of freedom to reduce PAPR. Due to relatively smaller RB size,
coding techniques may be used for PAPR reduction without signal
distortion. Coding has been shown to be an effective way to achieve
PAPR reduction, but the coding design is generally subject to high
complexity issues due to the large number of subcarriers. Some PAPR
reduction techniques, such as SLM and PTS, may be combined with the
RB based structure without incurring large additional complexity
because the number of per-RB signals is far less than the number of
subcarriers in the system.
[0107] FIG. 7 is a block diagram of one example of the RB-based MCM
transmitter using SLM or PTS 700. Referring to FIG. 7, the RB-based
MCM transmitter 400 arrangement of FIG. 4 is slightly altered to
comprise phase shift units 731a, 731b, . . . , 731k. Similar to the
RB-based MCM transmitter 400 arrangement of FIG. 4, the RB-based
MCM transmitter using SLM or PTS may comprise a serial-to-parallel
(S/P) converter 710, RB-MCM units 720a, 720b, . . . , 720k, RB
modulation units 725a, 725b, . . . , 725k, and mixers 730a, 730b, .
. . , 730k. SLM or PTS may be employed at the phase shift units
731a, 731b, . . . , 731k. Referring to FIG. 7, input symbols 705,
which may be QAM modulated, are partitioned into N RBs 715a, 715b,
. . . , 715k at a serial-to-parallel (S/P) converter 710 and are
input into the respective RB-MCM units 720a, 720b, . . . , 720k. A
baseband MCM scheme is applied to each resource block 715a, 715b, .
. . , 715k with a type of spectral leakage reduction technique
creating per-RB signals. Frequency shift (RB modulation) is then
applied for each per-RB signal at an RB modulation unit 725a, 725b,
. . . , 725k to modulate the per-RB signal from baseband to the
frequency band of each respective RB, outputting per-RB modulated
signals. The signal power for each RB may be different to combat
the fading of the channel or to control the interference to the
adjacent channel. As a result, the signal power of each RB may be
may be multiplied by a power factor .lamda.k at mixers 730a, 730b,
730k to scale the power of each per-RB modulated signal. PAPR
reduction techniques SLM or PTS may then be applied at phase shift
units 731a, 731b, . . . , 731k. f.sub.RB,k represents the center
frequency of k.sup.th RB, and .lamda..sub.k is the square root of
the power for k.sup.th RB. Note that the multiplication by a power
factor .lamda..sub.k at mixers 730a, 730b, . . . , 730k may take
place prior to the application of the baseband multicarrier
modulation scheme at RB-MCM units 720a, 720b, . . . , 720k; may
take place after the application of the baseband multicarrier
modulation scheme at RB-MCM units 720a, 720b, . . . , 720k, but
before RB modulation at the RB modulation units 725a, 725b, . . . ,
725k; or may take place after the RB modulation at the RB
modulation units 725a, 725b, . . . , 725k as shown in FIG. 7. Each
per-RB modulated signal is output by each respective phase shift
unit 731a, 731b, . . . , 731k. Each per-RB modulated signal is then
summed together at a summing unit 735 to form a transmit signal
which may be amplified at PA 740 for transmission, via antenna 745.
Besides the signal distortion-less PAPR reduction techniques,
certain signal distorted PAPR reduction techniques, such as
clipping and filtering, and companding may be used as well.
[0108] It should be appreciated that a high rate operation
transmitter arrangement using SLM or PTS may also be used. In such
a transmitter, the RB-MCM module 550 of FIG. 5 that performs the
MCM and RB modulation operations to generate per-RB modulated
signals for all RBs at a high rate may be slightly modified to
include a single phase shift unit with operations as described
above.
[0109] Referring ahead to FIG. 40, FIG. 40 is a block diagram of an
example RB-based MCM receiver using selective level mapping (SLM)
or partial transmit sequence (PTS) 4000 corresponding to the
RB-based MCM transmitter 700 depicted in FIG. 7. Referring to FIG.
40, a signal is received by an antenna 4005 connected to a
wide-band radio-frequency (RF) front-end unit 4010, which outputs
total received baseband signals to RB filters 4015a, 4015b, . . . ,
4015k which demodulate each respective RB from the frequency band
of that RB to a baseband frequency. The RB filter F.sub.RB,k, is
centered at f.sub.RB,k and has the bandwidth equal to N.sub.RB,k
subcarriers for k=1, . . . , K. The de-phase shift process is
applied at de-phase shift units 4017a, 4017b, . . . , 4017k. A
multicarrier demodulation process may be applied to the filtered
per-RB signals at de-MCM units 4020a, 4020b, . . . , 4020k. The
effective channels that each RB goes through are then estimated
using the demodulated per-RB signals and equalized at a channel
estimation (CHEST) and equalizer unit 4025a, 4025b, . . . , 4025k,
which outputs the estimated symbols for all K RBs 4030a, 4030b, . .
. , 4030k. Those estimated symbols are then demodulated and decoded
at a demodulating and decoding unit 4035 outputting demodulated
symbols. The demodulating unit may be a QAM demodulating unit if
the signal was modulated using QAM at the transmitter side. It
should be appreciated that a high rate operation receiver
arrangement may also be used. In such a high rate receiver, a
single module may contain a single RB filter, a single de-phase
shift unit, a single de-MCM unit, and a single CHEST and equalizer
unit. A demultiplexer (DMUX) may be added to output per-RB
estimated symbols.
[0110] Using MCM for each RB may be extended to single carrier
modulation, or coded single or multicarrier modulation and their
combination. Referring back to FIGS. 8A-8D, FIGS. 8A-8D show
example modulation schemes within each RB. It should be appreciated
that different RBs may use different modulation schemes. In FIGS.
8A-8D, QAM symbols are represented by each rounded rectangle. FIG.
8A shows an example of multicarrier modulation (MCM) within each
RB. FIG. 8B shows an example of single carrier modulation within
each RB. FIG. 8C shows an example of single carrier modulation with
CDMA within each RB. FIG. 8D shows an example of coded multicarrier
modulation within each RB.
[0111] Examples of RB-based MCM waveforms including the RB-based
filtered-OFDM (RB-F-OFDM), the RB-based single carrier modulation
(RC-SCM), and the RB based precoded OFDM (RB-P-OFDM) are disclosed
herein.
[0112] In one example of the RB-based filtered-OFDM (RB-F-OFDM)
waveform, the filtered-OFDM (F-OFDM) signal of each RB may be
generated independently, each with good spectral containment and
low latency.
[0113] In RB-F-OFDM, the signal of each RB is modulated and
filtered individually and then summed together to form a transmit
signal. It is assumed that each RB consists of M.sub.1 subcarriers.
The RBs are labeled 0, 1st, 2nd, . . . , kth, . . . , (K-1)th RBs,
where K is the maximum number of available RBs. The maximum number
of subcarriers M=KM.sub.1 The normalized center frequency of the
kth RB is
f k = m k L , Equation ( 1 ) ##EQU00001##
where m.sub.k is the center subcarrier of the kth RB and is not
necessarily an integer; and L is the number of samples per symbol
duration. The nth data symbol vector (an Mx1 vector) is defined
as:
S[n]=[S.sub.0[n]S.sub.1[n] . . . S.sub.K-1[n]].sup.T, Equation
(2)
where the nth data symbol vector for the kth RB is an
M.sub.1.times.1 vector defined as:
S.sub.k[n]=[S.sub.k0[n]S.sub.k1[n] . . .
S.sub.k,M.sub.1.sub.-1[n]].sup.T Equation (3)
When the kth RB is available for transmission and has data loaded,
S.sub.k[n].noteq.0. When the kth RB is not available for
transmission, S.sub.k[n]=0.
[0114] FIG. 9 is a representation of an RB based filtered-OFDM
transmitter (RB-F-OFDM Tx) 900. Referring to FIG. 9, the RB-F-OFDM
Tx 900 modulates each symbol vector s[n] 905 RB by RB,
independently, to form transmit signal x 990 which is the transmit
signal of sampling duration
T s = T L , ##EQU00002##
where T is the symbol duration and L is a power of 2.
[0115] FIG. 10 is a representation of a filtered-OFDM transmit
module (F-OFDM Tx) 1000 to be applied to each RB. Multiple per-RB
filtered-OFDM transmit modules may be comprised within the
RB-F-OFDM Tx 900, as described below and in FIG. 12. The F-OFDM Tx
1000 performs the MCM and RB modulation operations for each
respective RB. Thus, in the example RB-F-OFDM transmitter 900, each
respective symbol vector 905k represented as S.sub.k[n], k=0, 1, .
. . , K-1, is modulated and filtered by per-RB filtered-OFDM
transmit modules 1000 to form per-RB filtered modulated signals
x.sub.k 990k, that when summed together form transmit signal x 990.
Thus, the transmit signal x 990 is given as
x=.SIGMA..sub.k=0.sup.K-1.times..sub.k. Equation (4)
[0116] FIG. 11. illustrates the PSD of per-RB filtered modulated
signals forming the RB-F-OFDM transmit signal from values -120 dB
to 0 dB over a frequency range of -8 MHz to 8 MHz.
[0117] FIG. 12 is a block diagram of an example structure of an
RB-F-OFDM Tx 900. Referring to FIG. 12, the RB-F-OFDM Tx 900
comprises multiple filtered-OFDM transmit modules (F-OFDM Tx)
1000a, 1000b, . . . , 1000k, one for each RB, which output per-RB
multicarrier modulated signals X.sub.k 990a, 990b, . . . , 990k for
each RB from the respective symbol vectors 905a, 905b, . . . , 905k
of each RB. The per-RB multicarrier modulated signals 990a, 990b, .
. . , 990k form transmit signal x 990 by summing them together. The
RB-F-OFDM Tx 900 is different than the CP-OFDM or filtered-OFDM
transmitter, in that the per-RB filtered-OFDM transmit modules
1000a, 1000b, . . . , 1000k, comprised in the RB-F-OFDM transmitter
900, each only modulate the subcarriers in one RB and therefore,
the low-rate OFDM signal may be generated and then up-converted to
high rate, which will be explained in detail below.
[0118] Each per-RB multicarrier modulated signal x.sub.k 990a,
990b, . . . , 990k only has a signal overlapping its adjacent RBs
but not the RBs beyond its adjacent RBs, (i.e., x.sub.k overlaps
x.sub.k-1 and x.sub.k+1). It is assumed that a per-RB transmit
filter brings the signal leakage of a per-RB multicarrier modulated
signal to its non-adjacent RBs to be negligible. The signal overlap
between adjacent RBs may not create inter-subcarrier interference
due to orthogonality between subcarriers in different RBs. This is
shown in FIG. 13. More specifically, FIG. 13 shows power spectrum
density (PSD) of per-RB filtered-OFDM signal components from values
-80 dB to 0 dB over a frequency range of -4 MHz to -3 MHz.
[0119] FIG. 14 is a block diagram of an example RB-F-OFDM system
with a CP-OFDM/Filtered-OFDM receiver. Referring to FIG. 14, the
RB-F-OFDM transmitter 900, despite the filtering, modulates each
symbol vector 905 to the same subcarrier as it is modulated in
CP-OFDM using an L-point IFFT. Therefore, the CP-OFDM/Filtered-OFDM
receiver 1400 may be used to demodulate the transmit signal 990.
The transmit signal x 990 is sent over channel h 1401. Noise from
the environment may be added to the signal. The
CP-OFDM/Filtered-OFDM receiver 1400 receives the signal y 1402. As
shown in FIG. 14, the CP-OFDM/Filtered-OFDM receiver 1400 may
comprise a serial-to-parallel (S/P) converter 1410, a CP removal
unit 1420, a L-point FFT unit 1430 that outputs demodulated symbol
vectors 1435, a simple channel estimation (CHEST) unit 1440 as in
CP-OFDM/filtered-OFDM to estimate the equivalent channel of the
demodulated symbols including the RB-based transmit filter and
over-the-air-multipath channel, and a FDE unit 1450 as in
CP-OFDM/Filtered-OFDM to equalize the equivalent channel and yield
estimates of the original symbols s[n] 1490. The FDE unit 1450 may
be a one-tap FDE unit. Thus, the CP-OFDM/filtered-OFDM receiver
outputs estimates of the original symbols s[n] 1490.
[0120] Alternatively, a RB-F-OFDM specific receiver based on per-RB
matched filtering may be used to improve SINR. FIG. 15 is a
representation of a RB based filtered-OFDM receiver (RB-F-OFDM Rx)
1500 based on per-RB matched filtering. The RB-F-OFDM Rx 1500
demodulates the received multicarrier modulated signal y 1505 into
demodulated per-RB symbol vectors 1590.
[0121] FIG. 16 is a representation of a per-RB filtered-OFDM
receive module (F-OFDM Rx) 1600. Multiple per-RB F-OFDM receive
modules may be comprised within the RB-F-OFDM Rx 1500, as described
below and as shown in FIG. 17. The F-OFDM Rx 1600 performs the
deMCM and RB filtering operations for each received multicarrier
modulated signal y 1505 to generate per-RB demodulated symbol
vectors 1590k.
[0122] FIG. 17 is a block diagram of an example RB-F-OFDM receiver
(RB-F-OFDM Rx) 1500. Referring to FIG. 17, the RB-F-OFDM receiver
1500 comprises per-RB F-OFDM receive modules 1600a, 1600b, . . . ,
1600k which output per-RB demodulated symbol vectors 1590a, 1590b,
. . . , 1590k for each RB from the received multicarrier modulated
signal y 1505. The RB-F-OFDM receiver 1500, based on per-RB matched
filtering, is different than the CP-OFDM/Filtered-OFDM receiver
1400, because the per-RB F-OFDM receive modules 1600a, 1600b, . . .
, 1600k, comprised in the RB-F-OFDM receiver 1500, each only
demodulate the subcarriers in one RB and therefore, the signal may
be down-converted to low rate and then demodulated, which will be
explained in more detail below.
[0123] FIG. 18 is a block diagram of an example RB-F-OFDM system
that uses a RB-F-OFDM transmitter 1801 and a RB-F-OFDM receiver
1825. The RB-F-OFDM Tx 1801 may be the RB-F-OFDM 900 shown in FIG.
9. The RB-F-OFDM receiver 1825 may be the RB-F-OFDM receiver 1500
shown in FIG. 15. As shown in FIG. 18, the overall RB-F-OFDM
receiver structure 1820 comprises the RB-F-OFDM receiver 1825, a
FDE unit 1850, and a CHEST unit 1840. The FDE unit 1850 may be a
one-tap FDE unit. The RB-F-OFDM transmitter 1801 modulates each
data symbol vector 1805 RB by RB outputting a transmit signal x
1810. The transmit signal x 1810 is sent over channel h 1811. Noise
from the environment may be added to the signal. The RB-F-OFDM
receiver 1825 receives signal y 1812. The RB-F-OFDM receiver 1825
outputs demodulated symbol vectors S[n] 1830. The CHEST unit 1840
estimates the equivalent channel. The equivalent channel may
include Tx and Rx filters, for example the Tx filter 1925 as shown
in FIG. 19 and Rx filter 2215 as shown in FIG. 22 and propagation
channel h from the demodulated symbols S[n] 1830, and a FDE unit
1850 equalizes the equivalent channel to yield estimates of the
original symbol vectors 1890. The FDE unit 1850 may be derived
accordingly using zero-forcing (ZF) or minimum mean square error
(MMSE) criterion. The overall RB-F-OFDM receiver structure 1820
outputs estimates of the original symbol vectors s[n] 1890.
[0124] Embodiments for three types of RB-F-OFDM transmit and
receive modules, (Type-I, Type-II, and Type-III), are disclosed
hereafter.
[0125] FIG. 19 is a block diagram of an example Type-I per-RB
F-OFDM transmit module (F-OFDM Tx-I) 1900. The F-OFDM Tx-I 1900 may
be used as the per-RB filtered-OFDM transmit module 1000 in the
RB-F-OFDM transmitter 900 as shown in FIG. 12. Referring to FIG.
19, the symbol vectors of the kth RB 1901k are input into the
F-OFDM Tx-I 1900. The F-OFDM Tx-I 1900 comprises L.sub.1-point IFFT
unit 1905, a parallel-to-serial converter (P/S) 1910, a CP adder
unit 1915, an upsampling unit 1920, a baseband transmit filter p[n]
1925, and a RB modulation unit 1930.
[0126] The symbol vectors of the kth RB, S.sub.k[n] 1901k, are
first modulated and filtered to form a baseband filtered signal
q.sub.k 1940. Referring to FIG. 19, the nth data symbol vectors for
the kth RB S.sub.k[n] 1901k with zero-padding, first goes through
the L.sub.1-point IFFT unit 1905, where L.sub.1 is a power of 2 and
L.sub.1.ltoreq.L, to form the L.sub.1.times.1 vector as
follows:
s k [ n ] = [ s k 0 [ n ] s k 1 [ n ] s k , L 1 - 1 [ n ] ] T = 1 L
1 F L 1 H [ I M 1 0 ( L 1 - M 1 ) M 1 ] S k [ n ] , Equation ( 5 )
##EQU00003##
where an L-point discrete Fourier transform (DFT) matrix for a
general L is given as:
F L = [ 1 1 1 1 W L W L L - 1 1 W L L - 1 W L ( L - 1 ) ( L - 1 ) ]
, Equation ( 6 ) ##EQU00004##
and an L-point inverse DFT (IDFT) matrix for a general L is given
as:
F L H = [ 1 1 1 1 W L - 1 W L - ( L - 1 ) 1 W L - ( L - 1 ) W L - (
L - 1 ) ( L - 1 ) ] , Equation ( 7 ) ##EQU00005##
where
W L = e - j 2 .pi. L . Equation ( 8 ) ##EQU00006##
[0127] This vector S.sub.k[n] 1901k has sampling duration
T s 1 = T L 1 . ##EQU00007##
The elements, after parallel-to-serial (P/S) conversion at the
parallel-to-serial converter 1910, have the time indices nL.sub.1,
nL.sub.11, . . . , (n+1)L.sub.1-1. The transmit signal may have
sampling duration
T s = T L . ##EQU00008##
Therefore, the signal of sampling duration
T s 1 = T L 1 ##EQU00009##
needs to be upsampled, at the upsampling unit 1920, by an
integer
Q = L L 1 , Equation ( 9 ) ##EQU00010##
which may be a power of 2. However, before the upsampling, CP is
added at the CP adder unit 1915 and P/S conversion is performed at
the P/S converter 1910. After upsampling, the signal goes through
the baseband transmit filter p[n] 1925 which outputs the baseband
filtered signal q.sub.k 1940. The baseband filtered signal q.sub.k
1940 is modulated into the frequency band of the kth RB at the RB
modulation unit 1930 to form per-RB multicarrier modulated signal
x.sub.k 1990k for the respective RB. The sum of these signals form
the multicarrier modulated transmit signal x as shown in FIGS. 10
and 12. The baseband transmit filter p[n] 1925 may be chosen to be
a lowpass filter, wherein the passband is the bandwidth of M.sub.1
subcarriers in a single RB, (i.e.,
BW pass = M 1 T ) , ##EQU00011##
the transition band of each side has a bandwidth less than the
bandwidth of L.sub.1-M.sub.1 subcarriers, (i.e.,
BW trans .ltoreq. L 1 - M 1 T ) , ##EQU00012##
and the stopband may have attenuation of at least 55 dBm. FIG. 20
illustrates the PSD of the upsampled signal, the per-RB transmit
filter, and the per-RB filtered signal in Type-I per-RB F-OFDM Tx
from values -120 dB to 0 dB over a frequency range of -8 MHz to 8
MHz. The upsampling operation to the CP-OFDM signal may create
spectral images of the CP-OFDM signal as shown in FIG. 20. The
baseband transmit filter p[n] 1925 suppresses all created images in
its stopband, so that the inter-RB interference is negligible. The
passband of the whole RB-F-OFDM signal may go through the per-RB
transmit baseband filter 1925 that has the frequency response as in
FIG. 21. FIG. 21 is an illustration of the PSD of the passband of
the transmit filter in RB-F-OFDM. This filtering may be equalized
at the receiver.
[0128] FIG. 22 is a block diagram of an example Type-I per-RB
F-OFDM receive module (F-OFDM Rx-I) 2200. The F-OFDM Rx-I 2200 may
be used as the per-RB filtered-OFDM receive module 1600 in the
RB-F-OFDM receiver 1500 as shown in FIG. 17. The F-OFDM Rx-I 2200
shown in FIG. 22 has reverse operations of the F-OFDM Tx-I 1900
shown in FIG. 19. The F-OFDM Rx-I 2200 comprises a RB demodulation
unit 2210, a baseband receive filter p[n] 2215, a downsampling unit
2220, a serial-to-parallel converter (S/P) 2225, a CP removal unit
2230, and a L.sub.1-point FFT unit 2235. Referring to FIG. 22, for
the kth RB, the received signal y 2205 is demodulated from the
frequency band of the kth RB to baseband at the RB demodulation
unit 2210 to form a RB demodulated signal q.sub.k 2250. The RB
demodulated signal {circumflex over (q)}.sub.k 2250 is then
filtered at the baseband receive filter p[n] 2215, which could be,
but not necessary to, identical to the baseband transmit filter
p[n] 1925. The filtered signal goes through the downsampling unit
2220 with a downsampling factor Q, the S/P converter 2225, the CP
removal unit 2230, and the L.sub.1-point IFFT unit 2235. The first
M.sub.1 outputs from the L.sub.1-point FFT unit 2235 form the
demodulated symbol vectors S.sub.k[n] 2290. The demodulated symbol
vectors are obtained RB by RB in this way, similar to FIG. 17.
Similar to FIG. 18, CHEST may be obtained from the demodulated
symbol vectors at a CHEST unit and the demodulated symbol vectors
may go through a FDE unit, such as a one-tap FDE unit, to obtain
the estimate of the demodulated symbol vectors S.sub.k[n] 2290.
[0129] FIG. 23 is a block diagram of an example type-II per-RB
F-OFDM transmit module (F-OFDM Tx-II) 2300. The F-OFDM Tx-II 2300
may be used as the per-RB filtered-OFDM transmit module 1000 in the
RB-F-OFDM transmitter 900 as shown in FIG. 12. The F-OFDM Tx-II
2300 comprises a L.sub.1-point IFFT unit 2305, a RB modulation unit
2330, a parallel-to-serial converter (P/S) 2310, a CP adder unit
2315, an upsampling unit 2320, and a RB modulated transmit filter
2350.
[0130] In the F-OFDM Tx-II 2300, the symbol vectors of the kth RB,
S.sub.k[n] 2301k, may be modulated to the subcarriers' frequencies
and then filtered to form the per-RB multicarrier modulated signal
x.sub.k 2390k. The nth data symbol vector for the kth RB,
S.sub.k[n] 2301k with zero-padding, first goes through the
L.sub.1-point IFFT unit 2305, where L.sub.1 is a power of 2 and
L.sub.1.ltoreq.L, to form the L.sub.1.times.1 vector as
follows:
s k [ n ] = [ s k 0 [ n ] s k 1 [ n ] s k , L 1 - 1 [ n ] ] T = 1 L
1 F L 1 H [ I M 1 0 ( L 1 - M 1 ) M 1 ] S k [ n ] , Equation ( 10 )
##EQU00013##
where an L-point discrete Fourier transform (DFT) matrix for a
general L is given as:
F L = [ 1 1 1 1 W L W L L - 1 1 W L L - 1 W L ( L - 1 ) ( L - 1 ) ]
, Equation ( 11 ) ##EQU00014##
and an L-point inverse DFT (IDFT) matrix for a general L is given
as:
F L H = [ 1 1 1 1 W L - 1 W L - ( L - 1 ) 1 W L - ( L - 1 ) W L - (
L - 1 ) ( L - 1 ) ] , Equation ( 12 ) ##EQU00015##
where
W L = e - j 2 .pi. L . Equation ( 13 ) ##EQU00016##
[0131] This vector S.sub.k[n] has sampling duration
T s 1 = T L . ##EQU00017##
The elements after P/S conversion have the time indices nL.sub.1,
nL.sub.1+1, . . . , (n+1)L.sub.1-1. The transmit signal may have
the sampling duration
T s = T L . ##EQU00018##
Therefore, the signal of sampling duration
T s 1 = T L 1 ##EQU00019##
may be upsampled by an integer:
Q = L L 1 , Equation ( 14 ) ##EQU00020##
which may be a power of 2, before the upsampling, RB modulation,
adding CP, and P/S conversion are performed. The L.sub.1.times.1
vector S.sub.k[n] may go through the RB modulation unit 2330, to
form a RB modulated signal vector as follows:
a.sub.k[n]=[a.sub.k0[n]a.sub.k1[n] . . .
a.sub.k,L.sub.1.sub.-1[n]].sup.T=J.sub.k[n]s.sub.k[n], Equation
(15)
where the RB modulation diagonal matrix for the nth symbol is:
J k [ n ] = [ W L - m k nL 1 Q W L - m k ( nL 1 + l ) Q W L - m k (
( n + 1 ) L 1 - 1 ) Q ] = [ e j 2 .pi. m k nL 1 Q L e j 2 .pi. m k
( nL 1 + l ) Q L e j 2 .pi. m k ( ( n + 1 ) L 1 - 1 ) Q L ]
Equation ( 16 ) ##EQU00021##
And the integer Q is due to upsampling. Substituting equation (14)
into equation (16), after math manipulation, the RB modulation
diagonal matrix becomes a constant matrix for the kth RB as
follows:
J k [ n ] = [ e j 2 .pi. m k nL 1 L 1 e j 2 .pi. m k ( nL 1 + l ) L
1 e j 2 .pi. m k ( ( n + 1 ) L 1 - 1 ) L 1 ] = [ 1 e j 2 .pi. m k l
L 1 e j 2 .pi. m k ( L 1 - 1 ) L 1 ] = [ 1 W L 1 - m k l W L 1 - m
k ( L 1 - 1 ) ] = [ 1 W L 1 - d k l W L 1 - d k ( L 1 - 1 ) ] = J k
Equation ( 17 ) ##EQU00022##
where
d.sub.k=m.sub.k mod L.sub.1. Equation (18)
[0132] Then, equation (13) becomes
a.sub.k[n]=J.sub.ks.sub.k[n]. Equation (19)
[0133] Therefore, the RB modulation matrix J.sub.k performs, on
S.sub.k[n], a subcarrier shift of d.sub.k subcarriers. Substituting
equation (10) into equation (19), the following is obtained:
a k [ n ] = 1 L 1 J k F L 1 H [ I M 1 0 ( L 1 - M 1 ) .times. M 1 ]
S k [ n ] = L k [ I M 1 0 ( L 1 - M 1 ) .times. M 1 ] S k [ n ] ,
Equation ( 20 ) ##EQU00023##
where
L k = 1 L 1 J k F L 1 H = 1 L 1 [ 1 W L 1 - d k l W L 1 - d k ( L 1
- 1 ) ] [ 1 1 1 1 W L 1 - 1 W L 1 - ( L 1 - 1 ) 1 W L 1 - ( L 1 - 1
) W L 1 - ( L 1 - 1 ) ( L 1 - 1 ) ] . Equation ( 21 )
##EQU00024##
[0134] After multiplying two matrices in Equation (21), L.sub.k
becomes:
L k = 1 L 1 [ 1 1 1 W L 1 - d k W L 1 - ( d k + 1 ) W L 1 - ( d k +
L 1 - 1 ) W L 1 - d k ( L 1 - 1 ) W L 1 - ( d k + 1 ) ( L 1 - 1 ) W
L 1 - ( d k + L 1 - 1 ) ( L 1 - 1 ) ] = 1 L 1 F L 1 H [ 0 d k
.times. ( L 1 - d k ) I d k I L 1 - d k 0 ( L 1 - d k ) .times. d k
] = 1 L 1 F L 1 H P .pi. d k Equation ( 22 ) ##EQU00025##
where
P .pi. d k = [ 0 d k .times. ( L 1 - d k ) I d k I L 1 - d k 0 ( L
1 - d k ) .times. d k ] , Equation ( 23 ) ##EQU00026##
is a permutation that represents the circular shift d.sub.k
subcarriers. Substituting equation (22) into equation (20) the
following is obtained:
a k [ n ] = 1 L 1 F L 1 H P .pi. d k [ I M 1 0 ( L 1 - M 1 )
.times. M 1 ] S k [ n ] . Equation ( 24 ) ##EQU00027##
[0135] Alternatively, the F-OFDM Tx-II 2300 may be modified to use
circular shift (via permutation) as shown in FIG. 24. FIG. 24 is a
block diagram of the example type-II per-RB OFDM transmit module
using the circular shift (via permutation) 2400. The symbol vectors
of the kth RB, S.sub.k[n] 2401k are input into the type-II per RB
OFDM transmit module using the circular shift (via permutation)
2400, which comprises a permutation unit 2430 instead of the RB
modulation unit 2330 shown in FIG. 23. The type-II per-RB OFDM
transmit module using the circular shift (via permutation) 2400
also comprises a L.sub.1-point IFFT unit 2405, a parallel-to-serial
converter (P/S) 2410, a CP adder unit 2415, an upsampling unit
2420, and a RB modulated transmit filter 2450. The type-II per-RB
OFDM transmit module using the circular shift (via permutation)
2400 outputs per-RB multicarrier modulated signal x.sub.k 2490k.
The permutation unit 2430 lowers the complexity when compared to
the RB modulation unit 2330 via complex multiplications.
[0136] Referring back to FIG. 23, after the signal is modulated at
the RB modulation unit 2330, CP is added to the RB modulated signal
vector a.sub.k[n] at the CP adder unit 2315 to form an
(L.sub.1+L.sub.cp1).times.1 vector such that
b k [ n ] = [ b k 0 [ n ] b k 1 [ n ] b k , L 1 + L cp 1 - 1 [ n ]
] T = [ a k , L 1 - L cp 1 [ n ] a k , L 1 - 1 [ n ] a k 0 [ n ] a
k 1 [ n ] a k , L 1 - 1 [ n ] ] T = [ 0 L cp 1 .times. ( L 1 - L cp
1 ) I L cp 1 I L 1 ] a k [ n ] Equation ( 25 ) ##EQU00028##
[0137] After CP is added, P/S conversion is applied to b.sub.k[n]
at the P/S converter 2310 to form a 1.times.(L.sub.1+L.sub.cp1)
vector (b.sub.k[n]).sup.T. The vectors (b.sub.k[n]).sup.T for the
0th, 1st, 2nd, . . . , symbols form a sequence b.sub.k.sup.T
where
b.sub.k=[(b.sub.k[0]).sup.T(b.sub.k[1]).sup.T . . .
(b.sub.k[n]).sup.T . . . ].sup.T. Equation (26)
[0138] The sequence b.sub.k.sup.T goes through the upsampling unit
2320 by Q to form an upsampled sequence c.sub.k.sup.T where
c.sub.k=[(c.sub.k[0]).sup.T(c.sub.k[1]).sup.T . . .
(c.sub.k[n].sup.T . . . ].sup.T Equation (27)
where the 1.times.((L.sub.1+L.sub.cp1)Q) (vector) sequence for the
nth symbol in the kth RB is:
c k [ n ] = [ c k 0 [ n ] c k 1 [ n ] c k , ( L 1 + L cp 1 ) Q - 1
[ n ] ] T , and Equation ( 28 ) c k , lQ + d [ n ] = { b kl [ n ] ,
d = 0 0 , d = 1 , 2 , , Q - 1 . Equation ( 29 ) ##EQU00029##
[0139] The unsampled sequence c.sub.k.sup.T is of sampling rate
T s = T L . ##EQU00030##
Therefore, the CP duration is L.sub.cp1Q samples of sampling
rate
T s = T L . ##EQU00031##
The unsampled sequence c.sub.k then goes through the RB-modulated
transmit filter 2350 as follows:
p k [ n ] = p [ n ] W L - m k n = p [ n ] e j 2 .pi. m k n L ,
Equation ( 30 ) ##EQU00032##
where p[n] is the baseband per-RB transmit filter and it is the
sample index. For convenience, the RB-modulated transmit filter
2350 may be represented in a column vector form as p.sub.k. The RB
modulation in the RB-modulated transmit filter 2350 modulates the
signal to the same frequency band as the unfiltered signal of the
RB. The resulting signal:
x.sub.k.sup.T=p.sub.k.sup.T*c.sub.k.sup.T, Equation (31)
is the per-RB modulated signal x.sub.k 2390k for the kth RB, where
* is the convolution operation of two sequence to form one
sequence. The RB modulated transmit filter p[n] 2350 may be chosen
to be a lowpass filter, whose passband is the bandwidth of M.sub.1
subcarriers in a single RB, i.e.,
BW pass = M 1 T , ##EQU00033##
transition band each side has the bandwidth less than the bandwidth
of L.sub.1-M.sub.1 subcarriers, i.e.,
BW trans .ltoreq. L 1 - M 1 T , ##EQU00034##
and the stopband may have attenuation of at least 55 dBm. The sum
of the per-RB multicarrier modulated signals x.sub.k 2390k form the
transmit signal x 990 as shown in FIGS. 9 and 12.
[0140] FIG. 25 is a block diagram of an example type-II per-RB
F-OFDM receive module (F-OFDM Rx-II) 2500. The F-OFDM Rx-II 2500
may be used as the per-RB filtered-OFDM receive module 1600 in the
RB-F-OFDM receiver 1500 as shown in FIG. 17. The F-OFDM Rx-II 2500,
as shown in FIG. 25, has reverse operations of the F-OFDM Tx-II
2300, as shown in FIG. 23. Referring to FIG. 25, the F-OFDM Rx-II
2500 comprises a RB-modulated Rx filter 2505, a downsampling unit
2510, a serial-to-parallel converter (S/P) 2515, a CP removal unit
2520, a RB demodulation unit 2525 and a L.sub.1-point FFT unit
2530.
[0141] In the F-OFDM Rx-II 2500, for the kth RB, the received
signal y 2501 may be firstly filtered and then demodulated
according to the subcarriers' frequencies to obtain the demodulated
symbol vectors of the kth RB, {hacek over (S)}.sub.k[n] 2590k. The
demodulated symbol vectors are obtained RB by RB in this way, as in
FIG. 17.
[0142] Referring to FIG. 25, the received signal y 2501 goes
through the RB-modulated Rx filter p.sub.k[n] 2505, which may be
identical to the RB-modulated Tx filter 2350 as defined in equation
(30). Then, the filtered signal may go through the downsampling
unit 2510 of factor Q, the ST converter 2515, the CP removal unit
2520, the RB demodulation unit 2525, and the L.sub.1-point FFT unit
2530. The first M.sub.1 outputs from the L1-point FFT unit 2530
form the demodulated symbol vectors {hacek over (S)}.sub.k[n]
2590k. Alternatively, the F-OFDM Rx-II may be modified to use
circular shift (via permutation) to lower the complexity. FIG. 26
is a block diagram of an example type-II per-RB OFDM Rx module
using the circular shift (via permutation) 2600. Similar to the
Type-II per-RB F-OFDM Tx module 2400, the RB demodulation in the
Type-II per-RB F-OFDM Rx module may also be accomplished by a
permutation unit 2635 using a permutation
P.sub..pi.(L.sub.1.sub.-d.sub.k.sub.) that represents the circular
shift of L.sub.1-d.sub.k subcarriers, instead of the RB
demodulation unit 2525 as shown in FIG. 25, via complex
multiplications. The permutation matrix P.sub..pi.x. to circular
shift x subcarriers is defined in equation (23). Referring to FIG.
26, the received signal y 2601 goes through the RB-modulated Rx
filter p.sub.k[n] 2605, which is identical to the RB-modulated Tx
filter 2350 as defined in equation (30). Then, the filtered signal
may go through the downsampling unit 2610 of factor Q, the S/P
converter 2615, the CP removal unit 2620, the L.sub.1-point FFT
unit 2630, and the permutation unit 2635. The outputs of the
permutation unit 2635 form the demodulated symbol vectors {hacek
over (S)}.sub.k[n] 2690k.
[0143] Similar to FIG. 18, CHEST may be obtained from the
demodulated symbol vectors. The demodulated symbol vectors may also
go through an FDE unit, which may be a one-tap FDE unit, to obtain
estimates of the demodulated symbol vectors {hacek over
(S)}.sub.k[n] 2690k.
[0144] Type-I and type-II RB-F-OFDM transmit and receive modules
are equivalent up to a phase modulation. This difference in phase
modulation is introduced because CP is added before RB modulation
in type-I RB-F-OFDM transmit module 1900, but is added after RB
modulation in type-II RB-F-OFDM transmit module 2300. The phase
modulation for the nth data symbol in the kth RB is:
e - j 2 .pi. m k L cp n L Equation ( 32 ) ##EQU00035##
[0145] If the phase modulation is applied to the data symbols
before type-I RB-F-OFDM transmit module 1900, the transmit signal
is the same as the one generated by type-II RB-F-OFDM transmit
module 2300 using the data symbols without phase modulation. Then,
the transmit signal may be demodulated using the
CP-OFDM/filtered-OFDM receiver 1400 in FIG. 14. If the phase
modulation is not applied at the type-I RB-F-OFDM transmit module
1900, a phase modulation, which is the complex conjugate of the
phase modulation, may be applied to the output symbols at the
CP-OFDM/filtered-OFDM receiver 1400.
[0146] Referring ahead to FIG. 41, FIG. 41 is a block diagram of an
example Type-III per-RB F-OFDM transmit module (F-OFDM Tx-III)
4100. The F-OFDM Tx-III 4100 is similar to F-OFDM Tx-I, however in
the F-OFDM Tx-III, the upsampling unit 1920 and the baseband
transmit filter 1925, as shown in FIG. 19 are replaced with a low
sampling rate filter (p) 4120 and a cascaded integrator-comb (CIC)
combined with an interpolator unit 4125. The F-OFDM Tx-III 4100 may
be used as the per-RB filtered-OFDM transmit module 1000 in the
RB-F-OFDM transmitter 900 as shown in FIG. 12. The F-OFDM Tx-III
4100 comprises a L.sub.1-point IFFT unit 4105, a parallel-to-serial
converter (P/S) 4110, a CP adder unit 4115, a low sampling rate
filter p[n] 4120, a cascaded integrator-comb (CIC) combined with an
interpolator unit 4125 and a RB modulation unit 4130. Using CIC as
part of the filter may significantly reduce the implementation
complexity Referring to FIG. 41, the nth data symbol vectors for
the kth RB S.sub.k[n] 4101k first goes through the L.sub.1-point
IFFT unit 4105. CP is added at the CP adder unit 4115 and P/S
conversion is performed at the P/S converter 4110. The signal goes
through the low sampling rate filter p 4120 and the cascaded
integrator-comb (CIC) combined with an interpolator unit 4125 which
outputs the filtered signal q.sub.k 4127. The filtered signal
q.sub.k 4127 is modulated into the frequency band of the kth RB at
the RB modulation unit 4130 to form per-RB multicarrier modulated
signal x.sub.k 4190k for the respective RB.
[0147] FIG. 42 is a block diagram of an example Type-III per-RB
F-OFDM receive module (F-OFDM Rx-III) 4200. The Type III RB-F-OFDM
receiver model is similar to Type-I RB-F-OFDM receiver model 2200
as shown in FIG. 22. It replaces the baseband receive filter 2215
and down-sampling unit 2220 with a cascaded integrator-comb (CIC)
combined with a decimator 4215 and a low sampling rate filter p
4220. Using CIC as part of filter can significantly reduce the
implementation complexity. The F-OFDM Rx-III 4200 may be used as
the per-RB filtered-OFDM receive module 1600 in the RB-F-OFDM
receiver 1500 as shown in FIG. 17. The F-OFDM Rx-III 4200 has
reverse operations of the F-OFDM Tx-III 4100 shown in FIG. 41. The
F-OFDM Rx-III 4200 comprises a RB demodulation unit 4210, with a
cascaded integrator-comb (CIC) combined with a decimator 4215, a
low sampling rate filter p 4220, a serial-to-parallel converter
(S/P) 4225, a CP removal unit 4230, and a L.sub.1-point FFT unit
4235.
[0148] Referring to FIG. 42, for the kth RB, the received signal y
4205 is demodulated from the frequency band of the kth RB to
baseband at the RB demodulation unit 4210 to form a RB demodulated
signal {circumflex over (q)}.sub.k 4250. The RB demodulated signal
{circumflex over (q)}.sub.k 4250 is then filtered at the cascaded
integrator-comb (CIC) combined with a decimator 4215 and a low
sampling rate filter p 4220. The filtered signal goes through the
S/P converter 4225, the CP removal unit 4230, and the L.sub.1-point
IFFT unit 4235. The outputs from the L.sub.1-point FFT unit 4235
form the demodulated symbol vectors {hacek over (S)}.sub.k[n] 4290.
The demodulated symbol vectors are obtained RB by RB in this way,
similar to FIG. 17. Similar to FIG. 18, CHEST may be obtained from
the demodulated symbol vectors at a CHEST unit and the demodulated
symbol vectors may go through a FDE unit, such as a one-tap FDE
unit, to obtain the estimate of the demodulated symbol vectors
{hacek over (S)}.sub.k[n] 4290.
[0149] In practical systems with non-contiguous spectrum
allocation, it may be more computationally efficient to divide the
non-contiguous spectrum into non-uniform sized RBs. RB-F-OFDM may
be utilized for such non-uniform-sized RBs. One example
implementation of the per-RB F-OFDM transmit/receive module is to
use several per-RB IFFT/FFT components for different L.sub.1 values
and different pre-defined transmit/receive filters for RBs of
different sizes. Another example implementation is to use universal
per-RB IFFT/FFT components, (e.g., L.sub.1 is fixed), and have
different pre-defined transmit/receive filters for RBs of different
sizes.
[0150] Peak-to-average power ratio (PAPR) reduction techniques are
disclosed hereafter.
[0151] Since the per-RB F-OFDM transmit module deals with a smaller
number of subcarriers, and the signal of each RB is generated
independently, certain PAPR reduction techniques may be used in
conjunction with the RB-F-OFDM transmitter to reduce the PAPR to
improve energy efficiency.
[0152] In one example, coding may be used to reduce the PAPR in
each per-RB F-OFDM signal. Coding may have very high computational
complexity, due to the large number of subcarriers in MCM systems,
and due to the overlapping of symbols in some MCM systems. In
RB-F-OFDM, coding may be designed for one RB and used for all RBs.
This may highly reduce the offline design complexity.
[0153] In another example, phase rotation based techniques, such as
selective mapping (SLM) and partial transmit sequence (PTS), may be
applied to the per-RB F-OFDM signals to reduce the PAPR. Referring
back to FIG. 27, FIG. 27 is a block diagram of a RB-F-OFDM
transmitter with PAPR reduction 2700. Data symbol vectors 2705a,
2705b, . . . , 2705k are input into the RB-F-OFDM transmitter with
PAPR reduction 2700, which uses the RB-based SLM or PTS phase
vector [e.sup.j.phi..sup.0 e.sup.j.phi..sup.1 . . .
e.sup.j.phi..sup.K-1].sup.T at a selective mapping unit 2710. For
simplicity, FIG. 27 assumes that the RB-F-OFDM transmit module 1000
is used, however one skilled in the art would recognize that any
transmit module may be used. Each transmit module 1000a, 1000b, . .
. , 1000k outputs per-RB multicarrier modulated signals 2707a,
2707b, . . . , 2707k which are input into the selective mapping
unit 2710 where SLM or PTS may be applied. The RB-F-OFDM
transmitter with PAPR reduction 2700 outputs the transmit signal
2790.
[0154] FIG. 28 is a block diagram of a RB-F-OFDM receiver with PAPR
reduction 2800. The coefficients of the selective demapping unit
2810 at the RB-F-OFDM receiver are the complex conjugate of the
coefficients of the selective mapping unit 2710 at the RB-F-OFDM
transmitter to enable PAPR reduction. The RB-F-OFDM receiver with
PAPR reduction 2800 uses the RB-based SLM or PTS complex conjugate
phase vector [e.sup.j.phi..sup.0 e.sup.j.phi..sup.1 . . .
e.sup.j.phi..sup.K-1].sup.T at a selective demapping unit 2810. For
simplicity, FIG. 28 assumes that the RB-F-OFDM receive module 1600
is used, however one skilled in the art would recognize that any
receive module may be used. If the CP-OFDM/filtered-OFDM receiver
1400 is used, the configuration is identical to the SLM/PTS in
CP-OFDM. In practical implementation, the phase vector at the
receiver may be combined with an FDE unit, which may be a one-tap
FDE unit. Referring to FIG. 28, the received signal y 2805 is
received at the RB-F-OFDM receiver with PAPR reduction 2800. The
RB-based SLM or PTS complex conjugate phase vector is applied at
the selective demapping unit 2810, which outputs a per-RB signal
into the F-OFDM Rx module 1600a, 1600b, . . . 1600k, which in turn
outputs per-RB demodulated symbol vectors 2890a, 2890b, . . . ,
2890k. When the RB-based SLM/PTS is applied, the output signal
becomes
x=.SIGMA..sub.k+0.sup.K-1e.sup.j.phi..sup.kx.sub.k Equation
(33)
[0155] The RB-based phase rotation techniques may be combined with
the subcarrier-based phase rotation techniques in each RB. Fixed or
random phase rotations may also be used.
[0156] The above embodiments for PAPR reduction do not have signal
distortion that may result in spectral efficiency loss due to side
information of the parameters. Besides these techniques, PAPR
reduction techniques that have signal distortion may also be used
in conjunction with the RB-F-OFDM transmitter. For example,
clipping may be applied to the per-RB signal before the per-RB
transmit filter. Even though the PAPR is not yet evaluated at this
stage, the clipping threshold may be determined. For another
example, after the RB-F-OFDM signal is formed, clipping and
filtering, or companding may be applied to the RB-F-OFDM signal.
Such clipping and filtering may increase the transmitter
computational complexity and lead to in-band signal distortion. On
the other hand, companding may increase OOBE and degrade BER
performance due to signal distortion. RB-F-OFDM may not be very
sensitive to power amplifier (PA) nonlinearity. Therefore,
companding may be considered for RB-F-OFDM signal.
[0157] In SLM/PTS, while in general phase vectors are generated
from a given set of phases, phase vectors may be generated
randomly. The randomly generated phase vectors may have elements of
e.sup.j.phi..sup.k with arbitrary phase values or quantized phase
values from a candidate set.
[0158] In SLM, a number of phase vectors [e.sup.j.phi..sup.0
e.sup.j.phi..sup.1 . . . e.sup.j.phi..sup.K-1].sup.T may be
randomly generated. The output signal x for each phase vector and a
certain time period, (e.g., one subframe), may be obtained as in
equation (33). The maximum PAPR value in the subframe for each
phase vector may be calculated. The phase vector that results in
the lowest maximum PAPR in the time period, (e.g., one subframe),
may be chosen and the corresponding output signal may be the
transmit signal. The chosen phase vector, as side information,
needs to be made known to the receiver. Without loss of generality,
let .phi..sub.0=0 (so e.sup.j.phi..sup.0=1). The computational
complexity is added to the transmitter. If quantized phase values
are used, (e.g., the phase rotation comes from a candidate set
{1,j, -1, -j}), the additional computational complexity at the
transmitter may be reduced. At the receiver, the complex conjugate
of the chosen phase vector may be combined with an FDE unit, which
may be a one-tap FDE unit, and may not incur additional
computational complexity. Due to side information, there may be
spectral efficiency loss. The loss in data rate depends on the
number of unknown phase rotations in the chosen phase vector. If
SLM is used, there may be K-1 unknown phase rotations.
[0159] In PTS, instead of using a different phase rotation for each
entry, which is a per-RB modulated signal in the case of RB-F-OFDM,
the per-RB modulated signals may be divided into groups, and a
single phase rotation may be used for all per-RB modulated signals
in one group. The phase rotations for different groups may be
different. The K per-RB modulated signals may be divided into
groups, each having at most g RBs. It results in
K g ##EQU00036##
groups. There are
K g - 1 ##EQU00037##
unknown phase rotations, which is less than the case of SLM, when
g>1. Smaller group size may lead to larger spectral efficiency
loss, while larger group size may provide a smaller degree of
freedom in the PAPR reduction design. Without loss of generality,
SLM may be considered as a special case of PTS where the group size
g=1.
[0160] In one example, one chosen phase vector may be used for each
subframe. This may result in an additional inherent latency at the
transmitter. If one chosen phase vector is used for the duration to
yield one PAPR value, there may be no additional inherent latency
at the transmitter, because this may be accommodated due to the P/S
conversion at the transmitter. This may allow a greater degree of
freedom in PAPR reduction design. However, the spectral efficiency
loss may increase.
[0161] The side information for each subframe is the
K g - 1 ##EQU00038##
phase rotation values e.sup.j.phi..sup.1, . . . ,
e.sup.j.phi..sup.K-1 for the
K g - 1 ##EQU00039##
RB groups. such phase rotations may be seen as part of the
multipath channel and be included in RB-based pilot-based CHEST and
recovered in equalization. In this case, since the phase rotation
is different for each RB group, there should be enough pilot
symbols in each RB to enable reliable CHEST, and the CHEST for each
RB may be performed independently. On the other hand, if the pilot
density in each RB is not high enough, the CHEST of certain
subcarriers may rely on interpolation between channels of
subcarriers in the same RB as well as those in adjacent RBs. In
such a case, a pilot-based phase estimator may be needed for each
RB. In either case, there may be spectral efficiency loss due to
additional pilots. In all cases, since there may be reference
signals (RS) on each subcarrier in practical systems, SLM which
uses dedicated phase rotation for each subcarrier may not be
enabled.
[0162] In Type-I and Type-II RB-F-OFDM as described above, the
computational complexity may be high since it scales by the number
of available RBs. The RB-F-OFDM transmitter may alternatively be
represented as the per-RB CP-OFDM signals going through a synthesis
filter bank (SFB) of the filtered multitone (FMT) modulation.
Alternatively, the RB-F-OFDM receiver may be represented as the
received signals being passed through an analysis filter bank (AFB)
of the FMT demodulation followed by the per-RB CP-OFDM receive
modules.
[0163] In one embodiment, the RB-F-OFDM transmitter and receiver
may be implemented with a polyphase filter bank to further reduce
the complexity, for example, when the number of available RBs is
large. Without loss of generality, let m.sub.k=kM.sub.1, and it is
assumed that the number of subcarriers per RB M.sub.1 is a power of
2, so that Q.sub.1=L/M.sub.1 is a power of 2, and the RB modulation
and demodulation multipliers become
e .+-. j 2 .pi. kM 1 n L = e .+-. j 2 .pi. kn Q 1 .
##EQU00040##
[0164] Efficient polyphase implementations of the SFB and AFB are
described hereafter. FIG. 29 shows an example of a RB-F-OFDM
transmitter using polyphase implementation of a Synthesis Filter
Bank (SFB) 2900. The RB-F-OFDM transmitter using polyphase
implementation of a SFB 2900 may comprise OFDM transmit modules
2910a, 2910b, . . . , 2910k and a SFB 2920. Alternatively, CP-OFDM
transmit modules may be used. Referring to FIG. 29, the data symbol
vectors S[n] 2901a, 2901b, . . . , 2901k are input into the
respective OFDM transmit modules 2910a, 2910b, . . . , 2910k. A
block diagram of an example OFDM transmit module 2910 is shown in
FIG. 30. Referring to FIG. 30, the OFDM transmit module 2910 may
comprise an L.sub.1-point IFFT unit 3005, a CP adder unit 3010, and
a P/S converter 3015. Referring back to FIG. 29, the OFDM transmit
modules 2910a, 2910b, . . . , 2910k output signals b.sub.0[n],
b.sub.1[n], . . . , b.sub.K-1[n] 2915a, 2915b, . . . , 2915k which
are input into the SFB 2920. The SFB 2920 may comprise a
Q.sub.1-point IFFT unit 2930, polyphase filters 2935a, 2935b, . . .
, 2935k, and a multiplexer (MUX) 2940. The SFB 2920 outputs the
RB-F-OFDM transmit signal x[n] 2950. The RB-F-OFDM transmit signal
x[n] 2950 may be expressed as:
x [ n ] = k = 0 K - 1 x k [ n ] = k = 0 K - 1 ( m b k [ m ] p [ n -
mQ ] ) e j 2 .pi. kn Q 1 , Equation ( 34 ) ##EQU00041##
which is the expression of an FMT transmit signal, where
Q.sub.1.gtoreq.Q, Q.sub.1.gtoreq.K, and lcm(Q, Q.sub.1)=Q.sub.1.
After exchanging the order of summation, the following is
obtained:
x [ n ] = m ( k = 0 K - 1 b k [ m ] e j 2 .pi. kn Q 1 ) p [ n - mQ
] . Equation ( 35 ) ##EQU00042##
[0165] n may be expressed as:
n = n Q 1 Q 1 + v 1 , v 1 = 0 , 1 , , Q 1 - 1. Equation ( 36 )
##EQU00043##
[0166] Substituting equation (36) into (35), the following is
obtained:
x [ n ] = m ( k = 0 K - 1 b k [ m ] e j 2 .pi. kn Q 1 ) p [ n - mQ
] = m u v 1 [ m ] p [ n - mQ ] Equation ( 37 ) ##EQU00044##
where the signal
u v 1 [ m ] = k = 0 K - 1 b k [ m ] e j 2 .pi. kv 1 Q 1 , v 1 = 0 ,
1 , , Q 1 - 1 Equation ( 38 ) ##EQU00045##
may be obtained through a Q.sub.1-point IFFT. n may be expressed as
follows:
n = .rho. Q + v , .rho. = n Q , v = 0 , 1 , , Q - 1. Equation ( 39
) ##EQU00046##
[0167] The with polyphase component (with respect to Q) of filter
p[n] may be defined as:
p.sub.v[n]=p[nQ+v],v=0,1, . . . ,Q-1,n=0,1, . . . Equation (40)
and Q of such polyphase filters of p[n] may be obtained. Therefore,
substituting equation (39) into (37), the following is
obtained:
x[n]=.SIGMA..sub.mu.sub.v.sub.1[m]p[(.rho.-m)Q+v]=u.sub.v.sub.1[n]*p.sub-
.v[n], Equation (41)
which is the convolution of the inverse fast Fourier transform
(IFFT) output sequence u.sub.v.sub.1[n] and the v.sup.th polyphase
filter of p[n], and the values of v.sub.1 and v depend on n, and
are from equations (36) and (39), respectively. Since lcm(Q,
Q.sub.1)=Q.sub.1, the following is obtained:
v=v.sub.1 mod Q,v=0,1, . . . ,Q-1. Equation (42)
[0168] Therefore, each IFFT output sequence u.sub.v.sub.1[n]
corresponds to a unique polyphase filter p.sub.v[n]. The value of
the pair (v.sub.1, v) changes periodically with a period of
Q.sub.1. Thus, the multiplexer (MUX) 2940 may be used to choose the
samples from the polyphase filtered sequences based on the sample
index.
[0169] FIG. 31 shows an example RB-F-OFDM receiver using polyphase
implementation of an AFB 3100. The RB-F-OFDM receiver using
polyphase implementation of an AFB 3100 may comprise an Analysis
Filter Bank (AFB) unit 3110 that outputs vectors a.sub.0[n], . . .
, a.sub.k[n], . . . , and a.sub.K-1[n] 3112a, 3112b, . . . 3112k,
respectively, that are input into OFDM Rx modules 3120a, 3120b, . .
. , 3120k that output per-RB demodulated symbol vectors 3190a,
3190b, 3190k, respectively. Alternatively, the OFDM Rx modules may
be CP-OFDM receive modules. The AFB unit 3110 may comprise a S/P
converter 3130, periodic time-varying filters 3135a, 3135b, . . . ,
3135k, and a Q.sub.1-point FFT unit 3140. FIG. 32 is a block
diagram of the OFDM Rx module 3120. The OFDM Rx module 3120 may
comprise a S/P converter 3205, a CP removal unit 3210, and a
L.sub.1-point FFT unit 3215. The OFDM receive module outputs per-RB
demodulated symbol vectors 3190k. FIG. 33 is a block diagram of a
time-varying filter 3135 at the AFB. The time varying filter 3135
may comprise a multiplexer (MUX) unit 3305. The polyphase
implementation of the AFB may be derived similar to the SFB. Define
the Q.sub.1.times.1 output vector from the AFB in the mth instance
as:
a[m]=[a.sub.0[m]a.sub.1[m] . . . a.sub.K-1[m]a.sub.K[m] . . .
a.sub.Q.sub.1.sub.-1[m]].sup.T Equation (43)
where a.sub.k[m] is the input to the OFDM receive modules 3120a,
3120b, . . . , 3120k in the k.sup.th RB which is output from the
AFB unit 3110. The first K sequences are the inputs to the per-RB
OFDM Rx receive modules, and the last Q.sub.1-K sequences may be
discarded. It is assumed that the received signal is y[n] 3101. The
input to the OFDM receive module 3120 in the k.sup.th RB may be
expressed as:
a k [ m ] = [ ( y [ n ] e - j 2 .pi. kl Q 1 ) * p [ n ] ] | n = mQ
= l y [ l ] e - j 2 .pi. kl Q 1 p [ mQ - l ] . Equation ( 44 )
##EQU00047##
[0170] l may be expressed as:
l = .gamma. Q 1 + v 1 , .gamma. = l Q 1 , v 1 = 0 , 1 , , Q 1 - 1
Equation ( 45 ) ##EQU00048##
[0171] Substituting equation (45) into (44), the following is
obtained:
Equation ( 46 ) ##EQU00049## a k [ m ] = .gamma. v 1 = 0 Q 1 - 1 y
[ .gamma. Q 1 + v 1 ] e - j 2 .pi. kv 1 Q 1 p [ mQ - .gamma. Q 1 -
v 1 ] ##EQU00049.2##
Which is the expression of an FMT received signal at the kth
subcarrier. After exchanging the order of summation the following
is obtained:
Equation ( 47 ) ##EQU00050## a k [ m ] = v 1 = 0 Q 1 - 1 ( .gamma.
y [ .gamma. Q 1 + v 1 ] p [ m Q - .gamma. Q 1 - v 1 ] ) e - j 2
.pi. kv 1 Q 1 = v 1 = 0 Q 1 - 1 u ^ v 1 [ m ] e - j 2 .pi. kv 1 Q 1
##EQU00050.2##
where the signal
u.sub.v.sub.1[m]=.SIGMA..sub..gamma.y[.gamma.Q.sub.1+v.sub.1]p[mQ-.gamma-
.Q.sub.1-v.sub.1]=.SIGMA..sub..gamma.y.sub.v.sub.1[.gamma.]p[mQ-.gamma.Q.s-
ub.1-v.sub.1], Equation (48)
with the Q.sub.1-down-sampled received signal
y.sub.v.sub.1[.gamma.]=y[.gamma.Q.sub.1+v.sub.1],v.sub.1=0,1, . . .
,Q.sub.1-1. Equation (49)
[0172] Therefore, the output vector from the AFB may be obtained
through the Q.sub.1-point FFT 3140, where each FFT input sequence
u.sub.v.sub.1[m] comes from the convolution of
y.sub.v.sub.1[.gamma.] with a possible time varying filter. The
Q.sub.1 down-sampled received signal y.sub.v.sub.1[.gamma.] may be
obtained by using a serial-to-parallel (S/P) conversion at S/P
converter 3130 from y[n]. Let
m Q - v 1 = .rho. Q 1 + v , .rho. = m Q - v 1 Q 1 , v = ( m Q - v 1
) mod Q 1 , m .di-elect cons. Z . Equation ( 50 ) ##EQU00051##
[0173] The with polyphase component (with respect to Q.sub.1) of
filter p[n] is defined as:
p.sub.v[n]=p[nQ.sub.1+v],v=0,1, . . . ,Q.sub.1-1,n=0,1, . . . ,
Equation (51)
and Q.sub.1 of such polyphase filters of p[n] is obtained.
Therefore, substituting equations (50) and (51) into (48), after
some math manipulations, the following is obtained:
u.sub.v.sub.1[m]=.SIGMA..sub..gamma.y.sub.v.sub.1[.gamma.]p[(.rho.-.gamm-
a.)Q.sub.1+v]=y.sub.v.sub.1[.rho.]*P.sub.v[.rho.], Equation
(52)
where the values of .rho. and v depend on m, and are from equation
(50). Therefore, the polyphase filter p.sub.v[.rho.] for the fast
Fourier transform (FFT) input sequence u.sub.v.sub.1[m] is a
time-varying filter that changes periodically with a period of
q=Q.sub.1/Q. For notation simplicity, the periodic time-varying
filter for the FFT input sequence u.sub.v.sub.1[m] is defined
as:
g.sub.v.sub.1.sup.(.beta.)[n]=p.sub.(.beta.Q-v.sub.1.sub.)mod
Q.sub.1[n],.beta.=m mod q. Equation (53)
[0174] Single carrier modulation (SCM) with CP may be used as an
alternative waveform to OFDM in each RB. One advantage of SCM is
lower PAPR. In addition, since IFFT is not needed in the generation
of the SCM waveform, the complexity of the transmitter is
reduced.
[0175] In RB-based SCM (RB-SCM), the data symbols may be arranged
in a block and then CP may be added. As opposed to OFDM, an IFFT
unit may not be needed. FIG. 34 shows an example RB-SCM transmit
module 3400.
[0176] The RB-SCM transmit module 3400 has two differences from the
per-RB F-OFDM transmit module 1900 shown in FIG. 19. Zeros may not
be added to form the data block unless there is no symbol to
transmit, and the data block may not go through an IFFT unit.
[0177] Referring to FIG. 34, per-RB symbol vectors 3401k are input
into the RB-SCM transmit module 3400, which may comprise a P/S
converter 3405, a CP adder unit 3410, an upsampling unit 3415, a
baseband transmit filter 3420, and a RB modulation unit 3425.
[0178] After CP is added at the CP adder unit 3410, the data block
may go through P/S conversion at the P/S converter 3405, may be
upsampled at the upsampling unit 3415 with a ratio of Q=L/M, and
then may be filtered at the baseband transmit filter 3420. The
baseband transmit filter 3420 may be a lowpass filter. The lowpass
filter used in SCM may be sharper than the filter used in OFDM. In
RB-based OFDM, images created after upsampling are further away
from each other due to the zero-padding used in IFFT. Since IFFT
and zero-padding do not exist in the SCM case, the images may be
adjacent to each other and a sharper filter may be used to separate
them. As a result, the filter length may become longer. After
filtering, the filtered signal 3422 output by the baseband transmit
filter 3420, is RB modulated at the RB modulating unit 3425, which
outputs per-RB multicarrier modulated signal x.sub.k 3490.
[0179] FIG. 35 shows an example RB SCM receive module 3500. The
receive module for the kth RB is shown in FIG. 35. Referring to
FIG. 35, the RB SCM receive module 3500 may comprise a RB
demodulation unit 3505, a baseband receive filter 3510, a
downsampling unit 3515, a S/P converter 3520, a CP removal unit
3525, a M1-point FFT unit 3530, an equalizer 3535 and a M1-point
IFFT unit 3540. The received signal y 3501 is demodulated at the RB
demodulation unit 3505 from frequency band of the kth RB to
baseband to form a RB demodulated signal {circumflex over
(q)}.sub.k3502. Then, the RB demodulated signal {circumflex over
(q)}.sub.k3502 is filtered at the baseband receive filter 3510 and
downsampled at the downsampling unit 3515. CP is removed at the CP
removal unit 3525 and the signal is partitioned at the S/P
converter 3520. The data symbols may then be demodulated. With SCM,
frequency domain equalization may be performed. To achieve this,
FFT of the signal is taken at the M1-point FFT unit 3530. Then,
equalization is performed at the equalizer 3535 by using techniques
such as zero-forcing or MMSE equalization. Finally, IFFT is taken
to recover the data symbols at the M1-point IFFT unit 3540. The
RB-SCM receive module outputs demodulated symbols 3550a, . . . ,
3550k.
[0180] In a variation of the SCM, CP may be omitted. In this case,
at the receiver, time-domain equalization may be used, (i.e.,
equalization is performed on the signal after the S/P
operation).
[0181] In another example, RB-based precoded OFDM (RB-P-OFDM) may
be implemented. The precoding technique may be used to achieve
significant OOBE suppression with a relatively small spectral
efficiency loss. Moreover, precoding does not need signaling
overhead, has no bit error rate (BER) performance degradation, and
does not depend on the input data. However, there may be a few
drawbacks to the conventional precoding approach. Firstly, PAPR is
increased, since the symbols mapped to subcarriers are no longer
independent after precoding. Secondly, the design of the precoding
matrix depends on the assigned subcarriers and needs to be updated
each time when the assigned subcarriers are changed.
[0182] Examples for an individual precoded OFDM (I-P-OFDM) scheme
and two types of RB based precoded-OFDM (RB-P-OFDM) schemes, which
are named uniform resource block-based precoded-OFDM (U-RB-P-OFDM)
and non-uniform resource block-based precoded-OFDM (NU-RB-P-OFDM)
are disclosed hereafter.
[0183] In I-P-OFDM, each frequency band (FB) is precoded
independently. In RB-based OFDM systems, the FBs are divided into
RBs (where each RB is a group of subcarriers that form a contiguous
spectrum) and one RB is viewed as the smallest available
transmission band. In U-RB-P-OFDM, the signal of each RB is
precoded independently by a uniform precoding matrix and each has
good spectral containment. It has low complexity but the spectral
efficiency loss is relatively large, since it scales by the number
of available RBs. In NU-RB-P-OFDM, the signal of several contiguous
RBs is precoded together by pre-stored precoding matrices to reduce
the spectral efficiency loss caused by the increment of RBs. On the
other hand, the larger size of precoding matrices may cause the
larger complexity. Therefore, if the number of RBs in a contiguous
frequency band is larger than a certain value, the signal may be
divided into several sub blocks, and each sub block may be precoded
by the stored precoding matrices independently. In this way, a
tradeoff between spectral efficiency loss and complexity can be
made according to requirements of the systems.
[0184] A non-contiguous transmission band consisting of Q
contiguous frequency bands (FBs), e.g., 0.sup.th, 1.sup.st,
2.sup.nd, . . . , (Q-1).sup.th FB, is utilized by an OFDM-based
cognitive radio (CR) user. Each FB has N.sub.i contiguous
subcarriers. Thus, the total number of subcarriers used by the CR
user is N=.SIGMA..sub.i=0.sup.Q-1N.sub.i. Since each OFDM symbol is
precoded and decoded independent from other OFDM symbols, for
simplicity, the symbol index is ignored hereafter. The time-domain
OFDM transmit symbol .phi.(t), is expressed as:
.phi.(t)=.SIGMA..sub.i=0.sup.Q-1.phi..sub.i(t)=.SIGMA..sub.i=0.sup.Q-1.S-
IGMA..sub.j=0.sup.N.sup.i.sup.-1d.sub.i,jp.sub.i,j(t), Equation
(54)
where .phi..sub.i(t) is the OFDM transmit symbol modulated by the
i.sup.th FB. For the j.sup.th subcarrier in the i.sup.th FB,
d.sub.i,j is the data symbol and p.sub.i,j is the windowed
subcarrier waveform that is expressed as:
p i , j ( t ) = e j 2 .pi. i , j T s t g c ( t ) , Equation ( 55 )
##EQU00052##
with the pulse shape function as follows:
g c ( t ) = { 1 , - T CP .ltoreq. t < T s 0 , otherwise .
Equation ( 56 ) ##EQU00053##
[0185] In equations (55) and (56), T.sub.s is the effective symbol
duration and T.sub.CP is the cyclic prefix duration. The subcarrier
index N.sub.i,j is:
N.sub.i,j=N.sub.i,0+j,i=0,1, . . . ,Q-1,j=0,1, . . . ,N.sub.i-1,
Equation (57)
[0186] The frequency domain representation of the transmit signal
.phi.(t) in equation (54) at certain frequency f may be expressed
as follows:
X(f)=.SIGMA..sub.i=0.sup.Q-1X.sub.i(f)=.SIGMA..sub.i=0.sup.Q-1.SIGMA..su-
b.j=0.sup.N.sup.i.sup.-1d.sub.i,jP.sub.i,j(f), Equation (54)
with
P i , j ( f ) = 1 T .intg. - .infin. + .infin. p i , j ( t ) e - j
2 .pi. ft dt = e j .pi. ( i , j T s - f ) ( T s - T CP ) .pi. ( i ,
j T s - f ) T sin ( .pi. ( i , j T s - f ) T ) , Equation ( 59 )
##EQU00054##
where T=T.sub.CP+T.sub.s is the OFDM symbol duration, and
x.sub.i(f) is the frequency response of .phi..sub.i(t). To suppress
X(f) at notched frequency f.sub.k, k=0, 1, . . . , K-1 for the
user, a precoding process {G.sub.ij,mn} may be imposed on the data
symbol d.sub.m,n, that may be expressed as:
s.sub.i,j=.SIGMA..sub.m=0.sup.Q-1.SIGMA..sub.n=0.sup.M.sup.i.sup.-1G.sub-
.ij,mnd.sub.m,n Equation (60)
where M.sub.i is the number of data streams transmitted in the
i.sup.th FB. The symbols s.sub.i,j, instead of the data symbols,
are mapped to the subcarriers. Therefore, X(f.sub.k) in equation
(58) becomes as follows:
X(f.sub.k)=.SIGMA..sub.i=0.sup.Q-1.SIGMA..sub.j=0.sup.N.sup.i.sup.-1s.su-
b.i,jP.sub.i,j(f.sub.k). Equation (61)
[0187] Due to precoding, N.sub.i.gtoreq.M.sub.1, and the total
number of data streams is M=.SIGMA..sub.i=0.sup.Q-1M.sub.i. Using
matrix expressions, equations (60) and (61) become as follows:
x=Ps,s=Gd, Equation (62)
where
x = [ X ( f 0 ) X ( f 1 ) X ( f K - 1 ) ] , s = [ s 0 s 1 s Q - 1 ]
, s i = [ s i , 0 s i , 1 s i , N , - 1 ] , d = [ d 0 d 1 d Q - 1 ]
, d i = [ d i , 0 d i , 1 d i , M i - 1 ] Equation ( 63 ) P = [ P 0
P 1 P Q - 1 ] Equation ( 64 ) P i = [ P i , 0 ( f 0 ) P i , N i - 1
( f 0 ) P i , 0 ( f K - 1 ) P i , N i - 1 ( f K - 1 ) ] , G = [ G
ij , mn ] . Equation ( 65 ) ##EQU00055##
[0188] The precoding may be performed to suppress the OOBE of the
transmit signal, which may be done by suppressing the out-of-band
power leakage at the notched frequencies f.sub.k, k=0, 1, . . . ,
K-1, i.e., minimizing .parallel.x.parallel.. If the precoding
matrix G is chosen as a semi-unitary matrix such that
G.sup.HG=I.sub.m, at the receiver, after discrete Fourier transform
(DFT) or fast Fourier transform (FFT) and frequency domain
equalization, the received precoded data, denoted by s, may be
decoded by the decoding matrix G.sup.H. Thus, the estimated data
vector is represented as:
{circumflex over (d)}=G.sup.Hs. Equation (66)
[0189] The precoding may provide satisfying spectral containment
for non-contiguous spectrum. However, the design of the precoding
matrix depends on the assigned subcarriers and the chosen notched
frequencies according to equation (59). Furthermore, since G is an
N.times.(N-R) matrix, the complexity (evaluated in terms of the
number of real multiplications) of the precoding process in
equation (62) and decoding process in equation (66) has an order of
O(N.sup.2), which may be unacceptable when N is large.
[0190] In one example, a modified individual precoding scheme,
called I-P-OFDM, may be implemented to reduce the complexity by
precoding each FB independently. As shown in equations (58) and
(59), X(f) is the linear combination of P.sub.i,j(f), which is the
frequency response of the windowed subcarrier waveform at frequency
f and depends on the relative spacing between frequency f and the
subcarrier frequency
( N i , j T s ) . ##EQU00056##
For a chosen notched frequency f between the i.sup.th and
(i+1).sup.th FBs, the signals of the i.sup.th and (i+1).sup.th FBs
have a larger OOBE at f than the signals of other FBs have,
i.e.,
|x.sub.i(f)|.sup.2.gtoreq.|X.sub.l(f)|.sup.2,|X.sub.i+1(f)|.sup.2.gtoreq-
.|X.sub.l(f)|.sup.2,l=0,1, . . . ,Q-1,l.noteq.i,i+1. Equation
(67)
[0191] As a result, the problem of minimizing .parallel.x.parallel.
can be simplified to independently minimizing the OOBE at each
notched frequency brought by its two closest FBs on both sides as
follows:
x=[X.sub.0(f.sub.0)X.sub.1(f.sub.1), . . .
,X.sub.Q-1(f.sub.Q-1)].sup.T=[P.sub.0s.sub.0P.sub.1S.sub.1, . . .
,P.sub.Q-1s.sub.Q-1].sup.T,
s.sub.i=G.sub.id.sub.i,i=0,1, . . . ,Q-1, Equation (68)
where f.sub.i=[f.sub.i,0, f.sub.i,1, . . . ,
f.sub.i,K.sub.i.sub.-1] are the notched frequencies chosen for the
i.sup.th FB and
P i = [ P i , 0 ( f i , 0 ) P i , N i - 1 ( f i , 0 ) P i , 0 ( f i
, K i - 1 ) P i , N i - 1 ( f i , K i - 1 ) ] , G i = [ G ij , n ]
. Equation ( 69 ) ##EQU00057##
[0192] To minimize .parallel.x.parallel., the singular value
decomposition (SVD) may be performed on each P.sub.i as
follows:
P.sub.i=U.sub.i.SIGMA..sub.iV.sub.i.sup.H. Equation (70)
[0193] The N.sub.i.times.M.sub.1 precoding matrix for the i.sup.th
FB may be then chosen as:
G.sub.i=.left brkt-bot.v.sub.i,N.sub.i.sub.-M.sub.i.sub.+1 . . .
v.sub.i,N.sub.i.sub.-2v.sub.i,N.sub.i.sub.-1.right brkt-bot.
Equation (71)
[0194] Since M.sub.i.ltoreq.M, the complexity may be reduced
dramatically in most cases when M.sub.i<<M and the overall
coding rate is as follows:
.lamda. = 1 - i = 0 Q - 1 R i N . Equation ( 72 ) ##EQU00058##
[0195] FIG. 36 is a block diagram of an example I-P-OFDM
transceiver structure 3600. Referring to FIG. 36, the I-P-OFDM
transceiver may comprise a transmitter block 3601 and a receiver
block 3602. The transmitter block 3601 may comprise precoding units
3605a, 3605b, . . . , 3605k, a L-point IDFT unit 3610, a CP adder
unit 3615, and a P/S converter 3618. The receiver block 3602 may
comprise a CP removal unit 3620, a SIP converter 3625, a L-point
DFT unit 3630, and deprecoding units 3635a, 3635b, . . . ,
3635k.
[0196] Referring to FIG. 36, the symbol vector of each FB 3604a,
3604b, . . . , 3604k is precoded independently at the precoding
units 3605a, 3605b, . . . , 3605k, outputting precoded symbol
vectors 3606a, 3606b, . . . , 3606k. The output precoded symbol
vectors 3606a, 3606b, . . . , 3606k are then modulated together
through inverse discrete Fourier transform (IDFT) at the L-point
IDFT unit 3610. Some FBs may have the same number of subcarriers,
but the number of data streams and coding rates and the way of
choosing notched frequencies and precoding matrices may be
different. This scheme provides computational complexity reduction,
compared to the conventional precoding scheme. In some cases, an FB
may have a large number of subcarriers such that the precoding
matrix has a large dimension. Moreover, the design of the precoding
matrices still depends on the assigned subcarriers and the chosen
notched frequencies.
[0197] The transmitter block 3601 in FIG. 36 may be alternatively
represented similar to FIGS. 4 and 5 where the per-RB transmit
processing module is the transmit module shown in FIG. 37. FIG. 37
shows one example of the per-RB transmit processing module 3700
that comprises a precoding unit 3705, a L-point IDFT unit 3710, a
CP adder unit 3715, and a P/S converter 3720, and a RB modulation
unit 3725.
[0198] The receiver block 3602 in FIG. 36 may be alternatively
represented as in FIG. 6 where the per-RB receive processing module
is depicted in FIG. 38. FIG. 38 shows one example of the per-RB
receive processing module 3800. The per-RB receive processing
module 3800 comprises a RB demodulation unit 3805, a CP remover
unit 3810, a S/P converter 3815, a L-point DFT unit 3820, and a
deprecoding unit 3825. The implementation in FIG. 36 has much less
complexity because all FBs share the same IFFT operation at the
transmitter and same FFT operation at the receiver.
[0199] RB-P-OFDM may provide uniform or fixed precoding and
decoding matrices, which have relatively smaller sizes and do not
depend on the allocated subcarriers and notched frequencies, for
non-contiguous spectrum to obtain satisfying spectrum containment.
In one embodiment for the RB-P-OFDM, each FB may be divided into B
RBs (where each RB is a group of N.sub.B contiguous subcarriers and
is viewed as the smallest available transmission band) and the
signals are precoded by predetermined precoding matrices.
[0200] Referring back to FIG. 36, when the signal is received at
the I-P-OFDM receive block 3602, the signal may go through the CP
removal unit 3620, the S/P converter 3625, the L-point DFT unit
3630, and deprecoding units 3635a, 3635b, 3635k, which output
estimated data vectors 3640a, 3640b, . . . , 3640k.
[0201] Similar to I-P-OFDM, the implementation of the RB-P-OFDM
transmitters may use one IFFT operation for all RBs and the
implementation of the RB-P-OFDM receivers may use one FFT operation
for all RBs.
[0202] FIG. 39 is a block diagram of an example uniform resource
block-based precoded OFDM (U-RB-P-OFDM) structure 3900. Referring
to FIG. 39, the U-RB-P-OFDM structure includes a U-RB-P-OFDM
transmitter block 3901 and a U-RB-P-OFDM receiver block 3902. The
U-RB-P-OFDM transmitter block 3901 comprises precoding units 3910a,
3910b, . . . , 3910k, an L-point IDFT unit 3915, a CP adder unit
3920, and a P/S converter 3925. The U-RB-P-OFDM receiver block 3902
comprises a CP removal unit 3930, a S/P converter 3935, a L-point
DFT unit 3940, and deprecoding unit 3945a, 3945b, 3945k. In the
U-RB-P-OFDM transmitter block 3901, the symbol vector of each RB
3905a, 3905b, . . . , 3905k may be precoded at the precoding units
3910a, 3910b, . . . , 3910k by a uniform precoding matrix, G,
independently. The precoding units 3910a, 3910b, . . . , 3910k
output precoded symbol vectors 3906a, 3906b, . . . , 3906k.
According to equation (59), the matrix P.sub.i for the i.sup.th RB
depends on
( N i , j T s - f ) . ##EQU00059##
Once N.sub.B, the size of each RB, and the spacing between the
central frequency of each RB and the notched frequencies f are
determined, matrix P.sub.i may be fixed. Then a unique precoding
matrix G with coding redundancy RB may be obtained by equation
(71). In order to determine the precoding matrix for all the RBs,
different notched frequencies may be chosen for the RBs in a
contiguous band so that the spacing between each RB's central
frequency and the corresponding chosen notched frequency may be the
same as the others. This uniform regulation of same
( N i , j T s - f ) ##EQU00060##
for all i thus brings convenience for the design of the precoding
matrices. In addition, since the dimension of G is decreased from N
to N.sub.B, the computational complexity of the precoding and
decoding operation is also decreased. Since R.sub.B is the length
of redundancy for each RB, the total coding rate may be as
follows:
.lamda. = 1 - i = 0 B - 1 R B N = 1 - R B N B Equation ( 73 )
##EQU00061##
[0203] As shown in equation (73), the spectral efficiency of the
U-RB-P-OFDM system reduces as the number of RBs or the coding
redundancy increases. The spectral efficiency loss may be reduced
by precoding the signal of contiguous RBs together so that the
redundancy does not have to be reserved for every RB repetitively.
Denote the predetermined precoding matrix as G'.sub.i, i=1, 2, . .
. , W. The signal of contiguous i RBs may be precoded together by
the stored precoding matrix G'.sub.i, Here G'.sub.i is the
precoding matrix for an FB with i RBs, which depends on i. By this
method, the spectral efficiency loss may be reduced dramatically,
but when the maximum number of contiguous RBs, W, is too large, the
memory used to store the precoding matrices will be too large. On
the other hand, the complexity of precoding process, which is
proportional to (WN.sub.b).sup.2, may be too high. In light of
this, in one embodiment, an FB may be divided into several sub
blocks when the number of RBs in this FB is larger than a certain
value, W.sub.B, and the signal of each sub block may be precoded
independently. As a result, the complexity and the number of
precoding matrices that need to be stored may be reduced. Since the
sub blocks may be considered as virtual RBs of non-uniform sizes,
such a scheme is named NU-RB-P-OFDM. It provides a tradeoff between
the spectral efficiency loss and complexity by choosing W.sub.B
according to the system requirements.
[0204] Referring back to FIG. 39, after the symbol vector of each
RB 3905a, 3905b, . . . , 3905k is precoded at the precoding units
3910a, 3910b, . . . , 3910k, the signals go through the L-point
IDFT unit 3915, the CP adder unit 3920, and the P/S converter 3925,
which outputs the transmit signal. When the signal is received at
the U-RB-P-OFDM receiver block 3902, CP is removed at the CP
removal unit 3920, the signal goes through the S/P converter 3935,
an L-point DFT unit 3940 that outputs demodulated symbol vectors
3942a, 3942b, . . . , 3942k, and deprecoding units 3945a, 3945b, .
. . , 3945k which output estimated data vectors 3950a, 3950b, . . .
, 3950k.
[0205] Although features and elements are described above in
particular combinations, one of ordinary skill in the art will
appreciate that each feature or element can be used alone or in any
combination with the other features and elements. In addition, the
methods described herein may be implemented in a computer program,
software, or firmware incorporated in a computer-readable medium
for execution by a computer or processor. Examples of
computer-readable media include electronic signals (transmitted
over wired or wireless connections) and computer-readable storage
media. Examples of computer-readable storage media include, but are
not limited to, a read only memory (ROM), a random access memory
(RAM), a register, cache memory, semiconductor memory devices,
magnetic media such as internal hard disks and removable disks,
magneto-optical media, and optical media such as CD-ROM disks, and
digital versatile disks (DVDs). A processor in association with
software may be used to implement a radio frequency transceiver for
use in a WTRU, UE, terminal, base station, RNC, or any host
computer.
* * * * *