U.S. patent application number 14/980185 was filed with the patent office on 2017-06-29 for sensor systems, devices, and methods for continuous glucose monitoring.
The applicant listed for this patent is MEDTRONIC MINIMED, INC.. Invention is credited to Taly G. Engel, Jaeho Kim, Xiaolong Li, Bradley C. Liang, Mike C. Liu, Keith Nogueira, Rajiv Shah, Andy Y. Tsai, Andrea Varsavsky, Fei Yu.
Application Number | 20170184527 14/980185 |
Document ID | / |
Family ID | 57491744 |
Filed Date | 2017-06-29 |
United States Patent
Application |
20170184527 |
Kind Code |
A1 |
Nogueira; Keith ; et
al. |
June 29, 2017 |
SENSOR SYSTEMS, DEVICES, AND METHODS FOR CONTINUOUS GLUCOSE
MONITORING
Abstract
Electrochemical impedance spectroscopy (EIS) may be used in
conjunction with continuous glucose monitoring (CGM) to enable
identification of valid and reliable sensor data, as well
implementation of Smart Calibration algorithms.
Inventors: |
Nogueira; Keith; (Mission
Hills, CA) ; Engel; Taly G.; (Los Angeles, CA)
; Li; Xiaolong; (Porter Ranch, CA) ; Liang;
Bradley C.; (Bloomfield Hills, MI) ; Shah; Rajiv;
(Rancho Palos Verdes, CA) ; Kim; Jaeho; (Redmond,
WA) ; Liu; Mike C.; (Walnut, CA) ; Tsai; Andy
Y.; (Pasadena, CA) ; Varsavsky; Andrea; (Santa
Monica, CA) ; Yu; Fei; (Chatsworth, CA) |
|
Applicant: |
Name |
City |
State |
Country |
Type |
MEDTRONIC MINIMED, INC. |
Northridge |
CA |
US |
|
|
Family ID: |
57491744 |
Appl. No.: |
14/980185 |
Filed: |
December 28, 2015 |
Current U.S.
Class: |
1/1 |
Current CPC
Class: |
G01N 33/48707 20130101;
G01N 33/49 20130101; G01N 27/026 20130101 |
International
Class: |
G01N 27/02 20060101
G01N027/02; G01N 33/487 20060101 G01N033/487; G01N 27/416 20060101
G01N027/416; G01N 33/49 20060101 G01N033/49 |
Claims
1. A method for determining the validity of glucose sensor data,
said glucose sensor including physical sensor electronics, a
microcontroller, and a working electrode, and being in operational
contact with a display device configured to display said data to a
user, the method comprising: (a) performing, by said
microcontroller, an electrochemical impedance spectroscopy (EIS)
procedure to obtain real impedance values for said electrode; (b)
filtering, by said microcontroller, said real impedance values; (c)
analyzing said real impedance values by said microcontroller to
determine whether said values are stable; (d) if said real
impedance values are stable, comparing the most-recent real
impedance value to a first threshold value; and (e) based on said
comparison, determining whether said sensor data is valid.
2. The method of claim 1, wherein said real impedance values are 1
kHz real impedance values.
3. The method of claim 1, wherein, if the glucose sensor data is
determined to be valid, said data is transmitted to be displayed on
said display device.
4. The method of claim 1, wherein said first threshold value is
10,000.OMEGA..
5. The method of claim 4, wherein the sensor data is determined to
be valid if the most-recent real impedance value is less than the
first threshold value.
6. The method of claim 5, wherein, if the most-recent real
impedance value is greater than the first threshold value, the
method further includes determining, by the microcontroller,
whether said real impedance values have exceeded a second threshold
over a period of time.
7. The method of claim 6, wherein said period of time is the past 3
hours.
8. The method of claim 7, wherein if it is determined that the real
impedance values have exceeded said second threshold over the past
3 hours, then said sensor is terminated.
9. The method of claim 7, wherein, if it is determined that the
real impedance values have not exceeded said second threshold over
the past 3 hours, then said sensor data is determined to be invalid
and is not displayed on the display device.
10. The method of claim 9, further including periodically repeating
steps (a)-(e).
11. The method of claim 10, wherein, once the sensor data is
determined to be valid, said data is transmitted to be displayed on
said display device.
12. The method of claim 6, wherein said second threshold is between
about 10,000.OMEGA. and about 12,000.OMEGA..
13. A method for determining the validity of glucose sensor data,
said glucose sensor including physical sensor electronics, a
microcontroller, and a working electrode, the method comprising:
(a) performing, by said microcontroller, an electrochemical
impedance spectroscopy (EIS) procedure to obtain imaginary
impedance values for said electrode; (b) setting a threshold
reference for said imaginary impedance values; (c) calculating a
change value as a difference between the threshold reference and
the most-recent imaginary impedance value; (d) obtaining
measurements of the calibration factor for said sensor; (e)
comparing, by said microcontroller, said change value to a first
threshold and said calibration factor to a second threshold; and
(f) based on said comparison, determining whether said sensor data
is valid, such that the sensor can continue to operate, or said
data is invalid, such that the sensor should be terminated.
14. The method of claim 13, wherein said imaginary impedance values
are 8 kHz imaginary impedance values.
15. The method of claim 13, wherein said threshold reference is
calculated as the minimum 8 kHz imaginary impedance value since
sensor initialization.
16. The method of claim 15, wherein said threshold reference is
clipped so as to fall within the range -1,000.OMEGA. and
800.OMEGA..
17. The method of claim 13, wherein said change value is calculated
as the absolute difference between the threshold reference and the
most-recent imaginary impedance value.
18. The method of claim 13, wherein said first threshold value is
1,200.OMEGA. and said second threshold value is 14 mg/dL/nA.
19. The method of claim 13, wherein the sensor data is determined
to be valid if either the calibration factor is less than the
second lower threshold or the change value is less than the first
lower threshold for two consecutive measurements of the change
value.
20. The method of claim 13, further including terminating the
sensor if the change value is greater than the first threshold for
two consecutive measurements of the change value, and the
calibration factor is greater than said second threshold.
21. The method of claim 13, wherein the sensor includes a plurality
of working electrodes, and steps (a)-(f) are performed for each of
the plurality of electrodes.
Description
FIELD OF THE INVENTION
[0001] Embodiments of this invention are related generally to
subcutaneous and implantable sensor devices and, in particular
embodiments, to systems, devices, and methods for continuous
glucose monitoring (CGM).
BACKGROUND OF THE INVENTION
[0002] Over the years, a variety of sensors have been developed for
detecting and/or quantifying specific agents or compositions in a
patient's blood, which enable patients and medical personnel to
monitor physiological conditions within the patient's body.
Illustratively, subjects may wish to monitor blood glucose levels
in a subject's body on a continuing basis. Thus, glucose sensors
have been developed for use in obtaining an indication of blood
glucose levels in a diabetic patient. Such readings are useful in
monitoring and/or adjusting a treatment regimen which typically
includes the regular administration of insulin to the patient.
[0003] Presently, a patient can measure his/her blood glucose (BG)
using a BG measurement device (i.e., glucose meter), such as a test
strip meter, a continuous glucose measurement system (or a
continuous glucose monitor), or a hospital hemacue. BG measurement
devices use various methods to measure the BG level of a patient,
such as a sample of the patient's blood, a sensor in contact with a
bodily fluid, an optical sensor, an enzymatic sensor, or a
fluorescent sensor. When the BG measurement device has generated a
BG measurement, the measurement is displayed on the BG measurement
device.
[0004] Current continuous glucose measurement systems include
subcutaneous (or short-term) sensors and implantable (or long-term)
sensors. Sensors have been applied in a telemetered characteristic
monitor system. As described, e.g., in commonly-assigned U.S. Pat.
No. 6,809,653, the entire contents of which are incorporated herein
by reference, a telemetered system using an electrochemical sensor
includes a remotely located data receiving device, a sensor for
producing signals indicative of a characteristic of a user, and a
transmitter device for processing signals received from the sensor
and for wirelessly transmitting the processed signals to the
remotely located data receiving device. The data receiving device
may be a characteristic monitor, a data receiver that provides data
to another device, an RF programmer, a medication delivery device
(such as an infusion pump), or the like.
[0005] Regardless of whether the data receiving device (e.g., a
glucose monitor), the transmitter device, and the sensor (e.g., a
glucose sensor) communicate wirelessly or via an electrical wire
connection, a characteristic monitoring system of the type
described above is of practical use only after it has been
calibrated based on the unique characteristics of the individual
user. According to the current state of the art, the user is
required to externally calibrate the sensor. More specifically, and
in connection with the illustrative example of a diabetic patient,
the latter is required to utilize a finger-stick blood glucose
meter reading an average of two-four times per day for the duration
that the characteristic monitor system is used. Each time, blood is
drawn from the user's finger and analyzed by the blood glucose
meter to provide a real-time blood sugar level for the user. The
user then inputs this data into the glucose monitor as the user's
current blood sugar level which is used to calibrate the glucose
monitoring system.
[0006] Such external calibrations, however, are disadvantageous for
various reasons. For example, blood glucose meters are not
perfectly accurate and include inherent margins of error. Moreover,
even if completely accurate, blood glucose meters are susceptible
to improper use; for example, if the user has handled candy or
other sugar-containing substance immediately prior to performing
the finger stick, with some of the sugar sticking to the user's
fingers, the blood sugar analysis will result in an inaccurate
blood sugar level indication. Furthermore, there is a cost, not to
mention pain and discomfort, associated with each application of
the finger stick.
[0007] The current state of the art in continuous glucose
monitoring (CGM) is largely adjunctive, meaning that the readings
provided by a CGM device (including, e.g., an implantable or
subcutaneous sensor) cannot be used without a reference value in
order to make a clinical decision. The reference value, in turn,
must be obtained from a finger stick using, e.g., a BG meter. The
reference value is needed because there is a limited amount of
information that is available from the sensor/sensing component.
Specifically, the only pieces of information that are currently
provided by the sensing component for processing are the raw sensor
value (i.e., the sensor current or Isig) and the counter voltage.
Therefore, during analysis, if it appears that the raw sensor
signal is abnormal (e.g., if the signal is decreasing), the only
way one can distinguish between a sensor failure and a
physiological change within the user/patient (i.e., glucose level
changing in the body) is by acquiring a reference glucose value via
a finger stick. As is known, the reference finger stick is also
used for calibrating the sensor.
[0008] The art has searched for ways to eliminate or, at the very
least, minimize, the number of finger sticks that are necessary for
calibration and for assessing sensor health. However, given the
number and level of complexity of the multitude of sensor failure
modes, no satisfactory solution has been found. At most,
diagnostics have been developed that are based on either direct
assessment of the Isig, or on comparison of multiple Isigs, e.g.,
from redundant and/or orthogonally redundant, sensors and/or
electrodes. In either case, because the Isig tracks the level of
glucose in the body, by definition, it is not analyte independent.
As such, by itself, the Isig is not a reliable source of
information for sensor diagnostics, nor is it a reliable predictor
for continued sensor performance.
[0009] Another limitation that has existed in the art thus far has
been the lack of sensor electronics that can not only run the
sensor, but also perform real-time sensor and electrode
diagnostics, and do so for redundant electrodes, all while managing
the sensor's power supply. To be sure, the concept of electrode
redundancy has been around for quite some time. However, up until
now, there has been little to no success in using electrode
redundancy not only for obtaining more than one reading at a time,
but also for assessing the relative health of the redundant
electrodes, the overall reliability of the sensor, and the
frequency of the need, if at all, for calibration reference
values.
[0010] The art has also searched for more accurate and reliable
means for providing self-calibrating sensors, and for performing
sensor diagnostics by developing a variety of circuit models. In
such models, an attempt is generally made to correlate circuit
elements to parameters that may be used in intelligent diagnostics,
gross failure analysis, and real-time self-calibrations. However,
most such models have had limited success thus far.
[0011] For each of the short-term sensors and the long-term
sensors, a patient has to wait a certain amount of time in order
for the continuous glucose sensor to stabilize and to provide
accurate readings. In many continuous glucose sensors, the subject
must wait three hours for the continuous glucose sensor to
stabilize before any glucose measurements are utilized. This is an
inconvenience for the patient and in some cases may cause the
patient not to utilize a continuous glucose measurement system.
[0012] Further, when a glucose sensor is first inserted into a
patient's skin or subcutaneous layer, the glucose sensor does not
operate in a stable state. The electrical readings from the sensor,
which represent the glucose level of the patient, vary over a wide
range of readings. Thus, the sensor must first be stabilized. It is
also desirable to allow electrodes of the sensor to be sufficiently
"wetted" or hydrated before utilization of the electrodes of the
sensor. If the electrodes of the sensor are not sufficiently
hydrated, the result may be inaccurate readings of the patient's
physiological condition. A user of current blood glucose sensors is
instructed to not power up the sensors immediately. If they are
utilized too early, current blood glucose sensors do not operate in
an optimal or efficient fashion.
SUMMARY
[0013] According to an embodiment of the invention, a method for
real-time calibration of a glucose sensor for measuring the level
of glucose in a body of a user, the sensor having physical sensor
electronics, a microcontroller, and a working electrode, comprises:
measuring, by the physical sensor electronics, the electrode
current (Isig) for the working electrode; obtaining a blood glucose
(BG) value for the user; calculating, by the microcontroller, an
expected calibration factor (CF) value based on the glucose
sensor's age; and calculating, by the microcontroller, a calibrated
sensor glucose (SG) value associated with the Isig based on the CF
and BG values.
[0014] In accordance with another embodiment of the invention, a
method for determining the validity of glucose sensor data, the
glucose sensor having physical sensor electronics, a
microcontroller, and a working electrode, and being in operational
contact with a display device configured to display the data to a
user, comprises: performing, by the microcontroller, an
electrochemical impedance spectroscopy (EIS) procedure to obtain
real impedance values for the electrode; filtering, by the
microcontroller, the real impedance values; analyzing the real
impedance values by the microcontroller to determine whether the
values are stable; if the real impedance values are stable,
comparing the most-recent real impedance value to a first threshold
value; and based on the comparison, determining whether the sensor
data is valid.
[0015] In accordance with another embodiment of the invention, a
method for determining the validity of glucose sensor data, the
glucose sensor including physical sensor electronics, a
microcontroller, and a working electrode, comprises: performing, by
the microcontroller, an electrochemical impedance spectroscopy
(EIS) procedure to obtain imaginary impedance values for the
electrode; setting a threshold reference for the imaginary
impedance values; calculating a change value as a difference
between the threshold reference and the most-recent imaginary
impedance value; obtaining measurements of the calibration factor
for the sensor; comparing, by the microcontroller, the change value
to a first threshold and the calibration factor to a second
threshold; and based on the comparison, determining whether the
sensor data is valid, such that the sensor can continue to operate,
or the data is invalid, such that the sensor should be
terminated.
[0016] In yet another embodiment of the invention, a method for
signal dip detection during the first 4-12 hours of glucose sensor
data, the glucose sensor including physical sensor electronics, a
microcontroller, and a working electrode, and being in operational
contact with a display device configured to display the data to a
user, comprises: performing, by the microcontroller, an
electrochemical impedance spectroscopy (EIS) procedure to obtain
real impedance values for the electrode; periodically measuring, by
the physical sensor electronics, values of the electrode current
(Isig) for the working electrode; calculating, by the
microcontroller, sensor glucose (SG) values associated with the
Isig values; comparing a current value of the Isig to a first
threshold and the current value of the SG to a second threshold;
evaluating, by the microcontroller, a trend of the real impedance
values; and based on the comparison and the evaluation, determining
whether a dip event exists.
[0017] In a further embodiment of the invention, a method for
signal dip detection during the first 4 hours of glucose sensor
data, the glucose sensor including physical sensor electronics, a
microcontroller, and a working electrode, and being in operational
contact with a display device configured to display the data to a
user, comprises: performing, by the microcontroller, an
electrochemical impedance spectroscopy (EIS) procedure to obtain
real impedance values for the electrode; periodically measuring, by
the physical sensor electronics, values of the electrode current
(Isig) for the working electrode; comparing a current value of the
Isig to a first threshold; evaluating, by the microcontroller, a
trend of the real impedance values; and based on the comparison and
the evaluation, determining whether a dip event exists.
[0018] In another embodiment of the invention, a method for first
day calibration (FDC) of a glucose sensor for measuring the level
of glucose in a body of a user, the sensor including physical
sensor electronics, a microcontroller, and a working electrode,
comprises: measuring, by the physical sensor electronics, the
electrode current (Isig) for the working electrode; calculating, by
the microcontroller, a calibration ratio (CR); comparing the
calibration ratio to a threshold range; and based on the
comparison, calculating a time interval until the next
calibration.
BRIEF DESCRIPTION OF THE DRAWINGS
[0019] A detailed description of embodiments of the invention will
be made with reference to the accompanying drawings, wherein like
numerals designate corresponding parts in the figures.
[0020] FIG. 1 is a perspective view of a subcutaneous sensor
insertion set and block diagram of a sensor electronics device
according to an embodiment of the invention.
[0021] FIG. 2A illustrates a substrate having two sides, a first
side which contains an electrode configuration and a second side
which contains electronic circuitry.
[0022] FIG. 2B illustrates a general block diagram of an electronic
circuit for sensing an output of a sensor.
[0023] FIG. 3 illustrates a block diagram of a sensor electronics
device and a sensor including a plurality of electrodes according
to an embodiment of the invention.
[0024] FIG. 4 illustrates an alternative embodiment of the
invention including a sensor and a sensor electronics device
according to an embodiment of the invention.
[0025] FIG. 5 illustrates an electronic block diagram of the sensor
electrodes and a voltage being applied to the sensor electrodes
according to an embodiment of the invention.
[0026] FIG. 6A illustrates a method of applying pulses during a
stabilization timeframe in order to reduce the stabilization
timeframe according to an embodiment of the invention.
[0027] FIG. 6B illustrates a method of stabilizing sensors
according to an embodiment of the invention.
[0028] FIG. 6C illustrates utilization of feedback in stabilizing
the sensors according to an embodiment of the invention.
[0029] FIG. 7 illustrates an effect of stabilizing a sensor
according to an embodiment of the invention.
[0030] FIG. 8A illustrates a block diagram of a sensor electronics
device and a sensor including a voltage generation device according
to an embodiment of the invention.
[0031] FIG. 8B illustrates a voltage generation device to implement
this embodiment of the invention.
[0032] FIG. 8C illustrates a voltage generation device to generate
two voltage values according to an embodiment of the invention.
[0033] FIG. 8D illustrates a voltage generation device having three
voltage generation systems, according to embodiments of the
invention.
[0034] FIG. 9A illustrates a sensor electronics device including a
microcontroller for generating voltage pulses according to an
embodiment of the invention.
[0035] FIG. 9B illustrates a sensor electronics device including an
analyzation module according to an embodiment of the invention.
[0036] FIG. 10 illustrates a block diagram of a sensor system
including hydration electronics according to an embodiment of the
invention.
[0037] FIG. 11 illustrates an embodiment of the invention including
a mechanical switch to assist in determining a hydration time.
[0038] FIG. 12 illustrates a method of detection of hydration
according to an embodiment of the invention.
[0039] FIG. 13A illustrates a method of hydrating a sensor
according to an embodiment of the present invention.
[0040] FIG. 13B illustrates an additional method for verifying
hydration of a sensor according to an embodiment of the
invention.
[0041] FIGS. 14A, 14B, and 14C illustrate methods of combining
hydrating of a sensor with stabilizing a sensor according to an
embodiment of the invention.
[0042] FIG. 15A illustrates EIS-based analysis of system response
to the application of a periodic AC signal in accordance with
embodiments of the invention.
[0043] FIG. 15B illustrates a known circuit model for
electrochemical impedance spectroscopy.
[0044] FIG. 16A illustrates an example of a Nyquist plot where, for
a selected frequency spectrum from 0.1 Hz to 1000 Mhz, AC voltages
plus a DC voltage (DC bias) are applied to the working electrode in
accordance with embodiments of the invention.
[0045] FIG. 16B shows another example of a Nyquist plot with a
linear fit for the relatively-lower frequencies and the intercept
approximating the value of real impedance at the relatively-higher
frequencies.
[0046] FIGS. 16C and 16D show, respectively, infinite and finite
glucose sensor response to a sinusoidal working potential.
[0047] FIG. 16E shows a Bode plot for magnitude in accordance with
embodiments of the invention.
[0048] FIG. 16F shows a Bode plot for phase in accordance with
embodiments of the invention.
[0049] FIG. 17 illustrates the changing Nyquist plot of sensor
impedance as the sensor ages in accordance with embodiments of the
invention.
[0050] FIG. 18 illustrates methods of applying EIS technique in
stabilizing and detecting the age of the sensor in accordance with
embodiments of the invention.
[0051] FIG. 19 illustrates a schedule for performing the EIS
procedure in accordance with embodiments of the invention.
[0052] FIG. 20 illustrates a method of detecting and repairing a
sensor using EIS procedures in conjunction with remedial action in
accordance with embodiments of the invention.
[0053] FIGS. 21A and 21B illustrate examples of a sensor remedial
action in accordance with embodiments of the invention.
[0054] FIG. 22 shows a Nyquist plot for a normally-functioning
sensor where the Nyquist slope gradually increases, and the
intercept gradually decreases, as the sensor wear-time
progresses.
[0055] FIG. 23A shows raw current signal (Isig) from two redundant
working electrodes, and the electrodes' respective real impedances
at 1 kHz, in accordance with embodiments of the invention.
[0056] FIG. 23B shows the Nyquist plot for the first working
electrode (WE1) of FIG. 23A.
[0057] FIG. 23C shows the Nyquist plot for the second working
electrode (WE2) of FIG. 23A.
[0058] FIG. 24 illustrates examples of signal dip for two redundant
working electrodes, and the electrodes' respective real impedances
at 1 kHz, in accordance with embodiments of the invention.
[0059] FIG. 25A illustrates substantial glucose independence of
real impedance, imaginary impedance, and phase at relatively-higher
frequencies for a normally-functioning glucose sensor in accordance
with embodiments of the invention.
[0060] FIG. 25B shows illustrative examples of varying levels of
glucose dependence of real impedance at the relatively-lower
frequencies in accordance with embodiments of the invention.
[0061] FIG. 25C shows illustrative examples of varying levels of
glucose dependence of phase at the relatively-lower frequencies in
accordance with embodiments of the invention.
[0062] FIG. 26 shows the trending for 1 kHz real impedance, 1 kHz
imaginary impedance, and relatively-higher frequency phase as a
glucose sensor loses sensitivity as a result of oxygen deficiency
at the sensor insertion site, according to embodiments of the
invention.
[0063] FIG. 27 shows Isig and phase for an in-vitro simulation of
oxygen deficit at different glucose concentrations in accordance
with embodiments of the invention.
[0064] FIGS. 28A-28C show an example of oxygen deficiency-led
sensitivity loss with redundant working electrodes WE1 and WE2, as
well as the electrodes' EIS-based parameters, in accordance with
embodiments of the invention.
[0065] FIG. 28D shows EIS-induced spikes in the raw Isig for the
example of FIG. 28A-28C.
[0066] FIG. 29 shows an example of sensitivity loss due to oxygen
deficiency that is caused by an occlusion, in accordance with
embodiments of the invention.
[0067] FIGS. 30A-30C show an example of sensitivity loss due to
bio-fouling, with redundant working electrodes WE1 and WE2, as well
as the electrodes' EIS-based parameters, in accordance with
embodiments of the invention.
[0068] FIG. 30D shows EIS-induced spikes in the raw Isig for the
example of FIG. 30A-30C.
[0069] FIG. 31 shows a diagnostic procedure for sensor fault
detection in accordance with embodiments of the invention.
[0070] FIGS. 32A and 32B show another diagnostic procedure for
sensor fault detection in accordance with embodiments of the
invention.
[0071] FIG. 33A shows a top-level flowchart involving a current
(Isig)-based fusion algorithm in accordance with embodiments of the
invention.
[0072] FIG. 33B shows a top-level flowchart involving a sensor
glucose (SG)-based fusion algorithm in accordance with embodiments
of the invention.
[0073] FIG. 34 shows details of the sensor glucose (SG)-based
fusion algorithm of FIG. 33B in accordance with embodiments of the
invention.
[0074] FIG. 35 shows details of the current (Isig)-based fusion
algorithm of FIG. 33A in accordance with embodiments of the
invention.
[0075] FIG. 36 is an illustration of calibration for a sensor in
steady state, in accordance with embodiments of the invention.
[0076] FIG. 37 is an illustration of calibration for a sensor in
transition, in accordance with embodiments of the invention.
[0077] FIG. 38A is an illustration of EIS-based dynamic slope (with
slope adjustment) in accordance with embodiments of the invention
for sensor calibration.
[0078] FIG. 38B shows an EIS-assisted sensor calibration flowchart
involving low start-up detection in accordance with embodiments of
the invention.
[0079] FIG. 39 shows sensor current (Isig) and 1 kHz impedance
magnitude for an in-vitro simulation of an interferent being in
close proximity to a sensor in accordance with embodiments of the
invention.
[0080] FIGS. 40A and 40B show Bode plots for phase and impedance,
respectively, for the simulation shown in FIG. 39.
[0081] FIG. 40C shows a Nyquist plot for the simulation shown in
FIG. 39.
[0082] FIG. 41 shows another in-vitro simulation with an
interferent in accordance to embodiments of the invention.
[0083] FIGS. 42A and 42B illustrate an ASIC block diagram in
accordance with embodiments of the invention.
[0084] FIG. 43 shows a potentiostat configuration for a sensor with
redundant working electrodes in accordance with embodiments of the
invention.
[0085] FIG. 44 shows an equivalent AC inter-electrode circuit for a
sensor with the potentiostat configuration shown in FIG. 43.
[0086] FIG. 45 shows some of the main blocks of the EIS circuitry
in the analog front end IC of a glucose sensor in accordance with
embodiments of the invention.
[0087] FIGS. 46A-46F show a simulation of the signals of the EIS
circuitry shown in FIG. 45 for a current of 0-degree phase with a
0-degree phase multiply.
[0088] FIGS. 47A-47F show a simulation of the signals of the EIS
circuitry shown in FIG. 45 for a current of 0-degree phase with a
90-degree phase multiply.
[0089] FIG. 48 shows a circuit model in accordance with embodiments
of the invention.
[0090] FIGS. 49A-49C show illustrations of circuit models in
accordance with alternative embodiments of the invention.
[0091] FIG. 50A is a Nyquist plot overlaying an equivalent circuit
simulation in accordance with embodiments of the invention.
[0092] FIG. 50B is an enlarged diagram of the high-frequency
portion of FIG. 50A.
[0093] FIG. 51 shows a Nyquist plot with increasing Cdl in the
direction of Arrow A, in accordance with embodiments of the
invention.
[0094] FIG. 52 shows a Nyquist plot with increasing a in the
direction of Arrow A, in accordance with embodiments of the
invention.
[0095] FIG. 53 shows a Nyquist plot with increasing Rp in the
direction of Arrow A, in accordance with embodiments of the
invention.
[0096] FIG. 54 shows a Nyquist plot with increasing Warburg
admittance in the direction of Arrow A, in accordance with
embodiments of the invention.
[0097] FIG. 55 shows a Nyquist plot with increasing .lamda. in the
direction of Arrow A, in accordance with embodiments of the
invention.
[0098] FIG. 56 shows the effect of membrane capacitance on the
Nyquist plot, in accordance with embodiments of the invention.
[0099] FIG. 57 shows a Nyquist plot with increasing membrane
resistance in the direction of Arrow A, in accordance with
embodiments of the invention.
[0100] FIG. 58 shows a Nyquist plot with increasing Rsol in the
direction of Arrow A, in accordance with embodiments of the
invention.
[0101] FIGS. 59A-59C show changes in EIS parameters relating to
circuit elements during start-up and calibration in accordance with
embodiments of the invention.
[0102] FIGS. 60A-60C show changes in a different set of EIS
parameters relating to circuit elements during start-up and
calibration in accordance with embodiments of the invention.
[0103] FIGS. 61A-61C show changes in yet a different set of EIS
parameters relating to circuit elements during start-up and
calibration in accordance with embodiments of the invention.
[0104] FIG. 62 shows the EIS response for multiple electrodes in
accordance with embodiments of the invention.
[0105] FIG. 63 is a Nyquist plot showing the effect of Isig
calibration via an increase in glucose in accordance with
embodiments of the invention.
[0106] FIG. 64 shows the effect of oxygen (Vcntr) response on the
Nyquist plot, in accordance with embodiments of the invention.
[0107] FIG. 65 shows a shift in the Nyquist plot due to temperature
changes, in accordance with embodiments of the invention.
[0108] FIG. 66 shows the relationship between Isig and blood
glucose in accordance with embodiments of the invention.
[0109] FIGS. 67A-67B show sensor drift in accordance with
embodiments of the invention.
[0110] FIG. 68 shows an increase in membrane resistance during
sensitivity loss, in accordance with embodiments of the
invention.
[0111] FIG. 69 shows a drop in Warburg Admittance during
sensitivity loss, in accordance with embodiments of the
invention.
[0112] FIG. 70 shows calibration curves in accordance with
embodiments of the invention.
[0113] FIG. 71 shows a higher-frequency semicircle becoming visible
on a Nyquist plot in accordance with embodiments of the
invention.
[0114] FIGS. 72A and 72B show Vcntr rail and Cdl decrease in
accordance with embodiments of the invention.
[0115] FIG. 73 shows the changing slope of calibration curves in
accordance with embodiments of the invention
[0116] FIG. 74 shows the changing length of the Nyquist plot in
accordance with embodiments of the invention.
[0117] FIG. 75 shows enlarged views of the lower-frequency and the
higher-frequency regions of the Nyquist plot of FIG. 74.
[0118] FIGS. 76A and 76B show the combined effect of increase in
membrane resistance, decrease in Cdl, and Vcntr rail in accordance
with embodiments of the invention.
[0119] FIG. 77 shows relative Cdl values for two working electrodes
in accordance with embodiments of the invention.
[0120] FIG. 78 shows relative Rp values for two working electrodes
in accordance with embodiments of the invention.
[0121] FIG. 79 shows the combined effect of changing EIS parameters
on calibration curves in accordance with embodiments of the
invention.
[0122] FIG. 80 shows that, in accordance with embodiments of the
invention, the length of the Nyquist plot in the lower-frequency
region is longer where there is sensitivity loss.
[0123] FIG. 81 is a flow diagram for sensor self-calibration based
on the detection of sensitivity change in accordance with
embodiments of the invention.
[0124] FIG. 82 illustrates a horizontal shift in Nyquist plot as a
result of sensitivity loss, in accordance with embodiments of the
invention.
[0125] FIG. 83 shows a method of developing a heuristic EIS metric
based on a Nyquist plot in accordance with embodiments of the
invention.
[0126] FIG. 84 shows the relationship between Rm and Calibration
Factor in accordance with embodiments of the invention.
[0127] FIG. 85 shows the relationship between Rm and normalized
Isig in accordance with embodiments of the invention.
[0128] FIG. 86 shows Isig plots for various glucose levels as a
function of time, in accordance with embodiments of the
invention.
[0129] FIG. 87 shows Cdl plots for various glucose levels as a
function of time, in accordance with embodiments of the
invention.
[0130] FIG. 88 shows a second inflection point for the plots of
FIG. 86, in accordance with embodiments of the invention.
[0131] FIG. 89 shows a second inflection point for Rm corresponding
to the peak in FIG. 88, in accordance with embodiments of the
invention.
[0132] FIG. 90 shows one illustration of the relationship between
Calibration Factor (CF) and Rmem+Rsol in accordance with
embodiments of the invention.
[0133] FIG. 91A is a chart showing in-vivo results for MARD over
all valid BGs in approximately the first 8 hours of sensor life, in
accordance with embodiments of the invention.
[0134] FIG. 91B is a chart showing median ARD numbers over all
valid BGs in approximately the first 8 hours of sensor life, in
accordance with embodiments of the invention.
[0135] FIGS. 92A-92C show Calibration Factor adjustment in
accordance with embodiments of the invention.
[0136] FIGS. 93A-93C show Calibration Factor adjustment in
accordance with embodiments of the invention.
[0137] FIGS. 94A-94C show Calibration Factor adjustment in
accordance with embodiments of the invention.
[0138] FIG. 95 shows an illustrative example of initial decay in
Cdl in accordance with embodiments of the invention.
[0139] FIG. 96 shows the effects on Isig of removal of the
non-Faradaic current, in accordance with embodiments of the
invention.
[0140] FIG. 97A shows the Calibration Factor before removal of the
non-Faradaic current for two working electrodes, in accordance with
embodiments of the invention.
[0141] FIG. 97B shows the Calibration Factor after removal of the
non-Faradaic current for two working electrodes, in accordance with
embodiments of the invention.
[0142] FIGS. 98A and 98B show the effect on MARD of the removal of
the non-Faradaic current, in accordance with embodiments of the
invention.
[0143] FIG. 99 is an illustration of double layer capacitance over
time, in accordance with embodiments of the invention.
[0144] FIG. 100 shows a shift in Rmem+Rsol and the appearance of
the higher-frequency semicircle during sensitivity loss, in
accordance with embodiments of the invention.
[0145] FIG. 101A shows a flow diagram for detection of sensitivity
loss using combinatory logic, in accordance with an embodiment of
the invention.
[0146] FIG. 101B shows a flow diagram for detection of sensitivity
loss using combinatory logic, in accordance with another embodiment
of the invention.
[0147] FIG. 102 shows an illustrative method for using Nyquist
slope as a marker to differentiate between new and used sensors, in
accordance with embodiments of the invention.
[0148] FIGS. 103A-103C show an illustrative example of Nyquist
plots having different lengths for different sensor configurations,
in accordance with embodiments of the invention.
[0149] FIG. 104 shows Nyquist plot length as a function of time,
for the sensors of FIGS. 103A-103C.
[0150] FIG. 105 shows a flow diagram for blanking sensor data or
terminating a sensor in accordance with an embodiment of the
invention.
[0151] FIG. 106 shows a flow diagram for sensor termination in
accordance with an embodiment of the invention.
[0152] FIG. 107 shows a flow diagram for signal dip detection in
accordance with an embodiment of the invention.
[0153] FIG. 108A shows Isig and Vcntr as a function of time, and
FIG. 108B shows glucose as a function of time, in accordance with
an embodiment of the invention.
[0154] FIG. 109A calibration ratio as a function of time, and FIG.
109B show glucose as a function of time, in accordance with an
embodiment of the invention.
[0155] FIGS. 110A and 110B show calibration factor trends as a
function of time in accordance with embodiments of the
invention.
[0156] FIG. 111 shows a flow diagram for First Day Calibration
(FDC) in accordance with an embodiment of the invention.
[0157] FIG. 112 shows a flow diagram for EIS-based calibration in
accordance with an embodiment of the invention.
DETAILED DESCRIPTION
[0158] In the following description, reference is made to the
accompanying drawings which form a part hereof and which illustrate
several embodiments of the present inventions. It is understood
that other embodiments may be utilized and structural and
operational changes may be made without departing from the scope of
the present inventions.
[0159] The inventions herein are described below with reference to
flowchart illustrations of methods, systems, devices, apparatus,
and programming and computer program products. It will be
understood that each block of the flowchart illustrations, and
combinations of blocks in the flowchart illustrations, can be
implemented by programming instructions, including computer program
instructions (as can any menu screens described in the figures).
These computer program instructions may be loaded onto a computer
or other programmable data processing apparatus (such as a
controller, microcontroller, or processor in a sensor electronics
device) to produce a machine, such that the instructions which
execute on the computer or other programmable data processing
apparatus create instructions for implementing the functions
specified in the flowchart block or blocks. These computer program
instructions may also be stored in a computer-readable memory that
can direct a computer or other programmable data processing
apparatus to function in a particular manner, such that the
instructions stored in the computer-readable memory produce an
article of manufacture including instructions which implement the
function specified in the flowchart block or blocks. The computer
program instructions may also be loaded onto a computer or other
programmable data processing apparatus to cause a series of
operational steps to be performed on the computer or other
programmable apparatus to produce a computer implemented process
such that the instructions which execute on the computer or other
programmable apparatus provide steps for implementing the functions
specified in the flowchart block or blocks, and/or menus presented
herein. Programming instructions may also be stored in and/or
implemented via electronic circuitry, including integrated circuits
(ICs) and Application Specific Integrated Circuits (ASICs) used in
conjunction with sensor devices, apparatuses, and systems.
[0160] FIG. 1 is a perspective view of a subcutaneous sensor
insertion set and a block diagram of a sensor electronics device
according to an embodiment of the invention. As illustrated in FIG.
1, a subcutaneous sensor set 10 is provided for subcutaneous
placement of an active portion of a flexible sensor 12 (see, e.g.,
FIG. 2), or the like, at a selected site in the body of a user. The
subcutaneous or percutaneous portion of the sensor set 10 includes
a hollow, slotted insertion needle 14, and a cannula 16. The needle
14 is used to facilitate quick and easy subcutaneous placement of
the cannula 16 at the subcutaneous insertion site. Inside the
cannula 16 is a sensing portion 18 of the sensor 12 to expose one
or more sensor electrodes 20 to the user's bodily fluids through a
window 22 formed in the cannula 16. In an embodiment of the
invention, the one or more sensor electrodes 20 may include a
counter electrode, a reference electrode, and one or more working
electrodes. After insertion, the insertion needle 14 is withdrawn
to leave the cannula 16 with the sensing portion 18 and the sensor
electrodes 20 in place at the selected insertion site.
[0161] In particular embodiments, the subcutaneous sensor set 10
facilitates accurate placement of a flexible thin film
electrochemical sensor 12 of the type used for monitoring specific
blood parameters representative of a user's condition. The sensor
12 monitors glucose levels in the body, and may be used in
conjunction with automated or semi-automated medication infusion
pumps of the external or implantable type as described, e.g., in
U.S. Pat. Nos. 4,562,751; 4,678,408; 4,685,903 or 4,573,994, to
control delivery of insulin to a diabetic patient.
[0162] Particular embodiments of the flexible electrochemical
sensor 12 are constructed in accordance with thin film mask
techniques to include elongated thin film conductors embedded or
encased between layers of a selected insulative material such as
polyimide film or sheet, and membranes. The sensor electrodes 20 at
a tip end of the sensing portion 18 are exposed through one of the
insulative layers for direct contact with patient blood or other
body fluids, when the sensing portion 18 (or active portion) of the
sensor 12 is subcutaneously placed at an insertion site. The
sensing portion 18 is joined to a connection portion 24 that
terminates in conductive contact pads, or the like, which are also
exposed through one of the insulative layers. In alternative
embodiments, other types of implantable sensors, such as chemical
based, optical based, or the like, may be used.
[0163] As is known in the art, the connection portion 24 and the
contact pads are generally adapted for a direct wired electrical
connection to a suitable monitor or sensor electronics device 100
for monitoring a user's condition in response to signals derived
from the sensor electrodes 20. Further description of flexible thin
film sensors of this general type are be found in U.S. Pat. No.
5,391,250, entitled METHOD OF FABRICATING THIN FILM SENSORS, which
is herein incorporated by reference. The connection portion 24 may
be conveniently connected electrically to the monitor or sensor
electronics device 100 or by a connector block 28 (or the like) as
shown and described in U.S. Pat. No. 5,482,473, entitled FLEX
CIRCUIT CONNECTOR, which is also herein incorporated by reference.
Thus, in accordance with embodiments of the present invention,
subcutaneous sensor sets 10 may be configured or formed to work
with either a wired or a wireless characteristic monitor
system.
[0164] The sensor electrodes 20 may be used in a variety of sensing
applications and may be configured in a variety of ways. For
example, the sensor electrodes 20 may be used in physiological
parameter sensing applications in which some type of biomolecule is
used as a catalytic agent. For example, the sensor electrodes 20
may be used in a glucose and oxygen sensor having a glucose oxidase
(GOx) enzyme catalyzing a reaction with the sensor electrodes 20.
The sensor electrodes 20, along with a biomolecule or some other
catalytic agent, may be placed in a human body in a vascular or
non-vascular environment. For example, the sensor electrodes 20 and
biomolecule may be placed in a vein and be subjected to a blood
stream, or may be placed in a subcutaneous or peritoneal region of
the human body.
[0165] The monitor 100 may also be referred to as a sensor
electronics device 100. The monitor 100 may include a power source
110, a sensor interface 122, processing electronics 124, and data
formatting electronics 128. The monitor 100 may be coupled to the
sensor set 10 by a cable 102 through a connector that is
electrically coupled to the connector block 28 of the connection
portion 24. In an alternative embodiment, the cable may be omitted.
In this embodiment of the invention, the monitor 100 may include an
appropriate connector for direct connection to the connection
portion 104 of the sensor set 10. The sensor set 10 may be modified
to have the connector portion 104 positioned at a different
location, e.g., on top of the sensor set to facilitate placement of
the monitor 100 over the sensor set.
[0166] In embodiments of the invention, the sensor interface 122,
the processing electronics 124, and the data formatting electronics
128 are formed as separate semiconductor chips, however,
alternative embodiments may combine the various semiconductor chips
into a single or multiple customized semiconductor chips. The
sensor interface 122 connects with the cable 102 that is connected
with the sensor set 10.
[0167] The power source 110 may be a battery. The battery can
include three series silver oxide 357 battery cells. In alternative
embodiments, different battery chemistries may be utilized, such as
lithium based chemistries, alkaline batteries, nickel metalhydride,
or the like, and a different number of batteries may be used. The
monitor 100 provides power to the sensor set via the power source
110, through the cable 102 and cable connector 104. In an
embodiment of the invention, the power is a voltage provided to the
sensor set 10. In an embodiment of the invention, the power is a
current provided to the sensor set 10. In an embodiment of the
invention, the power is a voltage provided at a specific voltage to
the sensor set 10.
[0168] FIGS. 2A and 2B illustrate an implantable sensor and
electronics for driving the implantable sensor according to an
embodiment of the present invention. FIG. 2A shows a substrate 220
having two sides, a first side 222 of which contains an electrode
configuration and a second side 224 of which contains electronic
circuitry. As may be seen in FIG. 2A, a first side 222 of the
substrate comprises two counter electrode-working electrode pairs
240, 242, 244, 246 on opposite sides of a reference electrode 248.
A second side 224 of the substrate comprises electronic circuitry.
As shown, the electronic circuitry may be enclosed in a
hermetically sealed casing 226, providing a protective housing for
the electronic circuitry. This allows the sensor substrate 220 to
be inserted into a vascular environment or other environment which
may subject the electronic circuitry to fluids. By sealing the
electronic circuitry in a hermetically sealed casing 226, the
electronic circuitry may operate without risk of short circuiting
by the surrounding fluids. Also shown in FIG. 2A are pads 228 to
which the input and output lines of the electronic circuitry may be
connected. The electronic circuitry itself may be fabricated in a
variety of ways. According to an embodiment of the present
invention, the electronic circuitry may be fabricated as an
integrated circuit using techniques common in the industry.
[0169] FIG. 2B illustrates a general block diagram of an electronic
circuit for sensing an output of a sensor according to an
embodiment of the present invention. At least one pair of sensor
electrodes 310 may interface to a data converter 312, the output of
which may interface to a counter 314. The counter 314 may be
controlled by control logic 316. The output of the counter 314 may
connect to a line interface 318. The line interface 318 may be
connected to input and output lines 320 and may also connect to the
control logic 316. The input and output lines 320 may also be
connected to a power rectifier 322.
[0170] The sensor electrodes 310 may be used in a variety of
sensing applications and may be configured in a variety of ways.
For example, the sensor electrodes 310 may be used in physiological
parameter sensing applications in which some type of biomolecule is
used as a catalytic agent. For example, the sensor electrodes 310
may be used in a glucose and oxygen sensor having a glucose oxidase
(GOx) enzyme catalyzing a reaction with the sensor electrodes 310.
The sensor electrodes 310, along with a biomolecule or some other
catalytic agent, may be placed in a human body in a vascular or
non-vascular environment. For example, the sensor electrodes 310
and biomolecule may be placed in a vein and be subjected to a blood
stream.
[0171] FIG. 3 illustrates a block diagram of a sensor electronics
device and a sensor including a plurality of electrodes according
to an embodiment of the invention. The sensor set or system 350
includes a sensor 355 and a sensor electronics device 360. The
sensor 355 includes a counter electrode 365, a reference electrode
370, and a working electrode 375. The sensor electronics device 360
includes a power supply 380, a regulator 385, a signal processor
390, a measurement processor 395, and a display/transmission module
397. The power supply 380 provides power (in the form of either a
voltage, a current, or a voltage including a current) to the
regulator 385. The regulator 385 transmits a regulated voltage to
the sensor 355. In an embodiment of the invention, the regulator
385 transmits a voltage to the counter electrode 365 of the sensor
355.
[0172] The sensor 355 creates a sensor signal indicative of a
concentration of a physiological characteristic being measured. For
example, the sensor signal may be indicative of a blood glucose
reading. In an embodiment of the invention utilizing subcutaneous
sensors, the sensor signal may represent a level of hydrogen
peroxide in a subject. In an embodiment of the invention where
blood or cranial sensors are utilized, the amount of oxygen is
being measured by the sensor and is represented by the sensor
signal. In an embodiment of the invention utilizing implantable or
long-term sensors, the sensor signal may represent a level of
oxygen in the subject. The sensor signal is measured at the working
electrode 375. In an embodiment of the invention, the sensor signal
may be a current measured at the working electrode. In an
embodiment of the invention, the sensor signal may be a voltage
measured at the working electrode.
[0173] The signal processor 390 receives the sensor signal (e.g., a
measured current or voltage) after the sensor signal is measured at
the sensor 355 (e.g., the working electrode). The signal processor
390 processes the sensor signal and generates a processed sensor
signal. The measurement processor 395 receives the processed sensor
signal and calibrates the processed sensor signal utilizing
reference values. In an embodiment of the invention, the reference
values are stored in a reference memory and provided to the
measurement processor 395. The measurement processor 395 generates
sensor measurements. The sensor measurements may be stored in a
measurement memory (not shown). The sensor measurements may be sent
to a display/transmission device to be either displayed on a
display in a housing with the sensor electronics or transmitted to
an external device.
[0174] The sensor electronics device 360 may be a monitor which
includes a display to display physiological characteristics
readings. The sensor electronics device 360 may also be installed
in a desktop computer, a pager, a television including
communications capabilities, a laptop computer, a server, a network
computer, a personal digital assistant (PDA), a portable telephone
including computer functions, an infusion pump including a display,
a glucose sensor including a display, and/or a combination infusion
pump/glucose sensor. The sensor electronics device 360 may be
housed in a blackberry, a network device, a home network device, or
an appliance connected to a home network.
[0175] FIG. 4 illustrates an alternative embodiment of the
invention including a sensor and a sensor electronics device
according to an embodiment of the invention. The sensor set or
sensor system 400 includes a sensor electronics device 360 and a
sensor 355. The sensor includes a counter electrode 365, a
reference electrode 370, and a working electrode 375. The sensor
electronics device 360 includes a microcontroller 410 and a
digital-to-analog converter (DAC) 420. The sensor electronics
device 360 may also include a current-to-frequency converter (I/F
converter) 430.
[0176] The microcontroller 410 includes software program code,
which when executed, or programmable logic which, causes the
microcontroller 410 to transmit a signal to the DAC 420, where the
signal is representative of a voltage level or value that is to be
applied to the sensor 355. The DAC 420 receives the signal and
generates the voltage value at the level instructed by the
microcontroller 410. In embodiments of the invention, the
microcontroller 410 may change the representation of the voltage
level in the signal frequently or infrequently. Illustratively, the
signal from the microcontroller 410 may instruct the DAC 420 to
apply a first voltage value for one second and a second voltage
value for two seconds.
[0177] The sensor 355 may receive the voltage level or value. In an
embodiment of the invention, the counter electrode 365 may receive
the output of an operational amplifier which has as inputs the
reference voltage and the voltage value from the DAC 420. The
application of the voltage level causes the sensor 355 to create a
sensor signal indicative of a concentration of a physiological
characteristic being measured. In an embodiment of the invention,
the microcontroller 410 may measure the sensor signal (e.g., a
current value) from the working electrode. Illustratively, a sensor
signal measurement circuit 431 may measure the sensor signal. In an
embodiment of the invention, the sensor signal measurement circuit
431 may include a resistor and the current may be passed through
the resistor to measure the value of the sensor signal. In an
embodiment of the invention, the sensor signal may be a current
level signal and the sensor signal measurement circuit 431 may be a
current-to-frequency (I/F) converter 430. The current-to-frequency
converter 430 may measure the sensor signal in terms of a current
reading, convert it to a frequency-based sensor signal, and
transmit the frequency-based sensor signal to the microcontroller
410. In embodiments of the invention, the microcontroller 410 may
be able to receive frequency-based sensor signals easier than
non-frequency-based sensor signals. The microcontroller 410
receives the sensor signal, whether frequency-based or non
frequency-based, and determines a value for the physiological
characteristic of a subject, such as a blood glucose level. The
microcontroller 410 may include program code, which when executed
or run, is able to receive the sensor signal and convert the sensor
signal to a physiological characteristic value. In an embodiment of
the invention, the microcontroller 410 may convert the sensor
signal to a blood glucose level. In an embodiment of the invention,
the microcontroller 410 may utilize measurements stored within an
internal memory in order to determine the blood glucose level of
the subject. In an embodiment of the invention, the microcontroller
410 may utilize measurements stored within a memory external to the
microcontroller 410 to assist in determining the blood glucose
level of the subject.
[0178] After the physiological characteristic value is determined
by the microcontroller 410, the microcontroller 410 may store
measurements of the physiological characteristic values for a
number of time periods. For example, a blood glucose value may be
sent to the microcontroller 410 from the sensor every second or
five seconds, and the microcontroller may save sensor measurements
for five minutes or ten minutes of BG readings. The microcontroller
410 may transfer the measurements of the physiological
characteristic values to a display on the sensor electronics device
360. For example, the sensor electronics device 360 may be a
monitor which includes a display that provides a blood glucose
reading for a subject. In an embodiment of the invention, the
microcontroller 410 may transfer the measurements of the
physiological characteristic values to an output interface of the
microcontroller 410. The output interface of the microcontroller
410 may transfer the measurements of the physiological
characteristic values, e.g., blood glucose values, to an external
device, e.g., an infusion pump, a combined infusion pump/glucose
meter, a computer, a personal digital assistant, a pager, a network
appliance, a server, a cellular phone, or any computing device.
[0179] FIG. 5 illustrates an electronic block diagram of the sensor
electrodes and a voltage being applied to the sensor electrodes
according to an embodiment of the present invention. In the
embodiment of the invention illustrated in FIG. 5, an op amp 530 or
other servo controlled device may connect to sensor electrodes 510
through a circuit/electrode interface 538. The op amp 530,
utilizing feedback through the sensor electrodes, attempts to
maintain a prescribed voltage (what the DAC may desire the applied
voltage to be) between a reference electrode 532 and a working
electrode 534 by adjusting the voltage at a counter electrode 536.
Current may then flow from a counter electrode 536 to a working
electrode 534. Such current may be measured to ascertain the
electrochemical reaction between the sensor electrodes 510 and the
biomolecule of a sensor that has been placed in the vicinity of the
sensor electrodes 510 and used as a catalyzing agent. The circuitry
disclosed in FIG. 5 may be utilized in a long-term or implantable
sensor or may be utilized in a short-term or subcutaneous
sensor.
[0180] In a long-term sensor embodiment, where a glucose oxidase
(GOx) enzyme is used as a catalytic agent in a sensor, current may
flow from the counter electrode 536 to a working electrode 534 only
if there is oxygen in the vicinity of the enzyme and the sensor
electrodes 510. Illustratively, if the voltage set at the reference
electrode 532 is maintained at about 0.5 volts, the amount of
current flowing from the counter electrode 536 to a working
electrode 534 has a fairly linear relationship with unity slope to
the amount of oxygen present in the area surrounding the enzyme and
the electrodes. Thus, increased accuracy in determining an amount
of oxygen in the blood may be achieved by maintaining the reference
electrode 532 at about 0.5 volts and utilizing this region of the
current-voltage curve for varying levels of blood oxygen. Different
embodiments of the present invention may utilize different sensors
having biomolecules other than a glucose oxidase enzyme and may,
therefore, have voltages other than 0.5 volts set at the reference
electrode.
[0181] As discussed above, during initial implantation or insertion
of the sensor 510, the sensor 510 may provide inaccurate readings
due to the adjusting of the subject to the sensor and also
electrochemical byproducts caused by the catalyst utilized in the
sensor. A stabilization period is needed for many sensors in order
for the sensor 510 to provide accurate readings of the
physiological parameter of the subject. During the stabilization
period, the sensor 510 does not provide accurate blood glucose
measurements. Users and manufacturers of the sensors may desire to
improve the stabilization timeframe for the sensor so that the
sensors can be utilized quickly after insertion into the subject's
body or a subcutaneous layer of the subject.
[0182] In previous sensor electrode systems, the stabilization
period or timeframe was one hour to three hours. In order to
decrease the stabilization period or timeframe and increase the
timeliness of accuracy of the sensor, a sensor (or electrodes of a
sensor) may be subjected to a number of pulses rather than the
application of one pulse followed by the application of another
voltage. FIG. 6A illustrates a method of applying pulses during a
stabilization timeframe in order to reduce the stabilization
timeframe according to an embodiment of the present invention. In
this embodiment of the invention, a voltage application device
applies 600 a first voltage to an electrode for a first time or
time period. In an embodiment of the invention, the first voltage
may be a DC constant voltage. This results in an anodic current
being generated. In an alternative embodiment of the invention, a
digital-to-analog converter or another voltage source may supply
the voltage to the electrode for a first time period. The anodic
current means that electrons are being driven towards the electrode
to which the voltage is applied. In an embodiment of the invention,
an application device may apply a current instead of a voltage. In
an embodiment of the invention where a voltage is applied to a
sensor, after the application of the first voltage to the
electrode, the voltage regulator may wait (i.e., not apply a
voltage) for a second time, timeframe, or time period 605. In other
words, the voltage application device waits until a second time
period elapses. The non-application of voltage results in a
cathodic current, which results in the gaining of electrons by the
electrode to which the voltage is not applied. The application of
the first voltage to the electrode for a first time period followed
by the non-application of voltage for a second time period is
repeated 610 for a number of iterations. This may be referred to as
an anodic and cathodic cycle. In an embodiment of the invention,
the number of total iterations of the stabilization method is
three, i.e., three applications of the voltage for the first time
period, each followed by no application of the voltage for the
second time period. In an embodiment of the invention, the first
voltage may be 1.07 volts. In an embodiment of the invention, the
first voltage may be 0.535 volts. In an embodiment of the
invention, the first voltage may be approximately 0.7 volts.
[0183] The repeated application of the voltage and the
non-application of the voltage results in the sensor (and thus the
electrodes) being subjected to an anodic-cathodic cycle. The
anodic-cathodic cycle results in the reduction of electrochemical
byproducts which are generated by a patient's body reacting to the
insertion of the sensor or the implanting of the sensor. In an
embodiment of the invention, the electrochemical byproducts cause
generation of a background current, which results in inaccurate
measurements of the physiological parameter of the subject. In an
embodiment of the invention, the electrochemical byproduct may be
eliminated. Under other operating conditions, the electrochemical
byproducts may be reduced or significantly reduced. A successful
stabilization method results in the anodic-cathodic cycle reaching
equilibrium, electrochemical byproducts being significantly
reduced, and background current being minimized.
[0184] In an embodiment of the invention, the first voltage being
applied to the electrode of the sensor may be a positive voltage.
In an embodiment of the invention, the first voltage being applied
may be a negative voltage. In an embodiment of the invention, the
first voltage may be applied to a working electrode. In an
embodiment of the invention, the first voltage may be applied to
the counter electrode or the reference electrode.
[0185] In embodiments of the invention, the duration of the voltage
pulse and the non-application of voltage may be equal, e.g., such
as three minutes each. In embodiments of the invention, the
duration of the voltage application or voltage pulse may be
different values, e.g., the first time and the second time may be
different. In an embodiment of the invention, the first time period
may be five minutes and the waiting period may be two minutes. In
an embodiment of the invention, the first time period may be two
minutes and the waiting period (or second timeframe) may be five
minutes. In other words, the duration for the application of the
first voltage may be two minutes and there may be no voltage
applied for five minutes. This timeframe is only meant to be
illustrative and should not be limiting. For example, a first
timeframe may be two, three, five or ten minutes and the second
timeframe may be five minutes, ten minutes, twenty minutes, or the
like. The timeframes (e.g., the first time and the second time) may
depend on unique characteristics of different electrodes, the
sensors, and/or the patient's physiological characteristics.
[0186] In embodiments of the invention, more or less than three
pulses may be utilized to stabilize the glucose sensor. In other
words, the number of iterations may be greater than 3 or less than
three. For example, four voltage pulses (e.g., a high voltage
followed by no voltage) may be applied to one of the electrodes or
six voltage pulses may be applied to one of the electrodes.
[0187] Illustratively, three consecutive pulses of 1.07 volts
(followed by respective waiting periods) may be sufficient for a
sensor implanted subcutaneously. In an embodiment of the invention,
three consecutive voltage pulses of 0.7 volts may be utilized. The
three consecutive pulses may have a higher or lower voltage value,
either negative or positive, for a sensor implanted in blood or
cranial fluid, e.g., the long-term or permanent sensors. In
addition, more than three pulses (e.g., five, eight, twelve) may be
utilized to create the anodic-cathodic cycling between anodic and
cathodic currents in any of the subcutaneous, blood, or cranial
fluid sensors.
[0188] FIG. 6B illustrates a method of stabilizing sensors
according to an embodiment of the invention. In the embodiment of
the invention illustrated in FIG. 6B, a voltage application device
may apply 630 a first voltage to the sensor for a first time to
initiate an anodic cycle at an electrode of the sensor. The voltage
application device may be a DC power supply, a digital-to-analog
converter, or a voltage regulator. After the first time period has
elapsed, a second voltage is applied 635 to the sensor for a second
time to initiate a cathodic cycle at an electrode of the sensor.
Illustratively, rather than no voltage being applied, as is
illustrated in the method of FIG. 6A, a different voltage (from the
first voltage) is applied to the sensor during the second
timeframe. In an embodiment of the invention, the application of
the first voltage for the first time and the application of the
second voltage for the second time is repeated 640 for a number of
iterations. In an embodiment of the invention, the application of
the first voltage for the first time and the application of the
second voltage for the second time may each be applied for a
stabilization timeframe, e.g., 10 minutes, 15 minutes, or 20
minutes rather than for a number of iterations. This stabilization
timeframe is the entire timeframe for the stabilization sequence,
e.g., until the sensor (and electrodes) are stabilized. The benefit
of this stabilization methodology is a faster run-in of the
sensors, less background current (in other words a suppression of
some the background current), and a better glucose response.
[0189] In an embodiment of the invention, the first voltage may be
0.535 volts applied for five minutes, the second voltage may be
1.070 volts applied for two minutes, the first voltage of 0.535
volts may be applied for five minutes, the second voltage of 1.070
volts may be applied for two minutes, the first voltage of 0.535
volts may be applied for five minutes, and the second voltage of
1.070 volts may be applied for two minutes. In other words, in this
embodiment, there are three iterations of the voltage pulsing
scheme. The pulsing methodology may be changed in that the second
timeframe, e.g., the timeframe of the application of the second
voltage may be lengthened from two minutes to five minutes, ten
minutes, fifteen minutes, or twenty minutes. In addition, after the
three iterations are applied in this embodiment of the invention, a
nominal working voltage of 0.535 volts may be applied.
[0190] The 1.070 and 0.535 volts are illustrative values. Other
voltage values may be selected based on a variety of factors. These
factors may include the type of enzyme utilized in the sensor, the
membranes utilized in the sensor, the operating period of the
sensor, the length of the pulse, and/or the magnitude of the pulse.
Under certain operating conditions, the first voltage may be in a
range of 1.00 to 1.09 volts and the second voltage may be in a
range of 0.510 to 0.565 volts. In other operating embodiments, the
ranges that bracket the first voltage and the second voltage may
have a higher range, e.g., 0.3 volts, 0.6 volts, 0.9 volts,
depending on the voltage sensitivity of the electrode in the
sensor. Under other operating conditions, the voltage may be in a
range of 0.8 volts to 1.34 volts and the other voltage may be in a
range of 0.335 to 0.735. Under other operating conditions, the
range of the higher voltage may be smaller than the range of the
lower voltage. Illustratively, the higher voltage may be in a range
of 0.9 to 1.09 volts and the lower voltage may be in a range of
0.235 to 0.835 volts.
[0191] In an embodiment of the invention, the first voltage and the
second voltage may be positive voltages, or alternatively in other
embodiments of the invention, negative voltages. In an embodiment
of the invention, the first voltage may be positive and the second
voltage may be negative, or alternatively, the first voltage may be
negative and the second voltage may be positive. The first voltage
may be different voltage levels for each of the iterations. In an
embodiment of the invention, the first voltage may be a D.C.
constant voltage. In other embodiments of the invention, the first
voltage may be a ramp voltage, a sinusoid-shaped voltage, a stepped
voltage, or other commonly utilized voltage waveforms. In an
embodiment of the invention, the second voltage may be a D.C.
constant voltage, a ramp voltage, a sinusoid-shaped voltage, a
stepped voltage, or other commonly utilized voltage waveforms. In
an embodiment of the invention, the first voltage or the second
voltage may be an AC signal riding on a DC waveform. In an
embodiment of the invention, the first voltage may be one type of
voltage, e.g., a ramp voltage, and the second voltage may be a
second type of voltage, e.g., a sinusoid-shaped voltage. In an
embodiment of the invention, the first voltage (or the second
voltage) may have different waveform shapes for each of the
iterations. For example, if there are three cycles in a
stabilization method, in a first cycle, the first voltage may be a
ramp voltage, in the second cycle, the first voltage may be a
constant voltage, and in the third cycle, the first voltage may be
a sinusoidal voltage.
[0192] In an embodiment of the invention, a duration of the first
timeframe and a duration of the second timeframe may have the same
value, or alternatively, the duration of the first timeframe and
the second timeframe may have different values. For example, the
duration of the first timeframe may be two minutes and the duration
of the second timeframe may be five minutes and the number of
iterations may be three. As discussed above, the stabilization
method may include a number of iterations. In embodiments of the
invention, during different iterations of the stabilization method,
the duration of each of the first timeframes may change and the
duration of each of the second timeframes may change.
Illustratively, during the first iteration of the anodic-cathodic
cycling, the first timeframe may be 2 minutes and the second
timeframe may be 5 minutes. During the second iteration, the first
timeframe may be 1 minute and the second timeframe may be 3
minutes. During the third iteration, the first timeframe may be 3
minutes and the second timeframe may be 10 minutes.
[0193] In an embodiment of the invention, a first voltage of 0.535
volts is applied to an electrode in a sensor for two minutes to
initiate an anodic cycle, then a second voltage of 1.07 volts is
applied to the electrode for five minutes to initiate a cathodic
cycle. The first voltage of 0.535 volts is then applied again for
two minutes to initiate the anodic cycle and a second voltage of
1.07 volts is applied to the sensor for five minutes. In a third
iteration, 0.535 volts is applied for two minutes to initiate the
anodic cycle and then 1.07 volts is applied for five minutes. The
voltage applied to the sensor is then 0.535 during the actual
working timeframe of the sensor, e.g., when the sensor provides
readings of a physiological characteristic of a subject.
[0194] Shorter duration voltage pulses may be utilized in the
embodiment of FIGS. 6A and 6B. The shorter duration voltage pulses
may be utilized to apply the first voltage, the second voltage, or
both. In an embodiment of the present invention, the magnitude of
the shorter duration voltage pulse for the first voltage is -1.07
volts and the magnitude of the shorter duration voltage pulse for
the second voltage is approximately half of the high magnitude,
e.g., -0.535 volts. Alternatively, the magnitude of the shorter
duration pulse for the first voltage may be 0.535 volts and the
magnitude of the shorter duration pulse for the second voltage is
1.07 volts.
[0195] In embodiments of the invention utilizing short duration
pulses, the voltage may not be applied continuously for the entire
first time period. Instead, the voltage application device may
transmit a number of short duration pulses during the first time
period. In other words, a number of mini-width or short duration
voltage pulses may be applied to the electrodes of the sensor over
the first time period. Each mini-width or short duration pulse may
have a width of a number of milliseconds. Illustratively, this
pulse width may be 30 milliseconds, 50 milliseconds, 70
milliseconds or 200 milliseconds. These values are meant to be
illustrative and not limiting. In an embodiment of the invention,
such as the embodiment illustrated in FIG. 6A, these short duration
pulses are applied to the sensor (electrode) for the first time
period and then no voltage is applied for the second time
period.
[0196] In an embodiment of the invention, each short duration pulse
may have the same time duration within the first time period. For
example, each short duration voltage pulse may have a time width of
50 milliseconds and each pulse delay between the pulses may be 950
milliseconds. In this example, if two minutes is the measured time
for the first timeframe, then 120 short duration voltage pulses may
be applied to the sensor. In an embodiment of the invention, each
of the short duration voltage pulses may have different time
durations. In an embodiment of the invention, each of the short
duration voltage pulses may have the same amplitude values. In an
embodiment of the invention, each of the short duration voltage
pulses may have different amplitude values. By utilizing short
duration voltage pulses rather than a continuous application of
voltage to the sensor, the same anodic and cathodic cycling may
occur and the sensor (e.g., electrodes) is subjected to less total
energy or charge over time. The use of short duration voltage
pulses utilizes less power as compared to the application of
continuous voltage to the electrodes because there is less energy
applied to the sensors (and thus the electrodes).
[0197] FIG. 6C illustrates utilization of feedback in stabilizing
the sensor according to an embodiment of the present invention. The
sensor system may include a feedback mechanism to determine if
additional pulses are needed to stabilize a sensor. In an
embodiment of the invention, a sensor signal generated by an
electrode (e.g., a working electrode) may be analyzed to determine
if the sensor signal is stabilized. A first voltage is applied 630
to an electrode for a first timeframe to initiate an anodic cycle.
A second voltage is applied 635 to an electrode for a second
timeframe to initiate a cathodic cycle. In an embodiment of the
invention, an analyzation module may analyze a sensor signal (e.g.,
the current emitted by the sensor signal, a resistance at a
specific point in the sensor, an impedance at a specific node in
the sensor) and determine if a threshold measurement has been
reached 637 (e.g., determining if the sensor is providing accurate
readings by comparing against the threshold measurement). If the
sensor readings are determined to be accurate, which represents
that the electrode (and thus the sensor) is stabilized 642, no
additional application of the first voltage and/or the second
voltage may be generated. If stability was not achieved, in an
embodiment of the invention, then an additional anodic/cathodic
cycle is initiated by the application 630 of a first voltage to an
electrode for a first time period and then the application 635 of
the second voltage to the electrode for a second time period.
[0198] In embodiments of the invention, the analyzation module may
be employed after an anodic/cathodic cycle of three applications of
the first voltage and the second voltage to an electrode of the
sensor. In an embodiment of the invention, an analyzation module
may be employed after one application of the first voltage and the
second voltage, as is illustrated in FIG. 6C.
[0199] In an embodiment of the invention, the analyzation module
may be utilized to measure a voltage emitted after a current has
been introduced across an electrode or across two electrodes. The
analyzation module may monitor a voltage level at the electrode or
at the receiving level. In an embodiment of the invention, if the
voltage level is above a certain threshold, this may mean that the
sensor is stabilized. In an embodiment of the invention, if the
voltage level falls below a threshold level, this may indicate that
the sensor is stabilized and ready to provide readings. In an
embodiment of the invention, a current may be introduced to an
electrode or across a couple of electrodes. The analyzation module
may monitor a current level emitted from the electrode. In this
embodiment of the invention, the analyzation module may be able to
monitor the current if the current is different by an order of
magnitude from the sensor signal current. If the current is above
or below a current threshold, this may signify that the sensor is
stabilized.
[0200] In an embodiment of the invention, the analyzation module
may measure an impedance between two electrodes of the sensor. The
analyzation module may compare the impedance against a threshold or
target impedance value and if the measured impedance is lower than
the target or threshold impedance, the sensor (and hence the sensor
signal) may be stabilized. In an embodiment of the invention, the
analyzation module may measure a resistance between two electrodes
of the sensor. In this embodiment of the invention, if the
analyzation module compares the resistance against a threshold or
target resistance value and the measured resistance value is less
than the threshold or target resistance value, then the analyzation
module may determine that the sensor is stabilized and that the
sensor signal may be utilized.
[0201] FIG. 7 illustrates an effect of stabilizing a sensor
according to an embodiment of the invention. Line 705 represents
blood glucose sensor readings for a glucose sensor where a previous
single pulse stabilization method was utilized. Line 710 represents
blood glucose readings for a glucose sensor where three voltage
pulses are applied (e.g., 3 voltage pulses having a duration of 2
minutes each followed by 5 minutes of no voltage being applied).
The x-axis 715 represents an amount of time. The dots 720, 725,
730, and 735 represent measured glucose readings, taken utilizing a
finger stick and then input into a glucose meter. As illustrated by
the graph, the previous single pulse stabilization method took
approximately 1 hour and 30 minutes in order to stabilize to the
desired glucose reading, e.g., 100 units. In contrast, the three
pulse stabilization method took only approximately 15 minutes to
stabilize the glucose sensor and results in a drastically improved
stabilization timeframe.
[0202] FIG. 8A illustrates a block diagram of a sensor electronics
device and a sensor including a voltage generation device according
to an embodiment of the invention. The voltage generation or
application device 810 includes electronics, logic, or circuits
which generate voltage pulses. The sensor electronics device 360
may also include an input device 820 to receive reference values
and other useful data. In an embodiment of the invention, the
sensor electronics device may include a measurement memory 830 to
store sensor measurements. In this embodiment of the invention, the
power supply 380 may supply power to the sensor electronics device.
The power supply 380 may supply power to a regulator 385, which
supplies a regulated voltage to the voltage generation or
application device 810. The connection terminals 811 represent that
in the illustrated embodiment of the invention, the connection
terminal couples or connects the sensor 355 to the sensor
electronics device 360.
[0203] In an embodiment of the invention illustrated in FIG. 8A,
the voltage generation or application device 810 supplies a
voltage, e.g., the first voltage or the second voltage, to an input
terminal of an operational amplifier 840. The voltage generation or
application device 810 may also supply the voltage to a working
electrode 375 of the sensor 355. Another input terminal of the
operational amplifier 840 is coupled to the reference electrode 370
of the sensor. The application of the voltage from the voltage
generation or application device 810 to the operational amplifier
840 drives a voltage measured at the counter electrode 365 to be
close to or equal to the voltage applied at the working electrode
375. In an embodiment of the invention, the voltage generation or
application device 810 could be utilized to apply the desired
voltage between the counter electrode and the working electrode.
This may occur by the application of the fixed voltage to the
counter electrode directly.
[0204] In an embodiment of the invention as illustrated in FIGS. 6A
and 6B, the voltage generation device 810 generates a first voltage
that is to be applied to the sensor during a first timeframe. The
voltage generation device 810 transmits this first voltage to an op
amp 840 which drives the voltage at a counter electrode 365 of the
sensor 355 to the first voltage. In an embodiment of the invention,
the voltage generation device 810 also could transmit the first
voltage directly to the counter electrode 365 of the sensor 355. In
the embodiment of the invention illustrated in FIG. 6A, the voltage
generation device 810 then does not transmit the first voltage to
the sensor 355 for a second timeframe. In other words, the voltage
generation device 810 is turned off or switched off. The voltage
generation device 810 may be programmed to continue cycling between
applying the first voltage and not applying a voltage for either a
number of iterations or for a stabilization timeframe, e.g., for
twenty minutes. FIG. 8B illustrates a voltage generation device to
implement this embodiment of the invention. The voltage regulator
385 transfers the regulated voltage to the voltage generation
device 810. A control circuit 860 controls the closing and opening
of a switch 850. If the switch 850 is closed, the voltage is
applied. If the switch 850 is opened, the voltage is not applied.
The timer 865 provides a signal to the control circuit 860 to
instruct the control circuit 860 to turn on and off the switch 850.
The control circuit 860 includes logic which can instruct the
circuit to open and close the switch 850 a number of times (to
match the necessary iterations). In an embodiment of the invention,
the timer 865 may also transmit a stabilization signal to identify
that the stabilization sequence is completed, i.e., that a
stabilization timeframe has elapsed.
[0205] In an embodiment of the invention, the voltage generation
device generates a first voltage for a first timeframe and
generates a second voltage for a second timeframe. FIG. 8C
illustrates a voltage generation device to generate two voltage
values to implement this embodiment of the invention. In this
embodiment of the invention, a two position switch 870 is utilized.
Illustratively, if the first switch position 871 is turned on or
closed by the timer 865 instructing the control circuit 860, then
the voltage generation device 810 generates a first voltage for the
first timeframe. After the first voltage has been applied for the
first timeframe, the timer sends a signal to the control circuit
860 indicating the first timeframe has elapsed and the control
circuit 860 directs the switch 870 to move to the second position
872. When the switch 870 is at the second position 872, the
regulated voltage is directed to a voltage step-down or buck
converter 880 to reduce the regulated voltage to a lesser value.
The lesser value is then delivered to the op amp 840 for the second
timeframe. After the timer 865 has sent a signal to the control
circuit 860 that the second timeframe has elapsed, the control
circuit 860 moves the switch 870 back to the first position. This
continues until the desired number of iterations has been completed
or the stabilization timeframe has elapsed. In an embodiment of the
invention, after the sensor stabilization timeframe has elapsed,
the sensor transmits a sensor signal 350 to the signal processor
390.
[0206] FIG. 8D illustrates a voltage application device 810
utilized to perform more complex applications of voltage to the
sensor. The voltage application device 810 may include a control
device 860, a switch 890, a sinusoid voltage generation device 891,
a ramp voltage generation device 892, and a constant voltage
generation device 893. In other embodiments of the invention, the
voltage application may generate an AC wave on top of a DC signal
or other various voltage pulse waveforms. In the embodiment of the
invention illustrated in FIG. 8D, the control device 860 may cause
the switch to move to one of the three voltage generation systems
891 (sinusoid), 892 (ramp), 893 (constant DC). This results in each
of the voltage generation systems generating the identified voltage
waveform. Under certain operating conditions, e.g., where a
sinusoidal pulse is to be applied for three pulses, the control
device 860 may cause the switch 890 to connect the voltage from the
voltage regulator 385 to the sinusoid voltage generator 891 in
order for the voltage application device 810 to generate a
sinusoidal voltage. Under other operating conditions, e.g., when a
ramp voltage is applied to the sensor as the first voltage for a
first pulse of three pulses, a sinusoid voltage is applied to the
sensor as the first voltage for a second pulse of the three pulses,
and a constant DC voltage is applied to the sensor as the first
voltage for a third pulse of the three pulses, the control device
860 may cause the switch 890, during the first timeframes in the
anodic/cathodic cycles, to move between connecting the voltage from
the voltage generation or application device 810 to the ramp
voltage generation system 892, then to the sinusoidal voltage
generation system 891, and then to the constant DC voltage
generation system 893. In this embodiment of the invention, the
control device 860 may also be directing or controlling the switch
to connect certain ones of the voltage generation subsystems to the
voltage from the regulator 385 during the second timeframe, e.g.,
during application of the second voltage.
[0207] FIG. 9A illustrates a sensor electronics device including a
microcontroller for generating voltage pulses according to an
embodiment of the invention. The advanced sensor electronics device
may include a microcontroller 410 (see FIG. 4), a digital-to-analog
converter (DAC) 420, an op amp 840, and a sensor signal measurement
circuit 431. In an embodiment of the invention, the sensor signal
measurement circuit may be a current-to-frequency (I/F) converter
430. In the embodiment of the invention illustrated in FIG. 9A,
software or programmable logic in the microcontroller 410 provides
instructions to transmit signals to the DAC 420, which in turn
instructs the DAC 420 to output a specific voltage to the
operational amplifier 840. The microcontroller 410 may also be
instructed to output a specific voltage to the working electrode
375, as is illustrated by line 911 in FIG. 9A. As discussed above,
the application of the specific voltage to operational amplifier
840 and the working electrode 375 may drive the voltage measured at
the counter electrode to the specific voltage magnitude. In other
words, the microcontroller 410 outputs a signal which is indicative
of a voltage or a voltage waveform that is to be applied to the
sensor 355 (e.g., the operational amplifier 840 coupled to the
sensor 355). In an alternative embodiment of the invention, a fixed
voltage may be set by applying a voltage directly from the DAC 420
between the reference electrode and the working electrode 375. A
similar result may also be obtained by applying voltages to each of
the electrodes with the difference equal to the fixed voltage
applied between the reference and working electrode. In addition,
the fixed voltage may be set by applying a voltage between the
reference and the counter electrode. Under certain operating
conditions, the microcontroller 410 may generate a pulse of a
specific magnitude which the DAC 420 understands represents that a
voltage of a specific magnitude is to be applied to the sensor.
After a first timeframe, the microcontroller 410 (via the program
or programmable logic) outputs a second signal which either
instructs the DAC 420 to output no voltage (for a sensor
electronics device 360 operating according to the method described
in FIG. 6A) or to output a second voltage (for a sensor electronics
device 360 operating according to the method described in FIG. 6B).
The microcontroller 410, after the second timeframe has elapsed,
then repeats the cycle of sending the signal indicative of a first
voltage to be applied (for the first timeframe) and then sending
the signal to instruct no voltage is to be applied or that a second
voltage is to be applied (for the second timeframe).
[0208] Under other operating conditions, the microcontroller 410
may generate a signal to the DAC 420 which instructs the DAC to
output a ramp voltage. Under other operating conditions, the
microcontroller 410 may generate a signal to the DAC 420 which
instructs the DAC 420 to output a voltage simulating a sinusoidal
voltage. These signals could be incorporated into any of the
pulsing methodologies discussed above in the preceding paragraph or
earlier in the application. In an embodiment of the invention, the
microcontroller 410 may generate a sequence of instructions and/or
pulses, which the DAC 420 receives and understands to mean that a
certain sequence of pulses is to be applied. For example, the
microcontroller 410 may transmit a sequence of instructions (via
signals and/or pulses) that instruct the DAC 420 to generate a
constant voltage for a first iteration of a first timeframe, a ramp
voltage for a first iteration of a second timeframe, a sinusoidal
voltage for a second iteration of a first timeframe, and a
squarewave having two values for a second iteration of the second
timeframe.
[0209] The microcontroller 410 may include programmable logic or a
program to continue this cycling for a stabilization timeframe or
for a number of iterations. Illustratively, the microcontroller 410
may include counting logic to identify when the first timeframe or
the second timeframe has elapsed. Additionally, the microcontroller
410 may include counting logic to identify that a stabilization
timeframe has elapsed. After any of the preceding timeframes have
elapsed, the counting logic may instruct the microcontroller to
either send a new signal or to stop transmission of a signal to the
DAC 420.
[0210] The use of the microcontroller 410 allows a variety of
voltage magnitudes to be applied in a number of sequences for a
number of time durations. In an embodiment of the invention, the
microcontroller 410 may include control logic or a program to
instruct the digital-to-analog converter 420 to transmit a voltage
pulse having a magnitude of approximately 1.0 volt for a first time
period of 1 minute, to then transmit a voltage pulse having a
magnitude of approximately 0.5 volts for a second time period of 4
minutes, and to repeat this cycle for four iterations. In an
embodiment of the invention, the microcontroller 420 may be
programmed to transmit a signal to cause the DAC 420 to apply the
same magnitude voltage pulse for each first voltage in each of the
iterations. In an embodiment of the invention, the microcontroller
410 may be programmed to transmit a signal to cause the DAC to
apply a different magnitude voltage pulse for each first voltage in
each of the iterations. In this embodiment of the invention, the
microcontroller 410 may also be programmed to transmit a signal to
cause the DAC 420 to apply a different magnitude voltage pulse for
each second voltage in each of the iterations. Illustratively, the
microcontroller 410 may be programmed to transmit a signal to cause
the DAC 420 to apply a first voltage pulse of approximately 1.0
volt in the first iteration, to apply a second voltage pulse of
approximately 0.5 volts in the first iteration, to apply a first
voltage of 0.7 volts and a second voltage of 0.4 volts in the
second iteration, and to apply a first voltage of 1.2 volts and a
second voltage of 0.8 volts in the third iteration.
[0211] The microcontroller 410 may also be programmed to instruct
the DAC 420 to provide a number of short duration voltage pulses
for a first timeframe. In this embodiment of the invention, rather
than one voltage being applied for the entire first timeframe
(e.g., two minutes), a number of shorter duration pulses may be
applied to the sensor. In this embodiment, the microcontroller 410
may also be programmed to instruct the DAC 420 to provide a number
of short duration voltage pulses for the second timeframe to the
sensor. Illustratively, the microcontroller 410 may send a signal
to cause the DAC to apply a number of short duration voltage pulses
where the short duration is 50 milliseconds or 100 milliseconds. In
between these short duration pulses the DAC may apply no voltage or
the DAC may apply a minimal voltage. The microcontroller may cause
the DAC 420 to apply the short duration voltage pulses for the
first timeframe, e.g., two minutes. The microcontroller 410 may
then send a signal to cause the DAC to either not apply any voltage
or to apply the short duration voltage pulses at a magnitude of a
second voltage for a second timeframe to the sensor, e.g., the
second voltage may be 0.75 volts and the second timeframe may be 5
minutes. In an embodiment of the invention, the microcontroller 410
may send a signal to the DAC 420 to cause the DAC 420 to apply a
different magnitude voltage for each of the short duration pulses
in the first timeframe and/or in the second timeframe. In an
embodiment of the invention, the microcontroller 410 may send a
signal to the DAC 420 to cause the DAC 420 to apply a pattern of
voltage magnitudes to the short durations voltage pulses for the
first timeframe or the second timeframe. For example, the
microcontroller may transmit a signal or pulses instructing the DAC
420 to apply thirty 20-millisecond pulses to the sensor during the
first timeframe. Each of the thirty 20-millisecond pulses may have
the same magnitude or may have a different magnitude. In this
embodiment of the invention, the microcontroller 410 may instruct
the DAC 420 to apply short duration pulses during the second
timeframe or may instruct the DAC 420 to apply another voltage
waveform during the second timeframe.
[0212] Although the disclosures in FIGS. 6-8 disclose the
application of a voltage, a current may also be applied to the
sensor to initiate the stabilization process. Illustratively, in
the embodiment of the invention illustrated in FIG. 6B, a first
current may be applied during a first timeframe to initiate an
anodic or cathodic response and a second current may be applied
during a second timeframe to initiate the opposite anodic or
cathodic response. The application of the first current and the
second current may continue for a number of iterations or may
continue for a stabilization timeframe. In an embodiment of the
invention, a first current may be applied during a first timeframe
and a first voltage may be applied during a second timeframe. In
other words, one of the anodic or cathodic cycles may be triggered
by a current being applied to the sensor and the other of the
anodic or cathodic cycles may be triggered by a voltage being
applied to the sensor. As described above, a current applied may be
a constant current, a ramp current, a stepped pulse current, or a
sinusoidal current. Under certain operating conditions, the current
may be applied as a sequence of short duration pulses during the
first timeframe.
[0213] FIG. 9B illustrates a sensor and sensor electronics
utilizing an analyzation module for feedback in a stabilization
period according to an embodiment of the present invention. FIG. 9B
introduces an analyzation module 950 to the sensor electronics
device 360. The analyzation module 950 utilizes feedback from the
sensor to determine whether or not the sensor is stabilized. In an
embodiment of the invention, the microcontroller 410 may include
instructions or commands to control the DAC 420 so that the DAC 420
applies a voltage or current to a part of the sensor 355. FIG. 9B
illustrates that a voltage or current could be applied between a
reference electrode 370 and a working electrode 375. However, the
voltage or current can be applied in between electrodes or directly
to one of the electrodes and the invention should not be limited by
the embodiment illustrated in FIG. 9B. The application of the
voltage or current is illustrated by dotted line 955. The
analyzation module 950 may measure a voltage, a current, a
resistance, or an impedance in the sensor 355. FIG. 9B illustrates
that the measurement occurs at the working electrode 375, but this
should not limit the invention because other embodiments of the
invention may measure a voltage, a current, a resistance, or an
impedance in between electrodes of the sensor or directly at either
the reference electrode 370 or the counter electrode 365. The
analyzation module 950 may receive the measured voltage, current,
resistance, or impedance and may compare the measurement to a
stored value (e.g., a threshold value). Dotted line 956 represents
the analyzation module 950 reading or taking a measurement of the
voltage, current, resistance, or impedance. Under certain operating
conditions, if the measured voltage, current, resistance, or
impedance is above the threshold, the sensor is stabilized and the
sensor signal is providing accurate readings of a physiological
condition of a patient. Under other operating conditions, if the
measured voltage, current, resistance, or impedance is below the
threshold, the sensor is stabilized. Under other operating
conditions, the analyzation module 950 may verify that the measured
voltage, current, resistance, or impedance is stable for a specific
timeframe, e.g., one minute or two minutes. This may represent that
the sensor 355 is stabilized and that the sensor signal is
transmitting accurate measurements of a subject's physiological
parameter, e.g., blood glucose level. After the analyzation module
950 has determined that the sensor is stabilized and the sensor
signal is providing accurate measurements, the analyzation module
950 may transmit a signal (e.g., a sensor stabilization signal) to
the microcontroller 410 indicating that the sensor is stabilized
and that the microcontroller 410 can start using or receiving the
sensor signal from the sensor 355. This is represented by dotted
line 957.
[0214] FIG. 10 illustrates a block diagram of a sensor system
including hydration electronics according to an embodiment of the
invention. The sensor system includes a connector 1010, a sensor
1012, and a monitor or sensor electronics device 1025. The sensor
1012 includes electrodes 1020 and a connection portion 1024. In an
embodiment of the invention, the sensor 1012 may be connected to
the sensor electronics device 1025 via a connector 1010 and a
cable. In other embodiments of the invention, the sensor 1012 may
be directly connected to the sensor electronics device 1025. In
other embodiments of the invention, the sensor 1012 may be
incorporated into the same physical device as the sensor
electronics device 1025. The monitor or sensor electronics device
1025 may include a power supply 1030, a regulator 1035, a signal
processor 1040, a measurement processor 1045, and a processor 1050.
The monitor or sensor electronics device 1025 may also include a
hydration detection circuit 1060. The hydration detection circuit
1060 interfaces with the sensor 1012 to determine if the electrodes
1020 of the sensor 1012 are sufficiently hydrated. If the
electrodes 1020 are not sufficiently hydrated, the electrodes 1020
do not provide accurate glucose readings, so it is important to
know when the electrodes 1020 are sufficiently hydrated. Once the
electrodes 1020 are sufficiently hydrated, accurate glucose
readings may be obtained.
[0215] In an embodiment of the invention illustrated in FIG. 10,
the hydration detection circuit 1060 may include a delay or timer
module 1065 and a connection detection module 1070. In an
embodiment of the invention utilizing the short term sensor or the
subcutaneous sensor, after the sensor 1012 has been inserted into
the subcutaneous tissue, the sensor electronics device or monitor
1025 is connected to the sensor 1012. The connection detection
module 1070 identifies that the sensors electronics device 1025 has
been connected to the sensor 1012 and sends a signal to the timer
module 1065. This is illustrated in FIG. 10 by the arrow 1084 which
represents a detector 1083 detecting a connection and sending a
signal to the connection detection module 1070 indicating the
sensor 1012 has been connected to the sensor electronics device
1025. In an embodiment of the invention where implantable or
long-term sensors are utilized, a connection detection module 1070
identifies that the implantable sensor has been inserted into the
body. The timer module 1065 receives the connection signal and
waits a set or established hydration time. Illustratively, the
hydration time may be two minutes, five minutes, ten minutes, or 20
minutes. These examples are meant to be illustrative and not to be
limiting. The timeframe does not have to be a set number of minutes
and can include any number of seconds. In an embodiment of the
invention, after the timer module 1065 has waited for the set
hydration time, the timer module 1065 may notify the processor 1050
that the sensor 1012 is hydrated by sending a hydration signal,
which is illustrated by line 1086.
[0216] In this embodiment of the invention, the processor 1050 may
receive the hydration signal and only start utilizing the sensor
signal (e.g., sensor measurements) after the hydration signal has
been received. In another embodiment of the invention, the
hydration detection circuit 1060 may be coupled between the sensor
(the sensor electrodes 1020) and the signal processor 1040. In this
embodiment of the invention, the hydration detection circuit 1060
may prevent the sensor signal from being sent to signal processor
1040 until the timer module 1065 has notified the hydration
detection circuit 1060 that the set hydration time has elapsed.
This is illustrated by the dotted lines labeled with reference
numerals 1080 and 1081. Illustratively, the timer module 1065 may
transmit a connection signal to a switch (or transistor) to turn on
the switch and let the sensor signal proceed to the signal
processor 1040. In an alternative embodiment of the invention, the
timer module 1065 may transmit a connection signal to turn on a
switch 1088 (or close the switch 1088) in the hydration detection
circuit 1060 to allow a voltage from the regulator 1035 to be
applied to the sensor 1012 after the hydration time has elapsed. In
other words, in this embodiment of the invention, the voltage from
the regulator 1035 is not applied to the sensor 1012 until after
the hydration time has elapsed.
[0217] FIG. 11 illustrates an embodiment of the invention including
a mechanical switch to assist in determining a hydration time. In
an embodiment of the invention, a single housing may include a
sensor assembly 1120 and a sensor electronics device 1125. In an
embodiment of the invention, the sensor assembly 1120 may be in one
housing and the sensor electronics device 1125 may be in a separate
housing, but the sensor assembly 1120 and the sensor electronics
device 1125 may be connected together. In this embodiment of the
invention, a connection detection mechanism 1160 may be a
mechanical switch. The mechanical switch may detect that the sensor
1120 is physically connected to the sensor electronics device 1125.
In an embodiment of the invention, a timer circuit 1135 may also be
activated when the mechanical switch 1160 detects that the sensor
1120 is connected to the sensor electronics device 1125. In other
words, the mechanical switch may close and a signal may be
transferred to a timer circuit 1135. Once a hydration time has
elapsed, the timer circuit 1135 transmits a signal to the switch
1140 to allow the regulator 1035 to apply a voltage to the sensor
1120. In other words, no voltage is applied until the hydration
time has elapsed. In an embodiment of the invention, current may
replace voltage as what is being applied to the sensor once the
hydration time elapses. In an alternative embodiment of the
invention, when the mechanical switch 1160 identifies that a sensor
1120 has been physically connected to the sensor electronics device
1125, power may initially be applied to the sensor 1120. Power
being sent to the sensor 1120 results in a sensor signal being
output from the working electrode in the sensor 1120. The sensor
signal may be measured and sent to a processor 1175. The processor
1175 may include a counter input. Under certain operating
conditions, after a set hydration time has elapsed from when the
sensor signal was input into the processor 1175, the processor 1175
may start processing the sensor signal as an accurate measurement
of the glucose in a subject's body. In other words, the processor
1170 has received the sensor signal from the potentiostat circuit
1170 for a certain amount of time, but will not process the signal
until receiving an instruction from the counter input of the
processor identifying that a hydration time has elapsed. In an
embodiment of the invention, the potentiostat circuit 1170 may
include a current-to-frequency converter 1180. In this embodiment
of the invention, the current-to-frequency converter 1180 may
receive the sensor signal as a current value and may convert the
current value into a frequency value, which is easier for the
processor 1175 to handle.
[0218] In an embodiment of the invention, the mechanical switch
1160 may also notify the processor 1175 when the sensor 1120 has
been disconnected from the sensor electronics device 1125. This is
represented by dotted line 1176 in FIG. 11. This may result in the
processor 1170 powering down or reducing power to a number of
components, chips, and/or circuits of the sensor electronics device
1125. If the sensor 1120 is not connected, the battery or power
source may be drained if the components or circuits of the sensor
electronics device 1125 are in a power on state. Accordingly, if
the mechanical switch 1160 detects that the sensor 1120 has been
disconnected from the sensor electronics device 1125, the
mechanical switch may indicate this to the processor 1175, and the
processor 1175 may power down or reduce power to one or more of the
electronic circuits, chips, or components of the sensor electronics
device 1125.
[0219] FIG. 12 illustrates an electrical method of detection of
hydration according to an embodiment of the invention. In an
embodiment of the invention, an electrical detecting mechanism for
detecting connection of a sensor may be utilized. In this
embodiment of the invention, the hydration detection electronics
1250 may include an AC source 1255 and a detection circuit 1260.
The hydration detection electronics 1250 may be located in the
sensor electronics device 1225. The sensor 1220 may include a
counter electrode 1221, a reference electrode 1222, and a working
electrode 1223. As illustrated in FIG. 12, the AC source 1255 is
coupled to a voltage setting device 1275, the reference electrode
1222, and the detection circuit 1260. In this embodiment of the
invention, an AC signal from the AC source is applied to the
reference electrode connection, as illustrated by dotted line 1291
in FIG. 12. In an embodiment of the invention, the AC signal is
coupled to the sensor 1220 through an impedance and the coupled
signal is attenuated significantly if the sensor 1220 is connected
to the sensor electronics device 1225. Thus, a low level AC signal
is present at an input to the detection circuit 1260. This may also
be referred to as a highly attenuated signal or a signal with a
high level of attenuation. Under certain operating conditions, the
voltage level of the AC signal may be
Vapplied*(Ccoupling)/(Ccoupling+Csensor). If the detection circuit
1260 detects that a high level AC signal (lowly attenuated signal)
is present at an input terminal of the detection circuit 1260, no
interrupt is sent to the microcontroller 410 because the sensor
1220 has not been sufficiently hydrated or activated. For example,
the input of the detection circuit 1260 may be a comparator. If the
sensor 1220 is sufficiently hydrated (or wetted), an effective
capacitance forms between the counter electrode and the reference
electrode (e.g., capacitance C.sub.r-c in FIG. 12), and an
effective capacitance forms between the reference electrode and the
working electrode (e.g., capacitance C.sub.w-r in FIG. 12). In
other words, an effective capacitance relates to capacitance being
formed between two nodes and does not represent that an actual
capacitor is placed in a circuit between the two electrodes. In an
embodiment of the invention, the AC signal from the AC source 1255
is sufficiently attenuated by capacitances C.sub.r-c and C.sub.w-r
and the detection circuit 1260 detects the presence of a low level
or highly attenuated AC signal from the AC source 1255 at the input
terminal of the detection circuit 1260. This embodiment of the
invention is significant because the utilization of the existing
connections between the sensor 1120 and the sensor electronics
device 1125 reduces the number of connections to the sensor. In
other words, the mechanical switch, disclosed in FIG. 11, requires
a switch and associated connections between the sensor 1120 and the
sensor electronics device 1125. It is advantageous to eliminate the
mechanical switch because the sensor 1120 is continuously shrinking
in size and the elimination of components helps achieve this size
reduction. In alternative embodiments of the invention, the AC
signal may be applied to different electrodes (e.g., the counter
electrode or the working electrode) and the invention may operate
in a similar fashion.
[0220] As noted above, after the detection circuit 1260 has
detected that a low level AC signal is present at the input
terminal of the detection circuit 1260, the detection circuit 1260
may later detect that a high level AC signal, with low attenuation,
is present at the input terminal. This represents that the sensor
1220 has been disconnected from the sensor electronics device 1225
or that the sensor is not operating properly. If the sensor has
been disconnected from the sensor electronics device 1225, the AC
source may be coupled with little or low attenuation to the input
of the detection circuit 1260. As noted above, the detection
circuit 1260 may generate an interrupt to the microcontroller. This
interrupt may be received by the microcontroller and the
microcontroller may reduce or eliminate power to one or a number of
components or circuits in the sensor electronics device 1225. This
may be referred to as the second interrupt. Again, this helps
reduce power consumption of the sensor electronics device 1225,
specifically when the sensor 1220 is not connected to the sensor
electronics device 1225.
[0221] In an alternative embodiment of the invention illustrated in
FIG. 12, the AC signal may be applied to the reference electrode
1222, as is illustrated by reference numeral 1291, and an impedance
measuring device 1277 may measure the impedance of an area in the
sensor 1220. Illustratively, the area may be an area between the
reference electrode and the working electrode, as illustrated by
dotted line 1292 in FIG. 12. Under certain operating conditions,
the impedance measuring device 1277 may transmit a signal to the
detection circuit 1260 if a measured impedance has decreased to
below an impedance threshold or other set criteria. This represents
that the sensor is sufficiently hydrated. Under other operating
conditions, the impedance measuring device 1277 may transmit a
signal to the detection circuit 1260 once the impedance is above an
impedance threshold. The detection circuit 1260 then transmits the
interrupt to the microcontroller 410. In another embodiment of the
invention, the impedance measuring device 1277 may transmit an
interrupt or signal directly to the microcontroller.
[0222] In an alternative embodiment of the invention, the AC source
1255 may be replaced by a DC source. If a DC source is utilized,
then a resistance measuring element may be utilized in place of an
impedance measuring element 1277. In an embodiment of the invention
utilizing the resistance measuring element, once the resistance
drops below a resistance threshold or a set criteria, the
resistance measuring element may transmit a signal to the detection
circuit 1260 (represented by dotted line 1293) or directly to the
microcontroller indicating that the sensor is sufficiently hydrated
and that power may be applied to the sensor.
[0223] In the embodiment of the invention illustrated in FIG. 12,
if the detection circuit 1260 detects a low level or highly
attenuated AC signal from the AC source, an interrupt is generated
to the microcontroller 410. This interrupt indicates that sensor is
sufficiently hydrated. In this embodiment of the invention, in
response to the interrupt, the microcontroller 410 generates a
signal that is transferred to a digital-to-analog converter 420 to
instruct or cause the digital-to-analog converter 420 to apply a
voltage or current to the sensor 1220. Any of the different
sequence of pulses or short duration pulses described above in FIG.
6A, 6B, or 6C or the associated text describing the application of
pulses, may be applied to the sensor 1220. Illustratively, the
voltage from the DAC 420 may be applied to an op-amp 1275, the
output of which is applied to the counter electrode 1221 of the
sensor 1220. This results in a sensor signal being generated by the
sensor, e.g., the working electrode 1223 of the sensor. Because the
sensor is sufficiently hydrated, as identified by the interrupt,
the sensor signal created at the working electrode 1223 is
accurately measuring glucose. The sensor signal is measured by a
sensor signal measuring device 431 and the sensor signal measuring
device 431 transmits the sensor signal to the microcontroller 410
where a parameter of a subject's physiological condition is
measured. The generation of the interrupt represents that a sensor
is sufficiently hydrated and that the sensor 1220 is now supplying
accurate glucose measurements. In this embodiment of the invention,
the hydration period may depend on the type and/or the manufacturer
of the sensor and on the sensor's reaction to insertion or
implantation in the subject. Illustratively, one sensor 1220 may
have a hydration time of five minutes and one sensor 1220 may have
a hydration time of one minute, two minutes, three minutes, six
minutes, or 20 minutes. Again, any amount of time may be an
acceptable amount of hydration time for the sensor, but smaller
amounts of time are preferable.
[0224] If the sensor 1220 has been connected, but is not
sufficiently hydrated or wetted, the effective capacitances
C.sub.r-c and C.sub.w-r may not attenuate the AC signal from the AC
source 1255. The electrodes in the sensor 1120 are dry before
insertion and because the electrodes are dry, a good electrical
path (or conductive path) does not exist between the two
electrodes. Accordingly, a high level AC signal or lowly attenuated
AC signal may still be detected by the detection circuit 1260 and
no interrupt may be generated. Once the sensor has been inserted,
the electrodes become immersed in the conductive body fluid. This
results in a leakage path with lower DC resistance. Also, boundary
layer capacitors form at the metal/fluid interface. In other words,
a rather large capacitance forms between the metal/fluid interface
and this large capacitance looks like two capacitors in series
between the electrodes of the sensor. This may be referred to as an
effective capacitance. In practice, a conductivity of an
electrolyte above the electrode is being measured. In some
embodiments of the invention, the glucose limiting membrane (GLM)
also illustrates impedance blocking electrical efficiency. An
unhydrated GLM results in high impedance, whereas a high moisture
GLM results in low impedance. Low impedance is desired for accurate
sensor measurements.
[0225] FIG. 13A illustrates a method of hydrating a sensor
according to an embodiment of the present invention. In an
embodiment of the invention, the sensor may be physically connected
1310 to the sensor electronics device. After the connection, in one
embodiment of the invention, a timer or counter may be initiated to
count 1320 a hydration time. After the hydration time has elapsed,
a signal may be transmitted 1330 to a subsystem in the sensor
electronics device to initiate the application of a voltage to the
sensor. As discussed above, in an embodiment of the invention, a
microcontroller may receive the signal and instruct the DAC to
apply a voltage to the sensor or in another embodiment of the
invention, a switch may receive a signal which allows a regulator
to apply a voltage to the sensor. The hydration time may be five
minutes, two minutes, ten minutes and may vary depending on the
subject and also on the type of sensor.
[0226] In an alternative embodiment of the invention, after the
connection of the sensor to the sensor electronics device, an AC
signal (e.g., a low voltage AC signal) may be applied 1340 to the
sensor, e.g., the reference electrode of the sensor. The AC signal
may be applied because the connection of the sensor to the sensor
electronics device allows the AC signal to be applied to the
sensor. After application of the AC signal, an effective
capacitance forms 1350 between the electrode in the sensor that the
voltage is applied to and the other two electrodes. A detection
circuit determines 1360 what level of the AC signal is present at
the input of the detection circuit. If a low level AC signal (or
highly attenuated AC signal) is present at the input of the
detection circuit, due to the effective capacitance forming a good
electrical conduit between the electrodes and the resulting
attenuation of the AC signal, an interrupt is generated 1370 by the
detection circuit and sent to a microcontroller.
[0227] The microcontroller receives the interrupt generated by the
detection circuit and transmits 1380 a signal to a
digital-to-analog converter instructing or causing the
digital-to-analog converter to apply a voltage to an electrode of
the sensor, e.g., the counter electrode. The application of the
voltage to the electrode of the sensor results in the sensor
creating or generating a sensor signal 1390. A sensor signal
measurement device 431 measures the generated sensor signal and
transmits the sensor signal to the microcontroller. The
microcontroller receives 1395 the sensor signal from the sensor
signal measurement device, which is coupled to the working
electrode, and processes the sensor signal to extract a measurement
of a physiological characteristic of the subject or patient.
[0228] FIG. 13B illustrates an additional method for verifying
hydration of a sensor according to an embodiment of the present
invention. In the embodiment of the invention illustrated in FIG.
13B, the sensor is physically connected 1310 to the sensor
electronics device. In an embodiment of the invention, an AC signal
is applied 1341 to an electrode, e.g., a reference electrode, in
the sensor. Alternatively, in an embodiment of the invention, a DC
signal is applied 1341 to an electrode in the sensor. If an AC
signal is applied, an impedance measuring element measures 1351 an
impedance at a point within the sensor. Alternatively, if a DC
signal is applied, a resistance measuring element measures 1351 a
resistance at a point within the sensor. If the resistance or
impedance is lower than a resistance threshold or an impedance
threshold, respectively, (or other set criteria), then the
impedance (or resistance) measuring element transmits 1361 (or
allows a signal to be transmitted) to the detection circuit, and
the detection circuit transmits an interrupt to the microcontroller
identifying that the sensor is hydrated. The reference numbers
1380, 1390, and 1395 are the same in FIGS. 13A and 13B because they
represent the same action.
[0229] The microcontroller receives the interrupt and transmits
1380 a signal to a digital-to-analog converter to apply a voltage
to the sensor. In an alternative embodiment of the invention, the
digital-to-analog converter can apply a current to the sensor, as
discussed above. The sensor, e.g., the working electrode, creates
1390 a sensor signal, which represents a physiological parameter of
a patient. The microcontroller receives 1395 the sensor signal from
a sensor signal measuring device, which measures the sensor signal
at an electrode in the sensor, e.g., the working electrode. The
microcontroller processes the sensor signal to extract a
measurement of the physiological characteristic of the subject or
patient, e.g., the blood glucose level of the patient.
[0230] FIGS. 14A and 14B illustrate methods of combining hydrating
of a sensor with stabilizing of a sensor according to an embodiment
of the present invention. In an embodiment of the invention
illustrated in FIG. 14A, the sensor is connected 1405 to the sensor
electronics device. The AC signal is applied 1410 to an electrode
of the sensor. The detection circuit determines 1420 what level of
the AC signal is present at an input of the detection circuit. If
the detection circuit determines that a low level of the AC signal
is present at the input (representing a high level of attenuation
to the AC signal), an interrupt is sent 1430 to microcontroller.
Once the interrupt is sent to the microcontroller, the
microcontroller knows to begin or initiate 1440 a stabilization
sequence, i.e., the application of a number of voltage pulses to an
electrode of the sensors, as described above. For example, the
microcontroller may cause a digital-to-analog converter to apply
three voltage pulses (having a magnitude of +0.535 volts) to the
sensor with each of the three voltage pulses followed by a period
of three voltage pulses (having a magnitude of 1.07 volts to be
applied). This may be referred to transmitting a stabilization
sequence of voltages. The microcontroller may cause this by the
execution of a software program in a read-only memory (ROM) or a
random access memory. After the stabilization sequence has finished
executing, the sensor may generate 1450 a sensor signal, which is
measured and transmitted to a microcontroller.
[0231] In an embodiment of the invention, the detection circuit may
determine 1432 that a high level AC signal has continued to be
present at the input of the detection circuit (e.g., an input of a
comparator), even after a hydration time threshold has elapsed. For
example, the hydration time threshold may be 10 minutes. After 10
minutes has elapsed, the detection circuit may still be detecting
that a high level AC signal is present. At this point in time, the
detection circuit may transmit 1434 a hydration assist signal to
the microcontroller. If the microcontroller receives the hydration
assist signal, the microcontroller may transmit 1436 a signal to
cause a DAC to apply a voltage pulse or a series of voltage pulses
to assist the sensor in hydration. In an embodiment of the
invention, the microcontroller may transmit a signal to cause the
DAC to apply a portion of the stabilization sequence or other
voltage pulses to assist in hydrating the sensor. In this
embodiment of the invention, the application of voltage pulses may
result in the low level AC signal (or highly attenuated signal)
being detected 1438 at the detection circuit. At this point, the
detection circuit may transmit an interrupt, as is disclosed in
step 1430, and the microcontroller may initiate a stabilization
sequence.
[0232] FIG. 14B illustrates a second embodiment of a combination of
a hydration method and a stabilization method where feedback is
utilized in the stabilization process. A sensor is connected 1405
to a sensor electronics device. An AC signal (or a DC signal) is
applied 1411 to the sensor. In an embodiment of the invention, the
AC signal (or the DC signal) is applied to an electrode of the
sensor, e.g. the reference electrode. An impedance measuring device
(or resistance measuring device) measures 1416 the impedance (or
resistance) within a specified area of the sensor. In an embodiment
of the invention, the impedance (or resistance) may be measured
between the reference electrode and the working electrode. The
measured impedance (or resistance) may be compared 1421 to an
impedance or resistance value to see if the impedance (or
resistance) is low enough in the sensor, which indicates the sensor
is hydrated. If the impedance (or resistance) is below the
impedance (or resistance) value or other set criteria, (which may
be a threshold value), an interrupt is transmitted 1431 to the
microcontroller. After receiving the interrupt, the microcontroller
transmits 1440 a signal to the DAC instructing the DAC to apply a
stabilization sequence of voltages (or currents) to the sensor.
After the stabilization sequence has been applied to the sensor, a
sensor signal is created in the sensor (e.g., at the working
electrode), is measured by a sensor signal measuring device, is
transmitted by the sensor signal measuring device, and is received
1450 by the microcontroller. Because the sensor is hydrated and the
stabilization sequence of voltages has been applied to the sensor,
the sensor signal is accurately measuring a physiological parameter
(i.e., blood glucose).
[0233] FIG. 14C illustrates a third embodiment of the invention
where a stabilization method and hydration method are combined. In
this embodiment of the invention, the sensor is connected 1500 to
the sensor electronics device. After the sensor is physically
connected to the sensor electronics device, an AC signal (or DC
signal) is applied 1510 to an electrode (e.g., reference electrode)
of the sensor. At the same time, or around the same time, the
microcontroller transmits a signal to cause the DAC to apply 1520 a
stabilization voltage sequence to the sensor. In an alternative
embodiment of the invention, a stabilization current sequence may
be applied to the sensor instead of a stabilization voltage
sequence. The detection circuit determines 1530 what level of an AC
signal (or DC signal) is present at an input terminal of the
detection circuit. If there is a low level AC signal (or DC
signal), representing a highly attenuated AC signal (or DC signal),
present at the input terminal of the detection circuit, an
interrupt is transmitted 1540 to the microcontroller. Because the
microcontroller has already initiated the stabilization sequence,
the microcontroller receives the interrupt and sets 1550 a first
indicator that the sensor is sufficiently hydrated. After the
stabilization sequence is complete, the microcontroller sets 1555 a
second indicator indicating the completion of the stabilization
sequence. The application of the stabilization sequence voltages
results in the sensor, e.g., the working electrode, creating 1560 a
sensor signal, which is measured by a sensor signal measuring
circuit, and sent to the microcontroller. If the second indicator
that the stabilization sequence is complete is set and the first
indicator that the hydration is complete is set, the
microcontroller is able to utilize 1570 the sensor signal. If one
or both of the indicators are not set, the microcontroller may not
utilize the sensor signal because the sensor signal may not
represent accurate measurements of the physiological measurements
of the subject.
[0234] The above-described hydration and stabilization processes
may be used, in general, as part of a larger continuous glucose
monitoring (CGM) methodology. The current state of the art in
continuous glucose monitoring is largely adjunctive, meaning that
the readings provided by a CGM device (including, e.g., an
implantable or subcutaneous sensor) cannot be used without a
reference value in order to make a clinical decision. The reference
value, in turn, must be obtained from a finger stick using, e.g., a
BG meter. The reference value is needed because there is a limited
amount of information that is available from the sensor/sensing
component. Specifically, the only pieces of information that are
currently provided by the sensing component for processing are the
raw sensor value (i.e., the sensor current or Isig) and the counter
voltage, which is the voltage between the counter electrode and the
reference electrode (see, e.g., FIG. 5). Therefore, during
analysis, if it appears that the raw sensor signal is abnormal
(e.g., if the signal is decreasing), the only way one can
distinguish between a sensor failure and a physiological change
within the user/patient (i.e., glucose level changing in the body)
is by acquiring a reference glucose value via a finger stick. As is
known, the reference finger stick is also used for calibrating the
sensor.
[0235] Embodiments of the inventions described herein are directed
to advancements and improvements in continuous glucose monitoring
resulting in a more autonomous system, as well as related devices
and methodologies, wherein the requirement of reference finger
sticks may be minimized, or eliminated, and whereby clinical
decisions may be made based on information derived from the sensor
signal alone, with a high level of reliability. From a
sensor-design standpoint, in accordance with embodiments of the
invention, such autonomy may be achieved through electrode
redundancy, sensor diagnostics, and Isig and/or sensor glucose (SG)
fusion.
[0236] As will be explored further hereinbelow, redundancy may be
achieved through the use of multiple working electrodes (e.g., in
addition to a counter electrode and a reference electrode) to
produce multiple signals indicative of the patient's blood glucose
(BG) level. The multiple signals, in turn, may be used to assess
the relative health of the (working) electrodes, the overall
reliability of the sensor, and the frequency of the need, if at
all, for calibration reference values.
[0237] Sensor diagnostics includes the use of additional
(diagnostic) information which can provide a real-time insight into
the health of the sensor. In this regard, it has been discovered
that Electrochemical Impedance Spectroscopy (EIS) provides such
additional information in the form of sensor impedance and
impedance-related parameters at different frequencies. Moreover,
advantageously, it has been further discovered that, for certain
ranges of frequencies, impedance and/or impedance-related data are
substantially glucose independent. Such glucose independence
enables the use of a variety of EIS-based markers or indicators for
not only producing a robust, highly-reliable sensor glucose value
(through fusion methodologies), but also assessing the condition,
health, age, and efficiency of individual electrode(s) and of the
overall sensor substantially independently of the glucose-dependent
Isig.
[0238] For example, analysis of the glucose-independent impedance
data provides information on the efficiency of the sensor with
respect to how quickly it hydrates and is ready for data
acquisition using, e.g., values for 1 kHz real-impedance, 1 kHz
imaginary impedance, and Nyquist Slope (to be described in more
detail hereinbelow). Moreover, glucose-independent impedance data
provides information on potential occlusion(s) that may exist on
the sensor membrane surface, which occlusion(s) may temporarily
block passage of glucose into the sensor and thus cause the signal
to dip (using, e.g., values for 1 kHz real impedance). In addition,
glucose-independent impedance data provides information on loss of
sensor sensitivity during extended wear--potentially due to local
oxygen deficit at the insertion site--using, e.g., values for phase
angle and/or imaginary impedance at 1 kHz and higher
frequencies.
[0239] Within the context of electrode redundancy and EIS, as well
as other contexts, as will be described in further detail
hereinbelow, a fusion algorithm may be used to take the diagnostic
information provided by EIS for each redundant electrode and assess
the reliability of each electrode independently. Weights, which are
a measure of reliability, may then be added for each independent
signal, and a single fused signal may be calculated that can be
used to generate sensor glucose values as seen by the
patient/subject.
[0240] As can be seen from the above, the combined use of
redundancy, sensor diagnostics using EIS, and EIS-based fusion
algorithms allows for an overall CGM system that is more reliable
than what is currently available. Redundancy is advantageous in at
least two respects. First, redundancy removes the risk of a single
point of failure by providing multiple signals. Second, providing
multiple (working) electrodes where a single electrode may be
sufficient allows the output of the redundant electrode to be used
as a check against the primary electrode, thereby reducing, and
perhaps eliminating, the need for frequent calibrations. In
addition, EIS diagnostics scrutinize the health of each electrode
autonomously without the need for a reference glucose value (finger
stick), thereby reducing the number of reference values required.
However, the use of EIS technology and EIS diagnostic methods is
not limited to redundant systems, i.e., those having more than one
working electrode. Rather, as is discussed below in connection with
embodiments of the invention, EIS may be advantageously used in
connection with single- and/or multiple-electrode sensors.
[0241] EIS, or AC impedance methods, study the system response to
the application of a periodic small amplitude AC signal. This is
shown illustratively in FIG. 15A, where E is the applied potential,
I is the current, and impedance (Z) is defined as AE/AI. However,
although impedance, per se, may be mathematically simply defined as
AE/AI, heretofore, there has been no commercialization success in
application of EIS technology to continuous glucose monitoring.
This has been due, in part, to the fact that glucose sensors are
very complicated systems and, so far, no mathematical models have
been developed which can completely explain the complexity of the
EIS output for a glucose sensor.
[0242] One simplified electrical circuit model that has been used
to describe electrochemical impedance spectroscopy is shown in FIG.
15B. In this illustration, IHP stands for Inner Helmholtz Plane,
OHP stands for Outer Helmholtz Plane, CE is the counter electrode,
WE is the working electrode, C.sub.d is double layer capacitance,
R.sub.p is polarization resistance, Z.sub.w is Warburg impedance,
and R.sub.s is solution resistance. Each of the latter four
components--double layer capacitance (C.sub.d), Warburg impedance
(Z.sub.w), polarization resistance (R.sub.p), and solution
resistance (R.sub.s)--may play a significant role in sensor
performance, and can be measured separately by applying low- or
high-frequency alternating working potential. For example, Warburg
impedance is closely related to diffusional impedance of
electrochemical systems--which is primarily a low-frequency
impedance--and, as such, exists in all diffusion-limited
electrochemical sensors. Thus, by correlating one or more of these
components with one or more components and/or layers of a glucose
sensor, one may use EIS technology as a sensor-diagnostics
tool.
[0243] As is known, impedance may be defined in terms of its
magnitude and phase, where the magnitude (|Z|) is the ratio of the
voltage difference amplitude to the current amplitude, and the
phase (.theta.) is the phase shift by which the current is ahead of
the voltage. When a circuit is driven solely with direct current
(DC), the impedance is the same as the resistant, i.e., resistance
is a special case of impedance with zero phase angle. However, as a
complex quantity, impedance may also be represented by its real and
imaginary parts. In this regard, the real and imaginary impedance
can be derived from the impedance magnitude and phase using the
following equations:
Real
Impedance(.omega.)=Magnitude(.omega.).times.cos(Phase(.omega.)/180.-
times..pi.)
Imaginary
Impedance(.omega.)=Magnitude(.omega.).times.sin(Phase(.omega.)/180.times.-
.pi.)
where .omega. represents the input frequency at which the magnitude
(in ohms) and the phase (in degrees) are measured. The relationship
between impedance, on the one hand, and current and voltage on the
other--including how the former may be calculated based on
measurement of the latter--will be explored more fully below in
connection with the sensor electronics, including the Application
Specific Integrated Circuit (ASIC), that has been developed for use
in embodiments of the invention.
[0244] Continuing with the circuit model shown in FIG. 15B, total
system impedance may be simplified as:
Z t ( .omega. ) = Z w ( .omega. ) + R s + R p 1 + .omega. 2 R p 2 C
d 2 - j .omega. R p 2 C d 1 + .omega. 2 R p 2 C d 2
##EQU00001##
where Z.sub.w(.omega.) is the Warburg impedance, .omega. is the
angular velocity, j is the imaginary unit (used instead of the
traditional "i" so as not to be confused with electric current),
and C.sub.d, R.sub.p, and R.sub.s are the double layer capacitance,
the polarization resistance, and the solution resistance,
respectively (as defined previously). Warburg impedance can be
calculated as
Z w ( .omega. ) = Z 0 tanh ( ( js ) m ) ( js ) m ##EQU00002## s = L
2 .omega. / D = ( Membrane Thickness Frequency Dependent Diffusion
Length ) 2 ##EQU00002.2## Z 0 = RTL n 2 F 2 D C ##EQU00002.3##
where D is diffusivity, L is the sensor membrane thickness, C is
Peroxide concentration, and m: 1/2 corresponds to a 45.degree.
Nyquist slope.
[0245] A Nyquist plot is a graphical representation, wherein the
real part of impedance (Real Z) is plotted against its imaginary
part (Img Z) across a spectrum of frequencies. FIG. 16A shows a
generalized example of a Nyquist Plot, where the X value is the
real part of the impedance and the Y value is the imaginary part of
the impedance. The phase angle is the angle between the impedance
point (X,Y)--which defines a vector having magnitude |Z|--and the X
axis.
[0246] The Nyquist plot of FIG. 16A is generated by applying AC
voltages plus a DC voltage (DC bias) between the working electrode
and the counter electrode at selected frequencies from 0.1 Hz to
1000 MHz (i.e., a frequency sweep). Starting from the right, the
frequency increases from 0.1 Hz. With each frequency, the real and
imaginary impedance can be calculated and plotted. As shown, a
typical Nyquist plot of an electrochemical system may look like a
semicircle joined with a straight line at an inflection point,
wherein the semicircle and the line indicate the plotted impedance.
In certain embodiments, the impedance at the inflection point is of
particular interest since it is easiest to identify in the Nyquist
plot and may define an intercept. Typically, the inflection point
is close to the X axis, and the X value of the inflection point
approximates the sum of the polarization resistance and solution
resistance (R.sub.p+R.sub.b).
[0247] With reference to FIG. 16B, a Nyquist plot may typically be
described in terms of a lower-frequency region 1610 and a
higher-frequency region 1620, where the labels "higher frequency"
and "lower frequency" are used in a relative sense, and are not
meant to be limiting. Thus, for example, the lower-frequency region
1610 may illustratively include data points obtained for a
frequency range between about 0.1 Hz and about 100 Hz (or higher),
and the higher-frequency region 1620 may illustratively include
data points obtained for a frequency range between about 1 kHz (or
lower) and about 8 kHz (and higher). In the lower-frequency region
1610, the Nyquist slope represents the gradient of the linear fit
1630 of the lower-frequency data points in the Nyquist plot. As
shown, in the higher-frequencies region 1620, the value of
imaginary impedance is minimal, and may become negligible. As such,
the intercept 1600 is essentially the value of the real impedance
at the higher frequencies (e.g., approximately in the 1 kHz to 8
kHz range in this case). In FIG. 16B, the intercept 1600 is at
about 25 kOhms.
[0248] FIGS. 16C and 16D demonstrate how a glucose sensor responds
to a sinusoidal (i.e., alternating) working potential. In these
figures, GLM is the sensor's glucose limiting membrane, AP is the
adhesion promoter, HSA is human serum albumin, GOX is glucose
oxidase enzyme (layer), E.sub.de is DC potential, E.sub.ac is AC
potential, and C.sub.peroxide' is peroxide concentration during AC
application. As shown in FIG. 16C, if the sensor diffusion length,
which is a function of AC potential frequency, molecular
diffusivity, and membrane thickness, is small compared to the
membrane (GOX) length, the system gives a relatively linear
response with a constant phase angle (i.e., infinite). In contrast,
if the diffusion length is equal to the membrane (GOX) length, the
system response will become finite, resulting in a semi-circle
Nyquist plot, as shown in FIG. 16D. The latter usually holds true
for low-frequency EIS, where the non-Faradaic process is
negligible.
[0249] In performing an EIS analysis, an AC voltage of various
frequencies and a DC bias may be applied between, e.g., the working
and reference electrodes. In this regard, EIS is an improvement
over previous methodologies that may have limited the application
to a simple DC current or an AC voltage of single frequency.
Although, generally, EIS may be performed at frequencies in the
.mu.Hz to MHz range, in embodiments of the invention, a narrower
range of frequencies (e.g., between about 0.1 Hz and about 8 kHz)
may be sufficient. Thus, in embodiments of the invention, AC
potentials may be applied that fall within a frequency range of
between about 0.1 Hz and about 8 kHz, with a programmable amplitude
of up to at least 100 mV, and preferably at about 50 mV.
[0250] Within the above-mentioned frequency range, the
relatively-higher frequencies--i.e., those that fall generally
between about 1 kHz and about 8 kHz--are used to scrutinize the
capacitive nature of the sensor. Depending on the thickness and
permeability of membranes, a typical range of impedance at the
relatively-higher frequencies may be, e.g., between about 500 Ohms
and 25 kOhms, and a typical range for the phase may be, e.g.,
between 0 degrees and -40 degrees. The relatively-lower
frequencies--i.e., those that fall generally between about 0.1 Hz
and about 100 Hz--on the other hand, are used to scrutinize the
resistive nature of the sensor. Here, depending on electrode design
and the extent of metallization, a typical functioning range for
output real impedance may be, e.g., between about 50 kOhms and 300
kOhms, and a typical range for the phase may be between about -50
degrees to about -90 degrees. The above illustrative ranges are
shown, e.g., in the Bode plots of FIGS. 16E and 16F.
[0251] As noted previously, the phrases "higher frequencies" and
"lower frequencies" are meant to be used relative to one another,
rather than in an absolute sense, and they, as well as the typical
impedance and phase ranges mentioned above, are meant to be
illustrative, and not limiting. Nevertheless, the underlying
principle remains the same: the capacitive and resistive behavior
of a sensor can be scrutinized by analyzing the impedance data
across a frequency spectrum, wherein, typically, the lower
frequencies provide information about the more resistive components
(e.g., the electrode, etc.), while the higher frequencies provide
information about the capacitive components (e.g., membranes).
However, the actual frequency range in each case is dependent on
the overall design, including, e.g., the type(s) of electrode(s),
the surface area of the electrode(s), membrane thickness, the
permeability of the membrane, and the like. See also FIG. 15B
regarding general correspondence between high-frequency circuit
components and the sensor membrane, as well as between
low-frequency circuit components and the Faradaic process,
including, e.g., the electrode(s).
[0252] EIS may be used in sensor systems where the sensor includes
a single working electrode, as well those in which the sensor
includes multiple (redundant) working electrodes. In one
embodiment, EIS provides valuable information regarding the age (or
aging) of the sensor. Specifically, at different frequencies, the
magnitude and the phase angle of the impedance vary. As seen in
FIG. 17, the sensor impedance--in particular, the sum of Rp and
Rs--reflects the sensor age as well as the sensor's operating
conditions. Thus, a new sensor normally has higher impedance than a
used sensor as seen from the different plots in FIG. 17. In this
way, by considering the X-value of the sum of Rp and Rs, a
threshold can be used to determine when the sensor's age has
exceeded the specified operating life of the sensor. It is noted
that, although for the illustrative examples shown in FIGS. 17-21
and discussed below, the value of real impedance at the inflection
point (i.e., Rp+Rs) is used to determine the aging, status,
stabilization, and hydration of the sensor, alternative embodiments
may use other EIS-based parameters, such as, e.g., imaginary
impedance, phase angle, Nyquist slope, etc. in addition to, or in
place of, real impedance.
[0253] FIG. 17 illustrates an example of a Nyquist plot over the
life time of a sensor. The points indicated by arrows are the
respective inflection points for each of the sweeps across the
frequency spectrum. For example, before initialization (at time
t=0), Rs+Rp is higher than 8.5 kOhms, and after initialization (at
time t=0.5 hr), the value of Rs+Rp dropped to below 8 kOhms. Over
the next six days, Rs+Rp continues to decrease, such that, at the
end of the specified sensor life, Rs+Rp dropped to below 6.5 kOhms.
Based on such examples, a threshold value can be set to specify
when the Rs+Rp value would indicate the end of the specified
operating life of the sensor. Therefore, the EIS technique allows
closure of the loophole of allowing a sensor to be re-used beyond
the specified operating time. In other words, if the patient
attempts to re-use a sensor after the sensor has reached its
specified operating time by disconnecting and then re-connecting
the sensor again, the EIS will measure abnormally-low impedance,
thereby enabling the system to reject the sensor and prompt the
patient for a new sensor.
[0254] Additionally, EIS may enable detection of sensor failure by
detecting when the sensor's impedance drops below a low impedance
threshold level indicating that the sensor may be too worn to
operate normally. The system may then terminate the sensor before
the specified operating life. As will be explored in more detail
below, sensor impedance can also be used to detect other sensor
failure (modes). For example, when a sensor goes into a low-current
state (i.e., sensor failure) due to any variety of reasons, the
sensor impedance may also increase beyond a certain high impedance
threshold. If the impedance becomes abnormally high during sensor
operation, due, e.g., to protein or polypeptide fouling, macrophage
attachment or any other factor, the system may also terminate the
sensor before the specified sensor operating life.
[0255] FIG. 18 illustrates how the EIS technique can be applied
during sensor stabilization and in detecting the age of the sensor
in accordance with embodiments of the invention. The logic of FIG.
18 begins at 1800 after the hydration procedure and sensor
initialization procedure described previously has been completed.
In other words, the sensor has been deemed to be sufficiently
hydrated, and the first initialization procedure has been applied
to initialize the sensor. The initialization procedure may
preferably be in the form of voltage pulses as described previously
in the detailed description. However, in alternative embodiments,
different waveforms can be used for the initialization procedure.
For example, a sine wave can be used, instead of the pulses, to
accelerate the wetting or conditioning of the sensor. In addition,
it may be necessary for some portion of the waveform to be greater
than the normal operating voltage of the sensor, i.e., 0.535
volt.
[0256] At block 1810, an EIS procedure is applied and the impedance
is compared to both a first high and a first low threshold. An
example of a first high and first low threshold value would be 7
kOhms and 8.5 kOhms, respectively, although the values can be set
higher or lower as needed. If the impedance, for example, Rp+Rs, is
higher than the first high threshold, the sensor undergoes an
additional initialization procedure (e.g., the application of one
or more additional pulses) at block 1820. Ideally, the number of
total initialization procedures applied to initialize the sensor
would be optimized to limit the impact on both the battery life of
the sensor and the overall amount of time needed to stabilize a
sensor. Thus, by applying EIS, fewer initializations can be
initially performed, and the number of initializations can be
incrementally added to give just the right amount of
initializations to ready the sensor for use. Similarly, in an
alternative embodiment, EIS can be applied to the hydration
procedure to minimize the number of initializations needed to aid
the hydration process as described in FIGS. 13-14.
[0257] On the other hand, if the impedance, for example, Rp+Rs, is
below the first low threshold, the sensor will be determined to be
faulty and would be terminated immediately at block 1860. A message
will be given to the user to replace the sensor and to begin the
hydration process again. If the impedance is within the high and
low thresholds, the sensor will begin to operate normally at block
1830. The logic then proceeds to block 1840 where an additional EIS
is performed to check the age of the sensor. The first time the
logic reaches block 1840, the microcontroller will perform an EIS
to gauge the age of the sensor to close the loophole of the user
being able to plug in and plug out the same sensor. In future
iterations of the EIS procedure as the logic returns to block 1840,
the microprocessor will perform an EIS at fixed intervals during
the specified life of the sensor. In one preferred embodiment, the
fixed interval is set for every 2 hours, however, longer or shorter
periods of time can easily be used.
[0258] At block 1850, the impedance is compared to a second set of
high and low thresholds. An example of such second high and low
threshold values may be 5.5 kOhms and 8.5 kOhms, respectively,
although the values can be set higher or lower as needed. As long
as the impedance values stay within a second high and low
threshold, the logic proceeds to block 1830, where the sensor
operates normally until the specified sensor life, for example, 5
days, is reached. Of course, as described with respect to block
1840, EIS will be performed at the regularly scheduled intervals
throughout the specified sensor life. However, if, after the EIS is
performed, the impedance is determined to have dropped below a
second lower threshold or risen above a second higher threshold at
block 1850, the sensor is terminated at block 1860. In further
alternative embodiments, a secondary check can be implemented of a
faulty sensor reading. For example, if the EIS indicates that the
impedance is out of the range of the second high and low
thresholds, the logic can perform a second EIS to confirm that the
second set of thresholds is indeed not met (and confirm that the
first EIS was correctly performed) before determining the end of
sensor at block 1860.
[0259] FIG. 19 builds upon the above description and details a
possible schedule for performing diagnostic EIS procedures in
accordance with preferred embodiments of the present invention.
Each diagnostic EIS procedure is optional and it is possible to not
schedule any diagnostic EIS procedure or to have any combination of
one or more diagnostic EIS procedures, as deemed needed. The
schedule of FIG. 19 begins at sensor insertion at point 1900.
Following sensor insertion, the sensor undergoes a hydration period
1910. This hydration period is important because a sensor that is
not sufficiently hydrated may give the user inaccurate readings, as
described previously. The first optional diagnostic EIS procedure
at point 1920 is scheduled during this hydration period 1910 to
ensure that the sensor is sufficiently hydrated. The first
diagnostic EIS procedure 1920 measures the sensor impedance value
to determine if the sensor has been sufficiently hydrated. If the
first diagnostic EIS procedure 1920 determines impedance is within
a set high and low threshold, indicating sufficient hydration, the
sensor controller will allow the sensor power-up at point 1930.
Conversely, if the first diagnostic EIS procedure 1920 determines
impedance is outside a set high and low threshold, indicating
insufficient hydration, the sensor hydration period 1910 may be
prolonged. After prolonged hydration, once a certain capacitance
has been reached between the sensor's electrodes, meaning the
sensor is sufficiently hydrated, power-up at point 1930 can
occur.
[0260] A second optional diagnostic EIS procedure 1940 is scheduled
after sensor power-up at point 1930, but before sensor
initialization starts at point 1950. Scheduled here, the second
diagnostic EIS procedure 1940 can detect if a sensor is being
re-used prior to the start of initialization at 1950. The test to
determine if the sensor is being reused was detailed in the
description of FIG. 18. However, unlike the previous description
with respect to FIG. 18, where the aging test is performed after
initialization is completed, the aging test is shown in FIG. 19 as
being performed before initialization. It is important to
appreciate that the timeline of EIS procedures described in FIG. 19
can be rearranged without affecting the overall teaching of the
application, and that the order of some of the steps can be
interchanged. As explained previously, the second diagnostic EIS
procedure 1940 detects a re-used sensor by determining the sensor's
impedance value and then comparing it to a set high and low
threshold. If impedance falls outside of the set threshold,
indicating the sensor is being re-used, the sensor may then be
rejected and the user prompted to replace it with a new sensor.
This prevents the complications that may arise out of re-use of an
old sensor. Conversely, if impedance falls within a set threshold,
sensor initialization 1950 can start with the confidence that a new
sensor is being used.
[0261] A third optional diagnostic EIS procedure 1960 is scheduled
after initialization starts at point 1950. The third diagnostic EIS
procedure 1960 tests the sensor's impedance value to determine if
the sensor is fully initialized. The third diagnostic EIS procedure
1960 should be performed at the minimum amount of time needed for
any sensor to be fully initialized. When performed at this time,
sensor life is maximized by limiting the time a fully initialized
sensor goes unused, and over-initialization is averted by
confirming full initialization of the sensor before too much
initialization occurs. Preventing over-initialization is important
because over-initialization results in a suppressed current which
can cause inaccurate readings. However, under-initialization is
also a problem, so if the third diagnostic EIS procedure 1960
indicates the sensor is under-initialized, an optional
initialization at point 1970 may be performed in order to fully
initialize the sensor. Under-initialization is disadvantageous
because an excessive current results that does not relate to the
actual glucose concentration. Because of the danger of under- and
over-initialization, the third diagnostic EIS procedure plays an
important role in ensuring the sensor functions properly when
used.
[0262] In addition, optional periodic diagnostic EIS procedures
1980 can be scheduled for the time after the sensor is fully
initialized. The EIS procedures 1980 can be scheduled at any set
interval. As will be discussed in more detail below, EIS procedures
1980 may also be triggered by other sensor signals, such as an
abnormal current or an abnormal counter electrode voltage.
Additionally, as few or as many EIS procedures 1980 can be
scheduled as desired. In preferred embodiments, the EIS procedure
used during the hydration process, sensor life check,
initialization process, or the periodic diagnostic tests is the
same procedure. In alternative embodiments, the EIS procedure can
be shortened or lengthened (i.e., fewer or more ranges of
frequencies checked) for the various EIS procedures depending on
the need to focus on specific impedance ranges. The periodic
diagnostic EIS procedures 1980 monitor impedance values to ensure
that the sensor is continuing to operate at an optimal level.
[0263] The sensor may not be operating at an optimal level if the
sensor current has dropped due to polluting species, sensor age, or
a combination of polluting species and sensor age. A sensor that
has aged beyond a certain length is no longer useful, but a sensor
that has been hampered by polluting species can possibly be
repaired. Polluting species can reduce the surface area of the
electrode or the diffusion pathways of analytes and reaction
byproducts, thereby causing the sensor current to drop. These
polluting species are charged and gradually gather on the electrode
or membrane surface under a certain voltage. Previously, polluting
species would destroy the usefulness of a sensor. Now, if periodic
diagnostic EIS procedures 1980 detect impedance values which
indicate the presence of polluting species, remedial action can be
taken. When remedial action is to be taken is described with
respect to FIG. 20. Periodic diagnostic EIS procedures 1980
therefore become extremely useful because they can trigger sensor
remedial action which can possibly restore the sensor current to a
normal level and prolong the life of the sensor. Two possible
embodiments of sensor remedial actions are described below in the
descriptions of FIGS. 21A and 21B.
[0264] Additionally, any scheduled diagnostic EIS procedure 1980
may be suspended or rescheduled when certain events are determined
imminent. Such events may include any circumstance requiring the
patient to check the sensor reading, including for example when a
patient measures his or her BG level using a test strip meter in
order to calibrate the sensor, when a patient is alerted to a
calibration error and the need to measure his or her BG level using
a test strip meter a second time, or when a hyperglycemic or
hypoglycemic alert has been issued but not acknowledged.
[0265] FIG. 20 illustrates a method of combining diagnostic EIS
procedures with sensor remedial action in accordance with
embodiments of the present invention. The block 2000 diagnostic
procedure may be any of the periodic diagnostic EIS procedures 1980
as detailed in FIG. 19. The logic of this method begins when a
diagnostic EIS procedure is performed at block 2000 in order to
detect the sensor's impedance value. As noted, in specific
embodiments, the EIS procedure applies a combination of a DC bias
and an AC voltage of varying frequencies wherein the impedance
detected by performing the EIS procedure is mapped on a Nyquist
plot, and an inflection point in the Nyquist plot approximates a
sum of polarization resistance and solution resistance (i.e., the
real impedance value). After the block 2000 diagnostic EIS
procedure detects the sensor's impedance value, the logic moves to
block 2010.
[0266] At block 2010, the impedance value is compared to a set high
and low threshold to determine if it is normal. If impedance is
within the set boundaries of the high and low thresholds at block
2010, normal sensor operation is resumed at block 2020 and the
logic of FIG. 20 will end until a time when another diagnostic EIS
procedure is scheduled. Conversely, if impedance is determined to
be abnormal (i.e., outside the set boundaries of the high and low
thresholds) at block 2010, remedial action at block 2030 is
triggered. An example of a high and low threshold value that would
be acceptable during a sensor life would be 5.5 kOhms and 8.5
kOhms, respectively, although the values can be set higher or lower
as needed.
[0267] The block 2030 remedial action is performed to remove any of
the polluting species, which may have caused the abnormal impedance
value. In preferred embodiments, the remedial action is performed
by applying a reverse current, or a reverse voltage between the
working electrode and the reference electrode. The specifics of the
remedial action will be described in more detail with respect to
FIG. 21. After the remedial action is performed at block 2030,
impedance value is again tested by a diagnostic EIS procedure at
block 2040. The success of the remedial action is then determined
at block 2050 when the impedance value from the block 2040
diagnostic EIS procedure is compared to the set high or low
threshold. As in block 2010, if impedance is within the set
thresholds, it is deemed normal, and if impedance is outside the
set thresholds, it is deemed abnormal.
[0268] If the sensor's impedance value is determined to have been
restored to normal at block 2050, normal sensor operation at block
2020 will occur. If impedance is still not normal, indicating that
either sensor age is the cause of the abnormal impedance or the
remedial action was unsuccessful in removing the polluting species,
the sensor is then terminated at block 2060. In alternative
embodiments, instead of immediately terminating the sensor, the
sensor may generate a sensor message initially requesting the user
to wait and then perform further remedial action after a set period
of time has elapsed. This alternative step may be coupled with a
separate logic to determine if the impedance values are getting
closer to being within the boundary of the high and low threshold
after the initial remedial action is performed. For example, if no
change is found in the sensor impedance values, the sensor may then
decide to terminate. However, if the sensor impedance values are
getting closer to the preset boundary, yet still outside the
boundary after the initial remedial action, an additional remedial
action could be performed. In yet another alternative embodiment,
the sensor may generate a message requesting the user to calibrate
the sensor by taking a finger stick meter measurement to further
confirm whether the sensor is truly failing. All of the above
embodiments work to prevent a user from using a faulty sensor that
produces inaccurate readings.
[0269] FIG. 21A illustrates one embodiment of the sensor remedial
action previously mentioned. In this embodiment, blockage created
by polluting species is removed by reversing the voltage being
applied to the sensor between the working electrode and the
reference electrode. The reversed DC voltage lifts the charged,
polluting species from the electrode or membrane surface, clearing
diffusion pathways. With cleared pathways, the sensor's current
returns to a normal level and the sensor can give accurate
readings. Thus, the remedial action saves the user the time and
money associated with replacing an otherwise effective sensor.
[0270] FIG. 21B illustrates an alternative embodiment of the sensor
remedial action previously mentioned. In this embodiment, the
reversed DC voltage applied between the working electrode and the
reference electrode is coupled with an AC voltage. By adding the AC
voltage, certain tightly absorbed species or species on the
superficial layer can be removed since the AC voltage can extend
its force further from the electrode and penetrate all layers of
the sensor. The AC voltage can come in any number of different
waveforms. Some examples of waveforms that could be used include
square waves, triangular waves, sine waves, or pulses. As with the
previous embodiment, once polluting species are cleared, the sensor
can return to normal operation, and both sensor life and accuracy
are improved.
[0271] While the above examples illustrate the use, primarily, of
real impedance data in sensor diagnostics, embodiments of the
invention are also directed to the use of other EIS-based, and
substantially analyte-independent, parameters (in addition to real
impedance) in sensor diagnostic procedures. For example, as
mentioned previously, analysis of (substantially)
glucose-independent impedance data, such as, e.g., values for 1 kHz
real impedance and 1 kHz imaginary impedance, as well as Nyquist
slope, provide information on the efficiency of the sensor with
respect to how quickly it hydrates and is ready for data
acquisition. Moreover, (substantially) glucose-independent
impedance data, such as, e.g., values for 1 kHz real impedance,
provides information on potential occlusion(s) that may exist on
the sensor membrane surface, which occlusion(s) may temporarily
block passage of glucose into the sensor and thus cause the signal
to dip.
[0272] In addition, (substantially) glucose-independent impedance
data, such as, e.g., values for higher-frequency phase angle and/or
imaginary impedance at 1 kHz and higher frequencies, provides
information on loss of sensor sensitivity during extended wear,
which sensitivity loss may potentially be due to local oxygen
deficit at the insertion site. In this regard, the underlying
mechanism for oxygen deficiency-led sensitivity loss may be
described as follows: when local oxygen is deficient, sensor output
(i.e., Isig and SG) will be dependent on oxygen rather than glucose
and, as such, the sensor will lose sensitivity to glucose. Other
markers, including 0.1 Hz real impedance, the counter electrode
voltage (Vcntr), and EIS-induced spikes in the Isig may also be
used for the detection of oxygen deficiency-led sensitivity loss.
Moreover, in a redundant sensor system, the relative differences in
1 kHz real impedance, 1 kHz imaginary impedance, and 0.1 Hz real
impedance between two or more working electrodes may be used for
the detection of sensitivity loss due to biofouling.
[0273] In accordance with embodiments of the invention, EIS-based
sensor diagnostics entails consideration and analysis of EIS data
relating to one or more of at least three primary factors, i.e.,
potential sensor failure modes: (1) signal start-up; (2) signal
dip; and (3) sensitivity loss. Significantly, the discovery herein
that a majority of the impedance-related parameters that are used
in such diagnostic analyses and procedures can be studied at a
frequency, or within a range of frequencies, where the parameter is
substantially analyte-independent allows for implementation of
sensor-diagnostic procedures independently of the level of the
analyte in a patient's body. Thus, while EIS-based sensor
diagnostics may be triggered by, e.g., large fluctuations in Isig,
which is analyte-dependent, the impedance-related parameters that
are used in such sensor diagnostic procedures are themselves
substantially independent of the level of the analyte. As will be
explored in more detail below, it has also been found that, in a
majority of situations where glucose may be seen to have an effect
on the magnitude (or other characteristic) of an EIS-based
parameter, such effect is usually small enough--e.g., at least an
order of magnitude difference between the EIS-based measurement and
the glucose effect thereon--such that it can be filtered out of the
measurement, e.g., via software in the IC.
[0274] By definition, "start-up" refers to the integrity of the
sensor signal during the first few hours (e.g., t=0-6 hours) after
insertion. For example, in current devices, the signal during the
first 2 hours after insertion is deemed to be unreliable and, as
such, the sensor glucose values are blinded to the patient/user. In
situations where the sensor takes an extended amount of time to
hydrate, the sensor signal is low for several hours after
insertion. With the use of EIS, additional impedance information is
available (by running an EIS procedure) right after the sensor has
been inserted. In this regard, the total impedance equation may be
used to explain the principle behind low-startup detection using 1
kHz real impedance. At relatively higher frequencies--in this case,
1 kHz and above--imaginary impedance is very small (as confirmed
with in-vivo data), such that total impedance reduces to:
Z t ( .omega. ) = R s + R p 1 + .omega. 2 R p 2 C d 2
##EQU00003##
[0275] As sensor wetting is gradually completed, the double layer
capacitance (C.sub.d) increases. As a result, the total impedance
will decrease because, as indicated in the equation above, total
impedance is inversely proportional to C.sub.d. This is illustrated
in the form of the intercept 1600 on the real impedance axis shown,
e.g., in FIG. 16B. Importantly, the 1 kHz imaginary impedance can
also be used for the same purpose, as it also includes, and is
inversely proportional to, a capacitance component.
[0276] Another marker for low startup detection is Nyquist slope,
which relies solely on the relatively lower-frequency impedance
which, in turn, corresponds to the Warburg impedance component of
total impedance (see, e.g., FIG. 15B). FIG. 22 shows a Nyquist plot
for a normally-functioning sensor, where Arrow A is indicative of
the progression of time, i.e., sensor wear time, starting from t=0.
Thus, EIS at the relatively-lower frequencies is performed right
after sensor insertion (time t=0), which generates real and
imaginary impedance data that is plotted with a first linear fit
2200 having a first (Nyquist) slope. At a time interval after t=0,
a second (lower) frequency sweep is run that produces a second
linear fit 2210 having a second (Nyquist) slope larger than the
first Nyquist slope, and so on. As the sensor becomes more
hydrated, the Nyquist slope increases, and the intercept decrease,
as reflected by the lines 2200, 2210, etc. becoming steeper and
moving closer to the Y-axis. In connection with low startup
detection, clinical data indicates that there is typically a
dramatic increase of Nyquist slope after sensor insertion and
initialization, which is then stabilized to a certain level. One
explanation for this is that, as the sensor is gradually wetted,
the species diffusivity as well as concentration undergo dramatic
change, which is reflected in Warburg impedance.
[0277] In FIG. 23A, the Isig 2230 for a first working electrode WE1
starts off lower than expected (at about 10 nA), and takes some
time to catch up with the Isig 2240 for a second working electrode
WE2. Thus, in this particular example, WE1 is designated as having
a low start-up. The EIS data reflects this low start-up in two
ways. First, as shown in FIG. 23A, the real impedance at 1 kHz
(2235) of WE1 is much higher than the 1 kHz real impedance 2245 of
WE2. Second, when compared to the Nyquist slope for WE2 (FIG. 23C),
the Nyquist slope for WE1 (FIG. 23B) starts out lower, has a larger
intercept 2237, and takes more time to stabilize. As will be
discussed later, these two signatures--the 1 kHz real impedance and
the Nyquist slope--can be used as diagnostic inputs in a fusion
algorithm to decide which of the two electrodes can carry a higher
weight when the fused signal is calculated. In addition, one or
both of these markers may be used in a diagnostic procedure to
determine whether the sensor, as a whole, is acceptable, or whether
it should be terminated and replaced.
[0278] By definition, signal (or Isig) dips refer to instances of
low sensor signal, which are mostly temporary in nature, e.g., on
the order of a few hours. Such low signals may be caused, for
example, by some form of biological occlusion on the sensor
surface, or by pressure applied at the insertion site (e.g., while
sleeping on the side). During this period, the sensor data is
deemed to be unreliable; however, the signal does recover
eventually. In the EIS data, this type of signal dip--as opposed to
one that is caused by a glycemic change in the patient's body--is
reflected in the 1 kHz real impedance data, as shown in FIG.
24.
[0279] Specifically, in FIG. 24, both the Isig 2250 for the first
working electrode WE1 and the Isig 2260 for the second working
electrode WE2 start out at about 25 nA at the far left end (i.e.,
at 6 pm). As time progresses, both Isigs fluctuate, which is
reflective of glucose fluctuations in the vicinity of the sensor.
For about the first 12 hours or so (i.e., until about 6 am), both
Isigs are fairly stable, as are their respective 1 kHz real
impedances 2255, 2265. However, between about 12 and 18
hours--i.e., between 6 am and noon--the Isig 2260 for WE2 starts to
dip, and continues a downward trend for the next several hours,
until about 9 pm. During this period, the Isig 2250 for WE1 also
exhibits some dipping, but Isig 2250 is much more stable, and dips
quite a bit less, than Isig 2260 for WE2. The behavior of the Isigs
for WE1 and WE2 is also reflected in their respective 1 kHz real
impedance data. Thus, as shown in FIG. 24, during the time period
noted above, while the 1 kHz real impedance for WE1 (2255) remains
fairly stable, there is a marked increase in the 1 kHz real
impedance for WE2 (2265).
[0280] By definition, sensitivity loss refers to instances where
the sensor signal (Isig) becomes low and non-responsive for an
extended period of time, and is usually unrecoverable. Sensitivity
loss may occur for a variety of reasons. For example, electrode
poisoning drastically reduces the active surface area of the
working electrode, thereby severely limiting current amplitude.
Sensitivity loss may also occur due to hypoxia, or oxygen deficit,
at the insertion site. In addition, sensitivity loss my occur due
to certain forms of extreme surface occlusion (i.e., a more
permanent form of the signal dip caused by biological or other
factors) that limit the passage of both glucose and oxygen through
the sensor membrane, thereby lowering the number/frequency of the
chemical reactions that generate current in the electrode and,
ultimately, the sensor signal (Isig). It is noted that the various
causes of sensitivity loss mentioned above apply to both short-term
(7-10 day wear) and long term (6 month wear) sensors.
[0281] In the EIS data, sensitivity loss is often preceded by an
increase in the absolute value of phase (|phase|) and of the
imaginary impedance (|imaginary impedance|) at the relatively
higher frequency ranges (e.g., 128 Hz and above, and 1 kHz and
above, respectively). FIG. 25A shows an example of a
normally-functioning glucose sensor where the sensor current 2500
is responsive to glucose--i.e., Isig 2500 tracks glucose
fluctuations--but all relevant impedance outputs, such as, e.g., 1
kHz real impedance 2510, 1 kHz imaginary impedance 2530, and phase
for frequencies at or above about 128 Hz (2520), remain steady, as
they are substantially glucose-independent.
[0282] Specifically, the top graph in FIG. 25A shows that, after
the first few hours, the 1 kHz real impedance 2510 holds fairly
steady at about 5 kOhms (and the 1 kHz imaginary impedance 2530
holds fairly steady at about -400 Ohms). In other words, at 1 kHz,
the real impedance data 2510 and the imaginary impedance data 2530
are substantially glucose-independent, such that they can be used
as signatures for, or independent indicators of, the health,
condition, and ultimately, reliability of the specific sensor under
analysis. However, as mentioned previously, different
impedance-related parameters may exhibit glucose-independence at
different frequency ranges, and the range, in each case, may depend
on the overall sensor design, e.g., electrode type, surface area of
electrode, thickness of membrane, permeability of membrane,
etc.
[0283] Thus, in the example FIG. 25B--for a 90% short tubeless
electrode design--the top graph again shows that sensor current
2501 is responsive to glucose, and that, after the first few hours,
the 1 kHz real impedance 2511 holds fairly steady at about 7.5
kOhms. The bottom graph in FIG. 25B shows real impedance data for
frequencies between 0.1 Hz (2518) and 1 kHz (2511). As can be seen,
the real impedance data at 0.1 Hz (2518) is quite
glucose-dependent. However, as indicated by reference numerals
2516, 2514, and 2512, real impedance becomes more and more
glucose-independent as the frequency increases from 0.1 Hz to 1
kHz, i.e., for impedance data measured at frequencies closer to 1
kHz.
[0284] Returning to FIG. 25A, the middle graph shows that the phase
2520 at the relatively-higher frequencies is substantially
glucose-independent. It is noted, however, that "relatively-higher
frequencies" in connection with this parameter (phase) for the
sensor under analysis means frequencies of 128 Hz and above. In
this regard, the graph shows that the phase for all frequencies
between 128 Hz and 8 kHz is stable throughout the period shown. On
the other hand, as can be seen in the bottom graph of FIG. 25C,
while the phase 2522 at 128 Hz (and above) is stable, the phase
2524 fluctuates--i.e., it becomes more and more glucose-dependent,
and to varying degrees--at frequencies that are increasingly
smaller than 128 Hz. It is noted that the electrode design for the
example of FIG. 25C is the same as that used in FIG. 25B, and that
the top graph in the former is identical to the top graph in the
latter.
[0285] FIG. 26 shows an example of sensitivity loss due to oxygen
deficiency at the insertion site. In this case, the insertion site
becomes oxygen deprived just after day 4 (designated by dark
vertical line in FIG. 26), causing the sensor current 2600 to
become low and non-responsive. The 1 kHz real impedance 2610
remains stable, indicating no physical occlusion on the sensor.
However, as shown by the respective downward arrows, changes in the
relatively higher-frequency phase 2622 and 1 kHz imaginary
impedance 2632 coincide with loss in sensitivity, indicating that
this type of loss is due to an oxygen deficit at the insertion
site. Specifically, FIG. 26 shows that the phase at higher
frequencies (2620) and the 1 kHz imaginary impedance (2630) become
more negative prior to the sensor losing sensitivity--indicated by
the dark vertical line--and continue their downward trend as the
sensor sensitivity loss continues. Thus, as noted above, this
sensitivity loss is preceded, or predicted, by an increase in the
absolute value of phase (|phase|) and of the imaginary impedance
(|imaginary impedance|) at the relatively higher frequency ranges
(e.g., 128 Hz and above, and 1 kHz and above, respectively).
[0286] The above-described signatures may be verified by in-vitro
testing, an example of which is shown in FIG. 27. FIG. 27 shows the
results of in-vitro testing of a sensor, where oxygen deficit at
different glucose concentrations is simulated. In the top graph,
the Isig fluctuates with the glucose concentration as the latter is
increased from 100 mg/dl (2710) to 200 mg/dl (2720), 300 mg/dl
(2730), and 400 mg/dl (2740), and then decreased back down to 200
and/dl (2750). In the bottom graph, the phase at the
relatively-higher frequencies is generally stable, indicating that
it is glucose-independent. However, at very low oxygen
concentrations, such as, e.g., at 0.1% O.sub.2, the relatively
high-frequency phase fluctuates, as indicated by the encircled
areas and arrows 2760, 2770. It is noted that the magnitude and/or
direction (i.e., positive or negative) of fluctuation depend on
various factors. For example, the higher the ratio of glucose
concentration to oxygen concentration, the higher the magnitude of
the fluctuation in phase. In addition, the specific sensor design,
as well as the age of the sensor (i.e., as measured by time after
implant), affect such fluctuations. Thus, e.g., the older a sensor
is, the more susceptible it is to perturbations.
[0287] FIGS. 28A-28D show another example of oxygen deficiency-led
sensitivity loss with redundant working electrodes WE1 and WE2. As
shown in FIG. 28A, the 1 kHz real impedance 2810 is steady, even as
sensor current 2800 fluctuates and eventually becomes
non-responsive. Also, as before, the change in 1 kHz imaginary
impedance 2820 coincides with the sensor's loss of sensitivity. In
addition, however, FIG. 28B shows real impedance data and imaginary
impedance data (2830 and 2840, respectively) at 0.105 Hz. The
latter, which may be more commonly referred to as "0.1 Hz data",
indicates that, whereas imaginary impedance at 0.1 Hz appears to be
fairly steady, 0.1 Hz real impedance 2830 increases considerably as
the sensor loses sensitivity. Moreover, as shown in FIG. 28C, with
loss of sensitivity due to oxygen deficiency, V.sub.cntr 2850 rails
to 1.2 Volts.
[0288] In short, the diagrams illustrate the discovery that oxygen
deficiency-led sensitivity loss is coupled with lower 1 kHz
imaginary impedance (i.e., the latter becomes more negative),
higher 0.105 Hz real impedance (i.e., the latter becomes more
positive), and V.sub.cntr rail. Moreover, the oxygen-deficiency
process and V.sub.cntr-rail are often coupled with the increase of
the capacitive component in the electrochemical circuit. It is
noted that, in some of the diagnostic procedures to be described
later, the 0.105 Hz real impedance may not be used, as it appears
that this relatively lower-frequency real impedance data may be
analyte-dependent.
[0289] Finally, in connection with the example of FIGS. 28A-28D, it
is noted that 1 kHz or higher-frequency impedance measurement
typically causes EIS-induced spikes in the Isig. This is shown in
FIG. 28D, where the raw Isig for WE2 is plotted against time. The
drastic increase of Isig when the spike starts is a non-Faradaic
process, due to double-layer capacitance charge. Thus, oxygen
deficiency-led sensitivity loss may also be coupled with higher
EIS-induced spikes, in addition to lower 1 kHz imaginary impedance,
higher 0.105 Hz real impedance, and V.sub.cntr rail, as discussed
above.
[0290] FIG. 29 illustrates another example of sensitivity loss.
This case may be thought of as an extreme version of the Isig dip
described above in connection with FIG. 24. Here, the sensor
current 2910 is observed to be low from the time of insertion,
indicating that there was an issue with an insertion procedure
resulting in electrode occlusion. The 1 kHz real-impedance 2920 is
significantly higher, while the relatively higher-frequency phase
2930 and the 1 kHz imaginary impedance 2940 are both shifted to
much more negative values, as compared to the same parameter values
for the normally-functioning sensor shown in FIG. 25A. The shift in
the relatively higher-frequency phase 2930 and 1 kHz imaginary
impedance 2940 indicates that the sensitivity loss may be due to an
oxygen deficit which, in turn, may have been caused by an occlusion
on the sensor surface.
[0291] FIGS. 30A-30D show data for another redundant sensor, where
the relative differences in 1 kHz real impedance and 1 kHz
imaginary impedance, as well as 0.1 Hz real impedance, between two
or more working electrodes may be used for the detection of
sensitivity loss due to biofouling. In this example, WE1 exhibits
more sensitivity loss than WE2, as is evident from the higher 1 kHz
real impedance 3010, lower 1 kHz imaginary impedance 3020, and much
higher real impedance at 0.105 kHz (3030) for WE2. In addition,
however, in this example, V.sub.cntr 3050 does not rail. Moreover,
as shown in FIG. 30D, the height of the spikes in the raw Isig data
does not change much as time progresses. This indicates that, for
sensitivity loss due to biofouling, V.sub.cntr-rail and the
increase in spike height are correlated. In addition, the fact that
the height of the spikes in the raw Isig data does not change much
with time indicates that the capacitive component of the circuit
does not change significantly with time, such that sensitivity loss
due to biofouling is related to the resistance component of the
circuit (i.e., diffusion).
[0292] Various of the above-described impedance-related parameters
may be used, either individually or in combination, as inputs into:
(1) EIS-based sensor diagnostic procedures; and/or (2) fusion
algorithms for generating more reliable sensor glucose values. With
regard to the former, FIG. 31 illustrates how EIS-based data--i.e.,
impedance-related parameters, or characteristics--may be used in a
diagnostic procedure to determine, in real time, whether a sensor
is behaving normally, or whether it should be replaced.
[0293] The diagnostic procedure illustrated in the flow diagram of
FIG. 31 is based on the collection of EIS data on a periodic basis,
such as, e.g., hourly, every half hour, every 10 minutes, or at any
other interval--including continuously--as may be appropriate for
the specific sensor under analysis. At each such interval, EIS may
be run for an entire frequency spectrum (i.e., a "full sweep"), or
it may be run for a selected frequency range, or even at a single
frequency. Thus, for example, for an hourly data collection scheme,
EIS may be performed at frequencies in the .mu.Hz to MHz range, or
it may be run for a narrower range of frequencies, such as, e.g.,
between about 0.1 Hz and about 8 kHz, as discussed hereinabove. In
embodiments of the invention, EIS data acquisition may be
implemented alternatingly between a full sweep and an
narrower-range spectrum, or in accordance with other schemes.
[0294] The temporal frequency of EIS implementation and data
collection may be dictated by various factors. For example, each
implementation of EIS consumes a certain amount of power, which is
typically provided by the sensor's battery, i.e., the battery
running the sensor electronics, including the ASIC which is
described later. As such, battery capacity, as well as the
remaining sensor life, may help determine the number of times EIS
is run, as well as the breadth of frequencies sampled for each such
run. In addition, embodiments of the invention envision situations
that may require that an EIS parameter at a specific frequency
(e.g., real impedance at 1 kHz) be monitored based on a first
schedule (e.g., once every few seconds, or minutes), while other
parameters, and/or the same parameter at other frequencies, can be
monitored based on a second schedule (e.g., less frequently). In
these situations, the diagnostic procedure can be tailored to the
specific sensor and requirements, such that battery power may be
preserved, and unnecessary and/or redundant EIS data acquisition
may be avoided.
[0295] It is noted that, in embodiments of the invention, a
diagnostic procedure, such as the one shown in FIG. 31, entails a
series of separate "tests" which are implemented in order to
perform real-time monitoring of the sensor. The multiple tests, or
markers--also referred to as "multi markers"--are implemented
because each time EIS is run (i.e., each time an EIS procedure is
performed), data may be gathered about a multiplicity of
impedance-based parameters, or characteristics, which can be used
to detect sensor condition or quality, including, e.g., whether the
sensor has failed or is failing. In performing sensor diagnostics,
sometimes, there can be a diagnostic test that may indicate a
failure, whereas other diagnostic(s) may indicate no failure.
Therefore, the availability of multiple impedance-related
parameters, and the implementation of a multi-test procedure, are
advantageous, as some of the multiplicity of tests may act as
validity checks against some of the other tests. Thus, real-time
monitoring using a multi-marker procedure may include a certain
degree of built-in redundancy.
[0296] With the above in mind, the logic of the diagnostic
procedure shown in FIG. 31 begins at 3100, after the sensor has
been inserted/implanted, and an EIS run has been made, so as to
provide the EIS data as input. At 3100, using the EIS data as
input, it is first determined whether the sensor is still in place.
Thus, if the |Z| slope is found to be constant across the tested
frequency band (or range), and/or the phase angle is about
-90.degree., it is determined that the sensor is no longer in
place, and an alert is sent, e.g., to the patient/user, indicating
that sensor pullout has occurred. The specific parameters (and
their respective values) described herein for detecting sensor
pullout are based on the discovery that, once the sensor is out of
the body and the membrane is no longer hydrated, the impedance
spectrum response appears just like a capacitor.
[0297] If it is determined that the sensor is still in place, the
logic moves to step 3110 to determine whether the sensor is
properly initialized. As shown, the "Init. Check" is performed by
determining: (i) whether |(Z.sub.n-Z.sub.1)/Z.sub.1|>30% at 1
kHz, where Z.sub.1 is the real impedance measured at a first time,
and Z.sub.n is the measured impedance at the next interval, at
discussed above; and (2) whether the phase angle change is greater
than 10.degree. at 0.1 Hz. If the answer to either one of the
questions is "yes", then the test is satisfactory, i.e., the Test 1
is not failed. Otherwise, the Test 1 is marked as a failure.
[0298] At step 3120, Test 2 asks whether, at a phase angle of
-45.degree., the difference in frequency between two consecutive
EIS runs (f.sub.2-f.sub.1) is greater than 10 Hz. Again, a "No"
answer is marked as a fail; otherwise, Test 2 is satisfactorily
met.
[0299] Test 3 at step 3130 is a hydration test. Here, the inquiry
is whether the current impedance Z.sub.n is less than the
post-initialization impedance Z.sub.pi at 1 kHz. If it is, then
this test is satisfied; otherwise, Test 3 is marked as a fail. Test
4 at step 3140 is also a hydration test, but this time at a lower
frequency. Thus, this test asks whether Z.sub.n is less than 300
kOhms at 0.1 Hz during post-initialization sensor operation. Again,
a "No" answer indicates that the sensor has failed Test 4.
[0300] At step 3150, Test 5 inquires whether the low-frequency
Nyquist slope is globally increasing from 0.1 Hz to 1 Hz. As
discussed previously, for a normally-operating sensor, the
relatively lower-frequency Nyquist slope should be increasing over
time. Thus, this test is satisfied if the answer to the inquiry is
"yes"; otherwise, the test is marked as failed.
[0301] Step 3160 is the last test for this embodiment of the
diagnostic procedure. Here, the inquiry is whether real impedance
is globally decreasing. Again, as was discussed previously, in a
normally-operating sensor, it is expected that, as time goes by,
the real impedance should be decreasing. Therefore, a "Yes" answer
here would mean that the sensor is operating normally; otherwise,
the sensor fails Test 6.
[0302] Once all 6 tests have been implemented, a decision is made
at 3170 as to whether the sensor is operating normally, or whether
it has failed. In this embodiment, a sensor is determined to be
functioning normally (3172) if it passes at least 3 out of the 6
tests. Put another way, in order to be determined to have failed
(3174), the sensor must fail at least 4 out of the 6 tests. In
alternative embodiments, a different rule may be used to assess
normal operation versus sensor failure. In addition, in embodiments
of the invention, each of the tests may be weighted, such that the
assigned weight reflects, e.g., the importance of that test, or of
the specific parameter(s) queried for that test, in determining
overall sensor operation (normal vs. failed). For example, one test
may be weighted twice as heavily as another, but only half as
heavily as a third test, etc.
[0303] In other alternative embodiments, a different number of
tests and/or a different set of EIS-based parameters for each test
may be used. FIGS. 32A and 32B show an example of a diagnostic
procedure for real-time monitoring that includes 7 tests. Referring
to FIG. 32A, the logic begins at 3200, after the sensor has been
inserted/implanted, and an EIS procedure has been performed, so as
to provide the EIS data as input. At 3200, using the EIS data as
input, it is first determined whether the sensor is still in place.
Thus, if the |Z| slope is found to be constant across the tested
frequency band (or range), and/or the phase angle is about
-90.degree., it is determined that the sensor is no longer in
place, and an alert is sent, e.g., to the patient/user, indicating
that sensor pullout has occurred. If, on the other hand, the sensor
is determined to be in place, the logic moves to initiation of
diagnostic checks (3202).
[0304] At 3205, Test 1 is similar to Test 1 of the diagnostic
procedure discussed above in connection with FIG. 31, except that
the instant Test 1 specifies that the later measurement Z.sub.n is
taken 2 hours after the first measurement. As such, in this
example, Z.sub.n=Z.sub.2hr. More specifically, Test 1 compares the
real impedance 2 hours after (sensor implantation and)
initialization to the pre-initialization value. Similarly, the
second part of Test 1 asks whether the difference between the phase
2 hours after initialization and the pre-initialization phase is
greater than 10.degree. at 0.1 Hz. As before, if the answer to
either one of the inquiries is affirmative, then it is determined
that the sensor is hydrated normally and initialized, and Test 1 is
satisfied; otherwise, the sensor fails this test. It should be
noted that, even though the instant test inquires about impedance
and phase change 2 hours after initialization, the time interval
between any two consecutive EIS runs may be shorter or longer,
depending on a variety of factors, including, e.g., sensor design,
the level of electrode redundancy, the degree to which the
diagnostic procedure includes redundant tests, battery power,
etc.
[0305] Moving to 3210, the logic next performs a sensitivity-loss
check by inquiring whether, after a 2-hour interval (n+2), the
percentage change in impedance magnitude at 1 kHz, as well as that
in the Isig, is greater than 30%. If the answer to both inquiries
is "yes", then it is determined that the sensor is losing
sensitivity and, as such, Test 2 is determined to be failed. It is
noted that, although Test 2 is illustrated herein based on a
preferred percentage difference of 30%, in other embodiments, the
percentage differences in the impedance magnitude at 1 kHz and in
the Isig may fall within the range 10%-50% for purposes of
conducting this test.
[0306] Test 3 (at 3220) is similar to Test 5 of the algorithm
illustrated in FIG. 31. Here, as before, the question is whether
the low-frequency Nyquist slope is globally increasing from 0.1 Hz
to 1 Hz. If it is, then this test is passed; otherwise, the test is
failed. As shown in 3220, this test is also amenable to setting a
threshold, or an acceptable range, for the percent change in the
low-frequency Nyquist slope, beyond which the sensor may be deemed
to be failed or, at the very least, may trigger further diagnostic
testing. In embodiments of the invention, such threshold
value/acceptable range for the percent change in low-frequency
Nyquist slope may fall within a range of about 2% to about 20%. In
some preferred embodiments, the threshold value may be about
5%.
[0307] The logic next moves to 3230, which is another low-frequency
test, this time involving the phase and the impedance magnitude.
More specifically, the phase test inquires whether the phase at 0.1
Hz is continuously increasing over time. If it is, then the test is
failed. As with other tests where the parameter's trending is
monitored, the low-frequency phase test of Test 4 is also amenable
to setting a threshold, or an acceptable range, for the percent
change in the low-frequency phase, beyond which the sensor may be
deemed to be failed or, at the very least, raise a concern. In
embodiments of the invention, such threshold value/acceptable range
for the percent change in low-frequency phase may fall within a
range of about 5% to about 30%. In some preferred embodiments, the
threshold value may be about 10%.
[0308] As noted, Test 4 also includes a low-frequency impedance
magnitude test, where the inquiry is whether the impedance
magnitude at 0.1 Hz is continuously increasing over time. If it is,
then the test is failed. It is noted that Test 4 is considered
"failed" if either the phase test or the impedance magnitude test
is failed. The low-frequency impedance magnitude test of Test 4 is
also amenable to setting a threshold, or an acceptable range, for
the percent change in the low-frequency impedance magnitude, beyond
which the sensor may be deemed to be failed or, at the very least,
raise a concern. In embodiments of the invention, such threshold
value/acceptable range for the percent change in low-frequency
impedance magnitude may fall within a range of about 5% to about
30%. In some preferred embodiments, the threshold value may be
about 10%, where the range for impedance magnitude in normal
sensors is generally between about 100 KOhms and about 200
KOhms.
[0309] Test 5 (at 3240) is another sensitivity loss check that may
be thought of as supplemental to Test 2. Here, if both the
percentage change in the Isig and the percentage change in the
impedance magnitude at 1 kHz are greater than 30%, then it is
determined that the sensor is recovering from sensitivity loss. In
other words, it is determined that the sensor had previously
undergone some sensitivity loss, even if the sensitivity loss was
not, for some reason, detected by Test 2. As with Test 2, although
Test 5 is illustrated based on a preferred percentage difference of
30%, in other embodiments, the percentage differences in the Isig
and the impedance magnitude at 1 kHz may fall within the range
10%-50% for purposes of conducting this test.
[0310] Moving to 3250, Test 6 provides a sensor functionality test
with specific failure criteria that have been determined based on
observed data and the specific sensor design. Specifically, in one
embodiment, a sensor may be determined to have failed and, as such,
to be unlikely to respond to glucose, if at least two out of the
following three criteria are met: (1) Isig is less than 10 nA; and
(2) the imaginary impedance at 1 kHz is less than -1500 Ohm; and
(3) the phase at 1 kHz is less than -15.degree.. Thus, Test 6 is
determined to have been passed if any two of (1)-(3) are not met.
It is noted that, in other embodiments, the Isig prong of this test
may be failed if the Isig is less than about 5 nA to about 20 nA.
Similarly, the second prong may be failed if the imaginary
impedance at 1 kHz is less than about -1000 Ohm to about -2000
Ohms. Lastly, the phase prong may be failed if the phase at 1 kHz
is less than about -10.degree. to about -20.degree..
[0311] Lastly, step 3260 provides another sensitivity check,
wherein the parameters are evaluated at low frequency. Thus, Test 7
inquires whether, at 0.1 Hz, the magnitude of the difference
between the ratio of the imaginary impedance to the Isig (n+2), on
the one hand, and the pervious value of the ratio, on the other, is
larger than 30% of the magnitude of the previous value of the
ratio. If it is, then the test is failed; otherwise, the test is
passed. Here, although Test 7 is illustrated based on a preferred
percentage difference of 30%, in other embodiments, the percentage
difference may fall within the range 10%-50% for purposes of
conducting this test.
[0312] Once all 7 tests have been implemented, a decision is made
at 3270 as to whether the sensor is operating normally, or whether
an alert should be sent out, indicating that the sensor has failed
(or may be failing). As shown, in this embodiment, a sensor is
determined to be functioning normally (3272) if it passes at least
4 out of the 7 tests. Put another way, in order to be determined to
have failed, or to at least raise a concern (3274), the sensor must
fail at least 4 out of the 7 tests. If it is determined that the
sensor is "bad" (3274), an alert to that effect may be sent, e.g.,
to the patient/user. As noted previously, in alternative
embodiments, a different rule may be used to assess normal
operation versus sensor failure/concern. In addition, in
embodiments of the invention, each of the tests may be weighted,
such that the assigned weight reflects, e.g., the importance of
that test, or of the specific parameter(s) queried for that test,
in determining overall sensor operation (normal vs. failed).
[0313] As was noted previously, in embodiments of the invention,
various of the above-described impedance-related parameters may be
used, either individually or in combination, as inputs into one or
more fusion algorithms for generating more reliable sensor glucose
values. Specifically, it is known that, unlike a single-sensor
(i.e., a single-working-electrode) system, multiple sensing
electrodes provide higher-reliability glucose readouts, as a
plurality of signals, obtained from two or more working electrodes,
may be fused to provide a single sensor glucose value. Such signal
fusion utilizes quantitative inputs provided by EIS to calculate
the most reliable output sensor glucose value from the redundant
working electrodes. It is noted that, while the ensuing discussion
may describe various fusion algorithms in terms of a first working
electrode (WE1) and a second working electrode (WE2) as the
redundant electrodes, this is by way of illustration, and not
limitation, as the algorithms and their underlying principles
described herein are applicable to, and may be used in, redundant
sensor systems having more than 2 working electrodes.
[0314] FIGS. 33A and 33B show top-level flowcharts for two
alternative methodologies, each of which includes a fusion
algorithm. Specifically, FIG. 33A is a flowchart involving a
current (Isig)-based fusion algorithm, and FIG. 33B is a flowchart
directed to sensor glucose (SG) fusion. As may be seen from the
diagrams, the primary difference between the two methodologies is
the time of calibration. Thus, FIG. 33A shows that, for Isig
fusion, calibration 3590 is performed after the fusion 3540 is
completed. That is, redundant Isigs from WE1 to WEn are fused into
a single Isig 3589, which is then calibrated to produce a single
sensor glucose value 3598. For SG fusion, on the other hand,
calibration 3435 is completed for each individual Isig from WE1 to
WEn to produce calibrated SG values (e.g., 3436, 3438) for each of
the working electrodes. Thus, SG fusion algorithms provide for
independent calibration of each of the plurality of Isigs, which
may be preferred in embodiments of the invention. Once calibrated,
the plurality of calibrated SG values is fused into a single SG
value 3498.
[0315] It is important to note that each of flowcharts shown in
FIGS. 33A and 33B includes a spike filtering process (3520, 3420).
As was described above in the discussion relating to sensitivity
loss, 1 kHz or higher-frequency impedance measurements typically
cause EIS-induced spikes in the Isig. Therefore, once an EIS
procedure has been performed for each of the electrodes WE1 to WEn,
for both SG fusion and Isig fusion, it is preferable to first
filter the Isigs 3410, 3412, etc. and 3510, 3512, etc. to obtain
respective filtered Isigs 3422, 3424, etc. and 3522, 3524, etc. The
filtered Isigs are then either used in Isig fusion, or first
calibrated and then used in SG fusion, as detailed below. As will
become apparent in the ensuing discussion, both fusion algorithms
entail calculation and assignment of weights based on various
factors.
[0316] FIG. 34 shows the details of the fusion algorithm 3440 for
SG fusion. Essentially, there are four factors that need to be
checked before the fusion weights are determined. First, integrity
check 3450 involves determining whether each of the following
parameters is within specified ranges for normal sensor operation
(e.g., predetermined lower and upper thresholds): (i) Isig; (ii) 1
kHz real and imaginary impedances; (iii) 0.105 Hz real and
imaginary impedances; and (iv) Nyquist slope. As shown, integrity
check 3450 includes a Bound Check 3452 and a Noise Check 3456,
wherein, for each of the Checks, the above-mentioned parameters are
used as input parameters. It is noted that, for brevity, real
and/or imaginary impedances, at one or more frequencies, appear on
FIGS. 33A-35 simply as "Imp" for impedance. In addition, both real
and imaginary impedances may be calculated using impedance
magnitude and phase (which is also shown as an input on FIGS. 33A
and 33B).
[0317] The output from each of the Bound Check 3452 and the Noise
Check 3458 is a respective reliability index (RI) for each of the
redundant working electrodes. Thus, the output from the Bound Check
includes, e.g., RI_bound_We.sub.1 (3543) and RI_bound_We.sub.2
(3454). Similarly, for the Noise Check, the output includes, e.g.,
RI_noise_We.sub.1 (3457) and RI_noise_We.sub.2 (3458). The bound
and noise reliability indices for each working electrode are
calculated based on compliance with the above-mentioned ranges for
normal sensor operation. Thus, if any of the parameters falls
outside the specified ranges for a particular electrode, the
reliability index for that particular electrode decreases.
[0318] It is noted that the threshold values, or ranges, for the
above-mentioned parameters may depend on various factors, including
the specific sensor and/or electrode design. Nevertheless, in one
preferred embodiment, typical ranges for some of the
above-mentioned parameters may be, e.g., as follows: Bound
threshold for 1 kHz real impedance=[0.3e+4 2e+4]; Bound threshold
for 1 kHz imaginary impedance=[-2e+3, 0]; Bound threshold for 0.105
Hz real impedance=[2e+4 7e+4]; Bound threshold for 0.105 Hz
imaginary impedance=[-2e+5-0.25e+5]; and Bound threshold for
Nyquist slope=[2 5]. Noise may be calculated, e.g., using second
order central difference method where, if noise is above a certain
percentage (e.g., 30%) of median value for each variable buffer, it
is considered to be out of noise bound.
[0319] Second, sensor dips may be detected using sensor current
(Isig) and 1 kHz real impedance. Thus, as shown in FIG. 34, Isig
and "Imp" are used as inputs for dips detection 3460. Here, the
first step is to determine whether there is any divergence between
Isigs, and whether any such divergence is reflected in 1 kHz real
impedance data. This may be accomplished by using mapping 3465
between the Isig similarity index (RI_sim_isig12) 3463 and the 1
kHz real impedance similarity index (RI_sim_imp12) 3464. This
mapping is critical, as it helps avoid false positives in instances
where a dip is not real. Where the Isig divergence is real, the
algorithm will select the sensor with the higher Isig.
[0320] In accordance with embodiments of the invention, the
divergence/convergence of two signals (e.g., two Isigs, or two 1
kHz real impedance data points) can be calculated as follows:
diff_va1=abs(va1-(va1+va2)/2);
diff_va2=abs(va2-(va1+va2)/2);
RI_sim=1-(diff_va1+diff_va2)/(mean(abs(va1+va2))/4)
where va1 and va2 are two variables, and RI_sim (similarity index)
is the index to measure the convergence or divergence of the
signals. In this embodiment, RI_sim must be bound between 0 and 1.
Therefore, if RI_sim as calculated above is less than 0, it will be
set to 0, and if it is higher than 1, it will be set to 1.
[0321] The mapping 3465 is performed by using ordinary linear
regression (OLR). However, when OLR does not work well, a robust
median slope linear regression (RMSLR) can be used. For Isig
similarity index and 1 kHz real impedance index, for example, two
mapping procedures are needed: (i) Map Isig similarity index to 1
kHz real impedance similarity index; and (ii) map 1 kHz real
impedance similarity index to Isig similarity index. Both mapping
procedures will generate two residuals: res12 and res21. Each of
the dip reliability indices 3467, 3468 can then be calculated
as:
RI_dip=1-(res12+res21)/(RI_sim_isig+RI_sim_1K_real_impedance).
[0322] The third factor is sensitivity loss 3470, which may be
detected using 1 kHz imaginary impedance trending in, e.g., the
past 8 hours. If one sensor's trending turns negative, the
algorithm will rely on the other sensor. If both sensors lose
sensitivity, then a simple average is taken. Trending may be
calculated by using a strong low-pass filter to smooth over the 1
kHz imaginary impedance, which tends to be noisy, and by using a
correlation coefficient or linear regression with respect to time
during, e.g., the past 8 hours to determine whether the correlation
coefficient is negative or the slope is negative. Each of the
sensitivity loss reliability indices 3473, 3474 is then assigned a
binary value of 1 or 0.
[0323] The total reliability index (RI) for each of we1, we2, . . .
wen is calculated as follows:
RI_we 1 = RI_dip _we 1 .times. RI_sensitivity _loss _we 1 .times.
RI_bound _we 1 .times. RI_noise _we 1 RI_we 2 = RI_dip _we 2
.times. RI_sensitivity _loss _we 2 .times. RI_bound _we 2 .times.
RI_noise _we 2 RI_we 3 = RI_dip _we 3 .times. RI_sensitivity _loss
_we 3 .times. RI_bound _we 3 .times. RI_noise _we 3 RI_we 4 =
RI_dip _we 4 .times. RI_sensitivity _loss _we 4 .times. RI_bound
_we 4 .times. RI_noise _we 4 RI_we n = RI_dip _we n .times.
RI_sensitivity _loss _we n .times. RI_bound _we n .times. RI_noise
_we n ##EQU00004##
[0324] Having calculated the respective reliability indices of the
individual working electrodes, the weight for each of the
electrodes may be calculated as follow:
weight_we 1 = RI_we 1 / ( RI_we 1 + RI_we 2 + RI_we 3 + RI_we 4 + +
RI_we n ) ##EQU00005## weight_we 2 = RI_we 2 / ( RI_we 1 + RI_we 2
+ RI_we 3 + RI_we 4 + + RI_we n ) ##EQU00005.2## weight_we 3 =
RI_we 3 / ( RI_we 1 + RI_we 2 + RI_we 3 + RI_we 4 + + RI_we n )
##EQU00005.3## weight_we 4 = RI_we 4 / ( RI_we 1 + RI_we 2 + RI_we
3 + RI_we 4 + + RI_we n ) ##EQU00005.4## ##EQU00005.5## weight_we n
= RI_we n / ( RI_we 1 + RI_we 2 + RI_we 3 + RI_we 4 + + RI_we n )
##EQU00005.6##
[0325] Based on the above, the fused SG 3498 is then calculated as
follows:
SG=weight_we.sub.1.times.SG_we.sub.1+weight_we.sub.2.times.SG_we.sub.2+w-
eight_we.sub.3.times.SG_we.sub.3+weight_we.sub.4.times.SG_we.sub.4+
. . . +weight_we.sub.n.times.SG_we.sub.n
[0326] The last factor relates to artifacts in the final sensor
readout, such as may be caused by instant weight change of sensor
fusion. This may be avoided by either applying a low-pass filter
3480 to smooth the RI for each electrode, or by applying a low-pass
filter to the final SG. When the former is used, the filtered
reliability indices--e.g., RI_We1* and RI_We2* (3482, 3484)--are
used in the calculation of the weight for each electrode and,
therefore, in the calculation of the fused SG 3498.
[0327] FIG. 35 shows the details of the fusion algorithm 3540 for
Isig fusion. As can be seen, this algorithm is substantially
similar to the one shown in FIG. 34 for SG fusion, with two
exceptions. First, as was noted previously, for Isig fusion,
calibration constitutes the final step of the process, where the
single fused Isig 3589 is calibrated to generate a single sensor
glucose value 3598. See also FIG. 33B. Second, whereas SG fusion
uses the SG values for the plurality of electrodes to calculate the
final SG value 3498, the fused Isig value 3589 is calculated using
the filtered Isigs (3522, 3524, and so on) for the plurality of
electrodes.
[0328] In one closed-loop study involving a non-diabetic
population, it was found that the above-described fusion algorithms
provided considerable improvements in the Mean Absolute Relative
Difference (MARD) both on Day 1, when low start-up issues are most
significant and, as such, may have a substantial impact on sensor
accuracy and reliability, and overall (i.e., over a 7-day life of
the sensor). The study evaluated data for an 88% distributed layout
design with high current density (nominal) plating using three
different methodologies: (1) calculation of one sensor glucose
value (SG) via fusion using Medtronic Minimed's Ferrari Algorithm
1.0 (which is a SG fusion algorithm as discussed above); (2)
calculation of one SG by identifying the better ISIG value using 1
kHz EIS data (through the Isig fusion algorithm discussed above);
and (3) calculation of one SG by using the higher ISIG value (i.e.,
without using EIS). The details of the data for the study are
presented below:
TABLE-US-00001 (1) SG based on Ferrari 1.0 Alg for 88% distributed
layout with high current density (nominal) plating Day 1 2 3 4 5 6
7 Total Mean-ARD Percentage 040-080 19.39 17.06 22.27 17.50 37.57
11.43 19.69 080-120 19.69 09.18 09.34 08.64 10.01 08.31 11.33 11.56
120-240 19.01 17.46 12.44 07.97 11.75 08.82 12.15 12.92 240-400
10.25 08.36 14.09 10.86 12.84 22.70 12.88 Total 19.52 11.71 10.14
09.30 10.83 09.49 11.89 12.28 Mean-Absolute Bias (sg-bg) 040-080
14.86 11.78 15.81 11.07 29.00 07.26 14.05 080-120 19.53 09.37 09.49
08.78 09.88 08.44 11.61 11.62 120-240 30.04 29.73 19.34 14.45 18.25
12.66 18.89 20.60 240-400 26.75 22.23 39.82 29.00 33.00 61.36 35.19
Total 21.62 15.20 12.79 13.21 12.04 10.84 15.04 14.79 Mean-Signed
Bias (sg-bg) 040-080 12.15 09.78 15.81 11.07 29.00 07.26 13.01
080-120 -04.45 -04.92 -00.90 00.18 01.21 00.85 00.03 -01.44 120-240
-10.18 -27.00 -16.89 -02.91 -05.40 -01.24 -11.58 -10.71 240-400
11.25 02.23 -00.07 -27.00 -33.00 -61.36 -10.29 Total -04.81 -09.77
-05.09 -00.23 -00.22 00.67 -04.98 -03.56 Eval Points 040-080 007
004 000 002 006 003 004 026 080-120 090 064 055 055 067 056 047 434
120-240 028 025 022 021 016 032 026 170 240-400 000 002 004 008 003
001 002 020 Total 125 095 081 086 092 092 079 650
TABLE-US-00002 (2) SG based on better ISIG using 1 kHz EIS for 88%
distributed layout with high current density (nominal) plating Day
1 2 3 4 5 6 7 Total Mean-ARD Percentage 040-080 16.66 18.78 21.13
16.21 43.68 09.50 18.14 080-120 16.22 11.96 08.79 10.49 09.75 08.04
10.34 11.36 120-240 15.08 17.50 12.68 07.72 08.74 08.84 13.02 12.16
240-400 07.66 06.42 11.10 07.52 15.95 21.13 09.84 Total 15.96 13.70
09.92 09.95 09.96 09.40 11.31 11.83 Mean-Absolute Bias (sg-bg)
040-080 12.71 13.00 15.00 10.17 33.50 06.00 12.83 080-120 15.70
12.17 08.57 10.89 09.62 08.26 10.49 11.32 120-240 24.43 29.82 19.43
13.79 14.60 12.97 20.27 19.58 240-400 20.00 17.00 32.50 20.00 41.00
60.00 27.29 Total 17.72 17.20 12.56 13.55 10.95 11.21 14.12 14.20
Mean-Signed Bias (sg-bg) 040-080 08.71 13.00 15.00 10.17 33.50
06.00 11.67 080-120 -04.30 -08.62 -01.11 -03.64 02.52 00.40 -01.56
-02.52 120-240 -11.30 -29.64 -17.09 -08.74 -10.87 -07.23 -15.09
-14.05 240-400 20.00 00.50 09.50 -17.33 -41.00 -60.00 -03.18 Total
-05.30 -12.56 -06.20 -03.63 -00.10 -02.29 -06.35 -05.21 Eval Points
040-080 007 004 000 001 006 002 004 024 080-120 082 053 044 045 058
043 041 366 120-240 030 022 023 019 015 030 022 161 240-400 000 002
004 006 003 001 001 017 Total 119 081 071 071 082 076 068 568
TABLE-US-00003 (3) SG based on higher ISIG for 88% distributed
layout with high current density (nominal) plating Day 1 2 3 4 5 6
7 Total Mean-ARD Percentage 040-080 17.24 19.13 21.13 17.31 43.68
10.38 18.79 080-120 17.69 11.77 09.36 10.70 10.19 08.34 10.68 11.86
120-240 16.80 17.63 13.04 07.38 09.04 08.52 13.25 12.50 240-400
07.47 06.02 10.85 07.52 15.95 21.13 09.63 Total 17.44 13.60 10.37
10.00 10.40 09.36 11.66 12.26 Mean-Absolute Bias (sg-bg) 040-080
13.14 13.25 15.00 11.00 33.50 06.50 13.29 080-120 17.23 11.98 09.22
11.02 10.08 08.59 10.86 11.85 120-240 27.40 30.09 19.75 13.26 14.93
12.45 20.65 20.09 240-400 19.50 16.00 32.00 20.00 41.00 60.00 26.82
Total 19.53 17.09 13.00 13.35 11.37 11.18 14.53 14.67 Mean-Signed
Bias (sg-bg) 040-080 08.29 12.75 15.00 11.00 33.50 06.50 11.79
080-120 -04.72 -08.83 -02.35 -01.56 01.75 -00.18 -01.52 -02.70
120-240 -15.13 -29.73 -17.67 -08.42 -11.47 -07.03 -15.43 -14.86
240-400 19.50 01.50 06.33 -17.33 -41.00 -60.00 -04.12 Total -06.57
-12.70 -07.11 -02.46 -00.63 -02.56 -06.47 -05.57 Eval Points
040-080 007 004 000 001 006 002 004 024 080-120 083 054 046 048 060
044 042 377 120-240 030 022 024 019 015 031 023 164 240-400 000 002
004 006 003 001 001 017 Total 120 082 074 074 084 078 070 582
[0329] With the above data, it was found that, with the first
approach, the MARD (%) on Day 1 was 19.52%, with an overall MARD of
12.28%. For the second approach, the Day -1 MARD was 15.96% and the
overall MARD was 11.83%. Lastly, for the third approach, the MARD
was 17.44% on Day 1, and 12.26% overall. Thus, for this design with
redundant electrodes, it appears that calculation of SG based on
the better ISIG using 1 kHz EIS (i.e., the second methodology)
provides the greatest advantage. Specifically, the lower Day -1
MARD may be attributable, e.g., to better low start-up detection
using EIS. In addition, the overall MARD percentages are more than
1% lower than the overall average MARD of 13.5% for WE1 and WE2 in
this study. It is noted that, in the above-mentioned approaches,
data transitions may be handled, e.g., by a filtering method to
minimize the severity of the transitions, such as by using a
low-pass filter 3480 as discussed above in connection with FIGS.
33A-35.
[0330] It bears repeating that sensor diagnostics, including, e.g.,
assessment of low start-up, sensitivity-loss, and signal-dip events
depends on various factors, including the sensor design, number of
electrodes (i.e., redundancy), electrode
distribution/configuration, etc. As such, the actual frequency, or
range of frequencies, for which an EIS-based parameter may be
substantially glucose-independent, and therefore, an independent
marker, or predictor, for one or more of the above-mentioned
failure modes may also depend on the specific sensor design. For
example, while it has been discovered, as described hereinabove,
that sensitivity loss may be predicted using imaginary impedance at
the relatively higher frequencies--where imaginary impedance is
substantially glucose-independent--the level of glucose dependence,
and, therefore, the specific frequency range for using imaginary
impedance as a marker for sensitivity loss, may shift (higher or
lower) depending on the actual sensor design.
[0331] More specifically, as sensor design moves more and more
towards the use of redundant working electrodes, the latter must be
of increasingly smaller sizes in order to maintain the overall size
of the sensor. The size of the electrodes, in turn, affects the
frequencies that may be queried for specific diagnostics. In this
regard, it is important to note that the fusion algorithms
described herein and shown in FIGS. 33A-35 are to be regarded as
illustrative, and not limiting, as each algorithm can be modified
as necessary to use EIS-based parameters at frequencies that
exhibit the least amount of glucose dependence, based on the type
of sensor under analysis.
[0332] In addition, experimental data indicates that human tissue
structure may also affect glucose dependence at different
frequencies. For example, in children, real impedance at 0.105 Hz
has been found to be a substantially glucose-independent indicator
for low start-up detection. It is believed that this comes about as
a result of a child's tissue structure changing, e.g., the Warburg
impedance, which relates mostly to the resistive component. See
also the subsequent discussion relating to interferent
detection.
[0333] Embodiments of the invention herein are also directed to the
use of EIS in optimizing sensor calibration. By way of background,
in current methodologies, the slope of a BG vs. Isig plot, which
may be used to calibrate subsequent Isig values, is calculated as
follows:
slope = .alpha. .beta. ( isig - offset ) bg .alpha. .beta. ( isig -
offset ) 2 ##EQU00006##
where a is an exponential function of a time constant, .beta. is a
function of blood glucose variance, and offset is a constant. For a
sensor in steady condition, this method provides fairly accurate
results. As shown, e.g., in FIG. 36, BG and Isig follow a fairly
linear relationship, and offset can be taken as a constant.
[0334] However, there are situations in which the above-mentioned
linear relationship does not hold true, such as, e.g., during
periods in which the sensor experiences a transition. As shown in
FIG. 37, it is clear that Isig-BG pairs 1 and 2 are significantly
different from pairs 3 and 4 in terms of Isig and BG relationship.
For these types of conditions, use of a constant offset tends to
produce inaccurate results.
[0335] To address this issue, one embodiment of the invention is
directed to the use of an EIS-based dynamic offset, where EIS
measurements are used to define a sensor status vector as
follows:
V={real_imp_1K,img_imp_1K,Nyquist_slope,Nyquist_R_square}
where all of the elements in the vector are substantially BG
independent. It is noted that Nyquist_R_square is the R square of
linear regression used to calculate the Nyquist slope, i.e., the
square of the correlation coefficient between real and imaginary
impedances at relatively-lower frequencies, and a low R square
indicates abnormality in sensor performance. For each Isig-BG pair,
a status vector is assigned. If a significant difference in status
vector is detected--e.g., |V2-V3| for the example shown in FIG.
37--a different offset value is assigned for 3 and 4 when compared
to 1 and 2. Thus, by using this dynamic offset approach, it is
possible to maintain a linear relationship between Isig and BG.
[0336] In a second embodiment, an EIS-based segmentation approach
may be used for calibration. Using the example of FIG. 37 and the
vector V, it can be determined that sensor state during 1 and 2 is
significantly different from sensor state during 3 and 4.
Therefore, the calibration buffer can be divided into two segments,
as follows:
Isig_buffer1=[Isig1,Isig2];BG_buffer1=[BG1,BG2]
Isig_buffer2=[Isig3,Isig3];BG_buffer2=[BG3,BG3]
Thus, when the sensor operates during 1 and 2, Isig_buffer1 and
BG_buffer1 would be used for calibration. However, when the sensor
operates during 3 and 4, i.e., during a transition period,
Isig_buffer2 and BG_buffer2 would be used for calibration.
[0337] In yet another embodiment, an EIS-based dynamic slope
approach, where EIS is used to adjust slope, may be used for
calibration purposes. FIG. 38A shows an example of how this method
can be used to improve sensor accuracy. In this diagram, the data
points 1-4 are discrete blood glucose values. As can be seen from
FIG. 38A, there is a sensor dip 3810 between data points 1 and 3,
which dip can be detected using the sensor state vector V described
above. During the dip, slope can be adjusted upward to reduce the
underreading, as shown by reference numeral 3820 in FIG. 38A.
[0338] In a further embodiment, EIS diagnostics may be used to
determine the timing of sensor calibrations, which is quite useful
for, e.g, low-startup events, sensitivity-loss events, and other
similar situations. As is known, most current methodologies require
regular calibrations based on a pre-set schedule, e.g., 4 times per
day. Using EIS diagnostics, however, calibrations become
event-driven, such that they may be performed only as often as
necessary, and when they would be most productive. Here, again, the
status vector V may be used to determine when the sensor state has
changed, and to request calibration if it has, indeed, changed.
[0339] More specifically, in an illustrative example, FIG. 38B
shows a flowchart for EIS-assisted sensor calibration involving low
start-up detection. Using Nyquist slope, 1 kHz real impedance, and
a bound check 3850 (see, e.g., the previously-described bound check
and associated threshold values for EIS-based parameters in
connection with the fusion algorithms of FIGS. 33A-35), a
reliability index 3853 can be developed for start-up, such that,
when the 1 kHz real impedance 3851 and the Nyquist slope 3852 are
lower than their corresponding upper bounds, RI_startup=1, and
sensor is ready for calibration. In other words, the reliability
index 3853 is "high" (3854), and the logic can proceed to
calibration at 3860.
[0340] When, on the other hand, the 1 kHz real impedance and the
Nyquist slope are higher than their corresponding upper bounds (or
threshold values), RI_startup=0 (i.e., it is "low"), and the sensor
is not ready for calibration (3856), i.e., a low start-up issue may
exist. Here, the trend of 1 kHz real impedance and the Nyquist
slope can be used to predict when both parameters will be in range
(3870). If it is estimated that this will only take a very short
amount of time (e.g., less than one hour), then the algorithm waits
until the sensor is ready, i.e., until the above-mentioned
EIS-based parameters are in-bound (3874), at which point the
algorithm proceeds to calibration. If, however, the wait time would
be relatively long (3876), then the sensor can be calibrated now,
and then the slope or offset can be gradually adjusted according to
the 1 kHz real impedance and the Nyquist slope trend (3880). It is
noted that by performing the adjustment, serious over- or
under-reading caused by low start-up can be avoided. As noted
previously, the EIS-based parameters and related information that
is used in the instant calibration algorithm is substantially
glucose-independent.
[0341] It is noted that, while the above description in connection
with FIG. 38B shows a single working electrode, as well as the
calculation of a reliability index for start-up of that working
electrode, this is by way of illustration, and not limitation.
Thus, in a redundant sensor including two or more working
electrodes, a bound check can be performed, and a start-up
reliability index calculated, for each of the plurality of
(redundant) working electrodes. Then, based on the respective
reliability indices, at least one working electrode can be
identified that can proceed to obtain glucose measurements. In
other words, in a sensor having a single working electrode, if the
latter exhibits low start-up, actual use of the sensor (for
measuring glucose) may have to be delayed until the low start-up
period is over. This period may typically be on the order of one
hour or more, which is clearly disadvantageous. In contrast, in a
redundant sensor, utilizing the methodology described herein allows
an adaptive, or "smart", start-up, wherein an electrode that can
proceed to data gathering can be identified in fairly short order,
e.g., on the order of a few minutes. This, in turn, reduces MARD,
because low start-up generally provides about a 1/2% increase in
MARD.
[0342] In yet another embodiment, EIS can aid in the adjustment of
the calibration buffer. For existing calibration algorithms, the
buffer size is always 4, i.e., 4 Isig-BG pairs, and the weight is
based upon .alpha. which, as noted previously, is an exponential
function of a time constant, and .beta., which is a function of
blood glucose variance. Here, EIS can help to determine when to
flush the buffer, how to adjust buffer weight, and what the
appropriate buffer size is.
[0343] Embodiments of the invention are also directed to the use of
EIS for interferent detection. Specifically, it may be desirable to
provide a medication infusion set that includes a combination
sensor and medication-infusion catheter, where the sensor is placed
within the infusion catheter. In such a system, the physical
location of the infusion catheter relative to the sensor may be of
some concern, due primarily to the potential impact on (i.e.,
interference with) sensor signal that may be caused by the
medication being infused and/or an inactive component thereof.
[0344] For example, the diluent used with insulin contains m-cresol
as a preservative. In in-vitro studies, m-cresol has been found to
negatively impact a glucose sensor if insulin (and, therefore,
m-cresol) is being infused in close proximity to the sensor.
Therefore, a system in which a sensor and an infusion catheter are
to be combined in a single needle must be able to detect, and
adjust for, the effect of m-cresol on the sensor signal. Since
m-cresol affects the sensor signal, it would be preferable to have
a means of detecting this interferent independently of the sensor
signal itself.
[0345] Experiments have shown that the effect of m-cresol on the
sensor signal is temporary and, thus, reversible. Nevertheless,
when insulin infusion occurs too close to the sensor, the m-cresol
tends to "poison" the electrode(s), such that the latter can no
longer detect glucose, until the insulin (and m-cresol) have been
absorbed into the patient's tissue. In this regard, it has been
found that there is typically about a 40-minute time period between
initiation of insulin infusion and when the sensor has re-gained
the ability to detect glucose again. However, advantageously, it
has also been discovered that, during the same time period, there
is a large increase in 1 kHz impedance magnitude quite
independently of the glucose concentration.
[0346] Specifically, FIG. 39 shows Isig and impedance data for an
in-vitro experiment, wherein the sensor was placed in a 100 mg/dL
glucose solution, and 1 kHz impedance was measured every 10
minutes, as shown by encircled data points 3920. m-cresol was then
added to bring the solution to 0.35% m-cresol (3930). As can be
seen, once m-cresol has been added, the Isig 3940 initially
increases dramatically, and then begins to drift down. The
concentration of glucose in the solution was then doubled, by
adding an addition 100 mg/dL glucose. This, however, had no effect
on the Isig 3940, as the electrode was unable to detect the
glucose.
[0347] On the other hand, the m-cresol had a dramatic effect on
both impedance magnitude and phase. FIG. 40A shows a Bode plot for
the phase, and FIG. 40B shows a Bode plot for impedance magnitude,
for both before and after the addition of m-cresol. As can be seen,
after the m-cresol was added, the impedance magnitude 4010
increased from its post-initialization value 4020 by at least an
order of magnitude across the frequency spectrum. At the same time,
the phase 4030 changed completely as compared to its
post-initialization value 4040. On the Nyquist plot of FIG. 40C.
Here, the pre-initialization curve 4050 and the post-initialization
curve 4060 appear as expected for a normally-functioning sensor.
However, after the addition of m-cresol, the curve 4070 becomes
drastically different.
[0348] The above experiment identifies an important practical
pitfall of continuing to rely on the Isig after m-cresol has been
added. Referring back to FIG. 39, a patient/user monitoring the
sensor signal may be put under the mistaken impression that his
glucose level has just spiked, and that he should administer a
bolus. The user then administers the bolus, at which the Isig has
already started to drift back down. In other words, to the
patient/user, everything may look normal. In reality, however, what
has really happened is that the patient just administered an
unneeded dose of insulin which, depending on the patient's glucose
level prior to administration of the bolus, may put the patient at
risk of experiencing a hypoglycemic event. This scenario reinforces
the desirability of a means of detecting interferents that is as
glucose-independent as possible.
[0349] FIG. 41 shows another experiment, where a sensor was
initialized a 100 mg/dL glucose solution, after which glucose was
raised to 400 mg/dL for one hour, and then returned to 100 mg/dL.
m-cresol was then added to raise the concentration to 0.35%, with
the sensor remaining in this solution for 20 minutes. Finally, the
sensor was placed in a 100 mg/dL glucose solution to allow Isig to
recover after exposure to m-cresol. As can be seen, after
initialization, the 1 kHz impedance magnitude 4110 was at about 2
kOhms. When m-cresol was added, the Isig 4120 spiked, as did
impedance magnitude 4110. Moreover, when the sensor was returned to
a 100 md/dL glucose solution, the impedance magnitude 4110 also
returned to near normal level.
[0350] As can be seen from the above experiments, EIS can be used
to detect the presence of an interfering agent--in this case,
m-cresol. Specifically, since the interferent affects the sensor in
such a way as to increase the impedance magnitude across the entire
frequency spectrum, the impedance magnitude may be used to detect
the interference. Once the interference has been detected, either
the sensor operating voltage can be changed to a point where the
interferent is not measured, or data reporting can be temporarily
suspended, with the sensor indicating to the patient/user that, due
to the administration of medication, the sensor is unable to report
data (until the measured impedance returns to the pre-infusion
level). It is noted that, since the impact of the interferent is
due to the preservative that is contained in insulin, the impedance
magnitude will exhibit the same behavior as described above
regardless of whether the insulin being infused is fast-acting or
slow.
[0351] Importantly, as mentioned above, the impedance magnitude,
and certainly the magnitude at 1 kHz, is substantially
glucose-independent. With reference to FIG. 41, it can be seen
that, as the concentration of glucose is raised from 100 mg/dL to
400 mg/dL--a four-fold increase--the 1 kHz impedance magnitude
increase from about 2000 Ohms to about 2200 Ohms, or about a 10%
increase. In other words, the effect of glucose on the impedance
magnitude measurement appears to be about an order of magnitude
smaller compared to the measured impedance. This level of
"signal-to-noise" ratio is typically small enough to allow the
noise (i.e., the glucose effect) to be filtered out, such that the
resultant impedance magnitude is substantially glucose-independent.
In addition, it should be emphasized that the impedance magnitude
exhibits an even higher degree of glucose-independence in actual
human tissue, as compared to the buffer solution that was used for
the in-vitro experiments described above.
[0352] Embodiments of the invention are also directed to an Analog
Front End Integrated Circuit (AFE IC), which is a custom
Application Specific Integrated Circuit (ASIC) that provides the
necessary analog electronics to: (i) support multiple potentiostats
and interface with multi-terminal glucose sensors based on either
Oxygen or Peroxide; (ii) interface with a microcontroller so as to
form a micropower sensor system; and (iii) implement EIS
diagnostics, fusion algorithms, and other EIS-based processes based
on measurement of EIS-based parameters. More specifically, the ASIC
incorporates diagnostic capability to measure the real and
imaginary impedance parameters of the sensor over a wide range of
frequencies, as well as digital interface circuitry to enable
bidirectional communication with a microprocessor chip. Moreover,
the ASIC includes power control circuitry that enables operation at
very low standby and operating power, and a real-time clock and a
crystal oscillator so that an external microprocessor's power can
be turned off.
[0353] FIGS. 42A and 42B show a block diagram of the ASIC, and
Table 1 below provides pad signal descriptions (shown on the
left-hand side of FIGS. 42A and 42B), with some signals being
multiplexed onto a single pad.
TABLE-US-00004 TABLE 1 Pad signal descriptions Pad Name Functional
Description Power plane VBAT Battery power input 2.0 V to 4.5 V
VBAT VDDBU Backup logic power 1.4 to 2.4 V VDDBU VDD Logic power -
1.6-2.4 V VDD VDDA Analog power - 1.6-2.4 V VDDA VPAD Pad I/O power
- 1.8 V-3.3 V VPAD VSS Logic ground return and digital pad return
VSSA Analog ground return and analog pad return ADC_IN1, 2 ADC
Inputs, VDDA max input VDDA V1P2B 1.2 volt reference Bypass
capacitor VDDA nSHUTDN External VDD regulator control signal. Goes
low when battery is VBAT low. VPAD_EN Goes high when VPAD IOs are
active. Can control external VBAT regulator. DA1, 2 DAC outputs
VDDA TP_ANA_MUX Mux of analog test port - output or input VDDA
TP_RES External 1 meg ohm calibration resistor & analog test
port VDDA WORK1-5 Working Electrodes of Sensor VDDA RE Reference
Electrode of Sensor VDDA COUNTER Counter Electrode of Sensor VDDA
CMP1_IN General purpose Voltage comparator VDDA CMP2_IN General
purpose Voltage comparator VDDA WAKEUP Debounced interrupt input
VBAT XTALI, XTALO 32.768 kHz Crystal Oscillator pads VDDA
OSC_BYPASS Test clock control VDDA SEN_CONN_SW Sensor connection
switch input. Pulled to VSSA = connection VDDA VPAD_EN Enable the
VPAD power and VPAD power plane logic VBAT nRESET_OD Signal to
reset external circuitry such as a microprocessor SPI_CK, SPI
interface signals to microprocessor VPAD nSPI_CS, SPI_MOIS,
SPI_MISO UP_WAKEUP Microprocessor wakeup signal VPAD CLK_32KHZ
Gated Clock output to external circuitry microprocessor VPAD UP_INT
Interrupt signal to microprocessor VPAD nPOR1_OUT Backup Power on
reset, output from analog VBAT nPOR1_IN VBAT power plane reset,
input to digital in battery plane VBAT (VDDBU) nPOR2_OUT VDD POR
signal, output from analog VDD nPOR2_OUT_OD VDD POR signal open
drain (nfet out only), stretched output VBAT from digital nPOR2_IN
VDD power plane logic reset. Is level shifted to VDD inside the VDD
chip, input to digital VDD logic.
[0354] The ASIC will now be described with reference to FIGS. 42A
and 42B and Table 1.
[0355] Power Planes
[0356] The ASIC has one power plane that is powered by the supply
pad VBAT (4210), which has an operating input range from 2.0 volts
to 4.5 volts. This power plane has a regulator to lower the voltage
for some circuits in this plane. The supply is called VDDBU (4212)
and has an output pad for test and bypassing. The circuits on the
VBAT supply include an RC oscillator, real time clock (RC osc)
4214, battery protection circuit, regulator control, power on reset
circuit (POR), and various inputs/outputs. The pads on the VBAT
power plane are configured to draw less than 75 nA at 40.degree. C.
and VBAT=3.50V.
[0357] The ASIC also has a VDD supply to supply logic. The VDD
supply voltage range is programmable from at least 1.6 volts to 2.4
volts. The circuits on the VDD power plane include most of the
digital logic, timer (32 khz), and real time clock (32 khz). The
VDD supply plane includes level shifters interfacing to the other
voltage planes as necessary. The level shifters, in turn, have
interfaces conditioned so that any powered power plane does not
have an increase in current greater than 10 nA if another power
plane is unpowered.
[0358] The ASIC includes an onboard regulator (with shutdown
control) and an option for an external VDD source. The regulator
input is a separate pad, REG_VDD_IN (4216), which has electrostatic
discharge (ESD) protection in common with other I/Os on VBAT. The
onboard regulator has an output pad, REG_VDD_OUT (4217). The ASIC
also has an input pad for the VDD, which is separate from the
REG_VDD_OUT pad.
[0359] The ASIC includes an analog power plane, called VDDA (4218),
which is powered by either the VDD onboard regulator or an external
source, and is normally supplied by a filtered VDD. The VDDA
supplied circuits are configured to operate within 0.1 volt of VDD,
thereby obviating the need for level shifting between the VDDA and
VDD power planes. The VDDA supply powers the sensor analog
circuits, the analog measurement circuits, as well as any other
noise-sensitive circuitry.
[0360] The ASIC includes a pad supply, VPAD, for designated digital
interface signals. The pad supply has an operating voltage range
from at least 1.8 V to 3.3 V. These pads have separate supply
pad(s) and are powered from an external source. The pads also
incorporate level shifters to other onboard circuits to allow the
flexible pad power supply range independently of the VDD logic
supply voltage. The ASIC can condition the VPAD pad ring signals
such that, when the VPAD supply is not enabled, other supply
currents will not increase by more than 10 nA.
[0361] Bias Generator
[0362] The ASIC has a bias generator circuit, BIAS_GEN (4220),
which is supplied from the VBAT power, and which generates bias
currents that are stable with supply voltage for the system. The
output currents have the following specifications: (i) Supply
sensitivity: <.+-.2.5% from a supply voltage of 1.6 v to 4.5V;
and (ii) Current accuracy: <.+-.3% after trimming.
[0363] The BIAS_GEN circuit generates switched and unswitched
output currents to supply circuits needing a bias current for
operation. The operating current drain of the BIAS_GEN circuit is
less than 0.3 uA at 25.degree. C. with VBAT from 2.5V-4.5V
(excluding any bias output currents). Lastly, the temperature
coefficient of the bias current is generally between 4,000
ppm/.degree. C. and 6,000 ppm/.degree. C.
[0364] Voltage Reference
[0365] The ASIC, as described herein is configured to have a low
power voltage reference, which is powered from the VBAT power
supply. The voltage reference has an enable input which can accept
a signal from logic powered by VBAT or VDDBU. The ASIC is designed
such that the enable signal does not cause any increase in current
in excess of 10 nA from any supply from this signal interface when
VBAT is powered.
[0366] The reference voltage has the following specifications: (i)
Output voltage: 1.220.+-.3 mV after trimming; (ii) Supply
sensitivity: <.+-.6 mV from 1.6 V to 4.5V input; (iii)
Temperature sensitivity: <.+-.5 mV from 0.degree. C. to
60.degree. C.; and (iv) Output voltage default accuracy (without
trim): 1.220 V.+-.50 mV. In addition, the supply current is to be
less than 800 nA at 4.5V, 40.degree. C. In this embodiment, the
reference output will be forced to VSSA when the reference is
disabled so as to keep the VDD voltage regulator from overshooting
to levels beyond the breakdown voltage of the logic.
[0367] 32 kHz Oscillator
[0368] The ASIC includes a low power 32.768 kHz crystal oscillator
4222 which is powered with power derived from the VDDA supply and
can trim the capacitance of the crystal oscillator pads (XTALI,
XTALO) with software. Specifically, the frequency trim range is at
least -50 ppm to +100 ppm with a step size of 2 ppm max throughout
the trim range. Here, a crystal may be assumed with a load
capacitance of 7 pF, Ls=6.9512 kH, Cs=3.3952 fF, Rs=70 k, shunt
capacitance=1 pF, and a PC Board parasitic capacitance of 2 pF on
each crystal terminal.
[0369] The ASIC has a VPAD level output available on a pad, CLK 32
kHZ, where the output can be disabled under software and logic
control. The default is driving the 32 kHz oscillator out. An input
pin, OSC32K_BYPASS (4224), can disable the 32 kHz oscillator (no
power drain) and allows for digital input to the XTALI pad. The
circuits associated with this function are configured so as not add
any ASIC current in excess of 10 nA in either state of the
OSC32K_BYPASS signal other than the oscillator current when
OSC32K_BYPASS is low.
[0370] The 32 kHZ oscillator is required to always be operational
when the VDDA plane is powered, except for the bypass condition. If
the OSC32K_BYPASS is true, the 32 KHZ oscillator analog circuitry
is put into a low power state, and the XTALI pad is configured to
accept a digital input whose level is from 0 to VDDA. It is noted
that the 32 kHz oscillator output has a duty cycle between 40% and
60%.
[0371] Timer
[0372] The ASIC includes a Timer 4226 that is clocked from the 32
kHz oscillator divided by 2. It is pre-settable and has two
programmable timeouts. It has 24 programmable bits giving a total
time count to 17 minutes, 4 seconds. The Timer also has a
programmable delay to disable the clock to the CLK_32 KHz pad and
set the microprocessor (uP) interface signals on the VPAD plane to
a predetermined state (See section below on Microprocessor Wakeup
Control Signals). This will allow the microprocessor to go into
suspend mode without an external clock. However, this function may
be disabled by software with a programmable bit.
[0373] The Timer also includes a programmable delay to wakeup the
microprocessor by enabling the CLK_32 KHZ clock output and setting
UP_WAKEUP high. A transition of the POR2 (VDD POR) from supply low
state to supply OK state will enable the 32 kHz oscillator, the
CLK_32 KHZ clock output and set UP_WAKEUP high. The power shutdown
and power up are configured to be controlled with programmable
control bits.
[0374] Real Time Clock (RTC)
[0375] The ASIC also has a 48 bit readable/writeable binary counter
that operates from the ungated, free running 32 kHz oscillator. The
write to the real time clock 4228 requires a write to an address
with a key before the clock can be written. The write access to the
clock is configured to terminate between 1 msec and 20 msec after
the write to the key address.
[0376] The real time clock 4228 is configured to be reset by a
power on reset either by POR1_IN (the VBAT POR) or POR2_IN (the VDD
POR) to half count (MSB=1, all other bits 0). In embodiments of the
invention, the real time clock has programmable interrupt
capability, and is designed to be robust against single event
upsets (SEUs), which may be accomplished either by layout
techniques or by adding capacitance to appropriate nodes, if
required.
[0377] RC Oscillator
[0378] The ASIC further includes an RC clock powered from the VBAT
supply or VBAT derived supply. The RC Oscillator is always running,
except that the oscillator can be bypassed by writing to a register
bit in analog test mode (see section on Digital Testing) and
applying a signal to the GPIO_VBAT with a 0 to VBAT level. The RC
oscillator is not trimmable, and includes the following
specifications: (i) a frequency between 750 Hz and 1500 Hz; (ii) a
duty cycle between 50%.+-.10%; (iii) current consumption of less
than 200 nA at 25.degree. C.; (iv) frequency change of less than
.+-.2% from 1V to 4.5V VBAT supply and better than 1% from 1.8V to
4.5V VBAT supply; and (v) frequency change of less than +2, -2%
from a temperature of 15.degree. C. to 40.degree. C. with
VBAT=3.5V. The RC frequency can be measured with the 32 kHz crystal
oscillator or with an external frequency source (See Oscillator
Calibration Circuit).
[0379] Real Time RC Clock (RC Oscillator Based)
[0380] The ASIC includes a 48 bit readable/writeable binary ripple
counter based on the RC oscillator. A write to the RC real time
clock requires a write to an address with a key before the clock
can be written. The write access to the clock terminates between 1
msec and 20 msec after the write to the key address, wherein the
time for the protection window is configured to be generated with
the RC clock.
[0381] The real time RC clock allows for a relative time stamp if
the crystal oscillator is shutdown, and is configured to be reset
on POR1_IN (the BAT POR) to half count (MSB=1, all others 0). The
real time RC clock is designed to be robust against single event
upsets (SEUs) either by layout techniques or by adding capacitance
to appropriate nodes, where required. On the falling edge of
POR2_IN, or if the ASIC goes into Battery Low state, the RT real
time clock value may be captured into a register that can be read
via the SPI port. This register and associated logic are on the
VBAT or VDDBU power plane.
[0382] Battery Protection Circuit
[0383] The ASIC includes a battery protection circuit 4230 that
uses a comparator to monitor the battery voltage and is powered
with power derived from the VBAT power plane. The battery
protection circuit is configured to be always running with power
applied to the VBAT supply. The battery protection circuit may use
the RC oscillator for clocking signals, and have an average current
drain that is less than 30 nA, including a 3 MOhm total resistance
external voltage divider.
[0384] The battery protection circuit uses an external switched
voltage divider having a ratio of 0.421 for a 2.90V battery
threshold. The ASIC also has an internal voltage divider with the
ratio of 0.421.+-.0.5%. This divider is connected between
BATT_DIV_EN (4232) and VSSA (4234), and the divider output is a pin
called BATT_DIV_INT (4236). To save pins in the packaged part, the
BATT_DIV_INT in this embodiment is connected to BATT_DIV internally
in the package. Also in this configuration, BATT_DIV_EN does not
need to come out of the package, saving two package pins.
[0385] The battery protection circuit is configured to sample the
voltage on an input pin, BATT_DIV (4238), at approximately 2 times
per second, wherein the sample time is generated from the RC
Oscillator. The ASIC is able to adjust the divider of the RC
Oscillator to adjust the sampling time interval to 0.500 sec.+-.5
msec with the RC oscillator operating within its operating
tolerance. In a preferred embodiment, the ASIC has a test mode
which allows more frequent sampling intervals during test.
[0386] The comparator input is configured to accept an input from 0
to VBAT volts. The input current to the comparator input, BATT_DIV,
is less than 10 nA for inputs from 0 to VBAT volts. The comparator
sampling circuit outputs to a pad, BATT_DIV_EN, a positive pulse
which can be used by external circuitry to enable an off-chip
resistor divider only during the sampling time to save power. The
voltage high logic level is the VBAT voltage and the low level is
VSS level.
[0387] The output resistance of the BATT_DIV_EN pad shall be less
than 2 kOhms at VBAT=3.0V. This allows the voltage divider to be
driven directly from this output. After a programmable number of
consecutive samples indicating a low battery condition, the
comparator control circuitry triggers an interrupt to the interrupt
output pad, UP_INT. The default number of samples is 4, although
the number of consecutive samples is programmable from 4 to
120.
[0388] After a programmable number of consecutive samples
indicating a low battery after the generation of the UP_INT above,
the comparator control circuitry is configured to generate signals
that will put the ASIC into a low power mode: The VDD regulator
will be disabled and a low signal will be asserted to the pad,
VPAD_EN. This will be called the Battery Low state. Again, the
number of consecutive samples is programmable from 4 to 120
samples, with the default being 4 samples.
[0389] The comparator has individual programmable thresholds for
falling and rising voltages on BATT_DIV. This is implemented in the
digital logic to multiplex the two values to the circuit depending
on the state of the Battery Low state. Thus, if Battery Low state
is low, the falling threshold applies, and if the Battery Low state
is high, the rising threshold applies. Specifically, the comparator
has 16 programmable thresholds from 1.22 to 1.645.+-.3%, wherein
the DNL of the programmable thresholds is set to be less than 0.2
LSB.
[0390] The comparator threshold varies less than +/-1% from
20.degree. C. to 40.degree. C. The default threshold for falling
voltage is 1.44V (VBAT threshold of 3.41V with nominal voltage
divider), and the default threshold for rising voltage is 1.53V
(VBAT threshold of 3.63V with nominal voltage divider). After the
ASIC has been put into the Battery Low state, if the comparator
senses 4 consecutive indications of battery OK, then the ASIC will
initiate the microprocessor startup sequence.
[0391] Battery Power Plane Power on Reset
[0392] A power on reset (POR) output is generated on pad nPOR1_OUT
(4240) if the input VBAT slews more than 1.2 volt in a 50 usec
period or if the VBAT voltage is below 1.6.+-.0.3 volts. This POR
is stretched to a minimum pulse width of 5 milliseconds. The output
of the POR circuit is configured to be active low and go to a pad,
nPOR1_OUT, on the VBAT power plane.
[0393] The IC has an input pad for the battery power plane POR,
nPOR1_IN (4242). This input pad has RC filtering such that pulses
shorter than 50 nsec will not cause a reset to the logic. In this
embodiment, nPOR1_OUT is externally connected to the nPOR1_IN in
normal operation, thereby separating the analog circuitry from the
digital circuitry for testing. The nPOR1_IN causes a reset of all
logic on any of the power planes, and initializes all registers to
their default value. Thus, the reset status register POR bit is
set, and all other reset status register bits are cleared. The POR
reset circuitry is configured so as not to consume more than 0.1 uA
from VBAT supply for time greater than 5 seconds after power
up.
[0394] VDD Power On Reset (POR)
[0395] The ASIC also has a voltage comparator circuit which
generates a VDD voltage plane reset signal upon power up, or if the
VDD drops below a programmable threshold. The range is programmable
with several voltage thresholds. The default value is 1.8V-15%
(1.53V). The POR2 has a programmable threshold for rising voltage,
which implements hysteresis. The rising threshold is also
programmable, with a default value of 1.60V.+-.3%.
[0396] The POR signal is active low and has an output pad,
nPOR2_OUT (4244), on the VDD power plane. The ASIC also has an
active low POR open drain output, nPOR2_OUT OD (4246), on the VBAT
power plane. This could be used for applying POR to other system
components.
[0397] The VDD powered logic has POR derived from the input pad,
nPOR2_IN (4248). The nPOR2_IN pad is on the VDD power plane, and
has RC filtering such that pulses shorter than 50 nsec will not
cause a reset to the logic. The nPOR2_OUT is configured be
externally connected to the nPOR2_IN input pad under normal usage,
thereby separating the analog circuitry from the digital
circuitry.
[0398] The reset which is generated is stretched to at least 700
msec of active time after VDD goes above the programmable threshold
to assure that the crystal oscillator is stable. The POR reset
circuitry is to consume no more than 0.1 uA from the VDD supply for
time greater than 5 seconds after power up, and no more than 0.1 uA
from VBAT supply for time greater than 5 seconds after power up.
The register that stores the POR threshold value is powered from
the VDD power plane.
[0399] Sensor Interface Electronics
[0400] In an embodiment of the invention, the sensor circuitry
supports up to five sensor WORK electrodes (4310) in any
combination of peroxide or oxygen sensors, although, in additional
embodiments, a larger number of such electrodes may also be
accommodated. While the peroxide sensor WORK electrodes source
current, the oxygen sensor WORK electrodes sink current. For the
instant embodiment, the sensors can be configured in the
potentiostat configuration as shown in FIG. 43.
[0401] The sensor electronics have programmable power controls for
each electrode interface circuit to minimize current drain by
turning off current to unused sensor electronics. The sensor
electronics also include electronics to drive a COUNTER electrode
4320 that uses feedback from a RE (reference) electrode 4330. The
current to this circuitry may be programmed off when not in use to
conserve power. The interface electronics include a multiplexer
4250 so that the COUNTER and RE electrodes may be connected to any
of the (redundant) WORK electrodes.
[0402] The ASIC is configured to provide the following Sensor
Interfaces: (i) RE: Reference electrode, which establishes a
reference potential of the solution for the electronics for setting
the WORK voltages; (ii) WORK1-WORK5: Working sensor electrodes
where desired reduction/oxidation (redox) reactions take place; and
(iii) COUNTER: Output from this pad maintains a known voltage on
the RE electrode relative to the system VSS. In this embodiment of
the invention, the ASIC is configured so as to be able to
individually set the WORK voltages for up to 5 WORK electrodes with
a resolution and accuracy of better than or equal to 5 mV.
[0403] The WORK voltage(s) are programmable between at least 0 and
1.22V relative to VSSA in the oxygen mode. In the peroxide mode,
the WORK voltage(s) are programmable between at least 0.6 volt and
2.054 volts relative to VSSA. If the VDDA is less than 2.15V, the
WORK voltage is operational to VDDA-0.1V. The ASIC includes current
measuring circuits to measure the WORK electrode currents in the
peroxide sensor mode. This may be implemented, e.g., with
current-to-voltage or current-to-frequency converters, which may
have the following specifications: (i) Current Range: 0-300 nA;
(ii) Voltage output range: Same as WORK electrode in
peroxide/oxygen mode; (iii) Output offset voltage: .+-.5 mV max;
and (iv) Uncalibrated resolution: .+-.0.25 nA.
[0404] Current Measurement Accuracy after applying a calibration
factor to the gain and assuming an acquisition time of 10 seconds
or less is: [0405] 5 pA-1 nA: .+-.3%.+-.20 pA [0406] 1 nA-10 nA:
.+-.3%.+-.20 pA [0407] 10 nA-300 nA: .+-.3%.+-.0.2 nA
[0408] For current-to-frequency converters (ItoFs) only, the
frequency range may be between 0 Hz and 50 kHz. The current
converters must operate in the specified voltage range relative to
VSS of WORK electrodes in the peroxide mode. Here, the current
drain is less than 2 uA from a 2.5V supply with WORK electrode
current less than 10 nA per converter including digital-to-analog
(DAC) current.
[0409] The current converters can be enabled or disabled by
software control. When disabled, the WORK electrode will exhibit a
very high impedance value, i.e., greater than 100 Mohm. Again, for
ItoFs only, the output of the I-to-F converters will go to 32 bit
counters, which can be read, written to, and cleared by the
microprocessor and test logic. During a counter read, clocking of
the counter is suspended to ensure an accurate read.
[0410] In embodiments of the invention, the ASIC also includes
current measuring circuits to measure the WORK electrode currents
in the oxygen sensor mode. The circuit may be implemented as a
current-to-voltage or a current-to-frequency converter, and a
programmable bit may be used to configure the current converters to
operate in the oxygen mode. As before, the current converters must
operate in the specified voltage range of the WORK electrodes
relative to VSS in the oxygen mode. Here, again, the current range
is 3.7 pA-300 nA, the voltage output range is the same as WORK
electrode in oxygen mode, the output offset voltage is .+-.5 mV
max, and the uncalibrated resolution is 3.7 pA.+-.2 pA.
[0411] Current Measurement Accuracy after applying a calibration
factor to the gain and assuming an acquisition time of 10 seconds
or less is: [0412] 5 pA-1 nA: .+-.3%.+-.20 pA [0413] 1 nA-10 nA:
.+-.3%.+-.20 pA [0414] 10 nA-300 nA: .+-.3%.+-.0.2 nA
[0415] For current-to-frequency converters (ItoFs) only, the
frequency range may be between 0 Hz and 50 kHz, and the current
drain is less than 2 uA from a 2.5V supply with WORK electrode
current less than 10 nA per converter, including DAC current. The
current converters can be enabled or disabled by software control.
When disabled, the WORK electrode will exhibit a very high
impedance value, i.e., greater than 100 Mohm. Also, for ItoFs only,
the output of the I-to-F converters will go to 32 bit counters,
which can be read, written to, and cleared by the microprocessor
and test logic. During a counter read, clocking of the counter is
suspended to ensure an accurate read.
[0416] In embodiments of the invention, the Reference electrode
(RE) 4330 has an input bias current of less than 0.05 nA at
40.degree. C. The COUNTER electrode adjusts its output to maintain
a desired voltage on the RE electrode. This is accomplished with an
amplifier 4340 whose output to the COUNTER electrode 4320 attempts
to minimize the difference between the actual RE electrode voltage
and the target RE voltage, the latter being set by a DAC.
[0417] The RE set voltage is programmable between at least 0 and
1.80V, and the common mode input range of the COUNTER amplifier
includes at least 0.20 to (VDD-0.20)V. A register bit may be used
to select the common mode input range, if necessary, and to provide
for programming the mode of operation of the COUNTER. The WORK
voltage is set with a resolution and accuracy of better than or
equal to 5 mV. It is noted that, in the normal mode, the COUNTER
voltage seeks a level that maintains the RE voltage to the
programmed RE target value. In the force counter mode, however, the
COUNTER electrode voltage is forced to the programmed RE target
voltage.
[0418] All electrode driving circuits are configured to be able to
drive the electrode to electrode load and be free from oscillation
for any use scenario. FIG. 44 shows the equivalent ac
inter-electrode circuit according to the embodiment of the
invention with the potentiostat configuration as shown in FIG. 43.
The equivalent circuit shown in FIG. 44 may be between any of the
electrodes, i.e., WORK1-WORK5, COUNTER and RE, with value ranges as
follows for the respective circuit components: [0419] Ru=[200-5 k]
Ohms [0420] Cc=[10-2000] pF [0421] Rpo=[1-20] kOhms [0422]
Rf=[200-2000] kOhms [0423] Cf=[2-30] uF
[0424] During initialization, the drive current for WORK electrodes
and the COUNTER electrode need to supply higher currents than for
the normal potentiostat operation described previously. As such,
programmable register bits may be used to program the electrode
drive circuits to a higher power state if necessary for extra
drive. It is important to achieve low power operation in the normal
potentiostat mode, where the electrode currents are typically less
than 300 nA.
[0425] In preferred embodiments, during initialization, the WORK1
through WORK5 electrodes are programmable in steps equal to, or
less than, 5 mV from 0 to VDD volts, and their drive or sink
current output capability is a minimum of 20 uA, from 0.20V to
(VDD-0.20V). Also during initialization, the ASIC is generally
configured to be able to measure the current of one WORK electrode
up to 20 uA with an accuracy of .+-.2%.+-.40 nA of the measurement
value. Moreover, during initialization, the RE set voltage is
programmable as described previously, the COUNTER DRIVE CIRCUIT
output must be able to source or sink 50 uA minimum with the
COUNTER electrode from 0.20V to (VDD-0.20V), and the supply current
(VDD and VDDA) to the initialization circuitry is required to be
less than 50 uA in excess of any output current sourced.
[0426] Current Calibrator
[0427] In embodiments of the invention, the ASIC has a current
reference that can be steered to any WORK electrode for the purpose
of calibration. In this regard, the calibrator includes a
programmable bit that causes the current output to sink current or
source current. The programmable currents include at least 10 nA,
100 nA, and 300 nA, with an accuracy of better than .+-.1%.+-.1 nA,
assuming a 0 tolerance external precision resistor. The calibrator
uses a 1 MegOhm precision resistor connected to the pad, TP_RES
(4260), for a reference resistance. In addition, the current
reference can be steered to the COUNTER or RE electrodes for the
purpose of initialization and/or sensor status. A constant current
may be applied to the COUNTER or the RE electrodes and the
electrode voltage may be measured with the ADC.
[0428] High Speed RC Oscillator
[0429] With reference back to FIG. 42, the ASIC further includes a
high speed RC oscillator 4262 which supplies the analog-to-digital
converter (ADC) 4264, the ADC sequencer 4266, and other digital
functions requiring a higher speed clock than 32 kHz. The high
speed RC oscillator is phased locked to the 32 kHz clock (32.768
kHz) to give an output frequency programmable from 524.3 kHz to
1048 kHz. In addition, the high speed RC oscillator has a duty
cycle of 50%.+-.10%, a phase jitter of less than 0.5% rms, a
current of less than 10 uA, and a frequency that is stable through
the VDD operating range (voltage range of 1.6 to 2.5V). The default
of the high speed RC oscillator is "off" (i.e., disabled), in which
case the current draw is less than 10 nA. However, the ASIC has a
programmable bit to enable the High Speed RC oscillator.
[0430] Analog to Digital Converter
[0431] The ASIC includes a 12-bit ADC (4264) with the following
characteristics: (i) capability to effect a conversion in less than
1.5 msec with running from a 32 kHz clock; (ii) ability to perform
faster conversions when clocked from the high speed RC oscillator;
(iii) have at least 10 bits of accuracy (12 bit.+-.4 counts); (iv)
have a reference voltage input of 1.220V, with a temperature
sensitivity of less than 0.2 mV/.degree. C. from 20.degree. C. to
40.degree. C.; (v) full scale input ranges of 0 to 1.22V, 0 to
1.774V, 0 to 2.44V, and 0-VDDA, wherein the 1.774 and 2.44V ranges
have programmable bits to reduce the conversion range to lower
values to accommodate lower VDDA voltages; (vi) have current
consumption of less than 50 uA from its power supply; (vi) have a
converter capable of operating from the 32 kHz clock or the High
Speed RC clock; (vii) have a DNL of less than 1 LSB; and (viii)
issue an interrupt at the end of a conversion.
[0432] As shown in FIGS. 42A and 42B, the ASIC has an analog
multiplexer 4268 at the input of the ADC 4264, both of which are
controllable by software. In a preferred embodiment, at least the
following signals are connected to the multiplexer: [0433] (i)
VDD--Core Voltage and regulator output [0434] (ii) VBAT--Battery
source [0435] (iii) VDDA--Analog supply [0436] (iv) RE--Reference
Electrode of Sensor [0437] (v) COUNTER--Counter Electrode of Sensor
[0438] (vi) WORK1--WORK5--Working Electrodes of Sensor [0439] (vii)
Temperature sensor [0440] (viii) At least two external pin analog
signal inputs [0441] (ix) EIS integrator outputs [0442] (x) ItoV
current converter output.
[0443] The ASIC is configured such that the loading of the ADC will
not exceed .+-.0.01 nA for the inputs COUNTER, RE, WORK1-WORK5, the
temperature sensor, and any other input that would be adversely
affected by loading. The multiplexer includes a divider for any
inputs that have higher voltage than the input voltage range of the
ADC, and a buffer amplifier that will decrease the input resistance
of the divided inputs to less than 1 nA for load sensitive inputs.
The buffer amplifier, in turn, has a common mode input range from
at least 0.8V to VDDA voltage, and an offset less than 3 mV from
the input range from 0.8V to VDDA-0.1V.
[0444] In a preferred embodiment, the ASIC has a mode where the ADC
measurements are taken in a programmed sequence. Thus, the ASIC
includes a programmable sequencer 4266 that supervises the
measurement of up to 8 input sources for ADC measurements with the
following programmable parameters: [0445] (i) ADC MUX input [0446]
(ii) ADC range [0447] (iii) Delay time before measurement, wherein
the delays are programmable from 0 to 62 msec in 0.488 msec steps
[0448] (iv) Number of measurements for each input from 0 to 255
[0449] (v) Number of cycles of measurements: 0-255, wherein the
cycle of measurements refers to repeating the sequence of up to 8
input measurements multiple times (e.g., as an outer loop in a
program) [0450] (vi) Delay between cycles of measurement, wherein
the delays are programmable from 0 to 62 msec in 0.488 msec
steps.
[0451] The sequencer 4266 is configured to start upon receiving an
auto-measure start command, and the measurements may be stored in
the ASIC for retrieval over the SPI interface. It is noted that the
sequencer time base is programmable between the 32 kHz clock and
the High Speed RC oscillator 4262.
[0452] Sensor Diagnostics
[0453] As was previously described in detail, embodiments of the
invention are directed to use of impedance and impedance-related
parameters in, e.g., sensor diagnostic procedures and Isig/SG
fusion algorithms. To that end, in preferred embodiments, the ASIC
described herein has the capability of measuring the impedance
magnitude and phase angle of any WORK sensor electrode to the RE
and COUNTER electrode when in the potentiostat configuration. This
is done, e.g., by measuring the amplitude and phase of the current
waveform in response to a sine-like waveform superimposed on the
WORK electrode voltage. See, e.g., Diagnostic Circuitry 4255 in
FIG. 42B.
[0454] The ASIC has the capability of measuring the resistive and
capacitive components of any electrode to any electrode via, e.g.,
the Electrode Multiplexer 4250. It is noted that such measurements
may interfere with the sensor equilibrium and may require settling
time or sensor initialization to record stable electrode currents.
As discussed previously, although the ASIC may be used for
impedance measurements across a wide spectrum of frequencies, for
purposes of the embodiments of the invention, a relatively narrower
frequency range may be used. Specifically, the ASIC's sine wave
measurement capability may include test frequencies from about 0.10
Hz to about 8192 Hz. In making such measurements, the minimum
frequency resolution in accordance with an embodiment of the
invention may be limited as shown in Table 2 below:
TABLE-US-00005 TABLE 2 Min Frequency step [Hz] [Hz] .1 to 15 <1
16 to 31 1 32 to 63 2 64 to 127 4 128 to 255 8 256 to 511 16 512 to
1023 32 1024 to 2047 64 2048 to 4095 128 4096 to 8192 256
[0455] The sinewave amplitude is programmable from at least 10
mVp-p to 50 mVp-p in 5 mV steps, and from 60 mVp-p to 100 mVp-p in
10 mV steps. In a preferred embodiment, the amplitude accuracy is
better than .+-.5% or .+-.5 mV, whichever is larger. In addition,
the ASIC may measure the electrode impedance with accuracies
specified in Table 3 below:
TABLE-US-00006 TABLE 3 Impedance Phase Measurement Measurement
Frequency Range Impedance Range Accuracy Accuracy .1-10 Hz 2k to 1
Meg.OMEGA. .+-.5% .+-.0.5.degree. 10-100 Hz 1k to 100 k.OMEGA.
.+-.5% .+-.0.5.degree. 100 to 8000 Hz .5k to 20 k.OMEGA. .+-.5%
.+-.1.0.degree.
[0456] In an embodiment of the invention, the ASIC can measure the
input waveform phase relative to a time base, which can be used in
the impedance calculations to increase the accuracy. The ASIC may
also have on-chip resistors to calibrate the above electrode
impedance circuit. The on-chip resistors, in turn, may be
calibrated by comparing them to the known 1 MegOhm off-chip
precision resistor.
[0457] Data sampling of the waveforms may also be used to determine
the impedances. The data may be transmitted to an external
microprocessor with the serial peripheral interface (SPI) for
calculation and processing. The converted current data is
sufficiently buffered to be able to transfer 2000 ADC conversions
of data to an external device through the SPI interface without
losing data. This assumes a latency time of 8 msec maximum for
servicing a data transfer request interrupt.
[0458] In embodiments of the invention, rather than, or in addition
to, measuring electrode impedance with a sine wave, the ASIC may
measure electrode current with a step input. Here, the ASIC can
supply programmable amplitude steps from 10 to 200 mV with better
than 5 mV resolution to an electrode and sample (measure) the
resulting current waveform. The duration of the sampling may be
programmable to at least 2 seconds in 0.25 second steps, and the
sampling interval for measuring current may include at least five
programmable binary weighted steps approximately 0.5 msec to 8
msec.
[0459] The resolution of the electrode voltage samples is smaller
than 1 mV with a range up to .+-.0.25 volts. This measurement can
be with respect to a suitable stable voltage in order to reduce the
required dynamic range of the data conversion. Similarly, the
resolution of the electrode current samples is smaller than 0.04 uA
with a range up to 20 uA. The current measurements can be unipolar
if the measurement polarity is programmable.
[0460] In embodiments of the invention, the current measurement may
use an I-to-V converter. Moreover, the ASIC may have on-chip
resistors to calibrate the current measurement. The on-chip
resistors, in turn, may be calibrated by comparing them to the
known 1 MegOhm off-chip precision resistor. The current measurement
sample accuracy is better than .+-.3% or .+-.10 nA, whichever is
greater. As before, the converted current data is sufficiently
buffered to be able to transfer 2000 ADC conversions of data to an
external device through the SPI interface without losing data. This
assumes a latency time of 8 msec maximum for servicing a data
transfer request interrupt.
[0461] Calibration Voltage
[0462] The ASIC includes a precision voltage reference to calibrate
the ADC. The output voltage is 1.000V.+-.3% with less than .+-.1.5%
variation in production, and stability is better than 3 mV over a
temperature range of 20.degree. C. to 40.degree. C. This precision
calibration voltage may be calibrated, via the on-chip ADC, by
comparing it to an external precision voltage during manufacture.
In manufacturing, a calibration factor may be stored in a system
non-volatile memory (not on this ASIC) to achieve higher
accuracy.
[0463] The current drain of the calibration voltage circuit is
preferably less than 25 uA. Moreover, the calibration voltage
circuit is able to power down to less than 10 nA to conserve
battery power when not in use.
[0464] Temperature Sensor
[0465] The ASIC has a temperature transducer having a sensitivity
between 9 and 11 mV per degree Celsius between the range
-10.degree. C. to 60.degree. C. The output voltage of the
Temperature Sensor is such that the ADC can measure the
temperature-related voltage with the 0 to 1.22V ADC input range.
The current drain of the Temperature Sensor is preferably less than
25 uA, and the Temperature Sensor can power down to less than 10 nA
to conserve battery power when not in use.
[0466] VDD Voltage Regulator
[0467] The ASIC has a VDD voltage regulator with the following
characteristics: [0468] (i) Minimum input Voltage Range: 2.0V-4.5V.
[0469] (ii) Minimum output Voltage: 1.6-2.5V.+-.5%, with a default
of 2.0V. [0470] (iii) Dropout voltage: Vin-Vout<0.15V at
Iload=100 uA, Vin=2.0V. [0471] (iv) The output voltage is
programmable, with an accuracy within 2% of the indicated value per
Table 4 below:
TABLE-US-00007 [0471] TABLE 4 Hex vout hex vout 0 1.427 10 1.964 1
1.460 11 1.998 2 1.494 12 2.032 3 1.528 13 2.065 4 1.561 14 2.099 5
1.595 15 2.132 6 1.628 16 2.166 7 1.662 17 2.200 8 1.696 18 2.233 9
1.729 19 2.267 A 1.763 1A 2.300 B 1.796 1B 2.334 C 1.830 1C 2.368 D
1.864 1D 2.401 E 1.897 1E 2.435 F 1.931 1F 2.468
[0472] (v) The regulator can supply output of 1 mA at 2.5V with an
input voltage of 2.8V. [0473] (vi) The regulator also has input and
output pads that may be open circuited if an external regulator is
used. The current draw of the regulator circuit is preferably less
than 100 nA in this non-operational mode. [0474] (vii) The change
of output voltage from a load of 10 uA to 1 mA is preferably less
than 25 mV. [0475] (viii) Current Drain excluding output current @
1 mA load is less than 100 uA from source. [0476] (ix) Current
Drain excluding output current @ 0.1 mA load is less than 10 uA
from source. [0477] (x) Current Drain excluding output current @ 10
uA load is less than 1 uA from source.
[0478] General Purpose Comparators
[0479] The ASIC includes at least two comparators 4270, 4271
powered from VDDA. The comparators use 1.22V as a reference to
generate the threshold. The output of the comparators can be read
by the processor and will create a maskable interrupt on the rising
or falling edge determined by configuration registers.
[0480] The comparators have power control to reduce power when not
in use, and the current supply is less than 50 nA per comparator.
The response time of the comparator is preferably less than 50 usec
for a 20 mV overdrive signal, and the offset voltage is less than
.+-.8 mV.
[0481] The comparators also have programmable hysteresis, wherein
the hysteresis options include threshold=1.22V+Vhyst on a rising
input, threshold=1.22-Vhyst on a falling input, or no hysteresis
(Vhyst=25.+-.10 mV). The output from either comparator is available
to any GPIO on any power plane. (See GPIO section).
[0482] Sensor Connection Sensing Circuitry on RE
[0483] An analog switched capacitor circuit monitors the impedance
of the RE connection to determine if the sensor is connected.
Specifically, a capacitor of about 20 pF is switched at a frequency
of 16 Hz driven by an inverter with an output swing from VSS to
VDD. Comparators will sense the voltage swing on the RE pad and, if
the swing is less than a threshold, the comparator output will
indicate a connection. The above-mentioned comparisons are made on
both transitions of the pulse. A swing below threshold on both
transitions is required to indicate a connect, and a comparison
indicating high swing on either phase will indicate a disconnect.
The connect signal/disconnect signal is debounced such that a
transition of its state requires a stable indication to the new
state for at least 1/2 second.
[0484] The circuit has six thresholds defined by the following
resistances in parallel with a 20 pF capacitor: 500 k, 1 Meg, 2
MEG, 4 Meg, 8 Meg, and 16 Meg ohms. This parallel equivalent
circuit is between the RE pad and a virtual ground that can be at
any voltage between the power rails. The threshold accuracy is
better than .+-.30%.
[0485] The output of the Sensor Connect sensing circuitry is able
to programmably generate an interrupt or processor startup if a
sensor is connected or disconnected. This circuit is active
whenever the nPOR2_IN is high and the VDD and VDDA are present. The
current drain for this circuit is less than 100 nA average.
[0486] WAKEUP Pad
[0487] The WAKEUP circuitry is powered by the VDD supply, with an
input having a range from 0V to VBAT. The WAKEUP pad 4272 has a
weak pulldown of 80.+-.40 nA. This current can be derived from an
output of the BIAS_GEN 4220. The average current consumed by the
circuit is less than 50 nA with 0 v input.
[0488] The WAKEUP input has a rising input voltage threshold, Vih,
of 1.22.+-.0.1 V, and the falling input threshold is -25 mV.+-.12
mV that of the rising threshold. In preferred embodiments, the
circuit associated with the WAKEUP input draws no more than 100 nA
for any input whose value is from -0.2 to VBAT voltage (this
current excludes the input pulldown current). The WAKEUP pad is
debounced for at least 1/2 second.
[0489] The output of the WAKEUP circuit is able to programmably
generate an interrupt or processor startup if the WAKEUP pad
changes state. (See the Event Handler section). It is important to
note that the WAKEUP pad circuitry is configured to assume a low
current, <1 nA, if the Battery Protection Circuit indicates a
low battery state.
[0490] UART WAKEUP
[0491] The ASIC is configured to monitor the nRX_EXT pad 4274. If
the nRX_EXT level is continuously high (UART BREAK) for longer than
1/2 second, a UART WAKEUP event will be generated. The due to
sampling the UART WAKEUP event could be generated with a continuous
high as short as 1/4 second. The UART WAKEUP event can programmably
generate an interrupt, WAKEUP and/or a microprocessor reset
(nRESET_OD). (See the Event Handler section).
[0492] In preferred embodiments, the circuit associated with the
UART WAKEUP input draws no more than 100 nA, and the UART WAKEUP
pad circuitry is configured to assume a low current, <1 nA, if
the Battery Protection circuitry indicates a Battery Low state. The
UART Wakeup input has a rising input voltage threshold, Vih, of
1.22.+-.0.1 V. The falling input threshold is -25 mV.+-.12 mV that
of the rising threshold.
[0493] Microprocessor Wakeup Control Signals
[0494] The ASIC is able to generate signals to help control the
power management of a microprocessor. Specifically, the ASIC may
generate the following signals: [0495] (i) nSHUTDN--nSHUTDN may
control the power enable of an off chip VDD regulator. The nSHUTDN
pad is on the VBAT power rail. nSHUTDN shall be low if the Battery
Protection circuitry indicates a Battery Low state, otherwise
nSHUTDN shall be high.
[0496] (ii) VPAD_EN--VPAD_EN may control the power enable of an
external regulator that supplies VPAD power. An internal signal
that corresponds to this external signal ensures that inputs from
the VPAD pads will not cause extra current due to floating inputs
when the VPAD power is disabled. The VPAD_EN pad is an output on
the VBAT power rail. The VPAD_EN signal is low if the Battery
Protection signal indicates a low battery. The VPAD_EN signal may
be set low by a software command that starts a timer; the terminal
count of the timer forces VPAD_EN low. The following events may
cause the VPAD_EN signal to go high if the Battery Protection
signal indicates a good battery (see Event Handler for more
details): nPOR2_IN transitioning from low to high; SW/Timer
(programmable); WAKEUP transition; low to high, and/or high to low,
(programmable); Sensor Connect transition; low to high, and/or high
to low, (programmable); UART Break; and RTC Time Event
(programmable). [0497] (iii) UP_WAKEUP--UP_WAKEUP may connect to a
microprocessor wakeup pad. It is intended to wakeup the
microprocessor from a sleep mode or similar power down mode. The
UP_WAKEUP pad is an output on the VPAD power rail. The UP_WAKEUP
signal can be programmed to be active low, active high or a pulse.
The UP_WAKEUP signal may be set low by a software command that
starts a timer; the terminal count of the timer forces UP_WAKEUP
low. The following events may cause the UP_WAKEUP signal to go high
if the Battery Protection signal indicates a good battery (see
Event Handler for more details): nPOR2_IN transitioning from low to
high; SW/Timer (programmable); WAKEUP transition; low to high,
and/or high to low, (programmable); Sensor Connect transition; low
to high, and/or high to low, (programmable); UART Break; and RTC
Time Event (programmable). The WAKEUP signal may be delayed by a
programmable amount. If WAKEUP is programmed to be a pulse, the
pulse width may be programmed. [0498] (iv) CLK_32 KHZ--CLK_32 KHZ
pad may connect to a microprocessor to supply a low speed clock.
The clock is on-off programmable and programmably turns on to
wakeup events. The CLK_32 KHZ pad is an output on the VPAD power
rail. The CLK_32 KHZ signal is low if the Battery Protection signal
indicates a low battery. The CLK_32 KHZ output may be programmed
off by a programmable bit. The default is ON. The CLK_32 KHZ signal
may be disabled by a software command that starts a timer; The
terminal count of the timer forces CLK_32 KHZ low. The following
events may cause the CLK_32 KHZ signal to be enabled if the Battery
Protection signal indicates a good battery (see Event Handler for
more details): nPOR2_IN transitioning from low to high; SW/Timer
(programmable); WAKEUP transition; low to high, and/or high to low,
(programmable); Sensor Connect transition; low to high, and/or high
to low, (programmable); UART Break; RTC Time Event (programmable);
and Detection of low battery by Battery Protection Circuit. [0499]
(v) nRESET_OD--nRESET_OD may connect to a microprocessor to cause a
microprocessor reset. The nRESET_OD is programmable to wakeup
events. The nRESET_OD pad is an output on the VPAD power rail. This
pad is open drain (nfet output). The nRESET_OD signal is low if the
Battery Protection signal indicates a low battery. The nRESET_OD
active time is programmable from 1 to 200 msec. The default is 200
ms. The following events may cause the nRESET_OD signal to be
asserted low (see Event Handler for more details): nPOR2_IN;
SW/Timer (programmable); WAKEUP transition; low to high, and/or
high to low, (programmable); Sensor Connect transition; low to
high, and/or high to low, (programmable); UART Break; and RTC Time
Event (programmable). [0500] (vi) UP_INT--UP_INT may connect to a
microprocessor to communicate interrupts. The UP_INT is
programmable to wakeup events. The UP_INT pad is an output on the
VPAD power rail. The UP_INT signal is low if the Battery Protection
signal indicates a low battery. The UP_INT signal may be set high
by a software command that starts a timer; the terminal count of
the timer forces UP_INT high. The following events may cause the
UP_INT signal to be asserted high if the Battery Protection signal
indicates a good battery (see Event Handler for more details):
SW/Timer (programmable); WAKEUP transition; low to high, and/or
high to low, (programmable); Sensor Connect transition; low to high
and/or high to low, (programmable); UART Break; RTC Time Event
(programmable); Detection of low battery by Battery Protection
Circuit; and any of the ASIC interrupts when unmasked.
[0501] The ASIC has GPIO1 and GPIO0 pads able to act as boot mode
control for a microprocessor. A POR2 event will reset a 2 bit
counter whose bits map to GPIO1 & GPIO0 (MSB, LSB
respectively). A rising edge of UART break increments the counter
by one, wherein the counter counts by modulo 4, and goes to zero if
it is incremented in state 11. The boot mode counter is
pre-settable via SPI.
[0502] Event Handler/Watchdog
[0503] The ASIC incorporates an event handler to define the
responses to events, including changes in system states and input
signals. Events include all sources of interrupts (e.g. UART_BRK,
WAKE_UP, Sensor Connect, etc. . . . ). The event handler responses
to stimuli are programmable by the software through the SPI
interface. Some responses, however, may be hardwired
(non-programmable).
[0504] The event handler actions include enable/disable VPAD_EN,
enable/disable CLK_32 KHZ, assert nRESET_OD, assert UP_WAKEUP, and
assert UP_INT. The Event Watchdog Timer 1 through Timer 5 are
individually programmable in 250 msec increments from 250 msec to
16,384 seconds. The timeouts for Event Watchdog timers 6 through 8
are hardcoded. The timeout for Timer6 and Timer7 are 1 minute;
timeout for Timer8 is 5 minutes.
[0505] The ASIC also has a watchdog function to monitor the
microprocessor's responses when triggered by an event. The event
watchdog is activated when the microprocessor fails to acknowledge
the event induced activities. The event watchdog, once activated,
performs a programmable sequence of actions, Event Watchdog Timer
1-5, and followed by a hard-wired sequence of actions, Event
Watchdog Timer 6-8, to re-gain the response of the microprocessor.
The sequence of actions includes interrupt, reset, wake up, assert
32 KHz clock, power down and power up to the microprocessor.
[0506] During the sequences of actions, if the microprocessor
regains its ability to acknowledge the activities that had been
recorded, the event watchdog is reset. If the ASIC fails to obtain
an acknowledgement from the microprocessor, the event watchdog
powers down the microprocessor in a condition that will allow
UART_BRK to reboot the microprocessor and it will activate the
alarm. When activated, the alarm condition generates a square wave
with a frequency of approximately 1 kHz on the pad ALARM with a
programmable repeating pattern. The programmable pattern has two
programmable sequences with programmable burst on and off times.
The alarm has another programmable pattern that may be programmed
via the SPI port. It will have two programmable sequences with
programmable burst on and off times.
[0507] Digital to Analog (D/A)
[0508] In a preferred embodiment, the ASIC has two 8 bit D/A
converters 4276, 4278 with the following characteristics: [0509]
(i) The D/A settles in less than 1 msec with less than 50 pF load.
[0510] (ii) The D/A has at least 8 bits of accuracy. [0511] (iii)
The output range is programmable to either 0 to 1.22V or 0 to VDDA.
[0512] (iv) Temperature sensitivity of the D/A voltage reference is
less than 1 mV/.degree. C. [0513] (v) The DNL is less than 1 LSB.
[0514] (vi) Current consumed by the D/A is less than 2 uA from the
VDDA supply. [0515] (vii) Each D/A has an output 1 to a pad. [0516]
(viii) The D/A outputs are high impedance. Loading current must be
less than 1 nA. [0517] (ix) The D/A pads can be programmed to
output a digital signal from a register. The output swing is from
VSSA to VDDA.
[0518] Charger/Data Downloader Interface
[0519] The TX_EXT_OD 4280 is an open drain output whose input is
the signal on the TX UP input pad. This will allow the TX_EXT_OD
pad to be open in the UART idle condition. The TX_EXT_OD pad has a
comparator monitoring its voltage. If the voltage is above the
comparator threshold voltage for a debounce period (1/4 second),
the output, nBAT_CHRG_EN (4281), will go low. This comparator and
other associated circuitry with this function are on the VBAT
and/or VDDBU planes.
[0520] The circuitry associated with this function must allow lows
on TX_EXT_OD pad that result from normal communication with an
external device without disabling the assertion of nBAT_CHRG_EN. If
POR1 is active, nBAT_CHRG_EN will be high (not asserted). The
comparator's threshold voltage is between 0.50V and 1.2V. The
comparator will have hysteresis; The falling threshold is
approximately 25 mV lower than the rising threshold.
[0521] The nRX_EXT pad inverts the signal on this pad and output it
to RX_UP. In this way, the nRX_EXT signal will idle low. The
nRX_EXT must accept inputs up to VBAT voltage. The nRX_EXT
threshold is 1.22V.+-.3%. The output of this comparator will be
available over the SPI bus for a microprocessor to read.
[0522] The nRX_EXT pad also incorporates a means to programmably
source a current, which will be 80.+-.30 nA, with the maximum
voltage being VBAT. The ASIC layout has mask programmable options
to adjust this current from 30 nA to 200 nA in less than 50 nA
steps with a minimal number of mask layer changes. A programmable
bit will be available to block the UART break detection and force
the RX_UP high. In normal operation, this bit will be set high
before enabling the current sourcing to nRX_EXT and then set low
after the current sourcing is disabled to ensure that no glitches
are generated on RX_UP or that a UART break event is generated.
Note to implement a wet connector detector, while the current
source into nRX_EXT is active, an RX comparator output indicating a
low input voltage would indicate leakage current. The ASIC includes
a pulldown resistor approximately 100 k ohms on the nRX_EXT pad.
This pulldown will be disconnected when the current source is
active.
[0523] Sensor Connect Switch
[0524] The ASIC shall have a pad, SEN_CONN_SW (4282), which is able
to detect a low resistance to VSS (4284). The SEN_CONN_SW sources a
current from 5 to 25 uA with SEN_CONN_SW=0V and has a maximum open
circuit voltage of 0.4V. The ASIC layout has mask programmable
options to adjust this current from 1 uA to 20 uA in less than 5 uA
steps with a minimal number of mask layer changes. The SEN_CONN_SW
has associated circuitry that detects the presence of a resistance
between SEN_CONN_SW and VSSA (4234) whose threshold is between 2 k
and 15 k ohms. The average current drain of this circuit is 50 nA
max. Sampling must be used to achieve this low current.
[0525] Oscillator Calibration Circuit
[0526] The ASIC has counters whose inputs can be steered to
internal or external clock sources. One counter generates a
programmable gating interval for the other counter. The gating
intervals include 1 to 15 seconds from the 32 kHz oscillator. The
clocks that can be steered to either counter are 32 kHz, RC
oscillator, High Speed RC oscillator, and an input from any GPIO
pad.
[0527] Oscillator Bypassing
[0528] The ASIC can substitute external clocks for each of the
oscillators' outputs. The ASIC has a register that can be written
only when a specific TEST_MODE is asserted. This register has bits
to enable the external input for the RC Oscillator, and may be
shared with other analog test control signals. However, this
register will not allow any oscillator bypass bits to be active if
the TEST_MODE is not active.
[0529] The ASIC also has an input pad for an external clock to
bypass the RC Oscillator. The pad, GPIO_VBAT, is on the VBAT power
plane. The ASIC further includes a bypass enable pad for the 32 KHZ
oscillator, OSC32K_BYPASS. When high, the 32 KHZ oscillator output
is supplied by driving the OSC32 KHZ_IN pad. It is noted that,
normally, the OSC32 KHZ_IN pad is connected to a crystal.
[0530] The ASIC has inputs for an external clock to bypass the
HS_RC_OSC. The bypass is enabled by a programmable register bit.
The HS_RC_OSC may be supplied programmably by either the GPIO on
the VDD plane or by GPIOs on the VPAD plane.
[0531] SPI Slave Port
[0532] The SPI slave port includes an interface consisting of a
chip select input (SPI_nCS) 4289, a clock input (SPI_CK) 4286, a
serial data input (SPI_MOSI) 4287, and a serial data output
(SPI_MISO) 4288. The chip select input (SPI_nCS) is an active low
input, asserted by an off-chip SPI master to initiate and qualify
an SPI transaction. When SPI_nCS is asserted low, the SPI slave
port configures itself as a SPI slave and performs data
transactions based on the clock input (SPI_CK). When SPI_nCS is
inactive, the SPI slave port resets itself and remains in reset
mode. As this SPI interface supports block transfers, the master
should keep SPI_nCS low until the end of a transfer.
[0533] The SPI clock input (SPI_CK) will always be asserted by the
SPI master. The SPI slave port latches the incoming data on the
SPI_MOSI input using the rising edge of SPI_CK and driving the
outgoing data on the SPI_MISO output using the falling edge of
SPI_CK. The serial data input (SPI_MOSI) is used to transfer data
from the SPI master to the SPI slave. All data bits are asserted
following the falling edge of SPI_CK. The serial data output
(SPI_MISO) is used to transfer data from the SPI slave to the SPI
master. All data bits are asserted following the falling edge of
SPI_CK.
[0534] SPI_nCS, SPI_CK and SPI_MOSI are always driven by the SPI
master, unless the SPI master is powered down. If VPAD_EN is low,
these inputs are conditioned so that the current drain associated
with these inputs is less than 10 nA and the SPI circuitry is held
reset or inactive. SPI_MISO is only driven by the SPI slave port
when SPI_nCS is active, otherwise, SPI_MISO is tri-stated.
[0535] The chip select (SPI_nCS) defines and frames the data
transfer packet of an SPI data transaction. The data transfer
packet consists of three parts. There is a 4-bit command section
followed by a 12-bit address section, which is then followed by any
number of 8 bit data bytes. The command bit 3 is used as the
direction bit. A "1" indicates a write operation, and a "0"
indicates a read operation. The combinations of command bit 2, 1
and 0 have the following definitions. Unused combinations are
undefined. [0536] (i) 0000: read data and increment address. [0537]
(ii) 0001: read data, no change to address [0538] (iii) 0010: read
data, decrement address [0539] (iv) 1000: write data and increment
address [0540] (v) 1001: write data, no change to address [0541]
(vi) 1010: write data, decrement address [0542] (vii) x011: Test
Port Addressing
[0543] The 12-bit address section defines the starting byte
address. If SPI_nCS stays active after the first data byte, to
indicate a multi-byte transfer, the address is incremented by one
after each byte is transferred. Bit<11> of the address (of
address<11:0>) indicates the highest address bit. The address
wraps around after reaching the boundary.
[0544] Data is in the byte format, and a block transfer can be
performed by extending SPI_nCS to allow all bytes to be transferred
in one packet.
[0545] Microprocessor Interrupt
[0546] The ASIC has an output at the VPAD logic level, UP_INT, for
the purpose of sending interrupts to a host microprocessor. The
microprocessor interrupt module consists of an interrupt status
register, an interrupt mask register, and a function to logically
OR all interrupt statuses into one microprocessor interrupt. The
interrupt is implemented to support both edge sensitive and level
sensitive styles. The polarity of the interrupt is programmable.
The default interrupt polarity is TBD.
[0547] In a preferred embodiment, all interrupt sources on the AFE
ASIC will be recorded in the interrupt status register. Writing a
"1" to the corresponding interrupt status bit clears the
corresponding pending interrupt. All interrupt sources on the AFE
ASIC are mask-able through the interrupt mask register. Writing a
"1" to the corresponding interrupt mask bit enables the masking of
the corresponding pending interrupt. Writing a "0" to the
corresponding interrupt mask bit disables the masking of the
corresponding interrupt. The default state of the interrupt mask
register is TBD.
[0548] General Purpose Input/Outputs (GPIOs)/Parallel Test Port
[0549] In embodiments of the invention, the ASIC may have eight
GPIOs that operate on VPAD level signals. The ASIC has one GPIO
that operates on a VBAT level signal, and one GPIO that operates on
a VDD level signal. All off the GPIOs have at least the following
characteristics: [0550] (i) Register bits control the selection and
direction of each GPIO. [0551] (ii) The ASIC has a means to
configure the GPIOs as inputs that can be read over the SPI
interface. [0552] (iii) The ASIC has a means to configure the GPIOs
as input to generate an interrupt. [0553] (iv) The ASIC has a means
to configure each GPIO as an output to be controlled by a register
bit that can be written over the SPI interface. [0554] (v)
Programmably, the ASIC is able to output an input signal applied to
GPIO_VBAT or GPIO_VDD to a GPIO (on the VPAD power plane). (Level
shifting function). [0555] (vi) The ASIC has a means to configure
each GPIO as an input to the oscillator calibration circuit. [0556]
(vii) The ASIC has a means to configure each general purpose
comparator output to at least one GPIO on each power plane. The
polarity of the comparator output is programmable by a programmable
bit. [0557] (viii) The GPIOs have microprocessor interrupt
generating capability. [0558] (ix) The GPIOs are programmable to
open drain outputs. [0559] (x) The GPIOs on the VPAD power plane
are configurable to implement boot control of a microprocessor.
[0560] A Parallel Test Port shares the 8-bit GPIOs on the VPAD
voltage plane. The test port will be used for observing register
contents and various internal signals. The outputs of this port are
controlled by the port configuration register in the normal mode.
Writing 8'hFF to both GPIO_O1S_REG & GPIO_O2S_REG registers
will steer the test port data on the GPIO outputs, while writing
8'h00 to the GPIO_ON_REG register will disable the test port data
and enable the GPIO data onto the GPIO outputs.
[0561] Registers and pre-grouped internal signals can be observed
over this test port by addressing the target register through the
SPI slave port. The SPI packet has the command bits set to 4'b0011
followed by the 12-bit target register address. The parallel test
port continues to display the content of the addressed register
until the next Test Port Addressing command is received.
[0562] Analog Test Ports
[0563] The IC has a multiplexer feeding the pad, TP_ANAMUX (4290),
which will give visibility to internal analog circuit nodes for
testing. The IC also has a multiplexer feeding the pad, TP_RES
(4260), which will give visibility to internal analog circuit nodes
for testing. This pad will also accommodate a precision 1 meg
resistor in usual application to perform various system
calibrations.
[0564] Chip ID
[0565] The ASIC includes a 32 bit mask programmable ID. A
microprocessor using the SPI interface will be able to read this
ID. This ID is to be placed in the analog electronics block so that
changing the ID does not require a chip reroute. The design should
be such that only one metal or one contact mask change is required
to change the ID.
[0566] Spare Test Outputs
[0567] The ASIC has 16 spare digital output signals that can be
multiplexed to the 8 bit GPIO under commands sent over the SPI
interface. These signals will be organized as two 8 bit bytes, and
will be connected to VSS if not used.
[0568] Digital Testing
[0569] The ASIC has a test mode controller that uses two input
pins, TEST_CTL0 (4291) and TEST_CTL1 (4292). The test controller
generates signals from the combination of the test control signals
that have the following functionality (TEST CTL<1:0>): [0570]
(i) 0 is normal operating mode; [0571] (ii) 1 is Analog Test Mode;
[0572] (iii) 2 is Scan Mode; [0573] (iv) 3 is Analog Test mode with
the VDD EN controlled by an input to GPIO_VBAT.
[0574] The test controller logic is split between the VDD and VDDBU
power planes. During scan mode, testing LT_VBAT should be asserted
high to condition the analog outputs to the digital logic. The ASIC
has a scan chain implemented in as much digital logic as reasonably
possible for fast digital testing.
[0575] Leakage Test Pin
[0576] The ASIC has a pin called LT_VBAT that, when high, will put
all the analog blocks into an inactive mode so that only leakage
currents will be drawn from the supplies. LT_VBAT causes all
digital outputs from analog blocks to be in a stable high or low
state as to not affect interface logic current drain. The LT_VBAT
pad is on the VBAT plane with a pulldown with a resistance between
10 k and 40 k ohms.
[0577] Power Requirements
[0578] In embodiments of the invention, the ASIC includes a low
power mode where, at a minimum, the microprocessor clock is off,
the 32 kHz real time clock runs, and circuitry is active to detect
a sensor connection, a change of level of the WAKE_UP pin, or a
BREAK on the nRX_EXT input. This mode has a total current drain
from VBAT (VDDBU), VDD, and VDDA of 4.0 uA maximum. When the
Battery Protection Circuit detects a low battery (see Battery
Protection Circuit description), the ASIC goes to a mode with only
the VBAT and VDDBU power planes active. This is called Low Battery
state. The VBAT current in this mode is less than 0.3 uA.
[0579] With the ASIC programmed to the potentiostat configuration
with any one WORK electrode active in the H2O2 (peroxide) mode with
its voltage set to 1.535V, the COUNTER amplifier on with the VSET
RE set to 1.00V, a 20 MEG load resistor connected between WORK and
the COUNTER, the COUNTER and RE connected together and assuming one
WORK electrode current measurement per minute, the average current
drain of all power supplies is less than 7 uA. The measured current
after calibration should be 26.75 nA.+-.3%. Enabling additional
WORK electrodes increases the combined current drain by less than 2
uA with the WORK electrode current of 25 nA.
[0580] With the ASIC programmed to the potentiostat configuration
with the diagnostic function enabled to measure the impedance of
one of the WORK electrodes with respect to the COUNTER electrode,
the ASIC is configured to meet the following: [0581] (i) Test
frequencies: 0.1, 0.2, 0.3, 0.5 Hz, 1.0, 2.0, 5.0, 10, 100, 1000
and 4000 Hz. [0582] (ii) The measurement of the above frequencies
is not to exceed 50 seconds. [0583] (iii) The total charge supplied
to the ASIC is less than 8 millicoulombs.
[0584] Environment
[0585] In preferred embodiments of the invention, the ASIC: [0586]
(i) Operates and meets all specifications in the commercial
temperature range of 0 to 70.degree. C. [0587] (ii) Functionally
operates between -20.degree. C. and 80.degree. C., but may do so
with reduced accuracy. [0588] (iii) Is expected to operate after
being stored in a temperature range of -30 to 80.degree. C. [0589]
(iv) Is expected to operate in the relative humidity range of 1% to
95%. [0590] (v) ESD protection is greater than .+-.2 KV, Human Body
Model on all pins when packaged in a TBD package, unless otherwise
specified. [0591] (vi) Is configured such that the WORK1-WORK5,
COUNTER, RE, TX_EXT_OD, and nRX_EXT pads withstand greater than
.+-.4 KV Human Body Model. [0592] (vii) Is configured such that the
leakage current of the WORK1-WORK5 and RE pads is less than 0.05 nA
at 40.degree. C.
[0593] In embodiments of the invention, the ASIC may be fabricated
in 0.25 micron CMOS process, and backup data for the ASIC is on DVD
disk, 916-TBD.
[0594] As described in detail hereinabove, the ASIC provides the
necessary analog electronics to: (i) support multiple potentiostats
and interface with multi-terminal glucose sensors based on either
Oxygen or Peroxide; (ii) interface with a microcontroller so as to
form a micropower sensor system; and (iii) implement EIS
diagnostics based on measurement of EIS-based parameters. The
measurement and calculation of EIS-based parameters will now be
described in accordance with embodiments of the inventions
herein.
[0595] As mentioned previously, the impedance at frequencies in the
range from 0.1 Hz to 8 kHz can provide information as to the state
of the sensor electrodes. The AFE IC circuitry incorporates
circuitry to generate the measurement forcing signals and circuitry
to make measurements used to calculate the impedances. The design
considerations for this circuitry include current drain, accuracy,
speed of measurement, the amount of processing required, and the
amount of on time required by a control microprocessor.
[0596] In a preferred embodiment of the invention, the technique
the AFE IC uses to measure the impedance of an electrode is to
superimpose a sine wave voltage on the dc voltage driving an
electrode and to measure the phase and amplitude of the resultant
AC current. To generate the sine wave, the AFE IC incorporates a
digitally-synthesized sine wave current. This digital technique is
used because the frequency and phase can be precisely controlled by
a crystal derived timebase and it can easily generate frequencies
from DC up to 8 kHz. The sine wave current is impressed across a
resistor in series with a voltage source in order to add the AC
component to the electrode voltage. This voltage is the AC forcing
voltage. It is then buffered by an amplifier that drives a selected
sensor electrode.
[0597] The current driving the electrode contains the resultant AC
current component from the forcing sine wave and is converted to a
voltage. This voltage is then processed by multiplying it by a
square wave that has a fixed phase relative to the synthesized sine
wave. This multiplied voltage is then integrated. After the end of
a programmable number of integration intervals--an interval being
an integral number of 1/2 periods of the driving sine wave--the
voltage is measured by the ADC. By calculations involving the
values of the integrated voltages, the real and imaginary parts of
the impedance can be obtained.
[0598] The advantage of using integrators for the impedance
measurement is that the noise bandwidth of the measurement is
reduced significantly with respect to merely sampling the
waveforms. Also, the sampling time requirements are significantly
reduced which relaxes the speed requirement of the ADC.
[0599] FIG. 45 shows the main blocks of the EIS circuitry in the
AFE IC (designated by reference numeral 4255 in FIG. 42B). The IDAC
4510 generates a stepwise sine wave in synchrony with a system
clock. A high frequency of this system clock steps the IDAC through
the lookup table that contains digital code. This code drives the
IDAC, which generates an output current approximating a sine wave.
This sine wave current is forced across a resistor to give the AC
component, Vin_ac, with the DC offset, VSET8 (4520). When the IDAC
circuit is disabled, the DC output voltage reverts to VSET8, so the
disturbance to the electrode equilibrium is minimized. This voltage
is then buffered by an amplifier 4530 that drives the electrode
through a resistor in series, Rsense. The differential voltage
across Rsense is proportional to the current. This voltage is
presented to a multiplier 4540 that multiplies the voltage by
either +1 or -1. This is done with switches and a differential
amplifier (instrumentation amplifier). The system clock is divided
to generate the phase clock 4550 which controls the multiply
function and can be set to 0, 90, 180 or 270 degrees relative to
the sine wave.
[0600] The plots in FIGS. 46A-46F and 47A-47F show a simulation of
the signals of the circuit shown in FIG. 45 to a current that has 0
degree phase shift, which represents a real resistance. For these
example simulations, the simulation input values were selected to
give the current sense voltage equal to 0.150V. To obtain enough
information to derive the impedance and phase, two integrations are
required: one with a 0 degree phase multiply (FIGS. 46A-46F) and
one with a 90 degree phase multiply (FIGS. 47A-47F).
[0601] Calculation of Impedance
[0602] The equations describing the integrator output are provided
below. For simplicity, only 1/2 of a sine wave period is
considered. As can be seen from the plots of FIGS. 46A-46F and
47A-47F, total integrator output will be approximately the
integrated value of a 1/2 sine wave cycle multiplied by the number
of 1/2 cycles integrated. It is noted that the multiplying switches
in relation with the integrate time perform a "gating" function of
the signal to the integrator; this can be viewed as setting the
limits of integration. The multiplying signal has a fixed phase to
the generated sine wave. This can be set to 0, 90, 180, or 270
degrees with software. If the sine wave is in phase (0 degree
shift) with respect to the multiply square wave, the limits of
integration will be .pi.(180.degree.) and 0 (0.degree.). If the
sine wave is shifted by 90 degrees, the limits of integration can
be viewed as 3/4.pi. (270.degree.) and 1/4.pi. (90.degree.).
[0603] The formulas with the multiplying square wave in-phase
(0.degree.) with respect to the driving sine wave are shown below.
This will yield a voltage that is proportional to the real
component of the current. It is noted that .PHI. is the phase shift
of the sine wave relative to the multiplying square wave; Vout is
the integrator output, and Aamp1 is the current sine wave
amplitude. Also the period of the sine wave is 1/f, and RC is the
time constant of the integrator.
v out 0 = .intg. 0 1 2 f V i n RC .differential. t = A ampl RC
.intg. 0 1 2 f sin [ 2 .pi. f .differential. t + .phi. ] = - A ampl
2 .pi. fRC cos [ 2 .pi. f t + .phi. ] 0 1 2 f ##EQU00007## v out 0
= - A ampl 2 .pi. fRC [ cos [ .pi. + .phi. ] - cos [ .phi. ] ]
##EQU00007.2## cos ( .phi. + .PHI. ) = cos ( .phi. ) cos ( .PHI. )
- sin ( .phi. ) sin ( .PHI. ) ; cos ( .pi. + .phi. ) = - cos (
.phi. ) ; ##EQU00007.3## cos ( - .phi. ) = cos ( .phi. )
##EQU00007.4## v out 0 = - A ampl 2 .pi. fRC [ cos ( .pi. + .phi. )
- cos ( .phi. ) ] = A ampl 2 .pi. fRC [ cos ( .phi. ) + cos ( .phi.
) ] = A ampl .pi. fRC cos ( .phi. ) ##EQU00007.5##
[0604] If .PHI.=0,
v out 0 = A ampl .pi. fRC . ##EQU00008##
This corresponds to the real part of the current.
[0605] For the multiplying square wave quadrature phase
(90.degree.) with respect to the driving sine wave to yield an
output proportional to the imaginary component of the current:
v out 90 = .intg. 1 4 f 3 4 f V i n RC .differential. t = A ampl RC
.intg. 1 4 f 3 4 f sin [ 2 .pi. f .differential. t + .phi. ] = - A
ampl 2 .pi. fRC cos [ 2 .pi. f t + .phi. ] 3 4 f 1 4 f ##EQU00009##
v out 90 = - A ampl 2 .pi. fRC [ cos [ 3 2 .pi. + .phi. ] - cos [ 1
2 .pi. + .phi. ] ] ##EQU00009.2## cos ( .phi. + .PHI. ) = cos (
.phi. ) cos ( .PHI. ) - sin ( .phi. ) sin ( .PHI. ) ; cos [ 3 2
.pi. + .phi. ] = sin ( .phi. ) ; ##EQU00009.3## cos [ 1 2 .pi. +
.phi. ] = - sin ( .phi. ) ##EQU00009.4## v out 90 = - A ampl 2 .pi.
fRC [ sin ( .phi. ) + sin ( .phi. ) ] = - A ampl 2 .pi. fRC [ sin (
.phi. ) + sin ( .phi. ) ] = - A ampl .pi. fRC sin ( .phi. )
##EQU00009.5##
[0606] If .PHI.=0,
v out 90 = A ampl .pi. fRC sin ( .phi. ) = 0. ##EQU00010##
This corresponds to the imaginary part of the current.
[0607] In the first example plot shown in FIGS. 46A-46F, A.sub.amp1
is 0.150 v, the frequency is 1 kHz, .PHI.=0, the RC for the
integrator is 20M ohm and 25 pF which gives RC=0.5 msec. Plugging
in those numbers into the equations, gives 0.09549 v, which
favorably compares to the integrator output of the plot in FIG. 46.
It is noted that the integrator output over the period of
integration is the delta voltage from the start of integration to
the measurement.
[0608] For the 90.degree. square wave multiply, the result should
be 0 since sin(0)=0. The simulation result is close to this
value.
[0609] To calculate the phase:
since
v out 90 v out 0 = sin ( .phi. ) cos ( .phi. ) , ##EQU00011##
it follows:
.phi. = arctan sin ( .phi. ) cos ( .phi. ) = arctan v out 90 v out
0 ##EQU00012##
where V.sub.out90 is the integrator output with the 90.degree.
phase shift for the multiply, and V.sub.out0 is the integrator
output for the 0.degree. phase shift. The V.sub.out90 and
V.sub.out0 outputs must be integrated for the same number of 1/2
cycles or normalized by the number of cycles. It is important to
note that, in the actual software (e.g., ASIC) implementation, only
integral cycles (360.degree.) are allowed because an integral
number of cycles compensates for any offset in the circuitry before
the multiplier.
[0610] The magnitude of the current can be found from
I = A ampl R sense and A ampl = v out_ 90 .pi. fRC sin ( .phi. ) or
##EQU00013## A ampl = v out_ 0 .pi. fRC cos ( .phi. ) ,
##EQU00013.2##
or A.sub.amp1=.pi.fRC {square root over
(V.sub.out.sub._.sub.0.sup.2+V.sub.out.sub._.sub.90.sup.2)}. This
current has the phase angle as calculated above.
[0611] The above analysis shows that one can determine the current
amplitude and its phase with respect to the multiplying signal. The
forcing voltage is generated in a fixed phase (0, 90, 180 or 270
degrees) with respect to the multiplying signal--this is done
digitally so that it is precisely controlled. But there is at least
one amplifier in the path before the forcing sine wave is applied
to the electrode; this will introduce unwanted phase shift and
amplitude error. This can be compensated for by integrating the
forcing sine wave signal obtained electrically near the electrode.
Thus, the amplitude and any phase shift of the forcing voltage can
be determined. Since the path for both the current and voltage
waveform will be processed by the same circuit, any analog circuit
gain and phase errors will cancel.
[0612] Since the variable of interest is the impedance, it may not
be necessary to actually calculate the A.sub.amp1. Because the
current waveform and the voltage waveform are integrated through
the same path, there exists a simple relationship between the ratio
of the current and the voltage. Calling the integrated current
sense voltage V.sub.I.sub._.sub.out and the integrated electrode
voltage as V.sub.V.sub._.sub.out with the additional subscript to
describe the phase of the multiplying function:
I = A I_ampl R sense .angle..phi. = V I_out _ 0 .pi. fRC cos (
.phi. ) R sense .angle..phi. ; ##EQU00014## V = A V_ampl
.angle..theta. = V V_out _ 0 .pi. fRC cos ( .theta. )
.angle..theta. ##EQU00014.2##
[0613] The impedance will be the voltage divided by the current.
Thus,
Z = V .angle..theta. I .angle..phi. = V V_out _ 0 .pi. fRC
.angle..theta. cos ( .theta. ) V I_out _ 0 .pi. fRC .angle..phi.
cos ( .phi. ) R sense = R sense * V V_out _ 0 cos ( .phi. ) V I_out
_ 0 cos ( .theta. ) .angle. ( .theta. - .phi. ) ##EQU00015##
[0614] The magnitudes of the voltage and the current can also be
obtained from the square root of the squares of the 0 and 90 degree
phase integration voltages. As such, the following may also be
used:
Z = V .angle..theta. I .angle..phi. = V V_out _ 0 2 + V V_out _ 90
2 .angle..theta. V I _out _ 0 2 + V I _out _ 90 2 .angle..theta. =
R sense * V V_out _ 0 2 + V V_out _ 90 2 V I _out _ 0 2 + V I _out
_ 90 2 .angle. ( .theta. - .phi. ) ##EQU00016##
[0615] The integration of the waveforms may be done with one
hardware integrator for the relatively-higher frequencies, e.g.,
those above about 256 Hz. The high frequencies require four
measurement cycles: (i) one for the in-phase sensor current; (ii)
one for the 90 degree out of phase sensor current; (iii) one for
the in-phase forcing voltage; and (iv) one for the 90 degree out of
phase forcing voltage.
[0616] Two integrators may be used for the relatively-lower
frequencies, e.g., those lower than about 256 Hz, with the
integration value consisting of combining integrator results
numerically in the system microprocessor. Knowing how many
integrations there are per cycle allows the microprocessor to
calculate the 0 and 90 degree components appropriately.
[0617] Synchronizing the integrations with the forcing AC waveform
and breaking the integration into at least four parts at the lower
frequencies will eliminate the need for the hardware multiplier as
the combining of the integrated parts in the microprocessor can
accomplish the multiplying function. Thus, only one integration
pass is necessary for obtaining the real and imaginary current
information. For the lower frequencies, the amplifier phase errors
will become smaller, so below a frequency, e.g., between 1 Hz and
50 Hz, and preferably below about 1 Hz, the forcing voltage phase
will not need to be determined. Also, the amplitude could be
assumed to be constant for the lower frequencies, such that only
one measurement cycle after stabilization may be necessary to
determine the impedance.
[0618] As noted above, whereas one hardware integrator is used for
the relatively-higher frequencies, for the relatively-lower
frequencies, two integrators may be used. In this regard, the
schematic in FIG. 45 shows the EIS circuitry in the AFE IC as used
for the relatively-higher EIS frequencies. At these frequencies,
the integrator does not saturate while integrating over a cycle. In
fact, multiple cycles are integrated for the highest frequencies as
this will provide a larger output signal which results in a larger
signal to noise ratio.
[0619] For the relatively-lower frequencies, such as, e.g., those
below about 500 Hz, the integrator output can saturate with common
parameters. Therefore, for these frequencies, two integrators are
used that are alternately switched. That is, while a first
integrator is integrating, the second integrator is being read by
the ADC and then is reset (zeroed) to make it ready to integrate
when the integration time for first integrator is over. In this
way, the signal can be integrated without having gaps in the
integration. This would add a second integrator and associated
timing controls to the EIS circuitry shown in FIG. 45.
[0620] Stabilization Cycle Considerations
[0621] The above analysis is for steady state conditions in which
the current waveform does not vary from cycle to cycle. This
condition is not met immediately upon application of a sine wave to
a resistor-capacitor (RC) network because of the initial state of
the capacitor. The current phase starts out at 0 degrees and
progresses to the steady state value. However, it would be
desirable for the measurement to consume a minimum amount of time
in order to reduce current drain and also to allow adequate time to
make DC sensor measurements (Isigs). Thus, there is a need to
determine the number of cycles necessary to obtain sufficiently
accurate measurements.
[0622] The equation for a simple RC circuit--with a resistor and
capacitor in series--is
v a c = R * I ( t ) + 1 C .intg. I ( t ) .differential. t
##EQU00017##
[0623] Solving the above for I(t) gives:
I ( t ) = - 1 RC [ V c 0 C + .omega. V m R [ .omega. 2 + 1 R 2 C 2
] ] e - t RC + V m R [ 1 [ .omega. 2 + 1 R 2 C 2 ] ] [ .omega. 2
sin ( .omega. t ) + .omega. RC cos .omega. t ] ##EQU00018##
where V.sub.c0 is the initial value of the capacitor voltage,
V.sub.m is the magnitude of the driving sine wave, and .omega. is
the radian frequency (2.pi.f).
[0624] The first term contains the terms defining the non-steady
state condition. One way to speed the settling of the system would
be to have the first term equal 0, which may be done, e.g., by
setting
V cinit C = .omega. V m R [ .omega. 2 + 1 R 2 C 2 ] or V cinit = RC
.omega. V m [ R 2 C 2 .omega. 2 + 1 ] ##EQU00019##
[0625] While this may not be necessary in practice, it is possible
to set the initial phase of the forcing sine wave to jump
immediately from the DC steady state point to V.sub.cinit. This
technique may be evaluated for the specific frequency and
anticipated phase angle to find the possible reduction in time.
[0626] The non-steady state term is multiplied by the exponential
function of time. This will determine how quickly the steady state
condition is reached. The RC value can be determined as a first
order approximation from the impedance calculation information.
Given the following:
X c = 1 .omega. C = Z sin .phi. ##EQU00020##
and R=Z cos .phi., it follows that
RC = Z cos .phi. .omega. Z sin .phi. = 1 .omega. tan .phi.
##EQU00021##
[0627] For a sensor at 100 Hz with a 5 degree phase angle, this
would mean a time constant of 18.2 msec. For settling to less than
1%, this would mean approximately 85 msec settling time or 8.5
cycles. On the other hand, for a sensor at 0.10 Hz with a 65 degree
phase angle, this would mean a time constant of 0.75 sec. For
settling to less than 1%, this would mean approximately 3.4 sec
settling time.
[0628] Thus, in embodiments of the invention as detailed
hereinabove, the ASIC includes (at least) 7 electrode pads, 5 of
which are assigned as WORK electrodes (i.e., sensing electrodes, or
working electrodes, or WEs), one of which is labeled COUNTER (i.e.,
counter electrode, or CE), and one that is labeled REFERENCE (i.e.,
reference electrode, or RE). The counter amplifier 4321 (see FIG.
42B) may be programmably connected to the COUNTER, the REFERENCE,
and/or any of the WORK assigned pads, and in any combination
thereof. As has been mentioned, embodiments of the invention may
include, e.g., more than five WEs. In this regard, embodiments of
the invention may also be directed to an ASIC that interfaces with
more than 5 working electrodes.
[0629] It is important to note that, with the ASIC as described
herein, each of the above-mentioned five working electrodes, the
counter electrode, and the reference electrode is individually and
independently addressable. As such, any one of the 5 working
electrodes may be turned on and measure Isig (electrode current),
and any one may be turned off. Moreover, any one of the 5 working
electrodes may be operably connected/coupled to the EIS circuitry
for measurement of EIS-related parameters, e.g., impedance and
phase. In other words, EIS may be selectively run on any one or
more of the working electrodes. In addition, the respective voltage
level of each of the 5 working electrodes may be independently
programmed in amplitude and sign with respect to the reference
electrode. This has many applications, such as, e.g., changing the
voltage on one or more electrodes in order to make the electrode(s)
less sensitive to interference.
[0630] In embodiments where two or more working electrodes are
employed as redundant electrodes, the EIS techniques described
herein may be used, e.g., to determine which of the multiplicity of
redundant electrodes is functioning optimally (e.g., in terms of
faster start-up, minimal or no dips, minimal or no sensitivity
loss, etc.), so that only the optimal working electrode(s) can be
addressed for obtaining glucose measurements. The latter, in turn,
may drastically reduce, if not eliminate, the need for continual
calibrations. At the same time, the other (redundant) working
electrode(s) may be: (i) turned off, which would facilitate power
management, as EIS may not be run for the "off" electrodes; (ii)
powered down; and/or (iii) periodically monitored via EIS to
determine whether they have recovered, such that they may be
brought back on line. On the other hand, the non-optimal
electrode(s) may trigger a request for calibration. The ASIC is
also capable of making any of the electrodes--including, e.g., a
failed or off-line working electrode--the counter electrode. Thus,
in embodiments of the invention, the ASIC may have more than one
counter electrode.
[0631] While the above generally addresses simple redundancy,
wherein the redundant electrodes are of the same size, have the
same chemistry, the same design, etc., the above-described
diagnostic algorithms, fusion methodologies, and the associated
ASIC may also be used in conjunction with spatially distributed,
similarly sized or dissimilarly sized, working electrodes as a way
of assessing sensor implant integrity as a function of implant
time. Thus, in embodiments of the invention, sensors may be used
that contain electrodes on the same flex that may have different
shapes, sizes, and/or configurations, or contain the same or
different chemistries, used to target specific environments.
[0632] For example, in one embodiment, one or two working
electrodes may be designed to have, e.g., considerably better
hydration, but may not last past 2 or 3 days. Other working
electrode(s), on the other hand, may have long-lasting durability,
but slow initial hydration. In such a case, an algorithm may be
designed whereby the first group of working electrode(s) is used to
generate glucose data during early wear, after which, during
mid-wear, a switch-over may be made (e.g., via the ASIC) to the
second group of electrode(s). In such a case, the fusion algorithm,
e.g., may not necessarily "fuse" data for all of the WEs, and the
user/patient is unaware that the sensing component was switched
during mid-wear.
[0633] In yet other embodiments, the overall sensor design may
include WEs of different sizes. Such smaller WEs generally output a
lower Isig (smaller geometric area) and may be used specifically
for hypoglycemia detection/accuracy, while larger WEs--which output
a larger Isig--may be used specifically for euglycemia and
hyperglycemia accuracy. Given the size differences, different EIS
thresholds and/or frequencies must be used for diagnostics as among
these electrodes. The ASIC, as described hereinabove, accommodates
such requirements by enabling programmable, electrode-specific, EIS
criteria. As with the previous example, signals may not necessarily
be fused to generate an SG output (i.e., different WEs may be
tapped at different times).
[0634] As was noted previously, the ASIC includes a programmable
sequencer 4266 that commands the start and stop of the stimulus and
coordinates the measurements of the EIS-based parameters for
frequencies above about 100 Hz. At the end of the sequence, the
data is in a buffer memory, and is available for a microprocessor
to quickly obtain (values of) the needed parameters. This saves
time, and also reduces system power requirements by requiring less
microprocessor intervention.
[0635] For frequencies lower than about 100 Hz, the programmable
sequencer 4266 coordinates the starting and stopping of the
stimulus for EIS, and buffers data. Either upon the end of the
measurement cycle, or if the buffer becomes close to full, the ASIC
may interrupt the microprocessor to indicate that it needs to
gather the available data. The depth of the buffer will determine
how long the microprocessor can do other tasks, or sleep, as the
EIS-based parameters are being gathered. For example, in one
preferred embodiment, the buffer is 64 measurements deep. Again,
this saves energy as the microprocessor will not need to gather the
data piecemeal. It is also noted that the sequencer 4266 also has
the capability of starting the stimulus at a phase different from
0, which has the potential of settling faster.
[0636] The ASIC, as described above, can control the power to a
microprocessor. Thus, for example, it can turn off the power
completely, and power up the microprocessor, based on detection of
sensor connection/disconnection using, e.g., a mechanical switch,
or capacitive or resistive sensing. Moreover, the ASIC can control
the wakeup of a microprocessor. For example, the microprocessor can
put itself into a low-power mode. The ASIC can then send a signal
to the microprocessor if, e.g., a sensor connect/disconnect
detection is made by the ASIC, which signal wakes up the processor.
This includes responding to signals generated by the ASIC using
techniques such as, e.g., a mechanical switch or a capacitive-based
sensing scheme. This allows the microprocessor to sleep for long
periods of time, thereby significantly reducing power drain.
[0637] It is important to reiterate that, with the ASIC as
described hereinabove, both oxygen sensing and peroxide sensing can
be performed simultaneously, because the five (or more) working
electrodes are all independent, and independently addressable, and,
as such, can be configured in any way desired. In addition, the
ASIC allows multiple thresholds for multiple markers, such that EIS
can be triggered by various factors--e.g., level of V.sub.cntr,
capacitance change, signal noise, large change in Isig, drift
detection, etc.--each having its own threshold(s). In addition, for
each such factor, the ASIC enables multiple levels of
thresholds.
[0638] In yet another embodiment of the invention, EIS may be used
as an alternative plating measurement tool, wherein the impedance
of both the working and counter electrodes of the sensor substrate
may be tested, post-electroplating, with respect to the reference
electrode. More specifically, existing systems for performing
measurements of the sensor substrate which provide an average
roughness of the electrode surface sample a small area from each
electrode to determine the average roughness (Ra) of that small
area. For example, currently, the Zygo Non-contact Interferometer
is used to quantify and evaluate electrode surface area. The Zygo
interferometer measures a small area of the counter and working
electrodes and provides an average roughness value. This
measurement correlates the roughness of each sensor electrode to
their actual electrochemical surface area. Due to the limitations
of systems that are currently used, it is not possible, from a
manufacturing throughput point of view, to measure the entire
electrode surface, as this would be an extremely time-consuming
endeavor.
[0639] In order to measure the entire electrode in a meaningful and
quantitative manner, an EIS-based methodology for measuring surface
area has been developed herein that is faster than current, e.g.,
Zygo-based, testing, and more meaningful from a sensor performance
perspective. Specifically, the use of EIS in electrode surface
characterization is advantageous in several respects. First, by
allowing multiple plates to be tested simultaneously, EIS provides
a faster method to test electrodes, thereby providing for higher
efficiency and throughput, while being cost-effective and
maintaining quality.
[0640] Second, EIS is a direct electrochemical measurement on the
electrode under test, i.e., it allows measurement of EIS-based
parameter(s) for the electrode and correlates the measured value to
the true electrochemical surface area of the electrode. Thus,
instead of taking an average height difference over a small section
of the electrode, the EIS technique measures the double layer
capacitance (which is directly related to surface area) over the
whole electrode surface area and, as such, is more representative
of the properties of the electrode, including the actual surface
area. Third, EIS testing is non-destructive and, as such, does not
affect future sensor performance. Fourth, EIS is particularly
useful where the surface area to be measured is either fragile or
difficult to easily manipulate.
[0641] For purposes of this embodiment of the invention, the
EIS-based parameter of interest is the Imaginary impedance (Zim),
which may be obtained, as discussed previously, based on
measurements of the impedance magnitude (|Z|) in ohms and the phase
angle (.PHI.) in degrees of the electrode immersed in an
electrolyte. It has been found that, in addition to being a
high-speed process, testing using the electrochemical impedance of
both the Counter Electrode (CE) and the WE is an accurate method of
measuring the surface area of each electrode. This is also
important because, although the role of electrode size in glucose
sensor performance is dictated, at least in part, by the oxidation
of the hydrogen peroxide produced by the enzymatic reaction of
glucose with GOX, experiments have shown that an increased WE
surface area reduces the number of low start-up events and improves
sensor responsiveness--both of which are among the potential
failure modes that were previously discussed at some length.
[0642] Returning to the imaginary impedance as the EIS-based
parameter of interest, it has been found that the key parameters
that drive the electrode surface area, and consequently, its
imaginary impedance values are: (i) Electroplating conditions (time
in seconds and current in micro Amperes); (ii) EIS frequency that
best correlates to surface area; (iii) the number of measurements
conducted on a single electrode associated to the electrolyte used
in the EIS system; and (iv) DC Voltage Bias.
[0643] In connection with the above parameters, experiments have
shown that using Platinum plating solution as the electrolyte
presents a poor correlation between the imaginary impedance and
surface area across the entire spectrum. However, using Sulfuric
Acid (H2SO4) as the electrolyte presents very good correlation
data, and using Phosphate Buffered saline Solution with zero mg/ml
of Glucose (PBS-0) presents even better correlation data, between
imaginary impedance and Surface Area Ratio (SAR), especially
between the relatively-lower frequencies of 100 Hz and 5 Hz.
Moreover, fitted regression analysis using a cubic regression model
indicates that, in embodiments of the invention, the best
correlation may occur at a frequency of 10 Hz. In addition, it has
been found that reducing the Bias voltage from 535 mV to zero
dramatically reduces the day-to-day variability in the imaginary
impedance measurement.
[0644] Using the above parameters, the limits of acceptability of
values of imaginary impedance can be defined for a given sensor
design. Thus, for example, for the Comfort Sensor manufactured by
Medtronic Minimed, the imaginary impedance measured between the WE
and the RE (Platinum mesh) must be greater than, or equal to, -100
Ohms. In other words, sensors with an imaginary impedance value
(for the WE) of less than -100 Ohms will be rejected. For the WE,
an impedance value of greater than, or equal to, -100 Ohms
corresponds to a surface area that is equal to, or greater than,
that specified by an equivalent Ra measurement of greater than 0.55
um.
[0645] Similarly, the imaginary impedance measured between the CE
and the RE (Platinum mesh) must be greater than, or equal to, -60
Ohms, such that sensors with an imaginary impedance value (for the
CE) of less than -60 Ohms will be rejected. For the CE, an
impedance value of greater than, or equal to, -60 Ohms corresponds
to a surface area that is equal to, or greater than, that specified
by an equivalent Ra measurement greater than 0.50 um.
[0646] In accordance with embodiments of the invention, an
equivalent circuit model as shown in FIG. 48 may be used to model
the measured EIS between the working and reference electrodes, WE
and RE, respectively. The circuit shown in FIG. 48 has a total of
six (6) elements, which may be divided into three general
categories: (i) reaction-related elements; (ii) Membrane-related
elements; and (iii) solution-related elements. In the latter
category, Rsol is the solution resistance, and corresponds to the
properties of the environment external to the sensor system (e.g.,
interstitial fluid in vivo).
[0647] The reaction-related elements include R.sub.p, which is the
polarization resistance (i.e., resistance to voltage bias and
charge transfer between the electrode and electrolyte), and Cdl,
which is the double layer capacitance at the electrode-electrolyte
interface. It is noted that, while, in this model, the double layer
capacitance is shown as a constant phase element (CPE) due to
inhomogeneity of the interface, it can also be modeled as a pure
capacitance. As a CPE, the double layer capacitance has two
parameters: Cdl, which denotes the admittance, and a, which denotes
the constant phase of the CPE (i.e., how leaky the capacitor is).
The frequency-dependent impedance of the CPE may be calculated
as
Z CPE = 1 Cdl ( j .omega. ) .alpha. . ##EQU00022##
Thus, the model includes two (2) reaction-related elements--R.sub.p
and Cdl--which are represented by a total of three (3) parameters:
R.sub.p, Cdl, and .alpha..
[0648] The membrane-related elements include Rmem, which is the
membrane resistance (or resistance due to the chemistry layer), and
Cmem, which is the membrane capacitance (or capacitance due to the
chemistry layer). Although Cmem is shown in FIG. 48 as a pure
capacitance, it can also be modeled as a CPE in special cases. As
shown, W is the bounded Warburg element, and has two parameters:
Y.sub.0, which denotes the admittance of the Warburg element due to
glucose/H.sub.2O.sub.2 diffusion within the chemistry layer, and
.lamda., which denotes the diffusion time constant of the Warburg
element. It is noted that Warburg may also be modeled in other ways
(e.g., unbounded). The frequency-dependent impedance of the bounded
Warburg element may be calculated as
Z W = 1 Y 0 j .omega. .times. coth ( .lamda. j .omega. )
##EQU00023##
[0649] Thus, the model includes three (3) membrane-related
elements--Rmem, Cmem, and W--which are represented by a total of
four (4) parameters: Rmem, Cmem, Y.sub.0, and .lamda..
[0650] The top portion of FIG. 48 shows the overall structure of a
sensor in accordance with embodiments of the invention, where
Platinum Black refers to the electrode. Here, it is important to
note that, while a single electrode is depicted, this is by way of
illustration only, and not limitation, as the model may be applied
to sensors having a greater number of layers, and a larger number
of electrodes, than the illustrative 3-layer, single-electrode
structure shown in FIG. 48. As described previously herein, GLM is
the sensor's glucose limiting membrane, HSA is human serum albumin,
GOX is glucose oxidase enzyme (used as the catalyst), and Solution
refers to the environment in which the electrode is disposed, such
as, e.g., a user's bodily fluid(s).
[0651] In the ensuing discussion, the equivalent circuit model of
FIG. 48 will be used to explain some of the physical properties of
the sensor behavior. Nevertheless, it should be mentioned that,
depending on how the glucose diffusion is modeled, other circuit
configurations may also be possible. In this regard, FIGS. 49A-49C
show illustrations of some additional circuit models, some of which
include a larger number of elements and/or parameters. For purposes
of the invention, however, it has been discovered that the circuit
model of FIG. 48, wherein the mass transport limitation--i.e., the
Warburg component--is attributed to glucose diffusion through the
membrane, provides the best fit vis-a-vis empirical data. FIG. 50A
is a Nyquist plot showing that the equivalent circuit simulation
5020 fits the empirical data 5010 very closely. FIG. 50B is an
enlarged diagram of the high-frequency portion of FIG. 50A, showing
that the simulation tracks the actual sensor data quite accurately
in that region as well.
[0652] Each of the above-described circuit elements and parameters
affects the EIS output in various ways. FIG. 51 shows a Nyquist
plot, wherein Cdl increases in the direction of Arrow A. As can be
seen, as the value of Cdl increases, the length of the (lower
frequency) Nyquist plot decreases, and its slope increases. Thus,
the length of the Nyquist plot decreases from plot 5031 to plot
5039, with each of plots 5033, 5035, and 5037 having respective
lengths that progressively decrease as Cdl increases from plot 5031
to plot 5039. Conversely, the slope of the Nyquist plot increases
from plot 5031 to plot 5039, with each of plots 5033, 5035, and
5037 having respective slopes that progressively increase as Cdl
increases from plot 5031 to plot 5039. The higher-frequency region
of the Nyquist plot, however, is generally not affected.
[0653] FIG. 52 shows a Nyquist plot, wherein a increases in the
direction of Arrow A. Here, as a increases, the slope of the
Nyquist plot increases in the lower frequency region. In FIG. 53,
as R.sub.p increases in the direction of Arrow A, the length and
the slope of the lower-frequency Nyquist plot increase. The higher
the Rp, the higher the amount of resistance to the chemical
reaction and, therefore, the slower the rate of electron and ion
exchange. Thus, phenomenologically, FIG. 53 shows that the length
and the slope of the lower-frequency Nyquist plot increase as the
electron-ion exchange rate decreases--i.e., as the resistance to
the chemical reaction increases, which, in turn, means a lower
current (Isig) output. Again, there is minimal to no effect on the
higher-frequency region of the Nyquist plot.
[0654] The effect of change in the Warburg admittance is shown in
FIG. 54. As the Warburg admittance increases in the direction of
Arrow A, both the length and the slope of the lower-frequency
Nyquist plot increase. Phenomenologically, this means that the
length and the slope of the lower-frequency Nyquist plot tend to
increase as the influx of the reactant increases. In FIG. 55, as
.lamda. increases in the direction of Arrow A, the slope of the
Nyquist plot decreases.
[0655] In contrast to the above-described elements and parameters,
the membrane-related elements and parameters generally affect the
higher-frequency region of the Nyquist plot. FIG. 56 shows the
effect of the membrane capacitance on the Nyquist plot. As can be
seen from FIG. 56, changes in Cmem affect how much of the
high-frequency region's semi-circle is visible. Thus, as membrane
capacitance increases in the direction of Arrow A, progressively
less of the semi-circle can be seen. Similarly, as shown in FIG.
57, as the membrane resistance increases in the direction of Arrow
A, more of the high-frequency region semi-circle becomes visible.
In addition, as Rmem increases, the overall Nyquist plot shifts
from left to right. The latter parallel-shifting phenomenon also
holds true for Rsol, as shown in FIG. 58.
[0656] The above discussion in connection with the equivalent
circuit model of FIG. 48 may be summarized as follows. First, Cdl,
a, Rp, Warburg, and .lamda. generally control the low frequency
response. More specifically, the lower-frequency Nyquist
slope/Zimag primarily depends on Cdl, a, Rp, and .lamda., and the
lower-frequency length/Zmagnitude primarily depends on Cdl, Rp, and
Warburg Admittance. Second, Rmem and Cmem control the
higher-frequency response. In particular, Rmem determines the high
frequency semi-circle diameter, and Cmem determines the turning
point frequency, having minimal overall effect on the Nyquist plot.
Lastly, changes in Rmem and Rsol cause parallel shifts in the
Nyquist plot.
[0657] FIGS. 59A-59C, 60A-60C, and 61A-61C show results of in-vitro
experiments for changes in the above-described circuit elements
during sensor start-up and calibration. FIGS. 59A, 60A, and 61A are
identical. As shown in FIG. 59A, the experiments were generally run
with two redundant working electrodes 5050, 5060, and for a period
of (between 7 and) 9 days. A baseline glucose amount of 100 mg/dL
was used, although the latter was changed between zero and 400
mg/dL at various points throughout the experiment (5070). In
addition, the effects of a (solution) temperature change between
32.degree. C. and 42.degree. C. (5080) and a 0.1 mg/dL
acetaminophen response (5085) were explored. Lastly, the
experiments included an Oxygen stress test, where the supply of
Oxygen dissolved in the solution was varied (i.e., limited) between
0.1% and 5% (5075). For purposes of these experiments, a full EIS
sweep (i.e., from 0.1 Hz-8 kHz) was run, and the output data was
recorded (and plotted) about once every 30 minutes. However,
shorter or longer intervals may also be used.
[0658] In FIG. 59C, the sum of Rsol and Rmem--which, again, may be
estimated by the magnitude of real impedance at the inflection
point of the Nyquist plot--displays a general downwards trend as a
function of time. This is due primarily to the fact that the
membrane takes time to hydrate, such that, as time passes by, it
will become less resistant to the electrical charges. A slight
correlation can also be seen between the plot for Isig (FIG. 59A)
and that for Rsol+Rmem (FIG. 59C).
[0659] FIG. 60B shows the EIS output for Cdl. Here, there is
initially a relatively rapid drop (5087), over a period of several
hours, due to the sensor activation/sensor charge-up process.
Thereafter, however, Cdl remains fairly constant, exhibiting a
strong correlation with Isig (FIG. 60A). Given the latter
correlation, Cdl data, as an EIS parameter, may be less useful in
applications where glucose independence is desired. As shown in
FIG. 60C, the trend for Rp may be generally described as a mirror
image of the plot for Cdl. As the membrane becomes more hydrated,
the influx increases, which is reflected in the plot of Warburg
admittance in FIG. 61B. As shown in FIG. 61C, .lamda. remains
generally constant throughout.
[0660] FIGS. 62-65 show the actual EIS response for various parts
of the above-described experiments. Specifically, the changes that
were made during the first 3 days--i.e., glucose changes, Oxygen
stress, and temperature changes, as shown in FIGS. 59A, 60A, and
61A--are boxed (5091) in FIG. 62, with the Vcntr response 5093
being shown in the bottom portion of this Figure and in FIG. 59B.
FIG. 63 shows that an Isig calibration via an increase in glucose
caused the slope and length of the Nyquist plot to decrease. In
FIG. 64, the Oxygen (or Vcntr) response is shown in Day 2, where
Vcntr becomes more negative as the Oxygen content is decreased.
Here, the Nyquist plot becomes shorter in length, and its slope
decreases (5094), indicating a large decrease in imaginary
impedance. The plot length depends primarily on Cdl and Rp, and is
strongly correlated to Vcntr which, in turn, responds to changes in
glucose and Oxygen. In FIG. 65, the Isig changes negligibly from
Day 2 to Day 3. Nevertheless, the Nyquist plot shifts horizontally
(from the plot at 37.degree. C.) for data taken at 32.degree. C.
(5095) and at 42.degree. C. (5097). However, there is no
significant impact on Nyquist plot length, slope, or Isig.
[0661] Putting the above-described EIS output and signature
information together, it has been discovered that, during sensor
start-up, the magnitude of Rmem+Rsol decreases over time,
corresponding to a shift from right to left in the Nyquist plot.
During this period, Cdl decreases, and Rp increases, with a
corresponding increase in Nyquist slope. Finally, Warburg
admittance also increases. As noted previously, the foregoing is
consistent with the hydration process, with EIS plots and parameter
values taking on the order of 1-2 days (e.g., 24-36 hours) to
stabilize.
[0662] Embodiments of the invention are directed to real-time
self-calibration, and more particularly, to in-vivo
self-calibration of glucose sensors based on EIS data. Any
calibration algorithm, including self-calibration algorithms, must
address sensitivity loss. As discussed previously, two types of
sensitivity loss may occur: (1) Isig dip, which is a temporary loss
of sensitivity, typically occurring during the first few days of
sensor operation; and (2) permanent sensitivity loss, occurring
generally at the end of sensor life, and sometimes correlated with
the presence of a Vcntr rail.
[0663] It has been discovered that sensitivity loss can manifest
itself as an increase in Rsol or Rmem (or both), which can be
observed in the Nyquist plot as a parallel shift to the right, or,
if Rmem changes, a more visible start to a semicircle at the higher
frequencies (resulting in an increase in high-frequency imaginary
impedance). In addition to, or instead of, Rsol and Rmem, there
could be an increase in Cmem only. This can be observed as changes
in the high-frequency semicircle. Sensitivity loss will be
accompanied by a change in Cdl (by way of a longer tail in the
lower-frequency segment of the Nyquist plot). The foregoing
signatures provide a means for determining how different changes in
EIS output can be used to compensate for changes in
sensitivity.
[0664] For a normally operating glucose sensor, there is a linear
relationship between blood glucose (BG) and the sensor's current
output (Isig). Thus,
BG=CF.times.(Isig+c)
where "CF" is the Cal Factor, and "c" is the offset. This is shown
in FIG. 66, where the calibration curve is as shown by line 6005,
and "c" is the baseline offset 6007 (in nA). However, when there is
an increase in Rmem and/or a decrease in Cmem, then c will be
affected. Thus, line 6009 depicts a situation in which Rmem
increases and Cmem decreases--which signifies changes in the
membrane properties--thereby causing the offset "c" to move to
6011, i.e., a downward shift of the calibration curve. Similarly,
when there are (non-glucose related) changes in Cdl and increases
in Rp--with a resultant increase in the length of the
(lower-frequency) Nyquist plot--then the slope will be affected,
where the slope=1/CF. Thus, in FIG. 66, line 6013 has a different
(smaller) slope that line 6005. Combined changes can also occur,
which is illustrated by line 6015, indicating sensitivity loss.
[0665] The length of the lower-frequency segment of the Nyquist
plot (L.sub.nyquist)--which, for simplicity, may be illustratively
estimated as the length between 128 Hz and 0.105 Hz (real)
impedance--is highly correlated with glucose changes. It has been
discovered, through model fitting, that the only parameter that
changes during glucose changes is the double layer capacitance Cdl,
and specifically the double layer admittance. Therefore the only
Isig-dependent--and, by extension, glucose-dependent--parameter in
the equivalent circuit model of FIG. 48 is Cdl, with all other
parameters being substantially Isig-independent.
[0666] In view of the above, in one embodiment of the invention,
changes in Rmem and Cmem may be tracked to arrive at a readjustment
of the Cal Factor (BG/Isig) and, thereby, enable real-time
self-calibration of sensors without the need for continual
finger-stick testing. This is possible, in part, because changes in
Rmem and Cmem result in a change in the offset (c), but not in the
slope, of the calibration curve. In other words, such changes in
the membrane-related parameters of the model generally indicate
that the sensor is still capable of functioning properly.
[0667] Graphically, FIG. 67A shows actual blood glucose (BG) data
6055 that is being recorded, overlaid by the Isig output 6060 from
the working electrode. Comparing the data from a first period (or
time window) comprising approximately days 1-4 (6051) with the data
from a second period comprising approximately days 6-9 (6053), FIG.
67A shows that the sensor is drifting generally downwards during
the second time period, indicating perhaps a moderate sensitivity
loss in the sensor. There is also an increase in Vcntr during the
second time period, as shown in FIG. 67B.
[0668] With reference to FIGS. 68 and 69, it can be seen that the
sensitivity loss is clearly shown by a rather significant increase
in membrane resistance 6061, as well as a corresponding drop in
Warburg Admittance 6063, during the second time period between days
6 and 9. Accordingly, FIG. 70 shows that the calibration curve 6073
for the second time period 6053 is parallel to, but shifted down
from, the calibration curve 6071 for the first time period 6051.
Also, as discussed hereinabove in connection with FIG. 57, as the
membrane resistance (Rmem) increases, overall Nyquist plot shifts
from left to right, and more of the high-frequency region
semi-circle becomes visible. For the data of FIGS. 67A-70, this
phenomenon is shown in FIG. 71, where the enlarged higher-frequency
region of the Nyquist plot shows that the data from the second time
period 6053 moves the plot from left to right as compared with the
data from the first time period 6051, and that the semi-circle
becomes more and more visible (6080) as the shift in the Nyquist
plot progresses from left to right. In addition, the enlarged
lower-frequency region of the plot shows that there is no
significant change in L.sub.nyquist.
[0669] Changes in Cdl and Rp, on the other hand, generally indicate
that the electrode(s) may already be compromised, such that
recovery may no longer be possible. Still, changes in Cdl and Rp
may also be tracked, e.g., as a diagnostic tool, to determine,
based on the direction/trend of the change in these parameters,
whether, the drift or sensitivity loss has in fact reached a point
where proper sensor operation is no longer recoverable or
achievable. In this regard, in embodiments of the invention,
respective lower and/or upper thresholds, or ranges of thresholds,
may be calculated for each of Cdl and Rp, or for the change in
slope, such that EIS output values for these parameters that fall
outside of the respective threshold (range) may trigger, e.g.,
termination and/or replacement of the sensor due to unrecoverable
sensitivity loss. In specific embodiments, sensor-design and/or
patient-specific ranges or thresholds may be calculated, wherein
the ranges/thresholds may be, e.g., relative to the change in Cdl,
Rp, and/or slope.
[0670] Graphically, FIG. 72A shows actual blood glucose (BG) data
6155 that is being recorded, overlaid by the Isig output from two
working electrodes, WE1 6160 and WE2 6162. The graphs show data
from a first time window for day 1 (6170), a second time window for
days 3-5 (6172), a third time window for day 3 (6174), and a fourth
time window for days 51/2 to 91/2 (6176). Starting on Day 3, FIG.
72B shows that Vcntr rails at 1.2 volts. However, the decrease in
sensitivity occurs from about Day 5 or so (6180). Once the Vcntr
rails, the Cdl increases significantly, with a corresponding
decrease in Rp, signifying a higher resistance to the overall
electrochemical reaction. As expected, the slope of the calibration
curve also changes (decreases), and L.sub.nyquist becomes shorter
(see FIGS. 73-75). It is noted that, in embodiments of the
invention, the occurrence of a Vcntr rail may be used to trigger
termination of a sensor as unrecoverable.
[0671] The combined effect of the increase in membrane resistance,
the decrease in Cdl, and Vcntr rail is shown in FIGS. 76A-76B and
77-80. In FIG. 76A, actual blood glucose (BG) data 6210 is overlaid
by the Isig output from two working electrodes, WE1 6203 and WE2
6205. As can be seen, WE1 generally tracks the actual BG data
6210--i.e., WE1 is functioning normally. The Isig from WE2, on the
other hand, appears to start at a lower point, and continues a
downwards trend all the way from the beginning to Day 10, thus
signifying a gradual loss of sensitivity. This is consistent with
the Cdl for WE2 (6215) being lower than that for WE1 (6213), as
shown in FIG. 77, even though the Cdl for both working electrodes
generally exhibits a downward trend.
[0672] FIG. 79 shows the combined effect on the calibration curve,
where both the offset and the slope of the linear fit for the
period of sensitivity loss (6235) change relative to the
calibration curve 6231 for the normally-functioning time windows.
In addition, the Nyquist plot of FIG. 80 shows that, in the
lower-frequency region, the length of the Nyquist plot is longer
where there is sensitivity loss (6245), as compared to where the
sensor is functioning normally (6241). Moreover, near the
inflection point, the semicircles (6255) become more and more
visible where there is loss of sensitivity. Importantly, where
there is sensitivity loss, the Nyquist plot of FIG. 80 shifts
horizontally from left to right as a function of time. In
embodiments of the invention, the latter shift may be used as a
measure for compensation or self-correction in the sensor.
[0673] Thus, it has been discovered that, as an EIS signature, a
temporary dip may be caused by increased membrane resistance (Rmem)
and/or local Rsol increase. An increase in Rmem, in turn, is
reflected by increased higher-frequency imaginary impedance. This
increase may be characterized by the slope at high frequencies,
(S.sub.nyquist)--which, for simplicity, may be illustratively
estimated as the slope between 8 kHz and 128 Hz. In addition, Vcntr
railing increases Cdl and decrease Rp, such that the length and
slope decrease; this may be followed by gradual Cdl decrease and Rp
increase associated with sensitivity loss. In general, a decrease
in Cdl, combined with an increase in Rp (length increase) and in
Rmem may be sufficient to cause sensitivity loss.
[0674] In accordance with embodiments of the invention, an
algorithm for sensor self-calibration based on the detection of
sensitivity change and/or loss is shown in FIG. 81. At blocks 6305
and 6315, a baseline Nyquist plot length (L.sub.nyquist) and a
baseline higher frequency slope, respectively, are set, so as to be
reflective of the EIS state at the beginning of sensor life. As
noted, the Nyquist plot length is correlated to the Cdl, and the
higher frequency Nyquist slope is correlated to the membrane
resistance. The process then continues by monitoring the Nyquist
plot length (6335) and the higher frequency slope (6345), as well
as the Vcntr value (6325). When the Vcntr rails, the baseline
L.sub.nyquist is adjusted, or reset 6355, as the railing of the
Vcntr changes the Cdl significantly. There is therefore a feedback
loop 6358 to accommodate real-time changes in the monitored EIS
parameters.
[0675] As shown in block 6375, as the length of the Nyquist plot is
monitored, a significant increase in that length would indicate
reduced sensitivity. In specific embodiments, sensor-design and/or
patient-specific ranges or thresholds may be calculated, wherein
the ranges/thresholds may be, e.g., relative to the change in the
length of the Nyquist plot. Similarly, a more negative
higher-frequency slope S.sub.nyquist corresponds to an increased
appearance of the high-frequency semicircle and would be indicative
of a possible dip 6365. Any such changes in L.sub.nyquist and
S.sub.nyquist are monitored, e.g., either continuously or
periodically and, based on the duration and trend of the reduction
in sensitivity, a determination is made as to whether total (i.e.,
severe) sensitivity loss has occurred, such that specific sensor
glucose (SG) value(s) should be discarded (6385). In block 6395,
the Cal Factor may be adjusted based on the monitored parameters,
so as to provide a "calibration-free" CGM sensor. It is noted that,
within the context of the invention, the term "calibration-free"
does not mean that a particular sensor needs no calibration at all.
Rather, it means that the sensor can self-calibrate based on the
EIS output data, in real time, and without the need for additional
finger-stick or meter data. In this sense, the self-calibration may
also be referred to as "intelligent" calibration, as the
calibration is not performed based on a predetermined temporal
schedule, but on an as-needed basis, in real-time.
[0676] In embodiments of the invention, algorithms for adjustment
of the Cal Factor (CF) and/or offset may be based on the membrane
resistance which, in turn, may be estimated by the sum of Rmem and
Rsol. As membrane resistance is representative of a physical
property of the sensor, it generally cannot be estimated from EIS
data run for a single frequency. Put another way, it has been
observed that no single frequency will consistently represent
membrane resistance, since frequencies shift depending on sensor
state. Thus, FIG. 82, e.g., shows that, when there is some
sensitivity loss, there is a horizontal shift in the Nyquist plot,
and therefore, a shift in the inflection point that estimates the
value of Rmem+Rsol. In this case, the shift in the real component
of impedance is actually quite large. However, if only the
high-frequency (e.g., at 8 kHz) real impedance is monitored, there
is little to no shift at all, as indicated by the encircled region
in FIG. 82.
[0677] There is therefore a need to track membrane resistance in a
physically meaningful way. Ideally, this may be done through model
fitting, where Rmem and Rsol are derived from model fitting, and Rm
is calculated as Rm=Rmem+Rsol. However, in practice, this approach
is not only computationally expensive, as it may take an
unpredictably long amount of time, but also susceptible to not
converging at all in some situations. Heuristic metrics may
therefore be developed to approximate, or estimate, the value of
Rm=Rmem+Rsol. In one such metric, Rmem+Rsol is approximated by the
value of the real-impedance intercept at a fairly stable imaginary
impedance value. Thus, as shown in FIG. 83, for example, a region
of general stability for the imaginary impedance (on the Y axis)
may be identified at about 2000.OMEGA.. Taking this as a reference
value and traveling across, parallel to the X axis, a value
proportional to Rm may then be approximated as the real-impedance
value of where the reference line crosses the Nyquist plot. An
interpolation between frequencies may be performed to estimate
.DELTA.Rm.varies..DELTA. (Rmem+Rsol).
[0678] Having estimated the value of Rm as discussed above, the
relationship between Rm and the Cal Factor (CF) and/or Isig may
then be explored. Specifically, FIG. 84 shows the relationship
between the estimated Rm and CF, wherein the former is directly
proportional to the latter. The data points for purposes of FIG. 84
were derived for steady state sensor operation. FIG. 85 shows a
plot of normalized Isig vs. 1/Rm, where Isig has been normalized by
the BG range (of the Isig). As can be seen from the figure, Isig
can be adjusted based on changes in Rm. Specifically, an increase
in 1/Rm (i.e., reduced membrane resistance) will lead to a
proportional increase in Isig, as there is a linear relationship
between Isig and 1/Rm.
[0679] Thus, in one embodiment, an algorithm for adjustment of the
Cal Factor would entail monitoring the change in membrane
resistance based on a reference Cal Factor, and then modifying the
Cal Factor proportionally based on the correlation between Rm and
CF. In other words:
d ( CF ) d t .varies. d ( Rm ) d t Adjusted CF .varies. ( d ( Rm )
d t ) .times. CF ##EQU00024##
[0680] In another embodiment, a Cal Factor adjustment algorithm may
entail modification of Isig based on proportional changes in 1/Rm,
and independently of CF calculations. Thus, for purposes of such an
algorithm, the adjusted Isig is derived as
Adjusted Isig .varies. ( d ( 1 R m ) d t ) .times. Isig
##EQU00025##
[0681] Experiments have shown that the most dramatic CF changes
occur in first 8 hours of sensor life. Specifically, in one set of
in-vitro experiments, Isig was plotted as a function of time, while
keeping various glucose levels constant over the life of the
sensor. EIS was run every 3 minutes for the first 2 hours, while
all model parameters were estimated and tracked over time. As noted
previously, given a limited spectrum EIS, Rmem and Rsol cannot be
(independently) estimated robustly. However, Rm=Rmem+Rsol can be
estimated.
[0682] FIG. 86 shows the plots for Isig over time for various
glucose levels, including 400 mg/dL (6410), 200 mg/dL (6420), 100
mg/dL (6430), 60 mg/dL (6440), and 0 mg/dL (6450). At startup,
generally dramatic changes appear in all parameters. One example is
shown in FIG. 87, where Cdl is plotted as a function of time, with
plot 6415 corresponding to 400 mg/dL glucose, plot 6425
corresponding to 200 mg/dL glucose, plot 6435 corresponding to 100
mg/dL glucose, plot 6445 corresponding to 60 mg/dL glucose, and
plot 6455 corresponding to 0 mg/dL glucose. As is the case in the
illustrative example of FIG. 87, most parameters correlate well
with changes in the first 0.5 hour, but generally may not account
for changes in timeframes >0.5 hour.
[0683] It has been discovered, however, that Rm=Rmem+Rsol is the
only parameter that can account for changes in Isig over a similar
startup time frame. Specifically, FIG. 88 shows the same graph as
in FIG. 86, except for an indication that there is a peak, or
second inflection point, that occurs at about T=1 hour, especially
at low glucose levels, e.g., 100 mg/dL and lower. However, of all
the EIS parameters that were studied, membrane resistance was the
only one that exhibited a relationship to this change in Isig; the
other parameters generally tend to proceed fairly smoothly to
steady state. Thus, as shown in FIG. 89, Rm also exhibits a second
inflection point at about T=1 hour that corresponds to the peak in
Isig at the same time.
[0684] FIG. 90 shows the relationship between Cal Factor and Rm for
in-vivo data during the first 8 hours of sensor operation. Here,
EIS was run about once every 30 minutes at startup, and
interpolated for periods in between. As can be seen, Rm=Rmem+Rsol
correlates with Cal Factor (CF) during the first 8 hours of sensor
operation. For purposes of the diagram in FIG. 90, the baseline
offset was assumed to be 3 nA.
[0685] As noted above in connection with FIGS. 83-85, in one
embodiment of the invention, an algorithm for adjustment of the Cal
Factor at start up may include selecting a reference value for the
calibration factor (CF.sub.reference), estimating the value of
membrane resistance (R.sub.reference) for CF=CF.sub.reference,
monitoring the change in membrane resistance (Rm=Rmem+Rsol), and
based on the magnitude of that change, adjusting the calibration
factor in accordance with the relationship shown in FIG. 90.
Thus
CF(t)=CF.sub.reference-m(R.sub.reference-R.sub.m(t))
where m is the gradient of the correlation in FIG. 90. It is noted
that, for purposes of the above algorithm, the value of
CF.sub.reference is sensor-specific, to account for the differences
between sensors.
[0686] In another embodiment, the Cal Factor adjustment algorithm
may be modified by using a limited range of R.sub.m over which
adjustment occurs. This can help with small differences once
R.sub.m is smaller than .about.7000.OMEGA., as may happen due to
noise. The limited R.sub.m range can also help when R.sub.m is very
large, as may happen due to very slow sensor
hydration/stabilization. In yet another embodiment, the range of
allowable CF may be limited, such as, e.g., by setting a lower
limit of 4.5 for CF.
[0687] FIG. 91A is a chart showing in-vivo results for MARD over
all valid BGs in approximately the first 8 hours of sensor life. A
single (first) calibration is performed with the first BG at either
1 hour, 1.5 hours, or 2 hours after startup. As can be seen,
without any Cal Factor adjustment, the MARD for calibration at 1
hour is much higher than that for calibration performed at 2 hours
(22.23 vs. 19.34). However, with adjustment, or modified
adjustment, as described above, the difference between the
respective MARD numbers becomes smaller. Thus, for example, with
adjustment, the MARD for calibration at 1 hour is 16.98, as
compared to 15.42 for calibration performed at 2 hours. In
addition, the MARD with adjustment for calibration at 1 hour is
much less than the MARD without adjustment for calibration
performed at 2 hours (16.98 vs. 19.34). As such, in accordance with
embodiments of the invention, Cal Factor adjustments (and modified
adjustments) may be used to elongate the useable life of a
sensor--e.g., by starting the sensor one hour earlier, in this
example--while maintaining, or improving, the MARD. The chart in
FIG. 91B provides median ARD numbers over all valid BGs in
approximately the first 8 hours.
[0688] FIGS. 92A-92C, 93A-93C, and 94A-94C show examples of when
the above-described Cal Factor adjustment algorithms work better
than some current, non-EIS based, methods. In one such method,
generally referred to as "First Day Compensation" (or FDC), a first
Cal Factor is measured. If the measured Cal Factor falls outside of
a predetermined range, a constant linear decay function is applied
to bring the Cal Factor back to within normal range at a projected
time determined by the rate of the decay. As can be seen from FIGS.
92A-94C, the Cal Factor adjustment algorithms of the invention
(referred to in the diagrams as "Compensation") 6701, 6711, 6721
produce results that are closer to the actual blood glucose (BG)
measurements 6707, 6717, 6727 than results obtained by the FDC
method 6703, 6713, 6723.
[0689] Given the complexities of estimating the value of
EIS-related parameters, some of the current methods, including FDC,
may be computationally less complex than the EIS Cal Factor
adjustment algorithms described herein. However, the two approaches
may also be implemented in a complementary fashion. Specifically,
there may be situations in which FDC may be augmented by the
instant Cal Factor adjustment algorithms. For example, the latter
may be used to define the rate of change of the FDC, or to identify
the range for which FDC should be applied (i.e., other than using
CF alone), or to reverse the direction of FDC in special cases.
[0690] In yet other embodiments, the offset, rather than the Cal
Factor, may be adjusted. In addition, or instead, limits may be
imposed on applicable ranges of R.sub.m and CF. In a specific
embodiment, absolute, rather than relative, values may be used.
Moreover, the relationship between Cal Factor and membrane may be
expressed as multiplicative, rather than additive. Thus,
CF ( t ) CF reference = - m ( R ( t ) R reference )
##EQU00026##
[0691] In an embodiment using EIS-based dynamic offset, the total
current that is measured may be defined as the sum of the Faradaic
current and the non-Faradaic current, wherein the former is
glucose-dependent, while the latter is glucose-independent. Thus,
mathematically,
i.sub.total=i.sub.Faradaic+i.sub.non-Faradaic
[0692] Ideally, the non-Faradaic current should be zero, with a
fixed working potential, such that
i total = i Faradaic = A .times. Diffusivity .times. .differential.
C peroxide .differential. n ##EQU00027##
where A is the surface area, and
.differential. C peroxide .differential. n ##EQU00028##
is the gradient of Peroxide.
[0693] However, when the double layer capacitance in changing, the
non-Faradaic current cannot be ignored. Specifically, the
non-Faradaic current may be calculated as
q non - Faradaic = V .times. C = .intg. t 0 t 0 + .DELTA. t i non -
Faradaic d t ##EQU00029## d d t q non - Faradaic = i non - Faradaic
= d ( V .times. C ) d t = C d V d t + V d C d t ##EQU00029.2##
where q is the charge, V is the voltage, C is (double layer)
capacitance. As can be seen from the above, when both voltage (V)
and capacitance (C) are constant, both time-derivative values on
the right-hand side of the equation are equal to zero, such that
i.sub.non-Faradaic=0. In such an ideal situation, the focus can
then turn to diffusion and reaction.
[0694] When V and C are both functions of time (e.g., at sensor
initialization),
i non - Faradaic = d ( V .times. C ) d t = C d V d t + V d C d t
##EQU00030##
[0695] On the other hand, when V is constant, and C is a function
of time,
i non - Faradaic = V d C d t ##EQU00031##
Such conditions are present, for example, on day 1 of sensor
operation. FIG. 95 shows an example of a typical (initial) decay in
double layer capacitance during day 1, in this case, the first 6
hours after sensor insertion. As indicated on the graph, plot 6805
shows raw Cdl data based on EIS data obtained at half-hour
intervals, plot 6810 shows a spline fit on the raw Cdl data for
5-minute time intervals, plot 6815 shows the smoothed curve for
5-minute time intervals, and plot 6820 shows a polynomial fit on
the smoothed Cdl data for 5-minute time intervals.
[0696] It is noted that the Cdl decay is not exponential. As such,
the decay cannot be simulated with an exponential function. Rather,
it has been found that a 6.sup.th-order polynomial fit (6820)
provides a reasonable simulation. Thus, for the purposes of the
above-mentioned scenario, where V is constant and C is a function
of time, i.sub.non-Faradaic may be calculated if the polynomial
coefficients are known. Specifically,
C=P(1)t.sup.6+P(2)t.sup.5+P(3)t.sup.4+P(4)t.sup.3+P(5)t.sup.2+P(6)t.sup.-
1+P(7)
where P is the polynomial coefficient array, and t is time. The
non-Faradaic current can then be calculated as:
i non - Faradaic = V d C d t = V ( 6 P ( 1 ) t 5 + 5 P ( 2 ) t 4 +
4 P ( 3 ) t 3 + 3 P ( 4 ) t 2 + 2 P ( 5 ) t 1 + P ( 6 ) )
##EQU00032##
[0697] Finally, since i.sub.total=i.sub.Faradaic+.sub.non-Faradaic,
the non-Faradaic component of the current can be removed by
rearranging, such that
i.sub.Faradaic=i.sub.total-i.sub.non-Faradaic
[0698] FIG. 96 shows Isig based on the total current (6840), as a
function of time, as well as Isig after removal of the non-Faradaic
current based on the capacitance decay (6850). The non-Faradaic
component of the current may be as high as 10-15 nA. As can be seen
from the figure, removal of the non-Faradaic current helps remove a
large majority of the low start-up Isig data at the beginning of
sensor life.
[0699] It has been found that the above approach can be used to
reduce the MARD, as well as adjust the Cal Factor right at the
beginning of sensor life. With regard to the latter, FIG. 97A shows
the Cal Factor before removal of the non-Faradaic current for a
first working electrode (WE1) 6860, and a second working electrode
(WE2) 6870. FIG. 97B, on the other hand, shows the Cal Factor for
WE1 (6862) and WE2 (6872) after removal of the non-Faradaic
current. Comparing the Cal Factor for WE1 in FIG. 97A (6860) to
that for WE1 in FIG. 97B (6862), it can be seen that, with removal
of the non-Faradaic component, the Cal Factor (6862) is much closer
to the expected range.
[0700] In addition, the reduction in MARD can be seen in the
example shown in FIGS. 98A and 98B, where sensor glucose values are
plotted over time. As shown in FIG. 98A, before removal of the
non-Faradaic current, calibration at low startup causes significant
sensor over-reading at WE1 (6880), with a MARD of 11.23%. After
removal of the non-Faradaic current, a MARD of 10.53% is achieved
for WE1. It is noted that, for the illustrative purposes of FIGS.
97A-98B, the non-Faradaic current was calculated and removed in
pre-processing using the relation
i non - Faradaic = V d C d t = V ( 6 P ( 1 ) t 5 + 5 P ( 2 ) t 4 +
4 P ( 3 ) t 3 + 3 P ( 4 ) t 2 + 2 P ( 5 ) t 1 + P ( 6 ) ) ,
##EQU00033##
where P is the polynomial coefficient (array) used to fit the
double layer capacitance curve.
[0701] In real-time, separation of the Faradaic and non-Faradaic
currents may be used to automatically determine the time to conduct
the first calibration. FIG. 99 shows the double layer capacitance
decay over time. Specifically, over the constant time interval
.DELTA.T, the double layer capacitance undergoes a change from a
first value C.sub.T.sub.0.sub..DELTA.T (7005) to a second value
C.sub.T (7010). A first-order time difference method, e.g., can
then be used to calculate the non-Faradaic current as
i non - Faradaic = V d C d t .apprxeq. V C T 0 + .DELTA. T - C T
.DELTA. T ##EQU00034##
Other methods may also be used to calculate the derivative
d C d t , ##EQU00035##
such as, e.g., second-order accurate finite value method (FVM),
Savitzky-Golay, etc.
[0702] Next, the percentage of the total current, i.e., Isig, that
is comprised of the non-Faradaic current may be calculated simply
as the ratio i.sub.non-Faradaic/Isig. Once this ratio reaches a
lower threshold, a determination can then be made, in real-time, as
to whether the sensor is ready for calibration. Thus, in an
embodiment of the invention, the threshold may be between 5% and
10%.
[0703] In another embodiment, the above-described algorithm may be
used to calculate an offset value in real-time, i.e., an EIS-based
dynamic offset algorithm. Recalling that
i non - Faradaic = V d C d t = V ( 6 P ( 1 ) t 5 + 5 P ( 2 ) t 4 +
4 P ( 3 ) t 3 + 3 P ( 4 ) t 2 + 2 P ( 5 ) t 1 + P ( 6 ) )
##EQU00036##
and that sensor current Isig is the total current, including the
Faradaic and non-Faradaic components
i.sub.total=i.sub.Faradaic+i.sub.non-Faradaic
the Faradaic component is calculated as
i.sub.Faradaic=i.sub.total-i.sub.non-Faradaic
[0704] Thus, in an embodiment of the invention, the non-Faradaic
current, i.sub.non-Faradaic, can be treated as an additional offset
to Isig. In practice, when double layer capacitance decreases,
e.g., during the first day of sensor life, i.sub.non-Faradaic is
negative, and decreases as a function of time. Therefore, in
accordance with this embodiment of the invention, a larger
offset--i.e., the usual offset as calculated with current methods,
plus i.sub.non-Faradaic--would be added to the Isig at the very
beginning of sensor life, and allowed to decay following the
5.sup.th-order polynomial curve. That is, the additional offset
i.sub.non-Faradaic follows a 5.sup.th-order polynomial, the
coefficient for which must be determined. Depending on how dramatic
the change in double layer capacitance is, the algorithm in
accordance with this embodiment of the invention may apply to the
first few hours, e.g., the first 6-12 hours, of sensor life.
[0705] The polynomial fit may be calculated in various ways. For
example, in an embodiment of the invention, coefficient P may be
pre-determined based upon existing data. Then, the dynamic offset
discussed above is applied, but only when the first Cal Factor is
above normal range, e.g., .about.7. Experiments have shown that,
generally, this method works best when the real-time double layer
capacitance measurement is less reliable than desired.
[0706] In an alternative embodiment, an in-line fitting algorithm
is used. Specifically, an in-line double layer capacitance buffer
is created at time T. P is then calculated based on the buffer,
using a polynomial fit at time T. Lastly, the non-Faradaic current
(dynamic offset) at time T+.DELTA.T is calculated using P at time
T. It is noted that this algorithm requires double layer
capacitance measurements to be more frequent than their current
level (every 30 mins), and that the measurements be reliable (i.e.,
no artifacts). For example, EIS measurements could be taken once
every 5 minutes, or once every 10 minutes, for the first 2-3 hours
of sensor life.
[0707] In developing a real-time, self-calibrating sensor, the
ultimate goal is to minimize, or eliminate altogether, the reliance
on a BG meter. This, however, requires understanding of the
relationships between EIS-related parameters and Isig, Cal Factor
(CF), and offset, among others. For example, in-vivo experiments
have shown that there is a correlation between Isig and each of Cdl
and Warburg Admittance, such that each of the latter may be
Isig-dependent (at least to some degree). In addition, it has been
found that, in terms of factory calibration of sensors, Isig and Rm
(=Rmem+Rsol) are the most important parameters (i.e., contributing
factors) for the Cal Factor, while Warburg Admittance, Cdl, and
Vcntr are the most important parameters for the offset.
[0708] In in-vitro studies, metrics extracted from EIS (e.g., Rmem)
tend to exhibit a strong correlation with Cal Factor. However,
in-vivo, the same correlation can be weak. This is due, in part, to
the fact that patient-specific, or (sensor)
insertion-site-specific, properties mask the aspects of the sensor
that would allow use of EIS for self-calibration or factory
calibration. In this regard, in an embodiment of the invention,
redundant sensors may be used to provide a reference point that can
be utilized to estimate the patient-specific response. This, in
turn, would allow a more robust factory calibration, as well as
help identify the source of sensor failure mode(s) as either
internal, or external, to the sensor.
[0709] In general, EIS is a function of electric fields that form
between the sensor electrodes. The electric field can extend beyond
the sensor membrane, and can probe into the properties of the
(patient's) body at the sensor insertion site. Therefore, if the
environment in which the sensor is inserted/disposed is uniform
across all tests, i.e., if the tissue composition is always the
same in-vivo (or if the buffer is always the same in-vitro), then
EIS can be correlated to sensor-only properties. In other words, it
may be assumed that changes in the sensor lead directly to changes
in the EIS, which can be correlated with, e.g., the Cal Factor.
[0710] However, it is well known that the in-vivo environment is
highly variable, as patient-specific tissue properties depend on
the composition of the insertion site. For example, the
conductivity of the tissue around the sensor depends on the amount
of fat around it. It is known that the conductivity of fat is much
lower than that of pure interstitial fluid (ISF), and the ratio of
local fat to ISF can vary significantly. The composition of the
insertion site depends on the site of insertion, depth of
insertion, patient-specific body composition, etc. Thus, even
though the sensor is the same, the Rmem that is observed from EIS
studies varies much more significantly because the reference
environment is rarely, if ever, the same. That is, the conductivity
of the insertion site affects the Rmem of the sensor/system. As
such, it may not be possible to use the Rmem uniformly and
consistently as a reliable calibration tool.
[0711] As described previously, EIS can also be used as a
diagnostic tool. Thus, in embodiments of the invention, EIS may be
used for gross failure analysis. For example, EIS can be used to
detect severe sensitivity loss which, in turn, is useful for
determining whether, and when, to block sensor data, deciding on
optimal calibration times, and determining whether, and when, to
terminate a sensor. In this regard, it bears repeating that, in
continuous glucose monitoring and analysis, two major types of
severe sensitivity loss are typically considered: (1) Temporary
sensitivity loss (i.e., an Isig dip), which typically occurs early
in sensor life, and is generally believed to be a consequence of
external sensor blockage; and (2) Permanent sensitivity loss, which
typically occurs at the end of sensor life, and never recovers,
thus necessitating sensor termination.
[0712] Both in-vivo and in-vitro data show that, during sensitivity
loss and Isig dips, the EIS parameters that change may be any one
or more of Rmem, Rsol, and Cmem. The latter changes, in turn,
manifest themselves as a parallel shift in the higher-frequency
region of the Nyquist plot, and/or an increased appearance of the
high-frequency semicircle. In general, the more severe the
sensitivity loss, the more pronounced these symptoms are. FIG. 100
shows the higher-frequency region of the Nyquist plot for data at
2.6 days (7050), 3.5 days (7055), 6 days (7060), and 6.5 days
(7065). As can be seen, there may be a horizontal shift, i.e.,
Rmem+Rsol shifts, from left to right, during sensitivity loss
(7070), indicating an increase in membrane resistance. In addition,
the plot for 6 days, and especially that for 6.5 days (7065),
clearly show the appearance of the higher frequency semicircle
during sensitivity loss (7075), which is indicative of a change in
membrane capacitance. Depending on the circumstances and the
severity of the sensitivity loss, either or both of the
above-mentioned manifestations may appear on the Nyquist plot.
[0713] With specific regard to the detection of Isig dips, as
opposed to permanent sensitivity loss, some current methodologies
use the Isig only to detect Isig dips by, e.g., monitoring the rate
at which Isig may be dropping, or the degree/lack of incremental
change in Isig over time, thereby indicating that perhaps the
sensor is not responsive to glucose. This, however, may not be very
reliable, as there are instances when Isig remains in the normal BG
range, even when there is an actual dip. In such a situation,
sensitivity loss (i.e., the Isig dip) is not distinguishable from
hypoglycemia. Thus, in embodiments of the invention, EIS may be
used to complement the information that is derived from the Isig,
thereby increasing the specificity and sensitivity of the detection
method.
[0714] Permanent sensitivity loss may generally be associated with
Vcntr rails. Here, some current sensor-termination methodologies
rely solely on the Vcntr rail data, such that, e.g., when Vcntr
rails for one day, the sensor may be terminated. However, in
accordance with embodiments of the invention, one method of
determining when to terminate a sensor due to sensitivity loss
entails using EIS data to confirm whether, and when, sensitivity
loss happens after Vcntr rails. Specifically, the parallel shift in
the higher-frequency region of the Nyquist plot may be used to
determine whether permanent sensitivity loss has actually occurred
once a Vcntr rail is observed. In this regard, there are situations
in which Vcntr may rail at, e.g., 5 days into sensor life, but the
EIS data shows little to shift at all in the Nyquist plot. In this
case, normally, the sensor would have been terminated at 5-6 days.
However, with EIS data indicating that there was, in fact, no
permanent sensitivity loss, the sensor would not be terminated,
thereby saving (i.e., using) the remainder of the sensor's useful
life.
[0715] As mentioned previously, detection of sensitivity loss may
be based on change(s) in one or more EIS parameters. Thus, changes
in membrane resistance (Rm=Rmem+Rsol), for example, may manifest
themselves in the mid-frequency (.about.1 kHz) real impedance
region. For membrane capacitance (Cmem), changes may be manifested
in the higher-frequency (.about.8 kHz) imaginary impedance because
of increased semicircle. The double layer capacitance (Cdl) is
proportional to average Isig. As such, it may be approximated as
the length of lower-frequency Nyquist slope L.sub.nyquist. Because
Vcntr is correlated to oxygen levels, normal sensor behavior
typically entails a decrease in Vcntr with decreasing Isig.
Therefore, an increase in Vcntr (i.e., more negative), in
combination with a decrease in Isig may also be indicative of
sensitivity loss. In addition, average Isig levels, rates of
change, or variability of signal that are low or physiologically
unlikely may be monitored.
[0716] The EIS parameters must, nevertheless, be first determined.
As described previously in connection with Cal Factor adjustments
and related disclosure, the most robust way of estimating the EIS
parameters is to perform model fitting, where the parameters in
model equations are varied until the error between the measured EIS
and the model output are minimized. Many methods of performing this
estimate exist. However, for a real time application, model fitting
may not be optimal because of computational load, variability in
estimation time, and situations where convergence is poor. Usually,
the feasibility will depend on the hardware.
[0717] When the complete model fitting noted above is not possible,
in one embodiment of the invention, one method for real-time
application is through use of heuristic methodologies. The aim is
to approximate the true parameter values (or a corresponding metric
that is proportional to trends shown by each parameter) with simple
heuristic methods applied to the measured EIS. In this regard, the
following are implementations for estimating changes in each
parameter.
[0718] Double Layer Capacitance (Cdl)
[0719] Generally speaking, a rough estimate of Cdl can be obtained
from any statistic that measures the length of the lower-frequency
Nyquist slope (e.g., frequencies lower than .about.128 Hz). This
can be done, for example, by measuring L.sub.nyquist (the Cartesian
distance between EIS at 128 Hz and 0.1 Hz in the Nyquist plot).
Other frequency ranges may also be used. In another embodiment, Cdl
may be estimated by using the amplitude of the lower-frequency
impedance (e.g., at 0.1 Hz).
[0720] Membrane Resistance (Rmem) and Solution Resistance
(Rsol)
[0721] As has been discussed hereinabove, on the Nyquist plot,
Rmem+Rsol corresponds to the inflection point between the
lower-frequency and the higher-frequency semicircles. Thus, in one
embodiment, Rmem+Rsol may be estimated by localizing the inflection
point by detecting changes in directionality of the Nyquist slope
(e.g., by using derivatives and/or differences). Alternatively, a
relative change in Rmem+Rsol can be estimated by measuring the
shift in the Nyquist slope. To do this, a reference point in the
imaginary axis can be chosen (see FIG. 83) and interpolation can be
used to determine the corresponding point on the real axis. This
interpolated value can be used to track changes in Rmem+Rsol over
time. The chosen reference should lie within a range of values
that, for a given sensor configuration, are not overly affected by
large changes in the lower-frequency part of the Nyquist slope (for
example, because of Vcntr Rail). Typical values may be between 1
k.OMEGA. and 3 k.OMEGA.. In another embodiment, it may be possible
to use the real component of a single high frequency EIS (e.g., 1
kHz, 8 kHz). In certain sensor configurations, this may simulate
Rmem the majority of the time, though it is noted that a single
frequency may not be able to represent Rmem exactly in all
situations.
[0722] Membrane Capacitance (Cmem)
[0723] Increases in Cmem manifest as a more pronounced (or the more
obvious appearance of) a higher-frequency semicircle. Changes in
Cmem can therefore be detected by estimating the presence of this
semicircle. Thus, in one embodiment, Cmem may be estimated by
tracking the higher-frequency imaginary component of impedance. In
this regard, a more negative value corresponds to the increased
presence of a semicircle.
[0724] Alternatively, Cmem may be estimated by tracking the highest
point in the semicircle within a frequency range (e.g., 1 kHz-8
kHz). This frequency range can also be determined by identifying
the frequency at which the inflection point occurs, and obtaining
the largest imaginary impedance for all frequencies higher than the
identified frequency. In this regard, a more negative value
corresponds to an increased presence of the semicircle.
[0725] In a third embodiment, Cmem may be estimated by measuring
the Cartesian distance between two higher-frequency points in the
Nyquist plot, such as, e.g., 8 kHz and 1 kHz. This is the high
frequency slope (S.sub.nyquist) defined previously in the instant
application. Here, a larger absolute value corresponds to an
increased semicircle, and a negative slope (with negative imaginary
impedance on the y axis, and positive real impedance on the x)
corresponds to the absence of a semicircle. It is noted that, in
the above-described methodologies, there may be instances in which
some of the detected changes in the semicircle may also be
attributed to changes in Rmem. However, because changes in either
are indicative of sensitivity loss, the overlap is considered to be
acceptable.
[0726] Non-EIS Related Metrics
[0727] For context, it is noted that, prior to the availability of
EIS metrics, sensitivity loss was by and large detected according
to several non-EIS criteria. By themselves, these metrics are not
typically reliable enough to achieve perfect sensitivity and
specificity in the detection. They can, however, be combined with
EIS-related metrics to provide supporting evidence for the
existence of sensitivity loss. Some of these metrics include: (1)
the amount of time that Isig is below a certain threshold (in nA),
i.e., periods of "low Isig"; (2) the first order or second order
derivatives of Isig leading to a state of "low Isig", used as an
indication of whether the changes in Isig are physiologically
possible or induced by sensitivity loss; and (3) the
variability/variance of Isig over a "low Isig" period, which can be
indicative of whether the sensor is responsive to glucose or is
flat lining.
[0728] Sensitivity-Loss Detection Algorithms
[0729] Embodiments of the invention are directed to algorithms for
detection of sensitivity loss. The algorithms generally have access
to a vector of parameters estimated from EIS measurements (e.g., as
described hereinabove) and from non-EIS related metrics. Thus,
e.g., the vector may contain Rmem and or shift in horizontal axis
(of the Nyquist plot), changes in Cmem, and changes in Cdl.
Similarly, the vector may contain data on the period of time Isig
is in a "low" state, variability in Isig, rates of change in Isig.
This vector of parameters can be tracked over time, wherein the aim
of the algorithm is to gather robust evidence of sensitivity loss.
In this context, "robust evidence" can be defined by, e.g., a
voting system, a combined weighted metric, clustering, and/or
machine learning.
[0730] Specifically, a voting system may entail monitoring of one
or more of the EIS parameters. For example, in one embodiment, this
involves determining when more than a predetermined, or calculated,
number of the elements in the parameter vector cross an absolute
threshold. In alternative embodiments, the threshold may be a
relative (%) threshold. Similarly, the vector elements may be
monitored to determine when a particular combination of parameters
in the vector crosses an absolute or a relative threshold. In
another embodiment, when any of a subset of elements in the vector
crosses an absolute or a relative threshold, a check on the
remainder of the parameters may be triggered to determine if enough
evidence of sensitivity loss can be obtained. This is useful when
at least one of a subset of parameters is a necessary (but perhaps
insufficient) condition for sensitivity loss to be reliably
detected.
[0731] A combined weighted metric entails weighing the elements in
the vector according to, for example, how much they cross a
predetermined threshold by. Sensitivity loss can then be detected
(i.e., determined as occurring) when the aggregate weighted metric
crosses an absolute or a relative threshold.
[0732] Machine learning can be used as more sophisticated "black
box" classifiers. For example, the parameter vector extracted from
realistic in-vivo experimentation can be used to train artificial
neural networks (ANN), support vector machines (SVM), or genetic
algorithms to detect sensitivity loss. A trained network can then
be applied in real time in a very time-efficient manner.
[0733] FIGS. 101A and 101B show two illustrative examples of flow
diagrams for sensitivity-loss detection using combinatory logic. As
shown, in both methodologies, one or more metrics 1-N may be
monitored. In the methodology of FIG. 101A, each of the metrics is
tracked to determine if and when it crosses a threshold, and
described hereinabove. The output of the threshold-determination
step is then aggregated via a combinatory logic, and a decision
regarding sensitivity loss is made based on the output of the
combinatory logic. In FIG. 101B, values of the monitored metrics
1-N are first processed through a combinatory logic, and the
aggregate output of the latter is then compared to a threshold
value(s) to determine whether sensitivity loss has occurred.
[0734] Additional embodiments of the invention are also directed to
using EIS in intelligent diagnostic algorithms. Thus, in one
embodiment, EIS data may be used to determine whether the sensor is
new, or whether it is being re-used (in addition to methodologies
presented previously in connection with re-use of sensors by
patients). With regard to the latter, it is important to know
whether a sensor is new or is being re-used, as this information
helps in the determination of what type of initialization sequence,
if any, should be used. In addition, the information allows
prevention of off-label use of a sensor, as well as prevention of
sensor damage due to multiple reinitializations (i.e., each time a
sensor is disconnected and then re-connected, it "thinks" that it
is a new sensor, and therefore tries to reinitialize upon
re-connection). The information also helps in post-processing of
collected sensor data.
[0735] In connection with sensor re-use and/or re-connection, it
has been discovered that the lower-frequency Nyquist slope for a
new sensor before initialization is different from (i.e., lower
than) the lower-frequency Nyquist slope for a sensor that has been
disconnected, and then reconnected again. Specifically, in-vitro
experiments have shown that the Nyquist slope is higher for a
re-used sensor as opposed to a newly-inserted one. The Nyquist
slope, therefore, can be used as a marker to differentiate between
new and used (or re-used) sensors. In one embodiment, a threshold
may be used to determine, based on the Nyquist slope, whether a
specific sensor is being re-used. In embodiments of the invention,
the threshold may be a Nyquist slope=3. FIG. 102 shows the
low-frequency Nyquist plot with a reference slope=3 (8030), as well
as the plots for a new sensor (pre-initialization) 8010, a new
sensor (post-initialization) 8015, a reconnected sensor
(pre-initialization) 8020, and a reconnected sensor
(post-initialization) 8020. As noted, the slope for a new sensor
(pre-initialization) 8010 is lower than the reference, or threshold
(8030), while that for a reconnected sensor (pre-initialization)
8020 is higher than the threshold (8030).
[0736] Equivalently, lower-frequency phase measurements may be used
to detect sensors that have been previously initialized. Here, the
pre-initialization phase angle at 0.105 Hz, e.g., may be used to
differentiate between new and used (or re-used) sensors.
Specifically, a threshold may be set at a phase angle of about
-70.degree.. Thus, if the pre-initialization phase angle at 0.105
Hz is less than the threshold, then the sensor is considered to be
an old (i.e., previously-initialized) sensor. As such, no further
initialization pulses will be applied to the sensor.
[0737] In another embodiment, EIS data may be used to determine the
type of sensor being used. Here, it has been discovered that, if
the sensor designs are significantly different, the respective EIS
outputs should also be significantly different, on average.
Different sensor configurations have different model parameters. It
is therefore possible to use identification of these parameters at
any point during the sensor life to determine the sensor type
currently inserted. The parameters can be estimated, e.g., based on
methods described hereinabove in connection with gross
failure/sensitivity-loss analysis. Identification can be based on
common methods to separate values, for example, setting thresholds
on specific (single or multiple) parameters, machine learning (ANN,
SVM), or a combination of both methods.
[0738] This information may be used, e.g., to change algorithm
parameters and initialization sequences. Thus, at the beginning of
the sensor life, this can be used to have a single processing unit
(GST, GSR) to set optimal parameters for the calibration algorithm.
Offline (non real-time), the identification of sensor type can be
used to aid analysis/evaluation of on-the-field sensor
performance.
[0739] It has also been discovered that the length of the
lower-frequency Nyquist slope may be used to differentiate between
different sensor types. FIGS. 103A-103C show Nyquist plots for
three different sensors (i.e., different sensor configurations),
identified as Enlite (8050), Enlite 2 (i.e., "Enlite Enhanced")
(8060), and Enlite 3 (8070), all of which are manufactured by
Medtronic Minimed (Northridge, Calif.). As can be seen, for various
stages, including pre-initialization, post-initialization, and
second post-initialization (FIGS. 103A-103C, respectively), the
Enlite sensor has the shortest lower-frequency Nyquist slope length
(8050), followed by the Enlite 2 (8060), and the Enlite 3 (8070),
which has the longest length. The latter are also shown on FIG.
104, where Nyquist (slope) length, computed as the Cartesian
distance between EIS at 0.105 Hz and 1 Hz, is plotted against
time.
[0740] Embodiments of the invention are also directed to using
diagnostic EIS measurements as a guide in determining the type of
initialization that should be performed. As noted previously,
initialization sequences can be varied based on detected sensor
type (EIS-based or other), and/or detection of whether a new or old
sensor is inserted (EIS-based). In addition, however, EIS-based
diagnostics may also be used in determining a minimal hydration
state prior to initialization (e.g., by tracking Warburg
impedance), or in determining when to terminate initialization
(e.g., by tracking reaction-dependent parameter, such as, e.g., Rp,
Cdl, Alpha, etc.), so as to properly minimize sensor initialization
time.
[0741] More specifically, to minimize initialization response time,
additional diagnostics are required to control the processes that
occur during initialization. In this regard, EIS may provide for
the required additional diagnostics. Thus, for example, EIS may be
measured between each initialization pulse to determine if further
pulsing is required. Alternatively, or in addition, EIS may be
measured during high pulses, and compared to the EIS of optimal
initialization state to determine when the sensor is sufficiently
initialized. Lastly, as noted above, EIS may be used in estimating
a particular model parameter--most likely one or more
reaction-dependent parameters, such as Rp, Cdl, Alpha, etc.
[0742] As has been noted, sensor calibration in general, and
real-time sensor calibration in particular, is central to a robust
continuous glucose monitoring (CGM) system. In this regard,
calibration algorithms are generally designed such that, once a BG
is received by taking a fingerstick, the new BG value is used to
either generate an error message, or update the calibration factor
which, in turn, is used to calculate sensor glucose. In some
previous algorithms, however, a delay of 10-20 minutes may exist
between the time when a fingerstick is entered, and the time when
the user is notified of either the fingerstick being accepted or a
new fingerstick being required for calibration. This is burdensome,
as the user is left not knowing whether he/she will need his/her BG
meter again in a few minutes.
[0743] In addition, in some situations, the presence of older BG
values in the calibration buffer causes either perceived system
delay, due to the newest BG value carrying less than 100% weight,
or inaccuracy in the calculated SG (due to the older BG values no
longer being representative of the current state of the system).
Moreover, erroneous BG values are sometimes entered, but not caught
by the system, which may lead to large inaccuracies until the next
calibration.
[0744] In view of the above, embodiments of the invention seek to
address potential shortcomings in prior methodologies, especially
with regard to sensor performance for use with closed-loop systems.
For example, in order to make the system more predictable,
calibration errors may be notified only when the fingerstick (BG
value) is received by the transmitter (i.e., entered), rather than,
e.g., 10-15 minutes later. Additionally, in contrast to some
existing systems, where a constant calibration error (CE) threshold
is used, embodiments of the invention may utilize variable
calibration error thresholds when higher errors are expected (e.g.,
either due to lower reliability of the sensor, or high rates of
change), thereby preventing unnecessary calibration error alarms
and fingerstick requests. Thus, in one aspect, when the sensor is
in FDC mode, Isig dip calibration mode, or undergoing a high rate
of change (e.g., when 2-packet rate of change.times.CF>1.5
mg/dL/min.), a limit corresponding to 50% or 50 mg/dL may be
used.
[0745] On the other hand, when low error is expected, the system
may use a tighter calibration error limit, such as, e.g., 40% or 40
mg/dL. This reduces the likelihood that erroneous BG values may be
used for calibration, while also allowing the status of the
calibration attempt to be issued immediately (i.e., accepted for
calibration, or a calibration error). Moreover, in order to handle
situations where newer Isig values would cause a calibration error,
a check at calibration time (e.g., 5-10 minutes after fingerstick)
may select the most appropriate filtered Isig (fIsig) value to use
for calibration.
[0746] In connection with the aforementioned issues involving BG
values and the BG_buffer, embodiments of the invention aim to
reduce the delay, and the perceptions of delay, by assigning higher
weighting to the newer BG value than was assigned in previous
algorithms, and by ensuring that the early calibration update
occurs more frequently. In addition, in situations where there is a
confirmed sensitivity change (as confirmed, e.g., by the Smart
Calibration logic mentioned previously and to be explored
hereinbelow, and by recent calibration BG/Isig ratios), the
calibration buffer may undergo partial clearing. Lastly, whereas
prior algorithms may have employed an expected calibration factor
(CF) weight which was a constant, embodiments of the invention
provide for a variable CF value based on sensor age.
[0747] In short, embodiments of the invention provide for variable
calibration error thresholds based on expectation of error during
calibration attempt, as well as issuance of calibration error
message(s) without waiting for additional sensor data, less delay
in calibrating (e.g., 5-10 minutes), updated expected calibration
factor value based on sensor age, and partial clearing of the
calibration buffer as appropriate. Specifically, in connection with
First Day Compensation (FDC), embodiments of the invention provide
for requesting additional calibrations when higher Cal Factor
thresholds are triggered in order to more expeditiously correct
sensor performance. Such higher CF thresholds may be set at, e.g.,
between 7 and 16 mg/dL/nA, with the latter serving as the threshold
for indication of calibration error in embodiments of the
invention.
[0748] Thus, in one aspect, if a high CF threshold is triggered
after the first calibration, the system requires that the next
calibration be performed in 3 hours. However, if a high CF
threshold is triggered after the second, or subsequent,
calibration, the system requires that the next calibration be
performed in 6 hours. The foregoing procedure may be implemented
for a period of 12 hours from sensor connection.
[0749] In another aspect, the expected Cal Factor, which is used
during calibration to calculate the Cal Factor, is increased over
time so as to reduce the likelihood of under-reading. By way of
background, existing methodologies may use a fixed expected Cal
Factor throughout the sensor life, without accounting for possible
shifts in sensor sensitivity. In such methodologies, the expected
Cal Factor may be weighted in calculating the final Cal Factor, and
used to reduce noise.
[0750] In embodiments of the present invention, however, the
expected CF is calculated as a function of time, expressed in terms
of the age of the sensor. Specifically,
Expected CF = SensorAge .times. 0.109 mg / dL / nA day + 4.730 mg /
dL / nA ##EQU00037##
where Sensor Age is expressed in units of days. In further
embodiments, the expected Cal Factor may be calculated as a
function of the existing CF and impedance, such that any changes in
sensitivity may be reflected in the expected CF. In addition, in
aspects of the invention, expected CF may be calculated on every
Isig packet, rather than doing so only at a BG entry, so as to
gradually adjust the Cal Factor between calibrations.
[0751] In connection with calibration buffer and calibration error
calculations, embodiments of the invention provide for modification
of calibration buffer weights and/or clearing of the calibration
buffer. Specifically, when impedance measurements (e.g., through
EIS) indicate that the Cal Factor might have changed, and a
calibration attempt indicates that a change might have occurred,
the change in Cal Ratio (CR) is checked by comparing the CR of the
current BG to the most recent CR in the calibration buffer. Here,
such a change may be verified by, e.g., values of the 1 kHz
impedance, as detailed previously in connection with related EIS
procedures. In addition, weights may be added in the calibration
buffer calculation based on reliability indices, the direction in
which the Cal Factor is expected to change, and/or the rate of
change of calibration. In the latter situation, e.g., a lower
weight may be assigned, or CF only temporarily updated, if
calibration is on a high rate of change.
[0752] In embodiments of the invention, selection of filtered Isig
(fIsig) values for the calibration buffer may be initiated on the
second Isig packet after BG entry. Specifically, the most recent of
the past three (3) fIsig values that would not cause a calibration
error may be selected. Then, once accepted for calibration, the
calibration process will proceed without a calibration error being
issued. Such calibration error may be caused, e.g., by an invalid
Isig value, a Cal Ratio range check, a percentage error check,
etc.
[0753] In other embodiments, values of fIsig may be interpolated to
derive a one minute resolution. Alternatively, fIsig values may be
selected from recent values based on the rate of change in the
values (and accounting for delays). In yet another alternative
embodiment, fIsig values may be selected based on a value of CR
that is closest to a predicted CR value. The predicted CR value, in
turn, is closest to the current value of the Cal Factor, unless the
latter, or EIS data, indicate that CF should change.
[0754] As noted previously, in connection with FIGS. 24 and 34,
e.g., values for 1 kHz real impedance provide information on
potential occlusion(s) that may exist on the sensor membrane
surface, which occlusion(s) may temporarily block passage of
glucose into the sensor and thus cause the signal to dip. More
broadly, the 1 kHz real impedance measurement may be used to detect
sensor events that are typically sudden, and may indicate that the
sensor is no longer fully inserted. In this regard, FIG. 105 shows
a flow chart for a method of blanking sensor data or terminating
the sensor in accordance with an embodiment of the invention.
[0755] The methodology starts at block 9005, where 1 kHz real
impedance values are filtered using, e.g., a moving average filter,
and, based thereon, a determination is made as to whether the
EIS-derived values are stable (9010). If it is determined that the
EIS-derived values are not stable, the methodology proceeds to
block 9015, wherein a further determination is made based on the
magnitude of the 1 kHz impedance. Specifically, if both the
filtered and unfiltered values of 1 kHz real impedance are less
than 7,000.OMEGA., then EIS is set as stable (9020). If, on the
other hand, both the filtered and unfiltered values of 1 kHz real
impedance are not less than 7,000.OMEGA., then EIS is set as
unstable (9025). It is noted that the above-described 7,000.OMEGA.
threshold prevents data blanking or sensor termination for sensors
that have not stabilized.
[0756] When EIS is stable, the algorithm proceeds to block 9030.
Here, if the 1 kHz real impedance is less than 12,000.OMEGA.
(9030), and also less than 10,000.OMEGA. (9040), the algorithm
determines that the sensor is within normal operating range and, as
such, allows sensor data to continue to be displayed (9045). If, on
the other hand, the 1 kHz real impedance value is greater than
10,000.OMEGA. (i.e., when the 1 kHz real impedance is between 10
k.OMEGA. and 12 k.OMEGA.), the logic determines whether the 1 kHz
real impedance value has been high (i.e., greater than 10 k.OMEGA.)
for the past 3 hours (9050). If it is determined that the 1 kHz
real impedance value has been high for the past 3 hours, then the
sensor is terminated at 9060, as the sensor is assumed to have
pulled out, rendering sensor data invalid. Otherwise, the sensor is
not terminated, as the sensor signal may be simply drifting, which,
as discussed previously, may be a recoverable phenomenon.
Nevertheless, the sensor data is blanked (9055) while the sensor is
given a chance to recover.
[0757] It is noted that, in further embodiments, in determining
whether data should be blanked, or the sensor terminated, the logic
may also consider, in addition to the above-mentioned thresholds,
sudden increases in impedance by, e.g., comparing impedance
derivatives to historical derivatives. Moreover, the algorithm may
incorporate noise-based blanking or termination, depending on the
duration of high noise-low sensor signal combination. In this
regard, prior methodologies included termination of the sensor
after three (3) consecutive 2-hour windows of high noise and low
sensor signal. However, in order to prevent unreliable data from
being displayed to the user, embodiments of the invention employ
noise-based blanking, wherein the algorithm stops calculating SG
values after 2 consecutive 2-hour windows (i.e., at the start of
the third consecutive window) involving high noise and low signal.
In further aspects, the algorithm may allow further calculation and
display of the calculated SG values after one hour of blanking,
rather than two hours, where the sensor signal appears to have
recovered. This is an improvement over methodologies that blank
otherwise reliable data for longer periods of time.
[0758] Whereas 1 kHz real impedance may be used to detect sudden
sensor failures, measurements of imaginary impedance at higher
frequencies (e.g., 8 kHz) may be used to detect more gradual
changes, where sensor sensitivity has drifted significantly from
its typical sensitivity. In this regard, it has been discovered
that a large shift in 8 kHz imaginary impedance typically signifies
that the sensor has experienced a large change in glucose
sensitivity, or is no longer stable.
[0759] FIG. 106 shows a flow diagram for a method of sensor
termination in accordance with an embodiment of the invention. As
shown in FIG. 106, the algorithm employs a reference at 1.5 days
(since sensor start), as doing so provides for a more robust logic,
and ensures that the logic focuses on long-term sensitivity
changes. Thus, if the sensor has not been operating for at least
1.5 days (9002), no action is taken, and the algorithm "waits"
(9012), i.e., it periodically loops back to step 9002. Once the
condition in block 9002 is met, a determination is made as to
whether a reference imaginary impedance value is set (9022). If a
reference value has not been previously set, the algorithm proceeds
to set one by assigning the minimum 8 kHz imaginary impedance value
since sensor initialization as the reference value (9032), clipped
within the range -1,000.OMEGA.-800.OMEGA.. With the reference value
set, a change value is calculated as the absolute value of the
difference between the reference value and the current value of the
8 kHz imaginary impedance (9052). In block 9062, the algorithm
determines whether the change value is greater than 1,200.OMEGA.
for two consecutive measurements, as well as whether the Cal Ratio
is larger than 14. If at least one of the latter inquiries is
answered in the negative, then the sensor is allowed to continue
operating and display SG values (9072). However, if the change
value is greater than 1,200.OMEGA. for two consecutive
measurements, and the Cal Ratio is larger than 14, then the sensor
is terminated at block 9082.
[0760] Embodiments of the invention are also directed to assessment
of reliability of sensor glucose values, as well as estimation of
sensor-data error direction, in order to provide users and
automated insulin delivery systems--including those in closed-loop
systems--an indicator of how reliable the system is when SG is
displayed to the user. Depending on the reliability of sensor data,
such automated systems are then able to assign a corresponding
weight to the SG, and make a determination as to how aggressively
treatments should be provided to users. Additionally, the direction
of error can also be used to inform users and/or the insulin
delivery system in connection with SG being a "false low" or a
"false high" value. The foregoing may be achieved by, e.g.,
detecting dips in sensor data during the first day (EIS dip
detection), detecting sensor lag, and lower-frequency (e.g., 10 Hz)
impedance changes.
[0761] Specifically, in accordance with an embodiment of the
invention, it has been discovered that a Cal Factor (CF) of above
about 9 mg/dL/nA may be indicative of low sensor reliability and,
as such, a predictor of higher error. Thus, CF values outside of
this range may be generally indicative of one or more of the
following: abnormal glucose sensitivity; calibrations that occurred
during a dip in signal; delay in entering BG information, or high
rate of change when calibrating; BG error when calibrating; and
sensor with a transient change in glucose sensitivity.
[0762] FIG. 107 shows a flow diagram for a signal dip detection
methodology in accordance with an embodiment of the invention,
where increases in unfiltered real 1 kHz impedance may be used in
combination with low Isig values to identify the start of a dip. As
shown in the diagram, at block 9102, the logic determines whether
sensor data is currently being blanked due to signal dip. If data
is not being blanked, then the logic determines whether less than 4
hours have passed since sensor start (9104). If more than 4 hours
have elapsed since sensor start, the logic then determines whether
more than 12 hours have passed since sensor start (9106), in which
case there will be no dip detection or blanking of data (9108).
Thus, in this regard, the methodology is directed to identifying
transient dips during the first 12 hours of sensor data.
[0763] Returning to block 9106, if less than 12 hours have passed
since sensor start, then an inquiry is made regarding the recent
EIS, Isig, and SG values. Specifically, in block 9110, if the two
most-recent real impedance values (at 1 kHz) have been increasing,
Isig<18 nA, and SG<80 mg/dL, then the algorithm determines
that the start of a dip has been detected, and notifies the system
to stop displaying SG values (9112). On the other hand, if all of
the foregoing conditions are not met, then there will be no dip
detection or data blanking (9108).
[0764] When it is determined, at block 9104, that less than 4 hours
have passed since sensor start, then a sensor dip event may still
be encountered. Specifically, if the two most-recent EIS (i.e., 1
kHz impedance) values are increasing, and Isig<25 nA, then the
algorithm determines that the start of a dip has been detected, and
notifies the system to stop displaying SG values (9114, 9116). If,
however, the two most-recent 1 kHz impedance values are not
increasing, or Isig is not less than 25 nA, then there will be no
dip detection or data blanking (9108), as before.
[0765] Returning to block 9102, if it is determined that data is
currently being blanked due to a dip, there is still a possibility
that data will nevertheless be shown. That is, if Isig is greater
than about 1.2 times Isig at the start of the dip event (9118),
then it is determined that Isig has recovered, i.e., the dip event
is over, and data display will resume (9122). On the other hand, if
Isig is not greater than about 1.2 times Isig at the start of the
dip event (9118), then it is determined that Isig has not yet
recovered, i.e., the dip event is not over, and the system will
continue to blank sensor data (9120).
[0766] In accordance with embodiments of the invention, the
direction of error in SG (under-reading or over reading), in
general, may be determined by considering one or more factors
related to under- and/or over-reading. Thus, it has been discovered
that under-reading in sensors may occur when: (1) Vcntr is extreme
(e.g., Vcntr<-1.0 V); (2) CF is high (e.g., CF>9); (3) lower
frequency impedance (e.g., at 10 Hz) is high (e.g., real 10 Hz
impedance>10.2 k.OMEGA.); (4) FDC is in low CF mode; (5) sensor
lag suggests under-reading; (6) lower frequency impedance (e.g., at
10 Hz) increases (e.g., 10 Hz impedance increases over 700.OMEGA.);
and/or (7) EIS has detected a dip. Over-reading, on the other hand,
may occur when: (1) lower frequency impedance (e.g., 10 Hz)
decreases (e.g., lower frequency impedance<-200.OMEGA.); (2)
sensor lag suggests over-reading; and/or (3) FDC when CF is in
extreme mode.
[0767] Such under-reading or over-reading, especially in
closed-loop systems, can have a profound impact on patient safety.
For example, over-reading near the hypoglycemic range (i.e., <70
mg/dL) may cause an overdose of insulin to be administered to the
patient. In this regard, several indicators of error direction have
been identified, which may be used as test criteria, including: (1)
low sensitivity indicators; (2) sensor lag; (3) FDC mode; and (4)
loss/gain in sensitivity since calibration.
[0768] Two such low sensitivity indicators are high
(lower-frequency) real impedance (e.g., >10 k.OMEGA.) and high
Vcntr (e.g., Vcntr<-1.0V), both of which are, in general,
indicative of loss of sensitivity. FIG. 108A shows an example in
which Vcntr 9130 gradually increases (i.e., become more negative)
as a function of time. At about 115 hours, shown by line 9135,
Vcntr crosses -1.0V, as indicated by line 9137, and continues to
increase (i.e., Vcntr<-1.0V) to about -1.2V. As shown, prior to
about 115 hours, the Isig trend 9132 generally follows the Vcntr
trend. However, once Vcntr passes the threshold (i.e., to the right
of line 9135), the Isig departs from Vcntr, and continues to drop.
At the same time, as shown in FIG. 108B, glucose 9134 also has a
generally downward trend, with Cal errors 9136 being indicated at
about 130 hours and about 165 hours.
[0769] As discussed previously, (EIS) sensor dips are also
indicative of temporary sensitivity loss. Similarly, a high Cal
Factor is indicative of the sensor's attempt to compensate for
reduced sensitivity. In one example shown in FIGS. 109A and 109B,
the Cal Factor 9140 increases steadily as a function of time. At
about 120 hours (9145), the Cal Factor 9140 crosses a threshold
value of 9 (9147). As shown in FIG. 109B, once the Cal Factor
crosses the threshold, the glucose values 9142 show more frequent
departures from BG values, with several errors 9144 occurring
between about 135 hours and 170 hours.
[0770] As mentioned previously, sensor lag is another indicator of
error direction. Accordingly, in an embodiment of the invention,
the error that is caused by sensor lag is compensated for by
approximating what the glucose value will be. Specifically, in an
embodiment of the invention, the error from sensor lag may be
approximated by defining:
sg(t+h)=sg(t)+hsg'(t)+1/2h.sup.2sg''(t)
where sg(t) is the sensor glucose function, and "h" is the sensor
lag. The error may then be calculated as
Error = sg ( t + h ) - sg ( t ) sg ( t ) = ( hsg ' ( t ) + 1 2 h 2
sg '' ( t ) ) sg ( t ) ##EQU00038## or ##EQU00038.2## Error = k ( C
1 sg ' ( t ) + C 2 sg '' ( t ) ) sg ( t ) . ##EQU00038.3##
[0771] First day calibration (FDC) occurs when the Cal Factor (CF)
is not within the expected range. The CF is set to the value
indicated by the calibration, and then ramps up or down to the
expected range, as shown, e.g., in FIGS. 110A and 110B. During this
time, usually high, but generally predictable, errors may exist,
resulting in potential over-reads or under-reads. As can be seen
from FIGS. 110A and 110B, the CF changes at a generally constant
slope as it rises or falls, and then settles, in this case at 4.5
or 5.5.
[0772] Lastly, post-calibration sensitivity change, i.e., loss/gain
in sensitivity since calibration, is also an indicator of
error/error direction. Under normal circumstances, and except for
first day calibration as discussed hereinabove, the Cal Factor
remains generally constant until a new calibration is performed.
Shifts in sensitivity after calibration, therefore, can cause
over-reads and under-reads which, in turn, may be reflected by
values of lower-frequency (e.g., 10 Hz) real impedance.
[0773] Specifically, it has been discovered that a drop in
lower-frequency real impedance causes over-reading, with the
direction of error being indicated by the real impedance curve.
Conversely, lower-frequency real-impedance increases cause
under-reading, with the direction of error also being indicated by
the real impedance curve. However, current directionality tests may
be unable to readily decipher points at peaks and valleys of the
glucose profile. Thus, in one embodiment, the degree of sharpness
of such peaks and valleys may be reduced by filtering, such as,
e.g., by deconvolution with lowpass filtering.
[0774] As described previously in connection with FIG. 81, e.g.,
sensitivity change and/or loss may be used to inform proper sensor
calibration. In this regard, in a further aspect of the invention,
changes in sensor sensitivity may be predicted based on the
previous calibration factor or on impedance so as to enable
implementation of "smart calibrations", which help address
continued generation and/or display of inaccurate glucose data
when, e.g., sensor sensitivity has changed.
[0775] It is known that, in some existing continuous glucose
monitoring systems (CGMS), calibration fingersticks are required
every twelve hours. The calibration allows the CGMS to update the
function used to convert the measured sensor current into a
displayed glucose concentration value. In such systems, the 12-hour
calibration interval is selected as a balance between reducing the
user burden (of performing too many fingersticks) and using an
interval that is sufficient to adjust for changes in sensor
sensitivity before inaccuracies can cause too large of a problem.
However, while this interval may be appropriate in general, if the
sensor sensitivity has changed, 12 hours can be too long to wait if
a high level of accuracy (in support of closed loop insulin
delivery) is expected.
[0776] Embodiments of the invention, therefore, address the
foregoing issues by using the previous calibration factor (see
discussion of FDC below), or impedance (see discussion of EIS-based
"smart calibrations" below), to predict if sensitivity has changed.
Aspects of the invention also use time limits to maintain
predictability for users, as well as include steps (in the
associated methodology) to ensure that detection is robust to
variations between sensors.
[0777] FIG. 111 shows a flow diagram in accordance with an
embodiment of the invention for First Day Calibration (FDC).
Starting at block 9150, if FDC is not on after successful
calibration, there is simply no smart calibration request (9151).
However, if FDC is on, a determination is made at block 9153 as to
whether this is the first calibration and, if it is not, then a
smart calibration request is made, with the timer set for 6 hours,
i.e., it is requested that an additional calibration be made in 6
hours (9155). If, on the other hand, this is the first calibration,
then block 9157 determines whether the Cal Ratio is less than 4, or
greater than 7. If the condition in block 9157 is not met, then the
logic proceeds to block 9155 where, as noted above, a smart
calibration request is made, with the timer set for 6 hours.
However, if the criterion in block 9157 is not met, then a smart
calibration request is made, with the timer set for 3 hours, i.e.,
it is requested that an additional calibration be made in 3 hours
(9159). Thus, in order to improve accuracy for sensors which need
calibration adjusted, additional (smart) calibrations are requested
which, in turn, limit the amount of time where the adjustment is
incorrect.
[0778] In contrast with FDC mode, EIS-based smart calibration mode
provides for additional calibrations if impedance changes. Thus, in
an embodiment of the invention shown in FIG. 112, an allowed range
relating to impedance values (and as defined hereinbelow) is set in
the hour after calibration and, following the calibration, a
request for additional calibrations is made if impedance is outside
of range. Thus, if not within one hour since calibration, a
determination is made as to whether the filtered 1 kHz imaginary
impedance value is outside of range (9160, 9162). If the impedance
value is not outside of range, then no change is made (9164).
However, if the filtered 1 kHz imaginary impedance value is outside
of range, then the calibration timer is updated so that calibration
is requested to be performed at 6 hours from the previous
calibration (9168). It is noted that, while higher-frequency
imaginary impedance tends to better identify changes in glucose
sensitivity, towards the higher end of the frequency spectrum,
measurements are generally noisier and, as such, may require
filtering.
[0779] Returning to block 9160, if it is determined that less than
one hour has passed since calibration, then the range for impedance
values may be updated (9166). Specifically, in one embodiment, the
impedance range calculation is performed on the last EIS
measurement 1 hour after calibration. In a preferred embodiment,
the range is defined as
range=3.times.median(|x.sub.i-x.sub.j|)
where j is the current measurement, and i are the most recent 2
hours of values. In addition, the range may be limited to be values
between 50.OMEGA. and 100.OMEGA.. It is noted that the range as
defined above allows for 3 times median value. The latter has been
discovered to be more robust than the 2-standard-deviation approach
used in some prior algorithms, which allowed noise and outliers to
cause inconsistencies.
[0780] While the description above refers to particular embodiments
of the present invention, it will be understood that many
modifications may be made without departing from the spirit
thereof. Additional steps and changes to the order of the
algorithms can be made while still performing the key teachings of
the present invention. Thus, the accompanying claims are intended
to cover such modifications as would fall within the true scope and
spirit of the present invention. The presently disclosed
embodiments are, therefore, to be considered in all respects as
illustrative and not restrictive, the scope of the invention being
indicated by the appended claims rather than the foregoing
description. All changes that come within the meaning of, and range
of, equivalency of the claims are intended to be embraced
therein.
* * * * *