U.S. patent application number 15/381235 was filed with the patent office on 2017-04-27 for multi-stage permanent magnet structure and integrated power inductors.
The applicant listed for this patent is Board of Trustees of The University of Alabama. Invention is credited to Jaber A. Abu Qahouq.
Application Number | 20170117077 15/381235 |
Document ID | / |
Family ID | 57867639 |
Filed Date | 2017-04-27 |
United States Patent
Application |
20170117077 |
Kind Code |
A1 |
Abu Qahouq; Jaber A. |
April 27, 2017 |
MULTI-STAGE PERMANENT MAGNET STRUCTURE AND INTEGRATED POWER
INDUCTORS
Abstract
Apparatuses and methods directed to multi-stage permanent magnet
and implementations of a permanent magnet on-chip power inductor.
Various circuit models, design considerations and simulation
results are described. The multi-stage permanent magnet includes
layers with uniform or non-uniform magnets used to control the flux
distribution. The permanent magnet on-chip power converter for
DC-DC switching power converters that may include a top ferrite
layer, a spiral winding layer, a permanent magnet layer, a bottom
ferrite layer, and a substrate layer. The permanent magnet layer
may comprise a multi-stage structure wherein each stage has a
decreasing area as compared to an immediate lower stage. A method
of manufacturing a Permanent On-Chip Power Inductor (PMOI) is also
disclosed.
Inventors: |
Abu Qahouq; Jaber A.;
(Tuscaloosa, AL) |
|
Applicant: |
Name |
City |
State |
Country |
Type |
Board of Trustees of The University of Alabama |
Tuscaloosa |
AL |
US |
|
|
Family ID: |
57867639 |
Appl. No.: |
15/381235 |
Filed: |
December 16, 2016 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
|
|
14289161 |
May 28, 2014 |
9558878 |
|
|
15381235 |
|
|
|
|
61827851 |
May 28, 2013 |
|
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Current U.S.
Class: |
1/1 |
Current CPC
Class: |
H01F 2017/0073 20130101;
H01F 7/0278 20130101; H01F 17/0006 20130101; H01F 41/046 20130101;
H01F 2017/0066 20130101; H01F 27/24 20130101; B82Y 25/00 20130101;
Y10S 977/932 20130101; H01F 27/28 20130101 |
International
Class: |
H01F 7/02 20060101
H01F007/02; H01F 27/24 20060101 H01F027/24; H01F 27/28 20060101
H01F027/28 |
Claims
1. A power inductor for a DC-DC power converter, comprising: a
magnetic core; at least one winding wound about the magnetic core;
and a permanent magnet disposed within the magnetic core.
2. The power inductor of claim 1, wherein the magnetic core is
formed from a top ferrite layer, a winding layer and a bottom
ferrite layer, and wherein the permanent magnet is disposed between
the winding layer and the bottom ferrite layer.
3. The power inductor of claim 2, wherein the magnetic core is
formed on a substrate.
4. The power inductor of claim 1, wherein the magnetic core is
formed as a toroid, and wherein the permanent magnet is disposed in
a gap formed in the toroid.
5. The power inductor of claim 1, wherein the permanent magnetic is
coupled to the magnetic core having a plurality of legs each having
an associated winding, and wherein at least two windings are
coupled together.
6. The power inductor of claim 1, wherein the permanent magnetic is
uncoupled to the magnetic core.
7. A magnet, comprising: a plurality of magnetic layers arranged as
a multi-stage structure, wherein each stage has a decreasing area
as compared to an immediate lower stage.
8. The magnet of claim 7, wherein each stage is formed having a
generally square shape.
9. The magnet of claim 7, wherein the permanent magnet layer is
generally pyramid-shaped.
10. The magnet of claim 7, wherein the magnet comprises 10
stages.
11. The magnet of claim 7, wherein the magnet has a predetermined
magnetic field distribution designed to cancel winding flux.
12. The magnet of claim 11, wherein the magnet is provided in a
motor or generator in order to optimize the magnetic field
distribution.
Description
CROSS-REFERENCE TO RELATED APPLICATION
[0001] This application is a continuation of U.S. patent
application Ser. No. 14/289,161, filed May 28, 2014, entitled
"MULTI-STAGE PERMANENT MAGNET STRUCTURE AND INTEGRATED POWER
INDUCTORS," which claims priority to U.S. Provisional Patent
Application No. 61/827,851, filed May 28, 2013, entitled "PERMANENT
MAGNET INTEGRATED POWER INDUCTORS FOR DC-DC SWITCHING POWER
CONVERTERS." The respective disclosures of the above-referenced
applications are each incorporated herein by reference in its
entirety.
BACKGROUND
[0002] The performance of power devices such as power inductors and
power FETs (Field Effect Transistors) affects the performance of
switching power converter applications. The power inductor is one
of the largest and most lossy components in a switching power
converter, and it is accountable for much of the weight and the
size of switching power converters. Several figures of merit are
considered for power inductors including the inductance density,
the current/power density, the DC resistance, the AC
characteristics and the saturation current. In order to obtain
higher inductance density, in other words, achieving the required
inductance in as small area as possible, technologies like planner
inductor, integrated inductors, micro-fabricated inductor and on
chip inductor have emerged over the years. Saturation current is
related to the core structure as well as the magnetic core
material. For a given core structure and design, employing a
magnetic material with higher saturation flux density helps to
obtain a higher saturation current.
[0003] The power inductor, as a form of multiple winding coupled
magnetic structures, has been used in many applications, such as is
in multi-phase power converters. One of the main advantages of the
coupled power inductor used in DC-DC power converters is the
ability to obtain smaller equivalent transient inductance
(advantageous for lower output voltage dynamic deviation under
transients) with a larger equivalent steady-state inductance
(advantageous for smaller steady-state output voltage ripple and
higher power efficiency).
[0004] In a two-phase inductor, the two inductor windings can be
directly or inversely coupled. Inversely coupled power inductor was
employed in the multi-phase switching power converters to improve
both the steady-state and transient performances. Permanent magnet
power inductors (PMPI) utilize a permanent magnet (PM) to partially
offset the flux in the magnetic core due to the DC component of the
winding current, so that a higher saturation current could be
obtained by the same core structure power inductor.
SUMMARY
[0005] In accordance with some implementations described herein,
there is presented theory, apparatuses and methods directed to
implementations of a permanent magnet couple power inductor.
Various circuit models, design considerations and simulation
results are described. Also presented is an on-chip implementation
and fabrication techniques.
[0006] In accordance with an aspect, there is disclosed a permanent
magnet on-chip power converter for DC-DC switching power converters
that may include a top ferrite layer (or magnetic core), a spiral
winding layer, a permanent magnet layer, a bottom ferrite layer,
and a substrate layer. The permanent magnet layer may comprise a
multi-stage structure wherein each stage has a decreasing area as
compared to an immediate lower stage.
[0007] In accordance with other aspects, there is disclosed a
method of manufacturing a Permanent On-Chip Power Inductor (PMOI).
The method may include depositing of bottom ferrite layer on top of
Si wafer having a SiO2 layer; sputtering a first SiO2 insulation
layer on the bottom ferrite layer; depositing a multi-stage
permanent magnet on the SiO2 insulation layer; sputtering of a seed
layer for a winding layer; coating and pattering a photoresist mold
for a spiral winding in the winding layer; filing an isolation
material in between windings of the spiral winding; sputtering a
second SiO2 insulation layer on the winding layer; and depositing a
top ferrite layer.
[0008] In accordance with yet other aspects, there is disclosed a
Permanent Magnet Couple Power Inductor (PMCI) that may include a
first winding wound around a first leg, a second winding wound
around a second leg, and a permanent magnet disposed within a gap
central of a central leg. A first flux path associated with the
first winding and a second flux path associated with the second
winding interact with each other and are at least partially
canceled by the permanent magnet.
[0009] This summary is provided to introduce a selection of
concepts in a simplified form that are further described below in
the detailed description. This summary is not intended to identify
key features or essential features of the claimed subject matter,
nor is it intended to be used to limit the scope of the claimed
subject matter.
BRIEF DESCRIPTION OF THE DRAWINGS
[0010] The foregoing summary, as well as the following detailed
description of illustrative implementations, is better understood
when read in conjunction with the appended drawings. For the
purpose of illustrating the implementations, there are shown in the
drawings example constructions; however, the implementations are
not limited to the specific methods and instrumentalities
disclosed. In the drawings:
[0011] FIGS. 1A and 1B show a spiral conventional on-chip power
inductor (COPI) structure;
[0012] FIGS. 2A-2D illustrate examples permanent magnet (PM)
structures;
[0013] FIGS. 3A-3B illustrate operation points under different
input currents 0, I.sub.1, and I.sub.2
(0<I.sub.1<I.sub.2);
[0014] FIGS. 4A-4C illustrate an example spiral Permanent On-Chip
Power Inductor (PMOI);
[0015] FIGS. 5A-5B show ANSYS.RTM./Maxwell.RTM. 3-D physical model
of the example PMOI;
[0016] FIGS. 6A-6D show the B field of the spiral conventional
on-chip power inductor (COPI);
[0017] FIGS. 7A-7E show the B field of PMOI when the DC input
current increases from zero to 8 A;
[0018] FIG. 8 shows an example process of the PMOI fabrication;
[0019] FIG. 9 shows the process of FIG. 20 in more detail.
[0020] FIG. 10 illustrates a front view of a permanent magnet
coupled power inductor (PMCI) with an EI core structure;
[0021] FIG. 11 illustrates a magnet hysteresis loop and B-H
characteristic of a permanent magnet material;
[0022] FIGS. 12A-12B illustrate an operation region comparison of
(a) conventional coupled power inductor and (b) PMCI Magnetic
Circuit Model of Permanent Magnet Coupled Power Inductor;
[0023] FIG. 13 illustrates a magnetic circuit model of a proposed
EI core structure PMCI;
[0024] FIG. 14 illustrates a diagram of an example permanent magnet
coupled power inductor;
[0025] FIGS. 15A-15D show a ANSYS.RTM./Maxwell.RTM. 3-D physical
model of an example PMCI;
[0026] FIG. 16 illustrates a plot of the flux linkage;
[0027] FIGS. 17A-17E illustrate the flux density (B) vector changes
when the DC input current increases from zero to 25 A;
[0028] FIG. 18 illustrates a co-simulation model of an example
two-phase buck power converter with the PMCI;
[0029] FIGS. 19A-19B illustrate front views of a toroid power
inductor (TPI) and a permanent magnet toroid power inductor
(PMTPI);
[0030] FIG. 20 illustrates a diagram of an example PMTPI design;
and
[0031] FIG. 21 shows ANSYS.RTM./Maxwell.RTM. 3-D physical model of
the example PMTPI.
DETAILED DESCRIPTION
[0032] The present disclosure presents theory, apparatuses and
methods directed to implementations of a permanent magnet on-chip
power inductor. Various circuit models, design considerations and
simulation results are described.
[0033] Introduction to Permanent On-Chip Power Inductor (PMOI)
[0034] PMOI Structure and Operation Principle
[0035] The saturation current of a conventional power inductor can
be described as in equation (1),
I sat = N B sat A e L ( 1 ) ##EQU00001##
[0036] Where N is the winding turns, B.sub.sat is the saturation
flux density of the inductor core material, A.sub.e is the
effective cross section area of the flux path and L is inductance,
which is given by equation (2).
L = N 2 .mu. e .mu. 0 A e l e ( 2 ) ##EQU00002##
.mu..sub.0=4.pi.10.sup.-7 H/m is the vacuum permeability,
.mu..sub.e is the effective permeability of the flux path and
l.sub.e is the effective length of the flux path.
[0037] Due to the cancellation effect of the PM magnetic flux, the
saturation current of the PMOI can be described as equation
(3),
I.sub.sat-PMOI=I.sub.sat+I.sub.c (3)
Where, I.sub.c denotes the PM cancellation current, which is the
current value in the winding that results in zero net flux in the
PMOI core.
[0038] FIG. 1A shows a spiral conventional on-chip power inductor
(COPI) structure 100. There are three layers on top of the Si wafer
for this inductor: a spiral copper winding, the top and bottom
inductor core layers. FIG. 1B shows the PMOI structure diagram 102,
where there is one additional permanent magnet layer 104 in between
the bottom and core layer and spiral winding compared with the
structure in FIG. 1 (a).
[0039] In order to obtain higher saturation current for the PMOI
design, the PM in the PMOI (part of the invention) structure has to
cancel as much winding flux as possible. Therefore, PM structure
should carefully be designed for different shapes of windings in
order to cancel more winding flux and get higher cancellation
current. PM structure for spiral winding is investigated below.
[0040] An example spiral winding 200 is shown in FIG. 2A and its B
field (when the DC input current is 6 A) is shown in FIG. 2A. It
can be observed that B field of the winding is not uniformly
distributed. Thus, in order to cancel more winding flux, similar B
field distribution for PM is required.
[0041] There are several candidate structures for PM layer (part of
the invention). Three of them are shown in FIGS. 2B-2D. The
simplest structure is rectangular thin film PM 202, as shown in
FIG. 2B (conventional PM), whose B field distribution is shown in
FIG. 2B. In this case, too much flux is concentrated on the edges
of the PM film, which if used in the power inductor, it will cause
it to saturate. Meanwhile, there is almost no flux on top and
bottom of the central part of rectangular PM. Thus, this
configuration may not satisfy the design objective in terms of flux
cancellation.
[0042] FIG. 2C shows a cone PM film 204, and its B field
distribution is shown in FIG. 2C. FIG. 2D shows a multi-stage PM
structure 208 and the associated B field distribution is shown in
FIG. 2D. The PM structure 208 has a pyramid-like layered
configuration. These two PM structures 208 have the similar B field
distribution compared with spiral winding B field and therefore,
the configuration could be used in the PMOI. However, the thin film
cone structure is more difficult to fabricate compared to
multi-stage PM. Thus, the spiral PMOI uses multi-stage PM as shown
in FIG. 2C.
[0043] In addition to use in a power inductor, the multi-state PM
may be used in other applications to obtain desirable
characteristics, such as motors and generators in order to optimize
field distribution.
[0044] Operation Region Comparison of the Conventional on-Chip
Power Inductor (COPI) and the PMOI
[0045] The PMOI and the COPI operation regions on the BH curve of
magnetic core are compared in this subsection. In FIGS. 3A-3B,
P.sub.copi and P.sub.pmoi denote the operation points of COPI and
the PMOI, respectively. The operation points under different input
currents 0, I.sub.1, and I.sub.2 (0<I.sub.1<I.sub.2) are
shown in FIGS. 3A and 3B. When the input current is 0, P.sub.copi0
is located at origin (both B and H are zero) while P.sub.pmoi0 has
a negative B value. When the input current is I.sub.1, the B of
PMOI is zero while P.sub.copi1 is located in the linear region of
the curve. When the input current is I.sub.2, P.sub.pmoi2 is still
located in the linear region but P.sub.copi2 is already in the
saturation region. The operation region of COPI is represented by
dash-dotted line in FIG. 3A, and the operation region of PMOI is
represented by dashed line in FIG. 3B. It can be observed that PMOI
has a larger operation region compared with the COPI. It is mainly
because the PM opposing flux has a cancelation effect with the
windings flux.
[0046] ANSYS.RTM./Maxwell.RTM. 3-D Modeling and Simulation of
PMOI
[0047] The PMOI design as shown in FIG. 1B is also compared with
the air core spiral power inductor (API) and COPI as was shown in
FIG. 1A.
[0048] Physical Model
[0049] An example spiral PMOI design diagram 200 is shown in FIGS.
4A-4C and its design specifications are listed in Table I. A better
power inductor design requires higher inductance density, higher
frequency, higher saturation current and lower eddy current loss
and magnetic loss. Thus, inductor core material requires higher
permeability (.mu.), saturation flux density (Brat) and higher
resistivity (.rho.). The winding of the conductor placed with the
permanent magnet may be either copper or carbon nanotube (CNT). It
is noted that the winding may be made of other materials. NiCuZn
ferrite is used as the top and bottom inductor core layer.
NdFeBN45SH is used as the multi-stage permanent magnet in this
example design due to its high residual flux density (Br),
coercivity (Hc) and intrinsic coercivity (Hci), but other types can
also be used. The permanent magnet material can be part of the CNT,
such as coating the CNT with permanent magnet material or mixing
the permanent magnet and the CNT material. In some implementations,
10 stages may be provided in this PM structure, where each stage is
0.75 .mu.m thick. In some implementations, the magnetic core used
has non-uniform saturation levels and permeabilities (e.g., graded
with several core layers of different types, thicknesses, widths,
lengths, and areas) in order to match the flux distribution density
of the permanent magnet.
TABLE-US-00001 TABLE I PARAMETER VALUES Name dimension Material
Substrate 525 .mu.m thick Si Spiral winding 3 turns Cu, Carbon
Nonotube 5.2 mm .times. 5.2 mm (CNT) 100 .mu.m thick Wire width:
0.7 mm Winding gap: 50 .mu.m Bottom core 5.2 mm .times. 5.2 mm
NiCuZn ferrite layer 10 .mu.m thick Bsat = 0.46T Top core layer 5.2
mm .times. 5.2 mm .mu. = 120 11 .mu.m thick .rho. = 10.sup.8
.OMEGA.m PM layer Base: 5 mm .times. 5 mm NdFeBN45SH Top: 1 mm
.times. 1 mm Br = 1.32T 0.75 .mu.m .times. 10 thick Hc = 1003 kA/m
Hci = 1590 kA/m
[0050] FIGS. 17A-17E show ANSYS.RTM./Maxwell.RTM. 3-D physical
model of the designed PMOI according to the design specifications
in Table I. For comparison purposes, ANSYS.RTM./Maxwell.RTM. 3-D
physical models of an API and COPI without PM but having the same
design specifications with PMOI are also developed. Comparison is
shown in Table II. The dimensions specified in Tables I and II are
approximate values.
TABLE-US-00002 TABLE II ON-CHIP POWER INDUCTOR COMPARISON API COPI
PMOI Inductor dimensions 5.2 mm .times. 5.2 mm .times. 5.2 mm
.times. (without substrate) 5.2 mm .times. 5.2 mm .times. 5.2 mm
.times. 100 .mu.m 130 .mu.m 130 .mu.m Inductance (nH) 29.2 50.9
48.4 Inductance density 10.7 14.5 13.8 (nH/mm.sup.3) Saturation
current (A) -- 4 7
[0051] Simulation Results
[0052] The inductance of API and CPI measured from
ANSYS.RTM./Maxwell.RTM. are 29.2 nH and 50.9 nH, respectively.
FIGS. 5A-5B show B field and vector of the API when the input DC
current is 8 A. It can be observed that the maximum B is 0.019 T,
which is very low because there is no inductor core. The saturation
current of the API is very high but the inductance is low. FIGS.
6A-6D show the B field of the COPI under different input DC current
values. It can be observed from FIGS. 6A-6D that both the top and
bottom core layer is about to saturate when the DC input current is
4 A.
[0053] B Field of the PMOI
[0054] The inductance of PMOI measured from ANSYS.RTM./Maxwell.RTM.
is 48.4 nH, which is close to the inductance value of the CPI (50.9
nH). FIGS. 7A-7E show the B field of PMOI when the DC input current
increases from zero to 8 A. When the input current is 0, B field is
only generated by PM, as shown in FIG. 7A. The B value in FIG. 7A
is less than the Bsat (0.46 T) of both top and bottom ferrite core,
which means the inductor is not saturated by PM itself. From FIGS.
7A-7E, it can be observed that when the input DC current increases
from 0 to 8 A, B value first decreases and then increases. When the
input current is 3 A, the average B value gets the lowest value and
is approximately 0, as shown in FIG. 7B. Thus the cancellation
current of this PMOI design is 3 A. From equation (3), it can be
obtained that the saturation current of the designed PMOI is about
7 A, which is consistent with the observation in FIG. 7D. That is
to say, saturation current of the designed PMOI is 7 A. Thus, the
multi-stage PM layer helps to increase the saturation current from
4 A to 7 A in this PMOI design (theoretically the saturation
current can be doubled for the same size and inductance value).
[0055] Fabrication Process of the PMOI
[0056] FIG. 8 shows an example process of the PMOI fabrication. In
FIG. 8, the fabrication process for PMOI is shown as a series of
steps, including: (1) Depositing of bottom ferrite layer on top of
Si wafer with SiO2 layer; (2) Sputtering SiO2 insulation layer,
then deposit multi-stage permanent magnet on top; (3) Sputtering of
Cu seed layer for winding cupper; (4) coat and pattern photoresist
mold for winding; (5) Electroplate Cu winding; (6) etch away
photoresist mold and Cu seed layer; (7) fill isolation material in
between the Cu windings, then sputtering SiO2 layer on top for
insulation, (8) Top Ferrite deposition. For example, a 525
.mu.m-thick p-type (100) silicon wafers are used. The wafer has a
silicon dioxide layer on top.
[0057] The fabrication process starts with depositing of 10 .mu.m
bottom ferrite layer as show in FIG. 8 (1), then SiO.sub.2 layer is
sputtered for insulation. Next, multi-stage permanent magnet is
deposited in step (2), detailed PM deposition process is shown in
FIG. 9. After sputtering another SiO.sub.2 insulation layer, a Cu
seed layer for electroplating winding cupper is sputtered in step
(3). Then followed by step (4): coating and patterning photoresist
as a mold for spiral cupper winding, and electroplate winding
cupper in step (5). Next, etch away the photoresist mold and Cu
seed layer, then filled with insulation material, as shown in step
(6) and (7). Finally, in step (8), 11 .mu.m top ferrite layer is
deposited.
[0058] More detailed process of multi-stage permanent magnet
deposition is shown in FIG. 9, which shows a deposition process of
multi-stage PM, including: (1) Coating and patterning photoresist
mold for the base stage PM; (2) sputtering of base stage PM; (3)
Photoresist mold for the second stage; (4) sputtering of second
stage PM; (5) repeat (3) and (4) for Multi-stage PM. The process
starts with coating and patterning photoresist as a mold and
followed by sputtering the base stage PM, as shown in FIG. 9
(1)-(2). Then repeat the same process for sputtering the second
stage PM in step (3) and (4). Repeating step (3) and (4) for
multi-stage PM.
[0059] Thus, apparatuses and methods directed to implementations of
a permanent magnet couple power inductor are described. Various
circuit models, design considerations and simulation results are
presented. Also, the above describes an on-chip implementation and
fabrication techniques.
[0060] Introduction to a Permanent Magnet Coupled Power Inductor
for Multi-Phase DC-DC Switching Power Converters
[0061] To obtain larger saturation current and higher inductance
density for coupled power inductors, the present disclosure
describes a permanent magnet coupled power inductor (PMCI) for
multiphase power inductor by employing the operation principle of
PMPI in conventional coupled power inductors. PMCI enables the
reduction of the coupled power inductor size for multi-phase power
converters and therefore, contribute to the higher power density
system integration.
[0062] Permanent Magnet Coupled Power Inductor Structure and
Operation Principle
[0063] FIG. 10 illustrates the front view of a PMCI 1000 with an EI
core structure. In this PMCI structure, two windings 1002, 1004 are
wound around the two side legs 1006, 1008 and a relatively small
piece of permanent magnet (PM) 1010 is placed in the central leg
gap. The DC input current direction and the PM polarity are shown
in FIG. 10. The flux path and direction generated by the left
winding is represented by the solid arrowed line and the flux path
generated by the right winding is represented by dashed arrowed
line. The dash-dotted arrowed line denotes the flux path generated
by the PM 1010. The fluxes generated by the two windings 1002, 1004
interact with each other in FIG. 10, denoting the inverse coupling
effect between these two inductors. The PM flux loops and winding
flux loops are in opposite directions. Thus, the flux generated by
the windings could partially be canceled by the flux generated by
the PM 1010. This flux cancellation effect helps in increasing the
saturation current and inductance density of the power inductor, as
will be described below.
[0064] Permanent magnets have a B-H characteristic with a wide
hysteresis loop in order to prevent demagnetization of the material
as shown in FIG. 11. Generally, the second quadrant of the
hysteresis loop is used in analysis of permanent magnet behavior.
The relationship between flux density (B) and field intensity (H)
is known as the normal curve. The relationship between the
intrinsic magnetization (J) and H is known as the intrinsic curve.
The curves are related at every point by the equation B=J+H. In
FIG. 11, the residual flux density B.sub.r is the maximum flux
density of the PM in a closed loop configuration once the
magnetizing field has been removed. Hc is the demagnetizing force
which will reduce B to zero. Hci is demagnetizing force that
reduces J to zero.
[0065] Demagnetization occurs when a sufficient magnetic field is
applied across the magnet in the opposite direction of
magnetization. For PMCI design, the point on the curve is the
"knee" (as shown in FIG. 11), where the BH curve becomes nonlinear.
Whenever the field is driven at or past the point
(H.sub.x,B.sub.x), then the material will start to be irreversibly
demagnetized. Thus, the PMCI 1000 should avoid such demagnetization
under the maximum or rated input current. In FIG. 11, the dashed
part of the normal curve and dotted part of the intrinsic curve
indicate the operation region of the PM in a PMCI.
[0066] Operation Region Comparison of the Conventional Coupled
Power Inductor and the PMCI
[0067] With reference to FIGS. 12A-12B, there is illustrated an
operation region comparison of (a) conventional coupled power
inductor and (b) PMCI Magnetic Circuit Model of Permanent Magnet
Coupled Power Inductor. The PMCI and the conventional coupled power
inductor operation regions on the BH curve of magnetic core will
now be compared. In FIGS. 12A-12B, P.sub.cci and P.sub.pmci denote
the operation points of conventional coupled power inductor and the
PMCI, respectively. The operation points under different input
currents 0, I.sub.1, and I.sub.2 (0<I.sub.1<I.sub.2) are
shown in FIGS. 12A-12B curve (a) and curve (b). When the input
current is 0, P.sub.cci0 is located at origin (both B and H are
zero) while P.sub.pmci0 has a negative B value. When the input
current is I.sub.1, the B of PMCI is zero while P.sub.cci1 is
located in the linear region of the curve. When the input current
is I.sub.2, P.sub.pmci2 is still located in the linear region but
P.sub.cci2 is already in the saturation region. The operation
region of conventional coupled power inductor is represented by
dash-dotted line in FIG. 12A, and the operation region of PMCI is
represented by dashed line in FIG. 12B. It can be observed that
PMCI has a larger operation region compared with the conventional
coupled power inductor. It is mainly because the PM opposing flux
has a cancelation effect with the windings flux.
[0068] The magnetic circuit model 1300 of the proposed EI core
structure PMCI is shown in FIG. 13. The magnetic reluctance of the
three legs, namely R.sub.1, R.sub.2 and R.sub.c, may be determined
by the following equation:
R i = l .mu. A ( i = 1 , 2 , c ) ( 4 ) ##EQU00003##
Where l is the length of the magnetic flux path; .mu. is
corresponding permeability of materials in the flux path; and A is
corresponding cross-section area of the flux path. Compared to the
conventional coupled power inductor with the same core structure,
the side leg reluctances R.sub.1 and R.sub.2 of the PMCI remain
unchanged. According to equation (4), the permanent magnet piece in
the central leg gap in PMCI affects the central leg reluctance
R.sub.c. Usually, the permeability of PM material (e.g. the
permeability of SmCo28 is 1.038.mu..sub.0) has a value that is very
close to the air permeability (.mu..sub.0=4.pi.10.sup.-7 H/m).
[0069] Based on Ampere's law, for each flux loop in FIG. 13, there
exists
{ R 1 .0. 1 + R c ( .0. 1 + .0. 2 ) = N 1 i 1 - .xi. pm R 2 .0. 2 +
R c ( .0. 1 + .0. 2 ) = N 2 i 2 - .xi. pm ( 5 ) ##EQU00004##
Where O is the magnetic flux, N is the winding number of turns and
.xi.pmis the magnatomotive force of the permanent magnet piece.
[0070] The flux in each side leg can then be expressed by:
{ .0. 1 = N 1 ( R 2 + R c ) .DELTA. i 1 - N 2 R c .DELTA. i 2 - R 2
.DELTA. .xi. pm , .0. 2 = - N 1 R c .DELTA. i 1 + N 2 ( R 1 + R c )
.DELTA. i 2 - R 1 .DELTA. .xi. pm . ( 6 ) ##EQU00005##
[0071] Where:
A=R.sub.1R.sub.2+R.sub.1R.sub.c+R.sub.2R.sub.c (7)
[0072] The differential of equation (6) is
{ v 1 = N 1 .0. 1 t = N 1 2 ( R 2 R c ) .DELTA. i 1 t - N 1 N 2 R c
.DELTA. i 2 t , v 2 = N 2 .0. 2 t = - N 1 N 2 R c .DELTA. i 1 t + N
2 2 ( R 1 + R c ) .DELTA. i 2 t . ( 8 ) ##EQU00006##
[0073] The self-inductance, mutual inductance and coupling factor
may be found. The relationship between the inductance and the
magnetic reluctances are shown in (9).
{ L s 1 = N 1 2 ( R 2 + R c ) .DELTA. , L s 2 = N 2 2 ( R 1 + R c )
.DELTA. , M = - N 1 N 2 R c .DELTA. . ( 9 ) ##EQU00007##
Where L.sub.5 is self-inductance and M is mutual inductance of the
PMCI. Only symmetrical structure is further discussed for
simplification. That is:
{ N 1 = N 2 = N , R 1 = R 2 = R , L s 1 = L s 2 = L s . ( 10 )
##EQU00008##
[0074] Equation (9) can then be simplified as
{ L s = N 2 ( R + R c ) R ( R + 2 R c ) , M = - N 2 R c R ( R + 2 R
c ) , .alpha. = M L s = - R c R + R c . ( 11 ) ##EQU00009##
[0075] From equation (11), equation (12) is obtained, which gives
the PMCI magnetic design equations. For a given self-inductance
L.sub.s and a coupling factor .alpha., the magnetic reluctances of
the outer and center legs are given by (12). The major reluctances
of inductor cores are in the air gaps. Thus, the thickness of the
air gaps may be.
{ R = 1 1 - .alpha. N 2 L s , R c = - .alpha. 1 - .alpha. 2 N 2 L s
. ( 12 ) ##EQU00010##
[0076] It can be observed even though equations (5) and (6) account
for the PM effect (by .xi..sub.pm and R.sub.c). This indicates that
the utilization of PM does not have significant influence on the
inductances, coupling factor and core structure of the coupled
power inductor. This will be discussed further below.
[0077] ANSYS.RTM./Maxwell.RTM. 3-D Modeling and Simulation of
PMCI
[0078] The main objective of this section is to evaluate a PMCI
design, as shown in FIG. 10, where the coupling factor is -1/3 and
the self-inductance is about 480 nH. The measured inductance values
obtained from ANSYS.RTM./Maxwell physical simulation are shown in
Table IV, as compared to a reference coupled power inductor
(Original CI) design. The core size is 18 mm.times.10 mm.times.6.5
mm with the air gap length of 0.24 mm in each of the three
legs.
[0079] With the utilization of a PM, a coupled power inductor with
a smaller core is designed (New CI in Table IV) and modeled below.
The permanent magnet piece is placed in the central leg gap. FIG.
14 illustrates a diagram of a permanent magnet coupled power
inductor 500. The diagram of the PMCI is shown in FIG. 14 and the
design specifications are given in Table III. In Table III, d is
the depth of the core into the page, A.sub.w is the cross section
area of each copper winding, g is the air gap length in each of the
three legs, T.sub.pm and T.sub.core denotes material type of
permanent magnet and inductor core respectively, I.sub.pm is the
thickness of the PM, A.sub.pm is the cross section area of the PM,
and N denotes the winding number of turns for each of the side
legs.
[0080] FIGS. 15A-15A show ANSYS.RTM./Maxwell.RTM. 3-D physical
model of the designed PMCI according to the design specifications
in Table III. In particular, FIGS. 15A-15D show
ANSYS.RTM./Maxwell.RTM. 3-D model of the designed PMCI: (a) front
view (b) top view (c) E core and PM piece (d) 3-D view. The core
material used is 3F3 (which starts to saturate when the flux
density gets 0.35 T) and the permanent magnet material is
NdFeB-N38SH. The core size of the designed PMCI is 14.5 mm.times.8
mm.times.5.2 mm with one turn in each side leg.
TABLE-US-00003 TABLE III Parameter Values Parameter Value Units d 8
mm W.sub.e 1.6 mm W.sub.s 4 mm W.sub.c 3.2 mm W.sub.b 1.6 mm
D.sub.s 1.6 mm W.sub.t 1.92 mm A.sub.w 3.9 mm.sup.2 g 0.044 mm N 1
-- T.sub.core 3F3 -- T.sub.pm NdFeB - N38SH -- l.sub.pm g/2 mm
A.sub.pm W.sub.c .times. d mm.sup.2
[0081] In the inverse coupling case, without PM, the new designed
coupled power inductor (New CI) approximately starts to saturate
when the DC input current in each winding is 13 A. The inductance
measured from ANSYS.RTM./Maxwell.RTM. for the New CI is shown in
Table IV.
[0082] The designed PMCI is obtained by placing a PM in the center
leg of the NEW CI design. The inductance value measured in
ANSYS.RTM./Maxwell.RTM. is shown in Table IV. Flux linkages plots
versus the DC input current are shown in FIG. 16 for the PMCI's
winding1 and winding2. FIG. 16 illustrates a plot 1600 of the flux
linkage of winding1 1602 and winding2 1604 of the designed PMCI
when input current increase from zero to 28 A. It can be observed
that the flux linkages of the two windings first decrease to zero
when the input current is about 14 A, and then change the direction
and increase.
TABLE-US-00004 TABLE IV Power Inductor Comparison Original CI New
CI PMCI Core dimensions 18 .times. 10 .times. 14.5 .times. 8
.times. 5.2 = 14.5 .times. 8 .times. 5.2 = (mm.sup.3) 6.5 = 1170
603 603 Self-inductance 477 480 446 (nH) Mutual inductance -158
-162 -158 (nH) Coupling factor -0.33 -0.34 -0.35 Inductance density
0.41 0.80 0.75 (nH/mm.sup.3) Current density 6.68 6.41 6.41 under
25 A input (A/mm.sup.2) Saturation current 38 13 28 (A) per
winding
[0083] The changing tendency of B field is the same as the flux
linkages. When the input current is zero, the average flux density
is smaller than saturation flux density of the inductor core
material, which means that the inductor core will not be saturated
by the PM itself. It can be observed that the PMCI is about to
saturate when the input current is 28 A. This means that the
saturation current is higher by 15 A (more than doubled) compared
with the same core inversely coupled power inductor without
permanent magnet (the New CI), while keeping the same inductance
values. Detailed comparison is shown in Table IV.
[0084] Table V, below, demonstrates an alternative PMCI design
specification. Table VI compares the Original CPI, the New CPI and
the PMCI.
TABLE-US-00005 TABLE V The PMCI Design Specifications Parameter
Value Units Descriptions L.sub.e 18 mm Length of the core D.sub.c 5
mm Depth of the core H.sub.e 4 mm Height of the E core H.sub.w 2 mm
Height of the core window W.sub.c 4 mm Width of the central leg
W.sub.s 2 mm Width of the side leg L.sub.g 115 .mu.m Length of the
core gap L.sub.pm 47 .mu.m Thickness of the PM D.sub.pm D.sub.c mm
Depth of the PM W.sub.pm W.sub.c mm Width of the PM H.sub.i 2.4 mm
Height of the I core L.sub.PCB 0.5 mm Thickness of the PCB
TH.sub.cu 0.75 mm Thickness of the winding W.sub.cu 5 mm Width of
the winding N 2 -- Number of winding turns T.sub.core 3F3 --
Material type of the core T.sub.pm NdFeB - -- Material type of the
PM N38SH T.sub.PCB FR - 4 -- Material type of the PCB
TABLE-US-00006 TABLE VI Comparison of The Original CPI, The New CPI
and The PMCI Original CPI New CPI PMCI Total inductor dimensions 28
.times. 20 .times. 6.46 = 28 .times. 10 .times. 6.43 = 28 .times.
10 .times. 6.43 = with windings (mm.sup.3) 3618 1800 1800 Core
dimensions (mm.sup.3) 18 .times. 10 .times. 6.46 = 18 .times. 5
.times. 6.43 = 18 .times. 5 .times. 6.43 = 1163 579 579 Winding
turns/phase 2 2 2 Length of core gaps 245 .mu.m 115 .mu.m 115 .mu.m
Permanent Magnet -- -- 47 .mu.m thick L.sub.s/Phase (nH)* 478 444
481 M (nH)* -163 -144 -162 .alpha. = M/L.sub.s * -0.34 -0.34 -0.34
Inductance density 0.82 1.53 1.66 2L.sub.s/Core volume
(nH/mm.sup.3)* I.sub.sat (A)/phase* 30 15 30 Current density/phase
@ I = 8 4 8 I.sub.sat (A/mm.sup.2)*
[0085] FIGS. 17A-17E illustrate the flux density (B) vector changes
when the DC input current increases from zero to 25 A. The
magnitude of the B vector changes agree with the B field changes
and the direction changes agree with the direction changes of flux
linkages in the two windings as plotted in FIG. 16.
[0086] One of the design constrains of the power inductor is the
current density in the windings. The maximum current density value
is less than 9.3 A/mm.sup.2, which is acceptable for the PCB
winding. Moreover, a PMCI design should be such that the PM is
never demagnetized under the maximum input current. Demagnetization
of NdFeB-N38SH permanent magnet material occurs when a field
intensity (H), applied in the direction of demagnetization, is
larger than 12.75.times.10.sup.5 A/m at 20.degree. C.
[0087] The analysis of ANSYS.RTM./Maxwell.RTM. simulation results
above including inductance value, flux density (B) field, current
density (J) and field intensity (H) field verify the effectiveness
of the designed PMCI. Comparisons between the conventional coupled
power inductor and PMCI show the advantages of PMCI in increasing
the saturation current, reducing the power inductor core size and
increasing the inductance density.
[0088] Simplorer.RTM. and ANSYS.RTM./Maxwell.RTM. Co-simulation
Model and Results
[0089] Based on a two-phase DC-DC buck power converter with 50 A
load current (25 A in each phase), 5V input voltage and 1.5V output
voltage, the DC-DC buck converter operation waveforms with the PMCI
are obtained by using Simplorer.RTM. and ANSYS.RTM./Maxwell.RTM.
co-simulation model. The co-simulation model 1800 of the two-phase
buck power converter with the PMCI is shown in FIG. 18. The
ANSYS.RTM./Maxwell.RTM. physical simulation model of the PMCI is
used in the circuit model of the power converter in Simplorer.RTM..
In this co-simulation model, the switching frequency is 500 kHz,
output capacitor is 1 mF, load resistance is 0.03.OMEGA. (resulting
in 50 A total load current) and the duty cycle is 0.3
(D=1.5V/5V=0.3).
[0090] Using the information of the waveforms obtained from the
co-simulation model in equation (13), the steady-state inductance
can be obtained and is equal to 458 nH. This inductance affects the
steady-state performance (output voltage ripple and power
efficiency) of the power converter. Usually, the larger it is, the
better.
L w = ( V i n - V Trans - V out ) T on .DELTA. i = 3.5 .times. 6
.times. 10 - 7 4.5891 = 458 nH ( 13 ) ##EQU00011##
[0091] For the two-phase DC-DC buck converter with coupled power
inductor, the equivalent steady state inductance could also be
calculated from equation (14).
L ss = 1 - .alpha. 2 1 + D + .alpha. 1 - D L s = 1 - ( - 0.35 ) 2 1
+ 0.3 * ( - 0.35 ) 1 - 0.3 .times. 446 = 460 nH ( 14 )
##EQU00012##
Where .alpha. is coupling factor, D is duty cycle and L.sub.ss is
self-inductance.
[0092] From equation (14) and PMCI parameters in Table IV, the
steady-state equivalent inductance can be calculated as 460 nH,
which very closely agrees with the result obtained from the
co-simulation waveforms (458 nH).
[0093] The transient inductance could be calculated from equation
(15) and is found to be 290 nH.
L.sub.tr=(1+.alpha.)L.sub.s=(1-0.35).times.446=290 nH (15)
[0094] This equivalent transient inductance affects the dynamic
performance (output voltage deviation/overshoot/undershoot during
dynamic transients) of the power converter. Usually, the smaller it
is, the better.
[0095] Thus, as described above, a 25 A per phase, 14.5 mm.times.8
mm.times.5.2 mm two-phase PMCI which has .sup..about.460 nH
equivalent steady state inductance and .sup..about.290 nH
equivalent transient inductance is presented. The design and
simulation results based on a two-phase DC-DC buck power converter
with 50 A load current, 5V input voltage and 1.5V output voltage
show the effectiveness of designed PMCI. By using Simplorer.RTM.
and ANSYS.RTM./Maxwell.RTM. co-simulation model, the DC-DC buck
converter operation waveforms with the PMCI may be obtained. The
results show that the presented PMCI is able to increase the
saturation current by about 115% for the same size and inductance
value of the conventional coupled power inductor, or approximately
double the inductance density while maintaining high saturation
current compared with the conventional coupled power inductor.
[0096] Introduction to Permanent Magnet Toroid Power Inductor with
Increased Saturation Current
[0097] A toroid is a power inductor core structure which offers
high magnetic efficiency due to the uniformity of its
cross-sectional area. A gapped ferrite toroid core may be employed
to increase the saturation current by reducing the effective
permeability of the flux path. A permanent magnet toroid power
inductor (PMTPI) that utilizes a permanent magnet (PM) is described
to further increase the saturation current of the conventional
gapped toroid power inductor (TPI).
[0098] Structure and Operation Principle of the PMTPI
[0099] A PMTPI utilizes a PM to partially offset the flux in the
toroid magnetic core as a result of the winding current, such that
a higher saturation current could be obtained. The front view of a
TPI 1900 and a PMTPI 1902 diagrams are illustrated in FIGS. 19A and
19B, respectively. In the PMTPI structure, a winding is wound
around the toroid ferrite core and a small piece of PM is placed in
the gap. The DC input current direction and the PM polarity are
shown in FIGS. 19A-19B. In this diagram, the flux path generated by
winding is represented by the dotted arrowed (orange) line, while
the flux path generated by the PM is denoted by the dashed arrowed
(green) line. It can be observed from FIG. 19B that the two flux
loops are in opposite directions. Thus, the winding flux could
partially be canceled by the PM flux. This flux cancellation effect
helps in increasing the saturation current of the PMTPI.
[0100] Inductance (L) and saturation current (I.sub.sat) of TPI can
be described by equations (16) and (17), respectively:
L = N 2 .mu. e .mu. 0 A e l e ( 16 ) I sat = N B sat A e L ( 17 )
##EQU00013##
Where N is the number of winding turns, .mu..sub.e is the effective
permeability of the flux path, .mu..sub.0=4.pi.10.sup.-7 H/m is the
vacuum permeability, A, is the effective cross section area of the
toroid core, l.sub.e is the length of the flux path, and B.sub.sat
is the saturation flux density of the core material. Equation (16)
could also be used to calculate the inductance value of the
PMTPI.
[0101] Due to the cancellation effect of the PM magnetic flux, the
saturation current of the PMTPI can be described as equation
(18).
I.sub.sat.sub._.sub.PMTPI=I.sub.sat+I.sub.c (18)
Where, I.sub.c denotes the current at which net flux density of the
winding and the PM becomes zero in the PMTPI core. Accordingly, the
higher the cancellation current, the larger PMTPI saturation
current will be.
[0102] ANSYS.RTM./Maxwell.RTM. 3-D Modeling and Simulation Results
of the PMTPI
[0103] The PMTPI design is also compared with a conventional TPI
design with the same structure, size and inductance value.
[0104] Physical Model
[0105] The diagram of the PMTPI design with specifications is shown
in FIG. 20 and the corresponding parameter values are listed in
Table V. The design specifications include material and dimensions
of core, winding, insulator and PM.
TABLE-US-00007 TABLE V THE PMTPI DESIGN SPECIFICATIONS Parameter
Value Units Description OD 10 mm Outer diameter of core ID 6 mm
Inner diameter of core H 4 mm Height of core THg 0.3 mm Length of
gap THpm 0.08 mm Thickness of PM Wpm (OD - ID)/2 mm Width of PM Hpm
4 mm Height of PM Hi 4 mm Height of insulator Wi (OD - ID)/2 mm
Width of insulator THi THg - THpm mm Thickness of insulator N 2.75
-- Winding turns Dw 2.5 mm Diameter of copper wire Tcore 3C20 --
Material type of core Ti FR - 4 -- Material type of insulator Tpm
NdFeB - N45SH -- Material type of PM
[0106] FIG. 21 shows ANSYS.RTM./Maxwell.RTM. 3-D physical model of
the designed PMTPI according to the design specifications in FIG.
20 and Table V. In FIG. 21, (a) is a 3-D view, (b) is a front view,
(c) is a top view and (d) is a cross sectional view of the toroid
core along A-A' as in part (a) The PMTPI design uses the gapped
ferrite toroid TN10/6/4 as the inductor core. The core material
used is 3C20 (B.sub.sat is 0.47 T) and the PM material is
NdFeB-N45SH (Residual flux density Br=1.32 T, coercivity Hc=1003
kA/m and intrinsic coercivity Hci=1590 kA/m).
[0107] For comparison purposes, an ANSYS.RTM./Maxwell.RTM. 3-D
physical model of a conventional TPI (without PM) but having the
same design specifications as shown in Table V is also developed.
Results comparison between TPI and PMTPI is shown in Table VI. More
detailed descriptions for the ANSYS.RTM./Maxwell.RTM. simulation
results are given next.
TABLE-US-00008 TABLE VI Comparison Between TPI and PMTPI Based on
the ANSYS .RTM./Maxwell .RTM. 3-D Simulation TPI PMTPI Total
inductor 12.5 mm .times. 12.5 mm .times. 12.5 mm .times. 12.5 mm
.times. dimensions 6.5 mm 6.5 mm Core dimensions 10 mm/6 mm/4 mm 10
mm/6 mm/4 mm OD/ID/H Effective gap Air or Insulator Insulator and
PM 0.3 mm 0.3 mm Number of winding turns 2.75 2.75 Permanent Magnet
NO YES (YES/NO) Inductance (nH) 592 592 Saturation current 14 28
(A)
[0108] Simulation Results for the TPI
[0109] The inductance of the TPI measured from
ANSYS.RTM./Maxwell.RTM. is 592 nH. Results indicate that the
inductor core starts to saturate when the DC input current is 14 A,
i.e. I.sub.sat.sub._.sub.TPI'=14 A.
[0110] Field of the PMTPI
[0111] The inductance of PMTPI as measured from
ANSYS.RTM./Maxwell.RTM. is 592 nH. It can be observed that when the
input current is 0, the average net B value is less than B.sub.sat
of the inductor core material (0.47 T). This indicates that the
magnetic core is not saturated by PM itself. It could also be
observed that when the input DC current increases from 0 to 30 A,
the net B value first decreases to zero at 14 A, then increases to
B.sub.sat at 28 A. Thus the cancellation current of this PMTPI
design is 14 A. When the DC input current is 14 A, the fluxes of
winding and PM have the same values but in opposite directions,
which makes the net flux inside of the PMTPI core equals to zero.
It could be predicted from equation (5) that
I.sub.sat.sub._.sub.PMTPI=14+14=28 A. Observations indicate that
the PMTPI starts to saturate at 28 A. Simulation results show that
the saturation current in the PMTPI is twice of the saturation
current in the TPI with the same size and inductance value.
[0112] It can be observed that the net B vector changes when the DC
input current increases from zero to 28 A. The magnitude changes
are consistent with B field changes, and that the vector direction
becomes opposite when the DC input current increases from less than
the cancellation current (14 A) to higher than the cancellation
current.
[0113] Demagnetizing Field (H) of the PM
[0114] Demagnetization of NdFeB-N45SH PM material occurs when a
reverse field (H) larger than 12.97.times.10.sup.5 A/m (at
25.degree. C.) is applied to the PM. PMTPI design has to ensure
that the PM is never demagnetized under the maximum input current.
It can be observed from that the maximum H value is
11.33.times.10.sup.5 A/m. This indicates that the PM used in the
PMTPI design will not be demagnetized for an input current is as
high as 30 A.
[0115] The physical model simulation results of the permanent
magnet toroid power inductor (PMTPI) showed that the saturation
current can be doubled with the same size and inductance. The PMTPI
achieves these results with its relatively simple power inductor
structure and design.
[0116] The toroid core with permanent magnet can have one or more
windings that could be coupled or not coupled.
It should be emphasized that the above-described implementations
are merely possible examples of implementations set forth for a
clear understanding of the principles of this disclosure. Many
variations and modifications may be made to the above-described
implementations without departing substantially from the spirit and
principles of the disclosure. All such modifications and variations
are intended to be included herein within the scope of this
disclosure.
* * * * *