U.S. patent application number 15/347452 was filed with the patent office on 2017-03-30 for tunable compensation circuit for filter circuitry using acoustic resonators.
The applicant listed for this patent is Qorvo US, Inc.. Invention is credited to Robert Aigner, Nadim Khlat.
Application Number | 20170093370 15/347452 |
Document ID | / |
Family ID | 58406954 |
Filed Date | 2017-03-30 |
United States Patent
Application |
20170093370 |
Kind Code |
A1 |
Khlat; Nadim ; et
al. |
March 30, 2017 |
TUNABLE COMPENSATION CIRCUIT FOR FILTER CIRCUITRY USING ACOUSTIC
RESONATORS
Abstract
Tunable filter circuitry includes a series acoustic resonator
between first and second nodes and a compensation circuit in
parallel with the series acoustic resonator. The compensation
circuit includes first and second inductors coupled in series
between the first node and the second node, wherein the first
inductor and the second inductor are negatively coupled with one
another and a common node is provided between the first and second
inductors. The compensation circuit also includes first and second
shunt acoustic resonators, which are coupled in parallel with one
another between the common node and a fixed voltage node. A first
variable capacitor is also coupled between the common node and the
fixed voltage node, wherein changing a capacitance of the first
variable capacitor changes a bandwidth of a passband of the filter
circuitry.
Inventors: |
Khlat; Nadim; (Cugnaux,
FR) ; Aigner; Robert; (Ocoee, FL) |
|
Applicant: |
Name |
City |
State |
Country |
Type |
Qorvo US, Inc. |
Greensboro |
NC |
US |
|
|
Family ID: |
58406954 |
Appl. No.: |
15/347452 |
Filed: |
November 9, 2016 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
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15275957 |
Sep 26, 2016 |
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15347452 |
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62317675 |
Apr 4, 2016 |
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62232746 |
Sep 25, 2015 |
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Current U.S.
Class: |
1/1 |
Current CPC
Class: |
H03H 9/0004 20130101;
H03H 2210/015 20130101; H03H 9/542 20130101; H03H 9/02086 20130101;
H03H 9/02433 20130101; H03H 9/6406 20130101; H03H 2210/025
20130101; H03H 9/6489 20130101; H03H 9/6483 20130101; H03H 9/605
20130101; H03H 7/09 20130101 |
International
Class: |
H03H 9/54 20060101
H03H009/54; H03H 9/64 20060101 H03H009/64 |
Claims
1. Filter circuitry comprising: a first node and a second node; at
least one series acoustic resonator coupled between the first node
and the second node, wherein at least one main series resonance is
provided between the first node and the second node at a main
resonance frequency through the at least one series acoustic
resonator; and a compensation circuit comprising: a first inductor
and a second inductor coupled in series between the first node and
the second node, wherein the first inductor and the second inductor
are negatively coupled with one another and a common node is
provided between the first inductor and the second inductor; a
first shunt acoustic resonator coupled between the common node and
a fixed voltage node; a second shunt acoustic resonator coupled
between the common node and the fixed voltage node, wherein a first
series resonance at a first resonance frequency and a second series
resonance at a second resonance frequency, which is different from
the first resonance frequency and main resonance frequency, are
provided between the first node and the second node through the
compensation circuit; and a first variable capacitor coupled
between the common node and the fixed voltage node, wherein
changing a capacitance of the first variable capacitor changes a
bandwidth of a passband of the filter circuitry.
2. The filter circuitry of claim 1 further comprising a second
variable capacitor coupled in parallel with the at least one series
acoustic resonator, wherein changing a capacitance of the second
variable capacitor changes the bandwidth of the passband of the
filter circuitry.
3. The filter circuitry of claim 2 further comprising bias
circuitry configured to apply at least one bias signal to at least
one of the first shunt acoustic resonator, the second shunt
acoustic resonator, and the first variable capacitor, as well as to
at least one of the second variable capacitor and the at least one
series acoustic resonator, wherein changing the at least one bias
signal changes a bandwidth of a passband of the filter
circuitry.
4. The filter circuitry of claim 1 further comprising bias
circuitry configured to apply a bias signal to at least one of the
first shunt acoustic resonator, the second shunt acoustic
resonator, and the first variable capacitor, wherein changing the
bias signal changes the bandwidth of the passband of the filter
circuitry.
5. The filter circuitry of claim 1 further comprising a first
switch coupled in series with the second shunt acoustic resonator,
wherein the first switch and the second shunt acoustic resonator
are coupled between the common node and the fixed voltage node.
6. The filter circuitry of claim 5 further comprising a second
variable capacitor coupled in parallel with the at least one series
acoustic resonator, wherein changing a capacitance of the second
variable capacitor changes the bandwidth of the passband of the
filter circuitry.
7. The filter circuitry of claim 6 further comprising bias
circuitry configured to apply at least one bias signal to at least
one of the first shunt acoustic resonator, the second shunt
acoustic resonator, and the first variable capacitor, as well as to
at least one of the second variable capacitor and the at least one
series acoustic resonator, wherein changing the at least one bias
signal changes the bandwidth of the passband of the filter
circuitry.
8. The filter circuitry of claim 1 further comprising bias
circuitry configured to apply a bias signal to at least one of the
first shunt acoustic resonator, the second shunt acoustic
resonator, and the first variable capacitor, wherein changing the
bias signal changes the bandwidth of the passband of the filter
circuitry.
9. The filter circuitry of claim 1 wherein at least one of the
first resonance frequency and the second resonance frequency is
greater than the main resonance frequency.
10. The filter circuitry of claim 1 wherein the first resonance
frequency is less than the main resonance frequency, and the second
resonance frequency is greater than the main resonance
frequency.
11. The filter circuitry of claim 1 wherein the at least one series
acoustic resonator comprises a plurality of acoustic resonators
that are coupled in parallel with one another, and each of the
plurality of acoustic resonators has a different series resonance
frequency.
12. The filter circuitry of claim 1 wherein the compensation
circuit comprises at least one additional shunt acoustic resonator
coupled between the common node and the fixed voltage node.
13. The filter circuitry of claim 1 wherein: an equivalent .pi.
(pi) network of the compensation circuit comprises a series
equivalent impedance between the first node and the second node as
well as two shunt equivalent impedances; and the series equivalent
impedance exhibits negative capacitive behavior throughout multiple
frequency ranges.
14. The filter circuitry of claim 1 wherein: an equivalent .pi.
(pi) network of the compensation circuit comprises a series
equivalent impedance between the first node and the second node and
two shunt equivalent impedances; and the first series resonance at
the first resonance frequency and the second series resonance at
the second resonance frequency are provided through the series
equivalent impedance.
15. The filter circuitry of claim 1 wherein: an equivalent .pi.
(pi) network of the compensation circuit comprises a series
equivalent impedance between the first node and the second node and
two shunt equivalent impedances; the at least one series acoustic
resonator comprises a series impedance having a parallel resonance
at a first parallel resonance frequency; and the series impedance
of the at least one series acoustic resonator in parallel with the
series equivalent impedance of the equivalent .pi. (pi) network of
the compensation circuit provides an overall impedance having a
parallel resonance at a second parallel resonance frequency, which
is greater than the first parallel resonance frequency.
16. The filter circuitry of claim 1 wherein: an equivalent .pi.
(pi) network of the compensation circuit comprises a series
equivalent impedance between the first node and the second node and
two shunt equivalent impedances; the at least one series acoustic
resonator comprises a series impedance having a parallel resonance
at a first parallel resonance frequency; the series impedance of
the at least one series acoustic resonator in parallel with the
series equivalent impedance of the equivalent .pi. (pi) network of
the compensation circuit provides an overall impedance having a
parallel resonance at a second parallel resonance frequency, which
is greater than the first parallel resonance frequency; and the
series equivalent impedance exhibits negative capacitive behavior
throughout multiple frequency ranges.
17. The filter circuitry of claim 16 wherein at least one of the
first resonance frequency and the second resonance frequency is
greater than the main resonance frequency.
18. The filter circuitry of claim 16 wherein the first resonance
frequency is less than the main resonance frequency, and the second
resonance frequency is greater than the main resonance
frequency.
19. The filter circuitry of claim 1 wherein at least one of the
first shunt acoustic resonator and the second shunt acoustic
resonator has a series resonance at a third resonance frequency
wherein the third resonance frequency is greater than the main
resonance frequency.
20. The filter circuitry of claim 1 wherein the first inductor and
the second inductor have different inductances.
21. The filter circuitry of claim 1 wherein the at least one series
acoustic resonator, the first shunt acoustic resonator, and the
second shunt acoustic resonator are at least one of a bulk acoustic
wave (BAW) resonator and a surface acoustic wave (SAW)
resonator.
22. The filter circuitry of claim 1 wherein fc/BW*100 is between
3.5% and 12%, wherein fc is a center frequency of a passband of the
filter circuitry, and BW is a bandwidth of the passband.
23. The filter circuitry of claim 1 wherein a frequency response of
the filter circuitry comprises a plurality of passbands such that
adjacent passbands of the plurality of passbands are separated by a
stop band.
24. Filter circuitry comprising: a first node and a second node; at
least one series acoustic resonator coupled between the first node
and the second node, wherein at least one main series resonance is
provided between the first node and the second node at a main
resonance frequency through the at least one series acoustic
resonator; and a compensation circuit comprising: a first inductor
and a second inductor coupled in series between the first node and
the second node, wherein the first inductor and the second inductor
are negatively coupled with one another and a common node is
provided between the first inductor and the second inductor; a
first shunt acoustic resonator coupled between the common node and
a fixed voltage node; a second shunt acoustic resonator coupled
between the common node and the fixed voltage node; and a first
variable capacitor coupled between the common node and the fixed
voltage node, wherein changing a capacitance of the first variable
capacitor changes a bandwidth of a passband of the filter
circuitry.
25. The filter circuitry of claim 24 further comprising a second
variable capacitor coupled in parallel with the at least one series
acoustic resonator wherein changing a capacitance of the second
variable capacitor changes the bandwidth of the passband of the
filter circuitry.
Description
RELATED APPLICATIONS
[0001] This application claims the benefit of provisional patent
application Ser. No. 62/317,675, filed Apr. 4, 2016, the disclosure
of which is incorporated herein by reference in its entirety.
[0002] This application is a Continuation-in-Part of U.S. utility
patent application Ser. No. 15/275,957, filed Sep. 26, 2016, which
claims the benefit of provisional patent application Ser. No.
62/232,746, filed Sep. 25, 2015; the disclosures of which are
incorporated herein by reference in their entireties.
FIELD OF THE INVENTION
[0003] The present disclosure relates to acoustic resonators and in
particular to a tunable compensation circuit for filter circuitry
using acoustic resonators.
BACKGROUND
[0004] Acoustic resonators, such as Surface Acoustic Wave (SAW)
resonators and Bulk Acoustic Wave (BAW) resonators, are used in
many high-frequency communication applications. In particular, SAW
resonators are often employed in filter networks that operate
frequencies up to 1.8 GHz, and BAW resonators are often employed in
filter networks that operate at frequencies above 1.5 GHz. Such
filters need to have flat passbands, have steep filter skirts and
squared shoulders at the upper and lower ends of the passband, and
provide excellent rejection outside of the passband. SAW- and
BAW-based filters also have relatively low insertion loss, tend to
decrease in size as the frequency of operation increases, and are
relatively stable over wide temperature ranges. As such, SAW- and
BAW-based filters are the filter of choice for many 3rd Generation
(3G) and 4th Generation (4G) wireless devices and are destined to
dominate filter applications for 5th Generation (5G) wireless
devices. Most of these wireless devices support cellular, wireless
fidelity (Wi-Fi), Bluetooth, and/or near field communications on
the same wireless device and, as such, pose extremely challenging
filtering demands. While these demands keep raising the complexity
of wireless devices, there is a constant need to improve the
performance of acoustic resonators and filters that are based
thereon.
[0005] To better understand acoustic resonators and various
terminology associated therewith, the following provides an
overview of a BAW resonator. However, the concepts described herein
may employ any type of acoustic resonator and are not limited to
SAW- and BAW-based resonators. An exemplary BAW resonator 10 is
illustrated in FIG. 1. The BAW resonator 10 generally includes a
substrate 12, a reflector 14 mounted over the substrate 12, and a
transducer 16 mounted over the reflector 14. The transducer 16
rests on the reflector 14 and includes a piezoelectric layer 18,
which is sandwiched between a top electrode 20 and a bottom
electrode 22. The top and bottom electrodes 20 and 22 may be formed
of Tungsten (W), Molybdenum (Mo), Platinum (Pt), or like material,
and the piezoelectric layer 18 may be formed of Aluminum Nitride
(AlN), Zinc Oxide (ZnO), or other appropriate piezoelectric
material. Although shown in FIG. 1 as each including a single
layer, the piezoelectric layer 18, the top electrode 20, and/or the
bottom electrode 22 may include multiple layers of the same
material, multiple layers in which at least two layers are
different materials, or multiple layers in which each layer is a
different material.
[0006] The BAW resonator 10 is divided into an active region 24 and
an outside region 26. The active region 24 generally corresponds to
the section of the BAW resonator 10 where the top and bottom
electrodes 20 and 22 overlap and also includes the layers below the
overlapping top and bottom electrodes 20 and 22. The outside region
26 corresponds to the section of the BAW resonator 10 that
surrounds the active region 24.
[0007] For the BAW resonator 10, applying electrical signals across
the top electrode 20 and the bottom electrode 22 excites acoustic
waves in the piezoelectric layer 18. These acoustic waves primarily
propagate vertically. A primary goal in BAW resonator design is to
confine these vertically propagating acoustic waves in the
transducer 16. Acoustic waves traveling upward are reflected back
into the transducer 16 by the air-metal boundary at the top surface
of the top electrode 20. Acoustic waves traveling downward are
reflected back into the transducer 16 by the reflector 14 or by an
air cavity, which is provided just below the transducer in a Film
BAW Resonator (FBAR).
[0008] The reflector 14 is typically formed by a stack of reflector
layers (RL) 28, which alternate in material composition to produce
a significant reflection coefficient at the junction of adjacent
reflector layers 28. Typically, the reflector layers 28 alternate
between materials having high and low acoustic impedances, such as
tungsten (W) and silicon dioxide (SiO.sub.2). While only five
reflector layers 28 are illustrated in FIG. 1, the number of
reflector layers 28 and the structure of the reflector 14 varies
from one design to another.
[0009] The magnitude (Z) and phase (.phi.) of the electrical
impedance as a function of the frequency for a relatively ideal BAW
resonator 10 is provided in FIG. 2. The magnitude (Z) of the
electrical impedance is illustrated by the solid line, whereas the
phase (.phi.) of the electrical impedance is illustrated by the
dashed line. A unique feature of the BAW resonator 10 is that it
has both a resonance frequency and an anti-resonance frequency. The
resonance frequency is typically referred to as the series
resonance frequency (f.sub.s), the anti-resonance frequency is
typically referred to as the parallel resonance frequency
(f.sub.p). The series resonance frequency (f.sub.s) occurs when the
magnitude of the impedance, or reactance, of the BAW resonator 10
approaches zero. The parallel resonance frequency (f.sub.p) occurs
when the magnitude of the impedance, or reactance, of the BAW
resonator 10 peaks at a significantly high level. In general, the
series resonance frequency (f.sub.s) is a function of the thickness
of the piezoelectric layer 18 and the mass of the bottom and top
electrodes 20 and 22.
[0010] For the phase, the BAW resonator 10 acts like an inductance
that provides a 90.degree. phase shift between the series resonance
frequency (f.sub.s) and the parallel resonance frequency (f.sub.p).
In contrast, the BAW resonator 10 acts like a capacitance that
provides a -90.degree. phase shift below the series resonance
frequency (f.sub.s) and above the parallel resonance frequency
(f.sub.p). The BAW resonator 10 presents a very low, near zero,
resistance at the series resonance frequency (f.sub.s) and a very
high resistance at the parallel resonance frequency (f.sub.p). The
electrical nature of the BAW resonator 10 lends itself to the
realization of a very high Q (quality factor) inductance over a
relatively short range of frequencies, which has proved to be very
beneficial in high-frequency filter networks, especially those
operating at frequencies around 1.8 GHz and above.
[0011] Unfortunately, the phase (.phi.) curve of FIG. 2 is
representative of an ideal phase curve. In reality, approaching
this ideal is challenging. A typical phase curve for the BAW
resonator 10 of FIG. 1 is illustrated in FIG. 3A. Instead of being
a smooth curve, the phase curve of FIG. 3A includes ripple below
the series resonance frequency (f.sub.s), between the series
resonance frequency (f.sub.s) and the parallel resonance frequency
(f.sub.p), and above the parallel resonance frequency (f.sub.p).
The ripple is the result of spurious modes, which are caused by
spurious resonances that occur in corresponding frequencies. While
the vast majority of the acoustic waves in the BAW resonator 10
propagate vertically, various boundary conditions about the
transducer 16 result in the propagation of lateral (horizontal)
acoustic waves, which are referred to as lateral standing waves.
The presence of these lateral standing waves reduces the potential
Q associated with the BAW resonator 10.
[0012] As illustrated in FIG. 4, a border (BO) ring 30 is formed on
or within the top electrode 20 to suppress certain of the spurious
modes. The spurious modes that are suppressed by the BO ring 30 are
those above the series resonance frequency (f.sub.s), as
highlighted by circles A and B in the phase curve of FIG. 3B.
Circle A shows a suppression of the ripple, and thus of the
spurious mode, in the passband of the phase curve, which resides
between the series resonance frequency (f.sub.s) and the parallel
resonance frequency (f.sub.p). Circle B shows suppression of the
ripple, and thus of the spurious modes, above the parallel
resonance frequency (f.sub.p). Notably, the spurious mode in the
upper shoulder of the passband, which is just below the parallel
resonance frequency f.sub.p, and the spurious modes above the
passband are suppressed, as evidenced by the smooth or
substantially ripple free phase curve between the series resonance
frequency (f.sub.s) and the parallel resonance frequency (f.sub.p)
and above the parallel resonance frequency (f.sub.p).
[0013] The BO ring 30 corresponds to a mass loading of the portion
of the top electrode 20 that extends about the periphery of the
active region 24. The BO ring 30 may correspond to a thickened
portion of the top electrode 20 or the application of additional
layers of an appropriate material over the top electrode 20. The
portion of the BAW resonator 10 that includes and resides below the
BO ring 30 is referred to as a BO region 32. Accordingly, the BO
region 32 corresponds to an outer, perimeter portion of the active
region 24 and resides inside of the active region 24.
[0014] While the BO ring 30 is effective at suppressing spurious
modes above the series resonance frequency (f.sub.s), the BO ring
30 has little or no impact on those spurious modes below the series
resonance frequency (f.sub.s), as shown by the ripples in the phase
curve below the series resonance frequency (f.sub.s) in FIG. 3B. A
technique referred to as apodization is often used to suppress the
spurious modes that fall below the series resonance frequency
(f.sub.s).
[0015] Apodization tries to avoid, or at least significantly
reduce, any lateral symmetry in the BAW resonator 10, or at least
in the transducer 16 thereof. The lateral symmetry corresponds to
the footprint of the transducer 16, and avoiding the lateral
symmetry corresponds to avoiding symmetry associated with the sides
of the footprint. For example, one may choose a footprint that
corresponds to a pentagon instead of a square or rectangle.
Avoiding symmetry helps reduce the presence of lateral standing
waves in the transducer 16. Circle C of FIG. 3C illustrates the
effect of apodization in which the spurious modes below the series
resonance frequency (f.sub.s) are suppressed, as evidence by the
smooth or substantially ripple free phase curve below the series
resonance frequency (f.sub.s). Assuming no BO ring 30 is provided,
one can readily see in FIG. 3C that apodization fails to suppress
those spurious modes above the series resonance frequency
(f.sub.s). As such, the typical BAW resonator 10 employs both
apodization and the BO ring 30.
[0016] As noted previously, BAW resonators 10 are often used in
filter networks that operate at high frequencies and require high Q
values. A basic ladder network 40 is illustrated in FIG. 5A. The
ladder network 40 includes two series resonators B.sub.SER and two
shunt resonators B.sub.SH, which are arranged in a traditional
ladder configuration. Typically, the series resonators B.sub.SER
have the same or similar first frequency response, and the shunt
resonators B.sub.SH have the same or similar second frequency
response, which is different from the first frequency response, as
shown in FIG. 5B. In many applications, the shunt resonators
B.sub.SH are detuned versions of the series resonators B.sub.SER.
As a result, the frequency responses for the series resonators
B.sub.SER and the shunt resonators B.sub.SH are generally very
similar, yet shifted relative to one another such that the parallel
resonance frequency (f.sub.p,SH) of the shunt resonators
approximates the series resonance frequency (f.sub.s,SER) of the
series resonators B.sub.SER. Note that the series resonance
frequency (f.sub.s,SH) of the shunt resonators B.sub.SH is less
than the series resonance frequency (f.sub.s,SER) of the series
resonators B.sub.SER. The parallel resonance frequency (f.sub.p,SH)
of the shunt resonators B.sub.SH is less than the parallel
resonance frequency (f.sub.p,SER) of the series resonators
B.sub.SER.
[0017] FIG. 5C is associated with FIG. 5B and illustrates the
response of the ladder network 40. The series resonance frequency
(f.sub.s,SH) of the shunt resonators B.sub.SH corresponds to the
low side of the passband's skirt (phase 2), and the parallel
resonance frequency (f.sub.p,SER) of the series resonators
B.sub.SER corresponds to the high side of the passband's skirt
(phase 4). The substantially aligned series resonance frequency
(f.sub.s,SER) of the series resonators B.sub.SER and the parallel
resonance frequency (f.sub.p,SH) of the shunt resonators B.sub.SH
fall within the passband. FIGS. 6A through 6E provide circuit
equivalents for the five phases of the response of the ladder
network 40. During the first phase (phase 1, FIGS. 5C, 6A), the
ladder network 40 functions to attenuate the input signal. As the
series resonance frequency (f.sub.s,SH) of the shunt resonators
B.sub.SH is approached, the impedance of the shunt resonators
B.sub.SH drops precipitously such that the shunt resonators
B.sub.SH essentially provide a short to ground at the series
resonance frequency (f.sub.s,SH) of the shunt resonators (phase 2,
FIGS. 5C, 6B). At the series resonance frequency (f.sub.s,SH) of
the shunt resonators B.sub.SH (phase 2), the input signal is
essentially blocked from the output of the ladder network 40.
[0018] Between the series resonance frequency (f.sub.s,SH) of the
shunt resonators B.sub.SH and the parallel resonance frequency
(f.sub.p,SER) of the series resonators B.sub.SER, which corresponds
to the passband, the input signal is passed to the output with
relatively little or no attenuation (phase 3, FIGS. 5C, 6C). Within
the passband, the series resonators B.sub.SER present relatively
low impedance, whereas the shunt resonators B.sub.SH present a
relatively high impedance, wherein the combination of the two leads
to a flat passband with steep low- and high-side skirts. As the
parallel resonance frequency (f.sub.p,SER) of the series resonators
B.sub.SER is approached, the impedance of the series resonators
B.sub.SER becomes very high, such that the series resonators
B.sub.SER essentially present themselves as open at the parallel
resonance frequency (f.sub.p,SER) of the series resonators (phase
4, FIGS. 5C, 6D). At the parallel resonance frequency (f.sub.p,SER)
of the series resonators B.sub.SER (phase 4), the input signal is
again essentially blocked from the output of the ladder network 40.
During the final phase (phase 5, FIGS. 5C, 6E), the ladder network
40 functions to attenuate the input signal, in a similar fashion to
that provided in phase 1. As the parallel resonance frequency
(f.sub.p,SER) of the series resonators B.sub.SER is passed, the
impedance of the series resonators B.sub.SER decreases and the
impedance of the shunt resonators B.sub.SH normalizes. Thus, the
ladder network 40 functions to provide a high Q passband between
the series resonance frequency (f.sub.s,SH) of the shunt resonators
B.sub.SH and the parallel resonance frequency (f.sub.p,SER) of the
series resonators B.sub.SER. The ladder network 40 provides
extremely high attenuation at both the series resonance frequency
(f.sub.s,SH) of the shunt resonators B.sub.SH and the parallel
resonance frequency (f.sub.p,SER) of the series resonators. The
ladder network 40 provides good attenuation below the series
resonance frequency (f.sub.s,SH) of the shunt resonators B.sub.SH
and above the parallel resonance frequency (f.sub.p,SER) of the
series resonators B.sub.SER. As noted previously, there is a
constant need to improve the performance of acoustic resonators and
filters that are based thereon.
[0019] Those skilled in the art will appreciate the scope of the
present disclosure and realize additional aspects thereof after
reading the following detailed description of the preferred
embodiments in association with the accompanying drawing
figures.
SUMMARY
[0020] The present disclosure relates to tunable filter circuitry
including a series acoustic resonator between first and second
nodes and a compensation circuit in parallel with the series
acoustic resonator. The compensation circuit includes first and
second inductors coupled in series between the first node and the
second node, wherein the first inductor and the second inductor are
negatively coupled with one another and a common node is provided
between the first and second inductors. The compensation circuit
also includes first and second shunt acoustic resonators, which are
coupled in parallel with one another between the common node and a
fixed voltage node. A first variable capacitor is also coupled
between the common node and the fixed voltage node, wherein
changing a capacitance of the first variable capacitor changes a
bandwidth of a passband of the filter circuitry. A first switch may
be coupled in series with the second shunt acoustic resonator,
wherein the first switch and the second shunt acoustic resonator
are coupled between the common node and the fixed voltage node.
[0021] The compensation circuit may also include a second variable
capacitor, which is coupled in parallel with the at least one
series acoustic resonator, wherein changing a capacitance of the
second variable capacitor further changes the bandwidth of a
passband of the filter circuitry. Bias circuitry may be included to
provide a DC bias to one or more of the first and second shunt
acoustic resonators, the series acoustic resonators, the first
variable capacitor, and the second variable capacitor.
[0022] In one embodiment, a main series resonance is provided
between the first node and the second node at a main resonance
frequency through the series acoustic resonator. First and second
series resonances at first and second resonance frequencies are
provided between the first node and the second node through the
compensation circuit, wherein the first and second resonance
frequencies are different.
BRIEF DESCRIPTION OF THE DRAWING FIGURES
[0023] The accompanying drawing figures incorporated in and forming
a part of this specification illustrate several aspects of the
disclosure and, together with the description, serve to explain the
principles of the disclosure.
[0024] FIG. 1 illustrates a conventional Bulk Acoustic Wave (BAW)
resonator.
[0025] FIG. 2 is a graph of the magnitude and phase of impedance
over frequency responses as a function of frequency for an ideal
BAW resonator.
[0026] FIGS. 3A-3C are graphs of phase responses for various BAW
resonator configurations.
[0027] FIG. 4 illustrates a conventional BAW resonator with a
border ring.
[0028] FIG. 5A is a schematic of a conventional ladder network.
[0029] FIGS. 5B and 5C are graphs of a frequency response for BAW
resonators in the conventional ladder network of FIG. 5A and a
frequency response for the conventional ladder network of FIG.
5A.
[0030] FIGS. 6A-6E are circuit equivalents for the ladder network
of FIG. 5A at the frequency points 1, 2, 3, 4, and 5, which are
identified in FIG. 5C.
[0031] FIG. 7 illustrates an acoustic resonator in parallel with a
compensation circuit, which includes a single shunt acoustic
resonator.
[0032] FIG. 8 is a graph that illustrates exemplary frequency
responses for the acoustic resonator, compensation circuit, and
overall filter circuit of FIG. 7.
[0033] FIG. 9 illustrates an acoustic resonator in parallel with a
compensation circuit, which includes at least two shunt acoustic
resonators, according to a first embodiment.
[0034] FIG. 10 is a graph that illustrates exemplary frequency
responses for the acoustic resonator, compensation circuit, and
overall filter circuit of FIG. 9.
[0035] FIG. 11 is a graph that compares actual frequency responses
of the filter circuits of FIGS. 7 and 9.
[0036] FIG. 12 illustrates a plurality of parallel acoustic
resonators in parallel with a compensation circuit, which includes
at least two shunt acoustic resonators, according to a second
embodiment.
[0037] FIG. 13 is a graph that illustrates first exemplary
frequency responses for the acoustic resonator, compensation
circuit, and overall filter circuit of FIG. 12.
[0038] FIG. 14 is a graph that illustrates second exemplary
frequency responses for the acoustic resonator, compensation
circuit, and overall filter circuit of FIG. 12.
[0039] FIGS. 15A through 15D illustrate transformation of the
T-circuit impedance architecture of the compensation circuit of
FIG. 9 to a .pi. (pi) impedance model.
[0040] FIG. 16 illustrates the filter circuit of FIG. 9 using the
.pi. (pi) impedance model of FIG. 15D.
[0041] FIG. 17 is a graph illustrating the overall shunt impedance,
Zres, according to one embodiment.
[0042] FIG. 18 is a graph illustrating the series equivalent
impedance, ZA, according to one embodiment.
[0043] FIGS. 19A and 19B are graphs over different frequency ranges
illustrating the absolute or magnitude of series impedance ZS, the
series equivalent impedance ZA, and overall series impedance ZAs,
according to one embodiment.
[0044] FIG. 20 illustrates an acoustic resonator in parallel with a
compensation circuit including at least two shunt acoustic
resonators, according to a third embodiment.
[0045] FIG. 21 illustrates a series resonant inductor-capacitor
(L-C) circuit in parallel with a compensation circuit including at
least two shunt acoustic resonators, according to a fourth
embodiment.
[0046] FIG. 22 illustrates an acoustic resonator in parallel with a
compensation circuit including at least two shunt acoustic
resonators, according to a fifth embodiment.
[0047] FIG. 23 illustrates an acoustic resonator in parallel with a
compensation circuit including at least two shunt acoustic
resonators, according to a sixth embodiment.
[0048] FIG. 24 illustrates an acoustic resonator in parallel with a
compensation circuit including at least two shunt acoustic
resonators, according to a seventh embodiment.
[0049] FIG. 25 illustrates two series acoustic resonators in
parallel with a compensation circuit including at least two shunt
acoustic resonators, according to an eighth embodiment.
[0050] FIG. 26 illustrates two series acoustic resonators in
parallel with a compensation circuit including at least two shunt
acoustic resonators, according to a ninth embodiment.
[0051] FIG. 27 illustrates a communication circuit that is
configured to provide a tunable passband for the filter circuit,
according to a tenth embodiment.
[0052] FIG. 28 is a graph illustrating a variable frequency
response and the associated return loss for the embodiment of FIG.
27.
[0053] FIG. 29 illustrates a communication circuit that is
configured to provide a tunable passband for the filter circuit,
according to an eleventh embodiment.
[0054] FIG. 30 illustrates a communication circuit that is
configured to provide a tunable passband for the filter circuit,
according to a twelfth embodiment.
[0055] FIGS. 31 and 32 are graphs illustrating a variable frequency
response and the associated return loss for a first example of the
embodiment of FIG. 30.
[0056] FIGS. 33 and 34 are graphs illustrating a variable frequency
response and the associated return loss for a second example of the
embodiment of FIG. 30.
[0057] FIG. 35 illustrates a communication circuit that is
configured to provide a tunable passband for the filter circuit,
according to a thirteenth embodiment.
[0058] FIG. 36 illustrates a communication circuit that is
configured to provide a tunable passband for the filter circuit,
according to a fourteenth embodiment.
DETAILED DESCRIPTION
[0059] The embodiments set forth below represent the necessary
information to enable those skilled in the art to practice the
embodiments and illustrate the best mode of practicing the
embodiments. Upon reading the following description in light of the
accompanying drawing figures, those skilled in the art will
understand the concepts of the disclosure and will recognize
applications of these concepts not particularly addressed herein.
It should be understood that these concepts and applications fall
within the scope of the disclosure and the accompanying claims.
[0060] It will be understood that, although the terms first,
second, etc. may be used herein to describe various elements, these
elements should not be limited by these terms. These terms are only
used to distinguish one element from another. For example, a first
element could be termed a second element, and similarly, a second
element could be termed a first element, without departing from the
scope of the present disclosure. As used herein, the term "and/or"
includes any and all combinations of one or more of the associated
listed items.
[0061] It will be understood that when an element such as a layer,
region, or substrate is referred to as being "on" or extending
"onto" another element, it can be directly on or extend directly
onto the other element or intervening elements may also be present.
In contrast, when an element is referred to as being "directly on"
or extending "directly onto" another element, there are no
intervening elements present. Likewise, it will be understood that
when an element such as a layer, region, or substrate is referred
to as being "over" or extending "over" another element, it can be
directly over or extend directly over the other element or
intervening elements may also be present. In contrast, when an
element is referred to as being "directly over" or extending
"directly over" another element, there are no intervening elements
present. It will also be understood that when an element is
referred to as being "connected" or "coupled" to another element,
it can be directly connected or coupled to the other element or
intervening elements may be present. In contrast, when an element
is referred to as being "directly connected" or "directly coupled"
to another element, there are no intervening elements present.
[0062] Relative terms such as "below" or "above" or "upper" or
"lower" or "horizontal" or "vertical" may be used herein to
describe a relationship of one element, layer, or region to another
element, layer, or region as illustrated in the figures. It will be
understood that these terms and those discussed previously are
intended to encompass different orientations of the device in
addition to the orientation depicted in the figures.
[0063] The terminology used herein is for the purpose of describing
particular embodiments only and is not intended to be limiting of
the disclosure. As used herein, the singular forms "a," "an," and
"the" are intended to include the plural forms as well, unless the
context clearly indicates otherwise. It will be further understood
that the terms "comprises," "comprising," "includes," and/or
"including" when used herein specify the presence of stated
features, integers, steps, operations, elements, and/or components
but do not preclude the presence or addition of one or more other
features, integers, steps, operations, elements, components, and/or
groups thereof.
[0064] Unless otherwise defined, all terms (including technical and
scientific terms) used herein have the same meaning as commonly
understood by one of ordinary skill in the art to which this
disclosure belongs. It will be further understood that terms used
herein should be interpreted as having a meaning that is consistent
with their meaning in the context of this specification and the
relevant art and will not be interpreted in an idealized or overly
formal sense unless expressly so defined herein.
[0065] The present disclosure relates to tunable filter circuitry
including a series acoustic resonator between first and second
nodes and a compensation circuit in parallel with the series
acoustic resonator. The compensation circuit includes first and
second inductors coupled in series between the first node and the
second node, wherein the first inductor and the second inductor are
negatively coupled with one another and a common node is provided
between the first and second inductors. The compensation circuit
also includes first and second shunt acoustic resonators, which are
coupled in parallel with one another between the common node and a
fixed voltage node. A first variable capacitor is also coupled
between the common node and the fixed voltage node, wherein
changing a capacitance of the first variable capacitor changes a
bandwidth of a passband of the filter circuitry. A first switch may
be coupled in series with the second shunt acoustic resonator,
wherein the first switch and the second shunt acoustic resonator
are coupled between the common node and the fixed voltage node.
[0066] The compensation circuit may also include a second variable
capacitor, which is coupled in parallel with the at least one
series acoustic resonator wherein changing a capacitance of the
second variable capacitor further changes the bandwidth of a
passband of the filter circuitry. Bias circuitry may be provided to
provide a DC bias to one or more of the first and second shunt
acoustic resonators, the series acoustic resonators, the first
variable capacitor, and the second variable capacitor.
[0067] In various embodiments, the compensation circuit provides
three primary functions. The first is to provide a negative
capacitive behavior, such that a negative capacitance is presented
in parallel with the at least one series acoustic resonator. As
such, the effective capacitance of the at least one series acoustic
resonator is reduced, which functions to shift the parallel
resonance frequency f.sub.p higher. The second function is to add
one or more additional series resonances between the first and
second nodes. The combination of shifting the parallel resonance
frequency f.sub.p higher and adding additional series resonances
through the compensation circuit allows for passbands of greater
bandwidth while maintaining excellent out-of-band rejection. The
third function is to dynamically control the bandwidth of the
passbands. Details are provided below.
[0068] Turning now to FIG. 7, a series resonator B1 is shown
coupled between an input node I/P and an output node O/P. The
series resonator B1 has a series resonance frequency f.sub.s and
inherent capacitance, which generally limits the bandwidth of
filters that employ the series resonator B1. In the case of a Bulk
Acoustic Wave (BAW) resonator, the capacitance of the series
resonator B1 is primarily caused by its inherent structure, which
looks and acts like a capacitor, in part because the series
resonator includes the top and bottom electrodes 20, 22 (FIG. 1)
that are separated by a dielectric piezoelectric layer 18. While
BAW resonators are the focus of the example, other types of
acoustic resonators, such as Surface Acoustic Wave (SAW)
resonators, are equally applicable.
[0069] A compensation circuit 42 is coupled in parallel with the
series resonator B1 and functions to compensate for some of the
capacitance presented by the series resonator B1. The compensation
circuit 42 includes two negatively coupled inductors L1, L2 and a
shunt resonator B2. The inductors L1, L2 are coupled in series
between the input node VP and the output node O/P, wherein a common
node CN is provided between the inductors L1, L2. The inductors L1,
L2 are magnetically coupled by a coupling factor K, wherein the
dots illustrated in association with the inductors L1, L2 indicate
that the magnetic coupling is negative. As such, the inductors L1,
L2 are connected in electrical series and negatively coupled from a
magnetic coupling perspective. As defined herein, two (or more)
series-connected inductors that are negatively coupled from a
magnetic perspective are inductors that are: [0070] connected in
electrical series; and [0071] the mutual inductance between the two
inductors functions to decrease the total inductance of the two (or
more) inductors. The shunt resonator B2 is coupled between the
common node CN and ground, or other fixed voltage node.
[0072] To compensate for at least some of the capacitance of the
series resonator B1, the compensation circuit 42 presents itself as
a negative capacitance within certain frequency ranges, when
coupled in parallel with the series resonator B1. Since
capacitances in parallel are additive, providing a negative
capacitance in parallel with the (positive) capacitance of the
series resonator B1 effectively reduces the capacitance of the
series resonator B1. With the compensation circuit 42, the series
resonator B1 can actually function as a filter (instead of just a
resonator) and provide a passband, albeit a fairly narrow passband,
instead of a more traditional resonator response (solid line of
FIG. 2). FIG. 8 graphically illustrates the frequency responses of
the series resonator B1 (inside the block referenced B1), the
compensation circuit 42 (inside the block referenced 42), and the
filter circuit in which the compensation circuit 42 is placed in
parallel with the series resonator B1. As illustrated, the filter
circuit provides a relatively narrow passband. Further detail on
this particular circuit topology can be found in the co-assigned
U.S. patent application Ser. No. 15/004,084, filed Jan. 22, 2016,
and titled RF LADDER FILTER WITH SIMPLIFIED ACOUSTIC RF RESONATOR
PARALLEL CAPACITANCE COMPENSATION, and U.S. patent application Ser.
No. 14/757,651, filed Dec. 23, 2015, and titled SIMPLIFIED ACOUSTIC
RF RESONATOR PARALLEL CAPACITANCE COMPENSATION, which are
incorporated herein by reference.
[0073] While beneficial in many applications, the narrow passband
of the circuit topology of FIG. 7 has its limitations. With the
challenges of modern day communication systems, wider passbands and
the ability to provide multiple passbands within a given system are
needed. Fortunately, Applicants have discovered that certain
modifications to this topology provide significant and truly
unexpected increases in passband bandwidths and, in certain
instances, the ability to generate multiple passbands of the same
or varying bandwidths in an efficient and effective manner.
[0074] With reference to FIG. 9, a modified circuit topology is
illustrated wherein the circuit topology of FIG. 7 is modified to
include an additional shunt resonator B3, which is coupled between
the common node CN and ground. As such, a new compensation circuit
44 is created that includes the negatively coupled inductors L1 and
L2, which have a coupling coefficient K, and at least two shunt
resonators B2, B3. The compensation circuit 44 is coupled in
parallel with the series resonator B1. When the series resonance
frequencies f.sub.s of the shunt resonators B2, B3 are different
from one another, unexpectedly wide bandwidths passbands are
achievable while maintaining a very flat passbands, steep skirts,
and excellent cancellation of signals outside of the passbands.
[0075] FIG. 10 graphically illustrates the frequency responses of
the series resonator B1 (inside the block referenced B1), the
compensation circuit 44 (inside the block referenced 44), and the
filter circuit in which the compensation circuit 44 is placed in
parallel with the series resonator B1. As illustrated, the filter
circuit with compensation circuit 44 provides a much wider passband
(FIG. 10) than the filter circuit with compensation circuit 42
(FIG. 8).
[0076] While FIGS. 8 and 10 are graphical representations, FIG. 11
is an actual comparison of the frequency response of the filter
circuit using the different compensation circuits 42, 44, wherein
the filter circuit using compensation circuit 44 provides a
significantly wider and better formed passband (solid line) than
the filter circuit using compensation circuit 42 (dashed line).
[0077] As illustrated in FIG. 12, the concepts described herein not
only contemplate the use of multiple shunt resonators B2, B3, which
are coupled between the common node CN and ground, but also
multiple series resonators, such as series resonators B1 and B4,
which are coupled in parallel with one another between the input
node VP and the output node O/P. The series resonance frequencies
f.sub.s of the series resonators B1, B4 are different from one
another, and the series resonance frequencies f.sub.s of the shunt
resonators B2, B3 are also different from one another and different
from those of the series resonators. While only two series
resonators B1, B4 and two shunt resonators B2, B3 are illustrated,
any number of these resonators may be employed depending on the
application and the desired characteristics of the overall
frequency response of the circuit in which these resonators and
associated compensation circuits 44 are employed. While the theory
of operation is described further subsequently, FIGS. 13 and 14
illustrate just two of the many possibilities.
[0078] For FIG. 13, there are two series resonators B1, B4 and two
shunt resonators B2, B3, with different and relatively dispersed
series resonance frequencies f.sub.s. FIG. 13 graphically
illustrates the frequency responses of the combination of the two
series resonators B1, B4 (inside the block referenced BX), the
compensation circuit 44 with two shunt resonators B2, B3 (inside
the block referenced 44), and the filter circuit in which the
compensation circuit 44 is placed in parallel with the series
resonators B1, B4. As illustrated, the filter circuit in this
configuration has the potential to provide a passband that is even
wider than that for the embodiment of FIGS. 9 and 10. For example,
passbands of greater than 100 MHz, 150 MHz, 175 MHz, and 200 MHz
are contemplated at frequencies at or above 1.5 GHz, 1.75 GHz, and
2 GHz, respectively. In other words, center-frequency-to-bandwidth
ratios (fc/BW*100) of 3.5% to 9%, 12%, or greater are possible,
wherein fc is the center frequency of the passband and BW is the
bandwidth of the passband. If multiple passbands are provided, BW
may encompass all of the provided passbands. Further, when multiple
passbands are provided, the passbands may have the same or
different bandwidths or center-frequency-to-bandwidth ratios. For
example, one passband may have a relatively large
center-frequency-to-bandwidth ratio, such as 12%, and a second
passband may have a relatively small center-frequency-to-bandwidth
ratio, such as 2%. Alternatively, multiple ones of the passbands
may have a bandwidth of 100 MHz, or multiple ones of the passbands
may have generally the same center-frequency-to-bandwidth ratios.
In the latter case, the bandwidths of the passbands may inherently
be different from one another, even though the
center-frequency-to-bandwidth ratios are the same.
[0079] For FIG. 14, there are four series resonators, which are
coupled in parallel with one another (not shown), and two shunt
resonators (not shown) with different and more widely dispersed
series resonance frequencies f.sub.s. FIG. 14 graphically
illustrates the frequency responses of the combination of the four
series resonators (inside the block referenced BX), the
compensation circuit 44 with two shunt resonators B2, B3 (inside
the block referenced 44), and the filter circuit in which the
compensation circuit 44 is placed in parallel with four series
resonators. As illustrated, the filter circuit in this
configuration provides multiple passbands, which are separated by a
stop band. In this embodiment, two passbands are provided; however,
the number of passbands may exceed two. The number of passbands and
the bandwidth of each of the passbands is a function of the number
of shunt and series resonators B1-B4 and the series resonance
frequencies f.sub.s thereof.
[0080] The theory of the compensation circuit 44 follows and is
described in association with FIGS. 15A through 15D and 16. With
reference to FIG. 15A, assume the compensation circuit 44 includes
the two negatively coupled inductors L1, L2, which have an
inductance value L, and two or more shunt resonators BY, which have
an overall shunt impedance Zres presented between the common node
CN and ground. While the inductance values L of the negatively
coupled inductors L1, L2 are described as being the same, these
values may differ depending on the application. Also assume that
the one or more series resonators BX present an overall series
impedance ZS.
[0081] As shown in FIG. 15B, the two negatively coupled and
series-connected inductors L1, L2 (without Zres) can be modeled as
a T-network of three inductors L3, L4, and L5, wherein series
inductors L3 and L4 are connected in series and have a value of
L(1+K), and shunt inductor L5 has a value of -L*K, where K is a
coupling factor between the negatively coupled inductors L1, L2.
Notably, the coupling factor K is a positive number between 0 and
1. Based on this model, the overall impedance of the compensation
circuit 44 is modeled as illustrated in FIG. 15C, wherein the shunt
impedance Zres is coupled between the shunt inductor L5 and ground.
The resulting T-network, as illustrated in FIG. 15C, can be
transformed into an equivalent .pi. (pi) network, as illustrated in
FIG. 15D.
[0082] The .pi. network of FIG. 15D can be broken into a series
impedance ZA and two shunt equivalent impedances ZB. The series
equivalent impedance ZA is represented by two series inductances of
value L*(1+K), where K>0, and a special "inversion" impedance
Zinv. The inversion impedance Zinv is equal to
[L(1+K).omega.].sup.2/[Zres-jLK.omega.], where .omega.=2.pi.f and f
is the frequency. As such, the series equivalent impedance ZA
equals j*2*L(1+K).omega.+Zinv and is coupled between the input node
I/O and the output node O/P. Each of the two shunt equivalent
impedances ZB is represented by an inductor of value L(1-K) in
series with two overall shunt impedances Zres.
[0083] Notably, the series equivalent impedance ZA has a negative
capacitor behavior at certain frequencies at which broadband
cancellation is desired and has series resonance at multiple
frequencies. In general, the series equivalent impedance ZA has a
multiple bandpass-bandstop characteristic in that the series
equivalent impedance ZA will pass some frequencies and stop others.
When the series equivalent impedance ZA is placed in parallel with
the series impedance ZS of the series resonators BX, which can also
have a multiple bandpass-bandstop characteristic, a broadband
filter or a filter with multiple passbands may be created.
[0084] FIG. 16 illustrates the series impedance ZS of the series
resonators BX in parallel with the series equivalent impedance ZA
of the compensation circuit 44. The overall series impedance ZAs
represents the series impedance ZS in parallel with the series
equivalent impedance ZA. The two shunt impedances ZB are
respectively coupled between the input port VP and ground and the
output port O/P and ground. The primary focus for the following
discussion relates to the series equivalent impedance ZA and its
impact on the series impedance ZS when the series equivalent
impedance ZA is placed in parallel with the series impedance
ZS.
[0085] As noted previously, the series equivalent impedance ZA
provides two primary functions. The first provides a negative
capacitive behavior, and the second provides one or more additional
series resonances between the input node VP and the output node
O/P. These additional series resonances are provided through the
series equivalent impedance ZA and are in addition to any series
resonances that are provided through the series impedance ZS of the
series resonators BX. To help explain the benefits and concept of
the negative capacitive behavior provided by the series equivalent
impedance ZA, normal capacitive behavior is illustrated in
association with the overall shunt impedance Zres, which is
provided by the shunt resonators BY. FIG. 17 graphs the absolute
(magnitude) and imaginary components of the overall shunt impedance
Zres, which is formed by two shunt resonators BY, which are coupled
in parallel with one another.
[0086] The series resonance frequency f.sub.s for each of the two
shunt resonators BY occurs when the absolute impedance (abs(Zres))
is at or near zero. Since there are two shunt resonators BY, the
absolute impedance (abs(Zres)) is at or near zero at two
frequencies, and as such, there are two series resonance
frequencies f.sub.s. The parallel resonance frequencies f.sub.p
occur when the imaginary component (imag(Zres)) peaks. Again, since
there are two shunt resonators BY, there are two parallel resonance
frequencies f.sub.p provided by the overall shunt impedance
Zres.
[0087] Whenever the imaginary component (imag(Zres)) of the overall
shunt impedance Zres is less than zero, the overall shunt impedance
Zres has a capacitive behavior. The capacitive behavior is
characterized in that the reactance of the overall shunt impedance
Zres is negative and decreases as frequency increases, which is
consistent with capacitive reactance, which is represented by
1/j.omega.C. The graph of FIG. 17 identifies three regions within
the impedance response of the overall shunt impedance Zres that
exhibit capacitive behavior.
[0088] Turning now to FIG. 18, the series equivalent impedance ZA
is illustrated over the same frequency range as that of the overall
shunt impedance Zres, which was illustrated in FIG. 17. The series
equivalent impedance ZA has two series resonance frequencies
f.sub.s, which occur when the absolute impedance (abs(ZA)) is at or
near zero. The two series resonance frequencies f.sub.s for the
series equivalent impedance ZA are different from each other and
slightly different from those for the overall shunt impedance Zres.
Further, the number of series resonance frequencies f.sub.s
generally corresponds to the number of shunt resonators BY in the
compensation circuit 44, assuming the series resonance frequencies
f.sub.s are different from one another.
[0089] Interestingly, the imaginary component (imag(ZA)) of the
series equivalent impedance ZA is somewhat inverted with respect to
that of the overall shunt impedance Zres. Further, the imaginary
component (imag(ZA)) of the series equivalent impedance ZA has a
predominantly positive reactance. During the portions at which the
imaginary component (imag(ZA)) is positive, the reactance of the
series equivalent impedance ZA again decreases as frequency
increases, which is indicative of capacitive behavior. However, the
reactance is positive, whereas traditional capacitive behavior
would present a negative reactance. This phenomenon is referred to
as negative capacitive behavior. Those portions of the imaginary
component (imag(ZA)) of the series equivalent impedance ZA that are
positive and thus exhibit negative capacitive behavior are
highlighted in the graph of FIG. 18.
[0090] The negative capacitive behavior of the series equivalent
impedance ZA for the compensation circuit 44 is important, because
when the series equivalent impedance ZA is placed in parallel with
the series impedance ZS, the effective capacitance of the filter
circuit is reduced. Reducing the effective capacitance of the
filter circuit shifts the parallel resonance frequency f.sub.p of
the series impedance ZS higher in the frequency range, which is
described subsequently, and significantly increases the available
bandwidth for passbands while providing excellent out-of-band
rejection.
[0091] An example of the benefit is illustrated in FIGS. 19A and
19B. The solid line, which is labeled abs(VG), represents the
frequency response of the filter circuit illustrated in FIG. 12,
wherein there are two series resonators BX and two shunt resonators
BY in the compensation circuit 44. The frequency response has two
well-defined passbands, which are separated by a stop band. The
frequency response abs(VG) of the filter circuit generally
corresponds to the inverse of the overall series impedance ZAs,
which again represents the series impedance ZS in parallel with the
series equivalent impedance ZA, as provided in FIG. 16.
[0092] Notably, the parallel resonance frequencies f.sub.p(ZS) of
the series impedance ZS, in isolation, fall in the middle of the
passbands of frequency response abs(VG) of the filter circuit. If
the parallel resonance frequencies f.sub.p(ZS) of the series
impedance ZS remained at these locations, the passbands would be
severely affected. However, the negative capacitive behavior of the
series equivalent impedance ZA functions to shift these parallel
resonance frequencies f.sub.p(ZS) of the series impedance ZS to a
higher frequency and, in this instance, above the respective
passbands. This is manifested in the resulting overall series
impedance ZAs, in which the only parallel resonance frequencies
f.sub.p(ZAs) occur above and outside of the respective passbands.
An additional benefit to having the parallel resonance frequencies
f.sub.p(ZAs) occur outside of the respective passbands is the
additional cancellation of frequencies outside of the passbands.
Plus, the overall series impedance ZAs is lower than the series
impedance ZS within the respective passbands.
[0093] A further contributor to the exemplary frequency response
abs(VG) of the filter circuit is the presence of the additional
series resonance frequencies f.sub.s, which are provided through
the series equivalent impedance ZA. These series resonance
frequencies f.sub.s are offset from each other and from those
provided through the series impedance ZS. The series resonance
frequencies f.sub.s for the series equivalent impedance ZA in the
series impedance ZS occur when the magnitudes of the respective
impedances approach zero. The practical results are wider
passbands, steeper skirts for the passbands, and greater rejection
outside of the passbands, as evidenced by the frequency response
abs(VG) of the filter circuit.
[0094] Turning now to FIG. 20, another embodiment is provided
wherein the compensation circuit 44 is placed in parallel with one
or more series resonators BX. In this embodiment, shunt resonator
B2 is permanently coupled between the common node CN and ground.
Shunt resonators B3, B5, and B6 can be selectively coupled between
the common node CN and ground via respective switches S1, S2, and
S3. By using control circuitry (not shown) to selectively switch
the various shunt resonators B3, B5, and B6 into and out of the
compensation circuit 44, the passbands and stop bands provided by
the filter circuit can be dynamically adjusted for different modes
of operation. Again, the series resonance frequencies f.sub.s of
the shunt resonators B2, B3, B5, and B6 will generally differ from
one another. Resistors R1 and R2 are illustrated and may be coupled
between the input node I/P and ground and the output node O/P and
ground, respectively. In one embodiment, the series resonance
frequency f.sub.s of the series equivalent impedance ZA is greater
than the series resonance frequency f.sub.s of the series impedance
ZS. The series resonance frequency f.sub.s of at least one of the
shunt resonators B2, B3, B5, and B6 is greater than the series
resonance frequency f.sub.s of the series impedance ZS. As with any
of these embodiments, the number of shunt resonators BY and series
resonators BX may vary from embodiment to embodiment. The number
illustrated is merely for illustrative purposes.
[0095] FIG. 21 illustrates yet another embodiment, which is similar
to that illustrated in FIG. 20. The difference is that the series
resonators BX are replaced with a lumped series L-C circuit, which
is formed from a series capacitor CS and a series inductor LS that
are coupled in series between the input node I/P and the output
node O/P.
[0096] FIG. 22 illustrates an embodiment similar to that of FIG. 9,
except that at least one inductor L6 is coupled in series with
shunt resonator B3. As such, shunt resonator B2 is coupled between
the common node CN and ground without a series inductor, and shunt
resonator B2 and inductor L6 are coupled in series with one another
and between the common node CN and ground. In one embodiment, the
series resonance frequency f.sub.s of the series equivalent
impedance ZA is greater than the lowest series resonance frequency
f.sub.s of the series impedance ZS. FIG. 23 illustrates a further
modification to the embodiment of FIG. 22, wherein an inductor L7
is placed in series with the series resonators BX, such that the
inductor L7 and the series resonators BX are coupled in series with
one another between the input node I/P and the output node O/P.
[0097] With reference to FIG. 24, a more complex filter arrangement
is illustrated. In particular, resonators B7, B8, and B9 are
coupled in series between the input node I/P and the output node
O/P. The compensation circuits 44 of FIG. 22 are coupled across
resonator B7 and resonator B9. The resonators B7, B8, and B9 may
each represent a single acoustic resonator or multiple acoustic
resonators in parallel. Notably, the shunt resonators B2, B3, B5,
B6 in the embodiments of FIGS. 21 through 24 and 30 are considered
to be in parallel with one another between the common node CN and
ground, even if an additional switch, inductive, capacitive, or
like element is provided in series with one or more of the shunt
resonators.
[0098] FIG. 25 illustrates yet another embodiment wherein
resonators B10 and B11 are coupled in series between the input node
I/P and the output node O/P. An inductor L8 is coupled between node
N1 and ground. The compensation circuit 44 is coupled across both
of the resonators B10 and B11. Accordingly, the compensation
circuit 44 may be coupled across one or more acoustic resonators
along the series path that extends between the input node I/P and
the output node O/P.
[0099] FIG. 26 illustrates a modification to the embodiment of FIG.
25, wherein the inductor L8 is replaced with a shunt resonator
B12.
[0100] FIG. 27 illustrates an embodiment wherein the compensation
circuit 44 is tunable, such that the bandwidth of the passband for
the filter circuit can be varied, or tuned, in a dynamic fashion.
As illustrated, the compensation circuit 44 is provided in parallel
with the series resonator BX. The compensation circuit 44 differs
from the above-described embodiments in that a variable capacitor
C1, or varactor, is placed in parallel with at least two of the
shunt resonators B2, B3 between the common node CN and ground. As
the capacitance of the variable capacitor C1 varies, the bandwidth
of the passband for the filter circuit will vary. In this
embodiment, the upper skirt of the passband will remain relatively
constant, while the location of the lower skirt will vary. In
particular, as the capacitance of the variable capacitor C1
increases, the location of the lower skirt of the passband
increases, and vice versa.
[0101] The graph of FIG. 28 illustrates the frequency responses of
the filter circuit of FIG. 27 at three different capacitance levels
for the variable capacitor C1. Frequency response FR1 has the
broadest passband bandwidth and corresponds to the lower of the
three capacitance levels. Frequency response FR2 provides an
intermediate passband bandwidth and corresponds to an intermediate
capacitance level for the variable capacitor C1. Frequency response
FR3 provides the narrowest passband bandwidth and corresponds to a
higher capacitance level for the variable capacitor C1. As
depicted, the upper skirts of the three passbands remain relatively
constant, and the location of the lower skirts for the three
passbands progressively increase as the capacitance provided by the
variable capacitor C1 increases. The result is that the bandwidth
of the passbands decrease as the capacitance provided by the
variable capacitor C1 increases.
[0102] As illustrated in FIG. 28, the three frequency responses
FR1-3 exhibit flat passbands, steep side skirts, and excellent
out-of-band rejection above and below the passbands. A further
benefit of the described circuitry is that the return losses of
RL1-3 are excellent within the passband for each of the three
corresponding frequency responses FR1-3, especially for the two
wider-bandwidth passbands (RL1 and RL2).
[0103] FIG. 29 illustrates a modified version of the filter circuit
of FIG. 28. In this example, an additional variable capacitor C2 is
provided in parallel with at least two series resonators B1, B4.
The additional variable capacitor C2 provides additional tunability
of the overall frequency response.
[0104] Turning now to FIG. 30, another embodiment is provided where
the additional variable capacitor C2 is provided in parallel with
at least two series resonators B1, B4. In this embodiment, shunt
resonator B2 is permanently coupled between the common node CN and
ground. Shunt resonators B3 and B5 can be selectively coupled
between the common node CN and ground via respective switches S1
and S2. Variable capacitor C1 is provided in parallel with the
shunt resonators B3 and B5 between the common node CN and ground.
By using control circuitry 46 to selectively switch the various
shunt resonators B3 and B5 into and out of the compensation circuit
44 using switches S1 and S2 as well as varying the capacitance of
the variable capacitors C1 and C2, the bandwidths and locations of
the passbands provided by the filter circuit can be dynamically
adjusted for different modes of operation. Again, the series
resonance frequencies f.sub.s of the shunt resonators B2, B3 and B5
will generally differ from one another. In one embodiment, the
series resonance frequency f.sub.s of at least one of the shunt
resonators B2, B3, and B5 is greater than the series resonance
frequency f.sub.s of the series impedance ZS. As with any of these
embodiments, the number of shunt resonators BY and series
resonators BX may vary from embodiment to embodiment. The number
illustrated is merely for illustrative purposes.
[0105] FIGS. 31 and 32 are graphs of frequency responses and return
losses for an exemplary configuration of the filter circuit of FIG.
30. Assume that the shunt resonator B2 has a series resonance
frequency of 2.700 GHz, shunt resonator B3 has a series resonance
frequency of 2.470 GHz, and shunt resonator B5 as a series
resonance frequency of 2.490 GHz. For the filter circuit of FIG.
30, shunt resonator B2 is permanently coupled between the common
node CN and ground, and the shunt resonators B3 and B5 are
alternately switched into and out of the circuit using switches S1
and S2, respectively. The total inherent, or parasitic, capacitance
for either of the shunt resonators B3 or B5 in parallel with shunt
resonator B2 is approximately 0.65 pF. Assume that the variable
capacitor C1 has a capacitance that can vary between 0 pF and at
least 0.20 pF.
[0106] The frequency response FR4 and the return loss RL4
illustrated in FIG. 31 corresponds to the shunt resonator B3 being
switched into the circuit, shunt resonator B5 being switched out of
the circuit, and the capacitance for the variable capacitor C1
being set to zero. The passband provided by the frequency response
FR4 corresponds to the requisite passband for downlink band 41
(B41) of the LTE (Long Term Evolution) standard for cellular
communications. The passband for downlink LTE band B41 is 2.496 GHz
to 2.690 GHz.
[0107] The frequency response can be dynamically modified to the
requisite passband for downlink band 38X (B38X) of the LTE standard
by switching the shunt resonator B3 out of the circuit, switching
the shunt resonator B5 into the circuit, and adjusting the
capacitance for the variable capacitor C1 to approximately 0.20 pF,
such that the total capacitance between the common node CN and
ground is approximately 0.85 pF (0.20 pF+0.65 pF). The passband for
downlink LTE band B38X is 2.545 GHz to 2.655 GHz, wherein the upper
end of the passband is only 35 MHz lower that required for LTE band
B41. In FIG. 32, the frequency response FR5 and the return loss RL5
for LTE band B38X is illustrated along with the frequency response
FR4 and the return loss RL4 for LTE band B41. As illustrated, LTE
bands B41 and B38X have similar upper skirts but have significantly
different locations for the lower skirts. Notably, the return
losses RL4 and RL5 within the passbands are exceptionally low, and
in this example, less than -13 dB throughout a vast majority of the
passbands. In this example, assume that variable capacitor C2 is
not used or is set to approximately 0 pF.
[0108] For the example associated with graphs of FIGS. 33 and 34,
the variable capacitor C2 is employed along with the variable
capacitor C1. Assume that the shunt resonator B2 has a series
resonance frequency of 2.700 GHz, shunt resonator B3 has a series
resonance frequency of 2.470 GHz, shunt resonator B5 as a series
resonance frequency of 2.550 GHz, and series resonator B1 as a
series resonance frequency of 2.552 GHz. Again, shunt resonator B2
is permanently coupled between the common node CN and ground, and
the shunt resonators B3 and B5 are alternately switched into and
out of the circuit using switches S1 and S2, respectively. The
total inherent, or parasitic, capacitance for either of the shunt
resonators B3 or B5 in parallel with shunt resonator B2 is
approximately 0.95 pF. Assume that the variable capacitor C1 has a
capacitance that can vary between 0 pF and at least 1.20 pF, and
the variable capacitor C1 has a capacitance that can vary between 0
pF and at least 0.35 pF.
[0109] The frequency response FR6 and the return loss RL6
illustrated in FIG. 33 correspond to the shunt resonator B3 being
switched into the circuit, shunt resonator B5 being switched out of
the circuit, and the capacitances for the variable capacitor C1 and
the variable capacitor C2 being set to zero. The passband provided
by the frequency response FR4 again corresponds to the requisite
passband for LTE band B41, which is 2.496 GHz to 2.690 GHz.
[0110] The frequency response can be dynamically modified to the
requisite passband for downlink LTE band 7 (B7RX) of the LTE
standard by switching the shunt resonator B3 out of the circuit,
switching the shunt resonator B5 into the circuit, and adjusting
the capacitance for the variable capacitor C1 to approximately 1.20
pF and the capacitance for the variable capacitor C2 to
approximately 0.35 pF. The total capacitance between the common
node CN and ground is approximately 1.85 pF (1.20 pF+0.65 pF), and
the capacitance across the series resonators B1 and B4 is
approximately 1.3 pF (0.95 pF+0.35 pF). The passband for LTE band
B7RX is 2.620 GHz to 2.690 GHz, wherein the upper end of the
passband aligns precisely with that required for LTE band B41.
[0111] In FIG. 34, the frequency response FR7 and the return loss
RL7 for LTE band B7RX is illustrated along with the frequency
response FR6 and the return loss RL6 for LTE band B41. As
illustrated, LTE bands B41 and B7RX have upper skirts that align
with one another, but have significantly different locations for
the lower skirts. Notably, the return losses RL6 and RL7 within the
passbands are exceptionally low, and this example, less than -13 dB
throughout a vast majority of the passbands. While the above
examples relate to LTE bands B41, B41RX, and B38X, the concepts
above can apply to any bands in any communication standard. The
concepts above are particularly beneficial when implemented in
receive filters that rest between one or more antennas and
downconversion or other receiver circuitry in virtually any
wireless communication application. These concepts may also be
implemented in transmit filters.
[0112] With reference to FIG. 35, other techniques may be applied
to the compensation circuit 44 to facilitate tuning the passband
for the associated filter circuitry. As illustrated, bias circuitry
48 may be used to apply one or more DC bias signals to one or more
of the variable capacitor C1, shunt resonator B2, and shunt
resonator B3. In this example, the DC bias signal is a DC voltage
that is applied to the common node CN. Doing so adjusts the
effective capacitance between the common node CN and ground, and
thus, will shift at least the lower skirt of the passband up or
down.
[0113] As illustrated in FIG. 36, the same DC bias signal or
different DC bias signals may be provided to the variable capacitor
C2, series resonator B1, and/or series resonator B4, to adjust the
series capacitance presented between the input node VP and the
output node O/P. Adjusting the series capacitance also impacts the
center frequency, lower skirt, and/or upper skirt of the passband
provided by the filter circuit. The bias circuitry 48 may also be
applied to the embodiments of FIG. 30, wherein the DC bias signals
provided to the variable capacitor C1, variable capacitor C2,
and/or any of the series resonators B1, B4 and shunt resonators B2,
B3, and B5.
[0114] Those skilled in the art will recognize numerous
modifications and other embodiments that incorporate the concepts
described herein. These modifications and embodiments are
considered to be within scope of the teachings provided herein and
the claims that follow.
* * * * *