U.S. patent application number 15/145839 was filed with the patent office on 2016-08-25 for antenna structures and methods thereof for adjusting an operating frequency range of an antenna.
The applicant listed for this patent is SKYCROSS, INC.. Invention is credited to FRANK M. CAIMI, MARK T. MONTGOMERY.
Application Number | 20160248155 15/145839 |
Document ID | / |
Family ID | 52994784 |
Filed Date | 2016-08-25 |
United States Patent
Application |
20160248155 |
Kind Code |
A1 |
MONTGOMERY; MARK T. ; et
al. |
August 25, 2016 |
ANTENNA STRUCTURES AND METHODS THEREOF FOR ADJUSTING AN OPERATING
FREQUENCY RANGE OF AN ANTENNA
Abstract
A system that incorporates the subject disclosure may include,
for example, a circuit for obtaining a desired bandwidth of
operation of the antenna structure, adjusting a bandwidth of the
antenna to achieve the desired bandwidth of operation of the
antenna structure, and tuning the antenna to a new resonant
frequency to accommodate transmitting or receiving RF signals at a
desired band of operation. The antenna at the new resonant
frequency is at least approximately at the desired bandwidth of
operation of the antenna structure. Other embodiments are
disclosed.
Inventors: |
MONTGOMERY; MARK T.;
(Melbourne Beach, FL) ; CAIMI; FRANK M.; (Vero
Beach, FL) |
|
Applicant: |
Name |
City |
State |
Country |
Type |
SKYCROSS, INC. |
San Jose |
CA |
US |
|
|
Family ID: |
52994784 |
Appl. No.: |
15/145839 |
Filed: |
May 4, 2016 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
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14467915 |
Aug 25, 2014 |
9362619 |
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15145839 |
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61896233 |
Oct 28, 2013 |
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61932831 |
Jan 29, 2014 |
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61941888 |
Feb 19, 2014 |
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Current U.S.
Class: |
1/1 |
Current CPC
Class: |
H01Q 9/42 20130101; H01Q
21/29 20130101; H01Q 1/38 20130101; H03J 5/242 20130101; H04W 24/10
20130101; H04W 4/026 20130101; H01Q 1/2291 20130101; H01Q 21/28
20130101; H03J 5/0245 20130101; H03J 2200/15 20130101; H04B 1/0053
20130101; H04B 7/0608 20130101; H01Q 21/12 20130101; H04W 72/0486
20130101; H04W 36/08 20130101; H04W 64/003 20130101; H01Q 21/0006
20130101; H04B 7/0417 20130101; H04W 28/12 20130101; G01C 21/20
20130101; H01Q 1/243 20130101; H04B 1/38 20130101; H04W 88/06
20130101; H01Q 5/00 20130101; H01Q 5/371 20150115; H01Q 13/10
20130101; H01Q 3/24 20130101; H01Q 5/22 20150115; H01Q 9/30
20130101; H04B 1/401 20130101; H04W 36/18 20130101; H01Q 9/16
20130101; H03J 5/244 20130101; H04B 7/024 20130101; H01Q 1/50
20130101; H01Q 9/28 20130101; H01Q 1/521 20130101; H04W 24/08
20130101; H04W 36/32 20130101; H04W 88/08 20130101; H01Q 9/145
20130101; H04B 7/0691 20130101; H04W 40/12 20130101; G01R 29/0807
20130101; H01Q 9/04 20130101; H01Q 9/40 20130101; H04B 7/0628
20130101; H04B 7/0404 20130101; H04W 72/0453 20130101; H01Q 5/40
20150115; H04B 7/0632 20130101; G01R 23/02 20130101; H01Q 9/0442
20130101; G01S 5/0009 20130101; H01Q 3/34 20130101; H01Q 13/103
20130101; H01Q 5/48 20150115; H01Q 21/00 20130101; G01R 25/00
20130101; G01R 29/10 20130101; H04B 7/0413 20130101; H04W 72/085
20130101 |
International
Class: |
H01Q 3/24 20060101
H01Q003/24; H01Q 1/24 20060101 H01Q001/24; H01Q 1/50 20060101
H01Q001/50 |
Claims
1. A method, comprising: determining whether intraband carrier
aggregation is desired; adjusting, by a circuit comprising a
processor, a reactive element of an antenna to reduce an operating
bandwidth of the antenna responsive to a first determination that
intraband carrier aggregation is not desired; adjusting, by the
circuit, the reactive element of the antenna to increase the
operating bandwidth of the antenna responsive to a second
determination that intraband carrier aggregation is desired; and
tuning, by the circuit, the antenna to a new resonant frequency to
accommodate transmitting or receiving RF signals at a desired band
of operation, wherein the antenna at the new resonant frequency has
the reduced operating bandwidth or the increased operating
bandwidth of the antenna.
2. The method of claim 1, wherein the operating bandwidth is
reduced responsive to the first determination to reduce
harmonics.
3. The method of claim 2, wherein the harmonics are reduced due to
mixing when signals are being transmitted by the antenna.
4. The method of claim 1, wherein the operating bandwidth is
reduced responsive to the first determination to reduce
intermodulation distortion.
5. The method of claim 4, wherein the intermodulation distortion is
reduced due to mixing when signals are being transmitted by the
antenna.
6. The method of claim 1, wherein the operating bandwidth of the
antenna is reduced when signals are being transmitted by the
antenna to reduce spurious radiation outside a desired channel.
7. The method of claim 1, wherein the reactive element comprises
one of a tunable capacitor, a tunable inductor, or a combination
thereof.
8. The method of claim 1, further comprising detecting a frequency
offset in an operating frequency of the antenna, wherein the tuning
of the antenna is responsive to detecting the frequency offset.
9. The method of claim 8, further comprising determining a
magnitude difference between a first signal supplied to the antenna
and a second signal radiated by the antenna, wherein the frequency
offset is detected according to the magnitude difference.
10. The method of claim 8, further comprising determining a phase
difference between a first signal supplied to the antenna and a
second signal radiated by the antenna, wherein the frequency offset
is detected according to the phase difference.
11. The method of claim 8, further comprising measuring a change in
reactance of the antenna, wherein the frequency offset is detected
according to the change in reactance of the antenna.
12. A method, comprising: responsive to a first determination that
carrier aggregation is not desired, adjusting, by a circuit
comprising a processor and an antenna, a quality factor of the
antenna to reduce an operating bandwidth of the antenna and to
reduce spurious radiation due to transmission outside a passband of
the antenna; and tuning, by the circuit, the antenna to a new
resonant frequency to accommodate transmitting or receiving RF
signals at a desired band of operation, wherein the antenna at the
new resonant frequency has the reduced operating bandwidth of the
antenna.
13. The method of claim 12, further comprising: looking up LRC
model parameters from a table for the antenna; and adjusting a
reactive element of the antenna according to the LRC model
parameters.
14. The method of claim 13, wherein the reactive element comprises
one of a tunable capacitor, a tunable inductor, or a combination
thereof.
15. The method of claim 13, wherein the looking up is based on the
first determination.
16. The method of claim 12, further comprising detecting a
frequency offset in an operating frequency of the antenna, wherein
the tuning of the antenna is responsive to detecting the frequency
offset.
17. The method of claim 16, further comprising determining one of a
magnitude difference between a first signal supplied to the antenna
and a second signal radiated by the antenna, a phase difference
between the first signal supplied to the antenna and the second
signal radiated by the antenna, a change in reactance of the
antenna, or any combination thereof.
18. A communication device, comprising: an antenna structure; and a
circuit coupled to the antenna structure, wherein the circuit
performs operations comprising: responsive to a determination that
interband aggregation is not desired, narrowing of a bandwidth of
the antenna structure to achieve a desired bandwidth of operation
of the antenna structure to reduce harmonic distortion; and tuning
the antenna structure to a new resonant frequency to accommodate
transmitting or receiving RF signals at a desired band of
operation, wherein the antenna structure at the new resonant
frequency is at least approximately at the desired bandwidth of
operation of the antenna structure.
19. The communication device of claim 18, wherein the narrowing
further comprises obtaining a reactive parameter from a look-up
table to achieve the desired bandwidth of operation of the antenna
structure.
20. The communication device of claim 19, wherein the narrowing
further comprises adjusting a reactive element of the antenna
structure according to the reactive parameter to achieve the
desired bandwidth of operation of the antenna structure.
Description
CROSS REFERENCE TO RELATED APPLICATIONS
[0001] The present application is a continuation of U.S. patent
application Ser. No. 14/467,915 filed Aug. 25, 2014 by Montgomery
et al., entitled "Antenna Structures and Methods Thereof for
Adjusting an Operating Frequency Range of an Antenna," which claims
the benefit of priority to U.S. Provisional Application No.
61/896,233 filed on Oct. 28, 2013, 61/932,831 filed on Jan. 29,
2014, and 61/941,888 filed on Feb. 19, 2014, each of which is
hereby incorporated herein by reference in its entirety.
FIELD OF THE DISCLOSURE
[0002] The present disclosure relates generally to antenna
structures and methods thereof for adjusting an operating frequency
range of an antenna.
BACKGROUND
[0003] It is common for communications devices to have multiple
antennas that are packaged close together (e.g., less than a
quarter of a wavelength apart) and that can operate simultaneously
within the same frequency band. Common examples of such
communications devices include portable communications products
such as cellular handsets, personal digital assistants (PDAs), and
wireless networking devices or data cards for personal computers
(PCs). Many system architectures (such as Multiple Input Multiple
Output (MIMO)) and standard protocols for mobile wireless
communications devices (such as 802.11n for wireless LAN, and 3G
and 4G data communications such as 802.16e (WiMAX), HSDPA,
1.times.EVDO, and LTE) may require multiple antennas operating
simultaneously.
BRIEF DESCRIPTION OF THE DRAWINGS
[0004] Reference will now be made to the accompanying drawings,
which are not necessarily drawn to scale, and wherein:
[0005] FIG. 1A illustrates an antenna structure with two parallel
dipoles;
[0006] FIG. 1B illustrates current flow resulting from excitation
of one dipole in the antenna structure of FIG. 1A;
[0007] FIG. 1C illustrates a model corresponding to the antenna
structure of FIG. 1A;
[0008] FIG. 1D is a graph illustrating scattering parameters for
the FIG. 1C antenna structure;
[0009] FIG. 1E is a graph illustrating the current ratios for the
FIG. 1C antenna structure;
[0010] FIG. 1F is a graph illustrating gain patterns for the FIG.
1C antenna structure;
[0011] FIG. 1G is a graph illustrating envelope correlation for the
FIG. 1C antenna structure;
[0012] FIG. 2A illustrates an antenna structure with two parallel
dipoles connected by connecting elements in accordance with one or
more embodiments of the disclosure;
[0013] FIG. 2B illustrates a model corresponding to the antenna
structure of FIG. 2A;
[0014] FIG. 2C is a graph illustrating scattering parameters for
the FIG. 2B antenna structure;
[0015] FIG. 2D is a graph illustrating scattering parameters for
the FIG. 2B antenna structure with lumped element impedance
matching at both ports;
[0016] FIG. 2E is a graph illustrating the current ratios for the
FIG. 2B antenna structure;
[0017] FIG. 2F is a graph illustrating gain patterns for the FIG.
2B antenna structure;
[0018] FIG. 2G is a graph illustrating envelope correlation for the
FIG. 2B antenna structure;
[0019] FIG. 3A illustrates an antenna structure with two parallel
dipoles connected by meandered connecting elements in accordance
with one or more embodiments of the disclosure;
[0020] FIG. 3B is a graph showing scattering parameters for the
FIG. 3A antenna structure;
[0021] FIG. 3C is a graph illustrating current ratios for the FIG.
3A antenna structure;
[0022] FIG. 3D is a graph illustrating gain patterns for the FIG.
3A antenna structure;
[0023] FIG. 3E is a graph illustrating envelope correlation for the
FIG. 3A antenna structure;
[0024] FIG. 4 illustrates an antenna structure with a ground or
counterpoise in accordance with one or more embodiments of the
disclosure;
[0025] FIG. 5 illustrates a balanced antenna structure in
accordance with one or more embodiments of the disclosure;
[0026] FIG. 6A illustrates an antenna structure in accordance with
one or more embodiments of the disclosure;
[0027] FIG. 6B is a graph showing scattering parameters for the
FIG. 6A antenna structure for a particular dipole width
dimension;
[0028] FIG. 6C is a graph showing scattering parameters for the
FIG. 6A antenna structure for another dipole width dimension;
[0029] FIG. 7 illustrates an antenna structure fabricated on a
printed circuit board in accordance with one or more embodiments of
the disclosure;
[0030] FIG. 8A illustrates an antenna structure having dual
resonance in accordance with one or more embodiments of the
disclosure;
[0031] FIG. 8B is a graph illustrating scattering parameters for
the FIG. 8A antenna structure;
[0032] FIG. 9 illustrates a tunable antenna structure in accordance
with one or more embodiments of the disclosure;
[0033] FIGS. 10A and 10B illustrate antenna structures having
connecting elements positioned at different locations along the
length of the antenna elements in accordance with one or more
embodiments of the disclosure;
[0034] FIGS. 10C and 10D are graphs illustrating scattering
parameters for the FIGS. 10A and 10B antenna structures,
respectively;
[0035] FIG. 11 illustrates an antenna structure including
connecting elements having switches in accordance with one or more
embodiments of the disclosure;
[0036] FIG. 12 illustrates an antenna structure having a connecting
element with a filter coupled thereto in accordance with one or
more embodiments of the disclosure;
[0037] FIG. 13 illustrates an antenna structure having two
connecting elements with filters coupled thereto in accordance with
one or more embodiments of the disclosure;
[0038] FIG. 14 illustrates an antenna structure having a tunable
connecting element in accordance with one or more embodiments of
the disclosure;
[0039] FIG. 15 illustrates an antenna structure mounted on a PCB
assembly in accordance with one or more embodiments of the
disclosure;
[0040] FIG. 16 illustrates another antenna structure mounted on a
PCB assembly in accordance with one or more embodiments of the
disclosure;
[0041] FIG. 17 illustrates an alternate antenna structure that can
be mounted on a PCB assembly in accordance with one or more
embodiments of the disclosure;
[0042] FIG. 18A illustrates a three mode antenna structure in
accordance with one or more embodiments of the disclosure;
[0043] FIG. 18B is a graph illustrating the gain patterns for the
FIG. 18A antenna structure;
[0044] FIG. 19 illustrates an antenna and power amplifier combiner
application for an antenna structure in accordance with one or more
embodiments of the disclosure;
[0045] FIGS. 20A and 20B illustrate a multimode antenna structure
useable, e.g., in a WiMAX USB or ExpressCard/34 device in
accordance with one or more further embodiments of the subject
disclosure.
[0046] FIG. 20C illustrates a test assembly used to measure the
performance of the antenna of FIGS. 20A and 20B.
[0047] FIGS. 20D to 20J illustrate test measurement results for the
antenna of FIGS. 20A and 20B.
[0048] FIGS. 21A and 21B illustrate a multimode antenna structure
useable, e.g., in a WiMAX USB dongle in accordance with one or more
alternate embodiments of the subject disclosure.
[0049] FIGS. 22A and 22B illustrate a multimode antenna structure
useable, e.g., in a WiMAX USB dongle in accordance with one or more
alternate embodiments of the subject disclosure.
[0050] FIG. 23A illustrates a test assembly used to measure the
performance of the antenna of FIGS. 21A and 21B.
[0051] FIGS. 23B to 23K illustrate test measurement results for the
antenna of FIGS. 21A and 21B.
[0052] FIG. 24 is a schematic block diagram of an antenna structure
with a beam steering mechanism in accordance with one or more
embodiments of the subject disclosure.
[0053] FIGS. 25A to 25G illustrate test measurement results for the
antenna of FIG. 25A.
[0054] FIG. 26 illustrates the gain advantage of an antenna
structure in accordance with one or more embodiments of the subject
disclosure as a function of the phase angle difference between
feedpoints.
[0055] FIG. 27A is a schematic diagram illustrating a simple
dual-band branch line monopole antenna structure.
[0056] FIG. 27B illustrates current distribution in the FIG. 27A
antenna structure.
[0057] FIG. 27C is a schematic diagram illustrating a spurline band
stop filter.
[0058] FIGS. 27D and 27E are test results illustrating frequency
rejection in the FIG. 27A antenna structure.
[0059] FIG. 28 is a schematic diagram illustrating an antenna
structure with a band-rejection slot in accordance with one or more
embodiments of the subject disclosure.
[0060] FIG. 29A illustrates an alternate antenna structure with a
band-rejection slot in accordance with one or more embodiments of
the subject disclosure.
[0061] FIGS. 29B and 29C illustrate test measurement results for
the FIG. 29A antenna structure.
[0062] FIG. 30 depicts an illustrative embodiment of an antenna
structure in accordance with one or more embodiments;
[0063] FIG. 31 depicts an illustrative embodiment of a multiband
antenna structure in accordance with one or more embodiments;
[0064] FIGS. 32A, 32B and 32C illustrate tuning using discrete
selection of inductance to select antenna fundamental resonance
frequency in accordance with one or more embodiments;
[0065] FIGS. 33A, 33B and 33C illustrate tuning using discrete
selection of inductance to select fundamental resonance frequency
where a separate but co-located high band element is shown with
feed points F1H and F2H that allows for compatibility with RF
transceiver front end designs requiring separate low- and mid- or
low- and high-band connections to the antenna in accordance with
one or more embodiments;
[0066] FIGS. 34A, 34B and 34C illustrate tuning and filtering using
discrete selection of inductance to select antenna fundamental
resonance frequency in accordance with one or more embodiments;
[0067] FIGS. 35A, 35B and 35C illustrate tuning and filtering using
discrete selection of inductance to select fundamental resonance
frequency in accordance with one or more embodiments;
[0068] FIGS. 36A-36B depict illustrative embodiments of a near
field sensor;
[0069] FIG. 37 depicts illustrative embodiments for placement of
the near field sensor as a probe;
[0070] FIG. 38 depicts illustrative embodiments of environmental
use cases applied to the antenna structure of FIG. 27;
[0071] FIG. 39 depicts illustrative embodiments of return loss and
efficiency plots according to the environmental use cases of FIG.
38;
[0072] FIG. 40 depicts illustrative embodiments of Smith charts
according to the environmental use cases of FIG. 38;
[0073] FIG. 41 depicts illustrative embodiments of magnitude and
phase plots associated with one of the probes;
[0074] FIG. 42 depicts illustrative embodiments of magnitude and
phase plots associated with one of the probes;
[0075] FIG. 43 depicts illustrative embodiments of phase vs.
antenna frequency shift plots of the probes;
[0076] FIG. 44 depicts illustrative embodiments of probe power
shift vs. antenna radiated power shift plots of the probes;
[0077] FIG. 45 depicts illustrative embodiments of an antenna
structure and probe placements;
[0078] FIG. 46 depicts illustrative embodiments of a free space
resonance tuning use a Ltune (a tuning variable);
[0079] FIG. 47 depicts illustrative embodiments of efficiency gain
by tuning of the antenna structure based on measurements of one of
the probes of the antenna structure of FIG. 45;
[0080] FIG. 48 depicts illustrative embodiments of efficiency gain
by tuning of the antenna structure based on measurements of a
different one of the probes of the antenna structure of FIG.
45;
[0081] FIG. 49 depicts an illustrative embodiment of a near field
sensor;
[0082] FIG. 50 depicts an illustrative embodiment of the near field
sensor of FIG. 49 on a printed circuit board;
[0083] FIG. 51 depicts an illustrative embodiment of a first method
that can be applied to the subject disclosure;
[0084] FIG. 52 depicts an illustrative embodiment of a near field
sensor;
[0085] FIG. 53 depicts an illustrative embodiment of a phase
detector;
[0086] FIG. 54 depict illustrative phase error versus frequency and
power level plots resulting from the near field sensor embodiment
of FIG. 52;
[0087] FIG. 55 depicts an illustrative embodiment of a second
method that can be applied to the subject disclosure;
[0088] FIG. 56 depicts an illustrative embodiment of a reactance
sensor;
[0089] FIG. 57 depicts an illustrative embodiment of a third method
that can be applied to the subject disclosure;
[0090] FIG. 58 depicts an illustrative embodiment of a tunable
antenna;
[0091] FIG. 59 depicts illustrative embodiments of plots of the
tunable antenna of FIG. 58;
[0092] FIG. 60 depicts an illustrative embodiment of a fourth
method that can be applied to the subject disclosure;
[0093] FIG. 61 depicts an illustrative embodiment of a
communication device; and
[0094] FIG. 62 is a diagrammatic representation of a machine in the
form of a computer system within which a set of instructions, when
executed, may cause the machine to perform any one or more of the
methods described herein.
DETAILED DESCRIPTION
[0095] The subject disclosure describes, among other things,
illustrative embodiments for monitoring changes in an operating
frequency of an antenna and adjusting the operating frequency of
the antenna to mitigate such changes. Other embodiments are
described in the subject disclosure.
[0096] One embodiment of the subject disclosure includes a method
for measuring, by a circuit, from a first probe a first magnitude
of radiated energy by an antenna, where the first probe is placed
near the antenna, obtaining, by the circuit, a second magnitude of
a signal supplied to the antenna, comparing, by the circuit, the
first and the second magnitudes, detecting, by the circuit, an
offset in an operating frequency of the antenna based on a
difference between the first and the second magnitudes, and
adjusting, by the circuit, the operating frequency of the antenna
to mitigate the offset in the operating frequency of the
antenna.
[0097] One embodiment of the subject disclosure includes an antenna
structure having a first antenna element for receiving and
transmitting radio frequency signals within an operating frequency
range, a first aperture tuner for adjusting an operating frequency
of the antenna element, and a first near field sensor for sensing
radiated energy from the first antenna element. The first near
field sensor, the first antenna element, and the first aperture
tuner can be coupled to a circuit that performs operations
comprising measuring from the first near field sensor a first
magnitude of radiated energy by the first antenna element,
obtaining a second magnitude of a signal supplied to the first
antenna element, comparing the first and the second magnitudes,
detecting a change in an operating frequency of the first antenna
element based on a difference between the first and the second
magnitudes, and directing the first aperture tuner to adjust the
operating frequency of the first antenna element to counter the
change in the operating frequency of the first antenna element.
[0098] One embodiment of the subject disclosure includes a
communication device having a near field sensor coupled to the
antenna structure, and a circuit coupled to the antenna structure
and the near field sensor. The circuit can perform operations
including measuring from the near field sensor a first magnitude of
radiated energy by the antenna structure, obtaining a second
magnitude of a signal supplied to the antenna structure by a
transmitter circuit, comparing the first and the second magnitudes,
detecting an offset in an operating frequency of the antenna
structure based on a difference between the first and the second
magnitudes, and adjusting the operating frequency of the antenna
structure to mitigate the offset.
[0099] One embodiment of the subject disclosure includes a method
for measuring, by a circuit, from a first probe a first phase of
radiated energy by an antenna, wherein the first probe is placed
near the antenna, measuring, by the circuit, from a second probe a
second phase of a transmit signal supplied to the antenna, wherein
the second probe is placed in a transmission path of the transmit
signal, comparing, by the circuit, the first and the second phases
to generate a phase differential, detecting, by the circuit, an
offset in an operating frequency of the antenna based on the phase
differential, and adjusting, by the circuit, the operating
frequency of the antenna to mitigate the offset in the operating
frequency of the antenna.
[0100] One embodiment of the subject disclosure includes an antenna
structure including a first antenna element, a first aperture tuner
for adjusting an operating frequency of the antenna element, a
probe, and a first near field sensor for sensing radiated energy
from the first antenna element. The first near field sensor and the
first aperture tuner can be coupled to a circuit that performs
operations including measuring from the first near field sensor a
first phase of radiated energy by the first antenna element,
measuring from the probe a second phase of a first signal supplied
to the first antenna element, comparing the first and the second
phases to generate a first phase differential, detecting a change
in an operating frequency of the first antenna element based on the
phase differential, and directing the first aperture tuner to
adjust the operating frequency of the first antenna element
according to the first phase differential.
[0101] One embodiment of the subject disclosure includes a
communication device having an antenna structure, a near field
sensor, a probe, and a circuit coupled to the antenna structure and
the near field sensor. The circuit can perform operations including
measuring from the near field sensor a first signal representing
radiated energy from the antenna structure, measuring from the
probe a second signal supplied to the antenna structure,
determining a phase differential from a first phase of the first
signal and a second phase of the second signal, detecting a
frequency offset of the antenna structure based on the phase
differential, and adjusting an operating frequency of the antenna
structure to mitigate the frequency offset.
[0102] One embodiment of the subject disclosure includes a method
for measuring, by a circuit, a change in reactance of an antenna,
determining, by the circuit, a frequency offset of the antenna
based on a change in an operating frequency of the antenna
according to the change in reactance of the antenna, and adjusting,
by the circuit, the operating frequency of the antenna to mitigate
the frequency offset of the antenna.
[0103] One embodiment of the subject disclosure includes an antenna
structure having a first antenna element, and a sensor coupled to
the first antenna element. The sensor can be coupled to a circuit
that performs operations including measuring from the sensor a
change in a reactance of the antenna, obtaining impedance
characteristics of the antenna, determining a change in an
operating frequency of the antenna according to the change in
reactance of the antenna and the impedance characteristics of the
antenna, and adjusting the operating frequency of the antenna to
counteract the change in the operating frequency of the
antenna.
[0104] One embodiment of the subject disclosure includes a
communication device having an antenna structure, a sensor, and a
circuit coupled to the sensor. The circuit can perform operations
including measuring a change in a reactance of the antenna,
determining a frequency offset of the antenna according to the
change in reactance of the antenna, and adjusting an operating
frequency of the antenna to reduce the frequency offset of the
antenna.
[0105] One embodiment of the subject disclosure includes a method
for determining, by a circuit, a magnitude difference between a
first signal supplied to an antenna and a second signal radiated by
the antenna, determining, by the circuit, a phase difference
between the first signal supplied to the antenna and the second
signal radiated by the antenna, measuring, by the circuit, a change
in reactance of an antenna, detecting, by the circuit, an offset in
an operating frequency of the antenna based on one of the magnitude
difference, the phase difference, the change in reactance, or any
combination thereof, and adjusting, by the circuit, a resonant
frequency of the antenna to mitigate the offset in the operating
frequency of the antenna.
[0106] One embodiment of the subject disclosure includes an antenna
structure having a first antenna element, a first aperture tuner
for adjusting a resonant frequency of the antenna element,
a probe, a reactive sensor, and a first near field sensor for
sensing radiated energy from the first antenna element. The first
near field sensor and the first aperture tuner can be coupled to a
circuit that performs operations including measuring from the first
near field sensor a first phase and a first magnitude of radiated
energy by the first antenna element, measuring from the probe a
second phase and a second magnitude of a first signal supplied to
the first antenna element, comparing the first and the second
phases to generate a phase differential, comparing the first and
the second magnitudes to generate a magnitude differential,
measuring a change in reactance of the first antenna, detecting a
change in an operating frequency of the first antenna element based
on one of the phase differential, the magnitude differential, the
change in reactance of the antenna, or any combination thereof, and
causing the first aperture tuner to adjust the resonant frequency
of the first antenna element according to the change in the
operating frequency of the first antenna.
[0107] One embodiment of the subject disclosure includes a
communication device having an antenna structure, a near field
sensor, a probe, a reactive sensor, and a circuit coupled to the
near field sensor, probe and the reactive sensor. The circuit can
perform operations including measuring from the near field sensor a
first signal representing radiated energy from the antenna
structure, measuring from the probe a second signal supplied to the
antenna structure, measuring from the reactive sensor a reactive
load of the antenna structure, determining a phase differential
from a first phase of the first signal and a second phase of the
second signal, determining a magnitude differential from a first
magnitude of the first signal and a second magnitude of the second
signal, determining a change in reactance of the antenna structure
from the reactive load, detecting a frequency offset of the antenna
structure based on the phase differential, the magnitude
differential, the change in reactance, or any combination thereof,
and adjusting an operating frequency of the antenna structure to
mitigate the frequency offset.
[0108] One embodiment of the subject disclosure includes a method
for adjusting, by a circuit comprising a processor and an antenna,
a reactive element of the antenna to reduce or increase an
operating bandwidth of the antenna, and tuning, by the circuit, the
antenna to a new resonant frequency to accommodate transmitting or
receiving RF signals at a desired band of operation, wherein the
antenna at the new resonant frequency has the reduced or increased
operating bandwidth of the antenna.
[0109] One embodiment of the subject disclosure includes a method
for adjusting, by a circuit comprising a processor and an antenna,
a quality factor of the antenna to reduce or increase an operating
bandwidth of the antenna, and tuning, by the circuit, the antenna
to a new resonant frequency to accommodate transmitting or
receiving RF signals at a desired band of operation, wherein the
antenna at the new resonant frequency has the reduced or increased
operating bandwidth of the antenna.
[0110] One embodiment of the subject disclosure includes a
communication device having an antenna structure and a circuit
coupled to the antenna structure. The circuit can perform
operations including obtaining a desired bandwidth of operation of
the antenna structure, adjusting a bandwidth of the antenna to
achieve the desired bandwidth of operation of the antenna
structure, and tuning the antenna to a new resonant frequency to
accommodate transmitting or receiving RF signals at a desired band
of operation, wherein the antenna at the new resonant frequency is
at least approximately at the desired bandwidth of operation of the
antenna structure.
[0111] Antenna structures in accordance with various embodiments of
the disclosure are particularly useful in communications devices
that require multiple antennas to be packaged close together (e.g.,
less than a quarter of a wavelength apart), including in devices
where more than one antenna is used simultaneously and within the
same frequency band or multiple frequency bands in cases where
carrier aggregation is required. Common examples of such devices in
which the antenna structures can be used include portable
communications products such as cellular handsets, PDAs, smart
phones, tablets, and wireless networking devices or data cards for
PCs or other equipment integrated communication devices such as
automobiles, trucks, or other vehicle categories. The antenna
structures are also useful with system architectures such as MIMO
and standard protocols for mobile wireless communications devices
(such as 802.11n for wireless LAN, and 3G and 4G data
communications such as 802.16e (WiMAX), HSDPA, 1.times.EVDO, LTE)
that require multiple antennas operating simultaneously. The
embodiments of the subject disclosure can be applied to future
generations of wireless communication protocols such as 5G.
[0112] FIGS. 1A-1G illustrate the operation of an antenna structure
100. FIG. 1A schematically illustrates the antenna structure 100
having two parallel antennas, in particular parallel dipoles 102,
104, of length L. The dipoles 102, 104 are separated by a distance
d, and are not connected by any connecting element. The dipoles
102, 104 have a fundamental resonant frequency that corresponds
approximately to L=.lamda./2. Each dipole is connected to an
independent transmit/receive system, which can operate at the same
frequency. This system connection can have the same characteristic
impedance z.sub.0 for both antennas, which in this example is 50
ohms.
[0113] When one dipole is transmitting a signal, some of the signal
being transmitted by the dipole will be coupled directly into the
neighboring dipole. The maximum amount of coupling generally occurs
near the half-wave resonant frequency of the individual dipole and
generally increases as the separation distance d is made smaller.
For example, for d<.lamda./3, the magnitude of coupling is
greater than 0.1 or -10 dB, and for d<.lamda./8, the magnitude
of the coupling is greater than -5 dB.
[0114] It is desirable to have no coupling (i.e., complete
isolation) or to reduce the coupling (i.e., at least reduced
isolation) between the antennas. If the coupling is, e.g., -10 dB,
10 percent of the transmit power is lost due to that amount of
power being directly coupled into the neighboring antenna. There
may also be detrimental system effects such as saturation or
desensitization of a receiver connected to the neighboring antenna
or degradation of the performance of a transmitter connected to the
neighboring antenna. Currents induced on the neighboring antenna
distort the gain pattern compared to that generated by an
individual dipole. This effect is known to reduce the correlation
between the gain patterns produced by the dipoles. Thus, while
coupling may provide some pattern diversity, it has detrimental
system impacts as described above.
[0115] Because of the close coupling, the antennas do not act
independently and can be considered an antenna system having two
pairs of terminals or ports that correspond to two different gain
patterns. Use of either port involves substantially the entire
structure including both dipoles. The parasitic excitation of the
neighboring dipole enables diversity to be achieved at close dipole
spacing, but currents excited on the dipole pass through the source
impedance, and therefore manifest mutual coupling between
ports.
[0116] FIG. 1C illustrates a model dipole pair corresponding to the
antenna structure 100 shown in FIG. 1 used for simulations. In this
example, the dipoles 102, 104 have a square cross section of 1
mm.times.1 mm and length (L) of 56 mm. These dimensions yield a
center resonant frequency of 2.45 GHz when attached to a 50-ohm
source. The free-space wavelength at this frequency is 122 mm. A
plot of the scattering parameters S11 and S21 for a separation
distance (d) of 10 mm, or approximately .lamda./12, is shown in
FIG. 1D. Due to symmetry and reciprocity, S22=S11 and S12=S21. For
simplicity, only S11 and S21 are shown and discussed. In this
configuration, the coupling between dipoles as represented by S21
reaches a maximum of -3.7 dB.
[0117] FIG. 1E shows the ratio (identified as "Magnitude I2/I1" in
the figure) of the vertical current on dipole 104 of the antenna
structure to that on dipole 102 under the condition in which port
106 is excited and port 108 is passively terminated. The frequency
at which the ratio of currents (dipole 104/dipole 102) is a maximum
corresponds to the frequency of 180 degree phase differential
between the dipole currents and is just slightly higher in
frequency than the point of maximum coupling shown in FIG. 1D.
[0118] FIG. 1F shows azimuthal gain patterns for several
frequencies with excitation of port 106. The patterns are not
uniformly omni-directional and change with frequency due to the
changing magnitude and phase of the coupling. Due to symmetry, the
patterns resulting from excitation of port 108 would be the mirror
image of those for port 106. Therefore, the more asymmetrical the
pattern is from left to right, the more diverse the patterns are in
terms of gain magnitude.
[0119] Calculation of the correlation coefficient between patterns
provides a quantitative characterization of the pattern diversity.
FIG. 1G shows the calculated correlation between port 106 and port
108 antenna patterns. The correlation is much lower than is
predicted by Clark's model for ideal dipoles. This is due to the
differences in the patterns introduced by the mutual coupling.
[0120] FIGS. 2A-2F illustrate the operation of an exemplary two
port antenna structure 200 in accordance with one or more
embodiments of the disclosure. The two port antenna structure 200
includes two closely-spaced resonant antenna elements 202, 204 and
provides both low pattern correlation and low coupling between
ports 206, 208. FIG. 2A schematically illustrates the two port
antenna structure 200. This structure is similar to the antenna
structure 100 comprising the pair of dipoles shown in FIG. 1B, but
additionally includes horizontal conductive connecting elements
210, 212 between the dipoles on either side of the ports 206, 208.
The two ports 206, 208 are located in the same locations as with
the FIG. 1 antenna structure. When one port is excited, the
combined structure exhibits a resonance similar to that of the
unattached pair of dipoles, but with a significant reduction in
coupling and an increase in pattern diversity.
[0121] An exemplary model of the antenna structure 200 with a 10 mm
dipole separation is shown in FIG. 2B. This structure has generally
the same geometry as the antenna structure 100 shown in FIG. 1C,
but with the addition of the two horizontal connecting elements
210, 212 electrically connecting the antenna elements slightly
above and below the ports. This structure shows a strong resonance
at the same frequency as unattached dipoles, but with very
different scattering parameters as shown in FIG. 2C. There is a
deep drop-out in coupling, below -20 dB, and a shift in the input
impedance as indicated by S11. In this example, the best impedance
match (S11 minimum) does not coincide with the lowest coupling (S21
minimum). A matching network can be used to improve the input
impedance match and still achieve very low coupling as shown in
FIG. 2D. In this example, a lumped element matching network
comprising a series inductor followed by a shunt capacitor was
added between each port and the structure.
[0122] FIG. 2E shows the ratio (indicated as "Magnitude I2/I1" in
the figure) of the current on dipole element 204 to that on dipole
element 202 resulting from excitation of port 206. This plot shows
that below the resonant frequency, the currents are actually
greater on dipole element 204. Near resonance, the currents on
dipole element 204 begin to decrease relative to those on dipole
element 202 with increasing frequency. A point of low coupling
(2.44 GHz in this case) occurs near the frequency where currents on
both dipole elements are generally equal in magnitude. At this
frequency, the phase of the currents on dipole element 204 lag
those of dipole element 202 by approximately 160 degrees.
[0123] Unlike the FIG. 1C dipoles without connecting elements, the
currents on antenna element 204 of the FIG. 2B combined antenna
structure 200 are not forced to pass through the terminal impedance
of port 208. Instead a resonant mode is produced where the current
flows down antenna element 204, across the connecting element 210,
212, and up antenna element 202 as indicated by the arrows shown on
FIG. 2A. (Note that this current flow is representative of one half
of the resonant cycle; during the other half, the current
directions are reversed). The resonant mode of the combined
structure features the following: (1) the currents on antenna
element 204 largely bypass port 208, thereby allowing for high
isolation between the ports 206, 208, and (2) the magnitude of the
currents on both antenna elements 202,204 are approximately equal,
which allows for dissimilar and uncorrelated gain patterns as
described in further detail below.
[0124] Because the magnitude of currents is nearly equal on the
antenna elements, a much more directional pattern is produced (as
shown on FIG. 2F) than in the case of the FIG. 1C antenna structure
100 with unattached dipoles. When the currents are equal, the
condition for nulling the pattern in the x (or phi=0) direction is
for the phase of currents on dipole 204 to lag those of dipole 202
by the quantity .pi.-kd (where k=2.pi./.lamda., and .lamda. is the
effective wavelength). Under this condition, fields propagating in
the phi=0 direction from dipole 204 will be 180 degrees out of
phase with those of dipole 202, and the combination of the two will
therefore have a null in the phi=0 direction.
[0125] In the model example of FIG. 2B, d is 10 mm or an effective
electrical length of .lamda./12. In this case, kd equates .pi./6 or
30 degrees, and so the condition for a directional azimuthal
radiation pattern with a null towards phi=0 and maximum gain
towards phi=180 is for the current on dipole 204 to lag those on
dipole 202 by 150 degrees. At resonance, the currents pass close to
this condition (as shown in FIG. 2E), which explains the
directionality of the patterns. In the case of the excitation of
port 204, the radiation patterns are the mirror opposite of those
of FIG. 2F, and maximum gain is in the phi=0 direction. The
difference in antenna patterns produced from the two ports has an
associated low predicted envelope correlation as shown on FIG. 2G.
Thus the combined antenna structure has two ports that are isolated
from each other and produce gain patterns of low correlation.
[0126] Accordingly, the frequency response of the coupling is
dependent on the characteristics of the connecting elements 210,
212, including their impedance and electrical length. In accordance
with one or more embodiments of the disclosure, the frequency or
bandwidth over which a desired amount of isolation can be
maintained is controlled by appropriately configuring the
connecting elements. One way to configure the cross connection is
to change the physical length of the connecting element. An example
of this is shown by the multimode antenna structure 300 of FIG. 3A
where a meander has been added to the cross connection path of the
connecting elements 310, 312. This has the general effect of
increasing both the electrical length and the impedance of the
connection between the two antenna elements 302, 304. Performance
characteristics of this structure including scattering parameters,
current ratios, gain patterns, and pattern correlation are shown on
FIGS. 3B, 3C, 3D, and 3E, respectively. In this embodiment, the
change in physical length has not significantly altered the
resonant frequency of the structure, but there is a significant
change in S21, with larger bandwidth and a greater minimum value
than in structures without the meander. Thus, it is possible to
optimize or improve the isolation performance by altering the
electrical characteristic of the connecting elements.
[0127] Exemplary multimode antenna structures in accordance with
various embodiments of the disclosure can be designed to be excited
from a ground or counterpoise 402 (as shown by antenna structure
400 in FIG. 4), or as a balanced structure (as shown by antenna
structure 500 in FIG. 5). In either case, each antenna structure
includes two or more antenna elements (402, 404 in FIG. 4, and 502,
504 in FIG. 5) and one or more electrically conductive connecting
elements (406 in FIG. 4, and 506, 508 in FIG. 5). For ease of
illustration, only a two-port structure is illustrated in the
example diagrams. However, it is possible to extend the structure
to include more than two ports in accordance with various
embodiments of the disclosure. A signal connection to the antenna
structure, or port (418, 412 in FIGS. 4 and 510, 512 in FIG. 5), is
provided at each antenna element. The connecting element provides
electrical connection between the two antenna elements at the
frequency or frequency range of interest. Although the antenna is
physically and electrically one structure, its operation can be
explained by considering it as two independent antennas. For
antenna structures not including a connecting element such as
antenna structure 100, port 106 of that structure can be said to be
connected to antenna 102, and port 108 can be said to be connected
to antenna 104. However, in the case of this combined structure
such as antenna structure 400, port 418 can be referred to as being
associated with one antenna mode, and port 412 can be referred to
as being associated with another antenna mode.
[0128] The antenna elements are designed to be resonant at the
desired frequency or frequency range of operation. The lowest order
resonance occurs when an antenna element has an electrical length
of one quarter of a wavelength. Thus, a simple element design is a
quarter-wave monopole in the case of an unbalanced configuration.
It is also possible to use higher order modes. For example, a
structure formed from quarter-wave monopoles also exhibits dual
mode antenna performance with high isolation at a frequency of
three times the fundamental frequency. Thus, higher order modes may
be exploited to create a multiband antenna. Similarly, in a
balanced configuration, the antenna elements can be complementary
quarter-wave elements as in a half-wave center-fed dipole. However,
the antenna structure can also be formed from other types of
antenna elements that are resonant at the desired frequency or
frequency range. Other possible antenna element configurations
include, but are not limited to, helical coils, wideband planar
shapes, chip antennas, meandered shapes, loops, and inductively
shunted forms such as Planar Inverted-F Antennas (PIFAs).
[0129] The antenna elements of an antenna structure in accordance
with one or more embodiments of the disclosure need not have the
same geometry or be the same type of antenna element. The antenna
elements should each have resonance at the desired frequency or
frequency range of operation.
[0130] In accordance with one or more embodiments of the
disclosure, the antenna elements of an antenna structure have the
same geometry. This is generally desirable for design simplicity,
especially when the antenna performance requirements are the same
for connection to either port.
[0131] The bandwidth and resonant frequencies of the combined
antenna structure can be controlled by the bandwidth and resonance
frequencies of the antenna elements. Thus, broader bandwidth
elements can be used to produce a broader bandwidth for the modes
of the combined structure as illustrated, e.g., in FIGS. 6A, 6B,
and 6C. FIG. 6A illustrates a multimode antenna structure 600
including two dipoles 602, 604 connected by connecting elements
606, 608. The dipoles 602, 604 each have a width (W) and a length
(L) and are spaced apart by a distance (d). FIG. 6B illustrates the
scattering parameters for the structure having exemplary
dimensions: W=1 mm, L=57.2 mm, and d=10 mm. FIG. 6C illustrates the
scattering parameters for the structure having exemplary
dimensions: W=10 mm, L=50.4 mm, and d=10 mm. As shown, increasing W
from 1 mm to 10 mm, while keeping the other dimensions generally
the same, results in a broader isolation bandwidth and impedance
bandwidth for the antenna structure.
[0132] It has also been found that increasing the separation
between the antenna elements increases the isolation bandwidth and
the impedance bandwidth for an antenna structure.
[0133] In general, the connecting element is in the high-current
region of the combined resonant structure. It may therefore be
desirable for the connecting element to have a high
conductivity.
[0134] The ports are located at the feed points of the antenna
elements as they would be if they were operated as separate
antennas. Matching elements or structures may be used to match the
port impedance to the desired system impedance.
[0135] In accordance with one or more embodiments of the
disclosure, the multimode antenna structure can be a planar
structure incorporated, e.g., into a printed circuit board, as
shown as FIG. 7. In this example, the antenna structure 700
includes antenna elements 702, 704 connected by a connecting
element 706 at ports 708, 710. The antenna structure is fabricated
on a printed circuit board substrate 712. The antenna elements
shown in the figure are simple quarter-wave monopoles. However, the
antenna elements can be any geometry that yields an equivalent
effective electrical length.
[0136] In accordance with one or more embodiments of the
disclosure, antenna elements with dual resonant frequencies can be
used to produce a combined antenna structure with dual resonant
frequencies and hence dual operating frequencies. FIG. 8A shows an
exemplary model of a multimode dipole structure 800 where the
dipole antenna elements 802, 804 are split into two fingers 806,
808 and 810, 812, respectively, of unequal length. The dipole
antenna elements have resonant frequencies associated with each the
two different finger lengths and accordingly exhibit a dual
resonance. Similarly, the multimode antenna structure using
dual-resonant dipole arms exhibits two frequency bands where high
isolation (or small S21) is obtained as shown in FIG. 8B.
[0137] In accordance with one or more embodiments of the
disclosure, a multimode antenna structure 900 shown in FIG. 9 is
provided having variable length antenna elements 902, 904 forming a
tunable antenna. This may be done by changing the effective
electrical length of the antenna elements by a controllable device
such as an RF switch 906, 908 at each antenna element 902, 904. In
this example, the switch may be opened (by operating the
controllable device) to create a shorter electrical length (for
higher frequency operation) or closed to create a longer electrical
length (for lower frequency of operation). The operating frequency
band for the antenna structure 900, including the feature of high
isolation, can be tuned by tuning both antenna elements in concert.
This approach may be used with a variety of methods of changing the
effective electrical length of the antenna elements including,
e.g., using a controllable dielectric material, loading the antenna
elements with a variable capacitor such as a microelectromechanical
systems (MEMs) device, varactor, or tunable dielectric capacitor,
and switching on or off parasitic elements.
[0138] In accordance with one or more embodiments of the
disclosure, the connecting element or elements provide an
electrical connection between the antenna elements with an
electrical length approximately equal to the electrical distance
between the elements. Under this condition, and when the connecting
elements are attached at the port ends of the antenna elements, the
ports are isolated at a frequency near the resonance frequency of
the antenna elements. This arrangement can produce nearly perfect
isolation at particular frequency.
[0139] Alternately, as previously discussed, the electrical length
of the connecting element may be increased to expand the bandwidth
over which isolation exceeds a particular value. For example, a
straight connection between antenna elements may produce a minimum
S21 of -25 dB at a particular frequency and the bandwidth for which
S21<-10 dB may be 100 MHz. By increasing the electrical length,
a new response can be obtained where the minimum S21 is increased
to -15 dB but the bandwidth for which S21<-10 dB may be
increased to 150 MHz.
[0140] Various other multimode antenna structures in accordance
with one or more embodiments of the disclosure are possible. For
example, the connecting element can have a varied geometry or can
be constructed to include components to vary the properties of the
antenna structure. These components can include, e.g., passive
inductor and capacitor elements, resonator or filter structures, or
active components such as phase shifters.
[0141] In accordance with one or more embodiments of the
disclosure, the position of the connecting element along the length
of the antenna elements can be varied to adjust the properties of
the antenna structure. The frequency band over which the ports are
isolated can be shifted upward in frequency by moving the point of
attachment of the connecting element on the antenna elements away
from the ports and towards the distal end of the antenna elements.
FIGS. 10A and 10B illustrate multimode antenna structures 1000,
1002, respectively, each having a connecting element electrically
connected to the antenna elements. In the FIG. 10A antenna
structure 1000, the connecting element 1004 is located in the
structure such that the gap between the connecting element 1004 and
the top edge of the ground plane 1006 is 3 mm. FIG. 10C shows the
scattering parameters for the structure showing that high isolation
is obtained at a frequency of 1.15 GHz in this configuration. A
shunt capacitor/series inductor matching network is used to provide
the impedance match at 1.15 GHz. FIG. 10D shows the scattering
parameters for the structure 1002 of FIG. 10B, where the gap
between the connecting element 1008 and the top edge 1010 of the
ground plane is 19 mm. The antenna structure 1002 of FIG. 10B
exhibits an operating band with high isolation at approximately
1.50 GHz.
[0142] FIG. 11 schematically illustrates a multimode antenna
structure 1100 in accordance with one or more further embodiments
of the disclosure. The antenna structure 1100 includes two or more
connecting elements 1102, 1104, each of which electrically connects
the antenna elements 1106, 1108. (For ease of illustration, only
two connecting elements are shown in the figure. It should be
understood that use of more than two connecting elements is also
contemplated.) The connecting elements 1102, 1104 are spaced apart
from each other along the antenna elements 1106, 1108. Each of the
connecting elements 1102, 1104 includes a switch 1112, 1110. Peak
isolation frequencies can be selected by controlling the switches
1110, 1112. For example, a frequency f1 can be selected by closing
switch 1110 and opening switch 1112. A different frequency f2 can
be selected by closing switch 1112 and opening switch 1110.
[0143] FIG. 12 illustrates a multimode antenna structure 1200 in
accordance with one or more alternate embodiments of the
disclosure. The antenna structure 1200 includes a connecting
element 1202 having a filter 1204 operatively coupled thereto. The
filter 1204 can be a low pass or band pass filter selected such
that the connecting element connection between the antenna elements
1206, 1208 is only effective within the desired frequency band,
such as the high isolation resonance frequency. At higher
frequencies, the structure will function as two separate antenna
elements that are not coupled by the electrically conductive
connecting element, which is open circuited.
[0144] FIG. 13 illustrates a multimode antenna structure 1300 in
accordance with one or more alternate embodiments of the
disclosure. The antenna structure 1300 includes two or more
connecting elements 1302, 1304, which include filters 1306, 1308,
respectively. (For ease of illustration, only two connecting
elements are shown in the figure. It should be understood that use
of more than two connecting elements is also contemplated.) In one
possible embodiment, the antenna structure 1300 has a low pass
filter 1308 on the connecting element 1304 (which is closer to the
antenna ports) and a high pass filter 1306 on the connecting
element 1302 in order to create an antenna structure with two
frequency bands of high isolation, i.e., a dual band structure.
[0145] FIG. 14 illustrates a multimode antenna structure 1400 in
accordance with one or more alternate embodiments of the
disclosure. The antenna structure 1400 includes one or more
connecting elements 1402 having a tunable element 1406 operatively
connected thereto. The antenna structure 1400 also includes antenna
elements 1408, 1410. The tunable element 1406 alters the delay or
phase of the electrical connection or changes the reactive
impedance of the electrical connection. The magnitude of the
scattering parameters S21/S12 and a frequency response are affected
by the change in electrical delay or impedance and so an antenna
structure can be adapted or generally optimized for isolation at
specific frequencies using the tunable element 1406.
[0146] FIG. 15 illustrates a multimode antenna structure 1500 in
accordance with one or more alternate embodiments of the
disclosure. The multimode antenna structure 1500 can be used, e.g.,
in a WIMAX USB dongle. The antenna structure 1500 can be configured
for operation, e.g., in WiMAX bands from 2300 to 2700 MHz.
[0147] The antenna structure 1500 includes two antenna elements
1502, 1504 connected by a conductive connecting element 1506. The
antenna elements include slots to increase the electrical length of
the elements to obtain the desired operating frequency range. In
this example, the antenna structure is optimized for a center
frequency of 2350 MHz. The length of the slots can be reduced to
obtain higher center frequencies. The antenna structure is mounted
on a printed circuit board assembly 1508. A two-component lumped
element match is provided at each antenna feed.
[0148] The antenna structure 1500 can be manufactured, e.g., by
metal stamping. It can be made, e.g., from 0.2 mm thick copper
alloy sheet. The antenna structure 1500 includes a pickup feature
1510 on the connecting element at the center of mass of the
structure, which can be used in an automated pick-and-place
assembly process. The antenna structure is also compatible with
surface-mount reflow assembly.
[0149] FIG. 16 illustrates a multimode antenna structure 1600 in
accordance with one or more alternate embodiments of the
disclosure. As with antenna structure 1500 of FIG. 15, the antenna
structure 1600 can also be used, e.g., in a WIMAX USB dongle. The
antenna structure can be configured for operation, e.g., in WiMAX
bands from 2300 to 2700 MHz.
[0150] The antenna structure 1600 includes two antenna elements
1602, 1604, each comprising a meandered monopole. The length of the
meander determines the center frequency. The exemplary design shown
in the figure is optimized for a center frequency of 2350 MHz. To
obtain higher center frequencies, the length of the meander can be
reduced.
[0151] A connecting element 1606 electrically connects the antenna
elements. A two-component lumped element match is provided at each
antenna feed.
[0152] The antenna structure can be fabricated, e.g., from copper
as a flexible printed circuit (FPC) mounted on a plastic carrier
1608. The antenna structure can be created by the metalized
portions of the FPC. The plastic carrier provides mechanical
support and facilitates mounting to a PCB assembly 1610.
Alternatively, the antenna structure can be formed from
sheet-metal.
[0153] FIG. 17 illustrates a multimode antenna structure 1700 in
accordance with another embodiment of the disclosure. This antenna
design can be used, e.g., for USB, Express 34, and Express 54 data
card formats. The exemplary antenna structure shown in the figure
is designed to operate at frequencies from 2.3 to 6 GHz. The
antenna structure can be fabricated, e.g., from sheet-metal or by
FPC over a plastic carrier 1702.
[0154] FIG. 18A illustrates a multimode antenna structure 1800 in
accordance with another embodiment of the disclosure. The antenna
structure 1800 comprises a three mode antenna with three ports. In
this structure, three monopole antenna elements 1802, 1804, 1806
are connected using a connecting element 1808 comprising a
conductive ring that connects neighboring antenna elements. The
antenna elements are balanced by a common counterpoise, or sleeve
1810, which is a single hollow conductive cylinder. The antenna has
three coaxial cables 1812, 1814, 1816 for connection of the antenna
structure to a communications device. The coaxial cables 1812,
1814, and 1816 pass through the hollow interior of the sleeve 1810.
The antenna assembly may be constructed from a single flexible
printed circuit wrapped into a cylinder and may be packaged in a
cylindrical plastic enclosure to provide a single antenna assembly
that takes the place of three separate antennas. In one exemplary
arrangement, the diameter of the cylinder is 10 mm and the overall
length of the antenna is 56 mm so as to operate with high isolation
between ports at 2.45 GHz. This antenna structure can be used,
e.g., with multiple antenna radio systems such as MIMO or 802.11N
systems operating in the 2.4 to 2.5 GHz bands. In addition to port
to port isolation, each port advantageously produces a different
gain pattern as shown on FIG. 18B. While this is one specific
example, it is understood that this structure can be scaled to
operate at any desired frequency. It is also understood that
methods for tuning, manipulating bandwidth, and creating multiband
structures described previously in the context of two-port antennas
can also apply to this multiport structure.
[0155] While the above embodiment is shown as a true cylinder, it
is possible to use other arrangements of three antenna elements and
connecting elements that produce the same advantages. This
includes, but is not limited to, arrangements with straight
connections such that the connecting elements form a triangle, or
another polygonal geometry. It is also possible to construct a
similar structure by similarly connecting three separate dipole
elements instead of three monopole elements with a common
counterpoise. Also, while symmetric arrangement of antenna elements
advantageously produces equivalent performance from each port,
e.g., same bandwidth, isolation, impedance matching, it is also
possible to arrange the antenna elements asymmetrically or with
unequal spacing depending on the application.
[0156] FIG. 19 illustrates use of a multimode antenna structure
1900 in a combiner application in accordance with one or more
embodiments of the disclosure. As shown in the figure, transmit
signals may be applied to both antenna ports of the antenna
structure 1900 simultaneously. In this configuration, the multimode
antenna can serve as both antenna and power amplifier combiner. The
high isolation between antenna ports restricts interaction between
the two amplifiers 1902, 1904, which is known to have undesirable
effects such as signal distortion and loss of efficiency. Optional
impedance matching at 1906 can be provided at the antenna
ports.
[0157] FIGS. 20A and 20B illustrate a multimode antenna structure
2000 in accordance with one or more alternate embodiments of the
subject disclosure. The antenna structure 2000 can also be used,
e.g., in a WiMAX USB or ExpressCard/34 device. The antenna
structure can be configured for operation, e.g., in WiMAX bands
from 2300 to 6000 MHz.
[0158] The antenna structure 2000 includes two antenna elements
2001, 2004, each comprising a broad monopole. A connecting element
2002 electrically connects the antenna elements. Slots (or other
cut-outs) 2005 are used to improve the input impedance match above
5000 MHz. The exemplary design shown in the figure is optimized to
cover frequencies from 2300 to 6000 MHz.
[0159] The antenna structure 2000 can be manufactured, e.g., by
metal stamping. It can be made, e.g., from 0.2 mm thick copper
alloy sheet. The antenna structure 2000 includes a pickup feature
2003 on the connecting element 2002 generally at the center of mass
of the structure, which can be used in an automated pick-and-place
assembly process. The antenna structure is also compatible with
surface-mount reflow assembly. Feed points 2006 of the antenna
provide the points of connection to the radio circuitry on a PCB,
and also serve as a support for structural mounting of the antenna
to the PCB. Additional contact points 2007 provide structural
support.
[0160] FIG. 20C illustrates a test assembly 2010 used to measure
the performance of antenna 2000. The figure also shows the
coordinate reference for far-field patterns. Antenna 2000 is
mounted on a 30.times.88 mm PCB 2011 representing an ExpressCard/34
device. The grounded portion of the PCB 2011 is attached to a
larger metal sheet 2012 (having dimensions of 165.times.254 mm in
this example) to represent a counterpoise size typical of a
notebook computer. Test ports 2014, 2016 on the PCB 2011 are
connected to the antenna through 50-ohm striplines.
[0161] FIG. 20D shows the VSWR measured at test ports 2014, 2016.
FIG. 20E shows the coupling (S21 or S12) measured between the test
ports. The VSWR and coupling are advantageously low across the
broad range of frequencies, e.g., 2300 to 6000 MHz. FIG. 20F shows
the measured radiation efficiency referenced from the test ports
2014 (Port 1), 2016 (Port 2). FIG. 20G shows the calculated
correlation between the radiation patterns produced by excitation
of test port 2014 (Port 1) versus those produced by excitation of
test port 2016 (Port 2). The radiation efficiency is advantageously
high while the correlation between patterns is advantageously low
at the frequencies of interest. FIG. 20H shows far field gain
patterns by excitation of test port 2014 (Port 1) or test port 2016
(Port 2) at a frequency of 2500 MHz. FIGS. 201 and 20J show the
same pattern measurements at frequencies of 3500 and 5200 MHz,
respectively. The patterns resulting from test port 2014 (Port 1)
are different and complementary to those of test port 2016 (Port 2)
in the .phi.=0 or XZ plane and in the .theta.=90 or XY plane.
[0162] FIGS. 21A and 21B illustrate a multimode antenna structure
2100 in accordance with one or more alternate embodiments of the
subject disclosure. The antenna structure 2100 can also be used,
e.g., in a WiMAX USB dongle. The antenna structure can be
configured for operation, e.g., in WiMAX bands from 2300 to 2400
MHz.
[0163] The antenna structure 2100 includes two antenna elements
2102, 2104, each comprising a meandered monopole. The length of the
meander determines the center frequency. Other tortuous
configurations such as, e.g., helical coils and loops, can also be
used to provide a desired electrical length. The exemplary design
shown in the figure is optimized for a center frequency of 2350
MHz. A connecting element 2106 (shown in FIG. 21B) electrically
connects the antenna elements 2102, 2104. A two-component lumped
element match is provided at each antenna feed.
[0164] The antenna structure can be fabricated, e.g., from copper
as a flexible printed circuit (FPC) 2103 mounted on a plastic
carrier 2101. The antenna structure can be created by the metalized
portions of the FPC 2103. The plastic carrier 2101 provides
mounting pins or pips 2107 for attaching the antenna to a PCB
assembly (not shown) and pips 2105 for securing the FPC 2103 to the
carrier 2101. The metalized portion of 2103 includes exposed
portions or pads 2108 for electrically contacting the antenna to
the circuitry on the PCB.
[0165] To obtain higher center frequencies, the electrical length
of the elements 2102, 2104 can be reduced. FIGS. 22A and 22B
illustrate a multimode antenna structure 2200, the design of which
is optimized for a center frequency of 2600 MHz. The electrical
length of the elements 2202, 2204 is shorter than that of elements
2102, 2104 of FIGS. 21A and 21B because metallization at the end of
the elements 2202, 2204 has been removed, and the width of the of
the elements at feed end has been increased.
[0166] FIG. 23A illustrates a test assembly 2300 using antenna 2100
of FIGS. 21A and 21B along with the coordinate reference for
far-field patterns. FIG. 23B shows the VSWR measured at test ports
2302 (Port 1), 2304 (Port 2). FIG. 23C shows the coupling (S21 or
S12) measured between the test ports 2302 (Port 1), 2304 (Port 2).
The VSWR and coupling are advantageously low at the frequencies of
interest, e.g., 2300 to 2400 MHz. FIG. 23D shows the measured
radiation efficiency referenced from the test ports. FIG. 23E shows
the calculated correlation between the radiation patterns produced
by excitation of test port 2302 (Port 1) versus those produced by
excitation of test port 2304 (Port 2). The radiation efficiency is
advantageously high while the correlation between patterns is
advantageously low at the frequencies of interest. FIG. 23F shows
far field gain patterns by excitation of test port 2302 (Port 1) or
test port 2304 (Port 2) at a frequency of 2400 MHz. The patterns
resulting from test port 2302 (Port 1) are different and
complementary to those of test port 2304 (Port 2) in the .phi.=0 or
XZ plane and in the .theta.=90 or XY plane.
[0167] FIG. 23G shows the VSWR measured at the test ports of
assembly 2300 with antenna 2200 in place of antenna 2100. FIG. 23H
shows the coupling (S21 or S12) measured between the test ports.
The VSWR and coupling are advantageously low at the frequencies of
interest, e.g. 2500 to 2700 MHz. FIG. 23I shows the measured
radiation efficiency referenced from the test ports. FIG. 23J shows
the calculated correlation between the radiation patterns produced
by excitation of test port 2302 (Port 1) versus those produced by
excitation of test port 2304 (Port 2). The radiation efficiency is
advantageously high while the correlation between patterns is
advantageously low at the frequencies of interest. FIG. 23K shows
far field gain patterns by excitation of test port 2302 (Port 1) or
test port 2304 (Port 2) at a frequency of 2600 MHz. The patterns
resulting from test port 2302 (Port 1) are different and
complementary to those of test port 2304 (Port 2) in the .phi.=0 or
XZ plane and in the .theta.=90 or XY plane.
[0168] One or more further embodiments of the subject disclosure
are directed to techniques for beam pattern control for the purpose
of null steering or beam pointing. When such techniques are applied
to a conventional array antenna (comprising separate antenna
elements that are spaced at some fraction of a wavelength), each
element of the array antenna is fed with a signal that is a phase
shifted version of a reference signal or waveform. For a uniform
linear array with equal excitation, the beam pattern produced can
be described by the array factor F, which depends on the phase of
each individual element and the inter-element element spacing
d.
F = A 0 n = 0 N - 1 exp ( j n ( .beta. d cos .theta. + .alpha. ) ]
##EQU00001##
where .beta.=2.pi./.lamda., N=Total # of elements, .alpha.=phase
shift between successive elements, and .theta.=angle from array
axis
[0169] By controlling the phase .alpha. to a value .alpha..sub.i,
the maximum value of F can be adjusted to a different direction
.theta..sub.i, thereby controlling the direction in which a maximum
signal is broadcast or received.
[0170] The inter-element spacing in conventional array antennas is
often on the order of 1/4 wavelength, and the antennas can be
closely coupled, having nearly identical polarization. It is
advantageous to reduce the coupling between elements, as coupling
can lead to several problems in the design and performance of array
antennas. For example, problems such as pattern distortion and scan
blindness (see Stutzman, Antenna Theory and Design, Wiley 1998, pgs
122-128 and 135-136, and 466-472) can arise from excessive
inter-element coupling, as well as a reduction of the maximum gain
attainable for a given number of elements.
[0171] Beam pattern control techniques can be advantageously
applied to all multimode antenna structures described herein having
antenna elements connected by one or more connecting elements,
which exhibit high isolation between multiple feedpoints. The phase
between ports at the high isolation antenna structure can be used
for controlling the antenna pattern. It has been found that a
higher peak gain is achievable in given directions when the antenna
is used as a simple beam-forming array as a result of the reduced
coupling between feedpoints. Accordingly, greater gain can be
achieved in selected directions from a high isolation antenna
structure in accordance with various embodiments that utilizes
phase control of the carrier signals presented to its feed
terminals.
[0172] In handset applications where the antennas are spaced at
much less than 1/4 wavelength, mutual coupling effects in
conventional antennas reduce the radiation efficiency of the array,
and therefore reduce the maximum gain achievable.
[0173] By controlling the phase of the carrier signal provided to
each feedpoint of a high isolation antenna in accordance with
various embodiments, the direction of maximum gain produced by the
antenna pattern can be controlled. A gain advantage of, e.g., 3 dB
obtained by beam steering is advantageous particularly in portable
device applications where the beam pattern is fixed and the device
orientation is randomly controlled by the user. As shown, e.g., in
the schematic block diagram of FIG. 24, which illustrates a pattern
control apparatus 2400 in accordance with various embodiments, a
relative phase shift .alpha. is applied by a phase shifter 2402 to
the RF signals applied to each antenna feed 2404, 2408. The signals
are fed to respective antenna ports of antenna structure 2410.
[0174] The phase shifter 2402 can comprise standard phase shift
components such as, e.g., electrically controlled phase shift
devices or standard phase shift networks.
[0175] FIGS. 25A-25G provide a comparison of antenna patterns
produced by a closely spaced 2-D conventional array of dipole
antennas and a 2-D array of high isolation antennas in accordance
with various embodiments of the subject disclosure for different
phase differences a between two feeds to the antennas. In FIGS.
25A-25G, curves are shown for the antenna patterns at .theta.=90
degrees. The solid lines in the figures represents the antenna
pattern produced by the isolated feed single element antenna in
accordance with various embodiments, while the dashed lines
represent the antenna pattern produced by two separate monopole
conventional antennas separated by a distance equal to the width of
the single element isolated feed structure. Therefore, the
conventional antenna and the high isolation antenna are of
generally equivalent size.
[0176] In all cases shown in the figures, the peak gain produced by
the high isolation antenna in accordance with various embodiments
produces a greater gain margin when compared to the two separate
conventional dipoles, while providing azimuthal control of the beam
pattern. This behavior makes it possible to use the high isolation
antenna in transmit or receive applications where additional gain
is needed or desired in a particular direction. The direction can
be controlled by adjusting the relative phase between the
drivepoint signals. This may be particularly advantageous for
portable devices needing to direct energy toward a receive point
such as, e.g., a base station. The combined high isolation antenna
offers greater advantage when compared to two single conventional
antenna elements when phased in a similar fashion.
[0177] As shown in FIG. 25A, the combined dipole in accordance with
various embodiments shows greater gain in a uniform azimuth pattern
(.theta.=90) for .alpha.=0 (zero degrees phase difference).
[0178] As shown in FIG. 25B, the combined dipole in accordance with
various embodiments shows greater peak gain (at .phi.=0) with a
non-symmetric azimuthal pattern (.theta.=90 plot for .alpha.=30 (30
degrees phase difference between feedpoints).
[0179] As shown in FIG. 25C, the combined dipole in accordance with
various embodiments shows greater peak gain (at .phi.=0) with a
shifted azimuthal pattern (.theta.=90 plot for .alpha.=60 (60
degrees phase difference between feedpoints).
[0180] As shown in FIG. 25D, the combined dipole in accordance with
various embodiments shows even greater peak gain (at .phi.=0) with
a shifted azimuthal pattern (.theta.=90 plot for .alpha.=90 (90
degrees phase difference between feedpoints).
[0181] As shown in FIG. 25E, the combined dipole in accordance with
various embodiments shows greater peak gain (at .phi.=0) with a
shifted azimuthal pattern (.theta.=90 plot greater backlobe (at
.phi.=180) for .alpha.=120 (120 degrees phase difference between
feedpoints).
[0182] As shown in FIG. 25F, the combined dipole in accordance with
various embodiments shows greater peak gain (at .phi.=0) with a
shifted azimuthal pattern (.theta.=90 plot), even greater backlobe
(at .phi.=180) for .alpha.=150 (150 degrees phase difference
between feedpoints).
[0183] As shown in FIG. 25G, the combined dipole in accordance with
various embodiments shows greater peak gain (at .phi.=0&180)
with a double lobed azimuthal pattern (.theta.=90 plot) for
.alpha.=180 (180 degrees phase difference between feedpoints).
[0184] FIG. 26 illustrates the ideal gain advantage if the combined
high isolation antenna in accordance with one or more embodiments
over two separate dipoles as a function of the phase angle
difference between the feedpoints for a two feedpoint antenna
array.
[0185] Further embodiments of the subject disclosure are directed
to multimode antenna structures that provide increased high
isolation between multi-band antenna ports operating in close
proximity to each other at a given frequency range. In these
embodiments, a band-rejection slot is incorporated in one of the
antenna elements of the antenna structure to provide reduced
coupling at the frequency to which the slot is tuned.
[0186] FIG. 27A schematically illustrates a simple dual-band branch
line monopole antenna 2700. The antenna 2700 includes a
band-rejection slot 2702, which defines two branch resonators 2704,
2706. The antenna is driven by signal generator 2708. Depending on
the frequency at which the antenna 2700 is driven, various current
distributions are realized on the two branch resonators 2704,
2706.
[0187] The physical dimensions of the slot 2702 are defined by the
width Ws and the length Ls as shown in FIG. 27A. When the
excitation frequency satisfies the condition of Ls=lo/4, the slot
feature becomes resonant. At this point the current distribution is
concentrated around the shorted section of the slot, as shown in
FIG. 27B.
[0188] The currents flowing through the branch resonators 2704,
2706 are approximately equal and oppositely directed along the
sides of the slot 2702. This causes the antenna structure 2700 to
behave in a similar manner to a spurline band stop filter 2720
(shown schematically in FIG. 27C), which transforms the antenna
input impedance down significantly lower than the nominal source
impedance. This large impedance mismatch results in a very high
VSWR, shown in FIGS. 27D and 27E, and as a result leads to the
desired frequency rejection.
[0189] This band-rejection slot technique can be applied to an
antenna system with two (or more) antennas elements operating in
close proximity to each other where one antenna element needs to
pass signals of a desired frequency and the other does not. In one
or more embodiments, one of the two antenna elements includes a
band-rejection slot, and the other does not. FIG. 28 schematically
illustrates an antenna structure 2800, which includes a first
antenna element 2802, a second antenna element 2804, and a
connecting element 2806. The antenna structure 2800 includes ports
2808 and 2810 at antenna elements 2802 and 2804, respectively. In
this example, a signal generator drives the antenna structure 2802
at port 2808, while a meter is coupled to the port 2810 to measure
current at port 2810. It should be understood, however, that either
or both ports can be driven by signal generators. The antenna
element 2802 includes a band-rejection slot 2812, which defines two
branch resonators 2814, 2816. In this embodiment, the branch
resonators comprise the main transmit section of the antenna
structure, while the antenna element 2804 comprises a diversity
receive portion of the antenna structure.
[0190] Due to the large mismatch at the port of the antenna element
2802 with the band-reject slot 2812, the mutual coupling between it
and the diversity receive antenna element 2804, which is actually
matched at the slot resonant frequency will be quite small and will
result in relatively high isolation.
[0191] FIG. 29A is a perspective view of a multimode antenna
structure 2900 comprising a multi-band diversity receive antenna
system that utilizes the band-rejection slot technique in the GPS
band in accordance with one or more further embodiments of the
subject disclosure. (The GPS band is 1575.42 MHz with 20 MHz
bandwidth.) The antenna structure 2900 is formed on a flex film
dielectric substrate 2902, which is formed as a layer on a
dielectric carrier 2904. The antenna structure 2900 includes a GPS
band rejection slot 2906 on the primary transmit antenna element
2908 of the antenna structure 2900. The antenna structure 2900 also
includes a diversity receive antenna element 2910, and a connecting
element 2912 connecting the diversity receive antenna element 2910
and the primary transmit antenna element 2908. A GPS receiver (not
shown) is connected to the diversity receive antenna element 2910.
In order to generally minimize the antenna coupling from the
primary transmit antenna element 2908 and to generally maximize the
diversity antenna radiation efficiency at these frequencies, the
primary antenna element 2908 includes the band-rejection slot 2906
and is tuned to an electrical quarter wave length near the center
of the GPS band. The diversity receive antenna element 2910 does
not contain such a band rejection slot, but comprises a GPS antenna
element that is properly matched to the main antenna source
impedance so that there will be generally maximum power transfer
between it and the GPS receiver. Although both antenna elements
2908, 2910 co-exist in close proximity, the high VSWR due to the
slot 2906 at the primary transmit antenna element 2908 reduces the
coupling to the primary antenna element source resistance at the
frequency to which the slot 2906 is tuned, and therefore provides
isolation at the GPS frequency between both antenna elements 2908,
2910. The resultant mismatch between the two antenna elements 2908,
2910 within the GPS band is large enough to decouple the antenna
elements in order to meet the isolation requirements for the system
design as shown in FIGS. 29B and 29C.
[0192] In the antenna structures described herein in accordance
with various embodiments of the subject disclosure, the antenna
elements and the connecting elements can form a single integrated
radiating structure such that a signal fed to either port excites
the entire antenna structure to radiate as a whole, rather than
separate radiating structures. As such, the techniques described
herein provide isolation of the antenna ports without the use of
decoupling networks at the antenna feed points.
[0193] Other embodiments disclosed herein are directed to an
antenna that separates the fundamental (low band) resonance from
the high band resonance by using two separate structures, which are
connected at the feedpoint--thus accomplishing the goal of
achieving a MIMO or Diversity antenna with each feed exhibiting a
multiband capability, and whereby each feed is optimally isolated
from the opposite feed. By way of a non-limiting illustration, in
some implementations, high band frequencies can range from 1710 to
2170 MHz, and low band frequencies can range from 698 to 960
MHz.
[0194] In one or more embodiments of the antenna structures
described in the subject disclosure, electrical currents flowing
through neighboring antenna elements 3002 and 3004 (see FIG. 30)
can be configured to be similar in magnitude, such that an antenna
mode excited by one antenna port (e.g., Port 1) is approximately
electrically isolated from an antenna mode excited by another
antenna port (e.g., Port 2) at a given desired signal frequency
range. In one embodiment, this can be accomplished by configuring
antennas 3002 and 3004 with a connecting element 3006 to enable
common and difference mode currents, which when summed together
result in some or a substantial amount of isolation between ports 1
and 2. Configuring an antenna structure to control differential and
common mode current to cause isolation between any number of
antenna ports can be applied to any of the antenna embodiments
described herein.
[0195] FIG. 31 illustrates an exemplary multiband antenna 3100 in
accordance with one or more embodiments. The antenna 3100 can
include a low band structure comprising two low band antenna
elements 3102, 3104 connected by a connecting element 3106. A fixed
or variable reactive element 3126 such as a fixed or variable
inductor L is provided in the connecting element 3106 to provide
control (reduction) of the mutual coupling between feedpoints for
the low band element by varying the electrical length of the
connecting element 3106 in accordance with the disclosures of U.S.
Pat. No. 7,688,273, the disclosure of which is incorporated by
reference herein in its entirety. Similarly, a connecting element
3116 can be provided between the high band antenna elements 3112,
3114. A fixed or variable reactive element 3136 such as a fixed or
variable inductor L can be provided in the connecting element 3116
to provide control (reduction) of the mutual coupling between
feedpoints for the low band element by varying the electrical
length of the connecting element 3116 in accordance with the
disclosures of U.S. Pat. No. 7,688,273.
[0196] The high band structure comprising two high band antenna
elements 3112, 3114 can be connected to the low band structure at
feed points f1, f2. Two filters 3142 and 3144 are provided in the
high band antenna elements 3112, 3114 for blocking low band
frequencies, thereby isolating the high band antenna elements 3112,
3114 from the low band antenna elements 3102, 3104. The filters
3142 and 3144 can be passive or programmable pass band filters. In
the present illustration the filters 3142 and 3144 can represent
high pass filters implemented with a capacitor and/or other
components to achieve desired high pass filtering characteristics.
To achieve similar isolation with the low band structure, the low
band antenna elements 3102, 3104 can be configured with filters
3152, 3154 to block high band frequencies, thereby isolating the
high band antenna elements 3112, 3114 from the low band antenna
elements 3102, 3104. The filters 3152, 3154 can be passive or
programmable pass band filters. In the present illustration the
filters 3152, 3154 can represent low pass filters implemented with
reactive and passive components that achieve desired low pass
filtering characteristics.
[0197] By having a structure associated with low band resonance and
a separate structure associated with high band resonance, the low
band structure can be advantageously designed or optimized
independently of the high band structure and vice-versa. A further
advantage is that the low band or high band structures may
separately take on different antenna design realizations, e.g.,
monopole, loop, Planar Inverted "F" antenna (PIFA), etc. allowing
the designer to select the best option for the electrical and
mechanical design requirements. In one exemplary embodiment, the
low band structure may be a monopole, while the high band structure
may be a PIFA.
[0198] A separate network is provided for each structure. The low
band structure can use a fixed or variable inductive bridge 3126 as
an interconnecting element 3106. The high band element is fed from
the common feedpoint, but with a high pass network 3142, 3144--the
simplest being a series capacitor with low reactance at the high
band frequencies and higher reactance at the low band frequencies.
In addition, the low band antenna elements 3102, 3104 can be
configured with variable reactive components 3122, 3124 to perform
aperture tuning which enables shifting of the low band resonance
frequency of the low band structure. The reactive components 3122,
3124 can be independently controlled so that the resonance
frequency of low band antenna element 3102 can be independently
controlled from the low band resonance frequency of low band
antenna element 3104. The reactive components 3122, 3124 can be
represented by switched inductors which can be aggregated or
reduced to vary the electrical length of the low band antenna
elements 3102, 3104, respectively.
[0199] Similarly, the high band antenna elements 3112, 3114 can be
configured with variable reactive components 3132, 3134 to perform
aperture tuning which enables shifting of the high band resonance
frequency of the high band structure. The reactive components 3132,
3134 can be independently controlled so that the resonance
frequency of high band antenna element 3112 can be independently
controlled from the high band resonance frequency of high band
antenna element 3114. The reactive components 3132, 3134 can also
be represented by switched inductors which can be aggregated or
reduced to vary the electrical length of the high band antenna
elements 3112, 3114, respectively.
[0200] The aforementioned structures, enable high band tuning to be
performed relatively independent of low band tuning, providing a
simpler design process and better performance than antennas not
having such separate structures. Other more complex networks may
also be used advantageously to separate the interdependence of the
high and low band structures still using a common feedpoint for a
MIMO branch such as shown in FIG. 31. The method illustrated in
FIG. 31 is not limited to 2.times.2, 2.times.1 MIMO or 2 feed
antennas used for diversity applications, and may be extended to
higher branch order MIMO antennas, e.g., 3.times.3, etc.
[0201] A number of factors affect antenna performance in a hand
held mobile communication device. While these factors are related,
they generally fall into one of three categories; antenna size,
mutual coupling between multiple antennas, and device usage models.
The size of an antenna is dependent on three criteria; bandwidth of
operation, frequency of operation, and required radiation
efficiency. Bandwidth requirements have obviously increased as they
are driven by FCC frequency allocations in the US and carrier
roaming agreements around the world. Different regions use
different frequency bands, now with over 40 E-UTRA band
designations-many overlapping but requiring world capable wireless
devices to typically cover a frequency range from 698 to 2700
MHz.
[0202] A simple relationship exists between the bandwidth, size,
and radiation efficiency for the fundamental or lowest frequency
resonance of a physically small antenna.
.DELTA. f f .varies. ( a .lamda. ) 3 .eta. - 1 ( 1 )
##EQU00002##
[0203] Here a is the radius of a sphere containing the antenna and
its associated current distribution. Since a is normalized to the
operating wavelength, the formula may be interpreted as "fractional
bandwidth is proportional to the wavelength normalized modal
volume". The radiation efficiency .eta. is included as a factor on
the right side of (1), indicating that greater bandwidth, is
achievable by reducing the efficiency. Radio frequency currents
exist not only on the antenna element but also on the attached
conductive structure or "counterpoise". For instance, mobile phone
antennas in the 698-960 MHz bands use the entire PCB as a radiating
structure so that the physical size of the antenna according to (1)
is actually much larger than what appears to be the "antenna". The
"antenna" may be considered a resonator that is electromagnetically
coupled to the PCB so that it excites currents over the entire
conductive structure or chassis. Most smartphones exhibit
conductive chassis dimensions of approximately 70.times.130 mm,
which from an electromagnetic modal analysis predicts a fundamental
mode near 1 GHz suggesting that performance bandwidth degrades
progressively at lower excitation frequencies. The
efficiency-bandwidth trade-off is complex requiring E-M simulation
tools for accurate prediction. Results indicate that covering
698-960 MHz (Bands 12, 13, 17, 18, 19, 20, 5 and 8) with a
completely passive antenna with desirable antenna size and geometry
becomes difficult without making sacrifices in radiation
efficiency.
[0204] Factors determining the achievable radiation efficiency are
not entirely obvious, as the coupling coefficient between the
"antenna" and the chassis; radiative coupling to lossy components
on the PCB; dielectric absorption in plastic housing, coupling to
co-existing antennas; as well as losses from finite resistance
within the "antenna" resonator structure, all play a part. In most
cases, the requirements imposed by operators suggest minimum
radiation efficiencies of 40-50%, so that meeting a minimum TRP
requirement essentially requires tradeoffs between the power
amplifier (PA) output and the achievable antenna efficiency. In
turn, poor efficiency at the antenna translates to less battery
life, as the PA must compensate for the loss.
[0205] Prior to concerns over band aggregation, wireless devices
operated on one band at a time with need to change when roaming.
Consequently, the required instantaneous bandwidth would be
considerably less than that required to address worldwide
compatibility. Take a 3G example for instance, where operation in
band 5 from (824-894 MHz) compared to operation in bands 5 plus 8
(824-960 MHz). Then, add the requirements for band 13 and band 17
and the comparison becomes more dramatic--824-960 vs. 698-960 MHz.
This becomes a problematic as legacy phone antennas support
pentaband operation but only bands 5 and band 8. Given equation (1)
several choices exist. The most obvious would be to increase the
antenna system size, (i.e. the antenna and phone chassis footprint)
and/or to reduce the radiation efficiency. Since 4G smartphones
require 2 antennas, neither approach is necessarily desirable from
an industrial design standpoint, although it is possible to cover
the 700-2200 MHz bands with a completely passive antenna in a space
allocation of 6.5.times.10.times.60 mm.
[0206] Various alternative antenna configurations are the
following: limit the antenna(s) instantaneous bandwidth within
current antenna space allocations to allow use of 1 or more
antennas without compromising the industrial design (Antenna
Supplier motivation); make the antenna(s) smaller to achieve a
compact and sleek device with greater functionality by limiting the
instantaneous bandwidth with same or improved antenna efficiency
(OEM motivation); improve the antenna efficiency, and therefore the
network performance by controlling the antenna instantaneous
frequency/tuning (Operator motivation); make the antenna agile to
adapt to different usage models (OEM/User/Operator motivation); or
combinations of the above.
[0207] The simplest approach can be to limit the instantaneous
operation to a single band to satisfy the protocol requirements for
a single region. To satisfy the roaming requirements, the antenna
could be made frequency agile on a band-by-band basis. This
approach represents the most basic type of "state-tuned"
antenna.
[0208] Various embodiments disclosed herein are directed to an
antenna that separates the fundamental (low band) resonance from
the high band resonance by using two separate structures, which are
connected at the feedpoint--thus accomplishing the goal of
achieving a MIMO or Diversity antenna with each feed exhibiting a
multiband capability, and whereby each feed is optimally isolated
from the opposite feed. By way of non-limiting example, in some
implementations, high band frequencies can range from 1710 to 2700
MHz, and low band frequencies can range from 500 to 960 MHz.
[0209] The exemplary embodiments allow for tuning of the first
resonance of the antenna to accommodate multiple operational bands
depending on a tuning state, and broadband operation on the high
bands (e.g., 1710-2170 MHz, or 1710-2700 MHz) independent of the
low band tuning state.
[0210] Referring to FIG. 32A, an example is shown that is
illustrative of single low band-multiple high band aggregation
compatibility. The high band radiation efficiency in this case can
remain essentially the same independent of the low band tuning
state, but the low band resonance frequency is able to be tuned in
discrete frequency increments according to the equivalent
electrical length, as selected by the series inductance Lvar which
is shown in FIG. 32B. The variable inductance can be created using
discrete reactive elements such as inductors and a switching
mechanism such as an SP4T switch. The configuration as shown yields
3 different inductances depending on which state the switch is in:
(state 1) LVAR=L3+L4+L5 (switch connects to pole 1 or 4); (state 2)
LVAR=Lpath2.parallel.L3+L4+L5 or approximately L4+L5 (switch
connects to pole 2); or (state 3) LVAR=Lpath3.parallel.(L3+L4)+L5
OR approximately L5 (switch connects to pole 3). In this
embodiment, Lpath2 and Lpath3 refer to the equivalent inductances
of the circuit paths through the switch. Keeping the inductors
close to the switch can minimize or otherwise reduce the path
inductances such that the discrete inductors are essentially
shorted out by the switch.
[0211] The antenna incorporates a main structure that has a
fundamental resonance at the lowest frequency band. The solution
employs a multiband antenna having 3 low band tuning states as
shown in FIG. 32B. State 1 includes a low band (fundamental)
resonance suited for LTE 700 (698-742 MHz) operation: State 2
includes a low band resonance suited to GSM 850 (824-894 MHz)
operation, and state 3 a low band resonance suited to GSM 900
(880-960 MHz).
[0212] The high band resonance (1710-2170 MHz) can be reasonably
independent of the tuning state for the low band by nature of the
separation of the low and high band radiating elements from the
feedpoints. The low band tuning can be accomplished by switching
different reactive components in between the feedpoint and the
radiating structure. The high band operation of the antenna can be
governed primarily by the auxiliary radiating section at the
terminus of the capacitor opposite the feedpoint. The capacitor
functions primarily as a high pass filter to decouple the feedpoint
from the high and low bands portions of the antenna. In this way,
signals at different operating bands can be directed to the
appropriate radiating section of the combined antenna. The high
band resonance can be determined in part by the electrical length
of the high band portion of the antenna (indicated in the
illustration by horizontal conductive segments). In other
embodiments, the capacitor may be a highpass, bandpass, or tunable
filter. In a similar manner, the path from the feedpoint to the low
band radiating portion of the antenna may include a low pass,
bandpass or tunable filter.
[0213] Tuning can be accomplished using a switching device such one
capable of SP4T operation. In one embodiment, a solid state
silicon-based FET switch can be used in each leg of the antenna to
alter the series inductance presented to the antenna feedpoint,
thereby lowering the resonant frequency as a function of the amount
of inductance added. Although inductors are used in this
embodiment, other reactive components may also be used for the
purpose of altering the electrical length of the low band portion
of the antenna radiating structure including capacitive elements.
The switch may be of various types such as a mechanical MEMS type
device, a voltage/current controlled variable device, and so forth.
The switch may also be configured with multiple poles and with any
throw capability needed to select the number of tuning states
required for antenna operation. The number of throws can establish
the number of tuning states possible, which in turn is dictated by
the number of frequency bands to be supported. While three states
are shown in the illustrated embodiment, any number of states can
be utilized corresponding to any number of frequency bands or
ranges. In one embodiment, a pair of adjustable reactive elements
(e.g., fixed inductors coupled with switching mechanisms) can be
coupled with corresponding pairs of feedpoints, and the tuning can
be performed by settings each of the adjustable reactive elements
to the same tuning state among the group of tuning states.
[0214] Referring to FIG. 33A, a separate but co-located high band
element is shown with feed points F1H and F2H that allows for
compatibility with RF transceiver front end designs requiring
separate low- and mid- or low- and high-band connections to the
antenna. The variable inductance can be created using discrete
inductors and a SP4T switch as shown. The configuration as shown
yields 3 different inductances depending on which state the switch
is in: (state 1) LVAR=L3+L4+L5 (switch connects to pole 1 or 4);
(state 2) LVAR=Lpath2.parallel.L3+L4+L5 or approximately L4+L5
(switch connects to pole 2); or (state 3)
LVAR=Lpath3.parallel.(L3+L4)+L5 OR approx. L5 (switch connects to
pole 3). Lpath2 and Lpath3 refer to the equivalent inductances of
the circuit paths through the switch. Keeping the inductors close
to the switch minimizes the path inductances such that the discrete
inductors are essentially shorted out by the switch.
[0215] The exemplary antennas can provide better radiation
efficiency and/or smaller size compared to an untuned antenna by
nature of the tuning to each band of operation separately. The
reactive elements (e.g., inductors and their associated inductance)
can establish the electrical length of the low band elements, and
therefore can provide for adjusting the low band resonance
(tuning). Referring additionally to FIGS. 34A-35B, antenna
structures that enable tuning to each band of operation separately
while also providing for desired filtering through use of
low-pass-filters and high-pass-filters as illustrated. It should be
noted the embodiments for aperture tuning shown in FIGS. 32B, 33B,
34B and 35B can replaced with other suitable embodiments such as
those shown in FIGS. 32C, 33C, 34C and 35C.
[0216] Further, the fundamental mode associated of the antenna low
band resonance can be tuned by adjustment of the electrical length
of the low band portion of the antenna via reactive elements which
may exhibit either inductive or capacitive characteristics. As
illustrated in FIG. 32A, discrete inductors are shown in a series
connection between the antenna feed points and the radiating
element end plates on the each side of the antenna, thereby
increasing the equivalent electrical length. The use of separate or
discrete components is intended to be illustrative of the
principle, but by no means limiting to scope of the subject
disclosure. In one or more embodiments, the techniques and/or
components of the exemplary embodiments described herein that
provide for antenna tuning can be utilized in conjunction with
techniques and/or components described with respect to U.S. Pat.
No. 7,688,273. In one or more embodiments, the techniques and/or
components of the exemplary embodiments described herein that
provide for antenna tuning can be utilized in conjunction with
techniques and/or components described with respect to U.S. patent
application Ser. No. ______, entitled "Antenna Structures and
Methods Thereof", filed on May 22, 2014, under attorney docket
number 5000-0150 (2013-03).
[0217] FIG. 36A depicts an illustrative embodiment of a near field
sensor 3620 utilized with other RF components of a communication
device 3600. In one embodiment near field sensor 3620 can comprise
a first log detector 3622 that receives signals from a directional
coupler 3604, a second log detector 3626 that receives signals from
a near field probe 3624. In another embodiment, such as shown in
FIG. 36B, the near field sensor 3620 can comprise a single log
detector 3626 that receives signals from one or more near field
probes 3624 selectable by a switch 3625. Instead of relying on a
directional coupler 3604 such as shown in FIG. 36A, the controller
3632 of FIG. 36B can be configured to determine a magnitude of a
forward feed signal supplied by the transmitter 3602 based on a
known state of the transmitter 3602, which can be determined from a
look-up table. That is, the controller 3632 may be aware that the
transmitter 3602 has been configured to transmit at a desired
magnitude as a result of configuring components of the transmitter
3602 such as an amplifier or otherwise. The controller 3632 can
utilize a look-up table comprising magnitudes indexed according to
known states of the transmitter 3602, or can determine the
magnitude of a transmit signal based on an algorithm implemented by
the controller 3632. The look-up table can be created by
characterizing the transmitter 3602 in a lab or manufacturing
setting. Accordingly, the embodiments that follow can be based on
either the embodiments of FIG. 36A or 36B.
[0218] The near field probe 3624 can be a small trace of metal
serving as a miniature antenna that can receive radiated energy
from an adaptive antenna 3610 and which can have a very small
parasitic effect (if any) on the adaptive antenna 3610, thereby not
affecting the original operating characteristics of the adaptive
antenna 3610. The near field probe 3624 can be located on a printed
circuit board (PCB), a housing assembly component or some other
suitable location of the communication device 3600 that enables
placement of the near field probe 3624 at a particular perspective
of the adaptive antenna 3610. As depicted in FIGS. 36A and 36B,
more than on near field probe 3624 can be used at various location
of the communication device 3600 as will be described below.
[0219] The output signals of the first and second log detectors
3622 and 3626 can be supplied to a difference circuit 3628 that
produces a difference signal supplied to an analog to digital
converter (A2D) 3630, which in turn supplies a digital value to a
controller 3632 for processing. The difference signal represents a
difference in magnitude between the signals supplied by the first
and second log detectors 3622 and 3626. The signal supplied by the
first log detector 3622 can represent a measure of a forward feed
signal supplied by an RF transmitter 3602. The difference in
magnitude between the forward feed signal and the signal measured
by the near field probe 3624 can be used to detect a change in a
resonance frequency of the adaptive antenna 3610. In the embodiment
of FIG. 36B, the controller 3632 would be supplied a digital value
from the A2D 3630 that corresponds only to the magnitude of the
near field signal generated by the log detector 3626. The
controller 3632 in turn can obtain a digital representation of the
magnitude of the forward feed (or transmit) signal supplied by the
transmitter 3602 by retrieving a magnitude value from a look-up
table based on a known state of the transmitter 3602 or it can
obtain it by way of a computational approach that determines the
magnitude from the known state of the transmitter 3602. Once the
forward feed magnitude is determined, the controller 3632 can
determine a difference between the forward feed and near field
signal magnitudes using digital signal processing techniques, and
thereby detect a change in a resonance frequency of the adaptive
antenna 3610.
[0220] The adaptive antenna 3610 can be an antenna structure such
as any of the embodiments described above. For instance, the
adaptive antenna 3610 can be represented by one of the embodiments
of FIGS. 31-35, which enable programming of the resonant frequency
of the antenna. If the detected change of the resonance frequency
of the adaptive antenna 3610 is determined to be undesirable based
on the difference signals supplied by the A2D 3630, the adaptive
antenna 3610 can be tuned by the controller 3632 by changing the
electrical length of the antenna by reconfiguring the structure of
the antenna (e.g., adding or removing antenna components), by
adding or removing reactive components such as the embodiments of
FIGS. 32B, 33B, 34B and 35B, or changing the reactance of a
component of the antenna (e.g., a variable capacitor or variable
inductor). As the adaptive antenna 3610 is tuned by the controller
3632, the controller 3632 can also be configured to program a
matching network 3608 or supply the tuning data of the adaptive
antenna 3610 to a separate controller of the matching network to
enable programming of the match network to adapt to changes applied
to the adaptive antenna 3610 while tuning its resonant frequency.
It is noted that the near field sensor 3620 as described herein can
be used with other antenna structures not described in the subject
disclosure.
[0221] What follows is an illustrative algorithm for tuning the
adaptive antenna 3610 based on near field RF power measurements. We
begin by assuming the adaptive antenna 3610 is tunable with a total
of N states. The adaptive antenna 3610 can be set to a particular
state, k, based on operating conditions such as a band of
operation. The A2D 3630 provides the controller 3632 a relative
power measurement, Y, at a specific transmit frequency in use. In
one embodiment, an objective is to improve Y. To do this, the k is
stepped at time interval, .tau., and the response of Y is used to
determine if the state should be incremented, decremented, or kept
the same. The time .tau. may be chosen to be longer than a response
time of other power control loops used by the communication device
3600 and/or other network operations that may be dependent on the
transmit power level from the communication device 3600. With this
in mind, a tuning algorithm can be described as follows:
[0222] Let k0 be the nominal setting for the state value
[0223] Let the value of k alternate between k0, k0+1 and k0-1:
[0224] k(n.tau.)=k0+cos(n.pi./2), where n is the number of time
increments
[0225] For each state, measure the probe response Y(n.tau.)
[0226] Calculate the slope, m=dY/dk
[0227] Calculate the running average mbar, over M time
intervals,
m=.SIGMA..sub.i=n-M.sup.nm(n.tau.)
[0228] Compare mbar to a threshold 6 to determine whether to change
the state
[0229] If mbar.gtoreq..delta. k=k-1
[0230] If mbar.ltoreq..delta. k=k+1
[0231] If |mbar|<.delta. k is unchanged
[0232] In an embodiment where multiple near field probes 3624 are
used, the above algorithm can be changed so that delta measurements
of Y are averaged over time. Averaging delta Y readings can
increase the reliability of the measurement. In one embodiment, the
controller 3632 can select an individual near field probe 3624
using a programmable multiplexer (S) 3625. The controller 3632 can
then apply the above algorithm using only those near field probes
3624 that provide a delta Y with desirable results. This approach
can be applied to each near field probe 3624 until one or more near
field probes 3624 are identified as providing desirable tuning
results of the adaptive antenna 3610. The controller 3632 can then
average the delta Y magnitudes measured for each near field probe
3624, or select only one of the identified near field probes 3624
for tuning the adaptive antenna 3610. In a multi-antenna system
where the transmitter 3602 can transmit from any one of a plurality
of adaptive antennas 3610, the near field sensor 3620 can be
configured to choose a near field probe 3624 closest to the antenna
used during transmission for performing the above algorithm. In
another embodiment, if for a selected near field probe 3624 the
near field sensor 3620 is not performing due to unrepresentative
information supplied by the selected near field probe 3624 (e.g., a
user's finger is changing the performance of the selected near
field probe 3624) then the controller 3632 can be adapted to avoid
the identified near field probe 3624 and perform averaging from
other probes unaffected by the environment effect.
[0233] In other embodiments averaging can be performed for more
than one near field probe 3624 associated with a single adaptive
antenna. In yet another embodiment, multiple near field probes 3624
can be placed on each radiating element of an antenna (high band,
mid band, and low band near field probes). In another embodiment,
one near field probe can be used per radiating element of an
antenna. In yet another embodiment, a pad of an integrated circuit
(IC) can be used as a near field probe 3624 of an antenna.
[0234] Multiple near field probes 3624 can be used at different
locations of the communication device 3600 using a multiplexer 3625
to select between near field probes 3624 to get a measurement of
radiated energy of the adaptive antenna 3610 from different
perspectives (e.g., bottom, top, sides). FIG. 37 provides a
placement illustration of six near field probes 3624 on a printed
circuit board. The near field probes 3624 of FIG. 37 are sized to
yield -30 dB of coupling with the antenna structure shown. Low band
simulations were performed on an antenna structure with a PCB
ground of 110 mm.times.60 mm. FIG. 38 depicts six use cases applied
to the circuit board of FIG. 37. The gray sections of each use case
represent body tissue or steel. The simulation further assumed a 2
mm air gap between a furthest model element and a log material.
[0235] FIG. 39 depicts return loss plots (S11) for steel and body
tissue and their respective efficiencies for each use case.
Compared with free space, the resonant frequency of the antenna
shifts from free space from -150 to +60 MHz. The efficiency changes
from -18 to -2 dB. The Smith charts of FIG. 40 depict steel and
body tissue plots of all probes for all use cases. FIG. 41 depicts
magnitude and phase response of probe 2 which is closest to an end
of the antenna. FIG. 42 depicts the magnitude and phase response of
probe 4 which is to the left of the antenna. In this latter
embodiment, the response is not well behaved, which illustrates
that not all probe locations may provide optimal results. FIG. 43
depicts phase shift versus antenna frequency shift plots for probe
2. These plots show that the phase shift experienced at probe 2
correlates with an observed frequency shift. The sensitivity is
about 0.24 degrees per MHz. FIG. 44 depicts power shift versus
antenna radiated power shift plots for probe 2. These plots show
that the magnitude shift experienced at probe 2 correlates worst
with an observed power shift. Probe 7 (opposite end of ground)
correlates best with the radiated power shift. The sensitivity is
about 0.28 degrees per radiated power shift.
[0236] FIGS. 45-48 depict simulations for an aperture tuned antenna
model. FIG. 45 depicts two embodiments having near field probes
placed near and far from an antenna located at one end of a PCB
having dimensions of 60 mm.times.126 mm. The antenna is designed
for a low band nominal frequency of 860 MHz. A variable inductor
(Ltune) can be changed to shift the resonant frequency of the
antenna. The models shown in FIG. 45 can be used to determine how
much improvement can be made by changing Ltune in several loading
scenarios.
[0237] FIG. 46 depicts free space resonance tuning using Ltune.
Return loss (S11) is presented in the pots of FIG. 46 as a function
of the value of Ltune. The simulations assume in free space a
nominal setting of L=5.6 nH at a resonant frequency of 860 MHz. The
simulation methodology included running each of the models of FIG.
45 with all of the loading cases shown in FIG. 38. The value of
Ltune is then swept for each loading case. A determination is then
made of the maximum radiated power at the desired operating
frequency (fc) over the sweep of Ltune. The maximum radiated power
is then compared based on metrics such as maximizing a magnitude of
S21 (power coupled to the near field probe), minimizing a magnitude
of S11, and normalizing S21 phase to 0 degrees (the normalized
phase is the measured S21 phase minus the reference phase delay at
resonance center determined from the free space model or by the
actual phase when the antenna is tuned to the desired
frequency).
[0238] FIG. 47 depicts an efficiency gain by optimization of tuning
for the case of the near field probe nearest the antenna in FIG.
45. FIG. 47 shows an improvement in radiated power from 0 to 2.2 dB
depending on loading case. It is further observed that an ideal
optimization of S21 or S11 results in nearly the same gain as
optimizing on radiated power itself. It is also observed that
optimizing on S21 phase also works well except in five of the six
loading cases. FIG. 48 depicts an efficiency gain by optimization
of tuning for the case of the probe opposite the antenna in FIG.
45. FIG. 48 shows again that optimization of S21 or S11 are
reliable. However, in this case, optimizing on S21 phase works
poorly in the situation where near field probe is far from the
antenna.
[0239] FIG. 49 depicts an illustrative implementation of a near
field sensor. In one embodiment, the near field sensor can comprise
a transmission line coupled to a directional coupler, an RF IC
(e.g., an Analog Devices AD8302) powered by a battery for measuring
amplitude and phase between a forward feed signal supplied by the
directional coupler, and a signal supplied by the near field probe.
The magnitude and phase measured by the RF IC can be supplied to a
controller that performs algorithmic steps as described earlier for
detecting a change in a desire frequency of the antenna. RF paths
AB and AC can be configured to 90 degrees of path difference for
nominal free space, which is the center of the measurement range
for phase of the RF IC. RF paths AB and AC can be set to -30 dB
path loss, the center of the input range being -30 dBm. FIG. 50
depicts a PCB layout of the near field sensor.
[0240] It is noted that any of the adaptive antenna embodiments
described in the subject disclosure can be applied to multiple MIMO
configuration (2.times.2, 4.times.4, etc.) or diversity
configurations. It is further noted that adaptive antenna
embodiments described in the subject disclosure can be applied to
multiband antenna structures. In such configurations, the subject
disclosure enables support for carrier aggregation of multiple
bands for simultaneous transmission or reception in MIMO or
diversity configurations, while maintaining at least some isolation
between antenna ports.
[0241] FIG. 51 depicts an illustrative embodiment of a method 5100
that can be applied to the embodiments of the subject disclosure.
The method 5100 can begin at step S102 where circuitry (such as a
log detector) can be used to measure a magnitude of a forward feed
signal supplied by a directional coupler coupled to an adaptive
antenna by way of a matching network. At step 5104 a near field
probe can be used to measure a magnitude of radiated energy from
the adaptive antenna. At step 5106 a difference between the
magnitude of the forward feed signal and the radiated energy of the
antenna can be determined. At step 5110 a determination can be made
whether the difference in magnitude results in a change in the
operating frequency (fc) of the adaptive antenna. If there is no
change (or an insignificant change) in the operating frequency (fc)
is detected, method 5100 can be adapted for repeating subsequent
iterations of method 5100 beginning from step 5102. If the change
in the operating frequency (fc) is considered significant, at step
5112 a frequency offset error can be determined according to the
difference in magnitude determined at step 5106.
[0242] If the frequency offset error is considered insignificant,
method 5100 can be repeated in subsequent iterations beginning from
step 5102. If, however, the frequency offset error is considered
undesirable (or unacceptable), at step 5114 the adaptive antenna
can be tuned by, for example, varying the electrical length of the
antenna based on an Ltune value calculated from the difference in
magnitude between the forward feed signal and the radiated energy
of the antenna. The Ltune value can be used to configure a switched
array of inductors such as shown in FIGS. 32B, 33B, 34B and 35B.
Alternatively, a tuning value can be calculated from the difference
in magnitude between the forward feed signal and the radiated
energy of the antenna, which can be used to tune one or more
programmable variable reactive elements (such as one or more
variable capacitors, one or more variable inductors, or
combinations thereof) coupled to the antenna that reconfigure the
electrical length of the antenna, and thereby change the operating
frequency of the antenna to a desirable frequency.
[0243] FIG. 52 depicts another illustrative embodiment of a near
field sensor 5210. In this embodiment, the near field sensor 5210
can be configured to perform operations including measuring phase
changes in signals radiated by the adaptive antenna 3610 using one
or more near field probes 3624, measuring a phase of a forward feed
signal using a directional coupler 3604, determining a phase offset
between the phase radiated energy emitted by the adaptive antenna
3610 and the phase of the forward feed signal supplied to the
adaptive antenna 3610 using a phase detector 5212, and tuning the
operating frequency of the adaptive antenna 3610 according to the
phase offset using the controller 3632. FIG. 53 depicts a phase
detector from Analog Devices.TM. (AD8302), which can be used in the
subject disclosure. The phase detector of FIG. 53 can be used for
measuring gain/loss and phase up to a frequency of 2.7 GHz. The
phase detector includes dual demodulating log amps with a phase
detector input range of -60 dBm to 0 dBm in a 50 Ohm system.
[0244] In one embodiment, an inductive, capacitive, and resistive
(LRC) model (such as shown by reference 5214) of an unloaded
adaptive antenna 3610 can be characterized by an antenna supplier.
The LRC characteristics of the adaptive antenna 3610 can be
characterized at a time of manufacture of the adaptive antenna
3610. Characterization and/or factory measurements can be stored in
a look-up table (or hardcoded) in an algorithm executed by a
controller or ASIC design for calculating a frequency offset
measurement to perform tuning of the adaptive antenna 3610. A phase
calibration measurement can also be performed on a detector chain
(directional coupler 3604, phase detector 5212, near field antennas
3624) to remove phase measurement error in the algorithm. Phase
calibration may depend on frequency of operation of the adaptive
antenna 3610. Accordingly, a phase calibration look-up table can be
implemented that is frequency dependent to accommodate changes in
calibration based on frequency of operation of the adaptive antenna
3610.
[0245] In one embodiment, the impedance of the LRC model 5214 can
be described by the following equation:
Z = R + j .omega. L + 1 j .omega. C ##EQU00003##
[0246] where .omega.=2.pi.f and f is frequency. The complex power
delivered can be described by the equation:
S = V 2 R + j .omega. L + 1 j .omega. C = V 2 R + j ( .omega. L - 1
.omega. C ) = R - j ( .omega. L - 1 .omega. C ) R 2 + ( .omega. L -
1 .omega. C ) 2 * V 2 ##EQU00004##
[0247] Based on the above equation, phase of S can be determined
according to equation:
.0. = atan ( 1 .omega. C - .omega. L R ) ##EQU00005##
[0248] When O=0, the antenna model 5214 is at resonance.
[0249] Solving for .omega.:
.omega. 2 L + R * tan .0. * .omega. - 1 C = 0 ##EQU00006## .omega.
( .0. ) = - R * tan .0. + R 2 * tan 2 .0. + 4 L / C 2 L
##EQU00006.2## f ( .0. ) = .omega. ( .0. ) 2 .pi.
##EQU00006.3##
[0250] From the above equations, the following conditions can be
assessed: [0251] If f.sub.operation (op)=f.sub.resonance (r), then
O=0, and .omega.=1/ {square root over (LC)}=.omega..sub.0 [0252] If
f.sub.op<f.sub.r, then O>0, and .omega.<1/ {square root
over (LC)} [0253] If f.sub.op>f.sub.r, then O<0
[0254] Based on the above observations, one embodiment of an
algorithm for detecting a change in an operating frequency of the
adaptive antenna 3610 can comprise the following steps: [0255]
Phase detector 5212 measures a differential phase O [0256] The
controller 3632 can then calculate a measured frequency according
to .omega..sub.measured=f(O) [0257] The controller 3632 knows the
desired frequency .omega..sub.target [0258] The controller 3632 can
then calculate a relative scaling error in the measured versus the
target frequency
.varies..sub.offset=.omega..sub.measured/.omega..sub.target [0259]
As noted earlier, a phase calibration measurement can be performed
on the detector chain to remove phase measurement error. This can
be factored into calculating the .varies..sub.offset [0260]
.varies..sub.offset may be integrated or averaged over a time
interval or a number of samples to reduce noise or sensitivity to
transients shorter than desired time interval [0261] Also described
earlier, the antenna resonance can be modeled according to
.omega..sub.o=1/ {square root over (LC)} [0262] To correct a
detected offset in frequency back to a target frequency
(.omega..sub.target), the present operating frequency
(.omega..sub.measured) can be multiplied by 1/.varies..sub.offset
[0263] Removing the offset can be accomplished by tuning L or C of
the antenna. As noted earlier, the LC modeling of the antenna can
be characterized in a lab setting, during manufacturing, or by
other suitable means. An aperture tuner using a switch array of
inductors such as shown in FIGS. 32B-35B can be used to tune the
adaptive antenna 3610. Tunable capacitive devices can also be used
in place of the tunable inductors or combinations thereof. [0264]
To remove the detected frequency offset, the adaptive antenna 3610
can be tuned such that the product of LC is changed by a factor of
.varies..sup.2.sub.offset to return .omega..sub.measured to
.omega..sub.target. [0265] The plots of FIG. 54 depict phase error
versus frequency and power level of the phase detector (AD8302
shown in FIG. 53). To improve the accuracy of phase detection data
provided by the phase detector to the controller, the phase
detector can be utilized in a configuration whereby a phase delay
from a near field probe is set using a length of delay line such
that an RF signal reaching the phase detector input from the near
field probe is approximately 90 degrees out of phase with an RF
signal reaching another the phase detector input from the
directional coupler under nominal conditions, taken as a free space
condition. In another embodiment, the delay line can be instead
applied to the RF signal supplied by the directional coupler.
[0266] FIG. 55 depicts an illustrative embodiment of a method 5500
for frequency offset detection and tuning based on phase
measurements. Method 5500 can begin with step 5502 in which a
forward feed signal (e.g., a transmit signal supplied by the
transmitter 3602) is measured by way of the directional coupler
3604 and supplied to the phase detector 5212. Similarly, at step
5504, the near field probe 3624 can supply a measure of a signal
representing radiated energy from the adaptive antenna 3610. As
noted in other embodiments, a switch, such as shown in FIG. 52, can
be used to select from a plurality of near field probes 3624 to
provide measurements of radiated energy from the adaptive antenna
3610 from multiple positions of the probes. The controller 3632 can
selectively use multiple measurements from one or more near field
probes 3624 to improve the accuracy of the algorithm.
[0267] At step 5506, the phase detector 5212 can generate a phase
differential based on the phase of the signals measured at steps
S502 and S504. A digital representation of the phase differential
is then supplied by the A2D 3630 to the controller 3632 for
calculating a frequency offset (.varies..sub.offset). The
controller 3632 at step 5505 can retrieve phase calibration data
from a look-up table based on a target frequency of operation or a
measured frequency of operation. In one embodiment the look-up
table can further depend on an open loop state of the antenna and
use case information such as whether a communication device
utilizing the antenna structure of the subject disclosure is being
held by a user's hand, whether the communication device is in a
position next to a user's ear, and so on. The phase calibration
data can be used to adjust calibration errors in the phase
differential. To calculate a frequency offset as described above,
the controller 3632 can also retrieve from a look-up table at step
5507 an LRC model of the adaptive antenna 3610, which may also be
frequency dependent. The calibration data and the LRC model can be
stored in a memory of a communication device that integrates the
embodiments of FIG. 52. At step 5510, the controller 3632 can
determine whether a change in frequency from a desired target
frequency has occurred. A change in frequency can be determined
when, for example, .varies..sub.offset is greater than or less than
unity (i.e., 1).
[0268] At step 5512, the controller 3632 can be configured to make
a determination whether the frequency offset is significant enough
to warrant retuning of the measured operating frequency of the
adaptive antenna 3610. To avoid excessive retuning of the adaptive
antenna 3610, the controller 3632 can be configured to comp are
.varies..sub.offset to a range of thresholds. For example, when
.varies..sub.offset>1, .varies..sub.offset can be compared to a
first threshold. The first threshold can represent an acceptable
frequency overshoot range of the frequency measured
(w.sub.measured) above the target frequency (w.sub.target).
Similarly, when .varies..sub.offset<1, .varies..sub.offset can
be compared to a second threshold. The second threshold can
represent an acceptable frequency undershoot range of the frequency
measured (w.sub.measured) below the target frequency
(w.sub.target). The first and second thresholds can be based on
specifications provided by a network provider. Thus, when the
frequency measured exceeds the first threshold or is below the
second threshold, the controller 3632 can proceed to step 5514
where it retunes the adaptive antenna 3610 as previously described
in the subject disclosure to bring .varies..sub.offset closer to
unity.
[0269] It is further noted that more than two thresholds (e.g., two
thresholds for detecting large offsets and two additional
thresholds for detecting smaller offsets) can be used to enable a
determination when a coarse tuning versus a fine tuning of the
adaptive antenna 3610 is required.
[0270] In another embodiment of the subject disclosure, method 5500
of FIG. 55 can be adapted to use a first look-up table that maps a
detected phase output voltage to a frequency shift, and a second
lookup table that maps antenna tuning states to the frequency shift
determined from the first look-up table. Accordingly, when the
controller 3632 reads a voltage signal from the phase detector 5212
in digital format provided by the A2D 3630, the controller 3632
proceeds to look up a frequency shift in the first look-up table,
then utilizes the frequency shift to index through the second
look-up table to obtain one or more state increments to shift the
frequency of the antenna to a desired state. The one or more state
increments can represent an adjustment of the electrical length of
the adaptive antenna 3610 utilizing an array of selectable
inductors such as shown in FIGS. 32B-35B.
[0271] In also noted that the methods depicted in FIGS. 51 and 55
can be combined to further enhance frequency offset detection and
retuning. For example, in one embodiment delta averages of Y can be
compared to delta averages of .varies..sub.offset, and such
comparisons can be used to estimate a frequency offset. In another
embodiment, delta averages of Y can be correlated to expected delta
averages of .varies..sub.offset, which can be recorded in a look-up
table. Similarly, delta averages of a offset can be correlated to
expected delta averages of Y, which can be recorded in a look-up
table. The look-up tables can in turn be used for validation of a
frequency offset calculation. Other suitable techniques can be used
to compare the results of one algorithm against the other for
validation of measurements.
[0272] RF magnitude detection (embodiment of FIG. 51) and RF phase
detection (embodiment of FIG. 54) can be used together to provide
improved antenna frequency tuning. The RF magnitude detection
provides a direct measurement of radiated power. Phase detection
yields an instantaneous estimate of the amount of frequency error,
and hence the amount of correction needed. In one embodiment the
frequency error estimate determined from phase detection may be
used to calculate a specific amount of frequency shift and
direction for the next iterative step in the magnitude detection
algorithm. Thus phase detection algorithm of FIG. 54 can enable the
RF magnitude algorithm of FIG. 51 to close in on a desired
frequency of operation.
[0273] In another embodiment, an iterative algorithm for mitigating
a frequency offset can utilize both magnitude and phase readings as
described in the subject disclosure to determine an error estimate.
The error estimate can be based on a weighted combination of both
magnitude (mbar) and phase (.varies..sub.offset) offsets, with the
algorithm seeking to reduce the combined error estimate instead of
relying on either the magnitude or phase error estimates
individually.
[0274] FIG. 56 depicts an illustrative embodiment of a system 5600
having a reactance sensor 5608 for measuring a reactive load
applied to the antenna 3610. The reactive sensor 5608 can be used,
for example, to measure a load capacitance of the antenna 3610
utilizing a delta-sigma capacitance meter or other suitable
circuitry operating at a sampling frequency fs or at some band of
frequencies separate from the operating frequency band of the RF
and/or antenna system. The reactive sensor 5608 in turn supplies
the controller 3632 a digital signal representing the measured load
capacitance of the antenna 3610. An RF choke 5606 can be used to
restrict the flow of RF signals to the reactive sensor 5608 except
RF signals at or below the sampling frequency f.sub.s of the
reactive sensor 5608. The separation of the RF front end (i.e.,
3601 and/or 3602) from the reactive sensor 5608 may also be
accomplished with filters or other signal separation means,
including a diplexer, a duplexer, a multiplexer or other suitable
circuitry. In this illustration, the carrier frequency f.sub.c of
signals received by the receiver 3601 from the antenna 3610 or
supplied by the transmitter 3602 to the antenna 3610 are at a much
higher frequency than the sampling frequency f.sub.s of the
reactive sensor 5608, thereby enabling the RF choke 5606 to prevent
such signals from being leaked into the reactive sensor 5608. A
transient suppressor 5604 can be used for ESD protection, and can
utilize a diode circuit to prevent the sampling signal of the
reactive sensor 5608 from being dissipated to ground. It should be
noted that loading conditions contributed by the RF choke 5606, the
transient suppressor 5604 and the matching network 3608 any other
circuits of system 5600 can be pre-measured at various frequencies
in a controlled setting (e.g., a lab and/or manufacturing) based on
known loading conditions of the antenna 3610. Such loading
measurements can be tabulated in a look-up table at various
operating frequencies and utilized to improve the accuracy of
capacitive load readings of the antenna 3610 by removing known
errors from measurements made by the reactive sensor 5608.
[0275] In one embodiment, frequency tuning using the system 5600 of
FIG. 56 can be performed by measuring a change in reactance (e.g.,
capacitance) of the adaptive antenna 3610, converting the measured
change in reactance to a frequency offset, and tuning an operating
frequency of the adaptive antenna 3610 according to the frequency
offset. An LRC model of an unloaded antenna 5602 such as shown in
FIG. 56 (without Z.sub.IN and C.sub.L) can be characterized in a
lab setting, during manufacturing, or by an antenna supplier.
Characterizations of the LRC model at various frequencies can be
stored in a look-up table (or hardcoded) when performing frequency
tuning as will be described below.
[0276] In one embodiment, the impedance of the LRC model 5602 can
be described by the following equation, which was earlier
described:
Z = R + j .omega. L + 1 j .omega. C ##EQU00007##
[0277] where .omega.=2.pi.f and f is frequency,
R=R.sub.rad+R.sub.internal (radiation resistance+resistance of
antenna), L is the equivalent inductance of the antenna, and C is
the equivalent capacitance of the antenna 3610. The above equation
can be rewritten as:
Z = j [ .omega. L - 1 .omega. C ] + R ##EQU00008##
[0278] Resonance occurs when .omega.L=1/.omega.C or
.omega..sup.2=1/LC. It follows therefore that the resonance
frequency f.sub.r=1/2.pi. {square root over (LC)}.
[0279] Any external loading of the adaptive antenna 3610 may be
assumed to be (at first order) a parallel capacitance C.sub.L. The
load capacitance may also consist of other reactive components
associated with the circuit implementation, as indicated
previously. The resonance frequency can thus be determined from
f.sub.r=1/2.pi. {square root over (LC.sub.T)}, where
C.sub.T=C+C.sub.L. By measuring C.sub.L, and knowing LRC values
from a look-up table, a change in resonant frequency can be
calculated.
[0280] Considering the impedance Z.sub.IN 5603 of the matching
circuit 3608, which may be programmable, the impedance of the 5602
model can be further characterized as impedance Z.sub.A, which can
represent an impedance looking in from a voltage source of the LRC
model 5602. Z.sub.A can be described by the following
equations:
Z A = r + j b + j .omega. L - j .omega. * [ 1 C + CL ] + R
##EQU00009## Z A = r + R + j * ( b + .omega. L - 1 .omega. [ 1 C +
CL ] ) ##EQU00009.2##
[0281] where Z.sub.IN=r+jb, and where r is the real impedance and b
is the reactance of the matching network 3608 (whether fixed or
programmable), each which can be known to the controller 3632.
Resonance occurs when the imaginary quantity becomes zero, as
described by the following equation:
b + .omega. L = 1 .omega. [ 1 C + CL ] ##EQU00010##
[0282] which can be rewritten in quadratic form as:
.omega. 2 + b / L * .omega. - 1 L [ 1 C + CL ] = 0 ##EQU00011##
[0283] The variable .omega. can be solved for with the quadratic
formula based on known values for r, b, L, C and C.sub.L (which can
be measured by the reactive sensor 5606 as shown in FIG. 56). Once
co is solved, the new resonant frequency can be determined from
f.sub.r'=.omega./2.pi., which can be compared to an expected
resonant frequency (i.e., expected operating frequency of the
antenna 3610) to determine a frequency offset. Knowing the
frequency offset, the controller 3632 can be programmed to adjust
the resonant frequency of the antenna by programming an aperture
tuner of the antenna 3610 (such those shown in FIGS. 32B-32C,
33B-33C, 34B-34C, and 35B-35C). It is noted that the reactive
sensor 5608 can also be used to measure other parameters such as
inductance, or admittance.
[0284] FIG. 57 depicts an illustrative embodiment of a third method
5700 that can be applied to the subject disclosure. Method 5700 can
begin at step 5702 where the reactive sensor 5608 measures a load
capacitance of the antenna 3610. As noted earlier, the controller
3632 can retrieve capacitive offset values from a look-up table
based on a known operating frequency of the system 5600 to remove
sensing errors when measuring the capacitive load of the antenna
3610. Based on known tabulated values of the LRC model of the
antenna 3610 at different operating frequencies, the controller
3632 can retrieve such values based on an expected operating
frequency to determine at step 5706 a frequency offset of the
antenna 3610 (as described above) according to the load capacitance
measured at step 5702. If a frequency offset is detected at step
5708, the controller 3632 can proceed to step 5710 to assess if the
detected error is nominal and thus can be ignored, or if it
significant enough to warrant a retuning of the resonant frequency
of the antenna 3610 at step 5712 utilizing the aperture tuning
techniques described in the subject disclosure. The error
determination can be based on one or more thresholds applied to
step 5512.
[0285] The methods depicted in FIGS. 51, 55 and 57 can be combined
to improve the accuracy of detecting a frequency offset of the
antenna 3610 and mitigating such offsets through aperture tuning.
For example, in one embodiment delta averages of Y determined by
method 5100 can be compared to delta averages of
.varies..sub.offset determined by method 5500, and to delta
averages of frequency offsets determined by method 5700 (which will
be referred to as O.sub.offset) and such comparisons can be used to
estimate a collective frequency offset. In another embodiment,
delta averages of Y can be correlated to expected delta averages of
.varies..sub.offset, which can be recorded in a look-up table.
Similarly, delta averages of .varies..sub.offset can be correlated
to expected delta averages of Y, which can be recorded in a look-up
table. Similarly, delta averages of .varies..sub.offset can be
correlated to expected delta averages of O.sub.offset, which can be
recorded in a look-up table. Similarly, delta averages of
O.sub.offset can be correlated to expected delta averages of
.varies..sub.offset, which can be recorded in a look-up table.
Additional combinations of these look-up tables can be created to
account for all possible combinations or a subset thereof. The
collection of look-up tables in turn can be used for validation of
a frequency offset calculation by the controller 3632 by any one of
the algorithms described in the subject disclosure. Other suitable
techniques can be used to compare the results of one algorithm
against another for validation of measurements.
[0286] In one embodiment, an antenna can have a tunable quality
factor (Q) which can proportionally vary a bandwidth of the antenna
and can change the resonance of the antenna. A tunable capacitance
(C) and/or tunable inductance (L) can be used to vary Q factor of
the antenna. A closed loop tunable resonance technique can be used
to compensate for a change in bandwidth (BW) of the antenna.
Variable BW can be used for improving co-channel interference or
intermodulation distortion (IMD). Resonance tuning can also be used
for band and/or channel selectivity depending on how narrow the BW
of the antenna is. An antenna can be tuned to a narrower BW when
carrier aggregation (CA) is not needed. A narrower BW can also be
used to alleviate front end filter design.
[0287] FIG. 58 depicts an illustrative embodiment of an antenna
5802 having a tunable bandwidth. The antenna 5802 can be located on
an end of a 60.times.120 mm ground plane. The antenna 5802 can
include a variable inductor (L1) and variable capacitor (C1). L1
and C1 can be a part of a total equivalent L and C of the antenna
5802. C1 and/or L1 can be located at an end of the antenna 5802
near a feed point. Changes in L1 and C1 can alter the resonance
frequency of the antenna 5802, its resonance bandwidth (BW), and
input impedance. Accordingly, a closed loop resonance technique
such as described in the subject disclosure, such as by way of
illustration the methods of FIGS. 51, 55, and/or 57, can be used to
mitigate offsets caused in the resonance frequency of the antenna
5802 due to bandwidth tuning. It is noted that other configurations
of the antenna 5802 that are suitable for tuning a resonance
bandwidth can be applied to the subject disclosure.
[0288] A process for tuning a variable bandwidth antenna 5802 such
as illustrated in FIG. 58 can be described as follows. The antenna
5802 can be modeled as a single resonance antenna based on a series
RLC equivalent circuit--similar to the model 5602 shown in FIG. 56.
The equivalent inductance (L) of the antenna can be strongly
influenced by inductance in the high current region of the antenna
while the equivalent capacitance (C) of the antenna can be strongly
influenced capacitance at the high potential region of the antenna.
The resistance (R) is a combination of the radiation resistance and
an equivalent loss resistance. For a quarter-wave type antenna the
inductive portion can be considered near the feed point while the
capacitance portion can be considered near an end of the antenna
5802. The resonant frequency of the antenna 5802 can be expressed
as:
f r = 1 2 .pi. LC ##EQU00012##
[0289] f.sub.r can be raised or lowered by decreasing or increasing
L or C or both. The quality factor (Q) of the antenna can be
expressed as a function of RLC:
Q = 1 R L C ##EQU00013##
[0290] The fractional bandwidth BW=.DELTA.f/f of the antenna 5802
can be approximated to the inverse of Q: BW=1/Q. Q changes
proportionately to L and inversely with C. Hence, the bandwidth of
the antenna 5802 can be varied by changing the L/C ratio while
f.sub.r can be varied by changing the LC product. By using a
variable inductor (L) and variable capacitor (C) coupled to the
antenna 5802, the resonance bandwidth (L/C ratio) and the resonance
frequency (LC product) of the antenna 5802 can be tuned. FIG. 59
depicts plots resulting from tuning a bandwidth of the antenna 5802
of FIG. 58.
[0291] It is noted that the embodiment of antenna 5802 (and
variants thereof that support tuning a resonance bandwidth of an
antenna) can be used by more than one antenna of a communication
device in a MIMO or diversity configuration or other suitable
configurations. Additionally, the embodiment of antenna 5802 can be
combined with any of the embodiments of the subject disclosure for
improving isolation between antennas, among other desirable
improvements.
[0292] In one embodiment, antenna 5802 can be tuned to a narrower
bandwidth according to the aforementioned embodiments when
interband carrier aggregation (CA) is not required. Tuning the
antenna 5802 to a narrower bandwidth can increase isolation of the
antenna 5802 from other antennas operating outside the passband of
antenna 5802, and/or can increase isolation of the antenna 5802
from other antennas that are transmitting simultaneously in an
interband CA configuration. By tuning the antenna 5802 to a
narrower bandwidth, the narrower bandwidth can help reduce
desensitization due to transmissions outside the passband of the
antenna--from external or internal transmissions, the latter being
due to interband CA or noise sources.
[0293] In the case of noise sources, narrowing the resonance
bandwidth of the antenna 5802 can also potentially reduce total
isotropic sensitivity (TIS) thereby improving a performance of the
antenna 5802. Narrowing the bandwidth of the antenna 5802 can also
help reduce intermodulation distortion (IMD) and third order
intercept point (IP3/IIP3) issues. In another embodiment, when
intraband CA is required the bandwidth of the antenna 5802 can be
increased for simultaneous use of multiple channels within a
band.
[0294] When signals are being transmitted by the antenna 5802,
adjusting the bandwidth of the antenna 5802 can also reduce
harmonics and IMD byproducts due to mixing. Additionally, spurious
radiation outside a desired channel may be mitigated if the antenna
passband is sufficiently narrow.
[0295] FIG. 60 depicts an illustrative embodiment of a method 6000
that can be applied to the subject disclosure. Method 6000 can
begin at step 6002 where a communication device utilizing one or
more instances of the antenna 5802 of FIG. 58 determines a need to
adjust a bandwidth of operation of the antenna 5802. The need to
adjust the resonance bandwidth can represent a need to increase or
decrease the resonance bandwidth of the antenna 5802. For example,
the communication device can be configured to determine that
interband carrier aggregation is not required, and thereby proceed
to reduce the bandwidth of the antenna 5802 to improve isolation
and/or other issues discussed above. Alternatively, the
communication device can be configured to determine that intraband
carrier aggregation is required, and proceed to increase the
resonance bandwidth of the antenna 5802 to enable the communication
device to simultaneously use multiple channels for communication
purposes.
[0296] At step 6004, the communication device can obtain from a
look-up table the LRC model parameters of the antenna 5802, which
it can use for calculating with the above equations parameters for
tuning the variable inductance (L), variable capacitance (C), or
both to adjust the resonance bandwidth of the antenna 5802. In step
6006, the communication device can also determine a desired
bandwidth for tuning the antenna 5802. The desired bandwidth can be
determined based on whether interband CA is or is not being used,
intraband CA is being used, or any other circumstance where
adjusting the resonance bandwidth of the antenna 5802 can improve
the performance of the antenna 5802. To circumvent the need for
calculations by the communication device based on the above
equations, the communication device can utilize a look-up table
that cross-references a desired resonance bandwidth with
recommended tuning values for the variable inductor (L) of antenna
5802, variable capacitor (C) of antenna 5802, or both.
[0297] In addition, the look-up table can include recommended
resonance frequency tuning instructions to reduce a change in the
resonance frequency of the antenna 5802 resulting from tuning the
resonance fractional bandwidth of the antenna 5802. Resonance
tuning instructions can include among other things parameters for
adjusting an electrical length of the antenna 5802, parameters
adding or remove reactance from the antenna 5802 such as shown in
FIGS. 32B, 32C, 33B, 33C, 34B, 34C, 35B and/or 35C, or any other
suitable technique described in the subject disclosure for tuning
the resonant frequency of the antenna 5802.
[0298] Once the communication device has obtained tuning parameters
for adjusting the resonance fractional bandwidth of the antenna
5802, the communication device can proceed to step 6008 to tune the
variable inductor (L), variable capacitor (C), or both illustrated
in FIG. 58. The antenna 5802 can be combined with any of the
embodiments of the subject disclosure (e.g., FIGS. 36A, 36B, 49,
52, 56) for detecting and mitigating an offset in a desired
resonance frequency of the antenna. Accordingly, any offset in the
resonant frequency of the antenna 5802 caused by the tuning of the
resonance bandwidth can be mitigated by steps 6010, 6012, and 6014
using the techniques described above for determining and adjusting
a magnitude difference between a first signal supplied to the
antenna 5802 and a second signal radiated by the antenna 5802, a
phase difference between the first signal supplied to the antenna
5802 and the second signal radiated by the antenna 5802, a change
in reactance of the antenna 5802, or any combination thereof.
[0299] It is noted that the embodiments of the subject disclosure
can be applied to mobile or stationary communication devices.
Mobile communication devices can include without limitation
cellular phones, smartphones, tablets, laptop computers, and so on.
Stationary communication devices can include base stations such as
a cellular base station, a femto cell, a wireless fidelity access
point, a small cell, a micro cell, and so on.
[0300] FIG. 61 depicts an illustrative embodiment of a
communication device 6100. The communication device 6100 can
comprise a wireline and/or wireless transceiver 6102 (herein
transceiver 6102), a user interface (UI) 6104, a power supply 6114,
a location receiver 6116, a motion sensor 6118, an orientation
sensor 6120, and a controller 6106 for managing operations thereof.
The transceiver 6102 can support short-range or long-range wireless
access technologies such as Bluetooth, ZigBee, WiFi, DECT, or
cellular communication technologies, just to mention a few.
Cellular technologies can include, for example, CDMA-1.times.,
UMTS/HSDPA, GSM/GPRS, TDMA/EDGE, EV/DO, WiMAX, SDR, LTE, as well as
other next generation wireless communication technologies as they
arise. The transceiver 6102 can also be adapted to support
circuit-switched wireline access technologies (such as PSTN),
packet-switched wireline access technologies (such as TCP/IP, VoIP,
etc.), and combinations thereof. The transceiver 6102 can be
adapted to utilize any of the aforementioned adaptive antenna
embodiments described above singly or in any combination.
[0301] The UI 6104 can include a depressible or touch-sensitive
keypad 6108 with a navigation mechanism such as a roller ball, a
joystick, a mouse, or a navigation disk for manipulating operations
of the communication device 6100. The keypad 6108 can be an
integral part of a housing assembly of the communication device
6100 or an independent device operably coupled thereto by a
tethered wireline interface (such as a USB cable) or a wireless
interface supporting for example Bluetooth. The keypad 6108 can
represent a numeric keypad commonly used by phones, and/or a QWERTY
keypad with alphanumeric keys. The UI 6104 can further include a
display 6110 such as monochrome or color LCD (Liquid Crystal
Display), OLED (Organic Light Emitting Diode) or other suitable
display technology for conveying images to an end user of the
communication device 6100. In an embodiment where the display 6110
is touch-sensitive, a portion or all of the keypad 6108 can be
presented by way of the display 6110 with navigation features.
[0302] The display 6110 can use touch screen technology to also
serve as a user interface for detecting user input. As a touch
screen display, the communication device 6100 can be adapted to
present a user interface with graphical user interface (GUI)
elements that can be selected by a user with a touch of a finger.
The touch screen display 6110 can be equipped with capacitive,
resistive or other forms of sensing technology to detect how much
surface area of a user's finger has been placed on a portion of the
touch screen display. This sensing information can be used to
control the manipulation of the GUI elements or other functions of
the user interface. The display 6110 can be an integral part of the
housing assembly of the communication device 6100 or an independent
device communicatively coupled thereto by a tethered wireline
interface (such as a cable) or a wireless interface.
[0303] The UI 6104 can also include an audio system 6112 that
utilizes audio technology for conveying low volume audio (such as
audio heard in proximity of a human ear) and high volume audio
(such as speakerphone for hands free operation). The audio system
6112 can further include a microphone for receiving audible signals
of an end user. The audio system 6112 can also be used for voice
recognition applications. The UI 6104 can further include an image
sensor 6113 such as a charged coupled device (CCD) camera for
capturing still or moving images.
[0304] The power supply 6114 can utilize common power management
technologies such as replaceable and rechargeable batteries, supply
regulation technologies, and/or charging system technologies for
supplying energy to the components of the communication device 6100
to facilitate long-range or short-range portable applications.
Alternatively, or in combination, the charging system can utilize
external power sources such as DC power supplied over a physical
interface such as a USB port or other suitable tethering
technologies.
[0305] The location receiver 6116 can utilize location technology
such as a global positioning system (GPS) receiver capable of
assisted GPS for identifying a location of the communication device
6100 based on signals generated by a constellation of GPS
satellites, which can be used for facilitating location services
such as navigation. The motion sensor 6118 can utilize motion
sensing technology such as an accelerometer, a gyroscope, or other
suitable motion sensing technology to detect motion of the
communication device 6100 in three-dimensional space. The
orientation sensor 6120 can utilize orientation sensing technology
such as a magnetometer to detect the orientation of the
communication device 6100 (north, south, west, and east, as well as
combined orientations in degrees, minutes, or other suitable
orientation metrics).
[0306] The communication device 6100 can use the transceiver 6102
to also determine a proximity to a cellular, WiFi, Bluetooth, or
other wireless access points by sensing techniques such as
utilizing a received signal strength indicator (RSSI) and/or signal
time of arrival (TOA) or time of flight (TOF) measurements. The
controller 6106 can utilize computing technologies such as a
microprocessor, a digital signal processor (DSP), programmable gate
arrays, application specific integrated circuits, and/or a video
processor with associated storage memory such as Flash, ROM, RAM,
SRAM, DRAM or other storage technologies for executing computer
instructions, controlling, and processing data supplied by the
aforementioned components of the communication device 400.
[0307] Other components not shown in FIG. 61 can be used in one or
more embodiments of the subject disclosure. For instance, the
communication device 6100 can include a reset button (not shown).
The reset button can be used to reset the controller 6106 of the
communication device 6100. In yet another embodiment, the
communication device 6100 can also include a factory default
setting button positioned, for example, below a small hole in a
housing assembly of the communication device 6100 to force the
communication device 6100 to re-establish factory settings. In this
embodiment, a user can use a protruding object such as a pen or
paper clip tip to reach into the hole and depress the default
setting button. The communication device 400 can also include a
slot for adding or removing an identity module such as a Subscriber
Identity Module (SIM) card. SIM cards can be used for identifying
subscriber services, executing programs, storing subscriber data,
and so forth.
[0308] The communication device 6100 as described herein can
operate with more or less of the circuit components shown in FIG.
61. These variant embodiments can be used in one or more
embodiments of the subject disclosure.
[0309] It should be understood that devices described in the
exemplary embodiments can be in communication with each other via
various wireless and/or wired methodologies. The methodologies can
be links that are described as coupled, connected and so forth,
which can include unidirectional and/or bidirectional communication
over wireless paths and/or wired paths that utilize one or more of
various protocols or methodologies, where the coupling and/or
connection can be direct (e.g., no intervening processing device)
and/or indirect (e.g., an intermediary processing device such as a
router).
[0310] FIG. 62 depicts an exemplary diagrammatic representation of
a machine in the form of a computer system 6200 within which a set
of instructions, when executed, may cause the machine to perform
any one or more of the embodiments described above. One or more
instances of the machine can utilize the aforementioned adaptive
antenna embodiments singly or in any combination. In some
embodiments, the machine may be connected (e.g., using a network
6226) to other machines. In a networked deployment, the machine may
operate in the capacity of a server or a client user machine in
server-client user network environment, or as a peer machine in a
peer-to-peer (or distributed) network environment.
[0311] The machine may comprise a server computer, a client user
computer, a personal computer (PC), a tablet PC, a smart phone, a
laptop computer, a desktop computer, a control system, a network
router, switch or bridge, or any machine capable of executing a set
of instructions (sequential or otherwise) that specify actions to
be taken by that machine. It will be understood that a
communication device of the subject disclosure includes broadly any
electronic device that provides voice, video or data communication.
Further, while a single machine is illustrated, the term "machine"
shall also be taken to include any collection of machines that
individually or jointly execute a set (or multiple sets) of
instructions to perform any one or more of the methods discussed
herein.
[0312] The computer system 6200 may include a processor (or
controller) 6202 (e.g., a central processing unit (CPU), a graphics
processing unit (GPU, or both), a main memory 6204 and a static
memory 6206, which communicate with each other via a bus 6208. The
computer system 6200 may further include a display unit 6210 (e.g.,
a liquid crystal display (LCD), a flat panel, or a solid state
display. The computer system 6200 may include an input device 6212
(e.g., a keyboard), a cursor control device 6214 (e.g., a mouse), a
disk drive unit 6216, a signal generation device 6218 (e.g., a
speaker or remote control) and a network interface device 6220. In
distributed environments, the embodiments described in the subject
disclosure can be adapted to utilize multiple display units 6210
controlled by two or more computer systems 6200. In this
configuration, presentations described by the subject disclosure
may in part be shown in a first of the display units 6210, while
the remaining portion is presented in a second of the display units
6210.
[0313] The disk drive unit 6216 may include a tangible
computer-readable storage medium 6222 on which is stored one or
more sets of instructions (e.g., software 6224) embodying any one
or more of the methods or functions described herein, including
those methods illustrated above. The instructions 6224 may also
reside, completely or at least partially, within the main memory
6204, the static memory 6206, and/or within the processor 6202
during execution thereof by the computer system 6200. The main
memory 6204 and the processor 6202 also may constitute tangible
computer-readable storage media.
[0314] Dedicated hardware implementations including, but not
limited to, application specific integrated circuits, programmable
logic arrays and other hardware devices that can likewise be
constructed to implement the methods described herein. Application
specific integrated circuits and programmable logic array can use
downloadable instructions for executing state machines and/or
circuit configurations to implement embodiments of the subject
disclosure. Applications that may include the apparatus and systems
of various embodiments broadly include a variety of electronic and
computer systems. Some embodiments implement functions in two or
more specific interconnected hardware modules or devices with
related control and data signals communicated between and through
the modules, or as portions of an application-specific integrated
circuit. Thus, the example system is applicable to software,
firmware, and hardware implementations.
[0315] In accordance with various embodiments of the subject
disclosure, the operations or methods described herein are intended
for operation as software programs or instructions running on or
executed by a computer processor or other computing device, and
which may include other forms of instructions manifested as a state
machine implemented with logic components in an application
specific integrated circuit or field programmable gate array.
Furthermore, software implementations (e.g., software programs,
instructions, etc.) including, but not limited to, distributed
processing or component/object distributed processing, parallel
processing, or virtual machine processing can also be constructed
to implement the methods described herein. It is further noted that
a computing device such as a processor, a controller, a state
machine or other suitable device for executing instructions to
perform operations or methods may perform such operations directly
or indirectly by way of one or more intermediate devices directed
by the computing device.
[0316] While the tangible computer-readable storage medium 6222 is
shown in an example embodiment to be a single medium, the term
"tangible computer-readable storage medium" should be taken to
include a single medium or multiple media (e.g., a centralized or
distributed database, and/or associated caches and servers) that
store the one or more sets of instructions. The term "tangible
computer-readable storage medium" shall also be taken to include
any non-transitory medium that is capable of storing or encoding a
set of instructions for execution by the machine and that cause the
machine to perform any one or more of the methods of the subject
disclosure.
[0317] The term "tangible computer-readable storage medium" shall
accordingly be taken to include, but not be limited to: solid-state
memories such as a memory card or other package that houses one or
more read-only (non-volatile) memories, random access memories, or
other re-writable (volatile) memories, a magneto-optical or optical
medium such as a disk or tape, or other tangible media which can be
used to store information. Accordingly, the disclosure is
considered to include any one or more of a tangible
computer-readable storage medium, as listed herein and including
art-recognized equivalents and successor media, in which the
software implementations herein are stored.
[0318] Although the present specification describes components and
functions implemented in the embodiments with reference to
particular standards and protocols, the disclosure is not limited
to such standards and protocols. Each of the standards for Internet
and other packet switched network transmission (e.g., TCP/IP,
UDP/IP, HTML, and HTTP) represent examples of the state of the art.
Such standards are from time-to-time superseded by faster or more
efficient equivalents having essentially the same functions.
Wireless standards for device detection (e.g., RFID), short-range
communications (e.g., Bluetooth, WiFi, ZigBee), and long-range
communications (e.g., WiMAX, GSM, CDMA, LTE) can be used by
computer system 6200.
[0319] The illustrations of embodiments described herein are
intended to provide a general understanding of the structure of
various embodiments, and they are not intended to serve as a
complete description of all the elements and features of apparatus
and systems that might make use of the structures described herein.
Many other embodiments will be apparent to those of skill in the
art upon reviewing the above description. The exemplary embodiments
can include combinations of features and/or steps from multiple
embodiments. Other embodiments may be utilized and derived
therefrom, such that structural and logical substitutions and
changes may be made without departing from the scope of this
disclosure. Figures are also merely representational and may not be
drawn to scale. Certain proportions thereof may be exaggerated,
while others may be minimized. Accordingly, the specification and
drawings are to be regarded in an illustrative rather than a
restrictive sense.
[0320] Although specific embodiments have been illustrated and
described herein, it should be appreciated that any arrangement
calculated to achieve the same purpose may be substituted for the
specific embodiments shown. This disclosure is intended to cover
any and all adaptations or variations of various embodiments.
Combinations of the above embodiments, and other embodiments not
specifically described herein, can be used in the subject
disclosure.
[0321] The Abstract of the Disclosure is provided with the
understanding that it will not be used to interpret or limit the
scope or meaning of the claims. In addition, in the foregoing
Detailed Description, it can be seen that various features are
grouped together in a single embodiment for the purpose of
streamlining the disclosure. This method of disclosure is not to be
interpreted as reflecting an intention that the claimed embodiments
require more features than are expressly recited in each claim.
Rather, as the following claims reflect, inventive subject matter
lies in less than all features of a single disclosed embodiment.
Thus the following claims are hereby incorporated into the Detailed
Description, with each claim standing on its own as a separately
claimed subject matter.
[0322] It is to be understood that although the disclosure has been
described above in terms of particular embodiments, the foregoing
embodiments are provided as illustrative only, and do not limit or
define the scope of the disclosure.
[0323] Various other embodiments, including but not limited to the
following, are also within the scope of the claims. For example,
the elements or components of the various multimode antenna
structures described herein may be further divided into additional
components or joined together to form fewer components for
performing the same functions. For example, the antenna elements
and the connecting element or elements that are part of a multimode
antenna structure may be combined to form a single radiating
structure having multiple feed points operatively coupled to a
plurality of antenna ports or feed points.
[0324] It is further noted that the low band and high band antennae
structures described in the subject disclosure may be different or
dissimilar antenna types, such as, for example, monopole, PTA,
loop, dielectric or other structures known in the art. It is also
noted that the embodiments described herein may represent other
sub-frequency ranges such as, for example, low band, mid band, and
high band. Accordingly, the antenna structures described herein may
have differing antenna types, and differing frequency ranges.
[0325] Having described embodiments of the present disclosure, it
should be apparent that modifications can be made without departing
from the spirit and scope of the disclosure.
* * * * *