U.S. patent application number 14/794714 was filed with the patent office on 2016-01-14 for resonator balancing in wireless power transfer systems.
The applicant listed for this patent is WiTricity Corporation. Invention is credited to Morris P. Kesler, Andre B. Kurs, Guillaume Lestoquoy.
Application Number | 20160012967 14/794714 |
Document ID | / |
Family ID | 53673321 |
Filed Date | 2016-01-14 |
United States Patent
Application |
20160012967 |
Kind Code |
A1 |
Kurs; Andre B. ; et
al. |
January 14, 2016 |
Resonator Balancing in Wireless Power Transfer Systems
Abstract
The disclosure features systems for wireless power transfer that
include a resonator featuring a coil with at least two windings and
at least one inductor having an inductance value, where the at
least one inductor is connected in series to at least one of the
windings, and where the inductance value is selected so that when
the coil carries a current during operation of the system, the at
least one inductor maintains a distribution of current flows among
the at least two windings such that for each of the at least two
windings, an actual current flow in the winding differs from a
target current flow for the winding by 10% or less.
Inventors: |
Kurs; Andre B.; (Chestnut
Hill, MA) ; Lestoquoy; Guillaume; (Cambridge, MA)
; Kesler; Morris P.; (Bedford, MA) |
|
Applicant: |
Name |
City |
State |
Country |
Type |
WiTricity Corporation |
Watertown |
MA |
US |
|
|
Family ID: |
53673321 |
Appl. No.: |
14/794714 |
Filed: |
July 8, 2015 |
Related U.S. Patent Documents
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Application
Number |
Filing Date |
Patent Number |
|
|
62022133 |
Jul 8, 2014 |
|
|
|
62051647 |
Sep 17, 2014 |
|
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Current U.S.
Class: |
307/104 ;
336/182 |
Current CPC
Class: |
H02J 7/025 20130101;
H02J 50/70 20160201; B60L 53/122 20190201; H02J 50/12 20160201;
Y02T 90/12 20130101; H02J 50/005 20200101; H01F 27/2871 20130101;
B60L 2270/147 20130101; B60L 2210/30 20130101; H02J 7/0029
20130101; B60L 2210/10 20130101; B60L 53/126 20190201; H01F 38/14
20130101; Y02T 10/7072 20130101; Y02T 90/14 20130101; B60L 2210/40
20130101; H02J 50/80 20160201; Y02T 10/72 20130101; H02J 50/40
20160201; Y02T 10/70 20130101 |
International
Class: |
H01F 38/14 20060101
H01F038/14; H01F 5/00 20060101 H01F005/00; H02J 5/00 20060101
H02J005/00; H01F 27/28 20060101 H01F027/28 |
Claims
1. A system for wireless power transfer, comprising: a resonator
comprising a coil with at least two windings, each of the at least
two windings comprising a plurality of loops formed by a conductive
material and extending in a plane, wherein corresponding portions
of each of the at least two windings are oriented in parallel,
wherein at least one of the windings has a length that differs from
a length of another one of the windings, and wherein the at least
two windings are electrically connected in parallel; and at least
one inductor having an inductance value, wherein the at least one
inductor is connected in series to at least one of the windings,
wherein the inductance value is selected so that when the coil
carries a current during operation of the system, the at least one
inductor maintains a distribution of current flows among the at
least two windings such that for each of the at least two windings,
an actual current flow in the winding differs from a target current
flow for the winding by 10% or less.
2. The system of claim 1, wherein the at least one inductor
comprises an adjustable inductance value.
3. The system of claim 1, wherein corresponding portions of each of
the at least two windings are oriented in parallel along at least
80% of a length of at least one of the windings.
4. The system of claim 1, wherein the loops of each winding are
interleaved.
5. The system of claim 1, wherein the loops of each winding are
concentric and form a spiral.
6. The system of claim 1, further comprising an electronic
processor coupled to the at least two windings and configured to
control electrical currents in each of the windings based on the
target current flows for the at least two windings.
7. The system of claim 6, wherein the electronic processor is
configured to control electrical currents in each of the windings
by: determining a target inductance value for the at least one
inductor based on a figure of merit related to the target current
flows; and adjusting the inductance value of the at least one
inductor to match the target inductance value.
8. The system of claim 7, wherein the electronic processor is
configured to determine the target inductance value by: (i) for
each one of the windings: determining a self-inductance value of
the one winding based on a measurement of inductance of the one
winding when it is electrically disconnected from all other
windings; and determining a plurality of cross-inductance values of
the one winding, wherein each cross-inductance value is based on a
measurement of inductance of the one winding when it is
electrically disconnected from another one of the windings; (ii)
determining the target current flows for each of the windings based
on the self-inductance values and the cross-inductance values; and
(iii) determining the target inductance value based on the target
current flows for each of the windings.
9. The system of claim 8, wherein the electronic processor is
configured to determine the target current flows by: constructing
an inductance matrix based on the self-inductance values and the
cross-inductance values of each of the windings; calculating an
adjusted inductance matrix by adding to the inductance matrix an
inductance modification matrix comprising elements that correspond
to changes in inductance of each of the windings due to the at
least one inductor; calculating an inverse matrix of the adjusted
inductance matrix; and determining the target current flows based
on the inverse matrix.
10. The system of claim 9, wherein the inductance modification
matrix is a diagonal matrix, and wherein diagonal elements of the
inductance modification matrix are inductance values of respective
members of the at least one inductor connected to the windings.
11. A method, comprising: controlling electrical currents in each
of at least two windings of a resonator coil for wireless power
transfer, wherein each of the at least two windings comprises a
plurality of loops formed by a conductive material and extending in
a plane, wherein corresponding portions of each of the at least two
windings are oriented in parallel, wherein at least one of the
windings has a length that differs from a length of another one of
the windings, wherein the at least two windings are electrically
connected in parallel, and wherein at least one inductor having an
inductance value is connected in series to at least one of the
windings; and maintaining a distribution of current flows among the
at least two windings when the coil carries a current such that for
each of the at least two windings, an actual current flow in the
winding differs from a target current flow for the winding by 10%
or less.
12. The method of claim 11, wherein corresponding portions of each
of the at least two windings are oriented in parallel along at
least 80% of a length of at least one of the windings.
13. The method of claim 11, wherein the loops of each winding are
interleaved.
14. The method of claim 11, wherein the loops of each winding are
concentric and form a spiral.
15. The method of claim 11, further comprising controlling
electrical currents in each of the windings to maintain the
distribution of current flows by: determining a target inductance
value for the at least one inductor based on a figure of merit
related to the target current flows; and adjusting the inductance
value of the at least one inductor to match the target inductance
value.
16. The method of claim 15, further comprising determining the
target inductance value by: (i) for each one of the windings:
determining a self-inductance value of the one winding based on a
measurement of inductance of the one winding when it is
electrically disconnected from all other windings; and determining
a plurality of cross-inductance values of the one winding, wherein
each cross-inductance value is based on a measurement of inductance
of the one winding when it is electrically disconnected from
another one of the windings; (ii) determining the target current
flows for each of the windings based on the self-inductance values
and the cross-inductance values; and (iii) determining the target
inductance value based on the target current flows for each of the
windings.
17. The method of claim 16, further comprising determining the
target current flows by: constructing an inductance matrix based on
the self-inductance values and the cross-inductance values of each
of the windings; calculating an adjusted inductance matrix by
adding to the inductance matrix an inductance modification matrix
comprising elements that correspond to changes in inductance of
each of the windings due to the at least one inductor; calculating
an inverse matrix of the adjusted inductance matrix; and
determining the target current flows based on the inverse
matrix.
18. The method of claim 17, wherein the inductance modification
matrix is a diagonal matrix, and wherein diagonal elements of the
inductance modification matrix are inductance values of respective
members of the at least one inductor connected to the windings.
19. A resonator coil for wireless power transfer, the coil
comprising: a member formed of magnetic material; and at least two
windings electrically connected in parallel, each of the at least
two windings comprising a plurality of loops formed by a conductive
material, wherein the loops of each of the at least two windings
are interleaved so that corresponding portions of each of the at
least two windings are oriented in parallel along at least 80% of a
length of at least one of the windings; and wherein each one
winding of the at least two windings spatially overlaps at least
one other winding at one or more points along a length of the one
winding.
20. The coil of claim 19, further comprising at least one inductor
having an adjustable inductance connected in series to at least one
of the windings.
Description
CROSS-REFERENCE TO RELATED APPLICATIONS
[0001] This application claims priority to U.S. Provisional Patent
Application No. 62/022,133, filed on Jul. 8, 2014, and to U.S.
Provisional Patent Application No. 62/051,647, filed on Sep. 17,
2014, the entire contents of each of which are incorporated herein
by reference.
TECHNICAL FIELD
[0002] This disclosure relates to wireless power transfer systems
and methods.
BACKGROUND
[0003] Energy can be transferred from a power source to a receiving
device using a variety of known techniques such as radiative
(far-field) techniques. For example, radiative techniques using
low-directionality antennas can transfer a small portion of the
supplied radiated power, namely, that portion in the direction of,
and overlapping with, the receiving device used for pick up. In
this example, most of the energy is radiated away in directions
other than the direction of the receiving device, and typically the
transferred energy is insufficient to power or charge the receiving
device. In another example of radiative techniques, directional
antennas are used to confine and preferentially direct the radiated
energy towards the receiving device. In this case, an
uninterruptible line-of-sight and potentially complicated tracking
and steering mechanisms are used.
[0004] Another approach is to use non-radiative (near-field)
techniques. For example, techniques known as traditional induction
schemes do not (intentionally) radiate power, but use an
oscillating current passing through a primary coil, to generate an
oscillating magnetic near-field that induces currents in a near-by
receiving or secondary coil. Traditional induction schemes can
transfer modest to large amounts of power over very short
distances. In these schemes, the offset tolerances between the
power source and the receiving device are very small. Electric
transformers and proximity chargers use these traditional induction
schemes.
SUMMARY
[0005] In general, in a first aspect, the disclosure features
systems for wireless power transfer that include: a resonator
including a coil with at least two windings, each of the at least
two windings featuring a plurality of loops formed by a conductive
material and extending in a plane, where corresponding portions of
each of the at least two windings are oriented in parallel, where
at least one of the windings has a length that differs from a
length of another one of the windings, and where the at least two
windings are electrically connected in parallel; and at least one
inductor having an inductance value, where the at least one
inductor is connected in series to at least one of the windings,
and where the inductance value is selected so that when the coil
carries a current during operation of the system, the at least one
inductor maintains a distribution of current flows among the at
least two windings such that for each of the at least two windings,
an actual current flow in the winding differs from a target current
flow for the winding by 10% or less.
[0006] Embodiments of the systems can include any one or more of
the following features.
[0007] The at least one inductor can include an adjustable
inductance value. Corresponding portions of each of the at least
two windings can be oriented in parallel along at least 80% of a
length of at least one of the windings. The loops of each winding
can be interleaved. The loops of each winding can be concentric and
form a spiral.
[0008] The systems can include an electronic processor coupled to
the at least two windings and configured to control electrical
currents in each of the windings based on the target current flows
for the at least two windings. The electronic processor can be
configured to control electrical currents in each of the windings
by: determining a target inductance value for the at least one
inductor based on a figure of merit related to the target current
flows; and adjusting the inductance value of the at least one
inductor to match the target inductance value. The electronic
processor can be configured to determine the target inductance
value by: (i) for each one of the windings, determining a
self-inductance value of the one winding based on a measurement of
inductance of the one winding when it is electrically disconnected
from all other windings, and determining a plurality of
cross-inductance values of the one winding, where each
cross-inductance value is based on a measurement of inductance of
the one winding when it is electrically disconnected from another
one of the windings; (ii) determining the target current flows for
each of the windings based on the self-inductance values and the
cross-inductance values; and (iii) determining the target
inductance value based on the target current flows for each of the
windings. The electronic processor can be configured to determine
the target current flows by: constructing an inductance matrix
based on the self-inductance values and the cross-inductance values
of each of the windings; calculating an adjusted inductance matrix
by adding to the inductance matrix an inductance modification
matrix comprising elements that correspond to changes in inductance
of each of the windings due to the at least one inductor;
calculating an inverse matrix of the adjusted inductance matrix;
and determining the target current flows based on the inverse
matrix. The inductance modification matrix can be a diagonal
matrix, and diagonal elements of the inductance modification matrix
can be inductance values of respective members of the at least one
inductor connected to the windings.
[0009] In another aspect, the disclosure features methods that
include: controlling electrical currents in each of at least two
windings of a resonator coil for wireless power transfer, where
each of the at least two windings includes a plurality of loops
formed by a conductive material and extending in a plane, where
corresponding portions of each of the at least two windings are
oriented in parallel, where at least one of the windings has a
length that differs from a length of another one of the windings,
where the at least two windings are electrically connected in
parallel, and where at least one inductor having an inductance
value is connected in series to at least one of the windings; and
maintaining a distribution of current flows among the at least two
windings when the coil carries a current such that for each of the
at least two windings, an actual current flow in the winding
differs from a target current flow for the winding by 10% or
less.
[0010] Embodiments of the methods can include any one or more of
the following features.
[0011] Corresponding portions of each of the at least two windings
can be oriented in parallel along at least 80% of a length of at
least one of the windings. The loops of each winding can be
interleaved. The loops of each winding can be concentric and form a
spiral.
[0012] The methods can include controlling electrical currents in
each of the windings to maintain the distribution of current flows
by: determining a target inductance value for the at least one
inductor based on a figure of merit related to the target current
flows; and adjusting the inductance value of the at least one
inductor to match the target inductance value. The methods can
include determining the target inductance value by: (i) for each
one of the windings, determining a self-inductance value of the one
winding based on a measurement of inductance of the one winding
when it is electrically disconnected from all other windings, and
determining a plurality of cross-inductance values of the one
winding, where each cross-inductance value is based on a
measurement of inductance of the one winding when it is
electrically disconnected from another one of the windings; (ii)
determining the target current flows for each of the windings based
on the self-inductance values and the cross-inductance values; and
(iii) determining the target inductance value based on the target
current flows for each of the windings.
[0013] The methods can include determining the target current flows
by: constructing an inductance matrix based on the self-inductance
values and the cross-inductance values of each of the windings;
calculating an adjusted inductance matrix by adding to the
inductance matrix an inductance modification matrix featuring
elements that correspond to changes in inductance of each of the
windings due to the at least one inductor; calculating an inverse
matrix of the adjusted inductance matrix; and determining the
target current flows based on the inverse matrix. The inductance
modification matrix can be a diagonal matrix, and diagonal elements
of the inductance modification matrix can be inductance values of
respective members of the at least one inductor connected to the
windings.
[0014] In a further aspect, the disclosure features resonator coils
for wireless power transfer that include a member formed of
magnetic material and at least two windings electrically connected
in parallel, each of the at least two windings featuring a
plurality of loops formed by a conductive material, where the loops
of each of the at least two windings are interleaved so that
corresponding portions of each of the at least two windings are
oriented in parallel along at least 80% of a length of at least one
of the windings, and where each one winding of the at least two
windings spatially overlaps at least one other winding at one or
more points along a length of the one winding.
[0015] Embodiments of the coils can include any one or more of the
following features.
[0016] The loops of each winding can be oriented in a plane. The
loops of all windings can be oriented in a common plane. The loops
of each winding can be concentric and can form a spiral. The loops
of all windings can form a concentric spiral of loops.
[0017] The coils can include at least one inductor having an
adjustable inductance connected in series to at least one of the
windings. The coils can include at least one inductor having an
adjustable inductance connected in series to each of the at least
two windings. Each one winding of the at least two windings can
spatially overlap each of the other windings at one or more points
along the length of the one winding. For each one winding, the
points at which the one winding overlaps the at least some of the
other windings can be equally spaced along a circumference of the
one winding.
[0018] The at least two windings can include n windings, each one
winding of the n windings can spatially overlap each of the other
n-1 windings along the length of the one winding, and each one
winding of the n windings can include n-1 points of overlap with
the other windings, each one of the points corresponding to overlap
of the one winding with a different one of the other windings. The
quantity n can be greater than two (e.g., greater than three).
[0019] The at least two windings can include n windings, and at
least one winding of the n windings can include more than n-1
points of overlap with the other windings so that the at least one
winding spatially overlaps at least some of the other n-1 windings
more than once. The at least two windings can include n windings,
and at least one winding of the n windings can include fewer than
n-1 points of overlap with the other windings so that the at least
one winding does not spatially overlap all of the other n-1
windings.
[0020] Embodiments of the systems, methods, and coils can also
include any of the other features disclosed herein, including
features disclosed in connection with different embodiments, in any
combination as appropriate.
[0021] Unless otherwise defined, all technical and scientific terms
used herein have the same meaning as commonly understood by one of
ordinary skill in the art to which this disclosure belongs. In ease
of conflict with publications, patent applications, patents, and
other references mentioned or incorporated herein by reference, the
present disclosure, including definitions, will control. Any of the
features described above may be used, alone or in combination,
without departing from the scope of this disclosure. Other
features, objects, and advantages of the systems and methods
disclosed herein will be apparent from the following detailed
description and figures.
DESCRIPTION OF DRAWINGS
[0022] FIG. 1A is a schematic diagram of an embodiment of a
wireless power transfer system.
[0023] FIG. 1B is a schematic diagram of a resonator for wireless
power transfer.
[0024] FIGS. 2A-2C are schematic diagrams of a wireless power
transfer system integrated into a vehicle.
[0025] FIGS. 3A and 3B are schematic diagrams showing a source
resonator coil in proximity to a magnetic material.
[0026] FIG. 4 is a schematic diagram of a source resonator.
[0027] FIGS. 5A and 5B are schematic diagrams of devices configured
to wirelessly receiver power.
[0028] FIGS. 6A and 6B are plots showing measurements of coupling k
between a source resonator and a receiver resonator as a function
of relative displacement of the resonators.
[0029] FIGS. 7A and 7B are plots showing figure-of-merit as a
function of relative displacement between source and receiver
resonators.
[0030] FIGS. 8A and 8B are plots showing measurements of coupling k
as a function of relative displacement between source and receiver
resonators.
[0031] FIGS. 9A and 9B are plots showing figure-of-merit as a
function of relative displacement between source and receiver
resonators.
[0032] FIGS. 10A and 10B are plots showing measurements of coupling
k between a source resonator and a receiver resonator as a function
of relative displacement of the resonators.
[0033] FIGS. 11A and 11B are plots showing figure-of-merit as a
function of relative displacement between source and receiver
resonators.
[0034] FIGS. 12A and 12B are plots showing measurements of coupling
k between a source resonator and a receiver resonator as a function
of relative displacement of the resonators.
[0035] FIGS. 13A and 13B are plots showing figure-of-merit as a
function of relative displacement between source and receiver
resonators.
[0036] FIGS. 14A and 14B are plots showing measurements of coupling
k between a source resonator and a receiver resonator as a function
of relative displacement of the resonators.
[0037] FIGS. 15A and 15B are plots showing quality factor as a
function of relative displacement between source and receiver
resonators.
[0038] FIGS. 16A and 16B are plots showing figure-of-merit as a
function of relative displacement between source and receiver
resonators.
[0039] FIGS. 17A and 17B are schematic diagrams of impedance
matching networks.
[0040] FIGS. 18A and 18B are plots of device-side load impedance as
a function of output voltage.
[0041] FIGS. 19A and 19B are plots of amplifier-to-battery
efficiency as a function of output voltage.
[0042] FIGS. 20A and 20B are plots of power dissipated in a source
and a device as a function of output voltage.
[0043] FIGS. 21A and 21B are plots of voltage measured across one
or more capacitors in an impedance matching network.
[0044] FIGS. 22A and 22B are plots of the magnetic field amplitude
in a magnetic member of a source resonator.
[0045] FIG. 23 is a schematic circuit diagram of an embodiment of
device electronics.
[0046] FIGS. 24A and 24B are plots of a DC-DC boost conversion
ratio as a function of output voltage.
[0047] FIGS. 25A and 25B are plots of amplifier-to-converter
efficiency as a function of output voltage.
[0048] FIGS. 26A and 26B are plots of power dissipated in a source
as a function of output voltage.
[0049] FIGS. 27A and 27B are plots of measured voltages across one
or more matching network capacitors in a source.
[0050] FIGS. 28A and 28B are plots of the magnetic field in a
magnetic member attached to a source.
[0051] FIGS. 29A-29D are schematic diagrams showing embodiments of
resonators for wireless power transfer.
[0052] FIG. 30A is a plot of coupling rate as a function of source
resonator coil winding length.
[0053] FIG. 30B is a plot of figure-of-merit as a function of
source resonator coil winding length.
[0054] FIGS. 31A-31D are schematic diagrams of resonator coils.
[0055] FIG. 32A is a plot of coupling rate as a function of
resonator coil winding gap-to-edge distance and span.
[0056] FIG. 32B is a plot of figure-of-merit as a function of
resonator coil winding gap-to-edge distance and span.
[0057] FIGS. 33A and 33B are plots of coupling between source and
device resonators as a function of offset between the
resonators.
[0058] FIGS. 34A and 34B are plots of figure-of-merit for a
wireless power transfer system as a function of offset between
source and device resonators.
[0059] FIGS. 35A and 35B are schematic diagrams of impedance
matching networks.
[0060] FIG. 36 is a plot showing a minimum number of capacitors for
different impedance matching networks.
[0061] FIG. 37 is a schematic diagram of an impedance matching
network.
[0062] FIG. 38 is a plot of figure-of-merit as a function of output
voltage in an impedance-matched device.
[0063] FIG. 39 is a plot of power dissipated in source and receiver
resonators that are impedance-matched.
[0064] FIG. 40 is a plot of maximum magnetic field in source and
device resonators as a function of output voltage.
[0065] FIG. 41 is a plot of voltage across one or more capacitors
of an impedance matching network in source and receiver
resonators.
[0066] FIG. 42 is a schematic diagram showing an embodiment of a
magnetic member.
[0067] FIG. 43 is an image of an embodiment of a resonator.
[0068] FIG. 44 is an image of an embodiment of a source resonator
coil affixed to a magnetic member.
[0069] FIG. 45A is a schematic diagram of an embodiment of a
magnetic member.
[0070] FIG. 45B is an image of an embodiment of a magnetic
member.
[0071] FIG. 46 is an image of an embodiment of a device receiver
resonator coil.
[0072] FIG. 47A is a schematic diagram of an embodiment of a source
resonator coil.
[0073] FIG. 47B is a schematic diagram of an embodiment of a device
resonator coil.
[0074] FIGS. 48A and 48B are schematic diagrams of embodiments of
source resonator coils affixed to magnetic members.
[0075] FIG. 49 is an image of an embodiment of a resonator coil
with parallel windings.
[0076] FIG. 50 is a plot of the magnetic field in the magnetic
member of a source resonator.
[0077] FIG. 51 is a plot of the magnetic field in the magnetic
member of a source resonator.
[0078] FIG. 52 is a plot of the magnetic field in the magnetic
member of a source resonator.
[0079] FIG. 53 is a plot of the magnetic field in the magnetic
member of a device resonator.
[0080] FIGS. 54A and 54B are plots of the magnetic field above a
source resonator.
[0081] FIG. 55 is a plot of the magnetic field of FIGS. 54A and 54B
on a logarithmic scale.
[0082] FIG. 56 is an image of a coil with parallel windings.
[0083] FIGS. 57A and 57B are schematic diagrams showing inductors
connected in series with parallel windings of a coil.
[0084] FIG. 57C is a schematic diagram of a single inductor
connected in series with one of several parallel windings of a
coil.
[0085] FIGS. 58A and 58B are images showing temperature
measurements for parallel windings in a coil.
[0086] FIG. 59 is a plot showing the current carried in each of the
parallel windings of a coil as a function of time.
[0087] FIGS. 60A and 60B are plots showing measurements of coupling
k between a source resonator and a receiver resonator as a function
of relative displacement of the resonators.
[0088] FIGS. 61A and 61B are plots showing measurements of coupling
k between a source resonator and a receiver resonator as a function
of relative displacement of the resonators.
[0089] FIG. 62 is a schematic diagram of a receiver resonator.
[0090] FIG. 63 is a plot of Q-factor for source and receiver
resonators as a function of a side length of a shield.
[0091] FIGS. 64A and 64B are plots of peak current through the
diodes of a rectifier in an impedance matching network.
[0092] FIGS. 65A and 65B are plots of peak current through the
diodes of a rectifer in an impedance matching network.
[0093] FIGS. 66A and 66B are plots showing the number of capacitors
in a device's impedance matching network as a function of device
inductance.
[0094] FIG. 67 is a schematic diagram of an impedance matching
network.
[0095] FIG. 68 is a plot of figure of merit as a function of output
voltage for an impedance matching network.
[0096] FIG. 69 is a plot of power dissipation in various components
of a source.
[0097] FIG. 70 is a plot of power dissipation in various components
of a device.
[0098] FIG. 71 is a plot of the magnetic field in a magnetic member
as a function of output voltage.
[0099] FIG. 72 is a plot of voltages across capacitors in an
impedance matching network as a function of output voltage.
[0100] FIG. 73 is a plot of electrical current through the source
and device resonator coils as a function of output voltage.
[0101] FIG. 74 is a flow chart that includes a series of steps for
current partitioning and balancing among parallel windings of a
coil.
[0102] FIG. 75 is a schematic diagram of an electronic
controller.
DETAILED DESCRIPTION
[0103] The wireless power transfer systems disclosed herein use one
or more source resonators to generate oscillating magnetic fields.
The oscillating magnetic fields are captured by, and induce
electrical currents and voltages in, one or more receiving
resonators. The receiving resonators can be coupled to loads, and
the electrical currents and voltages can be used to drive the loads
to do useful work. The receiving resonators can also act as relay
resonators, further transmitting power wirelessly by generating
additional oscillating magnetic fields.
[0104] Wireless power transfer systems can be integrated into a
variety of devices and used for a wide range of power-demanding
applications. For example, such systems can be integrated into
electric vehicles and used to power and/or charge the vehicles.
Such systems can also be used to power electronic devices,
including fixed and portable devices, and can be integrated into a
diverse range of structures including furniture (e.g., desks,
tables) and structural features (e.g., floors, walls, columns,
streets). The systems can provide power in quantities that range
from very small amounts to significant quantities for high-power
applications. For example, the systems disclosed herein may provide
power greater than 1 kW, 3 kW, 5 kW, 10 kW, 20 kW, 50 kW, or more
from one or more source resonators to one or more receiving
resonators connected to electrical devices.
Introduction
[0105] FIG. 1A shows a schematic diagram of an embodiment of a
wireless power transfer system 100 that includes a wireless power
source 101 and device 107. Wireless power source 101 includes a
source resonator 102 coupled to source electronics 104, which are
connected to a power supply 106. Source electronics 104 can include
a variety of components including an AC/DC converter, an amplifier,
and an impedance matching network. Power supply 106 can include one
or more of AC mains, solar panels, and one or more batteries. Not
all of the components of power source 101 need to be present for
operation, and in some embodiments, certain components shown in
FIG. 1A can be integrated (source electronics 104 and power supply
106 can be integrated into a single component, for example).
[0106] Device 107 includes a device resonator 108 coupled to device
electronics 110 to provide power to a load 112. Device electronics
110 can include a variety of components, such as a rectifier and/or
an impedance matching network. Load 112 generally corresponds to
any of a variety of power-dissipating electrical components, such
as a battery and/or an electromechanical device. Not all of the
components of device 107 need to be present for operation, and in
some embodiments, certain components shown in FIG. 1A can be
integrated (device electronics 110 and load 112 can be integrated
into a single component, for example).
[0107] Source electronics 104 and device electronics 110 can each
include one or more electronic processors (processors 105 and 111,
respectively). Electronic processors 105 and 111 can perform a
variety of monitoring, computation, and control functions. For
example, as will be described in more detail subsequently,
processors 105 and/or 111 can measure electrical parameters of
various system components (by directing suitable control signals to
various sensors), calculate various performance-related metrics and
attributes based on measurement signals received from the sensors,
and transmit control signals to various system components based on
the calculated metrics and attributes. In general, processors 105
and 111 can be configured to perform any of the monitoring,
computational, and control functions disclosed herein. In addition,
or as an alternative, source electronics 104 and/or device
electronics 111 can include dedicated electrical circuits (e.g.,
application-specific integrated circuits) and logic units (e.g.,
programmable logic arrays) that can be configured to perform any
one or more of these functions.
[0108] Processors 105 and/or 111 can be coupled to one or more
components of system 100 in various configurations. In some
embodiments, processors 105 and/or 111 are coupled to system
components via a direct electrical connection. In certain
embodiments, processors 105 and/or 111 are coupled to system
components via wireless communication (e.g., radio-frequency,
Bluetooth communication). The coupling between the processors and
the system components can be different for different system
components. For example, processor 105 can be directly connected to
power supply 106 and source resonator 102, and coupled wirelessly
to device resonator 108 and/or device electronics 110.
[0109] Additional aspects and features of wireless power transfer
systems are disclosed, for example, in the following, the entire
contents of each of which are incorporated herein by reference:
U.S. Patent Application Publication No. 2012/0119569; U.S. Patent
Application Publication No. 2015/0051750; U.S. Pat. No. 8,772,973;
U.S. Patent Application Publication No. 2010/0277121; and U.S. Pat.
No. 8,598,743.
[0110] In some embodiments, processor 105 can direct power supply
106 to provide power to source resonator 102. For example,
processor 105 can increase the power output of power supply 106,
thereby increasing the power delivered to source resonator 102. The
power output can be delivered at an operating frequency
corresponding to a frequency of the oscillating magnetic field that
is generated by source resonator 102.
[0111] In certain embodiments, processor 105 (and/or processor 111)
can tune a resonant frequency of source resonator 102 and/or a
resonant frequency of device resonator 108. By tuning resonant
frequencies of the source and device resonators relative to the
operating frequency of power supply 106, the efficiency of power
transfer from power supply 106 to load 112 can be controlled. For
example, processor 105 (and/or processor 111) can tune the resonant
frequencies of source resonator 102 and/or device resonator 108 to
be substantially the same (e.g., within 0.5%, within 1%, within 2%)
to increase the efficiency of power transfer.
[0112] In some embodiments, processors 105 and/or 111 can tune the
resonant frequencies by adjusting capacitance values of components
in source resonator 102 and/or source electronics 104. Resonant
frequencies can also be tuned by adjusting capacitance values of
components in device resonator 108 and/or device electronics 110.
For example, to tune the resonance frequency of source resonator
102, processor 105 can adjust a capacitance of a capacitor
connected to a coil in source resonator 102. The adjustment can be
based on a measurement of the resonance frequency by processor 105
and/or based on a communication signal transmitted from source
resonator 102 and/or device resonator 108 to processor 105 (e.g.,
transmitted wirelessly). In certain embodiments, processor 105 can
tune the resonant frequency of source resonator 102 to be
substantially the same (e.g., within 0.5%, within 1%, within 2%) as
the operating frequency of power supply 106. In some embodiments,
processor 105 can tune the resonant frequency of source resonator
102 to be different from the operating frequency by 7% to 13%
(e.g., 10% to 15%, 13% to 19%). Similar considerations apply to the
tuning of the resonance frequency of device resonator 108 (e.g., by
processor 111 and/or processor 105).
[0113] In some embodiments, processors 105 and/or 111 can control
an impedance matching network in system 100 to adjust impedance
matching conditions in the system, and thereby control the
efficiency of power transfer. For example, processor 105 can tune
the capacitance of capacitors or networks of capacitors in an
impedance matching network connected between power supply 106 and
source resonator 102 (e.g., as part of source electronics 104).
Alternatively, or in addition, processor 105 can tune the
inductance of inductors or networks of inductors in an impedance
matching network. The optimum impedance conditions can be
calculated by processor 105 and/or can be received from an external
device.
[0114] Similarly, in certain embodiments, processor 111 can control
impedance matching conditions by tuning the capacitance and/or
inductance of capacitors and/or inductors, respectively, in an
impedance matching network connected between device resonator 108
and load 112 (e.g., as part of device electronics 110). Additional
aspects of frequency tuning and impedance matching networks are
disclosed, for example, in U.S. Patent Application Publication No.
2015/0051750, the entire contents of which are incorporated herein
by reference.
[0115] In this disclosure, "wireless energy transfer" from one coil
(e.g., a resonator coil) to another coil (e.g., another resonator
coil) refers to transferring energy to do useful work (e.g.,
electrical work, mechanical work, etc.) such as powering electronic
devices, vehicles, lighting a light bulb or charging batteries.
Similarly, "wireless power transfer" from one coil (e.g., resonator
coil) to another resonator (e.g., another resonator coil) refers to
transferring power to do useful work (e.g., electrical work,
mechanical work, etc.) such as powering electronic devices,
vehicles, lighting a light bulb or charging batteries. Both
wireless energy transfer and wireless power transfer refer to the
transfer (or equivalently, the transmission) of energy to provide
operating power that would otherwise be provided through a wired
connection to a power source, such as a connection to a main
voltage source. With the above understanding, the expressions
"wireless energy transfer" and "wireless power transfer" are used
interchangeably in this disclosure. It should also be understood
that, "wireless power transfer" and "wireless energy transfer" can
be accompanied by the transfer of information; that is, information
can be transferred via an electromagnetic signal along with the
energy or power to do useful work.
[0116] FIG. 1B is a schematic diagram showing a portion of a
resonator 150 used for wireless power transfer. Resonator 150
includes a coil 152, a magnetic member 154, and a shield 156. Coil
152 includes one or more loops and can be connected to one or more
capacitors and/or inductors, as well as other electrical components
(not shown). Coil 152 is formed of one or more conductive
materials, such as copper, silver, gold, and Litz wire. As an
example, Litz wire can be used for operation at frequencies of
lower than 1 MHz (e.g., 85 kHz). In certain embodiments, the coil
210 can be formed of a solid core wire, or one or more conducting
layers (e.g., copper layers) formed on a printed circuit board
(PCB). For example, solid core wire or conducting layers can be
used at operation frequencies of 1 MHz or higher.
[0117] Magnetic member 154 is positioned between coil 152 and
shield 154. That is, in FIG. 1A, coil 152 is positioned on one side
of magnetic member 154 and shield 156 is positioned on the opposite
side of magnetic member 156. In general, magnetic member 154 guides
magnetic flux induced by current flowing in the loops of coil 152.
The presence of magnetic member 154 can lead to an increase in the
magnetic flux density generated by coil 152 in a region adjacent to
coil 152 (i.e., in a plane above or below the plane of coil 152)
when oscillating electrical currents circulate in coil 152,
relative to the flux density in the absence of magnetic member
154.
[0118] In some embodiments, magnetic member 154 can include one or
more magnetic elements formed from magnetic materials such as
manganese-zinc (MnZn) and/or nickel-zinc (NiZn) ferrites. When
member 154 is formed from multiple magnetic elements, the gaps
between elements (not shown in FIG. 1B) can be filled with a
dielectric material such as an adhesive.
[0119] While magnetic materials are generally available in small
sizes, some applications for wireless power transfer utilize
magnetic members with a large areal size. For example, a car
battery charging application may use magnetic members of large
areal size (e.g., 30 cm.times.30 cm) to transfer high power of 1 kW
or more (e.g., 2 kW or more, 3 kW or more, 5 kW or more, 6 kW or
more). Magnetic members featuring a single monolithic piece of
material can be utilized when such a piece of material is
available. However, it can be difficult and/or expensive to
manufacture a monolithic piece of magnetic material such as MnZn or
NiZn ferrites with a large areal size (e.g., 30 cm.times.30 cm) for
high power transfer. Moreover, MnZn and NiZn ferrites can be
brittle, and accordingly, large-area pieces of these materials can
be highly susceptible to breakage.
[0120] To overcome such difficulties, ferrite materials can be
manufactured in pieces of small areal size (e.g., 5 cm.times.5cm),
and several such pieces can be joined together to form a larger
combined magnetic member. The smaller magnetic elements can behave
functionally in a collective manner very similar to a larger
magnetic member when they are joined. In certain embodiments, the
multiple magnetic elements can be contained in a holder made from
thermally conducting and electrically insulating materials (e.g.,
plastic, Teflon.RTM., aluminum oxide, aluminum nitride, etc.)
[0121] Shield 156, which generally corresponds to a sheet of
electrically conductive material, is typically positioned in
proximity to coil 152. Shield 156 can be formed from one or more
conductive materials, which can be the same as, or different from,
the conductive materials used to form coil 152. For example, shield
156 can be formed from a sheet of a material such as copper,
silver, gold, iron, steel, nickel and/or aluminum. Shield 156 acts
to shield coil 152 from loss-inducing objects (e.g., metallic
objects). Further, in some embodiments, shield 156 can increase
coupling of resonator 150 to another resonator by guiding magnetic
field lines in the vicinity of the resonator. For example, energy
loss from aberrant coupling to loss-inducing objects can be reduced
by using shield 156 to guide magnetic field lines away from the
loss-inducing objects.
[0122] FIGS. 2A-2C are schematic diagrams showing a wireless power
transfer system 204 integrated into a vehicle 202. FIG. 2A shows a
side view of vehicle 202 in the X-Z coordinate plane, FIG. 2B shows
a top view of vehicle 202 in the X-Y coordinate plane, and FIG. 2C
shows a front view of vehicle 202 in the Y-Z coordinate plane. For
purposes of the following discussion, the X-axis corresponds to the
"front-to-back" direction of the vehicle, the Y-axis corresponds to
the "side-to-side" direction of the vehicle, and the Z-axis
corresponds to the "top-to-bottom" direction of the vehicle.
[0123] For wireless power transfer in vehicle applications, source
and device resonators can be relatively large to accommodate
significant power transfer between the resonators. In some
embodiments, for example, the source resonator can have a maximum
dimension in the X-Y plane of 30 cm or more (e.g., 40 cm or more,
50 cm or more, 60 cm or more, 70 cm or more, 80 cm or more, 90 cm
or more, 100 cm or more). In certain embodiments, the device
resonator can have a maximum dimension in the X-Y plane of 20 cm or
more (e.g., 30 cm or more, 40 cm or more, 50 cm or more, 60 cm or
more, 70 cm or more, 80 cm or more, 90 cm or more, 100 cm or more).
In some embodiments, a maximum dimension of the source resonator
can be smaller than a maximum dimension of the device resonator by
10 cm or more (e.g., by 15 cm or more, by 20 cm or more, by 30 cm
or more).
[0124] The source and device resonator can each have a variety of
different cross-sectional shapes, including square, rectangular,
circular, elliptical, and more generally, regular polygonal. In
certain embodiments, the resonators can have different shapes. For
example, the source resonator can have a square cross-sectional
shape, while the device resonator can have a rectangular
cross-sectional shape.
[0125] The resonators (e.g., source resonators, receiving
resonators, repeater resonators) used in the wireless power
transfer systems disclosed herein can have a resonant frequency
f=.omega./2.pi., an intrinsic loss rate .GAMMA., and a Q-factor
Q=.omega./(2.GAMMA.) (also referred as "intrinsic" Q-factor in this
disclosure), where .omega. is the angular resonant frequency. The
resonant frequency f of a source or receiver resonator is typically
determined by the resonator's capacitance and inductance
values.
[0126] In some embodiments, any one of a source, receiver, and/or
repeater resonator can have a Q-factor that is a high Q-factor
where Q>100 (e.g., Q>100, Q>200, Q>300, Q>500,
Q>1000). For example, wireless power transfer systems can
include a power source having one or more source resonators, and at
least one of the source resonators can have a Q-factor of
Q.sub.1>100 (e.g., Q.sub.1>100, Q.sub.1>200,
Q.sub.1>300, Q.sub.1>500, Q.sub.1>1000). The wireless
power transfer system can include a power receiver having one or
more receiver resonators, and at least one of the receiver
resonators can have a Q-factor of Q.sub.2>100 (e.g.,
Q.sub.2>100, Q.sub.2>200, Q.sub.2>300, Q.sub.2>500,
Q.sub.2>1000). The system can include at least one repeater
resonator having a Q-factor of Q.sub.3>100 (e.g.,
Q.sub.3>100, Q.sub.3>200, Q.sub.3>300, Q.sub.3>500,
Q.sub.3>1000).
[0127] Utilizing high Q-factor resonators can lead to large energy
coupling between some or all of the resonators in a wireless power
transfer system. The high Q factors can lead to strong coupling
between resonators such that the "coupling time" between the
resonators is shorter than the "loss time" of the resonators. As a
consequence, energy can be transferred efficiently between
resonators at a faster rate than the energy loss rate due to losses
(e.g., heating loss, radiative loss) of the resonators. In certain
embodiments, a geometric mean, {square root over (Q.sub.iQ.sub.j)}
can be larger than 100 (e.g., {square root over
(Q.sub.iQ.sub.j)}>200, {square root over
(Q.sub.iQ.sub.j)}>300, {square root over
(Q.sub.iQ.sub.j)}>500, {square root over
(Q.sub.iQ.sub.j)}>1000) where i and j refer to a pair of
source-receiver resonators, source-repeater resonators, or
repeater-receiver resonators (e.g., i=1, j=2, or i=1, j=3, or i=2,
j=3.) Additional aspects of high-Q resonators are described, for
example, in U.S. Pat. No. 8,461,719, the entire contents of which
are incorporated herein by reference.
Resonator Configurations
[0128] The extent of coupling and the efficiency of wireless power
transfer between two resonators in a wireless power transfer system
depends upon a wide variety of different structural features of the
resonators. As such, different resonator configurations achieve
different power transfer efficiencies and rates; and thus,
different configurations are suitable for different types of power
transfer applications. In the following sections, a number of
different resonator configurations are shown, and the effect of
different structural features on wireless power transfer
performance will be discussed.
[0129] In some embodiments, a resonator coil can be offset from a
conductive shield (e.g., an aluminum shield) to decrease losses and
increase coupling to another resonator. FIGS. 3A-3B are schematic
diagrams showing a source resonator coil 302 in proximity to a
magnetic material 304, with a gap between magnetic material 304 and
a conductive shield 306. In FIG. 3A, there is no gap between
magnetic material 304 and shield 306. In FIG. 3B, there is a 40 mm
gap 308 between magnetic material 304 and shield 306 (an aluminum
shield). For a source resonator of dimensions 60 cm by 60 cm in the
X-Y plane, and offset in the Z-direction from a device resonator of
size 25 cm by 50 cm (not shown in FIGS. 3A-3B), with the offset
from the device resonator defined by the set of coordinates
(X,Y,Z)=(0,0,15) cm, a coupling rate k is measured to be
approximately 0.077 for source resonator 302 shown in FIG. 3A, and
approximately 0.083 for source resonator 302 shown in FIG. 3B.
[0130] In general, the thickness of magnetic material 304 in
proximity to (or even attached to) a resonator can be varied to
adjust the coupling k to another resonator. Table 1 summarizes
measurements of coupling rate k for a wireless transfer system that
includes a source resonator coil 402 of size 60 cm by 60 cm in the
X-Y plane, as shown in FIG. 4, with an offset defined by the set of
coordinates (X,Y,Z)=(10,10,150) cm from a device resonator coil 404
of size 25 cm by 50 cm in the X-Y plane. The source resonator coil
and magnetic material are spaced from the aluminum shield by a gap
of 40 mm, as in FIG. 3B discussed previously. Measurements of k are
taken for a source resonator coil 402 having different thicknesses
of magnetic material (e.g., "ferrite"), and under conditions when a
vehicle chassis is present and not present.
TABLE-US-00001 TABLE 1 Presence of Vehicle Chassis Ferrite
thickness Coupling k Chassis present 5 mm 0.060 Chassis not present
5 mm 0.075 Chassis not present 12 mm 0.083
[0131] In some embodiments, a device configured to receive power
wirelessly can be house both a device resonator and device
electronics an integrated manner. FIG. 5A is a schematic diagram
showing an embodiment of a device configured to wirelessly receive
power in which a device resonator coil 502, a magnetic material
504, and a conductive (e.g., aluminum) shield 506 are stacked onto
one another. FIG. 5B shows a schematic diagram of another
embodiment of a device configured to wirelessly receive power. The
device of FIG. 5B has a "top-hat" configuration, in which a center
portion of magnetic material 508 is stepped in the Z-direction to
form an empty region between magnetic material 508 and shield 506.
Device electronics 510 are positioned within the empty region and
coil 502 is wound around the stepped edges of magnetic material
508. By enclosing device electronics 510 within the device
resonator as shown in FIG. 5B, the compactness of the device can be
significantly increased.
[0132] The coupling k between source and receiver resonators in a
vehicle wireless power transfer system depends in part on the
presence and nature of the vehicle chassis in proximity to the
receiver resonator. FIGS. 6A and 6B are plots that show
measurements of the coupling k between a source resonator and a
receiver resonator 604 as a function of relative displacement
between the centers of the resonators in both the X- and
Z-directions. The receiver resonator is similar to the resonator
shown in FIG. 5A and the source resonator is similar to the
resonator shown in FIG. 3B. The plot in FIG. 6A shows measurements
of the coupling k in the absence of a vehicle chassis, while the
plot in FIG. 6B shows measurements in the presence of an aluminum
vehicle chassis. It is evident from FIGS. 6A and 6B that the
vehicle chassis reduces the value of the coupling k by
approximately 20%.
[0133] FIGS. 7A and 7B are plots showing figure-of-merit (U.sub.0)
measurements as a function of relative displacement between the
centers of a source resonator 602 and receiver resonator 604, in
the X- and Z-directions. The receiver resonator is similar to the
resonator shown in FIG. 5A and the source resonator is similar to
the resonator shown in FIG. 3B. The plot in FIG. 7A shows
measurements of U.sub.0 in the absence of a vehicle chassis, while
the plot in FIG. 7B shows measurements in the presence of an
aluminum vehicle chassis. In FIG. 7A, the quality factor of the
source resonator is approximately 1000 while the quality factor of
the receiver resonator is approximately 380. In FIG. 7B, the
quality factor of the source resonator is approximately 1000, while
the quality factor of the receiver resonator is approximately
460.
[0134] FIGS. 8A and 8B are plots showing measurements of the
coupling k as a function of relative displacement between the
centers of a source resonator 602 and a receiver resonator 604 in
the Y- and Z-directions. The receiver resonator is similar to the
resonator shown in FIG. 5A and the source resonator is similar to
the resonator shown in FIG. 3B. The plot in FIG. 8A shows
measurements of k in the absence of a vehicle chassis, while the
plot in FIG. 8B shows measurements in the presence of an aluminum
vehicle chassis. It is evident that the vehicle chassis reduces the
coupling k by approximately 20%.
[0135] FIGS. 9A and 9B are plots showing figure-of-merit (U.sub.0)
measurements as a function of relative displacement between the
centers of a source resonator 602 and a receiver resonator 604 in
the Y- and Z-directions. The receiver resonator is similar to the
resonator shown in FIG. 5A and the source resonator is similar to
the resonator shown in FIG. 3B. The plot in FIG. 9A shows
measurements of U.sub.0 in the absence of a vehicle chassis, while
the plot in FIG. 9B shows measurements in the presence of an
aluminum vehicle chassis.
[0136] FIGS. 10A and 10B are plots showing measurements of the
coupling k as a function of relative displacement between centers
of a source resonator 602 and receiver resonator 604 in the X- and
Z-directions. The receiver resonator is similar to the resonator
shown in FIG. 5B (i.e., a "top hat" configuration) and the source
resonator is similar to the resonator shown in FIG. 3B. The plot in
FIG. 10A shows measurements of k in the absence of a vehicle
chassis, while the plot in FIG. 10B shows measurements in the
presence of an aluminum vehicle chassis. The measured values of k
in FIGS. 10A and 10B do not differ substantially from the measured
values shown in the plots of FIGS. 6A and 6B, respectively.
[0137] FIGS. 11A and 11B are plots showing figure-of-merit
(U.sub.0) measurements as a function of relative displacement
between centers of a source resonator 602 and receiver resonator
604 in the X- and Z-directions. The receiver resonator is similar
to the resonator shown in FIG. 5B (i.e., a "top hat" configuration)
and the source resonator is similar to the resonator shown in FIG.
3B. The plot in FIG. 11A shows measurements of U.sub.0 in the
absence of a vehicle chassis, while the plot in FIG. 11B shows
measurements in the presence of an aluminum vehicle chassis. The
quality factor Q for the source resonator is 1000 while the quality
factor Q for the receiver resonator is 450.
[0138] FIGS. 12A and 12B are plots showing measurements of the
coupling k as a function of relative displacement between centers
of a source resonator 602 and receiver resonator 604 in the Y- and
Z-directions. The receiver resonator is similar to the resonator
shown in FIG. 5B (i.e., a "top hat" configuration) and the source
resonator is similar to the resonator shown in FIG. 3B. The plot in
FIG. 12A shows measurements of k in the absence of a vehicle
chassis, while the plot in FIG. 12B shows measurements in the
presence of an aluminum vehicle chassis.
[0139] FIGS. 13A and 13B are plots showing figure-of-merit
(U.sub.0) measurements as a function of relative displacement
between centers of a source resonator 602 and receiver resonator
604 in the Y- and Z-directions. The receiver resonator is similar
to the resonator shown in FIG. 5B (i.e., a "top hat" configuration)
and the source resonator is similar to the resonator shown in FIG.
3B. The plot in FIG. 13A shows measurements of U.sub.0 in the
absence of a vehicle chassis, while the plot in FIG. 13B shows
measurements in the presence of an aluminum vehicle chassis. The
quality factor Q for the source resonator is 1000 while the quality
factor Q for the receiver resonator is 450.
[0140] FIGS. 14A and 14B are plots showing measurements of the
coupling k as a function of relative displacement between centers
of a source resonator 602 and receiver resonator 604 in the X- and
Z-directions. The receiver resonator is similar to the resonator
shown in FIG. 5B (i.e., a "top hat" configuration) and the source
resonator is similar to the resonator shown in FIG. 3B. The plot in
FIG. 14A shows measurements of k in the presence of an aluminum
vehicle chassis, while the plot in FIG. 14B shows measurements of k
in the presence of a steel (e.g., ST1008 steel) vehicle chassis. It
is evident from FIGS. 14A and 14B that replacing the aluminum
chassis with a steel chassis does not have a significant effect on
the coupling k.
[0141] FIGS. 15A and 15B are plots showing measurements of the
quality factor Q.sub.0 as a function of relative displacement
between centers of a source resonator 602 and a receiver resonator
604 in the X- and Z-directions. The receiver resonator is similar
to the resonator shown in FIG. 5B (i.e., a "top hat" configuration)
and the source resonator is similar to the resonator shown in FIG.
3B. The plot in FIG. 15A shows source resonator Q.sub.0,src
measurements in the presence of a steel vehicle chassis. The plot
in FIG. 15B shows receiver ("device") resonator Q.sub.0,dev
measurements in the presence of a steel vehicle chassis. Both
source resonator Q.sub.0,src and receiver resonator Q.sub.0,dev are
significantly reduced compared to measured values in the presence
of an aluminum vehicle chassis or no vehicle chassis.
[0142] FIGS. 16A and 16B are plots showing figure-of-merit
(U.sub.0) measurements as a function of relative displacement
between centers of a source resonator 602 and a receiver resonator
604 in the X- and Z-directions. The receiver resonator is similar
to the resonator shown in FIG. 5B (i.e., a "top hat" configuration)
and the source resonator is similar to the resonator shown in FIG.
3B. The plot in FIG. 16A shows U.sub.0 measurements in the presence
of an aluminum vehicle chassis, while the plot in FIG. 16B shows
U.sub.0 measurements in the presence of a steel ("ST1008") vehicle
chassis.
[0143] As is evident from the foregoing discussion, the coupling k
between source and device resonators can be significantly affected
by the presence of a car chassis. FIGS. 60A and 60B are plots
showing measured values of the coupling k between source and device
resonators that are similar to those shown in FIGS. 44 and 46,
respectively, as a function of relative displacements between the
centers of the resonators in the X- and Y-directions. The source
and device resonators are spaced from one another by 15 cm in the
Z-direction. The plot in FIG. 60A shows measurements of k in the
presence of a vehicle chassis, while the plot in FIG. 60B shows
measurements of k with no vehicle chassis present. The presence of
the vehicle chassis reduces the coupling k by between 10% and 15%.
Allowing for maximum offsets of 10 cm in both the X- and
Y-directions (so that the maximum offset between the resonators
corresponds to the coordinate set (X,Y,Z)=(10,10,15) cm), the
system should be well matched for a minimum coupling k=0.07.
[0144] FIGS. 61A and 61B are plots showing measured values of the
coupling k between the same source and device resonators as in
FIGS. 60A and 60B as a function of relative displacements between
the centers of the resonators in the X- and Y-directions. The
source and device resonators are spaced from one another by 10 cm
in the Z-direction. The plot in FIG. 61A shows measurements of k in
the presence of a vehicle chassis, while the plot in FIG. 61B shows
measurements of k with no vehicle chassis present. The presence of
the vehicle chassis reduces the coupling k by between 1% and
8%.
[0145] FIG. 62 shows a schematic diagram of a receiver resonator
that includes a resonator coil 6202, a magnetic member 6204, a
first conductive shield 6206, and a second conductive shield 6208.
The receiver resonator is positioned in proximity to a vehicle
chassis 6210 formed of steel (e.g., ST1008). Second conductive
shield 6208 is formed of aluminum, and is square in shape with a
side length 6212.
[0146] To investigate the effect of the size of second shield 6208
on the mitigation of coupling losses due to chassis 6210, the side
length 6212 of second shield 6208 is varied from 50 cm to 150 cm,
and values of the Q-factor for both source and receiver resonators
in a wireless power transfer system are measured. The source and
receiver resonators are similar to those shown in FIGS. 44 and 46,
respectively. FIG. 63 is a plot showing measured Q-factor values
for the source resonator (curve 6302) and receiver resonator (curve
6304) as a function of the side length 6212 of second shield 6208.
The source and receiver resonators are displaced from one another
by 10 cm in both the X- and Y-directions, and by 10 cm in the
Z-direction, a relative offset at which the effect of the vehicle
chassis on the source resonator is greatest. As is evident from
FIG. 63, the side length 6212 of second shield 6208 is preferably
80 cm or larger to mitigate the lossy effect of the steel vehicle
chassis 6210.
Impedance Matching Networks and Electronic Components
[0147] Various impedance matching networks and configurations can
be used in the wireless power transfer systems disclosed herein to
ensure that power is transferred efficiently between source and
receiver resonators. Various features and aspects of impedance
matching networks are discussed, for example, in U.S. Patent
Application Publication No. 2012/0242225, the entire contents of
which are incorporated herein by reference.
[0148] FIG. 17A is a schematic diagram showing an example of an
impedance matching network for a source resonator that implements a
"balanced LCL" matching scheme. FIG. 17B is a schematic diagram
showing an example of an impedance matching network for a receiver
resonator that implements a "balanced series" matching scheme.
These impedance matching networks can be used, for example, at
power levels greater than 3 kW, and even greater than 7 kW.
[0149] In some embodiments, a wireless power transfer system can
include a source resonator with a quality factor Q.sub.0,src of
approximately 1000 and a receiver resonator with a quality factor
Q.sub.0,dev of approximately 450. The maximum coupling k value for
this system can be approximately 0.12. In the X-Z plane, the
minimum coupling k value for a source and receiver resonator of a
wireless power transfer system (referring to FIGS. 6A-6B, 7A-7B,
10A-10B, 11A-11B, and 14A-14B) can be approximately 0.08. The
minimum coupling k value for a source and receiver resonator of a
wireless power transfer system in the Y-Z plane (referring to FIGS.
8A-8B, 9A-9B, 12A-12B, and 13A-13B) can be approximately 0.06.
[0150] In certain embodiments, the impedance matching point of the
receiver resonator may be chosen such that the maximum power
dissipated in the device, including the impedance matching network
and diodes, is less than 300 W (e.g., less than 275 W, less than
250 W, less than 225 W, less than 200 W).
[0151] Impedance matching networks can generally include a variety
of different electronic components. For example, certain impedance
matching networks can include ceramic capacitors (for example,
capacitors from 800 E series, available from American Technical
Ceramics Corp., Huntington Station, N.Y.) rated for approximately
2000 V (peak voltage) and with quality factors of approximately
Q.sub.cap=2500. In certain embodiments, the capacitor voltage
rating can determine the target inductances of the resonator coils.
For example, the above capacitor rating can correspond to a source
resonator coil of inductance L=40 .mu.H and capacitance values of
C.sub.1a=C.sub.1b=C.sub.2=263 nF, and to a receiver resonator coil
of inductance L=100 .mu.H and capacitance values
C.sub.1a=C.sub.1b=70.1 nF. Other types of less expensive capacitors
can also be used in certain embodiments, including film capacitors
for example.
[0152] FIGS. 18A and 18B are plots of measured device-side load
impedance as a function of output voltage for a device with a
receiver resonator as shown in FIG. 5B, in the presence of an
aluminum vehicle chassis. FIG. 18A shows measured device-side load
impedance for a power level of 3.7 kW, while FIG. 18B shows
measured device-side load impedance for a power level of 7.4
kW.
[0153] FIGS. 19A and 19B are plots of amplifier-to-battery
efficiency as a function of output voltage for a device with a
receiver resonator as shown in FIG. 5B, in the presence of an
aluminum vehicle chassis. FIG. 19A shows amplifier-to-battery
efficiency for coupling k values of 0.12 (curve 1902), 0.08 (curve
1904), and 0.06 (curve 1906) for a power level of 3.7 kW. FIG. 19B
shows amplifier-to-battery efficiency for coupling k values of 0.12
(curve 1908), 0.08 (curve 1910), and 0.06 (curve 1912) for a power
level of 7.4 kW. In some embodiments, efficiency values at the
lower of the coupling k values can be improved by matching to a
lower figure-of-merit U.sub.d.
[0154] FIGS. 20A and 20B are plots of power dissipated in a source
(with a resonator corresponding to the source resonator shown in
FIG. 3B) and a device (with a receiver resonator corresponding to
the receiver resonator shown in FIG. 5B) as a function of output
voltage, in the presence of an aluminum vehicle chassis. FIG. 20A
shows power dissipated in the source for coupling k values of 0.12
(curve 2002), 0.08 (curve 2004), 0.06 (curve 2006), and in the
device (curve 2008) at a power level of 3.7 kW. FIG. 20B shows
power dissipated in the source for coupling k values of 0.12 (curve
2010), 0.08 (curve 2012), 0.06 (curve 2014) and in the device
(curve 2016) at a power level of 7.4 kW.
[0155] FIGS. 21A and 21B are plots of the voltage (V.sub.rms)
measured across one or more capacitors in an impedance matching
network for a system that includes a source (with a resonator
corresponding to the source resonator shown in FIG. 3B) and a
device (with a receiver resonator corresponding to the receiver
resonator shown in FIG. 5B) as a function of output voltage, in the
presence of an aluminum vehicle chassis. Voltages in FIGS. 21A and
21B are measured across capacitors C.sub.1a and C.sub.1b shown in
FIGS. 17A and 17B. FIG. 21A shows the RMS voltage across capacitor
C.sub.1 for a source with coupling k values of 0.12 (curve 2104),
0.08 (curve 2106), and 0.06 (curve 2108), and for a device (curve
2102) at a power level of 3.7 kW. FIG. 21B shows the RMS voltage
across capacitor C.sub.1 for a source with coupling k values of
0.12 (curve 2112), 0.08 (curve 2114), and 0.06 (curve 2116), and
for a device (curve 2110) at a power level of 7.4 kW.
[0156] FIGS. 22A and 22B are plots of the magnetic field (mT)
measured in the magnetic member attached to the resonators in a
wireless power transfer system (i.e., attached to a source
resonator such as the resonator shown in FIG. 3B and attached to a
receiver resonator as shown in FIG. 5B), as a function of output
voltage, in the presence of an aluminum vehicle chassis. For
measurements shown in FIGS. 22A and 22B, the magnetic member is
formed from 5 mm ferrite pieces. FIG. 22A shows the magnetic field
in the magnetic member of a source resonator for coupling k values
of 0.12 (curve 2204), 0.08 (curve 2206), and 0.06 (curve 2208), and
in a receiver resonator (curve 2202) at a power level of 3.7 kW.
FIG. 22BB shows the magnetic field in the magnetic member of a
source resonator for coupling k values of 0.12 (curve 2212), 0.08
(curve 2214), and 0.06 (curve 2216), and in a receiver resonator
(curve 2210) at a power level of 7.4 kW.
[0157] FIG. 23 shows a schematic circuit diagram of an embodiment
of device electronics 110. The device electronics include a device
resonator coil 2302 with series tuning, represented by series
capacitors 2304 and 2306. The device electronics can include a
half-wave or full-wave rectification stage 2310, one or more
filters 2312, and/or a DC-to-DC converter 2314. The DC-DC converter
can be used to tune the load impedance that the device sees to
achieve an improved and/or optimal impedance matching value. Load
112 can correspond to a variety of electronic devices such as, for
example, a battery 2316. In some embodiments, DC-DC converter 2314
can be a boost converter to minimize the voltage across capacitors
C.sub.1a, and C.sub.1b. In certain embodiments, DC-DC converter
2314 can be a buck converter to reduce losses in the rectification
diodes.
[0158] FIGS. 24A and 24B are plots showing the DC-DC boost
conversion ratio for a device with a receiver resonator (such as
the receiver resonator shown in FIG. 5B) as a function of output
voltage, in the presence of an aluminum vehicle chassis. FIG. 24A
shows the DC-DC conversion ratio for a device with a receiver
resonator having coupling k values of 0.12 (curve 2402), 0.08
(curve 2404), and 0.06 (curve 2406) at a power level of 3.7 kW.
FIG. 24B shows the DC-DC conversion ratio for a device with a
receiver resonator having coupling k values of 0.12 (curve 2402),
0.08 (curve 2404), and 0.06 (curve 2406) at a power level of 7.4
kW. In some embodiments, a DC-DC boost conversion ratio of
approximately 4:1 can be optimal for operation at both 3.7 kW and
7.4 kW for various positional offsets between the source and device
resonators as well as output voltages.
[0159] FIGS. 25A and 25B are plots showing the
amplifier-to-converter efficiency for a device with a receiver
resonator (such as the receiver resonator shown in FIG. 5B) as a
function of output voltage, in the presence of an aluminum vehicle
chassis. FIG. 25A shows the efficiency for a device with a receiver
resonator having coupling k values of 0.12 (curve 2502), 0.08
(curve 2504), and 0.06 (curve 2506) at a power level of 3.7 kW.
FIG. 25B shows the efficiency for a device with a receiver
resonator having coupling k values of 0.12 (curve 2502), 0.08
(curve 2504), and 0.06 (curve 2506) at a power level of 7.4 kW.
[0160] FIGS. 26A and 26B are plots showing power dissipated in a
source that includes a source resonator (such as the source
resonator shown in. FIG. 3B) and in a device that includes a
receiver resonator (such as the receiver resonator shown in FIG.
5B) as a function of output voltage in the presence of an aluminum
vehicle chassis. FIG. 26A shows the power dissipated in a source
for coupling k values of 0.12 (curve 2602), 0.08 (curve 2604), and
0.06 (curve 2606), and in a device for coupling k values of 0.12
(curve 2608), 0.08 (curve 2610), and 0.06 (curve 2612) at a power
level of 3.7 kW. FIG. 26B shows the power dissipated in a source
for coupling k values of 0.12 (curve 2614), 0.08 (curve 2616), and
0.06 (curve 2618), and in a device for coupling k values of 0.12
(curve 2620), 0.08 (curve 2622), and 0.06 (curve 2624) at a power
level of 7.4 kW.
[0161] FIGS. 27A and 27B are plots showing measured voltages across
one or more matching network capacitors in a source that includes a
source resonator (such as the source resonator shown in FIG. 3B)
and in a device that includes a receiver resonator (such as the
receiver resonator shown in FIG. 5B) as a function of output
voltage in the presence of an aluminum vehicle chassis. FIG. 27A
shows the voltage across one or more matching network capacitors in
a source for coupling k values of 0.12 (curve 2702), 0.08 (curve
2704), and 0.06 (curve 2706), and across one or more matching
network capacitors in a device for coupling k values of 0.12 (curve
2708), 0.08 (curve 2710), and 0.06 (curve 2712) at a power level of
3.7 kW. FIG. 27B shows the voltage across one or more matching
network capacitors in a source for coupling k values of 0.12 (curve
2714), 0.08 (curve 2716), and 0.06 (curve 2718), and across one or
more matching network capacitors in a device for coupling k values
of 0.12 (curve 2720), 0.08 (curve 2722), and 0.06 (curve 2724) at a
power level of 7.4 kW.
[0162] FIGS. 28A and 28B are plots of the magnetic field (mT)
measured in a magnetic member attached to the source and receiver
resonators in a wireless power transfer system, where the source
resonator is similar to the resonator shown in FIG. 3B and the
receiver resonator is similar to the resonator shown in FIG. 5B, as
a function of output voltage, in the presence of an aluminum
vehicle chassis. FIG. 28A shows the magnetic field measured in the
magnetic member of the source for coupling k values of 0.12 (curve
2802), 0.08 (curve 2804), and 0.06 (curve 2806), and in the
magnetic member of the device for coupling k values of 0.12 (curve
2808), 0.08 (curve 2810), and 0.06 (curve 2812) at a power level of
3.7 kW. FIG. 28B shows the magnetic field measured in the magnetic
member of the source for coupling k values of 0.12 (curve 2814),
0.08 (curve 2816), and 0.06 (curve 2818), and in the magnetic
member of the device for coupling k values of 0.12 (curve 2820),
0.08 (curve 2822), and 0.06 (curve 2824) at a power level of 7.4
kW.
[0163] In some embodiments, wireless power transfer systems can
include a switchable, multi-tapped transformer to variably tune the
impedance of source and/or receiver resonators. In some
embodiments, wireless power transfer systems can include a DC-DC
converter to modulate the output impedance.
Resonator Configurations
[0164] A wide variety of different resonator configurations can be
used in wireless power transfer systems. In this section, examples
of such configurations and certain performance characteristics of
the configurations will be discussed.
[0165] FIGS. 29A-29D are schematic diagrams showing exemplary
embodiments of resonator coils for wireless power transfer systems.
In each of FIGS. 29A-29D, the resonator winding length 2914 and
span 2916 varies. In each of FIGS. 29A-29D, the size of magnetic
member 2904 is 50 cm.times.50 cm.times.5 mm, and the size of shield
2902 is 60 cm.times.60 cm. Magnetic member 2904 is formed from
ferrite, and shield 2902 is formed from aluminum. FIG. 29A shows a
resonator coil 2906 with a minimum length of 400 mm and minimum
span of 50 mm. FIG. 29B shows a resonator coil 2908 with a maximum
length of 500 mm and minimum span of 50 mm. FIG. 29C shows a
resonator coil 2910 with a minimum length of 400 mm and maximum
span of 175 mm. FIG. 29D shows a resonator coil 2912 with a maximum
length of 500 mm and maximum span of 175 mm The resonators shown in
FIGS. 29A-29D can be used as source resonators (such as the source
resonator shown in FIG. 3B).
[0166] FIG. 30A is a plot of coupling rate k as a function of
source resonator coil winding length and span, measured at an
approximate offset of (X,Y,Z)=(10,10,15) cm relative to a receiver
resonator coil, where Z-offset is measured from coil surface to
surface. The receiver resonator dimensions are 25 mm by 50 mm. In
FIG. 30A, the dark region 3002 with greater coupling k indicates
that a winding length of 500 mm and winding span of 130 mm result
in higher coupling for certain source resonator coil dimensions.
FIG. 30B is a plot of figure-of-merit U.sub.0 as a function of
source resonator coil winding length and span, measured at an
approximate offset of (X,Y,Z)=(10,10,15) cm relative to a receiver
resonator coil, where Z-offset is measured from coil surface to
surface, for the same source and receiver resonators as in FIG.
30A.
[0167] FIGS. 31A-31D are schematic diagrams showing exemplary
embodiments of resonator coils for wireless power transfer systems.
In each of FIGS. 31A-31D, the winding gap-to-edge distance 3114 and
span 3116 vary. In FIGS. 31A-31D, the size of magnetic member 3104
is 20 cm.times.45 cm.times.5 mm, and the size of shield 3102 is 25
cm.times.50 cm. Magnetic member 3104 is formed from ferrite, and
shield 3102 is formed from aluminum. FIG. 31A shows a resonator
coil 3106 with a minimum gap-to-edge distance of 0 mm and a minimum
span of 25 mm. FIG. 31B shows a resonator coil 3108 with a maximum
gap-to-edge distance of 20 mm and a minimum span of 25 mm. FIG. 31C
shows a resonator coil 3110 with a minimum gap-to-edge distance of
0 mm and a maximum span of 50 mm. FIG. 31D shows a resonator coil
3112 with a maximum gap-to-edge distance of 20 mm and a maximum
span of 50 mm. The resonators shown in FIGS. 31A-31D can be used as
receiver resonators in devices, for example.
[0168] FIG. 32A is a plot of coupling k as a function of resonator
coil winding gap-to-edge distance and span for the resonators of
FIGS. 31A-31D. Highest coupling is achieved for resonator coils
with a winding gap-to-edge distance of 0 mm and a span of 50 mm.
FIG. 32B is a plot of figure-of-merit U.sub.0 as a function of
resonator coil winding gap-to-edge distance and span for the
resonators of FIGS. 31A-31D.
[0169] FIGS. 33A and 33B are plots of the coupling k between a
source resonator of the type shown in FIGS. 29A-29D and a device
resonator of the type shown in FIGS. 31A-31D, as a function of
relative offset between the resonators in the X- and Y-directions.
FIG. 33A shows the coupling k for a relative offset between the
resonators in the Z-direction of 10 cm, and FIG. 33B shows the
coupling k for a relative offset between the resonators in the
Z-direction of 15 cm.
[0170] FIGS. 34A and 34B are plots of the figure-of-merit U.sub.0
for a wireless power transfer system that includes a source
resonator of the type shown in FIGS. 29A-29D, and a device
resonator of the type shown in FIGS. 31A-31D, as a function of
relative offset between the resonators in the X- and Y-directions.
FIG. 34A shows the figure-of-merit U.sub.0 for a relative offset
between the resonators in the Z-direction of 10 cm, and FIG. 34B
shows the figure-of-merit U.sub.0 for a relative offset between the
resonators in the Z-direction of 15 cm.
Additional Impedance Matching Network Topologies
[0171] In addition to the impedance matching networks discussed
above, additional impedance matching network topologies can also be
used in the wireless power systems disclosed herein. FIGS. 35A and
35B are schematic diagrams of matching networks for use in device
electronics 110. FIG. 35A shows a delta capacitor matching network
3502 and FIG. 35B shows a wye capacitor matching network 3504. The
two capacitor networks are equivalent to each other through a
"delta-wye" transformation. Delta and wye networks that match a
device to an effective impedance that stays relatively flat as a
battery voltage and output power vary can be desirable.
Accordingly, wireless power transfer systems can include a matching
network of either topology. In some embodiments, the implementation
of either a delta or wye matching network is guided by the network
that uses the fewest capacitors of a given voltage rating, making
that network the cheaper of the two to implement.
[0172] FIG. 36 is a plot of the total minimum number of capacitors
for delta (3602) and wye (3604) impedance matching networks in a
device. The device matching point U.sub.d can affect the overall
efficiency and determine how the power dissipated is distributed
between the source and the device. A higher U.sub.d value means
less power is dissipated in the device and more power is dissipated
in the source. Delta and/or wye matching networks can be used to
match to U.sub.d=50 such that approximately equal power is
dissipated in the source and the device at maximum relative offset
between the resonators. In some embodiments, point 3606
(L.sub.d=37.5 .mu.H) can be chosen on the delta network 3602 to
minimize the number of capacitors and inductance. A lower
inductance may reduce the voltage across the winding.
[0173] FIG. 37 is a schematic diagram showing an embodiment of an
impedance matching network topology for use in device electronics
110. As one example, in FIG. 37, the various circuit component
positions can have the following values: L.sub.3a=L.sub.3b=25
.mu.H; C.sub.a-24.8 nF; C.sub.b1=C.sub.b2 =39 nF; and C.sub.c-71.8
nF. This topology provides additional degrees of freedom in the
impedance matching of the device as compared to topologies with
fewer component positions. Note that components in positions
C.sub.b1 and C.sub.b2 provide a balancing of the impedance matching
network shown in FIG. 35A. Additional aspects of the impedance
matching network topology shown in FIG. 37 are disclosed, for
example, in U.S. Pat. No. 8,461,719, the entire contents of which
are incorporated herein by reference.
[0174] FIG. 38 is a plot of the figure-of-merit U as function of
output voltage in a device that includes a receiving resonator with
a delta-type impedance matching network (as shown in FIG. 35A).
Curve 3802 shows the figure-of-merit U.sub.dR which is the
resistive component of the device matching impedance for power
output of 7.0 kW. Curve 3804 shows the figure of merit U.sub.dR
which is the resistive component of the device matching impedance
for power output of 3.5 kW. Curve 3806 shows the figure-of-merit
U.sub.dX which is the reactive component of the device matching
impedance for power output of 7.0 kW. Curve 3808 shows the figure
of merit U.sub.dX which is the reactive component of the device
matching impedance for power output of 3.5 kW. The reactive
component of the device matching impedance is generally smaller
than the resistive component, and thus the device resonator is not
significantly detuned off resonance. Furthermore, in these
conditions, current in the source does not increase excessively to
drive an off-resonance device resonator.
[0175] FIG. 39 is a plot showing the power dissipated in source and
receiver resonators that are matched using a delta-type impedance
matching network, as a function of output voltage. The system of
source and receiver resonators has a coupling k of 0.08. Curve 3902
shows the power dissipated in the source resonator coil and one or
more capacitors of the matching network at a power output of 7.0
kW. Curve 3904 shows the power dissipated in the source resonator
coil and one or more capacitors of the matching network at a power
output of 3.5 kW. Curve 3906 shows the power dissipated in the
receiver resonator coil and one or more source-side capacitors of
the matching network at a power output of 7.0 kW. Curve 3908 shows
the power dissipated in the receiver resonator coil and one or more
device-side capacitors of the matching network at a power output of
3.5 kW.
[0176] FIG. 40 is a plot showing the maximum magnetic field in the
source and device resonators as a function of output voltage for a
wireless power transfer system that includes source and device
resonators with a coupling k of 0.08. Curve 4002 shows the maximum
magnetic field in the source resonator at a power output of 7.0 kW.
Curve 4004 shows the maximum magnetic field in the source resonator
at a power output of 3.5 kW. Curve 4006 shows the maximum magnetic
field in the device resonator at a power output of 7.0 kW. Curve
4008 shows the maximum magnetic field in the device resonator at a
power output of 3.5 kW.
[0177] FIG. 41 is a plot showing the voltage across one or more
capacitors of a delta-matching network in a receiver resonator of a
device (curves 4102-4112) and in a source resonator (curves
4114-4116) as function of output voltage. Curve 4102 shows the
voltage across capacitor C.sub.a in FIG. 37 at a power output of
7.0 kW. Curve 4104 shows the voltage across capacitor C.sub.a in
FIG. 37 at a power output of 3.5 kW. Curve 4106 shows the voltage
across capacitor C.sub.b in FIG. 37 at a power output of 7.0 kW.
Curve 4108 shows the voltage across capacitor C.sub.b in FIG. 37 at
a power output of 3.5 kW. Curve 4110 shows the voltage across
capacitor C.sub.c in FIG. 37 at a power output of 7.0 kW. Curve
4112 shows the voltage across capacitor C.sub.c in FIG. 37 at a
power output of 3.5 kW. Curve 4114 shows the voltage across a
source capacitor at a power output of 7.0 kW. Curve 4116 shows the
voltage across a source capacitor at a power output of 3.5 kW.
[0178] Optimizing impedance matching networks for particular
resonator configurations and power delivery specifications involves
selecting electronic components for the network. For example, with
reference to the delta-matching network shown in FIG. 35A, one
component that is selected is the inductance value L.sub.3. If
L.sub.3 is too small, the diodes in the rectifier may conduct for
only a fraction of an oscillation period, so to transfer a fixed
amount of power, the peak current through the diodes would have to
be higher, leading to more power dissipation in the diodes. In
addition, the electrical current through L.sub.3 would also peak
higher and have more harmonic content, leading to losses in the
inductor. Conversely, large values of L.sub.3 can add too much ESR
to the impedance matching network. As such, the value of L.sub.3 is
chosen to balance these competing effects.
[0179] FIGS. 64A and 64B are plots showing the peak current through
the diodes and inductor L.sub.3 for delivery of 6.6 kW to a load of
V.sub.dc=420 V. For a capacitor with L.sub.3=20 .mu.H (FIG. 64A),
the peak current is 34.5 A. For a capacitor with L.sub.3=50 .mu.H
(FIG. 64B), the peak current is 26.7 A.
[0180] FIGS. 65A and 65B are plots showing the peak current through
the diodes and inductor L3 for delivery of 3.3 kW to a load of
V.sub.dc=420 V. For a capacitor with L.sub.3=20 .mu.H (FIG. 65A),
the peak current is 19.9 A. For a capacitor with L.sub.3=50 .mu.H
(FIG. 65B), the peak current is 15.3 A.
[0181] As discussed above, once the configuration of the impedance
matching network has been determined, the network configuration can
be optimized. In general, the device matching point U.sub.d affects
the overall efficiency of wireless power transfer and how power is
dissipated between the source and the device. Higher values of
U.sub.d mean that more power is dissipated in the source and less
is dissipated in the device. Impedance matching networks can be
optimized to satisfy the condition U.sub.d=50 (i.e., equal power
dissipation in the source and device) at the maximum offset between
the source and device resonators (i.e., where k=0.07). The optimum
configuration for an impedance matching network is generally the
configuration that uses the smallest number of capacitors, while
satisfying the optimization condition to within an acceptable
tolerance.
[0182] FIGS. 66A and 66B are plots showing the number of capacitors
in a device's impedance matching network as a function of the
inductance of the device's receiving resonator coil inductance, for
800 E series capacitors (American Technical Ceramics Corp.) (FIG.
66A) and film capacitors (available from EPCOS, Munich, Germany).
Each plot shows results for both delta- and wye-matching networks.
FIG. 66A corresponds to capacitors at 2000 V peak voltage and 6.6
kW, with a target U.sub.d of 50. FIG. 66B corresponds to capacitors
at 600 V RMS voltage and 6.6 kW, with a target U.sub.d of 50.
[0183] After choosing the type of capacitors to use (i.e., 800 E
series capacitors) and the inductance of the receiver resonator
coil (43.5 .mu.H, which can be achieved with 8 loops of conductive
material), the number and capacitance values of the different
capacitors in the impedance matching network are selected, subject
to the impedance matching condition (U.sub.d=50), and further
subject to the constraint that the number of capacitors used to
achieve the impedance matching condition should be as small as
possible.
[0184] FIG. 67 is a schematic diagram of an optimized device
impedance matching network. In the optimized network,
L.sub.3a=L.sub.3b=25 .mu.H, C.sub.a-28.6 nF (achieved with
5.times.5.1 nF+1.times.3.0 nF 800 E capacitors),
C.sub.b1=C.sub.b2=36.8 nF (achieved with 7.times.5.1 nF+1.times.1.0
nF 800 E series capacitors), and C.sub.3=51.0 nF (achieved with
10.times.5.1 nF 800 E series capacitors). The total number of
capacitors used in the optimized network is 32. The optimized
network is sufficiently small geometrically that it fits within the
empty volume of the "top hat" resonator shown in FIG. 5B.
[0185] FIG. 68 is a plot showing the figure of merit U.sub.d as a
function of output voltage for an optimized impedance matching
network, at a coupling k value of 0.07. The target U.sub.d=50 is
achieved between 300 V and 400 V (and specifically, at
approximately 350 V) for the resistive component of U.sub.d at 3.3
kW output (curve 6802) and 6.6 kW output (curve 6804), but not for
the reactive component of U.sub.d at either 6.6 kW output (curve
6806) or 3.3 kW output (curve 6808).
[0186] FIG. 69 is a plot showing the amount of power dissipated in
various components of the source for a wireless power transfer
system with an optimized device impedance matching network having a
coupling value k=0.07 between source and receiver resonators.
Curves 6902, 6904, 6906, and 6908 show the power dissipated in the
source's resonator coil windings, shield, ferrite magnetic member,
and capacitors at 7.0 kW output power. Curves 6910, 6912, 6914, and
6916 show the power dissipated in the source's resonator coil
windings, shield, ferrite magnetic member, and capacitors at 3.5 kW
output power.
[0187] FIG. 70 is a plot showing the amount of power dissipated in
various components of the device for a wireless power transfer
system with an optimized device impedance matching network having a
coupling value k=0.07 between source and receiver resonators.
Curves 7002, 7004, 7006, and 7008 show the power dissipated in the
device's receiver coil windings, shield, ferrite magnetic member,
and capacitors at 7.0 kW output power. Curves 7010, 7012, 7014, and
7016 show the power dissipated in the device's receiver coil
windings, shield, ferrite magnetic member, and capacitors at 3.5 kW
output power.
[0188] FIG. 71 is a plot showing the magnetic field in the ferrite
magnetic member as a function of output voltage for a wireless
power transfer system with an optimized device impedance matching
network having a coupling value k=0.07 between source and receiver
resonators. Curves 7102 and 7104 show the magnetic field in the
magnetic member of the source resonator at 6.6 kW and 3.3 kW power
output, respectively. Curves 7106 and 7108 show the magnetic field
in the magnetic member of the device's receiver resonator at 6.6 kW
and 3.3 kW power output, respectively.
[0189] FIG. 72 is a plot showing voltages across the capacitors as
a function of output voltage for a wireless power transfer system
with an optimized device impedance matching network having a
coupling value k=0.07 between source and receiver resonators.
Curves 7202 and 7204 show the voltages across C.sub.a at 6.6 kW and
3.3 kW output power, respectively. Curves 7206 and 7208 show the
voltages across C.sub.b at 6.6 kW and 3.3 kW output power,
respectively. Curves 7210 and 7212 show the voltages across C.sub.c
at 6.6 kW and 3.3 kW output power, respectively.
[0190] FIG. 73 is a plot showing the electrical current through the
source resonator coil and the device's receiver resonator coil as a
function of output voltage for a wireless power transfer system
with an optimized device impedance matching network having a
coupling value k=0.07 between source and receiver resonators.
Curves 7302 and 7304 show the current through the source resonator
coil at 6.6 kW and 3.3 kW output power, respectively. Curves 7306
and 7308 show the current through the device's receiver resonator
coil at 6.6 kW and 3.3 kW output power, respectively.
Additional Resonator Configurations
[0191] FIG. 42 is a schematic diagram showing an embodiment of a
magnetic member formed from an array of tiles 4202 of magnetic
material (e.g., ferrite). In some embodiments, the ferrite tiles
4202 can have dimensions of about 150 mm by 100 mm, and a thickness
of about 5 mm or 8 mm or greater. In certain embodiments, the
ferrite tiles 4202 can be arranged such that there are equal gaps
of about 0.4 mm between adjacent tiles. In some embodiments, the
maximum dimensions of the magnetic member can be approximately 500
mm by 500 mm.
[0192] In general, the central region of the magnetic member can be
left empty (as in FIG. 42) or filled with additional magnetic
material. In some embodiments, the magnetic member can be spaced
from an aluminum shield 4302 by about 40 mm in a resonator, as
shown in FIG. 43. The magnetic member shown in FIGS. 42 and 43 is
typically used in a source resonator.
[0193] FIG. 44 shows an image of an embodiment of a source
resonator coil 4402 affixed to a magnetic member 4202 positioned
over, and spaced from, an aluminum shield 4302. In FIG. 44, the
source resonator coil has similar outer dimensions to those of the
magnetic member (i.e., about 500 mm.times.500 mm), the resonator
coil windings span 4404 is approximately 130 mm, and the coil
windings have an inner dimension 406 of approximately 240 mm
square.
[0194] In general, higher resonator Q values can be achieved by
winding multiple resonator coils in parallel within a resonator. In
FIG. 44, three coils are wound in parallel with a minimum of 5
loops in each coil to achieve a targeted inductance value.
[0195] FIGS. 45A and 45B are a schematic diagram and an image,
respectively, that show an embodiment of a magnetic member formed
from an array of ferrite tiles 4502. In some embodiments, the
ferrite tiles 4502 can have dimensions of about 150 mm by 100 mm,
with a thickness of about 5 mm or 8 mm or greater. In certain
embodiments, the ferrite tiles 4502 can be arranged such that there
are equal gaps of about 0.4 mm between the tiles. In some
embodiments, the maximum dimensions of the magnetic member can be
approximately 200 mm by 450 mm. The magnetic member shown in FIGS.
45A and 45B is typically used in a device's receiver resonator.
[0196] FIG. 46 is an image of an embodiment of a device receiver
resonator coil 4602 affixed to a magnetic member 4502. In FIG. 46,
the device receiver resonator coil has similar outer dimensions to
the magnetic member shown in FIG. 42 (i.e., about 200 mm by 450
mm). The receiver resonator coil in FIG. 42 includes a single wire
that forms seven loops on the surface of the magnetic member of
FIG. 45. The inductance of the receiver resonator coil 4602 is
approximately 33.1 .mu.H and its quality factor is approximately
591.
[0197] FIG. 47A is a schematic diagram of an embodiment of a source
resonator coil 4706 affixed to a magnetic member 4704. In some
embodiments, for example, magnetic member 4704 can have dimensions
of about 50 cm by 50 cm, and a thickness of about 5 mm, about 8 mm,
or greater. Coil 4706 and magnetic member 4704 are positioned over,
and spaced from, a shield 4702. In certain embodiments, for
example, shield 4702 is formed from a conductive material such as
aluminum, and has dimensions of about 60 cm by 60 cm). In some
embodiments, gap 4708 between magnetic member 4704 and shield 4702
can be about 50 mm. In certain embodiments, the inductance of coil
4706 can be about 19.9 .mu.H and its quality factor can be about
1150. In certain embodiments, coil 4706 can include at least three
sets of coil windings wound in parallel, each formed from a
different wire or conductive material. The windings can be
connected in parallel to yield a high-Q resonator coil. As an
example, each coil winding can include at least five loops to
achieve the target inductance.
[0198] FIG. 47B is a schematic diagram showing an embodiment of a
device receiver resonator coil 4716 affixed to magnetic member
4712. In some embodiments, magnetic member 4712 can have dimensions
of about 45 cm by 20 cm, and a thickness of about 5 mm, about 8 mm,
or greater. In certain embodiments, a central region of magnetic
member 4712 can be stepped such that it protrudes into the region
internal to coil 4716, as shown in FIG. 5B. In some embodiments,
the thickness of the magnetic material in the center of the
resonator coil can be less than the thickness of the magnetic
material elsewhere.
[0199] Coil 4716 and magnetic member 4712 are affixed to a shield
4710 formed of conductive material (e.g., aluminum) and having
dimensions of, for example, about 50 cm by 25 cm. In some
embodiments, the inductance of coil 4716 can be 33.3 .mu.H and its
quality factor can be about 443.
[0200] For the source and device resonators shown in FIGS. 47A and
47B, coupling k values for a maximum relative offset of
(X,Y,Z)=(10,10,15) between the coils can be: for a source resonator
affixed to magnetic member of 5 mm thickness, k=0.0707; and for a
source resonator affixed to magnetic member of 8 mm thickness,
k=0.0710.
[0201] FIG. 48A is a schematic diagram showing an embodiment of a
source resonator affixed to a magnetic member 4802 formed from
ferrite tiles with dimensions of about 10 cm by 10 cm in each
corner, and tiles with dimensions of about 15 cm by 10 cm outside
the corners. The quality factor of this resonator is approximately
1220.
[0202] FIG. 48B is a schematic diagram showing an embodiment of a
source resonator affixed to a magnetic member 4804 formed from
ferrite tiles with dimensions of about 15 cm by 10 cm. The quality
factor of this resonator is approximately 1050.
Resonator Coils with Parallel Windings
[0203] FIG. 49 is an image showing an embodiment of a source
resonator 4902 that includes three parallel windings 4904, 4906,
and 4908 that are wound in a coil shape and electrically connected
in parallel over a magnetic member. In embodiments, windings may be
electrically connected in parallel instead of series in order to
reduce the overall voltage that can occur across the coil. While
three windings are shown in FIG. 49, more generally any number of
windings can wound in parallel to form a coil and electrically
connected in parallel. The windings can have the same or different
wire diameters, and the overall shape of the coil formed by the
parallel windings can be any of the different shapes disclosed
herein. For purposes of this disclosure, two sets of loops--each
corresponding to a coil winding--are physically "parallel" if the
sets of loops have complementary and corresponding shapes, and the
distance between the conductive material forming one set of loops
and the conductive material forming the other set of loops is the
same between corresponding portions of the sets of loops along 80%
or more of the lengths of the conductive materials. Two windings
may be considered to be physically parallel if the magnetic
coupling between the loops of the two windings is greater than 90%.
In some embodiments, sets of parallel loops are frequently
interleaved such that corresponding portions of the conductors that
form the loops are parallel to one another.
[0204] In some embodiments, where a coil includes multiple parallel
sets of loops (e.g., windings), excess current can flow in the
innermost coil winding. This can occur, for example, because the
innermost winding typically has a shorter overall length within the
coil than middle and outer windings. This is due to the geometry of
the windings within the coil--the innermost winding, because it
typically has a smaller average loop diameter than the other
windings, has a shorter total length, and therefore a smaller total
resistance and inductance than the other windings. As a result,
excess current can flow in the innermost coil winding relative to
the other windings. More generally, because each winding is
typically of a different length, the currents that flow in each of
the windings are different, and some (or even all) of these may
exceed design specifications and/or safety guidelines.
[0205] In general, excess current in any one coil winding may
result in decreased efficiency during wireless power transfer due
to greater heat dissipation in the winding with excess current.
Typically, heat dissipation increases proportionally with the
square of current flow in each parallelized coil winding. In this
section, various methods for balancing currents in multiple
parallel windings are disclosed. The general objective underlying
these methods is to control current flow in multiple coil windings
so that the actual currents that flow in the windings are equal to
a predetermined or selected distribution of currents, within an
acceptable tolerance range. Typically, the predetermined
distribution of currents is expressed as a percentage of total
current flow through the windings that make up the coil. For
example, for a coil that includes three parallel windings, the
predetermined distribution of current can correspond to 33.3%
(i.e., one third) of the total current carried by the coil flowing
through each winding.
[0206] In some embodiments, the predetermined current distribution
corresponds to an equal partitioning of the total current among the
windings. Thus, for a coil with four parallel windings for example,
the predetermined current distribution corresponds to a
partitioning of 25% of the total coil current carried by each of
the four windings.
[0207] In certain embodiments, the predetermined distribution does
not correspond to an equal partitioning of currents among the
windings. To achieve certain functionality, for example, it can be
advantageous to partition the total coil current unequally among
the coil's parallel windings. The methods and systems disclosed
herein can be used flexibly to achieve both equal and non-equal
predetermined current distributions among parallel windings of a
coil.
[0208] In some embodiments, to balance (i.e., nominally equalize)
the currents in each coil winding, coil windings may be twisted
"crossed-over" with one another. FIG. 49 shows three such twists at
locations 4910, 4912, and 4914. The coil windings are crossed in
the following manner: at location 4910, winding 4908 is crossed
with winding 4906; at location 4912, winding 4908 is crossed with
winding 4904; and at location 4914, winding 4906 is crossed with
winding 4904. In some embodiments, three such twists may be
sufficient to approximately balance currents throughout the coil.
More generally, however, the windings of the three coils may be
crossed or twisted throughout the coil (e.g., include any number of
crossings or twists) to further balance the current in the windings
as well as to ensure that each wire is of similar length. In some
embodiments, for a coil with three coil windings, such as 4904,
4906, and 4908 in FIG. 49, the twists in the windings may each be
separated by a 1/3 of the distance around the coil. More generally,
for a coil with n coil windings, the windings can be spaced by 1/n
of the total length of a single winding loop, measured along the
loop.
[0209] Typically, the crossings between the windings are used to
approximately equalize the lengths of the windings. By equalizing
the winding lengths, each winding has a similar resistance and
inductance value, and therefore, an approximately equal portion of
the total coil current flows through each of the windings.
[0210] FIG. 50 is a plot showing the magnetic field in the magnetic
member of a source resonator (such as the source resonator shown in
FIG. 47A). In FIG. 50, the magnetic member is formed of ferrite of
5 mm thickness. The maximum magnetic field is 170 mT at 198 A of
current.
[0211] FIG. 51 is a plot showing the magnetic field in the magnetic
member of a source resonator (such as the source resonator shown in
FIG. 47A). In FIG. 51, the magnetic member is formed of ferrite of
8 mm thickness. The maximum magnetic field is 107 mT at 198 A of
current.
[0212] FIG. 52 is a plot showing the magnetic field in the magnetic
member of a source resonator (such as the source resonator shown in
FIG. 47A). In FIG. 52, the magnetic member is formed from ferrite
tiles of dimension 15 cm by 10 cm by 8 mm. The maximum magnetic
field is 101 mT at 198 A of current.
[0213] FIG. 53 is a plot showing the magnetic field in the magnetic
member of a device resonator (such as the device resonator shown in
FIG. 47B). In FIG. 53, the magnetic member is formed of ferrite of
thickness 8 mm.
[0214] FIGS. 54A and 54B are plots showing the magnetic field at a
distance of 1 cm above a source resonator having the configuration
shown in FIG. 47A. FIG. 55 is a plot showing the magnetic field at
a distance of 1 cm above a source resonator having the
configuration shown in FIG. 47A, plotted on a logarithmic
scale.
[0215] As discussed above, in some embodiments, currents in
parallel windings that form a resonator coil can be balanced by
crossing the windings (e.g., using twists) at points along the
length of the windings. Other methods can also be used to balance
currents in multiple parallel coil windings. In particular,
referring to FIG. 56 for example, the currents in each of the
parallel windings 5602, 5604, 5606 can be balanced using one or
more inductors. FIGS. 57A and 57B show schematic circuit diagrams
of a set of inductors 5702, 5704, 5706 connected in series and used
to balance currents in a resonator coil with parallel windings
5602, 5604, and 5606. In some embodiments, as shown in FIG. 57B,
one or more (or even all) of inductors 5702, 5704, 5706 can be
tunable. In general, the series connected inductors add inductance
to the respective windings to which they are connected. By adding
suitable amounts of inductance to some or all of the windings, the
effective inductance of each winding can be adjusted to achieve a
predetermined or target current distribution among the various
windings of a coil.
[0216] To balance currents in the parallel windings 5602, 5604, and
5606 (henceforth referred to as windings 1, 2 and 3, respectively,
or as the innermost, middle, and outermost windings, respectively),
the inductances of each of the parallel windings can be measured,
and inductors 5702, 5704, and 5706 can be adjusted based on the
measured inductances to balance currents in the parallel windings.
Inductors 5702, 5704, are 5706 may be referred to as
L.sub.1.sup..DELTA., L.sub.2.sup..DELTA. and L.sub.3.sup..DELTA.,
respectively, in the following discussion. In general, a variety of
adjustable inductors can be used for inductors 5702, 5704, and
5706. Suitable adjustable inductors are disclosed, for example, in
U.S. Patent Application Publication No. 2015/0051750, the entire
contents of which are incorporated herein by reference.
[0217] The methods discussed below can be performed during the
manufacturing phase of a resonator, so that inductors with suitable
inductance values can be included in a resonator at time of
fabrication. That is, the methods can be performed with the goal of
selecting and/or adjusting inductors connected in series with
parallel coil windings so that when manufacturing of the resonator
is complete, the inductances of the coil windings have been
adjusted to achieve a particular predetermined partitioning of the
total coil current among the windings. As discussed above, the
partitioning can correspond to a nominally equal distribution of
current among the windings, but can also correspond to an unequal
distribution of current.
[0218] The methods can also be performed post-manufacture by an
electronic processor that is part of, or connected to, the
resonator. The electronic processor can be configured to determine
suitable inductance values of adjustable inductors once in a single
optimization sequence. Alternatively, the processor can be
configured to repeat the steps discussed below multiple times
(e.g., at predetermined time intervals and/or in response to a user
signal) for purposes such as calibration and re-calibration, in
response to changes in the operating environment and/or parameters
of a resonator. The discussion below describes various steps that
an electronic processor can perform. It should be appreciated that
these steps can be performed pre- and/or post-fabrication of a
resonator coil, and also that certain steps can be performed during
the design phase by a human.
[0219] In wireless power transfer systems, current balancing steps
and methods can be performed by one or more electronic processors
(e.g., processor 105 and/or 111) during operation of the system.
FIG. 74 is a flow chart 7400 that includes a series of steps for
balancing currents in parallel windings of a resonator coil. In the
discussion of FIG. 74 that follows, the three parallel windings 1,
2, and 3 above are referenced. More generally, however, it should
be understood that the methods disclosed herein can be used to
balance currents in any number of parallel windings that form a
resonator coil.
[0220] In a first step 7402, the electronic processor determines
the inductance matrix L of the three connected windings,
represented by inductance values L.sub.1, L.sub.2 and L.sub.3. In
some embodiments, L.sub.1<L.sub.2<L.sub.3 to accommodate
parallel windings in which L.sub.1 is the innermost winding,
L.sub.2 is the middle winding, and L.sub.3 is the outermost
winding. The components of the inductance matrix L can be obtained
directly by measuring the coupling between the windings (e.g.,
using an inductance sensor connected to the electronic processor,
not shown in FIG. 56). The electronic processor is configured to
receive coupling (i.e., inductance) measurements from the sensor,
and to determine values of the diagonal elements in matrix L based
on the measurements. Specifically, the diagonal elements in L
correspond to each winding's inductance measured when the other
windings are open-circuited by the electronic processor. The other
elements L.sub.i,j=L.sub.j,i are obtained by measuring the
inductance I.sub.i,j of a connected winding i while
short-circuiting connected winding j, since
l i , j = L i - L i , j 2 L j -> L i , j = L j ( L i - l i , j )
. ##EQU00001##
[0221] When a sinusoidal voltage of amplitude V, oscillating at the
angular frequency .omega., is applied to the windings connected in
parallel, the amplitude of the currents flowing in the three
windings is determined by:
[ I 1 I 2 I 3 ] = L - 1 .omega. [ V V V ] ##EQU00002##
[0222] For example, for L.sub.1=21.9 .mu.H, L.sub.2=23.0 .mu.H,
L.sub.3=23.7 .mu.H, I.sub.1,2=5.24 .mu.H, I.sub.1,3=6.35 .mu.H and
I.sub.2,3=5.61 .mu.H. In this example, the inductance measurements
received by the electronic processor are used by the processor to
determine the following inductance matrix:
L = ( 21.9 19.6 19.2 19.6 23.0 20.3 19.2 20.3 23.7 ) .mu.H .
##EQU00003##
[0223] The corresponding current repartition in the presence of a
shared voltage V across windings 1, 2, and 3 connected in parallel
is:
[ I 1 I 1 I 3 ] = ( 2.35 , 1.23 , 1.27 ) 10 4 V .omega.
##EQU00004##
[0224] This current repartition corresponds to the following
relative repartition of the total current among the three windings:
(48.45%, 25.36%, 26.19%). Thus, without balancing currents in such
a coil, current I.sub.1 is nearly twice as large as I.sub.2 or
I.sub.3.
[0225] In step 7404, the electronic processor determines the
desired or target current repartition among the windings. The
target current repartition can be a set of stored values retrieved
from a memory or data storage unit, a set of values supplied by a
user, or a hard-coded or fixed implementation in circuitry. The
target current repartition represents the performance condition
that the electronic processor attempts to achieve by adjusting
individual inductances coupled to the coil windings. As discussed
above, in some embodiments the target repartition corresponds to an
equal division of the total current among the windings. This
example will be discussed in more detail below. More generally,
however, the methods and systems disclosed herein can achieve any
target repartition of the total current among the windings.
[0226] The electronic processor then determines, in step 7406, the
correcting inductances L.sub.1.sup..DELTA. and L.sub.2.sup..DELTA.
to be added in series with respectively L.sub.1 and L.sub.2 to
achieve, e.g., an evenly split current repartition (i.e.
I.sub.1=I.sub.2=I.sub.3) with the minimum amount of added
inductance, before the process ends at step 7408.
L.sub.1.sup..DELTA. and L.sub.2.sup..DELTA. are given by:
L.sub.1.sup..DELTA.=L.sub.3-L.sub.1+L.sub.2,3-L.sub.1,2
L.sub.2.sup..DELTA.=L.sub.3-L.sub.2+L.sub.1,3-L.sub.1,2
[0227] In the previous example, this yields L.sub.1.sup..DELTA.=2.5
.mu.H and L.sub.2.sup..DELTA.=0.3 .mu.H, and the corrected current
repartition becomes:
[ I 1 .DELTA. I 2 .DELTA. I 3 .DELTA. ] = 1 .omega. [ L + [ L 1
.DELTA. 0 0 0 L 2 .DELTA. 0 0 0 0 ] ] - 1 [ V V V ] = ( 1.58 , 1.58
, 1.58 ) 10 4 V .omega. ##EQU00005##
[0228] As explained previously, the foregoing discussion of flow
chart 7400 involved three windings 1, 2, and 3 and an even current
repartition by way of example. More generally, however, the methods
for current balancing shown in flow chart 7400 can be applied to
coils with any number n of parallel windings, and to target a
desired current repartition between these windings. Thus, for a
coil with n parallel windings electrically connected in parallel,
the following steps can be performed by the electronic processor to
balance currents in each of the windings.
[0229] First, in step 7402, the electronic processor determines the
inductance matrix L of the n windings. The components of L can be
obtained directly by measuring the coupling (i.e., inductances)
between the windings. The diagonal elements in matrix L are each
connected winding's inductance measured when the other windings are
open-circuited. The off-diagonal matrix elements
L.sub.i,j=L.sub.j,i are determined by the electronic processor by
measuring the inductance I.sub.i,j of each winding i while
short-circuiting winding j, since
L i , j = L i - L i , j 2 L j -> L i , j = L j ( L i - l i , j )
. ##EQU00006##
[0230] Then, in step 7404, the electronic processor determines the
desired or target repartition of the total current among the coil
windings. Next, in step 7406, the electronic processor determines
appropriate inductance values of the inductors to effect the target
current repartition in the n windings from any combination of
L.sub.i.sup..DELTA., . . . L.sub.n.sup..DELTA. (i-1 to n) added in
series with respect to winding 1 through n. The added inductances
L.sub.i.sup..DELTA. can be implemented so that they do not couple
magnetically with any of the n windings nor with one another.
Because there is no cross-coupling, the inductances
L.sub.i.sup..DELTA. are added along the diagonal of the L matrix,
leaving the other elements of L unchanged. The repartition is
determined by the electronic processor by computing the inverse
matrix:
M .DELTA. = [ L + [ L 1 .DELTA. 0 0 L n .DELTA. ] ] - 1
##EQU00007##
which yields the corresponding current amplitude
I.sub.i.sup..DELTA. repartition:
[ I 1 .DELTA. I n .DELTA. ] = M .DELTA. .omega. [ V V ]
##EQU00008##
where V is the shared voltage across the windings connected in
parallel.
[0231] The electronic processor determines the combination of
inductances L.sub.1.sup..DELTA., . . . , L.sub.n.sup..DELTA. that
minimizes the figure of merit:
i = 1 n ( I i .DELTA. I - 1 n ) 2 ##EQU00009##
where .SIGMA.I=I.sub.i.sup..DELTA. for i=1 to n and where n equals
the number of connected coils.
[0232] In some embodiments, combinations of inductances
L.sub.i.sup..DELTA., . . . , L.sub.n.sup..DELTA. that yield
approximately equal current repartition in the n windings may be
more easily found with the addition of large inductor values. In
the absence of other constraints, some large values of
L.sub.i.sup..DELTA., . . . , L.sub.n.sup..DELTA. may naturally
minimize the figure of merit:
i = 1 n ( I i .DELTA. I - 1 n ) 2 . ##EQU00010##
[0233] For example, in certain embodiments, the added inductances
L.sub.1.sup..DELTA., . . . , L.sub.n.sup..DELTA. can be much larger
than the inductances of the windings. As an example,
L.sub.1.sup..DELTA.= . . . =L.sub.n.sup..DELTA.=.infin. would yield
an ideal current repartition among the windings. Similarly, if the
added inductances L.sub.1.sup..DELTA., . . . , L.sub.n.sup..DELTA.
are much larger than the most inductive winding, a near-perfect
current splitting can result. However, this may not be useful from
a practical standpoint. For practical reasons, considerations such
as space constraints, limiting additional losses, limiting the
additional amount of magnetic material, limiting unwanted
additional overall inductance, and limiting additional complexity
in wireless power transfer systems can be taken into account by the
electronic processor during the optimization. For example, the
number of windings n may be limited due to the space constraints
placed on the overall size of the source resonator coil. In some
embodiments, the range of values for L.sub.1.sup..DELTA., . . . ,
L.sub.n.sup..DELTA. may be constrained to be less than or equal to
the scaled inductance of the windings:
L.sub.i.sup..DELTA..ltoreq..alpha.L.sub.i
where .alpha. is a scalar between zero and the difference between
the maximum inductance of a winding and the minimum inductance of a
winding.
[0234] In certain embodiments, the number of windings m to which
additional inductance is added can be determined by the electronic
processor. The number of windings to which additional inductance is
added may be limited to be less than or equal to the total number
of windings n. In some embodiments, the number of windings m to
which additional inductance is added can be determined before
optimizing the current distribution over the n windings. In certain
embodiments, different combinations of the number of windings m to
which additional inductance is added can be iteratively selected
during the current distribution optimization.
[0235] In some embodiments, if m=n-1, the one winding where no
inductor is added is the one that has the largest inductance.
Conversely, if m<n-1, inductors can be added to the winding that
have the lowest inductance and leave the n-m windings with the
largest inductance without additional inductors for current
balancing. In certain embodiments, an electronic processor can
control the adding of inductors based on changes to the order of
inductance of the windings depending on external factors (presence
of a device, presence of lossy materials, etc.).
[0236] For example, for n=3 windings and m=1, three different
optimizations can be performed by the electronic processor, in each
optimization setting two of the three inductance values of the
additional inductors to zero, as follows:
L.sub.2.sup..DELTA.=L.sub.3.sup..DELTA.=0 (optimization 1)
L.sub.1.sup..DELTA.=L.sub.3.sup..DELTA.=0 (optimization 2)
L.sub.1.sup..DELTA.=L.sub.2.sup..DELTA.=0 (optimization 3)
[0237] The electronic processor can optimize the combination of
additional inductances (L.sub.1.sup..DELTA., L.sub.2.sup..DELTA.,
L.sub.3.sup..DELTA.) that minimizes:
( I 1 .DELTA. I - 1 3 ) 2 + ( I 2 .DELTA. I - 1 3 ) 2 + ( I 3
.DELTA. I - 1 3 ) 2 ##EQU00011##
where
.SIGMA.I=I.sub.1.sup..DELTA.+I.sub.2.sup..DELTA.+I.sub.3.sup..DELTA-
.. This yields three optimized configurations and corresponding
current distributions in the three windings. The electronic
processor can then compare the three configurations to determine
which additional inductor, L.sub.1.sup..DELTA.,
L.sub.2.sup..DELTA., or L.sub.3.sup..DELTA., yields the best
results in terms of current distribution (or another metric).
[0238] In some embodiments, as shown in FIG. 57C, a single inductor
5702 can be used to yield approximately equal current repartition
among multiple parallel coil windings. For example, a single
inductor L.sub.1.sup..DELTA.=2.18 .mu.H (i.e.
L.sub.2.sup..DELTA.=L.sub.3.sup..DELTA.=0) connected to winding
5602 (the innermost winding) can significantly improve current
repartition, with an approximately 33.5%-34.6%-31.9% predicted
distribution. These results are within 5% of a target of
approximately equal current distribution (e.g., 33.3%) in each
winding. In some embodiments, the optimization can be performed
iteratively by the electronic processor until the current
distribution in each of the windings is within 20% (e.g., within
15%, within 10%, within 5%) of a target current distribution, which
can be (but need not be) an equal current distribution among the
windings. The process shown in flow chart 7400 then ends at step
7408.
[0239] In certain embodiments, a slightly reduced current in
outermost winding 5606 (winding 3) can be advantageous, since the
longer length of winding 5606 yields a resistance that can be about
8% higher than the resistance of the other windings. In some
embodiments, a specific current distribution may be desired and the
electronic processor can minimize a more general
figure-of-merit
i = 1 n ( I i .DELTA. I - s i ) 2 ##EQU00012##
where s.sub.i is the targeted fraction of the total current flowing
in winding i (thus .SIGMA..sub.i=1.sup.ns.sub.i=1). For example,
one of the windings may be less effectively cooled compared to
other windings and thus, current repartition may be changed
accordingly. In another example, some windings may have a different
than expected resistance or inductance at the time of manufacture
and an electronic processor may be able to compensate for these
differences.
[0240] In some embodiments, inductor L.sub.1.sup..DELTA. can
include a pair of cores formed from magnetic material. For example,
an inductor with a planar E-core formed of ferrite can be used. For
an inductor with a 2 mm gap and 4 turns of 4200/44 Litz wire, the
inductor L.sub.1.sup..DELTA. can dissipate approximately 3.2 W at a
source current value I.sub.s=140 A.sub.RMS. Inductor
L.sub.1.sup..DELTA. can be wound with the same Litz wire as the
innermost winding (winding 5602) and can therefore simplify the
connection. In certain embodiments, with the addition of inductor
L.sub.1.sup..DELTA., the overall inductance measurements yield:
L = ( 24.2 19.6 19.2 19.6 23.0 20.3 19.2 20.3 23.7 ) .mu.H .
##EQU00013##
[0241] When a voltage V is applied across the three windings
connected in parallel, the following relationship is the
result:
( I 1 , I 2 , I 3 ) = ( 1.56 , 1.68 , 1.52 ) 10 4 V .omega. .
##EQU00014##
[0242] The current repartition among the windings can be calculated
by summing the elements in the above vector (1.56+1.68+1.52=4.76)
and dividing each element by the sum. This shows that the innermost
winding can carry 32.8% of the current, while the middle winding
carries 35.2% of the current and the outermost winding carries
32.0% of the current. For example, for a wireless power transfer
system that is tested at 2.5 kW output with 27.8 A.sub.RMS total
current in the source coil (with three parallel windings), the
following current repartition is achieved:
innermost winding: 8.4 A.sub.RMS (30.2%),
middle winding: 9.9 A.sub.RMS (35.7%),
outermost winding: 9.5 A.sub.RMS (34.1%).
This current repartition is close to the expected current
repartition based on low-power measurements.
[0243] In current embodiments, windings that carry larger currents
can reach higher operating temperatures and/or dissipate the most
power. FIGS. 58A and 58B are images showing temperature
measurements for windings in a coil carrying a total of 10 A of
current during operation. The middle winding 5604 heats up to a
temperature that is slightly larger (see FIG. 58A, measurement at
location 5802) than the temperature of innermost winding 5602 and
outermost winding 5606 (see FIG. 58B, measurement at location
5804), consistent with the predictions above. FIG. 59 is a plot
showing the current carried in each of windings 5602 (curve 5902),
5604 (curve 5904), and 5606 (curve 5906) as a function of time. As
shown in the figure, the current carried in the middle winding
(winding 5604) is larger than the currents carried in the innermost
and outermost windings, accounting for the greater heating of the
middle winding.
Hardware and Software Implementation
[0244] FIG. 75 shows an example of an electronic controller 7503,
which may be used with the systems and methods described herein. As
mentioned earlier, the electronic controller (and more
specifically, an electronic processor thereof such as processor 105
and/or 111) can be used to perform any of the control and/or
computation functions disclosed herein, including controlling power
transfer of a wireless power transfer system, for example, by
changing power output of a power source, adjusting operation and/or
resonant frequencies and adjusting impedance matching networks. The
electronic controller 7503 can be used to control the current
directions, magnitudes and phases of different coils relative to
other coils. In some embodiments, the electronic controller 7503
can be directly connected to, or wirelessly communicate with,
various elements of the system.
[0245] Electronic controller 7503 can include a processor 7502
(e.g., corresponding to processor 105 and/or 111), memory 7504, a
storage device 7506 and interfaces 7508 for interconnection. The
processor 7502 can process instructions for execution within the
electronic controller 7503, including instructions stored in the
memory 7504 or on the storage device 7506. For example, the
instructions can instruct the processor 7502 to determine
parameters of the system such as efficiency of power transfer,
operating frequency, resonant frequencies of resonators and
impedance matching conditions. The electronic controller 7503 can
determine type, size and alignment of a power receiving apparatus
based on detection signals from one or more sensors. In certain
embodiments, the processor 7502 is configured to send out control
signals to various elements (e.g., power source, power transmitting
apparatus, power receiving apparatus, power repeating apparatus,
impedance matching networks) to adjust the determined parameters.
For example, control signals can be used to tune capacitance values
of capacitors in an impedance matching network. In certain
embodiments, control signals can be used to adjust operation
frequency of a power source. Control signals can change capacitance
value of a capacitor in a resonator to tune its resonant frequency,
and/or change inductance values of tunable inductors to repartition
currents among parallel windings in a resonator coil.
[0246] The memory 7504 can store information about optimized
parameters of the system. For example, the information can include
optimized impedance matching conditions for various levels of power
output from the power source. In certain embodiments, the memory
7504 can store information such as resonant frequencies of
resonator and magnetic properties (e.g., magnetic permeability
depending on power levels) of magnetic components in the system,
which can be used by the processor 7502 for determining signals to
be sent out to control various elements in the system. The memory
can also store a set of values corresponding to a target current
repartition.
[0247] The storage device 7506 can be a computer-readable medium,
such as a floppy disk device, a hard disk device, an optical disk
device, or a tape device, a flash memory or other similar solid
state memory device, or an array of devices, including devices in a
storage area network or other configurations. The storage device
7506 can store instructions that can be executed by processor 7502
described above. In certain embodiments, the storage device 7506
can store information described in relation to memory 7504.
[0248] In some embodiments, electronic controller 7503 can include
a graphics processing unit to display graphical information (e.g.,
using a GUI or text interface) on an external input/output device,
such as display 7516. The graphical information can be displayed by
a display device (e.g., a CRT (cathode ray tube) or LCD (liquid
crystal display) monitor) for displaying information. A user can
use input devices (e.g., keyboard, pointing device, touch screen,
speech recognition device) to provide input to the electronic
controller 7503. In some embodiments, the user can monitor the
display 7516 to analyze the power transfer conditions of the
system. For example, when the power transfer is not in optimum
condition, the user can adjust parameters (e.g., power transfer
level, capacitor values in impedance matching networks, operation
frequency of power source, resonant frequencies of resonators) by
inputting information through the input devices. Based on the
receive input, the electronic controller 7503 can control the
system as described above.
[0249] In some embodiments, the electronic controller 7503 can
monitor hazardous conditions of the system. For example, the
electronic controller 7503 can detect over-heating in the system
and provide an alert (e.g., visual and/or audible alert) to the
user through its graphical display or audio device.
[0250] In certain embodiments, electronic controller 7503 can be
used to control magnitudes and phases of currents flowing in one or
more coils of the wireless power transfer system. For example,
processor 7502 can calculate and determine the magnitudes and phase
of currents to be supplied to coils in a power transmitting
apparatus. The determination can be based on the monitored power
transfer efficiency and information stored in memory 7504 or
storage unit 7506.
[0251] A feedback signal can be received and processed by the
electronic controller 7503. For example, the electronic controller
7503 can include a wireless communication device (e.g.,
radio-frequency, Bluetooth receiver) to receive information from
either or both of a power transmitting apparatus and a power
receiving apparatus (which can have its own wireless communication
device). In some embodiments, the received information can be
processed by processor 7502, which can further send out control
signals to adjust parameters of the system as described above. For
example, the control signals can be used to adjust the magnitudes
and phases of currents flowing in one or more coils of resonators
in the system to increase the power transfer efficiency.
[0252] Various embodiments of the systems and techniques described
here can be realized by one or more computer programs that are
executable and/or interpretable on the electronic controller 7503.
These computer programs (also known as programs, software, software
applications or code) include machine instructions for a
programmable processor, and can be implemented in a high-level
procedural and/or object-oriented programming language, and/or in
assembly/machine language. For example, computer programs can
contain the instructions that can be stored in memory 7504 and
storage unit 7506 and executed by processor 7502 as described
above. As used herein, the terms "computer-readable medium" refers
to any computer program product, apparatus and/or device (e.g.,
magnetic discs, optical disks, memory, Programmable Logic Devices
(PLDs)) used to provide machine instructions and/or data to a
programmable processor, including a machine-readable medium that
receives machine instructions.
[0253] Generally, electronic controller 7503 can be implemented in
a computing system to implement the operations described above. For
example, the computing system can include a back end component
(e.g., as a data server), or a middleware component (e.g., an
application server), or a front end component (e.g., a client
computer having a graphical user-interface), or any combination
therefor, to allow a user to utilized the operations of the
electronic controller 7503.
[0254] The electronic controller 7503 or one or more of its
elements can be integrated in a vehicle. The electronic controller
7503 can be utilized to control and/or monitor wireless power
charging of a battery installed in the vehicle. In some
embodiments, the display 7516 can be installed adjacent to the
driving wheel of the vehicle so that a user may monitor conditions
of the power charging and/or control parameters of the power
charging as described in relation to FIG. 75. The display 7516 can
also visualize information traffic information and road maps based
on Global Positioning System (GPS) information. Any of the elements
such as the processor 7502, memory 7504 and storage device 7506 can
be installed in the space behind the display 7516, which can
visualize the data process by those elements.
Other Embodiments
[0255] While this disclosure contains many specific implementation
details, these should not be construed as limitations on the scope
of the disclosure, but rather as descriptions of features in
connection with embodiments. Features that are described in the
context of separate embodiments can also generally be implemented
in combination in a single embodiment. Conversely, various features
that are described in the context of a single embodiment can also
be implemented in multiple embodiments separately or in any
suitable sub-combination. Moreover, although features may be
described above as acting in certain combinations and even
initially claimed as such, one or more features from a claimed
combination can generally be excised from the combination, and the
claimed combination may be directed to a sub-combination or
variation of a sub-combination.
[0256] In addition to the embodiments expressly disclosed herein,
other embodiments are within the scope of the disclosure.
* * * * *