U.S. patent application number 14/644308 was filed with the patent office on 2015-09-17 for time-of-flight (tof) receiver with high dynamic range.
The applicant listed for this patent is Texas Instruments Incorporated. Invention is credited to Sandeep Kesrimal Oswal, Raja Reddy Patukuri, Prabu Sankar Thirugnanam, Jagannathan Venkataraman.
Application Number | 20150260571 14/644308 |
Document ID | / |
Family ID | 54068546 |
Filed Date | 2015-09-17 |
United States Patent
Application |
20150260571 |
Kind Code |
A1 |
Venkataraman; Jagannathan ;
et al. |
September 17, 2015 |
TIME-OF-FLIGHT (TOF) RECEIVER WITH HIGH DYNAMIC RANGE
Abstract
The disclosure provides a receiver with high dynamic range. The
receiver includes a photodiode that generates a current signal. A
coupling capacitor is coupled to the photodiode, and generates a
modulation signal in response to the current signal received from
the photodiode. A sigma delta analog to digital converter (ADC) is
coupled to the coupling capacitor, and generates a digital data in
response to the modulation signal. A digital mixer is coupled to
the sigma delta ADC, and generates an in-phase component and a
quadrature component corresponding to the digital data. A processor
is coupled to the digital mixer, and processes the in-phase
component and the quadrature component corresponding to the digital
data.
Inventors: |
Venkataraman; Jagannathan;
(Bangalore, IN) ; Thirugnanam; Prabu Sankar;
(Chennai, IN) ; Patukuri; Raja Reddy; (Nizamabad,
IN) ; Oswal; Sandeep Kesrimal; (Bangalore,
IN) |
|
Applicant: |
Name |
City |
State |
Country |
Type |
Texas Instruments Incorporated |
Dallas |
TX |
US |
|
|
Family ID: |
54068546 |
Appl. No.: |
14/644308 |
Filed: |
March 11, 2015 |
Current U.S.
Class: |
250/206 |
Current CPC
Class: |
H03M 3/402 20130101;
H03M 3/494 20130101; H03M 3/30 20130101; H03M 3/43 20130101 |
International
Class: |
G01J 1/44 20060101
G01J001/44; H03M 3/00 20060101 H03M003/00; H01L 27/144 20060101
H01L027/144 |
Foreign Application Data
Date |
Code |
Application Number |
Mar 11, 2014 |
IN |
1245/CHE/2014 |
Claims
1. A receiver comprising: a photodiode configured to generate a
current signal; a coupling capacitor coupled to the photodiode, and
configured to generate a modulation signal in response to the
current signal received from the photodiode; a sigma delta analog
to digital converter (ADC) coupled to the coupling capacitor and
configured to generate a digital data in response to the modulation
signal; a digital mixer coupled to the sigma delta ADC and
configured to generate an in-phase component and a quadrature
component corresponding to the digital data; and a processor
coupled to the digital mixer and configured to process the in-phase
component and the quadrature component corresponding to the digital
data.
2. The receiver of claim 1 further comprising an ambient
cancellation circuit coupled to the photodiode, wherein the ambient
cancellation circuit comprising: a feedback circuit coupled to the
photodiode; and a first resistor coupled between an output node of
the feedback circuit and the photodiode.
3. The receiver of claim 2, wherein the feedback circuit comprises:
a first operational amplifier configured to receive a first common
mode voltage at a first input port; a second resistor coupled
between the photodiode and a second input port of the first
operational amplifier; and a first capacitor coupled between the
second input port of the first operational amplifier and the output
node of the feedback circuit, wherein the output node of the
feedback circuit receives an output of the first operational
amplifier.
4. The receiver of claim 1, wherein the sigma delta ADC comprises:
a first summer coupled to the coupling capacitor and configured to
generate an error signal in response to the modulation signal and a
feedback current signal; a loop filter coupled to the first summer
and configured to filter the error signal to generate a filtered
signal; a quantizer coupled to the loop filter and configured to
generate the digital data in response to the filtered signal; and a
digital to analog converter (DAC) configured to receive the digital
data as a feedback signal and configured to generate the feedback
current signal, wherein the first summer is configured to receive
the feedback current signal from the DAC.
5. The receiver of claim 4, wherein the loop filter comprises: a
continuous time resonator configured to generate a low pass voltage
and a band pass voltage in response to the error signal; a second
summer coupled to the continuous time resonator and configured to
receive the low pass voltage and the band pass voltage, the second
summer configured to generate a summed voltage; a discrete time
resonator coupled to the second summer and configured to generate a
discrete voltage in response to the summed voltage; and a third
summer coupled to the discrete time resonator, and configured to
receive the summed voltage and the discrete voltage, the third
summer configured to generate the filtered signal.
6. The receiver of claim 5, wherein the continuous time resonator
comprises: a second operational amplifier configured to receive a
second common mode voltage at a first input port, and configured to
receive the error signal at a second input port, the second
operational amplifier and configured to generate the band pass
voltage at a first output port; a second capacitor coupled between
a second input port of the second operational amplifier and the
first output port of the second operational amplifier; a third
operational amplifier configured to receive a third common mode
voltage at a first input port, the third operational amplifier
configured to generate the low pass voltage at a second output
port; a third capacitor coupled between a second input port of the
third operational amplifier and the second output port of the third
operational amplifier; a third resistor coupled between the first
output port of the second operational amplifier and the second
input port of the third operational amplifier; and a fourth
resistor coupled between the second input port of the second
operational amplifier and the second output port of the third
operational amplifier.
7. The receiver of claim 5, wherein the second summer comprises: a
fourth operational amplifier configured to receive a fourth common
mode voltage at a first input port, the fourth operational
amplifier configured to generate the summed voltage at a third
output port; a fourth capacitor coupled between the second output
port of the third operational amplifier and a second input port of
the fourth operational amplifier, the fourth capacitor configured
to receive the low pass voltage; a fifth capacitor coupled between
the first output port of the second operational amplifier and the
first input port of the fourth operational amplifier, the fifth
capacitor configured to receive the band pass voltage; a sixth
capacitor coupled between the second input port of the fourth
operational amplifier and the third output port of the fourth
operational amplifier; and a fifth resistor coupled between the
second input port of the fourth operational amplifier and the third
output port of the fourth operational amplifier.
8. The receiver of claim 1, wherein the sigma delta ADC is a band
pass sigma delta ADC.
9. A method comprising: generating a current signal from reflected
light pulses received by a photodiode, wherein the current signal
comprises a direct current (DC) signal; generating a modulation
signal from the current signal; generating a digital data in
response to the modulation signal; generating an in-phase component
and a quadrature component corresponding to the digital data; and
processing the in-phase component and the quadrature component
corresponding to the digital data.
10. The method of claim 9 further comprising cancelling the DC
signal in an ambient cancellation circuit, the ambient cancellation
circuit comprising: a first operational amplifier configured to
receive a first common mode voltage at a first input port; a second
resistor coupled between the photodiode and a second input port of
the first operational amplifier; a first capacitor coupled between
the second input port of the first operational amplifier and an
output node of the first operational amplifier; and a first
resistor coupled between the output node of the first operational
amplifier and the photodiode.
11. The method of claim 9, wherein generating the modulation signal
is by providing the current signal to a coupling capacitor.
12. The method of claim 9, wherein generating the digital data in
response to the modulation signal comprises: summing the modulation
signal and a feedback current signal from a digital to analog
converter (DAC) to generate an error signal; filtering the error
signal to generate a filtered signal; and quantizing the filtered
signal to generate the digital data, wherein the digital data is
provided as a feedback signal to the DAC.
13. The method of claim 9, wherein filtering the error signal
comprises: integrating the error signal to generate a band pass
voltage; integrating the band pass voltage to generate a low pass
voltage; summing the band pass voltage and the low pass voltage to
generate a summed voltage; generating a discrete voltage in
response to the summed voltage; and summing the discrete voltage
and the summed voltage to generate the filtered signal.
14. A time-of-flight (TOF) system comprising: a light source for
emitting light pulses at a clock frequency; a TOF sensor comprising
one or more TOF sensor pixels, each TOF sensor pixel comprising a
receiver, the receiver comprising: a photodiode configured to
generate a current signal from reflected light pulses, the light
pulses emitted by the light source are reflected from a target to
generate the reflected light pulses; a coupling capacitor coupled
to the photodiode, and configured to generate a modulation signal
in response to the current signal received from the photodiode; a
sigma delta analog to digital converter (ADC) coupled to the
coupling capacitor and configured to generate a digital data in
response to the modulation signal; and a digital mixer coupled to
the sigma delta ADC and configured to generate an in-phase
component and a quadrature component corresponding to the digital
data; a processor coupled to the TOF sensor and configured to
process the in-phase component and the quadrature component
received from the digital mixer in each receiver.
15. The TOF system of claim 14 further comprising an ambient
cancellation circuit coupled to the photodiode, wherein the ambient
cancellation circuit comprising: a feedback circuit coupled to the
photodiode; and a first resistor coupled between an output node of
the feedback circuit and the photodiode.
16. The TOF system of claim 15, wherein the feedback circuit
comprises: a first operational amplifier configured to receive a
first common mode voltage at a first input port; a second resistor
coupled between the photodiode and a second input port of the first
operational amplifier; and a first capacitor coupled between the
second input port of the first operational amplifier and the output
node of the feedback circuit, wherein the output node of the
feedback circuit receives an output of the first operational
amplifier.
17. The TOF system of claim 14, wherein the sigma delta ADC
comprises: a first summer coupled to the coupling capacitor and
configured to generate an error signal in response to the
modulation signal and a feedback current signal; a loop filter
coupled to the first summer and configured to filter the error
signal to generate a filtered signal; a quantizer coupled to the
loop filter and configured to generate the digital data in response
to the filtered signal; and a digital to analog converter (DAC)
configured to receive the digital data as a feedback signal and
configured to generate the feedback current signal, wherein the
first summer is configured to receive the feedback current signal
from the DAC.
18. The TOF system of claim 17, wherein the loop filter comprises:
a continuous time resonator configured to generate a low pass
voltage and a band pass voltage in response to the error signal; a
second summer coupled to the continuous time resonator and
configured to receive the low pass voltage and the band pass
voltage, the second summer configured to generate a summed voltage;
a discrete time resonator coupled to the second summer and
configured to generate a discrete voltage in response to the summed
voltage; and a third summer coupled to the discrete time resonator,
and configured to receive the summed voltage and the discrete
voltage, the third summer configured to generate the filtered
signal.
19. The TOF system of claim 18, wherein the continuous time
resonator comprises: a second operational amplifier configured to
receive a second common mode voltage at a first input port, the
second operational amplifier configured to generate the band pass
voltage at a first output port; a second capacitor coupled between
a second input port of the second operational amplifier and the
first output port of the second operational amplifier; a third
operational amplifier configured to receive a third common mode
voltage at a first input port, the third operational amplifier
configured to generate the low pass voltage at a second output
port; a third capacitor coupled between a second input port of the
third operational amplifier and the second output port of the third
operational amplifier; a third resistor coupled between the first
output port of the second operational amplifier and the second
input port of the third operational amplifier; and a fourth
resistor coupled between the second input port of the second
operational amplifier and the second output port of the third
operational amplifier.
20. The TOF system of claim 18, wherein the second summer
comprises: a fourth operational amplifier configured to receive a
fourth common mode voltage at a first input port, the fourth
operational amplifier configured to generate the summed voltage at
a third output port; a fourth capacitor coupled between the second
output port of the third operational amplifier and a second input
port of the fourth operational amplifier, the fourth capacitor
configured to receive the low pass voltage; a fifth capacitor
coupled between the first output port of the second operational
amplifier and the first input port of the fourth operational
amplifier, the fifth capacitor configured to receive the band pass
voltage; a sixth capacitor coupled between the second input port of
the fourth operational amplifier and the third output port of the
fourth operational amplifier; and a fifth resistor coupled between
the second input port of the fourth operational amplifier and the
third output port of the fourth operational amplifier.
Description
CROSS REFERENCES TO RELATED APPLICATIONS
[0001] This application claims priority from India provisional
patent application No. 1245/CHE/2014 filed on Mar. 11, 2014 which
is hereby incorporated by reference in its entirety.
TECHNICAL FIELD
[0002] The present disclosure is generally related to
time-of-flight (TOF) systems, and more particularly to achieving
high dynamic range in a receiver associated with a TOF system.
BACKGROUND
[0003] An emerging category of electronic devices is time-of-flight
(TOF) systems. The TOF systems find applications in accelerometers,
monolithic gyroscopes, light sensors, conveyor belts, depth
sensing, proximity sensing, gesture recognition and imagers. A TOF
system includes a light source that emits light pulses. The light
pulses are emitted towards a target, which reflects the light
pulses. The target is any object of interest which may include, but
not limited to, a human, an automated component, an animal, an
electronic device etc. A TOF sensor in the TOF system receives the
reflected light pulses. The TOF sensor receives the reflected light
pulses after a time of flight, which is proportional to a distance
of the target from the TOF system.
[0004] The TOF sensor includes one or more TOF sensor pixels. Each
TOF sensor pixel includes a receiver. The receiver processes the
reflected light pulses to estimate the distance of the target from
the TOF system. In addition to the reflected light pulses, the
receiver also receives other signals such as, but not limited to,
electrical interferences, crosstalk signals and ambient light. The
ambient light is due to one or more of the following, but not
limited to, florescent lamps, sunlight, bulbs etc.
[0005] The ambient light causes a direct current (DC) signal being
generated in the receiver. A high DC signal saturates the receiver.
This causes an error in estimating the distance of the target from
the TOF system. In one example, a signal strength of the reflected
light pulse is 30 dB whereas an interference generated is of the
order of 100 dB. Thus, a receiver with high dynamic range is
required.
SUMMARY
[0006] According to an aspect of the disclosure, a receiver is
disclosed. The receiver includes a photodiode that generates a
current signal. A coupling capacitor is coupled to the photodiode,
and generates a modulation signal in response to the current signal
received from the photodiode. A sigma delta analog to digital
converter (ADC) is coupled to the coupling capacitor, and generates
a digital data in response to the modulation signal. A digital
mixer is coupled to the sigma delta ADC, and generates an in-phase
component and a quadrature component corresponding to the digital
data. A processor is coupled to the digital mixer, and processes
the in-phase component and the quadrature component corresponding
to the digital data.
BRIEF DESCRIPTION OF THE VIEWS OF DRAWINGS
[0007] FIG. 1 illustrates a receiver;
[0008] FIG. 2 illustrates a receiver, according to an
embodiment;
[0009] FIG. 3 illustrates an ambient cancellation circuit,
according to an embodiment;
[0010] FIG. 4 illustrates a loop filter, according to an
embodiment;
[0011] FIG. 5 illustrates a continuous time resonator, according to
an embodiment;
[0012] FIG. 6 illustrates a summer, according to an embodiment;
and
[0013] FIG. 7 illustrates a time-of-flight (TOF) system, according
to an embodiment.
DETAILED DESCRIPTION OF THE EMBODIMENTS
[0014] FIG. 1 illustrates a receiver 100. The receiver 100 includes
a photodiode 102, a trans-impedance amplifier (TIA) 110, a mixer I
114, a mixer Q 116, a first filter and gain block 118, a second
filter and gain block 120, a first analog to digital converter
(ADC) 124, a second analog to digital converter (ADC) 126 and a
processor 130.
[0015] The photodiode 102 includes a sensor 104 and an associated
capacitance C.sub.D 106. The TIA 110 is coupled to the photodiode
102. The mixer I 114 and the mixer Q 116 are coupled to the TIA
110. The first filter and gain block 118 is coupled to the mixer I
114. The second filter and gain block 120 is coupled to the mixer Q
116.
[0016] The first ADC 124 is coupled to the first filter and gain
block 118. The second ADC 126 is coupled to the second filter and
gain block 120. The processor 130 is coupled to the first ADC 124
and the second ADC 126.
[0017] The operation of the receiver 100 illustrated in FIG. 1 is
explained now. The receiver in one example is used in a
time-of-flight (TOF) system. The TOF system includes a light source
that emits light pulses. The light pulses are emitted towards a
target, which reflects the light pulses. The receiver 100 in the
TOF system receives the reflected light pulses. The receiver 100
receives the reflected light pulses after a time of flight, which
is proportional to a distance of the target from the TOF
system.
[0018] The sensor 104 in the photodiode 102 receives the reflected
light pulses. The associated capacitance C.sub.D 106 stores a
charge corresponding to the reflected light pulses. The charge
represents a modulated signal received by the photodiode 102. The
charge stored in the associated capacitance C.sub.D 106 is received
by the TIA 110. The TIA 110 demodulates the modulated signal to
generate an in-phase voltage and a quadrature voltage.
[0019] The in-phase voltage is receiver by the mixer I 114, and the
quadrature voltage is received by the mixer Q 116. The mixer I 114
multiplies the in-phase voltage and a voltage corresponding to the
light pulses generated by the TOF system, to generate an in-phase
voltage component. The mixer Q 116 multiplies the quadrature
voltage and the voltage corresponding to the light pulses generated
by the TOF system, to generate a quadrature voltage component.
[0020] The in-phase voltage component is filtered and amplified in
the first filter and gain block 118. The quadrature voltage
component is filtered and amplified in the second filter and gain
block 120. The first ADC 124 generates a first digital data
corresponding to an output of the first filter and gain block 118.
The second ADC 126 generates a second digital data corresponding to
an output of the second filter and gain block 120.
[0021] The processor 130 processes the first digital data and the
second digital data to estimate a distance of the target from the
TOF system. The mixer I 114 and the mixer Q 116 provides an offset
to the in-phase voltage component and quadrature phase component.
This results in phase errors in the first digital data and the
second digital data. In addition, the filter and gain block 118 and
the filter and gain block 120, introduces gain error in the first
digital data and the second digital data respectively.
[0022] The receiver 100 also receives ambient light. The ambient
light is due to one or more of the following, but not limited to,
florescent lamps, sunlight, bulbs etc. The ambient light causes a
direct current (DC) being generated in the receiver 100. The
receiver 100 processes this direct current through the mixer,
filter and gain block and ADC. This current is cancelled in the
processor 130. However, when ambient light conditions are above a
threshold, it is required to cancel this direct current at the
input.
[0023] This is because if the direct current is cancelled by the
processor 130, a dynamic range of the receiver 100 gets limited as
it gets saturated by the direct current. Thus, the receiver 100
does not provide a high dynamic range in a TOF system in the
presence of ambient light. In one example, the direct current
generated in the receiver 100 is 30 uA while an alternating current
generated from the reflected light pulses is 100 pA. Thus, the
receiver 100 is required to detect the 100 pA alternating current
in presence of 30 uA direct current without reaching
saturation.
[0024] FIG. 2 illustrates a receiver 200, according to an
embodiment. The receiver 200 includes a photodiode 202, an ambient
cancellation circuit 208, a coupling capacitor Cc 210, a sigma
delta analog to digital converter (ADC) 220, a digital mixer 226
and a processor 230.
[0025] The photodiode 202 includes a sensor 204 and an associated
capacitance C.sub.D 206. The ambient cancellation circuit 208 is
coupled to the photodiode 202. The coupling capacitor Cc 210 is
coupled to the photodiode 202. The sigma delta ADC 220 is coupled
to the coupling capacitor Cc 210. The sigma delta ADC 220 includes
a first summer 212, a loop filter 214, a quantizer 216 and a
digital to analog converter (DAC) 218.
[0026] The first summer 212 is coupled to the coupling capacitor Cc
210. The loop filter 214 is coupled to the first summer 212. The
quantizer 216 is coupled to the loop filter 214. The DAC 218 is
coupled to the quantizer 216.
[0027] The digital mixer 226 is coupled to the quantizer 216 in the
sigma delta ADC 220.
[0028] The processor 230 is coupled to the digital mixer 226. The
receiver 200 may include one or more additional components known to
those skilled in the relevant art and are not discussed here for
simplicity of the description.
[0029] The operation of the receiver 200 illustrated in FIG. 2 is
explained now. The receiver 200 in one example is used in a
time-of-flight (TOF) system. The TOF system includes a light source
that emits light pulses. The light pulses are emitted towards a
target, which reflects the light pulses. The receiver 200 in the
TOF system receives the reflected light pulses. The receiver 200
receives the reflected light pulses after a time of flight, which
is proportional to a distance of the target from the TOF
system.
[0030] The sensor 204 in the photodiode 202 receives the reflected
light pulses. The associated capacitance C.sub.D 206 stores a
charge corresponding to the reflected light pulses. The photodiode
202 generates a current signal based on the reflected light pulses.
The current signal includes a direct current (DC) signal. The DC
signal is generated because of ambient light received by the
photodiode 202 in the receiver 200. The ambient light is due to one
or more of the following, but not limited to, florescent lamps,
sunlight, bulbs etc.
[0031] The ambient cancellation circuit 208 cancels the DC signal
in the current signal. A modulation signal is generated when the
current signal is provided to the coupling capacitor Cc 210. The
modulation signal is provided to the sigma delta ADC 220. The sigma
delta ADC 220 generates a digital data in response to the
modulation signal. In one example, the sigma delta ADC 220 is a
band pass sigma delta ADC.
[0032] The first summer 212 in the sigma delta ADC 220 generates an
error signal in response to the modulation signal and a feedback
current signal. The loop filter 214 filters the error signal to
generate a filtered signal. The quantizer 216 quantizes the
filtered signal to generate the digital data. In one version, the
quantizer 216 is a 1 bit quantizer. This eases the node scaling
requirements for the receiver 200. Also, this creates a low power
requirement for zero crossing detection in the quantizer 216. The
DAC 218 receives the digital data as a feedback signal. The DAC 218
generates the feedback current signal. The first summer 212
receives the feedback current signal from the DAC 218. In one
example, the first summer 212 subtracts the feedback current signal
from the modulation signal to generate the error signal.
[0033] The digital mixer 226 receives the digital data from the
sigma delta ADC 220. The digital mixer 226 generates an in-phase
component (I) and a quadrature component (Q) corresponding to the
digital data. The processor 230 processes the in-phase component
and the quadrature component corresponding to the digital data. The
processor 230 thereby estimates a distance of the target from the
TOF system.
[0034] In receiver 200, the current signal is directly provided for
processing and the processing in the sigma delta ADC 220 is
performed on the modulation signal. Hence, a trans-impedance
amplifier is not required as in the receiver 100. Also, the sigma
delta ADC 220 generates the digital data; hence the mixing is
performed digitally in the digital mixer 226. This avoids any phase
or gain errors that are prevalent when mixing is performed in
analog domain such as in the receiver 100.
[0035] The sigma delta ADC 220 is a current input band pass sigma
delta ADC which makes it a power efficient ADC. In addition, the
ambient cancellation circuit 208 is separated from a circuit in the
receiver 200 which processes the modulation signal. The ambient
cancellation circuit 208 is capable of cancelling the DC signal in
the current signal.
[0036] As the DC signal is canceled at an input of the receiver 200
and is not processed with the modulation signal, it results in high
dynamic range of the receiver 200. The ambient cancellation circuit
208 provides a good low frequency ambient rejection and it also
supports in relaxing the noise constraints of the receiver 200.
[0037] FIG. 3 illustrates an ambient cancellation circuit 300,
according to an embodiment. The ambient cancellation circuit 300 is
analogous to the ambient cancellation circuit 208 (illustrated in
FIG. 2) in connection and operation. The ambient cancellation
circuit 300 is coupled to a photodiode 302. The photodiode 302 is
similar in connection and operation to the photodiode 202.
[0038] The ambient cancellation circuit 300 is also coupled to a
coupling capacitor Cc 310 which is further coupled to a sigma delta
ADC 320. The coupling capacitor Cc 310 is analogous in connection
and operation to the coupling capacitor Cc 210. The sigma delta ADC
320 is similar in connection and operation to the sigma delta ADC
220.
[0039] The ambient cancellation circuit 300 includes a feedback
circuit 305 and a first resistor R1 324. The feedback circuit 305
is coupled to the photodiode 302, and the first resistor R1 324 is
coupled between an output node 316 of the feedback circuit 305 and
the photodiode 302.
[0040] The feedback circuit 305 includes a first operational
amplifier 308. The first operational amplifier 308 receives a first
common mode voltage V.sub.CM1 304 at a first input port 312. A
second resistor R2 306 is coupled between the photodiode 302 and a
second input port 314 of the first operational amplifier 308.
[0041] A first capacitor C1 322 is coupled between the second input
port 314 of the first operational amplifier 308 and the output node
316 of the feedback circuit 305. The output node 316 of the
feedback circuit 305 receives an output of the first operational
amplifier 308. The ambient cancellation circuit 300 may include one
or more additional components known to those skilled in the
relevant art and are not discussed here for simplicity of the
description.
[0042] The operation of the ambient cancellation circuit 300
illustrated in FIG. 3 is explained now. The photodiode 302
generates a current signal based on received light pulses. The
current signal includes a direct current (DC) signal. The DC signal
is generated because of ambient light received by the photodiode
302. The ambient light is due to one or more of the following, but
not limited to, florescent lamps, sunlight, bulbs etc.
[0043] A voltage generated at the second input port 314 of the
first operational amplifier 308 because of the DC signal is
compared with the first common mode voltage V.sub.CM1 304. A
current noise generated because of a voltage noise (Vnoise) of the
first operational amplifier 308 is given by
Inoise=Vnoise/[R.sub.1*(1+[sC.sub.1R.sub.2].sup.-1).parallel.R2]
(1)
Inoise.apprxeq.Snoise/(R.sub.1.parallel.R2) (2)
[0044] In receiver 100, the current noise generated is proportional
to the associated capacitance C.sub.D 106. Thus, the effect of
voltage noise in the first operational amplifier 308 is scaled with
the R1//R2 resistor instead of the associated capacitance C.sub.D
106 in receiver 100. This helps in relaxing the power specification
of the first operational amplifier 308. Also, appropriately
choosing the first resistor R1 324 and the second resistor R2 306
would further reduce the noise generated by the first operational
amplifier 308.
[0045] The first operational amplifier 308 is a low power and a low
noise operational amplifier. As the DC signal is canceled at an
input of a receiver (or a TOF receiver) and is not processed with
the modulation signal, it results in high dynamic range of the
receiver. The ambient cancellation circuit 300 provides a good low
frequency ambient rejection and it also supports in relaxing the
noise constraints of the receiver, for example receiver 200.
[0046] The ambient cancellation circuit 300 sets a bias of the
photodiode 302 and cancels any out of band interferers. The ambient
cancellation circuit 300 prevents saturation of the photodiode
302.
[0047] FIG. 4 illustrates a loop filter 400, according to an
embodiment. The loop filter 400 is analogous to the loop filter 214
(illustrated in FIG. 2) in connection and operation. The loop
filter 400 is explained in connection with the receiver 200
illustrated in FIG. 2. The loop filter 400 includes a continuous
time resonator 406, a second summer 410, a discrete time resonator
416 and a third summer 420.
[0048] The continuous time resonator 406 receives an error signal
402 from a first summer similar to the first summer 212. The second
summer 410 is coupled to the continuous time resonator 406. The
discrete time resonator 416 is coupled to the second summer 410,
and the third summer 420 is coupled to the discrete time resonator
416. The loop filter 400 may include one or more additional
components known to those skilled in the relevant art and are not
discussed here for simplicity of the description.
[0049] The operation of the loop filter 400 illustrated in FIG. 4
is explained now. The continuous time resonator 406 generates a low
pass voltage V.sub.LP and a band pass voltage V.sub.BP in response
to the error signal. The loop filter 400 utilizes anti-aliasing
property of the continuous time resonator 406. The second summer
410 receives the low pass voltage V.sub.LP and the band pass
voltage V.sub.BP, and generates a summed voltage 412. In one
example, the second summer 410 sums the low pass voltage V.sub.LP
and the band pass voltage V.sub.BP, to generate a summed voltage
412.
[0050] The discrete time resonator 416 generates a discrete voltage
418 in response to the summed voltage 412. In one implementation, a
transfer function of the discrete time resonator 416 is given by
following equation.
S = Z - 2 1 + Z - 2 ( 3 ) ##EQU00001##
[0051] The discrete time resonator 416 provides band pass shaping
to the summed voltage 412 to generate the discrete voltage 418.
Also, the band pass shaping provided by the discrete time resonator
416 is independent of process variations. In the above
implementation, a center frequency of the discrete time resonator
416 is selected at Fs/4 to ease the processing requirement, for
example, in the receiver 200, where Fs is a sampling frequency.
[0052] The third summer 420 receives the summed voltage 412 and the
discrete voltage 418, and generates a filtered output 424. In one
version, the third summer 420 sums the summed voltage 412 and the
discrete voltage 418 to generate the filtered output 424. The
operation of each of the continuous time resonator 406 and the
second summer 410 is further explained in detail in the subsequent
paragraphs.
[0053] FIG. 5 illustrates a continuous time resonator 500,
according to an embodiment. The continuous time resonator 500 is
analogous to the continuous time resonator 406 (illustrated in FIG.
4) in connection and operation. The continuous time resonator 406
includes a second operational amplifier 506 and a third operational
amplifier 520. The second operational amplifier 506 receives a
second common mode voltage V.sub.CM2 at a first input port 504. A
second input port 508 of the second operational amplifier 506
receives an error signal 502 (similar to the error signal 402).
[0054] A second capacitor C2 512 is coupled between the second
input port 508 of the second operational amplifier 506 and a first
output port 516 of the second operational amplifier 506. The third
operational amplifier 520 receives a third common mode voltage
V.sub.CM3 528 at a first input port 522. A third resistor R3 532 is
coupled between the first output port 516 of the second operational
amplifier 506 and a second input port 524 of the third operational
amplifier 520.
[0055] A third capacitor C3 534 is coupled between the second input
port 524 of the third operational amplifier 520 and a second output
port 540 of the third operational amplifier 520. A fourth resistor
R4 542 is coupled between the second input port 508 of the second
operational amplifier 506 and the second output port 540 of the
third operational amplifier 520.
[0056] In one example, the continuous time resonator 500 includes a
negative feedback amplifier 536 coupled between the fourth resistor
R4 542 and the second output port 540 of the third operational
amplifier 520. The continuous time resonator 500 may include one or
more additional components known to those skilled in the relevant
art and are not discussed here for simplicity of the
description.
[0057] The operation of the continuous time resonator 500
illustrated in FIG. 5 is explained now. The second operational
amplifier 506 integrates the error signal 502 to generate a band
pass voltage V.sub.BP 544. The band pass voltage V.sub.BP 544 is
generated at the first output port 516 of the second operational
amplifier 506. The third operational amplifier 520 receives the
band pass voltage V.sub.BP 544. The third operational amplifier 520
integrates the band pass voltage V.sub.BP 544 to generate a low
pass voltage V.sub.LP 550.
[0058] A noise contribution from the continuous time resonator 500
is highly dependent on the fourth resistor R4 542, the third
resistor R3 532 and the second capacitor C2 512. An input current
noise has contributions from the second operational amplifier 506,
the fourth resistor R4 542 and the third resistor R3 532.
[0059] A contribution of noise from the fourth resistor R4 542 is
inversely proportional to a square root of the fourth resistor R4
542. Thus, a higher value of the fourth resistor R4 542 results in
a lower noise. A contribution of noise from the third resistor R3
532 is directly proportional to a product of the second capacitor
C2 512 and a square root of the third resistor R3 532. Thus, a
lower value of the second capacitor C2 512 and the third resistor
R3 532 results in a lower noise.
[0060] A corner frequency of the continuous time resonator 500 is
defined as
Corner Frequency = 1 R 4 .times. R 3 .times. C 2 .times. C 3 ( 4 )
##EQU00002##
[0061] To combat the direct current (DC) signal because of ambient
light, the fourth resistor R4 542 is scaled in accordance with a
voltage swing constraints of the second operational amplifier 506
and the third operational amplifier 520. A large DC signal would
require a lower fourth resistor R4 542 which increases the noise
contribution as discussed above.
[0062] A lower fourth resistor R4 542 is compensated by increasing
the third resistor R3 532, the second capacitor C2 512 and the
third capacitor C3 534, to maintain the corner frequency desired
from the continuous time resonator 500. Increasing the third
resistor R3 532, the second capacitor C2 512 and the third
capacitor C3 534 would increase the noise. Since, the ambient
cancellation circuit 208 is separated in the receiver 200; it
removes the constraints of lower fourth resistor R4 542. Each of
the fourth resistor R4 542, the third resistor R3 532, the second
capacitor C2 512 and the third capacitor C3 534 is optimized to
achieve best noise performance from the continuous time resonator
500.
[0063] FIG. 6 illustrates a summer 600, according to an embodiment.
The summer 600 is analogous to the second summer 410 (illustrated
in FIG. 4) in connection and operation. The summer 600 is explained
in connection with the loop filter 400 and the continuous time
resonator 500. The summer 600 includes a fourth operational
amplifier 610. The fourth operational amplifier 610 receives a
fourth common mode voltage V.sub.CM4 at a first input port 602. The
fourth operational amplifier 610 generates a summed voltage V.sub.S
620 at a third output port 606.
[0064] A fourth capacitor C4 616 is coupled between a second input
port 604 of the fourth operational amplifier 610 and the second
output port 540 of the third operational amplifier 520. The fourth
capacitor C4 616 receives a low pass voltage V.sub.LP 612 from the
continuous time resonator 500. A fifth capacitor C5 618 is coupled
between the first input port 602 of the fourth operational
amplifier 610 and the first output port 516 of the second
operational amplifier 506.
[0065] The fifth capacitor C5 618 receives a band pass voltage
V.sub.BP 618 from the continuous time resonator 500. A sixth
capacitor C6 622 and a fifth resistor R5 624 are coupled between
the second input port 604 of the fourth operational amplifier 610
and the third output port 606 of the fourth operational amplifier
610.
[0066] The band pass voltage V.sub.BP 618 and the low pass voltage
V.sub.LP 612 are required to summed and gained with different
coefficients. A large gain is required to be provided because of
node scaling. Typically, the gains used for the band pass voltage
V.sub.BP 618 and the low pass voltage V.sub.LP 612 are high to
detect a low modulation signal received from the photodiode, for
example photodiode 202.
[0067] The high gain means that an offset of the second operational
amplifier 506 and the third operational amplifier 520 (in the
continuous time resonator 500) are also gained to a high value.
This can saturate later stages of the receiver 200. In the summer
600, the band pass voltage V.sub.BP 618 and the low pass voltage
V.sub.LP 612 are provided high gain without increasing the offsets
of the second operational amplifier 506 and the third operational
amplifier 520 in the continuous time resonator 500.
[0068] The summer 600 is a capacitive summer. The summer 600
cancels any offsets provided by the second operational amplifier
506 and the third operational amplifier 520. The summer 600 sums
the band pass voltage V.sub.BP 618 and the low pass voltage
V.sub.LP 612 to generate the summed voltage V.sub.S 620. The summer
600 improves a dynamic range of a receiver for example the receiver
200.
[0069] FIG. 7 illustrates a time-of-flight (TOF) system 700,
according to an embodiment. The TOF system 700 includes a light
source 702, an amplifier 704 and a timing generator 706. The
amplifier 704 is coupled to the timing generator 706, and the light
source 702 is coupled to the amplifier 704. In one example, the
light source 702 is an infrared (IR) light emitting diode (LED)
that transmits IR light.
[0070] The TOF system 700 also includes a TOF sensor 712. The TOF
sensor 712 includes one or more TOF sensor pixels illustrated as
714. Each TOF sensor pixel of the one or more TOF sensor pixels
includes a receiver. The receiver is analogous to the receiver 200
in connection and operation.
[0071] In one example, the TOF sensor 712 is coupled to a processor
720, and each receiver in the TOF sensor 712 is coupled to the
processor 720. In another example, each receiver of the one or more
receivers in a TOF sensor is associated with a processor (similar
to receiver 200) and together these processors form processor
720.
[0072] In yet another example, one or more processors are coupled
to the TOF sensor 712. The processor 720 can be, for example, a
CISC-type (Complex Instruction Set Computer) CPU, RISC-type CPU
(Reduced Instruction Set Computer), or a digital signal processor
(DSP).
[0073] The timing generator 706 generates a clock frequency. The
light source 702 emits light pulses at the clock frequency. The
light pulses emitted by the light source 702 are reflected from a
target 710 to generate reflected light pulses. A photodiode
associated with a receiver in the TOF sensor 712 receives the
reflected light pulses.
[0074] The receiver processes the reflected light pulses similar to
the processing performed in the receiver 200 illustrated in FIG. 2.
The processor 720 process an in-phase component and a quadrature
component received from a digital mixer associated with each
receiver, to estimate a distance of the target 710 from the TOF
system 700.
[0075] A trans-impedance amplifier (TIA) is not required as in the
receiver 100. The requirement of building an accurate voltage at
high modulating frequency using TIA is eliminated in the TOF system
700. This helps in eliminating high power TIA and associated phase
shift variation caused by TIA. The mixing is performed digitally in
the digital mixer associated with each receiver.
[0076] This avoids any phase or gain errors that are prevalent when
mixing is performed in analog domain such as in the receiver 100.
The sigma delta ADC used in the receiver is a current input band
pass sigma delta ADC which makes it a power efficient ADC. In
addition, the DC signal is canceled at an input of the receiver
which results in high dynamic range of the receiver.
[0077] The foregoing description sets forth numerous specific
details to convey a thorough understanding of the invention.
However, it will be apparent to one skilled in the art that the
invention may be practiced without these specific details.
Well-known features are sometimes not described in detail in order
to avoid obscuring the invention. Other variations and embodiments
are possible in light of above teachings, and it is thus intended
that the scope of invention not be limited by this Detailed
Description, but only by the following Claims.
* * * * *