U.S. patent application number 14/583179 was filed with the patent office on 2015-08-27 for receiving device.
The applicant listed for this patent is FUJITSU LIMITED. Invention is credited to Hideki Furudate.
Application Number | 20150244490 14/583179 |
Document ID | / |
Family ID | 53883296 |
Filed Date | 2015-08-27 |
United States Patent
Application |
20150244490 |
Kind Code |
A1 |
Furudate; Hideki |
August 27, 2015 |
RECEIVING DEVICE
Abstract
A receiving device includes: a propagation path compensation
unit that compensates a signal by using a propagation path
characteristic; a power arithmetic section that arithmetically
operates power of a signal; a first reciprocal processing section
that performs reciprocal processing on power to output a signal; an
error arithmetic section that arithmetically operates an error of
the compensated signal; a subtractor that subtracts the signal
output from the first reciprocal processing section from the error;
a second reciprocal processing unit that performs reciprocal
processing on the signal output from the subtractor to output a
signal; a first multiplier that multiplies the power and the signal
output from the second reciprocal processing unit together to
output a signal; and a second multiplier that multiplies the signal
compensated by the propagation path compensation unit and the
signal output from the first multiplier together to output a signal
to an adder.
Inventors: |
Furudate; Hideki; (Ota,
JP) |
|
Applicant: |
Name |
City |
State |
Country |
Type |
FUJITSU LIMITED |
Kawasaki-shi |
|
JP |
|
|
Family ID: |
53883296 |
Appl. No.: |
14/583179 |
Filed: |
December 25, 2014 |
Current U.S.
Class: |
375/349 |
Current CPC
Class: |
H04L 25/0224 20130101;
H04L 25/0204 20130101; H04L 27/2647 20130101; H04L 25/022 20130101;
H04B 7/0845 20130101; H04L 25/03159 20130101 |
International
Class: |
H04J 11/00 20060101
H04J011/00; H04L 27/26 20060101 H04L027/26; H04B 7/08 20060101
H04B007/08 |
Foreign Application Data
Date |
Code |
Application Number |
Feb 27, 2014 |
JP |
2014-037072 |
Claims
1. A receiving device, comprising: a plurality of antennas; each of
a plurality of receiving circuits that receive signal via one of
the plurality of antennas, respectively; and an adder that adds
signals output from the plurality of receiving circuits, wherein
each of the plurality of receiving circuits comprises: a Fourier
transformation unit that transforms a signal into a frequency
domain from a time domain; a propagation path estimation unit that
estimates a propagation path characteristic based on a known signal
in the signal in the frequency domain transformed by the Fourier
transformation unit; a propagation path compensation unit that
compensates the signal in the frequency domain transformed by the
Fourier transformation unit by using the propagation path
characteristic estimated by the propagation path estimation unit; a
power arithmetic section that arithmetically operates power of the
signal in the frequency domain transformed by the Fourier
transformation unit; a first reciprocal processing section that
performs reciprocal processing on the power arithmetically operated
by the power arithmetic section to output a signal; an error
arithmetic section that arithmetically operates an error of the
signal compensated by the propagation path compensation unit; a
subtractor that subtracts the signal output from the first
reciprocal processing section from the error arithmetically
operated by the error arithmetic section to output a signal; a
second reciprocal processing unit that performs reciprocal
processing on the signal output from the subtractor to output a
signal; a first multiplier that multiplies the power arithmetically
operated by the power arithmetic section and the signal output from
the second reciprocal processing unit together to output a signal;
and a second multiplier that multiplies the signal compensated by
the propagation path compensation unit and the signal output from
the first multiplier together to output a signal to the adder.
2. The receiving device according to claim 1, further comprising: a
first averaging section that averages the power arithmetically
operated by the power arithmetic section for each frequency to
output a signal; and a second averaging section that averages the
error arithmetically operated by the error arithmetic section for
each frequency to output a signal, wherein the first reciprocal
processing section performs reciprocal processing on the signal
output from the first averaging section, the subtractor subtracts a
signal output from the first reciprocal processing section from the
signal output from the second averaging section, and the first
multiplier multiplies the signal output from the first averaging
section and the signal output from the second reciprocal processing
unit together.
3. The receiving device according to claim 1, wherein the
subtractor adjusts a difference between the error arithmetically
operated by the error arithmetic section and a variance of the
signal output from the first reciprocal processing section and then
performs a subtraction.
4. The receiving device according to claim 1, wherein the
subtractor adjusts an average value of the signal output from the
first reciprocal processing section and then performs a
subtraction.
5. The receiving device according to claim 1, further comprising: a
nonlinear transformation section that nonlinearly transforms the
power arithmetically operated by the power arithmetic section or
the signal output from the first reciprocal processing section to
output a signal.
6. The receiving device according to claim 3, further comprising: a
peak-cut section that removes an error equal to or more than a
threshold value from errors arithmetically operated by the error
arithmetic section to output a signal, wherein the subtractor
adjusts a difference in variance between the signal output from the
peak-cut section and the signal output from the first reciprocal
processing section.
Description
CROSS-REFERENCE TO RELATED APPLICATION
[0001] This application is based upon and claims the benefit of
priority of the prior Japanese Patent Application No. 2014-037072,
filed on Feb. 27, 2014, the entire contents of which are
incorporated herein by reference.
FIELD
[0002] The embodiments discussed herein are directed to a receiving
device.
BACKGROUND
[0003] There has been known an on-vehicle apparatus that receives
in a diversity reception system a broadcasting signal modulated in
an orthogonal frequency division multiplexing system (see Patent
Document 1, for example). A plurality of receiving parts receive a
broadcasting signal. A weighting value setting part sets a
weighting value according to a signal level of each of the
broadcasting signals received by the receiving parts. A signal
combination part performs weighting processing by a weighting value
set by the weighting value setting part on the respective
broadcasting signals corresponding to respective carrier
frequencies received by the receiving parts, and
maximum-ratio-combines broadcasting signals obtained after the
weighting processing. The weighting value setting part includes: an
interference detection part; and a weighting value adjustment part.
The interference detection part detects a carrier frequency
containing noise from the received broadcasting signal. The
weighting value setting part sets, as a weighting value to be
applied to the carrier frequency detected by the interference
detection part, a weighting value lower than the weighting value
set according to the signal level.
[0004] There has been known an ICI amount estimation device that is
included in a receiving device of a multicarrier signal and
estimates an ICI amount in a carrier signal (see Patent Document 2,
for example). A propagation path variation estimation unit
calculates a variation amount of a propagation path frequency
characteristic to output a propagation path variation
characteristic. A fixity coefficient multiplying unit multiplies
the propagation path variation characteristic by a fixity
coefficient determined according to the predetermined number of
carriers. The ICI amount estimation device estimates an ICI amount
in each carrier based on the propagation path variation
characteristic.
[0005] There has been known a receiving device including a
plurality of antennas (see patent Document 3, for example). A
plurality of synthesizing units generate weighting coefficients
used for controlling amplitudes and phases of baseband signals only
by the number of baseband signals by using band components
different from one another out of individual baseband signals
obtained by the plurality of antennas and multiply the individual
baseband signals and the individual weighting coefficients together
respectively, and then add these. A plurality of demodulation
circuits, on synthesized signals output from the individual
synthesizing units, perform fast Fourier transformation and perform
demodulation processing based on an orthogonal frequency division
multiplexing system for each subcarrier, and thereby generate
amplitude and phase data. A carrier synthesizing unit synthesizes
data output from the individual demodulation circuits for each
subcarrier.
[0006] There has been known an OFDM diversity receiver having a
plurality of reception branches that receive orthogonal frequency
division multiplexing (OFDM) signals containing a plurality of
subcarriers orthogonal to each other and output the received
signals individually (see Patent Document 4, for example). An
interference wave detection unit determines the presence and
absence of an interference wave in each subcarrier of the received
signals and estimates a first subcarrier group where interference
waves exist and a second subcarrier group where no interference
waves exist. A multiplying unit multiplies the first subcarrier
group by a first weight used for eliminating the interference waves
and multiplies the second subcarrier group by a second weight used
for maximizing a signal-to-noise ratio. A combining unit combines
output signals from the multiplying unit.
[0007] There has been known a receiving device for combining OFDM
signals that receives an OFDM signal by a reception antenna
composed of a plurality of array elements (see Patent Document 5,
for example). A FFT unit transforms an OFDM signal received by the
reception antenna into a reception carrier symbol in a frequency
domain. An array combining unit weights and combines the reception
carrier symbol by a first weighting coefficient for each subcarrier
composing the OFDM signal to generate an array combined signal. A
weighting coefficient optimizing unit generates a reference signal
in which a transmission symbol has been estimated and generates a
second weighting coefficient so that an error between the reference
signal and the array combined signal may become minimum. A filter
processing unit filters the reciprocal of the second weighting
coefficient and then generates the re-reciprocal of the filtered
reciprocal of the second weighting coefficient as a first weighting
coefficient.
[0008] [Patent Document 1] Japanese Laid-open Patent Publication
No. 2010-226233
[0009] [Patent Document 2] Japanese Laid-open Patent Publication
No. 2009-141740
[0010] [Patent Document 3] Japanese Laid-open Patent Publication
No. 2006-217399
[0011] [Patent Document 4] Japanese Laid-open Patent Publication
No. 2006-186421
[0012] [Patent Document 5] Japanese Laid-open Patent Publication
No. 2011-188221
[0013] In radio communication, frequency selective fading caused by
multipath occurs and reception quality deteriorates. Further, a
spurious wave in a narrow-band is sometimes mixed in a frequency
band of a reception signal. When power of a spurious wave becomes
large to some extent with respect to a desired wave in a carrier
unit, reception quality in a carrier with spurious waves
deteriorates.
SUMMARY
[0014] A receiving device includes: a plurality of antennas; each
of a plurality of receiving circuits that receive signal via one of
the plurality of antennas, respectively; and an adder that adds
signals output from the plurality of receiving circuits, in which
each of the plurality of receiving circuits includes: a Fourier
transformation unit that transforms a signal into a frequency
domain from a time domain; a propagation path estimation unit that
estimates a propagation path characteristic based on a known signal
in the signal in the frequency domain transformed by the Fourier
transformation unit; a propagation path compensation unit that
compensates the signal in the frequency domain transformed by the
Fourier transformation unit by using the propagation path
characteristic estimated by the propagation path estimation unit; a
power arithmetic section that arithmetically operates power of the
signal in the frequency domain transformed by the Fourier
transformation unit; a first reciprocal processing section that
performs reciprocal processing on the power arithmetically operated
by the power arithmetic section to output a signal; an error
arithmetic section that arithmetically operates an error of the
signal compensated by the propagation path compensation unit; a
subtractor that subtracts the signal output from the first
reciprocal processing section from the error arithmetically
operated by the error arithmetic section to output a signal; a
second reciprocal processing unit that performs reciprocal
processing on the signal output from the subtractor to output a
signal; a first multiplier that multiplies the power arithmetically
operated by the power arithmetic section and the signal output from
the second reciprocal processing unit together to output a signal;
and a second multiplier that multiplies the signal compensated by
the propagation path compensation unit and the signal output from
the first multiplier together to output a signal to the adder.
[0015] The object and advantages of the invention will be realized
and attained by means of the elements and combinations particularly
pointed out in the claims.
[0016] It is to be understood that both the foregoing general
description and the following detailed description are exemplary
and explanatory and are not restrictive of the invention.
BRIEF DESCRIPTION OF DRAWINGS
[0017] FIG. 1 is a view illustrating a structure example of a
receiving device according to a first embodiment;
[0018] FIG. 2 is a diagram illustrating a configuration example of
a first receiving circuit in FIG. 1;
[0019] FIG. 3 is a view illustrating signal points;
[0020] FIG. 4 is a view illustrating a processing method of the
first receiving circuit in FIG. 2;
[0021] FIG. 5A and FIG. 5B are views illustrating examples of
signals when a subtractor is removed from FIG. 2;
[0022] FIG. 6 is a diagram illustrating a configuration example of
a first receiving circuit according to a second embodiment;
[0023] FIG. 7A and FIG. 7B are views for explaining a
transformation method of a nonlinear transformation section in FIG.
6;
[0024] FIG. 8 is a view for explaining processing methods of an
average value measuring section and a peak-cut section in FIG.
6;
[0025] FIG. 9 is a diagram illustrating a configuration example of
a first receiving circuit according to a third embodiment; and
[0026] FIG. 10 is a diagram illustrating a configuration example of
a first receiving circuit according to a fourth embodiment.
DESCRIPTION OF EMBODIMENTS
First Embodiment
[0027] FIG. 1 is a view illustrating a structure example of a
receiving device according to a first embodiment. The receiving
device includes: a first antenna 101a; a first receiving circuit
102a; a second antenna 101b; a second receiving circuit 102b; an
adder 103; and a decoding unit 104. A radio signal that a
transmitting device transmits propagates to a radio apparatus via a
plurality of paths having different propagation times due to
reflection from a building and the like (multipath). Thereby,
fading caused by interference, phase shift, or the like of the
radio signal occurs. The antennas 101a and 101b are disposed
mutually at an interval corresponding to wavelengths of radio
reception signals to receive radio signals. The first receiving
circuit 102a receives the signal received via the first antenna
101a and compensates the received signal according to a propagation
path characteristic to restore a transmitted signal. The second
receiving circuit 102b receives the signal received via the second
antenna 101b and compensates the received signal according to a
propagation path characteristic to restore a transmitted signal.
Further, the first receiving circuit 102a weights the received
signal of the first antenna 101a by a first weighting coefficient
according to a carrier-to-noise ratio (CNR) of the received signal
of the first antenna 101a to output a signal. The second receiving
circuit 102b weights the received signal of the second antenna 101b
by a second weighting coefficient according to a carrier-to-noise
ratio (CNR) of the received signal of the second antenna 101b to
output a signal. When the CNR is large, the weighting coefficient
is increased, and when the CNR is small, the weighting coefficient
is decreased. The adder 103 adds the received signal weighted by
the first receiving circuit 102a and the received signal weighted
by the second receiving circuit 102b to output a signal. Thereby,
it is possible to decrease an effect of fading caused by multipath
to improve reception quality. The decoding unit 104 decodes the
signal output from the adder 103.
[0028] Incidentally, the example where two pairs of a pair of the
first antenna 101a and the first receiving circuit 102a and a pair
of the second antenna 101b and the second receiving circuit 102b
are provided has been explained, but it is also possible to provide
three or more pairs and perform an addition in the adder 103.
[0029] FIG. 2 is a diagram illustrating a configuration example of
the first receiving circuit 102a in FIG. 1, and FIG. 4 is a view
illustrating a processing method of the first receiving circuit
102a in FIG. 2. In FIG. 4, the horizontal axis indicates a carrier,
and the vertical axis indicates the magnitude of signal. The first
receiving circuit 102a includes: a RF (Radio Frequency) processing
unit 201; an analog and digital (A and D) converting unit 202; a
guard interval (GI) removing unit 203; a Fast Fourier
Transformation (FFT) unit 204; a propagation path estimation unit
205; a propagation path compensation unit 206; a weighting
coefficient generation unit 219; a second reciprocal processing
unit 216; a first multiplier 217; and a second multiplier 218. The
weighting coefficient generation unit 219 includes: a modulation
error ratio (MER) measuring section 207; a second averaging section
208; a power arithmetic section 209; a first averaging section 210;
a first reciprocal processing section 211; a variance average value
measuring section 212; a variance measuring section 213; a variance
average value adjusting section 214; and a subtractor 215.
Hereinafter, the configuration of the first receiving circuit 102a
will be explained as an example, but the second receiving circuit
102b also has a configuration similar to that of the first
receiving circuit 102a.
[0030] The first antenna 101a receives radio signals in an
orthogonal frequency division multiplexing (OFDM) system. As for
the radio signals in the OFDM system, symbol signals are
transmitted at a predetermined time interval. Each symbol has a
plurality of carriers (frequencies). The RF processing unit 201
down-converts the frequency of the signal received via the first
antenna 101a and converts the signal into a baseband signal from a
RF signal to output a signal. The analog and digital converting
unit 202 analog-to-digital converts the signal output from the RF
processing unit 201 to output a signal. The GI removing unit 203
removes a guard interval of the signal output from the analog and
digital converting unit 202 to output a signal. The guard interval
is a redundant portion obtained by copying a rear portion of data
of a symbol to add the copied portion to the front of the date for
the purpose of preventing intersymbol interference in which data of
a symbol interferes with data of the previous symbol and data of
the subsequent symbol. The FFT unit 204 transforms the signal
output from the GI removing unit 203 into a frequency domain from a
time domain by Fourier transformation to output a signal A1. The
signal A1 contains an I channel signal and a Q channel signal in
each carrier as illustrated in FIG. 3.
[0031] The propagation path estimation unit 205 estimates a
propagation path characteristic A2 based on a known pilot signal in
the signal A1 in the frequency domain transformed by the FFT unit
204. The pilot signal is a known signal contained in a
predetermined carrier of each symbol, and communication data is
contained in the other carriers. The pilot signal contains known
data and is dispersively disposed in a symbol (time) direction and
in a carrier (frequency) direction. As illustrated in FIG. 3, a
transmission signal T of the transmitting device propagates to the
receiving device as a reception signal R through a propagation
path. The reception signal R of the receiving device results in a
signal obtained by multiplying the transmission signal T of the
transmitting device by the propagation path characteristic A2. The
propagation path estimation unit 205 estimates the propagation path
characteristic A2 by dividing a pilot signal R of the reception
signal by a pilot signal T of the known transmission signal. The
propagation path compensation unit 206 compensates the signal A1 by
dividing the signal A1 output from the FFT unit 204 by the
propagation path characteristic A2 to restore a signal A3
corresponding to the transmission signal.
[0032] The power arithmetic section 209 squares an I channel
component and a Q channel component of the known pilot signal in
the signal A1 to sum the results, to thereby arithmetically operate
power of the known pilot signal in the signal A1 to output the
power. Incidentally, it is also possible that the power arithmetic
section 209 squares I channel components and Q channel components
of signals of all the carriers in the signal A1 to sum the results,
to thereby arithmetically operate power of the signals of all the
carriers in the signal A1. The first averaging section 210 averages
the power output by the power arithmetic section 209 in the symbol
(time) direction for each carrier to output a signal A4. By the
averaging, an effect of noise such as additive white Gaussian noise
(AWGN) is decreased and only a frequency selective fading component
caused by multipath is left.
[0033] As illustrated in FIG. 4, the signal A4 contains notches
caused by frequency selective fading. The signal A4 becomes
substantially the same in all the carriers if there is no effect of
propagation paths. However, due to the effect of multipath of
propagation paths, frequency selective fading in which fading
variation differs in each carrier occurs. Thereby, the signal A4
differs in magnitude in each carrier, and a notch in which the
signal A4 becomes small in a predetermined carrier is
generated.
[0034] The first reciprocal processing section 211 performs
reciprocal processing on the signal output from the first averaging
section 210 to output a signal A5. The variance average value
measuring section 212 measures a variance 401 and an average value
402 of the signal A5.
[0035] The MER measuring section 207 is an error arithmetic section
that measures a MER being an error of the signal A3. In the case of
quadrature phase shift keying (GPSK), data of a symbol is expressed
by four ideal signal points 301 to 304 as illustrated in FIG. 3.
The four ideal signal points 301 to 304 are expressed by the I
channel signal and the Q channel signal. A reception signal point
305 in a first quadrant is estimated to contain an error with
respect to the ideal signal point 301 in the first quadrant. A
reception signal point in a second quadrant is estimated to contain
an error with respect to the ideal signal point 302 in the second
quadrant. A reception signal point in a third quadrant is estimated
to contain an error with respect to the ideal signal point 303 in
the third quadrant. A reception signal point in a fourth quadrant
is estimated to contain an error with respect to the ideal signal
point 304 in the fourth quadrant. For example, the ideal signal
point 301 is a signal point of the transmission signal T. The
reception signal point 305 is a signal point of the reception
signal R. The MER measuring section 207 subtracts the ideal signal
point 301 of the transmission signal T from the reception signal
point 305 of the reception signal R to arithmetically operate a
signal R-T as a MER (an error).
[0036] The second averaging section 208 averages a signal output
from the MER measuring section 207 in the symbol (time) direction
for each carrier to output a signal A6. In the signal A6, MERs 403
caused by spurious waves in a narrow-band are mixed in addition to
notches caused by frequency selective fading similar to those of
the signal A5. Incidentally, the signal A4 is on the same dimension
as that of the CNR, and the signal A5 is on the same dimension as
that of the MER of the signal A6. The variance measuring section
213 measures a variance 404 of the signal A6. An average value 405
is an average value of the signal A6.
[0037] The variance average value adjusting section 214 receives
the variance 401 and the average value 402 of the signal A5 and the
variance 404 of the signal A6 and adjusts the variance and the
average value of the signal A5 to output a signal A7. A variance of
the signal A7 is adjusted to be the same as the variance 404 of the
signal A6. An average value of the signal A7 is adjusted to be
"0."
[0038] Incidentally, the variance average value adjusting section
214 may be the one to adjust the variance 401 of the signal A5 and
the variance 404 of the signal A6 to be the same each other. That
is, the variance average value adjusting section 214 adjusts the
variance of the signal A5 and the variance of the signal A6 so as
to obtain a small difference in variance between the signal A5 and
the signal A6.
[0039] When the reception signal is automatically gain controlled
(AGC), the magnitude of the signal A5 varies according to a gain
value. Further, an arithmetic method of the power arithmetic
section 209 and a measurement method of the MER measuring section
207 are different, so that the variance 401 of the signal A5 and
the variance 404 of the signal A6 do not often agree with each
other. Further, there is also a method in which a gain value of AGC
is used to estimate the magnitude of the signal A5, but an AGC
amplifier often has nonlinear characteristics, and it is difficult
to estimate a correct gain value of AGC. Thus, in this embodiment,
the variances and the average values are adjusted by the variance
average value adjusting section 214.
[0040] The subtractor 215 subtracts the signal A7 from the signal
A6 for each carrier to output a signal A8. The signal A6 contains
an error caused by frequency selective fading and errors 403 caused
by spurious waves. The signal A7 contains an error caused by
frequency selective fading. Due to the subtraction, the signal A8
contains only the errors 403 caused by spurious waves. The second
reciprocal processing unit 216 performs reciprocal processing on
the signal A8 output from the subtractor 215 to output a signal A9.
The signal A9 is on the same dimension as that of the CNR and
contains a CNR component by spurious waves.
[0041] The first multiplier 217 multiplies the signal output from
the power arithmetic section 209 and the signal A9 together for
each carrier to output a signal A10. The signal A4 is a weighting
coefficient containing a CNR component by frequency selective
fading. The signal A9 is a weighting coefficient containing a CNR
component by spurious waves. The signal A10 is a weighting
coefficient containing a CNR component by frequency selective
fading and a CNR component by spurious waves.
[0042] The second multiplier 218 multiplies the signal A3
compensated by the propagation path compensation unit 206 by the
signal A10 to output a signal obtained by the multiplication to the
adder 103 in FIG. 1. For example, in FIG. 1, when the CNR in the
first receiving circuit 102a is large and the CNR in the second
receiving circuit 102b is small, the signal A10 being the weighting
coefficient of the first receiving circuit 102a increases and the
signal A10 being the weighting coefficient of the second receiving
circuit 102b decreases. Conversely, when the CNR in the first
receiving circuit 102a is small and the CNR in the second receiving
circuit 102b is large, the signal A10 being the weighting
coefficient of the first receiving circuit 102a decreases and the
signal A10 being the weighting coefficient of the second receiving
circuit 102b increases. The plural antennas 101a and 101b are
provided, the weighting coefficient of the receiving circuit 102a
or 102b having a larger CNR is increased, and multiplication is
performed, thereby making it possible to decrease effects of
frequency selective fading and spurious waves. This makes it
possible to improve reception quality.
[0043] Next, there will be explained an advantage obtained by
providing the subtractor 215. FIG. 5A and FIG. 5B are views
illustrating examples of the signals A4, A9, and A10 when the
subtractor 215 is removed from FIG. 2. In this case, the second
reciprocal processing unit 216 performs reciprocal processing on
the signal A6 output from the second averaging section 208. FIG. 5A
illustrates the examples of the signals A4, A9, and A10 of the
first receiving circuit 102a. FIG. 5B illustrates the examples of
the signals A4, A9, and A10 of the second receiving circuit 102b.
The signal A4 is the output signal of the first averaging section
210, and contains received power 501 by frequency selective fading
and received power 502 by spurious waves. The signal A9 is the
output signal of the second reciprocal processing unit 216, and
contains a CNR component 503 by frequency selective fading and a
CNR component 504 by spurious waves. The signal A10 is the output
signal of the first multiplier 217, and is a signal obtained by
multiplying the signal output from the power arithmetic section 209
and the signal A9 together. The first multiplier 217 multiplies the
received power 501 by frequency selective fading and the CNR
component 503 by frequency selective fading together in a carrier
not having the received power 502 by spurious waves and in a
carrier not having the CNR component 504 by spurious waves. As a
result, a frequency selective fading component is squared, as is a
signal 505 in the signal A10, the frequency selective fading
component is emphasized too much, and appropriate weighting cannot
be applied, resulting in that reception quality deteriorates.
[0044] That is, the signal A10 preferably contains only one of the
received power 501 by frequency selective fading in the signal A4
and the CNR component 503 by frequency selective fading in the
signal A9 in a carrier having no spurious waves. When the received
power 501 by frequency selective fading and the CNR component 503
by frequency selective fading are multiplied together, as is the
signal 505 in the signal A10, the frequency selective fading
component is emphasized too much and reception quality
deteriorates.
[0045] In contrast to this, in this embodiment, as illustrated in
FIG. 2, the subtractor 215 is provided and thereby the signal A7 is
subtracted from the signal A6. As illustrated in FIG. 4, the signal
A6 contains the error caused by frequency selective fading and the
errors 403 caused by spurious waves. By subtracting the signal A7
from the signal A6, in the signal A8, the error caused by frequency
selective fading is removed and only the errors 403 caused by
spurious waves remain. The first multiplier 217 multiplies the
signal output from the power arithmetic section 209 and the signal
A9 together, so that in the signal A10 being the weighting
coefficient, the frequency selective fading component in the signal
A6 is removed and the frequency selective fading component in the
signal A7 is contained. Thereby, it is possible to prevent that the
frequency selective fading is emphasized too much by the square of
the frequency selective fading component as is the signal 505 in
the signal A10 in FIG. 5A and FIG. 5B, and to generate the signal
A10 being an appropriate weighting coefficient. This makes it
possible to improve reception quality.
Second Embodiment
[0046] FIG. 6 is a diagram illustrating a configuration example of
a first receiving circuit 102a according to a second embodiment. A
second receiving circuit 102b also has a configuration similar to
that of the first receiving circuit 102a. This embodiment (FIG. 6)
is one in which a nonlinear transformation section 601, an average
value measuring section 602, and a peak-cut section 603 are added
to the first embodiment (FIG. 2). Hereinafter, there will be
explained points of which this embodiment is different from the
first embodiment.
[0047] The nonlinear transformation section 601 nonlinearly
transforms a signal A4 output from a first averaging section 210 to
output a signal to a first reciprocal processing section 211. The
first reciprocal processing section 211 performs reciprocal
processing on the signal output from the nonlinear transformation
section 601 to output a signal A5.
[0048] FIG. 7A and FIG. 7B are views for explaining a
transformation method of the nonlinear transformation section 601
in FIG. 6. As illustrated in FIG. 7A, the signal A4 has a linear
characteristic with respect to an actual CNR. In contrast to this,
as illustrated in FIG. 7B, a signal A9 has a nonlinear
characteristic with respect to an actual CNR. As for the signal A9,
the characteristic becomes nonlinear due to an effect of bit
precision of a digital signal in a region with a small CNR.
Further, as for the signal A9, in a region with a large CNR, the
characteristic becomes nonlinear in order to, as illustrated in
FIG. 3, deny the possibility that the reception signal point 305 in
the first quadrant is that an error is caused in the ideal signal
point 302 in the second quadrant and to arithmetically operate an
error R-T between the ideal signal point 301 and the reception
signal point 305 in the first quadrant. When a subtractor 215
performs a subtraction based on the signal A4 having a linear
characteristic in FIG. 7A and the signal A9 having a nonlinear
characteristic in FIG. 7B, the precision of a signal A8 decreases.
Thus, the nonlinear transformation section 601 nonlinearly
transforms the signal A4 so as to match the nonlinear
characteristic of the signal A9 in FIG. 7B. Concretely, in
consideration of the first reciprocal processing section 211
performing reciprocal processing, the nonlinear transformation
section 601 nonlinearly transforms the signal A4 so as to provide
an inverse characteristic to the nonlinear characteristic in FIG.
7B. For example, the nonlinear transformation section 601 performs
nonlinear transformation such that the characteristic becomes
inverse to nonlinear transformation with the horizontal axis in
FIG. 7B set as an input signal and the vertical axis in FIG. 7B set
as an output signal. The nonlinear transformation section 601
performs nonlinear transformation with a logarithmic transformation
function, a transformation function based on a square root, or a
transformation table. In the case of the transformation table, it
is only necessary that the nonlinear characteristic of FIG. 7B
should be measured beforehand and a transformation table that
provides an inverse characteristic of the nonlinear characteristic
should be used.
[0049] Incidentally, the nonlinear transformation section 601 may
also be provided at the subsequent stage of the first reciprocal
processing section 211. In the case, the first reciprocal
processing section 211 performs reciprocal processing on the signal
A4 to output a signal. The nonlinear transformation section 601,
similarly to the nonlinear characteristic in FIG. 7B, nonlinearly
transforms the signal output from the first reciprocal processing
section 211 to output a nonlinearly transformed signal to a
variance average value measuring section 212 and a variance average
value adjusting section 214.
[0050] FIG. 8 is a view for explaining processing methods of the
average value measuring section 602 and the peak-cut section 603 in
FIG. 6. In the first embodiment (FIG. 2), the variance measuring
section 213 measures a variance 802 of the signal A6. However, when
the number of carriers containing a spurious wave component is
increased in the signal A6, the variance 802 of the signal A6
increases. The variance measuring section 213 preferably removes
the spurious wave component from the signal A6 to obtain a variance
801 of only a propagation path component.
[0051] The average value measuring section 602 measures an average
value 805 of the signal A6 output from a second averaging section
208. The peak-cut section 603 sets a threshold value 804 obtained
by constant multiplying the average value 805 of the signal A6 and
sets an error 803 that is equal to or more than the threshold value
804 in the average value 805 to output a signal A11. That is, the
peak-cut section 603 removes the error 803 equal to or more than
the threshold value 804 from the signal A6 and sets an average
value of the signal A11 in the average value 805 to output the
signal A11. The variance measuring section 213 measures a variance
of the signal A11 output from the peak-cut section 603. Thereby,
the variance measuring section 213 can remove a spurious wave
component and obtain a variance of a propagation path
component.
Third Embodiment
[0052] FIG. 9 is a diagram illustrating a configuration example of
a first receiving circuit 102a according to a third embodiment. A
second receiving circuit 102b also has a configuration similar to
that of the first receiving circuit 102a. Hereinafter, there will
be explained points of which this embodiment (FIG. 9) is different
from the first embodiment (FIG. 2). A first multiplier 217 receives
a signal A4 output from a first averaging section 210 in place of a
signal output from a power arithmetic section 209. That is, the
first multiplier 217 multiplies the signal A4 output from the first
averaging section 210 and a signal A9 together for each carrier to
output a signal A10.
[0053] On the condition that a receiving device stands still or a
moving speed of the receiving device is slow, when the first
multiplier 217 receives the signal A4 output from the first
averaging section 210 rather than the signal output from the power
arithmetic section 209, it is sometimes possible to decrease an
effect of AWGN and to improve reception quality.
[0054] Incidentally, a spurious wave is normally generated at the
same frequency and with the same magnitude constantly, and in
contrast to this, the signal output from the power arithmetic
section 209 changes from moment to moment by the receiving device
moving mainly. Thus, when the first multiplier 217 receives the
signal A4 output from the first averaging section 210, changes of
the signal A4 become gentle to be difficult to be reflected in the
signal A10 being a weighting coefficient. Therefore, when the
receiving device moves, as is the first embodiment (FIG. 2), the
first multiplier 217 sometimes preferably receives the signal
output from the power arithmetic section 209.
Fourth Embodiment
[0055] FIG. 10 is a diagram illustrating a configuration example of
a first receiving circuit 102a according to a fourth embodiment. A
second receiving circuit 102b also has a configuration similar to
that of the first receiving circuit 102a. Hereinafter, there will
be explained points of which this embodiment (FIG. 10) is different
from the second embodiment (FIG. 6). Similarly to the third
embodiment, a first multiplier 217 receives a signal A4 output from
a first averaging section 210 in place of a signal output from a
power arithmetic section 209. That is, the first multiplier 217
multiplies the signal A4 output from the first averaging section
210 and a signal A9 together for each carrier to output a signal
A10. Thereby, this embodiment can obtain an effect similar to that
of the third embodiment.
[0056] As described above, according to the first to fourth
embodiments, even when a spurious wave is mixed in a reception
signal in addition to frequency selective fading caused by
multipath, weighting is applied by an optimized weighting
coefficient for each carrier and signals output from the plural
receiving circuits 102a and 102b are combined in the adder 103.
Thereby, it is possible to decrease the effect of frequency
selective fading caused by multipath and the effect by spurious
waves and to improve reception quality.
[0057] It should be noted that the above embodiments merely
illustrate concrete examples of implementing the present invention,
and the technical scope of the present invention is not to be
construed in a restrictive manner by these embodiments. That is,
the present invention may be implemented in various forms without
departing from the technical spirit or main features thereof.
[0058] The error arithmetic section outputs an error caused by an
effect of frequency selective fading caused by multipath and an
effect of spurious waves. The first reciprocal processing section
outputs an error caused by an effect of frequency selective fading
caused by multipath. The subtractor outputs an error caused by an
effect of spurious waves. The effect of spurious waves can be
decreased by the first multiplier and the effect of frequency
selective fading caused by multipath can be decreased by the second
multiplier, resulting in that it is possible to improve reception
quality.
[0059] All examples and conditional language provided herein are
intended for the pedagogical purposes of aiding the reader in
understanding the invention and the concepts contributed by the
inventor to further the art, and are not to be construed as
limitations to such specifically recited examples and conditions,
nor does the organization of such examples in the specification
relate to a showing of the superiority and inferiority of the
invention. Although one or more embodiments of the present
invention have been described in detail, it should be understood
that the various changes, substitutions, and alterations could be
made hereto without departing from the spirit and scope of the
invention.
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